CN102239633A - Wireless Energy Transfer System - Google Patents

Abstract

Described herein are methods for having a Q factor Q₁> 100 and characteristic dimension x₁A source resonator coupled to an energy source and having a Q factor Q₂> 100 and characteristic dimension x₂The improved capability of a second resonator coupled to an energy consuming device located at a distance D from the source resonator, wherein the source resonator and the second resonator are coupled to wirelessly exchange energy between the source resonator and the second resonator.

Description

Wireless Energy Transfer System

Cross References to Related Applications

This application claims priority to the following U.S. patent applications, each of which is hereby incorporated by reference in its entirety:

U.S. Application No. 61/100,721, filed September 27, 2008; U.S. Application No. 61/108,743, filed October 27, 2008; U.S. Application No. 61/147,386, filed January 26, 2009; 2009 U.S. Application No. 61/152,086, filed February 12; U.S. Application No. 61/178,508, filed May 15, 2009; U.S. Application No. 61/182,768, filed June 1, 2009; December 2008 U.S. Application No. 61/121,159 filed on 9th; U.S. Application No. 61/142,977 filed on January 7, 2009; U.S. Application No. 61/142,885 filed on January 6, 2009; January 6, 2009 U.S. Application No. 61/142,796 filed January 6, 2009; U.S. Application No. 61/142,880 filed January 6, 2009; U.S. Application No. 61/142,818; U.S. Application No. 61/142,887, filed January 6, 2009; U.S. Application No. 61/156,764, filed March 2, 2009; U.S. Application No. 6, filed January 7, 2009 No. 61/143,058; U.S. Application No. 61/152,390, filed February 13, 2009; U.S. Application No. 61/163,695, filed March 26, 2009; U.S. Application No. 6, filed April 24, 2009 61/172,633; US Application No. 61/169,240, filed April 14, 2009; and US Application No. 61/173,747, filed April 29, 2009.

background

technical field

The present disclosure relates to wireless energy transfer, also known as wireless power transfer.

Background technique

Energy or power can be transferred wirelessly using a variety of known radiating or far-field and non-radiating or near-field techniques. For example, radiative wireless information transfer using low-directivity antennas, such as those used in radio and cellular communication systems and home computer networks, can be considered wireless energy transfer. However, such radiative transfer is very inefficient since only a small fraction of the supplied or radiated power is harvested, ie the fraction that is along the direction of the receiver and overlaps it. Most of the power is radiated away along all other directions and lost in free space. Such inefficient power transfer is acceptable for data transmission, but impractical for transferring useful amounts of electrical energy for industrial purposes, such as to power or charge electrical devices. One way to improve the transfer efficiency of certain energy transfer schemes is to use directional antennas to confine and preferentially direct the radiated energy towards the receiver. However, these directional radiation schemes may require uninterrupted line of sight and potentially complex tracking and steering mechanisms with moving transmitters and/or receivers. Additionally, such schemes may pose a hazard to objects or persons passing through or traversing the beam while moderate or large amounts of power are being delivered. Known non-radiative or near-field wireless energy transfer schemes, often referred to as inductive or traditional inductive, do not (intentionally) radiate power, but instead use an oscillating current flowing through a primary coil to induce a current in a nearby receiving or secondary coil oscillating magnetic near-field. Traditional inductive schemes have demonstrated transfer of moderate to substantial power, however only over very short distances and with very small offset tolerances between the main power supply unit and the auxiliary receiver unit. Electric transformers and proximity chargers are examples of devices that utilize this known short-range, near-field energy transfer scheme.

Therefore, there is a need for a wireless power transfer scheme capable of transferring useful amounts of electrical power over mid-range distances or alignment offsets. Such wireless power transfer schemes should be able to transfer useful energy over greater distances and alignment offsets than those achieved by traditional inductive schemes, but without the limitations and risks inherent in radiative transfer schemes.

Contents of the invention

A non-radiative or near-field wireless energy transfer scheme capable of transferring useful amounts of power over medium-range distances and alignment offsets is disclosed herein. The technique of the present invention uses coupled electromagnetic resonators with long-lived oscillatory resonant modes to transfer power from a power source to a power drain. The technique is comprehensive and can be applied to a wide range of resonators, even in the case of the specific example disclosed herein involving electromagnetic resonators. If the resonator is designed such that the energy stored by the electric field is primarily confined within the structure and the energy stored by the magnetic field is primarily in the region around the resonator. The energy exchange is mediated mainly by the resonant magnetic near field. These types of resonators may be referred to as magnetic resonators. If the resonator is designed such that the energy stored by the magnetic field is mainly confined within the structure and the energy stored by the electric field is mainly in the region around the resonator. The energy exchange is mediated mainly by the resonant electric near field. These types of resonators may be referred to as electrical resonators. Either type of resonator may also be referred to as an electromagnetic resonator. Both types of resonators are disclosed herein.

The omnidirectional but fixed (lossless) nature of the near-field of our disclosed resonators enables efficient wireless energy transfer over medium-range distances over a wide range of directions and resonator orientations, suitable for charging, powering, or simultaneously Power and charge various electronic devices. As a result, a system can have multiple possible applications where a first resonator connected to a power source is in one location and a second resonator is potentially connected to electrical/electronic equipment, batteries, power supply and charging circuits, etc. In the second position, and wherein the distance from the first resonator to the second resonator is about a few centimeters to a few meters. For example, a first resonator connected to a wired power grid may be located on the ceiling of a room, while other resonators connected to devices such as robots, vehicles, computers, communication equipment, medical equipment, etc. move around in the room, and where , these devices receive power wirelessly from a source resonator either constantly or intermittently. For this one example, one can imagine many applications where the systems and methods disclosed herein can provide wireless power across medium-range distances, including consumer electronics devices, industrial applications, infrastructure power and lighting, transportation vehicles, Electronic games, military applications, etc.

Energy exchange between two electromagnetic resonators can be optimized when the resonators are tuned to substantially the same frequency and losses in the system are minimized. Wireless energy transfer systems can be designed such that the "coupling time" between resonators is much shorter than the "loss time" of the resonators. Accordingly, the systems and methods described herein can utilize high quality factor (high Q) resonators with low intrinsic loss rates. In addition, the systems and methods described herein may use sub-wavelength resonators with near-fields extending significantly longer than the characteristic dimensions of the resonators, such that the near-fields of the resonators exchanging energy overlap at mid-range distances. This is an area of operation that has not been practiced before and is distinctly different from traditional induction designs.

It is important to recognize the difference between the high-Q magnetic resonator scheme disclosed here and known proximity or proximity-sensing schemes, namely those known schemes that do not conventionally utilize high-Q resonators. Using coupled-mode theory (CMT) (see e.g. Waves and Fields in Optoelectronics , HAHaus, Prentice Hall, 1984), it can be shown that a high-Q resonator coupling mechanism enables resonators separated by mid-range distances than is achieved by conventional inductive schemes. Efficient power delivery that is several orders of magnitude higher in power delivery. Coupled high-Q resonators have demonstrated efficient energy transfer over medium-range distances and improved efficiency and offset tolerance in short-range energy transfer applications.

The systems and methods described herein can provide near-field wireless energy transfer via strongly coupled high-Q resonators, a method with the capability to transfer from picowatts to kilowatts safely and over much greater distances than can be achieved using conventional inductive techniques. power level potential of the technology. Efficient energy transfer can be achieved for a variety of general systems of strongly coupled resonators, such as strongly coupled acoustic resonators, atomic energy resonators, mechanical resonators, etc., as originally described by M.I.T. in its publication "Efficient wireless non-radiative mid-range energy transfer", Annals of Physics, vol.323, Issue 1, p.34 (2008) and "Wireless PowerTransfer via Strongly Coupled Magnetic Resonances", Science, vol.317, no.5834, p.83, ( 2007) as described. Disclosed herein are electromagnetic resonators and systems of coupled electromagnetic resonators, more specifically also referred to as coupled magnetic resonators and coupled electric resonators, having operating frequencies below 10 GHz.

This disclosure describes wireless energy transfer techniques, also referred to as wireless power transfer techniques. Throughout this disclosure, we will use the terms wireless energy transfer, wireless power transfer, wireless power transfer, etc. interchangeably. We may refer to the supply of energy or power from a source, AC or DC source, battery, source resonator, power supply, generator, solar panels and collectors, etc. to a device, remote device, multiple remote devices, one or multiple device resonators, etc. We can describe intermediate resonators by allowing energy to jump, transfer through, be temporarily stored, be partially dissipated, or in any way to mediate transfer from the source resonator to any combination of other devices and intermediate resonators Extend the range of wireless energy transfer systems so that energy transfer networks or strings or extended paths can be implemented. A device resonator can receive energy from a source resonator, convert a portion of that energy into electrical power for powering and charging a device, and simultaneously pass a portion of the received energy to other devices or to a mobile device resonator. Energy can be transferred from a source resonator to multiple device resonators, significantly extending the distance over which energy can be transferred wirelessly. Wireless power transfer systems can be implemented using a variety of system architectures and resonator designs. The system may include a single source or multiple sources delivering power to a single device or multiple devices. A resonator can be designed as a source or device resonator, or it can be designed as a repeater. In some cases, a resonator can be both a device and a source resonator, or it can be switched from operating as a source to operating as a device or repeater. Those skilled in the art will understand that a variety of system architectures can be supported by the wide range of resonator designs and functions described in this application.

In the wireless energy transfer system we describe, a remote device can be powered directly with wirelessly supplied power or energy, or the device can be coupled to an energy storage unit such as a battery, farad capacitor, supercapacitor, etc. (or other kind of power sink device), wherein the energy storage element can be charged or recharged wirelessly, and/or wherein the wireless power transfer mechanism is merely in addition to the main power source of the device. The device may be powered by a hybrid battery/energy storage device such as with an integrated storage capacitor. Additionally, novel batteries and energy storage devices can be designed to take advantage of the operational improvements enabled by wireless power transfer systems.

Other power management schemes include using wirelessly supplied power to recharge a battery or charge an energy storage unit while the device it is powering is turned off, in an idle state, in a sleep mode, and the like. The battery or energy storage unit can be charged or recharged at a high (fast) or low (slow) rate. Can trickle charge or float charge the battery or energy storage unit. Multiple devices may be charged or powered simultaneously in parallel, or power delivery to multiple devices may be serialized such that one or more devices receive power for a period of time after other power deliveries are switched to other devices. Multiple devices may be simultaneously or in time multiplexed or in frequency multiplexed or in space or in orientation multiplexed or in any combination of time and frequency and space and orientation multiplexed with one or more Other devices share power from one or more sources. Multiple devices may share power with each other, with at least one device being continuously, intermittently, periodically, occasionally or temporarily reconfigured to operate as a wireless power source. Those skilled in the art will understand that there are many ways to power and/or charge a device and that these ways can be applied to the techniques and applications described herein.

Wireless energy transfer has many possible applications including, for example, placing a source (e.g. one connected to a wired power grid) on the ceiling of a room, under the floor or in a wall, while placing a source such as a robot, vehicle, computer, PDA, etc. Place it indoors or move it freely indoors. Other applications may include powering or recharging electric engine vehicles, such as buses and/or hybrid cars, and medical devices, such as wearable or implantable devices. Additional exemplary applications include stand-alone electronic devices (e.g., laptop computers, cell phones, portable music players, household robots, GPS navigation systems, displays, etc.), sensors, industrial and manufacturing equipment, medical equipment and monitors, household appliances and implements (such as lamps, fans, drills, saws, heaters, monitors, televisions, over-the-counter appliances, etc.), military equipment, warming or lighting clothing, communication and navigation equipment, including embedded vehicles, clothing and protective clothing such as The ability to power or recharge devices such as helmets, body armor, and vests, and to deliver power to devices that are physically isolated, such as implanted medical devices, sensors or tags that are concealed, buried, implanted, or embedded, And/or from rooftop solar panels to indoor switchboards etc.

In one aspect, a system disclosed herein includes a source resonator having a quality factor _Q1 and a characteristic dimension _x1 coupled to a generator, and a quality factor _Q2 and a characteristic dimension _x2 coupled to a source resonator located at a second resonator of a load at a distance D, wherein the source resonator and the second resonator are coupled to wirelessly exchange energy between the source resonator and the second resonator, and wherein

且

Q₃可以小于100。Q ₁ can be less than 100. _Q2 can be less than 100. The system may include a third resonator having a quality factor _Q3 configured to non-radiatively transfer energy with the source and the second resonator, wherein

and

Q ₃ can be less than 100.

The source resonator can be coupled to the generator with a direct electrical connection. The system may include an impedance matching network, wherein the source resonator is coupled and impedance matched to the generator by a direct electrical connection. The system may include a tunable circuit, wherein the source resonator is coupled to the generator through the tunable circuit having a direct electrical connection. The tunable circuit may include a variable capacitor. The tunable circuit may include a variable inductor. The at least one direct electrical connection may be configured to substantially maintain a resonant mode of the source resonator. The source resonator may have a first terminal, a second terminal and a center terminal, and impedances between the first terminal and the center terminal and between the second terminal and the center terminal may be substantially equal. The source resonator may include a capacitively loaded loop having a first terminal, a second terminal and a center terminal, wherein impedances between the first terminal and the center terminal and between the second terminal and the center terminal are substantially equal. The source resonator may be coupled to an impedance matching network, and the impedance matching network may further include a first terminal, a second terminal, and a center terminal, wherein impedances between the first terminal and the center terminal and between the second terminal and the center terminal Basically equal.

The first and second terminals may be coupled directly to the generator and driven with oscillating signals approximately 180 degrees out of phase. The source resonator may have a resonant frequency ω ₁ , and the first and second terminals may be coupled directly to the generator and driven with an oscillating signal substantially equal to the resonant frequency ω ₁ . The center terminal may be connected to electrical ground. The source resonator may have a resonant frequency ω ₁ , and the first and second terminals may be directly coupled to the generator and driven at a frequency substantially equal to the resonant frequency. The system may include a plurality of capacitors coupled to a generator and a load. The source and second resonators may each be enclosed in a low loss tangent material. The system can include a power conversion circuit, wherein the second resonator is coupled to the power conversion circuit to deliver DC power to the load. The system can include a power conversion circuit, wherein the second resonator is coupled to the power conversion circuit to deliver AC power to the load. The system can include a power conversion circuit, wherein the second resonator is coupled to the power conversion circuit to deliver both AC and DC power to the load. The system may include a power conversion circuit and a plurality of loads, wherein the second resonator is coupled to the power conversion circuit and the power conversion circuit is coupled to the plurality of loads. The impedance matching network may include capacitors. The impedance matching network may include an inductor.

Throughout this disclosure, we may refer to certain circuit components such as capacitors, inductors, resistors, diodes, switches, etc. as circuit components or elements. We may also refer to series and parallel combinations of these components as elements, networks, topologies, circuits, etc. We can describe combinations of capacitors, diodes, varactors, transistors and/or switches as adjustable impedance networks, tuning networks, matching networks, tuning elements, etc. We may also refer to "self-resonant" objects that distribute both capacitance and inductance throughout the entire object (or partially, as opposed to being lumped individually). Those skilled in the art will appreciate that adjusting and controlling variable components within a circuit or network can adjust the performance of that circuit or network, and that those adjustments can be generally described as tuning, adjusting, matching, modifying, and the like. In addition to tuning tunable components such as inductors and capacitors or banks of inductors and capacitors, other methods of tuning or adjusting the wireless power transfer system may be used alone.

Unless defined otherwise, all technical and/or scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this disclosure belongs. In case of conflict with publications, patent applications, patents, and other references mentioned or incorporated herein by reference, the present specification, including definitions, will control.

Any of the above described features may be used alone or in combination without departing from the scope of the present disclosure. Other features, objects and advantages of the systems and methods disclosed herein will be apparent from the following detailed description and accompanying drawings.

Description of drawings

Figures 1(a) and (b) depict an exemplary wireless power system comprising a source resonator 1 and a device resonator 2 separated by a distance D.

FIG. 2 shows an example resonator tagged according to the tagging convention described in this disclosure. Note that extraneous objects or additional resonators are not shown in the vicinity of resonator 1 .

FIG. 3 shows an exemplary resonator tagged according to the tagging convention described in this disclosure in the presence of a "loading" object.

FIG. 4 illustrates an exemplary resonator tagged according to the tagging convention described in this disclosure in the presence of a "disturbing" object.

的绘图。Figure 5 shows the efficiency η versus the strong coupling coefficient

drawing.

Figure 6(a) shows a circuit diagram of an example of a resonator, (b) shows a diagram of an example of a capacitively loaded inductor loop magnetic resonator, and (c) shows self-resonance with distributed capacitance and inductance A diagram of a coil, (d) shows a simplified diagram of electric and magnetic field lines associated with an exemplary magnetic resonator of the present disclosure, and (e) shows a diagram of one example of an electric resonator.

Figure 7 shows a graph of the "quality factor" Q (solid line) as a function of frequency for an exemplary resonator that may be used for wireless power transfer at MHz frequencies. Absorptive Q (dashed line) increases with frequency, while radiative Q (dotted line) decreases with frequency, thus forcing the total Q to peak at a specific frequency.

Figure 8 shows a diagram of a resonator structure with its characteristic dimensions, thickness and width indicated.

Figures 9(a) and (b) show diagrams of exemplary induction loop elements.

Figures 10(a) and (b) show two examples of trace structures formed on a printed circuit board and used to implement inductive elements in a magnetic resonator structure.

Figure 11(a) shows a perspective view of a planar magnetic resonator, (b) shows a perspective view of two planar magnetic resonators with various geometries, and (c) shows two Perspective view of a planar magnetic resonator.

Fig. 12 is a perspective view of an example of a planar magnetic resonator.

Figure 13 is a perspective view of a planar magnetic resonator arrangement with circular resonator coils.

Fig. 14 is a perspective view of an active area of a planar magnetic resonator.

Figure 15 is a perspective view of an application of the wireless power transfer system where a source at the center of a table powers multiple devices placed around the source.

Figure 16(a) shows a 3D finite element model of a copper and magnetic material structure driven by a square loop of current around a choke point at its center. In this example, the structure could consist of two boxes made of a conductive material such as copper, covered by a layer of magnetic material, and connected by a piece of magnetic material. The interior of the two conductive boxes in this example will be shielded from the AC electromagnetic fields generated outside the boxes, and can accommodate lossy objects that may degrade the Q of the resonator or sensitive components that may be negatively affected by the AC electromagnetic field. Also shown are the calculated magnetic field streamlines generated by this structure, indicating that the magnetic field lines tend to follow lower reluctance paths in the magnetic material. Fig. 16(b) shows the interaction between two identical structures as shown in Fig. (a), as indicated by the calculated magnetic field streamlines. Due to symmetry and to reduce computational complexity, only one half of the system was modeled (however, the calculation assumed a symmetrical arrangement of the other half).

Figure 17 shows an equivalent circuit representation of a magnetic resonator comprising a wire wound N times around a structure, possibly containing a magnetically permeable material. The inductance is implemented using a conductive loop wound around a structure including a magnetic material, and the resistor represents the loss mechanism in the system (R _wire is for resistive losses in the loop, R _μ represents the equivalent series connection of the structure surrounded by the loop resistance). Losses can be minimized to achieve high-Q resonators.

Figure 18 shows a finite element method (FEM) simulation of two high conductivity surfaces above and below a disc composed of lossy dielectric material in an external magnetic field at a frequency of 6.78 MHz. Note that the magnetic field is uniform before the disk and conductive material is introduced into the simulated environment. Perform this simulation in a cylindrical coordinate system. The image is azimuthal symmetric about the r=0 axis. A lossy electrolyte disc has ε _r =1 and σ = 10 S/m.

Figure 19 shows a diagram of a magnetic resonator with a lossy object in its vicinity completely covered by a high conductivity surface.

Figure 20 shows a diagram of a magnetic resonator with a lossy object in its vicinity partially covered by a high conductivity surface.

Figure 21 shows a diagram of a magnetic resonator with in its vicinity a lossy object disposed above a high conductivity surface.

Figure 22 shows an illustration of a fully wireless projector.

Figure 23 shows the magnitudes of the electric and magnetic fields along the diameter containing the circular loop inductor and along the axis of the loop inductor.

Fig. 24 shows a magnetic resonator and its housing and the necessary but necessary but not limited to placement (a) in the corners of the housing, as far away from the resonator structure as possible or (b) on the center of the surface enclosed by the inductive elements in the magnetic resonator. A graph of lossy objects.

Figure 25 shows a diagram of a magnetic resonator with a high conductivity surface above it and a lossy object that can be brought into the vicinity of the resonator, but above the high conductivity sheet.

Figure 26(a) shows an axially symmetric FEM of a thin conductive (copper) cylinder or disk (20 cm in diameter and 2 cm in height) exposed to an initially uniform applied magnetic field (grey flux lines) along the z-axis simulation. The axis of symmetry is at r=0. The magnetic current lines shown originate at z=-∞, where they are spaced at 1 cm intervals from r=3 cm to r=10 cm. Axis scales are in meters. Figure 26(b) shows the same structure and applied field as in (a), except that the conductive cylinder has been modified to include on its outer surface a 0.25mm layer of magnetic material (not visible). Note that the magnetic current lines are deflected significantly less away from the cylinder than in (a).

FIG. 27 shows an axisymmetric view based on a variation of the system shown in FIG. 26 . Lossy materials have only one surface covered by a layered structure of copper and magnetic materials. As shown, the inductor loop is placed on the side of the copper and magnetic material structure opposite the lossy material.

Figure 28(a) depicts a general topology of a matching circuit including indirect coupling to a high-Q inductive element.

Figure 28(b) shows a block diagram of a magnetic resonator including a conductor loop inductor and a tunable impedance network. Physical electrical connections to this resonator may be made to terminal connections.

Figure 28(c) depicts the general topology of a matching circuit coupled directly to a high-Q inductive element.

Figure 28(d) depicts the general topology of a symmetric matching circuit coupled directly to a high-Q inductive element and driven anti-symmetrically (balanced drive).

Figure 28(e) depicts the general topology of a matching circuit coupled directly to a high-Q inductive element and grounded (unbalanced drive) at the symmetry point of the main resonator.

Figures 29(a) and 29(b) depict two topologies of a matching circuit transformer coupled (ie, indirectly or inductively) to a high-Q inductive element. The highlighted part of _{the Smith} chart in (c) depicts the complex impedance (from _the inductive element's L and R produced).

Figure 30(a), (b), (c), (d), (e), (f) depict six topologies of matching circuits that are directly coupled to a high-Q inductive element and include a capacitor in series with _Z structure. The topology shown in Figure 30(a),(b),(c) is driven with a common-mode signal at the input terminal, while the topology shown in Figure 30(d),(e),(f) is symmetrical , and receive balanced drive. The highlighted portion of the Smith chart in Figure 30(g) depicts complex impedances that can be matched by these topologies. Figures 30(h), (i), (j), (k), (l), (m) depict six of the matching circuits that are directly coupled to high-Q inductive elements and include an inductor in series with _Z0. Topology.

Figure 31 (a), (b), (c) depicts three topologies of matching circuits that are directly coupled to a high-Q inductive element and include a capacitor in series with _Z , grounded at the center point of the capacitor and receiving unbalanced drive. The highlighted portion of the Smith chart in Figure 31(d) depicts the complex impedances that can be matched by these topologies. 31(e), (f), (g) depict three topologies of matching circuits that are directly coupled to high-Q inductive elements and include an inductor in series with Z ₀ .

Figures 32(a), (b), (c) depict three topologies of matching circuits that are directly coupled to a high-Q inductive element and include a capacitor in series with Z ₀ . It is grounded through a tap at the center point of the inductor loop and receives an unbalanced drive. The highlighted portion of the Smith chart in (d) depicts complex impedances that can be matched by these topologies, and (e), (f), (g) depict _the Three topologies of inductor matching circuits.

Figure 33(a), (b), (c), (d), (e), (f) depict six topologies of matching circuits that are directly coupled to a high-Q inductive element and include a capacitor in parallel with _Z structure. The topology shown in Figure 33(a),(b),(c) is driven with a common-mode signal at the input terminal, while the topology shown in Figure 33(d),(e),(f) is symmetrical , and receive balanced drive. The highlighted portion of the Smith chart in Figure 33(g) depicts complex impedances that can be matched by these topologies. Figure 33(h), (i), (j), (k), (l), (m) depicts six of the matching circuits that are directly coupled to high-Q inductive elements and include an inductor in parallel with Z ₀ Topology.

34(a), (b), (c) depict three topologies of matching circuits that are directly coupled to high-Q inductive elements and include capacitors in parallel with Z ₀ . It is grounded at the center point of the capacitor and receives an unbalanced drive. The highlighted portion of the Smith chart in (d) depicts complex impedances that can be matched by these topologies. 34(e), (f), (g) depict three topologies of matching circuits that are directly coupled to high-Q inductive elements and include an inductor in parallel with Z ₀ .

Figures 35(a), (b), (c) depict three topologies of matching circuits that are directly coupled to a high-Q inductive element and include a capacitor in parallel with Z ₀ . It is grounded through a tap at the center point of the inductor loop and receives an unbalanced drive. The highlighted portions of the Smith charts in Figure 35(d), (e) and (f) depict complex impedances that can be matched by these topologies.

Figure 36(a), (b), (c), (d) depict fixed and variable capacitances designed to produce a total variable capacitance with finer tuning resolution and some with reduced voltage across the variable capacitors. Four network topologies of variable capacitors.

Figures 37(a) and 37(b) depict two network topologies of fixed capacitors and variable inductors designed to produce a total variable capacitance.

38 depicts a high-level block diagram of a wireless power transfer system.

39 depicts a block diagram of an exemplary wirelessly powered device.

40 depicts a block diagram of sources of an exemplary wireless power transfer system.

Fig. 41 shows an equivalent circuit diagram of a magnetic resonator. A dash through a capacitor symbol indicates that the represented capacitor may be fixed or variable. The port parameter measurement circuit can be configured to measure certain electrical signals, and can measure the magnitude and phase of the signal.

Figure 42 shows a circuit diagram of a magnetic resonator in which a tunable impedance network is implemented with voltage controlled capacitors. Such embodiments may be adjusted, tuned or controlled by circuitry including programmable or controllable voltage sources and/or computer processors. The voltage control capacitor may be adjusted in response to data measured by the port parameter measurement circuit and processed by the measurement analysis and control algorithm and hardware. The voltage controlled capacitor may be a switched capacitor bank.

Fig. 43 shows an end-to-end wireless power transfer system. In this example, both the source and the device contain port measurement circuits and processors. The box labeled "Coupler/Switch" indicates that the port measurement circuit can be connected to the resonator by a directional coupler or switch, enabling source and device resonator measurements, adjustments to be made in conjunction with the power transfer function or separately and control.

Fig. 44 shows an end-to-end wireless power transfer system. In this example, only the source contains the port measurement circuit and processor. In this case, the device resonator operating characteristics may be fixed, or may be adjusted by analog control circuitry, and no processor-generated control signals are required.

Fig. 45 shows an end-to-end wireless power transfer system. In this example, both the source and the device contain port measurement circuitry, but only the source contains the processor. Data from the device is communicated over a wireless communication channel which can be implemented with a separate antenna or through some modulation of the source driving signal.

Fig. 46 shows an end-to-end wireless power transfer system. In this example, only the source contains the port measurement circuit and processor. Data from the device is communicated over a wireless communication channel which can be implemented with a separate antenna or through some modulation of the source driving signal.

Figure 47 shows a coupled magnetic resonator whose frequency and impedance can be automatically adjusted using an algorithm implemented with a processor or computer.

Figure 48 shows a varactor array.

Figure 49 shows a device (laptop) wirelessly powered or charged by a source, where the source and device resonator is physically separate from, but electrically connected to, the source and device.

Figure 50(a) is an illustration of a laptop application being powered or charged wirelessly, with the device resonator inside the laptop housing and not visible.

Figure 50(b) is an illustration of a laptop application being powered and charged wirelessly, with the resonator under the laptop base and electrically connected to the laptop power input by a cable.

Figure 50(c) is an illustration of a laptop computer application being powered or charged wirelessly, where the resonator is attached to the laptop computer base.

Figure 50(d) is an illustration of a laptop computer application being powered and charged wirelessly, where the resonator is attached to the laptop computer display.

Figure 51 is an illustration of a rooftop PV panel with wireless power transfer.

Detailed ways

As noted above, the present disclosure relates to coupled electromagnetic resonators having long-lived oscillatory resonant modes that can wirelessly transfer power from a power source to a power sink. However, the present technique is not limited to electromagnetic resonators, but is general and can be applied to a variety of resonators and resonance objects. Therefore, we first describe the general technique and then disclose electromagnetic examples for wireless energy transfer.

resonator

A resonator can be defined as a system capable of storing energy in at least two different forms, and in which the stored energy oscillates between the two forms. A resonance has a specific mode of oscillation with a resonant (modal) frequency f and a resonant (modal) field. The angular resonance frequency ω can be defined as ω=2πf, the resonance wavelength λ can be defined as λ=c/f, where c is the speed of light, and the resonance period T can be defined as T=1/f=2π/ω. In the absence of loss mechanisms, coupling mechanisms, or external energy supply or consumption mechanisms, the total resonator stored energy W will remain fixed, and the two energy forms will oscillate, one of which will be a maximum while the other is a minimum value, and vice versa.

In the absence of extraneous materials or objects, energy in the resonator 102 shown in FIG. 1 may attenuate or be lost by inherent losses. The resonator field obeys the following linear equation:

$\frac{\mathrm{<span\; class="notranslate">\; da</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; dt</span>}}<span\; class="notranslate">\; =</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; \&omega;</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; i\&Gamma;</span>}<span\; class="notranslate">\; )</span>\mathrm{<span\; class="notranslate">\; a</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; ,</span>$

where the variable a(t) is the resonant field amplitude, defined such that the energy contained within the resonator is given by |a(t)| ² . Γ is the rate of intrinsic energy attenuation or loss (eg due to absorption and radiation losses).

The quality factor or Q-factor or Q of a resonator, which characterizes energy decay, is inversely proportional to these energy losses. It can be defined as Q=ω*W/P, where P is the time-averaged power lost in steady state. That is, a resonator 102 with a high Q has relatively low intrinsic losses and is capable of storing energy for a relatively long time. Since a resonator loses energy at its intrinsic decay rate 2Γ, its Q, also called its intrinsic Q value, is given by Q=ω/2Γ. The quality factor also expresses the number of oscillation periods T which cause the energy in the resonator to decay by a factor of e.

As mentioned above, we define the quality factor, or Q, of a resonator as being due only to intrinsic loss mechanisms. A subscript such as Q ₁ indicates the resonator that Q refers to (resonator 1 in this case). Figure 2 shows an electromagnetic resonator 102 labeled according to this convention. Note that in this figure, there are no extraneous objects or additional resonators in the vicinity of resonator 1.

Depending on various factors such as the distance between the resonator and the object or other resonator, the material composition of the object or other resonator, the structure of the first resonator, the power in the first resonator, etc., the Extraneous objects and/or additional resonators may perturb or load the first resonator, thereby perturbing or loading the Q of the first resonator. Unintentional external energy loss in the vicinity of a resonator or coupling mechanisms to extraneous materials and objects may be referred to as "perturbing" the Q of a resonator and may be indicated by a subscript within parentheses ( ). The expected external energy loss associated with energy transfer via coupling to other resonators and generators and loads in the wireless energy transfer system can be referred to as "loading" the Q of the resonator and can be denoted by the square brackets [ ] to indicate the subscript.

The Q of the resonator 102 connected or coupled to the generator g or the load 302l may be referred to as the "loaded figure of merit" or "loaded Q" and as shown in _{FIG .} To represent. Typically, more than one generator or load 302 may be connected to the resonator 102 . However, we do not list those generators or loads individually, but use "g" and "l" to refer to the equivalent circuit loading imposed by the combination of generator and load. In a general description, we can use the subscript "l" to refer to the generator or load that is connected to the resonator.

In some discussions in this paper, we define the "loaded figure of merit" or "loaded Q" due to a generator or load connected to a resonator as δQ _[l] , where 1/δQ _[l] ≡ 1 /Q _[l] -1/Q. Note that the greater the loaded Q of the generator or load, ie, δQ _[l] , the less the loaded Q, ie, Q _[l] , deviates from the unloaded Q of the resonator.

In the presence of extraneous objects 402p not intended to be part of the energy transfer system, the Q of the resonator may be referred to as the "perturbed figure of merit" or "perturbed Q", and as shown in _{FIG. )} to represent. In general, there may be many extraneous objects denoted p1 , p2 etc. or a set of extraneous objects {p} that perturb the Q of the resonator 102 . In this case, the perturbed Q can be denoted as Q _(p1+p2+...) or Q _({p}) . For example, Q1 _(brick+wood) may represent the perturbed figure of merit of the first resonator in a system for wireless power exchange in the presence of a brick or a piece of wood, and Q2 _({office}) may represent Perturbed figure of merit of a second resonator in a system for wireless power exchange in an office environment.

In some discussions in this paper, we define the "perturbation figure of merit" or "perturbation Q" due to an unrelated object p as δQ _(p) , where 1/δQ _(p) ≡ 1/Q _(p) -1 /Q. As mentioned above, the perturbed figure of merit can be due to a number of irrelevant objects pi, p2, etc. or a set of irrelevant objects {p}. The larger the perturbed Q of the object, ie δQ _(p) , the less the perturbed Q, ie Q _(p), deviates from the unperturbed Q of the resonator.

In some discussions in this paper, we also define Θ _(p) ≡ Q _(p) /Q and refer to it as the "quality factor insensitivity" or "Q insensitivity" of a resonator in the presence of extraneous objects ". A subscript such as Θ _1(p) indicates the resonator referred to by the perturbed or unperturbed figure of merit, ie Θ _1(p) ≡Q _1(p) /Q ₁ .

Note that it is also possible to characterize the figure of merit Q as "unperturbed" when necessary to distinguish it from the perturbed figure of merit Q _(p) , and as "unloaded" when necessary to distinguish it from the loaded The quality factor Q _[l] distinguishes it. Likewise, the disturbed figure of merit Q _(p) can also be characterized as “unloaded” when necessary to distinguish it from the loaded and disturbed figure of merit Q _(p)[l] .

coupled resonator

By any part of their near fields being coupled, resonators having substantially the same resonant frequency can interact and exchange energy. There are various physical pictures and models that can be used to understand, design, optimize and characterize this energy exchange. One way to describe two coupled resonators and model the energy exchange between them is to use coupled mode theory (CMT).

In coupled-mode theory, the resonator field obeys the following system of linear equations:

$\frac{{\mathrm{<span\; class="notranslate">\; da</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; dt</span>}}<span\; class="notranslate">\; =</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; i</span>}{\mathrm{<span\; class="notranslate">\; \&Gamma;</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}}<span\; class="notranslate">\; )</span>{\mathrm{<span\; class="notranslate">\; a</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; i</span>}\underset{\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; \&NotEqual;</span>\mathrm{<span\; class="notranslate">\; m</span>}}{\mathrm{<span\; class="notranslate">\; \&Sigma;</span>}}{\mathrm{<span\; class="notranslate">\; \&kappa;</span>}}_{\mathrm{<span\; class="notranslate">\; mn</span>}}{\mathrm{<span\; class="notranslate">\; a</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>$

出现，所以我们使用κ_mn来表示实耦合系数的绝对值。where the indices denote different resonators, and κ _mn is the coupling coefficient between resonators. For an inverse system, the coupling coefficient may obey the relation κ _mn =κ _nm . Note that for the purposes of this specification far-field radiation interference effects will be ignored and the coupling coefficients will therefore be considered real-valued. Furthermore, since in all subsequent calculations of system performance in this note, the coupling coefficient is almost squared by

appears, so we use κ _mn to denote the absolute value of the real coupling coefficient.

与谐振器m和n之间的所谓耦合因数κ_mn相关。我们用

将“强耦合系数”U_mn定义为谐振器m和n之间的耦合和损耗率的比率。Note that the coupling coefficient κ _mn from the above CMT via

Depends on the so-called coupling factor κ _mn between resonators m and n. we use

Define the "strong coupling coefficient" _Umn as the ratio of the coupling and loss rates between resonators m and n.

In a similar manner to the loading of a resonator by a connected power generating or consuming device, the resonator m in the presence of a resonator n or an additional resonator of similar frequency can be derived from this resonator n or an additional resonator The quality factor is loaded. The fact that resonator m can be loaded by resonator n and vice versa is just a different way of looking at the resonators being coupled.

The loaded Q of the resonator in these cases can be denoted as Q _m[n] and Q _n[m] . For multiple resonators or loading sources or devices, this can be determined by modeling each load as a resistive loss and summing the multiple loads in appropriate parallel and/or series combinations to determine the equivalent load of the ensemble The total loading of the resonator.

In some discussions in this paper, we define the "loaded quality factor" or "loaded Q _m " of resonator m due to resonator n as δQ _m[n] , where 1/δQ _m[n] ≡ 1 /Q _m[n] -1/Q _m . Note that resonator m also loads resonator n, and its "loading Q _n " is given by 1/δQ _n[m] ≡ 1/Q _n[m] - 1/Q _n .

When one or more resonators are connected to a generator or load, the linear equations are modified to:

$\frac{{\mathrm{<span\; class="notranslate">\; da</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; dt</span>}}<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; i</span>}<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; i</span>}{\mathrm{<span\; class="notranslate">\; \&Gamma;</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}}<span\; class="notranslate">\; )</span>{\mathrm{<span\; class="notranslate">\; a</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; i</span>}\underset{\mathrm{<span\; class="notranslate">\; no</span>}<span\; class="notranslate">\; \&NotEqual;</span>\mathrm{<span\; class="notranslate">\; m</span>}}{\mathrm{<span\; class="notranslate">\; \&Sigma;</span>}}{\mathrm{<span\; class="notranslate">\; \&kappa;</span>}}_{\mathrm{<span\; class="notranslate">\; mn</span>}}{\mathrm{<span\; class="notranslate">\; a</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; \&kappa;</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}}{\mathrm{<span\; class="notranslate">\; a</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\sqrt{\mathrm{<span\; class="notranslate">\; 2</span>}{\mathrm{<span\; class="notranslate">\; \&kappa;</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}}}{\mathrm{<span\; class="notranslate">\; the\; s</span>}}_{<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; m</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>$

${\mathrm{<span\; class="notranslate">\; the\; s</span>}}_{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; m</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\sqrt{\mathrm{<span\; class="notranslate">\; 2</span>}{\mathrm{<span\; class="notranslate">\; \&kappa;</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}}}{\mathrm{<span\; class="notranslate">\; a</span>}}_{\mathrm{<span\; class="notranslate">\; m</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; the\; s</span>}}_{<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; m</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; ,</span>$

where s _+m (t) and s _-m (t) are the amplitudes of the fields from the generator into the resonator m and out of the resonator m back toward the generator or into the load, respectively, defined such that by | s _+m (t)| ² and |s _-m (t)| ² to give the power it carries. The loading factor κ _m relates to the rate at which energy is exchanged between the resonator m and the generator or load connected to it.

Note that the loading factor κ _m from the CMT above is related to the loading figure of merit δQ _m[l] defined earlier by δQ m _[ l] = ω _m /2κ _m .

We define the "strong loading factor" U _m[l] as the ratio U _m[l] = κ _m /Γ _m = Q _m /δQ _m[l] of the loading and loss rates of a resonator m.

Fig. 1(a) shows an example of two coupled resonators 1000, namely a first resonator 102S configured as a source resonator and a second resonator 102D configured as a device resonator. Energy can be transferred over the distance D between the resonators. The source resonator 102S may be driven by a power source or a generator (not shown). Work may be extracted from the device resonator 102D by a power consuming device or load (eg, a load resistor, not shown). Let's use the subscripts "s" for source, "d" for device, "g" for generator, and "l" for load, and since there are only two resonances in this example and κ _sd = κ _ds , let us drop the subscripts on κ _sd , k _sd , and U _sd , and denote them as κ, k, and U, respectively.

The generator can constantly drive the source resonator at a constant drive frequency f corresponding to the angular drive frequency ω, where ω=2πf.

In this case, the efficiency of power transfer from generator to load (via source and device resonator) η = |s _−d | ² /|s _{+ s} | ² is maximized under the condition that the source must be made The resonant frequency, equipment resonant frequency and generator drive frequency match, that is

ω _s = ω _d = ω.

Furthermore, the loading QδQs _[g] of the source resonator due to the generator must be matched (equal) to the loaded QQs _[dl] of the source resonator due to the device resonator and the load, and conversely, The loaded Q δQ _d[l] of the device resonator due to the load must be matched (equal) to the loaded Q Q _d[sg] of the device resonator due to the source resonator and generator, i.e.

δQ _s[g] =Q _s[dl] and δQ _d[l] =Q _d[sg] .

These equations determine the optimal loading rate through the generator's source resonator, and through the load's plant resonator as

${\mathrm{<span\; class="notranslate">\; u</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; [</span>\mathrm{<span\; class="notranslate">\; l</span>}<span\; class="notranslate">\; ]</span>}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; \&kappa;</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}<span\; class="notranslate">\; /</span>{\mathrm{<span\; class="notranslate">\; \&Gamma;</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; Q</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; \&delta;</span>}{\mathrm{<span\; class="notranslate">\; Q</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}<span\; class="notranslate">\; [</span>\mathrm{<span\; class="notranslate">\; l</span>}<span\; class="notranslate">\; ]</span>}<span\; class="notranslate">\; =</span>\sqrt{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="u"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">u</figure-callout></span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; =</span>\sqrt{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>{<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; \&kappa;</span>}<span\; class="notranslate">\; /</span>\sqrt{{\mathrm{<span\; class="notranslate">\; \&Gamma;</span>}}_{\mathrm{<span\; class="notranslate">\; the\; s</span>}}{\mathrm{<span\; class="notranslate">\; \&Gamma;</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}}<span\; class="notranslate">\; )</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; Q</span>}}_{\mathrm{<span\; class="notranslate">\; the\; s</span>}}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; \&delta;</span>}{\mathrm{<span\; class="notranslate">\; Q</span>}}_{\mathrm{<span\; class="notranslate">\; the\; s</span>}<span\; class="notranslate">\; [</span>\mathrm{<span\; class="notranslate">\; g</span>}<span\; class="notranslate">\; ]</span>}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; \&kappa;</span>}}_{\mathrm{<span\; class="notranslate">\; the\; s</span>}}<span\; class="notranslate">\; /</span>{\mathrm{<span\; class="notranslate">\; \&Gamma;</span>}}_{\mathrm{<span\; class="notranslate">\; the\; s</span>}}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; u</span>}}_{\mathrm{<span\; class="notranslate">\; the\; s</span>}<span\; class="notranslate">\; [</span>\mathrm{<span\; class="notranslate">\; g</span>}<span\; class="notranslate">\; ]</span>}<span\; class="notranslate">\; .</span>$

Note that the above frequency matching and Q matching conditions are collectively referred to as "impedance matching" in electrical engineering.

并且由

给出，如图5所示。请注意，耦合效率η在U大于0.2时大于1％，在U大于0.7时大于10％，在U大于1时大于17％，在U大于3时大于52％，在U大于9时大于80％，在U大于19时大于90％，并且在U大于45时大于95％。在某些应用中，可以将其中U＞1的操作区称为“强耦合”区。Under the above conditions, maximizing efficiency is a monotonically increasing function of only the strong coupling factor between the source and device resonator,

and by

given, as shown in Figure 5. Note that the coupling efficiency η is greater than 1% when U is greater than 0.2, greater than 10% when U is greater than 0.7, greater than 17% when U is greater than 1, greater than 52% when U is greater than 3, and greater than 80% when U is greater than 9 , greater than 90% when U is greater than 19, and greater than 95% when U is greater than 45. In some applications, the region of operation where U > 1 may be referred to as a "strongly coupled" region.

所以可以使用高Q的谐振器。每个谐振器的Q可以是高的。谐振器的Q的几何平均数

也可以是或替代地是高的。Since in some cases expect large

So high-Q resonators can be used. The Q of each resonator can be high. The geometric mean of the Q of the resonator

It may also or alternatively be high.

可以是谐振频率的强函数。可以优选地将谐振器的谐振频率选择为实现高Q而不是实现低Γ，因为这两个目标在两个单独的谐振频率区处是可实现的。The coupling factor k is a number between 0≤k≤1, and it can be independent (or nearly independent) of the resonant frequency of the source and device resonators, to be precise, it can be mainly controlled by their relative geometry and their coupling determined by the laws of physical decay of the field. In contrast, the coupling coefficient

can be a strong function of the resonant frequency. It may be preferable to choose the resonant frequency of the resonator to achieve high Q rather than low Γ because these two goals are achievable at two separate resonant frequency regions.

可以将两个耦合谐振器称为高Q谐振器的系统。A high-Q resonator can be defined as a resonator with Q > 100. Two coupled resonators may be referred to as a system of high-Q resonators when each resonator has a Q greater than 100, Q _s >100, and Q _d >100. In other embodiments, when the geometric mean of Q of the resonator is greater than 100,

Two coupled resonators may be referred to as a system of high-Q resonators.

Resonators can be named or numbered. It may be called a source resonator, device resonator, first resonator, second resonator, repeater resonator, etc. It should be understood that while two resonators are shown in FIG. 1 , in many of the examples below, other implementations may include three (3) or more resonators. For example, a single source resonator 102S may transfer energy to multiple device resonators 102D or multiple devices. Energy can be transferred from a first device to a second, then from a second device to a third, and so on. Multiple sources may transfer energy to a single device or multiple devices connected to a single device resonator or multiple devices connected to multiple device resonators. The resonator 102 can act alternately or simultaneously as a source, a device, or it can be used to relay power from a source at one location to a device at another location. The distance range of the wireless energy transfer system can be extended using the intermediate electromagnetic resonator 102 . Multiple resonators 102 can be daisy-chained together to exchange energy with a wide range of sources and devices over extended distances. High power levels can be split among multiple sources 102S, diverted to multiple devices and recombined at a remote location.

The analysis of a single source and a single device resonator can be extended to multiple source resonators and/or multiple device resonators and/or multiple intermediate resonators. In such an analysis, it may be concluded that a large strong coupling factor _Umn between at least some or all of the plurality of resonators is preferred for high system efficiency in wireless energy transfer. Again, embodiments may use sources, devices and intermediate resonators with high Q. The Q of each resonator can be high. The geometric mean of Q for pairs of resonators m and n for which a large _Umn is expected It may also or alternatively be high.

Note that there can be independent The condition of the object perturbs the strength of any or all of these mechanisms.

Continuing the convention from the previous section for notation, we describe k as the coupling factor in the absence of extraneous objects or materials. We denote the coupling factor p in the presence of extraneous objects as k _(p) and call it the "perturbed coupling factor" or "perturbed k". Note that the coupling factor k can also be characterized as "unperturbed" when necessary to distinguish it from the perturbed coupling factor k _(p) .

We define δk _(p) ≡ k _(p) −k, and we call this the "perturbation of the coupling factor" or "perturbation of k" due to an unrelated object p.

We also define β _(p) ≡ k _(p) /k, and we refer to this as "coupling factor insensitivity" or "k insensitivity". Subscripts such as β _12(p) indicate the resonators involved in the perturbed and unperturbed coupling factors, ie β _12(p) ≡k _12(p) /k ₁₂ .

并且我们将其称为“被扰动强耦合因数”或“被扰动U”。请注意，必要时还可以将强耦合因数U表征为“未扰动”，以与被扰动强耦合因数U_(p)区别开。请注意，必要时还可以将强耦合因数U表征为“未扰动”，以与被扰动强耦合因数U_(p)区别开。Likewise, we describe U as a strong coupling factor in the absence of extraneous objects. We denote the strong coupling factor p in the presence of unrelated objects as U _(p) ,

And we call it the "perturbed strong coupling factor" or "perturbed U". Please note that the strong coupling factor U can also be characterized as "unperturbed" to distinguish it from the perturbed strong coupling factor U _(p) if necessary. Please note that the strong coupling factor U can also be characterized as "unperturbed" to distinguish it from the perturbed strong coupling factor U _(p) if necessary.

We define δU _(p) ≡U _(p) -U and call it the "perturbation of the strong coupling factor" or "perturbation of U" due to an unrelated object p.

We also define Ξ(p)≡U(p)/U and refer to it as "strong coupling factor insensitivity" or "U insensitivity". Subscripts such as Ξ _12(p) indicate the resonators involved in the perturbed and unperturbed coupling factors, ie Ξ _12(p) ≡U _12(p) /U ₁₂ .

因此，在被无关对象扰动的无线能量交换的系统中，对于无线能量转移中的高系统效率而言，可能期望至少某些或全部的多个谐振器之间的大的扰动强耦合因数U_mn(p)。源、设备和/或中间谐振器可以具有高Q_(p)。The efficiency of energy exchange in a perturbed system can be given by the same formula that gives the efficiency of an unperturbed system, where all parameters such as strong coupling factor, coupling factor and quality factor are replaced by their perturbed equivalent parameters. For example, in a system for wireless energy transfer including one source and one device resonator, the optimal efficiency can be calculated as

Therefore, in a system of wireless energy exchange perturbed by extraneous objects, a large perturbed strong coupling factor _Umn between at least some or all of the multiple resonators may be desired for high system efficiency in wireless energy transfer _(p) . Sources, devices and/or intermediate resonators may have high Q _(p) .

Certain irrelevant perturbations are sometimes unfavorable for the perturbed strong coupling factor (via a large perturbation of the coupling factor or quality factor). Therefore, techniques can be used to reduce the impact of extraneous perturbations on the system and maintain the insensitivity to large strong coupling factors.

energy exchange efficiency

So-called "useful energy" in useful energy exchange is energy or power that must be transferred to one or more devices in order to power or charge the devices. The transfer efficiency corresponding to useful energy exchange can be system or application dependent. For example, a high power vehicle charging application that diverts several kilowatts of power may need to have an efficiency of at least 80% in order to supply a useful amount of power, resulting in a useful energy exchange sufficient to recharge the vehicle battery without significantly affecting the transfer system. The various components are heated. In certain consumer electronic device applications, useful energy exchange may include any energy transfer efficiency greater than 10%, or any other acceptable amount to keep a rechargeable battery "topped off" and running for a long time . For some wireless sensor applications, transfer efficiencies much less than 1% may be suitable for powering multiple low-power sensors from a single source located at a considerable distance from the sensors. For other applications where wired power transfer is not possible or practical, a wide range of transfer efficiencies may be acceptable for useful energy exchange and may be considered to supply useful power to devices in those applications. In general, an operating distance is any distance over which useful power exchange is maintained or can be maintained in accordance with the principles described herein.

Useful energy exchange for wireless energy transfer in powering or recharging applications can be efficient, highly efficient or sufficiently efficient as long as wasted energy levels, heat dissipation and associated field strengths are within tolerable limits. Tolerable limits may depend on application, environment and system location. Useful energy exchange for wireless energy transfer in powering or recharging applications can be efficient, highly efficient, or sufficiently efficient as long as the desired system performance can be achieved for reasonable cost constraints, weight constraints, size constraints, etc. . Efficient energy transfer can be determined relative to what can be achieved using conventional inductive techniques for non-high-Q systems. Energy transfer can then be defined as efficient, highly efficient, or sufficiently efficient if more energy is delivered than can be delivered within a similar distance or alignment offset by a similarly sized coil structure in a conventional inductive scheme .

小于1/Q_m(p)，1/Q_n(p)和k_mn(p)之中的近似最大值，就可以实现高效的能量交换。对于高效能量交换而言，Q匹配条件可以不像频率匹配条件那么严格。由发电机和/或负载而引起的谐振器的强加载因数U_m[l]可以偏离其最优值但仍具有足够高效的能量交换的程度取决于特定的系统，是否所有或某些发电机和/或负载都是Q不匹配的等。Note that even though certain frequency and Q matching conditions can optimize the system efficiency of energy transfer, these conditions do not need to be fully satisfied in order to have sufficiently efficient energy transfer for useful energy exchange. As long as the relative shift of the resonant frequency

The approximate maximum value among 1/Q _m(p) , 1/Q _n(p) and k _mn(p) can realize efficient energy exchange. For efficient energy exchange, the Q matching condition may not be as strict as the frequency matching condition. The degree to which the strong loading factor U _m[l] of a resonator due to generators and/or loads can deviate from its optimal value and still have sufficiently efficient energy exchange depends on the particular system, whether all or some generators and/or loads are Q mismatched etc.

Thus, the resonant frequencies of the resonators may not be perfectly matched, but matched within the above tolerances. The strong loading factors of at least some resonators due to generators and/or loads may not exactly match their optimal values. Voltage levels, current levels, impedance values, material parameters, etc. may not be at the exact values described in this disclosure, but will be within some acceptable tolerance of those values. System optimization may include cost, size, weight, complexity, etc. considerations in addition to efficiency, Q, frequency, strong coupling factors, etc. considerations. Certain system performance parameters, specifications and designs may be far from optimal in order to optimize other system performance parameters, specifications and designs.

In some applications, at least some system parameters may change in time, for example because components such as sources or equipment may move or age, or because loads may be variable, or because disturbances or environmental conditions are changing, etc. In these cases, at least some system parameters may need to be dynamically adjustable or tuneable in order to achieve acceptable matching conditions. All system parameters may be dynamically adjustable or tuned to achieve near optimal operating conditions. However, based on the above discussion, sufficiently efficient energy exchange can be achieved even if some system parameters are not variable. In some instances, at least some devices may not be dynamically adjusted. In some examples, at least some sources may not be dynamically adjusted. In some examples, at least some of the intermediate resonators may not be dynamically adjusted. In some instances, none of the system parameters can be dynamically adjusted.

electromagnetic resonator

The resonators used to exchange energy may be electromagnetic resonators. In such resonators, the intrinsic energy decay rate Γ _m is given by the absorption (or resistive) losses and radiation losses of the resonator.

The resonator can be constructed such that the energy stored by the electric field is primarily confined within the structure and the energy stored by the magnetic field is primarily in the region around the resonator. The energy exchange is then mediated mainly by the resonant magnetic near-field. These types of resonators may be referred to as magnetic resonators.

The resonator can be constructed such that the energy stored by the magnetic field is primarily confined within the structure and the energy stored by the electric field is primarily in the region around the resonator. The energy exchange is mediated mainly by the resonant electric near field. These types of resonators may be referred to as electrical resonators.

Note that the total electrical and magnetic energy stored by the resonator must be equal, but its localization can be quite different. In some cases, a resonator may be characterized or described using the ratio of the average electric field energy to the average magnetic field energy specified at a distance from the resonator.

The electromagnetic resonator may include an inductance element, a distributed inductance, or a combination of inductance with an inductance L, and a capacitive element, a distributed capacitance, or a combination of capacitance with a capacitance C. A minimal circuit model of the electromagnetic resonator 102 is shown in Fig. 6a. The resonator may include an inductive element 108 and a capacitive element 104 . Provided with initial energy such as electric field energy stored in capacitor 104, the system will oscillate as the capacitor discharges, transferring energy to magnetic field energy stored in inductor 108, which in turn transfers energy back to the energy stored in The electric field energy in capacitor 104.

The resonator 102 shown in FIG. 6(b)(c)(d) can be called a magnetic resonator. For wireless energy transfer applications in residential environments, magnetic resonators may be preferable because most everyday materials including animals, plants, and humans are non-magnetic (i.e., μ _r ≈ 1), so their interaction with magnetic fields is Minimal, and mainly due to eddy currents (which are second order effects) induced by time-varying magnetic fields. This feature is important for safety reasons, and because it reduces the potential for interaction with extraneous environmental objects and materials that could alter system performance.

Figure 6d shows a simplified diagram of certain electric and magnetic field lines associated with the exemplary magnetic resonator 102B. The magnetic resonator 102B may include a conductor loop acting as an inductive element 108 and a capacitive element 104 (at the ends of the conductor loop). Note that this figure depicts most of the energy stored in the region around the resonator in the magnetic field and most of the energy stored in the resonator (between the capacitor plates) in the electric field. Some electric fields due to fringing fields, free charges, and time-varying magnetic fields can be stored in the region around the resonator, but magnetic resonators can be designed so that the electric field is localized as close as possible to the resonator or at the within itself.

The inductor 108 and capacitor 104 of the electromagnetic resonator 102 may be bulk circuit elements, or the inductance and capacitance may be distributed and result from the way the conductors are formed, shaped and positioned in the structure. For example, as shown in FIG. 6(b)(c)(d), the inductor 108 may be implemented by shaping the conductors into a closed surface area. Such a resonator 102 may be referred to as a capacitively loaded loop inductor. Note that we can use the terms "loop" or "coil" generically to refer to an electrically conducting structure (wire, tube, strip, etc.) that encloses a surface of any shape and size with any number of turns. In Figure 6b, the enclosed surface area is circular, but the surface can be any of a variety of other shapes and sizes, and can be designed to achieve certain system performance specifications. As an example indicating how inductance scales with physical size, the inductance for a length of circular conductor arranged to form a circular single-turn loop is approximately

$\mathrm{<span\; class="notranslate">\; L</span>}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; \&mu;</span>}}_{\mathrm{<span\; class="notranslate">\; 0</span>}}\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; ln</span>}\frac{\mathrm{<span\; class="notranslate">\; 8</span>}\mathrm{<span\; class="notranslate">\; x</span>}}{\mathrm{<span\; class="notranslate">\; a</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; ,</span>$

where _μ0 is the magnetic permeability of free space, x is the radius of the closed circular surface area, and a is the radius of the conductor used to form the inductor loop. A more precise inductance value for the loop can be calculated analytically or numerically.

The inductance for other cross-sectional conductors arranged to form other closed surface shapes, areas, dimensions, etc. and having any number of turns may be calculated analytically or numerically, or may be determined by measurement. Inductance may be implemented using inductor elements, distributed inductance, networks, arrays, series and parallel combinations of inductors and inductors, and the like. Inductance can be fixed or variable and can be used to vary impedance matching and resonant frequency operating conditions.

There are various ways to achieve the capacitance required to achieve the desired resonant frequency of the resonator structure. Capacitor plates 110 may be formed and utilized as shown in Figure 6b, or capacitance may be distributed and implemented between adjacent windings of the multi-loop conductor 114 as shown in Figure 6c. Capacitance may be implemented using capacitor elements, distributed capacitance, networks, arrays, series and parallel combinations of capacitance, and the like. Capacitance can be fixed or variable and can be used to vary impedance matching and resonant frequency operating conditions.

It should be understood that the inductance and capacitance in the electromagnetic resonator 102 can be lumped, distributed, or a combination of lumped and distributed inductance and capacitance, and can be achieved through combinations of the various elements, techniques, and effects described herein. Implement an electromagnetic resonator.

Electromagnetic resonator 102 may include inductors, inductors, capacitors, capacitors, and additional circuit elements such as resistors, diodes, switches, amplifiers, diodes, transistors, transformers, conductors, connectors, and the like.

The resonant frequency of an electromagnetic resonator

Electromagnetic resonator 102 may have a characteristic, natural or resonant frequency determined by its physical properties. This resonant frequency is the energy stored in the resonator in the electric field W _E ( _WE = q ² /2C, where q is the charge on the capacitor C) and the energy stored in the magnetic field of the resonator W _B (W _B = Li ² /2, where i is the frequency of oscillation between the current through the inductor L). In the absence of any losses in the system, energy will be continuously exchanged between the electric field in capacitor 104 and the magnetic field in inductor 108 . The frequency at which this energy is exchanged may be called the characteristic frequency, natural frequency, or resonance frequency of the resonator, and is given by ω,

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; \&pi;f</span>}<span\; class="notranslate">\; =</span>\sqrt{\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; LC</span>}}}<span\; class="notranslate">\; .</span><span\; class="notranslate">\; .</span>$

The resonant frequency of the resonator can be changed by tuning the inductance L and/or capacitance C of the resonator. The resonant frequency can be designed to operate at the so-called ISM (Industrial, Scientific and Medical) frequencies specified by the FCC. The resonator frequency may be selected to meet certain field limit specifications, specific absorption rate (SAR) limit specifications, electromagnetic compatibility (EMC) specifications, electromagnetic interference (EMI) specifications, component size, cost or performance specifications, and the like.

Quality Factor of Electromagnetic Resonators

Energy in the resonator 102 shown in FIG. 6 may decay or be lost due to inherent losses including absorptive losses (also known as ohmic or resistive losses) and/or radiative losses. The quality factor, or Q, of a resonator, which characterizes energy attenuation, is inversely proportional to these losses. Absorptive losses may be caused by the finite conductivity of the conductors used to form the inductor, as well as losses in other elements, components, connectors, etc. in the resonator. Inductors formed from low loss materials may be referred to as "high Q inductive elements", and elements, components, connectors, etc. having low losses may be referred to as having "high resistance Q". In general, the total absorptive loss of a resonator can be calculated as an appropriate series and/or parallel combination of the resistive losses of the various elements and components making up the resonator. That is, in the absence of any significant radiative or component/connection losses, the Q of the resonator can be given by Q _abs ,

${\mathrm{<span\; class="notranslate">\; Q</span>}}_{\mathrm{<span\; class="notranslate">\; abs</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; \&omega;\; L</span>}}{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; abs</span>}}}<span\; class="notranslate">\; ,</span>$

where ω is the resonant frequency, L is the total inductance of the resonator, and the resistance of the conductor used to form the inductor can be given, for example, by _Rabs = lρ/A (l is the length of the wire, ρ is the conductor material resistivity, and A is the cross-sectional area where current flows in the wire). For alternating current, the cross-sectional area over which the current flows can be smaller than the physical cross-sectional area of the conductor due to the skin effect. Thus, high-Q magnetic resonators can be composed of conductors with high conductivity, relatively large surface area, and/or with specially designed profiles (eg, litz wires) to minimize proximity effects and reduce AC resistance.

Magnetic resonator structures may include high-Q inductive elements composed of highly conductive wires, coated wires, litz wires, strips, strips or plates, tubes, paints, gels, traces, and the like. A magnetic resonator may be self-resonant, or it may include external coupling elements such as capacitors, inductors, switches, diodes, transistors, transformers, and the like. Magnetic resonators can include distributed and lumped capacitance and inductance. In general, the Q of a resonator will be determined by the Q of all the individual components of the resonator.

Since Q is proportional to the inductance L, the resonator can be designed to increase L within certain other constraints. For example, one way to increase L is to use more than one turn of the conductor to form the inductor in the resonator. Design techniques and tradeoffs can depend on the application, and there are a variety of structures, conductors, components, and resonant frequencies that can be chosen in the design of a high-Q magnetic resonator.

In the absence of significant absorption losses, the Q of a resonator can be determined primarily by radiation losses and is given by Q _rad =ωL/R _rad , where R _rad is the radiation loss of the resonator and can be determined by The size of a resonator relative to the frequency ω or wavelength λ of operation. For the magnetic resonators discussed above, the radiation loss can scale with R _rad ~ (x/λ) ⁴ (characteristic of magnetic dipole radiation), where x is the characteristic dimension of the resonator, such as Fig. 6b The radius of the inductive element shown, and where λ=c/f, where c is the speed of light and f is as defined above. The size of the magnetic resonator can be much smaller than the wavelength of operation, so radiation losses can be very small. Such structures may be referred to as sub-wavelength resonators. Radiation can be a loss mechanism for non-radiative wireless energy transfer systems, and design choices can be made to reduce or minimize R _rad . Note that for non-radiative wireless energy transfer schemes, a high Q _rad may be desirable.

Note also that resonators designed for nonradiative wireless energy transfer are different from antennas designed for communication or far-field energy transfer purposes. In particular, capacitively loaded conductive loops can be used as resonant antennas (e.g., in cellular telephones), but for those operating in the far-field region, where the radiation Q is deliberately designed to be small so that the antenna is under radiated energy is efficient. Such designs are not suitable for the efficient near-field wireless energy transfer techniques disclosed in this application.

The quality factor of a resonator including both radiation and absorption losses is Q=ωL(R _abs +R _rad ). Note that there may be a maximum Q value for a particular resonator, and that a resonator may be designed with special considerations for the size of the resonator, the materials and components used to construct the resonator, the frequency of operation, the connection mechanism, etc., so that Realize high-Q resonators. Figure 7 shows an exemplary magnetic resonator (in this case a coil with a diameter of 60 cm made of copper tubing with an outer diameter (OD) of 4 cm) that can be used for wireless power transfer at MHz frequencies A plot of Q. Absorptive Q (dashed line) 702 increases with frequency, while radiative Q (dotted line) 704 decreases with frequency, thus forcing the overall Q to peak 708 at a particular frequency. Note that the Q of this exemplary resonator is greater than 100 over a large frequency range. Magnetic resonators can be designed to have a high Q within a certain frequency range, and the system operating frequency can be set to any frequency within this range.

When describing a resonator in terms of loss rate, Q can be defined using the intrinsic decay rate, 2Γ, as mentioned earlier. The intrinsic decay rate is the rate at which an uncoupled and undriven resonator loses energy. For the magnetic resonator described above, the intrinsic loss can be given by Γ=(R _abs +R _rad )/2L, and the quality factor Q of the resonator can be given by Q=ω/2Γ.

Note that a quality factor that is only related to a specific loss mechanism can be expressed as Q _mechanism (if no resonator is specified) or Q1 _{, mechanism} (if a resonator is specified (for example, resonator 1)). For example, Q1 _,rad is the quality factor for resonator 1, related to its radiation loss.

Electromagnetic Resonator Near Field

High-Q electromagnetic resonators used in the near-field wireless energy transfer systems disclosed herein may be sub-wavelength objects. That is, the physical size of the resonator can be much smaller than the wavelength corresponding to the resonant frequency. Sub-wavelength magnetic resonators can store most of the energy in the region surrounding the resonator in their magnetic near-fields, and these fields can also be described as stationary or non-propagating because they do not radiate away from the resonator. The extent of the near-field in the region around the resonator is generally set by the wavelength, so for sub-wavelength resonators the extent of the near-field can extend well beyond the resonator itself. A confining surface in which the field properties change from near-field properties to far-field properties may be called a "radiation caustic".

The strength of the near field decreases the farther away from the resonator. Although the field strength of the near field of a resonator decays away from the resonator, the field can still interact with objects entering the general vicinity of the resonator. The extent to which a field interacts depends on a variety of factors, some of which can be controlled and designed, and some of which cannot. The wireless energy transfer scheme described herein can be achieved when the distance between the coupled resonators is such that one resonator is within the radiative caustic surface of the other.

The near-field distribution of an electromagnetic resonator may be similar to those generally associated with dipole resonators or oscillators. Such field distributions can be described as omnidirectional, meaning that the magnitude of the field is non-zero in all directions away from the object.

Characteristic Dimensions of Electromagnetic Resonators

Spatially separated and/or offset magnetic resonators of sufficient Q can enable efficient wireless energy transfer over much greater distances than seen in the prior art, even if the size and shape of the resonator structures are different. Such resonators can also be operated to achieve more efficient energy transfer over shorter range distances than is achievable with the aforementioned techniques. We describe such resonators as capable of mid-range energy transfer.

A mid-range distance may be defined as a distance larger than the characteristic dimension of the smallest one of the resonators involved in the transfer, where the distance is measured from the center of one resonator structure to the center of a second spatially separated resonator structure. In this definition, two-dimensional resonators are spatially separated when the regions bounded by their inductive elements do not intersect, and three-dimensional resonators are spatially separated when their volumes do not intersect. When the region bounded by the two-dimensional resonator is outside the volume of the three-dimensional resonator, the former is spatially separated from the latter.

Figure 8 shows some exemplary resonators with their characteristic dimensions marked. It will be appreciated that the characteristic dimension 802 of the resonator 102 may be defined in terms of the size of the conductor and the area bounded or enclosed by the inductive element in a magnetic resonator and the length of the conductor forming the capacitive element of an electric resonator. The characteristic dimension _802xchar of the resonator 102 may then be equal to the radius of the smallest sphere that can fit around the inductive or capacitive elements of the magnetic or electric resonator, respectively, and the center of the resonator structure is the center of the sphere. The characteristic thickness 804t _char of the resonator 102 may be the smallest possible height of the highest point of the respective inductive or capacitive element of the magnetic or capacitive resonator measured from the flat surface on which it rests. The characteristic width 808w _char of the resonator 102 may be the radius of the smallest possible circle through which the respective inductive or capacitive elements of a magnetic or electric resonator may pass while traveling along a straight line. For example, the characteristic width 808 of a cylindrical resonator may be the radius of the cylinder.

In the wireless energy transfer technique of the present invention, energy can be efficiently exchanged over a large distance range, but the technique is designed to exchange useful energy over medium-range distances and between resonators with different physical sizes, components, and orientations in order to Notable for its ability to power or recharge devices. Note that although k may be small in these cases, strong coupling and efficient energy transfer can be achieved by using high-Q resonators to achieve high U, That is, an increase in Q can be used to at least partially overcome a decrease in k to maintain useful energy transfer efficiencies.

Note also that while the near-field of a single resonator can be described as omnidirectional, the efficiency of energy exchange between two resonators can depend on the relative position and orientation of the resonators. That is, the efficiency of energy exchange can be maximized for a particular relative orientation of the resonators. The sensitivity of the transfer efficiency to the relative position and orientation of the two uncompensated resonators can be captured in the calculation of k or κ. While coupling can be achieved between resonators that are offset and/or rotated relative to each other, the efficiency of the exchange can depend on the details of the positioning and any feedback, tuning and compensation techniques implemented during operation.

High Q Magnetic Resonator

和R_rad＝π/6·η_oN²(ωx/c)⁴，其中，ρ是导体材料的电阻率且η_o≈120πΩ是自由空间的阻抗。用于此类N匝环路的电感L约为先前给出的单匝环路的电感的N²倍。此类谐振器的品质因数Q＝ωL/(R_abs+R_rad)对于由系统参数(图4)确定的特定频率而言是最高的。如前所述，在较低频率处，主要由吸收损耗来确定Q，并且在较高频率处，主要由辐射损耗来确定Q。In the near-field region (x ≤ λ) of a subwavelength capacitively loaded loop magnetic resonator, the resistance associated with a circular conductive loop inductor consisting of N turns of wire (whose radius is greater than the skin depth) is about

and R _rad =π/6·η _o N ² (ωx/c) ⁴ , where p is the resistivity of the conductor material and η _o ≈120πΩ is the impedance of free space. The inductance L for such an N-turn loop is approximately ^N2 times that of the single-turn loop given previously. The quality factor Q=ωL/(R _abs +R _rad ) of such a resonator is highest for a specific frequency determined by the system parameters ( FIG. 4 ). As mentioned earlier, at lower frequencies, Q is primarily determined by absorption losses, and at higher frequencies, Q is primarily determined by radiation losses.

Note that the formulas given above are approximate and are intended to illustrate the functional dependence of _Rabs , _Rrad and L on the physical parameters of the structure. More accurate numerical calculations of these parameters taking into account deviations from strict quasi-static limits (eg non-uniform current/charge distribution along conductors) may be useful for precise design of resonator structures.

Note that absorptive losses can be minimized by using low loss conductors to form the inductive element. For example, by using large surface area conductors such as inductor tubes, strips, strips, machined objects, plates, etc., by using special designs such as Litz wire, braided wire, wires of any cross-section and other conductors with low proximity losses Conductors (in which case the above frequency scaling properties may also be different) and minimize conductor losses by using low resistivity materials such as high purity copper and silver. One advantage of using conductive tubing as a conductor at higher operating frequencies is that it can be cheaper and lighter than a solid conductor of similar diameter, and can have similar resistance since most of the current flows along the outer surface of the conductor due to the skin effect. surface travel.

In order to obtain a rough estimate of the design of achievable resonators made of copper wire or tube and suitable for operation in the microwave region, ^it is possible to calculate ) optimal Q and resonant frequency of a resonator composed of a circular inductive element (N=1). Then, for an inductive element with characteristic dimension x = 1 cm and conductor diameter a = 1 mm (e.g. suitable for a cellular phone), the quality factor peaks at Q = 1225 when f = 380 MHz. For x = 30cm and a = 2mm, Q = 1103 at f = 17MHz for an inductive element size that might be suitable for a laptop computer or a home robot. For larger source inductive elements, eg possibly located in the ceiling, x = lm and a = 4mm, at f = 5MHz, Q can be as high as Q = 1315. Note that many practical examples give an expected figure of merit of Q ≈ 1000-1500 at λ/x ≈ 50-80. Measurements of a wider variety of coil shapes, sizes, materials and operating frequencies than described above show that Q > 100 can be achieved for a variety of magnetic resonator structures using commonly available materials.

As mentioned above, the rate for energy transfer between two resonators having characteristic dimensions _x1 and _x2 separated by a distance D between their centers can be given by κ. To give an example of how the definition parameters scale, consider the cell phone, laptop, and ceiling resonator example from above at three (3) distances; D/x=10,8,6. In the example considered here, the source and device resonators are the same size (x ₁ =x ₂ ) and shape, and are oriented as shown in Fig. 1(b). In the cellular phone example, ω/2κ=3033, 1553, 655, respectively. In the laptop example, ω/2κ=7131, 3651, 1540, and for the ceiling resonator example, ω/2κ=6481, 3318, 1400, respectively. The corresponding coupling loss ratios peak at frequencies where the inductive element Q peaks, and, for the three inductive element sizes and distances described above, κ/Γ = 0.4, 0.79, 1.97, and 0.15, 0.3, 0.72, and 0.2 , 0.4, 0.94. An example of using different sized inductive elements is an x ₁ =1 m inductor (eg a source in the ceiling) and an x ₂ =30 cm inductor (eg a home robot on the floor) separated by a distance D = 3m (eg room height). In this example, for an efficiency of approximately 14%, at an optimum operating frequency of f = 6.4 MHz, the figure of merit for strong coupling, Here, the optimum system operating frequency is between the peaks of the individual resonators Q.

Inductive elements used in high-Q magnetic resonators can be formed. We have demonstrated a variety of high-Q magnetic resonators based on copper conductors formed as closed-surface inductive elements. Inductive elements can be formed using a variety of conductors arranged in a variety of shapes (enclosing an area of any size or shape), and they can be single-turn or multi-turn elements. A diagram of exemplary inductive elements 900A-B is shown in FIG. 9 . Inductive elements may be formed to enclose circles, rectangles, squares, triangles, shapes with rounded corners, shapes that follow the contours of particular structures and devices, shapes that follow, fill, or utilize dedicated spaces within structures or devices, and the like. Designs can be optimized for size, cost, weight, appearance, performance, and the like.

These conductors can be bent or formed to a desired size, shape and number of turns. However, it can be difficult to accurately reproduce conductor shape and size using manual techniques. Additionally, it may be difficult to maintain a uniform or desired center-to-center spacing between conductor segments in adjacent turns of the inductive element. For example, accurate or uniform spacing may be important in determining the self-capacitance of the structure as well as any proximity effect induced increase in AC resistance.

Molds can be used to replicate inductor elements for high-Q resonator designs. Additionally, dies can be used to accurately form conductors into any kind of shape without creating kinks, buckles, or other potentially detrimental effects in the conductors. Molds can be used to form inductor elements, and the inductor elements can then be removed from these forms. Once removed, these inductive elements can be built into housings or devices that can house high-Q magnetic resonators. The formed elements may also or alternatively remain in the mold used to form them.

The mold may be formed using standard CNC (Computer Numerically Controlled) routing or milling tools or any other known technique for cutting or forming grooves in blocks. The mold may also or alternatively be formed using machining techniques, injection molding techniques, casting techniques, casting techniques, vacuum techniques, thermoforming techniques, in-situ cutting techniques, compression forming techniques, and the like.

The formed element can be removed from the mold, or it can remain in the mold. The mold can be modified with an internal inductive element. Molds can be covered, machined, attached, painted, etc. The die and conductor combination can be integrated into another housing, structure or device. The grooves cut into the mold can be of any size and can be designed to form conductive tubes, wires, strips, strips, blocks, etc. into the desired inductor shape and size.

Inductive elements used in magnetic resonators can contain more than one loop and can spiral inwards or outwards or up or down or in some combination of directions. In general, magnetic resonators can have a variety of shapes, sizes, and number of turns, and they can be composed of a variety of conductive materials.

The magnetic resonator may be freestanding, or it may be enclosed in a housing, container, sleeve or housing. A magnetic resonator may include a template used to fabricate an inductive element. These various forms and enclosures can be composed of almost any kind of material. For some applications, low loss materials such as Teflon, REXOLITE, Styrene, etc. may be preferred. These enclosures may contain fixtures that hold the inductive components.

A magnetic resonator can consist of a self-resonant coil of copper wire or tube. A magnetic resonator consisting of a self-resonant wire coil may comprise a wire of length l and cross-section of radius a wound into a helical coil of radius x, height h and number of turns N, which may for example be characterized as $N=\sqrt{l^2-{\mathrm{<figure-callout\; id="2"\; label="h"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">h</figure-callout>}}^2}/2\mathrm{\&pi;x}.$

The magnetic resonator structure can be configured such that x is about 30 cm, h is about 20 cm, a is about 3 mm and N is about 5.25, and during operation a power source coupled to the magnetic resonator can drive the resonator at a resonant frequency f , where f is about 10.6MHz. With x about 30 cm, h about 20 cm, a about 1 cm and N about 4, the resonator can be driven at a frequency f, where f is about 13.4 MHz. With x about 10 cm, h about 3 cm, a about 2 cm and N about 6, the resonator can be driven at a frequency f, where f is about 21.4 MHz.

High-Q inductive components can be designed using printed circuit board traces. Printed circuit board traces can have several advantages over mechanically formed inductive elements, including that they can be accurately reproduced and easily integrated using established printed circuit board manufacturing techniques, that custom designed conductor traces can be used wire to reduce its AC resistance, and can significantly reduce the cost of its mass production.

Can be used in such as FR-4 (epoxy E-glass), multifunctional epoxy resin, high performance epoxy resin, bismaleimide triazine resin/epoxy resin, polyimide, cyanate High-Q inductive components are manufactured using standard PCB techniques on any PCB material such as polyester, Teflon, FR-2, FR-3, CEM-1, CEM-2, Rogers, Resolute, etc. Conductor traces can be formed on printed circuit board materials that have a lower loss tangent.

The conductor traces may be composed of copper, silver, gold, aluminum, nickel, etc., and they may be composed of paint, ink or other cured material. The circuit board may be flexible, and it may be a flex circuit. Conductive traces may be formed by chemical deposition, etching, lithography, spray deposition, cutting, and the like. Conductive traces can be applied to form the desired pattern and can be formed using crystal and structure growth techniques.

The dimensions of the conductive traces, as well as the number of layers containing the conductive traces, the location, size and shape of those traces, and the architecture used to interconnect them can be designed to achieve or optimize certain system specifications, such as resonance Resonator Q, Q _(p) , resonator size, resonator material and manufacturing cost, U, U _(p) , etc.

As an example, as shown in FIG. 10( a ), a three-turn high-Q inductive element 1001A was fabricated on a four-layer printed circuit board using a rectangular copper trace pattern. The copper traces are shown in black and the PCB is shown in white. The width and thickness of the copper traces in this example are approximately 1 cm (400 mils) and 43 μm (1.7 mils), respectively. The edge spacing between turns of conductive traces on a single layer is about 0.75 cm (300 mils), and each ply is about 100 μm (4 mils) thick. The pattern shown in Figure 10(a) is repeated on each layer of the board, and the conductors are connected in parallel. The external dimensions of the 3-loop structure are approximately 30cm by 20cm. The measured inductance of this PCB loop is 5.3 μH. A magnetic resonator using this inductor element and tunable capacitor has a quality factor Q of 550 at its design resonance frequency of 6.78 MHz. The resonant frequency can be tuned by varying the inductance and capacitance values in the magnetic resonator.

As another example, a two-turn inductor 1001B is fabricated on a four-layer printed circuit board using a rectangular copper trace pattern as shown in FIG. 10(b). The copper traces are shown in black and the PCB is shown in white. The width and height of the copper traces in this example are approximately 0.75 cm (300 mils) and 43 μm (1.7 mils), respectively. The edge spacing between turns of conductive traces on a single layer is about 0.635 cm (250 mils), and each ply is about 100 μm (4 mils) thick. The pattern shown in Figure 10(b) is repeated on each layer of the board, and the conductors are connected in parallel. The outer dimensions of the double ring structure are approximately 7.62 cm by 26.7 cm. The measured inductance of this PCB loop is 1.3µH. Stacking two boards together with a vertical separation of about 0.635 cm (250 mils) and connecting the two boards together in series produces a PCB inductor with an inductance of about 3.4 μH. A magnetic resonator using this stacked inductor loop and tunable capacitor has a quality factor Q of 390 at its design resonance frequency of 6.78 MHz. The resonant frequency can be tuned by varying the inductance and capacitance values in the magnetic resonator.

The inductive element can be formed using magnetic materials of any size, shape, thickness, etc. and materials with a wide range of permeability and loss values. These magnetic materials can be solid blocks, which can enclose hollow volumes, which can be formed from many small pieces of magnetic material that are tiled or stacked together, and which can be combined with conductive sheets or housings made of highly conductive materials. integrated. A wire can be wrapped around a magnetic material to create a magnetic near field. These wires may be wound around one or more than one axis of the structure. Multiple wires can be wound around the magnetic material and combined in parallel or in series or via switches to form custom near-field patterns.

The magnetic resonator may comprise 15 turns of Litz wire wound around a 19.2 cm x 10 cm x 5 mm tiled block of 3F3 ferrite material. Litz wirewound ferrite material can be wound in any direction or combination of directions to achieve the desired resonator performance. The number of turns of wire, the spacing between turns, the type of wire, the size and shape of the magnetic material, and the type of magnetic material are all design parameters that can be changed or optimized for different applications.

High-Q magnetic resonators using magnetic material structures

The assembled magnetic materials can be used to form an open magnetic circuit (albeit one with a void approximately the size of the entire structure) to achieve a magnetic resonator structure. In these structures, a highly conductive material is wound around a structure made of magnetic material to form the inductive element of the magnetic resonator. The capacitive element can be connected to a highly conductive material and then the resonant frequency determined as described above. These magnetic resonators have their dipole moment in the plane of the two-dimensional resonator structure (rather than perpendicular to it as in the case of capacitively loaded inductor loop resonators).

A diagram of a single planar resonator structure is shown in Fig. 11(a). The planar resonator structure consists of a core of magnetic material 1121 , such as ferrite with one or more loops of conductive material 1122 wound around the core 1121 . This structure can be used as a source resonator to transfer power and as a device resonator to capture energy. When used as a source, the ends of the conductors can be coupled to a power source. An alternating current flowing through a conductor loop excites an alternating magnetic field. When using the structure to receive power, the ends of the conductors can be coupled to a power sink or load. Changing the magnetic field induces an electromotive force in one or more loops of the conductor wound around the core magnetic material. The dipole moment of these types of structures is in the plane of the structure and is oriented, for example, along the Y-axis of the structure as in Fig. 11(a). Two such structures have a strong coupling when placed substantially in the same plane (ie the X, Y planes of Figure 11). The structure of Figure 11(a) has an optimum orientation when the resonators are aligned in the same plane along their Y-axis.

For certain applications, the geometry and coupling orientation of the planar resonator may be preferred. Planar or planar resonator shapes may be easier to integrate into many electronic devices that are relatively flat and planar. Planar resonators can be integrated into the entire back or sides of a device without requiring changes in the geometry of the device. Due to the flat shape of many devices, the natural position of the device when placed on a flat surface is to lie flat in its largest dimension parallel to the surface on which it is placed. A planar resonator integrated into a flat device is naturally parallel to the plane of the surface and in a suitable coupling orientation relative to resonators of other devices or planar resonator sources placed on the flat surface.

As mentioned, the geometry of the planar resonator may allow for easier integration into devices. Its low profile may allow the resonator to be integrated into or as part of the entire side of the device. When the entire side of the device is covered by the resonator, magnetic flux can flow through the resonator core without being impeded by lossy material that may be part of the device or device circuitry.

The core of a planar resonator structure can have a variety of shapes and thicknesses, and it can be flat or planar such that the smallest dimension does not exceed 30% of the largest dimension of the structure. The core can have complex geometries and can have indentations, notches, ridges, and the like. Geometric enhancements can be used to reduce coupling dependence on orientation, and they can be used to facilitate integration into devices, packages, packaging, housings, covers, skins, etc. Two exemplary variations of the core geometry are shown in Fig. 11(b). For example, the planar core 1131 may be shaped such that the ends are much wider than the middle of the structure to create gaps for the conductor windings. The core material can be of varying thickness with the ends thicker and wider than the middle. The core material 1132 may have any number of notches or cutouts 1133 of various depths, widths, and shapes to accommodate conductor loops, housings, encapsulation, and the like.

The shape and dimensions of the core may also be dictated by the dimensions and properties of the device integrated into it. The core material may bend to follow the contours of the device, or may require asymmetric notches or cuts to allow clearance for various parts of the device. The core structure may be a single piece of monolithic magnetic material, or may consist of multiple tiles, blocks or sheets arranged together to form a larger structure. The different layers, tiles, blocks or sheets of the structure may be of similar material, or may be of different material. It may be desirable to use materials with different permeability at different locations in the structure. Core structures with different permeability can be useful to direct magnetic flux, improve coupling and influence the shape or extent of the active area of the system.

The conductor of the planar resonator structure may be wound at least once around the core. In some cases, it may be preferable to wind at least three times. The conductor can be any good conductor, including wire, litz wire, conductive tube, sheet, strip, gel, ink, trace, and the like.

The size, shape or dimensions of the active area of the source can also be further enhanced, altered or modified by the use of materials that block, shield or direct the magnetic field. To create an asymmetric active area around the source, the sides of the source can be covered with magnetic shielding to reduce the strength of the magnetic field in a particular direction. The shield may be a conductor or a layered combination of conductor and magnetic material that can be used to direct the magnetic field away from a particular direction. Structures consisting of layers of conductors and magnetic materials can be used to reduce energy losses that may occur due to shielding of the source.

The plurality of planar resonators may be integrated or combined into one planar resonator structure. One or more conductors may be wound around the core structure such that the loop formed by the two conductors is not coaxial. An example of such a structure is shown in Figure 12, where two conductors 1201, 1202 are wound around a planar rectangular core 1203 at orthogonal angles. The core may be rectangular, or it may be of various geometries with multiple extensions or projections. The protrusions may be useful for wrapping of conductors, reducing the weight, size or mass of the core, or may be used to enhance the directional or omnidirectional properties of the resonator. A multi-wound planar resonator with four protrusions is shown in Fig. 13 with an inner structure 1310, where four conductors 1301, 1302, 1303, 1304 are wound around the core. The core may comprise an extension 1305, 1306, 1307, 1308 having one or more conductor loops. Individual conductors may be wound around a core to form non-coaxial loops. For example, the four conductor loops of FIG. 13 can be formed with one continuous piece of conductor or with two conductors in which a single conductor is used to achieve all coaxial loops.

An inhomogeneous or asymmetrical field distribution around a resonator comprising multiple conductor loops can be generated by driving certain conductor loops with different parameters. Certain conductor loops of a source resonator with multiple conductor loops can be driven by power supplies with different frequencies, voltages, power levels, duty cycles, etc., all of which can be used to affect the magnitude of the magnetic field generated by each conductor. strength.

Planar resonator structures can be combined with capacitively loaded inductor resonator coils to provide a full omni-directional active area, including above and below the source, while maintaining a planar resonator structure. As shown in FIG. 13 , an additional resonator loop coil 1309 comprising one or more conductor loops may be placed in the same plane as the planar resonator structure 1310 . The outer resonator coil provides an active area substantially above and below the source. The resonator coils may be arranged with any number of planar resonator structures and arrangements described herein.

Planar resonator structures can be enclosed in magnetically permeable packages or integrated into other devices. The planar profile of the resonators within a single common plane allows packaging and integration into flat devices. A diagram illustrating the application of the resonator is shown in FIG. 14 . A planar source 1411 comprising one or more planar resonators 1414, each having one or more conductor loops, can be presented to a device 1412, integrated with other planar resonators 1415, 1416 and placed within the active region 1417 of the source. 1413 transfer power. The device may include multiple planar resonators such that the active area of the source is unchanged regardless of the orientation of the device relative to the source. Apart from invariance to rotational misalignment, a planar device including a planar resonator can be completely flipped over without substantially affecting the active area, since the planar resonator remains in the plane of the source.

Another diagram illustrating a possible use of a power transfer system using planar resonator structures is shown in FIG. 15 . A planar source 1521 placed above a surface 1525 can create an active area covering a substantial surface area, which creates an "energized surface" area. Devices such as computers 1524, mobile phones 1522, game consoles, and other electronic devices 1523 coupled to their respective planar device resonators can be accessed from The source receives energy. Multiple devices with different sizes can be placed in the active area and used normally while being charged or powered from a source without strict placement or alignment constraints. The source can be placed below the surface of a table, counter, desk, cabinet, etc., allowing it to be completely hidden while energizing the top surface of the table, counter, desk, cabinet, etc., creating a much larger surface than the source valid area.

The source may include a display or other visual, audible or vibratory indicators to show the direction of the charging device or what device is being charged, charging errors or problems, power level, charging time, etc.

The source resonator and circuitry can be integrated into any number of other devices. Sources can be integrated into devices such as clocks, keyboards, monitors, photo frames, etc. For example, a keyboard integrated with planar resonators and appropriate power and control circuitry can be used as a source for devices placed around the keyboard, such as computer mice, web cameras, mobile phones, etc., without taking up any additional desk space.

Although planar resonator structures have been described in the context of mobile devices, it should be clear to those skilled in the art that a planar source for wireless power transfer with an active area extending beyond its physical size has many other consumers and industrial applications. The structures and configurations can be useful for many applications in which electronic or electrical devices and power sources are positioned, positioned or manipulated, often in substantially the same plane and alignment. Some possible application scenarios include devices on walls, floors, ceilings, or any other substantially planar surface.

The planar source resonator can be integrated into a picture frame or hung on a wall, providing an active area within the plane of the wall where other electronic devices such as digital picture frames, TVs, lamps, etc. can be installed and powered without wires . Planar resonators can be integrated into the floor, resulting in an energized floor or active area on the floor on which devices can be placed to receive power. Audio speakers, lights, heaters, etc. can be placed within the active area and receive power wirelessly.

Planar resonators may have additional components coupled to conductors. Components such as capacitors, inductors, resistors, diodes, etc. can be coupled to the conductors and can be used to adjust or tune the resonant frequency and impedance matching for the resonator.

Planar resonator structures of the type described above and shown in Figure 11(a) can be produced, for example, with a quality factor Q of 100 or higher and even a Q of 1000 or higher. As shown in Fig. 11(c), energy can be wirelessly transferred from one planar resonator structure to another over a distance larger than the characteristic size of the resonator.

In addition to utilizing magnetic materials to achieve properties similar to those of inductive elements in magnetic resonators, combinations of good conductor materials and magnetic materials can also be used to achieve such inductive structures. Figure 16(a) shows a magnetic resonator structure 1602 which may comprise a high conductivity material surrounded by at least one layer of magnetic material and linked by blocks of magnetic material 1604 (the interior of which may be shielded from AC electromagnetic fields generated outside) One or more shells made.

The structure may comprise a sheet of highly conductive material covered on one side by a layer of magnetic material. The layered structure can instead be conformally applied to electronic devices, so that parts of the device can be covered by layers of highly conductive and magnetic material, while other parts that need to be easily accessible, such as buttons or screens, can be kept out of the way. is covered. The structure may also or alternatively comprise only layers or blocks of sheets of magnetic material. Thus, magnetic resonators can be incorporated into existing devices without significantly interfering with their existing functionality and requiring little extensive redesign. Furthermore, layers of good conductors and/or magnetic materials can be made thin enough (a millimeter or so or less) that they will add little additional weight and bulk to the finished device. The electromagnetic field associated with this structure can be excited using an oscillating current applied to a length of conductor wrapped around the structure as shown by the square loop on the center of the structure in FIG. 16 .

quality factor of the structure

Structures of the type described above can be produced with a quality factor Q of about 1000 or higher. This high Q is possible even if losses in the magnetic material are high if the fraction of magnetic energy within the magnetic material is small compared to the total magnetic energy associated with the object. For structures consisting of layers of conductive material and magnetic material, losses in the conductive material can be reduced by the presence of the magnetic material as previously described. In structures where the thickness of the layer of magnetic material is approximately 1/100 of the largest dimension of the system (e.g., the magnetic material is approximately 1 mm thick and the area of the structure is approximately 10 cm x 10 cm) and the relative permeability is approximately 1000, one can Such that the portion of the magnetic energy contained within the magnetic material is only a few percent of the total magnetic energy associated with the object or resonator. To see how that happens, note that the expression for the magnetic energy contained in the volume is U _m = ∫ _V drB(r) ² /(2μ _r /μ ₀ ), as long as B (rather than H) is across The dominant field maintained by the magnetic material-air interface (as is usually the case in open magnetic circuits), can significantly reduce the fraction of magnetic energy contained in the high μ _r region compared to that in air.

If frac is used to denote the fraction of magnetic energy in a magnetic material, and the loss tangent of the material is tanδ, then the Q of the resonator is Q = 1/(frac x tanδ), assuming the magnetic material is the only source of loss. Thus, even for loss tangents as high as 0.1, a Q of about 1000 can be achieved for these types of resonator structures.

If the structure is driven with N turns of wire wound around it, the losses in the excitation inductor loop can be ignored if N is high enough. Figure 17 shows a schematic diagram of an equivalent circuit 1700 for these structures and the scaling of loss mechanisms and inductance with the number of turns N wound around a structure made of conductive and magnetic materials. If proximity effects can be neglected (by using appropriate windings, or wires designed to minimize proximity effects, such as litz wire, etc.), the resistance 1702 due to the wire in the loop conductor scales linearly with the length of the loop The length of the loop is proportional to the number of turns. On the other hand, both the equivalent resistance 1708 and the equivalent inductance 1704 of these particular structures are proportional to the square of the magnetic field inside the structure. Since this magnetic field is proportional to N, both the equivalent resistance 1708 and the equivalent inductance 1704 are proportional to ^N2 . Thus, for sufficiently large N, the resistance 1702 of the wire is much smaller than the equivalent resistance 1708 of the magnetic structure, and the Q of the resonator is asymptotic to Q _max =ωL _μ /R _μ .

和σ＝0.5S/m的2mm磁性材料层完全覆盖的两个20cm×8cm×2cm空心区域。这两个平行六面体间隔开4cm，并被相同磁性材料的2cm×4cm×2cm块连接。激励环路被绕此块的中心缠绕。在300kHz的频率处，此结构具有计算的890的Q。可以将导体和磁性材料结构成形为使某些系统参数最优化。例如，被激励环路封闭的结构的尺寸可以是小的，以减小激励环路的电阻，或者其可以是大的，以减轻与大磁场相关联的磁性材料中的损耗。请注意，与由磁性材料组成的相同结构相关联的磁流线和Q将仅类似于这里所示的层导体和磁性材料设计。Figure 16(a) shows a diagram of a copper and magnetic material structure 1602 driven by a square current loop around a narrowed section at the center of the structure 1604 and the magnetic field streamlines 1608 produced by this structure. This exemplary structure consists of being capped with copper and then having the properties

Two 20 cm x 8 cm x 2 cm hollow areas completely covered by a 2 mm layer of magnetic material with σ = 0.5 S/m. These two parallelepipeds are spaced 4 cm apart and connected by a 2 cm x 4 cm x 2 cm block of the same magnetic material. An excitation loop is wound around the center of the block. At a frequency of 300 kHz, this structure has a calculated Q of 890. Conductor and magnetic material structures can be shaped to optimize certain system parameters. For example, the size of the structure enclosed by the excitation loop may be small to reduce the resistance of the excitation loop, or it may be large to mitigate losses in the magnetic material associated with large magnetic fields. Note that the magnetic flow lines and Q associated with the same structure composed of magnetic material will only be similar to the layered conductor and magnetic material design shown here.

Electromagnetic resonators interact with other objects

For electromagnetic resonators, extrinsic loss mechanisms that perturb the intrinsic Q-value can include absorption losses inside the material of nearby extraneous objects and radiation losses related to scattering from the resonant field from nearby extraneous materials. Absorption loss can be associated with a material that has a non-zero but finite conductivity σ (or equivalently a non-zero and finite imaginary part of the dielectric permittivity) in the frequency range of interest, such that electromagnetic fields can penetrate it and A current is induced therein, which then dissipates energy through resistive losses. An object may be described as lossy if it at least partially comprises lossy material.

可以根据P_d＝∫_Vdrσ|E|²＝∫_Vdr|J|²/σ来确定在对象内部耗散的功率P_d，其中，我们利用欧姆定律J＝σE，并且其中，E是电场且J是电流密度。Consider an object of homogeneous isotropic material comprising electrical conductivity σ and magnetic permeability μ. The penetration depth of the electromagnetic field inside the object is given by the skin depth

The power _Pd dissipated inside the object can be determined according to _Pd = ∫ _Vdrσ |E| ² = _∫Vdr |J| ² /σ, where we use Ohm's law J=σE, and where E is the electric field and J is the current density.

If, in the frequency range of interest, the conductivity σ of the material making up the object is low enough that the skin depth δ of the material can be considered long (i.e., δ is longer than the object's characteristic dimension, or δ is less lossy than the object's The characteristic dimension of the part is long), the electromagnetic fields E and H (where H is the magnetic field) can penetrate significantly into the object. These finite-valued fields can then generate dissipated power that scales with P _d ∼ σV _ol <|E| ² >, where, in the volume under consideration, V _ol is the volume of the lossy object and <|E | ² > is the spatial average of the square of the electric field. Thus, at the lower limit of conductivity, the dissipated power scales proportionally to the conductivity and returns to zero at the limit for non-conducting (purely dielectric) materials.

Electromagnetic fields E and H can only penetrate short distances into the object if the conductivity σ of the material making up the object is high enough in the frequency range of interest that the skin depth of the material can be considered short (i.e., It stays near the 'skin' of the material, where δ is less than the characteristic thickness of the lossy part of the object). In this case, the current induced inside the material can be concentrated very close to the material surface, approximately within the skin depth, and the surface current density (determined mainly by the shape of the incident electromagnetic field and provided that the thickness of the conductor is less than The skin depth is much larger, independent of frequency and conductivity of the first order) K(x,y) (where x and y are the coordinates that parameterize the surface) with a function that decays exponentially into the surface: exp(-z/δ)/δ (where z represents a coordinate locally perpendicular to the surface) to approximate its magnitude: J(x, y, z) = K(x, y) exp(-z/ δ)/δ. Then, the dissipated power P _d can be estimated by the following equation,

Thus, at the high conductivity limit, the dissipated power scales inversely with the square root of the conductivity, and returns to zero at the limit for perfectly conductive materials.

If the conductivity σ of the material making up the object is finite in the frequency range of interest, then the skin depth δ of the material can penetrate some distance into the object and potentially dissipate a certain amount of power within the object, Also depends on the size of the object and the strength of the electromagnetic field. This description can be generalized to also describe the general case of an object comprising a plurality of different materials with different properties and conductivities, such as an object having an arbitrary inhomogeneous and anisotropic distribution of conductivities inside the object.

Note that the magnitude of the loss mechanisms described above can depend on the position and orientation of the extraneous object relative to the resonator field and the material composition of the extraneous object. For example, high conductivity materials can shift the resonant frequency of a resonator and detune it from other resonant objects. This frequency shift can be fixed by applying a feedback mechanism to the resonator that modifies its frequency, such as through changes in the resonator's inductance and/or capacitance. These changes can be achieved using variable capacitors and inductors, and in some cases by changes in the geometry of the components in the resonator. Other novel tuning mechanisms described below can also be used to change the resonator frequency.

Where external losses are high, the perturbed Q can be low and steps can be taken to limit the absorption of resonator energy within such extraneous objects and materials. Due to the functional dependence of dissipated power on electromagnetic field strength, system performance can be maximized by designing the system such that the desired coupling is achieved with evanescent resonant field tails that are short at the source resonator and long at the device resonator. Optimizing so that the perturbed Q of the source is optimized in the presence of other objects (or vice versa if the perturbed Q of the device needs to be optimized.)

Note that many common extraneous materials and objects such as people, animals, plants, building materials, etc. may have low conductivity and thus may have little effect on the wireless energy transfer scheme disclosed herein. An important fact related to the magnetic resonator design we describe is that its electric field can be confined primarily within the resonator structure itself, so it should be possible to provide wireless power exchange over medium-range distances while being generally acceptable for human safety. operate within the policy.

Electromagnetic resonators with reduced interaction

One frequency range of interest for near-field wireless power transfer is between 10 kHz and 100 MHz. In this frequency range, many common non-metallic materials such as many types of wood and plastics can have relatively low electrical conductivity such that only small amounts of power can be dissipated within them. In addition, materials with low loss tangent tanΔ (where tanΔ=ε″/ε′, and ε″ and ε′ are the imaginary and real parts of permittivity, respectively) can also allow only a small amount of power to be dissipated inside . Metallic materials with relatively high electrical conductivity, such as copper, silver, gold, etc., may also have little power dissipated therein because, as previously discussed, electromagnetic fields do not penetrate significantly through these materials. These very low and very high conductivity materials and low loss tangent materials and objects can have negligible impact on the losses of the magnetic resonator.

However, within the frequency range of interest, there are materials and objects such as certain electronic circuits and certain low-conductivity metals, which can have moderate (often heterogeneous and anisotropic) conductivity and/or moderate highest loss tangent, and it can have relatively high dissipative losses. A relatively large amount of power can be dissipated inside it. These materials and objects can dissipate enough energy to reduce Q _(p) by a significant amount, and may be referred to as "lossy objects".

One way to reduce the effect of lossy materials on the Q _(p) of a resonator is to use high conductivity materials to shape the resonator field such that it avoids lossy objects. The process of using high conductivity materials to tune electromagnetic fields so that they avoid lossy objects in their vicinity can be understood by thinking of high conductivity materials as materials that deflect or reshape fields. This formulation is qualitatively correct as long as the thickness of the conductor is greater than the skin depth, because the boundary conditions for the electromagnetic field at the surface of a good conductor force the electric field to be nearly completely perpendicular to the plane of the conductor and the magnetic field to be nearly completely tangent to the plane of the conductor . Therefore, a perpendicular magnetic field or a tangential electric field will be "deflected away" from the conducting surface. Furthermore, it is even possible to force a tangential magnetic or perpendicular electric field to decrease in magnitude on one side and/or in particular the position of the conductive surface, depending on the relative position of the source of the field and the conductive surface.

As an example, Figure 18 shows a finite element method (FEM) simulation of two high conductivity surfaces 1802 above and below a lossy dielectric material 1804 in an external, initially uniform magnetic field at a frequency f = 6.78 MHz. The system is azimuthally symmetric about the r=0 axis. In this simulation, a lossy dielectric material 1804 is sandwiched between two conductors 1802 (shown as white lines at approximately z=±0.01 m). In the absence of conductive surfaces above and below the dielectric disk, the magnetic field (represented by the depicted magnetic field lines) will still be substantially uniform (field lines are straight and parallel to the z-axis), indicating that the magnetic field will pass straight through Lossy dielectric material. In this case the power would have been dissipated in the lossy dielectric disc. However, in the presence of conductive surfaces, this simulation shows that the magnetic field is reshaped. Forcing the magnetic field to be tangential to the surfaces of the conductors, and thus deflected around those conductive surfaces 1802, minimizes the amount of power that might be dissipated in the lossy dielectric material 1804 behind or between the conductive surfaces. As used herein, an axis of electrical symmetry refers to any axis about which a fixed or time-varying electric or magnetic field is substantially symmetric during energy exchange as disclosed herein.

Similar effects were observed even with only one conductive surface above or below the dielectric disc. If the dielectric disc is thin, the fact that the electric field is essentially zero at the surface and approaches it continuously and smoothly means that the electric field is very low anywhere close to the surface, ie inside the dielectric disc. A single surface implementation for deflecting the resonator field away from the lossy object may be preferred for applications where covering lossy material or both sides of the surface is not allowed (eg LCD screens). Note that even a very thin surface of conductive material with a few skin depths can suffice in the presence of lossy materials (the skin depth in pure copper at 6.78MHz is ~20μm, and at 250kHz is ~100 μm) significantly improves the Q _(p) of the resonator.

Loss-independent materials and objects can be part of devices in which high-Q resonators are to be integrated. A number of techniques can be used to reduce energy dissipation in these lossy materials and objects, including:

By positioning lossy materials and objects away from the resonator, or in a particular position and orientation relative to the resonator.

Partially or completely cover lossy materials and objects in the vicinity of the resonator by using high conductivity materials or structures.

The lossy object is completely covered by placing a closed surface of high conductivity material, such as a sheet or mesh, around the lossy object and shaping the resonator field so that it avoids the lossy object.

By placing a surface of high conductivity material, such as a sheet or mesh, around only a portion of the lossy object, such as along the top, bottom, along the sides, etc. of the object or material.

The strength of the field at the location of the lossy object is reduced by placing even a single surface of high conductivity material, such as a sheet or mesh, above or below or on one side of the lossy object.

FIG. 19 shows a capacitively loaded loop inductor forming the magnetic resonator 102 and a disc-shaped surface of high conductivity material 1802 completely surrounding a lossy object 1804 placed inside the loop inductor. Note that some lossy objects may be components such as electronic circuits, which may need to interact, communicate or be connected to the external environment and thus cannot be completely electromagnetically isolated. Partially covering a lossy material with a high conductivity material can still reduce additional losses while allowing the lossy material or object to function properly.

FIG. 20 shows a capacitively loaded loop inductor used as resonator 102 and a surface of high conductivity material 1802 placed inside the inductor loop surrounding only a portion of a lossy object 1804 .

By placing a single surface of high conductivity material above, below or to the side of a lossy object or material, etc., additional losses can be reduced but not completely eliminated. An example is shown in Figure 21, where a capacitively loaded loop inductor is used as the resonator 102, and a surface of high conductivity material 1802 is placed inside the inductor loop below a lossy object 1804 to reduce the lossy The strength of the field at the object's location. Due to considerations of cost, weight, assembly complication, airflow, visual accessibility, physical proximity, etc., it may be preferable to cover only one side of a material or object.

A single surface of high conductivity material can be used to avoid objects that cannot or should not be covered from both sides (such as LCD or plasma screens). Optically transparent conductors can be used to avoid such lossy objects. Instead of or in addition to the optically transparent conductor, a high conductivity optically opaque material may alternatively be placed on only a portion of the lossy object. The suitability of, and design trade-offs inherent in, single-sided versus multi-sided coverage implementations may depend on the details of the wireless energy transfer scenario and the nature of the lossy materials and objects.

Below, we describe an example of the use of high-conductivity surfaces to improve the Q-insensitivity Θ _(p) of integrated magnetic resonators used in wireless energy transfer systems. FIG. 22 shows a wireless projector 2200 . The wireless projector may include a device resonator 102C, a projector 2202, a wireless network/video adapter 2204, and a power conversion circuit 2208 arranged as shown. The device resonator 102C may include a three-turn conductor loop arranged as a closed surface, and a capacitor network 2210 . The conductor loop may be arranged such that the device resonator 102C has a high Q (eg, >100) at its operating resonant frequency. Prior to integration in the fully wireless projector 2200, this device resonator 102C had a Q of approximately 477 at the designed operating resonant frequency of 6.78 MHz. When integrating and placing the wireless network/video adapter card 2204 on the center of the resonator loop inductor, the resonator Q _(integrated) is reduced to about 347. At least some of the decrease from Q to Q _(integrated) is due to perturbing losses in the wireless network/video adapter card. As noted above, the electromagnetic field associated with the magnetic resonator 102C can induce currents in and on the wireless network/video adapter card 2204 that can be dissipated in resistive losses in the lossy materials making up the card . We observe that the Q _(integrated) of a resonator can be affected differently depending on the composition, position and orientation of objects and materials placed near the resonator.

In the completely wireless projector example, a thin copper pouch (the folded copper sheet that covers the top and bottom of the wireless network/video adapter card, but not the communications antenna) is used to cover the network/video adapter card to place the magnetic resonator's Q _{(integrated )} improved to a Q _{(integrated+copper pocket)} of about 444. In other words, using copper pockets to offset the resonator field away from the lossy material eliminates most of the reduction in Q _(integrated) due to perturbations caused by extraneous network/video adapter cards.

In another fully wireless projector example, covering the network/video adapter card with a single copper sheet placed under the card provides a Q _{(integrated + copper sheet)} approximately equal to Q _{(integrated + copper pocket)} . In this example, the perturbed Q of the system can be kept high with a single high conductivity sheet used to keep the resonator field away from the lossy adapter card.

It may be advantageous to locate or orient lossy materials and objects that are part of a device that includes a high-Q electromagnetic resonator where the field generated by the resonator is relatively weak so that little power is dissipated in these objects , so the Q insensitivity Θ _(p) can be large. As shown earlier, materials of different conductivity may respond differently to electric versus magnetic fields. Thus, localization techniques can be dedicated to one or the other field, depending on the conductivity of unrelated objects.

Figure 23 shows the electric field 2312 and magnetic field 2314 along a line containing the diameter of a circular loop inductor resonant at 10 MHz and the loop inductor along a capacitively loaded circular loop inductor for a wire of radius 30 cm The magnitudes of the electric field 2318 and magnetic field 2320 of the axis. It can be seen that the magnitude of the resonant near field reaches its maximum close to the wire and decays away from the loops 2312,2314. In the plane of the loop inductors 2318, 2320, the field reaches a local minimum at the center of the loop. Thus, given the finite size of the device, it is possible that the field is weakest at the extremes of the device, and that the field magnitude has a local minimum somewhere within the device. This argument applies to any other type of electromagnetic resonator 102 and any type of device. An example is shown in Figures 24a and 24b, where a capacitively loaded inductor loop forms the magnetic resonator 102, and an irrelevant lossy object 1804 is located where the electromagnetic field has a minimum magnitude.

In the exemplary example, a magnetic resonator is formed using a three-turn conductor loop and a capacitor network arranged to close a square surface (with rounded corners). The Q of the resonator is about 619 at the designed operating resonant frequency of 6.78 MHz. The perturbation Q of this resonator depends on the placement of the perturbing object (in this case a pocket projector) relative to the resonator. When the perturbation projector is placed inside the inductor loop and at its center or over the inductor turns, Q _(projector) is about 96, which is lower than when the perturbation projector is placed outside the resonator (in this case In the case, Q _(projector) is about 513). These measurements support the analysis showing that the fields inside the loop of an inductor can be larger than those outside it, and therefore a lossy object placed inside such a loop inductor versus a lossy object placed outside the loop inductor Compared with time, it can provide lower disturbance Q for the system. Depending on the resonator design and the material composition and orientation of the lossy objects, the arrangement shown in Figure 24b may provide a higher Q insensitivity Θ _(projector) than the arrangement shown in Figure 24a.

High-Q resonators can be integrated inside the device. Unrelated materials and objects of high dielectric permittivity, magnetic permeability or electrical conductivity can be part of the device into which the high-Q resonator will be integrated. For these extraneous materials and objects in the vicinity of a high-Q electromagnetic resonator, depending on their size, position and orientation relative to the resonator, the resonator field distribution can deform and deviate significantly from the original unperturbed field distribution of the resonator. Such deformation of the unperturbed field of the resonator can significantly reduce Q to a lower Q _(p) even if extraneous objects and materials are lossless.

High conductivity objects (which are part of a device comprising a high-Q electromagnetic resonator) are placed such that the surfaces of these objects are oriented as perpendicular as possible to the electric field lines generated by the unperturbed resonator and parallel to the electric field lines generated by the unperturbed resonator. The orientation of the resulting magnetic field lines, thus distorting the resonant field distribution by the smallest possible amount, may be advantageous. Other common objects that can be arranged perpendicular to the plane of the magnetic resonator loop include screens (LCDs, plasmas, etc.), batteries, housings, connectors, radiating antennas, etc. The Q insensitivity Θ _(p) of the resonator can be much larger than if the object were placed in a different orientation relative to the resonator field.

Lossy extraneous materials and objects that are not part of an integrated device that includes a high-Q resonator may be located or brought into the vicinity of the resonator, for example during use of the device. In some cases, high conductivity materials are used to tune the resonator field such that it avoids regions where lossy extraneous objects are located or introduced to reduce power dissipation in these materials and objects and increase Q insensitivity Θ _{( p)} may be advantageous. An example is shown in Figure 25, where a capacitively loaded loop inductor and capacitor are used as the resonator 102, and a surface of high conductivity material 1802 is placed over the inductor loop to reduce the area over the resonator With the magnitude of the field in , lossy irrelevant objects 1804 may be located or introduced into this region.

Note that a high conductivity surface brought into the vicinity of the resonator to reshape the field can also cause Q _{(cond.surface)} < Q. The reduction in perturbed Q can be due to energy dissipation inside the lossy conductor or deformation of the unperturbed resonator field associated with field boundary conditions at the surface of the matching conductor. Thus, while high conductivity surfaces can be used to reduce additional losses due to dissipation inside unrelated lossy objects, in some cases, especially in certain cases where this is achieved by significantly reshaping the electromagnetic field In some cases, using such high-conductivity surfaces to keep the field away from lossy objects can effectively yield Q _{(p+cond.surface)} < Q _(p) instead of the desired result Q _{(p+cond.surface)} > Q _(p) .

As described above, in the presence of loss-inducing objects, the perturbed figure of merit of a magnetic resonator can be improved if the electromagnetic field associated with the magnetic resonator is reshaped to avoid the loss-inducing objects. Another way to reshape the unperturbed resonator field is to use a high permeability material to completely or partially enclose or cover the loss-inducing object, thereby reducing the interaction of the magnetic field with the loss-inducing object.

Magnetic field shielding has been previously described eg in Electrodynamics 3rd Ed., Jackson, pp. 201-203. Therein, a spherical shell of magnetically permeable material shielding its interior from the external magnetic field is shown. For example, if a shell of inner diameter a, outer diameter b, and relative permeability μ _r is placed in an initially uniform magnetic field H ₀ , the field inside the shell will have a constant magnitude 9 μ _r H ₀ /[(2 μ _r +1 )(μ _r +2)-2(a/b) ³ (μ _r -1) ² ], if μ _r ＞＞1, then it tends to 9H ₀ /2μ _r (1-(a/b) ³ ) . This result shows that incident magnetic fields (but not necessarily incident electric fields) can be greatly attenuated inside the shell, even if the shell is rather thin, provided the magnetic permeability is sufficiently high. In some cases it may be advantageous to use a high permeability material to partially or completely cover lossy materials and objects such that they are shunned by the resonator magnetic field and thus little power is dissipated in these materials and objects . In this approach, the Q insensitivity Θ _(p) can be larger than if the material and object were not covered, possibly greater than 1.

It may be desirable to keep both electric and magnetic fields away from objects that induce losses. As mentioned above, one way to shape the field in this way is to use high conductivity surfaces to completely or partially enclose or cover loss-inducing objects. A layer of magnetically permeable material, also known as a magnetic material (any material or metamaterial with significant magnetic permeability), can be placed on or around a high-conductivity surface. This additional layer of magnetic material can present a lower reluctance path (compared to free space) for the deflected magnetic field to follow, and can partially shield the underlying electrical conductor from incident magnetic flux. This setting reduces losses due to induced currents in high conductivity surfaces. In some cases, the lower reluctance exhibited by magnetic materials can improve the perturbed Q of the structure.

Figure 26(a) shows an axially symmetric FEM simulation of a thin conductive 2604 (copper) disk (20 cm diameter, 2 cm height) exposed to an initial uniform, applied magnetic field (gray flux lines) along the z-axis . The axis of symmetry is at r=0. The magnetic current lines shown originate from z=-∞, where these magnetic current lines are spaced at 1 cm intervals from r=3 cm to r=10 cm. Axis scales are in meters. For example, imagine that the conductive cylinder encloses loss-inducing objects within the region bounded by the magnetic resonators in the wireless energy transfer system shown in FIG. 19 .

This high conductivity enclosure can increase the perturbed Q of the lossy object and thus the total perturbed Q of the system, but the perturbed Q may still be smaller than the unperturbed Q due to induced losses and changes in the electromagnetic field distribution in the conductive surface. The perturbed Q reduction associated with the high-conductivity enclosure can be at least partially restored by including a layer of magnetic material along one or more outer surfaces of the high-conductivity enclosure. Figure 26b shows an axially symmetric FEM simulation of the thin conductive 2604A (copper) disc (20 cm diameter, 2 cm height) from Figure 26a, but with additional magnetic material placed directly on the outer surface of the high conductivity enclosure layer. Note that the presence of magnetic material can provide a less reluctance path for the magnetic field, thereby at least partially shielding the underlying conductor and reducing losses due to eddy currents induced in the conductor.

FIG. 27 depicts a modification (in an axisymmetric view) to the system shown in FIG. 26 , where it is possible that not all of the lossy material 2708 is covered by the high conductivity surface 2706 . In some cases, it may be useful to cover only one side of a material or object, such as due to considerations of cost, weight, assembly complexity, airflow, visual accessibility, physical proximity, and the like. In the exemplary arrangement shown in Figure 27, only one surface of the lossy material 2708 is covered, and the resonator inductor loop is placed on the opposite side of the high conductivity surface.

和虚相对磁导率

的0.25cm厚的磁性材料层时，模拟显示扰动品质因数被增加至δQ_{(enclosure+magnetic material)}＝5,060。A mathematical model was used to simulate a high conductivity enclosure made of copper and shaped like a 20 cm diameter by 2 cm high cylindrical disk placed within the area defined by a magnetic resonator whose inductive element is Single-turn wire loop with loop radius r=11 cm and wire radius a=1 mm. Simulations for an applied 6.78 MHz electromagnetic field indicated that the perturbation figure of merit δQ _(enclosure) for this high conductivity enclosure was 1,870. When the high conductivity enclosure is modified to include a real relative permeability

and imaginary relative permeability

For a 0.25 cm thick magnetic material layer, simulations show that the disturbance quality factor is increased to δQ _{(enclosure+magnetic material)} = 5,060.

The performance improvement due to the addition of a thin layer of magnetic material 2702 can be even more pronounced if the high conductivity enclosure fills a larger portion of the area defined by the resonator's loop inductor 2704 . In the example above, if the radius of the inductor loop 2704 is reduced so that it is only 3 mm from the surface of the high conductivity enclosure, the perturbation figure of merit can be reduced from 670 to 670 by adding a thin layer of magnetic material 2702 outside the enclosure. (conductive shell only) improved to 2,730 (conductive shell with thin layer of magnetic material).

Resonator structures can be designed to have highly confined electric fields using eg shielding or distributed capacitors (eg, which yield high), even when the resonator is in close proximity to materials that would normally induce losses.

coupled electromagnetic resonator

使两个谐振器之间的耦合因数与谐振器中的每一个中的电感元件的电感L₁和L₂和其之间的互感M相关。请注意，此表达式假设存在通过电偶极子耦合的可忽略的耦合。对于其中由具有N匝、分开距离D并如图1(b)所示地取向的圆形导电环路来形成电感器环路的电容加载电感器环路谐振器而言，互感是M＝π/4·μ_oN₁N₂(x₁x₂)²/D³，其中，x₁、N₁和x₂、N₂分别是第一和第二谐振器的导体环路的特性尺寸和匝数。请注意，这是准静态结果，因此假设谐振器的尺寸比波长小得多，并且谐振器的距离比波长小得多，而且其距离是其尺寸的至少几倍。对于在准静态极限处和中程距离处操作的这些圆形谐振器而言，如上所述，当谐振器的品质因数大到足以补偿中程距离处的小k时，可以建立中程距离处的谐振器之间的强耦合(大U)。The efficiency of energy transfer between two resonators can be determined by the strong coupling quality factor, In a magnetic resonator implementation, one can use

The coupling factor between two resonators is related to the inductances _L1 and _L2 of the inductive elements in each of the resonators and the mutual inductance M between them. Note that this expression assumes negligible coupling via electric dipole coupling. For a capacitively loaded inductor loop resonator in which the inductor loop is formed by a circular conductive loop having N turns, separated by a distance D, and oriented as shown in Figure 1(b), the mutual inductance is M = π /4·μ _o N ₁ N ₂ (x ₁ x ₂ ) ² /D ³ , where x ₁ , N ₁ and x ₂ , N ₂ are the characteristic dimensions and number of turns. Note that this is a quasi-static result, so it assumes that the resonators are much smaller in size than the wavelength, and that the distance between the resonators is much smaller than the wavelength, and that the distance is at least several times their size. For these circular resonators operating at the quasi-static limit and at mid-range distances, as described above, Strong coupling (large U) between resonators at mid-range distances can be established when the quality factor of the resonators is large enough to compensate for small k at mid-range distances.

For electromagnetic resonators, if the two resonators include conductive parts, the coupling mechanism can be the induction of current on one resonator due to electric and magnetic fields generated from the other resonator. The coupling factor may be proportional to the flux of the magnetic field generated by the high-Q inductive element in one resonator across the enclosed region of the high-Q inductive element of the second resonator.

Coupled electromagnetic resonators with reduced interaction

As previously mentioned, a high-conductivity material surface can be used to shape the resonator field such that it avoids lossy objects p near the resonator, thereby reducing the overall excess loss and maintaining the high-Q insensitivity of the resonator Θ _{(p+ cond..surface)} . However, such surfaces may result in a perturbed coupling factor k _{(p+cond.surface)} between resonators, which is smaller than the perturbed coupling factor k _(p) and depends on the size, position of the high conductivity material relative to the resonators and orientation. For example, if a high-conductivity material is placed in the plane and within the region defined by the inductive element of at least one of the magnetic resonators in a wireless energy transfer system, it can block part of the magnetic flux through the region of the resonator (regulating the coupling ), and k can be reduced.

Consider the example of Figure 19 again. In the absence of a high-conductivity disk enclosure, a certain amount of external magnetic flux can pass through the defined area of the loop. In the presence of a high-conductivity disc enclosure, some of this flux may be deflected or blocked and may no longer pass through this region of the loop, thus resulting in a smaller perturbed coupling factor k _{12(p+ cond. surfaces)} . However, since deflection magnetic field lines may closely follow the edges of highly conductive surfaces, the reduction in flux through the loop defining the disk may be less than the ratio of the area of the face of the disk to the area of the loop.

High conductivity material structures may be used alone or in combination with magnetic materials to optimize perturbed figure of merit, perturbed coupling factor, or perturbed efficiency.

Consider the example of Figure 21. The lossy object is made to have a size equal to that of the capacitively loaded inductor loop resonator, thereby filling its area A 2102 . High conductivity surface 1802 may be placed beneath lossy object 1804 . Let this resonator 1 be in a system of two coupled resonators 1 and 2, and let us consider how U _{12(object+cond.surface)} scales compared to U ₁₂ as the area of the conductive surface _AS 2104 increases. In the absence of a conductive surface 1802 beneath a lossy object 1804, the k insensitivity β _12(object) can be approximately 1, but the Q insensitivity Θ _1(object) can be small, so U insensitivity Ξ _12(object) Can be small.

When the high-conductivity surface below the lossy object covers the entire area of the inductor loop resonator ( _AS = A), k _{12(object+cond.surface)} can be close to zero since little flux is allowed to pass through The inductor loops, so _{U12(object+cond.surface)} can be close to zero. The suppression of extrinsic losses and the associated Q insensitivity Θ _{1(object+cond.surface)} compared to Θ _1(object) can be sufficiently large for intermediate dimensions of high conductivity surfaces, while the reduction in coupling Probably not significant, and the associated k insensitivity β12 _{(object+cond.surface)} is probably not much smaller than _β12(object) , such that the total U12(object+cond.surface) can be increased compared to _{U12( object) cond. surface)} . The optimal degree of avoidance of extraneous lossy objects via high conductivity surfaces in a wireless energy transfer system may depend on the details of the system configuration and application.

We describe the use of high-conductivity materials to fully or partially enclose or cover induced loss objects near high-Q resonators as a potential way to achieve high perturbed Q for systems. However, covering the object with a good conductor alone can reduce the coupling of the resonators as described above, thereby reducing the efficiency of wireless power transfer. As the area of the conductive surface approaches that of the magnetic resonator, for example, the perturbed coupling factor k _(p) may approach zero, making the use of conductive surfaces incompatible with efficient wireless power transfer.

One way to solve the above problem is to place a layer of magnetic material around the high conductivity material, since this additional layer of magnetically permeable material can present a lower reluctance path (compared to free space) for the deflected magnetic field to follow, And can partially shield the electrical conductors beneath it from incident magnetic flux. In some cases, the lower reluctance paths presented by magnetic materials can improve the electromagnetic coupling of resonators to other resonators. The resonator field associated with using the conductive material to tune the resonator field so that it avoids lossy objects in and around the high-Q magnetic resonator can be at least partially restored by including a layer of magnetic material along one or more outer surfaces of the conductive material. The reduction of the disturbed coupling factor. The magnetic material can increase the perturbed coupling factor relative to its initial unperturbed value.

Note that the simulation results in Figure 26 show that layered magnetic materials and conductive structures can deflect the incident magnetic field less than conductive structures alone. If a magnetic resonator ring with a radius only slightly larger than that shown in Figure 26(a) and Figure 26(b) defines a disk, it is clear that, unlike the situation shown in Figure 26(a) Compared to the case shown in Figure 26(b), more flux lines will be trapped, so k _(disk) will be larger for the case shown in Figure 26(b). Therefore, including a layer of magnetic material over a conductive material can improve overall system performance. A system analysis can be performed to determine if these materials should be partially, fully or minimally integrated into the resonator.

和虚相对磁导率

的0.25cm厚的磁性材料层。使用上述相同的模型和参数，我们发现通过向导体的表面添加磁性材料，耦合因数不灵敏度被改善至0.64。As noted above, FIG. 27 depicts a layered conductor 2706 and magnetic material 2702 structure that may be suitable for use when not all of the lossy material 2708 may be covered by a conductor and/or magnetic material structure. It was previously shown that for a copper conductor disc with a 20 cm diameter and 2 cm height defined by a resonator with an inductor loop radius of 11 cm and a = 1 mm wire radius, the perturbation Q calculated for the copper cylinder is 1.870. If the resonator and conductive disk shell are placed in a uniform magnetic field (aligned along the axis of symmetry of the inductor loop), we calculated that the copper conductor has an associated coupling factor insensitivity of 0.34. For comparison, we model the same arrangement, but include

and imaginary relative permeability

A 0.25cm thick layer of magnetic material. Using the same model and parameters as above, we found that by adding magnetic material to the surface of the conductor, the coupling factor insensitivity was improved to 0.64.

Magnetic material can be placed within the region bounded by the magnetic resonators to increase coupling in wireless energy transfer systems. Consider a solid sphere of magnetic material with relative permeability μ _r placed in an initially uniform magnetic field. In this example, the lower reluctance path provided by the magnetic material can encourage the magnetic field to concentrate in the volume of the sphere. We found that by adding magnetic material, the magnetic flux through the area bounded by the equator of the ball is increased by a factor of _3μr /( _μr +2). This enhancement factor can be close to 3 if μ _r &gt;&gt;1.

It can also be shown that the dipole moment of a system comprising a magnetic sphere defined by an inductive element in a magnetic resonator will enhance its magnetic dipole by the same factor. Thus, magnetic spheres with high permeability effectively triple the dipole magnetic coupling of the resonator. Most of this increase in coupling can be maintained if we use a spherical housing of magnetic material with an inner diameter of a and an outer diameter of b, even if this housing is over a block or shell made of highly conductive material. In this case, the enhancement of the flux through the mid-dimensional wire is

$\frac{\mathrm{<span\; class="notranslate">\; 3</span>}{\mathrm{<span\; class="notranslate">\; \&mu;</span>}}_{\mathrm{<span\; class="notranslate">\; r</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>{<span\; class="notranslate">\; (</span>\frac{\mathrm{<span\; class="notranslate">\; a</span>}}{\mathrm{<span\; class="notranslate">\; b</span>}}<span\; class="notranslate">\; )</span>}^{\mathrm{<span\; class="notranslate">\; 3</span>}}<span\; class="notranslate">\; )</span>}{{\mathrm{<span\; class="notranslate">\; \&mu;</span>}}_{\mathrm{<span\; class="notranslate">\; r</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>{<span\; class="notranslate">\; (</span>\frac{\mathrm{<span\; class="notranslate">\; a</span>}}{\mathrm{<span\; class="notranslate">\; b</span>}}<span\; class="notranslate">\; )</span>}^{\mathrm{<span\; class="notranslate">\; 3</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}{<span\; class="notranslate">\; (</span>\frac{\mathrm{<span\; class="notranslate">\; a</span>}}{\mathrm{<span\; class="notranslate">\; b</span>}}<span\; class="notranslate">\; )</span>}^{\mathrm{<span\; class="notranslate">\; 3</span>}}<span\; class="notranslate">\; )</span>}<span\; class="notranslate">\; .</span><span\; class="notranslate">\; .</span>$

For _μr = 1,000 and (a/b) = 0.99, the enhancement factor is still 2.73, so even with thin layers of magnetic material, the coupling can be significantly improved.

As mentioned above, magnetic resonators can be realized using structures comprising magnetic materials. Figure 16(a) shows a three-dimensional model of a copper and magnetic material structure 1600 driven by a square current loop around a choke point at its center. Figure 16(b) shows the interaction indicated by the magnetic field streamlines between two identical structures 1600A-B having the same properties as the one shown in Figure 16(a). Due to symmetry, and to reduce computational complexity, only half of the system is modeled. If we fix the relative orientation between the two objects and vary their center-to-center distance (the images shown are at a relative separation of 50cm), we find that at 300kHz the coupling efficiency decreases as the separation between the structures changes from 30cm to 60cm From 87% to 55%. Each of the exemplary structures 1600A-B shown comprises a 4 cm x 4 cm x 2 cm block of magnetic material bonded and fully covered by a 2 mm layer of the same magnetic material (assumed to have μ _r =1,400+j5) made of copper. Two 20cm×8cm×2cm parallelepipeds. Neglect resistive losses in the drive loop. Each structure has a calculated Q of 815.

Electromagnetic Resonators and Impedance Matching

Impedance Matching Architecture for Low Loss Inductive Components

For the purposes of this discussion, an inductive element may be any coil or loop structure ('loop') of any conductive material with or without (gapped or ungapped) a core made of magnetic material, which may also be Coupling to other systems inductively or in any other contactless manner. The element is inductive in that its impedance (including both that of the loop and the so-called 'reflected' impedance of any potentially coupled systems) has a positive reactance X and resistance R.

Consider an external circuit such as a driver circuit or a driven load or a transmission line to which the inductive element may be connected. An external circuit (eg, a driver circuit) can deliver power to the inductive element, and the inductive element can deliver power to an external circuit (eg, a driven load). The efficiency and amount of power delivered between the inductive element and the external circuit at the desired frequency may depend on the impedance of the inductive element relative to the properties of the external circuit. Power delivery between the external circuit and the inductive element can be adjusted at a desired frequency f using impedance matching networks and external circuit control techniques.

的谐振网络时以最大效率递送功率(即，以驱动电路内的最小损耗)，其中，Z_o可以是复数，并且*表示复数共轭。外部电路可以是被配置为形成A、B、C、D、DE、E、F等类的整流器的被驱动负载，并且可以在其被具有特定阻抗

的谐振网络驱动时以最大的效率接收功率(即以被驱动负载内的最小损耗)，其中，Z_o可以是复数。外部电路可以是具有特性阻抗Z_o的传输线，并且可以在被连接到阻抗

时以最大效率(即以零反射)交换功率。我们将把外部电路的特性阻抗Z_o称为可以被与之相连以便以最大效率进行功率交换的阻抗的复数共轭。The external circuit may be a driver circuit configured to form an amplifier of class A, B, C, D, DE, E, F, etc., and may be driven with a specific impedance

The resonant network of , where Z _o can be a complex number, and * denotes the complex conjugate, delivers power with maximum efficiency (ie, with minimum losses within the drive circuit). The external circuit may be a driven load configured to form a rectifier of the class A, B, C, D, DE, E, F, etc., and may have a specific impedance across it

The resonant network of is driven to receive power with maximum efficiency (ie, with minimum loss in the driven load), where Z _o can be a complex number. The external circuit can be a transmission line with characteristic impedance Z _o and can be connected to the impedance

Power is exchanged with maximum efficiency (i.e., with zero reflections). We will refer to the characteristic impedance Z _o of the external circuit as the complex conjugate of the impedance that can be connected to it for power exchange with maximum efficiency.

相差悬殊。例如，如果电感元件具有低损耗(高X/R)，其电阻R可以比外部电路的特性阻抗Z₀的实部低得多。此外，电感元件本身可以不是谐振网络。连接到电感元件的阻抗匹配网络通常可以产生谐振网络，可以调节其阻抗。Generally, the impedance R+jX of the inductive element can be compared with

There is a huge difference. For example, if the inductive element has low losses (high X/R), its resistance R can be much lower than the real part of the external circuit's characteristic impedance Z ₀ . Furthermore, the inductive element itself may not be a resonant network. An impedance matching network connected to an inductive element usually creates a resonant network whose impedance can be adjusted.

Accordingly, the impedance matching network can be designed to maximize the efficiency of power delivered between the external circuit and the inductive element (including the reflected impedance of any coupled systems). The efficiency of delivering power can be maximized by matching the combined impedance of the impedance matching network and inductive element to the characteristic impedance of the external circuit (or transmission line) at the desired frequency.

The impedance matching network can be designed to deliver a specified amount of power (including the reflected impedance of any coupled systems) between the external circuit and the inductive element. The delivered power can be determined by adjusting the complex ratio of the combined impedance of the impedance matching network and inductive element to the impedance of the external circuit (or transmission line) at the desired frequency.

An impedance matching network connected to the inductive element can create a magnetic resonator. For certain applications such as wireless power transfer using strongly coupled magnetic resonators, a high Q of the resonators may be desired. Therefore, the inductive element can be chosen to have low losses (high X/R).

The components of the matching circuit can also be chosen to have low losses, since the matching circuit can often include additional sources of loss inside the resonator. Furthermore, in high power applications and/or due to the high resonator Q, large currents travel in part of the resonator circuit and large voltages exist across certain circuit elements within the resonator. Such currents and voltages may exceed specified tolerances for particular circuit elements, and may be too high for a particular component to withstand. In some cases, it may be difficult to find or achieve components with size, cost and performance (loss and current/voltage ratings) specifications sufficient to achieve high-Q and high-power resonators for certain tuning capacitor). We disclose designs, methods, implementations and techniques that can maintain high-Q matching circuits for magnetic resonators while reducing component requirements for low loss and/or high current/voltage ratings.

Matching circuit topologies can be designed that minimize loss and current rating requirements on certain elements of the matching circuit. The topology of the circuit matching the low-loss inductive element to impedance Z ₀ may be chosen such that some of its components are outside the associated high-Q resonator in series with the external circuit. Requirements for low series losses or high current ratings for these components can be reduced. Alleviating low series loss and/or high current rating requirements on circuit components is particularly useful when components need to be variable and/or have large voltage ratings and/or low parallel losses.

Matching circuit topologies can be designed that minimize voltage rating requirements on certain components of the matching circuit. The topology of a circuit matching a low-loss inductive element to impedance Z ₀ may be chosen such that some of its components are outside the associated high-Q resonator in parallel with Z ₀ . Requirements for low shunt losses or high voltage ratings for these components can be reduced. Alleviating low shunt losses and/or high voltage requirements on circuit components is particularly useful when the components need to be variable and/or have large current ratings and/or low series losses.

The topology of the circuit matching the low-loss inductive element to the external characteristic impedance _Z0 can be chosen such that the field diagram of the associated resonant mode and thus its high Q is preserved when coupling the resonator to the external impedance. Otherwise, inefficient coupling to the desired resonant mode may occur (possibly due to coupling to other undesired resonant modes), resulting in an effective reduction in the Q of the resonator.

For applications where the low loss inductive element or the external circuit may exhibit variations, it may be necessary to dynamically adjust the matching circuit to match the inductive element to the external circuit impedance Z ₀ at the desired frequency f. Since there can generally be two tuning objectives that match or control the real and imaginary parts of the impedance level Z ₀ at the desired frequency f, there can be two variable elements in the matching circuit. For inductive elements, the matching circuit may need to include at least one variable capacitance element.

The low loss inductive element can be matched by topology using two variable capacitors or a network of two variable capacitors. For example, the variable capacitor may be a tunable butterfly capacitor having a center terminal such as a ground or other lead for connection to a power supply or load and at least one other terminal across which the tunable butterfly capacitor can be changed or tuned. Capacitance of the capacitor, or can be any other capacitor with user-configurable, variable capacitance.

The low loss inductive elements can be matched by topology using a variable capacitor or network of variable capacitors and a variable inductor or network of variable inductors.

Low loss inductive elements can be matched by topology using a variable capacitor or network of variable capacitors and a variable mutual inductance or network of variable mutual inductances that transformer couple the inductive elements to external circuits or other systems .

In some cases, it may be difficult to find or implement a tunable lumped element with sufficient size, cost, and performance specifications for high-Q, high-power, and potentially high-speed, tunable resonator designs. The topology of the circuit that matches the variable inductance element with the external circuit can be designed so that by changing the frequency, amplitude, phase, waveform, duty factor, etc. of the drive signal applied to the transistor, diode, switch, etc. in the external circuit to give some variability to the external circuitry.

Changes in the resistance R and inductance L of the inductive element at the resonance frequency may be compensated only partially or not at all. Accordingly, proper system performance may be maintained by tolerances designed into other system components or specifications. Partial adjustments achieved using fewer tunable components or less capable tunable components may be sufficient.

Impedance matching circuits can be designed to implement impedance matching at high power conditions while minimizing voltage/current rating requirements on their tunable elements and enabling finer (i.e., more precise, with higher resolution) overall tunability The expected variability of the matching circuit architecture. The topology of the circuit that matches the variable inductive element to the impedance _Z can include an appropriate combination and placement of fixed and variable elements such that the voltage/current requirements on the variable components can be reduced and can be resolved with finer tuning rate to cover the desired tuning range. Can reduce voltage/current requirements on immutable components.

The following can be achieved using the disclosed impedance matching architecture and techniques:

The power delivered from the power driven generator to the source low loss inductive element (and any other system wirelessly coupled thereto) is maximized, or the impedance mismatch between them is minimized.

Maximize the power delivered from the device low-loss inductive element (and any other system wirelessly coupled thereto) to the power-driven load, or minimize the impedance mismatch therebetween.

A controlled amount of power is delivered from the power driven generator to the source low loss inductive element (and any other system wirelessly coupled thereto), or a certain impedance relationship is achieved between them.

A controlled amount of power is delivered from the device low loss inductive element (and any other system wirelessly coupled thereto) to the power driven load, or a certain impedance relationship is achieved therebetween.

Topology for mode distribution preservation (high Q)

The resonator structure can be designed to connect to the generator or load either wirelessly (indirectly) or with a hardwired connection (directly).

Consider a general indirect coupling matching topology such as shown by the block diagram in Figure 28(a). There, the inductive element 2802, labeled (R, L) and represented by the circuit symbol for inductor, may be any inductive element discussed in this disclosure or in the references provided herein, and wherein the impedance matching circuit 2402 includes or consists of parts A and B. B may be part of a matching circuit that connects impedance 2804, Z ₀ to the rest of the circuit (combination of A and inductive elements (A+(R,L))) via a wireless connection (inductive or capacitive coupling mechanism).

The combination of A and inductive element 2802 can form a resonator 102 with associated current and charge distributions that can alone support high-Q resonator electromagnetic modes. The absence of a wired connection between the external circuits Z ₀ and B and the resonators A+(R,L) guarantees that the high-Q resonator electromagnetic modes and their current/charge distributions can take the form of their intrinsic (isolated) distributions as long as the wireless coupling The extent is not too large. That is, using an indirect coupled matching topology, the electromagnetic modes, current/charge distribution and thus the high Q of the resonator can be automatically maintained.

In cases where inductive coupling is used between the external circuit and the inductor loop, this matching topology may be referred to as indirect coupling or transformer coupling or inductive coupling. In the demonstration of wireless energy transfer over medium range distances described in the referenced Science article, this type of coupling is used to couple the power supply to the source resonator and the device resonator to the light bulb.

Next consider an example of a system where the inductive element may include the inductive element and any indirect coupling. In this case, as disclosed above, and again due to the absence of an external circuit or wired connection between the coupling system and the resonator, the coupling system may not affect Resonator electromagnetic mode distribution and current/charge distribution of the resonator. Thus, an indirect coupling matching circuit, as defined herein, can function equally well for any general inductive element that is part of a resonator, as well as for inductive elements wirelessly coupled to other systems. Throughout this disclosure, our disclosed matching topology refers to a matching topology for such general inductive elements, that is, in which any additional system can be indirectly coupled to the low loss inductive element, and it is understood that Those additional systems do not greatly affect the resonator electromagnetic mode distribution and current/charge distribution of the resonator.

Based on the above discussion, in any number of wireless power transfer systems coupling source resonators, device resonators, and intermediate resonators, the wireless magnetic (inductive) coupling between resonators does not affect the electromagnetic mode distribution and current flow of each resonator / charge distribution. Therefore, while these resonators have high (unloaded and unperturbed) Q, their (unloaded and unperturbed) Q can be preserved in the presence of wireless coupling. (Note that the loaded Q of a resonator can be reduced in the presence of wireless coupling to another resonator, but we may be interested in keeping the unloaded Q, which only involves loss mechanisms and not coupling/loading mechanisms) .

Consider a matching topology such as that shown in Figure 28(b). The capacitor shown in Figure 28(b) may represent a capacitor circuit or network. The capacitors shown can be used to form the resonator 102 and adjust the frequency and/or impedance of the source and device resonators. This resonator 102 can be coupled directly to impedance Z ₀ using a port labeled “Terminal Connection” 2808 . Fig. 28(c) shows a generalized direct coupled matching topology, where impedance matching circuit 2602 includes or consists of parts A, B and C. Here, the circuit elements in A, B, and C can be considered part of the resonator 102 and part of the impedance matching 2402 (and frequency tuning) topology. B and C may be part of matching circuit 2402 connecting impedance Z ₀ 2804 (or network terminal) to the rest of the circuit (A and the inductive element) via each single wire connection. Note that B and C can be empty (short-circuited). If we disconnect or open circuit parts B and C (i.e. those single wire connections), the combination of A and the inductive elements (R, L) can form a resonator.

High-Q resonator electromagnetic modes can cause the distribution of the voltage distribution along the inductive element to have nodes, ie locations where the voltage is zero. One node may be approximately at the center of the length of the inductive element, such as the center of the conductor (with or without magnetic material) used to form the inductive element, and at least one other node may be within A. The voltage distribution may be approximately antisymmetric along the inductive element with respect to its voltage node. High Q can be maintained by designing the matching topology (A, B, C) and/or terminal voltages (V1, V2) such that this high Q resonator electromagnetic mode distribution can be approximately maintained across the inductive element. This high-Q resonator electromagnetic mode distribution can be approximately maintained on the inductive element by keeping the voltage node of the inductive element (approximately at the center). This article provides examples of achieving these design goals.

A, B and C can be arbitrary (ie not have any particular symmetry), and V1 and V2 can be chosen such that the voltage across the inductive element is symmetric (voltage node at the center inductor). These results can be achieved using simple matching circuits but potentially complex terminal voltages, since a topology-dependent common-mode signal (V1+V2)/2 may be required on both terminals.

Consider the 'axis' connecting all the voltage nodes of the resonator, where, again, one node is approximately at the center of the length of the inductive element and the other is within A. (Note that 'axis' is actually a set of points (voltage nodes) within the circuit topology and may not necessarily correspond to a straight line axis of the actual physical structure. 'Axis' can be compared to physical axis aligned.) If the impedance seen between each of the two points and the point on the 'axis' (i.e. the voltage node point of the resonator) is the same, then the two points of the resonator are relative to the ' axis' is electrically symmetrical.

B and C can be the same (C=B), and as shown in Figure 28(d), the two terminals can be connected to the 'axis' defined above and driven by opposite voltages (V2=-V1) ' for any two points of an electrically symmetric resonator (A+(R,L)). The two points of electrical symmetry of the resonator 102 may be the two points of electrical symmetry on the inductor loop. The two electrical symmetry points of the resonator may be the two electrical symmetry points inside A. If two points of electrical symmetry (to which equal parts B and C are each connected) are inside A, then A may need to be designed such that these points of electrical symmetry are accessible as connection points within the circuit. This topology may be referred to as a 'balanced drive' topology. These balanced drive examples may have the advantage that any common mode signal that may appear on ground (and may not reach the resonator) may be automatically rejected, eg due to disturbances at the external circuit or power grid. In some balanced drive examples, this topology may require more components than others.

In other examples, C may be chosen as a short circuit and the corresponding terminal connected to ground (V=0) and any point on the 'axis' of electrical symmetry (zero voltage) of the resonator, and B connected to a Any other point of the resonator on the 'axis' of symmetry, as shown in Figure 28(e). The ground point on the 'axis' of electrical symmetry may be a voltage node on the inductive element, approximately at the center of its conductor length. The grounding point on the 'axis' of electrical symmetry may be inside circuit A. Where the grounding point on the 'axis' of electrical symmetry is internal to A, it may be necessary to design A to include electrically accessible one such point on the 'axis' of electrical symmetry, i.e. where a connection can be made.

This topology may be referred to as an 'unbalanced drive' topology. An approximately anti-symmetrical voltage distribution along the electromagnetic modes of the inductive element may be approximately maintained even though the resonator may not be driven perfectly symmetrically. The reason is that the high Q and large associated R vs. _Z0 mismatch need allow a small current to flow through B and ground compared to the much larger current that can flow inside the resonator (A+(R,L)). In this case, the perturbation on the resonator mode can be weak and the position of the voltage node can be kept approximately at the center position of the inductive element. These unbalanced drive examples may have the advantage that they can be implemented using simple matching circuits and there is no limitation on the drive voltage at the V1 terminal. In some unbalanced drive examples, additional design may be required to reduce common-mode signals that may appear at the ground terminal.

A direct coupled impedance matching circuit generally comprising or consisting of parts A, B and C as shown in Figure 28(c) can be designed such that the wires and components of the circuit do not disturb the electromagnetic modes of the inductive element and/or resonator The electric and magnetic fields are distributed and thus keep the resonator Q high. The wires and metal components of the circuit can be oriented perpendicular to the electric field lines of the electromagnetic mode. The wires and components of the circuit can be placed in areas where the electric and magnetic fields of the electromagnetic modes are weak.

Topologies used to alleviate low series loss and high current rating requirements on components

并且流过端子的电流比流过电感元件的电流小得多。因此，与端子(诸如在直接耦合的B、C(图28(C))中)直接串联地连接的元件可以不载送高电流。然后，即使匹配电路具有有损耗元件，出现在与端子串联的元件中的电阻性损耗可以不导致谐振器的高Q的显著减小。也就是说，那些串联元件中的电阻性损耗可以不显著地降低从Z₀到电感元件或相反的功率传输的效率。因此，对于这些组件而言，可能不需要用对低串联损耗和/或高电流额定值的严格要求。通常，此类降低的要求可以得到可以被设计到高Q和/或高功率阻抗匹配和谐振器拓扑结构中的组件的更广泛选择。这些降低的要求在扩展可以在这些高Q和/或高功率阻抗匹配电路中使用的可变和/或高电压和/或低并联损耗组件的种类方面尤其有帮助。If the matching circuit used to match the small resistance R of the low-loss inductive element with the larger characteristic impedance Z ₀ of the external circuit can be considered lossless, then

And the current flowing through the terminal is much smaller than the current flowing through the inductive element. Thus, elements connected directly in series with terminals (such as in direct coupled B, C (FIG. 28(C))) may not carry high currents. Then, even if the matching circuit has lossy elements, resistive losses occurring in the elements in series with the terminals may not result in a significant reduction in the high Q of the resonator. That is, resistive losses in those series elements may not significantly reduce the efficiency of power transfer from Z ₀ to the inductive element or vice versa. Therefore, stringent requirements for low series losses and/or high current ratings may not be required for these components. In general, such reduced requirements can result in a wider selection of components that can be designed into high-Q and/or high-power impedance matching and resonator topologies. These reduced requirements are especially helpful in expanding the variety of variable and/or high voltage and/or low shunt loss components that can be used in these high Q and/or high power impedance matching circuits.

Topologies used to alleviate low shunt losses and high voltage rating requirements on components

If the matching circuit used to match the small resistance R of the low-loss inductive element with the larger characteristic impedance _Z of the external circuit as described above is lossless, then using the preceding analysis,

$<span\; class="notranslate">\; |</span>{\mathrm{<span\; class="notranslate">\; V</span>}}_{{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}}<span\; class="notranslate">\; /</span>{\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; load</span>}}<span\; class="notranslate">\; |</span><span\; class="notranslate">\; =</span><span\; class="notranslate">\; |</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}}{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}<span\; class="notranslate">\; /</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; R</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; R</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; wxya</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; |</span><span\; class="notranslate">\; \&ap;</span>\sqrt{\mathrm{<span\; class="notranslate">\; R</span>}<span\; class="notranslate">\; /</span>{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}}<span\; class="notranslate">\; \&Center\; Dot;</span>{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; =</span>\sqrt{{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; R</span>}}<span\; class="notranslate">\; /</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; /</span>\mathrm{<span\; class="notranslate">\; R</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; ,</span>$

Also, for low loss (high X/R) inductive elements, the voltage across the terminals can typically be much smaller than the voltage across the inductive element. Therefore, elements that are directly paralleled to the terminals may not need to withstand high voltages. Then, even if the matching circuit has lossy elements, resistive losses occurring in the elements in parallel with the terminals may not result in a significant reduction in the high Q of the resonator. That is, resistive losses in those parallel elements may not significantly reduce the efficiency of power transfer from Z ₀ to the inductive element or vice versa. Therefore, stringent requirements for low shunt losses and/or high voltage ratings may not be required for these components. In general, such reduced requirements can result in a wider selection of components that can be designed into high-Q and/or high-power impedance matching and resonator topologies. These reduced requirements are especially helpful in expanding the variety of variable and/or high current and/or low series loss components that can be used in these high Q and/or high power impedance matching and resonator circuits.

Note that the above design principles can reduce the current and voltage on the various components differently, as they suggest in different ways to use a network in series with Z ₀ (such as directly coupled B,C) or to use a network in parallel with Z ₀ . network. The preferred topology for a given application may depend on the availability of low series loss/high current rating or low parallel loss/high voltage rating components.

Fixtures for fine tunability and relief from high rating requirements on variable elements Combination with variable elements

circuit topology

It can be difficult or too expensive to obtain variable circuit elements with satisfactorily low losses and high voltage or current ratings. In this disclosure we describe that a combination of fixed and variable elements can be incorporated such that a large voltage or current can be distributed to fixed elements in a circuit (which are more likely to have appropriate voltage and current ratings) The voltage and current ratings on the variable elements require an impedance matching topology.

Variable circuit elements may have larger tuning ranges than those required for a given impedance matching application, and, in those cases, fine tuning resolution may be difficult to obtain using only such large-range elements. In this disclosure we describe impedance matching topologies that combine both fixed and variable elements such that finer tuning resolution can be achieved with the same variable elements.

Thus, a topology using a combination of both fixed and variable elements can yield both advantages: reduced voltage across or current through sensitive tuning components in the circuit and finer tuning resolution. Note that the maximum achievable tuning range can be related to the maximum reduction in voltage across or current passing through a tunable component in the circuit design.

Component topology

A single variable circuit element may be implemented by a topology that uses a combination of fixed and variable components connected in series or in parallel to achieve reduced rating requirements and finer tuning resolution for the variable components (same as above The network of elements in question is reversed). This can be proven mathematically by the fact that:

If x _|total| = x _|fixed| + x _|variable| ,

Then Δx _|total| /x _|total| ＝Δx _|variable| /(x _|fixed| +x _|variable| ),

And X _variable /X _total =X _variable /(X _fixed +X _variable ), where x _|subscript| is any component value (e.g. capacitance, inductance), X is voltage or current, and "+sign" indicates the appropriate combination of components (series addition or parallel addition). Note that the subscript format for x _|subscript| is chosen to easily distinguish it from the radius of the area enclosed by the circular inductive element (eg, x, _x1, etc.).

Furthermore, this principle can be used to realize a certain type of variable electrical element (eg capacitance or inductance) by using different types of variable elements, if the different types of variable elements are suitably combined with other fixed elements.

In summary, a topology optimization algorithm can be applied that determines the required number, location, placement, type and value of fixed and variable elements with the required tunable range as optimization constraints and determines the current and/or The minimization of the voltage is taken as the optimization goal.

example

In the following schematic diagrams we show different specific topology implementations for impedance matching of low loss inductive elements and resonator design for low loss inductive elements. In addition, we indicate for each topology: which of the above principles to use, equations giving the values of variable elements that can be used to achieve matching, the range of complex impedances that can be matched (described using inequalities and Smith charts). For these examples we assume that Z ₀ is a real number, but the extension to the characteristic impedance with a non-zero imaginary part is straightforward, since it only implies small adjustments in the required values of the components of the matching network. _{We will use the convention of normalization (divide by Z 0 )} implied by the subscript n on the volume.

Figure 29 shows two examples of transformer coupled impedance matching circuits where the two tunable elements are capacitors and the mutual inductance between two inductive elements. If we define X ₂ =ωL ₂ for Figure 29(a) and X ₂ =ωL ₂ -1/ωC ₂ for Figure 29(b), and X≡ωL, respectively, then the required value of the tunable element yes:

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="RX"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">RX</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; no</span>}}}$

$\mathrm{<span\; class="notranslate">\; \&omega;\; M</span>}<span\; class="notranslate">\; =</span>\sqrt{{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}\mathrm{<span\; class="notranslate">\; R</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; x</span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; no</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span>}<span\; class="notranslate">\; .</span>$

A particularly simple design could be to choose _X2 =0 for the topology of Figure 29(b). In this case, these topologies can match impedances satisfying the following inequalities:

R _n > 0, X _n > 0,

This is shown by the area enclosed by the thick line on the Smith chart of Fig. 29(c).

Given a pre-selected fixed M, the above mentioned matching topology with tunable _C2 can be used instead.

Figure 30 shows six examples (a)-(f) of direct coupled impedance matching circuits (where the two tunable elements are capacitors) and six examples (h)-(m) of direct coupled impedance matching circuits (where , the two tunable elements are a capacitor and an inductor). For the topology of Figure 30(a), (b), (c), (h), (i), (j), a common mode signal may be required at both terminals to maintain resonance at the center of the inductive element to the voltage node of the device and thus maintain a high Q. Note that these examples can be described as implementations of the general topology shown in Figure 28(c). For the symmetrical topologies of Figure 30(d), (e), (f), (k), (l), (m), it may be necessary to drive both terminals anti-symmetrically (balanced drive) to keep the inductance The voltage node of the resonator is at the center of the element and thus keeps the Q high. Note that these examples can be described as implementations of the general topology shown in Figure 28(d). It will be appreciated that a network of capacitors as used herein may generally refer to any circuit topology comprising one or more capacitors, including but not limited to any circuit specifically disclosed herein using capacitors or any other equivalent or different Circuit structure(s), unless otherwise specified or clear from the context.

Let us define Z=R+jωL for Fig. 30(a), (d), (h), (k), and Z=R+jωL for Fig. 30(b), (e), (i), (l), respectively. Z=R+jωL+1/jωC ₃ and Z=(R+jωL)∥(1/jωC ₃ ) for Fig. 30(c), (f), (j), (m), where the symbol " ‖ means "parallel combination of...", R≡Re{Z}, X≡Im{Z}. For Figure 30(a)~(f), the required value of the tunable element can be given by the following formula :

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; -</span>\sqrt{{\mathrm{<span\; class="notranslate">\; x</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>}}{{\mathrm{<span\; class="notranslate">\; x</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="R"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">R</figure-callout></span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; ,</span>$

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="C"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; =</span>\frac{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; X\&omega;</span>}{\mathrm{<span\; class="notranslate">\; C</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}<span\; class="notranslate">\; ,</span>$

And these topologies can match impedances satisfying the following inequalities:

${\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; \&le;</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; ,</span>{\mathrm{<span\; class="notranslate">\; x</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; \&Greater\; Equal;</span>\sqrt{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>}$

This is shown by the area enclosed by the thick line on the Smith chart of Figure 30(g).

For Figure 30(h)~(m), the required value of the tunable element can be given by the following formula:

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; +</span>\sqrt{{\mathrm{<span\; class="notranslate">\; x</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>}}{{\mathrm{<span\; class="notranslate">\; x</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="R"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">R</figure-callout></span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; ,</span>$

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="L"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">L</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; =</span><span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; X\&omega;</span>}{\mathrm{<span\; class="notranslate">\; C</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}<span\; class="notranslate">\; .</span>$

Figure 31 shows three examples (a)-(c) of direct coupled impedance matching circuits (where the two tunable elements are capacitors) and three examples (e)-(g) of direct coupled impedance matching circuits (where , the two tunable elements are a capacitor and an inductor). For the topologies of Figure 31 (a), (b), (c), (e), (f), (g), the ground terminal is connected between two capacitors 2C ₁ of equal value (i.e. at axis of symmetry of the main resonator) to keep the voltage node of the resonator at the center of the inductive element and thus keep the Q high. Note that these examples can be described as implementations of the general topology shown in Figure 28(e).

Let us define Z=R+jωL for Fig. 31(a), (e), Z=R+jωL+1/jωC for Fig. 31(b), (f) and for Fig. ₃₁ ( c), Z=(R+jωL)∥(1/jωC ₃ ) for (g), and then R≡Re{Z}, X≡Im{Z}. Then, for Figure 31(a)~(c), the required value of the tunable element can be given by the following formula:

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; -</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}\sqrt{{\mathrm{<span\; class="notranslate">\; x</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>}}{{\mathrm{<span\; class="notranslate">\; x</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="R"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">R</figure-callout></span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; ,</span>$

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="C"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; =</span>\frac{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; X\&omega;</span>}{\mathrm{<span\; class="notranslate">\; C</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; -</span>\frac{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; ,</span>$

And these topologies can match impedances satisfying the following inequalities:

${\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; \&le;</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; ,</span>{\mathrm{<span\; class="notranslate">\; x</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; \&Greater\; Equal;</span>\sqrt{\frac{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>$

This is shown by the area enclosed by the thick line on the Smith chart of Figure 31(d).

For Figure 31(e)~(g), the required value of the tunable element can be given by the following formula:

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}\sqrt{{\mathrm{<span\; class="notranslate">\; x</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>}}{{\mathrm{<span\; class="notranslate">\; x</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="R"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">R</figure-callout></span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; ,</span>$

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="L"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">L</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; X\&omega;</span>}{\mathrm{<span\; class="notranslate">\; C</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; -</span>\frac{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}{\mathrm{<span\; class="notranslate">\; 2</span>}}}{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}<span\; class="notranslate">\; .</span>$

Figure 32 shows three examples (a)-(c) of direct coupled impedance matching circuits (where the two tunable elements are capacitors) and three examples (e)-(g) of direct coupled impedance matching circuits (where , the two tunable elements are a capacitor and an inductor). For the topologies of Figure 32(a), (b), (c), (e), (f), (g), a ground terminal can be connected at the center of the inductive element to maintain the voltage node and thus keep Q high. Note that these examples can be described as implementations of the general topology shown in Figure 28(e).

Let us define Z=R+jωL for Fig. 32(a), Z=R+jωL+1/ _jωC3 for Fig. 32(b) and Z=(R+ jωL)‖(1/jωC ₃ ), then R≡Re{Z}, X≡Im{Z}. Then, for Figure 32(a)~(c), the required value of the tunable element can be given by the following formula:

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; -</span>\sqrt{\frac{{\mathrm{<span\; class="notranslate">\; x</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 2</span>}{\mathrm{<span\; class="notranslate">\; R</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}}}{{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="x"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">x</figure-callout></span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="R"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">R</figure-callout></span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; ,</span>$

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="C"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; =</span>\frac{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; X\&omega;</span>}{\mathrm{<span\; class="notranslate">\; C</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; -</span>\frac{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>\frac{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}\mathrm{<span\; class="notranslate">\; X\&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}{\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; )</span>}}<span\; class="notranslate">\; ,</span>$

where k is defined by M'=-kL', where L' is the inductance of each half of the inductor loop, and M' is the mutual inductance between the two halves, these topologies can match impedances satisfying the following inequalities:

${\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; \&le;</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; ,</span>{\mathrm{<span\; class="notranslate">\; x</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; \&Greater\; Equal;</span>\sqrt{\mathrm{<span\; class="notranslate">\; 2</span>}{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>}$

This is shown by the area enclosed by the thick line on the Smith chart of Figure 32(d).

For Figure 32(e)~(g), the required value of the tunable element can be given by the following formula:

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; +</span>\sqrt{\frac{{\mathrm{<span\; class="notranslate">\; x</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 2</span>}{\mathrm{<span\; class="notranslate">\; R</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}}}{{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="x"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">x</figure-callout></span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="R"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">R</figure-callout></span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; ,</span>$

In the circuits of Figures 30, 31, 32, capacitor _C2 or inductor _L2 (or two capacitors _2C2 or two inductors _L2 /2) are connected in series with the terminals and may not need to have very low series losses or withstand large currents.

Figure 33 shows six examples (a)-(f) of direct coupled impedance matching circuits (where the two tunable elements are capacitors) and six examples (h)-(m) of direct coupled impedance matching circuits (where , the two tunable elements are a capacitor and an inductor). For the topology of Figure 33(a), (b), (c), (h), (i), (j), a common mode signal may be required at both terminals to maintain resonance at the center of the inductive element voltage node of the device and therefore maintains a high Q. Note that these examples can be described as implementations of the general topology shown in Figure 28(c), where B and C are shorted and A is unbalanced. For the symmetrical topologies of Figure 33(d), (e), (f), (k), (l), (m), it may be necessary to drive both terminals anti-symmetrically (balanced drive) to keep the inductance The voltage node of the resonator is at the center of the element and therefore remains high Q. Note that these examples can be described as implementations of the general topology shown in Figure 28(d), where B and C are shorted and A is balanced.

Let us define Z=R+jωL for Fig. 33(a), (d), (h), (k) and for Fig. 33(b), (e), (i), (l) respectively Z=R+jωL+1/jωC ₃ and Z=(R+jωL)∥(1/jωC ₃ ) for Fig. 33(c), (f), (j), (m), and then R≡ Re{Z}, X≡Im{Z}. Then, for Figure 33(a)~(f), the required value of the tunable element can be given by the following formula:

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}\sqrt{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>}}<span\; class="notranslate">\; ,</span>$

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="C"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}}\sqrt{\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; ,</span>$

And these topologies can match impedances satisfying the following inequalities:

${\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; \&le;</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; ,</span>{\mathrm{<span\; class="notranslate">\; x</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; \&Greater\; Equal;</span>\sqrt{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>}$

This is shown by the area enclosed by the thick line on the Smith chart of Figure 33(g).

For Figure 35(h)~(m), the required value of the tunable element can be given by the following formula:

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}\sqrt{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>}}<span\; class="notranslate">\; ,</span>$

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="L"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">L</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; =</span>\frac{{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}}{\sqrt{\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}}<span\; class="notranslate">\; .</span>$

Figure 34 shows three examples (a)-(c) of direct coupled impedance matching circuits (where the two tunable elements are capacitors) and three examples (e)-(g) of direct coupled impedance matching circuits (where , the two tunable elements are a capacitor and an inductor). For the topologies of Figure 34(a), (b), (c), (e), (f), (g), the ground terminal is connected between two capacitors _2C of equal value (i.e. at axis of symmetry of the main resonator) to keep the voltage node of the resonator at the center of the inductive element and thus keep the Q high. Note that these examples can be described as implementations of the general topology shown in Figure 28(e).

Let us define Z=R+jωL for Fig. 34(a), (e), Z=R+jωL+1/jωC ₃ for Fig. 34(b), (f) and for Fig. 34( c), Z=(R+jωL)∥(1/jωC ₃ ) of (g), then R≡Re{Z}, X≡Im{Z}. For Figure 34(a)~(c), the required value of the tunable element can be given by the following formula:

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}\sqrt{\frac{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>}<span\; class="notranslate">\; ,</span>$

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="C"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{{\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}}\sqrt{\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; ,</span>$

And these topologies can match impedances satisfying the following inequalities:

${\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; \&le;</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; ,</span>{\mathrm{<span\; class="notranslate">\; x</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; \&Greater\; Equal;</span>\sqrt{\frac{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>$

This is shown by the area enclosed by the thick line on the Smith chart of Figure 34(d).

For Figure 34(e)~(g), the required value of the tunable element can be given by the following formula:

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}\sqrt{\frac{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>}}<span\; class="notranslate">\; ,</span>$

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="L"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">L</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 2</span>}{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}}{\sqrt{\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}}<span\; class="notranslate">\; .</span>$

Figure 35 shows three examples of direct coupled impedance matching circuits where the two tunable elements are capacitors. For the topology of Figure 35, a ground terminal can be connected at the center of the inductive element to maintain the voltage node of the resonator at this point and thus keep the Q high. Note that these examples can be described as implementations of the general topology shown in Figure 28(e).

Let us define Z=R+jωL for Fig. 35(a), Z=R+jωL+1/ _jωC3 for Fig. 35(b) and Z=(R+ jωL)‖(1/jωC ₃ ), then R≡Re{Z}, X≡Im{Z}. The required value of the tunable element can then be given by:

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="C"\; filenames="HPA00001374832500031.png,HPA00001374832500301.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 2</span>}}{\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; a</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\sqrt{{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}\mathrm{<span\; class="notranslate">\; R</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; a</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span>}}<span\; class="notranslate">\; ,</span>$

$\mathrm{<span\; class="notranslate">\; \&omega;</span>}{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="C"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">C</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 2</span>}}{\mathrm{<span\; class="notranslate">\; x</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; a</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; +</span>\sqrt{{\mathrm{<span\; class="notranslate">\; Z</span>}}_{\mathrm{<span\; class="notranslate">\; o</span>}}\mathrm{<span\; class="notranslate">\; R</span>}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; a</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; )</span>}}<span\; class="notranslate">\; ,</span>$

和k，其中，L’是每一半电感元件的电感，并且M’是两半之间的互感。这些拓扑结构可以匹配满足以下不等式的阻抗：Among them, it is defined by M'=-kL'

and k, where L' is the inductance of each half of the inductive element, and M' is the mutual inductance between the two halves. These topologies can match impedances satisfying the following inequalities:

${\mathrm{<span\; class="notranslate">\; x</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; \&Greater\; Equal;</span>\sqrt{\frac{{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}<span\; class="notranslate">\; )</span>}{\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; \&gamma;</span>}{\mathrm{<span\; class="notranslate">\; R</span>}}_{\mathrm{<span\; class="notranslate">\; no</span>}}}}<span\; class="notranslate">\; ,</span>$

in

$\mathrm{<span\; class="notranslate">\; \&gamma;</span>}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 6</span>}\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="k"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">k</figure-callout></span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; 2</span>}\mathrm{<span\; class="notranslate">\; k</span>}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="k"\; filenames="HPA00001374832500301.png,HPA00001374832500351.png"\; state="\{\{state\}\}">k</figure-callout></span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; \&le;</span>\mathrm{<span\; class="notranslate">\; 1</span>}$

Illustrated by the areas enclosed by thick lines on the three Smith charts shown in Figure 35(d) for k=0, Figure 35(e) for k=0.05, and Figure 35(f) for k=1 . Note that for 0<k<1, this topology is able to match the two disconnected regions of the Smith chart.

In the circuits of Figures 33, 34, 35, the capacitor _C2 or the inductor _L2 (or one of the two capacitors _2C2 or one of the two inductors _2L2 ) is connected in parallel with the terminals, so there may be no need to have High voltage rating. In the case of the two capacitors 2C ₂ or the two inductors 2L ₂ , both may not need to have high voltage ratings because approximately the same current flows through them, so they experience approximately the same voltage across them.

For the topologies of FIGS. 30-35 shown using capacitor C ₃ , the use of capacitor C ₃ can result in finer tuning of frequency and impedance. For the topologies of Figures 30-35, the use of a fixed capacitor _C3 in series with the inductive element ensures that most of the high inductive element voltage will be across this fixed capacitor _C3 , thus potentially relieving the impedance matching Voltage rating requirements for other components of the circuit (some of which may be variable). Whether such a topology is preferable depends on the availability, cost and size of suitably fixed, tunable components.

In all of the above examples, a pair of equal value variable capacitors with no common terminals can be implemented using a bank of capacitors or a varactor or diode bank or array that is biased and controlled to tune their value as an ensemble. A pair of peers with one common terminal can be implemented using tunable butterfly capacitors or any other tunable or variable capacitors or varactors or diode banks or arrays that are biased and controlled to tune their capacitance values as an ensemble variable value capacitors.

Another criterion that may be considered when selecting an impedance matching network is the response of the network to frequencies other than the desired operating frequency. The signal generated in the external circuit to which the inductive element is coupled may not be monochromatic at the desired frequency, but periodic with the desired frequency, eg a drive signal of a switching amplifier or a reflected signal of a switching rectifier. In some such cases, it may be desirable to suppress the amount of higher order harmonics entering an inductive element (eg, to reduce the radiation of these harmonics from this element). The choice of impedance matching network may then be one that sufficiently suppresses the amount of such harmonics entering the inductive element.

The impedance matching network can be such that when the external periodic signal is a signal that can be considered to function as a voltage source signal such as the driving signal of a class D amplifier with a series resonant load, the external circuit experiences The impedance is high such that at higher frequencies little current flows through the inductive element. Between the topologies of Figures 30-35, those using inductor _L2 may then be preferred since this inductor presents a high impedance at high frequencies.

The impedance matching network can be such that when the external periodic signal is a signal that can be considered to function as a current source signal, the impedance experienced by the external circuit at frequencies higher than the fundamental is low such that at higher frequencies the inductive element Almost no voltage is induced across them. Among the topologies of Figures 30-35, those using capacitor _C2 are preferred because this capacitor exhibits low impedance at high frequencies.

Figure 36 shows four examples of variable capacitance using a network of one variable capacitor and the rest fixed capacitors. Using these network topologies, fine tunability of the total capacitor value can be achieved. Furthermore, the topologies of Fig. 36(a), (c), (d) can be used to reduce the voltage across the variable capacitor since most of the voltage can be distributed across the fixed capacitor.

Figure 37 shows two examples of variable capacitance using a network of variable inductors and fixed capacitors. In particular, these networks can provide implementations for variable reactance, and at frequencies of interest, values of variable inductors can be used such that each network conforms to a net negative variable reactance, which can Effectively a variable capacitor.

Tunable elements such as tunable capacitors and tunable inductors may be mechanically tunable, electrically tunable, thermally tunable, and the like. The tunable elements may be variable capacitors or inductors, varactors, diodes, Schottky diodes, reverse biased PN diodes, varactor arrays, diode arrays, Schottky diode arrays, and the like. The diodes may be Si diodes, GaN diodes, SiC diodes, or the like. GaN and SiC diodes can be particularly attractive for high-power applications. The tunable element may be an electrically switched capacitor bank, an electrically switched mechanically tunable capacitor bank, an electrically switched varactor array bank, an electrically switched transformer coupled inductor bank, and the like. Tunable elements can be combinations of the above listed elements.

As noted above, the efficiency of power transfer between coupled high-Q magnetic resonators can be affected by how closely the resonators are matched at the resonant frequency and how well their impedances are matched to the power sources and power consumers in the system. Since a variety of external factors including the relative positions of extraneous objects or other resonators in the system or changes in those relative positions can change the resonant frequency and/or input impedance of a high-Q magnetic resonator, a tunable impedance network may be required at each maintain an adequate level of power transfer in such an environment or operating situation.

The capacitance values of the capacitors shown can be adjusted to adjust the resonant frequency and/or impedance of the magnetic resonator. Capacitors may be adjusted electrically, mechanically, thermally or by any other known method. It may be adjusted manually or automatically, such as in response to a feedback signal. It can be adjusted to achieve certain power transfer efficiencies or other operating characteristics between the power source and the power consuming device.

The inductance values of the inductors and inductive elements in the resonator can be adjusted to adjust the frequency and/or impedance of the magnetic resonator. The inductance can be adjusted using a coupling circuit including tunable components such as tunable capacitors, inductors and switches. The inductance can be tuned using a transformer-coupled tuning circuit. The inductance can be adjusted by switching different sections of conductors in the inductive element on and off and/or using ferromagnetic tuning and/or moving iron tuning, etc.

The resonant frequency of the resonator may be tuned or may be allowed to go to a lower or higher frequency. The input impedance of the resonator can be adjusted or allowed to go to lower or higher impedance values. The amount of power delivered by the source and/or received by the device may be adjusted or allowed to change to a lower or higher power level. The amount of power delivered to the source and/or received by the device from the device resonator may be adjusted or may be allowed to go to a lower or higher power level. The resonator input impedance, resonant frequency and power level can be adjusted according to the power consumers in the system and according to the objects or materials near the resonator. The resonator input impedance, frequency and power level can be adjusted manually or automatically, and can be made in response to feedback or control signals or algorithms.

The circuit element may be connected directly (ie by physical electrical contact) to the resonator, for example to the ends of conductors forming the inductive element and/or terminal connectors. Circuit elements may be soldered, welded, crimped, glued, clamped, or tightly positioned to conductors, or attached using a variety of electrical components, connectors, or connection techniques. Power sources and power consumers can be connected to the magnetic resonator directly or indirectly or inductively. Electrical signals can be supplied to or taken from the resonator via terminal connections.

Those skilled in the art will appreciate that in practical implementations of the principles described herein, there may be values of actual components (capacitors, inductors, resistors, etc.) , current, etc.) versus those proposed by symmetry or antisymmetry and the true geometric location of a point (such as the connection point or 'axis' point of a ground terminal close to the center of an inductive element) versus that proposed by symmetry or The associated tolerance or acceptable change in position proposed by the antisymmetry.

Example

System Block Diagram

We disclose embodiments of high-Q resonators for use in wireless power transfer systems that can wirelessly power or charge devices at medium-range distances. High-Q resonator wireless power transfer systems can also wirelessly power or charge devices with magnetic resonators that differ in size, shape, composition, arrangement, etc. from any source resonator in the system.

Figure 1(a)(b) shows high-level diagrams of two exemplary dual-resonator systems. Each of these exemplary systems has a single source resonator 102S or 104S and a single device resonator 102D or 104D. Figure 38 shows a high level block diagram of the system with some features highlighted. A wirelessly powered or charged device 2310 may include a device resonator 102D, a device power and control circuit 2304, etc., and one or more devices 2308 to which DC or AC or both AC and DC power is diverted, or by which composition. Energy or power sources for the system may include source power and control circuitry 2302, source resonator 102S, and the like. The one or more devices 2308 receiving power from the device resonator 102D and the power and control circuit 2304 may be any kind of device 2308 as previously described. The device resonator 102D and circuitry 2304, when in the vicinity of the source resonator 102S, deliver power to one or more devices 2308, which can be used to recharge the batteries of the one or more devices, directly charge the one or more one device or both simultaneously.

The source and device resonators can be separated by many meters, or they can be very close to each other, or they can be separated by any distance in between. The source and device resonators can be laterally or axially offset from each other. The source and device resonators may be directly aligned (no lateral offset), or they may be offset by a few meters, or anything in between. The source and device resonators can be oriented such that the surface areas enclosed by their inductive elements are approximately parallel to each other. The source and device resonators can be oriented such that the surface areas enclosed by their inductive elements are approximately perpendicular to each other, or they can be oriented for any relative angle therebetween (0 to 360 degrees).

The source and device resonators can be freestanding, or they can be enclosed in a housing, container, sleeve or case. These various enclosures can be composed of almost any kind of material. For some applications, low loss tangent materials such as Teflon, REXOLITE, Styrene, etc. may be preferred. Source and device resonators can be integrated in power supplies and power consumers. For example, source and device resonators can be integrated into keyboards, computer mice, displays, cell phones, etc. such that they are invisible outside of these devices. Source and device resonators can be separated from power sources and power consumers in the system and can be connected by standard or custom wires, cables, connectors or plugs.

Source 102S can be powered from many DC or AC voltage, current and power sources including a computer's USB port. The source 102S may be powered from the mains, from a wall plug, from a battery, from a power supply, from an engine, from a solar cell, from a generator, from another source resonator, or the like. Source power and control circuitry 2302 may include circuits and components that isolate the source electronics from the power supply so that any reflected power or signals are not coupled outside through the source input terminals. Source power and control circuitry 2302 may include power factor correction circuitry and may be configured to monitor power usage for monitoring accounting, billing, control, and similar functions.

The system can be bi-directional. That is, the energy or power generated or stored in the device resonator can be fed back to a power source including a power grid, a battery, any kind of energy storage unit, etc. The source power and control circuitry may include power factor correction circuitry and may be configured to monitor power usage for monitoring accounting, billing, control and similar functions for bi-directional energy flow. Wireless energy transfer systems can enable or facilitate vehicle-to-grid (V2G) applications.

The source and device may have tuning capabilities that allow adjustment of the operating point to compensate for varying environmental conditions, disturbances and load conditions (which can affect the operation of the source and device resonators and the efficiency of energy exchange). The tuning capability can also be used to multiplex power delivery to multiple devices, from multiple sources to multiple systems, to multiple repeaters or repeaters, and the like. The tuning capability may be controlled manually or automatically, and may be performed continuously, periodically, intermittently, or at predetermined times or intervals.

For example, device resonators and device power and control circuitry can be integrated into any part of the device, such as a battery compartment or device cover or sleeve or motherboard, and can be integrated alongside standard rechargeable batteries or other energy storage elements . A device resonator may include a device field reshaper, which may shield any combination of device resonator elements and device power and control electronics from electromagnetic fields used for power transfer, and which may deflect the resonator field away from lossy Device resonator components and device power and control electronics. Magnetic materials and/or high conductivity field reshapers can be used to increase the perturbed quality factor Q of the resonator and increase the perturbed coupling factor of the source and device resonators.

The source resonator and source power and control circuitry can be integrated into any type of furniture, structure, rug, carpet, photo frame (including digital photo frame, electronic frame), plug-in, electronic device, vehicle, etc. The source resonator may include a source field reshaper, which may shield any combination of source resonator elements and source power and control electronics from the electromagnetic fields used for power transfer, and which may deflect the resonator field away from lossy Source resonator elements and source power and control electronics. Magnetic materials and/or high conductivity field reshapers can be used to increase the perturbed quality factor Q of the resonator and increase the perturbed coupling factor of the source and device resonators.

A block diagram of subsystems in an example of a wireless power device is shown in FIG. 39 . The power and control circuitry can be designed to transform the AC power from the device resonator 102D and convert it to stable DC power suitable for powering or charging the device. The power and control circuitry can be designed to convert AC power at one frequency from the device resonator to AC power at a different frequency suitable for powering or charging the device. The power and control circuits may include impedance matching circuit 2402D, rectification circuit 2404, voltage limiting circuit (not shown), current limiting circuit (not shown), AC to DC converter 2408 circuit, DC to DC converter 2408 circuit, DC to AC converter 2408 circuit, AC to AC converter 2408 circuit, battery charging control circuit (not shown), etc., or consists of them.

The impedance matching 2402D network may be designed to maximize the power delivered between the device resonator 102D and the device power and control circuit 2304 at the desired frequency. Impedance matching components can be chosen and connected such that the high Q of the resonator is maintained. Depending on operating conditions, the impedance matching circuit 2402D may be changed or tuned to control the power delivered from the source to the device, from the source to the device resonator, between the device resonator and the device power and control circuitry, etc. Power, current and voltage signals can be monitored at any point in the device circuitry, and feedback algorithmic circuits and techniques can be used to control components to achieve desired signal levels and system operation. The feedback algorithm can be implemented using analog or digital circuit technology, and the circuit can include microprocessors, digital signal processors, field programmable gate array processors, and the like.

The third block of FIG. 39 shows a rectifier circuit 2404 that can rectify AC voltage power from the device resonator to a DC voltage. In this configuration, the output power of the rectifier 2404 may be the input to a voltage clamp circuit. A voltage clamp circuit (not shown) may limit the maximum voltage to the input of DC-to-DC converter 2408D or DC-to-AC converter 2408D. In general, it may be desirable to use a DC-to-DC/AC converter with a large input voltage dynamic range so that large variations in device location and operation can be tolerated while sufficient power is delivered to the device. For example, the voltage level at the output of the rectifier can fluctuate and reach high levels as the power input and load characteristics of the device vary. As a device performs different tasks, it may have varying power requirements. Changing power demands can cause high voltages at the output of the rectifier as the load characteristics vary. Likewise, as the device and the device resonator are brought closer and further away from the source, the power delivered to the device resonator may change and cause a change in the voltage level at the output of the rectifier. The voltage clamp circuit prevents the voltage output from the rectifier circuit from exceeding a predetermined value within the operating range of the DC to DC/AC converter. A voltage clamping circuit can be used to extend the operating mode and range of the wireless energy transfer system.

The next block of the device's power and control circuitry is the DC to DC converter 2408D that can generate a regulated DC output voltage. The DC-to-DC converter can be a boost converter, a buck converter, a buck-boost converter, a single-ended primary inductor converter (SEPIC), or any other DC-DC topology that meets the requirements of a particular application. If the equipment requires AC power, the DC-to-DC converter can be replaced by a DC-to-AC converter, or the DC-to-DC converter can be followed by a DC-to-AC converter. If the device contains a rechargeable battery, the final block of the device power and control circuitry can be a battery charge control unit that can manage the charging and maintenance of the battery in the battery powered device.

Device power and control circuitry 2304 may include a processor 2410D, such as a microcontroller, digital signal processor, field programmable gate array processor, microprocessor, or any other type of processor. A processor may be used to read or detect the state or operating point of the power and control circuits and device resonators. The processor can implement algorithms to interpret and adjust the operating points of circuits, elements, assemblies, subsystems, and resonators. The processor may be used to adjust impedance matching, resonators, DC-to-DC converters, DC-to-AC converters, battery charging units, rectifiers, etc. of the wireless powered device.

The processor may have wireless or wired data communication links to other devices or sources, and may transmit or receive data that may be used to adjust the operating point of the system. Any combination of power, voltage, and current signals at a single frequency or over a range of frequencies can be monitored at any point in the device circuit. These signals can be monitored using analog or digital or a combination of analog and digital techniques. These monitoring signals can be used in a feedback loop, or can be reported to the user in a variety of ways, or can be stored and retrieved at a later time. These signals can be used to warn the user of system failures, to indicate performance or to provide audio, visual, vibration, etc. feedback to the user of the system.

40 illustrates components of source power and control circuitry 2302 of an example wireless power transfer system configured to supply power to single or multiple devices. The source power and control circuitry 2302 of the exemplary system may be powered from an AC voltage source 2502 such as a household outlet, a DC voltage source such as a battery, a USB port of a computer, a solar cell, another wireless power source, or the like. Source power and control circuit 2302 may drive source resonator 102S with alternating current, such as with a frequency greater than 10 kHz and less than 100 MHz. The source power and control circuit 2302 may drive the source resonator 102S with alternating current at a frequency less than 10 GHz. Source power and control circuitry 2302 may include DC to DC converter 2408S, AC to DC converter 2408S, or both AC to DC converter 2408S and DC to DC 2408S converter, oscillator 2508, power amplifier 2504, impedance matching network 2402S etc.

The source power and control circuitry 2302 may be powered from a plurality of AC to DC voltage sources 2502 and may contain AC to DC and DC to DC converters 2408S to provide the required voltage levels for the circuit components and for the The DC voltage of the power amplifier that drives the source resonator. The DC voltage can be adjusted and it can be used to control the output power level of the power amplifier. The source may contain power factor correction circuitry.

The oscillator 2508 output may be used as an input to a power amplifier 2504 that drives the source resonator 102S. The oscillator frequency may be tunable, and the amplitude of the oscillator signal may be varied as a means of controlling the output power level from the power amplifier. The frequency, amplitude, phase, waveform and duty cycle of the oscillator signal can be controlled by analog circuitry, by digital circuitry, or by a combination of analog and digital circuitry. The control circuitry may include a processor 2410S, such as a microprocessor, digital signal processor, field programmable gate array processor, or the like.

Source and device tuner impedance matching block 2402 can be used to tune source and control circuits and source and device resonators. For example, the tuning of these circuits can be adjusted for perturbations in the quality factor Q of a source or device resonator due to unrelated objects or changes in the distance between the source and the device in the system. Tuning of these circuits can also be used to sense the operating environment, control power flow to one or more devices, control power to a wireless power network, reduce power when unsafe or failure mode conditions are detected, etc.

Any combination of power, voltage and current signals can be monitored at any point in the source circuit. These signals can be monitored using analog or digital or a combination of analog and digital techniques. These monitoring signals can be used in feedback circuits, or can be reported to the user in various ways, or can be stored and retrieved at a later time. These signals can be used to warn users of system failures, to warn users of exceeded safety thresholds, to indicate performance or to provide audio, visual, vibration, etc. feedback to users of the system.

The source power and control circuitry may include a processor. A processor can be used to read the state or operating point of the power and control circuits and the source resonator. The processor can implement algorithms to interpret and adjust the operating points of circuits, elements, assemblies, subsystems, and resonators. The processor may be used to tune impedance matching, resonators, DC-to-DC converters, AC-to-DC converters, oscillators, power amplifiers for sources, and the like. Frequency and/or temporal power delivery multiplexing schemes may be implemented using the processor and adjustable components of the system. The processor can have wireless or wired data communication links to devices and other sources, and can transmit or receive data that can be used to adjust the operating point of the system.

While a detailed and specific design has been shown in these block diagrams, it should be apparent to those skilled in the art that many different modifications and rearrangements of components and building blocks may be made within the spirit of the exemplary system. The division of circuitry is outlined for purposes of illustration, and it should be clear to those skilled in the art that the components of each block may also be further divided into smaller blocks or combined or shared. In equivalent examples, the power and control circuits could be composed of separate discrete components or larger integrated circuits. For example, the rectifier circuit can consist of discrete diodes, or use diodes integrated on a single chip. Depending on design criteria such as power or size or cost or application, many other circuits and integrated devices may be substituted in the design. The entire power and control circuit or any part of the source or device circuit can be integrated into a single chip.

Impedance matching networks for devices and/or sources may include capacitors or networks of capacitors, inductors or networks of inductors, or any combination of capacitors, inductors, diodes, switches, resistors, and the like. The components of the impedance matching network can be adjustable and variable, and can be controlled to affect the efficiency and operating point of the system. Impedance matching can be performed by controlling the connection points of the resonators, adjusting the permeability of the magnetic material, controlling the bias field, adjusting the frequency of the excitation, and the like. Impedance matching can use or include any number of varactors, varactor arrays, switching elements, capacitor banks, switches and tunable elements, reverse bias diodes, air gap capacitors, compression capacitors, BZT electrically tuned capacitors, MEMS tunable capacitors , voltage variable dielectrics, transformer coupled tuned circuits, etc. or combinations thereof. The variable assembly may be tuned mechanically, thermally, electrically, piezoelectrically, etc. The components for impedance matching may be silicon devices, gallium nitride devices, silicon carbide devices and the like. Components can be selected to withstand high current, high voltage, high power or any combination of current, voltage and power. The elements may be chosen to be high-Q elements.

Source matching and tuning calculations can be performed on an external device through the USB port that powers the device. A device may be a computer, PDA or other computing platform.

The exemplary system uses a source resonator coupled to a device resonator to wirelessly power/recharge multiple electronic consumer devices including, but not limited to, notebook computers, DVD players, projectors, cell phones, Monitors, TVs, Projectors, Digital Photo Frames, Lamps, TV/DVD Players, Portable Music Players, Circuit Breakers, Handheld Tools, Personal Digital Assistants, External Battery Chargers, Mice, Keyboards, Cameras, Active Loads, etc. Multiple devices can be powered simultaneously from a single device resonator. A device resonator can simultaneously operate as a source resonator. Power provided to a device resonator may pass through additional resonators before being diverted to its intended device resonator.

Monitoring, Feedback and Control

So-called port parameter measurement circuits can measure or monitor certain power, voltage and current signals in the system, and the processor or control circuit can adjust certain settings or operating parameters based on those measurements. In addition to these port parameter measurements, the magnitude and phase of voltage and current signals through the system and the magnitude of power signals can be accessed to measure or monitor system performance. The measured signals referred to throughout this disclosure may be port parameter signals as well as any combination of voltage signals, current signals, power signals, and the like. These parameters can be measured using analog or digital signals, sampled and processed, and digitized or converted using a number of known analog and digital processing techniques. The measured or monitored signal may be used in a feedback circuit or system to control the operation of the resonator and/or system. Typically, we can refer to these monitored or measured signals as reference signals or port parameter measurements or signals, although they are sometimes called error signals, monitor signals, feedback signals, etc. We will refer to the signal used to control a circuit element (eg, the voltage used to drive a voltage control capacitor) as a control signal.

In some cases, circuit elements may be adjusted to achieve specified or predetermined impedance values for the source and device resonators. In other cases, the impedance may be adjusted to achieve desired impedance values for the source and device resonator when the device resonator is connected to one or more power sinks. In other cases, the impedance can be adjusted to reduce changes in resonant frequency, changes in impedance or power levels due to movement of source and/or device resonators, or changes in the environment near the resonators (such as movement of interactive materials or objects). ). In other cases, the impedances of the source and device resonators may be tuned to different impedance values.

Coupled resonators can be made of different materials and can include different circuits, components and structural designs, or they can be the same. Coupled resonators may include performance monitoring and measurement circuitry, signal processing and control circuitry, or a combination of measurement and control circuitry. Some or all high-Q magnetic resonators may include tunable impedance circuits. Some or all of the high-Q magnetic resonators may include automatically controlled tunable impedance circuits.

Figure 41 shows a magnetic resonator with a port parameter measurement circuit 3802 configured to measure certain parameters of the resonator. The port parameter measurement circuit can measure the input impedance or reflected power of the structure. Port parameter measurement circuits can be included in source and/or device resonator designs and can be used to measure two port circuit parameters such as S-parameters (scattering parameters), Z-parameters (impedance parameters), Y-parameters (admittance parameters ), T parameters (transmission parameters), H parameters (mixing parameters), ABCD parameters (chain, cascade or transmission parameters), etc. These parameters can be used to describe the electrical performance of a linear power grid when various types of signals are applied.

Different parameters can be used to characterize the power network in different working or coupling schemes. For example, S-parameters can be used to measure matched and unmatched loads. Additionally, the magnitude and phase of voltage and current signals within the magnetic resonator and/or within the source and device itself can be monitored at various points to provide system performance information. This information can be presented to the user of the system via a user interface such as lights, readouts, horns, noise, vibration, etc., or it can be presented as a digital signal, or it can be provided to a processor in the system and in the automatic control of the system use. This information may be recorded, stored, or it may be used by advanced monitoring and control systems.

Figure 42 shows a circuit diagram of a magnetic resonator in which a tunable impedance network can be implemented with a voltage controlled capacitor 3902 or capacitor network. Such implementations may be adjusted, tuned or controlled by circuitry such as programmable voltage source 3908 and/or a computer processor. For example, voltage control capacitors may be adjusted in response to data acquired by port parameter measurement circuitry 3802 and processed by measurement analysis and control algorithm subsystem 3904 . The reference signal may be derived from port parameter measurement circuits or other monitoring circuits designed to measure deviations from expected system operating points. Measured reference signals may include voltages, currents, complex impedances, reflection coefficients, power levels, etc. at single or multiple points in the system and at single or multiple frequencies.

The reference signal can be provided to the measurement analysis and control algorithm subsystem module, which can generate control signals to change the values of various components in the tunable impedance matching network. The control signal may change the resonant frequency and/or input impedance of the magnetic resonator or the power level supplied by the source or drawn by the device to achieve the desired power exchange between the power source/generator and the power consumer/load.

Tuning algorithms may be used to tune the frequency and/or impedance of the magnetic resonator. The algorithm may accept a reference signal related to a degree of deviation from a desired operating point for the system and output correction or control signals related to the deviation to control variable or tunable elements of the system such that the system moves toward one or more The desired operating point is returned. The reference signal for the magnetic resonator can be acquired while the resonator is exchanging power in the wireless power transfer system, or it can be switched out of the circuit during system operation. Corrections to the system may be applied or performed continuously, periodically, upon limit violations, digitally, using analog methods, and the like.

Fig. 43 shows an end-to-end wireless power transfer system. Both the source and the device may include port measurement circuitry 3802 and processor 2410 . The box labeled "Coupler/Switch" 4002 indicates that the port measurement circuit 3802 can be connected to the resonator 102 by a directional coupler or switch, enabling the source and device resonator to be coupled in conjunction with the power transfer function or separately. Measure, adjust and control.

Port parameter measurement and/or processing circuitry may be co-located with some, any or all resonators in the system. Port parameter measurement circuits can use a portion of the power transfer signal or can use an excitation signal over a range of frequencies to measure the source/device resonator response (i.e., transmission and reflection between any two ports in the system) and can include amplitude and/or phase information. Such measurements can be made with a swept single-frequency signal or with a multi-frequency signal. can be generated by one or more processors and standard input/output (I/O) circuits including digital-to-analog converters (DACs), analog-to-digital converters (ADCs), amplifiers, signal generation chips, passive components, etc. to measure and monitor signals in resonators and wireless power transfer systems. Measurements can be achieved using test equipment such as a network analyzer or using custom circuits. The measured reference signal can be digitized by an ADC and processed using a custom algorithm running on a computer, microprocessor, DSP chip, ASIC, etc. The measured reference signal can be processed in an analog control loop.

The measurement circuit can measure any set of two port parameters, such as S-parameters, Y-parameters, Z-parameters, H-parameters, G-parameters, T-parameters, ABCD parameters, etc. Measurement circuits can be used to characterize the current and voltage signals at various points in the driver and resonator circuits, the impedance and/or admittance of the source and device resonators at opposite ends of the system (i.e. towards the device resonating towards the source ("Port 1" in Figure 43) and vice versa).

The device can measure relevant signal and/or port parameters, interpret the measured data, and adjust its matching network to optimize the impedance seen into the coupled system independent of source action. The source can measure relevant port parameters, interpret the measured data, and adjust its matching network to optimize the impedance seen into the coupled system independent of device action.

Figure 43 shows a block diagram of sources and devices in a wireless power transfer system. The system can be configured to execute a control algorithm that actively adjusts the tuning/matching network in either or both of the source and device tuners to optimize performance in the coupled system. Port measurement circuit 3802S may measure signals in the source and communicate those signals to processor 2410 . Processor 2410 may use the measured signals in performance optimization or stabilization algorithms and generate control signals based on the outputs of those algorithms. Control signals may be applied to variable circuit elements in tuning/impedance matching circuit 2402S to adjust operating characteristics of the source, such as power in the resonator and coupling to the device. Control signals can be applied to a power supply or generator to turn power on or off, increase or decrease power levels, modulate supply signals, and the like.

The power exchanged between the source and the device may depend on a variety of factors. These factors may include the effective impedance of the source and device, the Q of the source and device, the resonant frequency of the source and device, the distance between the source and device, the interaction of materials and objects near the source and device, etc. Port measurement circuitry and processing algorithms can operate concurrently to adjust resonator parameters to maximize power transfer, hold power transfer constant, controllably adjust power transfer, etc. under dynamic and steady state operating conditions.

Some, all, or none of the sources and devices in a system implementation may include port measurement circuitry 3802S and processing 2410 capabilities. Figure 44 shows an end-to-end wireless power transfer system in which only the source 102S contains the port measurement circuit 3802 and the processor 2410S. In this case, the device resonator 102D operating characteristics may be fixed, or may be adjusted by analog control circuitry, and no processor-generated control signals are required.

Fig. 45 shows an end-to-end wireless power transfer system. Both the source and the device may include the port measurement circuit 3802, but in the system of FIG. 45, only the source includes the processor 2410S. The source and device may communicate with each other, and the adjustment of certain system parameters may be in response to control signals that have been communicated wirelessly between the source and device, such as through wireless communication circuitry 4202 . Wireless communication channel 4204 may be separate from wireless power transfer channel 4208, or they may be the same. That is, the resonator 102 used for power exchange can also be used to exchange information. In some cases, information can be exchanged by modulating components of the source or device circuit and sensing changes in port parameters or other monitoring devices.

Implementations where the source contains only the processor 2410 may be beneficial for multi-device systems where the source can handle all tuning and adjustment "decisions" and simply transmit control signals back to the device(s). This implementation can make the device smaller and less expensive because it can eliminate the need for, or reduce the functionality required for, a processor in the device. A portion of the entire data set from each port measurement at each device can be sent back to the source microprocessor for analysis, and control instructions can be sent back to the device. These communications may be wireless communications.

Fig. 46 shows an end-to-end wireless power transfer system. In this example, the source only includes port measurement circuit 3802 and processor 2410S. The source and device may communicate with each other, such as via wireless communication circuitry 4202, and adjustments of certain system parameters may be in response to control signals that have been communicated wirelessly between the source and device.

Figure 47 shows a coupled electromagnetic resonator 102 whose frequency and impedance can be automatically adjusted using a processor or computer. The resonant frequency of the source and device resonators can be achieved with reverse biased diodes, Schottky diodes and/or varactor elements contained within the capacitor network shown as C1, C2 and C3 in FIG. 47 Tuning and continuous impedance adjustment. The circuit topologies that have been constructed and illustrated and described herein are exemplary and are not intended to limit the discussion of automated system tuning and control in any way. Other circuit topologies may be utilized with the measurement and control architectures discussed in this disclosure.

Device and source resonator impedance and resonant frequency can be measured with a network analyzer 4402A-B or by other means as described above and realized with a controller such as with Lab View 4404. The measurement circuit or device can output data to a computer or processor that implements a feedback algorithm and dynamically adjusts frequency and impedance via a programmable DC voltage source.

In one arrangement, the reverse-biased diode (Schottky, semiconductor junction, etc.) used to implement the tunable capacitance draws little DC current and can be reverse-biased by an amplifier with a large series output resistance. This implementation may enable direct application of DC control signals to controllable circuit elements in the resonator circuit while maintaining a very high Q in the magnetic resonator.

If the required DC bias voltages are different, the C2 bias signal can be isolated from the C1 and/or C3 bias signals with DC blocking capacitors as shown in Figure 47. The output of the bias amplifier can be bypassed to circuit ground to isolate the RF voltage from the bias amplifier and prevent non-fundamental RF voltages from being injected into the resonator. The reverse bias voltage for some capacitors may instead be applied through an inductive element in the resonator itself, since the inductive element acts as a short circuit at DC.

The port parameter measurement circuit may exchange signals with a processor (including any required ADCs and DACs) as part of a feedback or control system used to automatically adjust the resonant frequency, input impedance, stored by the resonator or Captured energy or power delivered by a source to a device load. The processor may also send control signals to tuning or adjustment circuitry within or attached to the magnetic resonator.

When utilizing varactors or diodes as tunable capacitors, it may be beneficial to have fixed capacitors in the tuning/matching circuit in parallel and in series with the tunable capacitor operating at high reverse bias. This arrangement can provide improvements in circuit and system stability and power handling capability by optimizing the operating voltage across the tunable capacitor.

Varactors or other reverse biased diodes can be used as voltage control capacitors. Varactor arrays can be used when higher voltage uniformity than a single varactor assembly or a different capacitance than that of a single varactor assembly is required. The varactors may be arranged as an N by M array of single two-terminal assemblies connected in series and in parallel and viewed as having different characteristics than the individual varactors in the array. For example, an N by N array of equal varactors (where components in each row are connected in parallel and components in each column are connected in series) may be used as having the same capacitance but with N times the voltage uniformity of a single varactor in the array. Depending on the variability and differences in the parameters of the individual varactors in the array, additional biasing circuits consisting of resistors, inductors, etc. may be required. A schematic diagram of a four by four array of unbiased varactors 4502 that may be suitable for magnetic resonator applications is shown in FIG. 48 .

Further improvements in system performance can be achieved by careful selection of fixed value capacitor(s) placed in parallel and/or in series with the tunable (varactor/diode/capacitor) elements. Multiple fixed capacitors switched in and out of the circuit may be able to compensate for variations in resonator Q, impedance, resonant frequency, power level, coupling strength, etc. that may be encountered in testing, developing and operating wireless power transfer systems . Switched capacitor banks and other switching element banks can be used to ensure convergence to the operating frequency and impedance values required by the system design.

Exemplary control algorithms for isolating and coupling magnetic resonators may be described for the circuit and system elements shown in FIG. 47 . A control algorithm first tunes each source and device resonator loop "in isolation", that is, the other resonators in the system are "shorted" and "removed" from the system. In effect, it is possible to "short circuit" the resonator by causing the resonator to resonate at a much lower frequency, such as by maximizing the values of C1 and/or C3. This step effectively reduces the coupling between resonators, effectively reducing the system to a single resonator at a specific frequency and impedance.

Tuning the magnetic resonator in isolation involves varying the tunable elements in the tuning and matching circuit until the values measured by the port parameter measurement circuit are at their predetermined, calculated or measured relative values. Desired values for the quantities measured by the port parameter measurement circuit may be selected based on desired matching impedance, frequency, strong coupling parameters, and the like. For the exemplary algorithms discussed below, the port parameter measurement circuitry measures S-parameters over a range of frequencies. The frequency range used to characterize the resonator may be a compromise between the obtained system performance information and calculation/measurement speed. For the algorithm described below, the frequency range may be approximately +/- 20% of the operating resonant frequency.

Each isolated resonator can be tuned as follows. First, short the resonator that is not being tuned. Next, minimize C1, C2 and C3 in the resonator being characterized and tuned. In most cases, there will be fixed circuit elements in parallel with C1, C2, and C3, so this step does not reduce the capacitance value to zero. Next, start increasing C2 until the resonator impedance matches the "target" real impedance at any frequency within the above measured frequency range. The initial "target" impedance may be less than the expected operating impedance for the coupled system.

C2 can be adjusted until the initial "target" impedance is achieved for frequencies within the measurement range. Then, C1 and/or C3 can be adjusted until the loop resonates at the desired operating frequency.

Each resonator can be tuned according to the algorithm described above. After tuning each resonator in isolation, a second feedback algorithm may be applied to optimize the resonant frequency and/or input impedance for wireless transfer of power in the coupled system.

C1 and/or C2 in each resonator in a coupled system can be determined by measuring and processing the values of the real and imaginary parts of the input impedance from either and/or both "ports" shown in Figure 43 and/or the required adjustment of C3. For coupled resonators, changing the input impedance of one resonator changes the input impedance of the other resonator. A control and tracking algorithm may adjust one port to a desired operating point based on measurements at the port, and then adjust the other port based on measurements at the other port. These steps can be repeated until both sides converge to the desired operating point.

S-parameters can be measured at both source and device ports, and the following series of measurements and adjustments can be made. In the description that follows, Z ₀ is the input impedance and may be the target impedance. In some cases, Z ₀ is 50 ohms or close to 50 ohms. Z ₁ and Z ₂ are intermediate impedance values that may be the same value as Z ₀ or may be different from Z ₀ . Re{value} means the real part of the value and Im{value} means the imaginary part of the value.

An algorithm that can be used to tune the input impedance and resonant frequency of the two coupled resonators is described below:

1) Tuning each resonator "in isolation" as described above.

2) Adjust the source C1/C3 until it is below, Re{S11}=(Z ₁ +/-ε _Re ) as follows:

- If Re {S11ω _o } > (Z ₁ +ε _Re ), then reduce C1/C3. If Re{S11ω _o }<(Zo-ε _Re ), then increase C1/C3.

3) Adjust source C2 until under ω _o , Im{S11}=(+/-ε _Im ) as follows:

- If Im{S11ω _o }>ε _Im , then decrease C2. If Im{S11ω _o }<-ε _Im , then increase C2.

4) Adjust equipment C1/C3 until under ω _o , Re{S22}=(Z ₂ +/-ε _Re ) is as follows:

- If Re {S22ω _o } > (Z ₂ +ε _Re ), then reduce C1/C3. If Re{S22ω _o }<(Zo-ε _Re ), then increase C1/C3.

5) Adjust device C2 until Im{S22}=0 under ω _o as follows:

- If Im{S22ω _o }>ε _Im , then decrease C2. If Im{S22ω _o }<-ε _Im , then increase C2.

We have converged to ((Z ₀ +/-ε Re ) under ω _o by repeating steps 1-4 until both (Re{S11}, Im{S11}) and (Re{S22}, Im{S22} ₎ , (+/-ε _Im )), where Z ₀ is the desired matching impedance and is the desired operating frequency. Here, ε _Im represents the maximum deviation of the imaginary part under ω _o from the expected value of 0, and ε _Re represents the maximum deviation of the real part from the expected value of Z ₀ . It should be understood that ε _Im and ε _Re can be adjusted to increase or decrease the number of steps to convergence at the potential cost of system performance (efficiency). It should also be understood that steps 1-4 may be performed in various sequences other than those described above and in various ways (i.e., adjust the source imaginary first, then the source real; or adjust the device real first, then the real device imaginary part, etc.). Intermediate impedances _Z1 and _Z2 can be adjusted during steps 1-4 to reduce the number of steps required for convergence. The desired or target impedance value may be complex and may change over time or in different operating schemes.

Steps 1-4 can be performed in any order, in any combination and any number of times. Having described the above algorithm, modifications to the steps or described implementations may be apparent to those skilled in the art. In the same way that a linear circuit can be analyzed using impedance or admittance alternatively to derive the same results, any equivalent linear network port parameter measurement (i.e., Z-parameters, Y-parameters, T-parameters, H-parameters, ABCD parameters, etc. ) or other monitoring signals mentioned above to implement the above algorithm.

Retuning of the resonators may be required due to changes in the "loading" resistances Rs and Rd caused by changes in the mutual inductance M (coupling) between the source and device resonators. Variations in the inductance Ls and Ld of the inductive element itself may be caused by the influence of external objects (as previously discussed) and may also require compensation. Such changes can be mitigated with the adjustment algorithm described above.

Directional couplers or switches may be used to connect the port parameter measurement circuit to the source resonator and tuning/adjustment circuit. The port parameter measurement circuit can measure properties of the magnetic resonator while it is exchanging power in the wireless power transfer system, or it can be switched out of the circuit during system operation. Port parameter measurement circuitry can measure parameters and the processor can control certain tunable elements of the magnetic resonator at startup or at certain intervals or in response to changes in certain system operating parameters.

Wireless power transfer systems may include circuitry to vary or tune the impedance and/or resonant frequency of source and device resonators. Note that while the tuning circuit is shown in both the source and device resonators, it may alternatively be included in only the source or device resonators, or may only be included in some sources and/or device resonators the circuit. Note also that although we refer to the circuit as "tuning" the impedance and/or resonant frequency of the resonator, this tuning operation simply means that various electrical parameters such as the inductance or capacitance of the structure are being changed. In some cases, these parameters may be varied to achieve specific predetermined values, in other cases they may be changed in response to a control algorithm or stabilize a changing target performance value. In some cases, parameters may vary depending on temperature, other sources or devices in the area, environment, and the like.

application

For each of the listed applications, those skilled in the art will recognize that there are numerous ways in which resonator structures used to effectuate wireless power transfer can be connected or integrated with objects being powered or powered. Resonators can be physically separated from source and device objects. Resonators may supply or remove power from the object using conventional inductive technology or by direct electrical connection with eg wires or cables. The electrical connection may be from the resonator output to an AC or DC power input port on the object. The electrical connection may be from the subject's output power port to the resonator input.

Figure 49 shows a source resonator 4904 physically separated from a power supply and a device resonator 4902 physically separated from a device 4900 (a laptop computer in this illustration). Through the electrical connection, power can be supplied to the source resonator and power can be taken directly from the device resonator. Those skilled in the art will understand from the materials incorporated by reference that the shape, size, material composition, arrangement, location and orientation of the resonators described above are provided by way of non-limiting examples and can be disclosed by technology to support wide variation of any and all of these parameters.

Continuing with the laptop computer example, and without limitation, a device resonator may be physically connected to the device it is powering or charging. For example, as shown in Figures 50a and 50b, the device resonator 5002 may (a) be integrated into the housing of the device 5000 or (b) may be attached by an adapter. The resonator 5002 may be (Fig. 50b-d) or may not (Fig. 50a) be visible on the device. The resonator may be attached to a device, integrated into a device, plugged into a device, and the like.

The source resonator may be physically connected to a source that supplies power to the system. As noted above for devices and device resonators, there are a variety of ways in which a resonator can be attached to, connected to, or integrated with a power supply. Those skilled in the art will understand that there are many ways in which resonators can be integrated in a wireless power transfer system, and that the sources and devices can utilize similar or different integration techniques.

Continuing again with the example of a technical laptop computer, and without limitation, the laptop computer may be powered, charged or recharged by the wireless power transfer system. Wireless power can be supplied using a source resonator, and wireless power can be captured using a device resonator. The device resonator 5002 can be integrated into the edge of the screen (display) as shown in Figure 50d, and/or into the base of a laptop computer as shown in Figure 50c. The source resonator 5002 can be integrated into the base of the laptop computer, and the device resonator can be integrated into the edge of the screen. The resonator may also or alternatively be attached to a power supply and/or laptop computer. The source and device resonators may also or alternatively be physically separated from the power supply and laptop and may be electrically connected with cables. The source and device resonators may also or alternatively be physically separated from the power supply and laptop and may be electrically coupled using conventional inductive techniques. Those skilled in the art will understand that while the previous examples relate to wireless power transfer for laptop computers, the methods and systems disclosed for this application may be suitably adapted for use with other electrical or electronic devices. Typically, the source resonator can be external to the source and supply power to the device resonator, which in turn supplies power to the device, or the source resonator can be connected to the source and supply power to the device resonator, which in turn supplies power to the device Power is supplied to a portion of the device, or the source resonator may be internal to the source and supply power to a device resonator which in turn supplies power to a portion of the device, and any combination of these.

The systems or methods disclosed herein can provide power to electrical or electronic devices such as, but not limited to, telephones, cellular telephones, cordless telephones, smartphones, PDAs, audio equipment, music players, MP3 players, wireless , portable radios and players, wireless headsets, wireless headsets, computers, laptops, wireless keyboards, wireless mice, televisions, monitors, flat screen displays, computer monitors, monitors built into furniture, digital photo frames, electronic Books (such as Kindle, e-ink books, magazines, etc.), remote control units (also known as controllers, game controllers, commanders, voters, etc., and used for remote control of multiple electronic devices, such as televisions, video games, monitors, computers, audio-visual equipment, lights, etc.), lighting equipment, cooling equipment, air circulation equipment, purification equipment, personal hearing aids, power tools, security systems, alarms, bells, strobe lights, sirens, sensors, loudspeakers, electronic locks, electronic Keypads, lighting switches, other electrical switches, etc. Here, the term electronic lock is used to refer to an electronically operated (eg, with electronic combination key, magnetic card, RFID card, etc.) door lock located on the door rather than a mechanical keyed lock. Such locks are often battery operated, with the risk that when the battery dies it may stop working, locking the user out. This can be avoided where the battery is recharged or completely replaced with wireless power transfer implementations as described herein.

Here, the term light switch (or other electronic switch) is intended to refer to any switch in one part of the room (for example, on the wall of the room) that turns on/off a device in another part of the room (for example, a lighting fixture in the center of the ceiling). ). To install such a switch with a direct connection, it would be necessary to run the wire all the way from the device to the switch. Once such switches are installed at a particular location, they can be very difficult to move. Alternatively, 'wireless switches' may be envisioned, where "wireless" means that the switch (on/off) command is transmitted wirelessly, but such switches traditionally require batteries for operation. In general, it may be impractical to have too many battery operated switches around the house, since so many batteries will need to be replaced periodically. Therefore, a wireless communication switch may be more convenient, provided that it is powered wirelessly. For example, battery powered communicating wireless doorbells already exist, but in these, the batteries in them must still be replaced periodically. The remote doorbell button could be made completely wireless, wherein the need for constant battery changes may not again be required. Note that the terms 'cordless' or 'wireless' or 'communication wireless' are used here to indicate that there is a cordless or wireless communication means between the device and another electrical component, such as a base station for a cordless phone, a base station for a wireless keyboard computer etc. Those skilled in the art will recognize that any electrical or electronic device may include wireless communication means, and that the systems and methods described herein may be used to add wireless power transfer to the device. As described herein, power to an electrical or electronic device may be delivered from an external or internal source resonator, to the device or a portion of the device. Wireless power transfer can significantly reduce the need to charge and/or replace batteries for devices that enter the vicinity of the source resonator and thereby can reduce downtime, cost, and disposal issues often associated with batteries.

The systems and methods described herein can provide power to lights without the need for corded power or batteries. That is, the systems and methods described herein can provide power to a light without a wired connection to any power source, and over medium-range distances (such as over distances of a quarter meter, one meter, three meters, etc.) ) provides energy to the lamp non-radiatively. As used herein, 'lamp' may refer to the light source itself, such as an incandescent bulb, fluorescent bulb lamp, halogen lamp, gas discharge lamp, fluorescent lamp, neon lamp, high intensity discharge lamp, sodium vapor lamp, mercury vapor lamp, electroluminescent lamp , light-emitting diode (LED) lights, etc.; lights that are part of a light fixture, such as table lamps, floor lamps, pendant lights, track lights, recessed light fixtures, etc.; light fixtures that are integrated with other functions, such as lamp/ceiling fan fixtures and Illuminated photo frames and more. Likewise, the systems and methods described herein can reduce the complexity for installing lights, such as by minimizing the installation of electrical wiring, and allow users to place or install lights with minimal attention to wired power sources. For example, a lamp can be placed anywhere near the source resonator, where the source resonator can be mounted in a number of different locations relative to the position of the lamp, such as on the floor of the room above (e.g., in the case of a chandelier). and especially if the room above is an attic); on the walls of the next room, on the ceiling of the room below (for example, in the case of floor lamps); in components within the room or in the room as described herein infrastructure; etc. For example, a light/ceiling fan combination is often installed in the master bedroom, and the master bedroom often has a loft above it. In this case, the user can more easily install the light fixture/ceiling fan combination in the master bedroom, such as by simply mounting the light/ceiling fan combination to the ceiling and placing the source coil (plugged into the house wired AC power supply) in the In the attic above the installed fixtures. In another example, the light may be an exterior light such as a floodlight or security light and a source resonator mounted inside the structure. This way of installing lighting may be particularly beneficial to renters as they may now be able to install lights and such other electrical components without the need to install new electrical wiring. Control of the lights may also be communicated by near field communication as described herein or by traditional wireless communication methods.

The systems and methods described herein may provide power from a source resonator to a device resonator embedded in or external to a device component, such that the device component may be a conventional electrical component or a fixture. For example, a pendant light could be designed or retrofitted with a device resonator integrated into the fixture, or the pendant light could be a traditional wired fixture and plugged into a separate electrical unit equipped with a device resonator. In an example, the electrical mechanism may be a wireless junction box designed with a device resonator for receiving wireless power, e.g. Many conventional power outlets powered by resonators. Ceiling-mounted wireless distribution boxes can now provide power to traditionally wired electrical components in the ceiling (e.g., pendant lights, track lights, ceiling fans). As a result, it is now possible to mount chandeliers to ceilings without the need to run wires through the building's infrastructure. Such device resonators that can be used to traditional electrical outlet distribution boxes can be used in a variety of applications, including being designed for use inside or outside buildings, made portable, made for vehicles, and the like. Wireless power can be transferred through common building materials such as wood, siding, insulation, glass, brick, stone, concrete, and more. The benefits of reduced installation cost, reconfigurability, and increased application flexibility can provide users with significant benefits over traditional wired installations. A device resonator for a conventional electrical outlet junction box may include a number of electrical components to facilitate power transfer from the device resonator to the conventional electrical outlet, such as converting specific frequencies required to achieve efficient power transfer to line voltage Power electronics for converting high frequency AC into usable voltage and frequency (AC and/or DC), power capture electronics for synchronizing capture devices and power output and ensuring consistent, safe and maximally efficient power transfer control etc.

The systems and methods described herein may be in a humid, harsh, controlled, etc. environment (which is outside and exposed to rain), in a swimming pool/sauna/shower, in marine applications, in a closed assembly, in a Lamps or electrical assemblies operating in explosion proof rooms, on exterior signs, in harsh industrial environments in volatile environments (for example from volatile vapors or airborne organics, such as in grain bins or bakeries) offer advantages. For example, lights installed below the water level in a swimming pool are normally difficult to wire up and need to be watertight despite the need for external wiring. However, swimming pool lights using the principles disclosed herein can be more easily made watertight because external wires may not be required. In another example, such as an explosion proof room containing volatile vapors may not only need to be airtight, but may need to have all electrical contacts (which can generate sparks) sealed. Again, the principles disclosed herein may provide a convenient way of supplying sealed electrical assemblies for such applications.

The systems and methods disclosed herein can provide power to a game controller application, such as to a remote handheld game controller. These game controllers may have traditionally been powered solely by batteries, where the use and power distribution of game controllers has resulted in frequent replacement of batteries, battery packs, rechargeable batteries, etc., which is critical for consistent use of game controllers. This may not be ideal, such as during extended game play. A device resonator can be placed into a game controller, and a source resonator connected to a power supply can be placed nearby. Additionally, a device resonator in a game controller can provide power directly to the game controller electronics without a battery; to a battery, battery pack, rechargeable battery, etc., which in turn provides power to the game controller electronics power; and so on. Game controllers can utilize multiple battery packs, each with a device resonator so that the game controller is constantly recharged while in proximity to the source resonator, whether the game controller is plugged in or not. The source resonator may be located in the main game controller device for the game, where power may be supplied to the main game controller device and the source resonator from an AC 'house' power source; in an AC power source in the form of an expansion device, such as in an integrated into the source resonator in the 'extension cord'; located in the gaming chair, which is at least one of plugged into the wall AC, plugged into the main game controller unit, powered by the battery pack in the gaming chair ;etc. The source resonator can be arranged and implemented in any of the configurations described herein.

The systems and methods disclosed herein may integrate device resonators into battery packs, such as battery packs that are interchangeable with other battery packs. For example, some portable devices may drain power at a high rate such that the user may need to have multiple interchangeable battery packs on hand for use, or the user may be operating the device outside of the range of the source resonator and require additional battery packs to continue operating, such as power tools, portable lights, remote control vehicles, etc. Using the principles disclosed herein provides a way not only for a device resonator enabled battery pack to be recharged while in use and within range, but also for batteries that are not in use and placed within range of a source resonator Group recharging mode. In this way, the battery pack can always be ready to be used when the user depletes the charge of the battery pack being used. For example, a user may be working with a cordless power tool where the demand at the time may be greater than can be achieved by powering directly from the source resonator. In such a case, although the systems and methods described herein may provide charging power to an active battery pack that is within range, the battery pack may still be depleted because the power usage may have exceeded the recharge rate. Also, while using the device, the user may simply move in and out of range, or be out of range entirely. However, the user can place an additional battery pack near the source resonator, which has been recharged when not in use, and is now fully charged for use. In another example, a user may be working with a power tool away from the vicinity of the source resonator, but leave a supplemental battery pack charged in the vicinity of the source resonator, such as in a room with a portable source resonator or an extension cord source resonator in the user's vehicle, in the user's toolbox, etc. This way, the user may not have to worry about spending time and/or remembering to plug their battery pack in for future use. The user may simply replace the used battery pack with a charged one and place the used one near the source resonator for recharging. Device resonators can be built into enclosures with known battery form factors and footprints, and can replace conventional battery chemistries in known devices and applications. For example, a device resonator can be built into an enclosure with mechanical dimensions equivalent to an AA battery, AAA battery, D battery, 9V battery, laptop computer battery, cell phone battery, and the like. In addition to the device resonator, the housing may include a small "button cell" to store power and provide extended operation over time or distance. In addition to or instead of a coin cell battery, other energy storage devices may be integrated with the device resonator or any associated power conversion circuitry. These new energy packs can provide voltage and current levels similar to those provided by conventional batteries, but can be composed of device resonators, energy conversion electronics, small batteries, and more. These new energy packs can last longer than traditional batteries because they can be recharged more easily and can be constantly recharged while they are in the wireless power zone. In addition, such energy packs could be lighter than conventional batteries, safer to use and store, operate over a wider range of temperature and humidity, cause less damage to the environment when discarded, and more. As described herein, when used in wireless power zones as described herein, these energy sets may exceed the lifetime of the product.

The systems and methods described herein may be used to power visual displays, such as is the case with laptop computer screens, but more generally will include the large variety of displays utilized in today's electrical and electronic assemblies, such as in In televisions, computer monitors, desktop computer monitors, laptop computer displays, digital photo frames, e-books, mobile device displays (eg on phones, PDAs, game consoles, navigation devices, DVD players), etc. Displays that may be powered by one or more of the wireless power transfer systems described herein may also include embedded displays, such as those embedded in electronic components (e.g., audio equipment, home appliances, automotive displays, entertainment equipment, cash registers, etc.) , remote control), in furniture, in building infrastructure, in a vehicle, on the surface of an object (eg, on a vehicle, building, clothing, sign, vehicle), etc. Displays can be very small, with tiny resonant devices, such as in smart cards as described herein, or very large, such as in billboards. Displays powered using the principles disclosed herein may also be any of a variety of imaging technologies, such as Liquid Crystal Displays (LCDs), Thin Film Transistor LCDs, Passive LCDs, Cathode Ray Tubes (CRTs), Plasma Displays, Projector Displays ( For example, LCD, DLP, LCOS), Surface Conduction Electron Emission Display (SED), Organic Light Emitting Diode (OLED), etc. Source coil configurations may include attachment to a mains power source, such as a building power supply, vehicle power supply, etc., via a wireless extension cord as described herein; box); intermediate relay source coil and so on. For example, hanging a digital display on a wall can be very attractive, as is the case with a digital photo frame that receives its information signal wirelessly or via a portable storage device, but can be rendered unsightly by requiring an unsightly power cord. However, the use of device coils embedded in the digital picture frame, such as wound within the frame portion, may allow the digital picture frame to be suspended without wires at all. The source resonator can then be placed near the digital photo frame, such as in an adjoining room on the other side of the wall, plugged directly into a conventional electrical outlet via a wireless extension cord such as described herein, via a central source resonator in the room, or the like.

The systems and methods described herein can provide wireless power transfer between different parts of an electronic device. Continuing with the laptop example, and without limitation, the screen of the laptop may require power from the base of the laptop. In this case, electrical power has traditionally been delivered through a direct electrical connection from the base of the laptop to the screen through the hinge connection portion of the laptop between the screen and the base. When utilizing a wired connection, the wired connection may be susceptible to wear and tear, the design functionality of the laptop may be limited by the required direct electrical connection, the design aesthetics of the laptop may be limited by the required direct electrical connection, etc. . However, a wireless connection between the base and the screen is possible. In this case, a device resonator could be placed in the screen portion to power the display, and the base could be powered by a second device resonator, by a traditional wired connection, by a hybrid resonator-battery-direct electrical connection, etc. powered by. This not only improves the reliability of the power connection due to the removal of physical wired connections, but also allows the designer to improve the functionality and/or aesthetics of the hinge portion of the laptop since there are no physical wires associated with the hinge. A laptop computer has been used here again to illustrate how the principles disclosed herein can improve the design of electrical or electronic equipment and should not be considered limiting in any way. For example, many other electrical devices that have separate physical parts can benefit from the systems and methods described herein, such as refrigerators that have electrical functions on the door, including ice makers, sensor systems, lights, etc.; Moving parts of robots; powertrains of cars, components in car doors, and more. Those skilled in the art will recognize that the ability to provide power to a device from an external source resonator via a device resonator, or to a portion of a device from an external or internal source resonator via a device resonator, can span a wide range of electrical and electronic devices applicable.

The systems and methods disclosed herein may provide for sharing of electrical power between devices, such as between charged and uncharged devices. For example, a charged device or appliance may act as a source and send a predetermined amount of energy, a dial-in amount of energy, a requested and approved amount of energy, etc. to nearby devices or appliances. For example, a user may have a cell phone and a digital camera, both capable of transmitting and receiving power through an embedded source and device resonator, and discover that one of the devices, such as the cell phone, is low on power. The user can then transfer power from the digital camera to the cell phone. The source and device resonators in these devices can transmit and receive with the same physical resonator, and with separate source and device resonators, one device can be designed to receive and transmit and the other can be designed to receive only, and the One device is designed to transmit only and another to receive only and so on.

To prevent completely draining the device's battery, there may be a setting that allows the user to specify how much power the receiving device is entitled to. For example, it may be useful to impose limits on the amount of power available to external devices and have the ability to turn off power delivery when battery power drops below a threshold.

The systems and methods described herein can provide wireless power transfer to nearby electrical or electronic components associated with an electrical device, where the source resonator is in the electrical device and the device resonator is in the electronic component. The source resonator may also be connected, plugged in, attached to an electrical device, such as through a common interface of the electrical device (eg, USB interface, PC card interface), additional power socket, universal attachment point, etc. For example, the source resonator may be inside the structure of a computer on a desk, or integrated into some object, mat or the like, which is connected to the computer, such as into one of the computer's USB ports. In the example of a source resonator embedded in an object, mat, etc. and powered via a USB interface, the source resonator can be easily added to a user's desktop computer without the need for integration into any other electronic device, thus, A wireless energy zone is conveniently provided around which a plurality of electrical and/or electronic devices can be powered. The electrical devices may be computers, lighting fixtures, dedicated source resonator electrical devices, etc., and the nearby components may be computer peripherals, peripheral electronic components, infrastructure equipment, etc., such as computer keyboards, computer mice, fax machines, Printers, speaker systems, cellular phones, audio equipment, walkie-talkies, music players, PDAs, lights, electric pencil sharpeners, fans, digital photo frames, calculators, video games, etc. For example, a computer system may be an electrical device with an integrated source resonator utilizing a 'wireless keyboard' and a 'wireless mouse', where the use of the term wireless is intended here to indicate that there is a wireless communication means between each device and the computer, and where , each device must still contain a separate battery power source. As a result, the batteries will need to be replaced periodically and, in large companies, can place a considerable burden on support personnel for battery replacement, battery cost and proper disposal of the batteries. Alternatively, the systems and methods described herein can provide wireless power transfer from the body of the computer to each of these peripheral devices, including power to not only the keyboard and mouse, but also to devices such as fax machines, Power to other peripheral components such as printers, speaker systems, etc. A source resonator integrated into an electrical device can provide wireless power transfer to multiple peripheral devices, user equipment, etc., making it possible to charge and/or replace batteries for devices in the vicinity of the source resonator integrated electrical device need to be significantly reduced. The electrical device may also provide tuning or auto-tuning software, algorithms, means, etc. for adjusting power transfer parameters between the electrical device and the wireless powered device. For example, the electrical device may be a computer on a user's desktop, and the source resonator may be integrated into or plugged into the computer (e.g., via a USB connection), wherein the computer provides means for providing a tuning algorithm (e.g., through a software program running on a computer).

The systems and methods disclosed herein can provide wireless power transfer to nearby electrical or electronic components associated with institutional infrastructure components where source resonators are mounted in or on institutional infrastructure components and device resonators are in electronic in the component. For example, an institutional infrastructure component could be a piece of furniture, a fixed wall, a movable wall or divider, a ceiling, a floor, and a source resonator attached to or integrated into a table or desk (e.g., just below/above the surface, on the sides, integrated into table tops or table legs), mats placed on the floor (e.g., under a desk, placed on top of a desk), mats on the garage floor (e.g. equipment charging), in parking lots/garages (e.g., on poles near parking spaces), televisions (e.g., to charge remote controls), computer monitors (e.g., to power wireless keyboards, wireless mice, cell phones/ charging), chairs (e.g. for powering electric blankets, medical equipment, personal health monitors), pictures, office furniture, common household appliances, etc. For example, an facility infrastructure component could be a lighting fixture in an office cubicle, where both the source resonator and the lamp within the fixture could be connected directly to the facility's wired power supply. However, with source resonators now provided in lighting fixtures, those nearby electrical or electronic components connected to or integrated with the device resonator will not need to have any additional wired connections. Additionally, as described herein, the need to replace batteries for devices having device resonators may be reduced.

Using the systems and methods described herein to supply power to electrical and electronic equipment from a central location (such as from a source resonator in an electrical installation, from institutional infrastructure components, etc.) can minimize electrical wiring infrastructure in surrounding work areas. For example, in a corporate office space there is often a large number of electrical and electronic devices that need to be powered by wired connections. With the systems and methods described herein, much of this wiring can be eliminated, saving the business installation costs, reducing the physical constraints associated with having electrical wiring in office walls, eliminating the need for electrical outlets and power strips minimize etc. The systems and methods described herein can save businesses money by reducing electrical infrastructure associated with installation, reinstallation (eg, reconfiguring office space), maintenance, and the like. In another example, the principles disclosed herein may allow for wireless placement of electrical outlets among rooms. Here, the source may be placed on the ceiling of the basement below the location where the floor on which the electrical outlet is expected to be placed. A device resonator can be placed on the floor of the room directly above it. It is now substantially easier to install new lighting fixtures (or any other electrical equipment for that matter, such as cameras, sensors, etc.) in the center of the ceiling for the same reason.

In another example, the systems and methods described herein can provide power "through" a wall. For example, suppose you have a power outlet in one room (eg, on the wall), but want to have a power outlet in the next room, without calling an electrician or drilling or running wires around the wall, etc. The source resonator can then be placed on a wall in one room, and the device resonator outlet/pickup placed on the other side of the wall. This can power things like a flat screen TV or stereo system (for example, one might not want to have ugly wires in the living room that climb up the wall, but don't mind having similar wires that run along the wall in the next room, e.g. storage room or closet, or a room with furniture that makes it impossible to see the wires running along the walls). The systems and methods described herein can be used to transfer power from indoor sources to various electrical devices outside a home or building without the need to drill holes through exterior walls or install conduits therein. In this case, the device can be powered wirelessly outside the building without aesthetic or structural damage or the risks associated with drilling through walls and paneling. Additionally, the systems and methods described herein may provide placement sensors to aid in the placement of internal source resonators for electrical components equipped with external device resonators. For example, a home owner may place a security light on the outside of their home, which includes a wireless device resonator, and now needs to properly or optimally place the source resonator inside the home. A placement sensor acting between the source and device resonator can make the placement better by indicating when or how good the placement is, such as with a visual indication, an audio indication, a display indication, and the like. In another example, and in a similar manner, the systems and methods described herein may provide for the installation of devices, such as radio transmitters and receivers, solar panels, etc., on the roof of a home or building. In the case of solar cells, a source resonator can be associated with the panel and power can be transferred wirelessly to a distribution panel inside the building without the need to drill holes through the roof. The systems and methods described herein may allow electrical or electrical equipment to be installed across the walls of vehicles, such as through the ceiling, without the need for drilling, such as for automobiles, boats, airplanes, trains, and the like. In this way, the walls of the vehicle can be kept intact without drilling holes, thus preserving the value of the vehicle, maintaining water tightness, eliminating the need for wiring, etc. For example, mounting a siren or light to the roof of a police car reduces future resale of the car, but with the systems and methods described herein, any light, horn, siren, etc. can be attached to the ceiling.

The systems and methods described herein can be used to transfer power wirelessly from solar photovoltaic (PV) panels. PV panels with wireless power transfer capability can have several benefits including simpler installation, more flexibility, reliable and weatherproof designs. Wireless power transfer can be used to transfer power from PV panels to equipment, houses, vehicles, etc. Solar PV panels may have wireless source resonators that allow the PV panels to directly power devices that are enabled to receive wireless power. For example, PV panels can be mounted directly to the roof of vehicles, buildings, and the like. The energy captured by the PV panels can be transferred wirelessly directly to devices inside the vehicle or under the roof of a building. A device with a resonator can receive power wirelessly from a PV panel. Wireless power from the PV panels can be used to transfer energy to a resonator coupled to a wired electrical system of a house, vehicle, etc., without requiring any direct contact between the external PV panels and the internal electrical system Allows traditional power distribution and power supply for conventional equipment.

With wireless power transfer, a significantly simpler installation of rooftop PV panels is possible, as power can be transferred wirelessly from the panels to the capture resonator in the house, eliminating all outdoor wiring, connectors and conduits and the roof through the structure or any hole in the wall. Wireless power transfer used with solar cells can have the benefit of reducing roof hazards because it eliminates the need for electricians to work on the roof to interconnect panels, wires and junction boxes. Installing solar panels integrated with wireless power transfer can require less skilled labor because fewer electrical contacts need to be made. With wireless power transfer, less site-specific design may be required, as the technology provides installers with the ability to optimize and position each solar PV panel individually, significantly reducing the need for costly engineering and battery board layout services as needed. Careful balancing of the solar load on each panel may be required and no dedicated DC wiring layout and interconnection is required.

For roof or wall mounting of PV panels, the capture resonator can be mounted on the underside of the roof, inside a wall, or any other easily accessible interior space within either the bottom or both of the solar PV panels . A diagram illustrating a possible typical rooftop PV panel installation is shown in FIG. 51 . Various PV solar collectors can be mounted on top of a roof with wireless power capture coils installed in the building below the roof. A resonator coil in a PV panel can transfer its energy wirelessly through the roof to a wireless capture coil. Captured energy from the PV cells can be harvested and coupled to the electrical system of the house to power electrical and electronic equipment or to the power grid when more power than required is produced. Energy is captured from the PV cells without requiring holes or wires penetrating the roof or walls of the building. Each PV panel may have a resonator coupled to a corresponding resonator on the vehicle or building interior. Multiple panels may utilize wireless power transfer between each other to transfer or harvest power to one or more designated panels coupled to a resonator on a vehicle or inside a house. A panel may have wireless power resonators on its side or in its perimeter that can couple to resonators located in other similar panels, allowing power to be transferred from panel to panel. An additional bus or connection structure may be provided that wirelessly couples and transfers power from multiple panels on the exterior of the building or vehicle to one or more resonators on the interior of the building or vehicle.

For example, as shown in FIG. 51 , a source resonator 5102 may be coupled to a PV cell 5100 mounted on a roof 5104 of a building. Corresponding trapping resonators 5106 are placed inside the building. Solar energy captured by the PV cells can then be transferred from the source resonator 5102 outside to the device resonator 5106 inside the building without direct holes and connections through the building.

Each solar PV panel with wireless power transfer can have its own inverter, significantly improving the economics of these solar systems by individually optimizing the power generation efficiency of each panel, supporting A mix of panel sizes and types, including single-panel "pay-as-you-go" system extensions. The reduction in installation costs can make a single panel economical to install. Eliminates the need for panel string designs and careful positioning and orientation of multiple panels and eliminates a single point of failure for the system.

Wireless power transfer in PV solar panels can enable more solar deployment scenarios as weatherproof solar PV panels eliminate the need to drill holes for wiring through sealed surfaces such as car roofs and boat decks and eliminate Requirements for installing panels in fixed locations. With wireless power transfer, PV panels can be deployed temporarily and then moved or removed without leaving permanent changes to the surrounding structure. It can be placed in the yard on a sunny day and moved back and forth with the sun, or taken inside for example for cleaning and storage. For backyard or mobile solar PV applications, an extension cord with a wireless energy capture device can be thrown in the ground or placed near the solar unit. The snap extension cord can be fully sealed and electrically isolated from the elements, allowing it to be used in any indoor or outdoor environment.

With wireless power transfer, there may be no need for wires or external connections, or the PV solar panel can be completely weatherproof. Significant improvements in the reliability and lifetime of electrical components in solar PV power generation and transmission circuits can be expected, as the weatherproof enclosure protects the components from UV radiation, humidity, weather, etc. With wireless power transfer and a weatherproof enclosure, less expensive components can be used as they will no longer be directly exposed to external elements and weather elements, and it can reduce the cost of PV panels.

Power transfer between PV panels and capture resonators inside buildings or vehicles can be bi-directional. Energy can be delivered from the premises electrical grid to the PV panels to provide power when the panels do not have sufficient energy to perform some of these tasks. The opposite power flow could be used to melt snow from the panels, or power an electric motor that would position the panels in a more favorable position relative to the sun's energy. Once the snow is melted or the panels are relocated and the PV panels are able to generate their own energy, the direction of power transfer can return to normal, delivering power from the PV panels to the building, vehicle or equipment.

PV panels with wireless power transfer may include automatic tuning at installation to ensure maximum and efficient power transfer to the wireless collector. Variations in the roofing material in different installations or in the distance between the PV panels and the wireless power harvester may affect the performance or perturb the properties of the resonator for wireless power transfer. To reduce installation complexity, the wireless power transfer assembly may include tuning capabilities that automatically adjust its operating point to compensate for any effects due to material or distance. Frequency, impedance, capacitance, inductance, duty cycle, voltage levels, etc. can be adjusted to ensure efficient and safe power transfer.

The systems and methods described herein may be used to provide a wireless power zone temporarily or in an extension of a conventional electrical outlet to a wireless power zone, such as through the use of a wireless power extension cord. For example, a wireless power extension cord can be configured with a plug for connection to a traditional electrical outlet, a long wire such as in a traditional power extension cord, and a resonator source coil at the other end (e.g., a replacement or otherwise). It is also possible to configure a wireless extension cord in which there are source resonators at multiple positions along the cordless extension cord. This configuration can then replace any traditional extension cords where there is a wireless power configuration device, such as providing wireless power to locations where no convenient electrical outlet exists, such as a location in a living room where no electrical outlet exists, where no wired Temporary wireless power to power infrastructure (e.g. construction sites), into yards where no power outlets exist (e.g. for parties or yard decorations that are powered wirelessly to reduce the chance of cutting traditional wires), etc. It is also possible to use a wireless extension cord as a drop in a wall or structure to provide a wireless power zone within the vicinity of the drop. For example, a wireless extension cord can be run within the walls of a new or renovated room to provide a wireless power zone without the need to install traditional electrical wiring and electrical outlets.

The systems and methods described herein can be used to control moving parts or rotating assemblies of vehicles, robots, mechanical equipment, wind turbines, or any other type of rotating device or structure that has moving parts (such as robotic arms, construction vehicles, active platform, etc.) to provide power between. Traditionally, power in such systems may be provided by eg slip rings or by rotary joints. Using wireless power transfer as described herein, the design simplicity, reliability, and longevity of these devices can be significantly improved by being able to switch between certain transfer power over a range of distances. In particular, the preferred coaxial and parallel alignment within the source and device coils can provide wireless power transfer that is not severely modulated by the relative rotational motion of the two coils.

The systems and methods described herein can be utilized to extend power requirements beyond the range of a single source resonator by providing a series of source-device-source-device resonators. For example, suppose an existing detached garage does not have power and the owner now wants to install a new power service. However, the owner may not want to run wires throughout the garage, or have to drive into walls to wire electrical outlets throughout the structure. In this case, the owner may choose to connect the source resonator to a new power service, enabling wireless power to be supplied to the facility resonator power outlets throughout the back of the garage. The owner can then install a device-source 'repeater' to supply wireless power to the device resonator power outlet in front of the garage. That is, the power repeater can now receive wireless power from the main source resonator, and then supply the available power to the second source resonator to supply power to the second set of device resonators in front of the garage. This configuration can be repeated iteratively to extend the effective range of supplied wireless power.

Multiple resonators can be used to expand the power requirements around the energy blocking material. For example, it may be desirable to integrate a source resonator into a computer or computer monitor so that the resonator can power devices placed around and especially in front of the monitor or computer, such as keyboards, computer mice, telephones, and the like. Due to aesthetics, space constraints, etc., the energy source available for the source resonator may only be located or connected to the back of a monitor or computer. In many designs of a computer or monitor, metal and metal components containing the circuit are used in the design and packaging, which can limit and prevent power transfer from a source resonator at the back of the monitor or computer to the front of the monitor or computer. An additional repeater resonator can be integrated into the base or base of the monitor or computer, which couples to the source resonator behind the monitor or computer and allows power transfer to the space in front of the monitor or computer. An intermediate resonator integrated into the base or pedestal of the monitor or computer requires no additional power supply, captures power from the source resonator and diverts it to the front around the blocking or power shielding metal components of the monitor or computer.

The systems and methods described herein can be built into, placed on, suspended from, embedded in, integrated into, etc., a structural portion of a space that Such as vehicles, offices, homes, rooms, buildings, outdoor structures, road infrastructure, etc. For example, one or more sources may be built into, placed on, suspended from, embedded in or integrated into walls, ceiling or ceiling panels, floors, partitions, doorways, stairs, compartments, road surfaces, sidewalks, ramps, Fences, external structures, etc. One or more sources can be built into entities in or around a structure, such as a bed, desk, chair, rug, mirror, clock, monitor, television, electronic device, counter, table, a piece of furniture, a piece of art, Shells, compartments, ceiling panels, floor or door panels, fenders, trunks, wheel wells, struts, beams, supports or any similar entity. For example, the source resonator may be integrated into the fender of a user's car so that any device equipped with or connected to the device resonator may be powered from the fender source resonator. In this way, devices that are brought into or integrated into the car can be constantly charged or powered while in the car.

The systems and methods described herein may provide power through the walls of vehicles such as boats, cars, trucks, buses, trains, airplanes, satellites, and the like. For example, a user may not want to drill holes through the wall of the vehicle in order to provide power to electrical equipment outside the vehicle. The source resonator may be placed inside the vehicle, and the device resonator may be placed outside the vehicle (eg, on the opposite side of a window, wall, or structure). In this way, the user can achieve greater flexibility in optimizing the placement, positioning and attachment of external equipment to the vehicle (such as regardless of electrical connections supplied or routed to the equipment). In addition, with wirelessly supplied electric power, the external device can be sealed so that it is watertight, making it safe in case the electrical device is exposed to weather such as rain, or even submerged in water. Similar techniques can be employed in a variety of applications, such as when charging or powering hybrid vehicles, navigation and communication equipment, construction equipment, remote control or robotic equipment, etc., where there is an electrical risk due to exposed conductors. The systems and methods described herein may provide power through the walls of vacuum chambers or other enclosed spaces, such as those used in semiconductor growth and processing, material coating systems, fish ponds, hazardous material handling systems, and the like. Power can be supplied to transforming stages, robotic arms, rotating stages, manipulation and collection equipment, cleaning equipment, etc.

The systems and methods described herein can provide wireless power to a kitchen environment, such as to countertop appliances, including mixers, coffee makers, toasters, toaster ovens, grills, griddles, skillets, electric kettles, Electric frying pans, waffle makers, blenders, food processors, crock pots, electric heating plates, induction hobs, lights, computers, monitors, etc. The present technology can improve the mobility and/or positioning flexibility of the equipment, reduce the number of power cords stored and spread across the countertop, improve the washability of the equipment, and the like. For example, an electric skillet may traditionally have separate parts, such as one that is submersible for cleaning and one that is not submersible because it includes external electrical connections (eg, cords or sockets for removable cords). However, with the device resonator integrated into the unit, all electrical connections can be sealed, so the whole device can now be submerged for cleaning. Additionally, the absence of external cords can eliminate the need for an available electrical wall outlet and eliminate the need to run power cords across the countertop or limit the location of the electric griddle to the location of an available electrical wall outlet.

The systems and methods described herein can provide continuous power/charging to a device equipped with a device resonator since the device does not leave the vicinity of the original resonator, such as stationary electrical equipment, personal computers, intercom systems, security systems, household Robots, lamps, remote control units, televisions, cordless phones, etc. For example, a home robot (eg, ROOMBA) can be powered/charged via wireless power, thus functioning arbitrarily without recharging. As such, power supply designs for home robots can be modified to take advantage of this continuous wireless power source, such as designing the robot to use only power from the source resonator without the need for a battery, using power from the source resonator to power the Recharging the battery of the robot, trickle charging the battery of the robot using power from the source resonator, charging the capacitive energy storage unit using power from the source resonator, etc. Similar optimizations of power supplies and power supply circuits can be enabled, designed, and implemented for any and all devices disclosed herein.

The systems and methods described herein can provide wireless power to electric blankets, heating pads/sheets, and the like. These electric heating devices can be used for a variety of indoor and outdoor applications. For example, a pair of resonators that can be supplied to the hands and The foot warmers are powered remotely.

The systems and methods described herein can be used to power a portable information device that includes a device resonator and that can be powered up when the information device is in the vicinity of an information source that includes a source resonator. For example, the information device may be a card (eg, credit card, smart card, electronic card, etc.) carried in a user's pocket, wallet, wallet, vehicle, bicycle, or the like. The portable information device may be powered on while it is in the vicinity of an information source, which then transmits information to the portable information device, which may include electronic logic, electronic processors, memory, display, LCD display, LEDs, RFID tags, etc. For example, a portable information device may be a credit card with a display that "turns on" when it is near a source of information, and provides the user with some information such as "You just received a coupon for 50% off your next Coca-Cola purchase." information. The information device may store information such as coupon or discount information that can be used on subsequent purchases. The portable information device can be programmed by the user to contain tasks, calendar appointments, reminders, alarms and reminders, and the like. The information device can receive current price information and inform the user of the location and price of previously selected and identified items.

The systems and methods described herein can provide wireless power transfer to directly power or charge batteries in sensors, such as environmental sensors, security sensors, agricultural sensors, appliance sensors, food spoilage sensors, power sensors, etc. Can be installed inside a structure, outside a structure, buried in the ground, installed in a wall, etc. For example, this capability could replace the need to dig up old sensors to physically replace batteries or bury new sensors because old sensors are incapable and no longer operational. These sensors can be recharged periodically by using a portable sensor source resonator charging unit. For example, a truck carrying an active resonator-equipped power supply (eg, providing ~kW of power) can provide enough power to a ~mW sensor within minutes to extend the sensor's operational duration for more than a year. It is also possible to power the sensor directly, such as in a location where it is difficult to wire it but it is still within the vicinity of the source resonator, such as a device outside the house (security camera), on the other side of a wall, Wait on the electric lock on the door. In another example, sensors that might otherwise need to be provided with a wired power connection can be powered by the systems and methods described herein. For example, ground fault interrupter circuit breakers combine residual current and overcurrent protection in one device for installation into service panels. However, sensors have traditionally had to be wired separately for power and this complicates installation. However, with the systems and methods described herein, it is possible to power a sensor with a device resonator, wherein a single source resonator is provided within the service panel, thereby simplifying the installation and wiring configuration within the service panel. Additionally, a single source resonator may power device resonators mounted on either side of a source resonator mounted within the service panel, across the service panel, mounted to an additional nearby service panel, or the like. The systems and methods described herein can be used to provide wireless power to any electrical component associated with a switchboard, electrical room, power distribution, etc., such as in a switchboard, switchboard, circuit breaker, transformer, backup battery, fire alarm control panel etc. By using the systems and methods described herein, power distribution and protection components and system installations can be more easily installed, maintained, and modified.

In another example, a battery-powered sensor can operate continuously without requiring battery replacement because wireless power can be supplied to periodically or continuously recharge or trickle charge the battery. In such applications, even low levels of power can sufficiently recharge or maintain a charge in a battery, significantly extending its life and usefulness. In some cases, battery life can be extended beyond the life of the device it powers, making it an essentially "endless" battery.

The systems and methods described herein can be used to charge implanted medical device batteries, such as in artificial hearts, pacemakers, heart pumps, insulin pumps, implanted coils for nerve or acupressure hemostasis/acupoint stimulation, etc. . For example, it may be inconvenient or unsafe to have a lead stick outside of the patient's body, since the lead can be a constant source of possible infection and is often very unpleasant for the patient. The systems and methods described herein may also be used to charge or power medical devices in or on a patient's body from an external source, such as from a hospital bed or a hospital wall or ceiling with a source resonator. Such medical devices may be easier to attach, read, use and monitor patients. The systems and methods described herein can alleviate the need to attach leads to the patient and to the patient's bed or bedside, making it easier for the patient to move around and get up out of bed without the risk of inadvertently disconnecting the medical device. This can be useful, for example, for a patient with multiple sensors to monitor it, such as for measuring pulse, blood pressure, glucose, etc. For medical and monitoring devices utilizing batteries, the batteries may need to be replaced quite frequently, perhaps multiple times a week. This can cause risks associated with people forgetting to change batteries, not noticing that a device or monitor is not working due to a dead battery, infection associated with improper cleaning of battery covers and compartments, and the like.

The systems and methods described herein can reduce the risk and complexity of medical device implantation procedures. Today, many implantable medical devices such as ventricular assist devices, pacemakers, defibrillators, etc. require surgical implantation due to their device form factors, which may be largely limited by the long-life batteries integrated into the devices. Effects of volume and shape. In one aspect, described herein is a non-invasive method of recharging a battery such that the battery size can be greatly reduced and the entire device can be implanted, such as via a catheter. Catheter implantable devices may include integrated capture or device coils. The catheter implantable capture or device coil can be designed such that it can be routed internally, such as after implantation. The capture or device coil can be deployed via the catheter as a coiled flexible coil (eg, coiled, eg, two spools, easily unrolled internally with a simple spreader mechanism). The power coil can be worn in a vest or clothing cut to fit the source in place; can be placed in a chair cushion or mattress; can be integrated into a bed or furniture, etc.

The systems and methods described herein may enable a patient to have a 'sensor vest', sensor patch, etc., which may include at least one of a device resonator and a plurality of medical sensors that may be powered or charged when in the vicinity of a source resonator. Traditionally, such medical surveillance agencies may have required batteries, thus making vests, patches, etc. heavy and possibly impractical. However, using the principles disclosed herein, a battery (or a lighter rechargeable battery) may not be required, thus making such devices more convenient and practical, especially in situations where straps may not be available (such as when there is no battery or a significantly Adhesives in the case of lighter batteries) to hold such medical devices in place. Medical institutions may be able to remotely read sensor data with the goal of anticipating strokes, heart attacks, etc. (eg, minutes in advance). When the vest is used by a person at a location away from a medical facility, such as in their home, the vest can then be integrated with a cell phone or communication device to call an ambulance in the event of an accident or medical event. The systems and methods described herein may be particularly useful in situations where the vest will be used by the elderly, where traditional non-wireless recharging practices may not be followed as required (eg, changing batteries, plugging in at night, etc.). The systems and methods described herein may also be used to charge devices that are used by or assist a disabled or handicapped person (who may have difficulty replacing or recharging batteries), or to reliably charge devices that are enjoyed or relied upon by The device supplies power.

The systems and methods described herein can be used for charging and powering prosthetics. Prosthetics have become very capable at replacing the function of original limbs such as arms, legs, hands and feet. However, powered prostheses may require considerable power (such as 10-20W), which may translate into relatively large batteries. In this case, the amputee can choose between a light battery that doesn't last very long, and a heavy battery that lasts much longer but is more difficult to 'carry' back and forth. The systems and methods described herein may enable prosthetics to be powered with a device resonator, where the source resonator is carried by the user and attached to a part of the body that can more easily support weight (e.g., such as a belt around the waist) or Located in an external location where the user will spend a significant amount of time keeping the device charged or powered, such as at his desk, in his car, in his bed, etc.

The systems and methods described herein can be used for charging and powering electric exoskeletons, such as those used in industrial and military applications, and for the elderly/infirm/ill. Motorized exoskeletons can provide up to a 10 to 20-fold increase in "strength" to a person, enabling that person to physically perform strenuous tasks repeatedly without much fatigue. However, the exoskeleton may require more than 100W of power in some use cases, so battery powered operation may be limited to 30 minutes or less. The delivery of wireless power described herein may provide the user of the exoskeleton with a continuous supply of power for powering the structure's motion of the exoskeleton and for powering various monitors and sensors distributed throughout the structure. For example, an exoskeleton with embedded device resonator(s) may be powered from a local source resonator. For industrial exoskeletons, the source resonator can be placed in the walls of the facility. For military exoskeletons, the source resonator could be carried by an armored vehicle. For exoskeletons used to assist elderly caregivers, the source resonator(s) may be installed or placed in room(s) of one's home.

The systems and methods described herein may be used to power/charge portable medical devices such as oxygen systems, ventilators, defibrillators, medication pumps, monitors, and devices in ambulances or mobile medical units, among others. Being able to transport a patient from the scene of an accident to a hospital, or in order to move a patient from his bed to another room or area, with all the equipment attached to it and powered at all times, provides great benefit to the patient's health and eventual recovery . Of course, the risks and problems posed by a medical device that stops working because its battery dies or because it has to be unplugged while transporting or moving a patient in any way is understandable. For example, an emergency medical team at the scene of an automobile accident may need to utilize a portable medical device in the emergency care of a patient at the scene. Such portable medical devices must be properly maintained such that there is sufficient battery life to power the device for the duration of the emergency. However, it is often the case that equipment is not properly maintained such that batteries are not fully charged and in some cases required equipment is not available to first responders. The systems and methods described herein can provide wireless power to portable medical devices (and associated sensor inputs on the patient's body) in a manner that provides charging and maintenance of batteries and power packs automatically and without human intervention . Such systems also benefit from improved patient mobility unencumbered by the various power cords attached to the many medical monitors and devices used in the patient's treatment.

The systems and methods described herein can be used to power/charge personal hearing aids. Personal hearing aids need to be small and lightweight to fit in or around a person's ear. Size and weight constraints limit the size of batteries that can be used. Likewise, the size and weight constraints of the device make battery replacement difficult due to the delicate nature of the components. The size and hygiene issues of the device make it difficult to integrate additional charging ports to allow recharging of the battery. The systems and methods described herein can be integrated into hearing aids and can reduce the size of batteries required, which can allow for even smaller hearing aids. Using the principles disclosed herein, the hearing aid's battery can be recharged without requiring an external connection and charging port. Charging and device circuitry and a small rechargeable battery can be integrated into the form factor of a conventional hearing aid battery, allowing retrofitting of existing hearing aids. The hearing aid can be recharged while it is being used and worn by a person. An energy source could be integrated into the pad or cup, allowing recharging while the hearing aid is placed in such a structure. A charging source can be integrated into the hearing aid dryer case, allowing wireless recharging while the hearing aid is being dried or sanitized. The source and device resonators can also be used to heat the device, reducing or eliminating the need for additional heating elements. A portable charging case powered by batteries or an AC adapter can be used as a storage and charging station.

A source resonator for the medical system described above could be in the body of some or all of the medical equipment, with the device resonator on the patient's sensor and device; the source resonator could be in the ambulance, and the device resonator on the patient's sensor and some on the body of some or all of the devices; the primary source resonator could be in the ambulance to divert wireless power to the device resonator on the medical device while the medical device is in the ambulance, and a second source resonator should be used when the device is away from the ambulance. The resonators may be in the main body of the medical device and a second device resonator on the patient sensor, among others. The systems and methods described herein can significantly improve the ease with which medical personnel can transport a patient from one location to another, wherein electrical wiring and the need to replace or manually recharge associated batteries can now be reduced.

The systems and methods described herein may be used for charging equipment inside military vehicles or facilities, such as tanks, armored vehicles, mobile shelters, and the like. For example, when a soldier returns to a vehicle after an "action" or mission, he can often begin charging his electronic devices. If its electronic device is equipped with a device resonator, and there is a source resonator inside the vehicle (for example, integrated in the seat or on the roof of the vehicle), its device will start charging immediately. In fact, the same vehicle can provide power to a soldier/robot (such as the packbot from iRobot) standing outside or walking next to the vehicle. This capability may be useful in: minimizing accidental battery swaps with other people (which can be an important issue since soldiers tend to trust only their own batteries); enabling quick exit from a vehicle under attack; Powering or charging laptops or other electronic devices inside the tank, as excess wires inside the tank can pose a hazard in reducing the ability to move around quickly in "trouble" and/or reduced visibility conditions. The systems and methods described herein may provide significant improvements related to powering portable electrical devices in a military environment.

The systems and methods described herein can be applied to vehicles such as golf carts or other types of carts, ATVs, electric bicycles, scooters, automobiles, lawn mowers, bobcats, and other vehicles commonly used in construction and landscaping, etc. mobile vehicles that provide wireless power and charging capabilities. The systems and methods described herein can provide wireless power or charging capabilities to micro-mobile vehicles, such as micro-helicopters, drones, remote-controlled aircraft, remote-controlled boats, remote-controlled or robotic aerial vehicles, remote-controlled or robotic lawnmowers or devices, bombs Detect robots, etc. For example, a micro-helicopter that flies above military vehicles to increase its field of view can fly for several minutes on standard batteries. If these micro-helicopters were equipped with device resonators, and the control vehicle had source resonators, the micro-helicopters might be able to fly indefinitely. The systems and methods described herein may provide an efficient alternative to recharging or replacing batteries for use in micro mobility vehicles. Additionally, the systems and methods described herein can provide power/charging to even smaller devices, such as microelectromechanical systems (MEMS), nanorobots, nanodevices, and the like. In addition, the present invention can be realized by installing the source device in a mobile vehicle or in-flight equipment so that it can act as a field or in-flight recharger (which can self-locate itself in the vicinity of a mobile vehicle equipped with a device resonator) The systems and methods described.

The systems and methods described herein can be used to provide electrical power grids for temporary facilities, such as barracks, oil rig installations, location filming locations, etc., where electrical power is required, such as for generators, and where power lines are typically run around temporary facilities cable. There are many situations when it is necessary to set up ad hoc agencies that require power. The systems and methods described herein may enable a more efficient way of quickly setting up and tearing down these devices, and may reduce the number of wires that must be run throughout the devices to supply power. For example, when special forces move into an area, they can pitch tents and pull many wires around the station to provide the required power. Alternatively, the systems and methods described herein may allow a military vehicle provided with power and source resonators to be parked at the center of the garrison and provide all power to a nearby tent where the device resonator may be integrated into the tent , or some other piece of equipment associated with each tent or area. A series of source-device-source-device resonators can be used to extend the power further into the tent. That is, the tent closest to the vehicle can then provide power to the tent behind it. The systems and methods described herein can provide significant improvements in the efficiency with which temporary installations can be set up and taken down, thereby increasing the mobility of associated equipment.

The systems and methods described herein may be used in a vehicle, such as for replacing wiring, installing new equipment, powering equipment brought into the vehicle, charging the vehicle's battery (e.g., for conventional gas-powered engine , for hybrid cars, for trams, etc.), to supply power to equipment installed inside or outside the vehicle, to supply power to equipment near the vehicle, etc. For example, the systems and methods described herein may be used to replace wires, such as to power lights, fans, and sensors throughout a vehicle. As an example, a typical car may have 50kg of wiring associated with it, and use of the systems and methods described herein may enable the elimination of a substantial amount of this wiring. The performance of larger and more weight sensitive vehicles such as airplanes or satellites can greatly benefit from reducing the number of cables that must be run throughout the vehicle. The systems and methods described herein may allow for the adaptation of removable or add-on portions of a vehicle with electric and electrical devices without the need for wires. For example, a motorcycle may have removable side boxes that act as temporary trunk volume when the rider is on a long road trip. These side boxes may have exterior lights, interior lights, sensors, automatic equipment, etc., and may require electrical connections and wiring if not equipped with the systems and methods described herein.

In-vehicle wireless power transfer systems can charge or power one or more mobile devices used in the car: hand-held mobile phones, Bluetooth headsets, Bluetooth hands-free speakerphones, GPS, MP3 players, for wireless communication via FM, Bluetooth, etc. Wireless audio transceiver for streaming MP3 audio through a car stereo. In-vehicle wireless power sources may utilize source resonators arranged in any of a number of possible configurations, including charging mats on the dashboard, charging mats mounted on the floor or between the seats and the center console, Charging "cups" or jacks that fit in cup holders or on the dashboard, etc.

The wireless power transfer source may utilize a rechargeable battery system such that the power battery is charged whenever the vehicle power is turned on, so that when the vehicle is turned off, the wireless power source can draw power from the power battery and continue wirelessly Charge or power mobile devices while still in the car.

Future plug-in electric cars, hybrid cars, etc. will need to be charged, and the user may need to plug in to charge when they get home or go to a charging station. Based on one overnight recharge, users may be able to drive up to 50 miles the next day. So, in the case of a hybrid car, if a person drives less than 50 miles on most days, they will be driving primarily on electric power. However, it would be beneficial if it didn't have to remember to plug the car in at night. That said, it would be nice to simply drive into the garage and let the car take care of its own charging. To this end, source resonators can be built into the garage floor and/or garage side walls, and device resonators can be built into the bottom (or sides) of the car. Even a transfer of a few kW may be enough to recharge the car overnight. In-vehicle device resonators can measure magnetic field properties to provide feedback to aid in the alignment of the vehicle (or any similar device) to a fixed resonant source. The vehicle can use this position feedback to automatically position itself for optimal alignment and therefore optimal power transfer efficiency. Another approach could be to use position feedback to help the human operator position the vehicle or equipment properly, such as by lighting up LEDs when it is well positioned, providing noise, etc. In cases where the amount of power being transferred may pose a safety threat to persons or animals that intrude into the active field volume, the source or receiver device may be equipped with a and alert the human operator to an active lighting curtain or some other external device. Additionally, the source device may be equipped with automatic sensing capabilities so that it can detect that its intended rate of power transfer has been interrupted by an intruding element, and in this case, switch off the source device and alert a human operator. Physical or mechanical structures such as hinged doors or inflatable air bags may be incorporated as physical barriers to prevent unwanted intrusion. Sensors such as optical, magnetic, capacitive, inductive, etc. can also be used to detect extraneous structures or disturbances between the source and the device resonator. The shape of the source resonator can be shaped to prevent the accumulation of water and debris. The source resonator can be placed in a conical housing, or it can have a housing with an angled top to allow water and debris to roll off. The source of the system may use the vehicle's battery power or its own battery power to communicate its presence to the source to initiate power transfer.

The source resonator may be mounted on a recessed or suspended post, on a wall, on a stand, etc. for coupling to a device resonator mounted on a shock absorber, cover, body panel, etc. of an electric vehicle. The source resonator can be enclosed or embedded in flexible housings such as cushions, cushions, bellows, spring-loaded housings, etc., allowing the electric vehicle to make contact with the structure containing the source coil without damaging the car in any way. The structure containing the source prevents objects from reaching between the source and the device resonator. Since wireless power transfer may be relatively insensitive to misalignment between the source and device coils, a variety of flexible source structures and parking procedures may be suitable for this application.

The systems and methods described herein may be used to trickle charge batteries of electric, hybrid or internal combustion engine vehicles. Vehicles may require small amounts of power to maintain or supplement battery power. Power can be transferred wirelessly from the source to a device resonator that can be incorporated into the front grill, roof, floor, or other portion of the vehicle. The device resonator can be designed to conform to the shape of the logo on the front of the vehicle or around the grille so as not to impede airflow through the radiator. A device or source resonator may have additional modes of operation that allow the resonator to be used as a heating element that can be used to melt snow or ice on a vehicle.

An electric or hybrid vehicle may require multiple device resonators, thereby increasing the ease with which the vehicle can approach the source resonator for charging (i.e., the greater the number and varying locations of the device resonators, the greater the vehicle's ability to pull in and greater opportunity to interface with a variety of charging stations), increasing the amount of power that can be delivered over a period of time (for example, additional device resonators may be required to prevent localized heating due to charging the current to acceptable levels), helping to The vehicle is parked/docked at a charging station, etc. For example, a vehicle may have multiple resonators (or a single resonator) with a feedback system that provides guidance to the driver or automated parking/docking structure for parking the vehicle for optimal charging conditions (i.e., traffic Optimal positioning of the tool's device resonator to the charging station's source resonator can provide greater power transfer efficiency). Automated parking/docking devices may allow automatic parking of vehicles based on how well the vehicles are coupled.

The power delivery system may be used to power equipment and peripherals of the vehicle. Power may be provided to peripheral devices while the vehicle is charging or not, or power may be delivered to a regular vehicle that does not require charging. For example, power can be transferred wirelessly to a conventional non-electric car to power the air conditioner, refrigeration unit, heater, lights, etc. while parked to avoid running the engine, which is essential to avoid running out of buildup in a garage parking spot or loading dock is very important. For example, power could be transferred wirelessly to the bus while it is parked to allow powering of lights, peripherals, passenger equipment, etc., avoiding the use of on-board engines or power supplies. Power can be transferred wirelessly to an aircraft while it is parked on the tarmac or in a pylon to power instruments, climate controls, de-icing equipment, etc. without having to use onboard engines or power supplies.

Wireless power transfer on vehicles can be used to implement vehicle-to-grid (V2G) concepts. Vehicle-to-grid is based on the use of electric vehicles and plug-in hybrid electric vehicles (PHEVs) as distributed energy storage for overnight charging when the grid is underutilized and available during peak demand periods that occur during the day discharge to the grid. The wireless power transfer system on vehicles and various infrastructures can be implemented in a way that enables bi-directional energy flow - allowing energy to flow back from the vehicle to the power grid - without requiring a connecting plug. Think of the huge fleet of vehicles parked in factories, offices, and parking lots as the "peak power capacity" of a smart grid. Wireless power transfer on vehicles could make such V2G scenarios a reality. By simplifying the process of connecting a vehicle to the grid (i.e., by simply parking it in a wireless charging-enabled parking lot), a certain number of vehicles will be "dispatchable" when the grid needs to tap their power. more likely. Without wireless charging, electric and PHEV owners will likely charge their vehicles at home and park them in regular parking lots at work. Who wants to have their vehicle plugged in at work if it doesn't need to be charged? With a wireless charging system capable of handling 3kW, 100,000 vehicles could return 300 megawatts to the grid—using energy generated the previous night at a cost-effective base load generation capacity. What makes it a viable V2G energy source is the silent self-charging PHEV and the streamlined ergonomics of electric vehicles.

The systems and methods described herein can be used to power sensors on a vehicle, such as sensors in tires to measure air pressure, or to operate peripherals in a vehicle, such as cellular phones, GPS devices, navigation devices, game consoles, audio Or video player, DVD player, wireless router, communication equipment, anti-theft equipment, radar equipment, etc. For example, the source resonator described herein can be built into the main cabin of a car to supply power to various devices located inside and outside the main cabin of the car. Where the vehicle is a motorcycle or the like, the devices described herein may be integrated into the body of the motorcycle, such as under the seat, and device resonators may be provided in the user's helmet, such as for communication, entertainment, signaling, etc., or a device resonator could be provided in the user's jacket, such as for displaying signals to the driver for safety reasons, etc.

The systems and methods described herein may be used in conjunction with transportation infrastructure, such as roads, trains, planes, ships, and the like. For example, source resonators can be built into roads, parking lots, rail lines, etc. Source resonators can be built into traffic lights, signals, etc. For example, using a source resonator embedded in the road and a device resonator built into the vehicle, it is possible to provide power to the vehicle as it travels along the road or while it is parked in a parking lot or on the side of the road. The systems and methods described herein may provide an efficient way to power and/or charge electrical systems in a vehicle while the vehicle traverses a road network or a portion of a road network. As such, the systems and methods described herein may contribute to the powering/charging of autonomous vehicles, automated guided vehicles, and the like. The systems and methods described herein may provide power to a vehicle in locations where the vehicle is normally idle or parked, such as near a traffic light or sign, on a highway ramp, or in a parking lot.

The systems and methods described herein may be used in an industrial setting, such as inside a factory to power machines, power/charge robots, power/charge wireless sensors on robotic arms, power/charge tools, and the like. For example, using the systems and methods described herein to supply power to devices on a robotic arm can help eliminate direct wire connections across the joints of the robotic arm. In this way, wear and tear on such direct wire connections is reduced and the reliability of the robot is increased. In this case, the device resonator could be outside on the robot arm, and the source resonator could be at the base of the robot, in a central location near the robot, integrated into the industrial facility the robot is servicing, and so on. Use of the systems and methods described herein can help eliminate wiring otherwise associated with power distribution within an industrial facility, and thus benefit the facility's overall reliability.

The systems and methods described herein may be used in subterranean applications such as drilling, mining, excavation, and the like. For example, electrical components and sensors associated with drilling or excavation can utilize the systems and methods described herein to eliminate cable runs associated with excavation mechanisms, drill bits, etc. its minimized. In another example, the systems and methods described herein can be used to provide power to mining equipment in mining applications, where the power requirements on the equipment can be high and the distances are large, but where no one will experience Related request fields. For example, a dig area may have resonator-powered dig equipment that has high power requirements and can dig relatively farther from the source resonator. As a result, the source resonator may need to provide high field strengths to meet these requirements, but personnel are sufficiently far outside these high strength fields. This high power unmanned solution can be adapted for many industrial applications.

The systems and methods described herein may also use near-field non-radiative resonance schemes for information transfer instead of power transfer (or in addition). For example, information conveyed by near-field non-radiative resonant techniques may be less susceptible to eavesdropping and thus may provide an increased level of security compared to conventional wireless communication schemes. In addition, information conveyed by near-field non-radiative resonance techniques may not interfere with the EM radiation spectrum and therefore may not be a source of EM interference, allowing communication over an extended frequency range and well within the limits set by any regulatory organization. Communication services may be provided between remote, inaccessible, or difficult-to-reach locations, such as between remote sensors, between parts of equipment or vehicles, in tunnels, caverns, and wells (e.g., oil wells, other drilling sites) and Between underwater or underground equipment, etc. Communication services can be provided where magnetic fields experience less loss than electric fields.

The systems and methods described herein may enable simultaneous transfer of power and communication signals between sources and devices in a wireless power transfer system, or they may enable transfer of power and communication signals during different time periods or at different frequencies . The performance characteristics of the resonator can be controllably varied to preferentially support or limit the efficiency or range of energy or information transfer. For example, the performance characteristics of the resonator can be controlled to improve security by reducing the range over which information can be transmitted. The performance characteristics of the resonator can be varied continuously, periodically, according to predetermined, calculated or automatically adjusted algorithms. For example, power and information transfer enabled by the systems and methods described herein may be provided in a time multiplexed or frequency multiplexed manner. The source and device can send signals to each other by tuning, changing, altering, dithering, etc., the impedance of the resonator, which can affect the reflected impedance of other resonators that can be detected. Information communicated as described herein may include information regarding device identification, device power requirements, handshaking protocols, and the like.

Sources and devices may sense, communicate, process and utilize location and positioning information about any other source and/or device in the power grid. Sources and devices may capture or use information such as altitude, inclination, longitude and latitude, etc., from a variety of sensors and sources that may be built into the source and device or may be components to which the source or device is connected. Position and orientation information may include sources such as global positioning sensors (GPS), compasses, accelerometers, pressure sensors, barometric barometric pressure sensors, positioning systems using Wi-Fi or cellular network signals, and the like. Sources and devices can use location and positioning information to find nearby wireless power transfer sources. Sources can broadcast or communicate with a central station or database that identifies their location. The device may obtain source location information from a central station or database, or from local broadcasts, and may guide the user or operator to the source by means of audible, vibratory or audio signals. Sources and devices can be nodes in power grids, communication networks, sensor networks, navigation networks, etc. or various combined functional networks.

Location and positioning information can also be used to optimize or coordinate power delivery. Additional information about the relative positions of the source and device can be used to optimize the magnetic field direction and resonator alignment. For example, the orientation of the device and source, which can be obtained from accelerometers and magnetic sensors, etc., can be used to identify the orientation of the resonator and the optimum direction of the magnetic field so that the flux is not blocked by the device circuitry. With such information, the source or combination of sources with the most suitable orientation can be used. Likewise, position and orientation information may be used to move or provide feedback to a user or operator of the device to place the device in an appropriate orientation or position to maximize power transfer efficiency, minimize losses, and the like.

Sources and devices may include power metering and measurement circuits and capabilities. Power metering can be used to track how much power is delivered to a device or delivered by a source. Power metering and power usage information may be used in a fee-based power delivery arrangement for billing purposes. Power metering can also be used to enable power delivery policies to ensure that power is allocated to multiple devices according to certain criteria. For example, power metering can be used to classify devices based on the amount of power received and prioritize those that have received the least power in terms of delivery. Power metering can be used to provide tiered delivery services that can be billed at separate rates, such as "guaranteed power" and "best effort power." Power metering can be used to compose and implement a hierarchical power delivery structure, and can enable priority devices to request and receive more power under certain circumstances or usage scenarios.

Power metering can be used to optimize power delivery efficiency and minimize absorption and radiation losses. The source may use information about the power received by the device in combination with information about the power output of the source to identify unsuitable operating environments or frequencies. For example, the source may compare the amount of power received by the device to the amount of power it transmitted to determine whether transmission losses may be unusually or unacceptably large. Substantial transmission losses may be due to unauthorized devices receiving power from the source, and the source and other devices may initiate frequency hopping of the resonant frequency or other countermeasures to prevent or deter unauthorized use. Large transmission losses can be due to, for example, absorption losses, and devices and sources can be tuned to the alternating resonant frequency to minimize such losses. Extensive transmission loss can also indicate the presence of unwanted or unknown objects or materials, and the source can turn down or turn off its power level until the unwanted or unknown object is removed or identified, at which point the source can resume communication with the remote device powered by.

Sources and devices can include authentication capabilities. Authentication can be used to ensure that only compatible sources and devices are able to transmit and receive power. Authentication can be used to ensure that only trusted devices of a particular manufacturer and not clones or devices and sources from other manufacturers or only devices that are part of a specific subscription or plan can receive power from the source. Authentication may be based on an encrypted request and response protocol, or it may be based on a perturbed unique signature of a particular device that allows it to be used and authenticated based on properties similar to physically unclonable functions. Authentication can be performed locally between each source and device with local communication, or it can be used with a third-person authentication method where the source and device communicate to a central authority. Authentication protocols can use location information to alert a local source of the real device.

Sources and devices can use frequency hopping techniques to prevent unauthorized use of wireless power sources. The source can continuously adjust or change the resonant frequency of power delivery. The frequency change may be performed in a pseudo-random or predetermined manner that is known, reproducible or communicated to authorized devices but difficult to predict. The rate of frequency hopping and the number of various frequencies used can be substantial and frequent enough to make unauthorized use difficult and impractical. Frequency hopping can be achieved by tuning the impedance network, tuning any drive circuits, using multiple resonators that are tuned or tunable to multiple resonant frequencies, etc.

The source may have user notification capabilities to show source status as to whether the source is coupled to the device resonator and delivering power, whether it is in standby mode, or whether the source resonator is detuned or disturbed by an external object. Notification capabilities may include visual, auditory and vibratory methods. This notification can be as simple as three colored lights, one for each state, and optionally a speaker to provide notification in case of operational error. Alternatively, the notification capability may involve an interactive display showing the status of the source and optionally providing instructions on how to fix or resolve any errors or problems identified.

As another example, wireless power transfer can be used to improve the safety of electronic detonators. The explosive device may be detonated with an electronic detonator, an electrical detonator or a shock tube detonator. With a low energy trigger signal transmitted conductively or by radio, electronic detonators use stored electrical energy (usually in a capacitor) to activate the igniter charge. Electrical detonators utilize a high energy conductive trigger signal to provide both the signal and energy needed to activate the igniter charge. Shock tubes send a controlled explosion from a generator to an igniter charge through a hollow tube coated with explosives. There are safety concerns associated with electrical and electronic detonators because of unintentional activation by accidental electromagnetic energy. Wireless power transfer via sharp resonant magnetic coupling can improve the safety of such systems.

Using the wireless power transfer methods disclosed herein, electronic initiation systems can be constructed without locally stored energy, thus reducing the risk of inadvertent activation. The wireless power source can be placed in close proximity (within a few meters) to the detonator. The detonator may be fitted with a resonant capture coil. Activation energy may be delivered when the wireless power source has been triggered. Triggering of the wireless power source can be initiated by any number of mechanisms: radio, magnetic near-field radio, conductive signaling, ultrasound, laser. Wireless power transfer based on resonant magnetic coupling also has the benefit of being able to transfer power through materials such as rock, soil, concrete, water, and other dense materials. The use of very high Q coils as receiver and source (with a very narrow band response and sharply tuned to the dedicated frequency) further ensures that the detonator circuit cannot catch accidental EMI and unintentional activation.

The resonator of a wireless powered device may be external or external to the device and wired to the device's battery. The device's battery can be modified to include appropriate rectification and control circuitry to receive AC power from the device's resonator. This may enable structures with larger external coils, such as may be built into the battery door of a keyboard or mouse or a digital still camera, or have a coil that is attached to the device but is back-wired to the battery/coil with a ribbon cable. An even larger coil of the converter. The battery door can be modified to provide interconnection from the external coil to the battery/converter (which will require exposed contacts to be able to contact the battery door contacts).

While this invention has been described in connection with certain preferred embodiments, those skilled in the art will recognize other embodiments and are intended to fall within the scope of this disclosure construed in the broadest sense permitted by law.

All documents referenced herein are hereby incorporated by reference.

Claims (26)

1. system comprises:

The source resonator, it has Q factor Q ₁With characteristic dimension x ₁, be coupled to generator, and second resonator, it has Q factor Q ₂With characteristic dimension x ₂, be coupled to the load that is positioned at resonator distance D place, source, wherein, the source resonator and second resonator are coupled with positive energy exchange wirelessly between the source resonator and second resonator, and wherein

2. the described system of claim 1, wherein, Q ₁＜100,

3. the described system of claim 1, wherein, Q ₂＜100.

4. the described system of claim 1 also comprises having Q factor Q ₃The 3rd resonator, it is configured to non-radiatively to shift energy with the source and second resonator, wherein

And

5. the described system of claim 4, wherein, Q ₃＜100.

6. the described system of claim 1, wherein, described source resonator is coupled to the generator with direct electrical connection.

7. the described system of claim 1 also comprises impedance matching network, and wherein, the source resonator is electrically connected coupling and is impedance-matched to generator with direct.

8. the described system of claim 1 also comprises tunable circuit, and wherein, the source resonator is coupled to generator with directly being electrically connected by tunable circuit.

9. claim 6,7 or 8 described system, wherein, at least one in directly being electrically connected is configured to keep basically the resonant mode of source resonator.

10. the described system of claim 6, wherein, the source resonator has the first terminal, second terminal and center terminal, and wherein, between the first terminal and the center terminal and second terminal equate basically with impedance between the center terminal.

11. the described system of claim 6, wherein, the source resonator comprise have the first terminal, the capacitive load loop of second terminal and center terminal, and wherein, between the first terminal and the center terminal and second terminal equate basically with impedance between the center terminal.

12. the described system of claim 6, wherein, the source resonator is coupled to impedance matching network and impedance matching network also comprises the first terminal, second terminal and center terminal, and wherein, between the first terminal and the center terminal and second terminal equate basically with impedance between the center terminal.

13. claim 10,11 or 12 described system, wherein, the first terminal and second terminal are directly coupled to generator, and drive with the oscillator signals of out-phase near 180 degree.

14. claim 10,11 or 12 described system, wherein, described source resonator has resonance frequency omega ₁, and the first terminal and second terminal be directly coupled to generator, and with being substantially equal to resonance frequency omega ₁Oscillator signal drive.

15. claim 10,11 or 12 described system, wherein, described center terminal is connected to electrical ground.

16. the described system of claim 15, wherein, described source resonator has resonance frequency omega ₁And the first terminal and second terminal are directly coupled to generator, and with being substantially equal to resonance frequency omega ₁Frequency drive.

17. the described system of claim 2 comprises a plurality of capacitors that are coupled to generator and load.

18. the described system of claim 1, wherein, the source resonator and second resonator all are closed in the low loss tangent material.

19. the described system of claim 1 also comprises circuit for power conversion, wherein, second resonator is coupled to circuit for power conversion to send DC power to load.

20. the described system of claim 1 also comprises circuit for power conversion, wherein, second resonator is coupled to circuit for power conversion to send AC power to load.

21. the described system of claim 1 also comprises circuit for power conversion, wherein, second resonator is coupled to circuit for power conversion to send AC and DC power to load.

22. the described system of claim 1 also comprises circuit for power conversion and a plurality of load, wherein, second resonator is coupled to circuit for power conversion, and circuit for power conversion is coupled to described a plurality of load.

23. the described system of claim 7, wherein, described impedance matching network comprises capacitor.

24. the described system of claim 7, wherein, described impedance matching network comprises inductor.

25. the described system of claim 8, wherein, described tunable circuit comprises variable capacitor.

26. the described system of claim 8, wherein, described tunable circuit comprises variable inductor.

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