200524265 (1) 九、發明說明 【發明所屬之技術領域】 本發明係關於同步電動機之控制裝置、電器及模組。 【先前技術】 交流電動機之速度或者不使用位置感測器之控制方 式，至目前爲止，公開有各種手法。例如，在以交流電動 機之代表例的永久磁鐵同步電動機爲對象之例子中，有曰 本專利特開2001-251889號公報等之方式爲所周知。此控 制方式係不使用位置感測器，代之在控制器內部進行磁極 位置的推測運算的方式。 另外，電動機之負載裝置所產生之週期性轉矩干擾的 控制方法，則有日本專利特開平1 0 - 1 7 4 4 8 8號公報、日本 專利特開 2 0 0 2 - 3 4 2 9 0號公報等。日本專利特開平1 Ο-ΐ 7 4 4 8 8 號公報 所記載 之方式 ，係 抽出 包含在 電動機 之速度 檢測値之脈動成分，在換流器輸出電壓加上補正以將其抵 消之方式。在此方式之實現上，需要速度資訊。 日本專利特開2002-3 4290號公報之方式，係檢測包 含在轉矩電流成分之脈動成分，藉由在旋轉速度加上補 正，以穩定地控制電動機之方式。 [專利文獻1]日本專利特開2 00 1 _2 5 1 8 89號公報 [專利文獻2]日本專利特開平ΐ〇·ΐ74488號公報 [專利文獻3]日本專利特開2 002-3 4290號公報 200524265 【發明內容】 [發明所欲解決之課題] 在日本專利特開2001-251889號公報之方式中，雖可 實現無位置感測器，但是，在負載裝置連接壓縮機等之週 期性干擾所伴隨之負載的情形，無法抑制該週期性干擾。 其結果爲，產生旋轉脈動，而有變成裝置之振動、噪音的 原因之課題。 日本專利特開平1 0- 1 744 8 8號公報之方式，雖可抑制 週期性干擾，但是，需要電動機之旋轉速度資訊。因此， 需要某種之速度檢測器。原理上，雖可安裝全1C等之位 置感測器，使用於電動機之速度檢測，但是，在負載裝置 爲空調等之壓縮機的情形，由於周圍環境之問題，感測器 之安裝有困難。 代替位置感測器，檢測電動機之中性點電位，由其之 變動成分以獲得速度資訊之方法雖也爲所周知，但是，速 度資訊以電氣角而言，只能每6 0度獲得，高速、高精度 之速度檢測有困難。特別是，基於驅動電動機之換流器之 導通延遲（空載期間）之影響所致之週期性干擾，對於電 動機之驅動頻率，係以6倍之頻率變動故，以電氣角6 0 度間隔之速度檢測，不可能抑制此干擾。另外，獲得中性 點電位用之配線，則有需要多1條線之課題。 日本專利特開2 0 0 2 - 3 4 2 9 0號公報之方式係因應包含 在轉矩電流之脈動，改變旋轉速度本身，以提升控制裝置 整體之穩定性之方式。因此，旋轉脈動進一步增加，振 -6- (3) (3)200524265 動、噪音的課題無法解決。另外，對象爲感應電動機故’ 照這樣，則難於適用在同步電動機。 本發明之目的在於提供：可以抑制週期性干擾所引起 之振動、噪音之電動機之控制裝震。 [解決課題用手段] 本發明之特徵之一爲具有，在同步電動機之控制裝置 中，依據軸誤差推算値，求得前述電動機或者負載之某一 方，或者雙方所產生的週期性干擾成分之週期性干擾推算 器。 [發明之效果] 如依據本發明，可以實現能抑制週期性干擾所引起之 振動、噪音之電動機之控制裝置。 【實施方式】 接著，參照第1圖至第1 5圖，說明依據本發明之交 流電動機之控制裝置的實施例。另外，在以下之實施例 中，電動機雖使用永久磁鐵型同步電動機（以下，省略爲 Ρ Μ電動機）做說明，但是’關於其他之同步電動機（例 如，繞線型同步電動機、磁阻電動機等），也可同樣地加 以實現。 [實施例Π 200524265 (4) 第1圖係顯示依據本發明之交流電動機控制 施例1之系統構造方塊圖。本實施例1之控制裝 藉由上位控制裝置之指令1 〇〇，對電動機給予旋 ω”之旋轉數指令產生器1，及運算電動機之交 壓，轉換爲脈波寬度調變波訊號（PWM訊號） 出之控制器2 ’及藉由此p w Μ訊號所驅動之換济 對換流器3供給電力之轉換器4，及控制對象之 機5 ’及ΡΜ電動機之負載之壓縮機6，及檢測 對換流器3供給之電流1 〇之電流檢測器7所形成 控制器2係依據藉由電流檢測器7所檢測之 在控制器內部運算流經ΡΜ電動機5之三相交流 Iv、Iw而加以再現之電流再現器8，及將所再現 流電流 Iuc、Ivc、Iwc藉由相位角（9 dc (在控制 假定之P Μ電動機的磁鐵磁通之位置）予以座標 軸上之成分Idc、Iqc之dq座標轉換器9，及對: 之電流成分，給予指令Iq*之Iq*產生器1〇，及 對於d軸上之電流成分，給予指令Id*之Id*產^ 及依據Id*、Iq*、以及電氣角頻率指令ωΐ*，運 令Vdc*、Vqc*之電壓指令運算器12，及將Vdc* 換爲三相交流電壓指令 Vu*、Vv*、Vw*之dq 1 3，及依據三相交流電壓指令，產生開關換流器 波寬調變訊號（PWM訊號）之PWM脈波產生器 算相當於P Μ電動機5之磁鐵磁通位置<9 d與在 內部所假定之位置0 dc之誤差的角度（軸誤差） 丨裝置之實 丨置係由： 轉數指令 流施加電 而予以輸 E器3，及 PM電動 轉換器4 • 〇 電流1 0， 電流Iu、 之三相交 器內部所 轉換爲各 於q軸上 同樣地， fe 器 11， 算電壓指 、V q c * 轉 逆轉換器 3用之脈 14，及推 控制器2 △ 0之 (5) (5)200524265 △ 0推算器1 5，及進行加法，以及減法之加減法器16, 及對於軸誤差推算値△ 0 dc，給予指令之零指令產生器 1 7，及爲了將△ 0控制爲零，對電氣角頻率指令ω ；[ *加上 補償之比例補償器1 8，及利用ρ Μ電動機之極數Ρ，將旋 轉數指令ω”轉換爲電動機之電氣頻率指令ω 1*之轉換增 益1 9，極積分電氣角頻率，運算磁鐵磁通位置β dc之積 分器2 0，極依據軸誤差推算値△ 0 d c，推算週期性干擾轉 矩成分之△ Tm之△ Tm推算器2 1 (週期性干擾推算器）， 極依據△ Tm之推算値△ Tmc，於q軸電流指令Iq*加上補 正之轉矩控制器（ATR ) 22所形成。 對換流器3供給電力之轉換器4係以：交流電源4 1， 及整流交流電之二極體橋4 2，及抑制包含在直流電壓之脈 動成分之平滑電容器43所構成。 . 接著，利用第1圖說明本實施例1之動作原理。轉換 增益1 9係依據來自旋轉數指令產生器1之旋轉數指令 ω”，運算PM電動機之電氣角頻率ωΐ*予以輸出。進 而，利用積分器20來積分ωΐ*，運算交流相位0dc。在 電流再現器8中，依據以電流檢測器7所檢測之電源電流 10，藉由日本專利特開平 8 - 1 92 3 6號公報等所記載之手 法，藉由運算PM電動機之三相交流電流而加以再現。接 著，在dq座標轉換器9中，藉由0 dc而將所再現之交流 電流Inc、Ivc、Iwc轉換爲以角頻率ω 1*旋轉之旋轉座標 軸（dq軸）上之電流成分Idc、Iqc。Iqc係在Iq*產生器 1〇中被處理，成爲q軸上之電流指令Iq*。另外，Id*產生 -9- 200524265 (6) 器1 1係產生d軸上之電流指令I d (在非凸極型轉1 電動機中，通常Id*=〇)。在電壓指令運算器12中 這些指令（Id*、Iq*)與角頻率指令ωΐ*，運算割 電動機之施加電壓Vdc*、Vqc。Vdc*、Vqc係藉由 轉換器13而再度被轉換爲交流量，進而，在PWM 生器1 4中，被轉換爲脈波寬調變波訊號，被送往 3。關於這些之基本動作，係與日本專利特開2 002-號公報所記載之手法相同。 在△ 0推算器15中，進行PM電動機內之磁 的位置0 d以及控制器內之位置0 dc之誤差△ 0之 △ Θ係藉由第2凸所示之向量圖所定義。設PM電 部之實際的磁鐵磁通P之位置爲d軸，與其正交之 軸。對於此，將在控制器內所假定之dq軸定義爲 軸，兩者之偏差係相當於軸誤差△ β。 如求得△ 0 ，藉由對其加以修正，可使d-q軸 qc軸一致，能實現PM電動機之無感測器控制。△ 算例如如第3圖所示般，在i q *與I q c之差乘上比 K0，可以當成△ 0之推算値△ 0dc。Iqc係依據負 等，於0d與0dc產生偏差而變動故，由Iqc之動 以反推△ β。但是，在第3圖之構造的情形’要高 得△ 0有困難。在提升精度上，例如，可以依據曰 特開2002-272194號公報之數學式（3)等加以運算 依據基於△ β推算器1 5所運算之軸誤差推算 dc，進行令其成爲零之反饋控制。藉由加減法器1 F之PM ，依據 :於 PM dq逆 脈波產 換流器 272194 鐵磁通 推算。 動機內 軸爲q d c - q c 與 dc- 0之推 例增益 載變動 向，可 精度求 本專利 〇 値△ Θ 6來求 -10 - 200524265 (7) 得零指令產生器17之指令（零）與ASdc之差，藉由比 例補償器1 8而在角頻率ω 1 *加上補償。如第2圖之向量 圖所示般，在△ 0爲正之情形，dc-qc軸比d-q軸更前進 故，藉由降低ω 1 *，可令△ 0減少。反之，在△ 0爲負之 情形，提升ω 1 *，可使d - q軸與d c - q c軸一致。藉由如此 控制，可以不使用P Μ電動機之磁極軸的位置感測器，使 控制器內部之相位角△ 0 dc與實際的ΡΜ電動機內之磁鐵 磁通位置0 d —致，能實現無位置感測器控制。 接著，詳細說明本發明之特徵部份之△ Tm推算器2 1 (週期性干擾推算器），及轉矩控制器22。關於圖之各區 塊，R係表示PM電動機之捲線電阻，L係PM電動機的 電感’ P係PM電動機的極數，Ke係PM電動機的發電常 數（磁鐵磁通），J係PM電動機與負載裝置之整體慣 量’ s係拉普拉斯轉換所使用之微分運算元。 如第4圖所示般，q軸電流I q係依據施加在p Μ電動 機之施加電壓Vq，及電壓干擾VD，及電動機之電氣常數 R、L之關係所產生。I q係正交於p Μ電動機的磁鐵磁通 (d軸）之成分，藉由乘以發電常數Ke，變成電動機轉矩 Tm。PM電動機之旋轉速度ω r係積分電動機轉矩Tm與負 載轉矩TL之差者。此處，負載轉矩Tl係依據負載裝置之 種類或用途，具有各種之特性。於ω r乘以極對數 (P/2) ’可以獲得電動機之電氣角頻率ωΐ，其之積分値 係變成P Μ電動機之位置β d。軸誤差△ β係作爲與控制 器內之相位β d c之差而可以獲得。 -11 - 200524265 (8) 此處，考慮在電壓干擾VD或者負載轉矩tl包含有週 期性成分。 週期性之電壓干擾V D，例如在ρ μ電動機之磁鐵磁 通不均勻’有導磁偏差之情形’或者捲線之相位間偏差之 情形’等效性成爲週期性電壓干擾而造成影響。或者基於 換流器之臂短路防止期間（空載時間）之影響所致之干擾 等，也以換流器之驅動頻率的6倍頻率而產生。 另外’週期性負載轉矩干擾例如可以思考在冷凍庫或 空調等所使用之往復式壓縮機，或單旋轉式壓縮機等之負 載。在往復式壓縮機之情形，以電動機之一旋轉爲一週 期，負載激烈變動。 爲了控制性地抑制這些振動、噪音，如構成前述之週 期性轉矩變動成爲零之控制系統即可。在習知的發明中， 係以某種手段檢測旋轉速度資訊，控制施加電壓令該旋轉 脈動成爲零而加以對應。在空調等之壓縮機中，難於直接 獲得速度資訊故，所以檢測電動機之中性點電位的變動， 獲得以電氣角60度刻度之資訊，以推算速度。 但是，在此方式中’對於電氣角週期，只能獲得6點 之資訊，作爲速度資訊並不充分。在此狀態下，產生60 度份之延遲的影響，或在速度檢測精度出現問題。或者對 於基於電動機之感應電動勢電壓之畸變所產生之脈動，成 爲比電氣角週期短之週期（主要爲1/6週期）故，要將 其抑制有其困難。 另外，雖也可考慮驅使控制理論，構築附在千擾觀測 -12- 200524265 (9) 器等，以推算脈動轉矩之手法，但是，在此 本身之響應頻率變成課題。脈動轉矩之頻率ϋ 應其’也需要提高觀測器之設定響應。脈動華 分變得愈高’則觀測器之高響應性更被要求， 要局速運算處理。因此，作爲目前爲止之週其 制方法，一般在低速領域之振動抑制雖屬可齡 速旋轉時之抑制很難。 舉其一例，考察利用泛用微電腦以構成 形。在設觀測器響應時間爲lms ( lOOOrad/s-之情形’可檢測之脈動轉矩爲3 0Hz之程度。 4極之電動機時，則變成 900 [r/min]。在壓縮 最局旋轉數多數在3000[r/min]以上故，如不右 速度以下，則變得無法適用。 在本發明中，著眼於第4圖之方塊圖，提 △ 0以推算轉矩脈動成分△ Tm之手法。軸誤 間可運算瞬間値故，可不受到運算延遲之影響 度之推算。另外，對於驅動頻率，對於高的頻 如’ 6倍之振動成分），也可以檢測爲其特 爲’與習知之週期性干擾控制方法相比，大幅 之推算變成可能。 在此種週期性干擾產生之情形，電動機轉 載轉矩TL之差，係變成週期性之轉矩變動， 噪音之原因。在抑制此振動、噪音上，例如， 材包圍裝置整體等之對策，變成裝置之大型化 |形，觀測器 E之情形，因 I矩之頻率成 結果爲，需 3性千擾之抑 ί，但是，局 觀測器之情 約 150Hz) 如將其設爲 !機之情形， :30%程度之 :出由軸誤差 差△ 0係瞬 ，而做局精 率成分（例 徵。此結果 至高速領域 矩Tm與負 成爲振動、 需要以吸音 ，以及成本 -13- 200524265 (10) 增加，係一種需要解決之課題。 在控制性地抑制振動、噪音上，如構成令前述之週期 性轉矩變動成爲零之控制系統即可。在習知的發明中，以 某種手段檢測旋轉速度，控制施加電壓以令該旋轉脈動成 爲零而加以對應。但是，在空調等之壓縮機中，電動機係 組裝在壓縮機內部故，難於簡單地獲得速度資訊，另外， 即使可以獲得，頂多只能獲得相當於電氣角6 0度刻度之 資訊。因此，局精度化有其困難。 在本發明中，著眼於第4圖之方塊圖，提出由軸誤差 △ <9以推算轉矩脈動成分△ T m之手法。軸誤差△ 0係瞬 間可運算瞬間値故，可不受到運算延遲之影響，而做高精 度之推算。另外，對於驅動頻率，對於高的頻率成分（例 如，6倍之振動成分），也可以檢測爲其特徵。 第5圖係分別顯示負載轉矩TL以角頻率ω d而在正 弦波狀包含振動之成分的情形之轉矩脈動成分（△ Tm )、 旋轉速度變動（△ ω r )、軸誤差（△ Θ )。如考慮穩定 狀,¾時’ Tm與TL之平均値係一致，ΔΤγπ只是振動成分 而已（第5(b)圖）。包含於旋轉速度之振動成分 係積分此ΔΤ!!!者，與△Tm相比，變成相位延遲90度之 波形。振動之大小本身雖依據慣量J而變化，但是，相位 可認爲幾乎延遲9 0度。軸誤差△ 0係變成進而積分 △ ω r ’令符號反轉者（以第2圖所示定義之關係，反轉 符號）故，相位變成前進9 0度（以積分而延遲9 0度，符 5虎反轉故’變成則進9 0度）。即A T m之變動成分，在 -14- (11) (11)200524265 △ 0中，係變成同相位之振動波形而被觀測到。如由方塊 線圖導出此關係時，變成如下。 第6(a)圖係顯示由ΔΤγπ至△ 0之方塊圖。藉由逆 轉換此方塊圖，可以求得由△ 0至△ Tm之傳達函數，變 成如同圖（c)般。 依據第6(c)圖而求得ΔΤιη時，由△edcCzXe之 推算値）可直接推算轉矩之脈動成分。但是，二階微分 △ 0dc實際上不可能。原本爲推算値，也多數包 含有檢測値之雜訊等故，使用微分會增加推算誤差，另 外，也有來自運算週期之界限。 因此，著眼於「千擾成分爲週期函數」之點，將s = j d代入第 6 ( c )圖。如此一來，如第 6 ( d )圖所示 般，變成可推算將△ β予以常數倍者會成爲△ Tm。此結果 爲，與第5圖之（b )與（d )之波形的關係一致。 具體化第6 ( d )圖之構造係第7圖所示之△ Tm推算 器2 1 (週期性干擾推算器）。△ Tm推算器2 1 (週期性干 擾推算器）係由將△ Θ dc予以2J/P倍之比例增益21 1，及 2個乘法器2 1 2所形成，實施第6 ( d )圖之運算。 依據第7圖，由△$(!(：可以推算包含在ΔΤιη之角速 度ω d之週期性干擾成分。 接著，說明抑制此△ Tm之轉矩控制器 22 (第 8 圖）。 轉矩控制器所必要之條件爲， (1 ) 對於週期性千擾成分，追從性高， -15- 200524265 (12) (2 ) 對於週期性干擾以外之成分，感度低 之2種。（1 )係爲了抑制週期性干擾，最爲重要之條 件。 (2 )係在過渡時，產生直流之△ β之情形，防止轉 矩控制器影響控制整體所必要之條件。如前述般，本補償 器係以「週期性干擾」爲前提，第6圖所示之等效轉換並 不適用於直流干擾等。因此，對於週期性千擾以外之成 分，需要降低感度。 第8圖所示之轉矩控制器係由：對於△ Tm之推算値 △ Tmc，給予零指令之零指令產生器17，及以角頻率ad 持有峰値之正弦波傳達函數221，及轉矩控制增益222所 構成。此時’令正弦波傳達函數221之角頻率ud與週期 性干擾轉矩之變動頻率一致。在壓縮機等之情形，週期性 干擾之變動頻率係成爲與驅動頻率一致之頻率故，容易使 ω d與脈動頻率一致。另外，對於週期性之電壓干擾，也 幾乎都成爲驅動頻率之整數倍故，此ω d之設定比較容 易。 另外’本轉矩控制器係滿足前述條件（1 ) 、 （ 2 )之 雙方。正弦波傳達函數在角頻率wd中，增益變成無限 大，可使只包含在ΔΤιη之yd成分之要素的偏差成爲零， 關於其以外之頻率成分’並不具有感度。關於正弦波傳達 函數之詳細’例如顯示在日本專利特開平7-2 0906號公報 藉由使用第8圖之轉矩控制器2 2，角頻率ω ^之正弦 -16- (13) 200524265 波狀的訊號IqS1N*被加在Iq*，可以抑制轉矩脈動成分 △ T m 〇 以上，雖以週期性干擾負載爲例而說明本發明之第1 實施例，但是，關於週期性電壓干擾，也可同樣地採取對 策。另外，電流檢測法雖使用由轉換器4之電流値10以 再現電動機電流之方法，但是也可直接以全C T或分激阻 抗來檢測各相電流。 [實施例2] 接著，利用第9圖，說明依據本發明之實施例2。 在實施例1中，導入了對於轉矩脈動之振動頻率 ω d，增益變成無限大之正弦波傳達函數。此結果爲，包 含在轉矩脈動之d成分雖被去除，代之，PM電動機之 驅動電流的畸變會變大。爲了瞬間追從負載轉矩變動，無 法避免瞬間電流之變動變大，此會成爲P Μ電動機之效率 劣化，或者基於峰値電流之過電流跳脫等之不良發生的原 因。因此，調整爲整體性緩和振動、噪音之降低效果，反 之，防止電流畸變之妥協點，可以說較爲實用。 實施例2係對於週期性干擾之抑制效果，附加調整抑 制能力之功能的實施例。 第9圖係顯示本實施例2之轉矩控制器22b之構造。 藉由代替第1圖之轉矩控制器2 2而使用本轉矩控制器 2 2，實施例得以實現。 在第9圖之轉矩控制器2 2 B中，係由：具有峰値抑制 -17- (14) (14)200524265 功能之正弦波傳達函數22 IB，及控制增益222B，及零指 令產生器1 7，及加減法器1 6所構成。具有峰値抑制功能 之正弦波傳達函數22 1B係在分母具有Ta · s項目，藉由 T a之大小，使得函數之峰値之數値可以改變。 此結果爲，變成可以調整ω d成分之千擾抑制效果， 在噪音、振動與P Μ電動機相電流之畸變的最佳點的驅動 變成可能。 [實施例3] 接著，利用第1 0圖，說明依據本發明之實施例3。 在實施例1及2中，作爲轉矩控制器，係導入對於轉 矩脈動之振動頻率ω d，增益變成最大（在實施例1中， 爲無限大）之傳達函數。在這些控制器中，最終，脈動成 分之收斂響應由輸出段之控制增益（控制增益222或者控 制增益2 2 2 B )的大小所決定。 但是，這些控制增一隻設定値以及實際響應時間之關 係複雜’變成要製作波得線圖而做檢討，或者藉由模擬、 實測而求得。因此，調整作業要非常多勞力。 在實施例3中，提供簡化調整作業用之轉矩控制器。 第1 0圖係顯示本實施例3之轉矩控制器22C的構造 例。藉由代替第1圖之轉矩控制器22而使用本轉矩控制 器22C，實施例3得以實現。— 在第10圖之轉矩控制器22C中，係由：單相-dq座標 轉換器2 2 3、一次延遲濾波器2 2 4、積分控制器2 2 5、dq- -18- 200524265 (15) 單相逆轉換器226、及積分器20、加減法器16、零指令產 生器1 7所構成。接著，說明此轉矩控制器2 2 C之動作。 將△ Tm推算器之輸出△ Tmc於單相-dq座標轉換器 223中分解爲SIN成分與COS成分。另外，單相-dq座標 轉換器2 23之轉換式係如下述： [數學式1] ATds cos(iy〆） ATqs -sin(^〇200524265 (1) IX. Description of the invention [Technical field to which the invention belongs] The present invention relates to a control device, an electric appliance and a module for a synchronous motor. [Prior art] Various methods have been disclosed so far for the speed of an AC motor or a control method that does not use a position sensor. For example, as an example of a permanent magnet synchronous motor which is a representative example of an AC motor, a method such as Japanese Patent Application Laid-Open No. 2001-251889 is known. This control method does not use a position sensor and instead performs a magnetic pole position estimation operation inside the controller. In addition, for the method of controlling the periodic torque interference generated by the load device of the motor, there are Japanese Patent Laid-Open No. 1 0-1 7 4 4 8 8 and Japanese Patent Laid-Open No. 2 0 0 2-3 4 2 9 0 Bulletin, etc. The method described in Japanese Patent Laid-Open No. 10-ΐ 7 4 4 8 8 is a method of extracting the pulsating component included in the speed detection of the motor and adding a correction to the inverter output voltage to cancel it out. Speed information is needed to implement this method. The method of Japanese Patent Laid-Open No. 2002-3 4290 is a method of detecting a pulsating component included in the torque current component and adding a correction to the rotation speed to control the motor stably. [Patent Literature 1] Japanese Patent Laid-Open No. 2 00 1 _2 5 1 8 89 [Patent Literature 2] Japanese Patent Laid-Open No. ΐ〇 · ΐ74488 [Patent Literature 3] Japanese Patent Laid-Open No. 2 002-3 4290 200524265 [Summary of the Invention] [Problems to be Solved by the Invention] In the method of Japanese Patent Laid-Open No. 2001-251889, although a position sensor can be realized, a periodic interference mechanism such as a compressor is connected to a load device. With the load situation, this periodic interference cannot be suppressed. As a result, there is a problem that a rotation pulsation occurs, which causes vibration and noise of the device. The method disclosed in Japanese Patent Laid-Open No. 10-1744744 can suppress periodic interference, but requires information on the rotation speed of the motor. Therefore, some kind of speed detector is needed. In principle, although all position sensors such as 1C can be installed and used for speed detection of motors, when the load device is a compressor such as an air conditioner, it is difficult to install the sensor due to the surrounding environment. Instead of the position sensor, the method of detecting the potential of the neutral point of the motor and obtaining the speed information from its changing components is well known, but the speed information can only be obtained every 60 degrees in terms of electrical angles. 2. High-precision speed detection is difficult. In particular, based on the periodic interference caused by the on-delay (no-load period) of the inverter that drives the motor, the driving frequency of the motor changes at a frequency of 6 times, so it is spaced at an electrical angle of 60 degrees. For speed detection, it is impossible to suppress this interference. In addition, there is a problem that an additional line is required to obtain a wiring for a neutral point potential. The method of Japanese Patent Laid-Open No. 2000-2-3 4 2 90 is a method of changing the rotation speed itself in accordance with the pulsation of the torque current to improve the overall stability of the control device. Therefore, the rotation pulsation further increases, and the problem of vibration and noise of vibration -6- (3) (3) 200524265 cannot be solved. In addition, since the object is an induction motor, it is difficult to apply it to a synchronous motor. An object of the present invention is to provide a control device for a motor that can suppress vibration and noise caused by periodic interference. [Means for solving problems] One of the features of the present invention is that the control device of the synchronous motor calculates the period of the periodic interference component generated by one of the motor or the load, or both, by estimating 値 based on the shaft error. Sexual interference estimator. [Effects of the Invention] According to the present invention, a control device for a motor capable of suppressing vibration and noise caused by periodic interference can be realized. [Embodiment] Next, an embodiment of a control device for an AC motor according to the present invention will be described with reference to Figs. 1 to 15. In addition, in the following embodiments, although the motor is described using a permanent magnet synchronous motor (hereinafter, omitted as a PM motor), 'about other synchronous motors (for example, a wound synchronous motor, a reluctance motor, etc.), The same can be achieved. [Embodiment Π 200524265 (4) Fig. 1 is a block diagram showing a system configuration of an AC motor control embodiment 1 according to the present invention. In the control device of the first embodiment, the instruction 1 of the host control device is used to give the motor a rotation number command generator 1 and calculate the cross voltage of the motor to convert it into a pulse width modulated wave signal (PWM Signal 2) and a converter 4 that supplies power to the inverter 3 by means of the drive driven by the pw M signal, and a control unit 5 'and a compressor 6 that is a load of a PM motor, and The controller 2 formed by the current detector 7 that detects the current 10 supplied to the inverter 3 is based on the calculation of the three-phase AC Iv, Iw flowing through the PM motor 5 inside the controller detected by the current detector 7. The reproduced current reproducer 8 and the reproduced currents Iuc, Ivc, Iwc give the components Idc, Iqc on the coordinate axis by a phase angle (9 dc (the position of the magnetic flux controlling the assumed PM motor). The dq coordinate converter 9 and the current component of: Iq * generator 10 which gives the instruction Iq *, and the current component on the d axis, which gives the instruction Id * of Id * production ^ and according to Id *, Iq * , And the electrical angular frequency command ωΐ *, the operation order Vdc *, Vqc * Voltage command calculator 12, and dq 1 3 for replacing Vdc * with three-phase AC voltage commands Vu *, Vv *, Vw *, and generating switching converter wave width modulation signals (PWM The signal of the PWM pulse generator is equivalent to the angle (axis error) of the magnetic flux position < 9 d of the PM motor 5 and the position 0 dc assumed internally (axis error). The rotation speed instruction stream is applied to the electric device 3 and the PM electric converter 4 • 〇 The current 10, the current Iu, and the internal current of three-phase inverter are converted into the same on the q-axis, and the voltage of the device 11 is calculated. Finger, V qc * pulse 14 for inverse converter 3, and push controller 2 △ 0 of (5) (5) 200524265 △ 0 estimator 15 and addition and subtraction device 16 for addition and subtraction, and For the estimation of the axis error 値 △ 0 dc, the command zero command generator 17 is given, and in order to control △ 0 to zero, the electrical angular frequency command ω is used; [* plus the compensated proportional compensator 18, and the use of ρ The number of poles P of the Μ motor converts the rotation number command ω ”into the electrical frequency command ω 1 * of the motor. Conversion gain 19, pole integral electrical angular frequency, integrator 2 0 for calculating magnetic flux position β dc, poles to calculate 値 △ 0 dc based on shaft error, to estimate △ Tm and △ Tm estimator 2 of periodic interference torque component 1 (periodic disturbance estimator), which is calculated based on △ Tm 値 △ Tmc, which is formed on the q-axis current command Iq * plus the corrected torque controller (ATR) 22. The conversion of the power supplied to inverter 3 The device 4 is composed of an AC power source 41, a diode bridge 4 2 that rectifies AC power, and a smoothing capacitor 43 that suppresses a pulsating component included in the DC voltage. Next, the operation principle of the first embodiment will be described using FIG. 1. The conversion gain 19 is based on the rotation number command ω "from the rotation number command generator 1, and calculates the electrical angular frequency ω 旋转 * of the PM motor and outputs it. Furthermore, the integrator 20 is used to integrate ωΐ * to calculate the AC phase 0dc. In the current The reproducer 8 calculates the three-phase AC current of the PM motor based on the power supply current 10 detected by the current detector 7 and by the method described in Japanese Patent Laid-Open No. 8-1 92 3 6 and the like. Then, in the dq coordinate converter 9, the reproduced AC currents Inc, Ivc, and Iwc are converted into current components Idc, on the rotating coordinate axis (dq axis) rotating at an angular frequency ω 1 * by 0 dc. Iqc. Iqc is processed in the Iq * generator 10 and becomes the current command Iq * on the q axis. In addition, Id * generates -9- 200524265 (6) Device 1 1 generates the current command I d on the d axis (In a non-salient pole type 1 motor, Id * = 〇). In the voltage command calculator 12, these commands (Id *, Iq *) and angular frequency command ωΐ * are used to calculate the applied voltage Vdc *, Vqc. Vdc * and Vqc are converted into AC again by converter 13. The amount is further converted into a pulse width modulated wave signal in the PWM generator 14 and sent to 3. The basic operations of these are the same as those described in Japanese Patent Laid-Open No. 2 002- In the △ 0 estimator 15, the error between the magnetic position 0 d in the PM motor and the position 0 dc in the controller Δ 0 Δ θ is defined by the vector diagram shown in the second convex. Let PM The position of the actual magnetic flux P of the electric part is the d-axis, which is orthogonal to the axis. For this, the dq-axis assumed in the controller is defined as the axis, and the difference between the two is equivalent to the axis error Δ β. If △ 0 is obtained, by modifying it, the dq axis and qc axis can be consistent, and sensorless control of PM motors can be realized. △ For example, as shown in Figure 3, between iq * and I qc The difference multiplied by the ratio K0 can be used as the calculation of △ 0 値 △ 0dc. Iqc is based on negative and equal, and varies between 0d and 0dc. Therefore, the movement of Iqc is used to reverse △ β. However, the structure in Figure 3 It is difficult to achieve a high value of △ 0. In terms of improving accuracy, for example, according to Japanese Patent Application Laid-Open No. 2002-272194 Equation (3) is calculated based on the axis error calculated by △ β estimator 15 to estimate dc and perform feedback control to make it zero. By adding and subtracting 1 F PM, the basis is: PM dq inverse The calculation of the ferromagnetic flux of the wave-converter 272194. The internal axis of the motive force is the decisive example of qdc-qc and dc-0. The gain and load change direction can be accurately calculated by this patent 〇 値 △ Θ 6 and -10-200524265 (7) The difference between the command (zero) of the zero command generator 17 and ASdc is compensated at the angular frequency ω 1 * by the proportional compensator 18. As shown in the vector diagram in Figure 2, in the case where Δ 0 is positive, the dc-qc axis advances more than the d-q axis. Therefore, Δ 0 can be reduced by reducing ω 1 *. Conversely, when Δ 0 is negative, raise ω 1 * to make the d-q axis coincide with the d c-q c axis. With this control, the position sensor of the magnetic pole axis of the PM motor can be omitted, and the phase angle △ 0 dc inside the controller can be matched with the magnetic flux position 0 d in the actual PM motor. Sensor control. Next, the Δ Tm estimator 2 1 (periodic disturbance estimator) and the torque controller 22 which are characteristic parts of the present invention will be described in detail. Regarding each block of the figure, R is the winding resistance of the PM motor, L is the inductance of the PM motor, P is the number of poles of the PM motor, Ke is the generation constant of the PM motor (magnet flux), and the J is the PM motor and the load. The device's overall inertia's are the differential operators used in Laplace transforms. As shown in Fig. 4, the q-axis current I q is generated based on the relationship between the applied voltage Vq and the voltage interference VD applied to the pM motor, and the electrical constants R and L of the motor. I q is a component of the magnet magnetic flux (d-axis) orthogonal to the p Μ motor, and is multiplied by the power generation constant Ke to become the motor torque Tm. The rotation speed ω r of the PM motor is the difference between the integral motor torque Tm and the load torque TL. Here, the load torque Tl has various characteristics depending on the type or application of the load device. By multiplying ω r by the number of pole pairs (P / 2) ′, the electrical angular frequency ωΐ of the motor can be obtained, and its integral 値 becomes the position β d of the PM motor. The axis error Δ β is obtained as the difference from the phase β d c in the controller. -11-200524265 (8) Here, it is considered that the voltage disturbance VD or the load torque t1 contains a periodic component. Periodic voltage interference V D, for example, in the case of non-uniform magnet magnetic flux of a ρ μ motor 'case of magnetic permeability deviation' or case of phase deviation of coils' equivalence becomes periodic voltage interference and affects it. Or the interference caused by the short-circuit prevention period (no-load time) of the inverter arm is also generated at 6 times the drive frequency of the inverter. In addition, the "cyclic load torque disturbance" can be considered, for example, the load of a reciprocating compressor used in a freezer or an air conditioner, or a single rotary compressor. In the case of a reciprocating compressor, one of the motors is rotated for one cycle, and the load is greatly changed. In order to control these vibrations and noises in a controlled manner, it is sufficient to construct a control system in which the aforementioned periodic torque fluctuations become zero. In the conventional invention, the rotation speed information is detected by some means, and the applied voltage is controlled so that the rotation pulsation becomes zero to correspond. In compressors such as air conditioners, it is difficult to obtain the speed information directly. Therefore, the change in the potential of the neutral point of the motor is detected to obtain the information on the 60-degree scale of the electrical angle to estimate the speed. However, in this method, only 6 points of information can be obtained for the electrical angular period, which is not sufficient as speed information. In this state, a delay of 60 degrees is affected, or a problem occurs in the accuracy of speed detection. Or the pulsation caused by the distortion of the induced electromotive force voltage of the motor becomes a cycle shorter than the electrical angular cycle (mainly 1/6 cycle), so it is difficult to suppress it. In addition, it is also possible to consider driving the control theory and construct a method attached to Sensing Observation -12-200524265 (9) to estimate the pulsating torque, but the response frequency itself becomes a problem here. In response to the frequency of the pulsating torque, it is also necessary to improve the setting response of the observer. The higher the pulsatile cents becomes, the more responsive the observer is required, and local speed calculation processing is required. Therefore, as a conventional weekly control method, it is difficult to suppress vibration in the low-speed range, although it is difficult to suppress the vibration at the age. As an example, consider the use of a general-purpose microcomputer to form a shape. When the observer response time is set to lms (1000 rad / s- ', the detectable pulsating torque is about 30Hz. For a four-pole motor, it becomes 900 [r / min]. Most compressions occur in the most local rotation of compression Above 3000 [r / min], it will not be applicable unless it is below the right speed. In the present invention, focusing on the block diagram of Fig. 4, the method of Δ0 is used to estimate the torque ripple component ΔTm. The axis error can be calculated instantaneously, and it can be estimated without being affected by the calculation delay. In addition, for the driving frequency, for the high frequency (such as '6 times the vibration component), it can also be detected as its special' and the known cycle Compared with the sexual interference control method, a large estimation is possible. In the case where such periodic interference occurs, the difference in the motor reload torque TL becomes a cause of periodic torque fluctuations and noise. In terms of suppressing this vibration and noise, for example, countermeasures such as surrounding the entire device with a material become a larger size of the device. In the case of the observer E, the frequency of the moment I results in the need to suppress the interference of three factors. However, the situation of the local observer is about 150Hz) If it is set to the machine, it is about 30%: the axis error difference △ 0 is instantaneous, and the local precision component (example. This result is high speed The field moment Tm becomes negative and vibration, need to absorb sound, and cost -13-200524265 (10) increase is a problem that needs to be solved. In order to control vibration and noise in a controlled manner, if the composition makes the aforementioned periodic torque change The control system may be a zero control system. In the conventional invention, the rotation speed is detected by some means, and the applied voltage is controlled so that the rotation pulsation becomes zero to cope with it. However, in a compressor such as an air conditioner, an electric motor system is assembled It is difficult to simply obtain the speed information inside the compressor. In addition, even if it can be obtained, at most, only the information corresponding to the 60-degree scale of the electrical angle can be obtained. Therefore, it is difficult to achieve local accuracy. In the Ming Dynasty, focusing on the block diagram of Fig. 4, a method for estimating the torque ripple component △ T m from the shaft error △ < 9 was proposed. The shaft error △ 0 is instantaneous and can be calculated instantly, so it is not affected by the delay In addition, for high-precision estimation. In addition, the driving frequency and high frequency components (for example, 6 times the vibration component) can also be detected as their characteristics. Figure 5 shows the load torque TL at the angular frequency ω d. When the sine wave contains vibration components, the torque pulsation component (△ Tm), the rotation speed variation (△ ω r), and the shaft error (△ Θ). If the stable state is considered, ¾ of Tm and TL The average system is the same, ΔΤγπ is only the vibration component (Figure 5 (b)). The vibration component included in the rotation speed is the integral of ΔΤ !!!, compared with △ Tm, it has a waveform with a phase delay of 90 degrees. Vibration Although the magnitude itself changes depending on the inertia J, the phase can be considered to be almost 90 degrees delayed. The axis error Δ 0 is changed to the integral △ ω r 'to reverse the sign (the relationship defined by the figure, Invert the sign) Therefore, the phase changes 90 degrees forward (90 degrees delayed by integral, 5 tigers reversed, so 'when it becomes 90 degrees). That is, the change component of AT m, in -14- (11) (11) 200524265 △ 0 It is observed as a vibration waveform of the same phase. When the relationship is derived from the block diagram, it becomes as follows. Figure 6 (a) shows a block diagram from ΔΤγπ to △ 0. This block diagram is converted by inverse , We can find the transfer function from △ 0 to △ Tm, as shown in Figure (c). When ΔTim is obtained according to Figure 6 (c), the pulsation component of the torque can be directly estimated by △ edcCzXe 値. . However, the second order differential Δ 0dc is practically impossible. Originally, it was estimated 値, and most of them contained noise such as detection ，. Therefore, using differential will increase the estimation error. In addition, there are limits from the calculation cycle. Therefore, focusing on the point that "the perturbation component is a periodic function", substituting s = j d into the 6th (c) graph. In this way, as shown in Fig. 6 (d), it can be estimated that Δβ can be multiplied by a constant multiple to become ΔTm. This result is consistent with the relationship between the waveforms (b) and (d) in FIG. 5. The structure of Fig. 6 (d) is the Δ Tm estimator 2 1 (periodic disturbance estimator) shown in Fig. 7. △ Tm estimator 2 1 (periodic interference estimator) is formed by adding △ Θ dc to a proportional gain of 2J / P 21 1 and two multipliers 2 1 2 to perform the calculation of the sixth (d) diagram. . According to FIG. 7, the periodic interference component included in the angular velocity ω d of Δτιη can be estimated from Δ $ (! (:). Next, the torque controller 22 (FIG. 8) that suppresses this ΔTm will be described. The torque controller The necessary conditions are: (1) High followability for periodic interference components, -15-200524265 (12) (2) Two types of low sensitivity for components other than periodic interference. (1) In order to The most important condition for suppressing periodic interference. (2) It is a condition necessary for the △ β of DC to prevent the torque controller from affecting the overall control during the transition. As mentioned above, this compensator is based on " "Periodic interference" is the prerequisite. The equivalent conversion shown in Figure 6 is not suitable for DC interference. Therefore, for components other than periodic interference, the sensitivity needs to be reduced. The torque controller shown in Figure 8 Composed of: △ ΔTmc for △ Tm, zero command generator 17 which gives zero command, and sine wave transfer function 221 which holds peak 値 at angular frequency ad, and torque control gain 222. Angular frequency ud and period of sine wave transfer function 221 The frequency of variation of the torque of the disturbance is the same. In the case of a compressor, the frequency of the fluctuation of the periodic interference is the same as the driving frequency, so it is easy to make the ω d coincide with the pulsation frequency. In addition, for the periodic voltage interference, It is almost an integer multiple of the driving frequency, so the setting of ω d is relatively easy. In addition, 'this torque controller satisfies both of the conditions (1) and (2). The sine wave transfer function is at the angular frequency wd, The gain becomes infinite, and the deviation of only the yd component included in ΔΤιn becomes zero, and the frequency components other than it have no sensitivity. The details of the sine wave transfer function are shown in Japanese Patent Laid-Open No. 7- 2 0906 By using the torque controller 2 in FIG. 8, the sine of the angular frequency ω ^ -16- (13) 200524265 The wave-like signal IqS1N * is added to Iq *, and the torque ripple component can be suppressed △ Above T m 〇, although the first embodiment of the present invention will be described by taking a periodic interference load as an example, the same measures can be taken for the periodic voltage interference. In addition, the current detection Although the method uses the current 値 10 of the converter 4 to reproduce the motor current, it is also possible to directly detect the current of each phase with full CT or divided impedance. [Embodiment 2] Next, using FIG. Embodiment 2 of the invention. In Embodiment 1, the vibration frequency ω d of the torque ripple is introduced, and the gain becomes an infinite sinusoidal wave transfer function. As a result, although the d component included in the torque ripple is removed, Instead, the distortion of the driving current of the PM motor will become larger. In order to follow the load torque fluctuations instantaneously, the transient current fluctuations cannot be avoided. This will result in the degradation of the efficiency of the PM motor or an overcurrent based on the peak current. Reasons for the occurrence of such problems as escape. Therefore, it is more practical to adjust the overall reduction effect of vibration and noise, and conversely, to prevent the current distortion. Embodiment 2 is an embodiment in which the function of adjusting the suppression capability is added to the suppression effect of periodic interference. Fig. 9 shows the structure of the torque controller 22b of the second embodiment. By using the torque controller 22 instead of the torque controller 22 shown in FIG. 1, the embodiment is realized. The torque controller 2 2 B in Fig. 9 is composed of a sine wave transmission function 22 IB with peak suppression -17- (14) (14) 200524265, and a control gain 222B, and a zero command generator. 1 7 and 16 are added and subtracted. The sine wave transmission function 22 1B with a peak-suppression function has a Ta · s term in the denominator. With the size of T a, the number of peaks 函数 of the function can be changed. As a result, the interference suppression effect of the ω d component can be adjusted, and driving at the optimum point of the distortion of noise, vibration, and phase current of the PM motor becomes possible. [Embodiment 3] Next, Embodiment 3 according to the present invention will be described with reference to Fig. 10. In the first and second embodiments, as a torque controller, a transfer function is introduced in which the vibration frequency ω d of the torque ripple is maximized (in the first embodiment, it is infinite). In these controllers, finally, the convergent response of the pulsation component is determined by the magnitude of the control gain (control gain 222 or control gain 2 2 2 B) of the output section. However, the relationship between the addition of these settings and the actual response time is complicated, and it becomes necessary to make a Bode line graph for review, or to obtain it through simulation or actual measurement. Therefore, adjustment work is very labor-intensive. In the third embodiment, a torque controller for simplifying the adjustment work is provided. Fig. 10 shows a configuration example of the torque controller 22C of the third embodiment. By using the present torque controller 22C instead of the torque controller 22 of Fig. 1, the third embodiment is realized. — In the torque controller 22C in Figure 10, it is composed of: single-phase-dq coordinate converter 2 2 3, primary delay filter 2 2 4, integral controller 2 2 5, dq--18- 200524265 (15 A single-phase inverse converter 226, an integrator 20, an adder-subtractor 16, and a zero instruction generator 17 are configured. Next, the operation of the torque controller 2 2 C will be described. The output Δ Tmc of the Δ Tm estimator is decomposed into a SIN component and a COS component in the single-phase-dq coordinate converter 223. In addition, the conversion formula of the single-phase-dq coordinate converter 2 23 is as follows: [Mathematical formula 1] ATds cos (iy〆) ATqs -sin (^ 〇
依據（數學式1 ) ，△ τ m如含有6； d之頻率成分，則 因應其量，ATds、ATqs之平均値變成非零之値。此平均 値分別與包含在△ Tmc之COS成分、以及SIN成分一 致。但是，ATcis、ATqs中多量含有之2倍成分故， 需要以一次延遲濾波器2 2 4消除交流成分。此結果爲， △ Tds、ATqs變成包含在ΔΊΓηιε之脈動成分的COS成分 以及SIN成分。接著，令此各成分成爲零，由零指令產生 器1 7給予指令「零」訊號，以加減法器！ 6運算與指令之 偏差。依據這些之偏差，積分控制器2 2 5進行積分補償， 將脈動成分控制爲零。最後，將Ids、IqS之値逆轉換爲單 相訊號，輸出I q s in *。此逆轉換係依據下述式子所運算。 [數學式2] 「/ώ =[cos(〜〇 —S1_〆)]_ …（數學式2 ) -19- (16) 200524265 脈動成分ATmc在以（數學式1)被座標轉換後，成 爲直流量故，可以積分控制器22 5去掉偏差。即此轉矩控 制器如來外部來看，在角頻率ω d中，與增益變成無限大 之補償要素爲等效。即變成具有與實施例1之轉矩控制器 2 2同等之頻率特性。 在轉矩控制器2 2 C之情形，與第8圖或第9圖之轉矩 控制器相比，調整處所成爲一次延遲濾波器之時間常數 Tatr與積分控制器225之增益KiATR之2處。但是，TAtr 對於ω d可以選擇充分大之時間常數故，調整方法並不特 別難。另外，KiATR之値係直接變成決定脈動成分抑制之 響應時間，控制響應時間對於KiATR之値，成爲線性。此 結果爲，可以獲得增益設定變得容易之效果。 [實施例4] 接著，利用第1 1圖，說明依據本發明之實施例4。 在竇施例3中，提供對於振動頻率ω d，增益變成無 限大之轉矩控制器。此係動作上與實施例1之轉矩控制器 (第8圖）等效。因此，產生與實施例2中所記載者相同 之課題。即包含在轉矩脈動之ω d成分雖被去除，代之， PM電動機的驅動電流之畸變變大，容易產生PM電動機 之效率劣化，或者基於峰値電流之過電流跳脫等之不良。 因此，與實施例2相同，提出將角頻率ω d之增益由 #卩艮大變成有限之方法。 第1 1圖係顯示實施例4之轉矩控制器22D的構造。 -20- 200524265 (17) 藉由代替第1圖之轉矩控制器22而使用本轉矩控制器 2 2 D，實施例4得以實現。 第1 1圖之轉矩控制器22D與第1 0圖之轉矩控制器 22C之差異爲，積分控制器225被變更爲不完全積分控制 器225D之點。依據不完全積分器225D內之時間常數Ti 與增益KiATR，峰値受到抑制。此結果爲，變成可調整ω d 成分之干擾抑制效果，在噪音、振動與PM電動機相電流 之畸變的最佳點之驅動變成可能。 [實施例5] 接著，利用第1 2圖，說明依據本發明之實施例5。 在實施例1〜4中，敘述了依據軸誤差△ 0之推算 値，以推算、抑制週期性轉矩脈動成分之方法。主要之脈 動成分雖出現於Iqc或軸誤差推算値，但是，對於Idc也 會產生影響。 d軸電流雖然對於轉矩沒有貢獻，但是，依據轉矩脈 動，旋轉軸產生偏差，在d軸方向也產生基於脈動之電 流。實施例5便是利用此以進而降低轉矩脈動之實施例。 弟1 2圖中’控制:ggp 2 E係與實施例1之控制器2幾乎 相同。新追加進行d軸（d c軸）之電流控制之d軸電流控 制器IdACR ( 22C ) 。22C例如係導入與第！ 〇圖所示之轉 矩控制器22C完全相同者（增益KiATR需要調整），輸入 I d c以代替△ T m c ’將輸出加在I d *。在電壓指令運算器！ 2 中’將I d * *當成新的指令値，進行電壓指令之運算。 -21 - 200524265 (18) 藉由追加IdACR，可去除包含在Idc之脈動成分，結 果爲，可以降低轉矩脈動成分。 [實施例6] 接著，利用第1 3圖說明依據本發明之實施例6。 第 13圖中，零件號碼 1、2、3、5、7、41、42、43 分別與實施例1的號碼之零件爲相同。在本實施例中，係 做成將換流器3、電流檢測器7、二極體橋42 —體化於功 率模組，使其小型化之實施例。作爲旋轉數指令產生器1 係使用外部之微電腦，藉由通訊而送來速度指令。其他， 藉由在功率模組配線交流電源4 1、平滑電容器43、PM電 動機5，可以實現能過抑制週期性轉矩脈動之同步電動機 的控制裝置。 本發明之目的係在於，藉由降低系統之噪音、振動， 以減少防音、防振材料，可以實現裝置的小型化。在實施 例6中，藉由將控制器或換劉器予以模組化，具有可進一 步實現裝置整體之小型化的效果。 [實施例7] 接著，利用第1 4圖說明依據本發明之實施例7。 第14圖中，零件號碼2、3、6、7、42、46係分別與 實施例1 (第1圖）以及實施例6 (第1 4圖）之相同號碼 之零件爲相同。本實施例係利用組裝有控制器2、換流器 3、電流檢測器7、二極體橋4 2之功率模組，構成空調之 -22- (19) 200524265 室外機3 0。在空調等之壓縮機中，係於密閉狀態之壓縮機 內部組裝有Ρ Μ電動機，Ρ Μ電動機的旋轉數或磁鐵磁通 之位置等之檢測有困難。 但是，藉由組裝有依據本發明之控制裝置，可不檢測 Ρ Μ電動機之旋轉數或位置而降低壓縮機所產生之振動、 噪音。 第1 5圖係顯示啓動依據本發明之空調的壓縮機，令 旋轉速度改變時，噪音的變化及電流波形之變化的一例。 第1 5圖中，令旋轉速度由高速往低速改變故，整體之噪 音會降低。 壓縮機之旋轉數改變化後，噪音、振動雖還殘留，但 是’噪音在數秒至數十秒以內降低。此時，在噪音降低之 前後，電流之畸變波形改變。此係壓縮機之旋轉數改變， 週期性干擾之發生條件改變故，而產生過渡現象。對於此 過渡現象，控制器逐漸加以反應，造後令噪音降低故，電 流之波形（畸變）也改變。 在本發明之情形，瞬間依據瞬間之軸誤差運算，週期 性干擾之抑制變成可能故，即使在旋轉速度高之情形，此 種現象也可觀測到。在射驅動壓縮機之頻率的最高頻率爲 100%之情形，在超過最高頻率之30%的範圍中，也可以低 噪音化、低振動化。 另外，如第1 5圖之電流波形般，不單比驅動頻率低 之頻率的干擾，對於比驅動頻率高之頻率成分，也可以獲 得同樣的效果。 -23- (20) (20)200524265 另外’實施例雖以空調爲例做說明，但是，在其他之 電器，例如套裝空調或冷凍庫等之情形，也可以獲得同樣 的效果。 如前述般’如依據本發明，不使用檢測同步電動機之 旋轉速度或旋轉軸位置之感測器，可以實現抑制負載裝置 或電動機本身所產生之週期性轉矩干擾之高性能的電動機 驅動。另外，即使在有檢測同步電動機之旋轉速度或旋轉 軸位置之感測器的情形，也可以同樣地加以實現。 【圖式簡單說明】 第1圖係顯示依據本發明之同步電動機控制裝置之實 施例1的系統構造方塊圖。 第2圖係顯示依據本發明之同步電動機控制裝置之實 施例的軸誤差△ 0之定義向量圖。 第3圖係顯示依據本發明之同步電動機控制裝置之實 施例1的軸誤差推算器之內部構造方塊圖。 第4圖係說明依據本發明之交流電動機控制裝置之實 施例1之由對於電動機之施加電壓至軸誤差發生之原理的 方塊圖。 第5圖係顯示依據本發明之交流電動機控制裝置之實 施例1的週期性干擾轉矩，以及其所引起產生之旋轉脈 動、軸誤差變動的原理之波形圖。 第6圖係說明依據本發明之交流電動機控制裝置之實 施例1的脈動轉矩成分之推算原理方塊圖。 -24- (21) (21)200524265 第7圖係顯示依據本發明之同步電動機控制裝置之實 施例1的△ T m推算器之內部構造方塊圖。 第8圖係顯示依據本發明之同步電動機控制裝置之實 施例1的轉矩控制器之內部構造方塊圖。 第9圖係顯示依據本發明之同步電動機控制裝置之實 施例2之轉矩控制器的內部構造方塊圖。 第1 〇圖係顯示依據本發明之同步電動機控制裝置之 實施例3之轉矩控制器的內部構造方塊圖。 第1 1圖係顯示依據本發明之同步電動機控制裝置之 實施例4之轉矩控制器的內部構造方塊圖。 第1 2圖係顯示依據本發明之同步電動機控制裝置之 實施例5的控制器之內部構造方塊圖。 第1 3圖係顯示適用依據本發明之同步電動機控制裝 置之實施例6的外觀構造圖。 第1 4圖係顯示將依據本發明之同步電動機控制裝置 適用於空調之實施例7之外觀構造圖。 第1 5圖係顯示在啓動依據本發明之空調的壓縮機， 令旋轉速度改變之情形，噪音的變化及電流波形之變化的 一例圖。 [主要元件符號說明】 1 :旋轉數指令產生器’ 2 :控制器，3 :換流器，4 ·· 轉換器，5 : P Μ電動機，6 :壓縮機，7 :電流檢測器， 8 :電流再現器，9 : dq座標轉換器，: iq*產生器， -25- 200524265 (22) 11: Id*產生器，12:電壓指令運算器，13: dq逆轉換 器，1 4 : P WM脈波產生器，1 5 : 0推算器，1 6 :加減 法器，1 7 :零指令產生器，1 8 :比例補償器，1 9 :轉換增 益，2 0 :積分器，2 1 : △ Tm推算器，2 2 :轉矩控制器， 41 :交流電源，42 :二極體橋，43 :平滑電容器.According to (Mathematical Formula 1), if △ τ m contains a frequency component of 6; d, according to its amount, the average 値 of ATds and ATqs becomes a non-zero 値. This average 値 corresponds to the COS component and the SIN component contained in ΔTmc, respectively. However, since ATcis and ATqs contain twice as many components, it is necessary to eliminate the AC components with the primary delay filter 2 2 4. As a result, ΔTds and ATqs become the COS component and the SIN component included in the pulsating component of ΔΊΓηιε. Then, make each component zero, and the zero command generator 17 gives a command "zero" signal to add and subtract! 6 Deviation between operation and instruction. Based on these deviations, the integral controller 2 2 5 performs integral compensation to control the pulsation component to zero. Finally, the inverse of Ids and IqS is converted into a single-phase signal, and I q s in * is output. This inverse conversion is calculated according to the following formula. [Mathematical formula 2] "/ FREE = [cos (~ 〇—S1_〆)] _… (Mathematical formula 2) -19- (16) 200524265 The pulsation component ATmc is transformed by the coordinates in (Mathematical formula 1) and becomes Therefore, it is possible to remove the deviation by integrating the controller 22 5. That is, the torque controller is equivalent to the compensation element at which the gain becomes infinite at the angular frequency ω d. That is to say, the torque controller has the same value as in Example 1. The torque controller 2 2 has the same frequency characteristics. In the case of the torque controller 2 2 C, compared with the torque controller of FIG. 8 or FIG. The gain KiATR of the integral controller 225 is 2. However, TAtr can choose a sufficiently large time constant for ω d, so the adjustment method is not particularly difficult. In addition, the system of KiATR directly becomes the response time that determines the suppression of the pulsation component. The control response time is linear with respect to KiATR. As a result, the effect of making gain setting easy can be obtained. [Embodiment 4] Next, Embodiment 4 according to the present invention will be described using FIG. 11. In Example 3, for the vibration frequency ω d, the gain becomes an infinite torque controller. This operation is equivalent to the torque controller of the first embodiment (Fig. 8). Therefore, the same problems as those described in the second embodiment arise. Although the ω d component of the torque ripple is removed, instead, the distortion of the drive current of the PM motor becomes large, and the efficiency of the PM motor is likely to be deteriorated, or defects such as overcurrent tripping based on the peak current are caused. The second embodiment is the same, and a method for changing the gain of the angular frequency ω d from # 卩 GEN to a limit is proposed. FIG. 11 shows the structure of the torque controller 22D of the fourth embodiment. -20- 200524265 (17) The present torque controller 2 2 D is used instead of the torque controller 22 of FIG. 1, and the fourth embodiment is realized. The difference between the torque controller 22D of FIG. 11 and the torque controller 22C of FIG. 10. For this reason, the integration controller 225 is changed to the point of the incomplete integration controller 225D. According to the time constant Ti and the gain KiATR in the incomplete integrator 225D, the peak chirp is suppressed. As a result, the interference becomes an adjustable ω d component Suppression effect, in the phase of noise and vibration with PM motor Driving of the optimal point of the distortion of the current becomes possible. [Embodiment 5] Next, Embodiment 5 according to the present invention will be described with reference to Fig. 12. In Embodiments 1 to 4, the axis error △ 0 will be described. Calculate 値 to estimate and suppress the periodic torque ripple component. Although the main pulsating component appears in Iqc or shaft error estimation 値, it also affects Idc. Although the d-axis current does not contribute to the torque, it does According to the torque pulsation, the rotation axis is deviated, and a pulsation-based current is also generated in the d-axis direction. Embodiment 5 is an embodiment using this to further reduce torque ripple. The control in the figure 12: ggp 2 E is almost the same as the controller 2 in the first embodiment. Newly added d-axis current controller IdACR (22C) for d-axis (d-axis) current control. 22C is for example the introduction and the first! 〇 The torque controller 22C shown in the figure is completely the same (the gain KiATR needs to be adjusted). Input I d c instead of △ T m c ′ and add the output to I d *. In voltage command calculator! In 2 ′, I d * * is regarded as a new instruction 値, and a voltage instruction operation is performed. -21-200524265 (18) By adding IdACR, the pulsation component included in Idc can be removed. As a result, the torque pulsation component can be reduced. [Embodiment 6] Next, Embodiment 6 according to the present invention will be described with reference to Figs. In Fig. 13, the part numbers 1, 2, 3, 5, 7, 41, 42, 43 are the same as the parts with the numbers of the first embodiment, respectively. In this embodiment, it is an embodiment in which the inverter 3, the current detector 7, and the diode bridge 42 are integrated into a power module to make it compact. As the rotation number command generator 1, an external microcomputer is used to send a speed command through communication. In addition, by wiring the AC power source 41, the smoothing capacitor 43, and the PM motor 5 to the power module, a control device for a synchronous motor capable of suppressing periodic torque ripple can be realized. The object of the present invention is to reduce the noise and vibration of the system to reduce the soundproof and vibration-proof materials, so that the device can be miniaturized. In the sixth embodiment, by modularizing the controller or the changer, there is an effect that the entire device can be further miniaturized. [Embodiment 7] Next, Embodiment 7 according to the present invention will be described using FIG. 14. In Fig. 14, the part numbers 2, 3, 6, 7, 42, and 46 are the same parts with the same numbers as those in Embodiment 1 (Fig. 1) and Embodiment 6 (Fig. 14), respectively. In this embodiment, a power module assembled with a controller 2, a converter 3, a current detector 7, and a diode bridge 42 is used to form an air conditioner -22- (19) 200524265 outdoor unit 30. In compressors for air conditioners, etc., the compressor in the closed state is equipped with a PM motor, and it is difficult to detect the number of rotations of the PM motor or the position of the magnetic flux of the magnet. However, by incorporating the control device according to the present invention, it is possible to reduce the vibration and noise generated by the compressor without detecting the rotation number or position of the PM motor. Fig. 15 shows an example of changes in noise and current waveforms when the compressor of the air conditioner according to the present invention is started and the rotation speed is changed. In Fig. 15, the rotation speed is changed from high speed to low speed, so the overall noise will be reduced. Although the number of rotations of the compressor is changed, noise and vibration remain, but the noise is reduced within seconds to tens of seconds. At this time, the distortion waveform of the current changes before and after the noise is reduced. This is because the rotation number of the compressor is changed, and the conditions for the occurrence of periodic interference are changed, resulting in a transient phenomenon. The controller gradually responds to this transient phenomenon, and the noise is reduced after it is made, so the current waveform (distortion) also changes. In the case of the present invention, the instantaneous calculation based on the instantaneous axis error makes it possible to suppress periodic interference. This phenomenon can be observed even in the case of high rotation speed. When the highest frequency of the radio-driven compressor is 100%, the noise and vibration can be reduced in a range exceeding 30% of the highest frequency. In addition, like the current waveform in Fig. 15, not only the interference at a frequency lower than the driving frequency, but also the frequency component higher than the driving frequency, the same effect can be obtained. -23- (20) (20) 200524265 In addition, although the embodiment is described using an air conditioner as an example, the same effect can be obtained in the case of other appliances such as a set of air conditioners or a freezer. As described above, according to the present invention, a high-performance motor drive capable of suppressing the periodic torque interference generated by the load device or the motor itself can be realized without using a sensor that detects the rotation speed or the position of the rotating shaft of the synchronous motor. It is also possible to implement the same in the case where there is a sensor that detects the rotation speed or the position of the rotating shaft of the synchronous motor. [Brief description of the drawings] Fig. 1 is a block diagram showing a system configuration of the first embodiment of the synchronous motor control device according to the present invention. Fig. 2 is a vector diagram showing the definition of the axis error Δ 0 of the embodiment of the synchronous motor control device according to the present invention. Fig. 3 is a block diagram showing the internal structure of a shaft error estimator of the first embodiment of the synchronous motor control device according to the present invention. Fig. 4 is a block diagram illustrating the principle of occurrence of an error from a voltage applied to a motor to a shaft according to Embodiment 1 of an AC motor control device according to the present invention. Fig. 5 is a waveform diagram showing the principle of the periodic disturbance torque of the AC motor control device according to the first embodiment of the present invention, and the principle of the variation of the rotation pulse and the shaft error caused by it. Fig. 6 is a block diagram illustrating an estimation principle of a pulsating torque component of the first embodiment of the AC motor control device according to the present invention. -24- (21) (21) 200524265 Fig. 7 is a block diagram showing the internal structure of the Δ T m estimator of the first embodiment of the synchronous motor control device according to the present invention. Fig. 8 is a block diagram showing the internal structure of a torque controller according to the first embodiment of the synchronous motor control device according to the present invention. Fig. 9 is a block diagram showing the internal structure of a torque controller of the second embodiment of the synchronous motor control device according to the present invention. Fig. 10 is a block diagram showing the internal structure of a torque controller according to the third embodiment of the synchronous motor control device according to the present invention. Fig. 11 is a block diagram showing the internal structure of a torque controller according to a fourth embodiment of a synchronous motor control device according to the present invention. Fig. 12 is a block diagram showing the internal structure of a controller according to a fifth embodiment of a synchronous motor control device according to the present invention. Fig. 13 is a diagram showing the appearance and construction of a sixth embodiment to which a synchronous motor control device according to the present invention is applied. Fig. 14 is a diagram showing the external appearance of a seventh embodiment in which a synchronous motor control device according to the present invention is applied to an air conditioner. Fig. 15 is a diagram showing an example of changes in noise and changes in current waveforms when the rotation speed is changed when the compressor of the air conditioner according to the present invention is started. [Description of main component symbols] 1: Rotation number command generator '2: Controller, 3: Inverter, 4 ... Converter, 5: PM motor, 6: Compressor, 7: Current detector, 8: Current reproducer, 9: dq coordinate converter ,: iq * generator, -25- 200524265 (22) 11: Id * generator, 12: voltage command calculator, 13: dq inverse converter, 1 4: P WM Pulse generator, 15: 0 estimator, 16: addition and subtraction, 17: zero command generator, 18: proportional compensator, 19: conversion gain, 2 0: integrator, 2 1: △ Tm estimator, 22: torque controller, 41: AC power, 42: diode bridge, 43: smoothing capacitor.
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