CN100338868C - Controller for synchromotor, electric equipment and module - Google Patents

Abstract

The present invention provides a control device for a synchronous motor. In a drive device without a device for directly detecting the rotor position and rotational speed of an AC synchronous motor, when a periodic disturbance occurs inside the drive device or a load device, the periodic disturbance is suppressed, Low vibration and low noise can be realized. It can be realized by setting the following device: By calculating the difference between the position of the magnetic flux axis of the AC synchronous motor and the position of the magnetic flux axis assumed in the controller (axis error), and correcting the rotational speed to make it zero, to realize sensorless drive , and, based on the calculated value of the shaft error, a device for extracting a pulsating component of a torque generated by a motor or a load device; and a device for compensating it.

Description

Synchronous motor control unit and system power module

technical field

The invention relates to a synchronous motor control device, electrical equipment and system power module.

Background technique

Various methods have been disclosed so far for the control method of the speed or position sensor not using the AC motor. For example, a method disclosed in JP-A-2001-251889 or the like is known for a permanent-magnet synchronous motor, which is a representative example of an AC motor. This control method does not use a position sensor, but estimates the magnetic pole position inside the controller.

In addition, there are JP-A-10-174488, JP-A-2002-34290, and the like as methods for controlling periodic torque disturbances generated in a load device of an electric motor. In the method described in Japanese Unexamined Patent Publication No. 10-174488, the pulsating portion included in the speed detection value of the motor is extracted, and correction is applied to the inverter output voltage to eliminate the pulsating portion. Implementation of this approach requires velocity information.

The system disclosed in JP-A-2002-34290 detects the pulsating portion included in the torque current component and applies correction to the rotational speed to stably control the motor.

[Patent Document 1] JP-A-2001-251889

[Patent Document 2] JP-A-10-174488

[Patent Document 3] JP-A-2002-34290

In the method of JP-A-2001-251889, position sensorless can be realized, but when a load accompanied by periodic disturbance, such as a compressor, is connected to the load device, the periodic disturbance cannot be suppressed. As a result, there is a problem that rotational pulsation occurs, which becomes a source of vibration and noise of the device.

In the system disclosed in JP-A-10-174488, periodic disturbances can be suppressed, but information on the rotational speed of the motor is required. Therefore, some speed detector is required. In principle, it is possible to install a position sensor such as a Hall IC to detect the speed of a motor, but when the load device is a compressor such as an air conditioner, it is difficult to install the sensor due to the ambient environment conditions.

Instead of a position sensor, there is also known a method of detecting the neutral point potential of the motor and obtaining speed information from its variable components, but the speed information can only be obtained every 60 degrees from the electrical angle, and high-speed and high-precision speed detection is difficult. In particular, periodic disturbances due to the influence of the conduction delay (delay time period) of the drive motor inverter fluctuate at six times the frequency of the drive frequency of the motor, so it is impossible to suppress the speed detection at intervals of 60 degrees in electrical angle. The disturbance. In addition, there is a problem that a spare wiring is required for obtaining the neutral point potential.

The method disclosed in JP-A-2002-34290 is to improve the overall stability of the control device by varying the rotational speed itself based on the ripples contained in the torque current. Therefore, if the rotational pulsation is further increased, the problems of vibration and noise cannot be solved. In addition, since the target is an induction motor, it has been difficult to apply it to a synchronous motor.

Contents of the invention

An object of the present invention is to provide a motor control device capable of suppressing vibration and noise caused by periodic disturbances.

One of the technical solutions of the present invention provides a control device for a synchronous motor, which has a controller for controlling the synchronous motor accompanying the load through the output voltage of the converter. The amount proportional to the deviation of the rotor position, that is, the Δθ estimator for estimation and calculation of the shaft error; and the periodic disturbance component that occurs on either or both of the motor or the load based on the output value of the Δθ estimator, that is, the estimated value of the shaft error Periodic disturbance estimator for .

The second technical solution of the present invention provides a refrigerator in which the control device for a synchronous motor described in the first technical solution is used to drive the compressor.

The third technical solution of the present invention provides an air conditioner, which uses the synchronous motor control device described in the first technical solution to drive the compressor.

The fourth technical solution of the present invention provides a system power module, which can be controlled by a converter circuit part that applies voltage to a synchronous motor connected to a load; a converter circuit part that provides power to the converter circuit part; The controller for the voltage of the synchronous motor is regarded as a single component as a whole. The controller has the function of estimating and calculating the shaft error, which is proportional to the deviation between the rotor position of the synchronous motor and the rotor position assumed in the control. The Δθ estimator; and the periodic disturbance estimator for solving the periodic disturbance component of either or both of the motor or the load based on the output value of the Δθ estimator, that is, the shaft error estimation value.

According to the present invention, a motor control device capable of suppressing vibration and noise caused by periodic disturbances can be realized.

Description of drawings

Fig. 1 is a block diagram showing the system configuration of Embodiment 1 of the synchronous motor control device of the present invention.

2 is a vector diagram showing the definition of an axis error Δθ in Embodiment 1 of the synchronous motor control device of the present invention.

3 is a block diagram showing the internal configuration of a shaft error estimator in Embodiment 1 of the synchronous motor control device of the present invention.

Fig. 4 is a block diagram illustrating the principles from the voltage application to the motor to the occurrence of an axis error in the first embodiment of the AC motor control device of the present invention.

5 is a waveform diagram showing the principle of periodic disturbance torque, rotation pulsation and shaft error variation caused by it in Embodiment 1 of the AC motor control device of the present invention.

6 is a block diagram illustrating the principle of estimating ripple torque components in the first embodiment of the AC motor control device of the present invention.

Fig. 7 is a block diagram showing the internal configuration of a ΔT _m estimator in Embodiment 1 of the synchronous motor control device of the present invention.

8 is a block diagram showing the internal configuration of a torque controller in Embodiment 1 of the synchronous motor control device of the present invention.

9 is a block diagram showing the internal configuration of a torque controller in Embodiment 2 of the synchronous motor control device of the present invention.

Fig. 10 is a block diagram showing the internal configuration of a torque controller in Embodiment 3 of the synchronous motor control device of the present invention.

Fig. 11 is a block diagram showing the internal configuration of a torque controller in Embodiment 4 of the synchronous motor control device of the present invention.

Fig. 12 is a block diagram showing the internal configuration of a torque controller in Embodiment 5 of the synchronous motor control device of the present invention.

Fig. 13 is a configuration diagram showing the appearance of Embodiment 6 of the synchronous motor control device to which the present invention is applied.

Fig. 14 is a configuration diagram showing an appearance of a seventh embodiment of a synchronous motor control device according to the present invention which is suitable for an air conditioner.

Fig. 15 is a graph showing an example of a change in noise and a change in a current waveform when the air conditioner compressor of the present invention is started and the rotational speed is changed.

Detailed ways

Next, an embodiment of a control device for an AC motor according to the present invention will be described with reference to FIGS. 1 to 15 . In addition, in the following embodiments, a permanent magnet synchronous motor (hereinafter, simply referred to as a PM motor) is used as the motor for description, but other synchronous motors (eg, coil type synchronous motors, reluctance motors, etc.) can be similarly implemented.

[Example 1]

Fig. 1 is a block diagram illustrating the system configuration of Embodiment 1 of an AC motor control device according to the present invention. The control device of the present embodiment 1 is composed of the following parts: the speed command generator 1 that gives the speed command ω _r ^* to the motor by the instruction 100 of the upper control device; calculates the AC applied voltage of the motor, and converts it into a pulse width modulation signal (PWM signal ) output controller 2; a converter (inverter) 3 driven by the PWM signal; a converter (converter) 4 that supplies power to the converter 3; a PM motor 5 as a control object; a compressor as a load of the PM motor 6; a current detector 7 for detecting the current I0 supplied by the converter 4 to the converter 3 .

The controller 2 is composed of the following parts: According to the current I0 detected by the current detector 7, the current reproducer that reproduces the three-phase alternating current I _u , I _v , and I _w flowing on the PM motor 5 through calculation inside the controller 8; Transform the reproduced three-phase AC currents I _uc , I _vc , and I _wc into components on each axis of d and q by using the phase angle θ _dc (the position of the magnet flux of the PM motor assumed inside the controller) dq coordinate converter 9 for I _dc and I _qc ; I q ^* generator 10 for giving command I _q ^* to the current component on the ^{q-axis; similarly, I d *} _{for giving the command I d * to} ^the current component on the d-axis generator 11; voltage command calculator 12 for calculating voltage commands V _dc ^{* and} V qc ^* according to I _d ^* , ^I _q ^* and electrical angular frequency command ω ₁ *; transforming V _dc ^* and _{V qc *} ^into three-phase AC A dq inverse converter 13 for voltage commands v _u ^* , v _v ^* , v _w ^* ; a PWM pulse generator for generating a pulse width modulation signal (PWM signal) for switching the converter 3 according to the three-phase AC voltage command 14. The Δθ estimator 15 for estimating and calculating Δθ corresponding to the angle (axis error) of the error between the magnet flux position θ _d of the PM motor 5 and the position θ _dc assumed inside the controller 2; performing addition and subtraction Addition and subtraction device 16; Zero command generator 17 that gives commands to the shaft error estimated value θ _dc ; In order to control Δ θ at zero, a proportional compensator 18 that increases compensation on the electrical angular frequency command ω _I ^* ; uses PM motor The number of poles P transforms the rotational speed command ω _r ^* into the electric angular frequency command ω ₁ ^* of the motor with a conversion gain of 19; integrates the electric angular frequency, and calculates the integrator 20 of the magnetic flux position θ _dc of the magnet; estimates the value Δθ _dc based on the shaft error, ΔT _m estimator 21 (periodic disturbance estimator) for estimating and calculating ΔT _m of ^the periodic disturbance torque component; based on the estimated value ΔT _mc of ΔT _m , a torque controller ( _ATR )twenty two.

The converter 4 that supplies power to the converter 3 is composed of an AC power source 41 , a diode bridge 42 that rectifies the AC, and a smoothing capacitor 43 that suppresses ripple components included in the DC voltage.

Next, the operating principle of the first embodiment will be described with reference to FIG. 1 . The conversion gain 19 calculates and outputs the electrical angular frequency ω ₁ ^* of the PM motor based on the rotation command ω _r ^* from the rotational speed command generator 1 . Furthermore, ω ₁ ^* is integrated by the integrator 20 to calculate the AC phase θ _dc . In the current reproducer 8, based on the power supply current I0 detected by the current detector 7, the three-phase AC current of the PM motor is reproduced by calculation by the method described in JP-A-8-19263 or the like. Next, in the dq coordinate converter 9, the reproduced AC currents I _uc , I _vc , and I _wc are converted by θ _dc into current components I _dc on the rotating coordinate axis (dq axis) rotating at the angular frequency ω ₁ ^* , I _qc . I _qc is processed in the I _q ^* generator 10 and becomes the current command I _q ^* on the q axis. Also, the I _d ^* generator 11 generates a current command I _d ^* on the d-axis (in a PM motor with a non-salient rotor, usually I _d ^* = 0). The voltage command calculator 12 calculates the applied voltages V _dc * ^and V _qc ^* to the PM motor based on these commands (I _d ^* , I _q ^* ) and the angular frequency command ω ₁ ^* . V _dc ^* and V _qc ^* are converted into AC quantities again by dq inverse converter 13 , and then converted into pulse width modulation signals in PWM pulse generator 14 and sent to converter 3 . These basic operations are the same as those described in JP-A-2002-272194.

In the Δθ estimator 15, an error Δθ between the magnet magnetic flux position θ _d in the PM motor and the position θ _dc in the controller is estimated. Δθ is defined using the vector diagram illustrated in Figure 2. Let the position of the actual magnetic flux Φ of the magnet inside the PM motor be the d-axis, and let the axis orthogonal to it be the q-axis. On the other hand, the dq axes assumed in the controller are defined as dc-qc, and the deviation between them corresponds to the axis error Δθ.

If Δθ is obtained, by correcting it, the dq axis and the dc-qc axis can be aligned, and sensorless control of the PM motor can be realized. The estimated calculation of Δθ can be performed by multiplying the difference between I _q ^* and I _qc by the proportional gain K0 as shown in FIG. 3 , and setting it as the estimated value Δθ _dc of Δθ. Since I _qc fluctuates due to variations in θ _d and θ _dc due to load fluctuations, etc., Δθ can be inversely estimated from the fluctuation of I _qc . However, with the configuration shown in FIG. 3, it is difficult to obtain Δθ with high precision. In order to improve the accuracy, it may be calculated according to the formula (3) in JP-A-2002-272194, for example.

Feedback control is performed based on the shaft error estimated value Δθ _dc calculated by the Δθ estimator 15 so that it becomes zero. The difference between the command (zero) of the zero command generator 17 and Δθ _dc is obtained by the adder-subtractor 16, and compensation is applied to the angular frequency ω ₁ ^* via the proportional compensator 18. As shown in the vector diagram of Figure 2, when Δθ is positive, the dc-qc axis is ahead of the dq axis, so by reducing ω ₁ ^* , Δθ can be reduced; on the contrary, when Δθ is negative, by increasing ω ₁ ^* , Make the dq axis coincide with the dc-qc axis. By controlling in this way, the phase angle _θdc inside the controller can be matched with the actual magnet flux position _θd in the PM motor without using a position sensor of the magnetic pole axis of the M motor, and position sensorless control can be realized.

Next, the ΔT _m estimator 21 (periodic disturbance estimator) and the torque controller 22, which are characteristic parts of the present invention, will be described in detail. First, the principle of torque generation of the PM motor and the relationship between the shaft error Δθ will be briefly described with reference to FIGS. 4 and 5 .

Fig. 4 is a block diagram showing the principle from the voltage applied to the PM motor to the occurrence of the shaft error Δθ. In each block of the figure, R is the coil resistance of the PM motor, L is the inductance of the PM motor, P is the number of poles of the PM motor, K _e is the power generation constant (magnet flux) of the PM motor, and J is the load of the PM motor The total inertia of the device, s is the differentiation operator used in the Laplace transform.

As shown in FIG. 4 , the q-axis current Iq is generated from the relationship between the voltage _Vq applied to the PM motor, the voltage disturbance VD, and the electrical constants R, L of the motor. I _q is a component orthogonal to the magnet flux (d-axis) of the PM motor, and is multiplied by the power generation constant _Ke to obtain the motor torque T _m . The rotational speed ω _r of the PM motor is a value integrating the difference between the motor torque T _m and the load torque T _L . Here, the load torque T _L has various characteristics depending on the type and use of the load device. Multiply the number of pole pairs (P/2) on ω _r to obtain the electrical angular frequency ω ₁ of the motor, and the integral value becomes the position θ _d of the PM motor. The axis error Δθ is obtained as the difference from the phase θ _dc in the controller.

Here, it is considered that a periodic component is included in the voltage disturbance VD or the load torque _TL .

As a periodic voltage disturbance VD, for example, when the magnet flux of the PM motor is uneven, the magnetization is discrete, or the coil phase is discrete, it is equivalently affected as a periodic voltage disturbance. Alternatively, a disturbance or the like due to the influence of the arm short-circuit prevention period (delay time) in the inverter also occurs at a frequency six times the drive frequency of the inverter.

In addition, as periodic load torque disturbances, for example, reciprocating compressors used in refrigerators, air conditioners, and the like, loads of single rotary compressors, and the like are considered. In the case of a reciprocating compressor, one rotation of the motor is regarded as one cycle, and the load fluctuates drastically.

In order to suppress vibration and noise in a controlled manner, it is only necessary to configure the control system so that the above-mentioned periodic torque variation becomes zero. The countermeasures in the conventional inventions are to detect rotational speed information with some means, and to control the applied voltage so that the rotational pulsation becomes zero. In compressors such as air conditioners, it is difficult to obtain speed information directly, so the change in the neutral point potential of the motor is detected, information is obtained at 60 degrees in electrical angle, and the speed is estimated.

However, in this method, only 6 points of information can be obtained with respect to the electrical angle cycle, which is insufficient as speed information. In this state, there are problems in the influence of the 60-degree delay, and the speed detection accuracy. Or, since the pulsation generated by the distortion of the induced voltage of the motor is shorter than the electrical angular period (mainly 1/6 period), it is difficult to suppress it.

In addition, a method of estimating and calculating the ripple torque by constructing a load disturbance observer using control theory is also conceivable, but in this case, the response frequency of the observer itself becomes a problem. When the frequency of the pulsating torque is high, corresponding to this, the setting response of the observer also needs to be improved. The higher the frequency component of the pulsating torque, the higher the responsiveness of the observer is required, and as a result, high-speed arithmetic processing is required. Therefore, as a conventional periodic disturbance suppression method, vibration suppression can generally be achieved in a low-speed range, but suppression during high-speed rotation is difficult.

As an example, consider the case of using a general-purpose microcomputer to form an observer. When the response time of the observer is set to 1ms (1,000rad/s→150Hz), the ripple torque that can be detected is about 30Hz. If it is set as a 4-pole motor, it will be 900[r/min]. In the case of a compressor, since the maximum rotation speed is above 3,000 [r/min], it cannot be applied at a speed below about 30%.

In the present invention, focusing on the block diagram in FIG. 4 , a method of estimating the torque ripple component ΔT _m from the axis error Δθ is proposed. Since the axial error Δθ can be calculated as an instantaneous value, it is possible to perform high-precision estimation calculation without being affected by calculation delay. In addition, it has a feature that it can also detect frequency components that are higher than the driving frequency (for example, vibration components that are 6 times higher). As a result, compared with the conventional periodic disturbance suppression method, it is possible to estimate and calculate in the high-speed region to a large extent.

When such a periodic disturbance occurs, the difference between the motor torque T _m and the load torque T _L becomes a periodic torque fluctuation, which causes vibration and noise. In order to suppress this vibration and noise, it is necessary to take countermeasures such as surrounding the entire device with a sound-absorbing material, and there are problems of increasing the size of the device and increasing the cost.

In order to suppress vibration and noise in a controlled manner, it is only necessary to configure the control system so that the above-mentioned periodic torque variation becomes zero. The countermeasures in the conventional inventions are to detect rotational speed information with some means, and to control the applied voltage so that the rotational pulsation becomes zero. However, in compressors such as air conditioners, since a motor is built into the compressor, it is difficult to obtain speed information easily, and even if obtained, at best, information corresponding to 60 degrees of electrical angle can be obtained. Therefore, it is difficult to increase the accuracy.

In the present invention, focusing on the block diagram in FIG. 4 , a method of estimating the torque ripple component ΔT _m from the axis error Δθ is proposed. Since the axial error Δθ can be calculated instantaneously, high-speed and high-precision estimation calculation is possible. In addition, it has a feature that it can also detect frequency components that are higher than the driving frequency (for example, vibration components that are 6 times higher).

Fig. 5 illustrates the torque ripple component (ΔT _m ), rotational speed variation (Δω _r ), and shaft error (Δθ) when the load torque T _L includes a component vibrating sinusoidally at an angular frequency ω _d . If the steady state is considered, the average values of T _m and _TL are the same, and ΔT _m is only a vibration component (Fig. 5(b)). The vibration component Δω _r included in the transfer is a component integrated with this ΔT _m , and has a waveform whose phase is delayed by 90 degrees compared with ΔT _m . The magnitude of the vibration itself changes depending on the inertia J, but it can also be considered that the phase is delayed by approximately 90 degrees. Axis error Δθ leads by 90 degrees in phase because further integration of Δω _r reverses the sign (in the defined relationship shown in Figure 2), so the phase leads by 90 degrees (after being delayed by 90 degrees in the integration, it leads by 90 degrees because the sign is reversed ). That is, the fluctuation component of ΔT _m is observed as a vibration waveform of the same phase in Δθ. If this relationship is derived from a box-and-wire diagram it is as follows.

Fig. 6(a) shows a block diagram from ΔT _m to Δθ. By inversely transforming this block diagram, the transfer function from Δθ to ΔT _m can be found, as shown in (c) of the same figure.

If ΔT _m is obtained from FIG. 6( c ), the ripple component of the torque can be directly estimated from Δθ _dc (estimated value of Δθ). However, two-stage differentiation Δθ _dc is not possible in the present. Δθ _dc is originally an estimated value, and since it often includes disturbances in detected values, the use of differentiation will increase the estimation error, and there is also a limitation in the calculation cycle.

Therefore, paying attention to the point that "the disturbance component is a periodic function", bring s= _jωd into Fig. 6(c). Then, as shown in FIG. 6( d ), the result of Δθ being amplified by a constant factor can be estimated as ΔT _m . As a result, the relationship between the waveforms of (b) and (d) in FIG. 5 agrees.

The configuration of Fig. 6(d) is the ΔT _m estimator 21 (periodic disturbance estimator) shown in Fig. 7 . The ΔT _m estimator 21 (periodic disturbance) consists of a proportional gain 211 that amplifies 3Δθ _dc by 2J/P times, and two multipliers 212, and implements the calculation in FIG. 6(d).

From FIG. 7 , the periodic disturbance component of the angular velocity ω _d included in ΔT _m can be estimated from Δθ _dc .

Next, the torque controller 22 ( FIG. 8 ) that suppresses this ΔT _m will be described.

The following two conditions are required as conditions necessary for the torque controller.

(1) High followability to periodic disturbances

(2) Sensitivity is low in components other than periodic disturbances. 3(1) is the most important condition for suppressing periodic disturbances. (2) is a necessary condition for preventing the torque controller from affecting the overall control when a direct current Δθ occurs during a transition or the like. As mentioned above, the premise of this compensator is "periodic disturbance", and the equivalent transformation shown in Fig. 6 cannot be applied to DC disturbance and the like. Therefore, it is necessary to reduce sensitivity in components other than periodic disturbances.

The torque controller illustrated in FIG. 8 is composed of a zero command generator 17 that gives a zero command to an estimated value ΔT _mc of ΔT _m ; a sine wave transfer function 221 having a peak at an angular frequency ω _d ; and a torque control gain 222 . At this time, the angular frequency ω _d of the sine wave transfer function 221 is made to coincide with the fluctuation frequency of the periodic disturbance torque. In the case of a compressor or the like, since the fluctuation period of the frequency disturbance is the same frequency as the drive frequency, it is easy to make ω _d coincide with the pulsation frequency. Also, even for periodic voltage disturbances, the setting of ω _d is relatively easy because it is almost an integer multiple of the driving frequency.

In addition, this torque controller satisfies both of the above-mentioned conditions (1) and (2). The sine wave transfer function has an infinite gain at the angular frequency ω _d , and only the component deviation of the ω _d component included in ΔT _mc can be set to zero, and has no sensitivity to other frequency components. Details of the sine wave transfer function are disclosed in, for example, JP-A-7-20906 or the like.

The torque ripple component ΔT _m can be suppressed by adding the sine wave signal I _qSIN of the angular frequency ω _d to I _q ^* using the torque controller 22 shown in FIG. 8 .

In the above, Embodiment 1 of the present invention has been described by taking a periodic disturbance load as an example, but it can also deal with periodic voltage disturbances in the same way. In addition, as a current detection method, a method of reproducing the motor current from the current value I0 of the converter 4 is used, but it is also possible to directly detect each phase current using a Hall CT and a shunt resistor.

[Example 2]

Next, Embodiment 2 of the present invention will be described with reference to FIG. 9 .

In Embodiment 1, for the torque ripple vibration frequency ω _d , a sine wave transfer function with an infinite gain is introduced. As a result, the _ωd component included in the torque ripple is removed, and the distortion of the drive current of the PM motor increases instead. Since it follows the load torque variation instantaneously, it is unavoidable that the instantaneous current fluctuates greatly. For this reason, the efficiency of the PM motor deteriorates, and abnormalities such as overcurrent trips due to peak currents occur. Therefore, it can be said that the reduction effect of vibration and noise is alleviated as a whole, and on the contrary, it is practical to adjust a compromise point for preventing current distortion.

Embodiment 2 is to add the function of adjusting the suppression ability to the suppression effect of the periodic disturbance.

FIG. 9 illustrates the configuration of the torque controller 22B in the second embodiment. Embodiment 2 is realized by using the present torque controller 22 instead of the torque controller 22 of FIG. 1 .

In the torque controller 22B in FIG. 9 , it is composed of a sine wave transfer function 221B having a peak suppression function, a control gain 222B, a zero command generator 17 , and an adder/subtractor 16 . The sine wave transfer function 221B with peak suppression function has a Ta·s term in the denominator, and the peak value of the function can be changed according to the magnitude of Ta.

As a result, the disturbance suppression effect of the ω _d component can be adjusted, and the motor can be driven at an optimum point for noise, vibration, and distortion of the phase current of the PM motor.

Next, Embodiment 3 of the present invention will be described with reference to FIG. 10 .

In Embodiments 1 and 2, as a torque controller, a transfer function having a maximum gain (infinity in Embodiment 1) is introduced for the vibration frequency ω _d of torque ripple. In these controllers, the convergence response of the ripple component is determined by the magnitude of the control gain (control gain 222, or control gain 222B) of the output stage at the final terminal.

However, the relationship between the set value of these control gains and the actual response time is complicated, and they are studied by making a board diagram, or obtained by simulation tests or actual measurements. Therefore, a lot of effort is required in the adjustment work.

In Embodiment 3, a torque control device for simplifying adjustment work is provided.

FIG. 10 shows a configuration example of a torque controller 22C in the third embodiment. Embodiment 3 can be realized by using the present torque controller 22C instead of the torque controller 22 of FIG. 1 .

In the torque controller 22C in Fig. 10, by monophase-dq coordinate converter 223, primary delay filter 224, integration controller 225, dq-monophase inverse transformer 226, and integrator 20, adder 16, zero Composition of instruction generator 17. The operation of this torque controller 22C will be described below.

The output ΔT _mc of the ΔT _m estimator is decomposed into a SIN component and a COS component in the one-way-dq coordinate converter 223 . Furthermore, the conversion formula of the one-way-dq coordinate converter 223 is as follows.

[Formula 1]

$\left[\begin{array}{c}\mathrm{<span\; class="notranslate">\; \&Delta;</span>}{\mathrm{<span\; class="notranslate">\; T</span>}}_{\mathrm{<span\; class="notranslate">\; ds</span>}}\\ {\mathrm{<span\; class="notranslate">\; \&Delta;T</span>}}_{\mathrm{<span\; class="notranslate">\; qs</span>}}\end{array}\right]<span\; class="notranslate">\; =</span>\left[\begin{array}{c}\mathrm{<span\; class="notranslate">\; cos</span>}<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>\\ <span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; sin</span>}<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span>\end{array}\right]\mathrm{<span\; class="notranslate">\; \&Delta;</span>}{\mathrm{<span\; class="notranslate">\; T</span>}}_{\mathrm{<span\; class="notranslate">\; mc</span>}}$

……(Formula 1)

According to (Equation 1), if the frequency component of ω _d is included in ΔT _mc , the average value of ΔT _ds and ΔT _qS is a non-zero value depending on the amount. This average value corresponds to the COS component contained in ΔT _mc , and the SIN component, respectively. However, since ΔT _ds and ΔT _qs contain a large amount of the double component of ω _d , it is necessary to delete the AC component in the primary delay filter 224 . As a result, ΔT _ds and ΔT _qs are pulsation components, COS components, and SIN components included in ΔT _mc . Next, since each component is set to zero, a "zero" signal is given as a command from the zero command generator 17, and the deviation from the command is calculated in the adder-subtractor 16. Based on these deviations, the integral controller 225 performs integral compensation to control the pulsation component to zero. Finally, invert the values of I _ds and I _qs into a single signal, and output I _qSIN . This inverse transformation is calculated according to the following formula.

[Formula 2]

${{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; wxya</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; =</span><span\; class="notranslate">\; [</span>\mathrm{<span\; class="notranslate">\; cos</span>}<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; sin</span>}<span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}\mathrm{<span\; class="notranslate">\; t</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; ]</span>\left[\begin{array}{c}{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; ds</span>}}\\ {\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; qs</span>}}\end{array}\right]$

... (Formula 2)

The pulsation component ΔT _mc becomes a DC amount after the coordinate conversion by (Formula 1), so the integral controller 225 can remove the deviation. That is, the torque controller is equivalent to a compensation element whose gain becomes infinite at the angular frequency ω _d when viewed from the outside. That is, it has the same frequency characteristics as the torque controller 22 of the first embodiment.

In the case of the torque controller 22C, compared with the torque controllers of FIGS. 8 and 9 , the adjustment position is at 2 of the time constant T _ATR of the primary delay filter and the gain K _iATR of the integral controller 225 . However, it is not particularly difficult to adjust T _ATR because it is only necessary to select a sufficiently large time constant for ω _d . In addition, the value of K _iATR directly determines the response time of pulsation component suppression, and the control response time is linear to the value of K _iATR . As a result, gain setting can be easily achieved.

[Example 4]

Next, Embodiment 4 of the present invention will be described using FIG. 11 .

In Embodiment 3, for the vibration frequency ω _d , a torque controller with an infinite gain is provided. This is equivalent to the operation of the torque controller ( FIG. 8 ) of the first embodiment. Therefore, the same problem as described in Example 2 arises. That is, the _ωd component included in the torque ripple is removed, and instead, the distortion of the driving current of the PM motor increases, the efficiency of the PM motor deteriorates, and an abnormality such as an overcurrent trip due to a peak current occurs.

Thus, as in Embodiment 2, a method of setting the gain in the angular frequency ω _d from infinity to finite is proposed.

FIG. 11 illustrates the configuration of a torque controller 22D in Embodiment 4. As shown in FIG. Embodiment 4 can be realized by using this torque controller 22D instead of the torque controller 22 of FIG. 1 .

The difference between the torque controller 22D in FIG. 11 and the torque controller 22C in FIG. 10 is that the integral controller 225 is changed to an incomplete integral controller 225D. The peak is suppressed with time constant Ti and gain K _iATR in incomplete integrator 225D. As a result, the disturbance suppression effect of the ω _d component can be adjusted, and the motor can be driven at an optimum point for noise, vibration, and distortion of the phase current of the PM motor.

[Example 5]

Next, Embodiment 5 of the present invention will be described using FIG. 12 .

In Embodiments 1 to 4, the method of estimating the periodic torque ripple component based on the estimated value of the shaft error Δθ and suppressing it was described. The main pulsation component appears as I _qc and the estimated value of axis error, but it also affects I _dc .

Although the d-axis current does not impart torque, the rotation axis deviates due to torque pulsation, and a current due to pulsation is generated in the d-axis direction. An example using this to further reduce torque ripple is Embodiment 5.

In FIG. 12, the controller 2E is substantially the same as the controller 2 in the first embodiment. A d-axis current controller I _dACR (22C) for performing d-axis (dc-axis) current control is newly added. For example, 22C imports exactly the same information as the torque controller 22C shown in FIG. 10 (the gain K _iATR needs to be adjusted), inputs I _dc instead of ΔT _mc , and adds I _d ^* to output. In the voltage command calculator 12, the calculation of the voltage command is performed using I _d ^** as a new command value.

By adding I _dACR , the ripple component included in I _dc can be removed, and as a result, the torque ripple component can be reduced.

[Example 6]

Next, Embodiment 6 of the present invention will be described with reference to FIG. 13 .

In FIG. 13, part numbers 1, 2, 3, 5, 7, 41, 42, and 43 are the same as those of the same number in Embodiment 1, respectively. In this embodiment, the controller 2, the converter 3, the current detector 7, and the diode bridge 42 are integrated on the power module to minimize the size. An external microcomputer is used as the rotational speed command generator 1, and a speed command is transmitted by communication. In addition, by wiring the AC power supply 41 , the smoothing capacitor 43 , and the PM motor 5 to the power module, a synchronous motor control device capable of suppressing periodic torque ripple can be realized.

The purpose of the present invention is to reduce the noise and vibration of the system, reduce the number of sound-proof and vibration-proof materials, and realize the miniaturization of the device. In Embodiment 6, by modularizing the controller and the inverter, there is an effect that the overall size of the device can be further realized.

[Example 7]

Next, Embodiment 7 of the present invention will be described using FIG. 14 .

In FIG. 14, part numbers 2, 3, 6, 7, 42, and 43 are the same as the parts of the same number in embodiment 1 (FIG. 1) and embodiment 6 (FIG. 14), respectively. In this embodiment, the outdoor unit 30 of the air conditioner is composed of a probability module assembled with the controller 2 , the converter 3 , the current detector 7 , and the diode bridge 42 . In a compressor such as an air conditioner, it is difficult to incorporate a PM motor inside a hermetically sealed compressor, and to detect the rotational speed of the PM motor, the position of magnetic flux, and the like.

However, by assembling the controller of the present invention, the vibration and noise generated by the compressor can be reduced without detecting the rotational speed and position of the PM motor.

Fig. 15 is a graph showing an example of changes in noise and changes in current waveforms when the air-conditioning compressor of the present invention is started and the rotational speed is changed. In Fig. 15, since the transfer is changed from high speed to low speed, the overall noise is reduced.

After the rotation speed of the compressor is changed, the noise and vibration remain, but the noise drops from several seconds to tens of seconds. At this time, the distortion waveform of the current changes before and after the noise reduction. This is due to the change in rotational speed of the compressor, which produces a transient phenomenon due to the changing conditions of the periodic disturbance. Even with this transient phenomenon, the waveform (distortion) of the current changes because the controller responds gradually and finally reduces the noise.

In the present invention, since periodic disturbances can be suppressed based on instantaneous shaft error calculations, this phenomenon can be observed even when the rotational speed is high. When the maximum frequency of the frequency for driving the compressor is set to 100%, even in a range exceeding 30% of the maximum frequency, noise reduction and vibration reduction can be realized.

In addition, as shown in the current waveform of FIG. 15 , the same effect can be obtained not only for disturbance components of frequencies lower than the driving frequency but also for frequency components higher than the driving frequency.

In addition, as an example, an air conditioner is described as an example, but the same effect can be obtained when other electrical equipment, for example, a compressor and a refrigerator are packaged.

As described above, according to the present invention, it is possible to realize a high-performance motor drive that suppresses the periodic disturbance of the load device or the motor itself without using a sensor for detecting the rotational speed and the position of the rotating shaft of the synchronous motor. Furthermore, the same can be realized even when there are sensors for detecting the rotational speed and the position of the rotating shaft of the synchronous motor.

Claims (12)

1, a kind of control device of synchronous motor has the controller of controlling the synchronous motor of following load by the output voltage of converter, it is characterized in that:

Above-mentioned controller have to the proportional amount of deviation of the rotor-position of above-mentioned synchronous motor and the rotor-position supposed in above-mentioned control inside, be the Δ θ estimator that axis error is inferred calculating; With according to the output valve of above-mentioned Δ θ estimator, be that the axis error presumed value is found the solution any one party of above-mentioned motor or load or the periodic perturbation estimator of the periodic perturbation composition that both sides are taken place.

2, the control device of synchronous motor as claimed in claim 1 is characterized in that:

Above-mentioned axis error presumed value is to calculate according to the detected value that flows through at least one side of the alternating current of above-mentioned synchronous motor or the electric current that power supply provides.

3, the control device of synchronous motor as claimed in claim 1 is characterized in that:

Have near the compensator that has the frequency characteristic of peak value variation frequency by periodic perturbation or the earthquake frequency and eliminate the torque controller of above-mentioned periodic perturbation.

4, the control device of synchronous motor as claimed in claim 1 is characterized in that above-mentioned load is a compressor.

5, the control device of synchronous motor as claimed in claim 1 is characterized in that:

Above-mentioned periodic perturbation estimator calculates above-mentioned periodic perturbation composition according to the constant of the variation frequency of above-mentioned axis error presumed value, periodic perturbation, above-mentioned synchronous motor and above-mentioned load device.

6, the control device of synchronous motor as claimed in claim 3 is characterized in that:

According to the output of above-mentioned torque controller, the output voltage of above-mentioned controller is carried out revisal.

7, the control device of synchronous motor as claimed in claim 3 is characterized in that:

Possesses the device that the above-mentioned peak value of change also can change the inhibition effect of periodic perturbation.

8, the control device of synchronous motor as claimed in claim 3 is characterized in that:

Above-mentioned torque controller,

Above-mentioned periodic perturbation as input,

SIN function and COS function with the frequency change of above-mentioned periodic perturbation are carried out multiplying,

Obtain each mean value, derive SIN composition, the COS composition of above-mentioned periodic perturbation,

Voltage to above-mentioned controller output applies based on integral control or the not exclusively revisal of integral control, so that above-mentioned SIN composition and above-mentioned COS composition are zero.

9, the control device of synchronous motor as claimed in claim 1 is characterized in that:

Possess: for the magnetic pole axle phase place of above-mentioned synchronous motor, calculate device as the exciting current composition of the electric current composition synchronous with it,

Possess and remove the device that is included in the ripple component in the above-mentioned exciting current composition.

10, a kind of freezer is characterized in that:

Control device with the described synchronous motor of claim 1 comes Driven Compressor.

11, a kind of air regulator is characterized in that:

Control device with the described synchronous motor of claim 1 comes Driven Compressor.

12, a kind of system power module can be by making the converter circuit portion that the synchronous motor that is connected with load is applied voltage; This converter circuit portion of giving provides the converter circuit portion of electric power; And the controller of the outer voltage that adds to above-mentioned synchronous motor of control is integrated and be considered as parts as assembly, it is characterized in that:

Above-mentioned controller have to the proportional amount of deviation of the rotor-position of above-mentioned synchronous motor and the rotor-position supposed in above-mentioned control inside, be the Δ θ estimator that axis error is inferred calculating; With according to the output valve of above-mentioned Δ θ estimator, be that the axis error presumed value is found the solution any one party of above-mentioned motor or load or the periodic perturbation estimator of the periodic perturbation composition that both sides are taken place.

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