JPS5925579A - Dc-dc converting circuit - Google Patents

Abstract

PURPOSE:To reduce generation of the same phase mode switching noise by a method wherein an electrostatical shielding layer is arranged between a primary winding and a secondary winding, and a voltage induced on one side of the primary winding and a voltage induced on another side of the primary winding are negated. CONSTITUTION:The neutral point of the primary winding 27 of a transformer 26 is opened, and a switching element 25 is connected between the open terminals 44, 45 thereof to construct a negating means. Accordingly, the primary winding 27 is divided into windings 27a, 27b. The electrostatically shielding layer 46 is interposed between the primary winding 27 and the secondary winding 28, and the electrostatically shielding layer 46 thereof is connected to a terminal 19a on the secondary side constituting the zero electric potential point (stationary terminal) of the AC current of the secondary winding 28. A terminal 19b on the secondary side is also the stationary terminal, and the same effect is obtained even when the electrostatically shielding layer 46 is connected to the terminal 19b on the secondary side.

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a DC-DC conversion circuit that switches DC power and supplies it to the primary winding of a transformer, rectifies and smoothes the output of the secondary winding of the transformer, and obtains a DC output. In particular, it relates to a circuit that separates input and output with high impedance and generates little common-mode switching noise.

<Background> This glazed DC-DC conversion circuit is used, for example, as a power source for a digital line termination device installed inside a subscriber's premises in a digital subscriber line transmission system. That is, as shown in FIG. 1, the digital line termination device 11 installed inside the subscriber's premises is generally configured to operate by distant power supply from the station, and the balanced type cable connected to the digital line termination device 11 is On the subscriber line 12, a digital signal and a feeding direct current are superimposed. subscriber line 1
The other end of 2 is led into the office and connected to a digital line termination device (not shown) inside the office.

The digital signal on subscriber M12 connects to device 11 at connection point 13,)? The signal is input to the pulse transmission circuit 5 via the pulse transmission circuit 14. The pulse transmission circuit 15 includes an equalization amplifier circuit, a pulse transmission circuit, and the like.

The supplied DC current on the subscriber line 12 is passed through the connection point 13 and the power separation filters 16a and 16b to the DC-DCC conversion circuit 17, and a DC voltage of 30 V, for example, is applied between these primary sides 18a and 18b. Ru. DC-1) The C conversion circuit 17 performs DC-DC conversion.
For example, a DC voltage of 5V is generated between the next sides 19a and 19b. The main part or the entire digital line termination device 11 operates as a DC-DCC conversion circuit 170. A DC blocking capacitor 21 inserted between the connection point 13 and the connection point between the filters 16a and 16b and the transformer 14 is provided to prevent the supply DC current from flowing through the transformer 14. The power separation filters 16a and 16b are constructed of, for example, coils so as to have low impedance for direct current and high impedance for alternating current. The primary 1i111 1 8 a , 1 8 of this uDc-DC conversion circuit 17
This is to avoid short-circuiting the digital signals since the AC impedance 1' 7 between them is low.

Now, in the DC-DC converter circuit 17 which is the power source of the digital line termination device 11 having the above configuration, the DC-DC converter circuit 17 having the conventional configuration is used.
When applying the C conversion circuit, the following drawbacks occur.

(a) Between the primary side and the secondary side of the DC-DC conversion circuit 17, that is, the primary side 18a and the secondary side 19
Switching noise v1, so-called common mode switching noise, occurs between a certain Fi1 order 'M'lJ 1 8 b and the secondary side 19b due to switching of the KDC-DC conversion circuit 17. This common mode switching noise is caused by the subscriber line 12 and the digital line termination device 1.
It is converted into differential mode noise according to the amount of unbalanced attenuation determined by
This can cause code errors in digital signals. For this reason, it is necessary to sufficiently reduce the generation of common mode switching noise, but conventional DC-DC conversion circuits have not been able to satisfy this requirement.

(b) Since vertical noise such as impulsive noise from the analog telephone line is induced onto the subscriber line 12 and causes code errors in digital signals, the unbalanced attenuation of the digital line termination device 11 is The amount needs to be high enough. In order to eliminate the drawbacks described in the taJ section, the second
As shown in the figure, an external capacitor 23 with sufficiently low impedance at the switching frequency of the DC-DC conversion circuit 17
Some devices suppress the common mode switching noise by connecting between the primary side 18a and the secondary side 191L (or between the primary side 18b and the secondary side 19b). However, if such a configuration is used and the DC-DCC conversion circuit 170 is connected to the ground with low impedance, resonance will occur in the vertical circuit from the coil for the power separation filter 16a (including Z6B) and the external capacitor 23. At this resonance point, the unbalanced attenuation of the digital line termination circuit 11 is extremely degraded, resulting in a code error. For this reason, it is necessary to set the resonance point at a frequency sufficiently higher than the pulse transmission band, and this requires an external capacitor. 23 and DC-D
The primary side 18a, 18b and the secondary side 19& of the C conversion circuit 17
! It is necessary to separate the transformer 19b from the transformer 19b with a high impedance equivalent to the stray capacitance of the transformer 14.

However, in this case, in the conventional circuit configuration, the above (a
The disadvantages mentioned in section 1 arise.

These points will be explained in more detail below.

FIG. 3 shows the basic configuration of a conventional DC-DC conversion circuit.

The DC input voltage E1 input from the DC power supply 24 through the primary sides 18a and 18b turns on/off the switch element 25.
Due to repeated OFF operation (hereinafter referred to as switching), υ
It is converted into an alternating voltage (hereinafter referred to as switching voltage),
A switching voltage el is generated between both ends 29,31 of the primary winding 27 of the transformer 26. Therefore, a switching voltage e2 is induced between both ends 32 and 33 of the secondary winding 28 of the transformer 26. This switching voltage e2 is half-wave rectified by a diode 34 and smoothed by an output capacitor 35 to obtain a DC output voltage E2, which is supplied to a load 36 through secondary sides 19a and 19b. Note that the switching voltage e1 induced between both ends 29.31 of the primary winding IvlJ270 and the both ends 32.3 of the secondary winding 28
The ratio between the switching voltage e2 induced between the primary winding 27 and the secondary winding m28 is determined by the winding ratio between the primary winding 27 and the secondary winding m28. An input capacitor 37\ is connected between both ends of the DC power supply 260,
Still, the switch element 25 is, for example, a transistor,
This transistor 25 is inserted in series with the primary winding 27, and the drive circuit 3 is inserted between the base and emitter of the transistor 25.
8 are connected.

The conventional DC-DC conversion circuit shown in FIG. 3 has a drawback that a large switching voltage is generated between the primary side and the secondary side, that is, between the primary side 18a and the secondary side 19a. This is called common mode switching noise, and will be explained below using FIG. 4. FIG. 4 shows the generation mechanism of common mode switching noise in the DC-DC conversion circuit shown in FIG. 3, focusing on the alternating current component. The symbols in FIG. 4 are the same as in FIG. 3, and el, effi, Vl, and v2 represent the switching voltages between the respective terminals at any given time, and the arrows represent the mutual relationship between the voltage polarities. Switch element 25
Connect the connection point with the power supply 24 to the primary side 18b1 diode 34
The connection point between the output capacitor 41 and the output capacitor 41 is the secondary side 19b. Capacitor 42 between winding ends 29 and 32, winding ends 3
Each of the capacitors 43 between 1.33 and 33 is a lumped constant circuit representing the stray capacitance distributed between the primary winding 27 and the secondary winding 28. The closed circuit consisting of the switch element 25, the primary winding 27, and the input capacitor 37 is named the closed circuit I, and the primary winding M27, the secondary winding 28, and the capacitor 42
．． The closed circuit consisting of the secondary winding 28, the diode 34, and the output capacitor 35 is called the closed circuit I.

In FIG. 4, attention is first paid to cycle I. Since the input capacitor 37 is short-circuited (sufficiently low impedance) to the switching frequency component, no switching voltage is generated across it. Therefore, from Kirchhoff's voltage law, the primary winding 270 is connected to both ends of the switch element 250.
．． A switching voltage with an amplitude equal to the switching voltage e1 induced between 31 and 31 is generated in opposite phase. Next, we will focus on the closed circuits. Since the output capacitor 35 is short-circuited for the switching frequency component, no switching voltage is generated across it. Therefore, according to Kirchhoff's voltage law, a switching voltage with an amplitude equal to the switching voltage e2 induced between both ends 32 and 33 of the secondary winding 28 is generated in opposite phase across the diode 340.

Next, let's focus on the cycle n. Normally, the capacitance values of the capacitors 42 and 43 are both small, and the impedance is sufficiently high over a high frequency range including the switching frequency. Now, closed circuit ■
, the switching voltages generated across the capacitors 42 and 43 are Vl and V!, respectively. Then, from Kirchhoff's voltage law, the relationship of V 1 +v s = e t - e t is satisfied. Further, the ratio between voltages v1 and v2 is inversely proportional to the capacitance value of each capacitor 42,43. The switching voltage v1 generated across the capacitor 42 is common mode switching noise. The following description will be given with reference to FIG. 5, which is an enlarged view of the closed circuit portion. The symbols in FIG. 5 correspond to those in FIG. The capacitance value of capacitor 42.43 is 01, Ct
shall be. From Figure 5, the amplitude value v1 of the common mode switching noise is equal to both the switching voltage et generated across the primary winding 27 and the switching voltage e2 induced across the secondary winding 28. Arises from influence.

In a normal DC-DC conversion circuit, it is e1\e2. Therefore, primary side 18a, 18b - secondary side 19all
In order to isolate Qb with high impedance and to reduce the occurrence of common mode switching noise, it is necessary to make the capacitance value C2 of the capacitor 43 sufficiently smaller than the capacitance value C1 of the capacitor 42. However, capacitor 43
The capacitance value C! is the primary winding 27 and secondary winding 28 of the transformer.
(and the shape of the core), the capacitance value C2 cannot be made small. On the other hand, if the capacitance value CI of the capacitor 42 is increased (for example, by externally attaching the capacitor), the common mode switching noise will be reduced, but the primary side 18a
, 18b, = -low impedance between the secondary sides 19a and 19b. From the above, the first order f! 11188, 18
It has been difficult in the prior art to isolate the b-secondary sides 19a and 19b with high impedance and to reduce common mode switching noise. - As an example, in the configuration shown in Fig. 3, the DC input voltage E1-30V, the DC output N,
Pressure E! = 5 V, output power of about 1 W DC-
In a DC conversion circuit, common mode switching noise is a ripple component of about 10 Vpp. However, 1
Next, this noise value differs slightly depending on the configuration of the secondary winding.

<Summary of the Invention> The object of the present invention is to provide an oc-DC converter which separates the primary side and the secondary side with high impedance over a high frequency range including direct current and switching frequencies, and which generates less common mode switching noise. The purpose is to provide circuits.

According to this invention, there is a silent insulator between the primary winding and the secondary winding.
! The electrostatic shielding layer is connected to the stationary end of the secondary winding, and the electrostatic shielding layer is connected to the static end of the secondary winding on one side of the primary winding when viewed from the midpoint of the primary winding. A canceling means is provided so that the voltage induced in the primary winding and the voltage induced in the electrostatic shielding layer on the other side of the primary winding are equal to each other and have opposite polarities.

<First Embodiment> FIG. 6 shows a first embodiment of the present invention, in which parts corresponding to those in FIG. 3 are given the same reference numerals. In this embodiment, the primary winding 27 is open at its midpoint, and the switch element 25 is connected between the open ends 44 and 45 to constitute canceling means. Therefore, the primary winding 27 is divided into windings 27a and 27b. An electrostatic shielding layer 46 is interposed between the primary winding 27 and the secondary winding 28. ) is 2
It is connected to the next side 19a. Secondary (111+19
b is also a stationary end, and the effect is the same even if the electrostatic shielding layer 46 is connected to the secondary side 19b. Since it has such a structure, the primary side 18a-secondary side 19 is connected as explained below.
This has the effect of reducing the switching voltage generated between terminals a, that is, common mode switching noise.

In the case of the conventional DC-DC conversion circuit shown in FIG.
The switching voltage e1 generated across the secondary winding 280
This occurs due to the influence of both the switching voltage e2 induced at both ends. However, in the embodiment shown in FIG.
1) A static shielding layer 46 between the primary winding 27 and the secondary winding 28
is provided, and this is the stationary end of the secondary winding 28.
The switching voltage e2 generated across the secondary winding 28 is generated as voltage (common mode switching noise) between the primary side 18a and the secondary side 19a. do not. (11) Also, since the switching element 25 is inserted and connected to the middle point of the primary winding 27, the potential distribution of the switching voltage between the electrostatic shielding layer 46 and the primary winding 27a and the electrostatic Shielding layers 46 and 1
The potential distribution of the switching voltage between the primary winding 27b and the primary winding 27b has opposite polarity and cancels each other out.
, 27b does not occur as a voltage (common mode switching noise) between the primary side 18a and the secondary side 19a.

These points will be explained in more detail using FIG. 7.

FIG. 7 shows the effect of reducing common mode switching noise in the DC-DC conversion circuit shown in FIG. 6, focusing on the alternating current component.

The symbols in FIG. 7 are the same as those in FIG. 6, and el, el, and vl represent the switching voltages between the respective terminals at any given time, and the arrows represent the mutual relationship between the voltage polarities. Capacitor 47.48
represents the stray capacitance distributed between the primary winding 27a and the electrostatic wall shield layer 46 using a lumped constant circuit, and the capacitor 47
．． 48 are both ends of the primary winding 27 & 29.44 and ¥ respectively
It is connected between the P electrode and the receiving layer 46. capacitor 4
9.51 represents the stray capacitance distributed between the primary winding 27b and the electrostatic shielding layer 46 using a lumped constant circuit, and capacitors 49.51 are connected to both ends 45.51 of the primary winding 27b, respectively.
39 and the Ft shielding layer 46. The capacitors 52 and 53 represent the stray capacitance distributed between the secondary winding 28 and the electrostatic shielding layer 46 using lumped constant circuits,
Capacitor 52.53 is connected to both ends of secondary winding 280 32.33
and the electrostatic shielding layer 46. capacitor 3
7. The closed circuit of the primary windings 27a, 27b and the switch element 25 is defined as a closed circuit I, and the primary windings 127a, 27b, switch f
'J child 25, the closed circuit of the electrostatic shielding layer 46 is a closed circuit ■,
Primary winding 27a1, silent shielding layer 46, capacitor 47.
The closed circuit of 48 is a closed circuit V, and the closed circuit of the switch element 25, electrostatic shielding layer 46, and capacitors 48 and 49 is a closed circuit ■
Then, the closed circuit of the primary winding 27b1, the static shield layer 46, and the capacitor 49.51 is a closed circuit ■, and the secondary winding i! 1128
, the capacitors 52, 53, and the electrostatic shielding layer 46 are defined as a closed circuit (2).

First, the middle part will be explained. In FIG. 7, attention is first paid to the cycle l. Since the output capacitor 35 is short-circuited (sufficiently low impedance) with respect to the switching frequency component, no switching voltage is generated across it.

Therefore, from the Kirchhoff voltage period, both ends 32 and 33 of the secondary winding 28 are connected to both ends of the diode 340 for half-wave rectification.
The switching voltage induced between e! Switching voltages of amplitude equal to are generated in opposite phases. Next, we will focus on the cycle ■. Since both ends of the capacitor 52 are short-circuited by the electrostatic shielding layer 46, no switching voltage is generated. Therefore, from the Kirchhoff voltage period in the closed circuit ■, both ends of the secondary winding 28 are 32 and 33 at both ends of the capacitor 53.
A switching voltage with an amplitude equal to the switching voltage e2 occurring between the two is generated in opposite phase. From the above, the switching voltage e2 generated on the secondary side is 9 due to the electrostatic shielding N446.
Closed only on the secondary side, primary (1tlllBa - secondary side 19
It was explained that it does not affect the voltage between a and a.

Next, the above item (11) will be explained. In FIG. 7, attention is paid to cycle 1. Since the input capacitor 37 is short-circuited for the switching frequency component, no switching voltage is generated across it. The primary winding 27 is at the midpoint 44
．． 45, the distance between both ends 29.44 of the primary winding 27a is equal to the distance between both ends 45.31 of the primary winding 27b. From the Kirchhoff voltage period in the closed circuit ■, the primary winding 27 is connected to both ends of the switch element 25.
between both ends 29.44 of the primary winding 27b, and between both ends 45.44 of the primary winding 27b.
A switching voltage having an amplitude equal to the sum e1 of the switching voltages induced between 31 and 31 is generated in opposite phase.

Next, we will focus on cycles ■, ■, and ■. Normal capacitor 47
．． The capacitance values of 48, 49 and 51 are both small, and the impedance is sufficiently high over a high frequency range including the switching frequency. In addition, by bifilar winding the normal primary windings 1J27a, 2jb and one end 29.45 of the same as the same example, between the electrostatic shielding layer 46 and the winding end 29,
The physical positional relationship between the electrostatic barrier layer 46 and the winding end 45 is equal. This useless capacitor 47 and capacitor 4
The capacitance values of 9 are approximately equal, and this is set as CI. Similarly, the physical positional relationship between the static shielding layer 46 and the winding end 44 and between the static shielding layer 46 and the winding end 31 are the same. Therefore, the capacitance values of capacitor 48 and capacitor 51 are approximately equal, and this is designated as C4.

Now, let's focus on the cycle ■. Assume that a switching voltage v1 is generated across the capacitor 47. As mentioned earlier, the sum of the voltages of the primary windings 27a and 27b and the switch element 2
Since the voltage of 51 is equal and in opposite phase, from the Kirchhoff voltage period, capacitor 47 is connected across capacitor 51.
A switching voltage with an amplitude equal to the switching voltage Vl generated across the zero terminal is generated in opposite phase. This switching voltage v1 is a common mode switching voltage. Next, it will be explained using FIG. Figure 8 shows the cycle ■ in Figure 7,
This is an abstraction of parts ① and ②, and the symbols follow those in FIG. The capacitance Jt value of the capacitor 47.49 is assumed to be C11, and the capacitance value C4 of the capacitor 48.51. Figure 8 cycle■
Solving the cycle equations for , ■ and ■, respectively, gives the following. From equation (2), the common mode switching noise is calculated by capacitor 47.49 (7) capacitance value C8 and capacitor 48
．． The smaller the difference from the capacitance value C4 of 51, the more the capacitance is reduced.

Now, in FIG. 6, the electrostatic shielding layer 46 - the winding end 29
(or 45) and the stray capacitance between the electrostatic shielding layer 4
6-The technique of equalizing the stray capacitance between the winding ends 4 and 4 (or 31) is relatively easy, and the electrostatic trace 1-
The physical position of the winding end 29 (or 45) with respect to 46 and the physical position of the electrostatic shielding layer 46-winding end 44 (or 31) may be made symmetrical. For example, primary winding 2
7 a and 27 b may be configured to be wound in one layer around the electrostatic shielding layer 46 . That is, a technique for making the capacitance value C8 of the capacitor 47 (or capacitor 49) and the capacitance value C4 of the capacitor 48 (or the condenser dirt 51) approximately equal in FIG. 8 is known. From the above, by inserting and connecting the switching element 25 to the midpoint of the primary winding 27, the switching voltage L induced across each end of the P primary windings 27a and 27b. It has been explained that the influence on the voltage between the next sides 19a can be reduced.

As described above, with the configuration shown in FIG. 6, the primary side 18a, 18b-2
DC-D that isolates the next side 19a and 19b with high impedance and generates little common mode switching noise
A C conversion circuit can be provided.

Second Embodiment> FIG. 9 shows a second embodiment of the present invention, in which the switch element 2
5a and 25b are connected in series to both ends of the primary winding 27, respectively, and the switch elements 25a and 25b are connected to the drive circuit 38.
Intermittent control is performed at the same timing. The other ends of the switch elements 25a and 25b are connected to both ends of the input capacitor 37. An electrostatic shielding layer 46 is interposed between the primary winding 27 and the secondary winding 28 . This electrostatic trace l-46
is connected to a winding end 32 which is an alternating current O potential point (hereinafter referred to as a static end) of the secondary winding 28. The secondary side 19b is also a stationary end, and the effect is the same even if the electrostatic shielding layer 46 is connected to the secondary side 19b. , and other symbols follow FIG. 3. This structure has the effect of reducing the switching voltage generated between the primary side 18a and the secondary side 19a, that is, the common mode switching noise, as described below.

According to the configuration shown in FIG. 9, +i) i-th winding 27-2
An electrostatic shield #46 is provided between the secondary windings 28 and is connected to the static end of the secondary winding 28, winding end 32, thereby causing the switching that occurs across the secondary winding 28. The voltage e2 is not generated as a voltage (common mode switching noise) between the primary side 18a and the secondary side 19JL. (11
) Switch element 25a.

25b are respectively connected to both ends of the primary winding 27, and the switches 27a and 27b are turned on and off at the same timing. The generated switching voltage and the winding end 29 (or 31)-1 which is both ends of the switch element 25a (or switch element 25b)
The switching voltage e1 generated across the primary winding 27 is caused by the switching voltage generated between the next side 18a (or 18b) having opposite polarity and canceling each other.
is not generated as a voltage (common mode switching noise) between the primary side 18a and the secondary side 19a.

These will be explained in detail using FIG. 10. 1st
FIG. 0 shows the effect of reducing common mode switching noise in the DC-DC conversion circuit shown in FIG. 9, focusing on the alternating current component. The symbols in FIG. 10 are the same as in FIG. 9, el, e2. Vl and v2 represent the switching voltages between the terminals at any given time, and the arrows represent the mutual relationship between the voltage polarities.

Also, in FIG. 10, parts corresponding to those in FIG. 7 are given the same reference numerals, and the closed circuits VX■ are the same.

The closed circuit I is composed of a capacitor 37, a primary winding 27, and switch elements 25a and 25b, and the closed circuit II is composed of a primary winding 27, a capacitor 47 and 51, and an electrostatic shielding layer 46.

First, the above item (1) will be explained. In FIG. 10, first focus on the cycle ■. Output capacitor 15 is short-circuited (sufficiently low impedance) for switching frequency components.
Therefore, no switching voltage is generated across it. Therefore, both ends 32.3 of the secondary winding 28 are connected to both ends of the diode 34 for half-wave rectification from the Kirchhoff voltage side.
The switching voltage induced between e! Switching voltages of amplitude equal to are generated in opposite phases. Next, we will focus on the cycle ■. Capacitors 52 and 53 are connected to the secondary winding 28.
The stray capacitance distributed between the thin layer 46 and the thin layer 46 is expressed by a lumped constant circuit. The capacitor 52 is connected to the winding end 32
and the electrostatic tracing layer 46, and the capacitor 53
is connected between the winding end 33 and the electrostatic shielding layer 46.

Here, since both ends of the capacitor 52 are short-circuited by the electrostatic shielding layer 46, no switching voltage is generated.

Therefore, from the Kirchhoff voltage side in the closed circuit (■) to both ends of the capacitor 53, both m32.3 of the secondary winding 28 are connected.
A switching voltage with an amplitude equal to the switching voltage e2 occurring between the two is generated in opposite phase. From the above, the switching voltage e2 generated on the secondary side is closed only to the secondary side by the electrostatic shielding layer 46, and does not affect the voltage between the primary side 18a and the secondary side 19a.

Next, the above item (11) will be explained. In FIG. 10, attention is paid to the cycle I. Since the input capacitor 37 is short-circuited for the switching frequency component, no switching voltage is generated across it. Further, a switching pressure e1 is induced between both ends 29 and 31 of the primary winding 270. Further, the switching elements 25a and 25b are turned on and off in the same phase.

Therefore, the primary winding 270 is connected to both ends of the switch elements 25a and 25b from the Kirchhoff voltage side in the closed circuit I.
A switching voltage having an opposite phase to the switching voltage e1 induced between both ends 29 and 31 is generated with the amplitude divided into two. Now, considering the variations in the characteristics of the switching elements, we will focus on the closed circuit (2) next to the switching voltage generated across the switching elements 25a and 25b. capacitor 47
．． 51 represents the stray capacitance distributed between the electrostatic shielding layer 46 and the primary winding 27 using a lumped constant circuit υ
, capacitor 47 is silent! A capacitor 51 is connected between the shielding layer 46 and the winding end 29, and a capacitor 51 is connected between the electrostatic shield M46 and the winding end 31.

Normally, the capacitance values of the capacitors 47 and 51 are both small, and the impedance is sufficiently high over the high frequency range including the switching frequency.
Let vx and vg be the switching voltages generated across the terminals, respectively, then vt from the Kirchhoff voltage side in the closed circuit
-1-vz=e1 is satisfied.

The following description will be made with reference to FIG. 11, which is an enlarged view of the closed circuit ■ portion. The symbol ti in FIG. 11 is shown in FIG. 10. Mfk value of each capacitor 47.51 is C1+.

Let's call it C4. A cycle equation is obtained for the cycle ■ in Fig. 11. Now, let's go back to FIG. 10 and explain. The common mode switching voltage is the switching voltage generated between the primary side 18a and the secondary side 19a, and when this is written as v8, the first
From figure 0, substituting equation (5) into equation (6) yields. In equation (7), the value of Δ is sufficiently small and the common mode switching noise is determined by the capacitance value C8 of the capacitor 47.
The smaller the difference between the capacitance value C4 and the capacitance value C4 of the capacitor 51, the more the difference is reduced. Now, in FIG. 9, the technique of making the stray capacitance between the electrostatic shielding layer 46 and the live wire end 29 equal to the stray capacitance between the electrostatic shielding layer 46 and the winding end 31 is relatively easy. The physical position of the winding end 29 with respect to the electrostatic shielding layer 46 and the physical position of the winding end 31 with respect to the electrostatic shielding layer 46 may be made symmetrical. For example, the primary winding 27 may be configured to have one layer of winding around the electrostatic shielding layer 46. That is, a technique for making the capacitance value C11 of the capacitor 47 and the capacitance value C4 of the capacitor 51 approximately equal in FIG. 10 is known. From the above, switch elements 25a, 2
5b on both sides of the primary winding 27, the switching voltage el generated across the primary winding 27 becomes 1.
It has been explained that the influence on the voltage between the secondary side 18a and the secondary side 19a can be reduced.

As described above, with the configuration shown in FIG. 9, the primary side 18a, 18b-2
DC-D that isolates the next side 19a and 19b with high impedance and generates little common mode switching noise
C conversion circuit can be provided.

Now, in Figures 6 and 9, when the secondary side is a DC-1)C conversion circuit with multiple outputs and multiple secondary windings, the primary winding and multiple secondary windings are sequentially coaxially connected. form on the heart, 2
By providing an electrostatic tracing layer between the next windings and connecting this electrostatic shielding layer to the electrostatic shielding layer 46 between the first winding and the second winding, common mode switching noise can be reduced. This is advantageous in reducing the occurrence of

<Third Embodiment> The above is a so-called current transmission type, that is, in FIGS. 6 and 9, when the primary side switch element 25, 258, 25b is off, the secondary side half-wave rectification third diode 34 is conductive. The present invention was applied to a DC-DC conversion circuit of the so-called voltage transmission type, that is, a DC-DC conversion circuit of the type in which the half-wave rectifier diode on the secondary side conducts when the switch element on the primary side is on. The present invention can also be applied to a DC conversion circuit.

An example thereof is shown in FIG. 12 corresponding to FIG. 6. The difference between these two figures is that the polarity of the rectifying diode 34 is reversed. Further, a voltage transmission type DC-DC conversion circuit corresponding to that shown in FIG. 9 is shown in FIG. In this case as well, only the polarity of the rectifying diode 34 is reversed.

Effects> As explained above, the present invention provides a DC system that isolates the input side and output side with high impedance over a high frequency range including DC and switching frequencies, and generates little common mode switching noise. - Since it can provide a DC conversion circuit, it can be used as a power receiving power source for a digital line termination device installed inside a subscriber's premises that operates by distant power supply from a station, for example in a digital subscriber line transmission system using balanced cables. There are advantages to the application of Specifically, D
Switching noise generated by the C-DC conversion circuit is less likely to be transmitted to the pulse transmission system circuit, and it is possible to separate the digital line termination device and the subscriber line by impedance. The effect of this is that a high unbalanced attenuation of the digital line termination device can be obtained in the pulse transmission band, and sign errors in the digital signal can be suppressed as much as possible against large vertical noise induced on the subscriber line.

A numerical example is shown next. In the configuration shown in Figures 6 and 12, the input voltage El is approximately 26V, and the manual current is approximately 24m.
1. Output voltage E2 is 5v±3.5%, output power is approximately 500
mW, the number of turns of the primary windings 27a and 27b is about 40 times each (bifilar winding), the number of turns of the secondary winding 28 is about 16 times, the thin layer 46 is made of copper foil, The switching element 25 is a MOS-FET, the rectifying diode 34 is a Schottky barrier diode, and the primary windings 27a, 27
b is a DC-DC conversion circuit in which one layer is wound around the electrostatic shielding layer 46 and the switching frequency is about 70 KHz, and the common mode switching noise is a ripple component of about 0.
It was 5Vpp. The common-mode switching noise was the same whether the switch element drive circuit 38 was a separately excited type or a self-excited type. Note that a shunt regulator is used on the secondary side to stabilize the output voltage. The power conversion efficiency was approximately 80%.

In addition, in the configurations shown in FIGS. 9 and 13, the number of turns of the primary winding 270 is approximately 80 times, and one no winding is applied to the electrostatic shielding layer 46, and the switch element +25b is a P-channel MO8-FET, Switch Yuko A to N channel MO
The same results as above were obtained except that 8-FET was used.

[Brief explanation of the drawing]

Figures 1 and 2 are diagrams each showing the configuration of a digital voice termination device installed inside a subscriber's premises; Figure 3 is a connection diagram showing the basic configuration of a conventional current transmission type DC-DC conversion circuit; Fig. 4 is an explanatory diagram of the common mode switching noise generation mechanism in the DC-DC conversion circuit of Fig. 3, Fig. 5 is an enlarged view of the closed circuit ■ part of Fig. 4, and Fig. 6 shows the present invention. A connection diagram showing an embodiment of a transmission type DC-DC converter circuit, FIG. 7 is an explanatory diagram of the common mode switching noise reduction effect in the DC-DC converter circuit of FIG. 6, and FIG.
The figure is an enlarged view of the closed circuit V, Vl, ■ part in Figure 7, Figure 9 is a connection diagram showing another embodiment in which the present invention is applied to a current transmission type DC-DC: & switching circuit, and Figure 10 is a DC- in M9 diagram
An explanatory diagram of the common mode switching noise reduction effect in a DC conversion circuit. Figure 411 is an enlarged view of the closed circuit ■ part in Figure 10, Figure 12 is a connection diagram showing an embodiment in which the present invention is applied to a voltage transmission type DC-DC conversion circuit, and Figure 13 is a voltage transmission type FIG. 3 is a connection diagram showing another embodiment applied to a DC-DC conversion circuit. 24: DC power supply, 25, 25a, 25b: Switch element, 26: Transformer, 27: Primary winding, 27'a, 27b
: Primary winding divided into two at the midpoint, 28: Secondary winding, 34
: Diode for half-wave rectification, 35: Output capacitor, 3
6: Load, 37: Input capacitor, 38: Switch confectionery driver 1 circuit, 46: Electrostatic shielding layer. Patent applicant: Takashi Kusano, agent of Nippon Telegraph and Telephone Public Corporation

Claims (3)

[Claims] (1) A DC-DC conversion circuit that switches DC power and supplies it to the primary winding of a transformer, rectifies and smoothes the output of the secondary winding of the transformer, and obtains a DC output.
a static shielding layer connected to the static end of the secondary winding between the primary winding and the secondary winding; and a static shielding layer connected to the static end of the secondary winding; The voltage induced on the shielding layer and the constant pressure induced on the shielding layer at the winding portion on the other side of the midpoint of the primary winding are of opposite polarity and A DC-DC conversion circuit characterized in that it has a canceling means for making the amplitude equal to so. (2) The canceling means is characterized in that the primary winding is opened at its midpoint, and a switch element is connected between the open terminals for the switching. The DC-DC conversion circuit described in Section 1. (3) The canceling means are provided at both ends of the primary winding, respectively.
The first of the switching reservoirs. A second switch element is connected in series with the first . 2. The DC-DC conversion circuit according to claim 1, wherein the second switch element is controlled on and off at the same timing.