JPH0295168A - Dc/dc converter circuit - Google Patents

Abstract

PURPOSE:To isolate between an input and an output by a high impedance, and to reduce the same phase mode switching noise by providing an electrostatic shielding layer between primary and secondary windings, and connecting a half-wave rectifying diode between both ends of the secondary winding. CONSTITUTION:An electrostatic shielding layer 46 is provided between a primary winding 27 and a secondary winding 28. This layer 46 is connected to a static end 29 of the winding 27, and cancelling means 34a, 34b for setting a voltage induced at one side of the winding 28 to the layer 46 as seen from the midpoint of the winding 28 and a voltage induced at the other side of the winding 27 with respect to the layer 46 to reverse polarities are provided. Thus, the primary side and the secondary side can be isolated by a high impedance over a high frequency range including a DC and a switching frequency, and the same phase mode switching noise can be reduced.

Description

DETAILED DESCRIPTION OF THE INVENTION This invention is a D-C-DC conversion method in which DC power is switched and supplied to the primary winding of a transformer, and the output of the secondary winding of the transformer is rectified and smoothed to obtain a DC output. The present invention relates to circuits, and particularly to circuits that separate input and output with high impedance and generate less common-mode switching noise.

<Background> This type of D(-DC conversion circuit) is used, for example, as a power source for a digital line termination device installed inside a subscriber's premises in a digital subscriber line transmission system. That is, as shown in FIG. The digital line termination device 11 installed inside the subscriber's premises is generally configured to operate by receiving power from a remote station, and the digital signal is transmitted on the subscriber line 12 of the balanced cable connected to the digital line termination device 11. and the feeding DC current are superimposed.Subscriber line 1
Although the other end of the line 2 is not shown in the figure, it is led into the office and connected to a digital line termination device inside the office.

A digital signal on the subscriber line 12 is input to a pulse transmission circuit 15 via a connection point 13 with the device 11 and a transformer 14. The pulse transmission circuit 15 includes an equalization amplifier circuit, a pulse transmission circuit, and the like.

The supplied DC current on the subscriber line 12 is passed through the connection point 13 and the power separation filters 16a and 16b to the DC-DC conversion circuit 1.
7 primary sides 18a and 18b, and these primary sides 1
For example, a DC voltage of 30 V is applied between 8a and 18b. The DC-DC conversion circuit 17 performs DC-DC conversion,
For example, a DC voltage of 5 V is generated between the secondary sides 19a and 19b of the DC-DC conversion circuit 17. The main part or the whole of the digital line termination device 11 is operated by the output of the secondary side 19a, 19b of the D(, -DC conversion circuit 17. DC blocking capacitor 21 inserted in
is provided to prevent the supply DC current from flowing through the transformer 14. The power separation filters 16a and 16b are constructed of, for example, coils so as to have low DC impedance and high AC impedance. This is to avoid short-circuiting the digital signals since the AC impedance between the primary sides 18a and 18b of the DC-DC conversion circuit 17 is low.

Now, the DC-DC conversion circuit 17, which is the power source of the digital line termination device 11 having the above configuration, is connected to the conventionally configured DC, -
When applying a DC conversion circuit, the following drawbacks occur.

(a) Between the primary side and the secondary side of the DC-DC conversion circuit 17, that is, between the primary side 18a and the secondary side 19a,
Or DC-D between the primary side 18b and the secondary side 19b.
Switching noise vI, so-called common mode switching noise, occurs due to switching of the C conversion circuit 17. This common mode switching noise
The noise is converted into differential mode noise according to the amount of unbalanced attenuation determined by 2 and the digital line termination device 11, and circulates between the secondary sides 22a and 22b of the transformer 14, causing code errors in the digital signal.

For this reason, it is necessary to sufficiently reduce the generation of common mode switching noise, but conventional DC/DC conversion circuits have not been able to satisfy this requirement.

(b) Various vertical noises such as impulsive noise from the analog telephone line are induced on the subscriber line 12 and cause code errors in the digital signal, so the unbalanced attenuation of the digital line termination device 11 is It needs to be high enough. In order to eliminate the drawbacks mentioned in paragraph (a) above, the second
As shown in the figure, an external capacitor 23 with sufficiently low impedance at the switching frequency of the DC-DC conversion circuit 17
Some devices suppress the common mode switching noise by connecting between the primary side 18a and the secondary side 19a (or between the primary side 18b and the secondary side 19b). However, when adopting such a configuration and connecting the secondary side 19a or 19b of the DC-DC conversion circuit 17 to the ground with low impedance, the power separation filter 16a (
Alternatively, a resonance point is formed in the vertical circuit by the coil for 16b) and the external capacitor 23, and at this resonance point, the unbalanced attenuation of the digital line termination circuit 11 is extremely degraded, causing a code error.

Therefore, it is necessary to set the resonance point at a frequency sufficiently higher than the pulse transmission band, and for this purpose, an external capacitor 23 is required.
is removed from the primary side 18a, 1 of the DC-DC conversion circuit 17.
It is necessary to separate the secondary side 8b and the secondary side 19a19b with a high impedance die approximately equal to the stray capacitance of the transformer 14.

However, in this case, in the conventional circuit configuration, the above (a
The disadvantages listed in section ) will occur.

These points will be explained in more detail below.

FIG. 3 shows the basic configuration of a conventional DC-DC conversion circuit.

The DC input voltage E input from the DC power supply 24 through the primary sides 18a and 18b is converted into an alternating voltage (hereinafter referred to as switching voltage) by the repeated on/off operation (hereinafter referred to as switching) of the switching element 25, A switching voltage e1 is generated between both ends 29,31 of the primary winding 27 of the transformer 26. Therefore, a switching voltage e2 is induced between both ends 32 and 33 of the secondary winding 28 of the transformer 26. This switching voltage e2 is applied to the diode 3
4, and is smoothed by the output capacitor 35 to obtain a DC output voltage E2, and this DC output is applied to the secondary side 19a.
, 19b to the load 36. Note that the switching voltage e induced between both ends 2931 of the primary winding 27
l and the switching voltage e2 induced between both ends 32 and 33 of the secondary winding 28 is the ratio between the primary winding 27 and the secondary winding 2.
It is determined by the winding ratio with 8. An input capacitor 37 is connected between both ends of the DC power supply 26, and the switching element 25 is, for example, a transistor, this transistor 25 is inserted in series with the primary winding 27, and a drive circuit is connected between the base and emitter of the transistor 25. 38 are connected.

The conventional DC-DC conversion circuit shown in FIG. 3 has a drawback in that a large switching voltage is generated between the primary side and the secondary side, that is, between the primary side 18a and the secondary side 19a. This is called common mode switching noise, and will be explained below using FIG. 4. FIG. 4 shows the generation mechanism of common mode switching noise in the DC-DC conversion circuit shown in FIG. 3, focusing on the alternating current component. The symbols in FIG. 4 correspond to those in FIG. 3, el + e2 + Vl and v2 indicate the switching voltages between the terminals at any given time, and the arrows indicate the mutual relationship between the voltage polarities. The connection point between the switch element 25 and the power supply 24 is the primary side 18b, and the connection point between the diode 34 and the output capacitor 35 is the secondary outside 19.
b. capacitor 42 between winding ends 29.32;
The ry capacitors 43 between the winding ends 31 and 33 each represent the stray capacitance distributed between the primary winding 27 and the secondary winding 28 using lumped constant circuits. Switch element 25
．． The closed circuit consisting of the primary winding 27 and the human power capacitor 37 is named closed circuit I, the closed circuit consisting of the primary winding 27, secondary winding 28, and capacitor 42.43 is named closed circuit ■, and the secondary winding 2
A closed circuit consisting of the diode 34, the output capacitor 35, and the output capacitor 35 is called a closed circuit (2).

In FIG. 4, attention is first paid to cycle I. Since the input capacitor 37 is short-circuited (sufficiently low impedance) to the switching frequency component, no switching voltage is generated across it. Therefore, from Kirchhoff's voltage law, both ends 29 of the primary winding 27 are connected to both ends of the switching element 25.
．． A switching voltage with an amplitude equal to the switching voltage e1 induced between 31 and 31 is generated in opposite phase. Next, the closed circuit ■
Focus on. Although the output capacitor 35 is short-circuited for the switching frequency component, no switching voltage is generated across it. Therefore, from Kirchhoff's voltage law, both ends 3 of the secondary winding 28 are connected to both ends of the diode 34.
A switching voltage with an amplitude equal to the switching voltage e2 induced between 2.3 and 33 is generated in opposite phase.

Next, we will focus on the cycle ■. Normally, the capacitance values of the capacitors 42 and 43 are both small, and the impedance is sufficiently high over high frequencies including the switching frequency. Now, closed circuit■
, the switching voltages generated across the capacitors 42 and 43 are V, respectively.

If v2, then from Kirchhoff's voltage law, V, +v2
=e, −e, is satisfied. Also, voltage V, and v2
The ratio of is inversely proportional to the capacitance value of each capacitor 42 and 43. The switching voltage (1) generated across the capacitor 42 is common mode switching noise. The following cycle■
This will be explained using FIG. 5, which is an enlarged view of the portion. The symbols in FIG. 5 correspond to those in FIG. Let the capacitance ilk values of capacitors 42 and 43 be C+ and Cz, respectively. From FIG. 5, there are C amplitude values v1 of the common mode switching noise, which are influenced by both the switching voltage e1 generated across the primary winding 27 and the switching voltage e2 induced across the secondary winding 28. arises from. normal DC-D
In the C conversion circuit, e, ≠ e2. Therefore, in order to isolate the primary sides 18a, 18b and the secondary sides 19a, 19b with high impedance and to reduce the occurrence of common mode switching noise, the capacitance value C2 of the capacitor 43 is required.
must be made sufficiently smaller than the capacitance value C1 of the capacitor 42. However, the capacitance value C2 of the capacitor 43 is determined by the shapes of the primary winding 27, secondary winding 28 and core of the transformer, and the capacitance value C2 cannot be made very small.

On the other hand, if the capacitance value CI of the capacitor 42 is increased (for example, by attaching an external capacitor), the common mode switching noise will be reduced;
The impedance between a19b is low. From the above, it is difficult with the conventional technology to isolate the primary sides 18a, 18b and the secondary sides 19a, 19b with high impedance and reduce common mode switching noise. - As an example, in the configuration shown in Fig. 3, the DC input voltage EI = 30V, the DC output voltage E2 = 5V, and the output voltage IW is about DC-DC.
In the conversion circuit, common-mode switching noise occurs as a ripple component of about 10 Vpp. However, first order.

This noise value differs slightly depending on the configuration of the secondary winding.

<Summary of the Invention> The purpose of the present invention is to provide a DC-DC conversion circuit that isolates the primary side and the secondary side with high impedance over a high frequency range including direct current and switching frequencies, and generates less common mode switching noise. It is about providing.

According to this invention, an electrostatic shielding layer is arranged between the primary winding and the secondary winding, and this electrostatic shielding layer is connected to the stationary end of the primary winding and inside the secondary winding. From the point of view, the voltage induced against the electrostatic shielding layer on one side of the secondary winding and the voltage induced against the electrostatic shielding layer on the other side of the primary winding have opposite polarity. A canceling means is provided.

<First Embodiment> FIG. 6 shows a first embodiment of the present invention, in which half-wave rectifying diodes 34a and 34b are connected in series to both ends of the secondary winding 28, respectively. The other ends of the diodes 34a and 34b are connected to both ends of the output capacitor 35. Primary winding 2
An electrostatic shielding layer 46 is interposed between the winding 7 and the secondary winding 2. This electrostatic shielding layer 46 serves as an alternating current zero of the primary winding 27.
It is connected to a winding end 29 which is a potential point (hereinafter referred to as a stationary end). The primary side 18b is also a stationary end, and the electrostatic shielding layer 4
6 is connected to the primary side 18b, the effect is the same. Other symbols follow FIG. 3. This structure has the effect of reducing the switching voltage generated between the primary side 18a and the secondary side 19a, that is, the common mode switching noise, as described below. According to the configuration shown in Part 6, (i) an electrostatic shielding layer 46 is provided between the primary winding 27 and the secondary winding 28, and this is the static end of the primary winding 27; Since the primary winding 27 is connected to the terminal 29, the switching voltage e generated across the primary winding 27 does not occur as a voltage (common mode switching noise) between the primary side 18a and the secondary side 19a. (ii) Diodes 34a and 34b for half-wave rectification are connected to both ends of the secondary winding 28, respectively, and the electrostatic shielding layer 46 - the winding end 32 (or 3
3) Switching voltage generated between the diode 34b
(or the winding ends 32 which are both ends of the diode 34a)
(or 33) - the switching voltage generated between the secondary side 19a (or 19b) has opposite polarity and cancels each other out, so that the switching voltage e2 generated across the secondary winding 28 is reduced to the primary side 18a-2. It does not occur as a voltage (common mode switching noise) between the next side 19a.

These will be explained in detail using FIG. 7.

FIG. 7 shows the effect of reducing common mode switching noise in the DC-DC conversion circuit shown in FIG. 6, focusing on the alternating current component. The symbols in FIG. 7 correspond to those in FIG. 6, e+ + e2+ v and v2 indicate the switching voltages between the terminals at any given time, and the arrows indicate the mutual relationship between the voltage polarities.

The capacitor 47 represents the stray capacitance distributed between the secondary winding 28 and the electrostatic shielding layer 46 using a lumped constant circuit.
It is connected between 6 and 6.

The capacitor 51 represents the stray capacitance distributed between the secondary winding 28 and the electrostatic shielding layer 46 using a lumped constant circuit, and the capacitor 51 is connected between the terminal 33 of the secondary winding 28 and the electrostatic shielding layer 46 Connected. Capacitors 52 and 53 represent the stray capacitance distributed between the primary winding 27 and the electrostatic shielding layer 46 using a lumped constant circuit, and the capacitors 52 and 53 are connected to both ends 29 and 31 of the primary winding 27 and the electrostatic shielding layer 46. layer 46. Capacitor 37. The closed circuit of the primary winding 27° switch element 25 is the closed circuit I, the closed circuit of the primary winding 27, the capacitors 52 and 53, and the electrostatic shielding layer 46 is the closed circuit ■, and the closed circuit ■ is the secondary winding. Wire 28, capacitors 47, 51
, the electrostatic shielding layer 46, and the closed circuit (■) is the secondary winding 28.
, diodes 34a, 34b and a capacitor 35. First, the above item (i) will be explained. In Figure 7,
First, let's focus on cycle I. Since the human power capacitor 37 is short-circuited (sufficiently low impedance) with respect to the switching frequency component, no switching voltage is generated across it. Therefore, according to Kirchhoff's voltage law, a switching voltage with an amplitude equal to the switching voltage e induced between both ends 29 and 31 of the primary winding 27 is generated at both ends of the switching element 25 in opposite phases. Next, we will focus on the cycle ■. Capacitors 52 and 53 connect the primary winding 27 and the electrostatic shielding layer 46.
The stray capacitance distributed between is expressed using a lumped constant circuit. Capacitor 52 is connected between winding end 29 and electrostatic shielding layer 46, and capacitor 53 is connected between winding end 3
1 and the electrostatic shield 1i46. Here, since both ends of the capacitor 52 are short-circuited by the electrostatic shielding layer 46, no switching voltage is generated. Therefore, from Kirchhoff's voltage law in the closed circuit ■, the capacitor 5
3, a switching voltage with an amplitude equal to the switching voltage e generated between the ends 29 and 31 of the primary winding 27 is generated in opposite phase. From the above, the switching voltage e1 generated on the primary side is closed only to the primary side by the electrostatic shielding layer 46, and does not affect the voltage between the primary side 18a and the secondary side 19a.

Next, the above item (ii) will be explained. In Fig. 7, focus on the cycle ■. Since the output capacitor 35 is short-circuited for the switching frequency component, no switching voltage is generated across it. Also, both ends 32 of the secondary winding 28.
A switching voltage e2 is induced between 33 and 33.

Here, from Kirchhoff's voltage law in the closed circuit ■, both ends of the secondary winding 2B are connected to both ends of the diodes 34a and 34b.
A switching voltage having an opposite phase to the switching voltage e2 induced between 2° and 33° is generated with the amplitude divided into two. Considering variations in diode characteristics, the amplitudes of the switching voltages generated across the diodes 34a and 34b are respectively □+△.

Let □−△.

Next, we will focus on the cycle ■. Capacitors 47 and 51 represent the stray capacitance distributed between the electrostatic shielding layer 46 and the secondary winding 28 using lumped constant circuits.
7 is connected between the electrostatic shielding layer 47 and the winding end 32,
A capacitor 51 is connected between the electrostatic shielding layer 46 and the winding end 33. Normally, the capacitance value of this capacitor 47 and 51 is the same.

It has a sufficiently high impedance over a high frequency range including the switching frequency. Now, capacitor 47,
The switching voltages generated across 51 are vl and v, respectively.
2, it satisfies the relationship V+ +Vz =ez from Kirchhoff's voltage law in the closed circuit ■. The explanation will be given below using FIG. 8, which is an enlarged view of the closed circuit (■) part. 8th
Symbols in the figure correspond to those in FIG. each capacitor 47.5
Let the capacitance values of 1 be C3 and C, respectively. Solving the cycle equation for the cycle ■ in Figure 8 yields C.

becomes. Now, let's go back to FIG. 7 and explain.

The common mode switching voltage is primary side 18a secondary side 19
This is the switching voltage that occurs between a and V3, and from FIG. In equation (4), the value of △ is sufficiently small and the common mode switching noise is determined by the capacitance value C3 of the capacitor 47.
The smaller the difference between the capacitance value C4 and the capacitance value C4 of the capacitor 51, the more the difference is reduced. Now, in FIG. 6, the technique of equalizing the stray capacitance between the electrostatic shielding layer 46 and the winding end 32 and the stray capacitance between the electrostatic shielding layer 46 and the winding end 33 is relatively easy; The physical position of winding end 32 with respect to layer 46 and the physical position of winding end 33 with respect to electrostatic shielding layer 46 may be symmetrical. For example, the secondary winding 28 may be configured to have one layer of winding around the electrostatic shielding layer 46. That is, in FIG. 8, the capacitance value C of the capacitor 47
3 and the capacitance value C4 of the capacitor 51,! Techniques for making : approximately equal are known. From the above, by connecting the half-wave rectifier diodes 34a and 34b to both ends of the secondary winding 2, the switching voltage e2 induced across the secondary winding 28 is changed from the primary side 18a to the secondary side 19a. I was able to explain that the effect on the voltage between the two can be reduced.

As described above, with the configuration shown in FIG. 6, the primary side 18a, 18b - the secondary side 1
It is possible to provide a DC-DC conversion circuit that isolates 9a and 19b with high impedance and generates less common mode switching noise.

In Figure 6, when the secondary side is a DC-DC conversion circuit with multiple outputs and multiple secondary windings, the primary winding and multiple secondary windings are sequentially formed on the coaxial center, and the secondary winding is An electrostatic shielding layer is also provided between the windings, and the electrostatic shielding layer in the parentheses is connected to the electrostatic shielding layer 46 between the primary winding and the secondary winding to reduce common mode switching noise. This is advantageous in reducing the occurrence of

<Second Embodiment> The above is a so-called current transmission type, that is, 1 in Fig. 6.
A type D in which the secondary side half-wave rectifier diodes 34a and 34b are conductive when the next side switch element 25 is off.
This invention was applied to a C-DC conversion circuit, but it is a so-called voltage transmission type, that is, when the primary side switch element is on, the
A type of DC- in which the diode for half-wave rectification on the next side is conductive.
The present invention can also be applied to a DC conversion circuit. FIG. 9 shows a voltage transmission type D(, -DC conversion circuit) of the present invention corresponding to FIG. have the same configuration.

Effects> As explained above, the present invention provides a DC-DC system that isolates the primary side and secondary side with high impedance over a high frequency range including direct current and switching frequencies, and generates little common-mode switching noise. Because it can provide a conversion circuit, it can be applied as a power receiving power source for digital line termination equipment installed inside the subscriber's premises that operates by distant power supply from the station in digital subscriber line transmission systems using balanced cables. There are advantages. Specifically, DC-DC
Switching noise generated by the conversion circuit is less likely to enter the pulse transmission system circuit, and high impedance separation between the digital line termination device and the subscriber line is possible. The effect of this is that a high unbalanced attenuation of the digital line termination device can be obtained in the pulse transmission band, and code errors in the digital signal can be suppressed as much as possible against large vertical noise induced on the subscriber line.

A numerical example is shown next. In the configuration shown in FIGS. 6 and 9, the input voltage E1 is approximately 26V, and the input current is approximately 24mA.
, output voltage E2 is 5■±3.5%, output power is approximately 500
mW, the number of turns of the primary winding 27 is about 80, the number of turns of the secondary winding 2
8, the number of windings is about 16, the electrostatic shielding layer 46 is copper foil, the switch element 25 is a MOS-FET, and the rectifier diode 3
4a and 34b are Schottky barrier diodes, the transformer secondary winding 28 is wound in one layer around the electrostatic shielding layer 46,
In a DC-DC conversion circuit with a switching frequency of approximately 70 KHz, the common mode switching noise was approximately 0.5 Vpp as a ripple component, and the power conversion efficiency was approximately 80%. Note that the common-mode switching noise was the same even if the switch element drive circuit 38 was a passive type or a self-excited type.

[Brief explanation of the drawing]

Figures 1 and 2 are diagrams showing the configuration of a digital line termination device installed inside a subscriber's premises, respectively. Figure 3 is a connection diagram showing the basic configuration of a conventional current transmission type DC-DC conversion circuit; 4 is an explanatory diagram of the common mode switching noise generation mechanism in the DC-DC conversion circuit of FIG. 3, FIG. 5 is an enlarged view of the closed circuit ■ part of FIG. 4, and FIG. A connection diagram showing an embodiment applied to a DC conversion circuit, FIG. 7 is an explanatory diagram of the common mode switching noise reduction effect in the DC-DC conversion circuit of FIG. 6, and FIG. The enlarged view, FIG. 9, is a connection diagram showing another embodiment in which the present invention is applied to a voltage transmission type DC-DC conversion circuit. 24... DC power supply, 25... Switch element, 26.
...Transformer, 27...Primary winding, 28...Secondary winding, 34a, 34b...Diode for half-wave rectification, 3
5... Output capacitor, 36... Load, 37...
Input capacitor, 38... switch element drive circuit,
46... Electrostatic shielding layer.

Claims (1)

[Claims] In a DC-DC conversion circuit that switches and supplies DC power to the primary winding of a transformer and rectifies and smoothes the output of the secondary winding of the transformer to obtain a DC output, the primary winding and the secondary winding an electrostatic shielding layer connected to the stationary end of the primary winding, and a first half-wave rectifier and a second half-wave rectifier at both ends of the secondary winding, respectively. D characterized in that diodes are connected in series.
C-DC conversion circuit.