JPH0113309B2 - - Google Patents

Description

[Detailed Description of the Invention] This invention relates to a DC-DC conversion circuit that switches DC power and supplies it to the primary winding of a transformer, rectifies and smoothes the output of the secondary winding of the transformer, and obtains a DC output. In particular, it relates to a circuit that separates input and output with high impedance and generates little common-mode switching noise.

<Background> This type of DC-DC conversion circuit is used, for example, as a power source for a digital line termination device installed inside a subscriber's premises in a digital subscriber line transmission system. That is, as shown in FIG. 1, the digital line termination device 11 installed inside the subscriber's premises is generally configured to operate by distant power supply from the station, and the balanced type connected to the digital line termination device 11 is A digital signal and a feeding direct current are superimposed on the subscriber line 12 of the cable. Although the other end of the subscriber line 12 is not shown in the figure, it is led into the office,
Connected to the digital line termination equipment inside the office.
A digital signal on the subscriber line 12 is input to a pulse transmission circuit 15 via a connection point 13 with the device 11 and a transformer 14. The pulse transmission circuit 15 includes an equalization amplifier circuit, a pulse transmission circuit, and the like.
The feed DC current on the subscriber line 12 is DC-DC through the connection point 13 and the power separation filters 16a, 16b.
The voltage is input to the primary sides 18a and 18b of the conversion circuit 17, and a DC voltage of 30 V, for example, is applied between the primary sides 18a and 18b. DC-DC conversion circuit 17
Now perform the DC-DC conversion and create the DC-DC conversion circuit 17.
For example, a DC voltage is applied between the secondary sides 19a and 19b of
Generates 5V. The main part or the whole of the digital line termination device 11 is operated by the outputs of the secondary sides 19a and 19b of the DC-DC conversion circuit 17.
A DC blocking capacitor 21 inserted between the connection point 13 and the connection point between the filters 16a and 16b and the transformer 14 is provided to prevent the supply DC current from flowing through the transformer 14. Power separation filter 16
a and 16b are constructed of, for example, coils so as to provide low impedance for direct current and high impedance for alternating current. This is to avoid short-circuiting the digital signals since the AC impedance between the primary sides 18a and 18b of the DC-DC conversion circuit 17 is low.

Now, the digital line termination device 1 with the above configuration.
When a conventionally configured DC-DC converter circuit is applied to the DC-DC converter circuit 17, which is the power source of No. 1, the following drawbacks occur.

(a) Between the primary side and the secondary side of the DC-DC conversion circuit 17, that is, the primary side 18a and the secondary side 19a
between the primary side 18b and the secondary side 19b
During this period, switching noise _v1 , so-called common mode switching noise, occurs due to switching of the DC-DC conversion circuit 17. This common mode switching noise is converted into differential mode sound according to the amount of unbalanced attenuation determined by the subscriber line 12, digital line termination device 11, etc.
It goes around between the secondary sides 22a and 22b of the transformer 14, causing a code error in the digital signal. For this reason, it is necessary to sufficiently reduce the generation of common-mode switching noise, but conventional
A DC-DC conversion circuit could not satisfy this requirement.

(b) Various types of vertical noise such as impulsive noise from the analog telephone line are induced on the subscriber line 12 and cause code errors in the digital signal, so the unbalanced attenuation of the digital line termination device 11 is It needs to be high enough. In order to eliminate the drawback described in the above item (a), as shown in FIG.
Some devices suppress the common-mode switching noise by connecting 3 between the primary side 18a and the secondary side 19a (or between the primary side 18b and the secondary side 19b). However, with such a configuration and the second part of the DC-DC conversion circuit 17.
When connecting the next side 19a or 19b to earth with low impedance, power isolation filter 1
A resonance point is formed in the vertical circuit from the coil for 6a (or 16b) and the external capacitor 23, and at this resonance point, the digital line termination device 11
The unbalanced attenuation of the signal is extremely degraded, resulting in code errors. For this reason, it is necessary to set the resonance point to a frequency sufficiently higher than the pulse transmission band, and for this purpose, the external capacitor 23 is removed and the primary side 18a, 18b and secondary side of the DC-DC conversion circuit 17 are 1
9a and 19b must be separated by a high impedance comparable to the stray capacitance of the transformer 14. However, in this case, the conventional circuit configuration suffers from the drawbacks described in item (a) above.

These points will be explained in more detail below.

FIG. 3 shows the basic configuration of a conventional DC-DC conversion circuit. The DC input voltage _E1 input from the DC power supply 24 through the primary sides 18a and 18b is converted into an alternating voltage (hereinafter referred to as switching voltage) by the repeated on/off operation (hereinafter referred to as switching) of the switch element 25, A switching voltage is applied between both ends 29 and 31 of the primary winding 27 of the transformer 26.
e ₁ occurs. Therefore, the secondary winding 2 of the transformer 26
A switching voltage e ₂ is induced between both ends 32 and 33 of 8. This switching voltage _e2 is half-wave rectified by a diode 34 and smoothed by an output capacitor 35 to obtain a DC output voltage _E2 , which is supplied to a load 36 through the secondary sides 19a and 19b. Note that the switching voltage _e 1 induced between both ends 29 and 31 of the primary winding 27 and the secondary winding 28
The ratio to the switching voltage e ₂ induced between both ends 32 and 33 of is determined by the winding ratio between the primary winding 27 and the secondary winding 28. An input capacitor 37 is connected between both ends of the DC power supply 24, and a switch element 2
5 is a transistor, for example, and this transistor 25 is inserted in series with the primary winding 27, and the drive circuit 3 is connected between the base and emitter of the transistor 25.
8 are connected.

In the conventional DC-DC conversion circuit shown in FIG. 3, between the primary side and the secondary side, that is, the primary side 18a,
The disadvantage was that a large switching voltage was generated between the next side 19a. This is called common mode switching noise, and will be explained below using FIG. 4. Fourth
The figure shows the DC-DC conversion circuit shown in Figure 3.
This figure shows the generation mechanism of common-mode switching noise, focusing on the AC component. The symbols in Fig. 4 are the same as in Fig. 3, e ₁ , e ₂ , v ₁ and v ₂ indicate the switching voltage between each terminal at any given time, and the arrows indicate the mutual relationship of each voltage polarity. . Switch element 25
The connection point between the diode 34 and the power supply 24 is the primary side 18b, and the connection point between the diode 34 and the output capacitor 35 is the secondary side 19b. Capacitor 42 between winding ends 29 and 32, capacitor 43 between winding ends 31 and 33
are the stray capacitances distributed between the primary winding 27 and the secondary winding 28, respectively, expressed by lumped constant circuits. switch element 25, primary winding 27,
The closed circuit consisting of the input capacitor 37 is called a closed circuit, the closed circuit consisting of the primary winding 27, the secondary winding 28, and the capacitors 42 and 43 is called a closed circuit, and the closed circuit consisting of the secondary winding 28, the diode 34, and the output capacitor 35 is called a closed circuit. Each closed circuit is called a closed circuit.

In FIG. 4, first focus on the closed circuit. Since the input capacitor 37 is short-circuited (sufficiently low impedance) for the switching frequency component, no switching voltage is generated across it. Therefore, from the Kirchoff voltage side, the switch element 25
A switching voltage with an amplitude equal to the switching voltage e ₁ induced between both ends 29 and 31 of the primary winding 27 is generated at both ends of the primary winding 27 in opposite phases. Next, we will focus on closed circuits. Since the output capacitor 35 is short-circuited for the switching frequency component, no switching voltage is generated across it. Therefore, a switching voltage with an amplitude equal to the switching voltage e ₂ induced between the ends 32 and 33 of the secondary winding 28 is generated in opposite phase across the diode 34 from the Kirchhoff voltage side.

Next, we will focus on closed circuits. Normal capacitor 42,
The capacitance values of 43 are both small, and the impedance is sufficiently high over a high frequency range including the switching frequency. Now, in a closed circuit, capacitor 4
If the switching voltages generated at both ends of 2 and 43 are v ₁ and v ₂ , respectively, then from the Kirchhoff voltage side
It satisfies the relationship v ₁ + v ₂ = e ₁ − e ₂ . Also the voltage v ₁
and v ₂ is inversely proportional to the capacitance value of each capacitor 42, 43. The switching voltage v ₁ generated across capacitor 42 is common mode switching noise. The following description will be made with reference to FIG. 5, which is an enlarged view of the closed circuit portion. The symbols in FIG. 5 correspond to those in FIG. 4. The capacitance values of capacitors 42 and 43 are C ₁ ,
Let it be C ₂ . From FIG. 5, the amplitude value v ₁ of the common mode switching noise is v ₁ = C ₂ /C ₁ +C ₂ (e ₂ - e ₁ ) (1), and the switching voltage e ₁ generated across the primary winding 27 is and the switching voltage e ₂ induced across the secondary winding 28.
In a normal DC-DC conversion circuit, e ₁ ≠ e ₂ . Therefore, the primary side 18a, 18b, the secondary side 19
In order to isolate between a and 19b with high impedance and to reduce the occurrence of common mode switching noise, it is necessary to make the capacitance value _C2 of the capacitor 43 sufficiently smaller than the capacitance value _C1 of the capacitor 42. . However, the capacitance value C ₂ of the capacitor 43
is determined by the shapes of the primary winding 27, secondary winding 28, and core of the transformer, and the capacitance value _C2 cannot be made very small. On the other hand, the capacitance value C ₁ of the capacitor 42
If you increase (for example, by adding an external capacitor),
Although the common mode switching noise becomes small, it forms a resonance point with the separation filter, and the primary side 18a, 1
8b and the secondary sides 19a and 19b have low impedance and have different unbalanced attenuation amounts. From the above, it has been difficult with the prior art to isolate the primary sides 18a, 18b and the secondary sides 19a, 19b with high impedance and reduce common mode switching noise. As an example, in a DC-DC converter circuit with the configuration shown in Figure 3, with a DC input voltage E ₁ = 30V, a DC output voltage E ₂ = 5V, and an output power of about 1W, the common-mode switching noise is a ripple component and is approximately
Approximately 10Vpp is generated. However, this noise value differs slightly depending on the configuration of the primary and secondary windings.

<Summary of the Invention> The purpose of the present invention is to provide a DC-DC conversion circuit that isolates the primary side and the secondary side with high impedance over a high frequency range including DC and switching frequencies, and that generates little common mode switching noise. Our goal is to provide the following.

According to this invention, an electrostatic shielding layer is arranged between the primary winding and the secondary winding, and this electrostatic shielding layer is connected to the stationary end of the primary winding and inside the secondary winding. From the point of view, the voltage induced against the electrostatic shielding layer on one side of the secondary winding and the voltage induced against the electrostatic shielding layer on the other side of the primary winding have opposite polarity. A canceling means is provided.

<First Embodiment> FIG. 6 shows a first embodiment of the present invention. In this embodiment, the secondary winding 28 is open at its midpoint and 2
The next windings 28a, 28b are the open ends 44,
A diode 34 for half-wave rectification is connected between 45 and 45 to constitute canceling means. Also, the primary windings 27 and 2
Electrostatic shielding layer 46 between the next windings 28a and 28b
is interposed, and this electrostatic shielding layer 46 is connected to the primary winding 2.
It is connected to the primary side 18a which is the AC zero potential point (hereinafter referred to as the stationary end) of No.7. The primary side 18b is also a stationary end, and the electrostatic shielding layer 46 is connected to the primary side 18b.
The effect is the same even if connected to . Other symbols follow Figure 3. Since it has such a configuration, as explained below, the primary side 18a-secondary side 19
This has the effect of reducing the switching voltage generated between terminals a, that is, common mode switching noise.

In the case of the conventional DC-DC conversion circuit shown in Figure 3, the common mode switching noise is calculated from equation (1) as 1
This occurs due to the effects of both the switching voltage e ₁ generated across the secondary winding 27 and the switching voltage e ₂ induced across the secondary winding 28 . However, in the embodiment shown in FIG. 6, (i) an electrostatic shielding layer 46 is provided between the primary winding 27 and the secondary windings 28a, 28b, and this is the stationary end of the primary winding 27; By being connected to the primary side 18a, the switching voltage _e1 generated across the primary winding 27 does not occur as a voltage between the primary side 18a and the secondary side 19a (common mode switching noise). (ii) Also, by inserting and connecting the half-wave rectifying diode 34 to the middle point of the secondary winding 28, the electrostatic shielding layer 46
The potential distribution of the switching voltage between the electrostatic shielding layer 46 and the secondary winding 28a and the potential distribution of the switching voltage between the electrostatic shielding layer 46 and the secondary winding 28b have opposite polarities, cancel each other out, and the secondary winding 28a, 28
The switching voltage e ₂ /2 generated across each end of the signal b does not occur as a voltage (common mode switching noise) between the primary side 18a and the secondary side 19a.

These points will be explained in more detail using FIG. 7. FIG. 7 shows the effect of reducing the common mode switching noise in the DC-DC conversion circuit shown in FIG. 6, focusing on the alternating current component. The symbols in FIG. 7 are the same as in FIG. 5, and e ₁ , e ₂ and v ₁ represent the switching voltages between the respective terminals at any given time, and the arrows represent the mutual relationship between the voltage polarities. The capacitors 47 and 48 are connected to the secondary winding 28a and the electrostatic shielding layer 46.
The stray capacitance distributed between is represented by a lumped constant circuit, and capacitors 47 and 48 are connected between both ends 32 and 44 of the secondary winding 28a and the electrostatic shielding layer 46, respectively. Capacitors 49 and 51 are 2
Stray capacitance distributed between the secondary winding 28b and the electrostatic shielding layer 46 is represented by a lumped constant circuit, and capacitors 49 and 51 are connected between both ends 45 and 33 of the secondary winding 28b and the electrostatic shielding layer 46, respectively. connected to. The capacitors 52 and 53 represent the stray capacitance distributed between the primary winding 27 and the electrostatic shielding layer 46 using lumped constant circuits.
Both ends 29 and 31 of the next winding 27 and the electrostatic shielding layer 46
connected between. The closed circuit of the capacitor 37, the primary winding 27, and the switch element 25 is a closed circuit,
The closed circuit of the primary winding 27, the capacitors 52 and 53, and the electrostatic shielding layer 46 is a closed circuit, and the secondary winding 28
a, the closed circuit of the capacitors 47, 48 and the electrostatic shielding layer 46 is a closed circuit, the closed circuit of the diode 34, the capacitors 48, 49, and the electrostatic shielding layer 46 is a closed circuit, the secondary winding 28b, the capacitors 49, 5
1. The closed circuit of the electrostatic shielding layer 46 is a closed circuit, and 2.
A closed circuit of the secondary windings 28a, 28b, the diode 34, the capacitors 47, 51, and the electrostatic shielding layer 46 is a closed circuit, and a closed circuit of the secondary windings 28a, 28b, the diode 34, and the output capacitor 35 is a closed circuit.

First, the above item (i) will be explained. In Figure 7,
First, let's focus on closed circuits. Since the input capacitor 37 is short-circuited (sufficiently low impedance) to the switching frequency component, no switching voltage is generated across it. Therefore, according to Kirchhoff's voltage law, a switching voltage with an amplitude equal to the switching voltage e ₁ induced between the ends 29 and 31 of the primary winding 27 is generated at both ends of the switch element 25 in opposite phases. Next, we will focus on closed circuits. Since both ends of the capacitor 52 are short-circuited by the electrostatic shielding layer 46, no switching voltage is generated. Therefore, according to Kirchhoff's voltage law in a closed circuit, a switching voltage with an amplitude equal to the switching voltage e ₁ generated between both ends 29 and 31 of the primary winding 27 is generated at both ends of the capacitor 53 in opposite phases. Switching voltage generated on the primary side from the above
e ₁ is closed only on the primary side by the electrostatic shielding layer 46 1
It was explained that the voltage between the secondary side 18a and the secondary side 19a is not affected.

Next, the above item (ii) will be explained. In FIG. 7, attention is paid to the closed circuit. Since the output capacitor 35 is short-circuited for the switching frequency component,
No switching voltage is generated across it. 2
Since the secondary winding 28 is divided into two parts at the midpoints 44 and 45, between both ends 32 and 44 of the secondary winding 28a,
Switching voltages e ₂ /2 of equal amplitude and in phase are induced between both ends 33 and 45 of the next winding 28b. Here, from the Kirchhoff voltage side in the closed circuit, between both ends 32 and 44 of the secondary winding 28a, between both ends 45 of the secondary winding 28b,
A switching voltage with an amplitude equal to the sum e ₂ of the switching voltages induced between 33 and 33 is generated in opposite phase.

Next, we will focus on cycles and. Normally, the capacitance values of capacitors 47, 48, 49, and 51 are all small, and the impedance is sufficiently high over a high frequency range including the switching frequency. Also, usually 2
By bifilar winding with the next windings 28a, 28b and their one ends 32, 45 on the same side,
The physical positional relationship between the electrostatic shielding layer 46 and the winding end 32 is equal to that between the electrostatic shielding layer 46 and the winding end 45. Therefore, the capacitance values of capacitor 47 and capacitor 49 are approximately equal, and this is designated as _C3 .
Similarly, the physical positional relationship between the electrostatic shielding layer 46 and the winding end 44 and between the electrostatic shielding layer 46 and the winding point 33 are the same. Therefore, the capacitance values of capacitor 48 and capacitor 51 are approximately equal, and this is designated as _C4 .

Here, we will focus on closed circuits. Assume that a switching voltage v ₁ is generated across the capacitor 47. Since the voltages of the secondary windings 28a and 28b mentioned above and the voltage of the diode 34 are equal and in opposite phases, from Kirchhoff's voltage law, the switching voltage v ₁ generated at both ends of the capacitor 47 is applied to both ends of the capacitor 51. Switching voltages of amplitude equal to are generated in opposite phases. This switching voltage v ₁ is the same layer mode switching voltage. Next, it will be explained using FIG. FIG. 8 is an abstraction of the circuit and parts in FIG. 7, and the symbols are the same as in FIG. 7. It is assumed that the capacitance values of the capacitors 47 and 49 are C ₃ and the capacitance values of the capacitors 47 and 51 are C ₄ . Solving the cycle equations for the cycles and in Figure 8, v ₁ = C ₄ − C 3 /C ₃ +C _{4 e 2} /4 (2) v ₂ =3C ₃ +C ₄ /C ₃ +C ₄ e ₂ / 4 (3) v ₃ =C ₃ +3C ₄ /C ₃ +C ₄ e ₂ /4 (4). From equation (2), the common mode switching noise is reduced as the difference between the capacitance value C ₃ of the capacitors 47 and 49 and the capacitance value C ₄ of the capacitors 48 and 51 is smaller. Now, in FIG. 6, the techniques for equalizing the stray capacitance between the electrostatic shielding layer 46 and the winding end 32 (or 45) and the stray capacitance between the electrostatic shielding layer 46 and the winding end 44 (or 33) are compared. The physical position of the winding end 32 (or 45) with respect to the electrostatic shielding layer 46 and the physical position of the electrostatic shielding layer 46 - the winding end 44 (or 33) may be made symmetrical. . For example, the secondary winding 28
a, 28b may be configured to be wound in one layer around the electrostatic shielding layer 46. That is, in FIG. 8, capacitor 47 (or capacitor 4
9) and _{the capacitance value C 4 of} the capacitor 48 (or capacitor 51) are known. From the above, by inserting and connecting the diode 34 for half-wave rectification to the middle point of the secondary winding 28, the switching voltage e ₂ /2 induced across the secondary windings 28a and 28b is changed to the primary side 18a- It has been explained that the influence on the voltage between the secondary side 19a can be reduced.

As described above, with the configuration shown in FIG. 6, the primary side 18
A, 18b and the secondary sides 19a, 19b are separated by high impedance, and a DC-DC conversion circuit that generates less common mode switching noise can be provided.

In FIG. 6, when the secondary side is a DC-DC converter circuit with multiple outputs and multiple secondary windings, the primary winding and multiple secondary windings are sequentially formed on the same axis,
By providing an electrostatic shielding layer between the secondary windings and connecting this electrostatic shielding layer to the electrostatic shielding layer 46 between the primary winding and the secondary winding, common mode This is advantageous in reducing the occurrence of switching noise.

<Second Embodiment> The above is a so-called current transmission type, that is, in FIG. 6, when the primary side switch element 25 is off,
The present invention was applied to a DC-DC conversion circuit in which the half-wave rectifier diode 34 on the secondary side conducts, but it is a so-called voltage transmission type, that is, when the primary side switch element is on, the The present invention can also be applied to a DC-DC conversion circuit in which a diode for wave rectification is conductive. FIG. 9 shows a voltage transmission type DC-DC conversion circuit of the present invention corresponding to FIG. 6. In these cases, the rectifier diode 34
The polarity is opposite to that in the case of FIG. 6, and the other configurations are the same.

<Effects> As explained above, the present invention provides a DC switching system that isolates the primary side and secondary side with high impedance over a high frequency range including DC and switching frequencies, and generates less common mode switching noise.
-Since we can provide DC conversion circuits, we can provide
It has an advantage in application as a power receiving power source for a digital line termination device installed inside a subscriber's premises that operates by supplying power from a remote station. Specifically, DC
- Switching noise generated by the DC conversion circuit is less likely to enter the pulse transmission system circuit, and high impedance separation between the digital line termination device and the subscriber line is possible. The effect of this is that a high unbalanced attenuation of the digital line termination device can be obtained in the pulse transmission band, and code errors in the digital signal can be suppressed as much as possible against large vertical noise induced on the subscriber line.

A numerical example is shown next. In the configuration shown in Figure 6, the input voltage _E1 is approximately 26V, the input current is approximately 24mA,
The output voltage _E2 is 5V±3.5%, the output power is about 500mW, the number of turns of the primary winding 27 is about 80, the secondary winding 28a,
The number of turns of the wire 28b is about 8 times each (bifilar winding), the electrostatic shielding layer 46 is copper foil, the switch element 25 is a MOS-FET, and the rectifier diode 3 is used.
4 is a shot key barrier diode, transformer 2
The next windings 28a and 28b are wound in one layer around the electrostatic shielding layer 46, and the switching frequency is approximately 70KHz.
In the DC-DC conversion circuit, the common mode switching noise was approximately 0.5Vpp as a ripple component, and the power conversion efficiency was approximately 80%. Note that the common mode switching noise was the same whether the switch element drive circuit 38 was a separately excited type or a self-excited type.

[Brief explanation of drawings]

Figures 1 and 2 are diagrams showing the configuration of a digital line termination device installed inside a subscriber's premises, respectively. Figure 3 is a connection diagram showing the basic configuration of a conventional current transmission type DC-DC conversion circuit. Figure 4 is the DC of Figure 3.
- An explanatory diagram of the common-mode switching noise generation mechanism in a DC conversion circuit, Figure 5 is an enlarged view of the closed circuit part of Figure 4, and Figure 6 shows the current transmission type of this invention.
A connection diagram showing an example applied to a DC-DC conversion circuit, Fig. 7 is an explanatory diagram of the common mode switching noise reduction effect in the DC-DC conversion circuit of Fig. 6, and Fig. 8 is a closed circuit diagram of Fig. 7. FIG. 9 is a connection diagram showing an embodiment in which the present invention is applied to a voltage transmission type DC-DC conversion circuit. 24: DC power supply, 25: Switch element, 26:
Transformer, 27: Primary winding, 28: Secondary winding, 2
8a, 28b: Secondary winding divided into two at the midpoint, 3
4: Diode for half-wave rectification, 35: Output capacitor, 36: Load, 37: Input capacitor, 3
8: Switch element drive circuit, 46: Electrostatic shielding layer.

Claims (1)

[Claims] 1. In a DC-DC conversion circuit that switches and supplies DC power to the primary winding of a transformer and rectifies and smoothes the output of the secondary winding of the transformer to obtain a DC output, the primary winding and the secondary an electrostatic shielding layer connected to the stationary end of the primary winding between the winding and the secondary winding being open at its midpoint, and having an electrostatic shielding layer between the open terminals for the rectification. It is characterized by a diode connected for half-wave rectification.
DC-DC conversion circuit.