JPH06284511A - Electric vehicle torque control method - Google Patents

Abstract

(57) [Summary] [Purpose] Stabilize driving even if torque fluctuations occur due to sudden acceleration, steep slope climbing, disturbances applied to the axle due to road surface irregularities, etc., and improve battery energy utilization efficiency per charge. We propose torque control for electric vehicles to improve mileage. [Structure] Torque command value calculation means 80, three-phase induction motor 40
Motor torque detection means 110 for detecting the generated torque, torque compensation means 90 for compensating the motor torque so that the motor torque matches the torque command value, torque current command value calculation means 100 for calculating the torque current command value from the torque compensation amount, torque current Slip frequency calculation means 10 for calculating the slip frequency from the command value
5, speed detection means 60 for measuring the angular velocity of the electric motor 40, secondary magnetic flux setting means for generating a secondary magnetic flux (rated value) corresponding to the angular velocity
130, secondary magnetic flux command value calculating means 140 for calculating the secondary magnetic flux command value from the secondary magnetic flux and the torque current command value;
A torque control method characterized by a secondary magnetic flux estimating means 170 and an exciting current command value determining means 160 for determining an exciting current so that they coincide with each other.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a torque control method for an electric vehicle, and more particularly to a torque control method for an electric vehicle suitable for driving a traveling drive induction motor mounted on the vehicle with a battery.

[0002]

2. Description of the Related Art For an electric vehicle, it is important to improve the energy utilization efficiency of the battery and increase the mileage for one charge. As a technique related to this, Japanese Patent Laid-Open No. 62-250803 is known. .

According to this content, a motor loss map for the motor magnetic flux is prepared in advance, and the motor loss is reduced based on the torque command value determined from the speed of the motor and the depression amount of the accelerator. It is known to determine a motor magnetic flux command value, calculate a torque current command value from the motor magnetic flux command value, and control the motor based on these command values to reduce the motor loss.

[0004]

The torque generated in the motor is proportional to the product of the secondary magnetic flux (motor magnetic flux) generated inside the motor (secondary circuit) and the torque current flowing in the secondary circuit. Therefore, when the magnetic flux command value is changed for the purpose of reducing the motor loss, the magnetic flux command value changes when the driving condition changes even if the torque command value is the same. Along with this, the torque current command value also changes, and the torque current flowing through the secondary circuit of the motor changes. In particular, when the load torque or the acceleration torque is large, the torque current is also large. Therefore, the sensitivity of the torque current to the disturbance is increased, and there is a concern that the amount of change in the torque current to the disturbance is also increased.

Further, when the magnetic flux (exciting current) command value is changed depending on driving conditions such as a torque command value and a motor speed,
Since the exciting current varies, the secondary magnetic flux generated in the secondary circuit is not constant and mutual interference with the torque current occurs. As a result, the torque generated by the motor vibrates, and this acts as a vibration source to vibrate the vehicle body.

It is an object of the present invention to generate a desired torque with a minimum power, and further, even if a torque current becomes excessively large, such as when climbing a slope, sudden acceleration or heavy load, the torque command value is not changed. A torque control method for an electric vehicle that can stably generate torque based on the above. Another object of the present invention is to provide a torque control method for an electric vehicle that can improve the efficiency of use of energy of a battery and improve the mileage per charge.

[0007]

The present invention provides a battery for supplying a DC voltage and a DC current, a PWM inverter for converting the DC voltage into an AC power source of variable frequency and variable voltage, and driven by the PWM inverter. An electric vehicle comprising an induction motor and having a control circuit for controlling a torque to be generated by the induction motor according to a torque command value determined based on an accelerator depression amount, and controlling a vehicle speed as desired. And a torque current command for the induction motor determined so as to compensate for the deviation between the torque command value and the torque generated by the induction motor obtained from the torque detection means. This is achieved by controlling the torque generated in the induction motor based on the value and the exciting current command value.

[0008]

The motor torque currently generated inside the motor is detected from the motor torque detecting means. When the secondary magnetic flux is not constant and the generated torque fluctuates, the detecting means detects the motor torque including the fluctuation component.
When the generated torque fluctuates and an error occurs between the torque command value and the generated torque, the motor torque compensating means compensates the torque command value so as to suppress the vibration of the generated torque, and generates the torque command value compensation value. . The torque command value compensation value and 2
The torque current command value is calculated from the torque current command value calculation unit based on the estimated secondary magnetic flux obtained from the secondary magnetic flux estimation unit. The secondary magnetic flux command calculation means obtains the load factor from the ratio of the torque current command value and the rated torque current, and the load factor is 2
A secondary magnetic flux command is obtained by multiplying the secondary magnetic flux setting value obtained from the secondary magnetic flux setting value determining means. The exciting current command value determining means performs a compensating operation for determining the exciting current command value such that the estimated secondary magnetic flux obtained from the secondary magnetic flux estimating means matches the secondary magnetic flux command value. The torque current command value and the exciting current command value obtained from the above operation are corrected so as to suppress the vibration of the generated torque and further generate the desired motor torque with the minimum primary current (motor current). The result of is reflected in the primary current command value and the primary current is controlled based on the primary current command value. Therefore, the torque control is stable even if the driving condition is changed, and the battery power is efficiently used. It can be performed.

[0009]

FIG. 1 shows an embodiment of the present invention. A drive system of an electric vehicle includes a battery 10, a PWM inverter 20 for converting a DC voltage of the battery 10 into an AC power source of variable frequency and variable voltage, a three-phase induction motor 40 driven by the AC power source, and an induction motor 40. The torque to be transmitted is transmitted to and driven by wheels (not shown in the figure). Induction motor 40
Is generated by the encoder 50 and the speed detecting means 60 connected to the rotor of the induction motor 40 and the three-phase AC current flowing in the primary winding of the induction motor 40 detected by the current sensors 30, 31, 32. The detected rotational angular velocity of the rotor is used for control by the following control means.

The torque command value calculating means 80 has an accelerator 7
Based on the amount output from 0, a torque command value τR for causing the three-phase induction motor 40 to generate the torque required to move the vehicle is calculated.

When the voltage of the battery is within the predetermined voltage range, τR is not corrected, but when the voltage drops below a predetermined voltage, τR is decreased according to the decrease. This is to prevent the battery from being over-discharged and at the same time prevent the PWM inverter 20 from being over-current. The predetermined voltage depends on the battery used, but usually,
A value larger than the final discharge voltage at which the capacity can be restored by charging is selected.

The voltage (primary voltage) obtained from the PWM inverter 20 is proportional to the battery voltage. On the other hand, the torque generated in the three-phase induction motor 40 is generated in proportion to the square of the primary voltage. Therefore, the torque command value τ R when the battery voltage decreases from the predetermined value is corrected so as to decrease in proportion to the square of the battery voltage decrease from the predetermined voltage.

Although not shown in the embodiment of FIG. 1,
If the magnitude of the primary voltage command value obtained from the current control means 180 described later is reduced to a predetermined value or less by the battery voltage, the same effect can be obtained even if it is corrected by a similar method.
Further, since the PWM inverter 20 becomes overcurrent in such a state, the magnitude of the primary current command value obtained from the primary current command value calculating means 120 may be reduced if such a state occurs.

The torque command value τR is input to the plus side terminal of the subtracter 150, and the motor torque τ obtained from the following calculation is input to the minus side terminal, and the difference Δτ between the two is taken. τ = m · p · {lm ′ / (lm ′ + l ₂ )} · {(lm ′ · Im) / (1 + T ₂ · s)} · It (1) where m: number of phases, p: induction motor The number of pole pairs of, lm ',
l ₂ : Excitation and secondary leakage inductance, T ₂ (= (lm '
+ L ₂ ) / r2, r2: secondary resistance), s: Laplace operator. Im and It are current sensors 30, 31, and 3 respectively.
The U-phase AC current, the V-phase AC current, and the W-phase AC currents iu, iv, and iw detected from the PWM inverter 20 are detected.
Is a d-axis component and a q-axis component obtained by conversion into a d-q axis coordinate system that rotates at a speed that is synchronized with the angular frequency ω 1. Im = iu · cos θ1 * + iv · cos (θ1 * −2π / 3) + iw · cos (θ1 * + 2π / 3) It = iu · sin θ1 * + iv · sin (θ1 * −2π / 3) + iw · sin (θ1 * + 2π / 3) (2) However, θ1 * = ∫ω1 · dt torque command value τR is compensated by the motor torque compensating means 90 so that the difference Δτ becomes zero, and as a result, the torque command value compensation value τ * is obtained. To be converted. The motor torque compensating means 90 may be composed of PI (proportional + integral) elements. The compensating means disclosed in the following embodiments is assumed to be composed of PI elements.

The torque current command value calculating means 100 calculates the torque current command value It * using τ * from the equation (3) and the estimated secondary magnetic flux φ2 obtained from the secondary magnetic flux estimating means 170. The torque current command value It * is input to the slip frequency calculation means 115 through the variable limiter 105.

The variable limiter 105 has an exciting current command value I issued from an exciting current determining means 160, which will be described later.
It is controlled so that it is inversely proportional to m *. That is, as the secondary magnetic flux weakens, the limiter value increases, and the movable range of the torque current command value expands. This is a measure for avoiding the fact that the capacity of the inverter is apparently reduced by reducing the exciting current command value Im * as the secondary magnetic flux weakens.

The slip angular frequency ωs is calculated by the slip frequency calculating means 115 based on the equation (4). It * = (1 / mp) · {(lm '+ l 2) / lm'} · (τM * / φ2) (3) ωs = r2 · {lm '/ (lm' + l 2)} · (It * / φ2) (4) The slip angular frequency ωs is added to the rotational angular velocity ωM of the induction motor obtained from the velocity detecting means 60 and the addition process of the equation (5) is performed by the adder 151 to obtain the angular frequency ω1 of the PWM inverter 20. This is taken into the primary current command value calculating means 120 and becomes the angular frequency of the primary current command value. ω1 = ωs + ωM (5) Ratio It * / of the above torque current command value It * output from the variable limiter 105 and the rated torque current (It) rate
The load factor α is obtained from the (It) rate by the load factor calculation means 145. α = It * / (It) rate (6) In order to generate a secondary magnetic flux corresponding to the load factor α in the secondary circuit of the induction motor 40, the secondary magnetic flux command value φ2 * is secondary in the following procedure. It is determined by the magnetic flux setting means 130 and the secondary magnetic flux command calculation means 140.

First, the secondary magnetic flux setting means 130 sets a secondary magnetic flux setting value φR corresponding to the rotational angular velocity ωM of the induction motor 40.
(Corresponding to the maximum magnetic flux at the corresponding rotation angular velocity) is given based on the equation (7). 0 ≤ ωM ≤ ωM0, φR = (φ2) rate (: rated secondary magnetic flux) ωM0 <ωM, φR = (ωM0 / ωM) · (φ2) rate (7) where ωM0: base rotational angular velocity The determined setting value φR of the secondary magnetic flux is input to the secondary magnetic flux command calculating means 140, and the secondary magnetic flux command value φ2 * is calculated by the equation (8). φ2 * = α · φR + φ20 (8) where φ20 = (Im0) · lm ′ where φ20 is given as a secondary magnetic flux command value to the induction motor 40 as a fixed component that is not affected by the torque current command value. Therefore, when there is no load, this fixed amount becomes the magnetic flux generated inside the electric motor 40. This value is selected so that the energy loss of the battery can be reduced as much as possible, and is usually selected to be 20% or less of the rated magnetic flux. It is selected so that the torque current flowing in the secondary circuit can be prevented from becoming excessive even if a sudden load is applied to the wheels in the no-load state. It should be set to zero from the viewpoint of energy saving before the vehicle is stopped or the accelerator is depressed.

Equation (8) is the principle shown in FIG. 2, that is, the desired torque is the minimum current (current flowing through the induction motor 40,
Hereinafter referred to as primary current. ), The secondary magnetic flux command value φ2 * is determined in association with the torque current command value. When the primary current I1 is expressed using the exciting current Im (= φ2 / lm ') for generating the secondary magnetic flux φ2 and the torque current It, the equation (9) is obtained. I ₁ · I ₁ = I _t · I _t + I _m · I _m (9) On the other hand, the torque τ generated in the induction motor 40 is expressed by the equation (1) as follows. It is proportional to the product of (exciting current in the steady state maintained). FIG. 2 shows the relationship between the torque current and the exciting current at a given motor torque, which can be represented by a hyperbola. Therefore, as shown in FIG. 2, when τM1, τM2, and τM3 increase, the minimum of primary currents that generate each torque corresponds to each contact of A, B, and C. This is the primary current.

That is, by controlling the torque current and the exciting current (or the secondary magnetic flux) in proportion to each other, it is possible to always generate a desired torque with a minimum primary current. Therefore, the electric vehicle can be driven with the minimum battery energy. Further, as the torque current decreases, the secondary magnetic flux is weakened, and the exciting current also decreases.

When the load suddenly increases in such a state, the torque current increases. However, if the limiter of the torque current command value remains at the value (It0) max determined at the rated point, the exciting current is increased. The reduced amount means that the required motor torque cannot be generated. Therefore, as shown in FIG. 3, the variable limiter 105 changes the limiter (It) max * of the torque current command value according to the equation (10) so that the torque current always flows up to the maximum value of the primary current. (It) max * = (I 1) max · sin (arccos (Im * / (I 1) max) (2 obtained in 10) above operation the rotor flux command value .phi.2 * and formula (11)
The exciting current determining means 16 is adapted to compensate for the deviation from the secondary magnetic flux φ2 estimated by the secondary magnetic flux estimating means 170 by
An exciting current command value Im * is generated from 0. φ2 = lm ′ · Im * / (1 + T ₂ · s) (11) where T ₂ = (lm ′ + l ₂ ) / r2, r2: secondary resistance primary current command value computing means 120 is a three-phase alternating current Generates command values iu *, iv *, iw *. iu * = I ₁ · cos (θ1 * + ψ) iv * = I ₁ · cos (θ1 * + ψ−2π / 3) iw * = I ₁ · cos (θ1 * + ψ + 2π / 3) (12) θ1 * = ∫ω1 Dt (13) ψ = arctan (It * / Im *) (14) The current control means 180 has the primary current command value calculation means 120.
AC current command values iu *, iv *, iw * of three phases obtained from the U-phase and V detected by the current sensors 30, 31, 32
AC currents iu, iv, and iw of phase W are input. The current control means 180 generates AC primary voltage command values Vu *, Vv *, Vw * so that the deviation between the command value and the current is eliminated.

The voltage command values Vu *, Vv *, Vw * are input to the PWM signal generating means 190 and become a modulated wave for generating the PWM signal. The modulated wave is generated by the PWM signal generating means 190
It is compared with a carrier wave (not shown) generated within to generate a PWM signal. The PWM signal is applied to the PWM inverter 20 to generate an AC voltage having a variable frequency and a variable voltage to drive the three-phase induction motor 40 at a variable speed.

According to this embodiment, since the exciting current commensurate with the generation of the desired torque with the minimum primary current always flows through the induction motor, the energy of the battery can be efficiently used and per charge. You can extend your mileage.

Further, even when a sudden disturbance is applied to the axle and the torque current command value increases in a state of almost no load, the secondary magnetic flux command value is increased at the same speed as the changing speed of the torque current command value. A stable torque control system can be realized because the exciting current is passed through the induction motor and the secondary magnetic flux generated inside the induction motor can be established at high speed.

FIG. 4 shows another embodiment. The difference from the embodiment of FIG. 1 is the configuration of the secondary magnetic flux estimating means 171 and the motor torque detecting means 110. Therefore, only this configuration will be described.

First, the secondary magnetic flux estimating means 171 will be described. U phase detected from the current sensors 30, 31, 32,
V-phase and W-phase alternating currents iu, iv, iw and current control means 1
The secondary magnetic flux φ2 is estimated from the voltage command values Vu *, Vv *, Vw * output from 80 using the equation (15). φ2 = φu · cosθ1 * + φv · cos (θ1 * -2π / 3) + φw · cos (θ1 * + 2π / 3) (15) where, φj = {∫ (Vj- r1 · ij) dt-l 1 · ij} (j
= u, v, w), Vj (j = u, v, w): Primary voltage command values Vu *, Vv
Primary voltage estimated from *, Vw * (= Kp · Vj *, Kp: proportional constant related to battery voltage) This method estimates the secondary magnetic flux from the primary voltage command value and AC currents iu, iv, iw. Since the detection can be performed with higher accuracy than the method of the embodiment shown in FIG. 1, the method may be applied to the secondary magnetic flux estimating means 170 in the embodiment of FIG.

Next, the motor torque detecting means 110 will be described. Using the DC current Id flowing through the PWM inverter 20 detected by the current sensor 33, the DC voltage Ed of the battery 10, and the rotational angular speed ωM of the induction motor 40 obtained from the speed detection means 60, the motor torque τ is calculated by the equation (16). To detect. τ = Ed · Id / ωM (16) This method is effective as a method that can easily detect torque.

FIG. 5 shows another embodiment. In the present embodiment, the torque control is further stabilized.
The torque current It and the exciting current Im are detected, and the torque control is performed based on these. It has a secondary resistance correction means and an excitation inductance correction means, and even if these parameters fluctuate, an appropriate excitation current command value Im
*, The torque current command value It *, and the slip angular frequency ωs are obtained.

The torque command value is once converted into a set value ItR of the torque current command value, and the manipulated variable It * of the torque current command value is set to match the set value ItR of the torque current command value with the torque current of the induction motor 40. It is a method of determining and performing torque control based on It *. Only parts different from the embodiment of FIG. 1 will be described.

First, using the torque command value τR output from the torque command value calculating means 80 and the secondary magnetic flux command value φ2 * obtained from the secondary command value calculating means 140, the torque current command value is calculated from the equation (17). The set value ItR of is calculated. ItR = kτR / φ 2 * (17) where k = (lm '+ l ₂ ) / (m ・ plm') where the exciting inductance lm 'is 2
A constant value may be used in the linear region where the secondary magnetic flux is not saturated. Generally, in an electric vehicle, since the weight of the electric motor is reduced, the size of the portion that generates magnetic flux tends to be smaller. For this reason, in a region where a large torque is required, the magnetic flux is saturated, and the exciting inductance is reduced from the normal value.

Therefore, the exciting inductance correcting means 11
In No. 1, the relationship between the exciting current Im and the inductance lm is made into a table in advance, and the excitation / excitation is performed based on this table.
Exciting current Im detected by the torque current detecting means 112
The exciting inductance lm 'corresponding to the magnitude of is determined. In this embodiment, the exciting inductance l
ItR is calculated using m'in equation (17).

ItR is introduced into the plus side terminal of the adder / subtractor 150, and the torque current It detected by the excitation / torque current detecting means 112 is introduced into the minus side terminal of the adder / subtractor 150. A torque current command value It * is generated from the torque current compensating means 91 to compensate the deviation ΔIt of the torque current command value obtained from the adder / subtractor 150. The torque control is performed using the torque current command value It * and the exciting current command value Im * obtained as follows.

Based on the exciting current Im obtained from the exciting / torque current detecting means 112, the secondary magnetic flux φ2 is estimated by using the equation (18). φ2 = lm ′ · Im / (1 + T2 · s) (18) Here, the secondary time constant T2 (= (lm ′ + l ₂ ) / r2) is related to the exciting inductance lm ′ and the secondary resistance r2. As the exciting inductance lm ', the value obtained from the exciting inductance correcting means 111 is used.

In an electric vehicle, overload or traveling at high ambient temperature may occur depending on the traveling state. As a result, when the temperature of the induction motor 40 increases, the secondary resistance r2 also changes. Therefore, the secondary resistance correction means 42 obtains the secondary resistance r2 with respect to the temperature of the induction motor 40, which has been measured and made into a table, based on the temperature tc of the induction motor 40 detected by the temperature sensor 41. ing. The values r2 and Im 'of the slip calculation means 115 are the values corrected by the above-mentioned means.

The secondary magnetic flux φ 2 is introduced to the minus side terminal of the adder / subtractor 152, and the secondary command value φ 2 * obtained by the same method as in the embodiment of FIG. 1 is introduced to the plus side terminal of the adder / subtractor 152. The deviation Δφ2 obtained from the adder / subtractor 152 is introduced into the exciting current determining means 160, and the exciting current determining means 160 determines the exciting current command value Im * to correct the deviation Δφ2.

In the present embodiment, the torque current It, obtained from the three-phase alternating currents iu, iv, iw flowing in the induction motor 40,
Since the torque is controlled by directly using the exciting current Im and by correcting the constant variation of the motor such as r2 and lm ', the torque is stable against the disturbance even if the constant of the motor changes depending on the running state. A control system can be realized. The correction of this parameter may be naturally performed in each of the embodiments shown in FIGS. 1 and 4 and the next FIG.

FIG. 6 shows another embodiment. The difference from the embodiment shown in FIG. 1 is that the deviation Δτ between the torque command value τ R and the motor torque τ is divided by the square of the secondary magnetic flux φ 2 obtained from the secondary magnetic flux estimating means 170 to calculate the slip angular frequency. The reason is as follows.

When there is no change in the secondary magnetic flux, the accuracy of the slip frequency does not change much even if the calculation is performed in the order of torque compensation, torque current command value, and slip frequency as in the embodiment shown in FIG. However, when the secondary magnetic flux is significantly reduced, the accuracy of the slip frequency finally obtained may be deteriorated due to the cancellation of digits in the calculation. Moreover, the slip angular frequency is generally the angular frequency ω of the PWM inverter 20.
1. The rotational speed is very small with respect to each value of ωM, etc., but in particular, as the load becomes lighter, the slip frequency decreases and becomes smaller. Therefore, in the light load state, the stability of torque control deteriorates.

Therefore, in the embodiment shown in FIG. 6, first, the construction is changed so that the slip angular frequency can be immediately obtained from the torque deviation so as to ensure the accuracy of the slip frequency. Figure 1 except that the order of slip angular frequency calculation is changed
The description is omitted because it is the same as the embodiment described above. According to the present embodiment, the calculation accuracy of the slip frequency is secured even if the secondary magnetic flux becomes small, and stable torque control can be realized.

FIG. 7 shows the relationship between the practical torque current and the exciting current. In an electric vehicle, in order to make the size of the electric motor as small as possible, the torque current is designed to be large without increasing the exciting current to generate the generated torque. Therefore, the ratio of the torque current to the exciting current is larger than 1. Usually it will be 4-6. Therefore, when the exciting current and the torque current are operated along the primary current minimization line as shown in FIG. 7, the exciting current has a rated value (Im) rate ('D') before the torque current. Correspondence). If the exciting current is larger than this value, it will be overexcited and cannot be increased.
Therefore, the exciting current and the torque current are controlled along the primary current minimization line until the rated exciting current is reached, and when the rated exciting current is reached, the rated exciting current is kept constant and only the torque current is allowed. Operate up to the current (It) max.

As another method, taking into consideration that the rated exciting current (Im) rate is smaller than the torque current (It) max, along the line A ₁ ... D ₁ shown in FIG. There is no big difference even if they are operated, and the same effect can be obtained.

In order to minimize the loss L of the motor, not only the primary copper loss as described above is minimized (the magnitude of the primary current is minimum), but also the secondary copper loss, the iron loss, etc. It is necessary to consider the loss of. Actually, it is necessary to determine the relationship between the exciting current and the torque current in consideration of these losses.

L = (r1 + rm) * (Im * Im) + (r1 + r2) * (It * It) (19) where rm: iron loss resistance where torque τ (proportional to It * Im) is constant. When the ratio γmin (= Im / It) between Im and It that minimizes the value of L in the original equation (19) is obtained, the equation (20) is obtained. (γmin) · (γmin) = (r1 + r2) / (r1 + rm) (20) The above value of (γmin) changes with changes in the motor temperature and rotational angular velocity, and the resistances r1, r2, rm, etc. fluctuate accordingly. Change. It depends on how the motor is designed,
In a practical range, this value is almost (Im) rate / (It)
It varies from the value determined by max to 1.

Therefore, it is most effective to operate in the triangular area filled in in FIG. When this is expressed by a mathematical expression, it becomes as shown in Expression (21). φ2 * = β · It * + φ20 (21) Im ′ ≧ β ≧ {(Im) rate · lm′−φ20} / (It) max In practical use, “characteristic of φ2 * and It” or “Im and It The characteristic 'is made into a table in advance, and when the rotation speed or temperature of the electric motor fluctuates, it may be obtained according to the fluctuation. Of course, a multidimensional table may be prepared corresponding to these parameters.

From the viewpoint of suppressing the overcurrent at the time of starting, it is preferable that the value of φ20 has a certain value (20% or less of the normally rated secondary magnetic flux) rather than zero. In order to improve efficiency, including waiting for traffic lights and temporary stoppages due to traffic congestion, etc.
Should be zero. In this case, the torque command value is delayed until the magnetic flux at the time of starting is established, or (It) max of the torque current as shown in formula (10) is compensated to increase φ2 *,
It is preferable to suppress the overcurrent (delay of generated torque) at the time of starting by making the exciting current flow quickly.

In the method described so far, the secondary magnetic flux command value φ2 * is determined from the torque current command value It * based on the equation (21), and the exciting current command value is generated so as to follow this. , The excitation current command value Im * was previously described (γmi
It may be obtained from the value of n). This embodiment is shown in FIGS.

In the embodiment shown in FIG. 8, the torque τ generated in the induction motor 40 is estimated by using the equations (22) and (23),
The estimated torque is used to correct the torque command value τR obtained from the torque command value calculation means 80 by the motor torque compensation means 90, and the resultant torque command value compensation value τ *
Is used to control the torque. In addition, the secondary magnetic flux generated at this time is converted into a three-phase alternating current (primary current).
It is also characterized in that it is estimated based on the excitation current Im detected from. τ = m · p · {lm ′ / (lm ′ + l ₂ )} · φ2 · It (22) φ2 = (Im · lm ′) / (1 + T2 · s) (23) Here, the exciting current Im and the torque current It is an amount detected by the excitation / torque current detection means 112 by the method already described. The secondary magnetic flux φ2 is obtained from the secondary magnetic flux estimating means 170 based on the equation (23). The secondary magnetic flux is saturated depending on the value of the exciting current Im. Therefore, (2
In the equation 3), in order to be able to estimate the secondary magnetic flux φ 2 even in the saturated region, the exciting inductance lm ′ is corrected by the exciting inductance correcting means 111 based on the actually detected exciting current. Use the value obtained.
The function of Im and lm 'in the correction means 111 is measured in advance,
Use the one obtained by calculation.

The motor torque compensating means 90 generates the torque command value compensation value τ * so that the deviation between the torque command value τR and the detected generated torque τ becomes zero. 90 is P
It is composed of I (proportional + integral) elements.

Using the torque command value compensation value τ * and the secondary magnetic flux φ2, the calculation of the equation (3) is performed by the torque command value calculation means 100 to obtain the torque current command value It *. Based on the torque current command value It *, the optimum exciting current command value determining means 146 determines the optimum exciting current command value from the function of Im and It stored in this means which minimizes the loss of the electric motor.

The exciting current command value is input to the limiter 132, and the limiter 132 determines the maximum value of the exciting current command value. This maximum value is the maximum secondary magnetic flux (φ) max determined by the secondary magnetic flux setting means 130 and is a value allowed by the drive system, and the characteristic with respect to the rotation angle ωM is the same as that shown in the equation (7). Is decided. The (φ) max thus determined is the maximum value (Im) max (= (φ) max of the exciting current command value by the exciting current converter 131).
/ Lm '). The exciting current command value is input to the plus terminal of the exciting current determining means 160 via the limiter 132 as an actual exciting current command value ImR. The exciting current Im actually flowing in the electric motor 40 detected by the detecting unit 112 is input to the negative terminal of the determining unit 160.

In the determining means 160, the exciting current command value ImR
So that the exciting current Im is equal to
* Decide. Therefore, the manipulated variable Im * is always determined in relation to the torque current command value It *.

Further, as for the torque current command value It *, the slip frequency calculating means 115 determines the slip angular frequency ωs based on the equation (4). The slip angular frequency ωs is input to the slip frequency limiter 116, and its operating range is expressed by the formula (2
4) and (25). 0 ≦ ωM ≦ ωM0, (ωs) max = ωs0 (24) ωs0: a value determined based on the maximum torque command value,
Normally, a value twice that of the rated slip angular frequency is selected. ωM0 ≤ ωM ≤ (ωM) max, (ωs) max = A · ωM + B (25) where A = [{r2 / (l ₁ + l ₂ )}-ωs0] / {(ωM) max- (ωM0 )} B = [{ωs0 · (ωM) max}-{r2 / (l ₁ + l ₂ )} · (ωM0)] / {(ωM) max- (ωM0)} Here, the equation (25) is When the rotational angular velocity ωM becomes higher than the base rotational angular velocity ωM0 and the secondary magnetic flux is weakened, the equation (4)
The slip angular frequency obtained from the above will increase. Therefore, the operating range of the slip angular frequency is expanded within the range where the stall torque is not exceeded at the maximum rotational angular velocity (ωM) max.

In this way, the operating range of the torque current can be widened at the predetermined target torque command value, and therefore the increase of the exciting current at the maximum motor torque at each rotational angular velocity can be suppressed. As a result, the rise of the induced voltage can be suppressed as much as possible at each rotational angular velocity ωM, so that the voltage that can be controlled by the PWM inverter 20 (the difference between the output voltage of the PWM inverter 20 and the induced voltage) increases, so that the high-speed range is increased. The motor can be driven stably up to.

The slip angular frequency ωs obtained by this processing is added to the rotational angular velocity ωM by the adder 151,
The primary angular frequency ω1 is obtained via the limiter 117. Here, the limiter 117 serves to prevent the rotation speed of the induction motor 40 from increasing above a predetermined value.

The torque current command value It *, the manipulated variable Im * of the exciting current, and the manipulated variables of the primary angular frequency ω1 obtained by the method described above are input to the primary current command computing means 120, and the Expressions (12) to (14) are calculated, and three-phase primary current command values iu *, iv *, and iw * are generated. Based on these command values, the current control means 180 constitutes a current control system and the primary voltage command values Vu *, Vv *, Vw *.
Generate.

A primary voltage command value is input to the PWM signal generating means 190, and a PWM signal is generated based on this command value. This PWM signal is applied to the PWM inverter 20 and controls the torque generated by the induction motor 40. Although not shown in the present embodiment, the output of the means 100 is further provided with a torque current determining means for correcting the torque current command value It * so that the torque current It obtained from the means 112 coincides with it. The primary current commands iu *, iv *, iw * may be obtained using the compensation value obtained by compensating the It * by the means. In this case, it is possible to perform more accurate torque control than the embodiment of FIG.

In the present embodiment, the torque command value issued via the accelerator is compensated by detecting the generated torque, and the torque current and the exciting current are controlled based on the compensated torque command value. The torque control with good tracking performance can be performed not only in the steady state but also in the transient state. For this reason, a feeling of acceleration can be obtained, and an electric vehicle that is comfortable to ride with little vibration against disturbance can be realized.

Further, the torque generated in the motor can be controlled while maintaining the relationship between the exciting current and the torque current that provides the highest efficiency including the transient state. As a result, in the electric vehicle in which the transient state is constantly maintained, the energy of the battery can be efficiently used at all times, and the traveling distance per charge can be increased.

Next, the embodiment shown in FIG. 9 will be described. This embodiment is the same as the embodiment shown in FIG. 8 until the torque current command value It *, the exciting current command value Im *, and the primary angular frequency ω1 which are the manipulated variables for torque control are obtained. Omit it. In FIG. 9, the d-axis voltage determining means 161 and the q-axis voltage determining means 16 have functions equivalent to those of the current control system without the current control system including the means 120 and 180.
2. The points obtained by the decoupling means 121 and the primary voltage command means 181 are different, and therefore these will be described.

The d-axis voltage determining means 161 adjusts the voltage compensation value V on the d-axis so that the exciting current Im obtained from the means 112 matches the exciting current command Im * obtained via the means 132.
Determine d '. The q-axis voltage determining means 162 is means 11
The voltage compensation value Vq 'on the q-axis is determined so that the torque current It obtained from 2 matches the torque current command value It * obtained from the means 100. The decoupling means 121 uses the secondary magnetic flux φ2 obtained from the detected exciting current Im in addition to the above-mentioned manipulated variable to calculate the voltage command values Vd * and Vq * on the dq axis coordinates. Obtained from (26). Vd * = r1 · Im−ω1 · σ · L1 · It + Vd ′ Vq * = r1 · It + ω1 {σ · L1 · Im + (lm ′ / L2) · φ2} + Vq ′ (26) However, σ = 1− (lm ′) / L1) ・ (lm '/ L2) L1 = lm' + l ₁ , L2 = lm '+ l ₂ Details are omitted, but by performing this operation, the state represented on the dq coordinates of the induction motor. From the equation, Im and It can be controlled without interfering with each other (transient terms of Im and It are zero).

Vd * and Vq * are input to the primary voltage command means 181 and the primary voltage command values Vu *, Vv *, Vw * are calculated based on the following equation.
To form. Vu * = V1 · cos (θ1 + δ) (27) Vv * = V1 · cos (θ1 + δ-2π / 3) (28) Vw * = V1 · cos (θ1 + δ + 2π / 3) (29) However, V1 · V1 = Vd *・ Vd * + Vd * ・ Vq * θ1 = ∫ω1 ・ dt, δ = arctan (Vq * / Vd *) In addition to the above, the above primary voltage command values Vu *, Vv *, Vw *
Can also be obtained by performing two-phase or three-phase conversion based on the above Vd * and Vq *. Primary voltage command value Vu *,
The method of obtaining the PWM signal from Vv * and Vw * is the same as in the embodiment of FIG.

As described above, according to this embodiment, the current control system can be omitted, so that the gain adjustment of the control system is eliminated and a simple controller can be realized. As a result, it is possible to realize torque control with an improved system reliability with a simple configuration.

[0063]

According to the present invention, the energy of the battery can be used efficiently, so that the traveling distance per charge of the electric vehicle can be improved. Further, even if the motor torque changes due to a disturbance applied to the axle, an uphill slope, a sudden acceleration, etc., smooth torque control corresponding to the accelerator amount can be performed, so that comfortable driving performance of the electric vehicle can be realized.

[Brief description of drawings]

FIG. 1 is a block diagram of an embodiment of the present invention.

FIG. 2 is a diagram showing a relationship between a torque current and an exciting current (secondary magnetic flux) for executing primary current minimization control.

FIG. 3 is an explanatory diagram of a method of varying the limiter of the torque current while keeping the primary current maximum value constant.

FIG. 4 is a block diagram of another embodiment of the present invention.

FIG. 5 is a block diagram when torque control is performed by using another method for estimating the secondary magnetic flux shown in FIG. 1.

FIG. 6 is a block diagram in the case of performing torque control for increasing the accuracy of a slip frequency.

FIG. 7 is a diagram illustrating a practical operating range of an exciting current and a torque current.

FIG. 8 is a block diagram in the case of performing torque control by having both a function of favorably controlling torque up to a transient state and a function of improving efficiency.

9 is a block diagram of an embodiment capable of simplifying the configuration of FIG.

[Explanation of symbols]

10 Battery 20 PWM Inverter 30, 31, 32 Current Sensor 40 Three-Phase Induction Motor 42 Secondary Resistance Correction Means 50 Encoder 60 Speed Detection Means 70 Accelerator 80 Torque Command Value Calculation Means 90 Motor Torque Compensation Means 91 Torque Current Compensation Means 100 Torque Current Command calculation means 101 Torque current command conversion means 105 Variable limiter 110 Motor torque detection means 111 Excitation inductance correction means 112 Excitation / torque current detection means 115 Slip frequency calculation means 116 Slip frequency limiter 117 PWM inverter angular frequency (primary angular frequency) Limiter 120 Primary current command calculation means 121 Decoupling means 130 Secondary magnetic flux setting means 140 Secondary magnetic flux command calculation means 145 Load factor calculation means 161 d-axis voltage determination means 162 q-axis voltage determination 181 primary voltage command calculation unit 150, 151 subtracter 152 subtractor 160 exciting current determining means 170 secondary magnetic flux estimator 180 current control unit 190 PWM signal generating means

Front page continuation (72) Inventor Taizo Miyazaki 7-1, 1-1 Omika-cho, Hitachi City, Ibaraki Prefecture Hitachi Research Laboratory, Hitachi Ltd. (72) Inventor Yusuke Takamoto 7-1-1, Omika-cho, Hitachi City, Ibaraki Prefecture Hitachi Ltd., Hitachi Research Laboratory (72) Inventor Riki Omae 7-1, 1-1 Omika-cho, Hitachi City, Ibaraki Prefecture Hitachi Ltd. Hitachi Research Laboratory (72) Inventor Sanshiro Ohara Katsuta City, Ibaraki Field 2520 Hitachi Automotive Systems Division

Claims (14)

[Claims] 1. A battery for supplying a DC voltage and a DC current, a PWM inverter for converting the DC voltage into an AC power source of a variable frequency and a variable voltage, an induction motor driven by the PWM inverter, and an accelerator depression amount. A torque control method for an electric vehicle having a control circuit for controlling the torque to be generated by the induction motor by a torque command value determined based on the torque command value, and controlling the speed of the vehicle as desired, wherein the torque generated by the induction motor is A torque current command value and an exciting current of the induction motor, which are provided with a torque detection means for detecting and are determined so as to compensate for a deviation between the torque generated by the induction motor and the torque command value obtained from the torque detection means. The torque of the electric motor is controlled based on a command value. Control method. 2. The torque control method for an electric vehicle according to claim 1, wherein the torque detecting means includes means for detecting at least a three-phase alternating current flowing in the primary winding of the induction motor, and the three-phase alternating current. Means for converting a current into a dq axis coordinate system that rotates at a synchronous speed with the angular frequency of the PWM inverter, and a product of a d axis component (excitation current) and a q axis component (torque current) obtained by the conversion means. A method of controlling torque of an electric vehicle, comprising: 3. The torque control method for an electric vehicle according to claim 1, wherein the torque detection means includes at least the P
Means for detecting a direct current flowing into the WM inverter,
An electric vehicle comprising: a unit that detects the battery voltage; a unit that detects the rotational angular velocity of the induction motor; and a unit that divides the product of the DC current and the battery voltage by the rotational angular velocity. Torque control method. 4. The torque control method for an electric vehicle according to claim 1, wherein the torque current command value is a new torque command value determined so that the generated torque detected by the torque detecting means matches the torque command value. The torque control method for an electric vehicle according to claim 1, wherein the operation amount is obtained by dividing the operation amount by a secondary magnetic flux generated in a secondary circuit of the induction motor. 5. The torque control method for an electric vehicle according to claim 1, wherein the torque command value is decreased in accordance with the decrease when the battery voltage drops below a predetermined value. Control method. 6. The torque control method for an electric vehicle according to claim 4, wherein the control circuit sets a secondary magnetic flux that sets a secondary magnetic flux generated in a secondary circuit of the induction motor according to a rotational angular velocity of the induction motor. A secondary magnetic flux setting value obtained from the secondary magnetic flux setting means is corrected by a torque current command value, and the correction amount and the secondary magnetic flux generated in the secondary circuit of the induction motor match. A torque control method for an electric vehicle, characterized in that an exciting current command value is determined. 7. The method for controlling torque of an electric vehicle according to claim 4, wherein the control circuit maintains the relationship between the torque current and the exciting current that is predetermined to optimize the efficiency of the induction motor. An optimum excitation current is provided corresponding to the torque current command value determined by the manipulated variable of the new torque command value, which is provided with the current command value determining means and is determined so that the generated torque detected by the torque detecting means matches the torque command value. Determine the reference value of the exciting current from the command value determining means,
Excitation current command value so that the excitation current of the d-axis component obtained by the conversion means for converting the three-phase alternating current flowing in the primary winding of the induction motor into the dq axis coordinate system matches the reference value. A method of controlling torque in an electric vehicle, comprising: 8. The torque control method for an electric vehicle according to claim 7, wherein the optimum exciting current command value determining means is a motor.
When the primary and secondary resistances of the motor are r1 and r2 and the iron loss resistance of the motor is rm, the value obtained by γmin = {(r1 + r2) / (r1 + rm)} ^1/2 is used. Torque control method for electric vehicles. 9. The torque control method for an electric vehicle according to claim 6, wherein the correction value of the secondary magnetic flux setting value is a load factor obtained by dividing the torque current command value by the rated torque current of the induction motor, A torque control method for an electric vehicle, comprising: multiplying the secondary magnetic flux setting value; and adding the value to a predetermined secondary magnetic flux setting value to obtain the value. 10. The torque control method for an electric vehicle according to claim 6, wherein the control circuit comprises an exciting inductance correction means having a characteristic for relating the exciting current and the exciting inductance to each other so as to correspond to the exciting current command value. A method of controlling torque in an electric vehicle, comprising: determining an excitation inductance by using the excitation inductance and the excitation current command value to estimate a secondary magnetic flux. 11. The torque control method for an electric vehicle according to claim 4, wherein the control circuit includes an excitation inductance correction unit having a characteristic that associates an excitation current with an excitation inductance, and the primary winding of the induction motor. A three-phase alternating current flowing in the dq axis coordinate system is converted into a dq axis coordinate system to obtain an exciting inductance corresponding to the exciting current of the d-axis component, and a secondary value is obtained by using the exciting inductance and the exciting current. A torque control method for an electric vehicle, characterized in that a magnetic flux is estimated. 12. The torque control method for an electric vehicle according to claim 4, wherein the correction amount of the secondary magnetic flux setting value is divided into a torque current command value It * and a term φ20 that is not related to the torque command value. , Φ2 * = β · It * + φ20 lm ′ ≧ β ≧ {(Im) rate · lm′−φ20} / (It) max lm ′: exciting inductance, (Im) rate: rated value of exciting current, ( It) max: the maximum value of the torque current, and the torque control method for an electric vehicle is determined. 13. The torque control method for an electric vehicle according to claim 7, wherein the PWM inverter comprises means for detecting a three-phase alternating current flowing in the primary winding of the induction motor, and the three inverters detected by the means. Phase alternating current follows the three-phase primary current command determined from the torque current command value, the excitation current command value, the secondary magnetic flux generated in the secondary circuit of the induction motor, and the rotational angular velocity of the induction motor. A method for controlling torque of an electric vehicle, wherein the method is controlled by a PWM signal emitted so as to control the electric vehicle. 14. The torque control method for an electric vehicle according to claim 4, wherein the control circuit determines the d-axis voltage Vd ′ so that the exciting current Im matches the exciting current command value Im *.
The PWM inverter comprises a shaft voltage determining means and a q-axis voltage determining means for determining a q-axis voltage Vq 'so that the torque current It matches the torque current command value It *.
q ', torque current command value It *, exciting current command value Im *, secondary magnetic flux φ2 generated in the secondary circuit of the induction motor, and rotational angular velocity ωM of the induction motor, the following equation, Vd * = r1・ Im * −ω1 ・ σ ・ L1 ・ It * + Vd 'Vq * = r1 ・ It * + ω1 {σ ・ L1 ・ Im * + (lm' / L2) ・ φ2} + Vq 'ω1 = ωM + {(lm' ・ Im *) / (T2 ・ φ2)} where σ = 1- (lm '/ L1) ・ (lm' / L2), L1
_{= Lm '+ l 1, L2} = lm' + l 2, T2 = L2 / r2, the voltage command value Vd on d-q-axis coordinate calculated from *, that is controlled by a PWM signal generated based on the Vq * A characteristic electric vehicle torque control method.