JP5639735B2 - Semiconductor device, display device, electronic device and display module - Google Patents

Description

  The present invention relates to a configuration of a light emitting device. The present invention particularly relates to a structure of an active matrix light-emitting device having a thin film transistor (hereinafter referred to as TFT) manufactured on an insulator such as glass or plastic. Further, the present invention relates to an electronic device using the light emitting device for a display portion.

  In recent years, development of display devices using light-emitting elements such as electroluminescence elements (EL elements) has been activated. Here, the EL element includes both those using light emission (fluorescence) from singlet excitons and those using light emission (phosphorescence) from triplet excitons. In this specification, an EL display device is given as an example of a light-emitting device, but a display device using another light-emitting element is also included.

  An EL element is configured such that a light emitting layer is sandwiched between a pair of electrodes (anode and cathode), and usually has a laminated structure. A typical example is a stacked structure of “hole transport layer / light emitting layer / electron transport layer” proposed by Tang et al. Of Eastman Kodak Company. This structure has very high luminous efficiency, and this structure is employed in almost all EL devices that are currently being studied.

  In addition to this, “hole injection layer / hole transport layer / light emitting layer / electron transport layer” or “hole injection layer / hole transport layer / light emitting layer / electron transport layer / electron injection layer” on the anode. Are stacked in the order of "." As the structure of the EL element in this specification, any of the above structures may be adopted. Further, a fluorescent pigment or the like may be doped into the light emitting layer.

  In this specification, all layers provided between the anode and the cathode are collectively referred to as an EL layer. Therefore, the above-described hole injection layer, hole transport layer, light emitting layer, electron transport layer, and electron injection layer are all included in the EL layer, and an element including the anode, the EL layer, and the cathode is referred to as an EL element. .

  FIG. 3A is a schematic view of a light emitting device. A pixel portion 301 is disposed at the center of the substrate 300. A source signal line driver circuit 302 for controlling the source signal lines and a gate signal line driver circuit 303 for driving the gate signal lines are arranged around the pixel portion 301. In FIG. 3A, the gate signal line driver circuit 303 is arranged symmetrically on both sides of the pixel portion 301, but may be arranged on only one side. However, considering the reliability and efficiency of the circuit operation, it is desirable to arrange both sides.

  A clock signal, a start pulse, a video signal, and the like are input to the source signal line driver circuit 302 and the gate signal line driver circuit 303 via a flexible printed circuit (FPC) or the like.

  The operation of the drive circuit will be described. In the gate signal line driving circuit, pulses for sequentially selecting the gate signal lines are output by the shift register 321 in accordance with the clock signal and the start pulse. Thereafter, after an operation such as conversion of the voltage amplitude of the signal by the level shifter 322, the signal is output to the gate signal line via the buffer 323, and the gate signal lines are sequentially selected.

  In the source signal line driver circuit, sampling pulses are sequentially output by the shift register 311 in accordance with the clock signal and the start pulse. The first latch circuit 312 holds the digital video signal according to the timing of the sampling pulse. When the operation for one horizontal period is completed, a latch pulse is input during the subsequent blanking period, and the digital video signals for one row held in the first latch circuit 312 are simultaneously latched in the second latch. The data is transferred to the circuit 313, and writing to the pixels in one row is performed simultaneously on the pixels in the row in which the gate signal line is selected.

  Subsequently, the operation of the pixel portion will be described. FIG. 3 (B) shows one pixel extracted by 310 in FIG. 3 (A). Source signal line (S (n)), gate signal line (G (m)), current supply line (Current), switching TFT 351, EL driving TFT 352, storage capacitor 353, EL element 354 which is a typical light emitting element Etc.

  When the gate signal line is selected, the switching TFT 351 is turned on. Meanwhile, the digital video signal held in the second latch circuit is output to the source signal line. The output digital video signal passes between the source and drain of the switching TFT 351, is input to the gate electrode of the EL driving TFT 352 and is turned ON, and a current flows through the EL element 354. On the other hand, charges are held in the storage capacitor 353. Even after the selection period of the gate signal line ends and the switching TFT 351 is turned off, the potential of the gate electrode of the EL driving TFT 352 is held by the charge held in the holding capacitor 353, and a current flows through the EL element 354. to continue.

  In this specification, when describing the circuit operation, the operation of the TFT may be described. When the TFT is turned on, the absolute value of the gate-source voltage of the TFT is the absolute value of the threshold voltage of the TFT. It means that the source and drain regions of the TFT become conductive through the channel formation region. When the TFT is turned off, the absolute value of the gate-source voltage of the TFT is the threshold voltage of the TFT. This means that the source region and the drain region of the TFT become nonconductive.

  Further, in this specification, in order to describe the connection of TFTs, “gate electrode, input electrode, output electrode” and “gate electrode, source region, drain region” are used differently. This is because the gate-source voltage is often considered when explaining the operation of the TFT, but it is difficult to clearly distinguish the source and drain regions of the TFT due to the structure of the TFT. When explaining input / output, it is called an input electrode and an output electrode, and when explaining the relationship between the potentials of the electrodes of a TFT, one of the input electrode and the output electrode is called a source region, and the other is called a drain region I will do it.

  In the case of expressing multiple gradations using a light emitting device, an analog gradation method and a digital gradation method can be given. In the case of the former analog gray scale, the current flowing through the EL element is controlled in an analog manner to control the luminance to obtain the gray scale, but there is a slight variation in the characteristics of the TFTs constituting the pixel portion. This greatly affects the variation in brightness. In other words, when the characteristics of the driving TFT 352 vary, even when the same potential is applied to the gate electrodes of different driving TFTs, the values of the source-drain currents in the two differ. That is, since the value of the current flowing through the EL element is different, the luminance varies.

  There is a digital gradation method as a method in which the variation in characteristics of the elements constituting the pixel hardly affects the image quality. In the digital gray scale method, the EL element is driven only in two states, an ON state (a state where the luminance is approximately 100%) and an OFF state (a state where the luminance is approximately 0%). That is, it can be said that it is a driving method in which it is difficult to discriminate the luminance variation of the EL element even if the current between the source and drain of the driving TFT varies.

  However, in the case of the digital gradation method, only two gradations can be displayed as it is, and a plurality of techniques for realizing multi-gradation in combination with another method have been proposed.

  One of the methods for realizing multi-gradation is a method that combines a digital gradation method and a time gradation method. The time gradation method is a method for performing gradation expression by controlling the time during which the EL element emits light. Specifically, one frame period is divided into a plurality of subframe periods having different lengths, and the light emission and non-light emission of the EL element in each period are selected, whereby the length of time during which light is emitted within one frame period. The gradation is expressed with the difference in height.

  As a method of combining the digital gradation method and the time gradation method, a method disclosed in Japanese Patent Laid-Open No. 2001-5426 will be described. Here, as an example, the case of 3-bit gradation expression will be described.

  Reference is made to FIGS. Usually, in a display device such as a liquid crystal display or an EL display, the frame frequency is about 60 [Hz]. That is, as shown in FIG. 9A, the screen is drawn about 60 times per second. Thereby, it is possible to prevent the human eye from feeling flicker (flickering of the screen). At this time, a period in which the screen is drawn once is referred to as one frame period.

  In the time gray scale method disclosed in Japanese Patent Laid-Open No. 2001-5426, one frame period is divided into a plurality of subframe periods. The number of divisions at this time is usually equal to the number of bits of the input digital video signal. Here, since it is a 3-bit gradation, it is divided into three subframe periods SF1 to SF3.

  However, among driving methods aimed at improving display quality, there is a method in which the number of divisions of the frame period is made larger than the number of bits of the digital video signal. An example of such a driving method is described in Japanese Patent Application No. 2000-267164.

Further, each subframe period has an address (writing) period Ta and a sustain (light emission) period Ts. An address (writing) period is a period during which a digital video signal is written to a pixel, and the length in each subframe period is equal. The sustain (light emission) period is a period during which the EL element emits light based on the digital video signal written to the pixel in the address (writing) period. At this time, the length ratio of the sustain (light emission) periods Ts1 to Ts3 is set to Ts1: Ts2: Ts3 = 4: 2: 1. That is, when expressing n-bit gradation, the ratio of lengths of n sustain (light emission) periods is 2 ^n-1 : 2 ^n-2 :...: 2 ¹ : 2 ⁰ . Then, depending on which sustain (light emission) period the EL element emits light, the length of the period during which each pixel emits light is determined per frame period, and gradation expression is thereby performed. That is, in FIG. 9B, by taking either the light emission state or the non-light emission state in each of the sustain (light emission) periods Ts1 to Ts3, the luminance is 0% by using the length of the total light emission time. , 14%, 28%, 43%, 57%, 71%, 86% and 100% can be expressed. When Ts1 emits light and Ts2 and Ts3 do not emit light, the luminance is 57%. When Ts1 and Ts3 emit light and Ts2 does not emit light, the luminance is 71%. That is, in the analog gray scale method, when it is desired to obtain 71% luminance, the luminance is controlled by a voltage or the like according to the luminance, and 71% luminance is maintained over one frame period. In this case, the same gradation is expressed by emitting light for 71% of the entire light emission period at 100% luminance.

  The operation will be specifically described. Continuing to refer to FIGS. 9A to 9C and FIG. First, when a selection pulse is input to the gate signal line, the switching TFT 351 is turned on. Next, a digital video signal is input from the source signal line, the ON / OFF of the driving TFT 352 is controlled by the potential, and the charge is held in the storage capacitor 353.

  At this time, even if the driving TFT 352 is turned on, light is not emitted by preventing voltage from being applied between the anode (cathode) 355 and the cathode (anode) 356 of the EL element 354. As a method, the potential of the cathode (anode) 356 is set equal to the potential of the anode (cathode) 355, that is, the potential of the current supply line (Current). Since the cathode (anode) 356 is normally short-circuited in all pixels, this operation is performed simultaneously in all pixels.

  From the first row to the last row, the address (write) period ends when the write operation is completed, and all the pixels simultaneously shift to the sustain (light emission) period. A voltage difference is generated between the anode (cathode) 355 and the cathode (anode) 356 of the EL element 354, and light is emitted when a current flows.

  By performing the above operation in all subframe periods, one frame period is formed. According to this method, in order to increase the number of display gradations, the number of divisions in the subframe period may be increased. Further, as shown in FIGS. 9B and 9C, the order of the subframe periods does not necessarily have to be the order of upper bits → lower bits, and may be arranged at random during one frame period. Further, the order may change within each frame period.

Problems to be solved by the invention

  By the way, in a display device manufactured by forming a TFT on an insulator, a complicated process leads to a decrease in yield and an increase in cost. Therefore, simplifying the process as much as possible is the main issue for cost reduction. Therefore, it is considered that the pixel portion and peripheral driving circuits (source signal line driving circuit, gate signal line driving circuit, and the like) are configured only by unipolar TFTs.

  Here, the operating voltages of the pixel and the driving circuit are considered again. Reference is now made to FIG. FIG. 5A shows the structure of the pixel portion of the EL element, and FIG. 5B schematically shows the connection of the switching TFT 501, the driving TFT 502, and the EL element 504. FIG.

First, in the EL element 504, a case where 505 is an anode and 506 is a cathode is considered. Now, assuming that the potential of the electrode 505 is V ₅₀₅ and the potential of the electrode 506 is V ₅₀₆ , it is necessary to give a potential difference between the two electrodes in order for the EL element 504 to emit light. Therefore, V ₅₀₅ > V ₅₀₆ . In order to surely turn on when the driving TFT 502 is an N-channel type and to normally apply a voltage between the electrodes of the EL element 504, the potential applied to the gate electrode of the driving TFT 502 is higher than V _505. (At least as much as the threshold value of the TFT 502). That is, it is necessary to increase the amplitude of the signal written from the source signal line. On the other hand, when the driving TFT 502 is a P-channel type, the potential is applied to the gate electrode of the driving TFT 502 to be higher than V ₅₀₅ in order to reliably turn on and apply a voltage normally between the electrodes of the EL element 504. It suffices that the threshold is lowered by at least the threshold value of TFT 502. Therefore, it is not necessary to widen the signal amplitude written from the source signal line. Therefore, in the case where the electrode 505 of the EL element 504 is an anode and the electrode 506 is a cathode, it is desirable to use a P-channel type for the driving TFT 502.

Subsequently, in the EL element 504, when 505 is a cathode and 506 is an anode, in order for the EL element 504 to emit light, it is necessary to apply a potential difference between both electrodes. Therefore, in this case, V ₅₀₅ <V ₅₀₆ . In order to surely turn on when the driving TFT 502 is an N-channel type and to normally apply a voltage between the electrodes of the EL element 504, the potential applied to the gate electrode of the driving TFT 502 is at least higher than V ₅₀₅ It only needs to be higher by the threshold value of the TFT 502. Therefore, the amplitude of the signal written from the source signal line does not need to be so wide. On the other hand, when the driving TFT 502 is a P-channel type, the potential is applied to the gate electrode of the driving TFT 502 to be higher than V ₅₀₅ in order to reliably turn on and apply a voltage normally between the electrodes of the EL element 504. It is necessary to further lower (at least by the threshold value of the TFT 502). That is, it is necessary to increase the amplitude of the signal written from the source signal line. Therefore, when the electrode 505 of the EL element 504 is a cathode and 506 is an anode, it is desirable to use an N-channel type for the driving TFT 502.

  Next, the relationship between the polarity of the driving TFT 502, the configuration of the EL element 504, and the emission direction will be described. 8A is a cross-sectional view of the structure of the EL element 504 when the driving TFT 502 is an N-channel type, and FIG. 8B is a cross-sectional view of the structure of the EL element 504 when the driving TFT 502 is a P-channel type. This is schematically shown in the figure.

In the cathode of the EL element 504, since the ability to inject electrons into the light emitting layer is required, it is desirable to use a metal material. Therefore, an electrode using a transparent electrode is usually an anode. Accordingly, in FIG. 8A, the driving TFT is an N-channel type, the current supply line is connected to the source region of the driving TFT 502, and the cathode of the EL element 504 is connected to the drain region. Therefore, since the light generated in the light emitting layer is emitted to the anode side which is a transparent electrode, the emission direction is the substrate on which the TFT is formed as shown in the figure (hereinafter referred to as the TFT substrate). On the other side.

  On the other hand, in FIG. 8B, the driving TFT 502 is a P-channel type, and a current supply line is connected to the source region of the driving TFT 502 and the anode of the EL element 504 is connected to the drain region. Therefore, since the light generated in the light emitting layer is emitted to the anode side which is a transparent electrode, the emission direction is the TFT substrate side as shown in the figure.

  In this specification, the emission direction shown in FIG. 8A is expressed as top emission, and the emission direction shown in FIG. 8B is expressed as bottom emission. In the case of bottom emission, the area occupied by the elements constituting the pixel portion affects the light emission area, whereas in the case of top emission, light can be extracted regardless of the area occupied by the elements constituting the pixel portion. It is advantageous for high aperture ratio. However, in the case of manufacturing a light emitting device with a top emission structure as shown in FIG. 8A, it is necessary to form an anode using a transparent electrode after forming an EL layer in the process. Since the EL layer is easily damaged and such a process is difficult at present, a bottom emission configuration as shown in FIG. 8B is generally employed.

Here, the polarity of the switching TFT 501 with respect to the polarity of the driving TFT 502 is considered. First, when the driving TFT 502 is a P-channel type, the condition for turning on the driving TFT 502 is that the absolute value of the gate-source voltage V _GS2 of the driving TFT 502 is the absolute value of the threshold voltage of the driving TFT 502. It is to exceed. That is, the L-level potential of the digital video signal input from the source signal line (here, the EL element emits light when the potential of the digital video signal is L level) becomes the potential of the source region of the driving TFT 502. On the other hand, it is lower than the threshold value.

At this time, when the switching TFT 501 has the same polarity as that of the driving TFT 502, that is, a P-channel type, the condition for turning on the switching TFT 501 is that the absolute value of the gate-source voltage V _GS1 of the switching TFT 501 is The absolute value of the threshold voltage is exceeded. That is, an L level potential of a pulse for selecting the gate signal line (here, since the switching TFT 501 is a P-channel type, when the L level is input to the gate signal line, the selected state is set. However, it is lower than the threshold value by the threshold value with respect to the potential of the source region of the switching TFT 501. Therefore, it is necessary to make the voltage amplitude on the gate signal line side wider than the voltage amplitude of the source signal line. This means that the operating voltage of the gate signal line driving circuit is increased.

  The same applies to the case where the switching TFT 501 and the driving TFT 502 are N-channel type. Therefore, in consideration of power consumption, it is desirable that the TFT of the pixel portion is configured using both an N channel type and a P channel type.

  For the above reasons, when the pixel portion and the drive circuit are configured by the unipolar TFT by the conventional method, the process can be reduced, but the power consumption is increased.

  The present invention has been made in view of the above-mentioned problems, and it is possible to reduce the number of steps by configuring a pixel portion and a drive circuit with a single polarity TFT, and to keep power consumption low with a novel circuit configuration. It is an object of the present invention to provide a realized light emitting device.

Means for solving the problem

  In a pixel having a conventional configuration, in order to reliably turn on the switching TFT, the switching TFT is larger than the voltage amplitude of the signal input to the input electrode of the switching TFT, that is, the signal output to the source signal line. Therefore, it is necessary to increase the voltage amplitude of the signal input to the gate electrode, that is, the signal for selecting the gate signal line.

  Here, consider a case where the voltage amplitude of the signal output to the source signal line is equal to the voltage amplitude of the signal for selecting the gate signal line. Refer to FIG. 5 again. Note that all the TFTs constituting the pixel are N-channel type.

The gate signal line is selected, and the switching TFT is turned on. At this time, the potential of the gate electrode of the switching TFT is V ₁ . The switching TFT is turned on, and the video signal output to the source signal line is input to the gate electrode of the driving TFT. At this time, if the video signal is at the H level and the potentials of the input electrode and the gate electrode of the switching TFT are equal to V ₁ , the potential of the signal appearing on the output electrode side of the switching TFT is changed from V ₁ to the switching potential. The potential (V ₁ −VthN) is obtained by subtracting the threshold value of the TFT. When the video signal is at the L level, the threshold value of the switching TFT is not affected, and the L level is equally input to the gate electrode of the driving TFT.

  That is, the voltage amplitude of the video signal is attenuated by the threshold value by the switching TFT. As a result, the potential of the gate electrode of the driving TFT does not rise sufficiently, and a desired drain current may not be obtained. As a result, the current flowing through the EL element is insufficient.

  Therefore, in the present invention, a voltage compensation circuit is provided between the output electrode of the switching TFT and the gate electrode of the driving TFT. The voltage compensation circuit is an application of a bootstrap circuit and has a role of returning the voltage amplitude of a signal attenuated by the threshold value of the switching TFT to the original amplitude.

  Thus, even when the voltage amplitude of the video signal output to the source signal line is equal to the voltage amplitude of the signal for selecting the gate signal line, a normal potential is applied to the gate electrode of the driving TFT. I can do it. Therefore, the driving voltage of the gate signal line driving circuit can be lowered, which contributes to lower power consumption of the light emitting device.

  In addition, the pixel having the voltage compensation circuit of the present invention is configured by a single polarity TFT, and the pixel portion of the light emitting device is configured by using this pixel, and the peripheral drive circuit is configured by the pixel portion. Constituting with a TFT having the same polarity as the TFT contributes to simplification of the manufacturing process.

  The configuration of the present invention will be described below.

The light emitting device of the present invention is
A light-emitting device configured using a plurality of transistors of one conductivity type,
Each pixel included in the light emitting device includes a source signal line, a gate signal line, a current supply line, a switching transistor, a driving transistor, a light emitting element, and a voltage compensation circuit.

The light emitting device of the present invention is
A light-emitting device configured using a plurality of transistors of one conductivity type,
Each pixel of the light emitting device includes a source signal line, a gate signal line, a current supply line, a switching transistor, a driving transistor, a light emitting element, and a voltage compensation circuit.
A gate electrode of the switching transistor is electrically connected to the gate signal line, an input electrode is electrically connected to the source signal line, and an output electrode is electrically connected to the gate electrode of the driving transistor. Connected,
An input electrode of the driving transistor is electrically connected to the current supply line, an output electrode is electrically connected to one electrode of the light emitting element,
The voltage compensation circuit is arranged between an output electrode of the switching transistor and a gate electrode of the driving transistor.

The light emitting device of the present invention is
A light-emitting device configured using a plurality of transistors of one conductivity type,
Among the pixels of the light emitting device, pixels scanned in the m-th row (m is a natural number, 1 ≦ m) are a source signal line, a gate signal line selected in the m-th row, and a current supply line, respectively. A switching transistor, a driving transistor, a light emitting element, and a voltage compensation circuit,
A gate electrode of the switching transistor is electrically connected to a gate signal line selected in the m-th row, an input electrode is electrically connected to the source signal line, and an output electrode is the driving transistor. Electrically connected to the gate electrode of
An input electrode of the driving transistor is electrically connected to the current supply line, an output electrode is electrically connected to one electrode of the light emitting element,
The voltage compensation circuit includes a refresh transistor, a compensation transistor, a first capacitor unit, and a second capacitor unit.
The first electrode of the first capacitor means is electrically connected to the output electrode of the switching transistor, and the second electrode is electrically connected to the first electrode of the second capacitor means. The second electrode is electrically connected to the current supply line;
A gate electrode of the refresh transistor is electrically connected to a gate signal line selected in the (m−1) th row, and an input electrode is electrically connected to a signal line or a power supply line for supplying a first power supply potential. The output electrode is electrically connected to the output electrode of the switching transistor and the gate electrode of the driving transistor,
A gate electrode of the compensation transistor is electrically connected to a first electrode of the first capacitor means, an output electrode of the switching transistor, and a gate electrode of the driving transistor, and the input electrode is a second electrode The output electrode is electrically connected to the second electrode of the first capacitor means and the first electrode of the second capacitor means. It is characterized by being connected.

The light emitting device of the present invention is
A light-emitting device configured using a plurality of transistors of one conductivity type,
Each pixel of the light emitting device is
A source signal line for inputting a video signal to the pixel;
A gate signal line for selecting any one of the pixels;
A light emitting element that emits light according to the input of the video signal;
A current supply line for supplying a current to the light emitting element;
A driving transistor for controlling a current supplied to the light emitting element;
A switching transistor that controls input of the video signal from a source signal line to the gate electrode of the driving transistor;
And a voltage compensation circuit for compensating or converting a voltage amplitude of a signal input to the gate electrode of the driving transistor in accordance with the video signal.

The light emitting device of the present invention is
A light-emitting device configured using a plurality of transistors of one conductivity type,
Each pixel of the light emitting device is
A source signal line for inputting a video signal to the pixel;
A gate signal line for selecting any one of the pixels;
A light emitting element that emits light according to the input of the video signal;
A current supply line for supplying a current to the light emitting element;
A driving transistor for controlling a current supplied to the light emitting element;
A switching transistor that controls input of the video signal from a source signal line to the gate electrode of the driving transistor;
A voltage compensation circuit that performs compensation or conversion of a voltage amplitude of a signal input to the gate electrode of the driving transistor according to the video signal;
A gate electrode of the switching transistor is electrically connected to the gate signal line, an input electrode is electrically connected to the source signal line, and an output electrode is electrically connected to the gate electrode of the driving transistor. Connected,
An input electrode of the driving transistor is electrically connected to the current supply line, an output electrode is electrically connected to one electrode of the light emitting element,
The voltage compensation circuit is arranged between an output electrode of the switching transistor and a gate electrode of the driving transistor.

The light emitting device of the present invention is
A light-emitting device configured using a plurality of transistors of one conductivity type,
Among the pixels of the light emitting device, the pixels scanned in the m-th row (m is a natural number, 1 ≦ m)
A source signal line for inputting a video signal to the pixel;
A gate signal line for selecting the m-th row of the pixels;
A light emitting element that emits light according to the input of the video signal;
A current supply line for supplying a current to the light emitting element;
A driving transistor for controlling a current supplied to the light emitting element;
A switching transistor that controls input of the video signal from a source signal line to the gate electrode of the driving transistor;
A voltage compensation circuit that performs compensation or conversion of a voltage amplitude of a signal input to the gate electrode of the driving transistor according to the video signal;
A gate electrode of the switching transistor is electrically connected to a gate signal line for selecting the m-th row, an input electrode is electrically connected to the source signal line, and an output electrode is connected to the driving transistor. Electrically connected to the gate electrode,
An input electrode of the driving transistor is electrically connected to the current supply line, an output electrode is electrically connected to one electrode of the light emitting element,
The voltage compensation circuit is:
A refresh transistor for applying a certain potential to the gate electrode of the driving transistor;
A compensating transistor for compensating for the voltage amplitude of the video signal input to the gate electrode of the driving transistor;
First capacitive means for forming capacitive coupling between the gate electrode and the output electrode of the compensating transistor;
Second capacitive means for forming capacitive coupling between the output electrode of the compensating transistor and the current supply line;
The first electrode of the first capacitor means is electrically connected to the output electrode of the switching transistor, and the second electrode is electrically connected to the first electrode of the second capacitor means. The second electrode is electrically connected to the current supply line;
A gate electrode of the refresh transistor is electrically connected to a gate signal line selected in the (m−1) th row, and an input electrode is electrically connected to a signal line or a power supply line for supplying a first power supply potential. The output electrode is electrically connected to the output electrode of the switching transistor and the gate electrode of the driving transistor,
A gate electrode of the compensation transistor is electrically connected to a first electrode of the first capacitor means, an output electrode of the switching transistor, and a gate electrode of the driving transistor, and the input electrode is a second electrode The output electrode is electrically connected to the second electrode of the first capacitor means and the first electrode of the second capacitor means. It is characterized by being connected.

In the light emitting device of the present invention,
The signal line or power supply line for supplying the first power supply potential is a gate signal line or the current supply line selected in the m-th row.

In the light emitting device of the present invention,
The signal line or power supply line for supplying the second power supply potential is a gate signal line or the current supply line selected in the m-th row.

In the light emitting device of the present invention,
The first capacitor means is a capacitor means having a capacity between a gate electrode of the compensation transistor and an input electrode or an output electrode of the compensation transistor.

In the light emitting device of the present invention,
The first and second capacitor means are capacitor means made of any two materials among an active layer material, a gate electrode material, and a wiring material, and an insulating layer between the two materials. Yes.

In the light emitting device of the present invention,
The one conductivity type is an N channel type.

In the light emitting device of the present invention,
The one conductivity type is a P-channel type.

The light emitting device of the present invention is
The present invention can be applied to electronic devices such as an OLED display, a video camera, a notebook personal computer, a portable information terminal, a sound reproducing device, a digital camera, and a mobile phone shown in FIG.

  FIG. 1 shows a configuration of a pixel having a voltage compensation circuit of the present invention. As shown in FIG. 1A, a switching TFT 101, a driving TFT 102, an EL element 104, a source signal line (S (n)), a gate signal line (G (m)), and a current supply line (Current) are conventionally used. Have as well. A feature of the present invention is that a voltage compensation circuit 110 is provided between the output electrode of the switching TFT 101 and the gate electrode of the driving TFT 102.

  FIG. 1B is a circuit diagram including the configuration of the voltage compensation circuit 110. The voltage compensation circuit 110 includes a first TFT 151, a second TFT 152, a first capacitor unit 153, and a second capacitor unit 154. In FIG. 1B, G (m) is a gate signal line scanned in the mth row, and G (m−1) is a gate signal line scanned in the m−1th row.

  The first capacitor means 153 and the second capacitor means 154 are arranged in series. The first electrode of the first capacitor unit 153 is connected to the output electrode of the switching TFT 101, and the second electrode of the first capacitor unit 153 is connected to the first electrode of the second capacitor unit 154. The second electrode of the second capacitor means 154 is connected to the current supply line.

The gate electrode of the first TFT 151 is connected to the gate signal line G (m−1), the input electrode is connected to a signal line or a power supply line that supplies the first power supply potential (V ₁ ), and the output electrode is , Connected to the output electrode of the switching TFT 101

The gate electrode of the second TFT 152 is connected to the output electrode of the switching TFT 101 and the first electrode of the first capacitor means, and the input electrode is a signal line that supplies the _second power supply potential (V ₂ ) or The output electrode is connected to the power line, and the output electrode is connected to the second electrode of the first capacitor means and the first electrode of the second capacitor means.

  From now on, regarding the two TFTs included in the voltage compensation circuit, the first TFT 151 will be referred to as a refresh TFT, and the second TFT 152 will be referred to as a compensation TFT.

  Note that the TFTs 101, 102, 151, and 152 constituting the pixel all use TFTs having the same polarity, and the polarity may be N-channel type or P-channel type.

However, the first power supply potential (V ₁ ) and the second power supply potential (V ₂ ) differ depending on the polarities of the TFTs constituting the pixel. When the TFT constituting the pixel is an N-channel type, V ₁ <V _2, and when the TFT constituting the pixel is a P-channel type, V ₁ > V ₂ .

When V ₁ <V ₂ , the potential of V ₁ is sufficiently lower than the threshold value of the N-channel TFT, and the potential of V ₂ is sufficiently higher than the threshold value of the N-channel TFT. For example, the potential of V ₁ is about the L level of the signal line, and the potential of V ₂ is about the H level of the signal line. If V ₁ > V ₂ , the potential may be reversed.

The operation of the circuit will be described. Here, as an example, the TFTs constituting the pixel are all N-channel type. The input signal is VDD when it is at H level and VSS when it is at L level for both the digital video signal output to the source signal line and the signal for selecting the gate signal line. Further, here, V ₁ = VSS and V ₂ = VDD. Further, the potential of the current supply line (Current) is V _C.

  FIG. 11 shows a timing chart for explaining the operation of the circuit of the present invention. (A) is the potential of the gate signal line (G (m-1)) of the m-1st row, (B) is the potential of the gate signal line (G (m)) of the mth row, and (C) is the source signal. The potential of the line (S (n)) and (D) indicate the potential of the gate electrode of the driving TFT 102. Further, a period 1101 from the selection of the m-th gate signal line to the selection of the m-th gate signal line again corresponds to the subframe period (SF #) shown in FIG. A period indicated by 1102 is one horizontal period. The operation will be described with reference to FIGS. 1 and 11 and a pixel in which the switching TFT 101 is controlled by the gate signal line selected in the m-th row.

First, in a period when the gate signal line in the (m-1) th row is selected, that is, in a period in which a video signal is written in the (m-1) th row, the gate signal line in the (m-1) th row is at the H level. The gate signal line in the m-th row is at the L level. Therefore, the switching TFT 101 is turned off and the refresh TFT 151 is turned on. At this time, V ₁ = VSS is input to the gate electrode of the driving TFT 102 and is turned OFF. In FIG. 11, the operation is performed during the period indicated by 1103.

  Subsequently, the horizontal period of the (m−1) th row ends, and the gate signal line (G (m−1)) becomes L level. As a result, the refresh TFT 151 is turned OFF. In the horizontal period of the m-th row, the gate signal line (G (m)) becomes H level. Along with this, the switching TFT 101 is turned ON. At this time, the digital video signal output to the source signal line is written to the pixel. When the digital video signal is at the H level, the switching TFT is turned on, so the potential of the gate electrode of the driving TFT 102 rises.

  However, since the gate signal line (G (m)) is now at the H level, the potential is VDD, and the digital video signal is at the H level, and the potential is also VDD, the potential appearing at the output electrode of the switching TFT. Therefore, the switching TFT is turned off when (VDD−VthN) is reached, and the output electrode of the switching TFT, that is, the gate electrode of the driving TFT 102 is in a floating state.

  On the other hand, since the potential of the output electrode of the switching TFT 101 rises to (VDD−VthN), the compensation TFT 152 is turned on, and the potential of the output electrode rises and approaches VDD. At this time, capacitive coupling by the first capacitive means 153 exists between the output electrode and the gate electrode of the compensation TFT 152. Now, since the gate electrode of the compensation TFT 152 is in a floating state with the potential of (VDD−VthN), it further rises as the potential of the output electrode of the compensation TFT 152 rises and becomes higher than VDD. .

  As a result, the digital video signal once attenuated by VthN through the switching TFT 10 is subjected to amplitude compensation by the voltage compensation circuit 110 and input to the gate electrode of the driving TFT 102. Therefore, the driving TFT 102 can be normally turned on, and a desired drain current can flow.

  Thereafter, the first and second capacitor means 153 and 154 hold the potential applied to the gate electrode of the driving TFT 102 even after the selection of the gate signal line is completed and the address (writing) period is completed. As a result, a drain current flows and the EL element 104 emits light. In the next subframe period, when the gate signal line (G (m−1)) of the (m−1) th row is selected and becomes H level, the refresh TFT 151 is turned on, and the potential of the gate electrode of the drive TFT 102 becomes Becomes L level and turns OFF. Thereafter, the above operation is repeated to draw the screen.

  Here, the first and second capacitor means 153 and 154 will be additionally described.

  The first capacitor unit 153 is a capacitor unit that is disposed between the gate electrode and the output electrode of the compensation TFT 152 and raises the potential of the gate electrode by capacitive coupling using the potential increase of the output electrode. The second capacitor means 154 is arranged in series with the first capacitor means 153, and capacitively couples between the current supply line having a constant potential and the gate electrode of the driving TFT 102, so that the gate electrode of the driving TFT 102 is connected. Capacitance means for holding a potential.

  Here, it is added that the second capacity means 154 is used as a load for reliably performing the bootstrap operation of the voltage guarantee circuit 110 as another role. In the absence of this load, the potential of the gate electrode of the compensation TFT 152 begins to rise due to the input of the digital video signal from the source signal line, and immediately after the threshold value is exceeded, the potential of the output electrode of the compensation TFT 152 rises. . If the potential of the output electrode rises too early, the bootstrap may not work properly. Therefore, by using the second capacitor means 154 as a load, the potential increase of the output electrode of the compensation TFT 152 is intentionally delayed, and the gate electrode is brought into a floating state before the increase of the potential of the output electrode is stopped. Thereby, the bootstrap operation can be performed more reliably.

  By the above method, the voltage amplitude of the gate signal line selection pulse that requires a voltage amplitude larger than the voltage amplitude of the digital video signal normally input to the source signal line can be made equal to the voltage amplitude of the digital video signal. It becomes possible. Therefore, power consumption of the gate signal line driver circuit can be reduced.

  Further, according to the present invention, the potential of the gate electrode of the driving TFT 102 can be made higher than the H level of the digital video signal input from the source signal line by the bootstrap operation. Since the potential of the gate electrode of the driving TFT 102 should normally rise to VDD at the H level, the voltage amplitude of the gate signal line selection pulse can be further reduced by performing detailed estimation of the potential rise due to capacitive coupling. I can do it.

  Examples of the present invention will be described below.

[Example 1]
In the present invention, regarding the first power supply potential (V ₁ ) and the second power supply potential (V ₂ ), V ₁ = VSS and V ₂ = VDD when the polarity of the TFT constituting the pixel is an N-channel type. In the case where the polarity of the TFT constituting the pixel is a P-channel type, V ₁ = VDD and V ₂ = VSS may be used to route the power supply line to the pixel portion, respectively. Will drop.

In this embodiment, connection between elements for supplying a desired power supply potential to V ₁ and V ₂ using existing signal lines such as a source signal line, a gate signal line, and a current supply line will be described.

  2A to 2C show examples of connections.

  In the case of FIG. 2A, the input electrode of the refresh TFT 151 is connected to the current supply line (Current), and the input electrode of the compensation TFT 152 is connected to the m-th gate signal line (G (m)). In the case of FIG. 2B, both the input electrode of the refresh TFT 151 and the input electrode of the compensation TFT 152 are connected to the m-th gate signal line (G (m)). In the case of FIG. 2C, the input electrode of the refresh TFT 151 is connected to the m-th gate signal line (G (m)), and the input electrode of the compensation TFT 152 is connected to the current supply line (Current).

  At this time, the polarity and emission direction of the EL element in each case will be described.

  First, in the pixel scanned in the m-th row, the refresh TFT 151 is turned on only during the selection period of the (m-1) -th row, and the potential of the gate electrode of the driving TFT 102 is lowered to the L level. In the case of FIG. 2A, since the current supply line has a substantially fixed potential, the potential may be set to the L level. In the case of FIGS. 2B and 2C, the gate signal line of the m-th row is at the L level in a period other than the selection period of the m-th row.

  On the other hand, in the pixel scanned in the m-th row, the compensation TFT 152 is turned on during the selection period of the m-th row, and the potential of the output electrode rises, so that the gate electrode of the driving TFT 102 is utilized using capacitive coupling. Increase the potential. Therefore, the signal line to which the input electrode of the compensation TFT 152 is connected needs to be at the H level during the selection period of the m-th row. 2A and 2B, the input electrode of the compensating TFT 152 is connected to the m-th gate signal line (G (m)), and the selection period of the m-th row is a gate signal line selection pulse. Is input, and the gate signal line (G (m)) in the m-th row is at the H level. In the case of FIG. 2C, the input electrode of the compensation TFT 152 is connected to a current supply line (Current). Since the current supply line has a substantially fixed potential, the potential may be set to be equivalent to the H level.

  The above is an example in the case where the TFT constituting the pixel is an N-channel type. However, even when the TFT constituting the pixel is a P-channel type, it is only necessary to reverse the H level and L level of each node and each power supply potential, so the connections shown in FIGS. Can be applied as is.

  From the above, in the case where the TFT constituting the pixel is an N-channel type, in FIGS. 2A and 2B, the potential of the current supply line (Current) may be in the vicinity of the L level. Therefore, since a low potential can be obtained in the circuit, the electrode connected to the driving TFT 102 in the EL element may be a cathode and the common electrode may be an anode. In this case, the emission direction is the anode side, that is, upward emission. On the other hand, in FIG. 2C, the potential of the current supply line (Current) may be in the vicinity of the H level. Accordingly, since a high potential can be obtained in the circuit, an electrode on the side connected to the driving TFT 102 in the EL element may be an anode and a common electrode may be a cathode. In this case, the emission direction is the anode side, that is, downward emission.

  On the other hand, when the TFT constituting the pixel is a P-channel type, the potential is opposite to that described above. That is, in the case of FIGS. 2A and 2B, an electrode on the side connected to the driving TFT 102 in the EL element may be an anode, and a common electrode may be a cathode. In the case of FIG. 2C, an electrode on the side connected to the driving TFT 102 in the EL element may be a cathode and a common electrode may be an anode.

[Example 2]
In this embodiment, a method for simultaneously manufacturing a pixel portion and a TFT of a driver circuit provided around the pixel portion over the same substrate will be described.

First, as shown in FIG. 6A, a silicon oxide film and silicon nitride are formed on a base 5001 made of barium borosilicate glass or aluminoborosilicate glass represented by Corning # 7059 glass or # 1737 glass. A base film 5002 made of an insulating film such as a film or a silicon oxynitride film is formed. Although not particularly illustrated, for the formation of the base film 5002, for example, a silicon oxynitride film formed from SiH ₄ , NH ₃ , and N ₂ O by a plasma CVD method is 10 to 200 [nm] (preferably 50 to 100 [nm]), and a silicon oxynitride silicon film similarly made of SiH ₄ and N ₂ O has a thickness of 50 to 200 [nm] (preferably 100 to 150 [nm]). Are laminated.

  Subsequently, the island-shaped semiconductor layers 5003 to 5005 are semiconductor films having an amorphous structure. A crystalline semiconductor film is formed using a laser crystallization method or a known thermal crystallization method. The island-shaped semiconductor layers 5003 to 5005 are formed to have a thickness of 25 to 80 [nm] (preferably 30 to 60 [nm]). There is no particular limitation on the material of the crystalline semiconductor layer, but it is preferably formed of silicon or a silicon germanium (SiGe) alloy.

In order to manufacture a crystalline semiconductor film by a laser crystallization method, a pulse oscillation type or continuous emission type excimer laser, YAG laser, or YVO ₄ laser is used. In the case of using these lasers, it is preferable to use a method in which laser light emitted from a laser oscillator is linearly collected by an optical system and irradiated onto a semiconductor film. Crystallization conditions are appropriately selected by the practitioner. When an excimer laser is used, the pulse oscillation frequency is set to 30 [Hz], and the laser energy density is set to 100 to 400 [mJ / cm ² ] (typically Is 200 to 300 [mJ / cm ² ]). When a YAG laser is used, the second harmonic is used, the pulse oscillation frequency is 1 to 10 [kHz], and the laser energy density is 300 to 600 [mJ / cm ² ] (typically 350 to 500 [ mJ / cm ² ]). Then, a laser beam condensed linearly with a width of 100 to 1000 [μm], for example, 400 [μm] is irradiated over the entire surface of the substrate, and the overlapping rate (overlap rate) of the linear laser at this time is 80 to It is performed as 98 [%].

Subsequently, a gate insulating film 5006 is formed to cover the island-shaped semiconductor layers 5003 to 5005. The gate insulating film 5006 is formed of an insulating film containing silicon with a thickness of 40 to 150 [nm] by using a plasma CVD method or a sputtering method. In this embodiment, a silicon oxynitride film is formed with a thickness of 120 [nm]. Needless to say, the gate insulating film is not limited to such a silicon oxynitride film, and another insulating film containing silicon may be used as a single layer or a stacked structure. For example, when silicon oxide is used, TEOS (Tetraethyl Orthosilicate) and O ₂ are mixed by a plasma CVD method to obtain a reaction pressure of 40 [Pa], a substrate temperature of 300 to 400 [° C.], and a high frequency (13.56 [13 [ NHz]) and can be formed by discharging at a power density of 0.5 to 0.8 [W / cm ² ]. The silicon oxide film thus produced can obtain good characteristics as a gate insulating film by subsequent thermal annealing at 400 to 500 [° C.].

  Then, a first conductive film 5007 and a second conductive film 5008 for forming a gate electrode are stacked over the gate insulating film 5006. In this embodiment, the first conductive layer 5007 is formed of tantalum (Ta) with a thickness of 50 to 100 [nm], and the second conductive layer 5009 is formed of tungsten (W) with a thickness of 100 to 300 [nm]. (FIG. 6A).

  The Ta film is formed by sputtering, and a Ta target is sputtered with Ar. In this case, when an appropriate amount of Xe or Kr is added to Ar, the internal stress of the Ta film can be relieved and peeling of the film can be prevented. The resistivity of the α-phase Ta film is about 20 [μΩcm] and can be used as a gate electrode, but the resistivity of the β-phase Ta film is about 180 [μΩcm] and is not suitable for the gate electrode. is there. In order to form an α-phase Ta film, tantalum nitride (TaN) having a crystal structure close to that of the α-phase of Ta is formed on the Ta base with a thickness of about 10 to 50 [nm]. A Ta film can be easily obtained.

When forming a W film, it is formed by sputtering using W as a target. In addition, it can be formed by a thermal CVD method using tungsten hexafluoride (WF ₆ ). In any case, in order to use as a gate electrode, it is necessary to reduce the resistance, and it is desirable that the resistivity of the W film be 20 [μΩcm] or less. The resistivity of the W film can be reduced by increasing the crystal grains. However, when there are many impurity elements such as oxygen in W, crystallization is hindered and the resistance is increased. From this, in the case of the sputtering method, by using a W target having a purity of 99.9999 [%] and further forming a W film with sufficient consideration so that impurities are not mixed in the gas phase during film formation, A resistivity of 9 to 20 [μΩcm] can be realized.

  Note that in this embodiment, the first conductive film 5007 is Ta and the second conductive film 5008 is W, but there is no particular limitation, and any element selected from Ta, W, Mo, Al, and Cu Alternatively, an alloy material or a compound material containing the above element as a main component may be used. Alternatively, a semiconductor film typified by a polycrystalline silicon film doped with an impurity element such as phosphorus may be used. As an example of another combination other than the present embodiment, a combination in which the first conductive film is TaN, the second conductive film is W, the first conductive film is TaN, and the second conductive film is Al. A combination of TaN as the first conductive film and Cu as the second conductive film is desirable.

Next, a resist mask 5009 is formed, and a first etching process is performed to form electrodes and wirings. In this embodiment, an ICP (Inductively coupled plasma) etching method is used, CF ₄ and Cl ₂ are mixed in an etching gas, and 500 [W] is applied to a coil-type electrode at a pressure of 1 [Pa]. RF (13.56 [MHz]) power is input to generate plasma. An RF power of 100 [W] is also applied to the substrate side (sample stage), and a substantially negative self-bias voltage is applied. When CF ₄ and Cl ₂ are mixed, the W film and the Ta film are etched to the same extent.

  Under the above etching conditions, the end portions of the first conductive film and the second conductive film are tapered due to a suitable mask shape by the resist and the effect of the bias voltage applied to the substrate side. The angle of the tapered portion is 15 to 45 °. In order to perform etching without leaving a residue on the gate insulating film, it is preferable to increase the etching time at a rate of 10 to 20%. Since the selection ratio of the silicon oxynitride film to the W film is 2 to 4 (typically 3), the surface where the silicon oxynitride film is exposed is etched by about 20 to 50 [nm] by the over-etching process. become. Thus, the first shape conductive layers 5010 to 5013 including the first conductive layers 5010a to 5013a and the second conductive layers 5010b to 5013b are formed by the first etching treatment. At this time, in the gate insulating film 5006, a region which is not covered with the first shape conductive layers 5010 to 5013 is etched and thinned by about 20 to 50 [nm] (FIG. 6B). ).

Then, a first doping process is performed, and an impurity element imparting N-type conductivity is added (FIG. 6B). The doping process may be performed by an ion doping method or an ion implantation method. The conditions for the ion doping method are a dose of 1 × 10 ^{13 to} 5 × 10 ¹⁴ [atoms / cm ² ] and an acceleration voltage of 60 to 100 [keV]. As the impurity element imparting N-type, an element belonging to Group 15, typically phosphorus (P) or arsenic (As), is used here, but P is used. In this case, the conductive layers 5010 to 5013 serve as a mask for the impurity element imparting N-type, and the first impurity regions 5014 to 5016 are formed in a self-aligning manner. An impurity element imparting N-type conductivity is added to the first impurity regions 5014 to 5016 in a concentration range of 1 × 10 ²⁰ to 1 × 10 ²¹ [atoms / cm ³ ].

Next, a second etching process is performed (FIG. 6C). Similarly, using an ICP etching method, CF ₄ , Cl _2, and O ₂ are mixed in an etching gas, and RF power of 500 [W] is supplied to the coil-type electrode at a pressure of 1 [Pa], and plasma is supplied. Generate and do. An RF power of 50 [W] is also applied to the substrate side (sample stage), and a lower self-bias voltage is applied than in the first etching process. Under such conditions, the second conductive layer W is anisotropically etched, and the first conductive layer Ta is anisotropically etched at a slower etching rate to form the second shape conductive layer. 5017 to 5020 (first conductive layers 5017a to 5020a and second conductive layers 5017b to 5020b) are formed. At this time, in the gate insulating film 5006, regions that are not covered with the second shape conductive layers 5017 to 5020 are further etched by about 20 to 50 [nm] to form thinned regions.

The etching reaction of the W film or Ta film with the mixed gas of CF ₄ and Cl ₂ can be estimated from the generated radicals or ion species and the vapor pressure of the reaction product. When the vapor pressures of W and Ta fluorides and chlorides are compared, the vapor pressure of WF ₆ , which is a fluoride of W, is extremely high, and the other WCl ₅ , TaF ₅ , and TaCl ₅ are similar. Therefore, both the W film and the Ta film are etched with a mixed gas of CF ₄ and Cl ₂ . However, when an appropriate amount of O ₂ is added to this mixed gas, CF ₄ and O ₂ react to form CO and F, and a large amount of F radicals or F ions are generated. As a result, the etching rate of the W film having a high fluoride vapor pressure is increased. On the other hand, even if Ta increases, the etching rate increases relatively little. Further, since Ta is more easily oxidized than W, the surface of Ta is oxidized by adding O ₂ . Since Ta oxide does not react with fluorine or chlorine, the etching rate of the Ta film is further reduced. Therefore, it becomes possible to make a difference in the etching rate between the W film and the Ta film.

Then, a second doping process is performed (FIG. 6D). In this case, doping is performed with an impurity element that imparts N-type as a condition of a high acceleration voltage by lowering the dose than in the first doping process. For example, the acceleration voltage is set to 70 to 120 [keV] with a dose of 1 × 10 ¹³ [atoms / cm ² ], and the first impurity region formed in the island-shaped semiconductor layer in FIG. A new impurity region is formed inside. Doping is performed by using the second conductive layers 5017b to 5020b as masks against the impurity elements and adding the impurity elements to the lower regions of the first conductive layers 5017a to 5020a.
Thus, second impurity regions 5021 to 5023 overlapping with the first conductive layer are formed.

Subsequently, a third etching process is performed (FIG. 7A). Here, Cl ₂ is used as an etching gas, and an ICP etching apparatus is used. In this embodiment, the gas flow rate ratio of Cl ₂ is set to 60 [sccm], RF power of 350 [W] is applied to the coil-type electrode at a pressure of 1 [Pa], plasma is generated, and etching is performed for 70 seconds. went. RF power is also applied to the substrate side (sample stage), and a substantially negative self-bias voltage is applied. The third etching causes the first conductive layer to recede to form third-shaped conductive layers 5024 to 5027 (first conductive layers 5024a to 5027a and second conductive layers 5024b to 5027b), and second The impurity regions 5021 to 5023 become second impurity regions 5028a to 5030a that overlap with the first conductive layer and third impurity regions 5028b to 5030b that do not overlap with the first conductive layer.

  Through the above steps, impurity regions are formed in each island-shaped semiconductor layer. The third shape conductive layers 5024 to 5026 overlapping with the island-shaped semiconductor layers function as gate electrodes of the TFTs. The third shape conductive layer 5027 functions as a source signal line.

  Subsequently, for the purpose of controlling the conductivity type, a step of activating the impurity element added to each island-shaped semiconductor layer is performed. This step is performed by a thermal annealing method using a furnace annealing furnace. In addition, a laser annealing method and a rapid thermal annealing method (RTA method) can be applied. In the thermal annealing method, oxygen concentration is 1 [ppm] or less, preferably 0.1 [ppm] or less in a nitrogen atmosphere at 400 to 700 [° C.], typically 500 to 600 [° C.], In this embodiment, heat treatment is performed at 500 [° C.] for 4 hours. However, when the wiring material used for 5024 to 5027 is vulnerable to heat, it is desirable to perform thermal activation after forming an interlayer insulating film (mainly silicon) in order to protect the wiring and the like.

  Further, a heat treatment is performed at 300 to 450 [° C.] for 1 to 12 hours in an atmosphere containing 3 to 100% hydrogen to perform a step of hydrogenating the island-shaped semiconductor layer. This step is a step of terminating dangling bonds in the semiconductor layer with thermally excited hydrogen. As another method of thermal hydrogenation for hydrogenation, plasma hydrogenation (using hydrogen excited by plasma) may be used.

  Next, as shown in FIG. 7B, a first interlayer insulating film 5031 is formed using a silicon oxynitride film with a thickness of 100 to 200 [nm]. After a second interlayer insulating film 5032 made of an organic insulating material is formed thereon, contact holes are opened in the first interlayer insulating film 5031, the second interlayer insulating film 5032, and the gate insulating film 5006. After forming a film of a wiring material and patterning each of the wirings 5033 to 5037 and the connection electrode 5038, the pixel electrode 5039 is formed by patterning so as to be in contact with the connection electrode 5038.

  In this specification, a substrate in which wirings 5033 to 5037 and connection electrodes 5038 are formed is referred to as an active matrix substrate.

  As the second interlayer insulating film 5032, a film made of an organic resin such as polyimide, polyamide, acrylic, or BCB (benzocyclobutene) is used. In particular, since the second interlayer insulating film 5032 has a strong meaning of flattening, acrylic having excellent flatness is desirable. In this embodiment, the acrylic film is formed with a film thickness that can sufficiently flatten the step formed by the TFT. Preferably, it may be 1 to 5 [μm] (more preferably 2 to 4 [μm]).

  The contact holes are formed by dry etching or wet etching, using N-type impurity regions 5014 to 5016, source signal lines 5027, gate signal lines (not shown), current supply lines (not shown), and gate electrodes. Contact holes reaching 5024 to 5026 (not shown) are formed.

  Further, as the wirings 5033 to 5038, a Ti film of 100 [nm], an Al film containing Ti of 300 [nm], a Ti film of 150 [nm], and a three-layer laminated film continuously formed by sputtering are formed in a desired shape. It is formed by patterning. Of course, other conductive materials may be used.

  In this embodiment, the pixel electrode (reflecting electrode) 5039 is formed and patterned to a thickness of 200 [nm] using MgAg or the like. Contact is made by arranging the pixel electrode 5039 so as to overlap the connection electrode 5038.

  Next, as illustrated in FIG. 7C, an insulating film is formed to a thickness of about 1 to 3 μm using an organic material such as acrylic, and an opening is formed at a position corresponding to the pixel electrode 5039. Then, a third interlayer insulating film 5040 is formed. When forming the opening, it is desirable to perform etching so that the side wall has a tapered shape. If the side wall is not sufficiently gentle, deterioration of the EL layer, step breakage, and the like due to the step become a significant problem.

  Subsequently, after the EL layer 5041 is formed by using a vacuum evaporation method, a counter electrode (transparent electrode) 5042 is formed. The film thickness of the EL layer may be 80 to 200 [um] (typically 100 to 120 [nm]), and the film thickness of the pixel electrode (transparent electrode) 5042 may be 110 [nm].

  In this step, an EL layer and a pixel electrode (transparent electrode) are sequentially formed for a pixel corresponding to red, a pixel corresponding to green, and a pixel corresponding to blue. However, since the EL layer has poor resistance to the solution, it must be formed individually for each color without using a photolithography technique. Therefore, it is desirable to cover other than the desired pixel with a metal mask or the like and selectively form the EL layer and the pixel electrode (transparent electrode) only at necessary portions.

  Here, a method of forming three types of EL elements corresponding to RGB is used, but a method of combining a white light emitting EL element and a color filter, a blue or blue green light emitting EL element, and a phosphor (fluorescent material). A method combining a color conversion layer (CCM) may be used.

  Note that a known material can be used for the EL layer 5041. As the known material, it is desirable to use an organic material in consideration of the driving voltage.

  Through the steps so far, a cathode made of MgAg, an EL layer, and an anode made of a transparent conductive film are formed. Next, a passivation film made of a silicon nitride film is formed as a protective film 5043 to a thickness of 50 to 300 [nm]. This protective film 5043 protects the EL layer from moisture and the like.

  In actuality, when the state shown in FIG. 7C is completed, a protective film (laminate film, ultraviolet curable resin film, etc.) or a light-transmitting material having high hermeticity and low outgassing is used so as not to be exposed to the outside air. It is preferable to package (enclose) with a sealing material. At this time, if the inside of the sealing material is made an inert atmosphere or a hygroscopic material (for example, barium oxide) is arranged inside, the reliability of the EL element is improved.

  In addition, when the airtightness is improved by processing such as packaging, a connector (flexible printed circuit: FPC) for connecting the terminal routed from the element or circuit formed on the substrate and the external signal terminal is attached. Completed as a product. In this specification, such a state that can be shipped is referred to as a light emitting device.

  Further, according to the steps shown in this embodiment, the number of photomasks necessary for the production of the active matrix substrate is four (an island semiconductor layer pattern, a first wiring pattern (a gate wiring, an island source wiring, a capacitor wiring). ), A contact hole pattern, and a second wiring pattern (including connection electrodes). As a result, the process can be shortened, and the manufacturing cost can be reduced and the yield can be improved.

[Example 3]
In the process shown in Embodiment 2, the TFT constituting the driver circuit and the pixel is a TFT having a normal single gate structure, but the present invention has an active layer sandwiched as shown in FIG. You may implement using the TFT of the structure which has several gate electrodes. The manufacturing process will be described below.

  A conductive film made of a conductive material is formed on a base 7001 made of barium borosilicate glass or alumino borosilicate glass typified by Corning # 7059 glass or # 1737 glass, and patterning is performed as shown in FIG. A lower gate electrode 7002 is formed as shown in FIG. The material constituting the lower gate electrode is not particularly limited as long as it is a conductive material, but typically Ta, W, or the like may be used.

  Next, a first insulating film 7003 is formed. The first insulating film 7003 is formed using silicon oxynitride to a thickness of 10 to 50 [nm].

  The surface at the time when the first insulating film 7003 is formed has unevenness caused by the lower gate electrode 7002 as shown in FIG. Considering the subsequent manufacturing process, it is desirable to flatten the unevenness. Here, CMP (Chemical Mechanical Polishing) is used as a planarization method. CMP is one of polishing methods for obtaining a precise smooth surface by applying a chemical treatment to the surface of an object to be polished so that the surface is easily polished and then mechanically polishing.

  A silicon oxide film or a silicon oxynitride film is formed as a planarization film 7004 to a thickness of 0.5 to 1 [μm] over the first insulating film 7003. As the CMP abrasive (slurry) for the planarizing film 7004, for example, fumed silica particles obtained by thermally decomposing silicon chloride gas in a KOH-added aqueous solution may be used. The planarizing film 7004 is polished and removed by about 0.5 to 1 [μm] by CMP to planarize the surface.

  In this way, a state in which the surface is flattened is obtained as shown in FIG. Thereafter, TFTs may be formed in accordance with Embodiment 4 to form peripheral circuits and pixels.

  The TFT manufactured here has a gate electrode and a lower gate electrode that overlap with each other with an active layer interposed therebetween. Here, when quick response is required, such as a switching circuit, a signal is input to both the lower gate electrode 7002 and the gate electrode 7006. By inputting the same signal to both gate electrodes, the depletion of the channel region in the active layer progresses quickly, the field effect mobility can be increased and the current capability can be increased, so that quick response can be expected. .

  On the other hand, when the characteristics are required to be uniform and the off-leakage current is reduced as in the driving TFT in the pixel portion, a signal is input to the gate electrode and the lower gate electrode is fixed to a certain potential. Used as described above. The certain constant potential at this time is a potential that reliably turns off when the potential is applied to the gate electrode of the TFT. Typically, when the TFT is an N-channel type, it may be connected to a low-potential side power source such as VSS, and when the TFT is a P-channel type, it may be connected to a high-potential power source such as VDD. In this case, variation in threshold voltage can be reduced as compared with a TFT having a structure having no lower gate electrode. Furthermore, it is effective because it can be expected to reduce the off-leakage current.

[Example 4]
In this example, an example of manufacturing a light-emitting device having the pixel described in the embodiment mode will be described.

  FIG. 4 shows a schematic diagram of the light emitting device. A pixel portion 401 is disposed at the center of the substrate 400. Although not particularly shown in FIG. 4, the configuration of one pixel is as shown in FIG. A source signal line driver circuit 402 for controlling the source signal line and a gate signal line driver circuit 407 for controlling the gate signal line are arranged around the pixel portion 401. The gate signal line driver circuit 407 is arranged symmetrically on both sides of the pixel portion 401 in FIG. 4, but may be arranged on only one side. However, it is desirable to arrange them symmetrically on both sides in terms of circuit operation efficiency and reliability.

  Signals input from the outside for driving the source signal line driver circuit 402 and the gate signal line driver circuit 407 are input via the FPC 410. In this embodiment, the signal input from the FPC 410 is in accordance with a commonly used IC drive voltage, and since the voltage amplitude is small, the voltage amplitude is converted by the level shifter 406. Thus, the signal is input to the source signal line driver circuit 402 and the gate signal line driver circuit 407.

  FIG. 13 shows the configuration of the source signal line driving circuit. A shift register 1303, a buffer 1304, a first latch circuit 1305, and a second latch circuit 1306 are included. Although the buffer is not shown in FIG. 20, when the load below the shift register is large, a buffer may be provided as shown in FIG.

  The source signal line driver circuit includes a source side clock signal (SCLK), a source side clock inverted signal (SCLKb), a source side start pulse (SSP), a scanning direction switching signal (LR), a scanning direction switching inverted signal (LRb), Digital video signals (Data 1 to 3) are input. Among these, the clock signal and the start pulse are input after undergoing amplitude conversion by the level shifters 1301 and 1302.

  FIG. 14 shows the structure of the shift register. In the block diagram shown in FIG. 14A, the block shown by 1400 is a pulse output circuit that outputs a sampling pulse for one stage, and the shift register of FIG. 14A has n stages (n is a natural number, 1 <n) pulse output circuit.

  FIG. 14B shows the configuration of the pulse output circuit in detail. Here, TFTs 1407, 1408, 1409, and 1410 are switching TFTs provided for switching the scanning direction. Switching between the left and right scanning directions is performed by a scanning direction switching signal (LR) and a scanning direction switching inversion signal (LRb). I do.

  In the case of forward scanning, the output of the sampling pulse is in the order of the first stage, the second stage,..., The (n−1) th stage, and the nth stage. The n-th stage, the (n-1) -th stage,..., the second stage, and the first stage.

  The pulse output circuit main body includes TFTs 1401 to 1406 and a capacitor 1411. In a pulse output circuit of a certain k-th stage (k is a natural number, 1 <k <n), the gate electrodes of the TFTs 1401 and 1404 and the gate electrodes of the TFTs 1402 and 1403 are respectively supplied from the k-1th stage pulse output circuit. Either an output pulse or an output pulse from the pulse output circuit at the k + 1 stage is input. Note that a start pulse (SP) is input to k = 1, that is, the gate electrodes of the TFTs 1401 and 1404 in the first-stage pulse output circuit and k = n, that is, the gate electrodes of the TFTs 1402 and 1403 in the last-stage pulse output circuit. The

  In the forward scanning direction, the scanning direction switching signal (LR) receives the Hi potential, and the scanning direction switching inversion signal (LRb) receives the Lo potential. Accordingly, the TFTs 1407 and 1410 are turned on, and the output pulses from the (k−1) th stage pulse output circuit are input to the gate electrodes of the TFTs 1401 and 1404. On the other hand, the output pulses from the (k + 1) th stage pulse output circuit are input to the gate electrodes of the TFTs 1402 and 1403.

  Here, a detailed circuit operation will be described by taking forward scanning as an example. Reference is made to the timing chart shown in FIG.

In a certain pulse output circuit of the k-th stage, an output pulse from the pulse output circuit of the (k−1) -th stage is input to the gate electrodes of the TFTs 1401 and 1404.
(k = 1, that is, in the case of the first stage, a start pulse is input) It becomes Hi potential, and the TFTs 1401 and 1404 are turned on (see 1501 in FIG. 15). Accordingly, the potential of the gate electrode of the TFT 1405 is raised to the VDD side (see FIG. 15 1502), and when the potential becomes VDD−VthN, the TFT 1401 is turned off and becomes a floating state. At this time, the gate-source voltage of the TFT 1405 exceeds the threshold value, and the TFT 1405 is turned on. On the other hand, the gate electrodes of the TFTs 1402 and 1403 have no pulse input yet and remain at the Lo potential. Therefore, since the potential of the gate electrode of the TFT 1406 is Lo and is OFF, the lock signal (either SCLK or SCLKb) input to the input electrode of the TFT 1405 is set to the Hi potential at the output terminal (SR Out). Accordingly, the potential of the output terminal (SR Out) of the pulse output circuit is raised to the VDD side (see 1503 in FIG. 15). However, in the state up to this point, the potential of the output terminal (SR Out) of the pulse output circuit is further lowered to VDD-2 (VthN), which is lower than the potential VDD-VthN of the gate electrode of the TFT 1405 by a threshold value. It can only rise.

  Here, since a capacitor 1411 is provided between the gate electrode and the output electrode of the TFT 1405, and the gate electrode of the TFT 1405 is in a floating state now, the potential of the output terminal (SR Out) of the pulse output circuit. Increases, that is, as the potential of the output electrode of the TFT 1405 increases, the potential of the gate electrode of the TFT 1405 is further raised from VDD-VthN by the action of the capacitor 1411. By this operation, the potential of the gate electrode of the TFT 1405 is finally higher than VDD + VthN (see FIG. 15 1502). The potential of the output terminal (SR Out) of the pulse output circuit normally rises to VDD without being affected by the threshold value of the TFT 1405 (see FIG. 15 1503).

  Similarly, a pulse is output from the pulse output circuit of the (k + 1) th stage (see 1504 in FIG. 15). The output pulse at the (k + 1) th stage is fed back to the kth stage and input to the gate electrodes of the TFTs 1402 and 1403. The potentials of the gate electrodes of the TFTs 1402 and 1403 are turned on as Hi, and the potential of the gate electrode of the TFT 1405 is lowered to the VSS side, so that the TFT 1405 is turned off. At the same time, the potential of the gate electrode of the TFT 1406 is turned ON as the Hi potential, and the potential of the output terminal (SR Out) of the k-th pulse output circuit becomes the Lo potential.

  Thereafter, pulses having an amplitude between VDD and VSS are sequentially output by the same operation up to the final stage. The circuit operation is the same in reverse scanning.

  In the final stage, since there is no pulse that is fed back from the next stage, the clock signal continues to be output through the TFT 1405 (see FIG. 15 1507). Therefore, the output pulse of the last pulse output circuit cannot be used as a sampling pulse. Similarly, in the case of reverse scanning, since the output pulse at the first stage becomes the final output, it cannot be used as a sampling pulse as well. Therefore, in the circuit shown in this embodiment, a shift register is configured by using a required number of stages + 2 stages of pulse output circuits, and both ends are treated as dummy stages (in FIG. 13, the buffer 1304 is not connected). The pulse output circuit at both ends corresponds to the dummy stage). Still, since the final output needs to be stopped in some way before the next horizontal period starts, the start pulse is input in the next horizontal period using the start pulse as the first stage input and the last stage input. At this point, the output of the final stage is stopped.

  FIG. 16 shows the configuration of the buffer 1304 used in the light emitting device of this embodiment. As shown in FIG. 16A, it has a four-stage configuration of 1601 to 1604, and the first stage is a 1-input 1-output type, and the second and subsequent stages are a 2-input 2-output type.

  FIG. 16B shows a circuit configuration of the first stage unit 1601. A signal is input to the gate electrodes of the TFTs 1652 and 1654. The gate electrode of the TFT 1651 is connected to the input electrode. When the Hi potential is input to the gate electrodes of the TFTs 1652 and 1654 and turned ON, the potential of the gate electrode of the TFT 1653 becomes the Lo potential, and as a result, the output terminal (Out) becomes the Lo potential. When the Lo potential is input to the gate electrodes of the TFTs 1652 and 1654 and the TFT 1651 is turned off because the gate electrode and the input electrode are connected to each other, the potential of the gate electrode of the TFT 1653 rises, and the shift described above is performed. As in the case of the register, the output becomes the Hi potential due to the coupling by the capacitor 1655.

  Note that as a relationship between the TFTs 1651 and 1652, since the gate electrode and the input electrode of the TFT 1651 are connected, when the TFT 1652 is turned on, both the TFT 1651 and the TFT 1652 are turned on. In this state, since the potential of the gate electrode of the TFT 1653 needs to be the Lo potential, the channel width of the TFT 1651 needs to be designed to be smaller than that of the TFT 1652. The channel width of the TFT 1651 may be minimized because it is sufficient to have the capacity to charge one gate electrode of the TFT 1653. Further, by reducing the TFT 1651, an increase in current consumption due to a through-path between VDD-TFT 1651-TFT 1652-VSS during a period in which the TFT 1652 is ON can be minimized.

  FIG. 16C shows a circuit configuration of units used in the second and subsequent stages. The input to the gate electrode of the TFT 1652 is the same as that of the first stage. In addition, the input of the previous stage is used as the inverting input for the gate electrode of the TFT 1651. In this way, the TFTs 1651 and 1652 are exclusively turned on and off, and the through-path between VDD-TFT1651 -TFT1652 -VSS in the configuration of FIG. 16B can be eliminated.

FIG. 17 shows the configuration of the clock signal level shifter (A) and the start pulse level shifter (B) used in the light emitting device of this embodiment. The basic configuration is a four-stage configuration in which the first stage is a level shifter and the second and subsequent stages are buffers, and is the same as the buffer circuit described above. A signal having an amplitude between VDD _{LO and} VSS is input, and an output signal having an amplitude between VDD and VSS is obtained (where | VDD _LO | <| VDD |).

  In the case of a clock signal level shifter, the first stage is a 1-input 1-output type, and the second and subsequent stages are 2-input 1-output types. For each input, each other's input is used as an inverting input.

  In the case of the start pulse level shifter, the configuration is the same as that of the buffer described above.

FIG. 17C shows the circuit configuration of the unit used in the first stage of the level shifter, and FIG. 17D shows the circuit configuration of the unit used in the second and subsequent stages.
Each circuit configuration and operation are the same as those shown in FIGS. 16B and 16C, except that the amplitude of the signal input to the first stage is between VDD _{LO and} VSS.

When the signal input to the gate electrode of the TFT 1752 is at the Hi potential, the TFT 1752 is turned on (however, the absolute value | VDD _LO −VSS | of the amplitude of the input signal is larger than the absolute value | VthN | of the threshold value of the TFT 1752). When it is surely large), the potential of the gate electrode of the TFT 1753 is lowered to the VSS side. Therefore, the Lo potential appears at the output terminal (Out). On the other hand, when the signal input to the gate electrode of the TFT 1752 is Lo potential, the TFT 1752 is turned off, and the potential of the gate electrode of the TFT 1753 is raised to the VDD side through the TFT 1751. Subsequent operations are the same as those of the buffer described above.

  A characteristic of the level shifter of this configuration is that an input signal is not directly input to the gate electrode in controlling the TFT 1751 connected to the high potential side (VDD side). Therefore, even when the amplitude of the input signal is small, the potential of the gate electrode of the TFT 1753 can be raised regardless of the threshold value of the TFT 1751, so that a high amplitude conversion gain can be obtained.

  FIG. 18 shows the configuration of the first and second latch circuits used in the light emitting device of this embodiment. As a configuration example of a latch circuit having a conventional CMOS structure, as shown in FIG. 21A, a latch circuit generally including a holding unit in which two inverters are connected in a loop and a switch for controlling holding timing is used. In addition, the configuration shown in FIG. 21B using a D-FF (flip-flop) circuit is also included. FIG. 21C is based on the simplest DRAM configuration, and the holding portion is configured by an inverter and a capacitor, and is input to the inverters of the first latch circuit (LAT1) and the second latch circuit (LAT2). The capacitor holds this potential. In this embodiment, the simplest configuration shown in FIG. 21C was used.

  The latch circuit shown in FIG. 18 has a configuration in which the analog switch in FIG. 21C is replaced with one N-channel TFT and the CMOS inverter is replaced with an NMOS inverter composed of four N-channel TFTs and a capacitor. Yes.

  When a digital video signal is input from the input electrode of the TFT 1850 (Data In), a sampling pulse is input to the gate electrode (Pulse In), and the TFT 1850 is turned on, the digital video signal is input to an inverter composed of TFTs 1851 to 1854 and a capacitor 1855. The polarity is inverted and output. The digital video signal is held using a capacitor 1856.

  In the second latch circuit, the digital video signal is written and held according to the input timing of the latch pulse (LAT) by the same operation.

  FIG. 12 shows a circuit configuration of the gate signal line driving circuit. A shift register 1203 and a buffer 1204 are included.

  A gate side clock signal (GCLK), a gate side clock inversion signal (GLKb), and a gate side start pulse (GSP) are input to the gate signal line driver circuit. These input signals are input after undergoing amplitude conversion by the level shifters 1201 and 1202.

  Note that the configuration and operation of the shift register 1203, the buffer 1204, the start pulse level shifter 1201, and the clock signal level shifter 1202 are the same as those used in the source signal line driver circuit, and thus description thereof is omitted here.

  In FIG. 19, since the gate signal line in the row indicated by α cannot obtain the gate signal line selection pulse input in the previous row in the pixel in the first row, the dummy stage (refresh operation in the pixel in the first row) Dedicated).

  A display device manufactured using the driving circuit introduced here and the pixel shown in the embodiment mode of the present invention is configured by using only a single-polarity TFT so that a part of the doping process in the process is performed. The number of photomasks can be reduced. Furthermore, the problem described in the above-mentioned problem section, such as an increase in current consumption by expanding the signal amplitude, can be solved by using a circuit to which the bootstrap method is applied.

[Example 5]
The light-emitting device of the present invention can be applied to manufacture of display devices used in various electronic devices. Examples of such electronic devices include portable information terminals (electronic notebooks, mobile computers, mobile phones, etc.), video cameras, digital cameras, personal computers, televisions, mobile phones, and the like. An example of these is shown in FIG.

  FIG. 19A illustrates an OLED display, which includes a housing 3001, a support base 3002, a display housing 3003, and the like. The present invention can be applied to the display portion 3003.

  FIG. 19B illustrates a video camera, which includes a main body 3011, a display portion 3012, an audio input portion 3013, operation switches 3014, a battery 3015, an image receiving portion 3016, and the like. The present invention can be applied to the display portion 3012.

  FIG. 19C illustrates a laptop personal computer, which includes a main body 3021, a housing 3022, a display portion 3023, a keyboard 3024, and the like. The present invention can be applied to the display portion 3023.

  FIG. 19D illustrates a portable information terminal which includes a main body 3031, a stylus 3032, a display portion 3033, operation buttons 3034, an external interface 3035, and the like. The present invention can be applied to the display portion 3033.

  FIG. 19E illustrates a sound reproducing device, specifically an in-vehicle audio device, which includes a main body 3041, a display portion 3042, operation switches 3043 and 3044, and the like. The present invention can be applied to the display portion 3042. In this embodiment, the in-vehicle audio device is taken as an example, but it may be used for a portable or home audio device.

  FIG. 19F illustrates a digital camera, which includes a main body 3051, a display portion (A) 3052, an eyepiece portion 3053, operation switches 3054, a display portion (B) 3055, a battery 3056, and the like. The present invention can be applied to the display portion (A) 3052 and the display portion (B) 3055.

  FIG. 19G illustrates a cellular phone, which includes a main body 3061, an audio output portion 3062, an audio input portion 3063, a display portion 3064, operation switches 3065, an antenna 3066, and the like. The present invention can be applied to the display portion 3064.

  It should be noted that the examples shown in the present embodiment are only examples and are not limited to these applications.

Effect of the invention

  In the light emitting device of the present invention, a pixel portion and a peripheral driving circuit are integrally formed using a single polarity TFT. As a result, part of the doping process is reduced, and the number of masks is also reduced, thereby contributing to yield improvement and cost reduction.

  Furthermore, the light-emitting device of the present invention can reduce the voltage amplitude of a signal for driving a pixel having a structure to which the bootstrap method is applied. This contributes to lower power consumption of the light emitting device.

  The figure which shows embodiment of this invention.   The figure which shows one Example of this invention.   The figure which shows the example of 1 structure of the light-emitting device used conventionally.   FIG. 6 illustrates a structural example of a light-emitting device of the present invention.   4A and 4B illustrate operation of a TFT and a light-emitting element in a pixel portion.   4A and 4B illustrate a manufacturing process of a light-emitting device of the present invention.   4A and 4B illustrate a manufacturing process of a light-emitting device of the present invention.   The figure which shows the pixel part cross section of the light-emitting device in the case of upward emission and downward emission.   FIG. 11 is a timing chart relating to driving of a light-emitting device.   8A and 8B show a cross section of a dual gate TFT and an example of a manufacturing process.   FIG. 6 shows potentials at respective nodes when a pixel of the light emitting device of the present invention is driven.   FIG. 3 is a configuration diagram of a gate signal line driving circuit constituting the light emitting device of the present invention.   FIG. 3 is a configuration diagram of a source signal line driving circuit constituting a light emitting device of the present invention.   FIG. 3 is a circuit configuration diagram of a shift register.   FIG. 9 is a timing chart relating to driving of a shift register.   The circuit block diagram of a buffer.   The circuit block diagram of a level shifter.   The circuit block diagram of a latch circuit.   FIG. 11 illustrates an example of an electronic device to which the present invention is applicable.

Claims (1)

A first transistor, a second transistor, a first switch, a second switch, and a capacitor;
One of a source and a drain of the first transistor is electrically connected to the first wiring;
The other of the source and the drain of the first transistor is electrically connected to an electrode that can function as a part of the light-emitting element;
One of a source and a drain of the second transistor is electrically connected to a second wiring;
A gate of the second transistor is electrically connected to a gate of the first transistor;
A first terminal of the first switch is electrically connected to the second wiring;
A second terminal of the first switch is electrically connected to a gate of the first transistor;
A first terminal of the second switch is electrically connected to a third wiring;
A second terminal of the second switch is electrically connected to a gate of the first transistor;
One electrode of the capacitor is electrically connected to the gate of the second transistor;
The other electrode of the capacitor is electrically connected to the other of the source and the drain of the second transistor;
The first transistor and the second transistor are N-channel type ,
A first period in which the first switch is conductive, the second switch is non-conductive, and the gate potential of the first transistor is at a low level;
Following said first period, said a first switch non-conducting state, the Ri second switch conductive state that is the potential of the second wiring is High level, the third A second period in which the potential of the wiring is at a high level;
After the second period, the first switch is non-conductive, the second switch is non-conductive, the gate potential of the first transistor is at a high level, and the first switch A third period in which the potential of the wiring 2 is at a high level;
After the third period, the a first switch non-conducting state, the second switch is nonconductive, the Ri second transistor is conductive state that is the second wiring The potential of the second electrode of the capacitor is at a high level, and the potential of the gate of the first transistor is higher than the potential of the other electrode of the capacitor. And a period of time .

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