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Topic Editors

School of Engineering and the Built Environment, Edinburgh Napier University, Edinburgh EH10 5DT, UK
School of Computing, Engineering and the Built Environment, Edinburgh Napier University, Edinburgh EH10 5DT, UK
Faculty of Engineering and Informatics, University of Bradford, Bradford BD7 1DP, UK

Antennas

Abstract submission deadline
closed (28 February 2023)
Manuscript submission deadline
closed (31 May 2023)
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Topic Information

Dear Colleagues,

Due to rapid growth in the area of modern wireless communication systems, the demand for different types of novel, multi-functional, and high-performance antennas is increasing exponentially. As a crucial part of the future communication system, breakthrough in the development of antennas will obviously improve the performance of the whole communication system. Over recent years, considerable research efforts have been directed toward the development of antennas. The adaptation of the antenna design and technologies for various wireless services requires careful consideration to meet demanding specifications. Having wide and multiple frequency coverage, compact size, well-defined radiation coverage, multi-mode operation, low-cost and ease of fabrication, energy-efficiency, ease of integration and assembly, and conformity are some examples of the key parameters that ensure the success of antenna systems for current and future wireless communication systems. In addition, MIMO and phased array arrangements of antenna systems with multiple adaptive and smart antennas can significantly enhance the system capacity toward meeting the requirements of future wireless networks. Therefore, advanced antenna design techniques that use novel approaches and address various aspects are required.

The objective of this Topic is to cover all aspects of antennas used in existing or future wireless communication systems. The aim is to highlight recent advances, current trends, and possible future developments of antennas. To further promote the development of this area, we invite researchers to submit their original research or review papers that are concerned with novel design techniques, analysis, signal processing, optimization, prototyping, and experimentation in this area.

Submissions can focus on conceptual and applied research in subjects including but not limited to the following:

  • Antenna Design
  • RFID Tag Antennas
  • 4G/5G/6G Antennas
  • Antenna Calibration
  • Integrated Antennas
  • Smartphone Antennas
  • Antenna Optimization
  • Antenna Miniaturization
  • Filtering Antenna Design
  • Reconfigurable Antennas
  • Beam Forming Techniques
  • Mutual Coupling Reduction
  • Energy Harvesting Antennas
  • Antenna Feeding Techniques
  • Distributed Antenna Systems
  • Adaptive and Smart Antennas
  • Electromagnetic Bandgap (EBG)
  • UWB and Multi-Band Antennas
  • Antenna Array Signal Processing
  • Diversity Techniques in Antennas
  • Reconfigurable Intelligent Surfaces
  • Decoupling Techniques of Antennas
  • In Situ Characterization of Antennas
  • MIMO and Massive MIMO Antennas
  • Antennas for Biomedical and WBAN
  • MM-Wave, THz, and Nano Antennas
  • Substrate-Integrated Waveguides (SIW)
  • Transmission and Detection Techniques
  • Textile, Wearable, and Flexible Antennas
  • Prototyping and Manufacturing Methods
  • Metasurfaces and Reflect Array Antennas
  • Phased Array and Beamforming Antennas
  • Automotive, Radar, and Satellite Antennas
  • Antenna Design for Massive MIMO and IoT
  • Dual-Polarized/Circular-Polarized Antennas
  • Angle of Arrival Estimation Using Antennas
  • Artificial Intelligence (AI) Empowered Antennas
  • Measurements and Experimentation of Antennas
  • Channel Capacity Estimation of Antenna Systems
  • Antenna on Chip (AoC) and Antenna in Package (AiP)
  • Advanced Algorithms of Array Analysis and Synthesis
  • Antennas for Complex Radio Wave Propagations Scenarios
  • Advanced Techniques for Numerical Modelling of Antennas.

Submissions should reflect the high quality of this international journal and should not have been submitted or published elsewhere. Extended versions of conference papers that show significant improvement (minimal of over 50%) can be considered for publication in this Topic.

Dr. Naser Ojaroudi Parchin
Dr. Chan Hwang See
Prof. Dr. Raed A. Abd-Alhameed
Topic Editors

Participating Journals

Journal Name Impact Factor CiteScore Launched Year First Decision (median) APC
Electronics
electronics
2.6 5.3 2012 16.8 Days CHF 2400
Future Internet
futureinternet
2.8 7.1 2009 13.1 Days CHF 1600
Sensors
sensors
3.4 7.3 2001 16.8 Days CHF 2600
Telecom
telecom
2.1 4.8 2020 22.7 Days CHF 1200

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Published Papers (47 papers)

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11 pages, 242 KiB  
Editorial
Editorial on Antennas
by Naser Ojaroudi Parchin, Chan Hwang See and Raed A. Abd-Alhameed
Sensors 2023, 23(24), 9643; https://doi.org/10.3390/s23249643 - 6 Dec 2023
Viewed by 1132
Abstract
In the ever-evolving landscape of modern wireless communication systems, the escalating demand for seamless connectivity has propelled the imperative for avant garde, versatile, and high-performance antennas to unprecedented heights [...] Full article
(This article belongs to the Topic Antennas)
18 pages, 7205 KiB  
Article
Optimal Pattern Synthesis of Linear Array Antennas Using the Nonlinear Chaotic Grey Wolf Algorithm
by Kunxia Zhao, Yan Liu and Kui Hu
Electronics 2023, 12(19), 4087; https://doi.org/10.3390/electronics12194087 - 29 Sep 2023
Cited by 5 | Viewed by 1228
Abstract
The grey wolf optimization (GWO) algorithm is a new nature-inspired meta-heuristic algorithm inspired by the social hierarchy and hunting behavior of grey wolves. In this paper, the GWO algorithm is improved to overcome previous shortcomings of being easily trapped in local optima and [...] Read more.
The grey wolf optimization (GWO) algorithm is a new nature-inspired meta-heuristic algorithm inspired by the social hierarchy and hunting behavior of grey wolves. In this paper, the GWO algorithm is improved to overcome previous shortcomings of being easily trapped in local optima and having a low convergence rate. The proposed enhancement of the GWO algorithm utilizes logistic-tent double mapping to generate initialized populations, which enhances its global search capability and convergence rate. This improvement is called the nonlinear chaotic grey wolf optimization (NCGWO) algorithm. The performance of the NCGWO algorithm was evaluated with four representative benchmark functions. Then, the NCGWO algorithm was applied to perform an optimal pattern synthesis of linear array antennas (LAAs) using two distinct approaches: optimizing the amplitudes of the antenna currents while preserving uniform spacing and optimizing the positions of the antennas while assuming uniform excitation. To validate the effectiveness of the proposed approach, the results obtained by the NCGWO algorithm were compared with those obtained by other intelligent algorithms. Additionally, the NCGWO algorithm was applied to a more complex planar antenna array to further validate its performance. Our results demonstrate that the NCGWO algorithm exhibits superior performance regarding electromagnetic optimization problems compared to widely recognized algorithms. Full article
(This article belongs to the Topic Antennas)
Show Figures

Figure 1

Figure 1
<p>Hierarchy of grey wolves.</p>
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<p>Wolves in 2D space.</p>
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<p>Population distribution of chaotic mapping.</p>
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<p>Convergence coefficients.</p>
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<p>Flow chart of the nonlinear chaotic grey wolf optimization (NCGWO) algorithm.</p>
Full article ">Figure 6
<p>Illustration of the two-dimensional versions of the test functions. (<b>a</b>,<b>b</b>) are unimodal functions; (<b>c</b>,<b>d</b>) are multimodal functions.</p>
Full article ">Figure 7
<p>(<b>a</b>) Uniform LAA geometry; (<b>b</b>) sparse LAA geometry.</p>
Full article ">Figure 8
<p>Planar array antennas geometry.</p>
Full article ">Figure 9
<p>(<b>a</b>) Radiation patterns of a 10-element linear array antenna SLL; (<b>b</b>) iteration curve.</p>
Full article ">Figure 10
<p>3D radiation patterns of a 10-element linear array antenna SLL near the main lobe minimization: (<b>a</b>) before and (<b>b</b>) after.</p>
Full article ">Figure 11
<p>(<b>a</b>) Radiation patterns of the SLL of a 20-element linear array antenna with a lower notch; (<b>b</b>) iteration curve.</p>
Full article ">Figure 12
<p>3D radiation patterns of the SLL of a 20-element linear array antenna with a lower notch: (<b>a</b>) before and (<b>b</b>) after.</p>
Full article ">Figure 13
<p>(<b>a</b>) Radiation patterns of the SLL of a 10-element sparse linear array antenna; (<b>b</b>) iteration curve.</p>
Full article ">Figure 14
<p>3D radiation patterns of the SLL of a 10-element sparse linear array antenna: (<b>a</b>) before and (<b>b</b>) after.</p>
Full article ">Figure 15
<p>(<b>a</b>) Radiation patterns of the SLL minimization and null of 32-element sparse linear array antennas; (<b>b</b>) iteration curve.</p>
Full article ">Figure 16
<p>3D radiation patterns of the SLL minimization and null of 32-element sparse linear array antennas: (<b>a</b>) before and (<b>b</b>) after.</p>
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<p>3D radiation patterns of the SLL minimization of planar array antennas: (<b>a</b>) side view and (<b>b</b>) current amplitude of each array element.</p>
Full article ">
11 pages, 7342 KiB  
Article
W-Band Broadband Circularly Polarized Reflectarray Antenna
by Zhicheng Wang, Rui Zhang, Wenke Song, Bingchuan Xie, Xiaobo Lin, Haixuan Li and Lu Tian
Electronics 2023, 12(18), 3849; https://doi.org/10.3390/electronics12183849 - 12 Sep 2023
Cited by 1 | Viewed by 1252
Abstract
We propose a W-band circularly polarized reflectarray antenna in this article, which contains a single-layer reflectarray and a linearly polarized horn feed. To realize the proposed antenna, we designed a novel W-band multi-resonant element containing a Rogers RT5880 substrate with copper patches printed [...] Read more.
We propose a W-band circularly polarized reflectarray antenna in this article, which contains a single-layer reflectarray and a linearly polarized horn feed. To realize the proposed antenna, we designed a novel W-band multi-resonant element containing a Rogers RT5880 substrate with copper patches printed on its both surfaces. Then, a circular reflectarray with 12.25 λ0 (39.1 mm) aperture diameter was designed based on the proposed multi-resonant element, whose center frequency is 94 GHz. We further fabricated the proposed circular reflectarray and tested its performance. In the measured results, we can see that the obtained 1 dB gain bandwidth is 19.1% (91~109 GHz) and the obtained 2 dB gain bandwidth can reach as wide as 27.6% (89~115 GHz). Moreover, the 3 dB axial ratio bandwidth is 13.8% (89~102 GHz). The measured gain of our proposed reflectarray antenna at 94 GHz is 29.1 dBi and the corresponding aperture efficiency can reach as high as 52.0%. Those results show that our proposed antenna may be prospective in wireless communication applications due to its strengths in broadband and high aperture efficiency. Full article
(This article belongs to the Topic Antennas)
Show Figures

Figure 1

Figure 1
<p>The structure of the element formed by four dipoles. (<b>a</b>) Perspective view, (<b>b</b>) top view.</p>
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<p>Simulation results of the mentioned element with E<sub>i</sub><sup>x</sup> at 94 GHz.</p>
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<p>(<b>a</b>) Perspective view and (<b>b</b>) top view of the single-layer multi-resonant element.</p>
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<p>Simulated surface current distributions of the multi-resonant element with E<sub>i</sub><sup>x</sup> at (<b>a</b>) 84 GHz, (<b>b</b>) 94 GHz, and (<b>c</b>) 104 GHz.</p>
Full article ">Figure 5
<p>The reflective phase shift characteristics of our proposed element with E<sub>i</sub><sup>x</sup> at 94 GHz.</p>
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<p>The characteristics of our proposed element when <span class="html-italic">l<sub>y</sub></span> is 1 mm. (<b>a</b>) Different incidence angles. (<b>b</b>) Different frequencies. (The blue arrow shows the coordinate axis which is applicable to the curve in the blue circle).</p>
Full article ">Figure 7
<p>The efficiencies versus the value of <span class="html-italic">F</span>/<span class="html-italic">D</span>.</p>
Full article ">Figure 8
<p>The required compensation phase of (<b>a</b>) the X-direction incident wave and (<b>b</b>) the Y-direction incident wave.</p>
Full article ">Figure 9
<p>(<b>a</b>) Measured setup of the proposed reflectarray antenna. (<b>b</b>) Top and (<b>c</b>) bottom layer of the fabricated antenna sample.</p>
Full article ">Figure 10
<p>Simulated and measured radiation patterns of the proposed RA. (<b>a</b>) E plane and (<b>b</b>) H plane at 89 GHz; (<b>c</b>) E plane and (<b>d</b>) H plane at 94 GHz; (<b>e</b>) E plane and (<b>f</b>) H plane at 102 GHz.</p>
Full article ">Figure 11
<p>Measured and simulated antenna gain or axial ratio within frequency band of 80–120 GHz. (The blue arrow shows the coordinate axis which is applicable to the curve in the blue circle).</p>
Full article ">
10 pages, 9444 KiB  
Communication
Design of a Crossed Dielectric Resonator-Loaded, Dual-Band Dual-Polarized Differential Patch Antenna with Improved Port Isolation and Gain
by Dongdong Wang, Yudong Liu and Jia Liang
Electronics 2023, 12(17), 3570; https://doi.org/10.3390/electronics12173570 - 24 Aug 2023
Cited by 1 | Viewed by 779
Abstract
To meet the urgent requirement for more channel capacity in modern wireless communication systems, antennas with more operation bands are demanded. However, large amounts of antennas suffer from low radiation gains and low port isolation levels. In view of this, a differentially fed, [...] Read more.
To meet the urgent requirement for more channel capacity in modern wireless communication systems, antennas with more operation bands are demanded. However, large amounts of antennas suffer from low radiation gains and low port isolation levels. In view of this, a differentially fed, dual-wideband, dual-polarized patch antenna is proposed in this paper. Compared with conventional crossed-feeding structures, the proposed crossed dielectric resonator (CDR) can provide extra resonances with improved isolation levels and radiation gain. Further, four shorting pins are introduced to the radiating patch to help improve the impedance-matching performance. In addition, the proposed antenna also has a very compact size of 0.46λ × 0.46λ × 0.12λ. Finally, a prototype of the proposed antenna is fabricated to validate the design concept. The measured results show that the proposed antenna generates dual wide bands of 1.86–2.52 GHz and 3.26–3.72 GHz for |S11| < −10 dB. High radiation gains of 8.9 ± 0.9 dBi and 10.8 ± 1.2 dBi are also obtained, as well as high port isolation levels of better than 38.4 dB and 36.2 dB at the two bands. The excellent performance of the proposed antenna makes it a promising candidate for 4G/5G wireless communication systems. Full article
(This article belongs to the Topic Antennas)
Show Figures

Figure 1

Figure 1
<p>Configuration of the proposed antenna: (<b>a</b>) top view, (<b>b</b>) back view, and (<b>c</b>) front view. The optimized parameters are as follows: L1 = 50.5, L2 = 30.2, W1 = 4.3, g = 2.1, Wg = 140.0, H1 = 15.0, H3 = 0.5 (Unit: mm).</p>
Full article ">Figure 2
<p>Evolution process of the proposed antenna.</p>
Full article ">Figure 3
<p>Simulated (<b>a</b>) S-parameters and (<b>b</b>) realized gains of Ant.1 and Ant.2.</p>
Full article ">Figure 4
<p>Current distributions on Ant.1 at 1.92 GHz and 3.96 GHz and on Ant.2 at 2.48 GHz and 3.64 GHz.</p>
Full article ">Figure 5
<p>Input impedance for Ant.1 and Ant.2.</p>
Full article ">Figure 6
<p>S-parameter for Ant.2 varied with length of the CDR.</p>
Full article ">Figure 7
<p>S-parameter for Ant.2 varied with width of the CDR.</p>
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<p>S-parameter and realized gains for Ant.2 and the proposed antenna.</p>
Full article ">Figure 9
<p>Input impedance for Ant.2 and the proposed antenna.</p>
Full article ">Figure 10
<p>S-parameter for the proposed antenna varied with distance from the shorting pin to patch edge.</p>
Full article ">Figure 11
<p>(<b>a</b>) testing environment, (<b>b</b>) 3 dB couplers, and (<b>c</b>) prototype of the proposed antenna.</p>
Full article ">Figure 12
<p>Simulated and measured S-parameters of the proposed antenna.</p>
Full article ">Figure 13
<p>Simulated and measured realized gains and radiation efficiencies of the proposed antenna.</p>
Full article ">Figure 14
<p>Simulated and measured radiation patterns of the antenna at different frequencies.</p>
Full article ">
34 pages, 23062 KiB  
Article
Linear Antenna Array Sectorized Beam Scanning Approaches Using Element Position Perturbation in the Azimuth Plane
by Safaa I. Abd Elrahman, Ahmed M. Elkhawaga, Amr H. Hussein and Abd Elhameed A. Shaalan
Sensors 2023, 23(14), 6557; https://doi.org/10.3390/s23146557 - 20 Jul 2023
Cited by 2 | Viewed by 1406
Abstract
In this paper, two sector beam scanning approaches (BSAs) based on element position perturbations (EPPs) in the azimuth plane are introduced. In EPP-BSA, the elements’ excitations are kept constant and the elements’ positions in the direction normal to the array line are changed [...] Read more.
In this paper, two sector beam scanning approaches (BSAs) based on element position perturbations (EPPs) in the azimuth plane are introduced. In EPP-BSA, the elements’ excitations are kept constant and the elements’ positions in the direction normal to the array line are changed according to a predetermined EPP pattern. The magnitude and repetition rate of the selected EPP pattern determines the steering angle of the main beam. However, EPP-BSA results in a wide scanning range with a significant increase in the side lobe level (SLL). To mitigate this drawback, a reduction in the SLL of the array pattern is firstly performed using the single convolution/genetic algorithm (SC/GA) technique and then perturbing the elements’ positions in the azimuth plane. This combination between SLL reduction and EPP-BSA (SLL/EPP-BSA) results in a smaller scanning range with a relatively constant half power beamwidth (HPBW) and a much lower SLL. In addition, keeping the synthesized excitation coefficients constant without adding progressive phase shifters facilitates the manufacturing process and reduces the cost of the feeding network. Furthermore, a planar antenna array thinning approach is proposed to realize the EPP-BSA. The results are realized using the computer simulation technology (CST) microwave studio software package, which provides users with an optimized modeling environment and results in realizable and realistic designs. Full article
(This article belongs to the Topic Antennas)
Show Figures

Figure 1

Figure 1
<p>Linear antenna array (<b>a</b>) without EPP and (<b>b</b>) with EPP.</p>
Full article ">Figure 2
<p>Geometry of a linear array of elements aligned along the <span class="html-italic">Z</span>-axis and has position perturbation along the <span class="html-italic">Y</span>-axis.</p>
Full article ">Figure 3
<p>The triangular relation between the element position in the <span class="html-italic">Y</span>-direction and the distance from the reference antenna element.</p>
Full article ">Figure 4
<p>Geometry of a linear array of elements aligned along the <span class="html-italic">Z</span>-axis and has position perturbation along the <span class="html-italic">Y</span>-axis indicating the far field radiation from the first antenna element.</p>
Full article ">Figure 5
<p>Geometry of a linear array of elements aligned along the <span class="html-italic">Z</span>-axis and has position perturbation along the <span class="html-italic">Y</span>-axis indicating the far field radiation from the second antenna element.</p>
Full article ">Figure 6
<p>The EPPs of the array elements at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>1</mn></mrow></semantics></math> for (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub></mrow></semantics></math> is changing from positive to negative and (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub></mrow></semantics></math> is changing from negative to positive.</p>
Full article ">Figure 7
<p>Polar plots of the synthesized radiation patterns using the EPP-BSA approach for 8-element LAA at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>1</mn></mrow></semantics></math> and (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0</mn></mrow></semantics></math>, (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.19</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>c</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>d</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>e</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>f</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mi>λ</mi></mrow></semantics></math>, (<b>g</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>h</b>)<math display="inline"><semantics><mrow><msub><mrow><mo> </mo><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>i</b>)<math display="inline"><semantics><mrow><mo> </mo><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, and (<b>j</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> and changing from positive to negative.</p>
Full article ">Figure 7 Cont.
<p>Polar plots of the synthesized radiation patterns using the EPP-BSA approach for 8-element LAA at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>1</mn></mrow></semantics></math> and (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0</mn></mrow></semantics></math>, (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.19</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>c</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>d</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>e</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>f</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mi>λ</mi></mrow></semantics></math>, (<b>g</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>h</b>)<math display="inline"><semantics><mrow><msub><mrow><mo> </mo><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>i</b>)<math display="inline"><semantics><mrow><mo> </mo><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, and (<b>j</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> and changing from positive to negative.</p>
Full article ">Figure 8
<p>Polar plot of the synthesized radiation pattern using the EPP-BSA approach for 8-element ULAA at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>1</mn></mrow></semantics></math> for <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> and changing from negative to positive.</p>
Full article ">Figure 9
<p>The EPPs of the array elements at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.875</mn></mrow></semantics></math> for (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub></mrow></semantics></math> is changing from positive to negative and (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub></mrow></semantics></math> is changing from negative to positive.</p>
Full article ">Figure 10
<p>Polar plots of the synthesized radiation patterns using the EPP-BSA approach for an 8-element ULAA at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.875</mn></mrow></semantics></math> and (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0</mn></mrow></semantics></math>, (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>c</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>d</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>e</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mi>λ</mi></mrow></semantics></math>, (<b>f</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>g</b>)<math display="inline"><semantics><mrow><msub><mrow><mo> </mo><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>h</b>)<math display="inline"><semantics><mrow><mo> </mo><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, and (<b>i</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> and changing from positive to negative.</p>
Full article ">Figure 11
<p>Relation between changes in the main beam scanning angle around the broadside direction and side lobe level changes.</p>
Full article ">Figure 12
<p>Polar plot of the synthesized radiation pattern using the EPP-BSA approach for 8-element ULAA at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.875</mn></mrow></semantics></math> for <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> and changing from negative to positive.</p>
Full article ">Figure 13
<p>The EPPs of the array elements at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.7778</mn></mrow></semantics></math> for (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub></mrow></semantics></math> is changing from positive to negative and (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub></mrow></semantics></math> is changing from negative to positive.</p>
Full article ">Figure 14
<p>Polar plots of the synthesized radiation patterns using the EPP-BSA approach for an 8-element ULAA at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.7778</mn></mrow></semantics></math> at (<b>a</b>)<math display="inline"><semantics><mrow><mo> </mo><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0</mn></mrow></semantics></math>, (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>c</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.27</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>d</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.3</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> (<b>e</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>f</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>g</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mi>λ</mi></mrow></semantics></math>, (<b>h</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>i</b>)<math display="inline"><semantics><mrow><msub><mrow><mo> </mo><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>j</b>)<math display="inline"><semantics><mrow><mo> </mo><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, and (<b>k</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> and changing from positive to negative.</p>
Full article ">Figure 14 Cont.
<p>Polar plots of the synthesized radiation patterns using the EPP-BSA approach for an 8-element ULAA at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.7778</mn></mrow></semantics></math> at (<b>a</b>)<math display="inline"><semantics><mrow><mo> </mo><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0</mn></mrow></semantics></math>, (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>c</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.27</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>d</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.3</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> (<b>e</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>f</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>g</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mi>λ</mi></mrow></semantics></math>, (<b>h</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>i</b>)<math display="inline"><semantics><mrow><msub><mrow><mo> </mo><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>j</b>)<math display="inline"><semantics><mrow><mo> </mo><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, and (<b>k</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> and changing from positive to negative.</p>
Full article ">Figure 15
<p>Polar plot of the synthesized radiation pattern using the EPP-BSA approach for 8-element ULAA at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.7778</mn></mrow></semantics></math> for <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> and changing from negative to positive.</p>
Full article ">Figure 16
<p>The EPPs of the array elements at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.7</mn></mrow></semantics></math> for (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub></mrow></semantics></math> is changing from positive to negative and (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub></mrow></semantics></math> is changing from negative to positive.</p>
Full article ">Figure 17
<p>Polar plots of the synthesized radiation patterns using the EPP techniques for 8-element ULAA at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.7</mn></mrow></semantics></math> at: (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0</mn></mrow></semantics></math>, (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>c</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.35</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>d</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.39</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>e</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>f</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>g</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mi>λ</mi></mrow></semantics></math>, (<b>h</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>i</b>)<math display="inline"><semantics><mrow><msub><mrow><mo> </mo><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>j</b>)<math display="inline"><semantics><mrow><mo> </mo><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, and (<b>k</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> and changing from positive to negative.</p>
Full article ">Figure 18
<p>Polar plot of the synthesized radiation pattern using the EPP-BSA approach for 8-element ULAA at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.7</mn></mrow></semantics></math> for <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> and changing from negative to positive.</p>
Full article ">Figure 19
<p>The EPPs of the array elements at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.5</mn></mrow></semantics></math> for (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub></mrow></semantics></math> is changing from positive to negative and (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub></mrow></semantics></math> is changing from negative to positive.</p>
Full article ">Figure 20
<p>Polar plots of the synthesized radiation patterns using the EPP-BSA approach for 8-element uniform LAA at compression ratio equal to 0.5 at: (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0</mn></mrow></semantics></math>, (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>c</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>d</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.6</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>e</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.65</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>f</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>g</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mi>λ</mi></mrow></semantics></math>, (<b>h</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>i</b>)<math display="inline"><semantics><mrow><msub><mrow><mo> </mo><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>j</b>)<math display="inline"><semantics><mrow><mo> </mo><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, and (<b>k</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> and changing from positive to negative.</p>
Full article ">Figure 21
<p>Polar plot of the synthesized radiation pattern using the EPP-BSA approach for 8-element uniform LAA due to compression ratio equal to <math display="inline"><semantics><mrow><mn>0.5</mn></mrow></semantics></math> for <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> and changing from negative to positive.</p>
Full article ">Figure 22
<p>(<b>a</b>) Dimensions of dipole antenna, (<b>b</b>) H-plane pattern, and (<b>c</b>) E-plane pattern.</p>
Full article ">Figure 23
<p>The scattering parameter (reflection coefficient) <math display="inline"><semantics><mrow><mfenced open="|" close="|" separators="|"><mrow><msub><mrow><mi>S</mi></mrow><mrow><mn>11</mn></mrow></msub></mrow></mfenced></mrow></semantics></math> against the frequency for the designed dipole antenna.</p>
Full article ">Figure 24
<p>The 3D radiation pattern plots of the synthesized antenna arrays using CST software package applying the proposed EPP-BSA approach for an 8-element ULAA for <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.875</mn></mrow></semantics></math> at: (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0</mn></mrow></semantics></math>, (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>c</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>d</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>e</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>f</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.251</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>g</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>h</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>i</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, and changing from positive to negative.</p>
Full article ">Figure 24 Cont.
<p>The 3D radiation pattern plots of the synthesized antenna arrays using CST software package applying the proposed EPP-BSA approach for an 8-element ULAA for <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.875</mn></mrow></semantics></math> at: (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0</mn></mrow></semantics></math>, (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>c</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>d</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>e</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>f</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.251</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>g</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>h</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>i</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, and changing from positive to negative.</p>
Full article ">Figure 25
<p>Polar plots of the synthesized antenna arrays radiation patterns using CST software package applying the proposed EPP-BSA approach for an 8-element ULAA for <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.875</mn></mrow></semantics></math> at: (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0</mn></mrow></semantics></math>, (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.25</mn><mi mathvariant="sans-serif">λ</mi></mrow></semantics></math>, (<b>c</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>d</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>e</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>f</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.251</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>g</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>h</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.75</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>i</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, and changing from positive to negative.</p>
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<p>The triangular relation between the element position <math display="inline"><semantics><mrow><msub><mrow><mi mathvariant="bold-italic">d</mi></mrow><mrow><msub><mrow><mi mathvariant="bold-italic">y</mi></mrow><mrow><mi mathvariant="bold-italic">n</mi></mrow></msub></mrow></msub></mrow></semantics></math> in the <span class="html-italic">Y</span>-direction and the distance from the reference antenna element.</p>
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<p>The synthesized array factor <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi><mi>F</mi></mrow><mrow><mi>S</mi><mi>O</mi></mrow></msub><mfenced separators="|"><mrow><mi>θ</mi></mrow></mfenced></mrow></semantics></math> using odd excitations compared to the original array factor <math display="inline"><semantics><mrow><mi>A</mi><mi>F</mi><mfenced separators="|"><mrow><mi>θ</mi></mrow></mfenced></mrow></semantics></math> of the 8-element ULAA.</p>
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<p>The synthesized array factor <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi><mi>F</mi></mrow><mrow><mi>S</mi><mi>E</mi></mrow></msub><mfenced separators="|"><mrow><mi>θ</mi></mrow></mfenced></mrow></semantics></math> using even excitations compared to the original array factor <math display="inline"><semantics><mrow><mi>A</mi><mi>F</mi><mfenced separators="|"><mrow><mi>θ</mi></mrow></mfenced></mrow></semantics></math> of the 8-element ULAA.</p>
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<p>The EPPs of the synthesized array elements at compression ratio <math display="inline"><semantics><mrow><mi mathvariant="bold-italic">R</mi><mo>=</mo><mn>1</mn></mrow></semantics></math> for <math display="inline"><semantics><mrow><msub><mrow><mi mathvariant="bold-italic">A</mi></mrow><mrow><mi mathvariant="bold-italic">c</mi></mrow></msub></mrow></semantics></math> is changing from positive to negative.</p>
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<p>Polar plot of the synthesized radiation pattern using the SLL/EPP-BSA approach at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>1</mn></mrow></semantics></math> and <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mi>λ</mi></mrow></semantics></math>.</p>
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<p>Polar plot of the synthesized radiation pattern using the SLL/EPP-BSA approach at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.875</mn></mrow></semantics></math> and <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>.</p>
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<p>Polar plot of the synthesized radiation pattern using the SLL/EPP-BSA approach at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.7778</mn></mrow></semantics></math> and <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.25</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>.</p>
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<p>Polar plot of the synthesized radiation pattern using the SLL/EPP-BSA approach at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.7</mn></mrow></semantics></math> and <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>1.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>.</p>
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<p>Polar plot of the synthesized radiation pattern using the SLL/EPP-BSA approach at compression ratio <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.5</mn></mrow></semantics></math> and <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>.</p>
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<p>Element distribution of the proposed <math display="inline"><semantics><mrow><mo>(</mo><mn>41</mn><mo>×</mo><mn>8</mn></mrow></semantics></math>) PAA configuration and the element distributions of the <math display="inline"><semantics><mrow><mi>M</mi><mo>=</mo><mn>8</mn></mrow></semantics></math> elements of the actual array using the proposed EPP-BSA approach at uniform element spacing <math display="inline"><semantics><mrow><msub><mrow><mi>d</mi></mrow><mrow><mi>z</mi></mrow></msub><mo>=</mo><mn>0.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math> on the <span class="html-italic">Z</span>-axis and <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub></mrow></semantics></math> changing from positive to negative.</p>
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<p>The implementation of the proposed <math display="inline"><semantics><mrow><mo>(</mo><mn>41</mn><mo>×</mo><mn>8</mn></mrow></semantics></math>) PAA configuration using CST microwave studio utilizing a dipole antenna with resonance frequency <math display="inline"><semantics><mrow><msub><mrow><mi>f</mi></mrow><mrow><mi>o</mi></mrow></msub><mo>=</mo><mn>1</mn><mo> </mo><mi mathvariant="normal">G</mi><mi mathvariant="normal">H</mi><mi mathvariant="normal">z</mi></mrow></semantics></math>.</p>
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<p>CST-simulated 3D radiation patterns using the PAA thinning for implementation of the proposed EPP-BSA approach for 8-element ULAA at <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.875</mn></mrow></semantics></math>: (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0</mn></mrow></semantics></math>, (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.5</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, (<b>c</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mi>λ</mi></mrow></semantics></math>, (<b>d</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, and changing from positive to negative.</p>
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<p>CST simulated polar plots of the radiation patterns using the PAA thinning for implementation of the proposed EPP-BSA approach for 8-element ULAA at <math display="inline"><semantics><mrow><mi>R</mi><mo>=</mo><mn>0.875</mn></mrow></semantics></math>: (<b>a</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0</mn></mrow></semantics></math>, (<b>b</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>0.5</mn><mo> </mo><mi mathvariant="sans-serif">λ</mi></mrow></semantics></math>, (<b>c</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mi>λ</mi></mrow></semantics></math>, (<b>d</b>) <math display="inline"><semantics><mrow><msub><mrow><mi>A</mi></mrow><mrow><mi>c</mi></mrow></msub><mo>=</mo><mn>2</mn><mo> </mo><mi>λ</mi></mrow></semantics></math>, and changing from positive to negative.</p>
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26 pages, 2757 KiB  
Article
Flexible Antenna with Circular/Linear Polarization for Wideband Biomedical Wireless Communication
by Mohammed E. Yassin, Khaled F. A. Hussein, Qammer H. Abbasi, Muhammad A. Imran and Shaimaa A. Mohassieb
Sensors 2023, 23(12), 5608; https://doi.org/10.3390/s23125608 - 15 Jun 2023
Cited by 10 | Viewed by 2349
Abstract
A wideband low-profile radiating G-shaped strip on a flexible substrate is proposed to operate as biomedical antenna for off-body communication. The antenna is designed to produce circular polarization over the frequency range 5–6 GHz to communicate with WiMAX/WLAN antennas. Furthermore, it is designed [...] Read more.
A wideband low-profile radiating G-shaped strip on a flexible substrate is proposed to operate as biomedical antenna for off-body communication. The antenna is designed to produce circular polarization over the frequency range 5–6 GHz to communicate with WiMAX/WLAN antennas. Furthermore, it is designed to produce linear polarization over the frequency range 6–19 GHz for communication with the on-body biosensor antennas. It is shown that an inverted G-shaped strip produces circular polarization (CP) of the opposite sense to that produced by G-shaped strip over the frequency range 5–6 GHz. The antenna design is explained and its performance is investigated through simulation, as well as experimental measurements. This antenna can be viewed as composed of a semicircular strip terminated with a horizontal extension at its lower end and terminated with a small circular patch through a corner-shaped strip extension at its upper end to form the shape of “G” or inverted “G”. The purpose of the corner-shaped extension and the circular patch termination is to match the antenna impedance to 50 Ω over the entire frequency band (5–19 GHz) and to improve the circular polarization over the frequency band (5–6 GHz). To be fabricated on only one face of the flexible dielectric substrate, the antenna is fed through a co-planar waveguide (CPW). The antenna and the CPW dimensions are optimized to obtain the most optimal performance regarding the impedance matching bandwidth, 3dB Axial Ratio (AR) bandwidth, radiation efficiency, and maximum gain. The results show that the achieved 3dB-AR bandwidth is 18% (5–6 GHz). Thus, the proposed antenna covers the 5 GHz frequency band of the WiMAX/WLAN applications within its 3dB-AR frequency band. Furthermore, the impedance matching bandwidth is 117% (5–19 GHz) which enables low-power communication with the on-body sensors over this wide range of the frequency. The maximum gain and radiation efficiency are 5.37 dBi and 98%, respectively. The overall antenna dimensions are 25 × 27 × 0.13 mm3 and the bandwidth-dimension ratio (BDR) is 1733. Full article
(This article belongs to the Topic Antennas)
Show Figures

Figure 1

Figure 1
<p>Wireless body area network in intensive care unit where the proposed G-shaped antenna is employed as central off-body antenna for transmitting the biotelemetry data to a nearby WiMAX/WLAN base station antenna.</p>
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<p>WiMAX/WLAN access point with antennas of arbitrary polarization.</p>
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<p>Geometry of the inverted G-shaped strip antenna showing the dimensional parameters.</p>
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<p>Evolution of the Inverted G-shape CP Antenna Design: (<b>a</b>) Evolution steps. (<b>b</b>) The reflection coefficient <math display="inline"><semantics> <mrow> <mo>∣</mo> <msub> <mi>S</mi> <mn>11</mn> </msub> <mo>∣</mo> </mrow> </semantics></math> graph of each design step. (<b>c</b>) The AR graph of each design step.</p>
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<p>Effect of changing the outer diameter, <span class="html-italic">D<math display="inline"><semantics> <msub> <mrow/> <mn>2</mn> </msub> </semantics></math></span>, of the semicircular strip on (<b>a</b>) The magnitude of the reflection coefficient, <math display="inline"><semantics> <mrow> <mo>∣</mo> <msub> <mi>S</mi> <mn>11</mn> </msub> <mo>∣</mo> </mrow> </semantics></math>, as a function of frequency (4–20 GHz). (<b>b</b>) The AR over the frequency range (4.8–6.8 GHz).</p>
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<p>Effect of changing the outer diameter, <span class="html-italic">D<math display="inline"><semantics> <msub> <mrow/> <mn>1</mn> </msub> </semantics></math></span>, of the semicircular strip on (<b>a</b>) the magnitude of the reflection coefficient, <math display="inline"><semantics> <mrow> <mo>∣</mo> <msub> <mi>S</mi> <mn>11</mn> </msub> <mo>∣</mo> </mrow> </semantics></math>, as a function of frequency (4–20 GHz), and (<b>b</b>) the AR over the frequency range (4.8–6.8 GHz).</p>
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<p>Effect of changing the vertical length, <span class="html-italic">L<math display="inline"><semantics> <msub> <mrow/> <mi>A</mi> </msub> </semantics></math></span>, of the corner-shaped extension of the curved strip on (<b>a</b>) The magnitude of the reflection coefficient, <math display="inline"><semantics> <mrow> <mo>∣</mo> <msub> <mi>S</mi> <mn>11</mn> </msub> <mo>∣</mo> </mrow> </semantics></math>, as a function of frequency (4–20 GHz). (<b>b</b>) The AR over the frequency range (4.8–6.8 GHz).</p>
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<p>Effect of changing the horizontal length, <span class="html-italic">L<math display="inline"><semantics> <msub> <mrow/> <mi>B</mi> </msub> </semantics></math></span>, of the corner-shaped extension of the curved strip on (<b>a</b>) The magnitude of the reflection coefficient, <math display="inline"><semantics> <mrow> <mo>∣</mo> <msub> <mi>S</mi> <mn>11</mn> </msub> <mo>∣</mo> </mrow> </semantics></math>, as a function of frequency (4–20 GHz). (<b>b</b>) The AR over the frequency range (4.8–6.8 GHz).</p>
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<p>The impact of altering the radius, <span class="html-italic">R</span>, of the circular patch termination on (<b>a</b>) The reflection coefficient magnitude, <math display="inline"><semantics> <mrow> <mo>∣</mo> <msub> <mi>S</mi> <mn>11</mn> </msub> <mo>∣</mo> </mrow> </semantics></math>, over the frequency range (4–20 GHz). (<b>b</b>) The AR over the frequency range (4.8–6.8 GHz).</p>
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<p>The impact of altering the length, <span class="html-italic">L<math display="inline"><semantics> <msub> <mrow/> <mi>f</mi> </msub> </semantics></math></span>, of the central strip extension on (<b>a</b>) The reflection coefficient magnitude, <math display="inline"><semantics> <mrow> <mo>∣</mo> <msub> <mi>S</mi> <mn>11</mn> </msub> <mo>∣</mo> </mrow> </semantics></math>, over the frequency range (4–20 GHz). (<b>b</b>) The AR as a function of frequency (4.8–6.8 GHz).</p>
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<p>The proposed antenna subjected to bend stresses in (<b>a</b>) The longitudinal plane, (<b>b</b>) The transverse plane.</p>
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<p>Frequency dependence of the reflection coefficient, <math display="inline"><semantics> <mrow> <mo>∣</mo> <msub> <mi>S</mi> <mn>11</mn> </msub> <mo>∣</mo> </mrow> </semantics></math>, for different values of the longitudinal bend angle, <math display="inline"><semantics> <msub> <mi>β</mi> <mi>L</mi> </msub> </semantics></math>.</p>
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<p>Frequency dependence of the AR for different values of the longitudinal bend angle, <math display="inline"><semantics> <msub> <mi>β</mi> <mi>L</mi> </msub> </semantics></math>.</p>
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<p>Frequency dependence of the reflection coefficient, <math display="inline"><semantics> <mrow> <mo>∣</mo> <msub> <mi>S</mi> <mn>11</mn> </msub> <mo>∣</mo> </mrow> </semantics></math>, for different values of the transverse bend bend angle, <math display="inline"><semantics> <msub> <mi>β</mi> <mi>T</mi> </msub> </semantics></math>.</p>
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<p>Frequency dependence of the AR for different values of the transverse bend angle, <math display="inline"><semantics> <msub> <mi>β</mi> <mi>T</mi> </msub> </semantics></math>.</p>
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<p>Dependence of the percentage (<b>a</b>) impedance matching bandwidth and (<b>b</b>) 3dB-AR bandwidth on the bend angles, <math display="inline"><semantics> <msub> <mi>β</mi> <mi>L</mi> </msub> </semantics></math> and <math display="inline"><semantics> <msub> <mi>β</mi> <mi>T</mi> </msub> </semantics></math>.</p>
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<p>Surface current distributions on the surface of the inverted G-shaped strip antenna at different frequencies range at sequential orthogonal phases (sequential time delays). (<b>a</b>) 5 GHz. (<b>b</b>) 5.5 GHz, (<b>c</b>) 6 GHz.</p>
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<p>Experimental setup for measurement of the gain, radiation patterns, and antenna efficiency.</p>
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<p>A photograph of the fabricated antennas. (<b>a</b>) G-shaped strip antenna. (<b>b</b>) Inverted G-shaped strip antenna.</p>
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<p>The fabricated prototype of the inverted G-strip antenna is connected to the VNA to measure the reflection coefficient magnitude, <math display="inline"><semantics> <mrow> <mo>∣</mo> <msub> <mi>S</mi> <mn>11</mn> </msub> <mo>∣</mo> </mrow> </semantics></math>, at the antenna feeding port.</p>
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<p>Frequency response as obtained by simulation compared to that obtained by measurement for the inverted G-shaped strip antenna (<b>a</b>) The <math display="inline"><semantics> <mrow> <mo>∣</mo> <msub> <mi>S</mi> <mn>11</mn> </msub> <mo>∣</mo> </mrow> </semantics></math>. (<b>b</b>) The AR.</p>
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<p>The inverted G-shaped antenna exhibits a dependency on frequency, as obtained by simulation and measurement over the frequency range (4–20 GHz) for (<b>a</b>) The Gain. (<b>b</b>) The radiation efficiency.</p>
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<p>Radiation patterns of (<b>a</b>) RHCP and (<b>b</b>) LHCP fields produced by the inverted G-shaped antenna in the plane <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <msup> <mn>0</mn> <mo>∘</mo> </msup> </mrow> </semantics></math> at the 5 GHz.</p>
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<p>Radiation patterns of (<b>a</b>) RHCP and (<b>b</b>) LHCP fields produced by the inverted G-shaped antenna in the plane <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <msup> <mn>0</mn> <mo>∘</mo> </msup> </mrow> </semantics></math> at the 5.5 GHz.</p>
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<p>Radiation patterns of (<b>a</b>) RHCP and (<b>b</b>) LHCP fields produced by the inverted G-shaped antenna in the plane <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <msup> <mn>0</mn> <mo>∘</mo> </msup> </mrow> </semantics></math> at the 6 GHz.</p>
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<p>Radiation patterns of the total electric field produced by the inverted G-shaped antenna (Linearly Polarized) in the plane <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <msup> <mn>0</mn> <mo>∘</mo> </msup> </mrow> </semantics></math> (<b>a</b>) 9 GHz, (<b>b</b>) 12 GHz, (<b>c</b>) 15 GHz, and (<b>d</b>) 18 GHz.</p>
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13 pages, 4369 KiB  
Article
A Patch Antenna with Enhanced Gain and Bandwidth for Sub-6 GHz and Sub-7 GHz 5G Wireless Applications
by Shehab Khan Noor, Muzammil Jusoh, Thennarasan Sabapathy, Ali Hanafiah Rambe, Hamsakutty Vettikalladi, Ali M. Albishi and Mohamed Himdi
Electronics 2023, 12(12), 2555; https://doi.org/10.3390/electronics12122555 - 6 Jun 2023
Cited by 14 | Viewed by 6779
Abstract
This paper presents a novel microstrip patch antenna design using slots and parasitic strips to operate at the n77 (3.3–4.2 GHz)/n78 (3.3–3.8 GHz) band of sub-6 GHz and n96 (5.9–7.1 GHz) band of sub-7 GHz under 5G New Radio. The proposed antenna is [...] Read more.
This paper presents a novel microstrip patch antenna design using slots and parasitic strips to operate at the n77 (3.3–4.2 GHz)/n78 (3.3–3.8 GHz) band of sub-6 GHz and n96 (5.9–7.1 GHz) band of sub-7 GHz under 5G New Radio. The proposed antenna is simulated and fabricated using an FR-4 substrate with a relative permittivity of 4.3 and copper of 0.035 mm thickness for the ground and radiating planes. A conventional patch antenna with a slot is also designed and fabricated for comparison. A comprehensive analysis of both designs is carried out to prove the superiority of the proposed antenna over conventional dual-band patch antennas. The proposed antenna achieves a wider bandwidth of 160 MHz at 3.45 GHz and 220 MHz at 5.9 GHz, with gains of 3.83 dBi and 0.576 dBi, respectively, compared to the conventional patch antenna with gains of 2.83 dBi and 0.1 dBi at the two frequencies. Parametric studies are conducted to investigate the effect of the parasitic strip’s width and length on antenna performance. The results of this study have significant implications for the deployment of high-gain compact patch antennas for sub-6 GHz and sub-7 GHz 5G wireless communications and demonstrate the potential of the proposed design to enhance performance and efficiency in these frequency bands. Full article
(This article belongs to the Topic Antennas)
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<p>Simulated and fabricated designs of the proposed patch antennas: (<b>a</b>) Simulated Antenna-1, (<b>b</b>) Simulated Antenna-2, (<b>c</b>) Fabricated Antenna-1, (<b>d</b>) Fabricated Antenna-2.</p>
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<p>Simulated and Measured |S<sub>11</sub>| and bandwidth results (<b>a</b>) Antenna-1 setup (<b>b</b>) Antenna-2 setup (<b>c</b>) Comparison between Antenna-1 and Antenna-2 obtained results.</p>
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<p>Radiation pattern measurement of the proposed antenna.</p>
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<p>Simulated antenna gains (<b>a</b>) Antenna-1 gain at 3.45 GHz (<b>b</b>) Antenna-1 gain at 5.9 GHz (<b>c</b>) Antenna-2 gain at 3.45 GHz (<b>d</b>) Antenna-2 gain at 5.9 GHz.</p>
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<p>Simulated antenna gains (<b>a</b>) Antenna-1 gain at 3.45 GHz (<b>b</b>) Antenna-1 gain at 5.9 GHz (<b>c</b>) Antenna-2 gain at 3.45 GHz (<b>d</b>) Antenna-2 gain at 5.9 GHz.</p>
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<p>Proposed antenna (Antenna-2) 2D polar radiation pattern (<b>a</b>) Simulated 2D polar pattern at 3.45 GHz (<b>b</b>) Measured 2D polar pattern at 3.45 GHz (<b>c</b>) Simulated 2D polar pattern at 5.9 GHz (<b>d</b>) Measured 2D polar pattern at 5.9 GHz.</p>
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<p>Simulated results of Antenna-1 and Antenna-2 (<b>a</b>) Gain over frequency (<b>b</b>) Efficiency over frequency.</p>
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<p>Surface current density of Antenna-2 (<b>a</b>) 3.45 GHz (<b>b</b>) 5.9 GHz band.</p>
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<p>Parametric studies (<b>a</b>) Case-1 with reduced length (<b>b</b>) Case-2 with reduced width.</p>
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<p>Simulated reflection coefficient and bandwidth for Case-1 and Case-2.</p>
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<p>Effect of parasitic strip length and width on gain (<b>a</b>) Case-1 at 3.45 GHz (<b>b</b>) Case-1 at 5.9 GHz (<b>c</b>) Case-2 at 3.45 GHz (<b>d</b>) Case-2 at 5.9 GHz.</p>
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<p>Effect of parasitic strip length and width on gain (<b>a</b>) Case-1 at 3.45 GHz (<b>b</b>) Case-1 at 5.9 GHz (<b>c</b>) Case-2 at 3.45 GHz (<b>d</b>) Case-2 at 5.9 GHz.</p>
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11 pages, 1718 KiB  
Communication
Maximizing Antenna Array Aperture Efficiency for Footprint Patterns
by Cibrán López-Álvarez, María Elena López-Martín, Juan Antonio Rodríguez-González and Francisco José Ares-Pena
Sensors 2023, 23(10), 4982; https://doi.org/10.3390/s23104982 - 22 May 2023
Viewed by 1658
Abstract
Despite playing a central role in antenna design, aperture efficiency is often disregarded. Consequently, the present study shows that maximizing the aperture efficiency reduces the required number of radiating elements, which leads to cheaper antennas with more directivity. For this, it is considered [...] Read more.
Despite playing a central role in antenna design, aperture efficiency is often disregarded. Consequently, the present study shows that maximizing the aperture efficiency reduces the required number of radiating elements, which leads to cheaper antennas with more directivity. For this, it is considered that the boundary of the antenna aperture has to be inversely proportional to the half-power beamwidth of the desired footprint for each ϕ-cut. As an example of application, it has been considered the rectangular footprint, for which a mathematical expression was deduced to calculate the aperture efficiency in terms of the beamwidth, synthesizing a rectangular footprint of a 2:1 aspect ratio by starting from a pure real flat-topped beam pattern. In addition, a more realistic pattern was studied, the asymmetric coverage defined by the European Telecommunications Satellite Organization, including the numerical computation of the contour of the resulting antenna and its aperture efficiency. Full article
(This article belongs to the Topic Antennas)
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<p>Circular aperture of radius <span class="html-italic">a</span>. The cylindrical coordinates <math display="inline"><semantics> <mrow> <mi>ρ</mi> <mo>,</mo> <mi>β</mi> </mrow> </semantics></math> refer to the antenna aperture, while the spherical coordinates <math display="inline"><semantics> <mrow> <mi>r</mi> <mo>,</mo> <mi>θ</mi> <mo>,</mo> <mi>ϕ</mi> </mrow> </semantics></math> are used for defining the field point <span class="html-italic">P</span>.</p>
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<p>Shape of the required antenna for a rectangular footprint pattern of 1:1 aspect ratio. The required antenna matches the available one. The contour is obtained from Equation (<a href="#FD10-sensors-23-04982" class="html-disp-formula">10</a>). The solid green line represents the initial antenna, while the solid blue line is the optimal antenna.</p>
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<p>Shape of the required antenna for a rectangular footprint pattern of 2:1 aspect ratio. The required antenna matches the available one. The contour is obtained from Equation (<a href="#FD10-sensors-23-04982" class="html-disp-formula">10</a>). The solid green line represents the initial antenna, while the solid blue line is the optimal antenna.</p>
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<p>Shape of the required antenna for a rectangular footprint pattern of 3:1 aspect ratio. The required antenna exceeds the available one. The contour is obtained from Equation (<a href="#FD10-sensors-23-04982" class="html-disp-formula">10</a>). The solid green line represents the initial antenna, while the dashed red line is the required antenna, and the solid blue line is the optimal antenna.</p>
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<p>Shape of the required antenna for a rectangular footprint pattern of 4:1 aspect ratio. The required antenna exceeds the available one. The contour is obtained from Equation (<a href="#FD10-sensors-23-04982" class="html-disp-formula">10</a>). The solid green line represents the initial antenna, while the dashed red line is the required antenna, and the solid blue line is the optimal antenna.</p>
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<p>Normalized aperture distribution and interpolated image of the reconstructed pattern with a threshold level set at −50 dB from the antenna contour shown in <a href="#sensors-23-04982-f003" class="html-fig">Figure 3</a>. (<b>a</b>) Normalized aperture distribution. (<b>b</b>) Reconstructed pattern.</p>
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<p>Patterns from (<b>a</b>) real and (<b>b</b>) complex roots. The HPBW for each set of roots is obtained at (<b>a</b>) <math display="inline"><semantics> <mrow> <msub> <mi>u</mi> <mn>0</mn> </msub> <mo>=</mo> <mn>4.54</mn> </mrow> </semantics></math> and (<b>b</b>) <math display="inline"><semantics> <mrow> <msub> <mi>u</mi> <mn>0</mn> </msub> <mo>=</mo> <mn>2.86</mn> </mrow> </semantics></math>. Produced with a side-lobe level <math display="inline"><semantics> <mrow> <mi>S</mi> <mi>L</mi> <mi>L</mi> <mo>=</mo> <mo>−</mo> </mrow> </semantics></math>25 dB, <math display="inline"><semantics> <mrow> <mover accent="true"> <mi>n</mi> <mo>¯</mo> </mover> <mo>=</mo> <mn>6</mn> </mrow> </semantics></math> inner roots, <math display="inline"><semantics> <mrow> <mi>M</mi> <mo>=</mo> <mn>2</mn> </mrow> </semantics></math> ripple cycles, and a ripple level of ±0.5 dB.</p>
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<p>Patterns from (<b>a</b>) real and (<b>b</b>) complex roots. The HPBW for each set of roots is obtained at (<b>a</b>) <math display="inline"><semantics> <mrow> <msub> <mi>u</mi> <mn>0</mn> </msub> <mo>=</mo> <mn>2.52</mn> </mrow> </semantics></math> and (<b>b</b>) <math display="inline"><semantics> <mrow> <msub> <mi>u</mi> <mn>0</mn> </msub> <mo>=</mo> <mn>1.75</mn> </mrow> </semantics></math>. Produced with a side-lobe level <math display="inline"><semantics> <mrow> <mi>S</mi> <mi>L</mi> <mi>L</mi> <mo>=</mo> <mo>−</mo> </mrow> </semantics></math>25 dB, <math display="inline"><semantics> <mrow> <mover accent="true"> <mi>n</mi> <mo>¯</mo> </mover> <mo>=</mo> <mn>5</mn> </mrow> </semantics></math> inner roots, <math display="inline"><semantics> <mrow> <mi>M</mi> <mo>=</mo> <mn>1</mn> </mrow> </semantics></math> ripple cycles, and a ripple level of ±0.5 dB.</p>
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<p>Shape of the required antenna for the case of the EuTELSAT antenna contour. The contour is obtained from Equation (<a href="#FD9-sensors-23-04982" class="html-disp-formula">9</a>), using as initial pattern both (<b>a</b>) real and (<b>b</b>) complex flat-topped beam pattern boundaries with two filled zeros. The dashed red line represents the minimum antenna, while the solid blue line is the optimal antenna.</p>
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<p>Shape of the required antenna for the case of the EuTELSAT antenna contour with one only zero. The contour is obtained from Equation (<a href="#FD9-sensors-23-04982" class="html-disp-formula">9</a>), using as initial pattern both (<b>a</b>) real and (<b>b</b>) complex flat-topped beam pattern boundaries with one filled zero. The dashed red line represents the minimum antenna, while the solid blue line is the optimal antenna.</p>
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<p>Contour of continental Europe covered by the EuTELSAT satellite.</p>
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16 pages, 6207 KiB  
Article
Multi-Layer Beam Scanning Leaky Wave Antenna for Remote Vital Signs Detection at 60 GHz
by Solomon Mingle, Despoina Kampouridou and Alexandros Feresidis
Sensors 2023, 23(8), 4059; https://doi.org/10.3390/s23084059 - 17 Apr 2023
Cited by 1 | Viewed by 1797
Abstract
A multi-layer beam-scanning leaky wave antenna (LWA) for remote vital sign monitoring (RVSM) at 60 GHz using a single-tone continuous-wave (CW) Doppler radar has been developed in a typical dynamic environment. The antenna’s components are: a partially reflecting surface (PRS), high-impedance surfaces (HISs), [...] Read more.
A multi-layer beam-scanning leaky wave antenna (LWA) for remote vital sign monitoring (RVSM) at 60 GHz using a single-tone continuous-wave (CW) Doppler radar has been developed in a typical dynamic environment. The antenna’s components are: a partially reflecting surface (PRS), high-impedance surfaces (HISs), and a plain dielectric slab. A dipole antenna works as a source together with these elements to produce a gain of 24 dBi, a frequency beam scanning range of 30°, and precise remote vital sign monitoring (RVSM) up to 4 m across the operating frequency range (58–66 GHz). The antenna requirements for the DR are summarised in a typical dynamic scenario where a patient is to have continuous monitoring remotely, while sleeping. During the continuous health monitoring process, the patient has the freedom to move up to one meter away from the fixed sensor position.The proposed multi-layer LWA system was placed at a distance of 2 m and 4 m from the test subject to confirm the suitability of the developed antenna for dynamic RVSM applications. A proper setting of the operating frequency range (58 to 66 GHz) enabled the detection of both heart beats and respiration rates of the subject within a 30° angular range. Full article
(This article belongs to the Topic Antennas)
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<p>Examples of RVSM applications.</p>
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<p>PRS and HIS unit cell structure where <math display="inline"><semantics> <mrow> <msub> <mi>L</mi> <mrow> <mi>o</mi> <mi>u</mi> <mi>t</mi> </mrow> </msub> <mo>=</mo> <mn>1.50</mn> </mrow> </semantics></math> mm, <math display="inline"><semantics> <mrow> <msub> <mi>L</mi> <mrow> <mi>i</mi> <mi>n</mi> </mrow> </msub> <mo>=</mo> <mn>0.60</mn> </mrow> </semantics></math> mm and periodicity <span class="html-italic">P</span> = 1.75 mm.</p>
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<p>(<b>a</b>) Magnitude and phase reflection coefficient of the PRS unit cell. (<b>b</b>) Phase reflection coefficient of the HIS unit cell with respect to <math display="inline"><semantics> <msub> <mi>L</mi> <mrow> <mi>i</mi> <mi>n</mi> </mrow> </msub> </semantics></math> with <math display="inline"><semantics> <msub> <mi>L</mi> <mrow> <mi>o</mi> <mi>u</mi> <mi>t</mi> </mrow> </msub> </semantics></math> = 1.50 mm.</p>
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<p>The feeding PDA structure. (<b>a</b>) dimensions in front view and back view; (<b>b</b>) simulated and measured <math display="inline"><semantics> <msub> <mi>S</mi> <mn>11</mn> </msub> </semantics></math> and realised gain.</p>
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<p>Proposed antenna structure: (<b>a</b>) the geometry of LWA; (<b>b</b>) view of the LWA along the <math display="inline"><semantics> <mrow> <mi>x</mi> <mi>y</mi> </mrow> </semantics></math> plane; (<b>c</b>) 3D view of the LWA with a frame where <span class="html-italic">y</span> = 5 mm, <span class="html-italic">n</span> = 5.2 mm, and <span class="html-italic">L</span> = 102 mm.</p>
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<p>Theoretically estimated beam scanning range of the LWA from 58 to 66 GHz for a fixed cavity height of (<b>a</b>) <math display="inline"><semantics> <msub> <mi>h</mi> <mrow> <mi>a</mi> <mn>1</mn> </mrow> </msub> </semantics></math> = 2.90 mm with HIS and (<b>b</b>) <math display="inline"><semantics> <msub> <mi>h</mi> <mrow> <mi>a</mi> <mn>1</mn> </mrow> </msub> </semantics></math> = 3.32 mm without HIS.</p>
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<p>LWA simulation results: (<b>a</b>) 2-D FRPs in <span class="html-italic">E</span>-plane at various frequencies within the operating band; (<b>b</b>) 3-D FRP of the proposed LWA at 62.5 GHz.</p>
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<p>Simulated LWA models (<b>a</b>) <math display="inline"><semantics> <msub> <mi>S</mi> <mn>11</mn> </msub> </semantics></math> and (<b>b</b>) realised gain for four different LWA configurations.</p>
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<p>(<b>a</b>) Fabricated LWA prototype with external connections (ant. and ant. B) (<b>b</b>) <math display="inline"><semantics> <msub> <mi>S</mi> <mn>11</mn> </msub> </semantics></math> test setup.</p>
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<p>Measured LWA results compared to simulated in (<b>a</b>) <math display="inline"><semantics> <msub> <mi>S</mi> <mn>11</mn> </msub> </semantics></math> and (<b>b</b>) Realised gain.</p>
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<p>Measured and simulated copolar and crosspolar (<b>a</b>) <span class="html-italic">E</span>-plane and (<b>b</b>) <span class="html-italic">H</span>-plane radiation patterns at 62.5 GHz.</p>
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<p>The measured and simulated (<b>a</b>) <span class="html-italic">E</span>-plane radiation patterns of the multi-layer LWA and (<b>b</b>) radiation efficiency.</p>
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<p>Demonstration of the RVSM experiment setup measurements.</p>
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<p>Measured RVSM results from 2 m distance at (<b>a</b>) 58 GHz, (<b>b</b>) 60 GHz, (<b>c</b>) 62.5 GHz, (<b>d</b>) 64 GHz, and (<b>e</b>) 66 GHz.</p>
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<p>Validation of RVSM using (LHS) non-contact device (RHS) traditional blood pressure machine.</p>
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<p>Measured RVSM results from a 4 m distance at 58 GHz, 62.5 GHz, and 66 GHz.</p>
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12 pages, 9281 KiB  
Communication
Experimental Demonstration of Beam Scanning of Dual-Metasurface Antenna
by Lucia Teodorani, Francesco Vernì, Giorgio Giordanengo, Rossella Gaffoglio and Giuseppe Vecchi
Electronics 2023, 12(8), 1833; https://doi.org/10.3390/electronics12081833 - 12 Apr 2023
Cited by 3 | Viewed by 1933
Abstract
Beam-scanning antennas are employed in a wide range of applications, such as in satellite communications and 5G networks. Current commercial solutions rely mostly on electronically reconfigurable phased arrays, which require complex feeding networks and are affected by high losses, high costs, and are [...] Read more.
Beam-scanning antennas are employed in a wide range of applications, such as in satellite communications and 5G networks. Current commercial solutions rely mostly on electronically reconfigurable phased arrays, which require complex feeding networks and are affected by high losses, high costs, and are often power-hungry. In this paper, a novel beam scanning architecture employing a pair of planar metasurfaces, for use in thin reconfigurable antennas, is presented and experimentally demonstrated. The structure consisted of a radiative passive (non-reconfigurable) modulated metasurface, and a second metasurface that controls beam pointing, operating as a variable-impedance ground plane. Unlike other existing approaches, surface impedance variation was obtained by on-plane varactor diodes, no vias and a single voltage bias. This paper presents a design procedure based on an approximate theoretical model and simulation verification; a prototype of the designed antenna was fabricated for operation in X band, and a good agreement between measured results and simulations was observed. In the presented simple embodiment of the concept, the angular scanning range was limited to 10°; this limitation is discussed in view of future applications. Full article
(This article belongs to the Topic Antennas)
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<p>Structure of the proposed antenna. (<b>a</b>) Schematic architecture. (<b>b</b>) Transverse equivalent network: <math display="inline"><semantics> <msub> <mi>Y</mi> <mi>MMTS</mi> </msub> </semantics></math> is the admittance of the upper metasurface, <math display="inline"><semantics> <msub> <mi>Y</mi> <mi>RMTS</mi> </msub> </semantics></math> is the admittance of the lower (reconfigurable) metasurface, <math display="inline"><semantics> <msub> <mi>d</mi> <mn>1</mn> </msub> </semantics></math> is the thickness of the dielectric layer, <math display="inline"><semantics> <msub> <mi>d</mi> <mn>0</mn> </msub> </semantics></math> indicates the air gap, <math display="inline"><semantics> <msub> <mi>Y</mi> <mi>UP</mi> </msub> </semantics></math> and <math display="inline"><semantics> <msub> <mi>Y</mi> <mi>DOWN</mi> </msub> </semantics></math> are the admittances looking up and down from the interface between the antenna and free space, <math display="inline"><semantics> <msub> <mi>Z</mi> <mi mathvariant="normal">s</mi> </msub> </semantics></math> is the equivalent impenetrable surface impedance that approximates the whole multilayer structure.</p>
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<p>Unit cell of the reconfigurable metasurface.</p>
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<p>Surface reactance of the whole structure vs gap width in the array of strips, for two different values of the varactors’ bias voltage.</p>
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<p>DC voltage distribution in the reconfigurable metasurface.</p>
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<p>Complete model of the designed antenna. (<b>a</b>) Upper sinusoidally modulated reactance surface. (<b>b</b>) Lateral view of the antenna. (<b>c</b>) Reconfigurable plane; the inset shows one of the two DC buses.</p>
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<p>Unit cell of the fully-stacked structure.</p>
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<p>Fabricated antenna prototype. (<b>a</b>) Top modulated metasurface. (<b>b</b>) Reconfigurable impedance plane. (<b>c</b>) Coaxial cable and matched tapered input section at the left end of the upper metasurface (same tapered section is present at the opposite end). (<b>d</b>) Zoom showing the soldered varactors in the lower reconfigurable metasurface.</p>
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<p>Measured and simulated radiation pattern in the E-plane at <math display="inline"><semantics> <mrow> <mn>10.65</mn> </mrow> </semantics></math> GHz for different values of bias voltage.</p>
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<p>Measured <math display="inline"><semantics> <msub> <mi>S</mi> <mn>11</mn> </msub> </semantics></math> and <math display="inline"><semantics> <msub> <mi>S</mi> <mn>21</mn> </msub> </semantics></math> for two different values of bias voltage.</p>
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<p>Simulated radiation pattern at <math display="inline"><semantics> <mrow> <mn>10.75</mn> </mrow> </semantics></math> GHz for different bias voltages for an alternative design with Rogers RO3006 substrate (<math display="inline"><semantics> <mrow> <msub> <mi>ε</mi> <mi>r</mi> </msub> <mo>=</mo> <mn>6.5</mn> </mrow> </semantics></math>).</p>
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19 pages, 12389 KiB  
Article
Design of Broadband Low-RCS Array Antennas Based on Characteristic Mode Cancellation
by Jialiang Han, Dan Jia, Biao Du, Guodong Han, Yongtao Jia and Zekang Zhao
Electronics 2023, 12(7), 1536; https://doi.org/10.3390/electronics12071536 - 24 Mar 2023
Cited by 2 | Viewed by 1714
Abstract
In this letter, a design method for low radar cross section (RCS) array antennas based on characteristic mode cancellation (CMC) is presented. Based on the characteristic mode theory (CMT), two novel microstrip elements are designed by introducing rectangular slots and cross slots, which [...] Read more.
In this letter, a design method for low radar cross section (RCS) array antennas based on characteristic mode cancellation (CMC) is presented. Based on the characteristic mode theory (CMT), two novel microstrip elements are designed by introducing rectangular slots and cross slots, which produce 180° scattering phase difference by adjusting the size of slots. The dominant characteristic modes of the two elements achieve broadband dual-linear polarization CMC and similar radiation performances. The 4 × 4 array antenna consisting of these two antenna elements is designed. The operating band is from 4.55 GHz to 5.49 GHz (relative bandwidth 18.7%). The gain loss of the proposed array is about 0.1 dB compared to the reference array. The monostatic RCS is reduced for dual−linear polarized waves, and the 6 dB radar cross section reduction (RCSR) bandwidths are 62.3% and 35.7%, respectively. The prototype is fabricated and measured. The measured results of radiation pattern and RCS are in good agreement with the simulated results. Full article
(This article belongs to the Topic Antennas)
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<p>Illustration of the array antenna composed of two kinds of elements.</p>
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<p>Geometry of the reference antenna and antenna A. (<b>a</b>) Reference antenna; (<b>b</b>) antenna A.</p>
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<p>Radiation performance simulation results of the reference antenna and antenna A. (<b>a</b>) S<sub>11</sub>; (<b>b</b>) far−field patterns.</p>
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<p><math display="inline"><semantics> <mrow> <msub> <mrow> <mi>A</mi> </mrow> <mrow> <mi>n</mi> </mrow> </msub> </mrow> </semantics></math> of the radiation characteristic modes of the reference antenna and antenna A. (<b>a</b>) Reference antenna; (<b>b</b>) antenna A.</p>
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<p>Radiation characteristic current distribution on the radiation patch at 5.1 GHz. (<b>a</b>) Reference antenna; (<b>b</b>) antenna A.</p>
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<p><math display="inline"><semantics> <mrow> <msub> <mrow> <mi>A</mi> </mrow> <mrow> <mi>n</mi> </mrow> </msub> </mrow> </semantics></math> of scattering characteristic modes of the reference antenna and antenna A. (<b>a</b>) Reference antenna; (<b>b</b>) antenna A.</p>
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<p>The <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="sans-serif">ϕ</mi> </mrow> <mrow> <mi>n</mi> </mrow> </msub> </mrow> </semantics></math> and the difference of <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="sans-serif">ϕ</mi> </mrow> <mrow> <mi>n</mi> </mrow> </msub> </mrow> </semantics></math> of the dominant scattering characteristic modes of the reference antenna and antenna A. (<b>a</b>) The <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="sans-serif">ϕ</mi> </mrow> <mrow> <mi>n</mi> </mrow> </msub> </mrow> </semantics></math> of the reference antenna and antenna A; (<b>b</b>) difference of <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="sans-serif">ϕ</mi> </mrow> <mrow> <mi>n</mi> </mrow> </msub> </mrow> </semantics></math> of the dominant scattering characteristic modes of the reference antenna and antenna A.</p>
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<p>Scattering characteristic current distribution and far−field patterns comparison at 4.4 GHz. (<b>a</b>) Mode 1 of the reference antenna; (<b>b</b>) mode 1 of antenna A.</p>
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<p>Scattering characteristic currents distribution and far−field patterns comparison for at 5.0 GHz. (<b>a</b>) Mode 2 of the reference antenna; (<b>b</b>) mode 2 of antenna A.</p>
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<p>Geometry of antenna B.</p>
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<p>Radiation characteristic modes analysis results of antenna B. (<b>a</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="normal">A</mi> </mrow> <mrow> <mi>n</mi> </mrow> </msub> </mrow> </semantics></math>; (<b>b</b>) characteristic current distribution at 5.1 GHz.</p>
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<p>Radiation performances of antenna A and antenna B. (<b>a</b>) S<sub>11</sub>; (<b>b</b>) far−field patterns.</p>
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<p>Scattering characteristic mode analysis results of antenna B. (<b>a</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>A</mi> </mrow> <mrow> <mi>n</mi> </mrow> </msub> </mrow> </semantics></math> of antenna B; (<b>b</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="sans-serif">ϕ</mi> </mrow> <mrow> <mi>n</mi> </mrow> </msub> </mrow> </semantics></math> of antenna A and antenna B; (<b>c</b>) phase difference of antenna A and antenna B.</p>
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<p>Scattering characteristic currents distribution and far−field patterns comparison for at 7.0 GHz. (<b>a</b>) Mode 2 of antenna A; (<b>b</b>) mode 2 of antenna B.</p>
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<p><math display="inline"><semantics> <mrow> <msub> <mrow> <mi>A</mi> </mrow> <mrow> <mi>n</mi> </mrow> </msub> </mrow> </semantics></math> of scattering characteristic modes of the reference antenna and antenna A. (<b>a</b>) Antenna A; (<b>b</b>) antenna B.</p>
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<p><math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="sans-serif">ϕ</mi> </mrow> <mrow> <mi>n</mi> </mrow> </msub> </mrow> </semantics></math> and difference of <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="sans-serif">ϕ</mi> </mrow> <mrow> <mi>n</mi> </mrow> </msub> </mrow> </semantics></math> of the dominant scattering characteristic modes of the reference antenna and antenna A. (<b>a</b>) <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="sans-serif">ϕ</mi> </mrow> <mrow> <mi>n</mi> </mrow> </msub> </mrow> </semantics></math> of antenna A and antenna B; (<b>b</b>) difference of <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="sans-serif">ϕ</mi> </mrow> <mrow> <mi>n</mi> </mrow> </msub> </mrow> </semantics></math> of the dominant scattering characteristic modes of the reference antenna and antenna A.</p>
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<p>Four forms of proposed arrays. (<b>a</b>) ABAB; (<b>b</b>) ABBA; (<b>c</b>) AABB; (<b>d</b>) checkerboard.</p>
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<p>RCS simulation results of different array forms.</p>
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<p>Geometry of reference array.</p>
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<p>Radiation far−field patterns of the reference and proposed array antenna.</p>
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<p>Monostatic RCS simulated results of the reference and proposed array antennas. (<b>a</b>) <span class="html-italic">x</span>−polarized; (<b>b</b>) <span class="html-italic">y</span>−polarized.</p>
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<p>Fabricated prototype of the proposed array antenna. (<b>a</b>) Structural diagram; (<b>b</b>) radiation far-field patterns test scene.</p>
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<p>Measured and simulated radiation performances of the proposed patch array antenna. (<b>a</b>) S<sub>11</sub>; (<b>b</b>) far−field patterns.</p>
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<p>The RCS test scenario of the array antenna prototype. (<b>a</b>) Array antenna prototype; (<b>b</b>) the RCS test environment.</p>
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<p>Measured and simulated monostatic RCS of the proposed array antenna for vertical incident plane waves. (<b>a</b>) <span class="html-italic">x</span>−polarized; (<b>b</b>) <span class="html-italic">y</span>−polarized.</p>
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<p>RCS of the array antenna in conditions of gap and displacement errors. (<b>a</b>) The space gap between the double PCB layers; (<b>b</b>) displacement between the double PCB layers.</p>
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14 pages, 3686 KiB  
Communication
Compact UWB MIMO Antenna for 5G Millimeter-Wave Applications
by Mohamed Atef Abbas, Abdelmegid Allam, Abdelhamid Gaafar, Hadia M. Elhennawy and Mohamed Fathy Abo Sree
Sensors 2023, 23(5), 2702; https://doi.org/10.3390/s23052702 - 1 Mar 2023
Cited by 22 | Viewed by 3200
Abstract
This paper presents a printed multiple-input multiple-output (MIMO) antenna with the advantages of compact size, good MIMO diversity performance and simple geometry for fifth-generation (5G) millimeter-wave (mm-Wave) applications. The antenna offers a novel Ultra-Wide Band (UWB) operation from 25 to 50 GHz, using [...] Read more.
This paper presents a printed multiple-input multiple-output (MIMO) antenna with the advantages of compact size, good MIMO diversity performance and simple geometry for fifth-generation (5G) millimeter-wave (mm-Wave) applications. The antenna offers a novel Ultra-Wide Band (UWB) operation from 25 to 50 GHz, using a Defective Ground Structure (DGS) technology. Firstly, its compact size makes it suitable for integrating different telecommunication devices for various applications, with a prototype fabricated having a total size of 33 mm × 33 mm × 0.233 mm. Second, the mutual coupling between the individual elements severely impacts the diversity properties of the MIMO antenna system. An effective technique of orthogonally positioning the antenna elements to each other increased their isolation; thus, the MIMO system provides the best diversity performance. The performance of the proposed MIMO antenna was investigated in terms of S-parameters and MIMO diversity parameters to ensure its suitability for future 5G mm-Wave applications. Finally, the proposed work was verified by measurements and exhibited a good match between simulated and measured results. It achieves UWB, high isolation, low mutual coupling, and good MIMO diversity performance, making it a good candidate and seamlessly housed in 5G mm-Wave applications. Full article
(This article belongs to the Topic Antennas)
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<p>The proposed antenna (<b>a</b>) Top view. (<b>b</b>) Bottom view.</p>
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<p>The fabricated prototype of 4 elements MIMO antenna (<b>a</b>) Top View (<b>b</b>) Bottom view.</p>
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<p>Measurement setup of proposed MIMO using ROHDE &amp; SCHWARZ ZVA67 Vector Network Analyzer.</p>
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<p>Steps of design (<b>a</b>) rectangular patch antenna with inset feed and full ground (<b>b</b>) single proposed antenna (top and bottom) (<b>c</b>) 2-port MIMO antenna with orthogonal orientation (<b>d</b>) Final step for 4-port MIMO antenna.</p>
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<p>The prototype of 4 elements MIMO antenna with connector: (<b>a</b>) Top View (<b>b</b>) Bottom view.</p>
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<p>S-parameters coefficient (<b>a</b>) Reflection coefficient (<b>b</b>) Transmission coefficient.</p>
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<p>ECC of the proposed four-element MIMO antenna between port 1 and 2.</p>
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<p>DG between ports 1 and 2 for the proposed four-element MIMO antenna.</p>
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<p>CCL of the proposed four-element MIMO antenna between port 1 and 2.</p>
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<p>ME of the proposed four-element MIMO antenna between port 1 and 2.</p>
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<p>TARC simulated and measured results of the proposed four-element MIMO antenna.</p>
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14 pages, 9989 KiB  
Article
Antipodal Linearly Tapered Slot Antenna with Quasi-Hemispherical Pattern Using Subwavelength Elements
by Rui Wang, Dashuang Liao and Feng Yang
Electronics 2023, 12(3), 628; https://doi.org/10.3390/electronics12030628 - 27 Jan 2023
Viewed by 1799
Abstract
Antennas with quasi-hemispherical radiation patterns are preferred in many wide−area wireless communication systems which require the signals to uniformly cover a wide two−dimensional region. In this work, a simple but effective beamwidth broadening technique based on an antipodal linearly tapered slot antenna (ALTSA) [...] Read more.
Antennas with quasi-hemispherical radiation patterns are preferred in many wide−area wireless communication systems which require the signals to uniformly cover a wide two−dimensional region. In this work, a simple but effective beamwidth broadening technique based on an antipodal linearly tapered slot antenna (ALTSA) is first proposed and then experimentally verified. Compared with most of the reported designs, the proposed antenna can significantly widen beamwidth and achieve a quasi-hemispherical radiation pattern without increasing the overall size and structural complexity. Only two rows of subwavelength metallic elements (eight elements in total) are simply and skillfully printed at specified positions on the dielectric substrate (relative permittivity εr = 2.94 and thickness h = 1.5 mm) of a general ALTSA whose peak gain is 11.7 dBi, approximately 200% half-power beamwidth (HPBW) enlargement can be obtained in all cut-planes containing the end-fire direction at the central frequency of 15 GHz, and the HPBW extensions in different cut-planes have good consistency. Thus, a quasi-hemispherical beam pattern can be acquired. Thanks to the simplicity of this method, the antenna size and structural complexity do not increase, resulting in the characteristics of easy fabrication and integration, being lightweight, and high reliability. This proposed method provides a good choice for wide−beam antenna design and will have a positive effect on the potential applications of wide-area wireless communication systems. Full article
(This article belongs to the Topic Antennas)
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<p>Configurations of the proposed wide-beam antenna. (<b>a</b>) Basic ALTSA. (<b>b</b>) Subwavelength units. (<b>c</b>) Proposed wide-beam antenna. (<b>d</b>) Exploded view. (<b>e</b>) Local geometries.</p>
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<p>Simulated reflection coefficients of the proposed wide−beam antenna and basic ALTSA.</p>
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<p>Simulated gain results of the proposed wide−beam antenna and basic ALTSA in different cut-planes.</p>
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<p>Simulated gain results of the proposed wide−beam antenna and basic ALTSA in different cut-planes.</p>
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<p>HPBW values of two antennas and their ratios in different cut-planes.</p>
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<p>The performances of antennas with different numbers of subwavelength elements in each row. (<b>a</b>) |S<sub>11</sub>|. (<b>b</b>) The amplification of HPBW in all cut-planes.</p>
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<p>The current distribution of the proposed antenna.</p>
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<p>(<b>a</b>) Top view of the fabricated antenna. (<b>b</b>) Bottom view of the fabricated antenna. (<b>c</b>) Photograph of the measurement scene.</p>
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<p>Measured reflection coefficients of the proposed antenna.</p>
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<p>Measured and simulated gain results of the proposed antenna in different cut−planes of (<b>a</b>) phi = 0°, (<b>b</b>) phi = 45°, (<b>c</b>) phi = 90°, and (<b>d</b>) phi = 135°.</p>
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16 pages, 8089 KiB  
Article
A Single-Fed Wideband Circularly Polarized Cross-Fed Cavity-Less Magneto-Electric Dipole Antenna
by Linyu Cai and Kin-Fai Tong
Sensors 2023, 23(3), 1067; https://doi.org/10.3390/s23031067 - 17 Jan 2023
Cited by 3 | Viewed by 2327
Abstract
In this paper, we proposed a new wideband circularly polarized cross-fed magneto-electric dipole antenna. Different from conventional cross-dipole or magneto-electric dipole antennas, the proposed simple geometry realizes a pair of complementary magnetic dipole modes by utilizing the two open slots formed between the [...] Read more.
In this paper, we proposed a new wideband circularly polarized cross-fed magneto-electric dipole antenna. Different from conventional cross-dipole or magneto-electric dipole antennas, the proposed simple geometry realizes a pair of complementary magnetic dipole modes by utilizing the two open slots formed between the four cross-fed microstrip patches for achieving circular polarization and high stable gain across a wide frequency band. No parasitic elements are required for extending the bandwidths; therefore, both the radiation patterns and in-band gain are stable. The simulated field distributions demonstrated the phase complementarity of the two pairs of magnetic and electric dipole modes. A parametric study was also performed to demonstrate the radiation mechanism between the electric and magnetic dipole modes. The radiating elements are realized on a piece of double-sided dielectric substrate fed and mechanically supported by a low-cost commercial semirigid cable. The overall thickness of the antenna is about 0.22λo at the center frequency of axial ratio bandwidth. The measured results show a wide impedance bandwidth (|S11| < −10 dB) of 70.2% from 2.45 to 5.10 GHz. The in-band 3-dB axial ratio bandwidth is 51.5% from 3.0 to 5.08 GHz. More importantly, the gain of the antenna is 9.25 ± 0.56 dBic across the 3-dB axial ratio bandwidth. Full article
(This article belongs to the Topic Antennas)
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<p>Geometry and fabricated prototype of the wideband CP cavity-less ME dipole antenna. <span class="html-italic">L<sub>ground</sub></span> = 80.0, <span class="html-italic">h<sub>sub</sub></span> = 1.27, <span class="html-italic">h<sub>air</sub></span> = 17.7 (0.24 λ<sub>0</sub>), <span class="html-italic">L<sub>patch</sub></span> = 18.5, <span class="html-italic">W<sub>gap</sub></span> = 2.1, <span class="html-italic">R<sub>ring</sub></span> = 5.2, <span class="html-italic">W<sub>ring</sub></span> = 1.5, <span class="html-italic">W<sub>con</sub></span> = 5.2. All dimensions are in millimeters.</p>
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<p>(<b>a</b>) The four fundamental modes of the proposed CP cavity-less ME dipole antenna, (<b>b</b>) <span class="html-italic">E</span>-field of the magnetic dipole modes at <span class="html-italic">t</span> = 0, <span class="html-italic">T</span>/4, <span class="html-italic">T</span>/2, and <span class="html-italic">t</span> = 3<span class="html-italic">T</span>/4 at 4 GHz.</p>
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<p>(<b>a</b>) The four fundamental modes of the proposed CP cavity-less ME dipole antenna, (<b>b</b>) <span class="html-italic">E</span>-field of the magnetic dipole modes at <span class="html-italic">t</span> = 0, <span class="html-italic">T</span>/4, <span class="html-italic">T</span>/2, and <span class="html-italic">t</span> = 3<span class="html-italic">T</span>/4 at 4 GHz.</p>
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<p>Simulated Z<sub>11</sub> of the proposed wideband CP cavity-less ME dipole antenna.</p>
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<p>The z-components of E-field of the proposed antenna (<b>a</b>) <span class="html-italic">t</span> = <span class="html-italic">T</span>/8; (<b>b</b>) <span class="html-italic">t</span> = 3<span class="html-italic">T</span>/8; in <span class="html-italic">xy</span>-plane; (<b>c</b>) <span class="html-italic">t</span> = 5<span class="html-italic">T</span>/8, (<b>d</b>) <span class="html-italic">t</span> = 7<span class="html-italic">T</span>/8, at 4 GHz.</p>
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<p>(<b>a</b>) Impedance (Z<sub>11</sub>), (<b>b</b>) reflection coefficient (S<sub>11</sub>), and (<b>c</b>) gain and axial ratio of the proposed antenna at different <span class="html-italic">L<sub>patch</sub></span>.</p>
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<p>(<b>a</b>) Impedance (Z<sub>11</sub>), (<b>b</b>) reflection coefficient (S<sub>11</sub>), and (<b>c</b>) gain and axial ratio of the proposed antenna at different <span class="html-italic">W<sub>con</sub></span>.</p>
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<p>(<b>a</b>) Impedance (Z<sub>11</sub>), (<b>b</b>) reflection coefficient (S<sub>11</sub>), and (<b>c</b>) gain and axial ratio of the proposed antenna at different <span class="html-italic">W<sub>ring</sub></span>.</p>
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<p>(<b>a</b>) Impedance (Z<sub>11</sub>), (<b>b</b>) reflection coefficient (S<sub>11</sub>), and (<b>c</b>) gain and axial ratio of the proposed antenna at different <span class="html-italic">R<sub>ring</sub></span>.</p>
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<p>(<b>a</b>) Impedance (Z<sub>11</sub>), (<b>b</b>) reflection coefficient (S<sub>11</sub>), and (<b>c</b>) gain and axial ratio of the proposed antenna at different <span class="html-italic">h<sub>air</sub></span>.</p>
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<p>(<b>a</b>) Impedance (Z<sub>11</sub>)<b>,</b> (<b>b</b>) reflection coefficient (S<sub>11</sub>), and (<b>c</b>) gain and axial ratio of the proposed antenna at different <span class="html-italic">L<sub>ground</sub></span>.</p>
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<p>Simulated and measured S<sub>11</sub> of the proposed antenna.</p>
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<p>Simulated and measured gain and axial ratio of the proposed antenna.</p>
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<p>Radiation patterns of the proposed antenna, (<b>a</b>–<b>d</b>) at 3.2 GHz, (<b>e</b>–<b>h</b>) at 4 GHz, and (<b>i</b>–<b>l</b>) at 4.8 GHz, in the four principal planes: <span class="html-italic">ϕ</span> = 0°, 45°, 90°, and 135°.</p>
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<p>Radiation patterns of the proposed antenna, (<b>a</b>–<b>d</b>) at 3.2 GHz, (<b>e</b>–<b>h</b>) at 4 GHz, and (<b>i</b>–<b>l</b>) at 4.8 GHz, in the four principal planes: <span class="html-italic">ϕ</span> = 0°, 45°, 90°, and 135°.</p>
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25 pages, 18266 KiB  
Article
A Novel Integrated UWB Sensing and 8-Element MIMO Communication Cognitive Radio Antenna System
by D Srikar, Anveshkumar Nella, Ranjith Mamidi, Ashok Babu, Sudipta Das, Sunil Lavadiya, Abeer D. Algarni and Walid El-Shafai
Electronics 2023, 12(2), 330; https://doi.org/10.3390/electronics12020330 - 8 Jan 2023
Cited by 6 | Viewed by 2700
Abstract
In this article, a cognitive radio (CR) integrated antenna system, which has 1 sensing and 24 communication antennas, is proposed for better spectrum utilization efficiency. In the 24 communication antennas, 3 different operating band antennas are realized with an 8-element MIMO configuration. The [...] Read more.
In this article, a cognitive radio (CR) integrated antenna system, which has 1 sensing and 24 communication antennas, is proposed for better spectrum utilization efficiency. In the 24 communication antennas, 3 different operating band antennas are realized with an 8-element MIMO configuration. The sensing antenna linked to port 1 is able to sense the spectrum that ranges from 2 to 12 GHz, whereas the communication 8-element MIMO antennas linked with ports 2 to 9, ports 10 to 17 and ports 18 to 25 perform operations in the 2.17–4.74 GHz, 4.57–8.62 GHz and 8.62–12 GHz bands, respectively. Mutual coupling is found to be less than −12 dB between the antenna elements. Peak gain and radiation efficiency of the sensing antenna are better than 2.25 dBi and 82%, respectively, whereas the peak gains and radiation efficiencies of all communication antennas are more than 2.5 dBi and 90%, respectively. Moreover, diversity characteristics of the MIMO antenna are assessed by parameters such as DG, ECC and CCL. It is found that ECC and CCL are less than 0.42 and 0.46 bits/s/Hz, respectively, and also DG is more than 9.1 dB. Full article
(This article belongs to the Topic Antennas)
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<p>Geometry of the 25-port CR MIMO antenna. (<b>a</b>) Structure of the antenna. (<b>b</b>) Top view. (<b>c</b>) Back view.</p>
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<p>Schematic of the antenna linked with port 1.</p>
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<p>Evolution of antenna linked with port 1.</p>
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<p>Evolution of the antenna linked with port 1.</p>
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<p>Schematic of the antennas linked with P2 to P9.</p>
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<p>Evolution of the antennas linked with P2 to P9.</p>
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<p>Reflection coefficients of the antennas in intermediate steps of the antenna linked with port 2.</p>
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<p>Reflection coefficients of the antenna linked with port 2 for different values of ln1.</p>
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<p>Schematic of the antennas linked with P10 to P17.</p>
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<p>Evolution of the antennas linked with P10 to P17.</p>
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<p>Reflection coefficients of the antennas in intermediate steps of the antenna linked with port 10.</p>
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<p>Reflection coefficients of the antenna linked with port 10 for different values of wn2.</p>
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<p>Schematic of the antennas linked with P18 to P24.</p>
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<p>Evolution of the antennas linked with P18 to P24.</p>
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<p>Reflection coefficients of the antennas in intermediate steps of the antenna linked with port 18.</p>
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<p>Reflection coefficient of the antenna linked with port 10 for different values of wf3.</p>
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<p>Inter-elemental spacing of the 25-port CR MIMO antenna.</p>
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<p>Plot of (<b>a</b>) S<sub>11,</sub> (<b>b</b>) S<sub>22,</sub> (<b>c</b>) S<sub>33,</sub> (<b>d</b>) S<sub>44,</sub> (<b>e</b>) S<sub>55,</sub> (<b>f</b>) S<sub>66,</sub> (<b>g</b>) S<sub>77,</sub> (<b>h</b>) S<sub>88,</sub> (<b>i</b>) S<sub>99</sub>.</p>
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<p>Plot of (<b>a</b>) S<sub>11,</sub> (<b>b</b>) S<sub>22,</sub> (<b>c</b>) S<sub>33,</sub> (<b>d</b>) S<sub>44,</sub> (<b>e</b>) S<sub>55,</sub> (<b>f</b>) S<sub>66,</sub> (<b>g</b>) S<sub>77,</sub> (<b>h</b>) S<sub>88,</sub> (<b>i</b>) S<sub>99</sub>.</p>
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<p>Plot of (<b>a</b>) S<sub>10 10,</sub> (<b>b</b>) S<sub>11 11,</sub> (<b>c</b>) S<sub>12 12,</sub> (<b>d</b>) S<sub>13 13,</sub> (<b>e</b>) S<sub>14 14,</sub> (<b>f</b>) S<sub>15 15,</sub> (<b>g</b>) S<sub>16 16,</sub> (<b>h</b>) S<sub>17 17</sub>.</p>
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<p>Plot of (<b>a</b>) S<sub>18 18,</sub> (<b>b</b>) S<sub>19 19,</sub> (<b>c</b>) S<sub>20 20,</sub> (<b>d</b>) S<sub>21 21,</sub> (<b>e</b>) S<sub>22 22,</sub> (<b>f</b>) S<sub>23 23,</sub> (<b>g</b>) S<sub>24 24,</sub> (<b>h</b>) S<sub>25 25</sub>.</p>
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<p>Plot of (<b>a</b>) S<sub>18 18,</sub> (<b>b</b>) S<sub>19 19,</sub> (<b>c</b>) S<sub>20 20,</sub> (<b>d</b>) S<sub>21 21,</sub> (<b>e</b>) S<sub>22 22,</sub> (<b>f</b>) S<sub>23 23,</sub> (<b>g</b>) S<sub>24 24,</sub> (<b>h</b>) S<sub>25 25</sub>.</p>
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<p>Mutual coupling between sensing and communication antennas (<b>a</b>) S<sub>21</sub>, S<sub>31,</sub> and S<sub>41</sub> (<b>b</b>) S<sub>10 1</sub>, S<sub>11 1</sub>, S<sub>12 1</sub>, and S<sub>18 1</sub>.</p>
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<p>Mutual coupling between identical communication antennas (<b>a</b>) S<sub>29</sub>, S<sub>34,</sub> S<sub>45</sub>, and S<sub>56</sub> (<b>b</b>) S<sub>67</sub>, S<sub>78</sub>, S<sub>89</sub>, and S<sub>10 11</sub> (<b>c</b>) S<sub>13 14</sub>, S<sub>19 20</sub>, S<sub>21 22</sub>, and S<sub>23 24</sub>.</p>
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<p>Mutual coupling between identical communication antennas (<b>a</b>) S<sub>29</sub>, S<sub>34,</sub> S<sub>45</sub>, and S<sub>56</sub> (<b>b</b>) S<sub>67</sub>, S<sub>78</sub>, S<sub>89</sub>, and S<sub>10 11</sub> (<b>c</b>) S<sub>13 14</sub>, S<sub>19 20</sub>, S<sub>21 22</sub>, and S<sub>23 24</sub>.</p>
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<p>Mutual coupling between non-identical communication antennas. (<b>a</b>) S<sub>2 10</sub>, S<sub>12 18,</sub> S<sub>3 12</sub>, and S<sub>4 13</sub> (<b>b</b>) S<sub>5 14</sub>, S<sub>5 19</sub>, S<sub>15 20</sub>, and S<sub>6 15</sub> (<b>c</b>) S<sub>21 6</sub>, S<sub>7 22</sub>, S<sub>7 16</sub>, and S<sub>16 23</sub> (<b>d</b>) S<sub>24 8</sub>, S<sub>8 25</sub>, S<sub>17 25</sub>, and S<sub>9 17</sub>.</p>
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<p>Far-field patterns of the sensing antenna in orthogonal planes at (<b>a</b>) 2.5 GHz (XZ), (<b>b</b>) 2.5 GHz (YZ), (<b>c</b>) 5 GHz (XZ), (<b>d</b>) 5 GHz (YZ), (<b>e</b>) 7.5 GHz (XZ), (<b>f</b>) 7.5 GHz (YZ), (<b>g</b>) 10 GHz (XZ), (<b>h</b>) 10 GHz (YZ).</p>
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<p>Far-field patterns of the communication antenna linked with port 2 at (<b>a</b>) 3 GHz (XZ) and (<b>b</b>) 3 GHz (YZ).</p>
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<p>Patterns of the communication antenna linked with port 10 at 6 GHz in (<b>a</b>) XZ and (<b>b</b>) YZ.</p>
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<p>Far-field patterns of the communication antenna linked with port 18 at (<b>a</b>) 9 GHz (XZ) and (<b>b</b>) 9 GHz (YZ).</p>
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<p>Radiation efficiency of the sensing antenna.</p>
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<p>Peak gain of the sensing antenna.</p>
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<p>Radiation efficiencies of the communication antennas.</p>
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<p>Peak gains of the communication antennas.</p>
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<p>Fabricated prototype of the 25-port CR MIMO antenna. (<b>a</b>) Top view. (<b>b</b>) Bottom view.</p>
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<p>ECC and DG of the 25-port CR MIMO antenna. (<b>a</b>) ECC 29, ECC 34, ECC 10 11 and ECC 19 20(<b>b</b>) ECC 45, ECC 56, ECC 13 14 and ECC 21 22 (<b>c</b>) ECC 67, ECC 78, ECC 89 and ECC 23 24 (<b>d</b>) DG 29, DG 34, DG 10 11 and DG 19 20 (<b>e</b>) DG 45, DG 56, DG 13 14 and DG 21 22 (<b>f</b>) DG 67, DG 78, DG 89 and DG 23 24.</p>
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<p>CCL of the 25-port CR MIMO antenna. (<b>a</b>) CCL 29, CCL 34, CCL 10 11 and CCL 19 20 (<b>b</b>) CCL 45, CC 56, CCL 13 14 and CCL 21 22 (<b>c</b>) CCL 67, CCL 78, CCL 89 and CCL 23 24.</p>
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<p>TARC of the 8-element communication antennas associated with (<b>a</b>) ports 2 to 9, (<b>b</b>) ports 10 to 17, and (<b>c</b>) ports 18 to 25.</p>
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<p>MEG of the 8-element communication antennas associated with (<b>a</b>) ports 2 to 9, (<b>b</b>) ports 10 to 17, and (<b>c</b>) ports 18 to 25.</p>
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<p>MEG of the 8-element communication antennas associated with (<b>a</b>) ports 2 to 9, (<b>b</b>) ports 10 to 17, and (<b>c</b>) ports 18 to 25.</p>
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12 pages, 2410 KiB  
Article
A Composite Right/Left-Handed Phase Shifter-Based Cylindrical Phased Array with Reinforced Particles Responsive to Magneto-Static Fields
by Muhammad Ayaz, Adnan Iftikhar, Benjamin D. Braaten, Wesam Khalil and Irfan Ullah
Electronics 2023, 12(2), 306; https://doi.org/10.3390/electronics12020306 - 6 Jan 2023
Cited by 9 | Viewed by 2081
Abstract
A conformal cylindrical phased array antenna excited with composite right/left-handed (CRLH) phase shifters is proposed. The phase tuning of the CRLH phase shifter is achieved by embedding novel magneto-static field-responsive micron-sized particles in its structure. It is shown that through the tiny magnet [...] Read more.
A conformal cylindrical phased array antenna excited with composite right/left-handed (CRLH) phase shifters is proposed. The phase tuning of the CRLH phase shifter is achieved by embedding novel magneto-static field-responsive micron-sized particles in its structure. It is shown that through the tiny magnet activation of these novel magneto-static particles at appropriate locations along the length of CRLH stub and inter-digital fingers, variable phase shifts are obtained. The proposed particle-based CRLH phase shifter operates in C-band (5–6) GHz with a low insertion loss and phase error. The 1 × 4 cylindrical phased array is excited with the four unit cells of the proposed particle-embedded CRLH transmission line phase shifters to scan the main beam at desired scan angles. A prototype of a 1 × 4 cylindrical phased array excited with the particle-based CRLH phase shifters was fabricated, and the results show that the simulated results are in close agreement with the measured results. The conformal cylindrical array with the proposed particle-based CRLH phase shifters has great potential for use in printed and flexible electronics design where commercially available phase shifters have a definite drawback. Full article
(This article belongs to the Topic Antennas)
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<p>(<b>a</b>) P-CRLH unit cell top view <math display="inline"><semantics> <mrow> <mo stretchy="false">(</mo> <mi>A</mi> <mi>l</mi> <mi>l</mi> <mtext> </mtext> <mi>d</mi> <mi>i</mi> <mi>m</mi> <mi>e</mi> <mi>n</mi> <mi>s</mi> <mi>i</mi> <mi>o</mi> <mi>n</mi> <mi>s</mi> <mtext> </mtext> <mi>a</mi> <mi>r</mi> <mi>e</mi> <mtext> </mtext> <mi>i</mi> <mi>n</mi> <mtext> </mtext> <mi>m</mi> <mi>m</mi> <mo>:</mo> <msub> <mi>L</mi> <mi>S</mi> </msub> <mo>=</mo> <mn>8</mn> <mo>,</mo> </mrow> </semantics></math><math display="inline"><semantics> <mrow> <mtext> </mtext> <msub> <mi>W</mi> <mi>S</mi> </msub> <mo>=</mo> <mn>1</mn> <mo>,</mo> <mtext> </mtext> </mrow> </semantics></math><math display="inline"><semantics> <mrow> <msub> <mi>W</mi> <mi>f</mi> </msub> <mo>=</mo> <mn>0.3</mn> <mo>,</mo> <mtext> </mtext> </mrow> </semantics></math><math display="inline"><semantics> <mrow> <msub> <mi>L</mi> <mi>f</mi> </msub> <mo>=</mo> <mn>6</mn> <mo>,</mo> <mtext> </mtext> </mrow> </semantics></math><math display="inline"><semantics> <mrow> <msub> <mi>L</mi> <mrow> <mi>μ</mi> <mi>s</mi> </mrow> </msub> <mo>=</mo> <mn>7.6</mn> <mo>,</mo> <mtext> </mtext> </mrow> </semantics></math><math display="inline"><semantics> <mrow> <msub> <mi>W</mi> <mrow> <mi>μ</mi> <mi>s</mi> </mrow> </msub> <mo>=</mo> <mn>1.8</mn> <mo>,</mo> <mtext> </mtext> </mrow> </semantics></math><math display="inline"><semantics> <mrow> <mi>d</mi> <mo>=</mo> <mn>0.6</mn> <mo>,</mo> <mtext> </mtext> </mrow> </semantics></math><math display="inline"><semantics> <mrow> <msub> <mi>W</mi> <mi>g</mi> </msub> <mo>=</mo> <mn>9</mn> <mo>,</mo> </mrow> </semantics></math> <math display="inline"><semantics> <mrow> <msub> <mi>L</mi> <mi>g</mi> </msub> <mo>=</mo> <mn>22</mn> <mo>,</mo> <mtext> </mtext> </mrow> </semantics></math><math display="inline"><semantics> <mrow> <mi>S</mi> <mo>=</mo> <mn>0.2</mn> <mo stretchy="false">)</mo> </mrow> </semantics></math>; (<b>b</b>) P-CRLH unit cell 3D view.</p>
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<p>Equivalent circuit model of the P-CRLH unit cell.</p>
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<p>(<b>a</b>) Reflection coefficient, insertion loss, and (<b>b</b>) phase response of the unit cell P-CRLH phase shifter with MRSs 1 &amp; 9 activation.</p>
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<p>Parametric study of particle activation in MRSs along the symmetrical stub transmission lines. (<b>a</b>) Reflection coefficient, insertion loss, and (<b>b</b>) phase response of the unit cell P-CRLH phase shifter.</p>
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<p>Schematic of the 1 × 4 P-CRLH excited cylindrical phased array.</p>
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<p>Simulation results of a 1 × 4 P-CRLH excited cylindrical phased array operating in C-band: (<b>a</b>–<b>c</b>) broadside patterns, (<b>d</b>–<b>f</b>) scanned patterns at <math display="inline"><semantics> <mrow> <msup> <mrow> <mn>15</mn> </mrow> <mo>°</mo> </msup> </mrow> </semantics></math> and (<b>g</b>–<b>i</b>) scanned patterns at <math display="inline"><semantics> <mrow> <msup> <mrow> <mn>30</mn> </mrow> <mo>°</mo> </msup> </mrow> </semantics></math>.</p>
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<p>Photographs of (<b>a</b>) A fabricated unit cell P-CRLH phase shifter; (<b>b</b>) A P-CRLH phase shifter-based 1 × 4 cylindrical antenna array in an anechoic chamber.</p>
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<p>Measured radiation patterns of a 1 × 4 P-CRLH-excited cylindrical phased array: (<b>a</b>) broadside pattern, (<b>b</b>) main beam scanning at <math display="inline"><semantics> <mrow> <msup> <mrow> <mn>15</mn> </mrow> <mo>°</mo> </msup> <mo>,</mo> </mrow> </semantics></math> and (<b>c</b>) main beam scanning at <math display="inline"><semantics> <mrow> <msup> <mrow> <mn>30</mn> </mrow> <mo>°</mo> </msup> </mrow> </semantics></math>.</p>
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20 pages, 18366 KiB  
Article
Design of 1 × 2 MIMO Palm Tree Coplanar Vivaldi Antenna in the E-Plane with Different Patch Structure
by Nurhayati Nurhayati, Eko Setijadi, Alexandre Maniçoba de Oliveira, Dayat Kurniawan and Mohd Najib Mohd Yasin
Electronics 2023, 12(1), 177; https://doi.org/10.3390/electronics12010177 - 30 Dec 2022
Cited by 2 | Viewed by 2483
Abstract
In this paper, 1 × 2 MIMO of Palm Tree Coplanar Vivaldi Antenna is presented that simulated at 0.5–4.5 GHz. Some GPR applications require wideband antennas starting from a frequency below 1 GHz to overcome high material loss and achieve deeper penetration. However, [...] Read more.
In this paper, 1 × 2 MIMO of Palm Tree Coplanar Vivaldi Antenna is presented that simulated at 0.5–4.5 GHz. Some GPR applications require wideband antennas starting from a frequency below 1 GHz to overcome high material loss and achieve deeper penetration. However, to boost the gain, antennas are set up in MIMO and this is costly due to the large size of the antenna. When configuring MIMO antenna in the E-plane, there is occasionally uncertainty over which antenna model may provide the optimum performance in terms of return loss, mutual coupling, directivity, beam squint, beam width, and surface current using a given substrate size. However, the configuration of E-plane antenna in MIMO has an issue of mutual coupling if the distance between elements is less than 0.5λ. Furthermore, it produces grating lobes at high frequencies.We implement several types of patch structures by incorporating the truncated, tilt shape, Hlbert and Koch Fractal, Exponential slot, Wave slot, the lens with elips, and metamaterial slot to the radiator by keeping the width of the substrate and the shape of the feeder. The return loss, mutual coupling, directivity, beam squint, beamwidth, and surface current of the antenna are compared for 1 × 2 MIMO CVA. A continuous patch MIMO has a spacing of 0.458λ at 0.5 GHz, which is equivalent to its element width. From the simulation, we found that Back Cut Palm Tree (BCPT) and Horizontale Wave Structure Palm Tree (HWSPT) got the best performance of return loss and mutual scattering at low-end frequency respectively. The improvement of directivity got for Metamaterial Lens Palm Tree (MLPT) of 4.453 dBi if compared with Regular Palm Tree-Coplanar Vivaldi Antena (RPT) at 4 GHz. Elips Lens Palm Tree (ELPT) has the best beam squint performance across all frequencies of 0°. It also gots the best beamwidth at 4.5 GHz of 3.320. In addition, we incorporate the MLPT into the radar application. Full article
(This article belongs to the Topic Antennas)
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<p>The 1 × 2 Coplanar Vivaldi Antenna of (<b>a</b>) Regular Palm Tree (RPT-CVA), (<b>b</b>) Front Cut Palm Tree (FCPT-CVA), (<b>c</b>) Middle Cut Palm Tree (MCPT-CVA), (<b>d</b>) Back Cut Palm Tree (BCPT-CVA), (<b>e</b>) Complete Cut Palm Tree (CCPT-CVA), (<b>f</b>) Left Tilt Palm Tree (LTPT-CVA), (<b>g</b>) Right Tilt Palm Tree (RTPT-CVA), (<b>h</b>) Hilbert Fractal Structure Palm Tree (HFSPT-CVA).</p>
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<p>The 1 × 2 MIMO Coplanar Vivaldi Antenna of (<b>a</b>) Koch Fractal Structure Palm Tree (KFSPT-CVA), (<b>b</b>) Exponential Slot Edge Palm Tree (ESEPT-CVA), (<b>c</b>) Vertical Wave Structure Palm Tree (VWPT-CVA), (<b>d</b>) Horizontale Wave Structure Palm Tree (HWPT-CVA), (<b>e</b>) Regular Lens Palm Tree (RLPT-CVA), (<b>f</b>) Elips Lens Palm Tree (ELPT-CVA), (<b>g</b>) Metamaterial Lens Palm Tree (MLPT-CVA).</p>
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<p><span class="html-italic">S</span><sub>11</sub> and <span class="html-italic">S</span><sub>21</sub> performance of 1 × 2 MIMO (<b>a</b>). Regular Palm Tree Coplanar Vivaldi Antena (RPT-CVA), Front Cut Palm Tree (FCPT-CVA), Middle Cut Palm Tree (MCPT-CVA), and (<b>b</b>). <span class="html-italic">S</span><sub>11</sub> and <span class="html-italic">S</span><sub>21</sub> of Regular Palm Tree Coplanar Vivaldi Antena (RPT-CVA), Back Cut Palm Tree (BCPT-CVA), Complete Cut Palm Tree (CCPT-CVA).</p>
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<p><span class="html-italic">S</span><sub>11</sub> and <span class="html-italic">S</span><sub>21</sub> performance of 1 × 2 MIMO (<b>a</b>). Regular Palm Tree-Coplanar Vivaldi Antena (RPT-CVA), Left Tilt Palm Tree (LTPT-CVA), Right Tilt Palm Tree (RTPT-CVA) and (<b>b</b>). Regular Palm Tree-Coplanar Vivaldi Antena (RPT-CVA), Hilbert Fractal Structure Palm Tree (HFSPT-CVA), Koch Fractal Structure Palm Tree (KFSPT-CVA).</p>
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<p><span class="html-italic">S</span><sub>11</sub> and <span class="html-italic">S</span><sub>21</sub> performance of 1 × 2 MIMO (<b>a</b>). Regular Palm Tree-Coplanar Vivaldi Antena (RPT-CVA), Exponential Slot Edge Palm Tree (ESEPT-CVA), Vertical Wave Structure Palm Tree (VWPT-CVA), and Horizontale Wave Structure Palm Tree (HWPTCVA) and (<b>b</b>) <span class="html-italic">S</span><sub>11</sub> and <span class="html-italic">S</span><sub>21</sub> of Regular Palm Tree-Coplanar Vivaldi Antena (RPT-CVA), Elips Lens Palm Tree (ELPT-CVA), and Metamaterial Lens Palm Tree (MLPT-CVA).</p>
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<p>Directivity of: (<b>a</b>). Single and 1 × 2 Regular Palm Tree (RPT-CVA), 1 × 2Front Cut Palm Tree (FCPT-CVA), 1 × 2 Middle Cut Palm Tree (MCPT-CVA), 1 × 2 Back Cut Palm Tree (BCPT-CVA), 1 × 2 Complete Cut Palm Tree (CCPT-CVA) and (<b>b</b>). Single and 1 × 2 of Regular Palm Tree (RPT-CVA), 1 × 2 Left Tilt Palm Tree (LTPT-CVA), 1 × 2 Right Tilt Palm Tree (RTPT-CVA), 1 × 2 Hilbert Fractal Structure Palm Tree (HFSPT-CVA), 1 × 2 Koch Fractal Structure Palm Tree (KFSPT-CVA).</p>
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<p>Directivity of (<b>a</b>). Element and 1 × 2 Regular Palm Tree- (RPT-CVA), 1 × 2 Exponential Slot Edge Palm Tree (ESE-CVA), 1 × 2 Vertical Wave Structure Palm Tree (VWPT-CVA), 1 × 2 Horizontale Wave Structure Palm Tree (HWPT-CVA), and (<b>b</b>). Element and 1 × 2 Regular Palm Tree-Coplanar Vivaldi Antena (RPT-CVA), 1 × 2 Regular Lens Palm Tree (RLPT-CVA), 1 × 2 Elips Lens Palm Tree (ELPT-CVA), And 1 × 2 Metamaterial Lens Palm Tree (MLPT-CVA).</p>
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<p>Side Lobe Level of (<b>a</b>). Element and 1 × 2 Regular Palm Tree (RPT-CVA), 1 × 2 Front Cut Palm Tree (FCPT-CVA), 1 × 2 Middle Cut Palm Tree (MCPT-CVA), 1 × 2 Back Cut Palm Tree (BCPT-CVA), 1 × 2 Complete Cut Palm Tree (CCPT-CVA), 1 × 2 Left Tilt Palm Tree (LTPT-CVA), 1 × 2 Right Tilt Palm Tree (RTPT-CVA), 1 × 2 Hilbert Fractal Structure Palm Tree (HFSPT-CVA), and (<b>b</b>). Element and 1 × 2 Regular Palm Tree (RPT-CVA), 1 × 2 Koch Fractal Structure Palm Tree (KFSPT-CVA), 1 × 2 Exponential Slot Edge Palm Tree (ESEPT-CVA), 1 × 2 Vertical Wave Structure Palm Tree (VWSPT-CVA), 1 × 2 Horizontale Wave Structure Palm Tree (HWSPT-CVA), 1 × 2 Regular Lens Palm Tree (RLPT-CVA), 1 × 2 Elips Lens Palm Tree (ELPT-CVA), and 1 × 2 Metamaterial Lens Palm Tree (MLPT-CVA).</p>
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<p>Beam Squint Performance of (<b>a</b>) Element and 1 × 2 Regular Palm Tree (RPT-CVA), 1 × 2 Front Cut Palm Tree (FCPT-CVA), 1 × 2 Middle Cut Palm Tree (MCPT-CVA), 1 × 2 Back Cut Palm Tree (BCPT-CVA), 1 × 2 Complete Cut Palm Tree (CCPT-CVA), 1 × 2 Left Tilt Palm Tree (LTPT-CVA), Right Tilt Palm Tree (RTPT-CVA), 1 × 2 Hilbert Fractal Structure Palm Tree (HFSPT-CVA), and (<b>b</b>). Element and 1 × 2 Regular Palm Tree (RPT-CVA), 1 × 2 Koch Fractal Structure Palm Tree (KFSPT-CVA), 1 × 2 Exponential Slot Edge Palm Tree (ESEPT-CVA), 1 × 2 Vertical Wave Structure Palm Tree (VWSPT-CVA), 1 × 2 Horizontale Wave Structure Palm Tree (HWSPT-CVA), 1 × 2 Regular Lens Palm Tree (RLPT-CVA), 1 × 2 Elips Lens Palm Tree (ELPT-CVA), and 1 × 2 Metamaterial Lens Palm Tree (MLPT-CVA).</p>
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<p>Beamwidth of (<b>a</b>). Element and 1 × 2 Regular Palm Tree (RPT-CVA), 1 × 2 Front Cut Palm Tree (FCPT-CVA), 1 × 2 Middle Cut Palm Tree (MCPT-CVA), 1 × 2 Back Cut Palm Tree (BCPT-CVA), 1 × 2 Complete Cut Palm Tree (CCPT-CVA), 1 × 2 Left Tilt Palm Tree (LTPT-CVA), 1 × 2 Right Tilt Palm Tree (RTPT-CVA), 1 × 2 Hilbert Fractal Structure Palm Tree (HFSPT-CVA), and (<b>b</b>). Element and 1 × 2 Regular Palm Tree (RPT-CVA), 1 × 2 Koch Fractal Structure Palm Tree (KFSPT-CVA), 1 × 2 Exponential Slot Edge Palm Tree (ESEPT-CVA), 1 × 2 Vertical Wave Structure Palm Tree (VWSPT-CVA), 1 × 2 Horizontale Wave Structure Palm Tree (HWSPT-CVA), 1 × 2 Regular Lens Palm Tree (RLPT-CVA), 1 × 2 Elips Lens Palm Tree (ELPT-CVA), and 1 × 2 Metamaterial Lens Palm Tree (MLPT-CVA).</p>
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<p>Radiation Pattern in the E-Plane of (<b>a</b>). RPT-CVA vs FCPT-CVA at 2 GHz, (<b>b</b>). RPT-CVA vs HFSPT-CVA at 2 GHz, (<b>c</b>) RPT-CVA vs ESE-CVA at 4 GHz, and (<b>d</b>). RPT-CVA vs MLPT-CVA at 4 GHz.</p>
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<p>Surface current performance of (<b>a</b>) Regular Palm Tree (RPT-CVA), (<b>b</b>) Front Cut Palm Tree (FCPT-CVA), (<b>c</b>) Back Cut Palm Tree (BCPT-CVA), (<b>d</b>) Hilbert Fractal Structure Palm Tree (HFSPT-CVA), (<b>e</b>) Vertical Wave Structure Palm Tree (VWSPT-CVA), (<b>f</b>) Horizontale Wave Structure Palm Tree (HWSPT-CVA), (<b>g</b>) Exponential Slot Edge Palm Tree (ESEPT-CVA), (<b>h</b>) Elips Lens Palm Tree (ELPT-CVA), and (<b>i</b>) Metamaterial Lens Palm Tree (MLPT-CVA).</p>
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<p>Surface current performance of (<b>a</b>) Regular Palm Tree (RPT-CVA), (<b>b</b>) Front Cut Palm Tree (FCPT-CVA), (<b>c</b>) Middle Cut Palm Tree (MCPT-CVA), (<b>d</b>) Back Cut Palm Tree (BCPT-CVA), (<b>e</b>) Complete Cut Palm Tree (CCPT-CVA), (<b>f</b>) Left Tilt Palm Tree (LTPT-CVA), (<b>g</b>) Right Tilt Palm Tree (RTPT-CVA), (<b>h</b>) Hilbert Fractal Structure Palm Tree (HFSPT-CVA), (<b>i</b>) Koch Fractal Structure Palm Tree (KFSPT-CVA), (<b>j</b>) Exponential Slot Edge Palm Tree (ESEPT-CVA), (<b>k</b>) Vertical Wave Structure Palm Tree (VWSPT-CVA), (<b>l</b>) Horizontale Wave Structure Palm Tree (HWSPT-CVA), (<b>m</b>) Regular Lens Palm Tree (RLPT-CVA), (<b>n</b>) Elips Lens Palm Tree (ELPT-CVA), and (<b>o</b>) Metamaterial Lens Palm Tree (MLPT-CVA).</p>
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<p>Simulation and measurement result of (<b>a</b>) ESEPT-CVA and (<b>b</b>) MLPT-CVA.</p>
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<p>Radar target measurement with MLPT-CVA prototype.</p>
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<p>Radar target detection in the xy-planes.</p>
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15 pages, 4190 KiB  
Article
Evolutionary Computation for Sparse Synthesis Optimization of CCAAs: An Enhanced Whale Optimization Algorithm Method
by Bohao Tang, Lihua Cai, Shuai Yang, Jiaxing Xu and Yi Yu
Future Internet 2022, 14(12), 347; https://doi.org/10.3390/fi14120347 - 22 Nov 2022
Cited by 1 | Viewed by 1342
Abstract
Concentric circular antenna arrays (CCAAs) can obtain better performance than other antenna arrays. However, high overhead and excessive sidelobes still make its application difficult. In this paper, we consider the sparse synthesis optimization of CCAAs. Specifically, we aim to turn off a specific [...] Read more.
Concentric circular antenna arrays (CCAAs) can obtain better performance than other antenna arrays. However, high overhead and excessive sidelobes still make its application difficult. In this paper, we consider the sparse synthesis optimization of CCAAs. Specifically, we aim to turn off a specific number of antennas while reducing the sidelobe of CCAAs. First, we formulate an optimization problem and present the solution space. Then, we propose a novel evolutionary method for solving the optimization problem. Our proposed method introduces hybrid solution initialization, hybrid crossover method, and hybrid update methods. Simulation results show the effectiveness of the proposed algorithm and the proposed improvement factors. Full article
(This article belongs to the Topic Antennas)
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<p>CCAA model.</p>
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<p>Inactivated or activated antenna model.</p>
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<p>Beam patterns obtained by different algorithms.</p>
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<p>Convergence rates obtained by different algorithms.</p>
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<p>State of the CCAA.</p>
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11 pages, 3874 KiB  
Article
Design of a Dual-Polarization Dipole Antenna for a Cylindrical Phased Array in Ku-Band
by Ning Zhang, Zhenghui Xue, Pei Zheng, Lu Gao and Jia Qi Liu
Electronics 2022, 11(22), 3796; https://doi.org/10.3390/electronics11223796 - 18 Nov 2022
Cited by 2 | Viewed by 3105
Abstract
This paper proposes a dual-polarization dipole antenna for a cylindrical phased array working in Ku-band. The dipole antenna is double-layer structured and is composed of two orthogonal butterfly shaped dipole radiators, two ground co-planar waveguide (GCPW) feeding structures and vias. Each dipole is [...] Read more.
This paper proposes a dual-polarization dipole antenna for a cylindrical phased array working in Ku-band. The dipole antenna is double-layer structured and is composed of two orthogonal butterfly shaped dipole radiators, two ground co-planar waveguide (GCPW) feeding structures and vias. Each dipole is in the shape of a butterfly. The dipole patch is grooved triangularly and one side of it is bent into an N shape, which effectively expands the working frequency band of the antenna. The double-layer structure improves the isolation between the antenna ports. The antenna works between 15 GHz to 16.2 GHz and the isolation between the antenna’s two feeding ports in this band is better than 20 dB. The proposed dipole antenna is applied in a 32-element cylinder array. The simulation and measured results show that the array can scan between −60° to +60° in the azimuth plane with a gain fluctuation less than 2.5 dB. Therefore, the proposed design is an attractive candidate for conformal devices at Ku-band frequencies, and it also has a great potential for application in larger antenna arrays. Full article
(This article belongs to the Topic Antennas)
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<p>Dipole Antenna model.</p>
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<p>Effect of side bending on antenna bandwidth. (<b>a</b>) |S11| and (<b>b</b>) |S12|.</p>
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<p>Detailed antenna structure: (<b>a</b>) side view; (<b>b</b>) top view; (<b>c</b>) isolation ground; and (<b>d</b>) bottom view.</p>
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<p>Simulation and measurement results of the dipole antenna: (<b>a</b>) |S11|; (<b>b</b>) |S12|; and (<b>c</b>) co/cross-pol radiation patterns.</p>
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<p>Current distributions of the proposed antenna fed by (<b>a</b>) Port 1 and (<b>b</b>) Port 2.</p>
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<p>Array antenna model of (<b>a</b>) overall view (<b>b</b>) top view.</p>
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<p>Simulation results of the array in the azimuthal plane.</p>
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<p>The prototype of the phased array. (<b>a</b>) Array photograph (<b>b</b>) the measurement scene.</p>
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<p>Simulated and measured results for different scan angles. (<b>a</b>) φ = 0°. (<b>b</b>) φ = −30°. (<b>c</b>) φ = 30°. (<b>d</b>) φ = −60°. (<b>e</b>) φ = 60°.</p>
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<p>Simulated radiation efficiency of the array at broadside.</p>
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16 pages, 11015 KiB  
Article
Radio Astronomical Antennas in the Central African Region to Improve the Sampling Function of the VLBI Network in the SKA Era?
by Marcellin Atemkeng, Patrice Okouma, Eric Maina, Roger Ianjamasimanana and Serges Zambou
Sensors 2022, 22(21), 8466; https://doi.org/10.3390/s22218466 - 3 Nov 2022
Cited by 1 | Viewed by 1979
Abstract
On the African continent, South Africa has world-class astronomical facilities for advanced radio astronomy research. With the advent of the Square Kilometre Array project in South Africa (SA SKA), six countries in Africa (SA SKA partner countries) have joined South Africa to contribute [...] Read more.
On the African continent, South Africa has world-class astronomical facilities for advanced radio astronomy research. With the advent of the Square Kilometre Array project in South Africa (SA SKA), six countries in Africa (SA SKA partner countries) have joined South Africa to contribute towards the African Very Long Baseline Interferometry (VLBI) Network (AVN). Each of the AVN countries aims to construct a single-dish radio telescope that will be part of the AVN, the European VLBI Network, and the global VLBI network. The SKA and the AVN will enable very high sensitivity VLBI in the southern hemisphere. In the current AVN, there is a gap in the coverage in the central African region. This work analyses the increased scientific impact of having additional antennas in each of the six countries in central Africa, i.e., Cameroon, Gabon, Congo, Equatorial Guinea, Chad, and the Central African Republic. A number of economic human capital impacts of having a radio interferometer in central Africa are also discussed. This work also discusses the recent progress on the AVN project and shares a few lessons from some past successes in ground stations retrofitting. Full article
(This article belongs to the Topic Antennas)
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<p>Decommissioned 32-m large satellite earth station antennas in Gabon.</p>
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<p>Green points: locations of the Kuntunse antenna in Ghana, the MeerKAT stations in South Africa, and the EVN. Red points: locations of abandoned old telecommunication satellite facilities in the ECCAS region and/or possible sites to build new radio telescopes.</p>
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<p>The EVN combined with the MeerKAT telescope (left panels) and the EVN combined with the MeerKAT telescope and the Kuntunse antenna in Ghana (right panels).</p>
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<p>ECCAS antennas <math display="inline"><semantics> <mrow> <mi>u</mi> <mi>v</mi> </mrow> </semantics></math>-coverage at 1.4 GHz at four declinations (<math display="inline"><semantics> <mrow> <mo>−</mo> <mn>20</mn> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <mo>+</mo> <mn>20</mn> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <mo>+</mo> <mn>45</mn> </mrow> </semantics></math>, and <math display="inline"><semantics> <mrow> <mo>+</mo> <mn>60</mn> </mrow> </semantics></math> deg), 10 h observation, and 16 MHz total bandwidth showing a lot of holes or gaps.</p>
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<p>Performance of the global VLBI <math display="inline"><semantics> <mrow> <mi>u</mi> <mi>v</mi> </mrow> </semantics></math>-coverage. The ECCAS antennas are correlated with the Kuntunse antenna, the MeerKAT telescope, and the EVN. The <math display="inline"><semantics> <mrow> <mi>u</mi> <mi>v</mi> </mrow> </semantics></math>-coverage is well-filled because of the extra medium-length baselines that relate the EVN and MeerKAt telescope to the ECCAS antennas.</p>
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<p>Simulated sky model with two point sources: a 1 Jy point source at the phase centre and a far-field point source with 5 Jy brightness located at 6 arcmin from the phase centre as seen by the EVN + MeerKAT + Kuntunse telescope (left) and by the EVN + MeerKAT + Kuntunse + ECCAS telescope (right) for a simulated observation at 16 GHz. To corrupt the simulation, 1 Jy Gaussian noise per visibility is used. The data are sampled during a total period of 10 h with 1 s integration time and using a total bandwidth of 16 MHz divided into 64 channels. Then, a few pixels are imaged to visualise only the source at the phase centre.</p>
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9 pages, 6093 KiB  
Article
A Simple and Effective Approach for Scattering Suppression in Multiband Base Station Antennas
by Madiha Farasat, Dushmantha Thalakotuna, Zhonghao Hu and Yang Yang
Electronics 2022, 11(21), 3423; https://doi.org/10.3390/electronics11213423 - 22 Oct 2022
Cited by 3 | Viewed by 2353
Abstract
The high band pattern distortions in an 1810–2690 MHz frequency band, introduced due to low band radiators working in 690–960 MHz, are mitigated by a simple yet effective change to the low band-radiating elements. A novel horizontal and vertical radiating element is designed [...] Read more.
The high band pattern distortions in an 1810–2690 MHz frequency band, introduced due to low band radiators working in 690–960 MHz, are mitigated by a simple yet effective change to the low band-radiating elements. A novel horizontal and vertical radiating element is designed instead of a conventional slant polarized low band-radiating element to reduce the scattering. The slant polarization is achieved from the horizontal and vertical dipoles, using a 180° hybrid coupler. The vertical dipole length is optimized to improve the high band patterns. The experimental results verified that the proposed horizontal and vertical low band dipole result in the reduction of high band pattern distortions. The low band-radiating elements provide >12 dB return loss over the entire frequency band 690–960 MHz and provide comparable pattern performance to a conventional slant low band dipole. Full article
(This article belongs to the Topic Antennas)
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Figure 1
<p>(<b>a</b>) Slant dipole configuration used in traditional interleaved scheme for dual-band dual-polarized BSA. (<b>b</b>) The proposed horizontal and vertical LB dipole configuration (LBHV) with HB subarrays. P1 and P3 refer to +45 polarizations; P2 and P4 refer to −45 polarization.</p>
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<p>High band-only and high band with slant LB antenna-measured azimuth patterns at (<b>a</b>) 1.8 GHz, (<b>b</b>) 2.4 GHz and (<b>c</b>) 2.6 GHz.</p>
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<p>High-band measured azimuth radiation patterns with a LBHV element at (<b>a</b>) 1.8 GHz, (<b>b</b>) 2.4 GHz, and (<b>c</b>) 2.6 GHz.</p>
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<p>Measured 10 dB azimuth beamwidth of HB arrays with (<b>a</b>) LBHV element (<b>b</b>) LB slant element, 10 dB azimuth beam peak of HB arrays with (<b>c</b>) LBHV element (<b>d</b>) LB slant element, 3 dB azimuth beam peak of HB arrays with (<b>e</b>) LBHV element (<b>f</b>) LB slant element. The P1, P2, P3, and P4 refer to polarizations indicated in <a href="#electronics-11-03423-f001" class="html-fig">Figure 1</a>.</p>
Full article ">Figure 4 Cont.
<p>Measured 10 dB azimuth beamwidth of HB arrays with (<b>a</b>) LBHV element (<b>b</b>) LB slant element, 10 dB azimuth beam peak of HB arrays with (<b>c</b>) LBHV element (<b>d</b>) LB slant element, 3 dB azimuth beam peak of HB arrays with (<b>e</b>) LBHV element (<b>f</b>) LB slant element. The P1, P2, P3, and P4 refer to polarizations indicated in <a href="#electronics-11-03423-f001" class="html-fig">Figure 1</a>.</p>
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<p>(<b>a</b>) Conversion of slant-polarized inputs to H and V polarized inputs, (<b>b</b>) creation of virtual slant polarizations from H and V feed signals.</p>
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<p>Circuit theory model of matching circuit.</p>
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<p>(<b>a</b>) Perspective view of the LBHV. The horizontal dipole is 135 mm long, and the vertical dipole is 60 mm long, (<b>b</b>) details of the LB horizontal dipole (LBHD) feed, (<b>c</b>) details of the LB vertical dipole (LBVD) feed.</p>
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<p>(<b>a</b>) The measured return loss of the short V-dipole and long H-dipole. (<b>b</b>) VSWR of proposed antenna LBHV.</p>
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<p>Fabricated prototype with (<b>a</b>) LBHV antenna element (<b>b</b>) LB slant element.</p>
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<p>Measured isolation of proposed antenna LBHV with slant.</p>
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18 pages, 2995 KiB  
Article
Beamforming with 1 × N Conformal Arrays
by Irfan Ullah, Benjamin D. Braaten, Adnan Iftikhar, Symeon Nikolaou and Dimitris E. Anagnostou
Sensors 2022, 22(17), 6616; https://doi.org/10.3390/s22176616 - 1 Sep 2022
Cited by 1 | Viewed by 2093
Abstract
The rapid growth of wireless spectrum access through cellular and IoT devices, for example, requires antennas with more capabilities such as being conformal and self-adapting beamforming. In this paper, the adaptive beamforming patterns of microstrip patch antenna arrays on changing flexible (or conformal) [...] Read more.
The rapid growth of wireless spectrum access through cellular and IoT devices, for example, requires antennas with more capabilities such as being conformal and self-adapting beamforming. In this paper, the adaptive beamforming patterns of microstrip patch antenna arrays on changing flexible (or conformal) curved surfaces are developed by deriving array coefficients based on the projection method that includes the mutual coupling between elements. A linear four-element microstrip patch antenna array is then embedded on two deformed conformal surfaces to investigate the projection method for desired beamforming patterns. The generated beamforming radiation patterns using the computed weighting coefficients are validated with theoretical equations evaluated in MATLAB, full-wave simulations in HFSS and measurement results. The measured results of the fabricated system agree with the simulated results. Furthermore, new guidelines are provided on the effects of mutual coupling and changing conformal surfaces for various beam-forming patterns. Such demonstrations pave the way to an efficient and robust conformal phased-array antenna with multiple beam forming and adaptive nulling capabilities. Full article
(This article belongs to the Topic Antennas)
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Figure 1
<p>(<b>a</b>) Illustration of the antenna elements on a singly curved (wedge) surface and (<b>b</b>) illustration of the antenna elements on a cylindrical-shaped surface.</p>
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<p>Geometrical illustration of the proposed projection method for (<b>a</b>) antenna array on a wedged-shaped surface with desired signal at angle <span class="html-italic">θ<sub>SOI</sub></span>; (<b>b</b>) antenna array on a wedged-shaped surface with undesired signal at angle <span class="html-italic">θ<sub>SNOI</sub></span> and (<b>c</b>) antenna array on a cylindrical curvature with desired (undesired) signal at angle <span class="html-italic">θ<sub>SOI(SNOI)</sub></span>.</p>
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<p><span class="html-italic">N</span>-port network illustration of conformal array with a signal of interest at angle <math display="inline"><semantics> <mrow> <msub> <mi>θ</mi> <mrow> <mi>S</mi> <mi>O</mi> <mi>I</mi> </mrow> </msub> </mrow> </semantics></math> or <math display="inline"><semantics> <mrow> <msub> <mi>θ</mi> <mrow> <mi>S</mi> <mi>N</mi> <mi>O</mi> <msub> <mi>I</mi> <mi>n</mi> </msub> </mrow> </msub> </mrow> </semantics></math>.</p>
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<p>(<b>a</b>) Topology of the four-element beamforming array, (<b>b</b>) a picture of the power divider, voltage-controlled phase shifters and voltage-controlled attenuators used for measurements and (<b>c</b>) a picture of the microstrip patch elements used for attachment to conformal surfaces.</p>
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<p>(<b>a</b>) Photograph of the four-element beamforming array being measured on the singly curved non-conducting wedge surface with <span class="html-italic">θ<sub>b</sub></span> = 30° and (<b>b</b>) photograph of the four-element beamforming array being measured on the cylindrical-shaped surface with <span class="html-italic">r</span> = 10 cm.</p>
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<p>(<b>a</b>) Pattern 1 beamforming results for the 1 × 4 microstrip patch array on the singly curved (wedge-shaped) surface with <span class="html-italic">θ<sub>b</sub></span> = 30°; (<b>b</b>) Pattern 2 beamforming results for the 1 × 4 microstrip patch array on the singly curved (wedge-shaped) surface with <span class="html-italic">θ<sub>b</sub></span> = 30°.</p>
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<p>(<b>a</b>) Pattern 1 beamforming results for the 1 × 4 microstrip patch array on the singly curved (wedge-shaped) surface with <span class="html-italic">θ<sub>b</sub></span> = 45°; and (<b>b</b>) Pattern 2 beamforming results for the 1 × 4 microstrip patch array on the singly curved (wedge-shaped) surface with <span class="html-italic">θ<sub>b</sub></span> = 45°.</p>
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<p>(<b>a</b>) Pattern 1 beamforming results for the 1 × 4 microstrip patch array on the cylindrical curvature surface with <span class="html-italic">r</span> = 10 cm; (<b>b</b>) Pattern 2 beamforming results for the 1 × 4 microstrip patch array on the cylindrical curvature surface with <span class="html-italic">r</span> = 10 cm.</p>
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<p>(<b>a</b>) Magnitude of the array weights for the 1 × 4 array on the singly curved (wedge) surface and (<b>b</b>) Phase of the array weights for the 1 × 4 array on the singly curved (wedge) surface.</p>
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18 pages, 19247 KiB  
Communication
Coplanar Meta-Surface-Based Substrate-Integrated Waveguide Antennas with Broadband and Low Reflections for K-Band Beam Scanning
by Chunli Wang, Dongxing Gao, Likai Liang and Yanling Wang
Sensors 2022, 22(17), 6353; https://doi.org/10.3390/s22176353 - 24 Aug 2022
Viewed by 2292
Abstract
Four novel substrate-integrated waveguide (SIW) antennas are proposed, in order to obtain K-band beam scanning through the coplanar meta-surfaces of properly devised complementary split-ring resonators. More specifically, coplanar rhombus- and hexagon-shaped meta-surfaces replace the metallized via holes in the traditional SIW structure, achieving [...] Read more.
Four novel substrate-integrated waveguide (SIW) antennas are proposed, in order to obtain K-band beam scanning through the coplanar meta-surfaces of properly devised complementary split-ring resonators. More specifically, coplanar rhombus- and hexagon-shaped meta-surfaces replace the metallized via holes in the traditional SIW structure, achieving low reflection and wide bandwidth, respectively. Another trapezoid-shaped meta-surface is introduced, in order to realize good leaky-wave radiation performance with high-gain beam scanning in both rhombus- and hexagon-shaped SIW components. These designs are further extended to two different mixed types of two-row meta-surfaces, with the rhombus and hexagon structures combined in different orders to enhance the complex SIW transmission lines and antennas, which can simultaneously obtain good reflection and bandwidth with different priority, depending on the arrangement. We explain the performance differences with rhombus and hexagon meta-surfaces through the analysis of relevant equivalent circuit models and extracting the effective medium parameters, and we verify the bandwidths and radiations of four SIW antennas both numerically and experimentally. The maximum gains of the four antennas are 18.1 dBi, 17.0 dBi, 18.8 dBi and 17.1 dBi, where the corresponding relative bandwidths are 10.74%, 19.42%, 14.13% and 18.38%. The maximum simulated radiation efficiency and aperture efficiency of the proposed antennas are 91.20% and 61.12%, respectively. Our approach for generating flexible and selectable tuned electromagnetic fields from SIWs is applicable for the development of mm-Wave antennas or sensors on PCB-integrated platforms for highly directive scanning radiation. Full article
(This article belongs to the Topic Antennas)
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<p>The Floquet model and corresponding equivalent circuits of the (<b>a</b>) rhombus- and (<b>b</b>) hexagon-shaped unit cells. Each unit has the dimensions of 2.5 mm × 2.5 mm × 0.787 mm and the substrate is chosen as RT5880 material, with relative permittivity of 2.2 and loss tangent of 0.001. Moreover, <span class="html-italic">a</span> = 2.5 mm, <span class="html-italic">h</span> = 0.787 mm, <span class="html-italic">H</span> = 3.75 mm, <math display="inline"><semantics> <msub> <mi>l</mi> <mn>1</mn> </msub> </semantics></math> = 1.58 mm, <math display="inline"><semantics> <msub> <mi>l</mi> <mn>2</mn> </msub> </semantics></math> = 1.2 mm, <math display="inline"><semantics> <msub> <mi>θ</mi> <mn>1</mn> </msub> </semantics></math> = <math display="inline"><semantics> <msup> <mn>90</mn> <mo>∘</mo> </msup> </semantics></math>, <math display="inline"><semantics> <msub> <mi>θ</mi> <mn>2</mn> </msub> </semantics></math> = <math display="inline"><semantics> <msup> <mn>120</mn> <mo>∘</mo> </msup> </semantics></math>, <math display="inline"><semantics> <msub> <mi>g</mi> <mn>1</mn> </msub> </semantics></math> = 0.16 mm, <math display="inline"><semantics> <msub> <mi>g</mi> <mn>2</mn> </msub> </semantics></math> = 0.5 mm, <math display="inline"><semantics> <msub> <mi>w</mi> <mn>1</mn> </msub> </semantics></math> = 0.2 mm, and <math display="inline"><semantics> <msub> <mi>w</mi> <mn>2</mn> </msub> </semantics></math> = 0.2 mm, based on the optimization results. The <span class="html-italic">S</span>-parameter of the (<b>c</b>) rhombus- and (<b>d</b>) hexagon-shaped unit cells is also provided.</p>
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<p>The coplanar (<b>a</b>) rhombus and (<b>b</b>) hexagon meta-surface-based SIW transmission lines compared with the traditional SIW structure, as well as the coplanar (<b>c</b>) rhombus–trapezoid and (<b>d</b>) hexagon–trapezoid meta-surface-based SIW leaky-wave antennas. The structural parameters are <math display="inline"><semantics> <msub> <mi>L</mi> <mn>1</mn> </msub> </semantics></math> = 50 mm, <span class="html-italic">W</span> = 22 mm, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>1</mn> </msub> </semantics></math> = 6 mm, <math display="inline"><semantics> <msub> <mi>d</mi> <mn>1</mn> </msub> </semantics></math> = 4 mm, <math display="inline"><semantics> <msub> <mi>w</mi> <mi>r</mi> </msub> </semantics></math> = 13 mm, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>2</mn> </msub> </semantics></math> = 5 mm, <math display="inline"><semantics> <msub> <mi>d</mi> <mn>2</mn> </msub> </semantics></math> = 2.5 mm, <math display="inline"><semantics> <msub> <mi>w</mi> <mi>h</mi> </msub> </semantics></math> = 13 mm, <math display="inline"><semantics> <msub> <mi>L</mi> <mn>2</mn> </msub> </semantics></math> = 200 mm, <span class="html-italic">m</span> = 9.4 mm, <span class="html-italic">p</span> = 8 mm, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>3</mn> </msub> </semantics></math> = 7 mm, <math display="inline"><semantics> <msub> <mi>d</mi> <mn>3</mn> </msub> </semantics></math> = 1.8 mm, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>4</mn> </msub> </semantics></math> = 5.7 mm, <math display="inline"><semantics> <msub> <mi>d</mi> <mn>4</mn> </msub> </semantics></math> = 1.7 mm, and <span class="html-italic">h</span> = 0.787 mm. For the trapezoid unit, the geometric sizes are set as: <math display="inline"><semantics> <msub> <mi>l</mi> <mn>3</mn> </msub> </semantics></math> = 2.9 mm, <math display="inline"><semantics> <msub> <mi>l</mi> <mn>4</mn> </msub> </semantics></math> = 3.9 mm, <math display="inline"><semantics> <msub> <mi>l</mi> <mn>5</mn> </msub> </semantics></math> = 4.9 mm, <math display="inline"><semantics> <msub> <mi>s</mi> <mn>1</mn> </msub> </semantics></math> = 0.9 mm, <math display="inline"><semantics> <msub> <mi>w</mi> <mn>3</mn> </msub> </semantics></math> = 0.2 mm, <math display="inline"><semantics> <msub> <mi>l</mi> <mn>6</mn> </msub> </semantics></math> = 3 mm, <math display="inline"><semantics> <msub> <mi>l</mi> <mn>7</mn> </msub> </semantics></math> = 4 mm, <math display="inline"><semantics> <msub> <mi>l</mi> <mn>8</mn> </msub> </semantics></math> = 5 mm, and <math display="inline"><semantics> <msub> <mi>s</mi> <mn>2</mn> </msub> </semantics></math> = 1 mm.</p>
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<p>The E-field distributions of the (<b>a</b>) rhombus and (<b>b</b>) hexagon meta-surface-based SIW transmission lines, as well as the corresponding <span class="html-italic">S</span>-parameters of the (<b>c</b>) rhombus and (<b>d</b>) hexagon meta-surface-based SIW transmission lines.</p>
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<p>The E-field of the (<b>a</b>) rhombus and (<b>b</b>) hexagon meta-surface-based SIW leaky-wave antennas, and the corresponding reflection coefficients of the (<b>c</b>) rhombus- and (<b>d</b>) hexagon-shaped SIW antenna designs with comparison groups. The 2D radiation patterns of the (<b>e</b>) rhombus and (<b>f</b>) hexagon meta-surface-based SIW leaky-wave antennas, and the corresponding maximum gains and 3D radiations of the (<b>g</b>) rhombus- and (<b>h</b>) hexagon-shaped designs.</p>
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<p>The coplanar (<b>a</b>) rhombus–hexagon and (<b>b</b>) hexagon–rhombus meta-surface-based SIW transmission lines, and the corresponding trapezoid-enabled meta-surface-based SIW leaky-wave antennas, as shown in (<b>c</b>,<b>d</b>). The geometric sizes are <math display="inline"><semantics> <msub> <mi>L</mi> <mn>1</mn> </msub> </semantics></math> = 50 mm, <span class="html-italic">W</span> = 22 mm, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>5</mn> </msub> </semantics></math> = 6 mm, <math display="inline"><semantics> <msub> <mi>d</mi> <mn>5</mn> </msub> </semantics></math> = 4 mm, <math display="inline"><semantics> <msub> <mi>w</mi> <mrow> <mi>r</mi> <mi>h</mi> </mrow> </msub> </semantics></math> = 13 mm, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>6</mn> </msub> </semantics></math> = 6 mm, <math display="inline"><semantics> <msub> <mi>d</mi> <mn>6</mn> </msub> </semantics></math> = 2.5 mm, <math display="inline"><semantics> <msub> <mi>w</mi> <mrow> <mi>h</mi> <mi>r</mi> </mrow> </msub> </semantics></math> = 13.1 mm, <math display="inline"><semantics> <msub> <mi>L</mi> <mn>2</mn> </msub> </semantics></math> = 200 mm, <span class="html-italic">m</span> = 9.4 mm, <span class="html-italic">p</span> = 8 mm, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>7</mn> </msub> </semantics></math> = 7 mm, <math display="inline"><semantics> <msub> <mi>d</mi> <mn>7</mn> </msub> </semantics></math> = 1.8 mm, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>8</mn> </msub> </semantics></math> = 5 mm and <math display="inline"><semantics> <msub> <mi>d</mi> <mn>8</mn> </msub> </semantics></math> = 2 mm. E-field distributions of the meta-surface-based SIW designs: (<b>e</b>) rhombus–hexagon and (<b>f</b>) hexagon–rhombus transmission lines and antennas. The corresponding reflection coefficients of the four SIW designs: (<b>g</b>) rhombus–hexagon and (<b>h</b>) hexagon–rhombus transmission lines and antennas. The far-field radiations of the meta-surface-based SIW leaky-wave antennas are demonstrated as follows: 2D patterns of (<b>i</b>) rhombus–hexagon and (<b>j</b>) hexagon–rhombus, and maximum gains of (<b>k</b>) rhombus–hexagon and (<b>l</b>) hexagon–rhombus with 3D patterns.</p>
Full article ">Figure 5 Cont.
<p>The coplanar (<b>a</b>) rhombus–hexagon and (<b>b</b>) hexagon–rhombus meta-surface-based SIW transmission lines, and the corresponding trapezoid-enabled meta-surface-based SIW leaky-wave antennas, as shown in (<b>c</b>,<b>d</b>). The geometric sizes are <math display="inline"><semantics> <msub> <mi>L</mi> <mn>1</mn> </msub> </semantics></math> = 50 mm, <span class="html-italic">W</span> = 22 mm, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>5</mn> </msub> </semantics></math> = 6 mm, <math display="inline"><semantics> <msub> <mi>d</mi> <mn>5</mn> </msub> </semantics></math> = 4 mm, <math display="inline"><semantics> <msub> <mi>w</mi> <mrow> <mi>r</mi> <mi>h</mi> </mrow> </msub> </semantics></math> = 13 mm, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>6</mn> </msub> </semantics></math> = 6 mm, <math display="inline"><semantics> <msub> <mi>d</mi> <mn>6</mn> </msub> </semantics></math> = 2.5 mm, <math display="inline"><semantics> <msub> <mi>w</mi> <mrow> <mi>h</mi> <mi>r</mi> </mrow> </msub> </semantics></math> = 13.1 mm, <math display="inline"><semantics> <msub> <mi>L</mi> <mn>2</mn> </msub> </semantics></math> = 200 mm, <span class="html-italic">m</span> = 9.4 mm, <span class="html-italic">p</span> = 8 mm, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>7</mn> </msub> </semantics></math> = 7 mm, <math display="inline"><semantics> <msub> <mi>d</mi> <mn>7</mn> </msub> </semantics></math> = 1.8 mm, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>8</mn> </msub> </semantics></math> = 5 mm and <math display="inline"><semantics> <msub> <mi>d</mi> <mn>8</mn> </msub> </semantics></math> = 2 mm. E-field distributions of the meta-surface-based SIW designs: (<b>e</b>) rhombus–hexagon and (<b>f</b>) hexagon–rhombus transmission lines and antennas. The corresponding reflection coefficients of the four SIW designs: (<b>g</b>) rhombus–hexagon and (<b>h</b>) hexagon–rhombus transmission lines and antennas. The far-field radiations of the meta-surface-based SIW leaky-wave antennas are demonstrated as follows: 2D patterns of (<b>i</b>) rhombus–hexagon and (<b>j</b>) hexagon–rhombus, and maximum gains of (<b>k</b>) rhombus–hexagon and (<b>l</b>) hexagon–rhombus with 3D patterns.</p>
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<p>Photos and experimental results of the manufactured meta-surface-based SIW leaky-wave antennas with comparison to the corresponding simulations. (<b>a</b>) Photos of the four SIW antennas. (<b>b</b>) The corresponding experiments in a microwave chamber. The measured radiation patterns of the (<b>c</b>) rhombus, (<b>d</b>) hexagon, (<b>e</b>) rhombus–hexagon and (<b>f</b>) hexagon–rhombus meta-surface-based SIW leaky-wave antennas, The measured reflection coefficients of the (<b>g</b>) rhombus and hexagon designs, and (<b>h</b>) rhombus–hexagon and hexagon–rhombus designs.</p>
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13 pages, 10638 KiB  
Communication
A Compact Four-Port MIMO Antenna for UWB Applications
by Aiting Wu, Mingyang Zhao, Pengquan Zhang and Zhonghai Zhang
Sensors 2022, 22(15), 5788; https://doi.org/10.3390/s22155788 - 3 Aug 2022
Cited by 17 | Viewed by 2725
Abstract
A compact four-port multiple-input multiple-output (MIMO) antenna for ultrawideband (UWB) applications is presented in this paper. The proposed antenna has four unit cell antennas. Each unit cell is placed orthogonal to its adjacent elements. The radiation element of each unit cell is composed [...] Read more.
A compact four-port multiple-input multiple-output (MIMO) antenna for ultrawideband (UWB) applications is presented in this paper. The proposed antenna has four unit cell antennas. Each unit cell is placed orthogonal to its adjacent elements. The radiation element of each unit cell is composed of a cut semicircular patch and a stepped microstrip feed line. The whole ground on the back side consists of four parts of defective ground and their extended branches, which are connected through a “卍” structure. The main decoupling technology used in the MIMO antenna is polarization diversity. In addition, protruded ground and parasitic elements are added to achieve a higher isolation. This compact antenna has a small area of 45 mm × 45 mm and is printed on a single layer substrate (FR4) with an εr = 4.4 and a thickness of 1.6 mm. This antenna has an impedance bandwidth (S11 < −10 dB) of 3.1–13.1 GHz (123%) and an isolation of less than −17 dB. The envelope correction coefficient (ECC) is less than 0.02 and the average gain is 4 dBi. The ultrawide bandwidth and compact size of the proposed antenna make it a promising candidate for UWB applications. Full article
(This article belongs to the Topic Antennas)
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Figure 1
<p>ant1 and ant2: (<b>a</b>) geometry; (<b>b</b>) S-parameters.</p>
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<p>ant1 and ant2: (<b>a</b>) geometry; (<b>b</b>) S-parameters.</p>
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<p>ant3: (<b>a</b>) geometry; (<b>b</b>) S-parameters.</p>
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<p>Surface current distribution at 6.5 GHz for ant3.</p>
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<p>ant4: (<b>a</b>) geometry; (<b>b</b>) S-parameters.</p>
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<p>ant5: (<b>a</b>) geometry; (<b>b</b>) S-parameters.</p>
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<p>The schematic diagram of the proposed antenna: (<b>a</b>) top; (<b>b</b>) bottom.</p>
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<p>Photograph of the fabricated antenna: (<b>a</b>) top; (<b>b</b>) bottom.</p>
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<p>Simulated and measured return loss (S11).</p>
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<p>Simulated and measured isolation (S21, S31).</p>
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<p>The surface current distribution at: (<b>a</b>) 3.5 GHz; (<b>b</b>) 7 GHz; (<b>c</b>) 10 GHz.</p>
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<p>The surface current distribution at: (<b>a</b>) 3.5 GHz; (<b>b</b>) 7 GHz; (<b>c</b>) 10 GHz.</p>
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<p>The ECC parameters between different ports.</p>
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<p>Diversity gain between different ports.</p>
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<p>Simulated and measured TARC.</p>
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<p>Radiation pattern at: (<b>a</b>) 3.5 GHz; (<b>b</b>) 7 GHz; (<b>c</b>) 10 GHz.</p>
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<p>Radiation pattern at: (<b>a</b>) 3.5 GHz; (<b>b</b>) 7 GHz; (<b>c</b>) 10 GHz.</p>
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<p>Antenna gain and radiation efficiency.</p>
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16 pages, 5442 KiB  
Article
Advanced Marine Predator Algorithm for Circular Antenna Array Pattern Synthesis
by Eunice Oluwabunmi Owoola, Kewen Xia, Samuel Ogunjo, Sandrine Mukase and Aadel Mohamed
Sensors 2022, 22(15), 5779; https://doi.org/10.3390/s22155779 - 2 Aug 2022
Cited by 12 | Viewed by 1849
Abstract
The pattern synthesis of antenna arrays is a substantial factor that can enhance the effectiveness and validity of a wireless communication system. This work proposes an advanced marine predator algorithm (AMPA) to synthesize the beam patterns of a non-uniform circular antenna array (CAA). [...] Read more.
The pattern synthesis of antenna arrays is a substantial factor that can enhance the effectiveness and validity of a wireless communication system. This work proposes an advanced marine predator algorithm (AMPA) to synthesize the beam patterns of a non-uniform circular antenna array (CAA). The AMPA utilizes an adaptive velocity update mechanism with a chaotic sequence parameter to improve the exploration and exploitation capability of the algorithm. The MPA structure is simplified and upgraded to overcome being stuck in the local optimum. The AMPA is employed for the joint optimization of amplitude current and inter-element spacing to suppress the peak sidelobe level (SLL) of 8-element, 10-element, 12-element, and 18-element CAAs, taking into consideration the mutual coupling effects. The results show that it attains better performances in relation to SLL suppression and convergence rate, in comparison with some other algorithms for the optimization case. Full article
(This article belongs to the Topic Antennas)
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<p>Asymmetric N isotropic circular antenna array (CAA).</p>
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<p>Selection of b1 and b2 for both 8–element and 16–element CAA.</p>
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<p>8−element CAA. (<b>a</b>) Radiation patterns obtained by different algorithms for reducing the PSLL. (<b>b</b>) Convergence rates of different algorithms for reducing the PSLL.</p>
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<p>10−element CAA. (<b>a</b>) Radiation patterns obtained by different algorithms for reducing the PSLL. (<b>b</b>) Convergence rates of different algorithms for reducing the PSLL.</p>
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<p>10−element CAA. (<b>a</b>) Radiation patterns obtained by different algorithms for reducing the PSLL. (<b>b</b>) Convergence rates of different algorithms for reducing the PSLL.</p>
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<p>3D radiation patterns of CAA. (<b>a</b>) 8−element CAA uniform array; (<b>b</b>) 8−element CAA for AMPA; (<b>c</b>) 10−element CAA uniform array; (<b>d</b>) 10−element CAA for AMPA.</p>
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<p>12−element CAA. (<b>a</b>) Radiation patterns obtained by different algorithms for reducing the PSLL. (<b>b</b>) Convergence rates of different algorithms for reducing the PSLL.</p>
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<p>12−element CAA. (<b>a</b>) Radiation patterns obtained by different algorithms for reducing the PSLL. (<b>b</b>) Convergence rates of different algorithms for reducing the PSLL.</p>
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<p>18−element CAA. (<b>a</b>) Radiation patterns obtained by different algorithms for reducing the PSLL. (<b>b</b>) Convergence rates of different algorithms for reducing the PSLL.</p>
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<p>3D radiation patterns of CAA. (<b>a</b>) 12−element CAA uniform array; (<b>b</b>) 12−element CAA for AMPA; (<b>c</b>) 18−element CAA uniform array; (<b>d</b>) 18−element CAA for AMPA.</p>
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<p>Bar graph for the computational time obtained by the algorithms for each CAA example.</p>
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19 pages, 10700 KiB  
Article
Portable Wideband Directional Antenna Scheme with Semicircular Corrugated Reflector for Digital Television Reception
by Bancha Luadang, Rerkchai Pukraksa, Pisit Janpangngern, Khanet Pookkapund, Sitthichai Dentri, Sompol Kosulvit and Chuwong Phongcharoenpanich
Sensors 2022, 22(14), 5338; https://doi.org/10.3390/s22145338 - 17 Jul 2022
Cited by 1 | Viewed by 1997
Abstract
This research proposed a portable wideband horizontally-polarized directional antenna scheme with a radome for digital terrestrial television reception. The operating frequency band of the proposed antenna scheme is 470–890 MHz. The portable antenna scheme was an adaptation of the Yagi-Uda antenna, consisting of [...] Read more.
This research proposed a portable wideband horizontally-polarized directional antenna scheme with a radome for digital terrestrial television reception. The operating frequency band of the proposed antenna scheme is 470–890 MHz. The portable antenna scheme was an adaptation of the Yagi-Uda antenna, consisting of a folded bowtie radiator, a semicircular corrugated reflector, and a V-shaped director. Simulations were carried out, and an antenna prototype was fabricated. To validate, experiments were undertaken to assess the antenna performance, including the impedance bandwidth (|S11| ≤ −10 dB), gain, and unidirectionality. The measured impedance bandwidth was 75.93%, covering 424–943 MHz, with a measured antenna gain of 2.69–4.84 dBi. The radiation pattern was of unidirectionality for the entire operating frequency band. The measured xz- and yz-plane half-power beamwidths were 150°, 159°, 160° and 102°, 78°, 102° at 470, 680, and 890 MHz, with the corresponding cross-polarization below −20 dB and −40 dB. The radome had a negligible impact on the impedance bandwidth, gain, and radiation pattern. The power obtained for the outdoor test, at 514 MHz, was 38.4 dBµV (−70.4 dBm) with a carrier-to-noise ratio (C/N) of 11.6 dB. In addition, the power obtained for the indoor test was 26.6 dBµV (−82.2 dBm) with a C/N of 10.9 dB. The novelty of this research lies in the concurrent use of the Yagi-Uda and bowtie antenna technologies to improve the impedance bandwidth and directionality of the antenna for digital terrestrial television reception. Full article
(This article belongs to the Topic Antennas)
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<p>Geometry of the portable wideband directional antenna: (<b>a</b>) top view, (<b>b</b>) perspective view, (<b>c</b>) side view, and (<b>d</b>) folded bowtie radiator.</p>
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<p>Geometry of the portable wideband directional antenna with radome: (<b>a</b>) top view, (<b>b</b>) side view.</p>
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<p>Conceptualization of the portable directional antenna scheme: (<b>a</b>) the Yagi-Uda antenna (model I), (<b>b</b>) the modified antenna with the U-shaped reflector and director (model II), and (<b>c</b>) the proposed portable antenna scheme (model III).</p>
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<p>Simulated impedance bandwidth (|<span class="html-italic">S</span><sub>11</sub>| ≤ −10 dB) of the antenna models I, II, and III.</p>
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<p>The simulated antenna gains of models I, II, and III.</p>
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<p>Simulated xz- and yz-plane radiation patterns of the antenna models I, II, and III.</p>
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<p>Evolutionary stages of the portable antenna scheme: (<b>a</b>) first generation, (<b>b</b>) second generation, and (<b>c</b>) third generation.</p>
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<p>Simulated impedance bandwidth (|<span class="html-italic">S</span><sub>11</sub>| ≤ −10 dB) of the first-, second-, and third-generation antenna schemes.</p>
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<p>Simulated antenna gains of the first-, second-, and third-generation antenna schemes.</p>
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<p>Simulated xz- and yz-plane radiation patterns of the first-, second-, and third-generation antenna schemes.</p>
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<p>Simulated surface current distribution of the proposed portable wideband directional antenna scheme at: (<b>a</b>) 470 MHz, (<b>b</b>) 680 MHz, and (<b>c</b>) 890 MHz.</p>
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<p>Simulated impedance bandwidth (|<span class="html-italic">S</span><sub>11</sub>| ≤ −10 dB) under variable parameters: (<b>a</b>) width of folded bowtie radiator (<span class="html-italic">W</span><sub>di</sub>), (<b>b</b>) angle of the folded bowtie radiator (<span class="html-italic">AG</span><sub>dp</sub>).</p>
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<p>Simulated impedance bandwidth under variable parameters: (<b>a</b>) distance from center of the supporting plate to the reflector (<span class="html-italic">D</span><sub>ref</sub>), (<b>b</b>) reflector length (<span class="html-italic">L</span><sub>ref</sub>), (<b>c</b>) reflector height (<span class="html-italic">h</span><sub>ref</sub>), and (<b>d</b>) radiation pattern at 890 MHz under variable <span class="html-italic">h</span><sub>ref</sub>.</p>
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<p>Simulated impedance bandwidth (|<span class="html-italic">S</span><sub>11</sub>| ≤ −10 dB) under variable parameters: (<b>a</b>) distance between the director and center of the supporting plate (<span class="html-italic">D</span><sub>di</sub>), (<b>b</b>) director’s arm length (<span class="html-italic">L</span><sub>di</sub>), and (<b>c</b>) angle of the director’s arm (<span class="html-italic">AG</span><sub>di</sub>).</p>
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<p>Simulated impedance bandwidth (|<span class="html-italic">S</span><sub>11</sub>| ≤ −10 dB) under variable parameters: (<b>a</b>) balun height (<span class="html-italic">H</span><sub>balun</sub>), (<b>b</b>) distance from center to center of the balun (<span class="html-italic">D</span><sub>balun</sub>).</p>
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<p>Prototype of the portable wideband directional antenna scheme: (<b>a</b>) without radome, (<b>b</b>) with radome.</p>
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<p>Simulated and measured impedance bandwidths of the portable wideband directional antenna scheme.</p>
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<p>Simulated and measured gains of the portable wideband directional antenna scheme.</p>
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<p>Simulated and measured xz- and yz-plane radiation patterns of the portable wideband directional antenna scheme: (<b>a</b>) 470 MHz, (<b>b</b>) 680 MHz, and (<b>c</b>) 890 MHz.</p>
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<p>Distance between the transmitting and receiving antennas without obstructions (line of sight).</p>
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<p>The portable wideband directional antenna scheme with a DVB signal receiver.</p>
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<p>The outdoor reception performance of the portable wideband directional antenna scheme.</p>
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<p>The indoor reception performance of the portable wideband directional antenna scheme.</p>
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<p>The experimental setup for the reception test in actual use.</p>
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13 pages, 3787 KiB  
Article
A Flower-Shaped Miniaturized UWB-MIMO Antenna with High Isolation
by Weidong Mu, Han Lin, Zhonggen Wang, Chenlu Li, Ming Yang, Wenyan Nie and Juan Wu
Electronics 2022, 11(14), 2190; https://doi.org/10.3390/electronics11142190 - 13 Jul 2022
Cited by 17 | Viewed by 2045
Abstract
An ultra-wideband (UWB) multiple-input, multiple-output (MIMO) antenna with a reasonably compact size of 30 × 18 × 1.6 mm3 is presented in this paper. The proposed antenna contains two radiating components, each of which is made up of three elliptically shaped patches [...] Read more.
An ultra-wideband (UWB) multiple-input, multiple-output (MIMO) antenna with a reasonably compact size of 30 × 18 × 1.6 mm3 is presented in this paper. The proposed antenna contains two radiating components, each of which is made up of three elliptically shaped patches situated 60 degrees apart, and resembles the shape of a flower. Moreover, the proposed antenna design incorporates a T-like ground branch that functions as a decoupling structure, and is composed of two modified inverted-L branches and an I-shaped stub, offering an isolation of more than 20 dB over the whole operation band (4.3–15.63 GHz). Furthermore, the proposed antenna system was fabricated and tested, and the envelope correlation coefficient (ECC), diversity gain (DG), and total active reflection coefficient (TARC), as well as the radiation characteristics and MIMO performance, were analyzed. The proposed UWB-MIMO antenna may be a suitable candidate for diverse UWB applications, based on the simulated and measured results of this study. Full article
(This article belongs to the Topic Antennas)
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<p>The proposed dual-port, flower-shaped UWB-MIMO antenna system structure.</p>
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<p>Fabricated prototype of the proposed dual-port, flower-shaped UWB-MIMO antenna: (<b>a</b>) front view, (<b>b</b>) back view.</p>
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<p>Evolution of the design process of UWB-MIMO system: (<b>a</b>) step 1, (<b>b</b>) step 2, (<b>c</b>) step 3, (<b>d</b>) step 4 (proposed MIMO system).</p>
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<p>Comparison of reflection coefficients of 4 proposed UWB-MIMO antenna designs.</p>
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<p>Comparison of transmission coefficients of 4 proposed UWB-MIMO antenna designs.</p>
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<p>Simulated S-parameters for tuning L<sub>g</sub>: (<b>a</b>) reflection coefficient, (<b>b</b>) transmission coefficient.</p>
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<p>Simulated S-parameters resulting from the tuning of L<sub>f</sub>: (<b>a</b>) reflection coefficient, (<b>b</b>) transmission coefficient.</p>
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<p>Surface current distribution when port 1 is stimulated at (<b>a</b>) 5.4 GHz, (<b>b</b>) 6 GHz, (<b>c</b>) 8 GHz, (<b>d</b>) 11.2 GHz, (<b>e</b>) 14.6 GHz.</p>
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<p>Simulated and measured S-parameters.</p>
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<p>Simulated and measured far-field patterns on XOZ and YOZ planes at (<b>a</b>) 5.4 GHz, (<b>b</b>) 6 GHz, (<b>c</b>) 8 GHz, (<b>d</b>) 11.2 GHz, and (<b>e</b>) 14.6 GHz.</p>
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<p>Simulated and measured far-field patterns on XOZ and YOZ planes at (<b>a</b>) 5.4 GHz, (<b>b</b>) 6 GHz, (<b>c</b>) 8 GHz, (<b>d</b>) 11.2 GHz, and (<b>e</b>) 14.6 GHz.</p>
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<p>Calculated radiation efficiency and peak gain.</p>
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<p>Simulated and measured ECC.</p>
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<p>Calculated DG.</p>
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<p>The comparison of simulated and measured TARC.</p>
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20 pages, 44792 KiB  
Article
Performance Analysis of Wearable Dual-Band Patch Antenna Based on EBG and SRR Surfaces
by Abdul Wajid, Ashfaq Ahmad, Sadiq Ullah, Dong-you Choi and Faiz Ul Islam
Sensors 2022, 22(14), 5208; https://doi.org/10.3390/s22145208 - 12 Jul 2022
Cited by 14 | Viewed by 3129
Abstract
This paper presents the performance comparison of a dual-band conventional antenna with a split-ring resonator (SRR)- and electromagnetic bandgap (EBG)-based dual-band design operating at 2.4 GHz and 5.4 GHz. The compactness and dual-frequency operation in the legacy Wi-Fi range of this design make [...] Read more.
This paper presents the performance comparison of a dual-band conventional antenna with a split-ring resonator (SRR)- and electromagnetic bandgap (EBG)-based dual-band design operating at 2.4 GHz and 5.4 GHz. The compactness and dual-frequency operation in the legacy Wi-Fi range of this design make it highly favorable for wearable sensor network-based Internet of Things (IoT) applications. Considering the current need for wearable antennas, wash cotton (with a relative permittivity of 1.51) is used as a substrate material for both conventional and metamaterial-based antennas. The radiation characteristics of the conventional antenna are compared with the EBG and SRR ground planes-based antennas in terms of return loss, gain, and efficiency. It is found that the SRR-based antenna is more efficient in terms of gain and surface wave suppression as well as more compact in comparison with its two counterparts. The compared results are found to be based on two distinct frequency ranges, namely, 2.4 GHz and 5.4 GHz. The suggested SRR-based antenna exhibits improved performance at 5.4 GHz, with gains of 7.39 dbi, bandwidths of 374 MHz, total efficiencies of 64.7%, and HPBWs of 43.2 degrees. The measurements made in bent condition are 6.22 db, 313 MHz, 52.45%, and 22.3 degrees, respectively. The three considered antennas (conventional, EBG-based, and SRR-based) are designed with a compact size to be well-suited for biomedical sensors, and specific absorption rate (SAR) analysis is performed to ensure user safety. In addition, the performance of the proposed antenna under bending conditions is also considered to present a realistic approach for a practical antenna design. Full article
(This article belongs to the Topic Antennas)
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<p>Geometrical representation of the patch antenna (conventional).</p>
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<p>Geometric model of the dual-band EBG.</p>
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<p>In-phase reflection of EBG unit cell.</p>
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<p>Simulated S11, S21 of EBG array.</p>
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<p>Geometric model of the dual-band SRR.</p>
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<p>In-phase reflection of SRR unit cell.</p>
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<p>Simulated S11, S21 of 5 × 5 SRR array.</p>
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<p>Geometric model of the dual-band EBG-based antenna.</p>
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<p>Geometric model (front view) of the dual-band SRR-based antenna.</p>
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<p>Geometric model of dual-band antennas in free space. (<b>a</b>) Conventional antenna. (<b>b</b>) EBG-based antenna. (<b>c</b>) SRR-based antenna.</p>
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<p>Geometric model of the dual-band antennas bent on human arm. (<b>a</b>) Conventional antenna. (<b>b</b>) EBG-based antenna. (<b>c</b>) SRR-based antenna.</p>
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<p>Reflection coefficient (S11) comparison of the conventional antenna with EBG- and SRR--based antennas.</p>
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<p>Gain patterns comparisons of conventional antenna with EBG- and SRR-based antennas under normal conditions. (<b>a</b>) E-plane at 2.4 GHz. (<b>b</b>) E-plane at 5.4 GHz. (<b>c</b>) H-plane at 2.4 GHz. (<b>d</b>) H-plane at 5.4 GHz.</p>
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<p>3D radiation pattern of normal scenario. (<b>a</b>) Conventional antenna at 2.4 GHz. (<b>b</b>) Conventional antenna at 5.4 GHz. (<b>c</b>) EBG-based antenna at 2.4 GHz. (<b>d</b>) EBG-based antenna at 5.4 GHz. (<b>e</b>) SRR-based antenna at 2.4 GHz. (<b>f</b>) SRR-based antenna at 5.4 GHz.</p>
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<p>S11 comparison of conventional antenna, antenna on EBG, and SRR/HIS in free space.</p>
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<p>Gain patterns comparisons of conventional antenna under bent condition (free space). (<b>a</b>) E-plane at 2.4 GHz. (<b>b</b>) E-plane at 5.4 GHz. (<b>c</b>) H-plane at 2.4 GHz. (<b>d</b>) H-plane at 5.4 GHz.</p>
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<p>3D radiation pattern of bent scenario (free space). (<b>a</b>) Conventional antenna at 2.4 GHz. (<b>b</b>) Conventional antenna at 5.4 GHz. (<b>c</b>) EBG-based antenna at 2.4 GHz. (<b>d</b>) EBG-based antenna at 5.4 GHz. (<b>e</b>) SRR-based antenna at 2.4 GHz. (<b>f</b>) SRR-based antenna at 5.4 GHz.</p>
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<p>Return loss comparison of conventional antenna, antenna on EBG, and SRR/HIS bent on human arm.</p>
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<p>Gain patterns comparisons of conventional antenna under bent condition (on-body). (<b>a</b>) E-plane at 2.4 GHz. (<b>b</b>) E-plane at 5.4 GHz. (<b>c</b>) H-plane at 2.4 GHz. (<b>d</b>) H-plane at 5.4 GHz.</p>
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<p>3D radiation patterns in bent scenarios (on-body). (<b>a</b>) Conventional antenna at 2.4 GHz. (<b>b</b>) Conventional antenna at 5.4 GHz. (<b>c</b>) EBG-based antenna at 2.4 GHz. (<b>d</b>) EBG-based antenna at 5.4 GHz. (<b>e</b>) SRR-based antenna at 2.4 GHz. (<b>f</b>) SRR-based antenna at 5.4 GHz.</p>
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<p>3D radiation patterns in bent scenarios (on-body). (<b>a</b>) Conventional antenna at 2.4 GHz. (<b>b</b>) Conventional antenna at 5.4 GHz. (<b>c</b>) EBG-based antenna at 2.4 GHz. (<b>d</b>) EBG-based antenna at 5.4 GHz. (<b>e</b>) SRR-based antenna at 2.4 GHz. (<b>f</b>) SRR-based antenna at 5.4 GHz.</p>
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<p>Conventional antenna bent around human arm: (<b>a</b>) SAR at 2.4 GHz. (<b>b</b>) SAR at 5.4 GHz, EBG-based antenna bent around human arm. (<b>c</b>) SAR at 2.4 GHz. (<b>d</b>) SAR at 5.4 GHz, SRR-based antenna bent around human arm. (<b>e</b>) SAR at 2.4 GHz. (<b>f</b>) SAR at 5.4 GHz.</p>
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10 pages, 72923 KiB  
Article
Dual-Band High-Gain Shared-Aperture Antenna Integrating Fabry-Perot and Reflectarray Mechanisms
by Xianjin Yi, Lin Zhou, Shuji Hao and Xing Chen
Electronics 2022, 11(13), 2017; https://doi.org/10.3390/electronics11132017 - 27 Jun 2022
Cited by 2 | Viewed by 2304
Abstract
This work presents a dual-band high-gain shared-aperture antenna. The proposed antenna integrates both the Fabry-Perot and reflectarray mechanisms; the antenna works as a Fabry-Perot cavity antenna (FPCA) in the S-band (2.45 GHz) and as a reflectarray antenna (RA) in the X-band [...] Read more.
This work presents a dual-band high-gain shared-aperture antenna. The proposed antenna integrates both the Fabry-Perot and reflectarray mechanisms; the antenna works as a Fabry-Perot cavity antenna (FPCA) in the S-band (2.45 GHz) and as a reflectarray antenna (RA) in the X-band (10 GHz). The antenna has a simple structure made up of only two printed circuit board layers. The bottom layer acts as a source antenna, a ground plane for the FPCA, and as a reflective surface for the RA. The upper layer contains the source antenna for the RA and serves as a partially reflective superstrate for the FPCA. The FPCA and RA thus share the same physical aperture but function independently. As an example, we design, fabricate, and characterize an antenna that operates at 2.45 and 10 GHz with an aperture size of 300 × 300 mm2. The measured results are found to be in good agreement with the simulations. We show that the proposed antenna achieves a gain of 16.21 dBi at 2.45 GHz and 21.6 dBi at 10 GHz with a −10 dB impedance bandwidths of 2.39–2.66 GHz and 9.40–10.28 GHz. The isolation between the two antenna ports is found to be larger than 30 dB. Full article
(This article belongs to the Topic Antennas)
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<p>Working principle of a (<b>a</b>) Fabry-Perot cavity antenna (FPCA), (<b>b</b>)reflectarray antenna (RA), and (<b>c</b>) shared-aperture antenna.</p>
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<p>Structure and simulated model of the superstrate unit cell.</p>
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<p>Simulated |<span class="html-italic">S</span><sub>11</sub>| and |<span class="html-italic">S</span><sub>21</sub>| of a superstrate unit cell.</p>
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<p>Structure and simulated model of the reflect surface unit cell.</p>
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<p>Simulated 10-GHz reflection phase distributions of a reflect surface unit cell.</p>
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<p>Simulated reflection amplitudes of a reflect surface unit cell with co-polarization and cross-polarization.</p>
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<p>Simulated cross-polarization reflection amplitude and phase of a reflect surface unit cell under different incident angles.</p>
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<p>Phase-shift distribution on the reflectarray surface.</p>
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<p>Photographs of (<b>a</b>) the fabricated superstrate, (<b>b</b>) reflect surface, and (<b>c</b>) antenna prototype.</p>
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<p>Simulated and measured <span class="html-italic">S</span> parameters.</p>
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<p>Simulated and measured gains of the proposed antenna at around 2.45 and 10 GHz.</p>
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<p>Radiation efficiency of the proposed antenna.</p>
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<p>Simulated and measured radiation patterns (dB) of (<b>a</b>) E plane and (<b>b</b>) H plane at 2.45 GHz.</p>
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<p>Simulated and measured radiation patterns (dB) of (<b>a</b>) E plane and (<b>b</b>) H plane at 10 GHz.</p>
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11 pages, 8661 KiB  
Article
On-Chip Polarization Reconfigurable Microstrip Patch Antennas Using Semiconductor Distributed Doped Areas (ScDDAs)
by Rozenn Allanic, Denis Le Berre, Cédric Quendo, Douglas Silva De Vasconcellos, Virginie Grimal, Damien Valente and Jérôme Billoué
Electronics 2022, 11(12), 1905; https://doi.org/10.3390/electronics11121905 - 17 Jun 2022
Cited by 1 | Viewed by 1675
Abstract
This paper presents two polarization reconfigurable patch antennas using semiconductor distributed doped areas (ScDDAs) as active components. One proposed antenna has a switching polarization between two linear ones, while the other one has a polarization able to commute from a linear to a [...] Read more.
This paper presents two polarization reconfigurable patch antennas using semiconductor distributed doped areas (ScDDAs) as active components. One proposed antenna has a switching polarization between two linear ones, while the other one has a polarization able to commute from a linear to a circular one. The antennas are designed on a silicon substrate in order to have the ScDDAs integrated in the substrate, overcoming the needs of classical PIN diodes. Therefore, the proposed co-design method between the antenna and the ScDDAs permits us to optimize the global reconfigurable function, designing both parts in the same process flow. Both demonstrators have a resonant frequency of around 5 GHz. The simulated results fit well with the measured ones. Full article
(This article belongs to the Topic Antennas)
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<p>The polarization reconfigurable antenna. (<b>a</b>) Top view. (<b>b</b>) Side view.</p>
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<p>The simulated reflection coefficient in both states of the two linear polarization antennae.</p>
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<p>A simulated EM field. (<b>a</b>) Horizontal Linear Polarization. (<b>b</b>) 45° Linear Polarization.</p>
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<p>Simulated radiation patterns of the realized gain. (<b>a</b>) In the OFF state at Phi = 0° and at Phi = 90°; (<b>b</b>) in the ON state at Phi = 45°.</p>
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<p>(<b>a</b>) The metallization mask. (<b>b</b>) The doping mask.</p>
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<p>A photograph of the first prototype.</p>
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<p>Measured results of the two linear polarization reconfigurable antenna in both states.</p>
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<p>(<b>a</b>) An antenna under measurement in the anechoic chamber. (<b>b</b>) A close-up of the positioner.</p>
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<p>(<b>a</b>) Measured radiation patterns of the realized gain. (<b>a</b>) In the OFF state at Phi = 0° and at Phi = 90°; (<b>b</b>) in the ON state at Phi = 45°.</p>
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<p>A top view of the circular to linear polarization switchable antenna design.</p>
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<p>S<sub>11</sub> simulated results of circular to linear polarization reconfigurable antennae in both states.</p>
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<p>Simulated radiation patterns of the realized gain. (<b>a</b>) In the OFF state in a circular polarization; (<b>b</b>) in the ON state in a linear polarization.</p>
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<p>A photograph of the second antenna.</p>
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<p>Measured results of the circular to linear polarization reconfigurable antenna.</p>
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<p>A comparison between simulated and measured results in both states.</p>
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<p>Measured results. (<b>a</b>) In the OFF state, in a circular polarization; (<b>b</b>) in the ON state, in a linear polarization.</p>
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14 pages, 17274 KiB  
Article
Electromechanical Coupling and Application of High-Frequency Communication Antenna Channel Capacity
by Yuefei Yan, Yan Wang, Baoqing Han, Xinlan Hu, Peiyuan Lian, Zhihai Wang, Kunpeng Yu, Meng Wang, Yang Wu, Guojun Leng and Congsi Wang
Electronics 2022, 11(12), 1857; https://doi.org/10.3390/electronics11121857 - 11 Jun 2022
Cited by 1 | Viewed by 1787
Abstract
The next-generation communication base station antennas represented by phased array antennas are towards high frequency, high gain, high density, and high pointing accuracy. The influence of mechanical structure factors on communication system channel quality is obviously increasing, and the electromechanical coupling problem is [...] Read more.
The next-generation communication base station antennas represented by phased array antennas are towards high frequency, high gain, high density, and high pointing accuracy. The influence of mechanical structure factors on communication system channel quality is obviously increasing, and the electromechanical coupling problem is becoming more prominent. To effectively guarantee the realization of 5G/6G communication in complex working environments and accelerate the commercial process of future communication systems, an electromechanical coupling channel capacity model is established in comprehensive consideration of the positional shift, attitude deflection, and temperature change of the communication base station phased array antennas. It can be used to rapidly evaluate the communication index degradation of RF devices within the heating environment. Moreover, a sensitivity model of the electric field strength and array antenna channel capacity to the random position error of each element is constructed. The influence of the random positioning error of each element on the communication indicators is analyzed and compared under different working conditions. The simulation results show that the proposed model can effectively provide a theoretical basis and guiding role for the design and manufacture of high-frequency array base station antennas. Full article
(This article belongs to the Topic Antennas)
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<p>A 5G/6G base station phased array antenna.</p>
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<p>Temperature field of base station array antenna: (<b>a</b>) Antenna element surface; (<b>b</b>) RF device surface.</p>
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<p>Thermal deformation of base station array antenna.</p>
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<p>Temperature field of base station array antenna: (<b>a</b>) Antenna element surface; (<b>b</b>) RF device surface.</p>
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<p>Sensitivity distribution in the <span class="html-italic">x</span> direction: (<b>a</b>) electric field strength of array factor; (<b>b</b>) channel capacity.</p>
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<p>Sensitivity distribution in the <span class="html-italic">y</span> direction: (<b>a</b>) electric field strength of array factor; (<b>b</b>) channel capacity.</p>
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<p>Sensitivity distribution in the <span class="html-italic">z</span> direction: (<b>a</b>) electric field strength of array factor; (<b>b</b>) channel capacity.</p>
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<p>Sensitivity distribution of random position error in <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <msup> <mn>0</mn> <mo>∘</mo> </msup> </mrow> </semantics></math> plane with scanning angle <math display="inline"><semantics> <mrow> <mi>θ</mi> <mo>=</mo> <msup> <mrow> <mn>15</mn> </mrow> <mo>∘</mo> </msup> </mrow> </semantics></math>: (<b>a</b>) <math display="inline"><semantics> <mi>x</mi> </semantics></math> direction; (<b>b</b>) <math display="inline"><semantics> <mi>y</mi> </semantics></math> direction; (<b>c</b>) <math display="inline"><semantics> <mi>z</mi> </semantics></math> direction.</p>
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<p>Sensitivity distribution of random position error in <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <msup> <mn>0</mn> <mo>∘</mo> </msup> </mrow> </semantics></math> plane with scanning angle <math display="inline"><semantics> <mrow> <mi>θ</mi> <mo>=</mo> <msup> <mrow> <mn>30</mn> </mrow> <mo>∘</mo> </msup> </mrow> </semantics></math>: (<b>a</b>) <math display="inline"><semantics> <mi>x</mi> </semantics></math> direction; (<b>b</b>) <math display="inline"><semantics> <mi>y</mi> </semantics></math> direction; (<b>c</b>) <math display="inline"><semantics> <mi>z</mi> </semantics></math> direction.</p>
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<p>Sensitivity distribution of random position error in <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <msup> <mrow> <mn>90</mn> </mrow> <mo>∘</mo> </msup> </mrow> </semantics></math> plane with scanning angle <math display="inline"><semantics> <mrow> <mi>θ</mi> <mo>=</mo> <msup> <mrow> <mn>15</mn> </mrow> <mo>∘</mo> </msup> </mrow> </semantics></math>: (<b>a</b>) <math display="inline"><semantics> <mi>x</mi> </semantics></math> direction; (<b>b</b>) <math display="inline"><semantics> <mi>y</mi> </semantics></math> direction; (<b>c</b>) <math display="inline"><semantics> <mi>z</mi> </semantics></math> direction.</p>
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<p>Sensitivity distribution of random position error in <math display="inline"><semantics> <mrow> <mi>ϕ</mi> <mo>=</mo> <msup> <mrow> <mn>90</mn> </mrow> <mo>∘</mo> </msup> </mrow> </semantics></math> plane with scanning angle <math display="inline"><semantics> <mrow> <mi>θ</mi> <mo>=</mo> <msup> <mrow> <mn>30</mn> </mrow> <mo>∘</mo> </msup> </mrow> </semantics></math>: (<b>a</b>) <math display="inline"><semantics> <mi>x</mi> </semantics></math> direction; (<b>b</b>) <math display="inline"><semantics> <mi>y</mi> </semantics></math> direction; (<b>c</b>) <math display="inline"><semantics> <mi>z</mi> </semantics></math> direction.</p>
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13 pages, 3974 KiB  
Article
Complementary Multi-Band Dual Polarization Conversion Metasurface and Its RCS Reduction Application
by Fengan Li and Baiqiang You
Electronics 2022, 11(10), 1645; https://doi.org/10.3390/electronics11101645 - 21 May 2022
Cited by 7 | Viewed by 2309
Abstract
In this paper, we present a metasurface composed of complementary units that can realize orthogonal linear and linear-to-circular polarization conversion in multi-band. Linear polarization conversion has seven high-conversion frequency bands: 9.1–9.7 GHz, 15.6–17.6 GHz, 19.4–19.7 GHz, 21.2–23.1 GHz, 23.5–23.8 GHz, 26.2 GHz, and [...] Read more.
In this paper, we present a metasurface composed of complementary units that can realize orthogonal linear and linear-to-circular polarization conversion in multi-band. Linear polarization conversion has seven high-conversion frequency bands: 9.1–9.7 GHz, 15.6–17.6 GHz, 19.4–19.7 GHz, 21.2–23.1 GHz, 23.5–23.8 GHz, 26.2 GHz, and 27.9 GHz. Linear-to-circular polarization conversion also has seven frequency bands with axial ratios (ARs) less than 3 dB: 8.9–9.0 GHz, 9.9–14.7 GHz, 19.1–19.3 GHz, 23.2–23.35 GHz, 23.4 GHz, 24.1–25.4 GHz, and 27.2–27.8 GHz, with the generation of multiple bands extended by the combination of complementary units. Then, we utilize the combined polarization conversion unit’s mirror placement to form a 4 × 4 array to realize the phase difference cancellation of the reflective field, giving the metasurface the radar cross section (RCS) reduction function and the dual-band 10-dB monostatic RCS reduction bandwidth: 8.9–9.7 GHz and 15.5–26.1 GHz. The measured and simulated results were essentially identical. Because the design uses the complementary units to form an array to expand the polarization conversion frequency bands, it provides a novel idea for future designs and can be applied to multiple microwave frequency bands. Full article
(This article belongs to the Topic Antennas)
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<p>Polarization conversion unit design. (<b>a</b>) Complementary unit. (<b>b</b>) Structure of unit 1. (<b>c</b>) Structure of unit 2.</p>
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<p>Simulation results of unit. (<b>a</b>) Reflection coefficient of unit 1. (<b>b</b>) Reflection coefficient of unit 2. (<b>c</b>) PCR for units 1 and 2. (<b>d</b>) AR for units 1 and 2.</p>
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<p>Simulation results of combined unit. (<b>a</b>) Reflection coefficient. (<b>b</b>) PCR. (<b>c</b>) Phase difference. (<b>d</b>) AR.</p>
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<p>Polarization conversion at oblique incidence: (<b>a</b>) PCR and (<b>b</b>) AR.</p>
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<p>Schematic diagram of PCM.</p>
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<p>RCS reduction metasurface-based PCM.</p>
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<p>(<b>a</b>) Monostatic RCS. (<b>b</b>) RCS reduction.</p>
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<p>3D scattering field of metal plane and RCS plane at 9.2 GHz and 16.5 GHz.</p>
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<p>Bistatic RCS reduction: (<b>a</b>) f = 9.2 GHz and (<b>b</b>) f = 16.5 GHz.</p>
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<p>Physical fabrication: (<b>a</b>) PCM and (<b>b</b>) RCS plane.</p>
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<p>Schematic diagram of measured device and environment.</p>
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<p>Comparison of measured and simulated results. (<b>a</b>) Measured and simulated reflection coefficients. (<b>b</b>) Measured and simulated RCS reduction.</p>
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13 pages, 5350 KiB  
Article
A Novel Meander Line Metamaterial Absorber Operating at 24 GHz and 28 GHz for the 5G Applications
by Syed Aftab Naqvi, Muhammad Abuzar Baqir, Grant Gourley, Adnan Iftikhar, Muhammad Saeed Khan and Dimitris E. Anagnostou
Sensors 2022, 22(10), 3764; https://doi.org/10.3390/s22103764 - 15 May 2022
Cited by 12 | Viewed by 2862
Abstract
Fifth generation (5G) communication systems deploy a massive MIMO technique to enhance gain and spatial multiplexing in arrays of 16 to 128 antennas. In these arrays, it is critical to isolate the adjacent antennas to prevent unwanted interaction between them. Fifth generation absorbers, [...] Read more.
Fifth generation (5G) communication systems deploy a massive MIMO technique to enhance gain and spatial multiplexing in arrays of 16 to 128 antennas. In these arrays, it is critical to isolate the adjacent antennas to prevent unwanted interaction between them. Fifth generation absorbers, in this regard, are the recent interest of many researchers nowadays. The authors present a dual-band novel metamaterial-based 5G absorber. The absorber operates at 24 GHz and 28 GHz and is composed of symmetric meander lines connected through a transmission line. An analytical model used to calculate the total number of required meander lines to design the absorber is delineated. The analytical model is based on the total inductance offered by the meander line structure in an impedance-matched electronic circuit. The proposed absorber works on the principal of resonance and absorbs two 5G bands (24 GHz and 28 GHz). A complete angular stability analysis was carried out prior to experiments for both transverse electric (TE) and transverse magnetic (TM) polarizations. Further, the resonance conditions are altered by changing the substrate thickness and incidence angle of the incident fields to demonstrate the functionality of the absorber. The comparison between simulated and measured results shows that such an absorber would be a strong candidate for the absorption in millimetre-wave array antennas, where elements are placed in proximity within compact 5G devices. Full article
(This article belongs to the Topic Antennas)
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<p>(<b>a</b>) Schematic representation of the finite absorber sheet consisting of the proposed meander-line-based 5G absorber. (<b>b</b>) Layout of the unit cell having optimized values as <span class="html-italic">l<sub>s</sub></span> = 8 mm, <span class="html-italic">l′</span> = 6 mm, <span class="html-italic">l<sub>w</sub></span> = 4.5 mm, <span class="html-italic">l</span> = 4 mm, <span class="html-italic">w</span> = 1 mm, <span class="html-italic">a</span> = 1.5 mm, <span class="html-italic">b</span> = 1 mm, <span class="html-italic">c</span> = 1 mm, <span class="html-italic">g</span> = 0.5 mm, and <span class="html-italic">s</span> = 1 mm.</p>
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<p>Absorption of the proposed MA keeping substrate thickness <span class="html-italic">t</span> = 0.8 mm (<b>a</b>) TE polarization and (<b>b</b>) TM polarization.</p>
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<p>Absorption of the proposed MA keeping substrate thickness <span class="html-italic">t</span> = 1.2 mm for (<b>a</b>) TE polarization (<b>b</b>) TM polarization.</p>
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<p>Absorption of the proposed PA keeping substrate thickness <span class="html-italic">t</span> = 1.6 mm for (<b>a</b>) TE polarization (<b>b</b>) TM polarization.</p>
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<p>Surface current density for 24.7 GHz (<b>a</b>) top metasurface and (<b>b</b>) bottom surface.</p>
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<p>Surface current density for 28.3 GHz (<b>a</b>) top metasurface (<b>b</b>) bottom surface.</p>
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<p>Effective impedance of the metamaterial absorber.</p>
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<p>Simulated electric field distribution for 24.5 GHz for (<b>a</b>) TE polarized wave and (<b>b</b>) TM polarized wave.</p>
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<p>A photograph of the fabricated prototype and experimental setup used for the measurement of the proposed MA.</p>
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<p>Performance comparison of the proposed MA for TE polarization at (<b>a</b>) 0°, (<b>b</b>) 10°, (<b>c</b>) 20° and (<b>d</b>) 30°.</p>
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<p>Performance comparison of the proposed MA for TM polarization at (<b>a</b>) 0°, (<b>b</b>) 10°, (<b>c</b>) 20° and (<b>d</b>) 30°.</p>
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15 pages, 2696 KiB  
Communication
Standing-Wave Feeding for High-Gain Linear Dielectric Resonator Antenna (DRA) Array
by Kerlos Atia Abdalmalak, Ayman Abdulhadi Althuwayb, Choon Sae Lee, Gabriel Santamaría Botello, Enderson Falcón-Gómez, Luis Emilio García-Castillo and Luis Enrique García-Muñoz
Sensors 2022, 22(8), 3089; https://doi.org/10.3390/s22083089 - 18 Apr 2022
Cited by 10 | Viewed by 3362
Abstract
A novel feeding method for linear DRA arrays is presented, illuminating the use of the power divider, transitions, and launchers, and keeping uniform excitation to array elements. This results in a high-gain DRA array with low losses with a design that is simple, [...] Read more.
A novel feeding method for linear DRA arrays is presented, illuminating the use of the power divider, transitions, and launchers, and keeping uniform excitation to array elements. This results in a high-gain DRA array with low losses with a design that is simple, compact and inexpensive. The proposed feeding method is based on exciting standing waves using discrete metallic patches in a simple design procedure. Two arrays with two and four DRA elements are presented as a proof of concept, which provide high gains of 12 and 15dBi, respectively, which are close to the theoretical limit based on array theory. The radiation efficiency for both arrays is about 93%, which is equal to the array element efficiency, confirming that the feeding method does not add losses as in the case of standard methods. To facilitate the fabrication process, the entire array structure is 3D-printed, which significantly decreases the complexity of fabrication and alignment. Compared to state-of-the-art feeding techniques, the proposed method provides higher gain and higher efficiency with a smaller electrical size. Full article
(This article belongs to the Topic Antennas)
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<p>Block diagram of DRA array based on the proposed feeding method.</p>
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<p>(<b>a</b>) 3D view of DRA Unit cell and (<b>b</b>) top view with transparent resonator.</p>
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<p>(<b>a</b>) Field distribution of <math display="inline"><semantics> <msub> <mi>TE</mi> <mn>113</mn> </msub> </semantics></math> mode at <math display="inline"><semantics> <mrow> <mn>3.9</mn> <mspace width="0.166667em"/> <mi>GHz</mi> </mrow> </semantics></math> and neighboring resonant modes inside DRA. (<b>b</b>) Simulated <math display="inline"><semantics> <msub> <mi mathvariant="normal">S</mi> <mn>11</mn> </msub> </semantics></math> and gain vs. frequency of antenna element.</p>
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<p>Snapshots of electric field distribution in proposed array demonstrating standing-wave excitation in the feed network.</p>
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<p>Simulated <math display="inline"><semantics> <msub> <mi mathvariant="normal">S</mi> <mn>11</mn> </msub> </semantics></math> and gain vs. frequency of the proposed two-element standing-wave DRA array.</p>
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<p>Schematic diagram of proposed four-element standing-wave DRA array: (<b>a</b>) 3D view and (<b>b</b>) top view with transparent resonators.</p>
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<p>Field distribution and radiation pattern of proposed four-element DRA array (<b>a</b>) without and (<b>b</b>) with shorting pins.</p>
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<p>Gain of four-element antenna vs. position of shorting pin along feed patch.</p>
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<p>Efficiency of proposed DRA array.</p>
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<p>(<b>a</b>) Theoretical array factor of four isotropic sources; (<b>b</b>) simulated element factor; and (<b>c</b>) simulated and estimated radiation pattern of proposed DRA array.</p>
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<p>(<b>a</b>) Manufactured prototype of four-element DRA array. (<b>b</b>) Measured and simulated <math display="inline"><semantics> <msub> <mi mathvariant="normal">S</mi> <mn>11</mn> </msub> </semantics></math> of manufactured four-element standing-wave DRA array prototype.</p>
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<p>Simulated and measured realized gain of four-element standing-wave DRA array.</p>
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<p>Simulated and measured radiation patterns of four-element standing-wave DRA array.</p>
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<p>Resimulated and measured: (<b>a</b>) realized gain; (<b>b</b>) radiation patterns after considering mismatching in dielectric characteristics.</p>
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14 pages, 5676 KiB  
Article
Design of a Low-Profile Wideband Magnetoelectric Dipole Antenna with Reduced Gain Drop
by Zhiyi Li, Xing Chen, Yuzhu Tang, Liangbing Liao, Linwan Deng and Zhifan Zhao
Electronics 2022, 11(7), 1156; https://doi.org/10.3390/electronics11071156 - 6 Apr 2022
Cited by 2 | Viewed by 2504
Abstract
In this paper, a novel low-profile magnetoelectric (ME) dipole antenna with wideband is presented. The conventional vertical fixing structure is bended four times from the center to the sides. The Γ-shaped feeding structure is bended two times to lower the height of the [...] Read more.
In this paper, a novel low-profile magnetoelectric (ME) dipole antenna with wideband is presented. The conventional vertical fixing structure is bended four times from the center to the sides. The Γ-shaped feeding structure is bended two times to lower the height of the antenna step by step. The effect of three kinds of vertical wall is discussed to show their influence on boresight gain. Through comparison, only one vertical wall is erected on the left side of the ground to decrease the boresight gain drop at 2.2 GHz. Both simulation and analysis are made to sufficiently explain the working principle. At last, the proposed ME dipole antenna has only 0.095λ00 is the center operating wavelength in free space) in height, and the wideband property is still maintained. By simulation, the relative bandwidth for VSWR < 2.0 is 47.9% (from 1.35 to 2.2 GHz). The boresight gain ranges from 8.1 to 9.6 dBi in the operating band. The measured relative bandwidth for VSWR < 2.0 is 50.3% (from 1.34 to 2.24 GHz), and the boresight gain ranges from 7.38 to 8.73 dBi. The gain drop on boresight is less than 1.4 dBi. Radiation patterns show a unidirectional characteristic in the whole operating band. Additionally, the cross-polarization level is less than −25 dB on boresight. The simulating and measuring results agree well with each other. Therefore, the proposed antenna is suitable for applications of limited height and wideband. Full article
(This article belongs to the Topic Antennas)
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<p>A depiction of the current low-profile techniques for an ME dipole antenna. (<b>a</b>) antenna 1; (<b>b</b>) antenna 2; (<b>c</b>) antenna 3; (<b>d</b>) antenna 4.</p>
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<p>The evolution of the proposed low-profile ME dipole.</p>
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<p>The structure of the proposed low-profile ME dipole. (<b>a</b>) Front view, (<b>b</b>) top view, (<b>c</b>) bended fixing structure, (<b>d</b>) feeding line.</p>
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<p>The structure of the proposed low-profile ME dipole. (<b>a</b>) Front view, (<b>b</b>) top view, (<b>c</b>) bended fixing structure, (<b>d</b>) feeding line.</p>
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<p>The schematic diagram of electric dipole and magnetic dipole.</p>
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<p>(<b>a</b>) The structure of ANT 1, ANT 2, and ANT3; (<b>b</b>) VSWR of ANT 1, ANT 2, and ANT3; (<b>c</b>) impedance of ANT 1, ANT 2, and ANT 3.</p>
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<p>(<b>a</b>) The structure of ANT 1, ANT 2, and ANT3; (<b>b</b>) VSWR of ANT 1, ANT 2, and ANT3; (<b>c</b>) impedance of ANT 1, ANT 2, and ANT 3.</p>
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<p>The structure of the antenna with (<b>a</b>) two symmetric vertical walls; (<b>b</b>) only flat ground plane; (<b>c</b>) one vertical wall; (<b>d</b>) the VSWR for different ground; (<b>e</b>) the radiating E field at 2.2 GHz for antenna 1; (<b>f</b>) the radiating E field at 2.2 GHz for the antenna with the single vertical wall on the left side; (<b>g</b>) the radiating E field at 2.2 GHz for antenna 3.</p>
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<p>The structure of the antenna with (<b>a</b>) two symmetric vertical walls; (<b>b</b>) only flat ground plane; (<b>c</b>) one vertical wall; (<b>d</b>) the VSWR for different ground; (<b>e</b>) the radiating E field at 2.2 GHz for antenna 1; (<b>f</b>) the radiating E field at 2.2 GHz for the antenna with the single vertical wall on the left side; (<b>g</b>) the radiating E field at 2.2 GHz for antenna 3.</p>
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<p>The distribution of electric current in one period. (<b>a</b>) at t = 0, (<b>b</b>) at t = T/4, (<b>c</b>) at t = T/2, (<b>d</b>) at t = 3T/4.</p>
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<p>The distribution of electric current in one period. (<b>a</b>) at t = 0, (<b>b</b>) at t = T/4, (<b>c</b>) at t = T/2, (<b>d</b>) at t = 3T/4.</p>
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<p>The parameter analysis for (<b>a</b>) W3, (<b>b</b>) L2, (<b>c</b>) a, and (<b>d</b>) b.</p>
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<p>The parameter analysis for (<b>a</b>) W3, (<b>b</b>) L2, (<b>c</b>) a, and (<b>d</b>) b.</p>
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<p>The measurement of proposed antenna.</p>
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<p>The comparison of simulated VSWR and boresight gain.</p>
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<p>The simulated and measured radiation patterns at (<b>a</b>) 1.4 GHz, (<b>b</b>) 1.8 GHz, (<b>c</b>) 2.2 GHz.</p>
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10 pages, 11431 KiB  
Communication
Wideband Filtering Slot Antenna Design with Stable Gain Using Characteristic Mode Analysis
by Chao Ni, Biyang Wen, Weijun Wu and Ping Ren
Sensors 2022, 22(7), 2780; https://doi.org/10.3390/s22072780 - 5 Apr 2022
Cited by 8 | Viewed by 2565
Abstract
A filtering slot antenna with a simple structure combination using characteristic mode analysis (CMA) is proposed. To realize filtering characteristics, characteristic magnetic currents of line and ring slots are analyzed and designed. Then, the folding-line slot and double-ring slot are selected to realize [...] Read more.
A filtering slot antenna with a simple structure combination using characteristic mode analysis (CMA) is proposed. To realize filtering characteristics, characteristic magnetic currents of line and ring slots are analyzed and designed. Then, the folding-line slot and double-ring slot are selected to realize radiation null separately and combined to construct the basic slot antenna. By properly exciting the selected characteristic modes, a wide filtering bandwidth and a stable gain are obtained. To validate the design process, a prototype antenna with a finite ground plane of about 1.1 λ × 1.1 λ is designed and fabricated. Simulated and measured results agree well, which both show a sharping roll rate in the lower and higher frequency and a flat gain realization in the pass band. The filtering bandwidth is 32.7%, the out-of-band suppression level at the higher frequency is over 20 dB, and the gain in the working frequency varies from 3.9 to 5.2 dB. Full article
(This article belongs to the Topic Antennas)
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<p>Antenna design stages. (<b>a</b>) antenna 1, (<b>b</b>) antenna 2, (<b>c</b>) antenna 3, (<b>d</b>) antenna 4.</p>
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<p>Geometry of the different line-slot antenna. (<b>a</b>) line slot antenna, (<b>b</b>) folding-line slot antenna.</p>
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<p>Modal significances of the folding-line slot antenna.</p>
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<p>Characteristic magnetic currents of the folding-slot antenna.</p>
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<p>Sing-ring/double-ring slot antenna with <span class="html-italic">R</span><sub>0</sub> = 32.3 mm, <span class="html-italic">W</span> = 3 mm, <span class="html-italic">R</span><sub>1</sub> = 28.3 mm, <span class="html-italic">G</span> = 1 mm, and corresponding characteristic magnetic current distribution of two typical CMs.</p>
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<p>Proposed filtering antenna. <span class="html-italic">R</span><sub>0</sub> = 21.8 mm, <span class="html-italic">R</span><sub>1</sub> = 17.5 mm, <span class="html-italic">W</span><sub>1</sub> = 3 mm, <span class="html-italic">W</span><sub>2</sub> = 2 mm, <span class="html-italic">W</span><sub>stub</sub> = 1.8 mm, <span class="html-italic">W</span><sub>3</sub> = 6 mm, <span class="html-italic">G</span><sub>stub</sub> = 1 mm, <span class="html-italic">G</span> = 1 mm, <span class="html-italic">W</span><sub>4</sub> = 3 mm, <span class="html-italic">θ</span><sub>1</sub> = 52°, <span class="html-italic">θ</span><sub>2</sub> = 10°, <span class="html-italic">θ</span><sub>3</sub> = 60°, <span class="html-italic">h</span> = 0.5 mm, <span class="html-italic">L<sub>f</sub></span> = 24 mm, <span class="html-italic">L<sub>f1</sub></span> = 8.2 mm, <span class="html-italic">W<sub>f</sub></span> = 0.92 mm, <span class="html-italic">W<sub>f1</sub></span> = 0.5 mm, <span class="html-italic">h</span> = 0.5 mm.</p>
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<p>Modal significances and modal weighting coefficients for the first five modes of the proposed antenna. (<b>a</b>) modal significances, (<b>b</b>) modal weighting coefficients.</p>
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<p>Magnetic eigencurrents of CMs.</p>
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<p>Simulated magnetic current distribution of the proposed antenna at different frequencies. (<b>a</b>) 2.08 GHz (<b>b</b>) 3.23 GHz.</p>
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<p>Simulated results for different parameter <span class="html-italic">θ</span><sub>2</sub>, (<b>a</b>) realized gain, (<b>b</b>) S11.</p>
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<p>Simulated results for different parameter <span class="html-italic">W</span><sub>4</sub>, (<b>a</b>) realized gain, (<b>b</b>) S11.</p>
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<p>Comparisons of the simulated/measured S11 and realized gain, prototype of the antenna.</p>
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<p>Simulated/measured radiation patterns of the proposed antenna, (<b>a</b>) xoz plane@2.5 GHz (<b>b</b>) yoz plane@2.5 GHz (<b>c</b>) xoz plane@3 GHz (<b>d</b>) yoz plane@3 GHz.</p>
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12 pages, 3935 KiB  
Communication
A Conformal Frequency Reconfigurable Antenna with Multiband and Wideband Characteristics
by Niamat Hussain, Adnan Ghaffar, Syeda Iffat Naqvi, Adnan Iftikhar, Dimitris E. Anagnostou and Huy H. Tran
Sensors 2022, 22(7), 2601; https://doi.org/10.3390/s22072601 - 29 Mar 2022
Cited by 28 | Viewed by 3467
Abstract
A compact flexible multi-frequency antenna for smart portable and flexible devices is presented. The antenna consists of a coplanar waveguide-fed slotted circular patch connected to a rectangular secondary resonator (stub). A thin low-loss substrate is used for flexibility, and a rectangular stub in [...] Read more.
A compact flexible multi-frequency antenna for smart portable and flexible devices is presented. The antenna consists of a coplanar waveguide-fed slotted circular patch connected to a rectangular secondary resonator (stub). A thin low-loss substrate is used for flexibility, and a rectangular stub in the feedline is deployed to attain wide operational bandwidth. A rectangular slot is etched in the middle of the circular patch, and a p-i-n diode is placed at its center. The frequency reconfigurability is achieved through switching the diode that distributes the current by changing the antenna’s electrical length. For the ON state, the antenna operates in the UWB region for −10 dB impedance bandwidth from 2.76 to 8.21 GHz. For the OFF state of the diode, the antenna operates at the ISM band (2.45/5.8 GHz), WLAN band (5.2 GHz), and lower X-band (8 GHz) with a minimum gain of 2.49 dBi and a maximum gain of 5.8 dBi at the 8 GHz band. Moreover, the antenna retains its performance in various bending conditions. The proposed antenna is suitable for modern miniaturized wireless electronic devices such as wearables, health monitoring sensors, mobile Internet devices, and laptops that operate at multiple frequency bands. Full article
(This article belongs to the Topic Antennas)
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<p>Schematic of proposed frequency reconfigurable antenna: (<b>a</b>) top-view, (<b>b</b>) bottom-view, and (<b>c</b>) side-view.</p>
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<p>Diode equivalent model: (<b>a</b>) ON-state, (<b>b</b>) OFF-state, and (<b>c</b>) biasing circuit.</p>
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<p>|S<sub>11</sub>| comparison among various steps included in the antenna design.</p>
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<p>Distribution of current density on the surface of antenna at (<b>a</b>) 3.75 GHz [diode-OFF], (<b>b</b>) 3.75 GHz [diode-ON], (<b>c</b>) 6.5 GHz [diode-OFF], and (<b>d</b>) 6.5 GHz [diode-ON]. Comparison among the various steps included in the antenna design.</p>
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<p>|S<sub>11</sub>| analysis for different (<b>a</b>) length of rectangular patch <span class="html-italic">Sy</span>, (<b>b</b>) width of rectangular patch <span class="html-italic">Sx</span>, and (<b>c</b>) width of slot <span class="html-italic">G</span><sub>3</sub>.</p>
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<p>Fabricated prototype: (<b>a</b>) top-view and (<b>b</b>) bottom view.</p>
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<p>Simulated and measured |S<sub>11</sub>|: (<b>a</b>) diode ON and (<b>b</b>) diode OFF.</p>
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<p>Antenna under conformal condition: (<b>a</b>) bending along the <span class="html-italic">x</span>-axis and (<b>b</b>) bending along the <span class="html-italic">y</span>-axis.</p>
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<p>Conformability analysis of the proposed antenna: (<b>a</b>) diode ON, (<b>b</b>) diode OFF.</p>
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<p>Far-field radiation pattern measurement setup.</p>
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<p>Radiation patterns of the proposed antenna for multiband mode: (<b>a</b>) 2.45 GHz, (<b>b</b>) 5.2 GHz, and (<b>c</b>) 8 GHz.</p>
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<p>The proposed antenna’s radiation patterns for UWB mode: (<b>a</b>) 3.2 GHz and (<b>b</b>) 5.8 GHz.</p>
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<p>(<b>a</b>) Gain and (<b>b</b>) radiation efficiency of the proposed antenna.</p>
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9 pages, 2708 KiB  
Communication
A Wideband Circularly Polarized Magnetoelectric Dipole Antenna for 5G Millimeter-Wave Communications
by Hussain Askari, Niamat Hussain, Md. Abu Sufian, Sang Min Lee and Nam Kim
Sensors 2022, 22(6), 2338; https://doi.org/10.3390/s22062338 - 17 Mar 2022
Cited by 22 | Viewed by 3698
Abstract
In this paper, a wideband circularly polarized (CP) magnetoelectric (ME) dipole antenna operating at 28 GHz band was proposed for 5G millimeter-wave (mm-wave) communications. The antenna geometry included two metallic plates with extended hook-shaped strips at its principal diagonal position, and two corners [...] Read more.
In this paper, a wideband circularly polarized (CP) magnetoelectric (ME) dipole antenna operating at 28 GHz band was proposed for 5G millimeter-wave (mm-wave) communications. The antenna geometry included two metallic plates with extended hook-shaped strips at its principal diagonal position, and two corners of truncated metallic plates at the secondary diagonal position. The pair of metallic vias connected the modified strips to the ground plane to create the magnetic dipole. The L-shaped probe feed between the strips was used to excite the antenna. The antenna showed stable gain and wideband characteristics. The simulated and measured results showed that the proposed CP ME dipole antenna had an overlapping (|S11|< −10 dB impedance and 3 dB axial ratio) bandwidth of 18.1% (25–30 GHz), covering the frequency bands dedicated for 5G new radio communications. Moreover, an average gain of 8 dBic was achieved by the antenna throughout the operating bandwidth. The measured data verified the design concept, and the proposed antenna had a small footprint of 0.83 λo × 0.83 λo × 0.125 λoo is free space wavelength at the lowest operating frequency), suitable for its application in 5G smart devices and sensors. Full article
(This article belongs to the Topic Antennas)
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<p>Global snapshot of the 5G millimeter-wave spectrum.</p>
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<p>The geometry of the proposed CP ME antenna: (<b>a</b>) 3D view, (<b>b</b>) top view, and (<b>c</b>) side view.</p>
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<p>Surface current distributions of the proposed CP ME dipole antenna for different time phases at 28 GHz.</p>
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<p>Photographs of the proposed CP ME dipole antenna: (<b>a</b>) fabricated prototype and (<b>b</b>) far field measurement setup.</p>
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<p>Proposed CP ME dipole antenna: (<b>a</b>) S-parameter |S<sub>11</sub>|and (<b>b</b>) axial ratio.</p>
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<p>Proposed CP ME dipole antennas: (<b>a</b>) broadside gain, (<b>b</b>) radiation efficiency and total efficiency.</p>
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<p>Radiation patterns of the proposed CP ME dipole antenna for different frequencies.</p>
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23 pages, 690 KiB  
Article
Hardware Efficient Massive MIMO Systems with Optimal Antenna Selection
by Shenko Chura Aredo, Yalemzewd Negash, Yihenew Wondie Marye, Hailu Belay Kassa, Kevin T. Kornegay and Feyisa Debo Diba
Sensors 2022, 22(5), 1743; https://doi.org/10.3390/s22051743 - 23 Feb 2022
Cited by 7 | Viewed by 2409
Abstract
An increase in the number of transmit antennas (M) poses an equivalent rise in the number of Radio Frequency (RF) chains associated with each antenna element, particularly in digital beamforming. The chain exhibits a substantial amount of power consumption accordingly. Hence, to alleviate [...] Read more.
An increase in the number of transmit antennas (M) poses an equivalent rise in the number of Radio Frequency (RF) chains associated with each antenna element, particularly in digital beamforming. The chain exhibits a substantial amount of power consumption accordingly. Hence, to alleviate such problems, one of the potential solutions is to reduce the number of RFs or to minimize their power consumption. In this paper, low-resolution Digital to Analogue Conversion (DAC) and transmit antenna selection at the downlink are evaluated to favour reducing the total power consumption and achieving energy efficiency in mMIMO with reasonable complexity. Antenna selection and low-resolution DAC techniques are proposed to leverage massive MIMO systems in free space and Close In (CI) path-loss models. The simulation results show that the power consumption decreases with antenna selection and low-resolution DAC. Then, the system achieves more energy efficiency than without low-resolution of DAC and full array utilization. Full article
(This article belongs to the Topic Antennas)
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<p>Digital beamforming URA configuration.</p>
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<p>Distortion versus number of symbols with <span class="html-italic">k</span> = 5 or 10 and <span class="html-italic">M</span> = 64.</p>
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<p>Quantization versus number of symbols with BPSK modulation.</p>
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<p>EE evaluation with different DAC power values, and evaluation of the total power consumption with and without low DAC resolution at randomly generated bits.</p>
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<p>Energy efficiency as a function of the number of BS antennas when the number of terminals <span class="html-italic">k</span> = 30.</p>
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<p>Energy efficiency versus spectral efficiency with without low resolution DAC.</p>
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<p>Energy efficiency as a function of <span class="html-italic">M</span> base station antennas with <math display="inline"><semantics> <mrow> <msub> <mi>P</mi> <mi>t</mi> </msub> <mo>=</mo> <mn>20</mn> </mrow> </semantics></math> mW.</p>
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<p>Energy efficiency evaluation as a function of the number of BS antennas with at mmWave frequency, <span class="html-italic">f</span> = 38 GHz and sub-6 GHz, and <span class="html-italic">f</span> = 2.5 GHz.</p>
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<p>Energy efficiency evaluation as a function of number of BS antennas with at mmWave frequency, <span class="html-italic">f</span> = 38 GHz, and <span class="html-italic">M</span> = 64.</p>
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<p>Energy efficiency evaluation with random and complex selection at <span class="html-italic">f</span> = 38 GHz and <span class="html-italic">M</span> = 100.</p>
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<p>Energy efficiency with and without low resolution DAC at <span class="html-italic">f</span> = 38 GHz and <span class="html-italic">M</span> = 100.</p>
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<p>Computationalx complexity of selection algorithms with adaptive <math display="inline"><semantics> <msup> <mo>ℓ</mo> <mo>⋆</mo> </msup> </semantics></math> and <span class="html-italic">M</span> = 64.</p>
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15 pages, 6604 KiB  
Article
A Study on Conformal Metasurface Influences on Passive Beam Steering
by Ruisi Ge, Ryan Striker and Benjamin Braaten
Electronics 2022, 11(5), 674; https://doi.org/10.3390/electronics11050674 - 22 Feb 2022
Cited by 2 | Viewed by 2373
Abstract
Beam-steering has drawn significant interest due to the expansion of network capacity. However, a traditional beam steering system involves active phase shifters and controlling networks which can be complex. This work studied the influence of passive conformal metasurface on conventional patch antenna. The [...] Read more.
Beam-steering has drawn significant interest due to the expansion of network capacity. However, a traditional beam steering system involves active phase shifters and controlling networks which can be complex. This work studied the influence of passive conformal metasurface on conventional patch antenna. The phase shifting was achieved by changing the curvature of a conformal metasurface. In addition, three low-cost conformal prototypes were fabricated and tested using different techniques such as 3D printing. The simulations and measurement results indicate up to 20° of beam shifting and reasonable gain increase. Compared with other research in the similar topic, the antenna system is completely passive, and the conformal metasurface is independent of the conventional patch antenna. Therefore, such study will be easy to implement with other antenna research especially for low power consumption beam steering systems. Full article
(This article belongs to the Topic Antennas)
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<p>Overview of the conformal metasurface.</p>
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<p>Conformal metasurface antenna proposal: (<b>a</b>) proposed conformal metasurface with radius of 40 mm. (<b>b</b>) proposed conformal metasurface with radius of 88 mm.</p>
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<p>Conformal metasurface antenna proposal: (<b>a</b>) proposed conformal metasurface with radius of 40 mm. (<b>b</b>) proposed conformal metasurface with radius of 88 mm.</p>
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<p>Patch Antenna layout: (<b>a</b>) Top view; (<b>b</b>) Side View.</p>
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<p>Conformal metasurface layout: (<b>a</b>) Top view; (<b>b</b>) Side View.</p>
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<p>Simulation result: comparison of different radii for Rogers 5870 conformal metasurface on patch antenna at 2.45 GHz. (<b>a</b>) r = 43 mm; (<b>b</b>) r = 50 mm; (<b>c</b>) r = 60 mm; (<b>d</b>) r = 75 mm; (<b>e</b>) r = 88 mm. (<b>f</b>) no conformal metasurface.</p>
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<p>Effect of different radii for Ninja Flex conformal metasurface on patch antenna at 2.45 GHz. (<b>a</b>) r = 43 mm; (<b>b</b>) r = 50 mm; (<b>c</b>) r = 60 mm; (<b>d</b>) r = 75 mm; (<b>e</b>) r = 88 mm. (<b>f</b>) no conformal metasurface.</p>
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<p>Effect of different radii for Panasonic Felios F775 conformal metasurface on patch antenna at 2.45 GHz. (<b>a</b>) r = 43 mm; (<b>b</b>) r = 50 mm; (<b>c</b>) r = 60 mm; (<b>d</b>) r = 75 mm; (<b>e</b>) r = 88 mm. (<b>f</b>) no conformal metasurface.</p>
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<p>Fabricated conformal metasurface: (<b>a</b>) Laser-direct engraving Rogers 5870; (<b>b</b>) 3D printed Ninja Flex; (<b>c</b>) The Panasoic Felios F775 flexible PCB.</p>
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<p>Measurement setup: (<b>a</b>) Conformal surface top view. (<b>b</b>) Conformal surface side view. (<b>c</b>) Measurement setup.</p>
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<p>Simulation radiation pattern of far field 0°: (<b>a</b>) Rogers 5870. (<b>b</b>) Ninja Flex 3D printing. (<b>c</b>) Panasoic Felios F775 flexible PCB.</p>
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<p>Simulation radiation pattern of far field in 90°: (<b>a</b>) Rogers 5870. (<b>b</b>) Ninja Flex 3D printing. (<b>c</b>) Panasoic Felios F775 flexible PCB.</p>
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<p>Measured radiation pattern polar plot in: (<b>a</b>) Rogers 5870. (<b>b</b>) Ninja Flex 3D printing. (<b>c</b>) Panasoic Felios F775 flexible PCB.</p>
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<p>Simulation and measurement results for Rogers 5870 conformal metasurface. (<b>a</b>) Simulation. (<b>b</b>) Measurement.</p>
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<p>Simulation and measurement results for Ninja Flex 3D printing conformal metasurface. (<b>a</b>) Simulation. (<b>b</b>) Measurement.</p>
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<p>Simulation and measurement results for Panasonic Felios F775 conformal metasurface: (<b>a</b>) Simulation. (<b>b</b>) Measurement.</p>
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12 pages, 18989 KiB  
Article
Research on Ultra-Low-Frequency Communication Based on the Rotating Shutter Antenna
by Faxiao Sun, Feng Zhang, Xiaoya Ma, Zhaoqian Gong, Yicai Ji and Guangyou Fang
Electronics 2022, 11(4), 596; https://doi.org/10.3390/electronics11040596 - 15 Feb 2022
Cited by 4 | Viewed by 2548
Abstract
This paper proposes a rotating shutter antenna that can directly generate 2FSK signals in the ULF band and it is expected to be used as the transmitter for magnetic induction (MI) underground communication systems. The antenna was modeled using ANSYS Maxwell and the [...] Read more.
This paper proposes a rotating shutter antenna that can directly generate 2FSK signals in the ULF band and it is expected to be used as the transmitter for magnetic induction (MI) underground communication systems. The antenna was modeled using ANSYS Maxwell and the magnetic field distribution was simulated. The results show that the interaction between the high-permeability shutter and the mutual cancellation of magnets decreased the transmitting magnetic moment of the antenna. A prototype antenna was manufactured and the time and frequency properties of the measured Bz field were the same as the simulated results, while the magnitude of the measured signal was larger. The propagation characteristics of the antenna in air–soil–rock were simulated using FEKO and the results show that the signal strength was greater than 1 fT at a depth of 450 m from the antenna whose magnetic moment as 1 Am2. The relationship between different magnetic components and azimuth could be used to enhance the signal strength. The formula of the Bz field was derived using the measured magnitude versus distance and the path loss was also analyzed. Finally, the 2FSK modulation property of the antenna was verified by indoor communication experiments with a code rate of 12.5 bps in the ULF band. Full article
(This article belongs to the Topic Antennas)
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<p>Block diagram of the rotating shutter antenna when <span class="html-italic">N</span> = 4.</p>
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<p>(<b>a</b>) Field at the point 3 m away from the shutter antenna. (<b>b</b>) Spectrogram of field.</p>
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<p>B-field distribution on the rotating shutter at different times: (<b>a</b>) t = 0 s; (<b>b</b>) t = 0.667 ms; (<b>c</b>) t = 1.67 ms.</p>
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<p>B-field distribution from different viewing angles: (<b>a</b>) side view; (<b>b</b>) top view.</p>
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<p>Change in magnetic flux density with the distance at different depths.</p>
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<p>Change in <math display="inline"><semantics> <msub> <mi>B</mi> <mi>r</mi> </msub> </semantics></math> and <math display="inline"><semantics> <msub> <mi>B</mi> <mi>φ</mi> </msub> </semantics></math> with azimuth at different depths: (<b>a</b>) depth = 0 m; (<b>b</b>) depth = 150 m; (<b>c</b>) depth = 300 m; (<b>d</b>) depth = 450 m.</p>
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<p>The rotating shutter antenna.</p>
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<p>ULF receiver: (<b>a</b>) magnetic sensor; (<b>b</b>) NI sampler.</p>
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<p>(<b>a</b>) Field at the point 3 m away from the shutter antenna. (<b>b</b>) Spectrogram of field.</p>
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<p>The change in <math display="inline"><semantics> <msub> <mi>B</mi> <mi>z</mi> </msub> </semantics></math>-field with distance.</p>
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<p>Path loss versus propagation range.</p>
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<p>(<b>a</b>) Modulated signals at the point 5.4 m away from the rotating shutter antenna; (<b>b</b>) spectrum of signals.</p>
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<p>(<b>a</b>) Spectrogram of 2FSK signal; (<b>b</b>) demodulated code.</p>
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12 pages, 3052 KiB  
Technical Note
Over-the-Air Testing of a Massive MIMO Antenna with a Full-Rank Channel Matrix
by Kazuhiro Honda
Sensors 2022, 22(3), 1240; https://doi.org/10.3390/s22031240 - 6 Feb 2022
Cited by 6 | Viewed by 2075
Abstract
This paper presents an over-the-air testing method in which a full-rank channel matrix is created for a massive multiple-input multiple-output (MIMO) antenna system utilizing a fading emulator with a small number of scatterers. In the proposed method, in order to mimic a fading [...] Read more.
This paper presents an over-the-air testing method in which a full-rank channel matrix is created for a massive multiple-input multiple-output (MIMO) antenna system utilizing a fading emulator with a small number of scatterers. In the proposed method, in order to mimic a fading emulator with a large number of scatterers, the scatterers are virtually positioned by rotating the massive MIMO antenna. The performance of a 64-element quasi-half-wavelength dipole circular array antenna was evaluated using a two-dimensional fading emulator. The experimental results reveal that a large number of available eigenvalues are obtained from the channel response matrix, confirming that the proposed method, which utilizes a full-rank channel matrix, can be used to assess a massive MIMO antenna system. Full article
(This article belongs to the Topic Antennas)
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<p>Configuration of the scatterers for evaluating the massive MIMO system.</p>
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<p>Massive MIMO OTA apparatus: (<b>a</b>) with the turn rail; (<b>b</b>) with the turntable.</p>
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<p>Massive MIMO OTA apparatus: (<b>a</b>) with the turn rail; (<b>b</b>) with the turntable.</p>
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<p>Number of channels vs. number of total scatterers.</p>
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<p>Average of the 64th eigenvalues.</p>
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<p>Configuration of the massive MIMO-OTA apparatus.</p>
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<p>CDF characteristics of the eigenvalues: (<b>a</b>) previous method; (<b>b</b>) proposed method.</p>
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<p>Eigenvalue distribution: (<b>a</b>) with the number of scatterer sets as a parameter; (<b>b</b>) with the rotation angle increment as a parameter.</p>
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<p>CDF characteristics of a system with 64 × 64 MIMO channel capacity.</p>
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11 pages, 2700 KiB  
Article
A Planar Four-Element UWB Antenna Array with Stripline Feeding Network
by Marek Garbaruk
Electronics 2022, 11(3), 469; https://doi.org/10.3390/electronics11030469 - 5 Feb 2022
Cited by 4 | Viewed by 2445
Abstract
This paper proposes a four-element ultrawideband (UWB) planar antenna array with elliptical-shaped radiators and a stripline excitation network designed for the 6–8.5 GHz UWB frequency band allowed in Europe by the European Commission. The designed antenna array has a symmetrical structure in which [...] Read more.
This paper proposes a four-element ultrawideband (UWB) planar antenna array with elliptical-shaped radiators and a stripline excitation network designed for the 6–8.5 GHz UWB frequency band allowed in Europe by the European Commission. The designed antenna array has a symmetrical structure in which the radiators are placed along one line in the central conducting layer, arranged between two layers of a dielectric. Radiating elements are fed by the stripline excitation network that provides uniform power distribution. The dimensions of the elliptical radiators’ axes are 14 mm × 16 mm. Two variants of array are proposed. The distance between the radiators’ centers is L = 19 mm for a shorter variant and L = 24 mm for a longer one. The presented antenna array structures have a size of 81 mm × 41 mm and 96 mm × 41 mm. These arrays present a measured gain of 6.4–10.6 dBi for the shorter variant and 8.5–10.8 dBi for the longer one and a fair impedance matching. The measured |S11| is less than −8.7 dB and −9.7 dB for the shorter and longer corresponding variants. Full article
(This article belongs to the Topic Antennas)
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<p>Geometry of proposed single prototype UWB antenna (dimensions in mm).</p>
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<p>Geometry of proposed four-element antenna arrays in <span class="html-italic">yz</span>-plane (dimensions in mm): (<b>a</b>) Shorter variant of UWB antenna array; (<b>b</b>) Longer variant of UWB antenna array.</p>
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<p>Photographs of proposed antennas: (<b>a</b>) Central metallic layer of shorter variant of four-element antenna array; (<b>b</b>) Central metallic layer of longer variant of four-element antenna array; (<b>c</b>) Fabricated antennas.</p>
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<p>Simulated and measured |<span class="html-italic">S</span><sub>11</sub>| of proposed antennas.</p>
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<p>Longer variant of four-element antenna array in anechoic chamber during measurements.</p>
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<p>Simulated and measured gain of proposed antennas in direction perpendicular to antenna surface (along +/− <span class="html-italic">x</span>-axis).</p>
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<p>Simulated and measured radiation pattern of the proposed arrays in <span class="html-italic">xy</span>-plane at different frequencies: (<b>a</b>) 6 GHz; (<b>b</b>) 7.2 GHz; (<b>c</b>) 8.4 GHz.</p>
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18 pages, 8738 KiB  
Article
Analysis and Design of an X-Band Reflectarray Antenna for Remote Sensing Satellite System
by Shimaa A. M. Soliman, Eman M. Eldesouki and Ahmed M. Attiya
Sensors 2022, 22(3), 1166; https://doi.org/10.3390/s22031166 - 3 Feb 2022
Cited by 13 | Viewed by 3517
Abstract
This paper presents the analysis and design of an X-band reflectarray. The proposed antenna can be used for a medium Earth orbit (MEO) remote sensing satellite system in the 8.5 GHz band. To obtain a nearly constant response along the coverage area of [...] Read more.
This paper presents the analysis and design of an X-band reflectarray. The proposed antenna can be used for a medium Earth orbit (MEO) remote sensing satellite system in the 8.5 GHz band. To obtain a nearly constant response along the coverage area of this satellite system, the proposed antenna was designed with a flat-top radiation pattern with a beam width of around 29° for the required MEO system. In addition, broadside pencil beam and tilted pencil beam reflectarrays were also investigated. The feeding element of the proposed reflectarray antennas is a Yagi–Uda array. The amplitude and phase distribution of the fields due to the feeding element on the aperture of the reflectarray antenna are obtained directly by numerical simulation without introducing any approximation. The required phase distribution along the aperture of the reflectarray to obtain the required flat-top radiation pattern is obtained using the genetic algorithm (GA) optimization method. The reflecting elements of the reflectarray are composed of stacked circular patches. This stacked configuration was found to be appropriate for obtaining a wide range of reflection phase shift, which is required to implement the required phase distribution on the reflectarray aperture. The antenna was fabricated and measured for verification. Full article
(This article belongs to the Topic Antennas)
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<p>A reflectarray antenna fed by a Yagi–Uda antenna.</p>
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<p>Proposed Yagi–Uda feeding antenna. (<b>a</b>) Geometry, (<b>b</b>) Simulated Reflection coefficient, (<b>c</b>) Simulated total gain pattern.</p>
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<p>Complex field distribution of the feeding element at the plane of the reflectarray: (<b>a</b>) 2D representation, (<b>b</b>) normalized amplitude and phase along the <math display="inline"><semantics> <mi>x</mi> </semantics></math>-axis.</p>
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<p>Phase distribution of the pencil beam reflectarray: (<b>a</b>) broadside beam, (<b>b</b>) tilted beam with a tilting angle of <math display="inline"><semantics> <mrow> <mn>15</mn> <mo>°</mo> </mrow> </semantics></math>.</p>
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<p>Geometry of the satellite coverage of the MEO satellite system.</p>
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<p>Required mask for the normalized radiation pattern.</p>
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<p>Flowchart of a general GA approach.</p>
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<p>Phase symmetry on half of the elements of the proposed reflectarray antenna.</p>
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<p>Optimized phase distribution of a flat-top radiation pattern.</p>
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<p>Flat-top radiation pattern obtained using a genetic algorithm.</p>
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<p>Geometry of the proposed unit cell.</p>
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<p>Reflection phase response of the unit cell.</p>
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<p>Simulation layout of the reflecting elements for the broadside pencil beam pattern.</p>
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<p>3D radiation pattern of the simulated broadside pencil beam reflectarray antenna.</p>
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<p>Simulation layout of the reflecting elements for the tilted beam pattern.</p>
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<p>3D radiation pattern of the simulated tilted beam reflectarray antenna.</p>
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<p>Reflecting elements for the flat-top pattern: (<b>a</b>) simulated layout, (<b>b</b>) fabricated upper layer, (<b>c</b>) fabricated bottom layer, and (<b>d</b>) fabricated ground plane.</p>
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<p>Measurement of the reflection coefficient of the fabricated Yagi–Uda antenna: (<b>a</b>) the fabricated Yagi–Uda antenna, (<b>b</b>) Yagi–Uda antenna connected to the Rhode and Schwartz model ZVA67 VNA, and (<b>c</b>) simulated and measured magnitude of the reflection coefficient.</p>
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<p>Fabricated prototype of the proposed reflectarray antenna.</p>
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<p>Fabricated antenna inside the anechoic chamber for the radiation pattern measurement.</p>
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<p>Measured and optimized radiation pattern of the proposed reflectarray antenna.</p>
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<p>Three-dimensional radiation pattern of the proposed reflectarray antenna.</p>
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21 pages, 3718 KiB  
Article
Optimal Field Sampling of Arc Sources via Asymptotic Study of the Radiation Operator
by Raffaele Moretta, Giovanni Leone, Fortuna Munno and Rocco Pierri
Electronics 2022, 11(2), 270; https://doi.org/10.3390/electronics11020270 - 14 Jan 2022
Cited by 1 | Viewed by 1385
Abstract
In this paper, the question of how to efficiently sample the field radiated by a circumference arc source is addressed. Classical sampling strategies require the acquisition of a redundant number of field measurements that can make the acquisition time prohibitive. For such reason, [...] Read more.
In this paper, the question of how to efficiently sample the field radiated by a circumference arc source is addressed. Classical sampling strategies require the acquisition of a redundant number of field measurements that can make the acquisition time prohibitive. For such reason, the paper aims at finding the minimum number of basis functions representing the radiated field with good accuracy and at providing an interpolation formula of the radiated field that exploits a non-redundant number of field samples. To achieve the first task, the number of relevant singular values of the radiation operator is computed by exploiting a weighted adjoint operator. In particular, the kernel of the related eigenvalue problem is first evaluated asymptotically; then, a warping transformation and a proper choice of the weight function are employed to recast such a kernel as a convolution and bandlimited function of sinc type. Finally, the number of significant singular values of the radiation operator is found by invoking the Slepian–Pollak results. The second task is achieved by exploiting a Shannon sampling expansion of the reduced field. The analysis is developed for both the far and the near fields radiated by a 2D scalar arc source observed on a circumference arc. Full article
(This article belongs to the Topic Antennas)
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<p>Geometry of the problem.</p>
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<p>Block diagram of the study.</p>
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<p>Diagram in degrees of the function <math display="inline"><semantics> <mi>C</mi> </semantics></math> in terms of <math display="inline"><semantics> <mrow> <mfrac> <mrow> <msub> <mi>r</mi> <mi>o</mi> </msub> </mrow> <mi>a</mi> </mfrac> </mrow> </semantics></math>.</p>
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<p>Comparison between the singular values of the radiation operator and those obtained by introducing the weighted adjoint. The diagram refers to the configuration <math display="inline"><semantics> <mrow> <mi>a</mi> <mo>=</mo> <mn>20</mn> <mi>λ</mi> <mo>,</mo> <mo> </mo> <mo> </mo> <msub> <mi>ϕ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>35</mn> <mo>°</mo> <mo>,</mo> <mo> </mo> <msub> <mi>θ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>50</mn> <mo>°</mo> </mrow> </semantics></math>.</p>
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<p>Optimal position of the far-field samples in the variable <math display="inline"><semantics> <mi>θ</mi> </semantics></math>. The diagram refers to the configuration <math display="inline"><semantics> <mrow> <mi>a</mi> <mo>=</mo> <mn>20</mn> <mi>λ</mi> <mo>,</mo> <mo> </mo> <msub> <mi>ϕ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>35</mn> <mo>°</mo> <mo>,</mo> <mo> </mo> <msub> <mi>θ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>50</mn> <mo>°</mo> </mrow> </semantics></math>.</p>
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<p>Comparison between the far field computed by the radiation model in (2) and the far field returned by the interpolation Formula (32). The diagram refers to the configuration <math display="inline"><semantics> <mrow> <mi>a</mi> <mo>=</mo> <mn>20</mn> <mi>λ</mi> <mo>,</mo> <mo> </mo> <msub> <mi>ϕ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>35</mn> <mo>°</mo> <mo>,</mo> <mo> </mo> <msub> <mi>θ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>50</mn> <mo>°</mo> </mrow> </semantics></math>.</p>
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<p>Comparison between the exact far-field, the far-field obtained by the interpolation of <math display="inline"><semantics> <mrow> <msub> <mi>N</mi> <mrow> <mi>S</mi> <mi>H</mi> </mrow> </msub> <mo>=</mo> <mn>35</mn> </mrow> </semantics></math> uniform samples and the far field obtained by the interpolation of <math display="inline"><semantics> <mrow> <msub> <mi>N</mi> <mrow> <mi>u</mi> <mi>p</mi> <mi>p</mi> <mi>e</mi> <mi>r</mi> </mrow> </msub> <mo>=</mo> <mn>71</mn> </mrow> </semantics></math> uniform samples. The diagram refers to the configuration <math display="inline"><semantics> <mrow> <mi>a</mi> <mo>=</mo> <mn>20</mn> <mi>λ</mi> <mo>,</mo> <mo> </mo> <msub> <mi>ϕ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>35</mn> <mo>°</mo> <mo>,</mo> <mo> </mo> <msub> <mi>θ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>50</mn> <mo>°</mo> </mrow> </semantics></math>.</p>
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<p>Comparison between the singular values of the radiation operator and those obtained by introducing the weighted adjoint. The diagrams refer to the configuration <math display="inline"><semantics> <mrow> <mi>a</mi> <mo>=</mo> <mn>20</mn> <mi>λ</mi> <mo>,</mo> <mo> </mo> <msub> <mi>ϕ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>25</mn> <mo>°</mo> <mo>,</mo> <mo> </mo> <msub> <mi>r</mi> <mi>o</mi> </msub> <mo>=</mo> <mn>40</mn> <mi>λ</mi> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <msub> <mi>θ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>35</mn> <mo>°</mo> </mrow> </semantics></math>.</p>
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<p>Optimal position of the far-field samples in the variable <math display="inline"><semantics> <mi>θ</mi> </semantics></math>. The diagram refers to the configuration <math display="inline"><semantics> <mrow> <mi>a</mi> <mo>=</mo> <mn>20</mn> <mi>λ</mi> <mo>,</mo> <mo> </mo> <msub> <mi>ϕ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>25</mn> <mo>°</mo> <mo>,</mo> <mo> </mo> <msub> <mi>θ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>35</mn> <mo>°</mo> </mrow> </semantics></math>.</p>
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<p>Comparison between the far-field computed by the radiation model in (3) and the far-field returned by the interpolation Formula (57). The diagram refers to the configuration <math display="inline"><semantics> <mrow> <mi>a</mi> <mo>=</mo> <mn>20</mn> <mi>λ</mi> <mo>,</mo> <mo> </mo> <msub> <mi>ϕ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>25</mn> <mo>°</mo> <mo>,</mo> <mo> </mo> <msub> <mi>r</mi> <mi>o</mi> </msub> <mo>=</mo> <mn>40</mn> <mi>λ</mi> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <msub> <mi>θ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>35</mn> <mo>°</mo> </mrow> </semantics></math>.</p>
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<p>Comparison between the exact near field, the near field obtained by the interpolation of <math display="inline"><semantics> <mrow> <msub> <mi>N</mi> <mrow> <mi>S</mi> <mi>H</mi> </mrow> </msub> <mo>=</mo> <mn>29</mn> </mrow> </semantics></math> uniform samples and the near field obtained by the interpolation of <math display="inline"><semantics> <mrow> <msub> <mi>N</mi> <mrow> <mi>u</mi> <mi>p</mi> <mi>p</mi> <mi>e</mi> <mi>r</mi> </mrow> </msub> <mo>=</mo> <mn>51</mn> </mrow> </semantics></math> uniform samples. The diagram refers to the configuration <math display="inline"><semantics> <mrow> <mi>a</mi> <mo>=</mo> <mn>20</mn> <mi>λ</mi> <mo>,</mo> <mo> </mo> <msub> <mi>ϕ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>25</mn> <mo>°</mo> <mo>,</mo> <mo> </mo> <msub> <mi>r</mi> <mi>o</mi> </msub> <mo>=</mo> <mn>40</mn> <mi>λ</mi> <mo>,</mo> <mo> </mo> <msub> <mi>θ</mi> <mrow> <mi>m</mi> <mi>a</mi> <mi>x</mi> </mrow> </msub> <mo>=</mo> <mn>35</mn> <mo>°</mo> </mrow> </semantics></math>.</p>
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17 pages, 7168 KiB  
Article
A Robust Design for Aperture-Level Simultaneous Transmit and Receive with Digital Phased Array
by Mingcong Xie, Xizhang Wei, Yanqun Tang and Dujuan Hu
Sensors 2022, 22(1), 109; https://doi.org/10.3390/s22010109 - 24 Dec 2021
Cited by 12 | Viewed by 2729
Abstract
Aperture-level simultaneous transmit and receive (ALSTAR) attempts to utilize adaptive digital transmit and receive beamforming and digital self-interference cancellation methods to establish isolation between the transmit and receive apertures of the single-phase array. However, the existing methods only discuss the isolation of ALSTAR [...] Read more.
Aperture-level simultaneous transmit and receive (ALSTAR) attempts to utilize adaptive digital transmit and receive beamforming and digital self-interference cancellation methods to establish isolation between the transmit and receive apertures of the single-phase array. However, the existing methods only discuss the isolation of ALSTAR and ignore the radiation efficiency of the transmitter and the sensitivity of the receiver. The ALSTAR array design lacks perfect theoretical support and simplified engineering implementation. This paper proposes an adaptive random group quantum brainstorming optimization (ARGQBSO) algorithm to simplify the array design and improve the overall performance. ARGQBSO is derived from BSO and has been ameliorated in four aspects of the ALSTAR array, including random grouping, initial value presets, dynamic probability functions, and quantum computing. The transmit and receive beamforming carried out by ARGQBSO is robust to all elevation angles, which reduces complexity and is conducive to engineering applications. The simulated results indicate that the ARGQBSO algorithm has an excellent performance, and achieves 166.8 dB of peak EII, 47.1 dBW of peak EIRP, and −94.6 dBm of peak EIS with 1000 W of transmit power in the scenario of an 8-element array. Full article
(This article belongs to the Topic Antennas)
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<p>Block diagram of the ALSTAR array cancellation architecture.</p>
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<p>Schematic diagram of random grouping.</p>
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<p>The flowchart of the ARGQBSO algorithm.</p>
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<p>Broadband antenna model: (<b>a</b>) 3D view; (<b>b</b>) gap structure; (<b>c</b>) feed-network; (<b>d</b>) metal reflector.</p>
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<p>Schematic diagram of the broadband digital phased array structure.</p>
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<p>(<b>a</b>) The port reflection parameters of the array. (<b>b</b>) Port isolation parameters of the array. (<b>c</b>) E-plane pattern of the first element. (<b>d</b>) H-plane pattern of the first element.</p>
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<p>(<b>a</b>) The port reflection parameters of the array. (<b>b</b>) Port isolation parameters of the array. (<b>c</b>) E-plane pattern of the first element. (<b>d</b>) H-plane pattern of the first element.</p>
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<p>(<b>a</b>) Comparison of the fitness value with/without preset initial value. (<b>b</b>) The influence curve of grouping method on algorithm running time. (<b>c</b>) Comparison of the fitness value of fixed probability and dynamic probability density function and quantum update.</p>
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<p>(<b>a</b>) Box plot of the three algorithms for EII, EIRP, EIS, and noise floor <math display="inline"><semantics> <mrow> <msub> <mi>P</mi> <mi>n</mi> </msub> </mrow> </semantics></math>. (<b>b</b>) Iterative curve and operation time of the four algorithms.</p>
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<p>(<b>a</b>) Box plot of the three algorithms for EII, EIRP, EIS, and noise floor <math display="inline"><semantics> <mrow> <msub> <mi>P</mi> <mi>n</mi> </msub> </mrow> </semantics></math>. (<b>b</b>) Iterative curve and operation time of the four algorithms.</p>
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<p>Transmit and receive beamforming vector.</p>
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<p>(<b>a</b>) EII with different transmit power. (<b>b</b>) EIRP with different transmit power. (<b>c</b>) EIS with different transmit power.</p>
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<p>(<b>a</b>) EII with different <math display="inline"><semantics> <mrow> <msub> <mi>w</mi> <mi>f</mi> </msub> </mrow> </semantics></math>. (<b>b</b>) EIRP with different <math display="inline"><semantics> <mrow> <msub> <mi>w</mi> <mi>f</mi> </msub> </mrow> </semantics></math>. (<b>c</b>) EIS with different <math display="inline"><semantics> <mrow> <msub> <mi>w</mi> <mi>f</mi> </msub> </mrow> </semantics></math>.</p>
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<p>(<b>a</b>) E-Pattern of the second element. (<b>b</b>) H-Pattern of the second element. (<b>c</b>) E-Pattern of the third element. (<b>d</b>) H-Pattern of the third element. (<b>e</b>) E-Pattern of the fourth element. (<b>f</b>) H-Pattern of the fourth element.</p>
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11 pages, 658 KiB  
Article
Floquet Spectral Almost-Periodic Modulation of Massive Finite and Infinite Strongly Coupled Arrays: Dense-Massive-MIMO, Intelligent-Surfaces, 5G, and 6G Applications
by Hamdi Bilel and Aguili Taoufik
Electronics 2022, 11(1), 36; https://doi.org/10.3390/electronics11010036 - 23 Dec 2021
Cited by 3 | Viewed by 3873
Abstract
In this study, we introduce a new formulation based on Floquet (Fourier) spectral analysis combined with a spectral modulation technique (and its spatial form) to study strongly coupled sublattices predefined in the infinite and large finite extent of almost-periodic antenna arrays (e.g., metasurfaces). [...] Read more.
In this study, we introduce a new formulation based on Floquet (Fourier) spectral analysis combined with a spectral modulation technique (and its spatial form) to study strongly coupled sublattices predefined in the infinite and large finite extent of almost-periodic antenna arrays (e.g., metasurfaces). This analysis is very relevant for dense-massive-MIMO, intelligent-surfaces, 5G, and 6G applications (used for very small areas with a large number of elements such as millimeter and terahertz waves applications). The numerical method that is adopted to model the structure is the method of moments simplified by equivalent circuits MoM GEC. Other numerical methods (such as the ASM-array scanning method and the windowing Fourier method) used this analysis in their kernel to treat periodic and pseudo-periodic (or quasi-periodic) arrays. Full article
(This article belongs to the Topic Antennas)
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<p>Construction of <math display="inline"><semantics> <mrow> <msub> <mi>f</mi> <mi>α</mi> </msub> <mrow> <mo>(</mo> <mi>x</mi> <mo>)</mo> </mrow> </mrow> </semantics></math> as given in Equations (<a href="#FD9-electronics-11-00036" class="html-disp-formula">9</a>)–(<a href="#FD11-electronics-11-00036" class="html-disp-formula">11</a>).</p>
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<p>Spectral representation of the interactions of a unit cell with its neighbors (infinite and finite cases) (valid for strong coupling interaction by using Floquet phases).</p>
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<p>Impedance variation against frequency band around 24 GHz: obtained by the MoM-GEC method.</p>
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<p>Impedance variation against frequency band around 77 GHz: obtained by the MoM GEC method.</p>
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<p><math display="inline"><semantics> <msub> <mi>S</mi> <mn>11</mn> </msub> </semantics></math>-parameter (dB) variation against frequency band around 24 GHz: a comparison between the MoM-GEC and a MATLAB tool (see references [<a href="#B41-electronics-11-00036" class="html-bibr">41</a>,<a href="#B42-electronics-11-00036" class="html-bibr">42</a>]).</p>
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<p>Variation of radiation pattern against Floquet states and application of superposition theorem for 10 elements of antenna array (uni-dimentionnal configuration) at 77 GHz: obtained by the MoM-GEC method.</p>
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<p>Variation of radiation pattern against Floquet states and application of superposition theorem for 100 elements of antenna array (uni-dimentionnal configuration) at 77 GHz: obtained by the MoM-GEC method.</p>
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