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Antenna Design and Sensors for Internet of Things

A special issue of Sensors (ISSN 1424-8220). This special issue belongs to the section "Physical Sensors".

Deadline for manuscript submissions: closed (31 October 2022) | Viewed by 39245
Please contact the Guest Editor or the Section Managing Editor at ([email protected]) for any queries.

Special Issue Editors


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Guest Editor
Department of Computer Science and Engineering, University of Quebec in Outaouais, Gatineau, QC J8X 3X7, Canada
Interests: antennas and propagation; radars; microwave circuits and systems; computational electrodynamics; FDTD method
Special Issues, Collections and Topics in MDPI journals

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Guest Editor
Laboratoire Électronique Ondes et Signaux pour les Transports (LEOST), University Gustave Eiffe, 59650 Villeneuve d’Ascq, France
Interests: miniature antennas; metamaterial antennas; metasurfaces; future railway communication systems; connected cars and cooperative ITS; communication systems and passive devices for vulnerable road users; electromagnetic compatibility of railway systems
Special Issues, Collections and Topics in MDPI journals

Special Issue Information

Dear Colleagues,

The emerging Internet of Things (IoT) paradigm is going to play an important role in modern and future communication systems. The main objective of IoT is to bring together people, data, processes, and things in order to fulfill the needs of human daily lives. It is a great opportunity for incorporating many different and heterogeneous systems in the development of a plethora of digital services for homes, health, agriculture, smart cities, factories, and transportation systems. Connecting all physical things in IoT will be done by wireless communication networks. The main challenges of communication systems for IoT include the need for reliable connectivity, the great number of frequency bands to support it, energy efficiency and cost. Thus, the antenna system will be a critical part of the smart devices. To make them intelligent, sensors will be added to IoT devices.

For each application, the choice of the antenna system presents a key design challenge.  Several factors need to be examined such as cost, antenna size, shape, and placement and other antenna performances (radiation efficiency, impedance matching, gain, etc.). IoT modules incorporate more and more wireless technologies. This makes the integration of antennas an increasingly significant challenge. Antenna designers face the constraints of maintaining reasonable performance in ever-shrinking footprints and under extreme interference conditions. The growth of Internet of Things and smart industrial applications creates many scientific and engineering challenges that call for innovative research efforts from both academia and industry in order to develop efficient, cost-effective, scalable, and reliable antenna systems for IoT. This Special Issue brings together researchers from academic and industrial domains to improve the field of IoT and its applications.

The main focus of this Special Issue is on the research challenges relating to the design and integration of antennas and sensors for Internet of Things.

Prof. Halim Boutayeb
Prof. Divitha Seetharamdoo
Guest Editors

Manuscript Submission Information

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Keywords

Potential topics include but are not limited to the following:

  • Antennas for IoT
  • Smart sensors
  • RFID
  • Miniaturized antennas
  • Millimeter wave antennas
  • Massive MIMO
  • Reconfigurable and smart antennas
  • Wearables antennas and devices
  • Antenna integration in complex media
  • Characteristic modes theory for antenna design
  • Base station
  • Handset antenna systems
  • Terminal antennas
  • Machine to machine communications
  • Driverless, connected vehicles
  • Autonomous sensors

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Related Special Issue

Published Papers (10 papers)

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Research

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14 pages, 2389 KiB  
Communication
A QoS-Adaptive Interference Alignment Technique for In-Band Full-Duplex Multi-Antenna Cellular Networks
by Ki-Hun Lee, Gyudong Park and Bang Chul Jung
Sensors 2022, 22(23), 9417; https://doi.org/10.3390/s22239417 - 2 Dec 2022
Cited by 1 | Viewed by 1847
Abstract
In this paper, we propose a novel interference alignment (IA) technique for an in-band full-duplex (IBFD) multiple-input multiple-output (MIMO) cellular network where a base station (BS) and user equipment (UE) are equipped with multiple antennas, and the local channel state information (CSI) is [...] Read more.
In this paper, we propose a novel interference alignment (IA) technique for an in-band full-duplex (IBFD) multiple-input multiple-output (MIMO) cellular network where a base station (BS) and user equipment (UE) are equipped with multiple antennas, and the local channel state information (CSI) is available at all nodes. Considering a practical IBFD MIMO cellular network, it is assumed that only the BS operates with full-duplex (FD) communication while UE operate in half-duplex (HD) mode. These IBFD networks introduce a new type of interference called cross-link interference (CLI), in which uplink UE affects downlink UE. The proposed IA technique consists of two symmetric IA schemes according to the number of antennas in the uplink and downlink UE, and both schemes effectively mitigate CLI in the IBFD MIMO network. It is worth noting that both IA schemes are adaptively applicable according to the network’s quality-of-service (QoS) requirements, such as uplink and downlink traffic demands. Furthermore, we theoretically characterize and prove the achievable sum-degrees-of-freedom (DoF) of the proposed IA technique. Simulation results show that the proposed IA technique significantly improves the sum rate performance compared to conventional HD communications (multi-user MIMO) while achieving the same achievable DoF as the interference-free IBFD MIMO network. Full article
(This article belongs to the Special Issue Antenna Design and Sensors for Internet of Things)
Show Figures

Figure 1

Figure 1
<p>System model of an in-band full-duplex (IBFD) MIMO cellular network.</p>
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<p>Geometric interpretation of the proposed IA schemes in IBFD MIMO networks.</p>
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<p>Sum rate performance of the proposed IA schemes, ideal IBFD MIMO (without CLI), and HD case (MU-MIMO) for uplink and downlink.</p>
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<p>Sum rate performance of the proposed IA schemes: <math display="inline"><semantics> <mrow> <mi>M</mi> <mo>=</mo> <mn>3</mn> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <msub> <mi>L</mi> <mi mathvariant="sans-serif">d</mi> </msub> <mo>=</mo> <mn>4</mn> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <msub> <mi>L</mi> <mi mathvariant="sans-serif">u</mi> </msub> <mo>=</mo> <mn>4</mn> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <msub> <mi>N</mi> <mi mathvariant="sans-serif">d</mi> </msub> <mo>=</mo> <mn>3</mn> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <msub> <mi>N</mi> <mi mathvariant="sans-serif">u</mi> </msub> <mo>=</mo> <mn>3</mn> </mrow> </semantics></math>.</p>
Full article ">
14 pages, 3226 KiB  
Article
Monopole Antenna with Enhanced Bandwidth and Stable Radiation Patterns Using Metasurface and Cross-Ground Structure
by Patrick Danuor, Kyei Anim and Young-Bae Jung
Sensors 2022, 22(21), 8571; https://doi.org/10.3390/s22218571 - 7 Nov 2022
Cited by 3 | Viewed by 4149
Abstract
In this paper, a printed monopole antenna with stable omnidirectional radiation patterns is presented for applications in ocean buoy and the marine Internet of Things (IoT). The antenna is composed of a rectangular patch, a cross-ground structure, and two frequency-selective surface (FSS) unit [...] Read more.
In this paper, a printed monopole antenna with stable omnidirectional radiation patterns is presented for applications in ocean buoy and the marine Internet of Things (IoT). The antenna is composed of a rectangular patch, a cross-ground structure, and two frequency-selective surface (FSS) unit cells. The cross-ground structure is incorporated into the antenna design to maintain consistent monopole-like radiation patterns over the antenna’s operating band, and the FSS unit cells are placed at the backside of the antenna to improve the antenna gain aiming at the L-band. In addition, the FSS unit cells exhibit resonance characteristics that, when incorporated with the cross-ground structure, result in a broader impedance bandwidth compared to the conventional monopole antenna. To validate the structure, a prototype is fabricated and measured. Good agreement between the simulated and measured results show that the proposed antenna exhibits an impedance bandwidth of 83.2% from 1.65 to 4 GHz, compared to the conventional printed monopole antenna. The proposed antenna realizes a peak gain of 4.57 dBi and a total efficiency of 97% at 1.8 GHz. Full article
(This article belongs to the Special Issue Antenna Design and Sensors for Internet of Things)
Show Figures

Figure 1

Figure 1
<p>Illustration of marine IoT for fishing gear automatic identification [<a href="#B10-sensors-22-08571" class="html-bibr">10</a>].</p>
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<p>Geometry of the conventional monopole antenna: (<b>a</b>) front view and (<b>b</b>) back view.</p>
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<p>Geometry of the conventional antenna incorporated with cross-ground structure: (<b>a</b>) front view, (<b>b</b>) back view, and (<b>c</b>) side view.</p>
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<p>Simulated results of the azimuth radiation patterns for (<b>a</b>) 1.7, (<b>b</b>) 1.8, (<b>c</b>) 1.86, and (<b>d</b>) 2 GHz.</p>
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<p>Simulated reflection coefficient of the conventional monopole only and with cross-ground structure.</p>
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<p>Geometry of proposed antenna with both cross-ground structure and FSS unit cells: (<b>a</b>) front view, (<b>b</b>) back view showing FSS unit cells, and (<b>c</b>) side view.</p>
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<p>Geometry of proposed FSS unit cell: (<b>a</b>) 2D view and (<b>b</b>) 3D view of simulation setup.</p>
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<p>Simulation results of (<b>a</b>) the reflection and transmission coefficient (S<sub>11</sub> and S<sub>21</sub>, respectively) of the proposed FSS unit cell and (<b>b</b>) gain results comparison after incorporation with FSS unit cells.</p>
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<p>Illustration of (<b>a</b>) FSS unit cell layer with inductive and capacitive components and (<b>b</b>) equivalent circuit model of proposed antenna structure with both cross-ground and FSS unit cells.</p>
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<p>Simulated results of the input reflection coefficient amplitude.</p>
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<p>Simulated results of the azimuth and elevation radiation patterns for (<b>a</b>) 1.7, (<b>b</b>) 1.8, (<b>c</b>) 1.86, and (<b>d</b>) 2 GHz.</p>
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<p>Photograph of the fabricated monopole antenna: (<b>a</b>) front view, (<b>b</b>) back view, and (<b>c</b>) far-field measurement set-up.</p>
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<p>Simulated and measured results of the azimuth and elevation radiation patterns for (<b>a</b>) 1.7 GHz, (<b>b</b>) 1.8 GHz, (<b>c</b>) 1.86 GHz, and (<b>d</b>) 2 GHz.</p>
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<p>Simulated and measured results of the input reflection coefficient amplitude.</p>
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<p>Results of the simulated and measured gain and antenna efficiency.</p>
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37 pages, 26894 KiB  
Article
A New Compact Triple-Band Triangular Patch Antenna for RF Energy Harvesting Applications in IoT Devices
by Chemseddine Benkalfate, Achour Ouslimani, Abed-Elhak Kasbari and Mohammed Feham
Sensors 2022, 22(20), 8009; https://doi.org/10.3390/s22208009 - 20 Oct 2022
Cited by 4 | Viewed by 2685
Abstract
This work proposes a new compact triple-band triangular patch antenna for RF energy harvesting applications in IoT devices. It is realized on Teflon glass substrate with a thickness of 0.67 mm and a relative permittivity of 2.1. Four versions of this antenna have [...] Read more.
This work proposes a new compact triple-band triangular patch antenna for RF energy harvesting applications in IoT devices. It is realized on Teflon glass substrate with a thickness of 0.67 mm and a relative permittivity of 2.1. Four versions of this antenna have been designed and realized with inclinations of 0°, 30°, 60° and 90° to study the impact of the tilting on their characteristics (S11 parameter, radiation pattern, gain) and to explore the possibilities of their implementation in the architectures of electronic equipment according to the available space. The antenna is also realized on waterproof paper with a thickness of 0.1 mm and a relative permittivity of 1.4 for biomedical domain. All the antennas (vertical antenna, tilted antennas and antenna realized on waterproof paper) have a size of 39 × 9 mm2 and cover the 2.45 GHz and 5.2 GHz Wi-Fi bands and the 8.2 GHz band. A good agreement is obtained between measured and simulated results. Radiation patterns show that all the antennas are omnidirectional for 2.45 GHz and pseudo-omnidirectional for 5.2 GHz and 8.2 GHz with maximum measured gains of 2.6 dBi, 4.55 dBi and 6 dBi, respectively. The maximum measured radiation efficiencies for the three antenna configurations are, respectively, of 75%, 70% and 72%. The Specific Absorption Rate (SAR) for the antenna bound on the human body is of 1.1 W/kg, 0.71 W/kg and 0.45 W/kg, respectively, for the three frequencies 2.45 GHz, 5.2 GHz and 8.2 GHz. All these antennas are then applied to realize RF energy harvesting systems. These systems are designed, realized and tested for the frequency 2.45 GHz, −20 dBm input power and 2 kΩ resistance load. The maximum measured output DC power is of 7.68 µW with a maximum RF-to-DC conversion efficiency of 77%. Full article
(This article belongs to the Special Issue Antenna Design and Sensors for Internet of Things)
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Figure 1

Figure 1
<p>Blocs schematic of RF energy harvesting system.</p>
Full article ">Figure 2
<p>Location of the RF-EH systems in the four examples (0°, 30°, 60° and 90°) of equipment architectures to be supplied.</p>
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<p>(<b>a</b>) Triangular, (<b>b</b>) rectangular and (<b>c</b>) circular antennas. <span class="html-fig-inline" id="sensors-22-08009-i001"><img alt="Sensors 22 08009 i001" src="/sensors/sensors-22-08009/article_deploy/html/images/sensors-22-08009-i001.png"/></span>: Bottom side.</p>
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<p>Simulated S<sub>11</sub> parameters and (electric, magnetic) fields intensities for 5 GHz.</p>
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<p>Two superposed resonators (<b>a</b>) Triangular and (<b>b</b>) circular antennas. <span class="html-fig-inline" id="sensors-22-08009-i002"><img alt="Sensors 22 08009 i002" src="/sensors/sensors-22-08009/article_deploy/html/images/sensors-22-08009-i002.png"/></span>: Bottom side.</p>
Full article ">Figure 6
<p>Simulated S<sub>11</sub> parameters and (electric, magnetic) fields intensities for 3.3 GHz.</p>
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<p>Triangular patch as resonant cavity.</p>
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<p>Proposed antenna shape. (<b>a</b>) Top side and (<b>b</b>) bottom side.</p>
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<p>Equivalent patch area of the proposed antenna. A<sub>i(i = 1, 2, 3)</sub> are the areas of each small triangular patch.</p>
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<p>The studied antenna shapes.</p>
Full article ">Figure 11
<p>Electrical equivalent circuit of the proposed antenna.</p>
Full article ">Figure 12
<p>Simulated S<sub>11</sub>-parameters of the three antennas; ∆f<sub>i(i = 1, 2, 3)</sub> are the frequency bandwidths.</p>
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<p>Simulated S<sub>11</sub> parameters of the antenna and electrical equivalent circuit on CST and ADS software, respectively.</p>
Full article ">Figure 14
<p>Comparison between E-field distribution of the antenna for (<b>a</b>) 2.45 GHz, (<b>b</b>) 5.2 GHz and (<b>c</b>) 8.2 GHz.</p>
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<p>Simulated 3D radiation pattern of the proposed antenna for 2.45 GHz, 5.2 GHz and 8.2 GHz frequencies.</p>
Full article ">Figure 16
<p>(<b>a</b>) Comparison between measured and simulated S<sub>11</sub> parameters, (<b>b</b>) representation of S<sub>11</sub> parameters on Smith chart, (<b>c</b>) realized antenna, (<b>d</b>) measurement prototype of S<sub>11</sub> parameters and (<b>e</b>) measurement prototype of radiation pattern.</p>
Full article ">Figure 17
<p>Measured 2D radiation pattern of the proposed antenna for 2.45 GHz, 5.2 GHz and 8.2 GHz frequencies.</p>
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<p>Different antenna tilting angles for RF-EH systems implementation.</p>
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<p>Simulated S<sub>11</sub> parameters of tilted antennas (30°, 60° and 90°).</p>
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<p>Antenna efficiency (<b>a</b>) and gain (<b>b</b>) as a function of frequency for the vertical antenna (0°) and the tilted antennas (30°, 60° and 90°).</p>
Full article ">Figure 21
<p>Simulated surface current density distribution for 2.45 GHz, 5.2 GHz and 8.2 GHz frequencies.</p>
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<p>Simulated 3D radiation patterns of the three tilted antennas for 2.45 GHz, 5.2 GHz and 8.2 GHz frequencies.</p>
Full article ">Figure 23
<p>(<b>a</b>) Measured S<sub>11</sub> parameters of the vertical antenna and the tilted antennas, (<b>b</b>) measurements prototype and (<b>c</b>) realized antennas.</p>
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<p>Measured and simulated radiation patterns (gain) of the 30° tilted antenna for the frequencies 2.45 GHz, 5.2 GHz and 8.2 GHz.</p>
Full article ">Figure 25
<p>Measured and simulated radiation patterns (gain) of the 60° tilted antenna for the frequencies 2.45 GHz, 5.2 GHz and 8.2 GHz.</p>
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<p>Measured and simulated radiation patterns (gain) of the 90° tilted antenna for the frequencies 2.45 GHz, 5.2 GHz and 8.2 GHz.</p>
Full article ">Figure 27
<p>Designed antenna on waterproof paper.</p>
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<p>Designed waterproof paper antenna on human body.</p>
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<p>Simulated S<sub>11</sub> parameters of the designed antenna on waterproof paper (undeformed and deformed cases) and bonded on human body.</p>
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<p>Simulated radiation patterns of the undeformed waterproof paper antenna, deformed one and of the bonded antenna on human body for the three resonant frequencies.</p>
Full article ">Figure 31
<p>(<b>a</b>) Radiation efficiency, (<b>b</b>) gain as function of frequency for the undeformed antenna, deformed one and antenna glued on the human body.</p>
Full article ">Figure 32
<p>(<b>a</b>) Measured S<sub>11</sub> parameters of the realized antenna on waterproof paper (undeformed, deformed and bonded on body cases), (<b>b</b>) measurement prototype and (<b>c</b>) realized antenna on waterproof paper (deformed).</p>
Full article ">Figure 33
<p>Measured radiation patterns of the undeformed waterproof paper antenna and deformed one for the three resonant frequencies 2.5 GHz, 5.2 GHz and 8.2 GHz.</p>
Full article ">Figure 34
<p>Specific absorption rate for 10 g of tissue and 100 mW input power at 2.5 GHz, 5.2 GHz and 8.2 GHz.</p>
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<p>Designed rectifier on ADS software.</p>
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<p>Representation of the exploited inner diode of the used nMOSFET transistor.</p>
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<p>(<b>a</b>) Transistor characterization on ADS software and (<b>b</b>) simulated and measured characteristics of the intrinsic diode of the used nMOS transistor.</p>
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<p>Designed matched rectifier for –20 dBm of input power, 2.45 GHz frequency and 2 kΩ resistance load.</p>
Full article ">Figure 39
<p>Equivalent capacitor and inductor circuit in microwave domain. L<sub>s</sub>, R<sub>s</sub>, R<sub>LS</sub> and C<sub>L</sub> are the series parasitic inductor and resistor of the selected capacitor, the parasitics resistor and capacitor of the selected inductor.</p>
Full article ">Figure 40
<p>Designed matched rectifier with parasitic elements of the impedance matching circuit for –20 dBm of input power, 2.45 GHz frequency and 2 kΩ resistance load.</p>
Full article ">Figure 41
<p>(<b>a</b>,<b>b</b>) Layout of the matched rectifiers on WP and Teflon glass substrates, respectively, with optimized dimensions, (<b>c</b>) simulated S<sub>11</sub> parameters of the matched rectifiers with and without parasitic elements, (<b>d</b>) co-simulated S<sub>11</sub> parameters and (<b>e</b>) simulated output DC voltages, all for −20 dBm input power, 2.45 GHz frequency and 2 kΩ resistance load.</p>
Full article ">Figure 42
<p>Measured S<sub>11</sub> parameters of the rectifier realized on (<b>a</b>) Teflon glass; (<b>b</b>) WP for input powers of −20 dBm, −10 dBm and 0 dBm and 2 kΩ resistance load; (<b>c</b>) realized rectifiers; and (<b>d</b>) measurement prototype.</p>
Full article ">Figure 43
<p>Realized RF energy harvesting systems on (<b>a</b>) Teflon glass, (<b>b</b>) WP substrate and (<b>c</b>,<b>d</b>) prototypes of measurement.</p>
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<p>Equivalent capacitance for each tilting angle.</p>
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<p>Simulation of the three tilted antennas taking into account the coupling capacitances.</p>
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<p>Simulated S<sub>11</sub> parameters of the tilted antennas (30°, 60° and 90°) with coupling capacitances.</p>
Full article ">
21 pages, 4510 KiB  
Article
Cost-Driven Design of Printed Wideband Antennas with Reduced Silver Ink Consumption for the Internet of Things
by Nicolas Claus, Jo Verhaevert and Hendrik Rogier
Sensors 2022, 22(20), 7929; https://doi.org/10.3390/s22207929 - 18 Oct 2022
Cited by 6 | Viewed by 1869
Abstract
The Internet of Things (IoT) accelerates the need for compact, lightweight and low-cost antennas combining wideband operation with a high integration potential. Although screen printing is excellently suited for manufacturing conformal antennas on a flexible substrate, its application is typically limited due to [...] Read more.
The Internet of Things (IoT) accelerates the need for compact, lightweight and low-cost antennas combining wideband operation with a high integration potential. Although screen printing is excellently suited for manufacturing conformal antennas on a flexible substrate, its application is typically limited due to the expensive nature of conductive inks. This paper investigates how the production cost of a flexible coplanar waveguide (CPW)-fed planar monopole antenna can be reduced by exploiting a mesh-based method for limiting ink consumption. Prototypes with mesh grids of different line widths and densities were screen-printed on a polyethylene terephthalate (PET) foil using silver-based nanoparticle ink. Smaller line widths decrease antenna gain and efficiency, while denser mesh grids better approximate unmeshed antenna behavior, albeit at the expense of greater ink consumption. A meshed prototype of 34.76×58.03mm with almost 80% ink reduction compared to an unmeshed counterpart is presented. It is capable of providing wideband coverage in the IMT/LTE-1/n1 (1.92–2.17 GHz), LTE-40/n40 (2.3–2.4 GHz), 2.45 GHz ISM (2.4–2.4835 GHz), IMT-E/LTE-7/n7 (2.5–2.69 GHz), and n78 5G (3.3–3.8 GHz) frequency bands. It exhibits a peak radiation efficiency above 90% and a metallized surface area of 2.46 cm2 (yielding an ink-to-total-surface ratio of 12.2%). Full article
(This article belongs to the Special Issue Antenna Design and Sensors for Internet of Things)
Show Figures

Figure 1

Figure 1
<p>(<b>a</b>) Test structures for the first (nonresonant) step in the characterization procedure: coplanar waveguide (CPW) transmission lines of length <math display="inline"><semantics> <mrow> <msub> <mi>l</mi> <mn>1</mn> </msub> <mo>=</mo> <mn>19</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math> and <math display="inline"><semantics> <mrow> <msub> <mi>l</mi> <mn>2</mn> </msub> <mo>=</mo> <mn>61</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>. (<b>b</b>) Test structures for the second (resonant) step in the characterization procedure: tapered coplanar resonators (CPRs) of length <math display="inline"><semantics> <mrow> <msub> <mi>L</mi> <mn>1</mn> </msub> <mo>=</mo> <mn>59.6</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math> and <math display="inline"><semantics> <mrow> <msub> <mi>L</mi> <mn>2</mn> </msub> <mo>=</mo> <mn>119.6</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>. All test structures are displayed on the same scale. The added insets provide more detail about the feed sections.</p>
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<p>Calculated (solid lines) and averaged (dashed lines) parameters for the polyethylene terephthalate (PET) substrate resulting from the first (nonresonant) characterization step: (<b>a</b>) effective permittivity and (<b>b</b>) loss tangent, as well as the real part of the complex propagation factor <math display="inline"><semantics> <mrow> <mi>γ</mi> <mo>=</mo> <mi>α</mi> <mo>+</mo> <mi>j</mi> <mi>β</mi> </mrow> </semantics></math>.</p>
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<p>Mapping between simulation and measurement for (<b>a</b>) the short resonator, with length <math display="inline"><semantics> <msub> <mi>L</mi> <mn>1</mn> </msub> </semantics></math>, and (<b>b</b>) the long resonator, with length <math display="inline"><semantics> <msub> <mi>L</mi> <mn>2</mn> </msub> </semantics></math>. The resulting values for the relative dielectric constant and loss tangent are <math display="inline"><semantics> <mrow> <msub> <mi>ε</mi> <mi>r</mi> </msub> <mo>=</mo> <mn>3.88</mn> </mrow> </semantics></math> and <math display="inline"><semantics> <mrow> <mo>tan</mo> <mo>(</mo> <mi>δ</mi> <mo>)</mo> <mo>=</mo> <mn>0.055</mn> </mrow> </semantics></math>, respectively.</p>
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<p>Theoretically calculated maximum radio frequency (RF) sheet resistance for the utilized ink layer.</p>
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<p>Topology of the unmeshed CPW-fed monopolar patch antenna. The inset shows more detail of the Hirose U.FL-R-SMT-1 U.FL connector footprint. The dimensions of Prototype I (P-I) are <math display="inline"><semantics> <mrow> <mi>W</mi> <mo>=</mo> <mn>36.10</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <mi>L</mi> <mo>=</mo> <mn>61.80</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <mi>w</mi> <mo>=</mo> <mn>23.50</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <mi>l</mi> <mo>=</mo> <mn>37.00</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <mi>d</mi> <mo>=</mo> <mn>2.80</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <msub> <mi>G</mi> <mi>w</mi> </msub> <mo>=</mo> <mn>16.75</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, and <math display="inline"><semantics> <mrow> <msub> <mi>G</mi> <mi>l</mi> </msub> <mo>=</mo> <mn>12.00</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>. The dimensions of the U.FL connector’s footprint are <math display="inline"><semantics> <mrow> <msub> <mi>F</mi> <mi>w</mi> </msub> <mo>=</mo> <mn>2.20</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <msub> <mi>f</mi> <mi>w</mi> </msub> <mo>=</mo> <mn>1.00</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <msub> <mi>P</mi> <mi>e</mi> </msub> <mo>=</mo> <mn>0.60</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <msub> <mi>P</mi> <mi>l</mi> </msub> <mo>=</mo> <mn>2.20</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <msub> <mi>P</mi> <mi>w</mi> </msub> <mo>=</mo> <mn>0.35</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <mi>s</mi> <mo>=</mo> <mn>0.45</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, and <math display="inline"><semantics> <mrow> <mi>g</mi> <mo>=</mo> <mn>0.20</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>.</p>
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<p>Topology of the meshed CPW-fed monopolar patch antenna, with a mesh grid (<math display="inline"><semantics> <mrow> <msub> <mi>N</mi> <mrow> <mi>p</mi> <mo>,</mo> <mi>x</mi> </mrow> </msub> <mo>×</mo> <msub> <mi>N</mi> <mrow> <mi>p</mi> <mo>,</mo> <mi>y</mi> </mrow> </msub> </mrow> </semantics></math>) and (<math display="inline"><semantics> <mrow> <msub> <mi>N</mi> <mrow> <mi>g</mi> <mo>,</mo> <mi>x</mi> </mrow> </msub> <mo>×</mo> <msub> <mi>N</mi> <mrow> <mi>g</mi> <mo>,</mo> <mi>y</mi> </mrow> </msub> </mrow> </semantics></math>) for the patch and planar monopole’s ground plane, respectively. The insets show more detailed dimensions of the mesh grid. Three prototypes were designed, denoted as Prototype II (P-II), Prototype III (P-III), and Prototype IV (P-IV). They share the same global dimensions: <math display="inline"><semantics> <mrow> <mi>W</mi> <mo>=</mo> <mn>34.76</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <mi>L</mi> <mo>=</mo> <mn>58.03</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <mi>w</mi> <mo>=</mo> <mn>23.28</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <mi>l</mi> <mo>=</mo> <mn>36.55</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <mi>d</mi> <mo>=</mo> <mn>2.98</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, <math display="inline"><semantics> <mrow> <msub> <mi>G</mi> <mi>w</mi> </msub> <mo>=</mo> <mn>16.08</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>, and <math display="inline"><semantics> <mrow> <msub> <mi>G</mi> <mi>l</mi> </msub> <mo>=</mo> <mn>8.50</mn> <mrow> <mi>mm</mi> </mrow> </mrow> </semantics></math>. The parameters of their mesh grids are different, as indicated in <a href="#sensors-22-07929-t001" class="html-table">Table 1</a>.</p>
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<p>Simulated (<b>a</b>) reflection coefficient <math display="inline"><semantics> <mrow> <mrow> <mo>|</mo> </mrow> <msub> <mi>S</mi> <mn>11</mn> </msub> <mrow> <mo>|</mo> </mrow> </mrow> </semantics></math> (in dB), (<b>b</b>) radiation efficiency (in %), and (<b>c</b>) peak gain (in dBi) for meshed antenna P-II with varying line width <span class="html-italic">t</span>.</p>
Full article ">Figure 8
<p>Photographs of the fabricated meshed antenna prototypes with different (<math display="inline"><semantics> <mrow> <msub> <mi>N</mi> <mrow> <mi>p</mi> <mo>,</mo> <mi>x</mi> </mrow> </msub> <mo>×</mo> <msub> <mi>N</mi> <mrow> <mi>p</mi> <mo>,</mo> <mi>y</mi> </mrow> </msub> </mrow> </semantics></math>) mesh grids: (<b>a</b>) P-II, with a (10 × 16) grid; (<b>b</b>) P-III, with a (16 × 20) grid; and (<b>c</b>) P-IV, with a (20 × 32) grid. As the antennas are largely transparent, they are shown with white styrofoam in the background.</p>
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<p>Simulated (solid lines) and measured (dashed lines) reflection coefficient <math display="inline"><semantics> <mrow> <mrow> <mo>|</mo> </mrow> <msub> <mi>S</mi> <mn>11</mn> </msub> <mrow> <mo>|</mo> </mrow> </mrow> </semantics></math> (in dB) for the different prototypes P-I, P-II, P-III, and P-IV (<b>a</b>) in the frequency range from 0 to 6 GHz and (<b>b</b>) zoomed in to the frequency range from 1.5 to <math display="inline"><semantics> <mrow> <mn>4.5</mn> </mrow> </semantics></math> GHz. The relevant frequency bands, namely, IMT/LTE-1/n1 (1.92–2.17 GHz), LTE-40/n40 (2.3–2.4 GHz), 2.45 GHz ISM (2.4–2.4835 GHz), IMT-E/LTE-7/n7 (2.5–2.69 GHz), and n78 5G (3.3–3.8 GHz), are indicated in blue, orange, yellow, purple, and green, respectively.</p>
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<p>Microscopic images of the CPW feed section with nominal gap width <math display="inline"><semantics> <mrow> <mi>g</mi> <mo>=</mo> <mn>200</mn> <mo> </mo> <mrow> <mo mathvariant="sans-serif">μ</mo> <mi mathvariant="normal">m</mi> </mrow> </mrow> </semantics></math> (see <a href="#sensors-22-07929-f005" class="html-fig">Figure 5</a>) for (<b>a</b>) P-I and (<b>b</b>) P-II; (<b>c</b>) mesh line with nominal width <math display="inline"><semantics> <mrow> <mi>t</mi> <mo>=</mo> <mn>100</mn> <mo> </mo> <mrow> <mo mathvariant="sans-serif">μ</mo> <mi mathvariant="normal">m</mi> </mrow> </mrow> </semantics></math> (see <a href="#sensors-22-07929-f006" class="html-fig">Figure 6</a>) for P-II.</p>
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<p>Simulated surface currents at <math display="inline"><semantics> <mrow> <mn>2.45</mn> </mrow> </semantics></math> GHz for prototypes (<b>a</b>) P-I, (<b>b</b>) P-II, (<b>c</b>) P-III, and (<b>d</b>) P-IV.</p>
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<p>Simulated (solid lines) and measured (dashed lines) radiation pattern for P-I at (<b>a</b>) <math display="inline"><semantics> <mrow> <mn>2.05</mn> </mrow> </semantics></math> GHz, (<b>b</b>) <math display="inline"><semantics> <mrow> <mn>2.45</mn> </mrow> </semantics></math> GHz, and (<b>c</b>) <math display="inline"><semantics> <mrow> <mn>3.55</mn> </mrow> </semantics></math> GHz. Left: XZ-plane; right: YZ-plane. To indicate the unreliability of the measured results around 180° due to the impact of the measurement setup, this region is shaded in gray.</p>
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<p>Radiation parameters for P-I: (<b>a</b>) simulated (solid lines) and measured (dashed lines) peak gain and (<b>b</b>) simulated radiation efficiency. The measured peak gain was corrected by characterizing the magnitude of the radiation pattern’s ripple through simulation of the non-idealities in the measurement setup.</p>
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<p>Simulated (solid lines) and measured (dashed lines) radiation pattern for P-IV at (<b>a</b>) <math display="inline"><semantics> <mrow> <mn>2.05</mn> </mrow> </semantics></math> GHz, (<b>b</b>) <math display="inline"><semantics> <mrow> <mn>2.45</mn> </mrow> </semantics></math> GHz, and (<b>c</b>) <math display="inline"><semantics> <mrow> <mn>3.55</mn> </mrow> </semantics></math> GHz. Left: XZ-plane; right: YZ-plane. To indicate the unreliability of the measured results around 180° due to the impact of the measurement setup, this region is shaded in gray.</p>
Full article ">Figure 15
<p>Radiation parameters for P-IV: (<b>a</b>) simulated (solid lines) and measured (dashed lines) peak gain and (<b>b</b>) simulated radiation efficiency. The measured peak gain was corrected by characterizing the magnitude of the radiation pattern’s ripple through simulation of the non-idealities in the measurement setup.</p>
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18 pages, 7606 KiB  
Article
Compact Quad Band MIMO Antenna Design with Enhanced Gain for Wireless Communications
by Sanjukta Nej, Anumoy Ghosh, Sarosh Ahmad, Adnan Ghaffar and Mousa Hussein
Sensors 2022, 22(19), 7143; https://doi.org/10.3390/s22197143 - 21 Sep 2022
Cited by 19 | Viewed by 2796
Abstract
In this paper, a novel microstrip line-fed meander-line-based four-elements quad band Multiple Input and Multiple Output (MIMO) antenna is proposed with a gain enhancement technique. The proposed structure resonates at four bands simultaneously, that is, 1.23, 2.45, 3.5 and 4.9 GHz, which resemble [...] Read more.
In this paper, a novel microstrip line-fed meander-line-based four-elements quad band Multiple Input and Multiple Output (MIMO) antenna is proposed with a gain enhancement technique. The proposed structure resonates at four bands simultaneously, that is, 1.23, 2.45, 3.5 and 4.9 GHz, which resemble GPS L2, Wi-Fi, Wi-MAX and WLAN wireless application bands, respectively. The unit element is extended to four elements MIMO antenna structure exhibiting isolation of more than 22 dB between the adjacent elements without disturbing the resonant frequencies. In order to enhance the gain, two orthogonal microstrip lines are incorporated between the antenna elements which result in significant gain improvement over all the four resonances. Furthermore, the diversity performance of the MIMO structure is analyzed. The Envelope Co-Relation Coefficient (ECC), Diversity Gain (DG), Channel Capacity Loss (CCL), Mean Effective Gain (MEG) and Multiplexing Efficiency are obtained as 0.003, 10 dB, 0.0025 bps/Hz, −3 dB (almost) and 0.64 (min.), respectively, which are competent and compatible with practical wireless applications. The Total Active Reflection Coefficient (TARC) resembles the characteristic of the individual antenna elements. The layout area of the overall MIMO antenna is 0.33 λ × 0.29 λ, where λ is the free-space wavelength corresponding to the lowest resonance. The advantage of the proposed structure has been assessed by comparing it with previously reported MIMO structures based on number of antenna elements, isolation, gain, CCL and compactness. A prototype of the proposed MIMO structure is fabricated, and the measured results are found to be aligned with the simulated results. Full article
(This article belongs to the Special Issue Antenna Design and Sensors for Internet of Things)
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Figure 1
<p>The evolution of meander-line antenna: (<b>a</b>) first step, (<b>b</b>) second step, (<b>c</b>) third step of antenna design and (<b>d</b>) |S11| characteristics according to the design evolution.</p>
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<p>Schematic diagram of single unit quad band meander-line antenna.</p>
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<p>Simulated current distribution at (<b>a</b>) 1st resonance, (<b>b</b>) 2nd resonance, (<b>c</b>) 3rd resonance, (<b>d</b>) 4th resonance of single unit quad band meander-line antenna.</p>
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<p>Schematic diagram of four element MIMO antenna (<b>a</b>) without parasitic cross microstrip line and (<b>b</b>) with parasitic cross microstrip line.</p>
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<p>|S11| graph of the single unit meander line antenna variation of length of (<b>a</b>) P<sub>1</sub> and (<b>b</b>) P<sub>5</sub>.</p>
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<p>|S11| graph of the single unit meander line antenna.</p>
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<p>(<b>a</b>) |Sii| graph and (<b>b</b>) |Sij| graph of the four element MIMO with and without parasitic cross microstrip line.</p>
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<p>Current distribution of MIMO antenna with cross parasitic microstrip line structure at (<b>a</b>) 1st resonance, (<b>b</b>) 2nd resonance, (<b>c</b>) 3rd resonance and (<b>d</b>) 4th resonance.</p>
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<p>Diversity performances of quad elements quadband MIMO antenna (<b>a</b>) ECC, (<b>b</b>) diversity gain, (<b>c</b>) CCL, (<b>d</b>) TARC, (<b>e</b>) MEG, (<b>f</b>) multiplexing efficiency plot.</p>
Full article ">Figure 9 Cont.
<p>Diversity performances of quad elements quadband MIMO antenna (<b>a</b>) ECC, (<b>b</b>) diversity gain, (<b>c</b>) CCL, (<b>d</b>) TARC, (<b>e</b>) MEG, (<b>f</b>) multiplexing efficiency plot.</p>
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<p>Image of fabricated four elements MIMO with parasitic cross microstrip line structure.</p>
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<p>Comparison scattering parameter graphs of the measurement results of the fabricated prototype and simulated results of the four element MIMO with parasitic microstrip line: (<b>a</b>) |Sii| graph and (<b>b</b>) Transmission coefficient graph.</p>
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<p>Normalized copolar and crosspolar radiation pattern of the four element MIMO at (<b>a</b>) 1st resonance, (<b>b</b>) 2nd resonance, (<b>c</b>) 3rd resonance and (<b>d</b>) 4th resonance.</p>
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<p>Normalized copolar and crosspolar radiation pattern of the four element MIMO at (<b>a</b>) 1st resonance, (<b>b</b>) 2nd resonance, (<b>c</b>) 3rd resonance and (<b>d</b>) 4th resonance.</p>
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18 pages, 4196 KiB  
Article
An On-Film AMC Antenna Insert-Molded in Earbuds with Enhancement in In-Ear and In Situ Received-Signal Sensing
by Yejune Seo, Inyeol Moon, Junghyun Cho, Yejin Lee, Jiyeon Jang, Morimoto Shohei, Kurosaki Toshifumi and Sungtek Kahng
Sensors 2022, 22(12), 4523; https://doi.org/10.3390/s22124523 - 15 Jun 2022
Cited by 2 | Viewed by 2103
Abstract
In this paper, a novel thin and flexible antenna is proposed for earbuds to gain an improvement in their wireless signal-sensing capability as a film-based artificial magnetic conductor (AMC) structure. As antenna designs for earbuds face challenges of being embedded beneath the top [...] Read more.
In this paper, a novel thin and flexible antenna is proposed for earbuds to gain an improvement in their wireless signal-sensing capability as a film-based artificial magnetic conductor (AMC) structure. As antenna designs for earbuds face challenges of being embedded beneath the top cover of the earbud, conformal to curved surfaces, and very close to metallic ground and touch-panel parts, as well as scarce degrees of freedom from feeding conditions and functional degradation by human tissue, unlike conventional techniques such as quasi quarter-wavelength radiators on LDS and epoxy molding compounds (relatively thick and pricy), an antenna of a metal pattern on a film is made with another film layer as the AMC to mitigate problems of the antenna in a small and curved space of an insert-molded wireless device. The antenna was designed, fabricated, and embedded in earbud mockups to work for the 2.4 GHz Bluetooth RF link, and its functions were verified by RF and antenna measurement, showing that it could overcome the limitations in impedance matching with only lumped elements and poor radiation by the ordinary schemes. The input reflection coefficient and antenna efficiency were 10 dB and 9% better than other methods. In particular, the on-film AMC antenna (OFAA) presents robustness against deterioration by the human tissue, when it is placed in the ear phantom at the workbench and implemented in an in situ test using a large zorb ball mimicking a realistic sensing environment. This yielded an RSSI enhancement of 20–30 dB. Full article
(This article belongs to the Special Issue Antenna Design and Sensors for Internet of Things)
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Figure 1
<p>The earphone placed in the ear of the SAM: (<b>a</b>) overall view; (<b>b</b>) antenna in the earbud.</p>
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<p>Investigating the conventional antenna: (<b>a</b>) geometry; (<b>b</b>) attached to the human phantom; (<b>c</b>) S<sub>11</sub>; (<b>d</b>) radiated field pattern of the modified monopole antenna in the free space; (<b>e</b>) radiated field pattern of the modified monopole antenna with the phantom; (<b>f</b>) SAR.</p>
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<p>On−film antenna earphone in free space: (<b>a</b>) initial geometry of the proposed antenna; (<b>b</b>) layer structure; (<b>c</b>) S<sub>11</sub>; (<b>d</b>) radiated field pattern in free space; (<b>e</b>) radiated field attached to the human phantom; (<b>f</b>) SAR.</p>
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<p>On−film antenna earphone in free space: (<b>a</b>) initial geometry of the proposed antenna; (<b>b</b>) layer structure; (<b>c</b>) S<sub>11</sub>; (<b>d</b>) radiated field pattern in free space; (<b>e</b>) radiated field attached to the human phantom; (<b>f</b>) SAR.</p>
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<p>On−film antenna earbud in free space: (<b>a</b>) combined with the AMC; (<b>b</b>) layer structure; (<b>c</b>) geometry of the AMC; (<b>d</b>) S<sub>11</sub>; (<b>e</b>) radiated field pattern in free space; (<b>f</b>) radiated field pattern attached to the human phantom; (<b>g</b>) SAR; (<b>h</b>) comparison of upward near-field strength.</p>
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<p>On−film antenna earbud in free space: (<b>a</b>) combined with the AMC; (<b>b</b>) layer structure; (<b>c</b>) geometry of the AMC; (<b>d</b>) S<sub>11</sub>; (<b>e</b>) radiated field pattern in free space; (<b>f</b>) radiated field pattern attached to the human phantom; (<b>g</b>) SAR; (<b>h</b>) comparison of upward near-field strength.</p>
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<p>Fabricated antennas and measured S<sub>11</sub> and far-field patterns for the conventional antenna (CA) and OFAA: (<b>a</b>) photographs and S<sub>11</sub> of (<b>b</b>) CA; (<b>c</b>) initial geometry (short radiator without the AMC); (<b>d</b>) OFAA; (<b>e</b>) beam pattern of the CA (<b>left</b>) in free space and (<b>right</b>) in the ear; (<b>f</b>) beam pattern of the initial geometry (<b>left</b>) in free space and (<b>right</b>) in the ear; (<b>g</b>) beampattern of the OFAA (<b>left</b>) in free space and (<b>right</b>) in the ear.</p>
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<p>Fabricated antennas and measured S<sub>11</sub> and far-field patterns for the conventional antenna (CA) and OFAA: (<b>a</b>) photographs and S<sub>11</sub> of (<b>b</b>) CA; (<b>c</b>) initial geometry (short radiator without the AMC); (<b>d</b>) OFAA; (<b>e</b>) beam pattern of the CA (<b>left</b>) in free space and (<b>right</b>) in the ear; (<b>f</b>) beam pattern of the initial geometry (<b>left</b>) in free space and (<b>right</b>) in the ear; (<b>g</b>) beampattern of the OFAA (<b>left</b>) in free space and (<b>right</b>) in the ear.</p>
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<p>POPEYE-V10 [<a href="#B35-sensors-22-04523" class="html-bibr">35</a>].</p>
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<p>Antennas in the earbuds and RSSI tests using the head phantom: (<b>a</b>) plastic zorb ball mimicking a sensing sphere; (<b>b</b>) configuration of the measurement; (<b>c</b>) human phantom centered in the zorb ball as a TX in the form of a BLE module; (<b>d</b>) scheme to collect data from grid of sensors on the sphere; (<b>e</b>) device positioned on the SAM for the RSSI test; (<b>f</b>) measured points on the latitudes; (<b>g</b>) received signal strength distribution plot on the ball about the conventional antenna earbud; (<b>h</b>) received signal strength distribution plot on the ball about the proposed antenna earbud.</p>
Full article ">Figure 7 Cont.
<p>Antennas in the earbuds and RSSI tests using the head phantom: (<b>a</b>) plastic zorb ball mimicking a sensing sphere; (<b>b</b>) configuration of the measurement; (<b>c</b>) human phantom centered in the zorb ball as a TX in the form of a BLE module; (<b>d</b>) scheme to collect data from grid of sensors on the sphere; (<b>e</b>) device positioned on the SAM for the RSSI test; (<b>f</b>) measured points on the latitudes; (<b>g</b>) received signal strength distribution plot on the ball about the conventional antenna earbud; (<b>h</b>) received signal strength distribution plot on the ball about the proposed antenna earbud.</p>
Full article ">Figure 7 Cont.
<p>Antennas in the earbuds and RSSI tests using the head phantom: (<b>a</b>) plastic zorb ball mimicking a sensing sphere; (<b>b</b>) configuration of the measurement; (<b>c</b>) human phantom centered in the zorb ball as a TX in the form of a BLE module; (<b>d</b>) scheme to collect data from grid of sensors on the sphere; (<b>e</b>) device positioned on the SAM for the RSSI test; (<b>f</b>) measured points on the latitudes; (<b>g</b>) received signal strength distribution plot on the ball about the conventional antenna earbud; (<b>h</b>) received signal strength distribution plot on the ball about the proposed antenna earbud.</p>
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16 pages, 3912 KiB  
Article
Developing Broadband Microstrip Patch Antennas Fed by SIW Feeding Network for Spatially Low Cross-Polarization Situation
by Farzad Karami, Halim Boutayeb, Ali Amn-e-Elahi, Alireza Ghayekhloo and Larbi Talbi
Sensors 2022, 22(9), 3268; https://doi.org/10.3390/s22093268 - 24 Apr 2022
Cited by 15 | Viewed by 5445
Abstract
A stacked multi-layer substrate integrated waveguide (SIW) microstrip patch antenna with broadband operating bandwidth and low cross-polarization radiation is provided. A complete study on the propagating element bandwidth and cross polarization level is presented to demonstrate the importance of the design. The proposed [...] Read more.
A stacked multi-layer substrate integrated waveguide (SIW) microstrip patch antenna with broadband operating bandwidth and low cross-polarization radiation is provided. A complete study on the propagating element bandwidth and cross polarization level is presented to demonstrate the importance of the design. The proposed antenna includes three stacked printed circuit board (PCB) layers, including one layer for the radiating 2 × 2 rectangular patch elements and two SIW PCB layers for the feeding network. There are two common methods for excitation in cavity-backed patch antennas: probe feeding (PF) and aperture coupling (AC). PF can be used to increase the bandwidth of the antenna. Although this method increases the antenna’s bandwidth, it produces a strong cross-polarized field. The AC method can be used to suppress cross-polarized fields in microstrip patch antennas. As microstrip patch antennas are inherently narrowband, the AC method has little effect on their bandwidth. This paper proposes an antenna that is simultaneously fed by AC and PF. As a result of this innovation, the operating bandwidth of the antenna has increased, and cross-polarization has been reduced. Actually, the combination of probe feeding and aperture coupling schemes leads to achieving a broadband operating bandwidth. The arrangement of radiator elements and cavities implements a mirrored excitation technique while maintaining a low cross-polarization level. In both numerical and experimental solutions, a less than −30 dB cross-polarization level has been achieved for all of the main directions. A fractional impedance bandwidth of 29.8% (10.55–14.25 GHz) for S11 < −10-dB is measured for the proposed array. Simulated and measured results illustrate good agreement. Having features like low cost, light weight, compactness, broadband, integration capabilities, and low cross-polarization level makes the designed antenna suitable for remote-sensing and satellite applications. Full article
(This article belongs to the Special Issue Antenna Design and Sensors for Internet of Things)
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Figure 1
<p>Configuration of the proposed 2 × 2 patch array. (<b>a</b>) Exploded view, (<b>b</b>) top view, (<b>c</b>), back view, and top views of the (<b>d</b>) top layer, (<b>e</b>) middle layer, and (<b>f</b>) bottom layer.</p>
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<p>Configuration of the proposed 2 × 2 patch array. (<b>a</b>) Exploded view, (<b>b</b>) top view, (<b>c</b>), back view, and top views of the (<b>d</b>) top layer, (<b>e</b>) middle layer, and (<b>f</b>) bottom layer.</p>
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<p>Configuration of the proposed 2 × 2 patch array. (<b>a</b>) Exploded view, (<b>b</b>) top view, (<b>c</b>), back view, and top views of the (<b>d</b>) top layer, (<b>e</b>) middle layer, and (<b>f</b>) bottom layer.</p>
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<p>Simulated electric field distributions of the proposed array. (<b>a</b>) Top layer, (<b>b</b>) middle layer, and (<b>c</b>) bottom layer.</p>
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<p>Photographs of the proposed fabricated array.</p>
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<p>Simulated and measured S<sub>11</sub> and the realized gain of the fabricated patch array.</p>
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<p>Simulated and measured radiation pattern of the proposed array for different frequencies of (<b>a</b>) 10.55, (<b>b</b>) 12.5, and (<b>c</b>) 14.25 GHz. (<b>d</b>) Simulated cross-polarization level of the proposed array versus Ɵ and ???? angles.</p>
Full article ">Figure 5 Cont.
<p>Simulated and measured radiation pattern of the proposed array for different frequencies of (<b>a</b>) 10.55, (<b>b</b>) 12.5, and (<b>c</b>) 14.25 GHz. (<b>d</b>) Simulated cross-polarization level of the proposed array versus Ɵ and ???? angles.</p>
Full article ">Figure 5 Cont.
<p>Simulated and measured radiation pattern of the proposed array for different frequencies of (<b>a</b>) 10.55, (<b>b</b>) 12.5, and (<b>c</b>) 14.25 GHz. (<b>d</b>) Simulated cross-polarization level of the proposed array versus Ɵ and ???? angles.</p>
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16 pages, 6095 KiB  
Article
Design and Realization of an UHF Frequency Reconfigurable Antenna for Hybrid Connectivity LPWAN and LEO Satellite Networks
by Abdellatif Bouyedda, Bruno Barelaud and Laurent Gineste
Sensors 2021, 21(16), 5466; https://doi.org/10.3390/s21165466 - 13 Aug 2021
Cited by 10 | Viewed by 4817
Abstract
UHF satellite communication for Internet of Things (IoT) technology is rapidly emerging in monitoring applications as it offers the possibility of lower-costs and global coverage. At the present time, Low Power Wide Area Network (LPWAN) solutions offer low power consumption, but still suffer [...] Read more.
UHF satellite communication for Internet of Things (IoT) technology is rapidly emerging in monitoring applications as it offers the possibility of lower-costs and global coverage. At the present time, Low Power Wide Area Network (LPWAN) solutions offer low power consumption, but still suffer from white zones. In this paper, the authors propose an UHF frequency reconfigurable Antenna for hybrid connectivity LoRaWAN (at 868 MHz) and UHF satellite communication (Tx at 401 MHz and Rx at 466 MHz) with the Low Earth Orbit (LEO) Kineis constellation. The antenna is based on a meandered line structure loaded with lumped components and a PIN diode to control the antenna resonant frequencies. It resonates at 401 and 868 MHz when the PIN diode is forward-biased (ON state) and 466 MHz in reverse-biased configuration (OFF state). The antenna is designed inside the enclosure with the presence of all the parts of the connected device. The results of EM simulations and parametric studies on the values of the lumped components and the PIN diode equivalent model, which are obtained with HFSS, are presented. The antenna is prototyped and has dimensions of 78 mm × 88 mm × 1.6 mm. The paper proposes a fast and practical method to reduce time development and compensate the frequency shift between measurement and simulation. Full article
(This article belongs to the Special Issue Antenna Design and Sensors for Internet of Things)
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Figure 1

Figure 1
<p>Designed tracker: (<b>a</b>) exploded view of the tracker device; (<b>b</b>) assembled parts of the tracker device.</p>
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<p>Proposed antenna: (<b>a</b>) top view: structure reserved for GPS and BLE (black color); (<b>b</b>) lumped components, UFL connector and PIN diode placements; (<b>c</b>) bottom view; (<b>d</b>) top view with geometric key parameters.</p>
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<p>Schematic of the EM circuit co-simulation (HFSS) of the modeled antenna within enclosure, the PIN diode and the loaded lumped components: (<b>a</b>) PIN diode equivalent in ON state; (<b>b</b>) PIN diode equivalent model in OFF state.</p>
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<p>Simulated input reflection coefficient of the antenna for varied parameter L3 when PIN diode is OFF (Cp = 0 fF, Rp = 15 KΩ and Ls = 0.6 nH): (<b>a</b>) at lower frequency; (<b>b</b>) at the higher frequency.</p>
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<p>Simulated S11 for varied parameter L4 when the PIN diode is ON (Ls = 0.6 nH and Rs = 1 Ω).</p>
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<p>Simulated input reflection coefficient of the antenna with variated parameters of the PIN diode: (<b>a</b>) variation of Rp; (<b>b</b>) variation of Cp.</p>
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<p>Simulated S11 of the antenna in OFF configuration with and without matching network: (<b>a</b>) at 401 MHz; (<b>b</b>) at 868 MHz.</p>
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<p>Simulated S11 of the antenna in ON configuration with and without matching network.</p>
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<p>Current distributions: (<b>a</b>) at 401 MHz (OFF state); (<b>b</b>) at 868 MHz (OFF state); (<b>c</b>) at 466 MHz (ON state).</p>
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<p>Simulated radiation patterns: (<b>a</b>) at 401 MHz (OFF state); (<b>b</b>) at 868 MHz (OFF state); (<b>c</b>) at 466 MHz (ON state).</p>
Full article ">Figure 10 Cont.
<p>Simulated radiation patterns: (<b>a</b>) at 401 MHz (OFF state); (<b>b</b>) at 868 MHz (OFF state); (<b>c</b>) at 466 MHz (ON state).</p>
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<p>Prototyped tracker device: (<b>a</b>) prototyped antenna and the different parts of the connected device; (<b>b</b>) assembled connected device (tracker).</p>
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<p>Simulated and measured S11 of the antenna in OFF configuration: (<b>a</b>) at 401 MHz; (<b>b</b>) at 868 MHz.</p>
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<p>Simulated and measured S11 of the antenna in ON configuration.</p>
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<p>Simulated and measured S11: step 1.</p>
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<p>Simulated and measured S11: step 2.</p>
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<p>Simulated and measured S11: step 3.</p>
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<p>Measured S11 in steps 3 (with PIN diode) and 4.</p>
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Review

Jump to: Research

18 pages, 1866 KiB  
Review
Circularly Polarized Hybrid Dielectric Resonator Antennas: A Brief Review and Perspective Analysis
by Rajasekhar Nalanagula, Naresh K. Darimireddy, Runa Kumari, Chan-Wang Park and R. Ramana Reddy
Sensors 2021, 21(12), 4100; https://doi.org/10.3390/s21124100 - 15 Jun 2021
Cited by 22 | Viewed by 4999
Abstract
Recently, it has been a feasible approach to build an antenna, in view of the potential advantages they offer. One of the recent trends in dielectric resonator antenna research is the use of compound and hybrid structures. Several considerable investigations have been already [...] Read more.
Recently, it has been a feasible approach to build an antenna, in view of the potential advantages they offer. One of the recent trends in dielectric resonator antenna research is the use of compound and hybrid structures. Several considerable investigations have been already underway showing quite interesting and significant features in bandwidth, gain, and generation of circular polarization. A critical review on a journey of circularly polarized hybrid dielectric resonator antennas is presented in this article. A general discussion of circular polarization and DR antennas are provided at the forefront. Evolution, significant challenges, and future aspects with new ideas in designing hybrid dielectric resonator antennas are indicated at the end of the review. State-of-the-art advances and associated design challenges of circularly polarized hybrid DR antennas and related empirical formulas used to find resonance frequency of different hybrid modes produced are discussed in this paper. Full article
(This article belongs to the Special Issue Antenna Design and Sensors for Internet of Things)
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<p>Venn Diagram representation for dielectric antenna technology.</p>
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<p>A few examples of hybrid dielectric resonator antenna technology. (<b>a</b>) microstrip disc loaded dielectric antennas, and (<b>b</b>) dielectric loaded microstrip patch antennas.</p>
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<p>(<b>a</b>) 3D-view and (<b>b</b>) Top-view of hybrid structure (Reference [<a href="#B116-sensors-21-04100" class="html-bibr">116</a>]).</p>
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<p>Representation of simulated and measured (<b>a</b>) S11 (<b>b</b>) AR plots, along with orthogonal modes. (Reference [<a href="#B116-sensors-21-04100" class="html-bibr">116</a>]).</p>
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<p>Representation of simulated and measured (<b>a</b>) S11 (<b>b</b>) AR plots, along with orthogonal modes. (Reference [<a href="#B116-sensors-21-04100" class="html-bibr">116</a>]).</p>
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<p>Diagonally arranged two-element multiple inputs multiple output (MIMO) hybrid structure (<b>a</b>) Side view, (<b>b</b>) Top view (Reference [<a href="#B117-sensors-21-04100" class="html-bibr">117</a>]).</p>
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<p>Configuration of dual-sense polarized hybrid DRA. (<b>a</b>) Top view, and (<b>b</b>) side view (Reference [<a href="#B128-sensors-21-04100" class="html-bibr">128</a>]).</p>
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32 pages, 12665 KiB  
Review
Liquid-Based Reconfigurable Antenna Technology: Recent Developments, Challenges and Future
by Habshah Abu Bakar, Rosemizi Abd Rahim, Ping Jack Soh and Prayoot Akkaraekthalin
Sensors 2021, 21(3), 827; https://doi.org/10.3390/s21030827 - 26 Jan 2021
Cited by 15 | Viewed by 5591
Abstract
Advances in reconfigurable liquid-based reconfigurable antennas are enabling new possibilities to fulfil the requirements of more advanced wireless communication systems. In this review, a comparative analysis of various state-of-the-art concepts and techniques for designing reconfigurable antennas using liquid is presented. First, the electrical [...] Read more.
Advances in reconfigurable liquid-based reconfigurable antennas are enabling new possibilities to fulfil the requirements of more advanced wireless communication systems. In this review, a comparative analysis of various state-of-the-art concepts and techniques for designing reconfigurable antennas using liquid is presented. First, the electrical properties of different liquids at room temperature commonly used in reconfigurable antennas are identified. This is followed by a discussion of various liquid actuation techniques in enabling high frequency reconfigurability. Next, the liquid-based reconfigurable antennas in literature used to achieve the different types of reconfiguration will be critically reviewed. These include frequency-, polarization-, radiation pattern-, and compound reconfigurability. The current concepts of liquid-based reconfigurable antennas can be classified broadly into three basic approaches: altering the physical (and electrical) dimensions of antennas using liquid; applying liquid-based sections as reactive loads; implementation of liquids as dielectric resonators. Each concept and their design approaches will be examined, outlining their benefits, limitations, and possible future improvements. Full article
(This article belongs to the Special Issue Antenna Design and Sensors for Internet of Things)
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Figure 1
<p>Illustrative comparison between (<b>a</b>) electrocapillary actuation (ECA) and (<b>b</b>) continuous electrowetting (CEW) [<a href="#B20-sensors-21-00827" class="html-bibr">20</a>].</p>
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<p>Illustration of the mechanism of (<b>a</b>) continuous electrowetting (CEW) and (<b>b</b>) electrochemically controlled capillary (ECC) [<a href="#B37-sensors-21-00827" class="html-bibr">37</a>].</p>
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<p>(<b>a</b>) Liquid metal in an electrolyte. (<b>b</b>) Potential gradient across the electrolyte [<a href="#B42-sensors-21-00827" class="html-bibr">42</a>].</p>
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<p>Fluidic slot antenna. (<b>a</b>) Liquid metal slugs in aperture. (<b>b</b>) Liquid metal slug in microstrip line [<a href="#B36-sensors-21-00827" class="html-bibr">36</a>].</p>
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<p>Frequency reconfigurable antenna. (<b>a</b>) Top view. (<b>b</b>) Side view [<a href="#B42-sensors-21-00827" class="html-bibr">42</a>].</p>
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<p>Frequency reconfigurable antenna. (<b>a</b>) Top view. (<b>b</b>) Side view [<a href="#B42-sensors-21-00827" class="html-bibr">42</a>].</p>
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<p>Planar inverted-F antenna (PIFA) [<a href="#B22-sensors-21-00827" class="html-bibr">22</a>].</p>
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<p>Patch antenna with two slotted patches [<a href="#B45-sensors-21-00827" class="html-bibr">45</a>].</p>
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<p>The proposed frequency tunable antenna. (<b>a</b>) Schematic antenna. (<b>b</b>) Fabricated antenna [<a href="#B32-sensors-21-00827" class="html-bibr">32</a>].</p>
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<p>Dual patch antenna. (<b>a</b>) OFF configuration. (<b>b</b>) ON configuration [<a href="#B46-sensors-21-00827" class="html-bibr">46</a>].</p>
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<p>The proposed wearable antenna. (<b>a</b>) Galinstan and silicon tubing. (<b>b</b>) Loop antenna [<a href="#B47-sensors-21-00827" class="html-bibr">47</a>].</p>
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<p>The CPW folded slot antenna with Galinstan bridges [<a href="#B23-sensors-21-00827" class="html-bibr">23</a>].</p>
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<p>(<b>a</b>) Top view of the proposed switchable dual band slot antenna. (<b>b</b>) A-A’ cross section of the antenna [<a href="#B35-sensors-21-00827" class="html-bibr">35</a>].</p>
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<p>(<b>a</b>) Frequency-tunable antenna. (<b>b</b>) Cross section of the microfluidic channel structure [<a href="#B48-sensors-21-00827" class="html-bibr">48</a>].</p>
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<p>(<b>a</b>) The pixelated dipole antenna. (<b>b</b>) Top side pixel: ‘on’ state on the left side. (<b>c</b>) Bottom side pixel: ‘off’ state on the left side. (<b>d</b>) Side view [<a href="#B43-sensors-21-00827" class="html-bibr">43</a>].</p>
Full article ">Figure 14 Cont.
<p>(<b>a</b>) The pixelated dipole antenna. (<b>b</b>) Top side pixel: ‘on’ state on the left side. (<b>c</b>) Bottom side pixel: ‘off’ state on the left side. (<b>d</b>) Side view [<a href="#B43-sensors-21-00827" class="html-bibr">43</a>].</p>
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<p>The proposed reconfigurable quarter-mode substrate integrated waveguide (QMSIW) antenna. (<b>a</b>) Top view. (<b>b</b>) A-A’ cross section view. (<b>c</b>,<b>d</b>) ON and OFF states of via [<a href="#B49-sensors-21-00827" class="html-bibr">49</a>].</p>
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<p>The proposed reconfigurable quarter-mode substrate integrated waveguide (QMSIW) antenna. (<b>a</b>) Top view. (<b>b</b>) A-A’ cross section view. (<b>c</b>,<b>d</b>) ON and OFF states of via [<a href="#B49-sensors-21-00827" class="html-bibr">49</a>].</p>
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<p>The meander antenna. (<b>a</b>) Perspective view. (<b>b</b>) Meander patch [<a href="#B50-sensors-21-00827" class="html-bibr">50</a>].</p>
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<p>Design principles of the proposed DRA in [<a href="#B51-sensors-21-00827" class="html-bibr">51</a>]: (<b>a</b>) DRA without polarizer; (<b>b</b>) DRA with polarizer; (<b>c</b>) distribution of electric-field into DRA; (<b>d</b>) distribution of electric-field into DRA with polarizer; (<b>e</b>) working angle before and after the liquid-metal injection.</p>
Full article ">Figure 17 Cont.
<p>Design principles of the proposed DRA in [<a href="#B51-sensors-21-00827" class="html-bibr">51</a>]: (<b>a</b>) DRA without polarizer; (<b>b</b>) DRA with polarizer; (<b>c</b>) distribution of electric-field into DRA; (<b>d</b>) distribution of electric-field into DRA with polarizer; (<b>e</b>) working angle before and after the liquid-metal injection.</p>
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<p>(<b>a</b>) Antenna structure. (<b>b</b>) Electric field of the dielectric resonator antenna (DRA) without the liquid metal. (<b>c</b>) Electric field of the dielectric resonator antenna (DRA) with the liquid metal [<a href="#B52-sensors-21-00827" class="html-bibr">52</a>].</p>
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<p>Liquid metal actuation within the fluidic channels for the antenna in [<a href="#B40-sensors-21-00827" class="html-bibr">40</a>]: (<b>a</b>) one-sided channel structure. (<b>b</b>) State 1 with liquid metal (black) and electrolyte solution (green) in the channels. (<b>c</b>) Withdrawing the liquid metal into reservoir. (<b>d</b>) Reversing the polarity to actuate the liquid metal into State 2. (<b>e</b>) The cross section of each arm structure. (<b>f</b>) Front and back of the assembled antenna.</p>
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<p>Schematic of antenna proposed in [<a href="#B53-sensors-21-00827" class="html-bibr">53</a>]: (<b>a</b>) upper layer, (<b>b</b>) top view, (<b>c</b>) bottom view of the lower layer, (<b>d</b>) top view of fabricated antenna.</p>
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<p>(<b>a</b>) Truncated-corner patch antenna. (<b>b</b>) Annular slot antenna [<a href="#B54-sensors-21-00827" class="html-bibr">54</a>].</p>
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<p>Extended E-shaped patch antenna [<a href="#B45-sensors-21-00827" class="html-bibr">45</a>].</p>
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<p>States of the Galinstan slug. (<b>a</b>) The 0 mm offset state. (<b>b</b>–<b>d</b>) The <span class="html-italic">X</span>-mm-offset state [<a href="#B17-sensors-21-00827" class="html-bibr">17</a>].</p>
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<p>Fabricated prototype. (<b>a</b>) Photograph of antenna. (<b>b</b>) Proposed antenna [<a href="#B55-sensors-21-00827" class="html-bibr">55</a>].</p>
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<p>(<b>a</b>) The directivity-reconfigurable wideband antenna assembly. (<b>b</b>) Design parameters of the two-arm spiral antenna [<a href="#B16-sensors-21-00827" class="html-bibr">16</a>].</p>
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<p>Unit cell of the nested ring-split ring transmitarray. (<b>a</b>) Double layered. (<b>b</b>) One layer for the rotation angle of 20°. (<b>c</b>) Fabricated double layered antenna [<a href="#B56-sensors-21-00827" class="html-bibr">56</a>].</p>
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<p>Polarization and pattern reconfigurable dipole antenna [<a href="#B41-sensors-21-00827" class="html-bibr">41</a>].</p>
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<p>Schematic of the (<b>a</b>) reconfigurable crossed dipole, and (<b>b</b>) feed detail [<a href="#B18-sensors-21-00827" class="html-bibr">18</a>].</p>
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<p>The proposed antenna in [<a href="#B39-sensors-21-00827" class="html-bibr">39</a>]: (<b>a</b>) side view, (<b>b</b>) top view, and (<b>c</b>) antenna prototype.</p>
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<p>3D printed antenna tree [<a href="#B57-sensors-21-00827" class="html-bibr">57</a>].</p>
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<p>Substrate stack-up of the liquid-metal monopole antenna [<a href="#B38-sensors-21-00827" class="html-bibr">38</a>].</p>
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<p>(<b>a</b>) Bottom and top layer of the fluidically switched antenna. (<b>b</b>) Fluid switch. (<b>c</b>) Cross sectional view of the fluid switch [<a href="#B58-sensors-21-00827" class="html-bibr">58</a>].</p>
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<p>(<b>a</b>) Plastic antenna box, latex diaphragm, and syringe. (<b>b</b>) Bent dipole antenna and coaxial cable arrangement. (<b>c</b>) Dimensions of the two bent dipole antennas with castor oil (on the left) and ethyl acetate (on the right) [<a href="#B59-sensors-21-00827" class="html-bibr">59</a>].</p>
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<p>Microfluidic frequency reconfigurable patch antenna. (<b>a</b>) 3D view. (<b>b</b>) Side view [<a href="#B60-sensors-21-00827" class="html-bibr">60</a>].</p>
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<p>Liquid dielectric resonator antenna (DRA) [<a href="#B61-sensors-21-00827" class="html-bibr">61</a>].</p>
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<p>Reconfigurable dielectric resonator antenna (DRA) presented in [<a href="#B62-sensors-21-00827" class="html-bibr">62</a>].</p>
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