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WO2022166566A1 - 一种开关管的控制方法、装置及直流变换器 - Google Patents

一种开关管的控制方法、装置及直流变换器 Download PDF

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Publication number
WO2022166566A1
WO2022166566A1 PCT/CN2022/072280 CN2022072280W WO2022166566A1 WO 2022166566 A1 WO2022166566 A1 WO 2022166566A1 CN 2022072280 W CN2022072280 W CN 2022072280W WO 2022166566 A1 WO2022166566 A1 WO 2022166566A1
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WIPO (PCT)
Prior art keywords
current
transformer
time
phase
switch
Prior art date
Application number
PCT/CN2022/072280
Other languages
English (en)
French (fr)
Inventor
杨永春
何硕祖
罗宇浩
Original Assignee
昱能科技股份有限公司
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Filing date
Publication date
Application filed by 昱能科技股份有限公司 filed Critical 昱能科技股份有限公司
Priority to US18/710,200 priority Critical patent/US20250030349A1/en
Priority to EP22748845.9A priority patent/EP4485781A1/en
Publication of WO2022166566A1 publication Critical patent/WO2022166566A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
    • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
    • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
    • H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/3353—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having at least two simultaneously operating switches on the input side, e.g. "double forward" or "double (switched) flyback" converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
    • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
    • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
    • H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33573—Full-bridge at primary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0009—Devices or circuits for detecting current in a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0016—Control circuits providing compensation of output voltage deviations using feedforward of disturbance parameters
    • H02M1/0019—Control circuits providing compensation of output voltage deviations using feedforward of disturbance parameters the disturbance parameters being load current fluctuations
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0043—Converters switched with a phase shift, i.e. interleaved
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/083—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the ignition at the zero crossing of the voltage or the current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/01—Resonant DC/DC converters
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention relates to the technical field of power electronics, in particular to a control method and device of a switch tube and a DC converter.
  • a DC converter includes a DC power supply, a phase-shifted full-bridge circuit, a transformer and a rectifier circuit.
  • FIG. 1 is a schematic structural diagram of a DC converter in the prior art. After the switch Q1 in the leading bridge arm in the phase full bridge circuit is turned off, the corresponding switch Q4 in the lagging bridge arm is controlled to be turned off; after the switch Q2 in the leading bridge arm in the phase-shifted full bridge circuit is turned off, the control lag The corresponding switch tube Q3 in the bridge arm is turned off.
  • the switch on the lagging bridge arm usually works in a hard switching state, that is, when the control switch is turned off, the current flowing through the switch is still very large, so that the loss of the switch is large.
  • the switch tube works in a hard switching state, the anti-interference ability of the switch tube is low, that is, EMC (Electromagnetic Compatibility, electromagnetic compatibility) noise will be generated.
  • the purpose of the present invention is to provide a control method, device and DC converter for a switch tube, which can ensure that the current flowing through the switch tube on the hysteresis bridge arm is small when the switch tube on the hysteresis bridge arm is disconnected, so as to realize the switch tube on the hysteresis bridge arm.
  • the soft turn-off of the switch reduces the loss, improves the anti-interference ability of the switch, and improves the EMS performance of the switch.
  • the present invention provides a control method of a switch tube, which is applied to a DC converter, the DC converter includes a DC power supply, a phase-shifted full-bridge circuit, a transformer and a rectifier circuit provided with an LC resonance circuit, include:
  • the DC converter further includes a current transformer, an integrating circuit, an amplifier and a comparator connected in sequence, the input end of the current transformer is connected to the transformer, and the current transformer is used for collecting the data on the transformer.
  • the current on the transformer the integrating circuit integrates the current on the transformer to obtain a voltage signal
  • the input positive terminal of the amplifier is provided with a bias voltage for amplifying the voltage signal and adding a bias voltage, outputting the amplified voltage signal
  • the comparator is used for comparing the amplified voltage signal with the maximum integration threshold and the minimum integration threshold, and when the amplified voltage signal is greater than the maximum integration threshold or smaller than the minimum integration threshold
  • Output a first level for a limited time output a second level when the amplified voltage signal is between the maximum integration threshold and the minimum integration threshold, and the first level is opposite to the second level ;
  • the time from the above-mentioned integration start time to the integration end time is taken as the integration time
  • the switch tube on the leading bridge arm is turned off as the starting time to judge whether the phase-shift angle time is reached, so as to realize the judgment of the transformer. Whether the current is not greater than the current preset value;
  • phase shift angle time based on the integration time and the resonance parameters of the LC resonant circuit, it also includes:
  • the switch on the leading bridge arm in the phase-shifted full-bridge converter is turned off, the switch on the leading bridge arm is turned off as the starting time to determine whether the maximum phase shift angle is reached time;
  • the step of controlling the corresponding switch on the hysteresis bridge arm in the phase-shifted full-bridge converter to turn off is entered.
  • determining the maximum phase shift angle time includes:
  • the primary leakage inductance parameter of the transformer Based on the capacitance value of the junction capacitance of each switch tube in the phase-shifted full-bridge circuit, the primary leakage inductance parameter of the transformer, the resonance parameter of the LC resonant circuit, the output voltage and output current of the transformer, and the The integration time determines the maximum phase shift angle time.
  • the method further includes:
  • the time when the corresponding switch tube on the lag bridge arm in the phase-shifted full-bridge converter is turned off is used as the start time to judge whether the preset dead time is reached;
  • the time when the corresponding switch tube on the lag bridge arm in the phase-shifted full-bridge converter is turned off is used as the starting time to determine whether the preset dead time is reached, further comprising:
  • the preset dead time is set based on the maximum dead time, and the preset dead time is less than the maximum dead time;
  • the DC converter further comprises a current transformer, an amplifier and a comparator connected in sequence, the input end of the current transformer is connected to the transformer for collecting the current of the transformer, and the amplifier is used for Amplify the current of the transformer collected by the current transformer, output an amplified current signal, and the comparator is used for comparing the amplified current signal with the current preset value, and When the amplified current signal is not greater than the current preset value, a third level is output, and when the amplified current signal is greater than the current preset value, a fourth level is output, and the third level is output opposite to the fourth level;
  • the method further includes:
  • the present invention provides a control device for a switch tube, comprising:
  • the processor is configured to implement the steps of the above-mentioned control method of the switch tube when executing the computer program.
  • the present invention provides a DC converter, which includes the control device of the switch tube as described above, a DC power supply, a phase-shifted full-bridge circuit, a transformer and a rectifier circuit with an LC resonance circuit connected in sequence.
  • a DC converter which includes the control device of the switch tube as described above, a DC power supply, a phase-shifted full-bridge circuit, a transformer and a rectifier circuit with an LC resonance circuit connected in sequence.
  • the input end of the current transformer is connected to the transformer for collecting the current of the transformer
  • the amplifier is used for amplifying the current of the transformer collected by the current transformer, and outputting the amplified current signal;
  • the comparator is used for comparing the amplified current signal with the current preset value, and outputting a third level when the amplified current signal is not greater than the current preset value, and when the amplified current signal is not greater than the current preset value When the amplified current signal is greater than the preset current value, a fourth level is output, and the third level is opposite to the fourth level.
  • the present application provides a control method, device and DC converter for a switch tube.
  • the current on the transformer is not greater than the preset current
  • the corresponding switch on the hysteresis bridge arm is controlled by the value to be disconnected, which can ensure that the current flowing through the switch on the hysteresis bridge arm is small when the switch on the hysteresis bridge arm is disconnected, so as to realize the soft turn-off of the switch on the hysteresis bridge arm and reduce the loss.
  • improve the anti-interference ability of the switch tube improve the EMS performance of the switch tube.
  • FIG. 1 is a schematic structural diagram of a DC converter in the prior art
  • FIG. 2 is a schematic flowchart of a control method for a switch tube provided by the present invention
  • Fig. 3 is a specific structural schematic diagram of a DC converter provided by the present invention.
  • FIG. 4 is a specific structural schematic diagram of another DC converter provided by the present invention.
  • Fig. 5 is the flow chart of controlling the turn-on and turn-off of the switch in the phase-shifted full-bridge circuit in the prior art
  • FIG. 6 is a waveform diagram of key parameters in a DC converter provided by the present invention.
  • Fig. 7 is a kind of waveform diagram about current preset value provided by the present invention.
  • Fig. 8 is a kind of waveform diagram about integral threshold value provided by the present invention.
  • Fig. 9 is a kind of waveform diagram about the voltage on the switch tube provided by the present invention.
  • Fig. 11 is another waveform diagram of key parameters in the prior art when the switch tube in the lead bridge arm is controlled;
  • FIG. 12 is a schematic structural diagram of a DC converter provided with a current transformer provided by the present invention.
  • FIG. 13 is a waveform diagram of a current and a digital voltage signal of a transformer after adding a bias voltage provided by the present invention
  • FIG. 15 is a schematic structural diagram of a control device for a switch tube provided by the present invention.
  • the core of the present invention is to provide a switch tube control method, device and DC converter, which can ensure that the current flowing through the switch tube on the hysteresis bridge arm is small when the switch tube on the hysteresis bridge arm is disconnected, so as to realize the switch tube on the hysteresis bridge arm.
  • the soft turn-off of the switch reduces the loss, improves the anti-interference ability of the switch, and improves the EMS performance of the switch.
  • FIG. 2 is a schematic flowchart of a control method of a switch tube provided by the present invention.
  • the method is applied to a DC converter.
  • the DC converter includes a DC power supply, a phase-shifted full-bridge circuit, a transformer T and a rectifier circuit provided with an LC resonance circuit.
  • the method includes:
  • the DC converter in the prior art includes a DC power supply E, a phase-shifted full-bridge circuit, a transformer T and a rectifier circuit.
  • the rectifier circuit in the prior art DC converter has four Diodes D5, D6, D7 and D8 and an output filter inductor L2, when the primary side current of the transformer T is positive from the direction and negative from the undershoot to the direction, or negative from the direction and positive from the undershoot to the direction, the output filter The current of the inductor L2 cannot be abruptly changed, so that when the output filter inductor L2 is freewheeling, the transformer T is short-circuited with the diode.
  • the output filter inductor L2 the load, the diode D5 and the diode D6 form a loop, or the output filter inductor L2, the load, Diode D7 and diode D8 form a loop, so that the voltage of the secondary side of the transformer T is 0. Even if the voltage of the primary side of the transformer T is 0, the voltage output by the DC power supply E is all added to the primary side resonant inductance of the primary side of the transformer T. On L1, the output duty cycle of the transformer T is lost, which cannot meet the normal power demand of the load.
  • the rectifier circuit in the present application is provided with an LC resonant circuit, as shown in FIG. 3 , which is a specific structural schematic diagram of a DC converter provided by the present invention.
  • the resonant inductance Lrs of the secondary side can resonate , and the resonant inductors C5 and C6 on the secondary side can store and discharge when the direction of the current on the transformer T changes, so as to ensure that the transformer T can output power to the load normally, so as to avoid the loss of the output duty cycle of the transformer T, thereby It is ensured that the voltage on the secondary side of the transformer T is not 0, which provides conditions for the subsequent control of the switching tubes in the phase-shifted full-bridge circuit.
  • Fig. 5 is a flow chart of controlling the turn-on and turn-off of switches in a phase-shifted full-bridge circuit in the prior art.
  • the two switches on different bridge arms cross The turn-on time and turn-off time are separated by the phase-shift angle time, that is, the first switch tube Q1 is turned on and the fourth switch tube Q4 is turned on after the phase-shift angle time, and the first switch tube Q1 is turned off and the phase-shift tube is turned on.
  • the fourth switch Q4 is turned off, the second switch Q2 is turned on and the third switch Q3 is turned on after the phase shift time, and the second switch Q2 is turned off after the phase shift time. Then, the third switch tube Q3 is controlled to be turned off.
  • the bridge arms where the first switch transistor Q1 and the second switch transistor Q2 are located are the leading bridge arms
  • the bridge arms where the third switch transistor Q3 and the fourth switch transistor Q4 are located are the lagging bridge arms. Since the primary side of the transformer T is provided with the primary side resonant capacitor C7 and the primary side resonant inductor L1, please refer to FIG. 1, it can ensure that the first switch transistor Q1 and the second switch transistor Q2 on the leading bridge arm can realize zero voltage conduction or zero
  • the voltage is turned off, that is, when the voltage across the first switch tube Q1 or the second switch tube Q2 satisfies the zero-voltage turn-off condition, it is turned on or off.
  • the phase-shift angle time is preset and unchanged.
  • the switch on the lagging bridge arm is controlled to be turned on or off.
  • the current flowing through the switch is large at this time, For example, when the DC converter is lightly loaded, the energy stored in the primary resonant inductance L1 of the primary side of the transformer T cannot completely drain the charge of the drain-source capacitance of the switch on the lagging bridge arm, resulting in a higher current flowing through the switch. If it is large, the loss of the switch tube will be larger when it is disconnected, that is, the switch tube on the hysteresis bridge arm works in a hard switching state. When the switch tube works in a hard switching state, the EMC performance is poor, and the anti-interference of the switch tube is lower capacity.
  • the present application judges whether the current on the transformer T is not greater than the current preset value after the switch on the leading bridge arm of the phase-shift full-bridge converter is disconnected, instead of only based on the phase-shift angle time to the switch.
  • the present application does not limit the method for judging whether the current on the transformer T is not greater than the preset value of the current.
  • the switch on the lagging bridge arm is disconnected, so as to ensure that when the current flowing through the switch on the lagging bridge arm is small, the switch on the lagging bridge arm will be switched off. disconnected, so as to realize the soft turn-off of the switch tube on the hysteresis bridge arm, so as to reduce the loss of the switch tube.
  • the fourth switch tube Q4 is controlled to be turned off, the second switch tube Q2 is turned off, and the current flowing through the third switch tube Q3 is relatively small. After hours, the third switch tube Q3 is controlled to be disconnected.
  • the switch on the lagging bridge arm is turned off, and the current on the transformer T also changes direction, increasing or decreasing from 0. Therefore, It can also ensure that the rectifier diode in the rectifier circuit on the secondary side of the transformer T is turned off softly, so as to reduce the loss of the rectifier diode, improve the efficiency, and improve the EMS performance of the rectifier diode.
  • the phase-shift angle time can be determined according to the expected output power of the DC converter, so that when the phase-shift angle time is reached, the current flowing through the switch on the lagging bridge arm also satisfies the soft-off condition; Or combine the relationship between the current on the transformer T and the current preset value to control the switch tube to realize variable frequency control, so that the switching frequency of the switch tube can be reduced at low power, which is not only conducive to soft switching control, but also reduces switching loss, To achieve better efficiency and EMC performance at low power.
  • the dual-resonance phase-shift full-bridge control of the primary side and the secondary side can effectively utilize the two resonance points of the primary side and the secondary side of the transformer T, so as to control the switch tube based on the optimal resonant frequency and realize the optimal soft switching control.
  • FIG. 6 is a waveform diagram of key parameters in a DC converter provided by the present invention.
  • the figure includes the drive signals PWM_Q1 ⁇ Q4 of each switch tube in the phase-shifted full-bridge circuit, the primary current of the transformer T, the excitation voltage of the primary side of the transformer T, the excitation current of the primary side of the transformer T, the leakage inductance voltage of the transformer T, and the secondary
  • the voltage of the side capacitor and the current of the rectifier diode of the secondary side, the switching period in the figure is the time from when the first switch Q1 is turned on this time to the next time the first switch Q1 is turned on.
  • the control cycle of the four switches is divided into 12 processes, namely t0-t1, t1-t2, t2-t3, t3-t4, t4-t5, t5-t6, t6-t7, t7-t8, t8-t9, t9-t10, t10-t11, t11-t0.
  • the first switch transistor Q1 and the fourth switch transistor Q4 are both in the conducting state.
  • the excitation current on the primary side of the transformer T will first decrease in the negative direction and then increase in the positive direction.
  • the side current gradually increases from 0, the second resonant capacitor C6 on the secondary side of the transformer T is a charging capacitor at this time, the voltage across the second resonant capacitor C6 gradually rises, the first resonant capacitor C5 discharges, and the voltage across the first resonant capacitor C5
  • the voltage is gradually decreasing, the second diode D6 is turned on, and the current flowing through the second diode D6 first increases and then gradually decreases.
  • the first switch tube Q1 is turned off, the fourth switch tube Q4 is turned on, the first capacitor C1 connected in parallel at both ends of the first switch tube Q1 is charged, and the two ends of the second switch tube Q2 are connected in parallel.
  • the second capacitor C2 discharges, the excitation current of the primary side of the transformer T increases slowly in the positive direction, the primary side current begins to decrease, the second resonant capacitor C6 on the secondary side of the transformer T acts as a charging capacitor, and the voltage across the second resonant capacitor C6 continues to gradually increase When it rises, the first resonant capacitor C5 is discharged, the voltage across the first resonant capacitor C5 gradually decreases, and the second diode D6 outputs a current and gradually decreases.
  • the first switch tube Q1 is in an off state at this time, and the fourth switch tube Q4 continues to be turned on.
  • the body diode D2 of the second capacitor C2 conducts freewheeling, creating conditions for the zero-voltage conduction of the second switch tube Q2.
  • the transformer T is The excitation current of the side increases slowly in the positive direction, the current of the primary side continues to decrease, the second resonant capacitor C6 on the secondary side is still a charging capacitor, the voltage across the second resonant capacitor C6 rises slowly, the first resonant capacitor C5 continues to discharge, and the first resonant capacitor C5 continues to discharge. The voltage across the resonant capacitor C5 decreases gradually, and the second diode D6 outputs the current and decreases gradually.
  • the second switch transistor Q2 is turned on at t3, and the fourth switch transistor Q4 is still in the on state.
  • the time period from t1 to t3 is the dead time between the first switch transistor Q1 and the second switch transistor Q2.
  • the second switch transistor Q2 is turned on at zero voltage.
  • the excitation current of the primary side of the transformer T increases slowly in the positive direction, the primary side current continues to decrease, the secondary side second resonant capacitor C6 is still a charging capacitor, and the second The voltage across the resonant capacitor C6 rises slowly, the first resonant capacitor C5 continues to discharge, the voltage across the first resonant capacitor C5 gradually decreases, and the second diode D6 outputs a current that gradually decreases.
  • the fourth switch Q4 is turned off at time t4, the second switch Q2 continues to be turned on, and the fourth switch Q4 is turned off at zero voltage (the voltage across the fourth switch Q4 starts to rise from 0).
  • the current on the transformer T is also close to 0, that is, less than the preset value of the current, and the fourth switch Q4 is also turned off at approximately zero current)
  • the fourth capacitor C4 connected in parallel across the fourth switch Q4 is charged
  • the third capacitor C3 connected in parallel at both ends of the third switch tube Q3 discharges
  • the excitation current of the primary side of the transformer T increases slowly in the positive direction
  • the primary side current continues to decrease
  • the secondary side second resonant capacitor C6 is still a charging capacitor
  • the voltage across the second resonant capacitor C6 rises slowly
  • the first resonant capacitor C5 continues to discharge
  • the voltage across the first resonant capacitor C5 gradually decreases
  • the second diode D6 outputs a current that gradually decreases.
  • the second switch Q2 is in an on state, and the fourth switch Q4 is in an off state.
  • the body diode D3 of the third switch transistor Q3 When the voltage of the third capacitor C3 connected in parallel at the terminal is 0, the body diode D3 of the third switch transistor Q3 will be turned on, creating conditions for the third switch transistor Q3 to realize zero-voltage conduction, and the second resonant capacitor C6 on the secondary side is still a charging capacitor. , the voltage across the second resonant capacitor C6 rises slowly, the first resonant capacitor C5 continues to discharge, the voltage across the first resonant capacitor C5 gradually decreases, and the second diode D6 outputs a current that gradually decreases to 0.
  • the second switch Q2 and the third switch Q3 are both in the conducting state.
  • the excitation current on the primary side of the transformer T will first decrease in the forward direction and then increase in the reverse direction (the direction of the excitation current is defined by
  • the first resonant capacitor C5 on the secondary side is a charging capacitor, the voltage across the first resonant capacitor C5 gradually increases, the second resonant capacitor C6 discharges, and the second resonant capacitor C6
  • the voltage at both ends gradually drops, and the first diode D5 outputs the current and drops gradually.
  • the second switch tube Q2 is turned off, the third switch tube Q3 is turned on, the second capacitor C2 connected in parallel at both ends of the second switch tube Q2 is charged, and the two ends of the first switch tube Q1 are connected in parallel.
  • the first capacitor C1 discharges, the current on the primary side of the transformer T increases slowly in the reverse direction, and the excitation current on the primary side begins to decrease in the reverse direction.
  • the first resonant capacitor C5 on the secondary side of the transformer T acts as a charging capacitor, and the voltage across the first resonant capacitor C5 continues. Gradually rising, the second resonant capacitor C6 discharges, the voltage across the second resonant capacitor C6 gradually decreases, the first diode D5 outputs a current, and gradually decreases.
  • the body diode D1 of the first capacitor C1 conducts freewheeling, which creates conditions for the zero-voltage conduction of the first switch tube Q1, and the transformer T original
  • the excitation current of the side increases slowly in the reverse direction, the current of the primary side continues to decrease in the reverse direction, the first resonant capacitor C5 on the secondary side is still a charging capacitor, the voltage across the first resonant capacitor C5 rises slowly, and the second resonant capacitor C6 continues to discharge, The voltage across the second resonant capacitor C6 gradually decreases, and the first diode D5 outputs the current and gradually decreases.
  • the first switch tube Q1 is turned on at this moment, the first switch tube Q1 and the third switch tube Q3 are both in the conduction state, and the excitation current on the primary side of the transformer T increases in reverse.
  • the body diode D1 of the tube Q1 conducts freewheeling, and the first switch tube Q1 is turned on at zero voltage.
  • the excitation current increases slowly in the reverse direction, the primary current continues to decrease in the reverse direction, and the first resonant capacitor C5 on the secondary side is still In order to charge the capacitor, the voltage across the first resonant capacitor C5 rises slowly, the second resonant capacitor C6 discharges, the voltage across the second resonant capacitor C6 gradually decreases, and the first diode D5 outputs current and gradually decreases.
  • the third switch tube Q3 is turned off, the first switch tube Q1 continues to be turned on, and the third switch tube Q3 realizes zero-voltage turn-off (the voltage across the third switch tube Q3 starts to rise from 0).
  • the current on the transformer T is also close to 0, that is, less than the current preset value, the third switch Q3 is also turned off at approximately zero current)
  • the fourth capacitor C4 connected in parallel at both ends of the fourth switch Q4 discharges , the third capacitor C3 connected in parallel at both ends of the third switch tube Q3 is charged.
  • the excitation current of the primary side of the transformer T increases slowly in the reverse direction, the current of the primary side continues to decrease in the reverse direction, and the first resonant capacitor C5 of the secondary side is still charged.
  • the voltage across the first resonant capacitor C5 rises slowly, the second resonant capacitor C6 continues to discharge, the voltage across the second resonant capacitor C6 gradually decreases, and the first diode outputs current and gradually decreases.
  • the first switch Q1 is in an on state, and the third switch Q3 is in an off state.
  • the body diode D4 of the fourth switch Q4 When the voltage of the fourth capacitor C4 connected in parallel at the terminal is 0, the body diode D4 of the fourth switch Q4 will be turned on, creating conditions for the fourth switch Q4 to achieve zero-voltage conduction, and the secondary side first resonant capacitor C5 is still charging capacitor, the voltage across the first resonant capacitor C5 rises slowly, the second resonant capacitor C6 continues to discharge, the voltage across the second resonant capacitor C6 gradually decreases, and the main power current gradually decreases to zero, charging the first resonant capacitor C5 And the discharge of the second resonance capacitor C6 is gradually completed.
  • capacitors and diodes connected in parallel at both ends of each switch tube in the phase-shifted full-bridge current in this application may be the junction capacitor and body diode of each switch tube itself, which is not limited in this application.
  • the corresponding switch on the lagging bridge arm is controlled to be disconnected, which can ensure that the lagging bridge arm is turned off.
  • the switch tube is disconnected, the current flowing through the switch tube is small, so that the soft turn-off of the switch tube on the lagging bridge arm is realized, the loss is reduced, the anti-interference ability of the switch tube is improved, and the EMS performance of the switch tube is improved.
  • the DC converter further includes a current transformer CT, an integrating circuit, an amplifier U1 and a comparator connected in sequence.
  • the input end of the current transformer CT is connected to the transformer T, and the current transformer CT is used to collect the transformer.
  • the current on T, the integrating circuit integrates the current on the transformer T to obtain a voltage signal
  • the input positive terminal of the amplifier U1 is provided with a bias voltage Vref, which is used to amplify the voltage signal and add the bias voltage Vref
  • the output is amplified
  • the comparator is used to compare the amplified voltage signal with the maximum integration threshold and the minimum integration threshold, and output the first level when the amplified voltage signal is greater than the maximum integration threshold or smaller than the minimum integration threshold.
  • a second level is output, and the first level is opposite to the second level;
  • the time from the above-mentioned integration start time to the integration end time is taken as the integration time
  • the switch on the leading bridge arm is turned off as the starting time to judge whether the phase-shift angle time is reached, so as to determine whether the current on the transformer T is not greater than the current preset value;
  • the phase shift angle time can be calculated, so as to realize that the switch tube on the leading bridge arm is turned off as the starting time, and the phase shift time is reached.
  • the angular time is reached, the current on the transformer T is already less than the preset current value, wherein the phase-shift angular time in this embodiment is not fixed, but is calculated in real time based on the current on the transformer T.
  • the amplified voltage signal at each moment is compared with the maximum integration threshold Vth1 and the minimum integration threshold Vth2.
  • the first level is output.
  • the amplified voltage signal is The second level is output between the maximum integration threshold and the minimum integration threshold, and the integration time is determined when the comparator outputs the first level as the integration end time.
  • FIG. 7 is a waveform diagram of a current preset value provided by the present invention. Izcs in the figure is the current preset value, and the time between t0 and t4 is the integration time.
  • FIG. 8 which is a waveform diagram of the integration threshold provided by the present invention.
  • Vpc1 in the figure is the maximum integration threshold
  • Vpc2 is the minimum integration threshold
  • the bias voltage Vref is 1.65V.
  • the phase shift angle time can be calculated, so that the switch on the lagging bridge arm is turned off and the phase shift is reached after the corresponding switch on the leading bridge arm is turned off. It is turned off at the corner time, that is, the soft turn-off of the switch tube on the lagging bridge arm can be realized.
  • the method further includes:
  • the step of controlling the corresponding switch on the lagging bridge arm in the phase-shifted full-bridge converter to turn off is entered.
  • FIG. 9 is a waveform diagram of a voltage on a switch tube provided by the present invention.
  • the applicant considers that at time t4 in FIG. 9 , the fourth switch tube Q4 on the hysteresis bridge arm is turned off at zero current, that is, to realize ZCS (Zero Current Switch, zero current turn off), and the hysteresis bridge at time t6
  • the third switch tube Q3 on the arm is turned on at zero voltage, that is, ZVS is realized.
  • the voltage Vds_Q3 across the third switch tube Q3 is at the bottom end, and the third switch tube Q3 is turned on at this time to realize ZVS (Zero Voltage Switching, zero voltage turn-on).
  • ZVS Zero Voltage Switching, zero voltage turn-on
  • Uc is the input voltage of the DC converter
  • Io is the current on the transformer T
  • n is the turns ratio between the primary and secondary sides of the transformer T
  • C is the junction capacitance of each switch in the phase-shifted full-bridge circuit or The capacitors connected in parallel at both ends of each switch tube
  • Lr is the leakage inductance parameter of the primary side of the transformer T
  • tonv is the time when the switch tubes on the leading bridge arm and the corresponding switch tubes on the lagging bridge arm are turned on at the same time.
  • the dead time that is, the period of t4-t6 remains unchanged, when the phase shift time Tps>Tpsmax, due to the small leakage inductance current of the primary side of the transformer T, the resonance energy is not enough, the voltage Vds_Q3 across the third switch tube Q3 Or the voltage Vds_Q4 across the fourth switch tube Q4 may only resonate to t5, and the voltage across the switch tube is too high, that is, ZVS cannot be achieved. Therefore, the maximum phase-shift angle time is a critical condition that enables the switch on the lagging bridge arm to achieve ZVS.
  • phase-shift angle time is used as the phase-shift angle time to control the switch on the lag bridge arm; when the phase-shift angle time is less than the maximum phase-shift angle time, the phase-shift angle time is not changed, and the lag bridge arm is adjusted based on the determined phase-shift angle time. on the switch tube for control.
  • the method in this embodiment can not only enable the switches on the hysteresis bridge arm to implement ZCS, but also ensure that the switches on the hysteresis bridge arm implement ZVS.
  • determining the maximum phase shift angle time includes:
  • the maximum phase-shift angle time is determined based on the capacitance of the junction capacitance of each switch in the phase-shifted full-bridge circuit, the primary leakage inductance parameter of the transformer T, the resonance parameter of the LC resonant circuit, the output voltage and output current of the transformer T, and the integration time. .
  • Uc is the input voltage of the DC converter
  • Io is the current on the transformer T
  • n is the turns ratio of the primary and secondary sides of the transformer T
  • C is the junction capacitance of each switch in the phase-shifted full-bridge circuit or The capacitors connected in parallel at both ends of each switch tube
  • Lr is the leakage inductance parameter of the primary side of the transformer T
  • tonv is the time when the switch tubes on the leading bridge arm and the corresponding switch tubes on the lagging bridge arm are turned on at the same time.
  • the method further includes:
  • the time when the corresponding switch tube on the lagging bridge arm in the phase-shifted full-bridge converter is turned off is used as the starting time to judge whether the preset dead time is reached;
  • the switches on the hysteresis bridge arm are staggered and turned on, even if the two switches on the hysteresis bridge arm have no time to be turned on at the same time, after the switch tube in the conduction state is turned off, after a preset dead time
  • the second switch on the hysteresis bridge arm is controlled to be turned on after the time zone, that is, after the third switch Q3 is turned off and the preset dead time has passed, the fourth switch Q4 is controlled to be turned on, and the fourth switch Q4 is turned off.
  • the third switch Q3 is controlled to be turned on, so as to prevent the switches on the lagging bridge arm from being turned on at the same time and causing a short circuit, so that the DC converter cannot work normally.
  • the preset dead time when the preset dead time is short, it can be ensured that the switch tubes on the lagging bridge arm cannot participate in the freewheeling, that is, it can be ensured that the switching tubes on the lagging bridge arm work in a soft switching state.
  • the time when the corresponding switch tube on the lagging bridge arm in the phase-shifted full-bridge converter is turned off is used as the starting time to determine whether the preset dead time is reached, and further includes:
  • the preset dead time is set based on the maximum dead time, and the preset dead time is less than the maximum dead time;
  • the applicant considers that if the dead time Tdt is greater than the time between t4 and t6, at t6, it can be seen from FIG. 9 that at the dotted line, the voltage Vds_Q3 across the third switch tube Q3 changes from 0 At this time, the current on the transformer T also begins to increase in reverse, which will make the switch tube on the lagging bridge arm unable to achieve ZVS.
  • the dead time Tdt is determined based on the capacitance value of the junction capacitance of each switch tube on the lagging bridge arm in the phase-shifted full-bridge circuit and the primary leakage inductance parameter of the transformer T.
  • the LC resonance time of the primary side of the transformer T determines the maximum dead time Tdtmax, and ensuring that the dead time Tdt is less than the maximum dead time Tdtmax can make the switch on the lagging bridge arm realize ZVS.
  • the DC converter further includes a current transformer CT, an amplifier U1 and a comparator connected in sequence.
  • the input end of the current transformer CT is connected to the transformer T for collecting the current of the transformer T.
  • the amplifier U1 uses It is used to amplify the current of the transformer T collected by the current transformer CT, and output the amplified current signal.
  • the comparator is used to compare the amplified current signal with the current preset value, and the amplified current signal is not greater than outputting a third level when the current preset value, outputting a fourth level when the amplified current signal is greater than the current preset value, and the third level being opposite to the fourth level;
  • the current on the transformer T when judging whether the current on the transformer T is less than the preset current value, the current on the transformer T can be directly detected, and the amplified current signal is compared with the preset current value, and when the comparator outputs The third signal, that is, when the current on the transformer T is less than the preset current, the switch tube on the lagging bridge arm can be turned off, so that the switch tube on the lagging bridge arm can realize ZCS.
  • the method before judging whether the current on the transformer T is not greater than the preset current value after the switch on the leading bridge arm in the phase-shift full-bridge converter is turned off, the method further includes:
  • the switch on the leading bridge arm of the phase-shifted full-bridge circuit is disconnected, it is usually controlled by setting a voltage threshold or directly by current.
  • a current transformer CT, an integrating circuit, an amplifier U1 and an analog-to-digital conversion module are usually set at the output end of the transformer T, so that the current on the transformer T is integrated into a value that lags the current on the transformer T by 90 degrees.
  • FIG. 10 is the prior art.
  • the waveform diagram of the key parameters including the switching period of each switch tube in the phase-shifted full-bridge circuit, the drive signal PWM_Q1 ⁇ Q4, the current of the transformer T and the digital voltage signal, it can be seen that the switch tube on the lagging bridge arm is turned on Until it is turned off, the digital voltage signal is in a monotonically increasing or monotonically decreasing state, so that when the digital voltage signal reaches the preset voltage threshold, the corresponding switch on the leading bridge arm can be controlled to be turned off.
  • the digital voltage signal is close to 0 in the time period between time t0 and time ta, and the resolution is poor.
  • FIG. 11 is another method of controlling the switch tube in the lead bridge arm in the prior art.
  • the waveform diagram of the key parameters when the current on the transformer T reaches the preset current threshold value in the prior art, the corresponding switch tube on the leading bridge arm is controlled to be disconnected. It can be seen from Figure 11 that the current gradually increases between the time t0 and the time t1, and the current decreases after the time t1.
  • the current threshold in the prior art may be calculated based on the output current, output voltage or power of the DC converter.
  • the switch tube on the leading bridge arm is controlled by the current on the transformer T, the current of the transformer T first rises and then falls from the time t0 to the time t4 in FIG.
  • the LC resonant circuit will The current can be controlled from 0 degrees to 90 degrees, and if the DC converter outputs high power, the current on the LC resonant circuit enters the range of 90 degrees to 180 degrees, and the current in the phase from 0 degrees to 90 degrees increases monotonically, 90 degrees.
  • the current in the phase from 180 degrees to 180 degrees decreases monotonically, and the same current value corresponds to two time points.
  • a bias voltage Vref is added to the input positive end of the amplifier U1 of the DC converter, please refer to FIG. 12 , which is a specific structural schematic diagram of a DC converter provided with a current transformer provided by the present invention , wherein the bias voltage Vref can be but is not limited to 1.65V. Please refer to FIG. 13 for the waveform diagram of the current of the transformer T and the digital voltage signal after adding the bias voltage Vref.
  • FIG. 14 is the waveform diagram of the key parameters in another DC converter provided by the present invention. It includes the switching period of each switch tube in the phase-shifted full-bridge circuit, the driving signals PWM_Q1 to Q4, the current of the transformer T and the voltage signal output by the amplifier U1. Due to the addition of the bias voltage Vref, the voltage signal output by the amplifier U1 is positive. voltage, which solves the problem in the prior art that when the voltage signal is less than 0, it cannot be used as the basis for controlling the switch tube on the leading bridge arm.
  • the switch on the leading bridge arm when the switch on the leading bridge arm is controlled to be turned off, it is necessary to judge whether the digital voltage signal reaches the voltage threshold after the corresponding switch on the lagging bridge arm is turned on, and when the threshold is reached, the leading The corresponding switch tube on the bridge arm is disconnected to ensure that the power input to the load can make it work normally.
  • the fourth switch tube Q4 After the fourth switch tube Q4 is turned on, it is judged whether the digital voltage signal reaches the voltage threshold, and when it is judged that the digital voltage signal reaches the voltage threshold, the first switch tube Q1 is controlled to turn off; after the third switch tube Q3 is turned on It is judged whether the voltage signal of the digital quantity reaches the voltage threshold, and when it is judged that the voltage signal of the digital quantity reaches the voltage threshold, the second switch tube Q2 is controlled to be turned off. Since the first switch transistor Q1 and the second switch transistor Q2 are connected to the DC power supply E, the input of the DC power supply E can be stopped after the first switch transistor Q1 and the second switch transistor Q2 are disconnected.
  • FIG. 15 is a schematic structural diagram of a control device for a switch tube provided by the present invention.
  • the device includes:
  • memory 1 for storing computer programs
  • the processor 2 is configured to implement the steps of the above-mentioned control method of the switch tube when executing the computer program.
  • control device for a switch tube For the introduction of the control device for a switch tube provided by the present invention, please refer to the above method embodiments, which will not be repeated in the present invention.
  • the present invention provides a DC converter, including the control device of the switch tube as described above, a DC power supply, a phase-shifted full-bridge circuit, a transformer T and a rectifier circuit provided with an LC resonant circuit connected in sequence, Also includes:
  • the current transformer CT, amplifier U1 and comparator connected in sequence;
  • the input end of the current transformer CT is connected to the transformer T for collecting the current of the transformer T;
  • the amplifier U1 is used to amplify the current of the transformer collected by the current transformer CT, and output the amplified current signal;
  • the comparator is used to compare the amplified current signal with the current preset value, and output a third level when the amplified current signal is not greater than the current preset value, and output when the amplified current signal is greater than the current preset value
  • the fourth level, the third level is opposite to the fourth level.

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Abstract

一种开关管的控制方法、装置及直流变换器，该方案中，在移相全桥变换器中超前桥臂上的开关管断开之后且变压器上的电流不大于电流预设值控制滞后桥臂上相应的开关管断开，能够保证滞后桥臂上的开关管断开时流经该开关管的电流较小，从而实现滞后桥臂上开关管的软关断，降低损耗并提高开关管的抗干扰能力，改善开关管的EMS性能。

Description

一种开关管的控制方法、装置及直流变换器
本申请要求于2021年2月8日提交至中国专利局、申请号为202110182359.5、发明名称为“一种开关管的控制方法、装置及直流变换器”的中国专利申请的优先权，其全部内容通过引用结合在本申请中。
技术领域
本发明涉及电力电子技术领域，特别是涉及一种开关管的控制方法、装置及直流变换器。
背景技术
现有技术中，直流变换器包括直流电源、移相全桥电路、变压器及整流电路，具体结构请参照图1，图1为现有技术中的一种直流变换器的结构示意图，其中，移相全桥电路中超前桥臂中的开关管Q1关断后，控制滞后桥臂中相应的开关管Q4关断；移相全桥电路中超前桥臂中的开关管Q2关断后，控制滞后桥臂中相应的开关管Q3关断。但是，移相全桥电流中滞后桥臂上的开关管通常工作在硬开关状态，即控制开关管关断时流经开关管的电流仍旧很大，从而使开关管的损耗较大，此外，开关管工作在硬开关状态时开关管的抗干扰能力较低，也即会产生EMC(Electromagnetic Compatibility，电磁兼容性)噪声。
发明内容
本发明的目的是提供一种开关管的控制方法、装置及直流变换器，能够保证滞后桥臂上的开关管断开时流经该开关管的电流较小，从而实现滞后桥臂上开关管的软关断，降低损耗并提高开关管的抗干扰能力，改善开关管的EMS性能。
为解决上述技术问题，本发明提供了一种开关管的控制方法，应用于直流变换器，所述直流变换器包括直流电源、移相全桥电路、变压器和设 有LC谐振电路的整流电路，包括：
在移相全桥变换器中超前桥臂上的开关管断开之后判断所述变压器上的电流是否不大于电流预设值；
若是，则控制所述移相全桥变换器中滞后桥臂上相应的开关管关断。
优选地，所述直流变换器还包括依次连接的电流互感器、积分电路、放大器及比较器，所述电流互感器的输入端与所述变压器连接，所述电流互感器用于采集所述变压器上的电流，所述积分电路对所述变压器上的电流进行积分，得到电压信号，所述放大器的输入正端设有偏置电压，用于将所述电压信号进行放大处理并加入偏置电压，输出放大后的电压信号，所述比较器用于将放大后的电压信号与最大积分门限及最小积分门限比较，并当所述放大后的电压信号大于所述最大积分门限或小于所述最小积分门限时输出第一电平，当所述放大后的电压信号在所述最大积分门限及所述最小积分门限之间时输出第二电平，所述第一电平与所述第二电平相反；
在移相全桥变换器中超前桥臂上的开关管断开之后判断变压器上的电流是否不大于电流预设值，包括：
将所述移相全桥变换器的滞后桥臂上的开关管导通时刻作为积分开始时刻控制所述积分电路对所述电流互感器采集到的所述变压器上的电流进行积分，得到所述电压信号；
将所述比较器输出所述第二电平时的时刻作为积分结束时刻；
将从上述积分开始时刻至所述积分结束时刻之间的时间作为积分时间；
基于所述积分时间及所述LC谐振电路的谐振参数确定移相角时间；
在所述移相全桥变换器中超前桥臂上的开关管关断之后以所述超前桥臂上的开关管关断时作为开始时间判断是否到达所述移相角时间，以实现判断变压器上的电流是否不大于电流预设值；
若是，则进入控制所述移相全桥变换器中滞后桥臂上相应的开关管关断的步骤。
优选地，基于所述积分时间及所述LC谐振电路的谐振参数确定移相 角时间之后，还包括：
确定最大移相角时间；
判断所述移相角时间是否小于所述最大移相角时间；
若小于，则进入在所述移相全桥变换器中超前桥臂上的开关管关断之后以所述超前桥臂上的开关管关断时作为开始时间判断是否到达所述移相角时间的步骤；
若不小于，则在所述移相全桥变换器中超前桥臂上的开关管关断之后以所述超前桥臂上的开关管关断时作为开始时间判断是否到达所述最大移相角时间；
若到达，则进入控制所述移相全桥变换器中滞后桥臂上相应的开关管关断的步骤。
优选地，确定最大移相角时间，包括：
基于所述移相全桥电路中各个开关管的结电容的容值、所述变压器的原边漏感参数、所述LC谐振电路的谐振参数、所述变压器的输出电压及输出电流及所述积分时间确定所述最大移相角时间。
优选地，控制所述移相全桥变换器中滞后桥臂上相应的开关管关断之后，还包括：
将所述移相全桥变换器中滞后桥臂上相应的开关管关断时的时刻作开始时刻判断是否到达预设死区时间；
若是，则控制所述移相全桥变换器中滞后桥臂上相应的另一开关管导通。
优选地，将所述移相全桥变换器中滞后桥臂上相应的开关管关断时的时刻作开始时刻判断是否到达预设死区时间之前，还包括：
基于所述移相全桥电路中滞后桥臂上各个开关管的结电容的容值及所述变压器的原边漏感参数确定所述变压器原边的LC谐振时间；
基于所述变压器原边的LC谐振时间确定最大死区时间；
基于所述最大死区时间设定所述预设死区时间，所述预设死区时间小于所述最大死区时间；
进入将所述移相全桥变换器中滞后桥臂上相应的开关管关断时的时刻 作开始时刻判断是否到达预设死区时间的步骤。
优选地，所述直流变换器还包括依次连接的电流互感器、放大器及比较器，所述电流互感器的输入端与所述变压器连接，用于采集所述变压器的电流，所述放大器用于对所述电流互感器采集到的所述变压器的电流进行放大处理，输出放大后的电流信号，所述比较器用于将所述放大后的电流信号与所述电流预设值进行比较，并在所述放大后的电流信号不大于所述电流预设值时输出第三电平，在所述放大后的电流信号大于所述电流预设值时输出第四电平，所述第三电平与所述第四电平相反；
在移相全桥变换器中超前桥臂上的开关管断开之后判断变压器上的电流是否不大于电流预设值，包括：
在移相全桥变换器中超前桥臂上的开关管断开之后判断所述比较器输出的信号是否为所述第三电平；
若是，则进入控制所述移相全桥变换器中滞后桥臂上相应的开关管关断的步骤。
优选地，在移相全桥变换器中超前桥臂上的开关管断开之后判断变压器上的电流是否不大于电流预设值之前，还包括：
基于所述直流变换器的输出端连接的负载确定所述变压器上的期望电流值；
所述滞后桥臂上相应的开关管导通后判断所述变压器上的电流是否到达所述期望电流值；
若是，则控制所述移相全桥电路中超前桥臂上相应的开关管断开；
进入在移相全桥变换器中超前桥臂上的开关管断开之后判断变压器上的电流是否不大于电流预设值的步骤。
为解决上述技术问题，本发明提供了一种开关管的控制装置，包括：
存储器，用于存储计算机程序；
处理器，用于执行所述计算机程序时实现如上述所述开关管的控制方法的步骤。
为解决上述技术问题，本发明提供了一种直流变换器，包括如上述所述的开关管的控制装置和依次连接的直流电源、移相全桥电路、变压器及 设有LC谐振电路的整流电路，还包括：
依次连接的电流互感器、放大器及比较器；
所述电流互感器的输入端与所述变压器连接，用于采集所述变压器的电流；
所述放大器用于对所述电流互感器采集到的所述变压器的电流进行放大处理，输出放大后的电流信号；
所述比较器用于将所述放大后的电流信号与所述电流预设值进行比较，并在所述放大后的电流信号不大于所述电流预设值时输出第三电平，在所述放大后的电流信号大于所述电流预设值时输出第四电平，所述第三电平与所述第四电平相反。
本申请提供了一种开关管的控制方法、装置及直流变换器，该方案中，在移相全桥变换器中超前桥臂上的开关管断开之后且变压器上的电流不大于电流预设值控制滞后桥臂上相应的开关管断开，能够保证滞后桥臂上的开关管断开时流经该开关管的电流较小，从而实现滞后桥臂上开关管的软关断，降低损耗并提高开关管的抗干扰能力，改善开关管的EMS性能。
附图说明
为了更清楚地说明本发明实施例中的技术方案，下面将对现有技术和实施例中所需要使用的附图作简单地介绍，显而易见地，下面描述中的附图仅仅是本发明的一些实施例，对于本领域普通技术人员来讲，在不付出创造性劳动的前提下，还可以根据这些附图获得其他的附图。
图1为现有技术中的一种直流变换器的结构示意图；
图2为本发明提供的一种开关管的控制方法的流程示意图；
图3为本发明提供的一种直流变换器的具体的结构示意图；
图4为本发明提供的另一种直流变换器的具体的结构示意图；
图5为现有技术中控制移相全桥电路中开关管导通与关断的流程图；
图6为本发明提供的一种直流变换器中关键参数的波形图；
图7为本发明提供的一种关于电流预设值的波形图；
图8为本发明提供的一种关于积分门限值的波形图；
图9为本发明提供的一种关于开关管上的电压的波形图；
图10为现有技术中对超前桥臂中的开关管进行控制时关键参数的波形图；
图11为现有技术中另一种对超前桥臂中的开关管进行控制时关键参数的波形图；
图12为本发明提供的一种设有电流互感器的直流变换器的具体的结构示意图；
图13为本发明提供的一种加入偏置电压后的变压器的电流和数字量的电压信号的波形图；
图14为本发明提供的另一种直流变换器中关键参数的波形图；
图15为本发明提供的一种开关管的控制装置的结构示意图。
具体实施方式
本发明的核心是提供一种开关管的控制方法、装置及直流变换器，能够保证滞后桥臂上的开关管断开时流经该开关管的电流较小，从而实现滞后桥臂上开关管的软关断，降低损耗并提高开关管的抗干扰能力，改善开关管的EMS性能。
为使本发明实施例的目的、技术方案和优点更加清楚，下面将结合本发明实施例中的附图，对本发明实施例中的技术方案进行清楚、完整地描述，显然，所描述的实施例是本发明一部分实施例，而不是全部的实施例。基于本发明中的实施例，本领域普通技术人员在没有做出创造性劳动前提下所获得的所有其他实施例，都属于本发明保护的范围。
请参照图2，图2为本发明提供的一种开关管的控制方法的流程示意图。
该方法应用于直流变换器，直流变换器包括直流电源、移相全桥电路、变压器T和设有LC谐振电路的整流电路，该方法包括：
S11：在移相全桥变换器中超前桥臂上的开关管断开之后判断变压器T上的电流是否不大于电流预设值，并在判定不大于时进入S12；
现有技术中的直流变换器包括直流电源E、移相全桥电路、变压器T及整流电路，但是，如图1所示，现有技术中的直流变换器中的整流电路中设 有四个二极管D5、D6、D7及D8和一个输出滤波电感L2，在变压器T的原边电流从方向为正，下冲到方向为负，或从方向为负，下冲到方向为正时，输出滤波电感L2的电流无法突变，从而使输出滤波电感L2续流时和二极管将变压器T短路，具体地，输出滤波电感L2、负载、二极管D5和二极管D6构成一个回路，或者输出滤波电感L2、负载、二极管D7和二极管D8构成一个回路，从而使变压器T的副边电压为0，也即使变压器T的原边电压为0，直流电源E输出的电压全部加在了变压器T原边的原边谐振电感L1上，导致变压器T的输出占空比丢失，无法满足负载正常的用电需求。
为了解决上述问题，本申请中的整流电路中设有LC谐振电路，如图3所示，图3为本发明提供的一种直流变换器的具体的结构示意图，副边的谐振电感Lrs能够谐振，且副边的谐振电感C5及C6能够在变压器T上的电流的方向变化时进行储能及放电，以保证变压器T能够正常输出电能至负载，以避免变压器T的输出占空比丢失，从而保证变压器T副边的电压不为0，为后续对移相全桥电路中的开关管进行控制时提供条件。当然，由于原边谐振电感Lrp和各个开关管两端并联的电容可以谐振，因此，为了节省成本，图3中的原边谐振电容C7也可以省略，请参照图4，图4为本发明提供的另一种直流变换器的具体的结构示意图。
现有技术中对移相全桥电路中的开关管进行控制时，通常控制同一桥臂中的两个开关管错开导通，具体地，各个开关管导通与关断的过程如图5所示，图5为现有技术中控制移相全桥电路中开关管导通与关断的流程图，可见，同一桥臂中的两个开关管的动作之间相隔死区时间，即第一开关管Q1关断且经过死区时间后再控制第二开关管Q2导通，第二开关管Q2关断且经过死区时间后再控制第一开关管Q1导通，第三开关管Q3关断且经过死区时间后再控制第四开关管Q4导通，第四开关管Q4关断且经过死区时间后再控制第三开关管Q3导通，不同桥臂上交叉的两个开关管的导通时间及关断时间相隔移相角时间，即第一开关管Q1导通且经过移相角时间后再控制第四开关管Q4导通，第一开关管Q1关断且经过移相角时间后再控制第四开关管Q4关断，第二开关管Q2导通且经过移相角时间后再控制第三开关管Q3导通，第二开关管Q2关断且经过移相角时间后再控制第三开关管Q3关 断。综上，第一开关管Q1及第二开关管Q2所在的桥臂为超前桥臂，第三开关管Q3及第四开关管Q4所在的桥臂为滞后桥臂。由于变压器T的原边设有原边谐振电容C7及原边谐振电感L1，请参照图1，所以能够保证超前桥臂上第一开关管Q1及第二开关管Q2实现零电压导通或零电压关断，即在第一开关管Q1或第二开关管Q2两端电压满足零电压关断的条件时将其导通或关断。然而，移相角时间是预先设定且不变的，当到达移相角时间便控制滞后桥臂上的开关管导通或关断，但是，若此时流经开关管的电流较大，例如，当直流变换器轻载时，变压器T原边的原边谐振电感L1所储存的能量无法将滞后桥臂上开关管的漏源电容电荷抽完，从而导致流过开关管两端的电流较大，则会导致开关管断开时的损耗也更大，也即滞后桥臂上的开关管工作在硬开关状态，开关管工作在硬开关状态时，EMC性能较差，开关管的抗干扰能力较低。
为了解决上述问题，本申请在移相全桥变换器中超前桥臂上的开关管断开之后判断变压器T上的电流是否不大于电流预设值，而不是仅基于移相角时间对开关管进行控制，此外，本申请对判断变压器T上的电流是否不大于电流预设值的方法不作限定。
S12：控制移相全桥变换器中滞后桥臂上相应的开关管关断。
通过保证变压器T上的电流不大于电流预设值时将滞后桥臂上的开关管断开，即可保证在流经滞后桥臂上的开关管的电流较小时将滞后桥臂上的开关管断开，从而实现滞后桥臂上的开关管的软关断，以实现减小开关管的损耗。
具体地，第一开关管Q1断开且流过第四开关管Q4的电流较小时再控制第四开关管Q4断开，第二开关管Q2断开且流过第三开关管Q3的电流较小时再控制第三开关管Q3断开。
此外，本申请中变压器T上的电流小于电流预设值时，将滞后桥臂上的开关管断开，变压器T上的电流也随之改变方向，从0开始增大或减小，因此，还能够保证变压器T副边的整流电路中的整流二极管软关断，以减小整流二极管的损耗，提升效率，并改善整流二极管的EMS性能。
本实施例中，可以根据直流变换器的期望输出功率确定移相角时间， 从而使在到达移相角时间时，流经滞后桥臂上的开关管的电流也已满足软关断的条件；或结合变压器T上的电流和电流预设值之间的关系控制开关管，以实现变频控制，从而可以在低功率下降低开关管的开关频率，既有利于软开关控制，又降低开关损耗，以达到低功率下更优的效率和EMC性能。原边和副边的双谐振移相全桥控制，可以有效利用变压器T原边和副边的两个谐振点，从而基于最佳谐振频率对开关管进行控制，实现最优软开关控制。
各个开关管具体的导通与关断的过程及各个电流的波形图请参照图6，图6为本发明提供的一种直流变换器中关键参数的波形图。图中包括移相全桥电路中各个开关管的驱动信号PWM_Q1～Q4，变压器T的原边电流、变压器T原边的励磁电压、变压器T原边的励磁电流、变压器T的漏感电压、副边电容的电压及副边整流二极管的电流，图中的开关周期即为从本次第一开关管Q1导通至下次第一开关管Q1导通之间的时间。四个开关管的控制周期分为12个过程，即t0-t1，t1-t2，t2-t3，t3-t4，t4-t5，t5-t6，t6-t7，t7-t8，t8-t9，t9-t10，t10-t11，t11-t0。
在t0-t1时间段内，第一开关管Q1与第四开关管Q4共同处于导通状态，此时变压器T原边的励磁电流会先负向减小再正向增大，变压器T的原边电流从0开始逐步增大，变压器T副边的第二谐振电容C6此时为充电电容，第二谐振电容C6两端的电压逐步上升，第一谐振电容C5放电，第一谐振电容C5两端的电压在逐渐下降，第二二极管D6导通，流过第二二极管D6的电流先上升然后逐步下降。
t1-t2时间段内，在t1时刻第一开关管Q1关断，第四开关管Q4导通，第一开关管Q1两端并联的第一电容C1充电，第二开关管Q2两端并联的第二电容C2放电，变压器T原边的励磁电流正向缓慢增大，原边电流开始减小，变压器T副边的第二谐振电容C6作为充电电容，第二谐振电容C6两端的电压继续逐步上升，第一谐振电容C5放电，第一谐振电容C5两端的电压逐渐下降，第二二极管D6输出电流，并逐步下降。
t2-t3时间段内，此时第一开关管Q1处于关断状态，第四开关管Q4继续导通，原边当第一开关管Q1两端并联的第一电容C1充满，第二开关管Q2 两端并联的第二电容C2放完电，即VC1＝Vin，此时第二电容C2的体二极管D2续流导通，为第二开关管Q2的零电压导通创造条件，变压器T原边的励磁电流正向缓慢增大，原边电流继续减小，副边第二谐振电容C6仍为充电电容，第二谐振电容C6两端的电压缓慢上升，第一谐振电容C5继续放电，第一谐振电容C5两端的电压逐渐下降，第二二极管D6输出电流，并逐步下降。
t3-t4时间段内，在t3时刻第二开关管Q2导通，第四开关管Q4仍处于导通状态，t1-t3时间段为第一开关管Q1及第二开关管Q2之间的死区时间，第二开关管Q2为零电压导通，此时变压器T原边的励磁电流正向缓慢增大，原边电流继续减小，副边第二谐振电容C6仍为充电电容，第二谐振电容C6两端的电压缓慢上升，第一谐振电容C5继续放电，第一谐振电容C5两端的电压逐渐下降，第二二极管D6输出电流，并逐步下降。
t4-t5时间段内，在t4时刻第四开关管Q4关断，第二开关管Q2继续导通，第四开关管Q4实现零电压关断(第四开关管Q4两端的电压从0开始上升，同时，变压器T上的电流也接近0，即小于电流预设值，第四开关管Q4也就近似零电流关断)，此过程中第四开关管Q4两端并联的第四电容C4充电，第三开关管Q3两端并联的第三电容C3放电，此时变压器T原边的励磁电流正向缓慢增大，原边电流继续减小，副边第二谐振电容C6仍为充电电容，第二谐振电容C6两端的电压缓慢上升，第一谐振电容C5继续放电，第一谐振电容C5两端的电压逐渐下降，第二二极管D6输出电流，并逐步下降。
t5-t6时间段内，第二开关管Q2处于导通状态，第四开关管Q4处于关断状态，此时变压器T原边的励磁电流正向缓慢增大，原边电流继续减小至近似到0，此过程第四开关管Q4两端并联的第四电容C4充满电，即VC4＝Vin，第三开关管Q3两端并联的第三电容C3放完电，当第三开关管Q3两端并联的第三电容C3的电压为0时，第三开关管Q3的体二极管D3会打开，为第三开关管Q3实现零电压导通创造条件，副边第二谐振电容C6仍为充电电容，第二谐振电容C6两端的电压缓慢上升，第一谐振电容C5继续放电，第一谐振电容C5两端的电压逐渐下降，第二二极管D6输出电流，并逐步下降到0。
t6-t7时间段内，第二开关管Q2及第三开关管Q3均处于导通状态，此时变压器T原边的励磁电流会先正向减小再反向增大(定义励磁电流方向由上到下为正向)，原边电流反向增大，副边第一谐振电容C5为充电电容，第一谐振电容C5两端的电压逐步上升，第二谐振电容C6放电，第二谐振电容C6两端的电压逐渐下降，第一二极管D5输出电流，并逐步下降。
t7-t8时间段内，在t7时刻第二开关管Q2关断，第三开关管Q3导通，第二开关管Q2两端并联的第二电容C2充电，第一开关管Q1两端并联的第一电容C1放电，变压器T原边电流反向缓慢增大，原边励磁电流开始反向减小，变压器T副边的第一谐振电容C5作为充电电容，第一谐振电容C5两端的电压继续逐步上升，第二谐振电容C6放电，第二谐振电容C6两端的电压逐渐下降，第一二极管D5输出电流，并逐步下降。
t8-t9时间段内，此时第二开关管Q2处于关断状态，第三开关管Q3继续导通，原边当第二开关管Q2两端并联的第二电容C2充满，第一开关管Q1两端并联的第一电容C1放完电，即VC2＝Vin，此时第一电容C1的体二极管D1续流导通，为第一开关管Q1的零电压导通创造条件，变压器T原边的励磁电流反向缓慢增大，原边电流继续反向减小，副边第一谐振电容C5仍为充电电容，第一谐振电容C5两端的电压缓慢上升，第二谐振电容C6继续放电，第二谐振电容C6两端的电压逐渐下降，第一二极管D5输出电流，并逐步下降。
t9-t10时间段内，此时刻第一开关管Q1导通，第一开关管Q1和第三开关管Q3均处于导通状态，变压器T原边的励磁电流反向增大，由于第一开关管Q1的体二极管D1续流导通，第一开关管Q1即为零电压导通，此时励磁电流反向缓慢增大，原边电流继续反向减小，副边第一谐振电容C5仍为充电电容，第一谐振电容C5两端的电压缓慢上升，第二谐振电容C6放电，第二谐振电容C6两端的电压在逐渐下降，第一二极管D5输出电流，并逐步下降。
t10-t11时间段内，在t10时刻第三开关管Q3关断，第一开关管Q1继续导通，第三开关管Q3实现零电压关断(第三开关管Q3两端的电压从0开始上升，同时，变压器T上的电流也接近0，即小于电流预设值，第三开关管 Q3也就近似零电流关断)，此过程中第四开关管Q4两端并联的第四电容C4放电，第三开关管Q3两端并联的第三电容C3充电，此时变压器T原边的励磁电流反向缓慢增大，原边电流继续反向减小，副边第一谐振电容C5仍为充电电容，第一谐振电容C5两端的电压缓慢上升，第二谐振电容C6继续放电，第二谐振电容C6两端的电压逐渐下降，第一二极管输出电流，并逐步下降。
t11-t0时间段内，第一开关管Q1处于导通状态，第三开关管Q3处于关断状态，此时变压器T原边的励磁电流反向缓慢增大，原边电流继续减小至近似到0，此过程第三开关管Q3两端并联的第三电容C3充满电，即VC3＝Vin，第四开关管Q4两端并联的第四电容C4放完电，当第四开关管Q4两端并联的第四电容C4的电压为0时，第四开关管Q4的体二极管D4会导通，为第四开关管Q4实现零电压导通创造条件，副边第一谐振电容C5仍为充电电容，第一谐振电容C5两端的电压缓慢上升，第二谐振电容C6继续放电，第二谐振电容C6两端的电压逐渐下降，主功率电流也逐渐减小到零，对第一谐振电容C5的充电及第二谐振电容C6的放电逐渐完成。
需要说明的是，本申请中移相全桥电流中各个开关管两端并联的电容和二极管可以为各个开关管自身的结电容和体二极管，本申请对此不做限定。
综上，在移相全桥变换器中超前桥臂上的开关管断开之后且变压器上的电流不大于电流预设值控制滞后桥臂上相应的开关管断开，能够保证滞后桥臂上的开关管断开时流经该开关管的电流较小，从而实现滞后桥臂上开关管的软关断，降低损耗并提高开关管的抗干扰能力，改善开关管的EMS性能。
在上述实施例的基础上：
作为一种优选的实施例，直流变换器还包括依次连接的电流互感器CT、积分电路、放大器U1及比较器，电流互感器CT的输入端与变压器T连接，电流互感器CT用于采集变压器T上的电流，积分电路对变压器T上的电流进行积分，得到电压信号，放大器U1的输入正端设有偏置电压Vref， 用于将电压信号进行放大处理并加入偏置电压Vref，输出放大后的电压信号，比较器用于将放大后的电压信号与最大积分门限及最小积分门限比较，并当放大后的电压信号大于最大积分门限或小于最小积分门限时输出第一电平，当放大后的电压信号在最大积分门限及最小积分门限之间时输出第二电平，第一电平与第二电平相反；
在移相全桥变换器中超前桥臂上的开关管断开之后判断变压器T上的电流是否不大于电流预设值，包括：
将移相全桥变换器的滞后桥臂上的开关管导通时刻作为积分开始时刻控制积分电路对电流互感器CT采集到的变压器T上的电流进行积分，得到电压信号；
将比较器输出第二电平时的时刻作为积分结束时刻；
将从上述积分开始时刻至积分结束时刻之间的时间作为积分时间；
基于积分时间及LC谐振电路的谐振参数确定移相角时间；
在移相全桥变换器中超前桥臂上的开关管关断之后以超前桥臂上的开关管关断时作为开始时间判断是否到达移相角时间，以实现判断变压器T上的电流是否不大于电流预设值；
若是，则进入控制移相全桥变换器中滞后桥臂上相应的开关管关断的步骤。
本实施例中，在判断变压器T上的电流是否小于电流预设值时，可以通过计算移相角时间，从而实现在以超前桥臂上的开关管断开为起始时间，到该移相角时间时，变压器T上的电流已小于电流预设值，其中，本实施例中的移相角时间不是固定的，而是基于变压器T上的电流实时计算而得。
具体地，基于电流预设值设定对应的电压阈值Vpc，从而可以通过电压阈值Vpc和偏置电压Vref获取最大积分门限及最小积分门限，其中，最大积分门限Vth1＝偏置电压Vref+电压阈值Vpc，最小积分门限Vth2＝偏置电压Vref-电压阈值Vpc，积分电路从滞后桥臂上的开关管导通时刻为积分开始时刻对变压器T上的电流进行积分，也即以超前桥臂上的开关管及滞后桥臂上相应的开关管同时闭合时为积分开始时刻，放大器U1将积分电路在把每一个时刻积分得到的电压值加上偏置电压Vref后进行放大，比较器对放 大器U1输出的每一个时刻的放大后的电压信号与最大积分门限Vth1和最小积分门限Vth2进行比较，当放大后的电压信号大于最大积分门限或小于最小积分门限时输出第一电平，当放大后的电压信号在最大积分门限及最小积分门限之间时输出第二电平，并将比较器输出第一电平时作为积分结束时刻，从而能够确定积分时间。需要说明的是，本申请中的电压阈值Vpc＝func(I，a，R，T，Rc，C)，其中，i为变压器T上的电流，a为电流互感器CT的比例系数，R为电流互感器CT的电阻，Rc为积分电流中的电阻，C为积分电流中的电容，T为上述计算出的积分时间。请参照图7，图7为本发明提供的一种关于电流预设值的波形图，图中的Izcs即为电流预设值，t0-t4之间的时间即为积分时间。请参照图8，图8为本发明提供的一种关于积分门限值的波形图，图中的Vpc1即为最大积分门限，Vpc2即为最小积分门限，偏置电压Vref为1.65V。计算出积分时间T后，基于积分时间T及LC谐振电路的谐振参数确定移相角时间，具体地，移相角时间Tps＝Ts/2-T，Ts为开关周期时间，谐振角频率w根据LC谐振电路的谐振参数确定。
综上，通过设定与电流预设值对应的电压阈值Vpc，能够计算出移相角时间，从而使滞后桥臂上的开关管在超前桥臂上相应的开关管关断后并到达移相角时间时关断，即能够实现滞后桥臂上开关管的软关断。
作为一种优选的实施例，基于积分时间及LC谐振电路的谐振参数确定移相角时间之后，还包括：
确定最大移相角时间；
判断移相角时间是否小于最大移相角时间；
若小于，则进入在移相全桥变换器中超前桥臂上的开关管关断之后以超前桥臂上的开关管关断时作为开始时间判断是否到达移相角时间的步骤；
若不小于，则在移相全桥变换器中超前桥臂上的开关管关断之后以超前桥臂上的开关管关断时作为开始时间判断是否到达最大移相角时间；
若到达，则进入控制移相全桥变换器中滞后桥臂上相应的开关管关断的步骤。
请参照图9，图9为本发明提供的一种关于开关管上的电压的波形图。本实施例中，申请人考虑到在图9中的t4时刻，滞后桥臂上的第四开关管Q4零电流关断，即实现ZCS(Zero Current Switch，零电流关断)，t6时刻滞后桥臂上的第三开关管Q3零电压开通，即实现ZVS。其中，在t6时刻，第三开关管Q3两端的电压Vds_Q3在最底端，这时候将第三开关管Q3开通，可以实现ZVS(Zero Voltage Switching，零电压开通)。当ZCS点很接近ZVS点的时候会影响ZVS的实现，本发明提出设计可变的移相角时间Tps来消除ZCS对ZVS的影响，且移相角时间Tps必须保证在小于一个阈值，也即小于最大移相角时间Tpsmax，最大移相角时间Tpsmax可以根据Tpsmax＝func(w,Uc,Io,n,C,Lr,tonv)计算得出，其中，谐振角频率w根据LC谐振电路的谐振参数确定，Uc为直流变换器的输入电压，Io为变压器T上的电流，n为变压器T原边和副边的匝数比，C为移相全桥电路中各个开关管的结电容或者在各个开关管两端并联的电容，Lr为变压器T原边的漏感参数，tonv为超前桥臂上的开关管和滞后桥臂上相应的开关管同时导通的时间。如果死区时间，也即t4-t6这段时间不变，当移相角时间Tps>Tpsmax时，由于变压器T原边的漏感电流小，谐振能量不够，第三开关管Q3两端的电压Vds_Q3或第四开关管Q4两端的电压Vds_Q4可能只能谐振到t5处，开关管两端的电压太高，也即无法实现ZVS。因此，最大移相角时间是能够使滞后桥臂上的开关管实现ZVS的临界条件。
为了解决上述技术问题，本申请在确定了移相角时间之后，还需将移相角时间和最大移相角时间进行比较，当移相角时间大于最大移相角时间时，将最大移相角时间作为移相角时间对滞后桥臂上的开关管进行控制；当移相角时间小于最大移相角时间时，不对移相角时间进行改变，基于确定的移相角时间对滞后桥臂上的开关管进行控制。
综上，本实施例中的方法不仅能够使滞后桥臂上的开关管实现ZCS，也能够保证滞后桥臂上的开关管实现ZVS。
作为一种优选的实施例，确定最大移相角时间，包括：
基于移相全桥电路中各个开关管的结电容的容值、变压器T的原边漏感参数、LC谐振电路的谐振参数、变压器T的输出电压及输出电流及积分 时间确定最大移相角时间。
本实施例中，在确定最大移相角时间时，可以通过根据Tpsmax＝func(w,Uc,Io,n,C,Lr,tonv)计算得出，其中，谐振角频率w根据LC谐振电路的谐振参数确定，Uc为直流变换器的输入电压，Io为变压器T上的电流，n为变压器T原边和副边的匝数比，C为移相全桥电路中各个开关管的结电容或者在各个开关管两端并联的电容，Lr为变压器T原边的漏感参数，tonv为超前桥臂上的开关管和滞后桥臂上相应的开关管同时导通的时间。确定最大移相角时间后，不仅能够使滞后桥臂上的开关管实现ZCS，也能够保证滞后桥臂上的开关管实现ZVS。
作为一种优选的实施例，控制移相全桥变换器中滞后桥臂上相应的开关管关断之后，还包括：
将移相全桥变换器中滞后桥臂上相应的开关管关断时的时刻作开始时刻判断是否到达预设死区时间；
若是，则控制移相全桥变换器中滞后桥臂上相应的另一开关管导通。
为了保证使均在滞后桥臂上的开关管错开导通，也即使滞后桥臂上的两个开关管无同时导通的时间，在处于导通状态的开关管关断之后，经过预设死区时间再控制滞后桥臂上的另一开关管导通，即在第三开关管Q3关断且经过预设死区时间后再控制第四开关管Q4导通，在第四开关管Q4关断且经过预设死区时间后再控制第三开关管Q3导通，避免滞后桥臂上的开关管同时导通而导致短路，使直流变换器无法正常工作。
此外，当预设死区时间较短时，可以保证滞后桥臂上的开关管无法参与续流，也即能够保证滞后桥臂上的开关管工作在软开关状态。
作为一种优选的实施例，将移相全桥变换器中滞后桥臂上相应的开关管关断时的时刻作开始时刻判断是否到达预设死区时间之前，还包括：
基于移相全桥电路中滞后桥臂上各个开关管的结电容的容值及变压器T的原边漏感参数确定变压器T原边的LC谐振时间；
基于变压器T原边的LC谐振时间确定最大死区时间；
基于最大死区时间设定预设死区时间，预设死区时间小于最大死区时间；
进入将移相全桥变换器中滞后桥臂上相应的开关管关断时的时刻作开始时刻判断是否到达预设死区时间的步骤。
本实施例中，申请人考虑到若死区时间Tdt大于t4-t6之间的时间，在t6时，从图9中可以看出，在虚线处，第三开关管Q3两端的电压Vds_Q3从0开始增大，此时变压器T上的电流也开始反向增大，会使滞后桥臂上的开关管无法实现ZVS。
为解决上述技术问题，本申请在对死区时间Tdt进行设定前，先基于移相全桥电路中滞后桥臂上各个开关管的结电容的容值及变压器T的原边漏感参数确定变压器T原边的LC谐振时间，从而确定最大死区时间Tdtmax，保证死区时间Tdt小于最大死区时间Tdtmax即可使滞后桥臂上的开关管实现ZVS。
作为一种优选的实施例，直流变换器还包括依次连接的电流互感器CT、放大器U1及比较器，电流互感器CT的输入端与变压器T连接，用于采集变压器T的电流，放大器U1用于对电流互感器CT采集到的变压器T的电流进行放大处理，输出放大后的电流信号，比较器用于将放大后的电流信号与电流预设值进行比较，并在放大后的电流信号不大于电流预设值时输出第三电平，在放大后的电流信号大于电流预设值时输出第四电平，第三电平与第四电平相反；
在移相全桥变换器中超前桥臂上的开关管断开之后判断变压器T上的电流是否不大于电流预设值，包括：
在移相全桥变换器中超前桥臂上的开关管断开之后判断比较器输出的信号是否为第三电平；
若是，则进入控制移相全桥变换器中滞后桥臂上相应的开关管关断的步骤。
本实施例中，在判断变压器T上的电流是否小于电流预设值时，可以通过直接对变压器T上的电流进行检测，将放大后的电流信号与电流预设值进行比较，当比较器输出第三信号，也即变压器T上的电流小于预设电流时，即可对滞后桥臂上的开关管进行关断控制，以使滞后桥臂上的开关管实现ZCS。
作为一种优选的实施例，在移相全桥变换器中超前桥臂上的开关管断开之后判断变压器T上的电流是否不大于电流预设值之前，还包括：
基于直流变换器的输出端连接的负载确定变压器T上的期望电流值；
滞后桥臂上相应的开关管导通后判断变压器T上的电流是否到达期望电流值；
若是，则控制移相全桥电路中超前桥臂上相应的开关管断开；
进入在移相全桥变换器中超前桥臂上的开关管断开之后判断变压器T上的电流是否不大于电流预设值的步骤。
申请人考虑到现有技术中对移相全桥电路中超前桥臂上的开关管断开时，通常通过设定电压阈值或直接通过电流进行控制。具体地，现有技术中通常在变压器T的输出端设置电流互感器CT、积分电路、放大器U1及模数转换模块，从而将变压器T上的电流积分为滞后于变压器T上的电流90度的电压信号，经放大器U1放大并进行模数转换后的得到的数字量的电压信号与电流信号的波形图请参照图10，图10为现有技术中对超前桥臂中的开关管进行控制时关键参数的波形图，其中包括移相全桥电路中各个开关管的开关周期、驱动信号PWM_Q1～Q4、变压器T的电流及数字量的电压信号，可见，从滞后桥臂上的开关管导通至其关断，数字量的电压信号均呈单调递增或单调递减的状态，从而能够在数字量的电压信号到达预设电压阈值时控制超前桥臂上相应的开关管关断，但是，从图10可知，数字量的电压信号在t0时刻至ta时刻之间的时间段内接近于0，分辨率较差。而通过电路进行控制也即直接通过变压器T上的电流对超前桥臂上的开关管进行控制，请参照图11，图11为现有技术中另一种对超前桥臂中的开关管进行控制时关键参数的波形图，当变压器T上的电流到达预先设定的现有技术中的电流阈值时，控制超前桥臂上相应的开关管断开。从图11可知，t0时刻至t1时刻之间电流逐步上升，而t1时刻后电流下降，由于t1时刻至t4时刻之间电流在减小，但变压器T实际输出的电流在增加，电流的方向是反向的，因此，若通过t1时刻至t4时刻之间变压器T上的电流对超前桥臂上的开关管进行控制，控制过程会非常复杂，因此，通常只能对t0-tb这一个范围做检测控制。其中，现有技术中的电流阈值可以基于直流变换器的输出电流、 输出电压或功率计算得出。此外，由于通过变压器T上的电流对超前桥臂上的开关管进行控制时，图11中从t0时刻至t4时刻变压器T的电流先上升后下降，从t4时刻至t10时刻变压器T的电流信号线下降再上升，若仅通过变压器T的电流信号对超前桥臂中的开关管进行控制，由于变压器T上的电流是正弦电流，所以在直流变换器输出中低功率时，LC谐振电路上的电流可以控制在0度至90度，而若直流变换器输出高功率时，LC谐振电路上的电流进入90度至180度的范围，0度至90度的相位内的电流呈单调递增，90度至180度的相位内的电流呈单调递减，同一电流值对应了两个时间点，而若在错误的时间点对超前桥臂中的开关管进行控制，会导致直流变换器输出的电流与期望电流存在偏差，也即会导致对直流变换器的最大功率跟踪出现偏差，而导致直流变换器不断重启，系统无法正常工作。
本实施例中，考虑到直流变换器输出功率至负载上时，负载的承受能力是有限的，因此，为了避免直流变换器的输出功率过大，从而导致无法正常工作，且为了解决上述技术问题，本实施例中直流变换器的放大器U1的输入正端加入了偏置电压Vref，请参照图12，图12为本发明提供的一种设有电流互感器的直流变换器的具体的结构示意图，其中，偏置电压Vref可以但不限定为1.65V，加入偏置电压Vref后变压器T的电流和数字量的电压信号的波形图请参照图13，图13为本发明提供的一种加入偏置电压后的变压器的电流和数字量的电压信号的波形图，放大器U1输出的电压的波形请参照图14，图14为本发明提供的另一种直流变换器中关键参数的波形图，图中包括移相全桥电路中各个开关管的开关周期、驱动信号PWM_Q1～Q4、变压器T的电流及放大器U1输出的电压信号，由于加入了偏置电压Vref，放大器U1输出的电压信号均为正压，解决了现有技术中的电压信号小于0时无法用于对超前桥臂上的开关管进行控制的依据的问题。
本实施例中，为了保证直流变换器输出的功率为负载所需的期望功率，本申请中在控制超前桥臂上的开关管关断前，需先判断变压器T上的电流是否达到期望电流值，达到期望电流值时控制超前桥臂上的开关管关断，从而实现直流变换器为负载提供期望功率，满足负载的用电需求。
具体地，本申请中在控制超前桥臂上的开关管断开时，需要在滞后桥 臂上相应的开关管导通后判断数字量的电压信号是否到达电压阈值，并在到达阈值时将超前桥臂上相应的开关管断开，以保证输入到负载上的功率能够使其正常工作。
在第四开关管Q4导通后判断数字量的电压信号是否到达电压阈值，并在判断数字量的电压信号到达电压阈值时控制第一开关管Q1断开；在第三开关管Q3导通后判断数字量的电压信号是否到达电压阈值，在判断数字量的电压信号到达电压阈值时控制第二开关管Q2断开。由于第一开关管Q1及第二开关管Q2与直流电源E连接，将第一开关管Q1及第二开关管Q2断开后即可停止直流电源E的输入。
请参照图15，图15为本发明提供的一种开关管的控制装置的结构示意图，该装置包括：
存储器1，用于存储计算机程序；
处理器2，用于执行计算机程序时实现如上述开关管的控制方法的步骤。
对于本发明提供的一种开关管的控制装置的介绍请参照上述方法实施例，本发明在此不再赘述。
为解决上述技术问题，本发明提供了一种直流变换器，包括如上述的开关管的控制装置和依次连接的直流电源、移相全桥电路、变压器T及设有LC谐振电路的整流电路，还包括：
依次连接的电流互感器CT、放大器U1及比较器；
电流互感器CT的输入端与变压器T连接，用于采集变压器T的电流；
放大器U1用于对电流互感器CT采集到的变压器的电流进行放大处理，输出放大后的电流信号；
比较器用于将放大后的电流信号与电流预设值进行比较，并在放大后的电流信号不大于电流预设值时输出第三电平，在放大后的电流信号大于电流预设值时输出第四电平，第三电平与第四电平相反。
对于本发明提供的一种直流变换器的介绍请参照上述方法实施例，本 发明在此不再赘述。
还需要说明的是，在本说明书中，诸如第一和第二等之类的关系术语仅仅用来将一个实体或者操作与另一个实体或操作区分开来，而不一定要求或者暗示这些实体或操作之间存在任何这种实际的关系或者顺序。而且，术语“包括”、“包含”或者其任何其他变体意在涵盖非排他性的包含，从而使得包括一系列要素的过程、方法、物品或者设备不仅包括那些要素，而且还包括没有明确列出的其他要素，或者是还包括为这种过程、方法、物品或者设备所固有的要素。在没有更多限制的情况下，由语句“包括一个……”限定的要素，并不排除在包括所述要素的过程、方法、物品或者设备中还存在另外的相同要素。
对所公开的实施例的上述说明，使本领域专业技术人员能够实现或使用本发明。对这些实施例的多种修改对本领域的专业技术人员来说将是显而易见的，本文中所定义的一般原理可以在不脱离本发明的精神或范围的情况下，在其他实施例中实现。因此，本发明将不会被限制于本文所示的这些实施例，而是要符合与本文所公开的原理和新颖特点相一致的最宽的范围。

Claims (10)

  1. 一种开关管的控制方法，应用于直流变换器，所述直流变换器包括直流电源、移相全桥电路、变压器和设有LC谐振电路的整流电路，其特征在于，包括：
    在移相全桥变换器中超前桥臂上的开关管断开之后判断所述变压器上的电流是否不大于电流预设值；
    若是，则控制所述移相全桥变换器中滞后桥臂上相应的开关管关断。
  2. 如权利要求1所述的开关管的控制方法，其特征在于，所述直流变换器还包括依次连接的电流互感器、积分电路、放大器及比较器，所述电流互感器的输入端与所述变压器连接，所述电流互感器用于采集所述变压器上的电流，所述积分电路对所述变压器上的电流进行积分，得到电压信号，所述放大器的输入正端设有偏置电压，用于将所述电压信号进行放大处理并加入偏置电压，输出放大后的电压信号，所述比较器用于将放大后的电压信号与最大积分门限及最小积分门限比较，并当所述放大后的电压信号大于所述最大积分门限或小于所述最小积分门限时输出第一电平，当所述放大后的电压信号在所述最大积分门限及所述最小积分门限之间时输出第二电平，所述第一电平与所述第二电平相反；
    在移相全桥变换器中超前桥臂上的开关管断开之后判断变压器上的电流是否不大于电流预设值，包括：
    将所述移相全桥变换器的滞后桥臂上的开关管导通时刻作为积分开始时刻控制所述积分电路对所述电流互感器采集到的所述变压器上的电流进行积分，得到所述电压信号；
    将所述比较器输出所述第二电平时的时刻作为积分结束时刻；
    将从上述积分开始时刻至所述积分结束时刻之间的时间作为积分时间；
    基于所述积分时间及所述LC谐振电路的谐振参数确定移相角时间；
    在所述移相全桥变换器中超前桥臂上的开关管关断之后以所述超前桥臂上的开关管关断时作为开始时间判断是否到达所述移相角时间，以实现 判断变压器上的电流是否不大于电流预设值；
    若是，则进入控制所述移相全桥变换器中滞后桥臂上相应的开关管关断的步骤。
  3. 如权利要求2所述的开关管的控制方法，其特征在于，基于所述积分时间及所述LC谐振电路的谐振参数确定移相角时间之后，还包括：
    确定最大移相角时间；
    判断所述移相角时间是否小于所述最大移相角时间；
    若小于，则进入在所述移相全桥变换器中超前桥臂上的开关管关断之后以所述超前桥臂上的开关管关断时作为开始时间判断是否到达所述移相角时间的步骤；
    若不小于，则在所述移相全桥变换器中超前桥臂上的开关管关断之后以所述超前桥臂上的开关管关断时作为开始时间判断是否到达所述最大移相角时间；
    若到达，则进入控制所述移相全桥变换器中滞后桥臂上相应的开关管关断的步骤。
  4. 如权利要求3所述的开关管的控制方法，其特征在于，确定最大移相角时间，包括：
    基于所述移相全桥电路中各个开关管的结电容的容值、所述变压器的原边漏感参数、所述LC谐振电路的谐振参数、所述变压器的输出电压及输出电流及所述积分时间确定所述最大移相角时间。
  5. 如权利要求3所述的开关管的控制方法，其特征在于，控制所述移相全桥变换器中滞后桥臂上相应的开关管关断之后，还包括：
    将所述移相全桥变换器中滞后桥臂上相应的开关管关断时的时刻作开始时刻判断是否到达预设死区时间；
    若是，则控制所述移相全桥变换器中滞后桥臂上相应的另一开关管导通。
  6. 如权利要求5所述的开关管的控制方法，其特征在于，将所述移相全桥变换器中滞后桥臂上相应的开关管关断时的时刻作开始时刻判断是否到达预设死区时间之前，还包括：
    基于所述移相全桥电路中滞后桥臂上各个开关管的结电容的容值及所述变压器的原边漏感参数确定所述变压器原边的LC谐振时间；
    基于所述变压器原边的LC谐振时间确定最大死区时间；
    基于所述最大死区时间设定所述预设死区时间，所述预设死区时间小于所述最大死区时间；
    进入将所述移相全桥变换器中滞后桥臂上相应的开关管关断时的时刻作开始时刻判断是否到达预设死区时间的步骤。
  7. 如权利要求1所述的开关管的控制方法，其特征在于，所述直流变换器还包括依次连接的电流互感器、放大器及比较器，所述电流互感器的输入端与所述变压器连接，用于采集所述变压器的电流，所述放大器用于对所述电流互感器采集到的所述变压器的电流进行放大处理，输出放大后的电流信号，所述比较器用于将所述放大后的电流信号与所述电流预设值进行比较，并在所述放大后的电流信号不大于所述电流预设值时输出第三电平，在所述放大后的电流信号大于所述电流预设值时输出第四电平，所述第三电平与所述第四电平相反；
    在移相全桥变换器中超前桥臂上的开关管断开之后判断变压器上的电流是否不大于电流预设值，包括：
    在移相全桥变换器中超前桥臂上的开关管断开之后判断所述比较器输出的信号是否为所述第三电平；
    若是，则进入控制所述移相全桥变换器中滞后桥臂上相应的开关管关断的步骤。
  8. 如权利要求1至7任一项所述的开关管的控制方法，其特征在于，在移相全桥变换器中超前桥臂上的开关管断开之后判断变压器上的电流是否不大于电流预设值之前，还包括：
    基于所述直流变换器的输出端连接的负载确定所述变压器上的期望电流值；
    所述滞后桥臂上相应的开关管导通后判断所述变压器上的电流是否到达所述期望电流值；
    若是，则控制所述移相全桥电路中超前桥臂上相应的开关管断开；
    进入在移相全桥变换器中超前桥臂上的开关管断开之后判断变压器上的电流是否不大于电流预设值的步骤。
  9. 一种开关管的控制装置，其特征在于，包括：
    存储器，用于存储计算机程序；
    处理器，用于执行所述计算机程序时实现如权利要求1至8任一项所述开关管的控制方法的步骤。
  10. 一种直流变换器，其特征在于，包括如权利要求9所述的开关管的控制装置和依次连接的直流电源、移相全桥电路、变压器及设有LC谐振电路的整流电路，还包括：
    依次连接的电流互感器、放大器及比较器；
    所述电流互感器的输入端与所述变压器连接，用于采集所述变压器的电流；
    所述放大器用于对所述电流互感器采集到的所述变压器的电流进行放大处理，输出放大后的电流信号；
    所述比较器用于将所述放大后的电流信号与所述电流预设值进行比较，并在所述放大后的电流信号不大于所述电流预设值时输出第三电平，在所述放大后的电流信号大于所述电流预设值时输出第四电平，所述第三电平与所述第四电平相反。
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