KR20120036246A - Motor control system - Google Patents

Abstract

An object of the present invention is to always maintain the one-phase peak current of the recovery current flowing through the diode by one phase without synchronizing the turn-on of two or more switching elements, thereby reducing the reliability, low loss, and low noise of the device and the mounted components. It is an object of the present invention to provide a motor control device, a motor control system, and a motor control module that can realize low noise in a relatively inexpensive configuration.
The PWM controller receives a control signal of each phase from a PWM control signal converter for calculating a PWM control signal for driving an inverter and a PWM control signal converter, and when the control signals of the respective phases do not have a predetermined time interval, respectively. And a PWM control signal corrector for correcting the control signal of the at least one phase to space the predetermined time.

Description

Motor Control System {MOTOR CONTROL SYSTEM}

The present invention relates to a motor control technique.

In recent years, the position sensorless driving device of a permanent magnet synchronous motor (hereinafter referred to as a "motor") is required to operate with high efficiency, and the shaft error between the motor rotor shaft and the control shaft is determined from voltage and current information applied to the motor. Optimum control is performed by PWM-controlling the switching element which comprises an inverter based on the frequency which is a speed command, adjusting the voltage and electric current applied to a motor so that an estimated axis error may be controlled to predetermined value. In addition, the switching elements constituting the inverter include an insulated gate bipolar transistor (IGBT) of an insulated gate type bipolar or a metal oxide film type metal oxide semiconductor (M0SFET) of a field effect transistor (hereinafter abbreviated as a switching element). It is common to configure a diode by connecting (parallel connection) between a collector and an emitter.

As for the operating temperature characteristic of the switching element, it is necessary to limit the heat generation due to the switching loss of the element itself, and since a large amount of ripple current remains during the output, there is a problem such as deterioration of operating efficiency or noise.

As for the diode, a current called a recovery current Irr or a reverse recovery current Trr (hereinafter abbreviated as recovery current) flows in a period in which the switching element is turned off. There was a problem that a harmonic current (ripple current) included in the noise and noise were generated due to it.

Regarding the switching loss of the switching element, Patent Document 1 discloses a method of applying silicon carbide (SiC) for the purpose of "reducing switching loss when operated at light load", and in recent years, high efficiency and miniaturization have been proposed. For the purpose, a switching element employing a superjunction structure (SJ) has also been developed.

Regarding the recovery current flowing through the diode, Patent Document 2 discloses a method of synchronizing the turn-on timing of two or more switching elements for the purpose of "suppressing the recovery current to protect the switching element and the diode and reducing the cost".

Japanese Patent Application Publication No. 2010-115110 Japanese Patent Application Publication No. 2004-215357

In the technique of Patent Literature 1, the loss of the switching element is determined by the size of "Vce x Ic" similarly to a general transistor, and the improvement of the operating efficiency (inverter efficiency) in the light load region where Vce changes to a constant value is almost constant. From this difficulty, it has been described that silicon carbide (SiC) is applied to the switching element and that the loss can be reduced in the light load region of the air conditioner. However, the improvement regarding noise is not described. In addition, the device itself is expensive. In addition, the harmonic current (ripple current) included in the recovery current flowing through the diode in the period in which the switching element is turned off and the reduction of noise due to it are not considered.

In the technique of Patent Document 2, it is described that the floating inductance is increased by simultaneously generating the recovery current for the diode by synchronizing the turn-on timing of two or more switching elements, thereby reducing the recovery current peak value of each phase. As a result, an increase in the current capacity of the switching element and the diode of each phase can be suppressed, but since the peak current of the recovery current returned to the DC power supply side is doubled in one phase (total of two phases), the current of the power supply device. Increasing capacity or adding parts may increase costs.

An object of the present invention is to reduce the peak of the recovery current.

The object of the present invention,

In a motor control system for outputting a PWM control signal from the PWM controller to the inverter,

The PWM controller,

A PWM control signal converter for calculating the PWM control signal for driving the inverter;

A PWM control signal corrector for receiving control signals of each phase from the PWM control signal converter and correcting the control signals of the at least one phase so as to space the predetermined time when the control signals of the respective phases do not have a predetermined time interval. It is achieved by a motor control system having a.

In addition, the object of the present invention,

A motor control system having a control device for controlling through a inverter such that the rotational speed of the permanent magnet synchronous motor matches the speed command value,

Converter (rectifier) which converts AC power supplied from commercial power supply into DC power, DC power supply which supplies electric power to DC side of inverter with battery, and parallel arrangement (three phases) in which two switching elements are connected in series By using an inverter including a diode connected to the collector and the emitter of each switching element, and using a semiconductor computing element such as a microcomputer or a digital signal processor (DSP), a DC voltage detector and a DC current detector A control device for inputting a detection signal and driving a switching element constituting the inverter based on the speed command ω i,

The said control apparatus is achieved by the motor control system characterized by always making only one phase, without making the switching element in the said inverter turn-on control which synchronized two or more phases.

According to the present invention, the peak of the recovery current can be reduced.

1 is a configuration diagram of a motor control system.
2 is a functional block diagram of a control device according to a first embodiment.
3 is a diagram for explaining a control system estimation axis and a rotor axis of the motor control system according to the first embodiment.
4 is a configuration diagram of a voltage command controller of the motor control device according to the first embodiment.
5 is a view for explaining a PWM controller of a conventional motor control system.
FIG. 6 is a diagram for explaining a PWM control signal corrector of the motor control system according to the first embodiment. FIG.
7 is a diagram for explaining a processing flow of a PWM control signal corrector;
8 is a diagram of modulation rates and respective phase voltages in the PWM control signal conversion method.
Fig. 9 is a diagram when PWM control signal correction is not required.
10 is a diagram when PWM control signal correction is required.
Fig. 11A is a diagram showing the driving of the switching element and the current of the diode (phase 1).
Fig. 11B is a diagram showing the driving of the switching element and the current of the diode (two phases).
12 is a real operation waveform diagram showing a suppression result of a recovery current;
13 is a real operation waveform diagram showing a suppression result of a recovery current;
14 is a functional block diagram of a control device according to a second embodiment.
15 is a configuration diagram of a motor control system according to a third embodiment.
16 is an external view of a module used in the motor control device according to the fourth embodiment.

Hereinafter, description will be given using the drawings.

[First Embodiment]

1 is a configuration diagram of a motor control system according to a first embodiment of the present invention.

The motor control system 100 includes a permanent magnet synchronous motor 1, a DC power supply 2, an inverter 3 for converting DC power to AC power, and a DC voltage detector for detecting voltages of the DC power supply 2. (6), a direct current detector 7 for detecting the direct current side current of the inverter 3, and a control device 8 are provided.

The motor 1 is a permanent magnet synchronous motor. The DC power supply 2 is a converter (rectifier) or a battery that converts AC power supplied from a commercial power source into DC power, and provides electric power to the DC side of the inverter 3. The inverter 3 is provided with six switching elements which are the switching elements 4, and the diode 5 connected to the collector and emitter of each switching element.

In addition, the control apparatus 8 is comprised with semiconductor arithmetic elements, such as a microcomputer or a digital signal processor (DSP). The detection signals of the DC voltage detector 6 and the DC current detector 7 are input to the control device 8, and the speed command ω i is input, so that the control device 8 uses the PWM control signal ( 9) is calculated and output to the inverter 3. The PWM control signal 9 is a signal for ON / OFF control of a switching element which is a semiconductor power element constituting the inverter 3.

Fig. 2 is a functional block diagram of the control device 8 (8a) of Fig. 1 which is the first embodiment of the present invention, and each function is realized by a CPU (Central Processing Unit) and a program which is a computer.

The control device 8a generates a PWM control signal based on the speed command ω i and controls the inverter by dq coordinate system vector control. The control device 8a includes a dq vector control unit 60, a phase calculator 10, a PLL controller 11, a speed controller 13, an adder 14, and a d-axis current command generator 15. And an axis error calculator 16.

The dq vector control unit 60 controls the voltage command controller 12, the two-axis three-phase converter 17, the PWM controller 18, the three-phase two-axis converter 19, and the current reproduction calculator 20. And a PWM control signal using a current command value (dc axis current command value Idc *, qc axis current command value Iqc *) and the phase? Dc of the control shaft.

The current reproduction calculator 20 uses the bus current Ish output by the DC current detector 7 (FIG. 1), and the three-phase motor currents Iu, Iv, and Iw using the three-phase voltage command values Vu *, Vv *, and Vw *. To reproduce.

The three-phase two-axis converter 19 calculates the dc-axis current detection value Idc and qc-axis current detection value Iqc based on the reproduced three-phase motor currents Iu, Iv, Iw and the estimated phase θdc of the control shaft. Calculate based on In addition, the dc? Qc axis is defined as the control axis, the d? Q axis is defined as the rotor axis of the motor 1, and the axis errors of the dc? Qc axis and the d? Q axis are defined as Δθc (see FIG. 3).

The voltage command controller 12 includes a dc axis current command value Idc * calculated by the d-axis current command generator 15, a qc axis current command value Iqc * calculated by the speed controller 13, and a three-phase two-axis converter. Using dc-axis current detection value Idc and qc-axis current detection value Iqc calculated by (19), speed command ωi, and motor constant set values (r *, Ld *, Lq *, Ke *, not shown) The dc axis voltage command value Vdc * and the qc axis voltage command value Vqc * are calculated.

4 is a detailed functional block diagram of the voltage command controller 12 (FIG. 2). The voltage command controller 12 includes adders 24 and 25, current controllers 21 and 22, and a vector operator 23.

The current controller 21 calculates the second dc axis current command value Idc ** based on the output of the adder 24 (deviation between the dc axis current command value Idc * and the dc axis current detection value Idc).

The current controller 22 calculates the second qc axis current command value Iqc ** based on the output of the adder 25 (deviation of qc axis current command value Iqc * and qc axis current detection value Iqc).

In the vector calculator 23, as shown in equation (2) using the second dc axis current command value Idc **, the second qc axis current command value Iqc **, the speed command ω i and the motor constant setting value, The voltage command value Vdc * and the qc-axis voltage command value Vqc * are calculated and output to the 2-axis three-phase converter 17.

In Equation 2, r * is the motor winding resistance setting of the control system, Ld * is the d-axis inductance setting of the motor, Lq * is the q-axis inductance setting of the motor, Ke * is the motor induced voltage constant setting of the control system, and ωi is the speed It is order.

The two-axis three-phase converter 17, based on the dc-axis voltage command value Vdc * and qc-axis voltage command value Vqc *, the estimated phase θdc of the control shaft, from the equation (3) three-phase voltage command value Vu *, Outputs Vv * and Vw *.

Next, a velocity and phase estimation method for realizing position sensorless control will be described.

The axis error calculator 16 calculates an axis error Δθc from the dc axis voltage command value Vdc *, the qc axis voltage command value Vqc *, the dc axis current value Idc, the qc axis current value Iqc, and the set value of the motor constant.

The PLL controller 11 processes the deviation between the axis error [Delta] [theta] c and the axis error command value [Delta] [theta] c * output by the axis error calculator 16 by using the PI controller, and outputs an estimated value? 1 * of the motor rotational speed. Axis error command value (DELTA) (theta) c * is the information which the PLL controller 11 hold | maintains, and is normally set to 0 vicinity.

Here, the PI controller controls the estimated axis error Δθc between the rotor axis d? Q axis and the control system axis dc? Qc axis of the motor 1 so as to coincide with the axis error command value Δθc * (normally near 0). will be.

In the phase calculator 10, the estimated motor rotational speed ω 1 * is integrated to calculate the phase θ dc of the control axis.

FIG. 5 is a diagram showing a detailed functional block configuration of a conventional PWM controller 18a (for example, corresponding to the PWM controller 18 in FIG. 2). The PWM controller 18a includes a voltage modulation rate calculator 26 and a PWM control signal converter 27. In the voltage modulation rate calculator 26, a voltage modulation rate (expressed as a ratio of the relationship between the three-phase voltage command values Vu *, Vv *, Vw * output from the two-axis three-phase converter 17 and the DC voltage Vd ( khu *, khv *, khw *) are calculated by the equation (5), and output to the PWM control signal converter 27.

The PWM control signal converter 27 includes the time per PWM cycle (1 / PWM frequency) at the PWM frequency and the voltage modulation rate (khu *, khv *, khw *) calculated by the voltage modulation rate calculator 26. Is converted to the on time of the PWM control signal as shown in Equation 6, and the PWM control signal 9 is calculated.

The switching element which is the switching element 4 which mounts the computed PWM control signal 9 in the inverter 3 is controlled ON / OFF, and electric power is supplied to a motor. At this time, no relationship is given to the information corresponding to each phase of the PWM control signal 9. This point is mentioned later.

The above is the basic operation of the position sensorless operation in the control device of the present embodiment. Next, the recovery current will be described.

As shown in Fig. 11A, a device for supplying electric power to a motor by turning on / off a switching element, etc., has a PWM control signal on (upper U phase arm, lower V phase) of a switching element connected in series in the inverter 3. Arm), and once the PWM control signal is turned off (current flows in the forward direction of the diode on the U-phase lower arm side) and the PWM control signal is turned on again (turn-on), the U-phase lower arm Recovery current flows through the diode. This recovery current is generated because the carriers generated inside the diode tube disappear. The magnitude of the current is determined to some extent by the structure of the diode.

In the conventional PWM controller 18a (FIG. 5), as described above, no relationship is given to information corresponding to each phase of the PWM control signal 9. In other words, the relationship between the three-phase PWM control signals is not monitored. For this reason, the switching element output of each phase may turn on / off simultaneously.

11B shows a current flowing by turning on the upper arms two phases (U phase, V phase), the lower arm, and the first phase (W phase), and once stopped (current flows in the forward direction of the diode of the lower arm), again, It is a figure in the case of output (turning on).

FIG. 12 is a diagram corresponding to FIG. 11B.

12 (1) shows a case where the switching elements of the upper arm are turned on at the same time in two phases, and the peak value of the recovery current becomes very high. In the conventional PWM controller 18a (FIG. 5), even in such a state, a current-capacitance element or the like in which the switching element or the diode is not damaged is selected and applied.

12 (2) shows a case where simultaneous two-phase on is avoided by shifting the timing for a predetermined time without turning on the switching element of the upper arm at the same time. By doing in this way, the peak value of a recovery current can be reduced compared with FIG. The peak of (1) is generated two times at the same time, whereas the peak (maximum value) of (2) is shifted by a predetermined time and is generated twice. That is, the peak current (maximum value) of the recovery current is reduced to one phase.

6 shows a structure for reducing peak current. The structure is a PWM controller 18b (for example, corresponding to the PWM controller 18 in FIG. 2) provided with the PWM control signal corrector 28. As shown in FIG. The functions of the voltage modulation rate calculator 26 and the PWM control signal converter 27 are the same as those of the conventional PWM controller 18a described above. The PWM control signal corrector 28 corrects the PWM control signal based on the PWM control signals PWMu *, PWMv *, and PWMw * calculated by the PWM control signal converter 27 to avoid turn-on of two-phase synchronization. It has a structure.

7 shows the processing flow of the PWM control signal corrector 28.

In process 29, the deviation of each PWM control signal with respect to the intermediate phase is calculated. In the maximum phase,

ΔP_max = maximum phase-intermediate phase

Is computed and for the minimum phase,

ΔP_min = Minimum to Medium Phase

Calculate

Next, in decision 30, it is determined whether the three phases are in an approaching state. When both ΔP_max and ΔP_min are less than any predetermined time P_lmt, it is determined that the three phases are approaching. In this case, namely, in the case where the judgment is made in the judgment 30, the processing ends without performing the correction for shifting the on timing of the intermediate phase by a predetermined time (P_lmt).

In practice, as shown in the relationship between the PWM control signal conversion method and the respective phase voltages in FIG. Even in the case where the tuning is low, there are few cases in which the maximum phase and the minimum phase approach each other, and it is considered that very few cases are determined in the judgment 30 on the practical operation. In addition, since the current flowing through the switching element is small because the applied voltage is low, it is considered that there is no problem even if the correction process described later is not executed.

If ΔP_max or ΔP_min is equal to or greater than a predetermined time P_lmt, that is, in the case of a determination of NO in the decision 30, the correction process is executable, and it is determined that the PWM control signal correction is feasible and proceeds to the next step.

In the determination 31, it is determined whether the maximum phase and the intermediate phase are in the approaching state, and when ΔP_max is less than P_lmt (YES), the determination is made as the approaching state so that the on timing of the intermediate phase is the predetermined time P_lmt so as to be corrected. Correction is performed by using P_max =? P_max? P_lmt. If no access is made (no judgment), the process proceeds to the next step.

In the determination 33, it is determined whether the minimum phase and the intermediate phase are in an approaching state, and when ΔP_max is less than? P_lmt (Yes determination), the determination is made as the approaching state and the correction process is performed so that the on timing of the intermediate phase is a predetermined time? P_lmt. 34) (P_min = ΔP_min + P_lmt). If it is not approaching (no judgment), since the maximum phase and the minimum phase are not approaching the intermediate phase, the process ends without doing anything.

9 illustrates a case where the PWM signal on timing based on the intermediate phase does not approach in the triangular wave PWM control scheme. Since both the maximum phase and the minimum phase are secured for more than a predetermined time (± P_lmt) with the intermediate phase, no correction is necessary.

10 illustrates a case where the on timing approaches. FIG. 10 (1) shows a case where the maximum phase and the intermediate phase need correction in the approach state (P_max <P_lmt), and the minimum phase and the intermediate phase have a deviation of more than a predetermined time (P_min ≧? P_lmt) and do not need correction. . Therefore, as shown in (2) of FIG. 10, the ON timing of the intermediate phase is corrected so as to secure the deviation of the maximum phase and the intermediate phase for a predetermined time period P_lmt.

FIG. 13 shows the recovery current when the avoidance time of the PWM control signal in which the PWM control signal corrector 28 described in FIG. 6 avoids turning-on of two-phase synchronization is changed. As can be seen from the waveform, it can be seen that the peak current is divided into two phases and reduced as the predetermined time is taken large. If it is increased to 0.2us, the recovery currents of each divided phase are almost the same. According to experimental examination, this value is about 0.2us to 0.3us. Therefore, the peak current of the recovery current can be made the same by adjusting the turn-on avoidance time of the two-phase synchronization to about 0.2 to 0.3 us in accordance with the switching element and the diode to be applied.

Second Embodiment

Although the structure of the motor control apparatus of the second embodiment is the same as that shown in FIG. 1, the vector control method inside the control apparatus 8 is different.

14 is a functional block diagram of the control device 8 (8b) in the form of the second embodiment. In addition, the same code | symbol as FIG. 2 performs the same operation.

A part different from FIG. 2 is a PLL controller 11 which calculates the qc axis current command value Iqc * in the position sensorless mode from the low pass filter 52 and estimates the rotational speed ω1 * of the motor 1 ( 2) is changed to the speed error calculator 50 that calculates the speed error, and the adder 51 that adds the speed error and the speed command.

That is, the speed error calculator 50 calculates the speed error Δωm by proportionally calculating the axis error Δθc calculated by the axis error calculator 16, and the adder 51 adds the speed command ω i and the speed error Δωm and adds it. The result is input to the phase calculator 10.

As a result, the arithmetic processing in the voltage command controller 12a is simplified as in equation (7).

Current reproduction, axis error calculation, and phase calculation processing are performed in the same manner as in the first embodiment.

In this embodiment, the qc-axis current detection value Iqc reproduced by the current reproduction calculator 20 and the three-phase two-axis converter 19 is passed through the low pass filter 52 to the qc-axis current command value Iqc * (observed current average. ), The arithmetic unit (speed controller 13, adder 14, PLL controller 11) for controlling the speed in the control device 8a of FIG. 2 is simplified. Thereby, the number of parameters, such as a gain, in an arithmetic machine can be reduced, and the practicality (universal) can be improved.

Third Embodiment

Fig. 15 is a configuration diagram of the motor control system of the third embodiment, in which the upper three in parallel arrangements (three phases) of two series connection of switching elements mounted on the inverter 3 of the first embodiment are insulated gate bipolars. Is an example of a metal oxide film type SJMOS having a superjunction structure (SJ) for the purpose of high efficiency and miniaturization of the IGBT and the lower three. In addition, the same code | symbol in a block diagram performs the same operation | movement as FIG. 1, and you may use the motor control apparatus structure of 2nd Example with respect to a control apparatus.

SJ-MOS has high efficiency at low current, while the reverse recovery time of the parasitic diode is slow, so that the Irr flowing through the reflux diode is known to be large. Therefore, in combination with the "two-phase modulation based on the lower arm" shown in FIG. 8 (3), the lower arm is centered at the time of low current, and switching can be achieved with high efficiency. At high current, the switching operation rate of the IGBT of the upper arm is increased, and the operation rate of the lower arm side is lowered, so that the change in efficiency can be suppressed.

[Example 4]

FIG. 16: is an external view of the motor drive module 200 of a 4th Example, and shows one form of a final product.

The module 200 is a module for a motor control apparatus in which the semiconductor element 202 is mounted on the control board 201. The control board 201 includes a DC current detector 7 and a DC voltage detector (see FIG. 1). 5) and the control device 8 are directly mounted, and the inverter 3 is mounted as a semiconductor element 202 having a single chip. By modularization, miniaturization can be achieved and the apparatus cost can be reduced. In addition, a module means "standardized structural unit", and is comprised from the component of detachable hardware / software. In addition, although it is preferable to be comprised on the same board | substrate in manufacture, it is not limited to the same board | substrate. From this, you may be comprised on the some circuit board built in the same housing.

According to the above embodiments, the peak current of the recovery current flowing through the diode can be suppressed to one phase by always making it into one phase without synchronizing the turn-on of two or more switching elements. As a result, the reliability, low loss, low noise, and low noise of the element and the component to be mounted can be realized with a relatively inexpensive configuration.

1: Motor (permanent magnet synchronous motor, compressor motor)
2: DC power
3: inverter
4, 4b: switching element
5: diode
6: DC voltage detector
7: DC current detector
8, 8a, 8b: control unit
9: PWM control signal
10: phase calculator
11: PLL Controller
12: voltage command controller
13: speed controller
14, 24, 25, 51: adder
15: d-axis current command generator
16: axis error calculator
17: 2-axis 3-phase converter
18: PWM controller
19: 3-phase 2-axis converter
20: current reproduction calculator
21, 22: current controller
23: vector operator
26: voltage modulation rate calculator
27: PWM control signal converter
28: PWM control signal compensator
50: speed error calculator
52: low pass filter
60: vector control unit
100, 10Ob: Motor Control System
200: module
201: control board
202: semiconductor device (power module)

Claims (8)

In a motor control system for outputting a PWM control signal from the PWM controller to the inverter,
The PWM controller,
A PWM control signal converter for calculating the PWM control signal for driving the inverter;
A PWM control signal corrector for receiving control signals of each phase from the PWM control signal converter and correcting the control signals of the at least one phase so as to space the predetermined time when the control signals of the respective phases do not have a predetermined time interval. Having a motor control system. The motor control system according to claim 1, wherein when there is no predetermined interval between the maximum phase and the intermediate phase, the control signal of the maximum phase is corrected so as to have a predetermined time interval. The motor control system according to claim 1, wherein when there is no predetermined interval between the minimum phase and the intermediate phase, the control signal of the minimum phase is corrected so as to have a predetermined time interval. A motor control system having a control device for controlling through a inverter such that the rotational speed of the permanent magnet synchronous motor matches the speed command value,
Parallel arrangement of two connections in series with a DC power supply that supplies power to the DC side of the inverter by a converter (rectifier) or a battery that converts AC power supplied from commercial power to DC power, and six switching elements in series (three-phase A DC voltage detector and a DC current detector using an inverter including a diode connected to the collector and emitter of each switching element, and a semiconductor computing element such as a microcomputer or a digital signal processor (DSP). And a control device for inputting a detection signal of the device and driving a switching element constituting the inverter based on the speed command ωi.
The control device is a motor control system, characterized in that the switching device in the inverter is always in one phase without turning-on control in which two or more phases are synchronized. The motor control system according to claim 4, wherein the control device generates a PWM control signal for driving the switching element based on the triangular wave. 5. The motor control system according to claim 4, wherein in one phase only, the time for one phase can be arbitrarily adjusted. 7. The motor control system according to claim 6, wherein any time can be adjusted by an external terminal or an external storage device of a semiconductor computing element such as a microcomputer or a digital signal processor (DSP) mounted in the control device. The motor control system according to claim 4, wherein a diode is connected to the collector and the emitter of the switching element.

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