JPH03245789A - Vector control system of induction motor - Google Patents

Abstract

PURPOSE:To reduce a control error, and to maintain the relationship of a torque command and a torque output in a linear type at all times by controlling vector by primary resistance as a measurable constant, set secondary resistance, leakage reactance and exciting inductance. CONSTITUTION:When speed omegar is input to a function arithmetic means 1, a function arithmetic circuit 1a outputs M'.lambda2/M, a function arithmetic circuit 1b outputs M', and a current arithmetic circuit 1c outputs currents iO* from these circuits 1a, 1b. Currents iO* are input to a coordinate transformation means 3 through a command correction means 2. A speed command omegar* is converted into a torque command currents iT* by a speed control means 6 after detecting speed omegar is subtracted, and input to the coordinate transformation means 3 and a slip-frequency arithmetic means 5. Slip frequency omegas is computed by the slip-frequency arithmetic means 6. Slip frequency omegas is added to the detecting speed omegar, and input to the coordinate transformation means 3 through an integrating means 7, and two control currents are output to a two phase/ three phase conversion means 4.

Description

[Detailed Description of the Invention] A. Industrial Application Field The present invention relates to a vector control method that uses an inverter to perform variable speed control and torque control of an induction motor, and in particular, vector control that takes into account changes in the inductance of an induction motor. Regarding the method.

B9 Summary of the Invention The present invention provides a vector control method that uses an inverter to perform variable speed control and torque control of an induction motor, in which the primary resistance R1 and the set secondary resistance R1 are measurable constants.
', leakage reactance Lσ and exciting inductance M'
By performing control using It provides technology that can be used.

C1 Prior Art A vector control method using a slip frequency is known for variable speed drive of an induction motor.

FIG. 4 is a configuration diagram showing an example of a slip frequency vector control method for an induction motor. In the figure, 40 is the induction N motive.
41 is a PWM inverter that controls the induction motor 40;
2 is a three-phase current control means (A
CR), 43 is a coordinate conversion means, 44 is a two-phase/three-phase conversion means, 45 is a speed control means (ASR), 46 is a slip frequency calculation means, 47 is an integration means, 48 is a motor current detection means, 49 is a This is speed detection means (TG) coaxially attached to the induction motor 40.

In the figure, the motor speed command ω,' is inputted to the speed control means 45 after subtracting the motor speed ω, detected by the speed detection means 49, and a torque command current iT'' is generated, and the coordinates are exchanged. The current control means 42 is driven through the means 43 and the phase conversion means 44.

In order to output the required torque while keeping the secondary magnetic flux λI constant, the above device gives the current amplitude, phase, and slip frequency as command values to the coordinate conversion means 43, but the slip frequency ωS is Various constants of the motor are used in the calculation, and for example, in the above-mentioned shear frequency calculation means 46, ω5-(R2/L2) ・(iT"/i0")
Perform the calculation. Here, the secondary resistance R and the secondary reactance L are motor constants. Conventionally, motor constants were considered to be constant even when the frequency and magnetic flux density changed, and values calculated from measurement conditions near the motor rating or design values were used. It is also known that the resistance component changes with temperature, and various methods of temperature compensation for the resistance component are being studied.

D9 Problem to be solved by the invention In recent years, with improvements in the output voltage accuracy of inverters and the processing speed of detected data, the inverter itself can freely output the voltage and frequency required for the motor constant to measure the motor constant. Now I can do it.

However, there are the following three problems when performing control using this measurement data.

(1) Conventional vector control consists of -order resistance Rr, secondary resistance R7, secondary inductance L9. Mutual inductance M
It is actually used as a beam.

Since it is impossible to separate M and M, the negative-order resistance R1, set secondary resistance R2', leakage reactance Lσ, and excitation inductance M' are measured.

Therefore, it is necessary to replace the constants used in conventional vector control with measurable constants.

(2) Also, if M' is constant, vector control that keeps the magnetic flux constant is established, but if M' changes, there is a question as to whether or not it is appropriate to keep the magnetic flux constant, so the torque command A new magnetic flux command must be considered from the perspective of the linear relationship between torque and torque output.

(3) The above M' is thought to change depending on two factors, frequency and magnetic flux, but if this is the case, the processor (
In order to store the data as a table inside the CPU (hereinafter referred to as CPU), a two-dimensional table is required, and the memory capacity becomes enormous.

The present invention was created in view of these problems, and
A vector control method for induction motors that has little control error, always maintains a linear relationship between torque command and torque output, improves calculation accuracy when controlling by estimating induced electromotive force, and requires less memory capacity. is intended to provide.

90 Means for Solving the Problems Means for solving the above problems in the present invention is a vector control method for an induction motor that controls the current of an inverter by calculating a slip frequency using a required motor constant. The vector control method of the induction motor is controlled by the primary resistance R8, the set secondary resistance R7', the leakage reactance Lσ, and the excitation inductance M', and the torque transfer function remains constant even if the excitation inductance M' changes. holding and controlling;
It is preferable to store the excitation inductance M' in a one-dimensional table for the frequency f.

F1 Effect The vector control system of the present invention consists of the following three stages.

(2) Derive the voltage-current equation of the induction motor using measurable constants M', R,', Lσ, and RI, and considering that M' changes.

■From the above formula, set the torque transfer function to a constant.

■Create a table to be used for the calculation of M'.

■Voltage-current equation when M' changes The instantaneous voltage-current equation for an induction motor is known as (p=d/dt
) treats θ as a scalar quantity, so we have to use the conceptless quantity 10m. Therefore, θ is treated as a vector quantity as shown below.

The velocity of (υx13)·12=H and each vector component of the electromotive force have a relationship as shown in FIG.

In the figure, if the vector corresponding to υ is defined as the angular velocity pe-me, then - (pθ×Φ) (1) holds true.

This results in The known equation is ・・・・・・(2) Then, Kome Using equation (2), we can calculate the primary current and secondary magnetism. Derive the equation represented by the bundle.

The secondary magnetic flux Φ is Φ,=M・ Well, +L1・d.

(3) Here, the tensor transformation matrix C is α-t, t/M If we transform equation (2) using this matrix using the tensor matrix, we get [Z]' = C'' [Z C-1. Become.

here, (4) ...(5) ...(6) M' M2/L.

R2′ R7・(M/L2)”, then by tensor transformation, it becomes ・・・・・・(7). However, since there is actually no power source in the secondary conductor, (
L, /M) can be replaced with υt-0, and (L4
/M)L can be replaced with (L t/M) (12=^, /M-4. Therefore, equation (7) becomes the following margin (8), and if this is separated into each component, is obtained, means , means , means .

・・・・・・・・・・・・(9) The first term is the voltage drop component due to resistance The second term is the transformer electromotive force due to magnetic flux change The third term is the electromotive force due to the rotation of the rotor. The torque T is Electric energy that changes due to rotor rotation obtained from θP　θO θt θt ...(10) 12 If λ, /M, then ■ XL (11) becomes.

To keep the torque current and torque torque linear, (l ■) From the formula, Lxλ, 30 λ2 constant must be kept, need to be retained.

■Slip frequency type vector control Slip frequency type vector control is one in which the control circuit defines virtual rotational coordinates and performs feedforward so that the magnetic flux coincides with these coordinates. Therefore, the slip frequency is the relative velocity of the rotor to the primary angular velocity arbitrarily set by the control circuit.

If this -order angular velocity is defined as ω1, the above equation (8) can be converted into a voltage-current equation on this virtual rotational coordinate. In order to avoid confusion, variables are indicated with "e" as shown below.

Margin below fixed coordinate system ω rotating coordinate system υ υ1゜ 1゜ Sa1・ ^.

^?・ ex (ωwork ω,) The formula for conversion is It is as follows.

The above equation (8) can be rewritten as follows.

(13) here, Lσ and ωS are Lσ=L.

M' L.

M”/Lt ω　S ω ωr and Baek To achieve constant magnetic flux with torque control teeth, p4゜7M R,'a,.

(R2'+M'ωsx)^,. /M ■ It suffices if in, (d.

λ2. /M) L, 7M-d2 i) T Defined as torque current.

This torque current dT is always ^, in order to satisfy the torque linearization condition of equation (12). An orthogonal relationship with 7M must be maintained.

If the magnetic flux is on the control axis, it will be controlled under two conditions of orthogonal relationship, λ and fiT, and the torque will be (ffi'r-t, t/ML) with a coefficient (M'・^, /M). Note that this is based on the premise that the control axis and the magnetic flux match, and is established by maintaining vector control conditions.

Incidentally, in reality, Lt, Lx, M, and Rt cannot be measured, and L and M cannot be measured separately, and Lσ and M' are not constant values but are variable amounts. Conventionally, this was expressed using a constant in a limited measurable range, but in the present invention, for M, it is expressed as M'.

Replace with and consider it as a setting constant. The reason for this is that M'・dO is always constant even if M changes, and the actual secondary magnetic flux changes, but conversely, even if M' changes, the linearity with the torque current does not change. This is because the torque linearity always holds true because the secondary magnetic flux is changed accordingly.

■Table of M' Since M' is a function of frequency and magnetic flux, these two
If you try to provide a table covering the entire range of combinations of two, the data will become two-dimensional and the memory capacity will become enormous. However, in the constant torque range of the motor, the magnetic flux is constant, and in the constant output range, it is expressed as the reciprocal of the output frequency, and the magnetic flux with respect to frequency is uniquely determined. In the present invention, by utilizing this characteristic, only the M' table for frequencies is provided, and the memory capacity is saved by organizing the data into -dimensional data.

G. Embodiments Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

FIG. 1 is a block diagram of an embodiment of the present invention, and shows an example of vector control using a new constant and taking changes in M' into consideration. The figure shows the application of the present invention to the slip frequency type vector control circuit of the induction motor explained in Fig. 4.
This figure shows the present invention and parts that have been changed in accordance with the present invention. In the figure, 1 is a function calculation means of the present invention, 2 is a command correction means, 3 is a coordinate conversion means, 4 is a two-phase/three-phase conversion means, 5 is a speed control means, 6 is a slip frequency calculation means,
7 is an integrating means, and 8 and 9 are adders.

The function calculation means I is disposed in the CPU, divides the function into two elements, and calculates the first function by the function of ωa→(M'・12/M).
It is composed of a function calculation circuit 1a based on a function of ωA, a second function calculation circuit 1b based on a function of ωA M', and a current calculation circuit 1c based on these, and inputs the speed ωA from a motor speed detection means (not shown). Then, the first function calculation circuit 1a performs M'.
/112/M, and the second function calculation circuit 1b outputs M'
The current calculation circuit 1c outputs a current of 10" from them.
Output. This is ``(
11). The current 101 is input to the coordinate conversion means 3 via the command correction means 2. The command correction means 2 adjusts 1 when the command (M'・^, /M) changes.
+(M'/R1')S correction is performed.

On the other hand, the speed command ωa' is converted into a torque command current iT1 by the speed control means 5 after subtracting the detected speed ωa, and is input to the coordinate conversion means 3 and the slip frequency calculation means 6. The current j01 is also input to the slip frequency calculation means 6. The slip frequency calculation means 6 calculates ωs= (Rp'/M') (i T"/i 0" based on the formula (13) in ■ in the "Effect" column.
), the slip frequency ωS is calculated. ωS is added to the detected speed ωa, is inputted to the coordinate conversion means 3 via the integration means 7, and outputs the two-phase control current to the two-phase/three-phase conversion means 4. .

FIGS. 2(a), (b), and (c) are graphs showing the characteristics of the example. Figure (a) shows the relationship between frequency f and magnetic flux Φ, and is a characteristic diagram with induced voltage added for reference. Φ = Φmax, and in the constant output range (fT~), the relationship is Φ = Φmax (fT/f),
Φ is obtained. Actually, the magnetic flux is calculated using fy=ω, /2π based on the detected speed ωa as an approximate value instead of f.

Next, a locus on the measurement results of M' that maintains the above-mentioned relationship between frequency and magnetic flux is obtained as shown in FIG. 2(b). Figure (c) shows the data of M' at this time with respect to frequency f.
It will look like this. In addition, in Figure (c), for convenience of illustration, 3
Although only points are shown, it goes without saying that any point can be set. Furthermore, if the frequency is too low, the voltage output is small and the accuracy of M' is poor due to the influence of voltage error, so it is assumed that there is no change below a predetermined frequency.

As described above, the M' value can be tabulated as one-dimensional data for the frequency f and stored in the CPU.

FIGS. 3(a) and 3(b) are explanatory diagrams of table contents of the present invention. Figure (a) shows an example using broken line approximation, in which points with predetermined frequencies are memorized, and the frequencies between them are calculated by approximating them with broken lines. Figure (b) shows an example by specifying a range, where the frequency It is assumed that the value within the predetermined interval is the average value of M' at the two ends, and it is preferable to use either one of these 0 deviations depending on the purpose. In any case, it is possible to create a table of functions that output M′ by inputting frequency f,
In addition, vector control conditions can be established using the amount of change in M'.

As described above, the following effects are evident in this embodiment.

(1) Since vector control using constants obtained as measurement results becomes possible, control errors are reduced compared to when design values are used.

(2) By controlling M'·φ/M to be constant, the relationship between the torque command and the torque output maintains a linear relationship even if the value of M' changes.

(3) In recent years, a method of controlling the current control system by estimating the induced electromotive force of the motor has become popular.
′ can be accurately measured and used, which improves the calculation accuracy of the induced voltage.
As a result, current control accuracy is improved.

(4) Since M' is simplified as a one-dimensional table for frequencies, the memory capacity can be reduced.

H1 Effects of the Invention As explained above, according to the present invention, the control error is small, the relationship between the torque command and the torque output always maintains a linear relationship, and even when control is performed by estimating the induced electromotive force, calculation is possible. It is possible to provide a vector control method for an induction motor that improves accuracy and requires less memory capacity.

[Brief explanation of drawings]

Fig. 1 is a block diagram of an embodiment of the present invention, Fig. 2 is a graph of characteristics of the embodiment, Fig. 3 is an explanatory diagram of table contents, Fig. 4 is a block diagram of a conventional example, and Fig. 5 is a vector diagram. It is an explanatory diagram of components. ! ... Function calculation means, 2 ・Command correction means, 3, 43
Coordinate conversion means, 4.44...Phase conversion means, 5゜4
5... Speed control means, 6, 46. Slip frequency calculation means, 7.47... Integrating means, 8, 9. Adder, 40
...Motor, 41...Inverter, 42...Current control means. 2 other people (a) Figure 2 Graph of characteristics of the example Frequency diagram (a) (b)

Claims (3)

[Claims] (1) In the induction motor vector control method that controls the current of the inverter by calculating the slip frequency using the required motor constants, the primary resistance R_1, which is a measurable constant, and the set secondary resistance R
A vector control system for an induction motor, characterized in that control is performed using __2', leakage reactance L_σ, and excitation inductance M'. (2) The vector control system according to claim (1), characterized in that control is performed while keeping the torque transfer function constant even if the excitation inductance M' changes. (3) Claim (1) characterized in that the excitation inductance M' is stored in a one-dimensional table for the frequency f.
) Vector control method described in