JP6422796B2 - Synchronous machine control device and drive system - Google Patents

Description

  Embodiments described herein relate generally to a synchronous machine control device and a drive system.

  Conventionally, as a control method of a synchronous machine, sensorless control that does not use a rotation sensor such as a resolver encoder has been proposed in order to reduce the size and weight, reduce the cost, and improve the reliability. In general, in sensorless control, it is difficult to improve control accuracy in a low speed region, and various developments are being made.

  For example, as a control method in a low speed region, a method using PWM harmonics, a method of superposing a high frequency voltage on a voltage command, or a method of superposing a high frequency current on a current command is used. In either case, the rotational phase angle and angular frequency of the synchronous machine can be estimated based on the high-frequency components of the current and voltage flowing through the synchronous machine. In addition, a method for reducing noise by making the superposed high-frequency signal variable has also been proposed. By using these methods, sensorless control of the synchronous machine in the low speed range is possible.

  However, when a high-frequency signal is superimposed on a command, an error occurs in the output high-frequency component due to the effect of slot harmonics generated according to the angular frequency of the synchronous machine, and the rotational phase angle and angular frequency of the synchronous machine are accurately estimated. There was a fear that I could not. When the estimation accuracy of the rotational phase angle is lowered, there is a problem that the control becomes unstable and the step-out occurs.

JP 2009-153347 A JP 2001-339999 A JP 2011-172324 A

  Provided are a synchronous machine control device and a drive system capable of suppressing the influence of slot harmonics and accurately estimating the rotational phase angle of the synchronous machine.

  A synchronous machine control device according to an embodiment includes an inverter, a voltage command generation unit, a high frequency superimposition unit, and an estimation unit. The inverter drives the synchronous machine. The voltage command generation unit generates a voltage command based on the current command and the output current of the inverter. The high frequency superimposing unit can superimpose a predetermined high frequency signal on the current command or the voltage command. The estimation unit calculates an estimated rotational phase angle of the synchronous machine. The high frequency superimposing unit superimposes the high frequency signal when the difference between the slot harmonic frequency and the frequency of the high frequency signal is equal to or greater than a predetermined value, and does not superimpose the high frequency signal when the difference is less than the predetermined value.

The figure which shows the synchronous machine drive system which concerns on 1st Embodiment. The figure explaining a three-phase fixed coordinate system and a dcqc axis | shaft rotation coordinate system. The figure which shows the structure of the high frequency superimposition part of FIG. The figure which shows the structure of the speed and rotational phase angle estimation part 29 of FIG. The figure which shows the structure of the 1st error estimation part of FIG. The figure which shows the structure of the amplitude detection part of FIG. The figure explaining the function of the band pass filter of FIG. The figure explaining the detection method of the amplitude by the FFT analysis part of FIG. The figure which shows the structure of the 2nd error estimation part of FIG. The figure explaining a slot and a slot harmonic. The figure which shows the structure of the control system switching part of FIG. The figure explaining the influence by a slot harmonic. The figure explaining the influence by a slot harmonic. The figure explaining the influence by a slot harmonic. The figure explaining the influence by a slot harmonic. The figure explaining the influence by a slot harmonic. The figure explaining the influence by a slot harmonic. The figure explaining the influence by a slot harmonic. The figure explaining the influence by a slot harmonic. The figure which shows the synchronous machine drive system which concerns on 2nd Embodiment. The figure which shows the structure of the high frequency superimposition part of FIG. The figure explaining the change method of the frequency and amplitude of a high frequency voltage. The figure explaining the change method of the frequency and amplitude of a high frequency voltage. The figure explaining the change method of the frequency and amplitude of a high frequency voltage. The figure explaining the change method of the frequency and amplitude of a high frequency voltage. The figure explaining the change method of the frequency and amplitude of a high frequency voltage.

  Embodiments of the present invention will be described below with reference to the drawings.

(First embodiment)
A synchronous machine drive system (hereinafter referred to as “system”) according to the first embodiment will be described with reference to FIGS. FIG. 1 is a diagram illustrating a configuration of a system according to the present embodiment. As shown in FIG. 1, the system according to the present embodiment includes a synchronous machine 1 and a synchronous machine control device 2.

  The synchronous machine 1 is a motor including a stator and a rotor. The synchronous machine 1 is a permanent magnet synchronous machine including a permanent magnet, a reluctance synchronous machine that uses magnetic saliency such as a synchronous reluctance motor (SynRM), or a winding field that supplies field magnetic flux by a secondary winding. Although it is a magnetic synchronous machine, it is not restricted to this. Hereinafter, it is assumed that the synchronous machine 1 is a permanent magnet synchronous machine.

  The stator of the synchronous machine 1 has three excitation phases (U phase, V phase, and W phase), and generates a magnetic field by a three-phase alternating current flowing in each excitation phase. The rotor is rotated by the magnetic field generated by the stator.

  The synchronous machine control device 2 (hereinafter referred to as “control device 2”) according to the present embodiment uses two types of control methods, a control method using a high-frequency signal and a control method using no high-frequency signal. The synchronous machine 1 is controlled without a sensor. Hereinafter, a case where a high-frequency voltage is used as the high-frequency signal will be described, but a high-frequency current may be used as the high-frequency signal.

  As shown in FIG. 1, the control device 2 includes an inverter 21, a current detector 22, a coordinate conversion unit 23, a voltage command generation unit 24, a coordinate conversion unit 25, a modulation unit 26, and a voltage detector. 27, a high frequency superimposing unit 28, a speed / rotation phase angle estimating unit 29, and a control method switching unit 30.

  The inverter 21 is a circuit including a switching element. The inverter 21 converts the power from a power source (not shown) into alternating current by switching ON / OFF of the switching element, and supplies it to the synchronous machine 1. The inverter 21 receives a control signal for controlling ON / OFF of each switching element from the modulation unit 26.

  The current detector 22 detects a two-phase or three-phase current among the three-phase alternating current flowing through the stator of the synchronous machine 1. FIG. 1 shows a configuration for detecting two-phase (U-phase and W-phase) currents iu and iw. The three-phase alternating current flowing through the stator of the synchronous machine 1 may be obtained by calculation based on the direct current on the inverter 21. In this case, the control device 2 may not include the current detector 22.

  The coordinate conversion unit 23 converts the currents iu and iw detected by the current detector 22 from the three-phase fixed coordinate system to the dcqc axis rotation coordinate system, and calculates the currents idc and iqc. The current idc is the dc axis component of the current flowing through the stator (output current of the inverter 21), and the current iqc is the qc axis component of the current flowing through the stator (output current of the inverter 21). Here, the three-phase fixed coordinate system and the dcqc axis rotation coordinate system will be described with reference to FIG.

  As shown in FIG. 2, the three-phase fixed coordinate system is a fixed coordinate system composed of an α axis and a β axis. In FIG. 2, the α axis is set in the U-phase direction, and the β axis is set in a direction perpendicular to the α axis. The currents iu and iw detected by the current detector 22 are represented on such three-phase fixed coordinates.

  On the other hand, the dcqc axis rotation coordinate system is a rotation coordinate system including a dc axis and a qc axis. The dc axis is set in the direction estimated by the control device 2 as the d-axis direction (the direction in which the rotor inductance is minimum), and the qc axis is the q-axis direction in the q-axis direction (the direction in which the rotor inductance is maximum). Is set in the estimated direction. The inductance ellipse in FIG. 2 indicates the inductance of the rotor.

  As shown in FIG. 2, the dcqc axis and the dq axis do not always coincide with each other. The actual rotational phase angle θ of the rotor is represented by an angle from the α axis to the d axis. Further, the estimated rotational phase angle θest of the rotor estimated by the control device 2 is represented by an angle from the α axis to the dc axis. Hereinafter, an error between the rotational phase angle θ and the estimated rotational phase angle θest is referred to as an error Δθ.

  The coordinate conversion unit 23 can convert the three-phase fixed coordinate system into the dcqc axis rotation coordinate system by using the estimated rotation phase angle θest output from the speed / rotation phase angle estimation unit 29.

The voltage command generation unit 24 (current control unit) causes the current flowing through the synchronous machine 1 to be the current commands idc ^* and iqc ^* based on the currents idc and iqc, the current commands idc ^* and iqc ^* , and the estimated speed ωest. Then, voltage commands vdc ^* and vqc ^* are calculated.

The current command idc ^* is a dc axis component of the current flowing through the synchronous machine 1. The current command iqc ^* is the qc axis component of the current that flows through the synchronous machine 1. The estimated speed ωest is the rotor speed (angular frequency) ω estimated by the control device 2. The voltage command vdc ^* is a dc axis component of the voltage applied to the stator of the synchronous machine 1. The voltage command vqc ^* is a qc-axis component of a voltage applied to the stator of the synchronous machine 1.

In the following, it is assumed that the current commands idc ^* and iqc ^* are input from an external device, but the control device 2 further includes a current command generation unit that generates the current commands idc ^* and iqc ^* based on a torque command or the like. It is possible to provide a configuration.

The coordinate conversion unit 25 converts the voltage commands vdc ^* and vdc ^* output from the voltage command generation unit 24 from a dcqc axis rotation coordinate system to a three-phase fixed coordinate system. Similar to the coordinate conversion unit 23, the coordinate conversion unit 25 converts the dcqc-axis rotation coordinate system into a three-phase fixed coordinate system by using the estimated rotation phase angle θest.

Hereinafter, the voltage commands vdc ^* and vdc ^* coordinate-converted by the coordinate conversion unit 25 are referred to as voltage commands vu ^* , vv ^* and vw ^* . The voltage command vu ^* is a voltage applied to the U phase of the stator, the voltage command vv ^* is a voltage applied to the V phase of the stator, and the voltage command vw ^* is applied to the W phase of the stator. Voltage.

The modulation unit 26 generates control signals vu ′, vv ′, vw ′ for controlling ON / OFF of each switching element of the inverter 21 based on the voltage commands vu ^* , vv ^* , vw ^* . The control signal vu ′ is a binary signal corresponding to ON / OFF of the U-phase switching element. The same applies to the control signals vv ′ and vw ′.

The modulation unit 26 generates the control signals vu ′, vv ′, vw ′ by modulating the voltage commands vu ^* , vv ^* , vw ^* by PWM (Pulse-Width Modulation) using a triangular wave, for example. Can do. The modulation unit 26 inputs the generated control signals vu ′, vv ′, vw ′ to the inverter 21.

  The voltage detector 27 detects the control signals vu ′, vv ′, vw ′ output from the modulation unit 27 and inputs them to the speed / rotation phase angle estimation unit 29.

The high frequency superimposing unit 28 superimposes the high frequency voltage vdh on the voltage command vdc ^* according to the switching signal input from the control method switching unit 30. The high frequency voltage vdh is a voltage value superimposed on the voltage command vdc ^* .

  Here, FIG. 3 is a diagram illustrating the high frequency superimposing unit 28. As shown in FIG. 3, the high frequency superimposing unit 28 includes a high frequency voltage calculating unit 31.

When 1 is input as the switching signal, the high-frequency voltage calculation unit 31 calculates vdh = Vdh ^* × sin (2fdh × t) and outputs the high-frequency voltage vdh. Vdh ^* is a set value of the amplitude of the high frequency voltage. fdh is the frequency of the high frequency voltage. The high frequency voltage vdh output from the high frequency voltage calculation unit 31 is superimposed on the voltage command vdc ^* and input to the coordinate conversion unit 25. The switching signal will be described later.

On the other hand, when 0 is input as the switching signal, the high-frequency voltage calculation unit 31 calculates vdh = 0 × sin (2fdh × t) and outputs 0. Therefore, the voltage command vdc ^* is input to the coordinate conversion unit 25 as it is.

In addition, when a high frequency current is superimposed as a high frequency signal, the high frequency superimposing unit 28 may superimpose the high frequency current on the current commands idc ^* and iqc ^* according to the switching signal. For example, when the high frequency current idh is superimposed on the current command idc ^* , the high frequency current idh is idh = Idh ^* × sin (2fdh × t). Idh ^* is a set value of the amplitude of the high-frequency current idh.

  The speed / rotation phase angle estimation unit 29 (hereinafter referred to as “estimation unit 29”) estimates the rotor speed ω and the rotation phase angle θ of the synchronous machine 1, and calculates the estimated speed ωest and the estimated rotation phase angle θest. . The estimated speed ωest output from the estimation unit 29 is input to the voltage command generation unit 24 and the control method switching unit 30. The estimated rotational phase angle θest is input to the coordinate conversion units 23 and 26 and used for coordinate conversion.

  FIG. 4 is a diagram illustrating a configuration of the estimation unit 29. As shown in FIG. 4, the estimation unit 29 includes a first error estimation unit 32, a second error estimation unit 33, a PLL control unit 34, and an integrator 35.

  The first error estimating unit 32 calculates an error Δθ1 between the rotational phase angle θ and the estimated rotational phase angle θest by an arbitrary method using the high-frequency voltage vh. The second error estimating unit 33 calculates an error Δθ2 between the rotational phase angle θ and the estimated rotational phase angle θest by an arbitrary method that does not use the high-frequency voltage vh.

  The PLL controller 34 receives the error Δθ1 or the error Δθ2 according to the switching signal. The PLL control unit 34 receives an error Δθ1 when the switching signal is 1, and receives an error Δθ2 when the switching signal is 0. The PLL control unit 34 performs PLL control so that the input error Δθ becomes zero, and calculates the estimated speed ωest. The estimated speed ωest output from the PLL control unit 34 is input to the integrator 35.

  The integrator 35 integrates the estimated speed ωest to calculate the estimated rotational phase angle θest.

  That is, when the switching signal is 1, the estimation unit 29 calculates the estimated speed ωest and the estimated rotational phase angle θest using the high-frequency voltage vh. When the switching signal is 0, the estimation unit 29 performs estimation using a method that does not use the high-frequency voltage vh. The speed ωest and the estimated rotational phase angle θest are calculated.

  Here, each of the first error estimation unit 32 and the second error estimation unit 33 will be described in detail. First, an example of a method for calculating the error Δθ1 by the first error estimating unit 32 will be described.

  Generally, when the error Δθ is 0, that is, when the actual dq axis matches the estimated dcdq axis, the dq axis voltage equation is expressed by the following equation.

  In Equation (1), ωe is the electrical angular rotation frequency, vd is the d-axis voltage, vq is the q-axis voltage, id is the d-axis current, iq is the q-axis current, R is the winding resistance of the synchronous machine 1, and Ψ is the magnetic flux , Ω is the speed (angular frequency) of the synchronous machine 1, Ld is a d-axis inductance, Lq is a q-axis inductance, and p is a differential operator (d / dt).

  On the other hand, when the error Δθ is not 0, that is, when an error occurs between the actual dq axis and the estimated dcdq axis, the dq axis voltage equation is expressed by the following equation.

  In Expression (2), vdc is a dc-axis voltage (estimated value of d-axis voltage), vqc is a qc-axis voltage (estimated value of q-axis voltage), idc is a dc-axis current (estimated value of d-axis current), and iqc is qc-axis current (estimated value of q-axis current).

When the high-frequency voltage vdh is superimposed on the dc-axis voltage (voltage command value vdc ^* ) as in the present embodiment, the differential term of Equation (2) is expressed by the following equation.

  From equation (7), it can be seen that when the error Δθ is 0, even if the high frequency voltage vdh is superimposed on the dc axis voltage, sin2Δθ = 0, and no high frequency current flows through the qc axis (piqc = 0). On the other hand, when the error Δθ is not 0, when the high frequency voltage vdh is superimposed on the dc axis voltage, sin2Δθ ≠ 0, and it can be seen that a high frequency current flows in the qc axis (piqc ≠ 0).

  When the error Δθ is sufficiently small, it can be approximated as cos2Δθ = 1 and sin2Δθ = 2Δθ. Therefore, equation (7) can be modified as follows.

  In equation (8), dt is the current sampling interval, and iqc′_pp is the amplitude of the qc-axis high-frequency current (the high-frequency component of the qc-axis current). The first error estimator 32 can calculate the error Δθ1 by the above equation (8).

  Here, FIG. 5 is a diagram illustrating the first error estimation unit 32 that calculates the error Δθ1 by the above method. As shown in FIG. 5, the first error estimation unit 32 includes an amplitude detection unit 36.

  The amplitude detector 36 detects the amplitude iqc′_pp of the high-frequency current from the qc-axis current iqc input from the coordinate converter 23. The first error estimating unit 32 calculates the error Δθ1 by substituting the amplitude iqc′_pp output from the amplitude detecting unit 36 and the high frequency voltage vdh input from the high frequency superimposing unit 28 into Expression (8). . The inductances Ld and Lq may be acquired with reference to an inductance table associated with the current iqc, for example. The sampling interval dt is a set value.

  Here, FIG. 6 is a diagram illustrating the amplitude detector 36. As shown in FIG. 6, the amplitude detector 36 includes a bandpass filter 37 and an FFT analyzer 38.

  As shown in FIG. 7, the band-pass filter 37 passes a frequency component in a predetermined range including the frequency fdh of the high-frequency voltage vdh in the input current iqc, and attenuates a frequency component outside the range. Thereby, the band pass filter 37 detects the high frequency current iqc ′ having the frequency fdh from the current iqc.

  The cut-off frequency of the bandpass filter 37 may be constant or variable as long as the high-frequency current iqc ′ can be detected. The high frequency current iqc ′ output from the bandpass filter 37 is input to the FFT analysis unit 38.

  The FFT analysis unit 38 calculates the amplitude iqc′_pp of the high-frequency current iqc ′ detected by the bandpass filter 37. For example, as shown in FIG. 8, the FFT analysis unit 38 samples the high-frequency current iqc ′ four times during one cycle (= 1 / fdh) of the high-frequency voltage vdh, and four sampled current values Is used to calculate the amplitude iqc′_pp.

  Since the excessive frequency component is removed from the high-frequency current iqc ′ by the band-pass filter 37, the FFT analysis unit 38 can calculate the amplitude iqc′_pp with high accuracy as shown in FIG.

  The calculation method of the error Δθ1 by the first error estimation unit 32 is not limited to the above method, and can be arbitrarily selected from known methods using the high-frequency voltage vh.

  Next, the second error estimation unit 33 will be described. FIG. 9 is a diagram illustrating a configuration of the second error estimation unit 33. As shown in FIG. 9, a coordinate conversion unit 39, a harmonic observation unit 40, an inductance distribution approximation calculation unit 41, and a Δθ2 calculation unit 42 are provided.

  The coordinate conversion unit 39 converts the currents iu and iw detected by the current detector 22 into the αβ axis coordinate system of FIG. 2 and calculates the currents Iα and Iβ. Similarly, the coordinate conversion unit 39 converts the control signals vu ′, vv ′, vw ′ input from the voltage detector 27 into the αβ axis coordinate system, and calculates the voltages Vα, Vβ.

  The high frequency observation unit 40 calculates PWM harmonic components dIαfh / dt and dIβfh / dt of the currents Iα and Iβ calculated by the coordinate conversion unit 39. Similarly, the high frequency observation unit 40 calculates PWM harmonic components ΔVαhf and ΔVβhf of the voltages Vα and Vβ calculated by the coordinate conversion unit 39.

  The inductance distribution approximate calculation unit 41 calculates an inductance matrix L based on the PWM harmonic components dIαfh / dt, dIβfh / dt, ΔVαhf, and ΔVβhf.

  The Δθ2 calculation unit 42 calculates the error Δθ2 based on the inductance matrix L.

  Note that the method of calculating the error Δθ2 by the second error estimating unit 33 is not limited to the method using the PWM harmonics, and can be arbitrarily selected from known methods that do not use the high-frequency voltage vh.

  The control method switching unit 30 compares the frequency of the slot harmonics with the frequency of the high frequency voltage vhf, and outputs a 1 or 0 switching signal according to the comparison result.

  Here, the slot and the slot harmonic will be described. As shown in FIG. 10, the synchronous machine 1 includes a stator 101 and a rotor 102. The stator 101 includes a plurality of salient poles 103. The salient poles 103 are arranged at predetermined intervals so as to face the rotor 102. A slot 104 is formed between adjacent salient poles 103.

  The ease with which the magnetic flux passes varies depending on the positional relationship between the rotor 102, the salient pole 103, and the slot 104. As shown in FIG. 10, the salient pole 103 is easy to pass magnetic flux (large inductance), and the slot 104 is difficult to pass magnetic flux (small inductance). In such a synchronous machine 1, when the rotor 102 rotates, slot harmonics resulting from a change in inductance with the stator 101 are generated and superimposed on the current and voltage of the stator 101. In general, in the dq axis coordinate system, the slot harmonic frequency fslot is expressed by the following equation.

  In Equation (9), SL is the number of slots, fe is the electrical angular rotation frequency, fmech is the rotation frequency of the synchronous machine 1, and Pp is the number of pole pairs. As can be seen from the equation (9), the frequency fslot of the slot harmonic changes according to the frequency fmech of the synchronous machine 1. The frequency fslot of the slot harmonic increases as the frequency fmech of the synchronous machine 1 increases, and becomes closer to the frequency fdh of the high-frequency voltage vdh.

  When the frequency of the slot harmonic is close to the frequency of the high-frequency voltage vdh, a current corresponding to the slot harmonic passes through the bandpass filter 37 and an error occurs in the amplitude ipc′_pp calculated by the FFT analysis unit 38. Arise. For this reason, an error occurs in the rotational phase angle θ and the speed ω estimated using the high-frequency voltage vhf, causing the control of the synchronous machine 1 to become unstable and step out. Details of the influence of the slot harmonics on the amplitude ipc′_pp will be described later.

  Therefore, the control method switching unit 30 switches the control method of the synchronous machine 1 according to the switching signal depending on whether the frequency of the slot harmonics and the frequency of the high-frequency voltage vdh are close or separate.

  More specifically, the control method switching unit 30 outputs 1 as a control signal when the difference between the slot harmonic frequency and the frequency of the high-frequency voltage vdh is greater than or equal to a predetermined value. The deviation here refers to the difference or ratio between the frequency of the slot harmonic and the frequency of the high-frequency voltage vdh. The shift decreases as the frequency of the slot harmonic and the frequency of the high-frequency voltage vdh approach each other, and increases as the distance increases. In the following, it is assumed that the deviation is a difference.

Accordingly, when the frequency of the slot harmonic and the frequency of the high frequency voltage vdh are separated, the high frequency voltage vhf is superimposed on the voltage command vdc ^* , and the estimated speed is calculated based on the error Δθ1 calculated using the high frequency voltage vhf. ωest and the estimated rotational phase angle θest are calculated.

  When the frequency of the slot harmonics and the frequency of the high-frequency voltage vdh are far from each other, the current corresponding to the slot harmonics does not pass through the bandpass filter 37, so the influence of the slot harmonics is reduced. Therefore, the speed ω and the rotational phase angle θ can be accurately estimated using the high-frequency voltage vhf.

On the other hand, when the difference between the slot harmonic frequency and the frequency of the high-frequency voltage vdh is less than a predetermined value, the control method switching unit 30 outputs 0 as a control signal. Accordingly, when the frequency of the slot harmonic and the frequency of the high-frequency voltage vdh are close, the high-frequency voltage vhf is not superimposed on the voltage command vdc ^* , and the estimated speed is calculated based on the error Δθ2 calculated by the method not using the high-frequency voltage vhf. ωest and the estimated rotational phase angle θest are calculated.

  When the frequency of the slot harmonic and the frequency of the high-frequency voltage vdh are close, as described above, the influence of the slot harmonic on the error Δθ1 increases, but the influence on the error Δθ2 calculated by the method not using the high-frequency voltage vhf is small. . Therefore, the speed ω and the rotational phase angle θ can be accurately estimated based on the error Δθ2.

  Thus, by switching the control method of the synchronous machine 1 depending on whether the frequency of the slot harmonic and the frequency of the high-frequency voltage vdh are close to each other, the speed ω and the rotation phase angle caused by the slot harmonic are switched. The estimation error of θ can be suppressed, and the rotational phase angle θ of the synchronous machine 1 can be estimated with high accuracy. Thereby, instability and step-out of the control of the synchronous machine 1 can be suppressed.

  FIG. 11 is a diagram illustrating a configuration of the control method switching unit 30. The control method switching unit 30 first calculates the frequency of the slot harmonic from the estimated speed ωest input from the estimation unit 29 by the following formula.

  In Expression (10), ωslot_est is an estimated value of the angular frequency of the slot harmonic. Next, the control method switching unit 30 calculates a ratio α between the angular frequency ωdh of the high-frequency voltage vhf and the estimated angular frequency ωslot_est.

  The angular frequency ωdh may be set in advance, may be input from the high frequency superimposing unit 28, or may be calculated from the frequency fdh by the equation (12).

  Then, the control method switching unit 30 compares the ratio α with a predetermined threshold β and outputs a switching signal corresponding to the comparison result. The threshold value β may be set in advance or may be calculated during the control of the synchronous machine 1.

  When β ≧ α, that is, when the difference (ωdh−ωslot_est) between the frequency of the slot harmonic and the frequency of the high frequency voltage vdh is equal to or greater than a predetermined value (ωdh (1-β)), the control method switching unit 30 Is output.

  On the other hand, when β <α, that is, when the difference (ωdh−ωslot_est) between the frequency of the slot harmonic and the frequency of the high frequency voltage vdh is less than a predetermined value (ωdh (1−β)), the control method switching unit 30 , 0 is output.

  Here, the influence of the slot harmonics on the amplitude ipc′_pp will be described in detail. First, the case where there is no influence of slot harmonics will be described with reference to FIGS.

  When there is no influence of slot harmonics and the error Δθ is 0, the high frequency current iqc ′ does not flow even if the high frequency voltage vdh is superimposed as shown in FIG. On the other hand, when the error Δθ is not 0, as shown in FIG. 13, when the high frequency voltage vdh is superimposed, a high frequency current iqc ′ having the same frequency as the frequency of the high frequency voltage vdh flows. As described with reference to Expression (7), the error Δθ can be calculated from the amplitude iqc′_pp of the high-frequency current iqc ′.

  Next, a case where there is an influence of slot harmonics will be described with reference to FIGS.

  When there is an influence of slot harmonics and the error Δθ is 0, as shown in FIG. 14, even if the high frequency voltage vdh is superimposed, the high frequency current iqc ′ does not flow, but the current iqs corresponding to the slot harmonics is qc. It flows as an axial current. Therefore, current iqc includes current iqs.

  However, when the rotation frequency fmech of the synchronous machine 1 is small, the frequency fslot of the slot harmonic is also small from the equation (9). When the frequency fslot of the slot harmonic is smaller than the frequency fdh of the high frequency voltage vdh by a predetermined value or more, the current iqs is cut by the band pass filter 37 when the current iqc is input to the estimation unit 36 as shown in FIG. .

  When the error Δθ is not 0, as shown in FIG. 16, when the high-frequency voltage vdh is superimposed, a high-frequency current iqc ′ having the same frequency as the frequency of the high-frequency voltage vdh and a current iqs corresponding to the slot harmonic are qc-axis current. Flowing as. Therefore, the current iqc includes a high-frequency current iqc ′ and a current iqs.

  However, when the rotation frequency fmech of the synchronous machine 1 is small and the frequency fslot of the slot harmonic is smaller than the frequency fdh of the high frequency voltage vdh by a predetermined value or more, the current iqc is input to the estimation unit 36 as shown in FIG. The current iqs is cut by the band pass filter 37. On the other hand, the high frequency current iqc ′ passes through the band pass filter 37. Therefore, the error Δθ can be calculated from the amplitude iqc′_pp of the high-frequency current iqc ′ as in the case where there is no influence of the slot harmonics.

  Thus, when the frequency of the slot harmonic is small and the frequency of the slot harmonic and the frequency of the high-frequency voltage vdh are separated, the current iqs is cut, so that the influence of the slot harmonic is suppressed. Therefore, the speed ω and the rotational phase angle θ can be accurately estimated using the high-frequency voltage vdh. This is the same even when the slot harmonic frequency is higher than the high-frequency voltage vdh.

  On the other hand, when the frequency of the slot harmonic is large and the frequency of the slot harmonic is close to the frequency of the high-frequency voltage vdh, the influence of the slot harmonic becomes large. When the error Δθ is not 0, as shown in FIG. 18, when the high frequency voltage vdh is superimposed, the high frequency current iqc ′ and the current iqs flow as the qc axis current. This is the same as when the slot harmonic frequency is small.

  However, when the frequency of the slot harmonic is large, as shown in FIG. 19, even if the current iqc is input to the bandpass filter 37, the current iqs cannot be sufficiently reduced, and the high-frequency current iqc ′ on which the current iqs is superimposed. Is output. For this reason, an error occurs in the amplitude iqc′_p−p of the high-frequency current iqc ′, and the estimation accuracy of the speed ω and the rotational phase angle θ decreases.

  As described above, the effect of the slot harmonics increases as the frequency of the slot harmonics approaches the frequency of the high-frequency voltage vdh. Since the control device 2 according to the present embodiment switches the control method depending on whether the influence of the slot harmonics is large or small, the speed ω and the rotation phase angle θ can be accurately estimated.

  In this embodiment, the speed ω and the rotational phase angle θ are estimated by two types of methods, but may be estimated by three or more types of methods. In this case, for example, a plurality of threshold values β to be compared with the ratio α may be set, and an estimation method may be selected according to the comparison result.

  In this embodiment, the estimated speed ωest and the estimated rotational phase angle θest are calculated by PLL control, but a method of directly calculating the estimated speed ωest and the estimated rotational phase angle θest may be used.

Furthermore, in the present embodiment, the high frequency voltage vdh is superimposed on the voltage command vdc ^* . However, the high frequency voltage may be superimposed on both the voltage command vqc ^* and the voltage commands vdc ^* and vqc ^* . Even if the high-frequency voltage vqh is superimposed on the voltage command vqc ^* , the phase angle error can be calculated by a known method using the high-frequency voltage vqh. In the present embodiment, the high frequency voltage vdh is superimposed on the voltage command vdc ^* . However, as described above, the high frequency current is superimposed on the current command and the voltage applied to the synchronous machine 1 (the output voltage of the inverter 21) is added. Based on this, the phase angle error may be calculated.

(Second Embodiment)
A system according to the second embodiment will be described with reference to FIGS. FIG. 20 is a diagram illustrating a configuration of a system according to the present embodiment. The control device 2 according to the present embodiment controls the synchronous machine 1 sensorlessly using a control method using the high-frequency voltage vdh without switching the control method. For this reason, as shown in FIG. 20, the control device 2 does not include the voltage detector 27 or the control method switching unit 30.

  And this control apparatus 2 changes the amplitude and frequency of the high frequency voltage vdh instead of switching a control system. Here, FIG. 21 is a diagram illustrating a configuration of the high frequency superimposing unit 28 of the control device 2 according to the present embodiment. As shown in FIG. 21, the high frequency superimposing unit 28 includes a slot harmonic frequency calculating unit 43, a high frequency voltage frequency determining unit 44, a high frequency voltage amplitude determining unit 45, and a high frequency voltage calculating unit 31.

  The slot harmonic frequency calculation unit 43 (hereinafter referred to as “calculation unit 43”) calculates the frequency of the slot harmonics and inputs it to the high frequency voltage frequency determination unit 44 and the high frequency voltage amplitude determination unit 45. The calculation unit 43 may calculate the estimated angular frequency ωslot_est or the frequency fslot (= ωslot_est / 2π) as the slot harmonic frequency. The estimated angular frequency ωslot_est can be calculated from the estimated speed ωest using equation (10). Hereinafter, it is assumed that the calculation unit 43 calculates the frequency fslot.

  The high frequency voltage frequency determination unit 44 (hereinafter referred to as “frequency determination unit 44”) determines the frequency fdh of the high frequency voltage vdh based on the slot harmonic frequency fslot input from the calculation unit 43. The frequency determination unit 44 changes the frequency fdh according to the frequency fslot, and determines the frequency fdh so that the difference between the frequency fslot and the frequency fdh is equal to or greater than a predetermined value. The frequency fdh determined by the frequency determination unit 44 is input to the high frequency voltage calculation unit 31.

  In the present embodiment, the band-pass filter 37 of the estimation unit 29 preferably has a variable cutoff frequency so that the frequency fdh determined by the frequency determination unit 44 can pass.

  The high frequency voltage amplitude determining unit 45 (hereinafter referred to as “amplitude determining unit 45”) determines the amplitude Vdh of the high frequency voltage vdh based on the slot harmonic frequency fslot input from the calculating unit 43. The amplitude determination unit 45 changes the amplitude Vdh according to the frequency fslot, and increases the amplitude Vdh as the difference between the frequency fslot and the frequency fdh is smaller. The amplitude Vdh determined by the amplitude determination unit 45 is input to the high frequency voltage calculation unit 31.

The high frequency voltage calculation unit 31 calculates vdh = Vdh × sin (2fdh × t) and outputs the high frequency voltage vdh. The high frequency voltage vdh output from the high frequency voltage calculation unit 31 is superimposed on the voltage command vdc ^* and input to the coordinate conversion unit 25.

  As described above, when the slot harmonic frequency fslot is close to the frequency fdh of the high-frequency voltage vdh, the estimation accuracy of the speed ω and the rotation phase angle θ decreases due to the effect of the slot harmonic. However, the control device 2 according to the present embodiment changes the frequency fdh of the high-frequency voltage vdh so that the difference between the frequency fslot and the frequency fdh is equal to or greater than a predetermined value. Thereby, the influence of the slot harmonic can be suppressed and the speed ω and the rotational phase angle θ can be estimated with high accuracy.

  Further, the control device 2 increases the amplitude Vdh of the high-frequency voltage vdh as the frequency fdh is closer to the frequency vdh. As can be seen from equation (8), the higher the high-frequency voltage vdh, the smaller the influence of the amplitude ipc′_pp that causes an error due to the slot harmonics. Therefore, the influence of the slot harmonics can be made relatively small, and the estimation accuracy of the speed ω and the rotation phase angle θ can be improved.

  In the present embodiment, the high frequency superimposing unit 28 may keep the amplitude Vdh constant and change only the frequency fdh. For example, the frequency fdh may be changed stepwise as shown in FIG. 22, or may be changed continuously as shown in FIG.

  Further, the high frequency superimposing unit 28 may keep the frequency fdh constant and change only the amplitude Vdh. For example, the amplitude Vdh may be changed stepwise as shown in FIG. 24, or may be changed continuously as shown in FIG.

  Furthermore, the high frequency superimposing unit 28 may change both the frequency fdh and the amplitude Vdh, as shown in FIG. Thereby, the torque ripple of the synchronous machine 1 can be reduced and drive efficiency can be improved. The reason is as follows.

  In general, there is an upper limit (carrier frequency) for the frequency fdh. For this reason, when the frequency fslot of the slot harmonic is high, there is a possibility that the frequency fdh larger than the frequency fslot by a predetermined value or more cannot be set. If a sufficiently large frequency fdh cannot be set, the current iqs corresponding to the slot harmonic cannot be cut by the bandpass filter 37, and a ripple occurs in the estimated rotational phase angle θest. The synchronous machine 1 generates torque ripple (torque fluctuation). The torque ripple T of the synchronous machine 1 is expressed by the following equation.

  As can be seen from Equation (13), the torque ripple T increases as the error Δθ increases. As shown in FIGS. 22 and 23, when only the frequency fdh is changed, the torque ripple T of the synchronous machine 1 may increase due to the above reason. When the frequency fdh is increased, there is a problem that the design of the bandpass filter 37 becomes difficult. Furthermore, as the amplitude Vdh is larger, the iron loss and the copper loss due to the high-frequency current iqc ′ are increased, and the driving efficiency of the synchronous machine 1 is lowered.

  On the other hand, when both the frequency fdh and the amplitude Vdh are changed, the amount of change of each of the frequency fdh and the amplitude Vdh can be reduced as compared with the case where only one of them is changed. Therefore, an increase in torque ripple and a decrease in driving efficiency as described above can be suppressed.

  The method for determining the frequency fdh and the amplitude Vdh is arbitrary. The frequency fdh and the amplitude Vdh may be set in advance in a table, a value that minimizes torque ripple and loss may be calculated, or may be changed according to inductance saturation.

1: synchronous machine, 2: synchronous machine control device, 21: inverter, 22: current detector, 23: coordinate converter, 24: voltage command generator, 25: coordinate converter, 26: modulator, 27: voltage detection 28: high frequency superposition unit 29: speed / rotation phase angle estimation unit 30: control method switching unit 31: high frequency voltage calculation unit 32: first error estimation unit 33: second error estimation unit 34: PLL control unit, 35: integrator, 36: amplitude detection unit, 37: bandpass filter, 38: FFT analysis unit, 39 coordinate conversion unit, 40: harmonic observation unit, 41: inductance distribution approximation calculation unit, 42: Δθ2 Calculation unit, 43: Slot harmonic frequency calculation unit, 44: High frequency voltage frequency determination unit, 45: High frequency voltage amplitude determination unit, 101: Stator, 102: Rotor, 103: Salient pole, 104: Slot

Claims (9)

An inverter that drives the synchronous machine;
A voltage command generator for generating a voltage command based on the current command and the output current of the inverter;
A high frequency superimposing unit capable of superimposing a predetermined high frequency signal on the current command or the voltage command;
An estimation unit for calculating an estimated rotational phase angle of the synchronous machine;
With
The high-frequency superimposing unit superimposes the high-frequency signal when the difference between the slot harmonic frequency and the frequency of the high-frequency signal is equal to or greater than a predetermined value, and does not superimpose the high-frequency signal when less than the predetermined value. Machine control device. The estimation unit calculates the estimated rotational phase angle based on a high-frequency component of the output current of the inverter when a difference between the frequency of the slot harmonic and the frequency of the high-frequency signal is a predetermined value or more. Item 2. A synchronous machine control device according to Item 1. When the deviation between the frequency of the slot harmonic and the frequency of the high-frequency signal is less than a predetermined value, the estimation unit calculates the estimated rotational phase angle based on at least one of PWM harmonics, magnet magnetic flux, and inductance. The synchronous machine control device according to claim 1 or claim 2 to be calculated. An inverter that drives the synchronous machine;
A voltage command generator for generating a voltage command based on the current command and the output current of the inverter;
A high frequency superimposing unit that superimposes a high frequency signal on the current command or the voltage command;
An estimation unit that calculates an estimated rotational phase angle of the synchronous machine based on a high-frequency component of the output current of the inverter;
With
The high frequency superimposing unit is a synchronous machine control device that changes at least one of a frequency and an amplitude of the high frequency signal according to a frequency of a slot harmonic. The high-frequency superimposing unit changes the frequency of the high-frequency signal so that a difference between the frequency of the slot harmonic, the frequency of the high-frequency signal, and the frequency of the slot harmonic becomes a predetermined value or more. 4. The synchronous machine control device according to 4. The synchronous machine control device according to claim 4 or 5, wherein the high-frequency superimposing unit increases the amplitude of the high-frequency signal as the difference between the frequency of the high-frequency signal and the frequency of the slot harmonic is small. The synchronous machine control apparatus according to any one of claims 1 to 6, wherein the synchronous machine is a synchronous reluctance motor or a permanent magnet synchronous machine. A synchronous machine,
An inverter for driving the synchronous machine;
A voltage command generator for generating a voltage command based on the current command and the output current of the inverter;
A high frequency superimposing unit capable of superimposing a predetermined high frequency signal on the current command or the voltage command;
An estimation unit for calculating an estimated rotational phase angle of the synchronous machine;
With
The high-frequency superimposing unit superimposes the high-frequency signal when the difference between the slot harmonic frequency and the frequency of the high-frequency signal is equal to or greater than a predetermined value, and does not superimpose the high-frequency signal when less than the predetermined value. Machine drive system. A synchronous machine,
An inverter for driving the synchronous machine;
A voltage command generator for generating a voltage command based on the current command and the output current of the inverter;
A high-frequency superimposing unit that superimposes a high-frequency signal on the current command and the voltage command;
An estimation unit that calculates an estimated rotational phase angle of the synchronous machine based on a high-frequency component of the output current of the inverter;
With
The high frequency superimposing unit is a synchronous machine driving system that changes at least one of a frequency and an amplitude of the high frequency signal according to a frequency of a slot harmonic.

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