JP5618854B2 - Synchronous motor drive system - Google Patents

Description

  Embodiments described herein relate generally to a synchronous motor drive system that drives a synchronous motor by a sensorless control method.

  An electric motor using a field winding, a permanent magnet, or an iron core as a rotor is called a synchronous motor because it energizes the stator in synchronization with the rotation of the electric motor and generates torque. In more detail, what generates torque by using a field winding and obtains torque is simply called a synchronous motor, and a permanent magnet is used for the rotor to generate rotor flux, and magnet torque is What is obtained is called a permanent magnet synchronous motor, and what obtains reluctance torque using only an iron core as a rotor is called a reluctance motor or the like.

  In recent years, a diode is inserted into an embedded magnet type permanent magnet synchronous motor combining these, or a coil short-circuited in a rotor as a field winding as in Non-Patent Document 1, and a high-frequency field is inserted into this coil. A half-wave rectifying brushless synchronous motor that obtains a field by applying a permanent magnet synchronous motor, and a half-wave rectifying brushless permanent motor that combines this motor with a permanent magnet synchronous motor have been developed.

  In motors belonging to the category of “synchronous motors” as described above, control is generally performed to obtain high torque with high efficiency by measuring the rotation angle during operation and energizing the appropriate phase angle. . However, in order to measure the rotation angle, it is necessary to install a rotation angle sensor typified by a resolver, encoder (pulse generator), etc., and the rotation angle sensor is not required because of the need for saving installation space and cost reduction. A control scheme has been developed. This control method is generally referred to as “sensorless control”.

  In sensorless control, the rotation angle is estimated by the rotation angle estimator from the voltage and current observed by the motor control device, utilizing the fact that the rotation angle information is included in the relationship between the voltage and current applied to the motor. The method is general and is described in detail in, for example, Non-Patent Document 2.

  Sensorless control can be roughly divided into two categories. "High-speed sensorless control" used in the high-speed rotation range and "Low-speed sensorless control" used in the low-speed rotation range. For example, in the case of a permanent magnet synchronous motor, the high speed sensorless control uses the fact that the counter electromotive voltage induced in the stator by the permanent magnet magnetic flux of the rotor is synchronized with the rotation angle, and the counter electromotive voltage component from the voltage and current of the motor. The rotation angle is estimated by extracting. Although various methods have been proposed for low-speed sensorless control, the basic concept is that a high-frequency voltage or high-frequency current is applied based on the electrical saliency of the rotor, and the rotation included in the response. The rotation angle estimation is realized by extracting the angle information using a bandpass filter or the like. The low speed sensorless control is described in Patent Documents 1 and 2.

  The rotation angle estimated in the low-speed sensorless control has an estimation error of 0 ° or 180 ° in principle, and cannot be distinguished. This is because the estimation principle in low-speed sensorless control is based on electrical saliency. The electrical saliency is a function of 2Δθ with respect to the estimation error Δθ. Therefore, when the estimation is performed so that 2Δθ becomes zero, the estimation converges to either rotation Δθ = 0 ° or 180 °. . As a technique for dealing with such a problem, a technique called “polarity discrimination” has been developed and is described in detail in Non-Patent Document 3.

  The basic principle of polarity discrimination is that a stator DC current is applied in the same direction as the field magnetic flux, and magnetic saturation is generated by the magnetic flux generated by this current and the field magnetic flux, thereby detecting that the inductance is reduced. Is.

  When a stator DC current is applied in the direction opposite to the field magnetic flux, magnetic saturation does not occur and the inductance does not decrease. Therefore, polarity determination can determine whether the estimated rotation angle error is 0 ° or 180 ° by applying positive and negative stator DC currents and measuring the inductance in each case.

  Further, as a sensorless starting position detection method of the “half-wave rectifying brushless synchronous motor” described in Non-Patent Document 1, a method described in Non-Patent Document 4 has been proposed. Non-Patent Document 4 describes the realization of low-speed sensorless control including driving and magnetic pole discrimination in a half-wave rectifying brushless synchronous motor having a rotor coil with a diode inserted therein.

  In Non-Patent Document 4, on the premise that magnetic flux generated by a rotor coil inserted by a diode is used as a driving magnetic flux, a triangular wave excitation current is passed to excite the rotor coil. At this time, the rotation angle is estimated using the rotation angle information appearing in the control stator voltage.

  However, even in Non-Patent Document 4, since the rotation angle is estimated based on the term of 2Δθ as in the case of the low-speed sensorless described above, it is necessary to determine the magnetic pole, and after estimating the phase angle of 0 ° or 180 ° The magnetic pole discrimination is realized by estimating the on / off state of the stator winding diode. That is, the procedure is “initial phase estimation”, “magnetic pole discrimination”, and “drive start” in this order. In Non-Patent Document 4, estimation is possible even during driving, and on the premise that the phase angle does not reverse after the magnetic pole discrimination is performed, 2Δθ is obtained by a calculation method almost similar to the initial phase estimation, and the rotation angle is estimated. Yes.

  Further, Non-Patent Document 5 is an application of the concept of Non-Patent Documents 1 and 4. The method described in Non-Patent Document 5 uses a half-wave rectifying brushless synchronous motor, applies a rotating high-frequency voltage that rotates forward and backward in the dq axis coordinate system used for vector control of the synchronous motor, and In the forward high-frequency current and the reverse high-frequency current, the phase angle where the amplitude is spatially maximized is searched for, and the average value is used as the estimated rotation angle. According to this method, the resistance of the rotor winding It is stated that it is possible to remove estimation errors caused by components.

Japanese Patent Laid-Open No. 11-150983 JP 7-245981 A

IEEJ Transaction D, Vol.107, No.10, 1987 "Principle and basic characteristics of half-wave rectifying brushless synchronous motor" Koyama, Toba, Higuchi, Yamada (Nagasaki University) Design and Control of Embedded Magnet Synchronous Motor Yoji Takeda, Nobuyuki Matsui, Shigeo Morimoto, Yukio Honda Ohm Corporation IEICE Technical Committee Automotive Society, March 8, 2002, VT-02-12, p67 "Rotational Sensorless Control of Permanent Magnet Reluctance Motor" Yosuke Nakazawa (Toshiba) Proceedings of Industrial Conference of the Institute of Electrical Engineers of Japan, 1, pp.589-598 (2004) “Modeling of sensorless driving position estimation system for half-wave rectifying brushless synchronous motor” Oyama, Abe, Higuchi, Left village, Kono (Nagasaki University) IEEJ Transactional D, Vol. 128, No. 10, 2008 "New Initial Phase Estimation Method for Self-Excited Hybrid Field Synchronous Motor" Shinnaka, Yashiro (Kanagawa University, Bosch)

  As described above, there are a number of methods for sensorless control of a synchronous motor, and not only a method proposal but also a drive with an actual device has been put into practical use. However, the following problems still exist in the current system, particularly in the low speed sensorless control.

  First, there is a problem of necessity of magnetic pole discrimination due to restrictions on the rotation angle estimation principle. As described above, a normal synchronous motor that does not use a rotor coil with a diode inserted has limitations on the rotation angle estimation principle, and it is necessary to separately perform rotation angle estimation and magnetic pole discrimination. In particular, the magnetic pole discrimination needs to ignite a large positive / negative direct current compared to the high frequency current for estimation, and it may be adversely affected on the driving torque, so that it is not realistic to carry out the driving. Therefore, it is usually necessary to determine the magnetic pole at the time of starting, and thereafter control so that the estimated rotation angle error is within ± 90 °. Conversely, when the estimated rotation angle error exceeds ± 90 °, it is necessary to detect the situation, perform the start sequence again, and perform the magnetic pole determination again.

  For this reason, at the time of starting and restarting, it takes about 0.1 seconds at the shortest time for magnetic pole discrimination, and it is used for driving applications (for example, hybrid cars, trains, etc.) that are operated by inputting a torque command or the like from a human system. In some cases, this may lead to a driver's discomfort. This problem can also be said for the sensorless control method of the synchronous motor without a half-wave rectifying brush described in Non-Patent Document 4.

  Further, there is a problem of calculation complexity in the sensorless control system described in Non-Patent Document 5. Here, it can be said that the method described in Non-Patent Document 5 is a method corresponding to the problem of necessity of magnetic pole discrimination due to the restriction of the rotation angle estimation principle described above. That is, it can be said that the method described in Non-Patent Document 5 performs magnetic pole discrimination and rotation angle estimation simultaneously by searching for a phase angle that maximizes the amplitude of the rotating high-frequency current.

  However, the method described in Non-Patent Document 5 applies a high-frequency voltage of forward rotation and reverse rotation, observes the rotation high-frequency current, searches for a current having the maximum amplitude, and performs each of forward rotation and reverse rotation. A complicated process of averaging the phase angle of the maximum amplitude current is required. Such computational complexity generally increases the proportion of computation for this processing relative to the control cycle of an electric motor processed at a frequency of several hundreds to several kilohertz, and thus other advanced controls. There is a possibility that it becomes a restriction that the operation cannot be incorporated.

  From the opposite perspective, the method described in Non-Patent Document 5 requires an expensive and high-performance computing device (for example, a high-performance microcomputer or DSP: Digital Signal Processor) in order to incorporate this processing within the constraints of computation time. It can also be said.

  Further, in principle, the method described in Non-Patent Document 5 can update the estimated value for the first time when the forward rotation and the reverse rotation high-frequency voltage are applied at least one period at a time and both response current values are obtained. However, this lengthens the update period of the estimated value and may lead to deterioration of the controllability of the motor that should be capable of high-speed control response. Although this point may be improved by increasing the frequency of the applied high-frequency voltage, the controllability is degraded due to the limitations of the estimation method despite the fact that a relatively faster response is possible. Is the same.

  The present invention solves the above-mentioned problems of the prior art, and in the low-speed sensorless control of a synchronous motor having a rotor coil with a diode inserted, the estimation of the rotation angle and the magnetic pole discrimination can be performed without the need for complicated processing. It is an object of the present invention to provide a synchronous motor drive system based on a sensorless control method that can be performed simultaneously, in other words, the estimated range is not −90 ° ≦ Δθ ≦ + 90 ° but −180 ° ≦ Δθ ≦ + 180 °. .

In order to solve the above problems, a synchronous motor drive system according to an embodiment includes a synchronous motor and a control device that controls the synchronous motor. The synchronous motor includes a stator having a stator coil, and a diode inserted therein. A rotor having a rotor coil formed, and the control device includes a high frequency magnetic flux generation control unit that generates a high frequency magnetic flux command for generating a high frequency magnetic flux in the stator, and the high frequency magnetic flux generation control unit. Based on at least one of the high-frequency voltage and the high-frequency current generated in the stator coil according to the generated high-frequency magnetic flux command, the difference in the high-frequency voltage according to the presence or absence of the current flowing through the diode or the high-frequency current It calculates the differential rotation of estimating the rotation angle of the synchronous motor based on the difference of the difference or the high frequency current calculated the high-frequency voltage And having a degree estimation unit.

It is a block diagram which shows the structure of the synchronous motor drive system of the form of Example 1. FIG. It is a figure which shows the equivalent circuit of the rotor coil in the synchronous motor of the synchronous motor drive system of the form of Example 1. FIG. It is a figure which shows the equivalent circuit of the rotor coil in the synchronous motor of the synchronous motor drive system of the form of Example 1. FIG. It is a figure which shows the example of a waveform of the high frequency magnetic flux command by the high frequency magnetic flux generation control part of the synchronous motor drive system of the form of Example 1. FIG. It is a figure which shows the definition of a synchronous motor model and coordinates. It is a schematic diagram of the synchronous motor of the synchronous motor drive system of the form of Example 1. FIG. It is a figure which shows the difference of (delta) axis | shaft voltage used as the parameter | index of the estimation error of the synchronous motor drive system of the form of Example 1. FIG. It is a figure which shows the difference of (delta) axis | shaft voltage used as the parameter | index of the estimation error of the synchronous motor drive system of the form of Example 2. FIG. It is a PLL block diagram of rotation angle estimation. It is a figure which shows the difference of the (gamma) -axis voltage used as the parameter | index of the magnetic pole position of the synchronous motor drive system of the form of Example 3. FIG. It is a block diagram which shows the structure of the synchronous motor drive system of the form of Example 4. FIG. It is a wave form diagram which shows the high frequency current response with respect to a high frequency voltage command.

  Embodiments of a synchronous motor drive system according to the present invention will be described below in detail with reference to the drawings.

  Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is a block diagram illustrating a configuration of a synchronous motor drive system according to a first embodiment of the present invention. As shown in FIG. 1, the synchronous motor drive system according to the present embodiment includes a synchronous motor control device 1, an inverter 2, a current detector 6, and a synchronous motor 7.

  The control device 1 that controls the synchronous motor 7 via the inverter 2 includes a high-frequency magnetic flux generation control unit 11, an electric motor control unit 12 a, a PWM processing unit 13, and a rotation angle estimation unit 14. Further, the motor control unit 12a includes a torque / current conversion unit 121, a current control unit 122, a dq / 3 phase conversion unit 123, and a three phase / dq conversion unit 124. Detailed description of each component will be described later.

  The inverter 2 converts the DC voltage into a three-phase AC voltage by turning on and off the built-in switching element based on the gate command output by the PWM processing unit 13 in the control device 1.

  The synchronous motor 7 generates a magnetic field when a three-phase AC current based on the three-phase AC voltage output from the inverter 2 flows, and generates a torque by a magnetic interaction with the rotor 3. The synchronous motor 7 includes a rotor 3 and a stator 4. The stator 4 includes an armature coil as a stator coil. Further, the rotor 3 has a coil (rotor coil 5) of one turn or more arranged so that the exciting magnetic flux from the stator 4 is interlinked, and a diode is inserted into the rotor coil 5. ing.

  FIG. 2 is a diagram showing an equivalent circuit of the rotor coil 5 in the synchronous motor 7 of the synchronous motor drive system of the present embodiment. The rotor coil 5 has at least one diode in the circuit, and is configured such that the diode is turned on / off according to the direction of the current flowing through the coil. In FIG. 2, the coil and the diode are connected in series, but other electrical components such as a resistor and a capacitor can be inserted. FIG. 3 is a diagram showing another example of an equivalent circuit of the rotor coil 5 in the synchronous motor 7 of the synchronous motor drive system of the present embodiment, where the voltage due to the change in the interlinkage magnetic flux of the coil is in the direction in which the diode is turned on. The total resistance of the circuit is different depending on whether the circuit is turned off. On the other hand, in the example of FIG. 2, when the diode is on, the resistance value of the coil is a circuit resistance, and when the diode is off, the resistance is substantially infinite.

  The PWM processing unit 13 generally uses triangular wave comparison PWM (Pulse Width Modulation), and generates an on / off gate command for each switching element of the inverter by comparing the voltage command with the triangular wave carrier. Other PWM processing methods include hysteresis PWM and space vector PWM.

  Hysteresis PWM receives a three-phase current command as input and generates a switching element on / off gate command depending on whether the three-phase current response is within the error width (hysteresis width) specified for the current command. To do. The space vector PWM receives a three-phase voltage command as an input, calculates the voltage vector to be output by the inverter and its output time according to the spatial position when the voltage command is considered as a vector, Generate on / off gate command.

  The current detection unit 6 detects a two-phase or three-phase current response value in the three-phase AC current flowing through the synchronous motor 7. The current response value indicates a response value with respect to a command value such as a three-phase voltage command, and here indicates the magnitude of the current flowing through the synchronous motor 7.

The high frequency magnetic flux generation control unit 11 generates a high frequency magnetic flux command for causing the stator 4 to generate a high frequency magnetic flux. That is, the high frequency magnetic flux generation control unit 11 calculates and outputs a high frequency magnetic flux command so that the stator coil of the synchronous motor 7 generates a high frequency magnetic flux. The physical meaning of the high frequency magnetic flux command is the synchronous motor. 7 is the voltage or current that is energized. Since the actual magnetic flux is generated in the stator coil by the current flowing through the stator coil, it is a direct stator current command, but indirectly, a high frequency magnetic flux is used as a voltage command for controlling the stator current. Directives may be included. The high frequency magnetic flux generation control unit 11 of this embodiment superimposes the high frequency magnetic flux command on the current command I _δ ^ref output by the torque / current conversion unit 121.

  FIG. 4 is a diagram illustrating a waveform example of a high frequency magnetic flux command by the high frequency magnetic flux generation control unit 11 of the synchronous motor drive system of the present embodiment. Although FIG. 4 shows an example in which the high frequency magnetic flux command is given as a triangular wave current command, it may be a rectangular wave voltage command, or may be a sine wave or other high frequency waveform. The high frequency here refers to a sufficiently high frequency component that does not cause disturbance such as torque with respect to the frequency synchronized with the rotation of the synchronous motor 7. In general, in the case of the rotation angle estimation at a low speed which is the object of the present proposal, the rotation frequency is 0 to several tens Hz, and therefore, a frequency component of several hundreds to several kiloHz is often set to a high frequency.

  Normally, a frequency equal to or higher than the control calculation frequency in the control device is not selected, but a higher frequency can be selected by using a hardware such as FPGA or providing a dedicated calculation unit separately from the calculation unit performing the control calculation. Sometimes it becomes possible.

  The electric motor control unit 12a performs a calculation for controlling the rotation of the electric motor. In FIG. 1, the motor control unit 12 a receives the torque command of the synchronous motor 7 as an input and outputs a voltage command applied to the synchronous motor 7. However, other configurations can be adopted, for example, the rotation speed It is also possible to input a command.

  As a content of the control calculation, a control method called vector control is adopted as a method for controlling the response of the synchronous motor 7 such as torque and rotational speed at high speed. The configuration of this embodiment shown in FIG. 1 is a configuration example using vector control. This vector control will be described using a permanent magnet synchronous machine using a permanent magnet for the rotor 3 as an example.

FIG. 5 is a diagram showing the definition of the synchronous motor model and coordinates. First, as shown in FIG. 5, as a coordinate system that rotates in synchronization with the rotation of the permanent magnet synchronous motor, the direction of the magnetic flux of the permanent magnet is defined as the d axis and the axis orthogonal to the d axis is defined as the q axis. Further, the U-phase winding direction is defined as the α axis, and the direction perpendicular thereto is defined as the β axis, and the angle up to the d axis direction with respect to the α axis direction is defined as the rotational phase angle θ of the synchronous motor. Based on this definition, the relationship between the voltage and current of the permanent magnet synchronous motor is expressed by equation (1).

Here, V _d is a d-axis voltage, and V _q is a q-axis voltage. Further, I _d is a d-axis current, and I _q is a q-axis current. R is a resistance. L _d is a d-axis inductance, and L _q is a q-axis inductance. Further, Φ is a permanent magnet magnetic flux, ω is a rotational speed, and p is a differential operator.

However, since the control device 1 does not have a rotation angle sensor and the rotation angle θ itself cannot be detected, the phase angle estimated by the control device 1 is used instead. Therefore, as shown in FIG. 5, the estimated phase angle is defined as θ _est, and the corresponding coordinate system is defined as γ-axis and δ-axis. When the estimation error Δθ occurs, the γδ axis is a position rotated by the estimation error Δθ from the dq axis.

The torque command in FIG. 1 is given by the host control system. The torque / current conversion unit 121 generates the γ-axis current command I _γ ^ref and the δ-axis current command I _δ ^ref by using the equation (2) based on the torque command.

Here, Trq ^ref is a torque command. Further, k is a constant and can be said to represent a conversion ratio between torque and current. Θ _i is a current phase angle with respect to the γ axis in the γδ axis coordinate system.

The torque / current conversion unit 121 may generate a current command I _γ ^ref or I _δ ^ref by preparing a table that can refer to the torque command as a parameter in advance and generating the current command by referring to this table. Is possible. A method using such a table is effective when it is not preferable to formulate the relationship between torque and current as in the above formula.

The current control unit 122 generates the current commands I _γ ^ref and I _δ ^ref generated by the torque / current conversion unit 121, and the γ-axis response value I _γ ^res and the δ-axis response value I _δ ^{res of the} current flowing through the synchronous machine. As inputs, for example, the γ-axis voltage command V _γ ^ref and the δ-axis voltage command V _δ ^ref are calculated and output by proportional-integral control as shown in Equation (3).

Here, K _p is a proportional gain, K _i is an integral gain, and s is a Laplace operator. Thereby, the current control unit 122 performs control so that the stator current follows the input current command value.

The dq / 3-phase conversion unit 123 generates a γ-axis voltage command V _γ generated by the current control unit 122 based on the rotation phase angle estimation value θ _est output from the rotation angle estimation unit 14 that is a rotation phase angle estimation unit. ^{The ref} and δ-axis voltage command V _δ ^ref are coordinate-converted using the equation (4) to generate three-phase voltage commands V _u ^ref , V _v ^ref , and V _w ^ref .

  The dq / 3-phase converter 123 outputs the three-phase voltage command obtained in this way to the PWM processor 13.

The three-phase / dq conversion unit 124 generates the current response used in the vector control, and thus the three-phase detected by the current detection unit 6 based on the rotation phase angle estimated value θ _est output by the rotation angle estimation unit 14. The coordinate transformation of the current response is performed by the equation (5) to generate the γ-axis current response value I _γ ^res and the δ-axis current response value I _δ ^res .

Here, using the fact that the sum of the three-phase currents flowing through the permanent magnet synchronous motor is 0, the three-phase / dq conversion unit 124 can convert two phases out of the three-phase currents, as shown by the equation (6). The γ-axis current response value I _γ ^res and the δ-axis current response value I _δ ^res can be obtained from the current values. In this case, it is only necessary to provide the current detector 6 for two phases, and the apparatus can be simplified as compared with the case of detecting for three phases.

  The rotation angle estimation unit 14 estimates the rotation angle based on the voltage and current that control the synchronous motor 7. Here, although not shown in FIG. 1, the rotation angle estimation unit 14, for example, a voltage command input to the PWM processing unit 13, and a three-phase current value flowing through the synchronous motor 7 detected by the current detection unit 6. Based on this, the rotation angle can be estimated. Moreover, the rotation angle estimation part 14 can also use the value which measured the three-phase terminal voltage actually applied to the synchronous motor 7, for example using measuring means, such as a voltage sensor, as a voltage command value.

  Next, the operation of the present embodiment configured as described above will be described. The rotation angle estimator 14 in the present embodiment is a current that flows through the diode based on at least one of a high-frequency voltage and a high-frequency current generated in the stator coil in accordance with the high-frequency magnetic flux command generated by the high-frequency magnetic flux generation controller 11. An index corresponding to the presence or absence of is calculated, and the rotation angle of the synchronous motor 7 is estimated based on the calculated index.

  Here, the principle of estimation of the rotation angle in the present proposal will be described. Here, the case where a permanent magnet synchronous motor is used is illustrated.

FIG. 6 is a schematic diagram of a synchronous motor. In FIG. 6, it is assumed that the diode is turned on when the current i _rd flowing through the rotor coil flows in the positive direction (the direction of the arrow), and the magnetic flux generated by the rotor coil at this time coincides with the magnetic flux. However, the interlinkage magnetic flux direction of the rotor coil, the current direction, and the forward / reverse direction of the diode can be freely set without departing from the gist of the present proposal.

Here, in the dq-axis coordinate system shown in FIG. 5, the voltage v, current i, and magnetic flux λ are expressed as in equations (7) and (8).

Where R _a is an armature resistance, R _rd is a rotor coil resistance, ω is a rotating electrical angular frequency of the motor, and L _d , L _q , and L _rd are self-inductances of the dq axis and the rotor coil. M _rd is the mutual inductance of the rotor coil and the stator coil. The subscripts d, q, and rd represent variables related to the d axis, the q axis, and the rotor coil, respectively. Λ _PM is a permanent magnet magnetic flux.

By substituting equation (8) into equation (7), equation (9) can be obtained.

In Equation (9), considering that the high-frequency current is superimposed on the dq-axis current, the differential term becomes larger than the other terms. Therefore, if the terms of the resistance component and the ω component can be ignored, Equation (10) As shown in FIG. However, p is a differential operator.

When the diode of the rotor coil is on, the terminal voltage v _{rd of the} coil becomes zero, and the following expressions (11) and (12) are obtained from the third expression of the expression (10).

By substituting equation (12) into the first equation of equation (10), equation (13) shown below is obtained.

On the other hand, considering the case where the diode is off, the following equation (14) is obtained similarly.

The rotation angle estimation unit 14 according to the present embodiment calculates, based on the high frequency voltage, an inductance difference between when the current flowing through the rotor coil 5 is rectified by the diode and when it is not rectified as an index. When the equations (13) and (14) are viewed from the γδ axis, which is the control coordinate in FIG. 5, the diode-on time can be represented as the equation (15), and the diode-off time can be represented by the equation (16). Can be expressed as:

Here, it can be seen from equation (12) that when i _d is increased in the minus direction, i _rd becomes positive and the diode is turned on.

Considering the case of the estimation error Δθ≈90 °, since i _d ≈−i _δ holds, the on / off state of the diode can be determined based on the direction of increase / decrease of i _δ . For example, the diode is turned on when i _δ increases and turned off when i _δ decreases.

Therefore, the high-frequency magnetic flux generation control unit 11 generates a high-frequency magnetic flux command for generating a high-frequency magnetic flux in a direction electrically orthogonal to the interlinkage magnetic flux of the rotor coil 5, and a triangular wave having a frequency at which current control is sufficiently achieved. The current is superimposed on the command value of i _δ . In other words, it can be said that the high-frequency magnetic flux generation control unit 11 superimposes a triangular wave-shaped high-frequency current command corresponding to the generated high-frequency magnetic flux command on the current command value.

If the superimposed voltage of the triangular wave current is detected and added, the current differential term has the opposite sign when it rises and falls, so the sum of (15) and (16) Is a voltage difference as shown in equation (17).

Δv _δ is expressed as (18) by the equation (17).

The equation (18) indicates that any Δθ is a negative value. However, considering the case where the estimation error is Δθ = −90 °, for example, the diode ON / OFF relationship is reversed. ) And (18) are also reversed, and the expression (18) becomes the expression (19).

That is, FIG. 7 is a graph of equations (18) and (19) taking into account the diode on / off relationship. FIG. 7 is a diagram showing a difference in the δ-axis voltage that is an index of estimation error when estimating the rotation angle of the rotor in the synchronous motor drive system of the present embodiment, and the horizontal axis is the estimation error Δθ. In FIG. 7, Δv _δ also becomes zero at a point where the estimation error is 0, so that this can be used as an evaluation function to bring the estimated angle closer to the true angle by using a PLL or the like.

Summarizing the above, the rotation angle estimation unit 14 in the present embodiment flows a high-frequency current i _δ that generates a magnetic flux orthogonal to the interlinkage magnetic flux direction of the rotor coil 5, and a voltage for flowing the current, From the same direction component v _{δ as the} high-frequency current, a difference component caused by on / off of the diode is detected and used as an index to estimate the rotation angle. That is, the rotation angle estimator 14 in the present embodiment uses an index indicating the angle difference between the estimated rotation angle and the actual rotation angle due to the presence or absence of current flowing in the diode based on the high-frequency voltage generated in the stator coil. calculate. Specifically, the rotation angle estimator 14 calculates the index based on the difference in the high-frequency voltage v _δ corresponding to each timing when the triangular wave-shaped high-frequency current command rises and falls.

  As described above, according to the synchronous motor drive system according to the first embodiment of the present invention, in the low-speed sensorless control of the synchronous motor 7 including the rotor coil 5 in which the diode is inserted, there is no need to perform complicated processing. It is possible to perform the estimation of the rotation angle and the magnetic pole discrimination at the same time. In other words, the synchronous motor by the sensorless control method in which the estimation range is not −90 ° ≦ Δθ ≦ + 90 ° but −180 ° ≦ Δθ ≦ + 180 ° A drive system can be provided.

  That is, with the configuration as described above, the synchronous motor drive system described in the present embodiment can estimate the rotation angle of the rotor at a low rotation by a simple calculation without using a rotation angle sensor. In addition to miniaturization, cost reduction, and ease of maintenance, magnetic pole polarity discrimination, which is a restriction on the principle of low-speed sensorless control, can be made unnecessary.

  In this embodiment, the rotor 3 is configured such that the direction of the interlinkage magnetic flux to the rotor coil 5 is the d-axis direction defined when the synchronous motor 7 is vector-controlled, and high-frequency magnetic flux generation is performed. The control unit 11 generates the high frequency magnetic flux command so that the high frequency magnetic flux is generated in the q-axis direction, but another method is also conceivable.

  That is, the rotor 3 is configured such that the direction of the flux linkage to the rotor coil 5 is the q-axis direction defined when the synchronous motor 7 is vector-controlled, and the high-frequency magnetic flux generation control unit 11 is The high frequency magnetic flux command may be generated so that the high frequency magnetic flux is generated in the direction of the d axis. Thereby, since a magnetic flux does not pass in the magnetic flux direction of the rotor coil 5, it can be made hard to receive the influence of a magnetic flux.

  Alternatively, the rotor 3 is configured such that the direction of the interlinkage magnetic flux to the rotor coil 5 is the same as the magnetic flux generated by the torque current component that is normally used by the synchronous motor 7, and the high-frequency magnetic flux generation control unit 11 It is good also as a structure which produces | generates a high frequency magnetic flux command so that a high frequency magnetic flux may generate | occur | produce in the direction orthogonal to the direction of an alternating magnetic flux.

Next, a synchronous motor drive system according to Embodiment 2 of the present invention will be described. In the first embodiment, the rotation angle is estimated by superimposing a high frequency current on i _δ in the direction orthogonal to the interlinkage magnetic flux direction of the rotor coil 5 and detecting v _{δ in the} same direction component as this. The synchronous motor drive system of the embodiment detects a voltage difference of v _δ in a direction orthogonal to the high frequency current by superimposing a high frequency on i _γ which is the same direction as the interlinkage magnetic flux direction. Therefore, the configuration of the synchronous motor drive system of the present embodiment is almost the same as that of the first embodiment, and the superposition destination of the high frequency current command by the high frequency magnetic flux generation control unit 11 is the γ current command. That is, the high frequency magnetic flux generation control unit 11 according to the present embodiment generates a high frequency magnetic flux command that generates a high frequency magnetic flux in the same electrical direction with respect to the interlinkage magnetic flux of the rotor coil 5.

  Other configurations are the same as those of the first embodiment, and redundant description is omitted.

Next, the operation of the present embodiment configured as described above will be described. Since the detected Δv _δ is expressed by the equation (20), the rotation angle estimator 14 in the present embodiment can obtain characteristics as shown in FIG. 8 by using this as an index. The index of FIG. 8 can be converged to a point where the estimation error is zero similarly to FIG.

The method shown in Example 1 and the method shown in Example 2 can be used in combination. Specifically, by superimposing a spatially rotating high-frequency current command on the γδ axis, it is possible to superimpose a high frequency on both i _γ and i _δ , and the phase is shifted by 90 °. It is possible to detect the corresponding voltage without interfering with.

For example, the high frequency magnetic flux generation control unit 11 generates a high frequency magnetic flux command for superimposing a rotational high frequency with an amplitude I and an angular frequency ω _h as shown in the equation (21).

At this time, the high-frequency voltage is expressed by the equation (22), and if a sin (ω _h t) component and cos (ω _h t) are extracted using a process such as Fourier series expansion, the first matrix on the right side It is possible to calculate each element of an inductance matrix.

The diode on / off switches in response to positive and negative high frequency current differential, since the _L'd and L _d in response to this operation are switched, by calculating the inductance matrix for each half cycle of the high frequency current (17) If the voltage difference corresponding to the equation is obtained, the rotation angle estimation unit 14 can obtain the amount resulting from the on / off of the diode as in equation (23).

  If the elements of the inductance matrix can be obtained from equation (23), the rotation angle estimation unit 14 can obtain an index related to the estimation error Δθ.

  In the first embodiment, the high-frequency current is superimposed in the direction orthogonal to the interlinkage magnetic flux of the rotor coil 5, but in the synchronous motor having the configuration of the rotor coil shown in FIG. This could cause torque pulsations. However, in this embodiment, it is possible to reduce torque pulsation by superimposing a high-frequency current in the same direction as the interlinkage magnetic flux of the rotor coil 5.

  As described above, according to the synchronous motor drive system according to the second embodiment of the present invention, the rotation angle of the rotor at a low rotation can be estimated with a simple calculation without using a rotation angle sensor. In addition to miniaturization, cost reduction, and ease of maintenance, it is possible to eliminate magnetic pole polarity discrimination, which is a restriction on the principle of low-speed sensorless control, and to reduce torque pulsation. .

  Next, a synchronous motor drive system according to Embodiment 3 of the present invention will be described. The configuration of the synchronous motor drive system of the present embodiment may be substantially the same as that of the first embodiment, and an aspect in which the proposed estimation method is used in combination with general low-speed sensorless control will be described. That is, as shown in the equations (15) and (16), the non-diagonal component used in the general low speed sensorless control also appears in the synchronous motor including the proposed rotor coil. It is also possible to use the proposed estimation method together while performing sensorless control.

Here, general low-speed sensorless control will be described. The basic equation for estimating the rotation angle is based on the equation (16). When the equation (16) is arranged, it is expressed as the equation (16a).

  The expression (16a) is an expression that focuses on the current differential term. In the case of a general low-speed sensorless, in addition to the voltage / current for controlling the motor (herein referred to as a steady term), the high-frequency voltage or the high-frequency current is By superimposing, the differential term becomes relatively larger than the steady term, and the component of the equation (16a) is easily detected.

For example, when a triangular wave current is superimposed on i _γ as a high-frequency current, pi _γ has a substantially rectangular waveform. At this time, since no high-frequency current is superimposed on i _δ , pi _γ is almost zero, and v _{δ in} equation (16a) is expressed by equation (16b).

In the equation (16b), if attention is paid to the sign of pi _γ , v _δ can be regarded as a function of sin (2Δθ), and Δθ can be obtained as an index that increases or decreases in accordance with Δθ when the zero cross matches.

  Based on this index, by configuring the PLL as shown in FIG. 9 so that the index becomes zero, a feedback loop for rotation angle estimation is configured via the current control system shown in FIG. That is, the actual rotation angle and the estimated rotation angle can be matched.

  The proposed method will be described for the low-speed sensorless control exemplified above. The off-diagonal component of the equation (16a) includes Ld, and when the rotor diode is turned on, it is replaced by Ld ′ as in the equation (15), but there is no influence of the diode while the diode is off. As in Expressions (16) and (16a), a general low-speed sensorless principle is maintained. Therefore, if it is assumed that on / off of the diode can be almost detected in the direction of current increase / decrease, it is possible to implement general low-speed sensorless by calculating the equation (16b) during the diode off period. Furthermore, since the equation (23) can be obtained by taking the difference between ON / OFF of the diode, the proposed estimation method can also be implemented.

Further, as a combination method, not only the rotation angle estimation but also the magnetic pole polarity determination may be used. For example, if v _γ is detected by superimposing a high frequency on i _γ , the component of cos 2Δθ can be obtained in the equation (17). When Δθ = 0 ° and Δθ = 180 °, the diode is turned on / off. It can be obtained as a value with a different sign.

FIG. 10 is a diagram illustrating a γ-axis voltage difference Δv _{γ serving} as an index of the magnetic pole position of the synchronous motor drive system of the present embodiment. Based on this information, when the cos2Δθ component is positive, it can be determined as a normal magnetic pole position, and when it is negative, it can be determined as a reverse magnetic pole position. Therefore, it is possible to always perform magnetic pole polarity determination while performing low-speed sensorless control. It is.

  As described above, according to the synchronous motor drive system according to the third embodiment of the present invention, it is possible to always determine the magnetic pole polarity while performing the conventional low-speed sensorless control. For example, sudden acceleration / deceleration of the synchronous motor, etc. Thus, it is possible to prevent the estimation error from expanding instantaneously and, in the estimation principle, the situation where the magnetic poles are reversed to the position where the estimation error is converged (position where the estimation error is 180 °) is prevented.

  Next, a synchronous motor drive system according to Embodiment 4 of the present invention will be described. In the forms of the first to third embodiments, the high-frequency current is superimposed and the index is detected from the high-frequency voltage. However, in practice, the control band of the current control unit is often not sufficiently fast, for example, When the current control period is 500 μsec, the high frequency that can be selected is 4 to 8 times or more, and the frequency is 1/4 to 1/8 or less. In such a case, since the frequency of the superimposed high-frequency current is lowered, the convergence when an estimation error occurs is delayed, and it can be considered that torque pulsation that cannot be ignored appears.

  Therefore, in this embodiment, it is considered that the high frequency voltage is superimposed and the high frequency current generated thereby is detected to calculate the index. Since it is common to use an AD converter with a sampling frequency that is much higher than the current control frequency to detect the current flowing through the motor, a high-frequency voltage with a frequency equivalent to the current control frequency is superimposed. However, the high-frequency current generated thereby can be detected with high accuracy.

  FIG. 11 is a block diagram showing the configuration of the synchronous motor drive system according to the fourth embodiment of the present invention. The difference from the configuration of the first embodiment shown in FIG. 1 is that the high-frequency magnetic flux generation control unit 11 superimposes the high-frequency voltage command on the γ voltage command, and the rotation angle estimation unit 14 is based on the three-phase / dq conversion unit 124. The rotation angle is estimated based on the output γδ current response. Other configurations are the same as those of the first embodiment, and redundant description is omitted.

Next, the operation of the present embodiment configured as described above will be described. The high frequency magnetic flux generation control unit 11 superimposes a high frequency voltage command corresponding to the generated high frequency magnetic flux command on a voltage command value (γ voltage command) for controlling the stator 4. The response regarding the high frequency component when the high frequency voltage is superimposed on the voltage command can be derived from the equation (15) or (16) as the following equation (24).

However, Formula (24) is a case where the diode of the rotor coil 5 is off, and when it is on, Ld may be replaced with L′ d. Although diode on / off depending on the increase / decrease of i _d, if the estimation error Δθ is sufficiently small, can be said that the diode of the on / off is determined by the substantially i _gamma. Therefore, if Δθ = 0 in equation (24) and a rectangular high frequency voltage is superimposed on v _γ , i _γ will be the response of equation (25), resulting in the waveform shown in FIG.

  However, 1 / p means an integral operation. Therefore, the high frequency magnetic flux generation control unit 11 of the present embodiment superimposes the rectangular wave high frequency voltage command on the voltage command value (γ voltage command) for controlling the stator 4. Since there is actually a resistance component, the triangular wave as shown in FIG. 12 is not obtained. However, the effect of the resistance component is relatively reduced by increasing the frequency of the superimposed high-frequency voltage, and can be regarded as a substantially triangular wave. become able to.

From FIG. 12, it can be seen that i _γ increases on the positive side of the high-frequency voltage and decreases on the negative side, and the diodes of the rotor coil are turned off and on, respectively. Therefore, when the current changes on the positive and negative sides of the rectangular wave are calculated and added, equation (26) is derived from equation (24).

However, Δt is a time corresponding to a half period of the rectangular wave. When Δi _δ is detected in the equation (26), the same characteristics as in FIG. 8 are obtained, so that the rotation angle estimation unit 14 can estimate the rotation angle by using this as an index. That is, the rotation angle estimator 14 in the present embodiment uses an index indicating an angle difference between the estimated rotation angle and the actual rotation angle due to the presence or absence of the current flowing through the diode based on the high-frequency current generated in the stator coil. Based on the high-frequency current, the difference between the inductance when the current flowing through the rotor coil 5 is rectified by the diode and when the current is not rectified can be calculated as an index.

  Specifically, the rotation angle estimation unit 14 calculates the index based on the difference in the slope of the high-frequency current corresponding to the positive and negative timings of the rectangular wave-shaped high-frequency voltage command.

  As described above, according to the synchronous motor drive system according to the fourth embodiment of the present invention, the high-frequency current obtained by using the high-frequency voltage command even when the current control response cannot be sufficiently high compared to the superimposed frequency. The index can be calculated from the above, and the rotation angle can be estimated stably. Furthermore, since the frequency to be superimposed can be made higher than when the high frequency command is superimposed on the current, the frequency of the torque pulsation generated thereby is also increased, and the influence of disturbance on a constant torque command can be reduced. Is possible.

  Although several embodiments of the present invention have been described, these embodiments are presented by way of example and are not intended to limit the scope of the invention. These novel embodiments can be implemented in various other forms, and various omissions, replacements, and changes can be made without departing from the scope of the invention. These embodiments and modifications thereof are included in the scope and gist of the invention, and are included in the invention described in the claims and the equivalents thereof.

DESCRIPTION OF SYMBOLS 1 Control apparatus 2 Inverter 3 Rotor 4 Stator 5 Rotor coil 6 Current detection part 7 Synchronous motor 11 High frequency magnetic flux generation control part 12a, 12b Motor control part 13 PWM process part 14 Rotation angle estimation part 121 Torque / current conversion part 122 Current controller 123 dq / 3-phase converter 124 3-phase / dq converter

Claims (12)

A synchronous motor,
A control device for controlling the synchronous motor,
The synchronous motor is
A stator having a stator coil;
A rotor having a rotor coil in which a diode is inserted;
The controller is
A high-frequency magnetic flux generation control unit that generates a high-frequency magnetic flux command for generating high-frequency magnetic flux in the stator;
The high- frequency voltage corresponding to the presence or absence of a current flowing through the diode based on at least one of a high-frequency voltage and a high-frequency current generated in the stator coil in response to the high-frequency magnetic flux command generated by the high-frequency magnetic flux generation control unit And a rotation angle estimation unit for estimating a rotation angle of the synchronous motor based on the calculated difference in the high frequency voltage or the difference in the high frequency current. Synchronous motor drive system.   2. The synchronous motor drive according to claim 1, wherein the high-frequency magnetic flux generation control unit generates a high-frequency magnetic flux command for generating a high-frequency magnetic flux in a direction electrically orthogonal to the interlinkage magnetic flux of the rotor coil. system.   2. The synchronous motor drive system according to claim 1, wherein the high-frequency magnetic flux generation control unit generates a high-frequency magnetic flux command for generating a high-frequency magnetic flux in the same electrical direction with respect to the interlinkage magnetic flux of the rotor coil. . The rotation angle estimation unit calculates an index indicating an angle difference between an estimated rotation angle and an actual rotation angle caused by the presence or absence of a current flowing through the diode based on the difference in the high-frequency voltage or the difference in the high-frequency current. The synchronous motor drive system according to any one of claims 1 to 3, wherein the synchronous motor drive system is provided. The rotational angle estimation unit, on the basis of the difference between the difference or the high frequency current of the high-frequency voltage, the inductance difference of the indices of the case where the current flowing through the rotor coil is not rectified and if it is rectified by the diode The synchronous motor drive system according to claim 1, wherein the synchronous motor drive system is calculated as follows. The control device has a current control unit that controls the stator current to follow the input current command value,
The synchronous motor drive according to any one of claims 1 to 5, wherein the high-frequency magnetic flux generation control unit superimposes a high-frequency current command corresponding to the generated high-frequency magnetic flux command on the current command value. system. The high frequency magnetic flux generation control unit superimposes a triangular wave high frequency current command on the current command value,
The said rotation angle estimation part calculates the said parameter | index based on the difference of the said high frequency voltage corresponding to each timing at the time of the rise of the high frequency current command of the triangular wave shape, and a fall. Synchronous motor drive system.   6. The high frequency magnetic flux generation control unit superimposes a high frequency voltage command corresponding to the generated high frequency magnetic flux command on a voltage command value for controlling a stator. The synchronous motor drive system according to the item. The high-frequency magnetic flux generation control unit superimposes a rectangular wave-shaped high-frequency voltage command on the voltage command value,
The said rotation angle estimation part calculates the said parameter | index based on the difference of the inclination of the said high frequency current corresponding to each timing of the positive and negative of the said high frequency voltage instruction | command of rectangular wave shape. Synchronous motor drive system. The rotor is configured such that the direction of the flux linkage to the rotor coil is the d-axis direction defined when the synchronous motor is vector-controlled,
The synchronous motor drive system according to any one of claims 1 to 9, wherein the high-frequency magnetic flux generation control unit generates a high-frequency magnetic flux command so that a high-frequency magnetic flux is generated in a q-axis direction. The rotor is configured such that the direction of the flux linkage to the rotor coil is a q-axis direction defined when the synchronous motor is vector-controlled,
The synchronous motor drive system according to any one of claims 1 to 9, wherein the high-frequency magnetic flux generation control unit generates a high-frequency magnetic flux command so that a high-frequency magnetic flux is generated in a d-axis direction. The rotor is configured such that the direction of the interlinkage magnetic flux to the rotor coil is the same as the magnetic flux due to the torque current component commonly used by the synchronous motor,
10. The high frequency magnetic flux generation control unit generates a high frequency magnetic flux command so that a high frequency magnetic flux is generated in a direction orthogonal to the direction of the interlinkage magnetic flux. Synchronous motor drive system.

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