JP4945307B2 - Twin drive control device and torsional vibration suppression method - Google Patents

Description

  The present invention relates to a system for driving a load having a torsional vibration system such as a rotary shear or a flying shear from both sides by a motor (hereinafter referred to as twin drive), and in particular, a twin drive system for driving a load by a synchronous motor. The present invention relates to a twin drive control device and a torsional vibration suppression method.

  Conventionally, a twin drive control device that drives a load having a torsional vibration system from both sides is known. Hereinafter, four forms are shown as a conventional twin drive control apparatus, and each will be described.

  First, a conventional first twin drive control device will be described. As a conventional twin drive control device, one shown in FIG. 12 is known. In this twin drive control device, for example, as shown in FIG. 11, the rotary cutter 2 is vector-controlled for the motors 6A and 6B via the speed reducers 4A and 4B. A typical example of the load device to which such a twin drive control device is applied is the twin drive system of the rotary cutter 2. Here, the motors 6A and 6B are synchronous motors suitable for low moment of inertia, power saving, and miniaturization.

  In general, the load device is driven from one side like a rotary cutter or a high-speed crank shear that cuts a sheet of paper, cardboard, steel plate or the like that is continuously fed to a certain size. However, in recent years, as the sheet width of paper, corrugated board, steel plate, etc. becomes wider, the dimensions of the rotary cutter, crank shear, etc. have increased, and a large capacity motor capable of rapid acceleration / deceleration has been required. Here, a large-capacity motor is large and expensive, and the entire machine becomes large, which is disadvantageous. Therefore, as shown in FIG. 11, a twin drive system in which two conventional motors are used for driving is employed.

  Referring to FIG. 11, the rotary cutter 2 has a twin drive system in which the rotary cutter 2 is connected to a first motor (corresponding to a first synchronous motor) via speed reducers 4 </ b> A and 4 </ b> B constituted by gears from both sides. .) Driven by 6A and a second motor (corresponding to a second synchronous motor) 6B. On the other hand, the rotary cutter 2 is partially hollow inside the rotary cutter 2 in order to reduce the weight of the rotary cutter 2. Further, the circumferential surface of the rotary cutter 2 has a structure in which torsional vibration is likely to occur because the cutter (blade) is processed parallel or obliquely to the central axis of the rotary cutter 2. For this reason, in the twin drive system in which both sides of the rotary cutter 2 are driven, the motors are controlled so that both motors 6A and 6B are driven with the same torque command. Similarly, in the case of a high-speed crank shear (not shown), both ends of the shear (blade) are driven by the crank, so that it is easy to generate torsional vibration. For this reason, the same control as in the case of the rotary cutter 2 is performed.

  Referring to FIG. 12, in the twin drive control device, a position command (hereinafter simply referred to as a speed command) VC is generated by a speed command generation circuit 10 according to a sheet cutting length L0, a sheet traveling speed from a pulse generator 12, and the like. And a speed obtained by inputting a speed feedback pulse signal of the first motor 6A from the pulse generator 14. On the other hand, a speed feedback pulse signal from the pulse generator 14 is converted into a speed feedback signal VA by an FVC (frequency / voltage conversion circuit) 16.

  A speed deviation (VC-VA) is calculated by the subtracter 18 in the speed loop, and a torque command τ * A is created by the speed amplifier 20 and the clamp circuit 22 based on the speed deviation (VC-VA). The This torque command τ * A is supplied to the vector control circuit 24A of the first motor 6A and the vector control circuit 24B of the second motor 6B, respectively. Each vector control circuit 24A, 24B supplies a current command to power units 26A, 26B composed of transistors. At this time, the output currents of the power units 26A and 26B are fed back to the vector control circuits 24A and 24B through the current loop, and the current command is set so that the deviation between the feedback value and the current command value becomes zero. Is created.

  As described above, the conventional first twin drive control device creates a single torque command using the speed deviation of the first motor 6A, and sends the single torque command to the vector control circuits 24A and 24B. By supplying each, the 1st motor 6A and the 2nd motor 6B are each driven.

  Next, a conventional second twin drive control device will be described. As a conventional twin drive control device, the one described in Patent Document 1 is known. As shown in FIG. 13, the twin drive control device performs vector control of the first motor 6A and the second motor 6B. By this vector control, the torsional vibration of the load is suppressed with respect to the system in which the load having the torsional vibration system is driven by the first and second motors 6A and 6B from both sides.

  Specifically, the twin drive control device generates a basic torque command τ * A to the first motor 6A by the speed command generation circuit 10, the FVC 16A, the subtractor 18, the speed amplifier 20, and the clamp circuit 22, The subtractor 32 of the antiphase compensation circuit 30 calculates the difference between the speed feedback of the first motor 6A and the speed feedback of the second motor 6B, and this speed difference is compensated by the speed amplifier 34 and the differentiation circuit 36. The command is converted to a command, and the adder 40 adds a compensation torque command to the basic torque command τ * A to create a torque command τ * B1 to the second motor 6B.

  In other words, this twin drive control device calculates the difference between the means for creating the basic torque command τ * A for the first motor 6A and the speed feedback of the first motor 6A and the speed feedback of the second motor 6B. And a means for converting the speed difference into a compensation torque command and a means for adding the compensation torque command to the basic torque command τ * A to create a torque command τ * B1 for the second motor 6B. Is.

  As described above, the conventional second twin drive control device creates the basic torque command using the speed deviation of the first motor 6A, and the difference between the speed feedbacks of the first motor 6A and the second motor 6B. Is used to create a compensation torque command. The basic torque command is a torque command to the first motor 6A, and a command obtained by adding the basic torque command and the compensation torque command is a torque command to the second motor 6B, and is supplied to the vector control circuits 24A and 24B, respectively. By doing so, the first motor 6A and the second motor 6B are respectively driven. In this case, the torque command to the second motor 6B is based on the difference in speed feedback between the two motors 6A and 6B, and is controlled so that this speed difference becomes zero. And torsional vibration of the load can be suppressed.

  Next, a conventional third twin drive control device will be described. As a conventional twin drive control device, the one described in Patent Document 2 is known. This twin drive control device drives a load with extremely low rigidity by vector control of the first motor 6A and the second motor 6B, and suppresses torsional vibration of the load.

  Specifically, as shown in FIG. 14, the twin drive control device uses a subtractor 105 and a speed regulator (ASR) 104 to send a basic speed command to the first motor 6A and the speed of the first motor 6A. A torque command τ * A is created from the difference from the feedback signal VF1 (corresponding to the actual speed ω) and is used as a torque command to the first motor 6A. The difference between the rotation angle from the position detector (AE) 101A and the rotation angle from the absolute position detector 101B of the second motor 6B is calculated, the differentiator 108 of the phase adjuster (APR) 106, and the phase error amplifier 109. The rotation angle difference is converted into a compensation torque command by the adder 110 and the speed error amplifier 111, the compensation torque command is added to the torque command τ * A by the adder 112, and the second motor Characterized by creating a torque command tau * B2 to B.

  As described above, the conventional third twin drive control device creates the basic torque command using the speed deviation of the first motor 6A, and the difference in rotation angle between the first motor 6A and the second motor 6B. Is used to create a compensation torque command. The basic torque command is a torque command to the first motor 6A, and a command obtained by adding the basic torque command and the compensation torque command is a torque command to the second motor 6B, which is supplied to the vector control inverters 103A and 103B, respectively. By doing so, the first motor 6A and the second motor 6B are respectively driven. In this case, the torque command to the second motor 6B is based on the difference between the rotation angles of the two motors 6A and 6B, and is controlled so that the phase difference between the rotation angles becomes zero. It works in the direction of suppression, and the torsional vibration of the load can be suppressed.

  Next, a conventional fourth twin drive control device will be described. As a conventional twin drive control device, the one shown in FIG. 15 is known. The twin drive control device 52 suppresses torsional vibration of the load by vector control in consideration of the initial magnetic phase of the first motor 6A and the second motor 6B. Compared with the above-described conventional first to third twin drive control devices, although common in that vector control is used for power control of the first motor 6A and the second motor 6B, 4 is different in that power control is performed in consideration of the initial magnetic pole phases of the first motor 6A and the second motor 6B.

  Hereinafter, in the twin drive control device 52 shown in FIG. 15, the case where the first motor 6A is driven and the case where the second motor 6B is driven will be described separately. First, when driving the first motor 6A, in order to realize generally known vector control, the coordinate converter 55 includes a torque divided voltage command V * q, an excitation divided voltage command V * d, A new initial phase obtained by subtracting the first initial magnetic pole phase (hereinafter referred to as the first magnetic pole phase) θ1 which is the initial magnetic pole phase of the first motor 6A from the rotational phase γ of the first motor 6A by the subtractor 59. Is input, and the input torque divided voltage command V * q and excitation divided voltage command V * d are coordinate-converted according to the new rotation phase Φ1, and the resulting three-phase sine wave voltage command V * is obtained. u1, V * v1, and V * w1 are output to the PWM controller 56.

  Here, the rotational phase γ of the first motor 6A is obtained by differentiating the pulse signal from the pulse generator 14 mechanically connected to the rotation shaft of the first motor 6A by a differentiator (not shown). This value is obtained by integrating the feedback signal (actual speed) ω by the integrator 58. The first magnetic pole phase θ1 is the magnetic pole phase of the first motor 6A, and as shown in FIG. 4, the first motor 6A is DC-excited and the rotor of the first motor 6A (not shown). Is the magnetic pole phase when is held in the locked state.

  Further, as shown in FIG. 15 and FIG. 17B (described later), the PWM controller 56 has three-phase sine wave voltage commands V * u1, V * v1, and V * w1 (in FIG. 17B, a sine wave). Voltage command) and a carrier wave signal (not shown in FIG. 15 and a carrier wave (triangular wave signal from FIG. 17B) in FIG. 17B), and a three-phase sine wave voltage command V * u1, V *. The pulse width of v1, V * w1 is controlled by a carrier wave signal, and three-phase PWM signals (pulse signals) Vu1p, Vv1p, Vw1p are converted into drive signals by a gate drive circuit (not shown) and output to the power converter 57. . Here, the gate drive circuit (not shown) receives three-phase PWM signals Vu1p, Vv1p, and Vw1p from the PWM controller 56, generates a drive signal based on these signals, and outputs the drive signal to the power converter 57. To do. In FIG. 15, for the convenience of explanation, the output of the PWM controller 56 is a three-phase PWM signal Vu1p, Vv1p, Vw1p.

  As shown in FIG. 17A, which will be described later, the power converter 57 inputs a DC voltage Ed as a main circuit, and turns on and off the gates of the drive signals of IGBTs (six) to form a three-phase pulse shape. Sinusoidal voltage signals U1, V1, W1 are output. That is, when a drive signal (three-phase PWM signals Vu1p, Vv1p, Vw1p in FIG. 15) is input to the power converter 57, the power converter 57 causes the drive signal to be a switching element in the power converter 57, such as an IGBT ( The gate of the Insulated Gate Bipolar Transistor) is turned on and off, and three-phase pulsed sine wave voltage signals U1, V1, W1 for driving the first motor 6A are supplied to the windings of the first motor 6A.

  Next, when driving the second motor 6B, as in the case of driving the first motor 6A, the coordinate converter 61 generates a torque voltage command V * q, A second initial magnetic pole phase (hereinafter referred to as a second magnetic pole phase) θ2, which is an initial magnetic pole phase of the second motor 6B, is subtracted from the divided voltage command V * d and the rotational phase γ of the first motor 6A. A new rotation phase Φ2 obtained by subtraction by 60 is input, and the input torque component voltage command V * q and excitation component voltage command V * d are coordinate-converted according to the new rotation phase Φ2 to obtain 3 Phase sine wave voltage commands V * u2, V * v2, and V * w2 are output to the PWM controller 62.

  The PWM controller 62 inputs a three-phase sine wave voltage command V * u2, V * 2, V * w2 and a carrier wave signal (not shown in FIG. 15), and inputs a three-phase sine wave voltage command V *. The pulse widths of u2, V * v2, and V * w2 are controlled by a carrier wave signal, and three-phase PWM signals Vu2p, Vv2p, and Vw2p are converted into drive signals by a gate drive circuit (not shown) and output to the power converter 63. . Here, a gate drive circuit (not shown) receives three-phase PWM signals Vu2p, Vv2p, and Vw2p from the PWM controller 62, generates a drive signal based on these signals, and outputs the drive signal to the power converter 63. To do. In FIG. 15, for the convenience of explanation, the output of the PWM controller 62 is a three-phase PWM signal Vu2p, Vv2p, Vw2p.

  As in the case of the power converter 57 in FIG. 17A, the power converter 63 receives the DC voltage Ed as the main circuit, and turns on and off the gates of the IGBTs (six) of the drive signals to 3 Phase sine wave voltage signals U2, V2 and W2 are output. That is, when a drive signal (three-phase PWM signals Vu2p, Vv2p, Vw2p in FIG. 15) is input to the power converter 63, the power converter 63 causes the drive signal to be a switching element that is inherent in the power converter 63, for example, an IGBT. The three-phase pulse sine wave voltage signals U2, V2, and W2 for driving the second motor 6B by turning on and off the gate are supplied to the windings of the second motor 6B.

  The twin drive control device 52 of FIG. 15 has been described separately for the case where the first motor 6A is driven and the case where the second motor 6B is driven. In either case, the coordinate converter 55, The torque divided voltage command V * q and the excitation divided voltage command V * d which are input signals 61 are generated by the speed and current control unit 54. FIG. 16 is a control block diagram of the speed and current control unit 54 shown in FIG. As shown in FIG. 16, the torque divided voltage command V * q is coordinate-transformed by the torque command τ * (equivalent to the torque current command i * 1) controlled by the speed controller 81 and the coordinate converter 85. This is a command that is current-controlled by a current controller 87 with respect to a torque component current deviation Δiq, which is a difference from the torque component current feedback signal iq. Further, the excitation voltage command V * d includes an excitation current feedback signal id coordinate-converted by the coordinate converter 85, and a zero current command 0 * necessary for zero current control on the d axis in the current controller 90. The excitation current deviation Δid, which is the difference between the two, is a command that is current-controlled by the current controller 90 (zero current control on the d-axis). Here, the torque command τ * is a command in which the speed controller 81 speed-controls the speed deviation Δω of the first motor 6A (the difference between the speed command ω * and the speed feedback signal ω). Further, the torque component current feedback signal iq and the excitation component current feedback signal id are converted from the U-phase current feedback signal iu and the W-phase current feedback signal iw of the first motor 6A in the coordinate converter 85 by the rotation of the first motor 6A. It is a signal that has been coordinate-transformed by the phase γ.

  As described above, the conventional fourth twin drive control device obtains a new rotation phase Φ1 by using the first magnetic pole phase θ1 that is the initial magnetic pole phase of the first motor 6A, and this new rotation phase Φ1. Is used to convert the torque divided voltage command V * q and the excitation divided voltage command V * d into three-phase sine wave voltage commands V * u1, V * v1, V * w1, and convert the PWM signals Vu1p, Vv1p, Vw1p to Then, the first motor 6A is driven via the power converter 57. Similarly, a new rotation phase Φ2 is obtained using the second magnetic pole phase θ2 that is the initial magnetic pole phase of the second motor 6B, and the torque component voltage command V * q and the excitation component are calculated using the new rotation phase Φ2. The voltage command V * d is coordinate-transformed into three-phase sine wave voltage commands V * u2, V * v2, V * w2, and PWM signals Vu2p, Vv2p, Vw2p are generated, and the second voltage is converted via the power converter 63. The motor 6B is driven. In this case, since the first motor 6A and the second motor 6B are driven in consideration of the first initial phase θ1 and the second initial phase θ2, respectively, it works in a direction to suppress torsional vibrations, Torsional vibration of the load can be suppressed.

JP-A-8-174481 JP 2003-116292 A

  As described above, the conventional first twin drive control device creates a single torque command using the speed deviation of the first motor 6A, and uses the single torque command as the vector control circuits 24A, 24B. Are respectively supplied to drive the first motor 6A and the second motor 6B.

  Further, the conventional second twin drive control device creates a basic torque command using the speed deviation of the first motor 6A, and uses the difference in speed feedback between the first motor 6A and the second motor 6B. To create a compensation torque command. The basic torque command is a torque command to the first motor 6A, and a command obtained by adding the basic torque command and the compensation torque command is a torque command to the second motor 6B (the torque to the second motor 6B). The command is based on the difference in speed feedback between the motors 6A and 6B, and is controlled so that the speed difference becomes zero.) By supplying the command to the vector control circuits 24A and 24B, respectively, The motor 6A and the second motor 6B are each driven.

  Further, the conventional third twin drive control device creates a basic torque command using the speed deviation of the first motor 6A, and uses the difference in rotation angle between the first motor 6A and the second motor 6B. To create a compensation torque command. The basic torque command is a torque command to the first motor 6A, and a command obtained by adding the basic torque command and the compensation torque command is a torque command to the second motor 6B (a torque command to the second motor 6B). Is based on the difference between the rotation angles of the motors 6A and 6B, and is controlled so that the phase difference between the rotation angles becomes zero.) By supplying the vector control inverters 103A and 103B, respectively, The first motor 6A and the second motor 6B are each driven.

  Further, the conventional fourth twin drive control device includes coordinate converters 55 and 61, PWM controllers 56 and 62, and power converters 57, 62 for controlling the first motor 6 </ b> A and the second motor 6 </ b> B. 63, each of which includes a first initial phase θ1 that is an initial magnetic pole phase of the first motor 6A and a second initial phase θ2 that is an initial magnetic pole phase of the second motor 6B. 6A and the second motor 6B are each driven. Here, even when the first initial phase θ1 and the second initial phase θ2 are taken into consideration, if the initial phases θ1 and θ2 are different, the magnetic pole phase of the first motor 6A and the magnetic pole of the second motor 6B are used. There will be a difference in phase. This twin drive control device performs vector control of the first motor 6A and the second motor 6B as described above even if there is a magnetic pole phase between the motors, and PWM signals Vu1p, Vv1p, Vw1p and Power control is performed based on Vu2p, Vv2p, and Vw2p.

  However, in the conventional first to fourth twin drive control devices, the initial phases θ1 and θ2 are different, and there is a difference between the magnetic phase of the first motor 6A and the magnetic phase of the second motor 6B. Even in this case, since the first motor 6A and the second motor 6B are driven by vector control, torsional vibration of the load based on the difference in torque generated between the motors 6A and 6B occurs. In order to suppress this torsional vibration, the initial magnetic pole phases θ1 and θ2 of both motors 6A and 6B must be matched. That is, there is a problem that mechanical adjustment work for matching the initial magnetic pole phases θ1 and θ2 of both the motors 6A and 6B must be performed, and the adjustment work is difficult.

  Hereinafter, the reason why torsional vibration occurs when there is a magnetic pole phase difference angle δ (corresponding to (θ2−θ1) / 2) between the first magnetic pole phase θ1 and the second magnetic pole phase θ2. This will be described with reference to FIGS.

  FIG. 17 is a diagram illustrating an example of input / output signals in the PWM controller and the power converter. FIG. 17A shows an example of input / output signals in the power converter 57. A bus voltage (DC voltage) Ed is connected to the input portion of the main circuit of the power converter 57, and a three-phase pulse shape is output as the output portion. Sinusoidal voltage signals U1, V1, W1 are shown. FIG. 17B shows an example of input / output signals in the PWM controller 56. In FIG. 17B, a PWM controller 56 inputs a sine wave voltage command and a carrier wave signal from a triangular wave generator, compares and calculates the level of the sine wave voltage command and the level of the carrier wave signal, and outputs a PWM signal. To do. Based on this PWM signal, the drive signal is output to the gates of the IGBTs (six) in the power converter 57 via a gate drive circuit (not shown).

  FIG. 18 is a diagram illustrating a high-frequency excitation voltage based on a sinusoidal voltage command that is not in the same phase in the conventional fourth twin drive control device. FIG. 18A is a diagram showing respective sine wave voltage commands V * u1 (M1) and V * u2 (M2) input to the PWM controllers 56 and 62 for the U phase, for example, and the horizontal axis ωt is The phase angle and the vertical axis indicate the signal level of the sine wave voltage command, respectively. Here, the sine wave voltage commands M1, M2 are the sine wave voltage commands V * u1, V * to the first motor 6A and the second motor 6B having the first magnetic pole phase θ1 and the second magnetic pole phase θ2. Each of u2 is shown, and the phase angle θn is a phase angle at an arbitrary time tn.

  FIG. 18B shows an enlarged comparison of the signal levels of the sine wave voltage command M1, the sine wave voltage command M2 and the carrier wave signal at the phase angle θn, where the horizontal axis t is the elapsed time and the vertical axis is the signal. Each level is shown.

  FIG. 18C shows the PWM signal Vu2p generated by the PWM controller 62 based on the sine wave voltage command M2, the horizontal axis t shows the elapsed time, and the vertical axis shows the signal level of the PWM signal Vu2P. According to FIG. 18C, the PWM controller 62 compares the signal level of the sine wave voltage command M2 with the signal level of the carrier wave signal, and the signal level of the sine wave voltage command M2 is equal to the signal level of the carrier wave signal. When the signal level of the PWM signal Vu2p is set to the DC voltage value + Ed / 2 when the signal level of the sine wave voltage command M2 is lower than the level of the carrier wave signal, the signal level of the PWM signal Vu2p is set to the DC voltage. It can be seen that the value is set to -Ed / 2.

  FIG. 18D shows the PWM signal Vu1p generated by the PWM controller 56 based on the sine wave voltage command M1, the horizontal axis t is the elapsed time, and the vertical axis is the signal level of the PWM signal Vu1P. According to FIG. 18 (D), the PWM controller 56 compares the signal level of the sine wave voltage command M1 with the signal level of the carrier wave signal in the same manner as the PWM controller 62 described above, and the sine wave voltage command M1. When the signal level is equal to or greater than the signal level of the carrier wave signal, the signal level of the PWM signal Vu1p is set to the DC voltage value + Ed / 2, and the signal level of the sine wave voltage command M1 is smaller than the signal level of the carrier wave signal Further, it can be seen that the level of the PWM signal Vu1p is set to the DC voltage value -Ed / 2.

  FIG. 18E shows a difference signal (high-frequency excitation voltage) obtained by subtracting the signal level of the PWM signal Vu1p based on the sine wave voltage command M1 from the signal level of the PWM signal Vu2p based on the sine wave voltage command M2. The horizontal axis t represents the elapsed time, and the vertical axis represents the signal level of the difference signal.

  18A to 18E, when a phase difference (θ1−θ2) exists between the sine wave voltage command M1 and the sine wave voltage command M2, the PWM signal Vu2p and the PWM control by the PWM controller 62 are obtained. The difference signal which is the level difference from the PWM signal Vu1p by the device 56 does not become zero. That is, the fourth twin drive control device 52 performs the initial operation of the second motor 6B in order to output the electric power for driving the first motor 6A and the second motor 6B by turning on and off the gate of the IGBT. If there is a magnetic pole phase difference angle δ = (θ2−θ1) / 2 between the magnetic pole phase θ2 and the initial magnetic pole phase θ1 of the first motor 6A, the width of each PWM signal which is a pulse wave-modulated rectangular wave is Since they do not match, the difference in generated power for driving the first motor 6A and the second motor 6B (pulse sine wave voltage signals U1, V1, W1 for the first motor 6A and pulse sine for the second motor 6B) A power difference between the wave voltage signals U2, V2, and W2. Therefore, the twin drive control device 52 causes a difference in generated torque for driving the first motor 6A and the second motor 6B, resulting in a problem that the torsional vibration of the load cannot be suppressed.

  The present invention has been made to solve the above-described problems. When the load is driven by two synchronous motors from both sides, the initial magnetic pole phases of the two synchronous motors are different. Another object of the present invention is to provide a twin drive control device capable of suppressing torsional vibration of a load and a method for suppressing torsional vibration.

  In order to achieve the above object, a twin drive control device according to the present invention provides a load that is rotatable in a system in which a rotatable load is driven by a first synchronous motor and a second synchronous motor, respectively, from both sides of the rotating shaft. A twin drive control device that suppresses torsional vibrations, the rotational phase correcting means for correcting the rotational phase of the first synchronous motor based on the initial magnetic phase of each of the first and second synchronous motors; The speed deviation between the speed command for the synchronous motor and the actual speed of the first synchronous motor is controlled, and the torque current command obtained by the speed control and the actual current flowing through the winding of the first synchronous motor The current deviation between the two phases is subjected to current control, and the two-phase DC voltage command obtained by the current control is coordinate-converted according to the rotational phase corrected by the rotational phase correcting means, and the coordinate conversion is performed. Based on a single three-phase PWM signal common to the first and second synchronous motors obtained by controlling the pulse width of the obtained three-phase AC voltage command with a predetermined carrier wave signal and by the pulse width control, And control means for controlling power supplied to the first and second synchronous motors.

  Further, in the twin drive control device according to the present invention, the control means speed-controls a speed deviation between the speed command for the first synchronous motor and the actual speed of the first synchronous motor, and obtains the speed control. Current and current difference between the torque current command to be generated and the actual current flowing through the winding of the first synchronous motor, and a speed and current for generating a torque divided voltage command and an excitation divided voltage command as a two-phase DC voltage command A torque divided voltage command and an excitation divided voltage command as a two-phase DC voltage command generated by the control means and the speed and current control means are coordinate-converted according to the rotation phase corrected by the rotation phase correction means. Coordinate conversion means for generating a three-phase AC voltage command, and pulse width control of the three-phase AC voltage command generated by the coordinate conversion means with a predetermined carrier wave signal, and the first and second synchronization PWM control means for generating a single three-phase PWM signal common to the motor, power conversion based on the single three-phase PWM signal generated by the PWM control means, and first and second synchronous motors Power conversion means for generating a pulsed AC voltage signal to be supplied to the power supply.

  In the twin drive control device according to the present invention, the rotational phase correction means provided in the control means adds the initial magnetic pole phase of the first synchronous motor and the initial magnetic pole phase of the second synchronous motor, and adds the addition. The result is multiplied by a half, and the rotation phase of the first synchronous motor is corrected by subtracting the rotation phase of the multiplication result from the rotation phase of the first synchronous motor.

  In the twin drive control device according to the present invention, the speed and current control means of the control means controls the speed deviation between the speed command for the first synchronous motor and the actual speed of the first synchronous motor. The torque command obtained by the speed control is subjected to field control in accordance with the actual speed of the first synchronous motor, and the current between the torque current command obtained by the field control and the actual current flowing through the winding of the first synchronous motor Dividing the deviation and the magnetic flux by the first synchronous motor by its inductance, and adding the field control current command obtained by field control of the division result according to the actual speed of the first synchronous motor, current control, A torque voltage command and an excitation voltage command are generated as a two-phase DC voltage command.

  A torsional vibration suppression method according to the present invention is a method for suppressing torsional vibration of a load in a system in which a rotatable load is driven by a first synchronous motor and a second synchronous motor from both sides of the rotation shaft. Rotation phase correction step for correcting the rotation phase of the first synchronous motor based on the initial magnetic pole phases of the first and second synchronous motors, and a speed detection step for detecting the actual speed of the first synchronous motor A speed control step for speed-controlling a speed deviation between the speed command for the first synchronous motor and the actual speed, a torque current command obtained by the speed control, and an actual current flowing through the first synchronous motor A current control step for controlling the current deviation between them, and a coordinate conversion for converting the two-phase DC voltage command obtained by the current control according to the corrected rotation phase Common to the first and second synchronous motors obtained by the pulse width control and the PWM control step for controlling the pulse width of the three-phase AC voltage command obtained by the coordinate conversion by a predetermined carrier wave signal. And a power control step for controlling power supplied to the first and second synchronous motors by a single three-phase PWM signal.

  As described above, according to the twin drive control device and the torsional vibration suppression method of the present invention, the single magnetic field corrected by using the initial magnetic pole phase of the first synchronous motor and the initial magnetic pole phase of the second synchronous motor. A single AC voltage command is generated based on the rotational phase of the first PWM signal, a single PWM signal is generated, and electric power supplied to the first synchronous motor and the second synchronous motor is generated based on the single PWM signal. Each was controlled. That is, even if the initial magnetic pole phase is different between the first synchronous motor and the second synchronous motor, a single PWM signal is generated from a single rotational phase, and a common single PWM signal is generated. Since the first synchronous motor and the second synchronous motor are subjected to power control based on the above, the torque generated by both synchronous motors becomes the same, and the torsional vibration of the load can be suppressed.

Hereinafter, the best mode for carrying out the present invention will be described in detail with reference to the drawings.
〔Constitution〕
First, the configuration of the twin drive control device according to the present invention will be described. FIG. 1 is a control block diagram when the twin drive control device according to the present invention is applied to a rotary cutter device. FIG. 2 is a control block diagram of the speed and current control unit in the twin drive control device shown in FIG. FIG. 3 is a continuation of the control block diagram of FIG.

  As shown in FIG. 1, the rotary cutter device 71A includes a first synchronous motor 6A, a second synchronous motor 6B, a pulse generator 14, a rotary cutter 2, and a twin drive control device 72A. Since the first synchronous motor 6A, the second synchronous motor 6B, and the rotary cutter 2 have been described above, description thereof will be omitted. In general, the rotary cutter device 71A generates a single PWM signal Vup, Vvp, Vwp by the twin drive control device 72A, and a power converter is generated by the common single PWM signal Vup, Vvp, Vwp. The gates of the switching elements 57 and 76 are turned on and off to supply desired pulsed sine wave voltage signals U1, V1, W1 and U2, V2, and W2 to the first synchronous motor 6A and the second synchronous motor 6B, respectively. Then, the rotary cutter 2 is driven to rotate.

  The pulse generator 14 is a rotation sensor that detects the rotation angle of the first synchronous motor 6A, and an absolute type or an increment type is used. However, in the case of the increment type, when the first synchronous motor 6A makes one rotation, a one-pulse signal is generated.

  As shown in FIG. 1, the twin drive control device 72A includes current detectors 64 and 65, a subtractor 53, a speed and current control unit 73, an integrator 58, a rotational phase correction means 74, a coordinate converter 75, and PWM control. 56 and power converters 57 and 76. Since the integrator 58 and the power converter 57 have been described above, a description thereof will be omitted. The twin drive control device 72A is roughly described. The speed command ω * for the first synchronous motor 6A and the second synchronous motor 6B, the speed feedback signal ω that is the actual speed by the pulse generator 14, and will be described later. The U-phase current feedback signal iu from the current detector 64 and the W-phase current feedback signal iw from the current detector 65 are input and newly set using a preset magnetic pole phase correction angle μ (details will be described later). The rotational phase Φ is obtained, the speed control, the current control and the pulse width control are performed, and the power converters 57 and 76, for example, the gates of the IGBTs are turned on and off by the PWM signals Vup, Vvp and Vwp, and the three-phase pulse sine wave voltage The signals U1, V1, W1 and U2, V2, W2 are supplied to the first synchronous motor 6A and the second synchronous motor 6B, respectively.

  The current detector 64 is a sensor that detects a current to be measured in a non-contact manner using, for example, a Hall element that is a magnetoelectric conversion element, and is a U-phase winding between the power converter 57 and the first synchronous motor 6A. U is an output current that is connected in series to the line to control the gate of the IGBT inherent in the power converter 57 and to form a pulsed sine wave voltage signal U1 that is commensurate with the load of the first synchronous motor 6A. It is a detector that detects the phase current feedback signal iu. Similarly to the current detector 64, the current detector 65 is connected in series to the W-phase winding between the power converter 57 and the first synchronous motor 6A and detects the W-phase current feedback signal iw. is there. The subtractor 53 differentiates the speed command ω * for the first synchronous motor 6A and the second synchronous motor 6B and the pulse signal from the pulse generator 14 with a differentiator (not shown), and is obtained as a result of the calculation. The speed feedback signal ω is input, the speed feedback signal ω is subtracted from the speed command ω *, and the obtained speed deviation Δω is output to the speed and current control unit 73 based on the calculation result.

  The speed and current control unit 73 includes a speed deviation Δω by the subtractor 53, a new rotational phase Φ of the first synchronous motor 6A by the rotational phase correcting means 74 described later, and a U-phase current feedback signal iu by the current detector 64. And a W-phase current feedback signal iw from the current detector 65, a torque voltage command V * q and an excitation voltage command V * d, which are two-phase voltage commands, are generated and output to the coordinate converter 75. To do. Details of the speed and current control unit 73 will be described later.

The rotational phase correction means 74 includes a subtractor 78. The subtractor 78 inputs the rotational phase γ of the first synchronous motor 6A and the magnetic pole phase correction angle μ by the integrator 58 described above, and subtracts the magnetic pole phase correction angle μ from the rotational phase γ of the first synchronous motor 6A. Then, the new rotation phase Φ of the first synchronous motor 6A obtained is output to the coordinate converter 75 based on the calculation result. Here, the magnetic pole phase correction angle μ is calculated by the following equation (1).
μ = (θ1 + θ2) / 2 (1)
The new rotational phase Φ of the first synchronous motor 6A is calculated by the following equation (2).
Φ = γ−μ = γ− (θ1 + θ2) / 2 (2)
In other words, the magnetic pole phase correction angle μ indicates an intermediate angle between the first magnetic pole phase θ1 and the second magnetic pole phase θ2, and the first magnetic pole phase θ1 and the second magnetic pole phase θ2 are set by the operator. By measuring, it is preset in the twin drive controller 72A. Thus, the rotational phase correction means 74 outputs a new rotational phase Φ obtained by correcting the rotational phase γ of the first synchronous motor 6A to the coordinate converter 75.

  FIG. 4 shows the first and second magnetic pole phases θ1 and θ2 of the permanent magnets in the first and second synchronous motors 6A and 6B and the torque current (corresponding to the primary current) of the first synchronous motor 6A. ) Is separated into a torque component and an excitation component in the rotating magnetic field coordinate system, with the horizontal axis indicating the d-axis (excitation axis) and the vertical axis indicating the q-axis (torque axis). Here, Φm1 and Φm2 indicate magnetic fluxes of the permanent magnets of the first synchronous motor 6A and the second synchronous motor 6B, respectively. Further, i1 indicates a primary current of the first synchronous motor 6A, and iq and id indicate a q component and a d component of the primary current i1 of the first synchronous motor 6A, respectively. As shown in FIG. 4, the magnetic pole phase correction angle μ in the equation (1) indicates an intermediate angle between the angles formed by the first magnetic pole phase θ1 and the second magnetic pole phase θ2.

  The coordinate converter 75 has the same function as the coordinate converter 55 of FIG. 15 described above, inputs a new rotation phase Φ instead of the new rotation angle Φ1 shown in FIG. V * q and excitation voltage command V * d are respectively input, and torque voltage command V * q and excitation voltage command V * d are fixed to the fixed axis coordinates from the rotation axis coordinate system according to the new rotation angle Φ. Coordinate conversion calculation is performed on the system, and the obtained three-phase sine wave voltage commands V * u, V * v, and V * w are output to the PWM controller 56 based on the calculation result.

  As described above, the PWM controller 56 inputs the three-phase sine wave voltage commands V * u, V * v, V * w and the carrier wave signal (not shown in FIG. 1), and inputs the three-phase sine wave. The pulse voltage control of the wave voltage commands V * u, V * v, V * w is carried out by a carrier wave signal such as a triangular wave signal, and PWM signals Vup, Vvp, Vwp which are three-phase pulse signals are generated to generate a power converter 57. , 76.

  The power converter 76 receives the three-phase PWM signals Vup, Vvp, and Vwp in the same manner as the power converter 57 described above, and the three-phase PWM signals Vup, Vvp, and Vwp serve as a gate drive circuit (not shown). The three-phase pulse sine wave voltage for driving the second synchronous motor 6B by turning on and off the switching elements, for example, the gates of the IGBT (six), which are present in the power converter 76 by the generated drive signal. The signals U2, V2, and W2 are supplied to the UVW phase windings of the second synchronous motor 6B.

  In this case, as shown in FIG. 1, the connection method between the power converter 76 that supplies power to the second synchronous motor 6B and the PWM controller 56 is such that the phase order is the UVW phase in the case of the first synchronous motor 6A. On the other hand, in the case of the second synchronous motor 6B, the order is reversed as in the WVU phase. Thereby, when the twin drive control device 72A controls the first synchronous motor 6A and the second synchronous motor 6B, the twin drive control device 72A can rotate in the opposite directions at the same rotational coordinate. Therefore, the twin drive control device 72A can connect the first synchronous motor 6A and the second synchronous motor 6B to face each other and rotate the rotary cutter 2 in the same direction.

  According to the above description, the twin drive control device 72A has the first synchronous motor 6A and the second synchronous motor 6A and the second synchronous motor 6A and the second synchronous motor 6B even when the initial magnetic pole phase between the first synchronous motor 6A and the second synchronous motor 6B is different. A single three-phase sinusoidal voltage command V * u, V * v, V * is used as a signal for turning on and off the gates of the IGBTs inherent in the power converters 57 and 76 for supplying power to the synchronous motor 6B. A single PWM signal Vup, Vvp, Vwp is used based on w. In FIG. 18A, the phase difference between the sine wave voltage command M1 for the first synchronous motor 6A and the sine wave voltage command M2 for the second synchronous motor 6B is equal to zero because it never exists. Thus, the difference signal (high-frequency excitation signal) in FIG. 18E becomes zero. Therefore, since the power control of the first synchronous motor 6A and the second synchronous motor 6B is performed based on a single PWM signal, the torque generated by both the synchronous motors 6A and 6B can be made the same. Torsional vibration of the load can be suppressed.

  Next, the speed and current control unit 73 shown in FIG. 1 will be described in detail. 2 and 3, the speed and current control unit 73 includes a speed controller 81, a field controller 82, a cosine calculator 83, a sine calculator 88, a coordinate converter 85, adders / subtracters 84 and 89, and current control. Devices 87 and 90, a field controller 91, a sine calculator 93, and a cosine calculator 92.

  The speed controller 81 subtracts the speed deviation Δω obtained by subtracting the speed feedback signal ω of the first synchronous motor 6A from the speed command ω * for the first synchronous motor 6A and the second synchronous motor 6B by the subtractor 53. Then, the speed deviation Δω is speed-controlled (the speed deviation Δω is multiplied by a speed amplifier gain constant), and a torque command τ * is output to the field controller 82 as an operation amount.

  The field controller 82 receives the torque command τ *, multiplies the torque command τ * by the base speed ω0 of the first synchronous motor 6A, and returns the speed feedback of the first synchronous motor 6A to the value obtained as a result of the calculation. The inverse of the signal ω is multiplied, and the obtained torque current command i * 1 is output to the cosine calculator 83 and the sine calculator 88 according to the calculation result.

  FIG. 5 shows the relationship between the magnetic pole phase difference angle and the torque current command of the first and second synchronous motors, with the horizontal axis indicating the d-axis (excitation axis) and the vertical axis indicating the q-axis (torque axis). Here, i * d and i * q indicate a d-axis component and a q-axis component of the torque current command i * 1. According to FIG. 5, the magnetic flux Φm1 exists at a position advanced from the d-axis coordinate by the magnetic pole phase difference angle δ. This is because the magnetic flux Φm1 generated by the first synchronous motor 6A is shifted from the d-axis coordinate by the magnetic pole phase difference angle δ, so that the torque current command i * 1 is orthogonal to the magnetic flux Φm1 according to the torque. This is because it is necessary to provide the d-axis torque current command (hereinafter referred to as excitation current command) i * d and the q-axis torque current command (hereinafter referred to as torque current command) i * q.

  Returning to FIG. 2, the cosine calculator 83 receives the torque current command i * 1, and multiplies the torque current command i * 1 by a cosine value COSδ determined according to the magnetic pole phase difference angle δ shown in FIG. Based on the result of the calculation, a torque component current command i * q, which is an obtained value, is output to the adder / subtractor 84. The sine calculator 88 receives the torque current command i * 1, multiplies the torque current command i * 1 by a sine value SINδ determined according to the magnetic pole phase difference angle δ shown in FIG. The obtained value is multiplied by “−1”, and the excitation current command i * d, which is the obtained value, is output to the adder / subtracter 89 based on the result of the calculation.

  On the other hand, the coordinate converter 85 includes a U-phase current feedback signal iu that is a current feedback signal flowing in the U-phase winding of the first synchronous motor 6A, and a W-phase current feedback signal that is a current feedback signal flowing in the W-phase winding. iw and a new rotation phase Φ of the synchronous motor 6A are input, and a V-phase current feedback signal iv is calculated from the U-phase current period signal iu and the W-phase current feedback signal iv, and the obtained UVW three-phase is obtained according to the calculation result Current feedback signals iu, iv, and iw are subjected to coordinate transformation calculation according to the new rotation phase Φ of the first synchronous motor 6A, and the obtained torque component current feedback signal iq and excitation component current feedback signal are obtained according to the calculation result. The id is output to the adder / subtractor 84 and the adder / subtractor 89, respectively.

  The adder / subtractor 84 inputs a torque component current command i * q, a torque component current feedback signal iq, and a field control torque component current command i * δq to be described later for field control. The current feedback signal iq is subtracted, the field control torque component current command i * δq is added to the current deviation obtained from the calculation result, and the corrected torque component current deviation Δiq is obtained from the calculation result as the current controller. Output to 87. As in the case of the adder / subtractor 84, the adder / subtracter 89 inputs an excitation current command i * d, an excitation current feedback signal id, and a field control excitation current command i * δd to be described later for field control. The excitation current feedback signal id is subtracted from the current division command i * d, and the field control excitation current command i * δd is added to the current deviation obtained from the calculation result. The excitation current deviation Δid is output to the current controller 90.

  The current controller 87 inputs the torque component current deviation Δiq, performs current control on the torque component current deviation Δiq (multiplies the torque component current deviation Δiq by the current amplifier gain constant), and obtains the obtained torque component voltage command V * q. The operation amount is output to the coordinate converter 75. As in the case of the current controller 87, the current controller 90 inputs the excitation component current deviation Δid and controls the excitation component current deviation Δid (multiplies the field component current deviation Δid by the current amplifier gain constant). The obtained excitation voltage command V * d is output to the coordinate converter 75 as an operation amount.

Next, means for generating the field control torque component current command i * δq and the field control excitation component current command i * δd will be described with reference to FIG. First, the field control current command i * δ output by the field controller 91 is a value obtained by multiplying the current I1 flowing through the winding of the first synchronous motor 6A by (1-ω0 / ω) as a field control component. is there. Φm1 / L1 is used in place of the current I1 (see the following formula (3)). Here, Φm1 indicates the magnetic flux generated by the permanent magnet of the first synchronous motor 6A, and L1 indicates the inductance of the winding in the first synchronous motor 6A. I1 is a current that flows in the motor winding for generating a magnetic flux equivalent to the magnetic flux Φm1 by the permanent magnet of the first synchronous motor 6A.
I1 = Φm1 / L1 (3)

  That is, the field controller 91 inputs Φm1 / L1 set in advance, multiplies it by (1-ω0 / ω) as a field control component, and signs the field control current command i * δ obtained as a result of the calculation. The result is output to the calculator 93 and the cosine calculator 92. The sine calculator 93 receives the field control current command i * δ, multiplies the field control current command i * δ by a sine value SINδ that changes in accordance with the magnetic pole phase difference angle δ, and obtains a value obtained as a result of the calculation. Is multiplied by “−1”, and the obtained field control torque component current command i * δq is output to the adder / subtractor 84 based on the result of the calculation. The cosine calculator 92 receives the field control current command i * δ, multiplies the field control current command i * δ by a cosine value COSδ that changes in accordance with the magnetic pole phase difference angle δ, and obtains a value obtained as a result of the calculation. Is multiplied by “−1”, and the field control excitation current command i * δd obtained is output to the adder / subtractor 89 based on the result of the calculation. As a result, the twin drive control device 72A allows the first synchronous motor 6A to match the speed command ω * in the range where the speed feedback signal ω of the first synchronous motor 6A ranges from “0” to the maximum speed. And the second synchronous motor 6B can be controlled.

  As described above, the twin drive control device 72A can suppress the torsional vibration of the load even when the initial magnetic pole phases of the first synchronous motor 6A and the second synchronous motor 6B are different. In addition, the first synchronous motor 6A and the second synchronous motor 6B are controlled so as to coincide with the speed command ω * when the speed feedback signal ω of the first synchronous motor 6A ranges from “0” to the maximum speed. can do.

[Operation]
Next, the operation of the twin drive control device 72A will be described. The twin drive control device 72A is roughly described. The speed command ω * for the first synchronous motor 6A and the second synchronous motor 6B, the speed feedback signal ω of the first synchronous motor 6A, the first drive motor 72A, The current feedback signals iu and iw flowing in the UW-phase windings of the synchronous motor 6A are input, and a single three-phase PWM signal Vup, Vvp, and Vwp are generated using a preset magnetic phase correction angle μ. It outputs to the power converters 57 and 76. Further, when a drive signal generated through a gate drive circuit (not shown) by the three-phase PWM signals Vup, Vvp, Vwp is input to the gates of the IGBTs of the power converters 57, 76, a pulsed sine wave voltage signal U1, V1, W1 and U2, V2, W2 are supplied to the first synchronous motor 6A and the second synchronous motor 6B, respectively.

  As shown in FIG. 1, the speed deviation Δω between the speed command ω * and the speed feedback signal ω, the new rotational phase Φ of the first synchronous motor 6A, and the U-phase current feedback signal by the current detector 64 When iu and the W-phase current feedback signal iw from the current detector 65 are input to the speed and current control unit 73, the torque and voltage control command V * q, which is a two-phase voltage command, and excitation are generated by the speed and current control unit 73. The divided voltage command V * d is output to the coordinate converter 75, respectively. Since the speed and current control unit 73 differs in part in the control method depending on the magnitude of the speed feedback signal ω of the first synchronous motor 6A, the details of the operation will be described later.

  When the magnetic pole phase correction angle μ and the rotational phase γ of the first synchronous motor 6A are input to the rotational phase correction means 74, the rotational phase correction means 74 sets the magnetic pole phase correction angle μ from the rotational phase γ of the first synchronous motor 6A. The new rotation phase Φ obtained by the subtraction is output to the coordinate converter 75. When the torque divided voltage command V * q, the excitation divided voltage command V * d, and the new rotation angle Φ are input to the coordinate converter 75, the coordinate converter 75 causes the torque divided voltage command V * q and the excitation divided voltage command V to be input. * D is coordinate-converted according to the new rotation phase Φ, and UVW three-phase sine wave voltage commands V * u, V * v, and V * w are output to the PWM controller 56.

  Three-phase sinusoidal voltage commands V * u, V * v, V * w and a carrier wave signal (not shown in FIG. 1) by the triangular wave generator shown in FIG. Then, the PWM controller 56 controls the pulse width of the three-phase sinusoidal voltage commands V * u, V * v, and V * w with a carrier wave signal, and uses the UVW three-phase PWM signals Vup, Vvp, and Vwp for gates. Output to a drive circuit (not shown). When the three-phase PWM signals Vup, Vvp, Vwp are input to a gate drive circuit (not shown), a drive for turning on and off the gate of the IGBT in the power converter 57 by a gate drive circuit (not shown). The signal is output to the power converter 57.

  When the drive signal is input to the power converter 57, the power converter 57 turns on and off the gate of the IGBT of the power converter 57, and the three-phase pulsed sine wave voltage signals U1, V1, and W1 are generated. 1 is supplied to the UVW phase windings of one synchronous motor 6A. In the case of the second synchronous motor 6B, the three-phase PWM signals Vup, Vvp, Vwp are shared by the PWM controller 56 in the case of the first synchronous motor 6A, and the three-phase PWM signals Vup, Vvp, Vwp are shared. When the three-phase PWM signals Vup, Vvp, and Vwp by the PWM controller 56 are respectively input to the WVU phase of the power converter 76, the power converter 76 As in the case of the first synchronous motor 6, the three-phase pulsed sine wave voltage signals U 2, V 2, and W 2 are supplied to the UVW phase windings of the second synchronous motor 6 B by 76.

  Next, details of the operation of the speed and current control unit 73 will be described. FIG. 6 shows the relationship between the speed and torque characteristics of the synchronous motor by the twin drive control device according to the present invention, wherein the horizontal axis ω is the speed feedback signal ω of the first synchronous motor 6A, and the vertical axis is ω0 / ω. Each is shown. According to FIG. 6, the twin drive controller 72A sets ω0 / ω to “1” in the range from the speed feedback signal ω of the first synchronous motor 6A from “0” to the base speed ω0, that is, a constant value. It can be seen that torque control is performed, and constant output control, that is, field weakening control is performed in a range where the speed feedback signal ω exceeds the base speed ω0 and reaches the maximum speed.

  First, the case where the speed feedback signal ω of the first synchronous motor 6A is in the range from “0” to the base speed ω0 will be described. In this range, constant torque control is performed on the first synchronous motor 6A and the second synchronous motor 6B. That is, the field controllers 82 and 91 shown in FIGS. 2 and 3 output signals obtained by multiplying the input signals by “1” and “0”, respectively.

  As shown in FIG. 2, the speed and current control unit 73 subtracts the speed feedback signal ω from the speed command ω * of the first synchronous motor 6A and the second synchronous motor 6B by the subtractor 53 and obtains a speed deviation. When Δω is input to the speed controller 81, the speed controller 81 outputs a torque command τ * (equal to the torque current command τ * 1) as an operation amount to the field controller 82, and the cosine is directly passed through the field controller 82. The result is output to the calculator 83 and the sine calculator 88. When the torque current command i * 1 is input to the cosine calculator 83, the cosine calculator 83 multiplies the torque current command i * 1 by the cosine value COSδ, and the torque component current command i * q is input to the adder / subtractor 84 as an operation amount. Output. Similarly, when the torque current command i * 1 is input to the sine calculator 88, the sine calculator 88 multiplies the torque current command i * 1 by the sine value SINδ, and the value obtained by the calculation is “−1”. And the obtained excitation current command i * d is output to the adder / subtractor 89 as an operation amount.

  When the current feedback signals iu and iw flowing through the U-phase and W-phase windings of the first synchronous motor 6A and the new rotation angle Φ of the first synchronous motor 6A are input to the coordinate converter 85. The V-phase current feedback signal iv is calculated from the U-phase and W-phase current feedback signals iu and iw. Then, the coordinate converter 85 performs coordinate conversion of the current feedback signals iu, iw, iv of each phase of the UVW according to the new rotation angle Φ, and outputs the torque component current feedback signal iq that is a two-phase DC current feedback signal and excitation. The divided DC feedback signal id is output to the adders / subtracters 84 and 89, respectively.

  Further, when the torque component current command i * q and the torque component current feedback signal iq are input to the adder / subtractor 84, the torque subtractor current feedback signal iq is subtracted from the torque component current command i * q by the adder / subtractor 84, and the calculation result is obtained. As a result, the obtained torque component current deviation Δiq is output to the current controller 87. When the excitation current command i * d and the excitation current feedback signal id are input to the adder / subtractor 89, the adder / subtractor 89 subtracts the excitation current feedback signal id from the excitation current command i * d to calculate the current. As a result, the obtained excitation current deviation Δid is output to the current controller 90. Here, the field control torque component current command i * δq and the field control excitation component current command i * δd are each zero because the field controller 91 shown in FIG. 3 multiplies the input signal by “0” and outputs it. The adder / subtractor 84 and the adder / subtractor 89 are not used for the calculation.

  When the torque component current deviation Δiq is input to the current controller 87, the current controller 87 performs current control on the torque component current deviation Δiq, and the coordinate shown in FIG. Output to the converter 75. When the excitation current deviation Δid is input to the current controller 90, the current controller 90 performs current control on the excitation current deviation Δid and uses the excitation voltage command V * d as the manipulated variable to adjust the coordinates shown in FIG. Output to the converter 75.

  Next, the case where the speed feedback signal ω of the first synchronous motor 6A is in the range from the base speed ω0 to the maximum speed will be described. In this range, field control is performed on the first synchronous motor 6A and the second synchronous motor 6B. That is, the field controllers 82 and 91 shown in FIGS. 2 and 3 output signals obtained by multiplying the input signals by “ω0 / ω” and “1-ω0 / ω”, respectively.

  As shown in FIG. 2, the speed and current control unit 73 subtracts the speed feedback signal ω from the speed command ω * of the first synchronous motor 6A and the second synchronous motor 6B by the subtractor 53 and obtains a speed deviation. When Δω is input to the speed controller 81, the speed controller 81 outputs a torque command τ * as an operation amount to the field controller 82. When the torque command τ * is input to the field controller 82, the field controller 82 outputs the torque current command i * 1 to the cosine calculator 83 and the sine calculator 88 as an operation amount. Here, the torque current command i * 1 is a value obtained by multiplying the torque command τ * by “ω0 / ω” because the field controller 82 outputs the input signal multiplied by “ω0 / ω”.

  When the torque current command i * 1 is input to the cosine calculator 83, the cosine calculator 83 multiplies the torque current command i * 1 by the cosine value COSδ, and the obtained torque component current command i * is obtained according to the calculation result. q is output to the adder / subtractor 84 as an operation amount. Similarly, when the torque current command i * 1 is input to the sine calculator 88, the sine calculator 88 multiplies the torque current command i * 1 by the sine value SINδ, and the value obtained by the calculation is “−1. And the obtained excitation current command i * d is output to the adder / subtractor 89 as an operation amount.

  When the current feedback signals iu and iw flowing through the U-phase and W-phase windings of the first synchronous motor 6A and the new rotation angle Φ of the first synchronous motor 6A are input to the coordinate converter 85. The V-phase current feedback signal iv is calculated from the U-phase and W-phase current feedback signals iu and iw. Then, the coordinate converter 85 performs coordinate conversion of the current feedback signals iu, iw, iv of each phase of the UVW according to the new rotation angle Φ, and outputs the torque component current feedback signal iq that is a two-phase DC current feedback signal and excitation. The divided DC feedback signal id is output to the adders / subtracters 84 and 89, respectively.

  On the other hand, the current I1 flowing through the winding of the first synchronous motor 6A for applying the magnetic flux equivalent to the magnetic flux Φm1 by the permanent magnet of the first synchronous motor 6A is Φm1 / L1 as described above. When the current I1 (Φm1 / L1) flowing through the winding of the first synchronous motor 6A is input to the field controller 91, the field controller 91 adds a function “(1−ω0 / ω)” to the current I1 flowing through the motor winding. The field control current command i * δ obtained by the multiplication is output to the sine calculator 93 and the cosine calculator 92.

  When the field control current command i * δ is input to the sine calculator 93, the sine calculator 93 multiplies the field control current command i * δ by a sine value SINδ determined in accordance with the magnetic pole phase difference angle δ. Then, the obtained value is multiplied by “−1”, and the obtained field control torque component current command i * δq is output to the adder / subtractor 84 shown in FIG. When the field control current command i * δ is input to the cosine calculator 92, the cosine calculator 92 multiplies the field control current command i * δ by the cosine value COSδ determined according to the magnetic pole phase difference angle δ. As a result, the obtained value is multiplied by “−1”, and the obtained field control excitation current command i * δd is output to the adder / subtractor 89 shown in FIG.

  Returning to FIG. 2, when the torque component current command i * q, the torque component current feedback signal iq, and the field control torque component current command i * δq are input to the adder / subtractor 84, the adder / subtractor 84 causes the torque component current command i * Subtract the torque component current feedback signal iq from q, add the field control torque component current command i * δq to the value obtained from the calculation result, and output the obtained torque component current deviation Δiq to the current controller 87. . Further, when the excitation current command i * d, the excitation current feedback signal id, and the field control excitation current command i * δd are input to the adder / subtractor 89, the adder / subtractor 89 converts the excitation current command i * d from the excitation current command i * d. The current feedback signal id is subtracted, the field control excitation current command i * δd is further added to the obtained value based on the calculation result, and the obtained excitation current deviation Δid is output to the current controller 90.

  When the torque component current deviation Δiq is input to the current controller 87, the current controller 87 performs current control on the torque component current deviation Δiq, and the coordinate converter shown in FIG. Output to 75. When the excitation current deviation Δid is input to the current controller 90, the current controller 90 performs current control on the excitation current deviation Δid, and the coordinate converter shown in FIG. Output to 75.

  FIG. 10 is a diagram showing an experimental example of the relationship between the frequency and the signal level when the twin drive control device 72A shown in FIG. 1 is applied to the rotary cutter device 71A. FIG. 10A shows the frequency and signal level when the rotary cutter 2 is of the one-side drive system, the horizontal axis shows the frequency and the vertical axis shows the signal level. FIG. 10B shows the frequency and signal level when the rotary cutter 2 is of the double-side drive system, the horizontal axis shows the frequency and the vertical axis shows the signal level. Here, the frequency on the horizontal axis indicates a frequency obtained by performing a differential operation on the pulse signal from the pulse generator 14 to obtain a speed and analyzing the frequency based on the speed. The peripheral speed of the rotary cutter 2 is 10 mpm (meter / min). As can be seen from FIG. 10A, in the case of the one-side drive method, the rotary cutter 2 exhibits a resonance phenomenon in the vicinity of 115 Hz. Further, according to FIG. 10B, it can be seen that the rotary cutter 2 does not cause a resonance phenomenon in the case of the double-side drive method. Here, in both figures, the signal level is high in the region where the frequency is near 1 Hz. This is due to uneven rotation of the rotary cutter 2. Further, the signal level is high even in the region where the frequency is around 5 Hz. This is due to the rotational speed of the rotary cutter 2 itself.

[Modification]
Next, a modification of the twin drive control device according to the present invention will be described. FIG. 19 is a control block diagram when another twin drive control device according to the present invention is applied to a rotary cutter device. The twin drive control device 72B includes a power converter 77 instead of the power converters 57 and 76 in the twin drive control device 72A shown in FIG. It is assumed that power converter 77 has a current capacity that is at least twice that of power converter 57 or 76. Further, the UVW three-phase output line of the power converter 77 is connected to the first synchronous motor 6A and the second synchronous motor 6B so as to be in the normal phase order and the reverse phase order, respectively. Since the twin drive control device 72B is basically the same as the twin drive control device 72A, detailed description thereof is omitted.

  As described above, the twin drive control device 72B is similar to the twin drive control device 72A even if the initial magnetic pole phase between the first synchronous motor 6A and the second synchronous motor 6B is different. A single three-phase sinusoidal voltage command V * is used as a signal for turning on and off the gate of each IGBT in the power converter 77 that supplies power to the first synchronous motor 6A and the second synchronous motor 6B. A single PWM signal Vup, Vvp, Vwp based on u, V * v, V * w is used. In FIG. 18A, the phase difference between the sine wave voltage command M1 for the first synchronous motor 6A and the sine wave voltage command M2 for the second synchronous motor 6B is equal to zero because it never exists. Thus, the difference signal (high-frequency excitation signal) in FIG. 18E becomes zero. Therefore, since the power control of the first synchronous motor 6A and the second synchronous motor 6B is performed based on a single PWM signal, the torque generated by both the synchronous motors 6A and 6B can be made the same. Torsional vibration of the load can be suppressed. Further, the first synchronous motor 6A and the second synchronous motor 6B are controlled so that the speed feedback signal ω of the first synchronous motor 6A matches the speed command ω * in the range from “0” to the maximum speed. be able to.

[Torsional vibration suppression method]
Next, the torsional vibration suppressing method according to the present invention will be described. FIG. 7 is a flowchart showing the procedure of the torsional vibration suppression method according to the present invention. FIG. 8 is a continuation of the flowchart of FIG. FIG. 9 is a continuation of the flowchart of FIG.

  First, the torsional vibration suppression method will be described briefly. Referring to FIG. 1, in controlling the first synchronous motor 6A and the second synchronous motor 6B, a three-phase sine wave voltage command V * u , V * v, V * w are three-phase sinusoidal voltage commands V * u, V * v, V * so that PWM signals Vup, Vvp, Vwp obtained by pulse width modulation become a single signal. Based on w, the PWM controller 56 generates PWM signals Vup, Vvp, Vwp, and on / off operation of the gates of the IGBTs included in the power converters 57, 76 based on the single PWM signals Vup, Vvp, Vwp. Thus, the same electric power is supplied to the first synchronous motor 6A and the second synchronous motor 6B. Here, the connection between the PWM controller 56 and the power converters 57 and 76 is connected to the power converters 57 and 76 in the order of normal phase and reverse phase, respectively.

  With reference to FIG. 7, first, the magnetic pole phase θ1 of the first synchronous motor 6A and the magnetic pole phase θ2 of the second synchronous motor 6B are detected in advance by the operator's work (step S-1). Then, the twin drive control device 72A sets the first magnetic pole phase θ1 and the second magnetic pole phase θ2, and when the twin drive control is started, inputs the speed command ω * (step S-2), The speed feedback signal ω, which is the actual speed of the first synchronous motor 6A, is input (step S-3), and the speed feedback signal ω is integrated to output the rotational phase γ of the first synchronous motor 6A (step S-4). ), A new rotational phase Φ of the first synchronous motor 6A and the second synchronous motor 6B is obtained by the above-described equation (1) (step S-5).

  On the other hand, the speed feedback signal ω is subtracted from the speed command ω *, and the obtained speed deviation Δω is output based on the calculation result (step S-6), and the speed deviation Δω is speed-controlled as shown in FIG. The torque command τ * is output (step S-7). When the speed feedback signal ω of the first synchronous motor 6A is in the range from “0” to the base speed ω0, the torque command τ * is output as the torque current command i * 1, and the speed feedback signal ω is the base speed. When it is within the range from ω0 to the maximum speed, the torque command τ * is subjected to field control, and the torque command τ * is multiplied by the base speed ω0 as the torque current command i * 1. The inverse value of the feedback signal ω is multiplied and the resulting value ω0 / ω × τ * is output (step S-8). Further, the torque current command i * 1 is decomposed into a torque component current command i * q and an excitation component current signal i * d by the magnetic pole phase difference angle δ (step S-9).

  Also, the V-phase current feedback signal iv is calculated from the U-phase and W-phase current feedback signals iu, iw (step S-10), and the three-phase current feedback signals iu, iv, iw is subjected to coordinate conversion in accordance with the new rotation phase Φ of the first synchronous motor 6A, and a torque component current feedback signal iq and an excitation component current feedback signal id are output (step S-11).

  On the other hand, referring to FIG. 9, when the speed feedback signal ω is in the range from “0” to the base speed ω0, the field control current command i * δ is set to “0”, and the speed feedback signal ω sets the base speed ω0. If it is in the range up to the maximum speed, the magnetic flux Φm1 is multiplied by the reciprocal of the inductance L1 of the motor winding, the resulting current value I1 is field-controlled, and a field control current command i * δ is output (step S-12). ). When the speed feedback signal ω is in the range from the base speed ω0 to the maximum speed, the field control current command i * δ is decomposed into a torque component and an excitation component based on the magnetic pole phase difference angle δ, and the field control torque is The divided current command i * δq and the exciting current command i * δd for field control are output (step S-13).

  Returning to FIG. 8, the torque component current command i * q, the torque component current feedback signal iq, and the field control torque component current command i * δq (applicable only when the field control is performed) are input, and they are added and subtracted. Torque current deviation Δiq is output (step S-14A). Then, current control is performed on the torque component current deviation Δiq, and a torque component voltage command V * q is output (step S-15A). Similarly to the case of step S-14A, the excitation current command i * d, the excitation current feedback signal id, and the field control excitation current command i * δd (applicable only for field control) are input. Are added and subtracted to output an excitation current deviation Δid (step S-14B). Then, as in the case of step S-15A, the excitation component current deviation Δid is subjected to current control, and the excitation component voltage command V * d is output (step S-15B).

  Returning to FIG. 7, the torque divided voltage command V * q and the excitation divided voltage command V * d are coordinate-converted according to the new rotation angle Φ, and the three-phase sine wave voltage commands V * u, V * v are converted. , V * w is output (step S-16). Then, the single PWM controller 56 outputs three-phase PWM signals Vup, Vvp, Vwp from the three-phase sinusoidal voltage commands V * u, V * v, V * w and the carrier wave signal (step S− 17) Three-phase PWM signals Vup, Vvp, and Vwp are input by the power converters 57 and 76 in the order of the normal phase and the reverse phase, respectively, and the three-phase pulsed sine wave voltage signals U1, V1, W1, and U2, V2 and W2 are supplied to the respective windings of the first synchronous motor 6A and the second synchronous motor 6B (step S-18). Then, the first synchronous motor 6A and the second synchronous motor 6B are driven (step S-19).

  From the above description, the torsional vibration suppression method can be applied to the first synchronous motor 6A and the second synchronization motor even when the initial magnetic pole phase between the first synchronous motor 6A and the second synchronous motor 6B is different. In the power converters 57 and 76 for supplying power to the motor 6B, the signals of the IGBTs included in each of the power converters 57 and 76 are turned on and off as single three-phase sinusoidal voltage commands V * u, V * v, and V * w. Based on a single PWM signal Vup, Vvp, Vwp. That is, since the power control of the first synchronous motor 6A and the second synchronous motor 6B is performed based on a single PWM signal, the torque generated by both the synchronous motors 6A and 6B can be made the same. Torsional vibration of the load can be suppressed. Further, the first synchronous motor 6A and the second synchronous motor 6B are controlled so that the speed feedback signal ω of the first synchronous motor 6A matches the speed command ω * in the range from “0” to the maximum speed. be able to.

  The above-described torsional vibration suppression method has been described using the rotary cutter device 71A shown in FIG. 1, but can be similarly applied to the rotary cutter device 71B shown in FIG.

  The present invention has been described with reference to the embodiment. However, the present invention is not limited to the above embodiment, and various modifications can be made without departing from the technical idea thereof. For example, the rotary cutter 2 has been described as an example of the load. However, the present invention is not limited to this, and the present invention is a load driven from both sides of the rotary shaft by the first synchronous motor 6A and the second synchronous motor 6B. If so, it can be applied.

  In the rotary cutter devices 71A and 71B shown in FIGS. 1 and 19, the speed deviation Δω and the rotational phase γ are calculated from the speed feedback signal ω of the first synchronous motor 6A. You may make it calculate from the speed feedback signal of the motor 6B.

It is a control block diagram at the time of applying the twin drive control apparatus by this invention to a rotary cutter apparatus. It is a control block diagram of the speed and current control means in the twin drive control device according to the present invention. FIG. 3 is a continuation of the control block diagram of FIG. 2. It is a figure which shows the relationship between each initial magnetic pole phase of a 1st and 2nd synchronous motor, the magnetic flux in a synchronous motor, and a primary current. It is a figure which shows the relationship between the magnetic pole phase difference angle of 1st and 2nd synchronous motor, and a torque electric current command. It is a figure which shows the relationship between the speed and torque characteristic of a synchronous motor in the twin drive control apparatus by this invention. It is a flowchart which shows the procedure of the torsional vibration suppression method by this invention. It is a continuation of the flowchart of FIG. It is a continuation of the flowchart of FIG. It is a figure which shows the experimental example of the relationship between the frequency and signal level at the time of applying the twin drive control apparatus by this invention to a rotary cutter apparatus. It is a figure which shows the example of the rotary cutter of a twin drive system. It is a control block diagram in the conventional first twin drive control device. It is a control block diagram in the conventional 2nd twin drive control apparatus. It is a control block diagram in the conventional 3rd twin drive control apparatus. It is a control block diagram in the conventional 4th twin drive control apparatus. It is a control block diagram of a speed and current control unit in a conventional fourth twin drive control device. It is a figure which shows the example of the input-output signal in a power converter and a PWM controller. It is a figure which shows the high frequency excitation voltage by the sine wave voltage command in case the magnetic pole phases of both synchronous motors differ in the conventional 4th twin drive control apparatus. It is a control block diagram at the time of applying the other twin drive control apparatus by this invention to a rotary cutter apparatus.

Explanation of symbols

2 Rotary cutters 4A, 4B Reducers 6A, 6B Motor 10 Speed command generation circuits 12, 14, 14A, 14B Pulse generators 16, 16A, 16B FVC (frequency / voltage conversion circuit)
18 Subtractor 20 Speed amplifier 22 Clamp circuit 24A, 24B Vector control circuit 26A, 26B Power unit 30 Reverse phase compensation circuit 32 Subtractor 34 Speed amplifier 36 Differentiation circuit 38 Gain circuit 40 Adders 51, 71A, 71B Rotary cutter devices 52, 72A , 72B Twin drive controller 53 Subtractor 54, 73 Speed and current controller 55, 61, 75 Coordinate converter 56, 62 PWM controller 57, 63, 76, 77 Power converter 58 Integrator 59, 60, 78 Subtractors 64, 65 Current detector 74 Rotation phase correction means 81 Speed controller 82 Field controller 83 Cosine calculator 84, 89 Adder / subtractor 85 Coordinate converter 87, 90 Current controller 88 Sine calculator 91 Field controller 92 Cosine calculator 93 Sine calculator 101A, 101B Absolute position detector ( E)
102A, 102B Pulse / speed converter 103A, 103B Vector control inverter 104 Speed regulator (ASR)
105, 107 Subtractor 106 Phase adjuster (APR)
108 Differentiator 109 Phase error amplifier 110, 112 Adder 111 Speed error amplifier ω * Speed command ω Speed feedback signal (actual speed)
ω0 Base speed τ * Torque command i * 1 Torque current command 0 * Zero current command δ Magnetic pole phase difference angle μ Magnetic pole phase correction angle γ First motor rotational phase Φ First motor new rotational phase i * q Torque Divided current command i * d Excited divided current command iu U phase current feedback signal iv V phase current feedback signal iw W phase current feedback signal i * δ Field control current command i * δq Field control torque component current command i * δd Field control excitation Divided current command iq Torque divided current feedback signal id Excitation current feedback signal L1 Inductance Φm1 of first motor winding Magnetic flux Φm2 by first motor Magnetic flux Φ1 by second motor First magnetic pole phase θ2 Second magnetic pole Phase V * q Torque voltage command V * d Excitation voltage command V * u U phase sine wave voltage command V * v V phase sine wave voltage command V * w W phase sine wave voltage command Vup U phase PWM signal Vvp V-phase PWM signal
Vwp W-phase PWM signal U1 U-phase pulsed sine wave voltage signal (for first motor)
V1 V-phase pulsed sinusoidal voltage signal (for first motor)
W1 W-phase pulsed sine wave voltage signal (for first motor)
U2 U-phase pulsed sine wave voltage signal (for second motor)
V2 V-phase pulsed sine wave voltage signal (for second motor)
W2 W-phase pulsed sine wave voltage signal (for second motor)

Claims (5)

A twin drive control device for suppressing torsional vibration of the load in a system in which a rotatable load is driven by a first synchronous motor and a second synchronous motor from both sides of the rotation shaft,
Rotational phase correction means for correcting the rotational phase of the first synchronous motor based on the initial magnetic pole phase of each of the first and second synchronous motors;
The speed deviation between the speed command for the first synchronous motor and the actual speed of the first synchronous motor is controlled, and the torque current command obtained by the speed control and the actual current flowing through the winding of the first synchronous motor 3 is obtained by performing coordinate conversion on the two-phase DC voltage command obtained by the current control according to the rotation phase corrected by the rotation phase correcting means, The first and second AC voltage commands for the phase are controlled by pulse width control using a predetermined carrier wave signal, and based on a single three-phase PWM signal common to the first and second synchronous motors obtained by the pulse width control. Control means for controlling the power supplied to the two synchronous motors;
A twin drive control device comprising: The control means includes
The speed deviation between the speed command for the first synchronous motor and the actual speed of the first synchronous motor is controlled, and the torque current command obtained by the speed control and the actual current flowing through the winding of the first synchronous motor A speed and current control means for current-controlling the current deviation between the two and generating a torque divided voltage command and an excitation divided voltage command as a two-phase DC voltage command;
The torque divided voltage command and the excitation divided voltage command as the two-phase DC voltage command generated by the speed and current control unit are coordinate-converted according to the rotation phase corrected by the rotation phase correction unit, and three-phase Coordinate conversion means for generating an AC voltage command;
PWM control means for controlling the pulse width of the three-phase AC voltage command generated by the coordinate conversion means with a predetermined carrier wave signal and generating a single three-phase PWM signal common to the first and second synchronous motors When,
Power conversion means for converting power based on a single three-phase PWM signal generated by the PWM control means and generating a pulsed AC voltage signal to be supplied to the first and second synchronous motors;
The twin drive control device according to claim 1, comprising: The rotational phase correction means provided in the control means is
The initial magnetic pole phase of the first synchronous motor and the initial magnetic pole phase of the second synchronous motor are added, the addition result is multiplied by a half, and the rotation phase of the multiplication result is rotated by the rotation of the first synchronous motor. 3. The twin drive control device according to claim 1, wherein the rotational phase of the first synchronous motor is corrected by subtracting from the phase. The speed and current control means of the control means are:
Speed-controlling the speed deviation between the speed command for the first synchronous motor and the actual speed of the first synchronous motor;
Field control of the torque command obtained by the speed control according to the actual speed of the first synchronous motor;
The current deviation between the torque current command obtained by the field control and the actual current flowing through the winding of the first synchronous motor and the magnetic flux generated by the first synchronous motor are divided by their inductances, and the division result is divided into the first Current control is performed by adding a field control current command obtained by field control according to the actual speed of the synchronous motor,
The twin drive control device according to any one of claims 1 to 3, wherein a torque divided voltage command and an excitation divided voltage command are generated as a two-phase DC voltage command. A method for suppressing torsional vibration of the load in a system in which a rotatable load is driven by a first synchronous motor and a second synchronous motor, respectively, from both sides of the rotating shaft,
A rotational phase correcting step for correcting the rotational phase of the first synchronous motor based on the initial magnetic pole phase of each of the first and second synchronous motors;
A speed detecting step for detecting an actual speed of the first synchronous motor;
A speed control step for controlling a speed deviation between a speed command for the first synchronous motor and the actual speed;
A current control step for controlling the current deviation between the torque current command obtained by the speed control and the actual current flowing through the first synchronous motor;
A coordinate conversion step of converting the two-phase DC voltage command obtained by the current control according to the corrected rotation phase;
PWM control step for controlling the pulse width of a three-phase AC voltage command obtained by the coordinate conversion using a predetermined carrier wave signal;
A power control step of controlling power supplied to the first and second synchronous motors by a single three-phase PWM signal common to the first and second synchronous motors obtained by the pulse width control;
A method for suppressing torsional vibration, comprising:

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