JP4929212B2 - FM receiver, method thereof, program thereof and recording medium - Google Patents

Description

  The present invention relates to an FM receiver, an FM reception processing method, an FM reception processing program, and a recording medium on which the FM reception processing program is recorded.

  2. Description of the Related Art Conventionally, an FM receiver that receives and processes an FM broadcast wave or the like obtained by FM (Frequency Modulation) modulation of a composite signal including a pilot signal serving as a reference for demodulation processing in addition to a main channel signal and a subchannel signal Etc. are widely spread. In such a device, in synchronization with the pilot signal in the detection signal, a reference signal used to convert the subchannel signal in the detection signal into a baseband signal having the same frequency band as the main channel signal in the detection signal is generated. In order to achieve this, a phase-locked loop (PLL) technique is generally employed (see Patent Document 1; hereinafter referred to as “Conventional Example 1”).

  In addition to the technique of Conventional Example 1, a technique of an FM receiver that generates the reference signal without using the PLL technique is also proposed (see Patent Document 2; hereinafter referred to as “Conventional Example 2”). . In the technique of Conventional Example 2, the phase difference between the internally generated base signal and the pilot signal in the detected signal is obtained. The obtained phase difference is added to the base signal.

JP 2000-13339 A JP 2006-528451 Gazette

  Although the technique of the conventional example 1 employs the PLL technique, the PLL is generally difficult to design. In addition, when the PLL method is used, it is difficult to synchronize with the pilot signal with good follow-up in a situation where the reception environment changes, such as in the case of mobile reception of FM broadcast waves. .

  Further, in the technique of Conventional Example 2 described above, a reference signal for matching the phase of the predetermined signal is necessary. However, when the frequency and phase of the predetermined signal fluctuate due to the occurrence of multipath, it is not always easy to generate a reference signal for synchronizing with the predetermined signal with high accuracy.

  For this reason, there is a need for a technique that can easily and quickly generate a reference signal that is synchronized with a pilot signal and reproduce high-quality audio from the received FM broadcast wave. Meeting this requirement is one of the problems to be solved by the present invention.

  The present invention has been made in view of the above, and an object of the present invention is to provide an FM receiver and an FM reception processing code that can perform high-quality sound reproduction with a simple configuration.

According to the first aspect of the present invention, the first signal and the second signal that are orthogonal to each other are reflected on the composite signal obtained by FM-detecting the FM broadcast wave by reflecting the phase of the pilot signal in the composite signal. A quadrature signal generating means for generating; a phase calculating means for calculating a phase reflecting the pilot signal based on the first signal and the second signal; and the pilot signal based on a calculation result by the phase calculating means A reference signal generating means for generating a reference signal having a predetermined relationship with a signal corresponding to a subchannel signal that is a signal in a frequency band higher than the frequency of the pilot signal in the composite signal, using the reference signal A baseband for converting to a baseband subchannel signal that is a signal in a frequency band lower than the frequency of the pilot signal Converting means; wherein the quadrature signal generating means performs orthogonalization of the composite signal, the orthogonalization means for generating a first orthogonal signal and the second quadrature signal are orthogonal to each other; said first orthogonal Filter means for selectively passing the first signal included in the signal and the second signal included in the second orthogonalized signal; and the orthogonal signal generating means in addition to the orthogonalization An FM receiver characterized by performing frequency shift .

According to the seventh aspect of the present invention, the first signal and the second signal which are made orthogonal to each other by reflecting the phase of the pilot signal in the composite signal with respect to the composite signal obtained by FM detecting the FM broadcast wave. A quadrature signal generation step to generate; a phase calculation step to calculate a phase reflecting the pilot signal based on the first signal and the second signal; and a pilot signal based on a calculation result in the phase calculation step A reference signal generating step for generating a reference signal having a predetermined relationship with the reference signal; and a signal corresponding to a sub-channel signal that is a signal in a frequency band higher than the frequency of the pilot signal in the composite signal, using the reference signal Baseband sub-channel signal that is a signal in a frequency band lower than the frequency of the pilot signal. And de conversion step; wherein the quadrature signal generating step performs orthogonalization of the composite signal, the orthogonalization step of generating a first orthogonal signal and the second quadrature signal are orthogonal to each other; said first quadrature And a filtering step of selectively passing the first signal included in the converted signal and the second signal included in the second orthogonalized signal. In the orthogonal signal generating step, in addition to the orthogonalization Then, an FM reception processing method characterized by performing frequency shift .

The invention according to claim 8 is an FM reception processing program characterized by causing an arithmetic means to execute the FM reception processing method according to claim 7 .

The invention according to claim 9 is a recording medium in which the FM reception processing program according to claim 8 is recorded so as to be readable by a calculation means.

  Hereinafter, embodiments of the present invention will be described with reference to the accompanying drawings. In the following description, the same or equivalent elements are denoted by the same reference numerals, and redundant description is omitted.

[First Embodiment]
First, a first embodiment of the present invention will be described with reference to FIGS.

<Configuration>
FIG. 1 is a block diagram illustrating a schematic configuration of an FM receiver 500A according to the first embodiment. As shown in FIG. 1, the FM receiver 500 </ b> A includes an antenna 510, a control unit 520 </ b> A, and a front end unit 530. The FM receiver 500A includes a detection unit 540, a stereo demodulation unit 550A, and an audio processing unit 560. Furthermore, the FM receiver 500A includes an operation input unit 570 and speakers 580L and 580R.

  The antenna 510 receives broadcast waves. A reception result by the antenna 510 is output to the front end unit 530 as a reception signal RFS.

  The controller 520A controls the overall operation of the FM receiver 500A. The control unit 520A will be described later.

  The front end unit 530 selects a reception desired station according to the channel selection command CSL from the control unit 520A, and outputs the channel selection processing result to the detection unit 540 as an intermediate frequency signal IFD of a predetermined intermediate frequency. The front end unit 530 includes a tuning circuit (not shown) and a local oscillation circuit (OSC).

  The tuning circuit selects the signal of the desired reception station based on the local oscillation signal from the local oscillation circuit. The extraction result by the tuning circuit is sent to the detection unit 540 as an intermediate frequency signal IFD having a predetermined intermediate frequency.

  The local oscillation circuit includes an oscillator that can control the oscillation frequency by voltage control or the like. This local oscillation circuit generates a local oscillation signal having a frequency corresponding to a reception desired station in the tuning circuit in accordance with the channel selection command CSL supplied from the control unit 520A, and supplies the local oscillation signal to the tuning circuit.

  Note that the above tuning circuits all include an input filter (not shown), a radio frequency amplifier (RF-AMP), a band pass filter (RF filter), a mixer (mixer), an intermediate frequency filter (IF (Intermediate)). Frequency) filter), AD (Analogue to Digital) converter (ADC), and the like.

  The detection unit 540 performs digital detection processing on the intermediate frequency signal IFD from the front end unit 530 by a predetermined method to generate a detection signal DCD (hereinafter also simply referred to as “signal DCD”) as a composite signal. The signal DCD generated in this way is output toward the stereo demodulator 550A.

In the first embodiment, as shown in FIG. 2, the signal DCD includes a left channel signal (hereinafter also referred to as “L component” or simply “L”) and a right channel signal (hereinafter referred to as “R component”). or simply referred to as "R") and the sum of (L + R) the angular frequency 0～Omega _c of the frequency band of the signal component SG1 consisting of, i.e., a main channel signal SG1 (hereinafter, "main channel signal (L + R)" also referred to as) Is included. Further, the signal DCD is a signal component SG2 in the frequency band of angular frequencies ω _{c to} 3ω _c composed of the difference between the L component and the R component (L _M −R _M ), that is, the sub channel signal SG2 (hereinafter referred to as “sub channel”). Signal (L _M -R _M ) ”). Further, the signal DCD includes a pilot signal PS having an angular frequency ω _c .

Here, the sub-channel signal SG2 is the frequency band of the signal of the angular frequency 0~ω _c, 2 times the angular frequency of the signal _{(αsin [2 (ω c ·} t + φ 0)]) of the pilot signal PS amplitude modulation Signal.

  Returning to FIG. 1, stereo demodulation section 550A performs stereo demodulation processing on signal DCD from detection section 540, and generates demodulated signals LDA and RDA. As shown in FIG. 3, the stereo demodulator 550A having such a function includes low-pass filters (LPF) 551M and 551S, a subchannel processor 552A, an adder 553L, and a subtractor 553R.

The LPF 551M receives the signal DCD from the detection unit 540. Then, the LPF 551M selectively allows the components of the angular frequency 0 to ω _c to pass through. As a result, the main channel signal (L + R) selectively passes through the LPF 551M. Thus, the main channel signal (L + R) that has passed through the LPF 551M is output to the adder 553L and the subtractor 553R as the signal MCA.

In the first embodiment, in order to simplify the description, it is assumed that the pilot signal PS included in the signal DCD is expressed by the following equation (1).
PS (t) ∝sin [θ _C (t)] = sin (ω _C t + φ ₀ ) (1)
Here, the value φ ₀ is the phase of pilot signal PS when time t = 0.

The LPF 551S is configured in the same manner as the LPF 551M. The LPF 551S receives the signal SCP from the subchannel processing unit 552A. Then, the LPF 551S selectively allows the components of the angular frequency 0 to ω _c to pass through. As a result, the sub-channel signal (LR) selectively passes through the LPF 551S. The subchannel signal (LR) thus passed through the LPF 551S is output as the signal SCA to the adder 553L and the subtractor 553R.

The subchannel processing unit 552A receives the signal DCD from the detection unit 540. Then, the subchannel processing unit 552A processes the subchannel signal (L _M −R _M ) in synchronization with the pilot signal PS included in the signal DCD to generate the signal SCP. As shown in FIG. 4, the sub-channel processing unit 552A having such a function includes an orthogonal signal generation unit 110A as an orthogonal signal generation unit and a phase calculation unit 120A as a phase calculation unit. The sub-channel processing unit 552A includes a reference signal generation unit 130A as a reference signal generation unit and a signal processing unit 140 as a baseband conversion unit.

The orthogonal signal generator 110A generates two signals OSA ₁ and OSA ₂ whose components of the angular frequency ω _C are orthogonal to each other from the signal DCD received from the detector 540, and then the pilot signal PS from the signals OSA ₁ and OSA ₂ The two signals PSA ₁ and PSA ₂ reflecting the phase θ _C (t) of are extracted. As shown in FIG. 5, the orthogonal signal generation unit 110A having such a function includes an orthogonalization unit 112A as orthogonalization means and filters (FIL) 113A ₁ and 113A ₂ as filter means.

As shown in FIG. 6, the orthogonalizing unit 112A includes a 90 ° phase shift unit 119 related to a signal having an angular frequency ω _C. In the orthogonalizing unit 112A configured as described above, a signal in phase with the received signal DCD is output to the FIL 113A ₁ and the signal processing unit 140 as the signal OSA ₁ . On the other hand, the orthogonalizing unit 112A shifts the phase of the received signal DCD by 90 ° with respect to the component of the angular frequency ω _C and outputs the signal OSA ₂ to the FIL 113A ₂ and the signal processing unit 140.

Returning to FIG. 5, the FIL 113A ₁ is configured as a digital filter such as an infinite impulse response filter (IIR). The FIL 113A ₁ receives the signal OSA ₁ from the orthogonalizing unit 112A. Then, FIL 113A ₁ selectively passes the component of angular frequency ω _C in signal OSA ₁ , that is, the signal component corresponding to pilot signal PS, and outputs the signal as signal PSA ₁ toward phase calculation unit 120A.

Note that a phase shift Δθ due to filter delay may occur through the FIL 113A ₁ . In the first embodiment, it is assumed that such a phase shift Δθ occurs. This phase shift Δθ is determined by the configuration of the FIL 113A ₁ and is determined at the design stage of the FIL 113A ₁ .

The signal PSA ₁ output from the FIL 113A ₁ is expressed by the following equation (2).
PSA ₁ (t) = A (t) · sin [θ _S (t)]
= A (t) · sin [θ _C (t) −Δθ]
= A (t) · sin [(ω _C t + φ ₀ ) −Δθ] (2)
Here, A (t) represents the amplitude value of the pilot signal PS.

The FIL 113A ₂ is configured in the same manner as the FIL 113A ₁ . The FIL 113A ₂ receives the signal OSA ₂ from the orthogonalizing unit 112A. Then, FIL 113A ₂ selectively passes the component of angular frequency ω _C in signal OSA ₂ , that is, the signal component corresponding to pilot signal PS, and outputs signal PSA ₂ .

The signal PSA ₂ output from the FIL 113A ₂ is expressed by the following equation (3).
PSA ₂ (t) = A (t) · cos [θ _S (t)]
= A (t) · cos [θ _C (t) −Δθ]
= A (t) · cos [(ω _C t + φ ₀ ) −Δθ] (3)

Returning to FIG. 4, the phase calculation unit 120A calculates the phase θ _C (t) of the pilot signal PS based on the signal PSA ₁ and the signal PSA ₂ from the orthogonal signal generation unit 110A. At the time of such calculation, the phase calculation unit 120A calculates, for example, the phase θ _C (t) by performing an arctan calculation on the signal PSA ₁ and the signal PSA ₂ and correcting the amount of the phase shift Δθ described above. To do.

Θ _C (t) calculated in this way is output as a signal PHA from the phase calculation unit 120A to the reference signal generation unit 130A.

The reference signal generation unit 130A generates reference signals BSA ₁ and BSA ₂ having an angular frequency of 2ω _C based on the signal PHA from the phase calculation unit 120A. As shown in FIG. 7, the reference signal generation unit 130A having such a function includes a phase processing unit 131A and signal generation units 132 ₁ and 132 ₂ .

The phase processing unit 131A receives the signal PHA from the phase calculation unit 120A and processes the phase θ _C (t) indicated by the signal PHA. In the first embodiment, the phase processing unit 131A calculates the phase θ _M (t) by the following equation (4).
θ _M (t) = 2θ _C (t) = 2 (ω _C t + φ ₀ ) (4)

That is, in the first embodiment, the phase processing unit 131A calculates the phase θ _M (t) that changes at the angular frequency 2ω _C. The phase θ _M (t) calculated in this way is output to the signal generators 132 ₁ and 132 ₂ as the phase processing signal MPA.

The signal generator 132 ₁ generates the reference signal BSA ₁ based on the phase processing signal MPA from the phase processing unit 131A. In the first embodiment, the signal generator 132 ₁ includes a sine value table in which amplitude values corresponding to the phase values are registered, and a sine wave having a phase θ _M (t) indicated by the phase processing signal MPA. The signal is generated as the reference signal BSA ₁ .

This reference signal BSA ₁ is expressed by the following equation (5).
BSA ₁ (t) = C ₀ · sin [θ _M (t)]
= C ₀ · sin [2θ _C (t)]
= C ₀ · sin [2 (ω _C t + φ ₀ )] (5)

The signal generator 132 ₂ generates the reference signal BSA ₂ based on the phase processing signal MPA from the phase processing unit 131A. In the first embodiment, signal generator 132 ₂ includes a cosine value table amplitude value corresponding to the phase value is registered, the cosine wave phase theta _M indicated by the phase processing signal MPA (t) A signal is generated as a reference signal BSA ₂ .

This reference signal BSA ₂ is expressed by the following equation (6).
BSA ₂ (t) ≈C ₀ · cos [θ _M (t)]
= C ₀ · cos [2θ _C (t)]
= C ₀ · cos [2 (ω _C t + φ ₀ )] (6)

The reference signals BSA ₁ and BSA ₂ generated in this way are output toward the signal processing unit 140.

Returning to FIG. 4, the signal processing unit 140 receives the signals OSA ₁ and OSA ₂ from the orthogonal signal generation unit 110A and the reference signals BSA ₁ and BSA ₂ from the reference signal generation unit 130A. Then, the signal processing unit 140 processes the signals OSA ₁ and OSA ₂ using the reference signals BSA ₁ and BSA ₂ , and a baseband subchannel signal (LR) in the frequency band of the angular frequency 0 to ω _c. Is generated. The signal processing unit 140 having such a function includes processing units 141 ₁ and 141 ₂ and an addition unit 142 as shown in FIG.

The processing unit 141 ₁ receives the signal OSA ₁ from the orthogonal signal generation unit 110A and the signal BSA ₁ from the reference signal generation unit 130A. Then, the processing unit 141 _1, processes the signal OSA ₁ using a signal BSA _1, generates a signal MSA ₁ in the frequency band of the angular frequency 0~ω _c.

The processing unit 141 ₂ receives the signal OSA ₂ from the orthogonal signal generation unit 110A and the signal BSA ₂ from the reference signal generation unit 130A. Then, the processing unit 141 _2, processes the signals OSA ₂ using a signal BSA _2, generates a signal MSA ₂ frequency bands of the angular frequency 0~ω _c.

Each processing unit 141 _j (j = 1, 2) having such a function includes a multiplication unit 146 as shown in FIG.

Multiplier 146 receives signal OSA _j at the A terminal and receives reference signal BSA _j at the B terminal. Then, the multiplication result of the reception result at the A terminal and the reception result at the B terminal is output from the C terminal to the adding unit 142 as the signal MSA _j . Here, the signal corresponding to the signal component SG2 (that is, the subchannel signal (L _M −R _M )) in the signal OSA _j has an angular frequency component of the angular frequency ω _{C to} 3ω _C , and the reference signal BSA _j is It has only an angular frequency 2ω _C component. For this reason, the signal component corresponding to the signal component SG2 in the mixed signal MXA _j is frequency-converted into a component having an angular frequency of 0 to ω _{C and} a component having an angular frequency of 3ω _{C to} 5ω _C.

Returning to FIG. 8, the adding unit 142 calculates the sum of the signal received at the A terminal and the signal received at the B terminal, and outputs the sum from the C terminal. In the addition unit 142 receives a signal MSA ₁ from the processing unit 141 ₁ in the A terminal receives a signal MSA ₂ from the processing unit 141 ₂ at the B terminal. Then, the addition result of the signal MSA ₁ and the signal MSA ₂ is output from the C terminal to the LPF 551S as the signal SCP. Although description using mathematical expressions is omitted, the LPF 551S is supplied as a signal SCP obtained by frequency-shifting the composite signal with a subcarrier so as to be the baseband subchannel component (LR) shown in FIG. It has become so.

  Returning to FIG. 3, the adding unit 553L receives the signal MCA from the LPF 551M and the signal SCA from the LPF 551S. The adder 553L relates only to the L channel by calculating the sum of the main channel signal (L + R) supplied as the signal MCA and the baseband subchannel signal (LR) supplied as the signal SCA. To generate a signal. The generation result is output to the audio processing unit 560 as a signal LDA.

  Subtractor 553R receives signal MCA from LPF 551M and signal SCA from LPF 551S. The subtracting unit 553R relates only to the R channel by calculating a difference between the main channel signal (L + R) supplied as the signal MCA and the baseband subchannel signal (LR) supplied as the signal SCA. To generate a signal. The generation result is output to the audio processing unit 560 as a signal RDA.

  Returning to FIG. 1, the audio processing unit 560 includes a digital-analog converter, an electronic volume, a power amplifier, and the like (all not shown) prepared for the L channel. The audio processing unit 560 includes a digital-analog converter, an electronic volume, a power amplifier, and the like (all not shown) prepared for the R channel.

  Audio processing unit 560 converts each of signals LDA and RDA received from stereo demodulation unit 550A into an analog signal. Then, the volume and the like are adjusted in accordance with the audio processing command VLC from the control unit 520A. The results thus obtained are output to the L-channel speaker 580L and the R-channel speaker 580R as the L-channel audio signal LSA and the R-channel audio signal RSA.

  The operation input unit 570 includes a key unit provided in the main body of the FM receiver 500A and / or a remote input device including the key unit. Here, as a key part provided in the main body part, a touch panel provided in the display part can be used. Moreover, it can replace with the structure which has a key part, or can also employ | adopt the structure input with a sound using a voice recognition technique in combination. The result of operation input to the operation input unit 570 is sent to the control unit 520A as operation input data IPD.

  The speaker 580L reproduces and outputs the L channel sound in accordance with the L channel audio signal LSA from the audio processing unit 560. The speaker 580R reproduces and outputs the R channel sound in accordance with the R channel audio signal RSA from the audio processing unit 560.

  As described above, the control unit 520A controls the overall operation of the FM receiver 500A. This control unit 520A issues a channel selection command CSL for selecting a reception desired station to the front end unit 530 in accordance with the channel selection designation from the operation input unit 570. Further, the control unit 520A issues an audio processing command VLC to the audio processing unit 560 for setting the volume or the like corresponding to the designation of the volume or the like according to the designation of the volume or the like from the operation input unit 570.

<Operation>
The operation of FM receiving apparatus 500A configured as described above will be described mainly focusing on stereo demodulation processing.

  As a premise, in FM receiving apparatus 500A, it is assumed that the desired station has been selected according to the channel selection designation from operation input unit 570. That is, it is assumed that control unit 520A generates a channel selection command CSL corresponding to the desired station and outputs it to front end unit 530.

  In FM receiving apparatus 500A, it is assumed that the output volume from speakers 580L and 580R is adjusted in accordance with the designation of the volume from operation input unit 570. That is, it is assumed that the control unit 520A generates an audio processing command VLC corresponding to the designation of the volume and outputs the audio processing command VLC to the audio processing unit 560.

  When broadcast waves from various broadcast stations are received by the antenna 510, the reception result is sent from the antenna 510 to the front end unit 530 as a reception signal RFS (see FIG. 1). The front end unit 530 that has received the reception signal RFS extracts a signal corresponding to the desired station from the reception signal RFS, and sends the extraction result to the detection unit 540 as an intermediate frequency signal IFD of an intermediate frequency (see FIG. 1).

Receiving the intermediate frequency signal IFD, the detection unit 540 performs digital detection processing on the intermediate frequency signal IFD by a predetermined method to generate a signal DCD that is a composite signal. As described above, the signal DCD includes the main channel signal (L + R), the pilot signal PS, and the subchannel signal (L _M −R _M ) in the first embodiment (see FIG. 2). The signal DCD generated in this way is output toward the stereo demodulator 550A.

Upon receiving the signal DCD, the stereo demodulation unit 550A supplies the signal DCD to the LPF 551M and the subchannel processing unit 552A (see FIG. 3). LPF551M which has received the signal DCD sends a main channel signal is a component of the angular frequency 0～Omega _c a (L + R) selectively passes, as a signal MCA to the adder 553L and the subtraction unit 553R.

On the other hand, in the subchannel processing unit 552A, the orthogonal signal generation unit 110A receives the signal DCD (see FIG. 4). In the orthogonal signal generation unit 110A that receives the supply of the signal DCD, first, the orthogonalization unit 112A generates _two signals OSA ₁ and OSA _{2 in} which the components of the angular frequency ω _C included in the signal DCD are orthogonal to each other, and the FIL 113A. ₁ and 113A _{2 as} well as the signal processing unit 140 (see FIG. 5).

Subsequently, FIL113A ₁ which has received the signal OSA ₁ are components of the angular frequency omega _C in the signal OSA _1, i.e., selectively pass a signal component corresponding to the pilot signal PS, for a signal PSA ₁ to phase calculation section 120A Output. The FIL 113A ₂ that has received the signal OSA ₂ selectively allows the component of the angular frequency ω _C in the signal OSA ₂ , that is, the signal component corresponding to the pilot signal PS, to pass to the phase calculation unit 120A as the signal PSA _2. (See FIG. 5).

Here, the signal PSA ₁ has a waveform represented by the above-described equation (2). On the other hand, the signal PSA ₂ has a waveform represented by the above-described equation (3).

The phase calculator 120A that receives the signals PSA ₁ and PSA ₂ operates as described above to calculate the phase θ _C (t) of the pilot signal PS. The phase θ _C (t) calculated in this way is sent to the reference signal generator 130A as a signal PHA (see FIG. 4).

The reference signal generator 130A that has received the signal PHA generates reference signals BSA ₁ and BSA ₂ having an angular frequency of 2ω _C based on the signal PHA. In generating the reference signals BSA ₁ and BSA ₂ , in the reference signal generation unit 130A, first, the phase processing unit 131A calculates the phase θ _M (t) that changes at the angular frequency 2ω _C by the above-described equation (4). And sent to the signal generators 132 ₁ and 132 ₂ (see FIG. 7). Signal generating unit 132 _1, 132 ₂ having received the phase theta _M a (t) refers to the internal table on the basis of the phase theta _M (t), the above-described (5), represented by the formula (6) The reference signals BSA ₁ and BSA ₂ which are signals are generated. The reference signals BSA ₁ and BSA ₂ generated in this way are sent to the signal processing unit 140 (see FIG. 7).

The signal processing unit 140 that receives the reference signals BSA ₁ and BSA ₂ from the reference signal generation unit 130A and the signals OSA ₁ and OSA ₂ from the orthogonal signal generation unit 110A uses the reference signals BSA ₁ and BSA _2. The signals OSA ₁ and OSA ₂ are processed to generate a baseband subchannel signal (LR) in the frequency band of the angular frequency 0 to ω _c . Upon generation of such baseband sub-channel signal (L-R), the signal processing unit 140, first, the processing unit 141 _1, and calculates the product of the signal OSA ₁ and the signal BSA _1, to generate a signal MSA ₁ , the processing unit 141 _2, and calculates the product of the signal OSA ₂ and the signal BSA _2, generates a signal MSA _2. Then, the adding unit 142 calculates the sum of the signal MSA ₁ and the signal MSA ₂ to generate the signal SCP. The signal SCP generated in this way is sent to the LPF 551S.

LPF551S which receives the signal SCP from subchannel processor 552A selectively passes the component of the angular frequency 0~ω _c. As a result, the sub-channel signal (LR) selectively passes through the LPF 551S. The subchannel signal (LR) thus passed through the LPF 551S is output as the signal SCA to the adder 553L and the subtractor 553R.

  The adder 553L, which receives the signal MCA from the LPF 551M and the signal SCA from the LPF 551S, receives the main channel signal (L + R) supplied as the signal MCA and the baseband subchannel signal (LR) supplied as the signal SCA. To generate a signal related only to the L channel. The generation result is sent to the audio processing unit 560 as a signal LDA (see FIG. 3).

  The subtractor 553R that has received the signal MCA from the LPF 551M and the signal SCA from the LPF 551S receives the main channel signal (L + R) supplied as the signal MCA and the baseband subchannel signal (LR) supplied as the signal SCA. The signal related to only the R channel is generated by calculating the difference. The generation result is sent to the audio processing unit 560 as a signal RDA (see FIG. 3).

  The audio processing unit 560 that has received the signals LDA and RDA from the stereo demodulation unit 550A converts each of the signals LDA and RDA into an analog signal. Then, the audio processing unit 560 adjusts the volume and the like according to the audio processing command VLC from the control unit 520A. The results thus obtained are output to the L channel speaker 580L and the R channel speaker 580R as an L channel audio signal LSA and an R channel audio signal RSA (see FIG. 1).

  Upon receiving the audio signal LSA from the audio processing unit 560, the speaker 580L reproduces and outputs L channel audio in accordance with the audio signal LSA. Further, the speaker 580R that has received the audio signal RSA from the audio processing unit 560 reproduces and outputs the R channel audio in accordance with the audio signal RSA.

As described above, in the first embodiment, the pilot signal can be obtained from the detection signal DCD including the pilot signal PS of the angular frequency ω _C without using a feedback loop in the PLL system or a base signal that is a reference for phase measurement. The phase θ _C (t) of PS is derived. Therefore, it is possible to generate the reference signals BSA ₁ and BSA ₂ for generating the baseband subchannel signal (LR) that is easily and quickly synchronized with the pilot signal PS. The sound contained in the waves can be played faithfully. As a result, it is possible to reproduce the sound included in the FM broadcast wave with good quality.

Further, in this first embodiment, in consideration of the phase shift Δθ in FIL113A _1, 113A _2, since the calculation of the phase theta _C (t) of the pilot signal PS, calculates accurately the phase theta _C a (t) be able to. Therefore, it is possible to generate the reference signals BSA ₁ and BSA ₂ that are accurately synchronized with the pilot signal PS.

In the first embodiment, the sine value table is used when generating the reference signal BSA _{1 and} the cosine value table is used when generating the reference signal BSA _2. However, only one table is prepared. The other may be referred to with a shift of π / 2.

  In the first embodiment, the LPFs 551M and 551S are arranged before the adding unit 553L and the subtracting unit 553R, but may be arranged after the adding unit 553L and the subtracting unit 553R.

[Second Embodiment]
Next, a second embodiment of the present invention will be described with reference mainly to FIGS.

  FIG. 11 is a block diagram showing a schematic configuration of an FM receiver 500B according to the second embodiment. As shown in FIG. 11, the FM receiving device 500B includes a control unit 520B instead of the control unit 520A, and a stereo demodulation unit 550A compared to the FM receiving device 500A of the first embodiment. Instead, only the stereo demodulation unit 550B is provided. Hereinafter, description will be made mainly focusing on these differences.

  In addition to the function of the control unit 520A described above, the control unit 520B has a function of specifying the offset phase Δφ with respect to the stereo demodulation unit 550B. In order to implement such an offset phase designation function, the control unit 520B receives the intermediate frequency signal IFD from the front end unit 530. Then, control unit 520B calculates offset phase Δφ according to a predetermined algorithm based on the signal level of signal IFD, the level of signal components of adjacent interfering stations, and the like. The offset phase Δφ thus calculated is output to the stereo demodulator 550B as the offset phase signal POS.

  The stereo demodulator 550B differs from the stereo demodulator 550A described above only in that it includes a subchannel processor 552B instead of the subchannel processor 552A. As shown in FIG. 12, the subchannel processing unit 552B is different from the above-described subchannel processing unit 552A only in that a reference signal generation unit 130B is provided instead of the reference signal generation unit 130A.

The reference signal generation unit 130B generates reference signals BSB ₁ and BSB ₂ having an angular frequency of 2ω _C based on the signal PHA from the phase calculation unit 120A. As shown in FIG. 13, the reference signal generation unit 130B having such a function is arranged between the phase processing unit 131A and the signal generation units 132 ₁ and 132 _{2 as} compared to the reference signal generation unit 130A described above. The difference is that an adder 135 is further provided.

The adding unit 135 receives the phase processing signal MPA from the phase processing unit 131A and the offset phase signal POS from the control unit 520B. Then, the adding unit 135 adds the phase θ _M (t) reported by the phase processing signal MPA and the offset phase Δφ specified by the offset phase signal POS, and calculates the phase θ _P (t). .

Here, the phase θ _P (t) is expressed by the following equation (7).
θ _P (t) = θ _M (t) + Δφ = 2θ _C (t) + Δφ
= 2 (ω _C t + φ ₀ ) + Δφ (7)

The phase θ _P (t) calculated in this way is output to the signal generators 132 ₁ and 132 ₂ as the phase processing signal MPB. As a result, the signal generators 132 ₁ and 132 ₂ generate the reference signals BSB ₁ and BSB ₂ expressed by the following equations (8) and (9).
BSB ₁ (t) = C ₀ · sin [θ _P (t)]
= C ₀ · sin [2θ _C (t) + Δφ]
= C ₀ · sin [2 (ω _C t + φ ₀ ) + Δφ] (8)
BSB ₂ (t) = C ₀ · cos [θ _P (t)]
= C ₀ · cos [2θ _C (t) + Δφ]
= C ₀ · cos [2 (ω _C t + φ ₀ ) + Δφ] (9)

The reference signals BSB ₁ and BSB ₂ thus generated are output from the reference signal generation unit 130B to the signal processing unit 140.

<Operation>
Next, the operation of FM receiving apparatus 500B configured as described above will be described mainly focusing on stereo demodulation processing.

  As a premise, similarly to the case of the first embodiment described above, in FM receiving apparatus 500B, a station desired to receive corresponding to the channel selection is selected according to the channel selection from operation input unit 570. To do. In FM receiving apparatus 500B, it is assumed that audio processing such as output volume from speakers 580L and 580R is adjusted in accordance with designation of volume and the like from operation input unit 570.

  When broadcast waves from various broadcast stations are received by the antenna 510, the reception result is sent to the front end unit 530 as a reception signal RFS. The front end unit 530 that has received the reception signal RFS extracts a signal corresponding to the desired station from the reception signal RFS, and sends it to the control unit 520B and the detection unit 540 as an intermediate frequency signal IFD of the intermediate frequency (see FIG. 11).

  As described above, the control unit 520B that has received the intermediate frequency signal IFD calculates the offset phase Δφ based on the signal level and the interference signal level of the intermediate frequency signal IFD, and uses the stereo demodulation unit 550B (from Specifically, it is sent to the reference signal generator 130B) (see FIG. 11).

  The detection unit 540 that has received the intermediate frequency signal IFD performs digital detection processing on the intermediate frequency signal IFD by a predetermined method in the same manner as in the first embodiment, and generates a signal DCD that is a composite signal. The signal DCD generated in this way is sent to stereo demodulation section 550B.

In stereo demodulation section 550B that receives signal DCD, signal DCD is supplied to LPF 551M and subchannel processing section 552B. LPF551M which has received the signal DCD, as in the case of the first embodiment, selectively pass a main channel signal is a component of the angular frequency 0~ω _c (L + R), the addition unit 553L and subtraction as a signal MCA Send to part 553R.

On the other hand, in the subchannel processing unit 552B, the orthogonal signal generation unit 110A receives the signal DCD (see FIG. 12). In the orthogonal signal generation unit 110A that receives the supply of the signal DCD, the signals PSA ₁ and PSA ₂ are generated using the orthogonalization unit 112A and the FILs 113A ₁ and 113A ₂ as in the case of the first embodiment. . The signals PSA ₁ and PSA ₂ generated in this way are sent to the phase calculation unit 120A.

The phase calculation unit 120A that receives the signals PSA ₁ and PSA ₂ calculates the phase θ _C (t) of the pilot signal PS in the same manner as in the first embodiment. The phase θ _C (t) calculated in this way is sent as a signal PHA to the reference signal generator 130B (see FIG. 12).

The reference signal generator 130B that has received the signal PHA generates reference signals BSB ₁ and BSB ₂ having an angular frequency 2ω _C based on the signal PHA from the phase calculator 120A and the offset phase signal POS from the controller 520B. When generating the reference signals BSB ₁ and BSB ₂ , in the reference signal generation unit 130B, the phase processing unit 131A performs the angular frequency 2ω according to the above-described equation (4), as in the case of the reference signal generation unit 130A. _The phase θ _M (t) having _C is calculated and sent to the adder 135 as the phase processing signal MPA (see FIG. 13).

Upon receiving the phase processing signal MPA, the adding unit 135 adds the offset phase Δφ specified by the offset phase signal POS to the phase θ _M (t) according to the above-described equation (7), and adds the phase θ _P (t). calculate. The phase θ _P (t) calculated in this way is sent to the signal generators 132 ₁ and 132 ₂ (see FIG. 13).

Signal generating unit 132 _1, 132 ₂ having received the phase theta _P (t) refers to the internal table on the basis of the phase theta _P (t), represented by the above-described (8) and (9) The reference signals BSB ₁ and BSB ₂ which are signals are generated. The reference signals BSB ₁ and BSB ₂ generated in this way are sent to the signal processing unit 140 (see FIG. 13).

The signal processing unit 140 that receives the reference signals BSB ₁ and BSB ₂ from the reference signal generation unit 130B and the signals OSA ₁ and OSA ₂ from the orthogonal signal generation unit 110A uses the reference signals BSB ₁ and BSB _2. processing the signal OSA _1, OSA _2, generates a corresponding signal SCP to the baseband sub-channel signal in the frequency band of the angular frequency _{0~ω c (α (L-R} )). Here, the value α varies depending on the value of the offset phase Δφ.

The signal SCP generated in this way is proportional to the value of cos (Δφ) as represented by the following equation (10).
SCP (t) ∝cos (Δφ) (10)

For this reason, as shown in FIG. 14, the signal SCP has a signal component in the frequency band of the angular frequency 0 to ω _c and a signal whose amplitude value changes by changing the set value of the offset phase Δφ. It has become. The signal SCP generated in this way is sent to the LPF 551S.

Receiving the signal SCP from the subchannel processing unit 552B, the LPF 551S selectively passes the component of the angular frequency 0 to ω _c . As a result, the sub-channel signal (LR) selectively passes through the LPF 551S. The subchannel signal (LR) thus passed through the LPF 551S is output as the signal SCB to the adder 553L and the subtractor 553R.

  The adder 553L that has received the signal MCA from the LPF 551M and the signal SCB from the LPF 551S receives the main channel signal (L + R) supplied as the signal MCA and the baseband sub-channel signal (α (LR) supplied as the signal SCB. )) To calculate a signal mainly related to the L channel. The generation result is sent to the audio processing unit 560 as a signal LDB (see FIG. 11).

  Receiving the signal MCA from the LPF 551M and the signal SCB from the LPF 551S, the subtractor 553R receives the main channel signal (L + R) supplied as the signal MCA and the baseband subchannel signal (α (LR) supplied as the signal SCB. )) To generate a signal mainly related to the R channel. The generation result is sent to the audio processing unit 560 as a signal RDB (see FIG. 11).

  The audio processing unit 560 that has received the signals LDB and RDB from the stereo demodulation unit 550B converts each of the signals LDB and RDB into an analog signal in the same manner as in the first embodiment, and then receives the signal from the control unit 520B. The volume and the like are adjusted according to the audio processing command VLC. The results thus obtained are output to the L channel speaker 580L and the R channel speaker 580R as the L channel audio signal LSB and the R channel audio signal RSB (see FIG. 11).

  Upon receiving the audio signal LSB from the audio processing unit 560, the speaker 580L reproduces and outputs L channel audio in accordance with the audio signal LSB. In addition, the speaker 580R that has received the audio signal RSB from the audio processing unit 560 reproduces and outputs the R channel audio in accordance with the audio signal RSB.

As described above, in the second embodiment, as in the case of the first embodiment, from the detection signal DCD including the pilot signal PS of the angular frequency ω _C , the feedback loop in the PLL system and the reference for phase measurement Without using the base signal, the phase θ _C (t) of the pilot signal PS is derived. Therefore, the reference signals BSB ₁ and BSB ₂ for generating the baseband subchannel signal (LR) that is synchronized with the pilot signal PS can be generated easily and quickly. The sound contained in the wave can be played. As a result, it is possible to reproduce the sound included in the FM broadcast wave with good quality.

In the second embodiment, as in the case of the first embodiment, the phase θ _C (t) of the pilot signal PS is calculated in consideration of the phase shift Δθ in the FILs 113A ₁ and 113A ₂ . The phase θ _C (t) can be calculated well. Therefore, it is possible to generate the reference signals BSB ₁ and BSB ₂ that are accurately synchronized with the pilot signal PS.

In the second embodiment, the offset phase Δφ corresponding to the signal level of the intermediate frequency signal IFD from the front end unit 530 and the interference signal level is set, and the reference signals BSB ₁ and BSB ₂ reflecting the offset phase Δφ are set. Therefore, the amplitude of the baseband subchannel signal can be changed in accordance with the signal level of the signal DCD and the interference signal level. As a result, the degree of separation between the L channel sound and the R channel sound can be easily controlled according to the signal level of the desired station signal and the interference signal level in the received signal.

  In the second embodiment, as shown in FIG. 13, the offset phase signal POS is added to the phase processing signal MPA that is an output signal of the phase processing unit 131A. On the other hand, the offset phase signal POS may be input to the phase processing unit, and the offset phase signal POS may be added to the phase processing signal MPA and then multiplied by a predetermined value.

[Third Embodiment]
Next, a third embodiment of the present invention will be described with reference mainly to FIGS.

  FIG. 15 is a block diagram illustrating a schematic configuration of an FM receiver 500C according to the third embodiment. As shown in FIG. 15, the FM receiver 500C is different from the FM receiver 500A of the first embodiment only in that a stereo demodulator 550C is provided instead of the stereo demodulator 550A. Hereinafter, description will be made mainly focusing on these differences.

  Stereo demodulator 550C differs from stereo demodulator 550A described above only in that it includes subchannel processor 552C instead of subchannel processor 552A. As shown in FIG. 16, the subchannel processing unit 552C includes an orthogonal signal generation unit 110C instead of the orthogonal signal generation unit 110A, as compared with the above-described subchannel processing unit 552A, and is replaced with a phase calculation unit 120A. The difference is that the phase calculation unit 120C is provided and the reference signal generation unit 130C is provided instead of the reference signal generation unit 130A.

The orthogonal signal generation unit 110C generates two signals OSC ₁ and OSC ₂ whose components corresponding to the pilot signal PS are orthogonal to each other from the signal DCD received from the detection unit 540, and then the pilot signal from the signals OSC ₁ and OSC ₂ Two signals PSC ₁ and PSC ₂ reflecting the phase θ _C (t) of PS are extracted. As shown in FIG. 17, the orthogonal signal generating unit 110C having such a function includes a band limiting filter 111 as a composite band limiting unit, an orthogonalizing unit 112C as an orthogonalizing unit, and a filter (FIL) as a filtering unit. 113C ₁ and 113C ₂ are provided.

In the third embodiment, the limiting filter 111 is a digital filter such as a finite impulse response filter (FIR), and there is no phase shift in a required band. The band limiting filter 111 selectively passes a signal component of a predetermined frequency band including the angular frequencies ω _{C to} 3ω _C in the signal DCD, and is directed to the orthogonalization unit 112C as a band limited signal LSI whose frequency band is limited. Output.

In the third embodiment, the band limiting filter 111 has, for example, the filtering characteristics shown in FIG. That is, the band limiting filter 111, together with pass without being substantially attenuated relative to the signal component of the angular frequency 0～3Omega _C, the angular frequency is configured as a low pass filter for blocking a large excess signal component 3 [omega] _C .

Returning to FIG. 17, the orthogonalizing unit 112C generates _two signals OSC ₁ and OSC ₂ which are orthogonal to each other and whose components corresponding to the pilot signal PS are orthogonal to each other based on the band limited signal LSI. The orthogonalizing unit 112C having such a function includes an oscillating unit 210C and multiplying units 220 ₁ and 220 ₂ as shown in FIG.

The oscillator 210C generates a signal OTS ₁ to be supplied to the multiplier 220 ₁ and a signal OTS ₂ to be supplied to the multiplier 220 ₂ . In the third embodiment, the signal OTS ₁ and the signal OTS ₂ are expressed by the following equations (11) and (12).
OTS ₁ (t) = B ₀ · cos (ω _SH · t) (11)
OTS ₂ (t) = B ₀ · sin (ω _SH · t) (12)

In the third embodiment, the angular frequency ω _SH is set to a predetermined value larger than 3ω _C. The value of the angular frequency ω _SH is determined from a comprehensive viewpoint together with the band limitation specification by the band limitation filter 111 described above from the viewpoint of preventing the noise component from being mixed into the frequency shift result of the pilot signal PS.

Multiplier 220 ₁ receives band-limited signal LSI at the A terminal and receives signal OTS ₁ from oscillator 210C at the B terminal. Then, a reception result in the A terminal, the multiplication result between the reception result at the B terminal is outputted to the C terminal as signal OSC ₁ to FIL113C _1. Here, pilot signal PS in signal DCD has an angular frequency ω _C , and signal OTS ₁ has only a component of angular frequency ω _SH . For this reason, the signal component corresponding to the pilot signal PS in the signal OSC ₁ is frequency-converted into two components of an angular frequency (ω _SH −ω _C ) and an angular frequency (ω _SH + ω _C ).

Multiplier 220 ₂ receives band-limited signal LSI at terminal A and receives signal OTS ₂ from oscillator 210C at terminal B. Then, a reception result in the A terminal, the multiplication result between the reception result at the B terminal is outputted to the C terminal as a signal OSC ₂ to FIL113C _2. Here, pilot signal PS in signal DCD has an angular frequency ω _C , and signal OTS ₂ has only a component of angular frequency ω _SH . For this reason, the signal component corresponding to the pilot signal PS in the signal OSC ₂ is divided into _two components of the angular frequency (ω _SH −ω _C ) and the angular frequency (ω _SH + ω _C ) as in the case of the signal OSC _1. Converted.

Returning to FIG. 17, the FIL 113C ₁ is configured as a digital filter such as an infinite impulse response filter (IIR). The FIL 113C ₁ receives the signal OSC ₁ from the orthogonalizing unit 112C. The FIL 113C ₁ selectively passes _one of the components of the angular frequency (ω _SH −ω _C ) in the signal OSC ₁ , that is, the signal component corresponding to the pilot signal PS, to the phase calculation unit 120C as the signal PSC ₁ . Output toward.

Note that a fixed phase shift Δθ associated with the filter delay may occur through the FIL 113C ₁ . In the third embodiment, it is assumed that such a phase shift Δθ occurs. This phase shift Δθ is determined by the configuration of the FIL 113C ₁ and is determined at the design stage of the FIL 113C ₁ .

The signal PSC ₁ output from the FIL 113C ₁ is expressed by the following equation (13).
PSC ₁ (t) = − A (t) · sin [θ _S (t)]
= −A (t) · sin [ω _SH t−θ _C (t) −Δθ]
= −A (t) · sin [ω _SH t− (ω _C t + φ ₀ ) −Δθ] (13)
Here, A (t) represents the amplitude value of the pilot signal PS.

The FIL 113C ₂ is configured in the same manner as the FIL 113C ₁ . The FIL 113C ₂ receives the signal OSC ₂ from the orthogonalizing unit 112C. Then, FIL 113C ₂ selectively passes one of the components of angular frequency (ω _SH −ω _C ) in signal OSC ₂ , that is, the signal component corresponding to pilot signal PS, and outputs signal PSC ₂ .

The signal PSC ₂ output from the FIL 113C ₂ is expressed by the following equation (14).
PSC ₂ (t) = A (t) · cos [θ _S (t)]
= A (t) · cos [ω _SH t−θ _C (t) −Δθ]
= A (t) · cos [ω _SH t− (ω _C t + φ ₀ ) −Δθ] (14)

Returning to FIG. 16, the phase calculation unit 120C calculates the phase θ _C (t) of the pilot signal PS based on the signal PSC ₁ and the signal PSC ₂ from the orthogonal signal generation unit 110C. In this calculation, the phase calculation unit 120C performs, for example, an arctan operation on the signal PSC ₁ and the signal PSC ₂ and then considers the angular frequency ω _SH and the phase shift Δθ to reflect the pilot signal PS. Is calculated.

  The phase thus calculated is output as a signal PHA from the phase calculation unit 120C to the reference signal generation unit 130C.

The reference signal generation unit 130C generates the reference signals BSC ₁ and BSC ₂ of the angular frequency (ω _SH −2ω _C ) based on the signal PHA from the phase calculation unit 120C. As shown in FIG. 20, the reference signal generation unit 130C having such a function includes a phase processing unit 131C and signal generation units 132 ₁ and 132 ₂ .

The phase processing unit 131C receives the signal PHA from the phase calculation unit 120C and processes the phase θ _C (t) indicated by the signal PHA. In the third embodiment, the phase processing unit 131C calculates the phase θ _M (t) by the following equation (15).
θ _M (t) = ω _SH t−2θ _C (t) = ω _SH t−2 (ω _C t + φ ₀ ) (15)

That is, in the third embodiment, the phase processing unit 131C calculates the phase θ _M (t) that changes at the angular frequency (ω _SH -2ω _C ). The phase θ _M (t) calculated in this way is output to the signal generators 132 ₁ and 132 ₂ as the phase processing signal MPC.

As a result, in the third embodiment, the signal generators 132 ₁ and 132 ₂ are based on the phase processing signal MPC from the phase processing unit 131C and the reference signals BSC ₁ , BSC ₂ is generated.
BSC ₁ (t) = − C ₀ · sin [θ _M (t)]
= −C ₀ · sin [ω _SH t−2θ _C (t)]
= −C ₀ · sin [ω _SH t−2 (ω _C t + φ ₀ )] (16)
BSC ₂ (t) = C ₀ · cos [θ _M (t)]
= C ₀ · cos [ω _SH t−2θ _C (t)]
= C ₀ · cos [ω _SH t−2 (ω _C t + φ ₀ )] (17)

The reference signals BSC ₁ and BSC ₂ generated in this way are output toward the signal processing unit 140. Therefore, the signal processing unit 140 processes the signals OSC ₁ and OSC ₂ using the signals BSC ₁ and BSC ₂ , and generates a signal SCA similar to that in the first embodiment. As a result, as shown in FIG. 15, the stereo demodulation unit 550C outputs the same signals LDA and RDA as those of the stereo demodulation unit 550A of the first embodiment.

<Operation>
Next, the operation of FM receiving apparatus 500C configured as described above will be described mainly focusing on stereo demodulation processing.

  As a premise, as in the case of the first embodiment described above, in FM receiving apparatus 500C, the station desired to receive corresponding to the channel selection is selected according to the channel selection from operation input unit 570. To do. In FM receiving apparatus 500C, it is assumed that output volume etc. from speakers 580L and 580R are adjusted in accordance with designation of volume etc. from operation input unit 570.

  When broadcast waves from various broadcast stations are received by the antenna 510, the reception results are sequentially processed by the front end unit 530 and the detection unit 540, as in the first embodiment. Then, the detection signal DCD is sent to the stereo demodulation unit 550C.

In stereo demodulation section 550C that has received signal DCD, signal DCD is supplied to LPF 551M and subchannel processing section 552C. LPF551M which has received the signal DCD, as in the case of the first embodiment, selectively pass a main channel signal is a component of the angular frequency 0~ω _c (L + R), the addition unit 553L and subtraction as a signal MCA Send to part 553R.

On the other hand, in the subchannel processing unit 552C, the orthogonal signal generation unit 110C receives the signal DCD (see FIG. 16). In the orthogonal signal generation unit 110C that receives the supply of the signal DCD, first, the band limiting filter 111 selectively passes a signal component of a predetermined frequency band including the angular frequencies ω _{C to} 3ω _C in the signal DCD, Is sent to the orthogonalization unit 112C as a band-limited signal LSI with limited (see FIG. 17). As a result, even if the signal DCD has a signal component widely in an angular frequency region larger than the angular frequency 3ω _C as shown by a two-dot chain line in FIG. 21, as shown in FIG. The frequency band is limited.

Upon receiving the band limited signal LSI, the orthogonalizing unit 112C generates _two signals OSC ₁ and OSC ₂ that are orthogonal to each other based on the band limited signal LSI, and sends them to the FILs 113C ₁ and 113C _{2 and} to the signal processing unit 140. Send (see FIG. 17). Here, as shown in FIG. 23, each of the signals OSC _j (j = 1, 2) has a signal component PSM _j having an angular frequency (ω _SH −ω _C ) as a signal component corresponding to the pilot signal PS and The signal component PSP _j of the angular frequency (ω _SH + ω _C ) is included.

In the third embodiment, the orthogonalizing unit 112C performs frequency conversion of the band limited signal LSI. Therefore, the signal component PSM _j and the signal component PSP that can be generated when the frequency conversion of the signal DCD having a signal component widely in the angular frequency region larger than the angular frequency 3ω _C indicated by the two-dot chain line in FIG. 21 is performed. It is possible to prevent noise from being mixed into _j (see the alternate long and short dash line in FIG. 23).

Subsequently, FIL113C ₁ which has received the signal OSC ₁ is the component of the angular frequency of the signal _{OSC 1 (ω SH -ω C)} , i.e., selectively pass a signal component corresponding to the pilot signal PS, phase as the signal PSC ₁ Output to the calculation unit 120C. Further, FIL113C ₂ which has received the signal OSC ₂ is the component of the angular frequency of the signal _{OSC 2 (ω SH -ω C)} , i.e., selectively pass a signal component corresponding to the pilot signal PS, phase as the signal PSC ₂ It outputs towards the calculation part 120C (refer FIG. 17).

Here, the signal PSC ₁ has a waveform represented by the above-described equation (13). On the other hand, the signal PSC ₂ has a waveform represented by the above-described equation (14).

The phase calculation unit 120C that receives the signals PSC ₁ and PSC ₂ operates as described above to calculate a phase reflecting the pilot signal PS. The phase calculated in this way is sent to the reference signal generation unit 130C as a signal PHA (see FIG. 16).

The reference signal generator 130C that has received the signal PHA generates the reference signals BSC ₁ and BSC ₂ based on the signal PHA. In generating the reference signals BSC ₁ and BSC ₂ , in the reference signal generation unit 130C, first, the phase processing unit 131C calculates the phase θ _M (t) by the above-described equation (15), and the signal generation unit 132 _1. , 132 ₂ (see FIG. 20). Signal generating unit 132 _1, 132 ₂ having received the phase theta _M a (t) refers to the internal table on the basis of the phase theta _M (t), the above-described (16), represented by equation (17) The reference signals BSC ₁ and BSC ₂ which are signals are generated. The reference signals BSC ₁ and BSC ₂ generated in this way are sent to the signal processing unit 140 (see FIG. 20).

The signal processing unit 140 that receives the reference signals BSC ₁ and BSC ₂ from the reference signal generation unit 130C and the signals OSC ₁ and OSC ₂ from the orthogonal signal generation unit 110C uses the reference signals BSC ₁ and BSC _2. The signals OSC ₁ and OSC ₂ are processed to generate a signal SCP. The signal SCP generated in this way is sent to the LPF 551S.

Receiving the signal SCP from the subchannel processing unit 552C, the LPF 551S selectively passes the component of the angular frequency 0 to ω _c . As a result, the sub-channel signal (LR) selectively passes through the LPF 551S. The subchannel signal (LR) thus passed through the LPF 551S is output as the signal SCB to the adder 553L and the subtractor 553R.

  Thereafter, as in the case of the first embodiment, the adder 553L and the subtractor 553R generate the signal LDA related only to the L channel and the signal RDA related only to the R channel, and send them to the audio processor 560. (See FIG. 15).

  The audio processing unit 560 that receives the signals LDA and RDA from the stereo demodulation unit 550C converts each of the signals LDA and RDA into an analog signal in the same manner as in the first embodiment, and then converts the signals from the control unit 520A. The volume and the like are adjusted according to the audio processing command VLC. The results thus obtained are output to the L-channel speaker 580L and the R-channel speaker 580R as the L-channel audio signal LSA and the R-channel audio signal RSA (see FIG. 15).

  Upon receiving the audio signal LSA from the audio processing unit 560, the speaker 580L reproduces and outputs L channel audio in accordance with the audio signal LSA. Further, the speaker 580R that has received the audio signal RSA from the audio processing unit 560 reproduces and outputs the R channel audio in accordance with the audio signal RSA.

As described above, in the third embodiment, as in the case of the first embodiment, from the detection signal DCD including the pilot signal PS of the angular frequency ω _C , the feedback loop in the PLL system and the reference for phase measurement The phase reflecting the pilot signal PS is derived without using the base signal. Therefore, it is possible to generate the reference signals BSC ₁ and BSC ₂ for generating the baseband subchannel signal (LR) that is easily and quickly synchronized with the pilot signal PS. The sound contained in the waves can be played faithfully. As a result, it is possible to reproduce the sound included in the FM broadcast wave with good quality.

In the third embodiment, the phase θ reflecting the pilot signal PS is calculated in consideration of the phase shift Δθ in the FILs 113C ₁ and 113C ₂ , so that the phase can be calculated with high accuracy. Therefore, it is possible to generate the reference signals BSC ₁ and BSC ₂ that are accurately synchronized with the pilot signal PS.

  In the third embodiment, since orthogonalization is performed while performing frequency conversion in the orthogonalization unit 112C, the amount of calculation can be reduced compared to the case of orthogonalization using Hilbert transform or the like.

  In the third embodiment, since the band limited signal LSI obtained by band limiting the signal DCD by the band limiting filter 111 is orthogonalized while performing frequency conversion, noise mixing in the frequency conversion result of the pilot signal PS is reduced. Can be made.

[Fourth Embodiment]
Next, a fourth embodiment of the present invention will be described with reference mainly to FIGS.

  FIG. 24 is a block diagram showing a schematic configuration of an FM receiver 500D according to the fourth embodiment. As shown in FIG. 24, the FM receiver 500D includes a control unit 520D instead of the control unit 520A, and a stereo demodulator 550C compared to the FM receiver 500C of the third embodiment. Instead, only the stereo demodulation unit 550D is provided. Hereinafter, description will be made mainly focusing on these differences.

In addition to the function of the control unit 520A described above, the control unit 520D has a function of specifying the shift angular frequency ω _SHM for the stereo demodulation unit 550D. In order to implement such a shift angular frequency designation function, the control unit 520D receives the intermediate frequency signal IFD from the front end unit 530. Then, control unit 520D calculates shift angular frequency ω _SHM according to a predetermined algorithm based on the signal level of signal IFD and the level of a predetermined noise component in signal DCD that is the FM detection output. The shift angular frequency ω _SHM calculated in this way is output to the stereo demodulator 550D as a frequency conversion control signal FTC.

The stereo demodulator 550D differs from the stereo demodulator 550C described above only in that it includes a subchannel processor 552D instead of the subchannel processor 552C. As shown in FIG. 25, the subchannel processing unit 552D includes an orthogonal signal generation unit 110D instead of the orthogonal signal generation unit 110C, as compared with the above-described subchannel processing unit 552C, and is replaced with a phase calculation unit 120C. The difference is that a phase calculation unit 120D is provided and a reference signal generation unit 130D is provided instead of the reference signal generation unit 130C. Also, the subchannel processing unit 552D is different from the subchannel processing unit 552C in that it further includes high-pass filters (HPF) 115 ₁ and 115 ₂ as orthogonal band limiting means.

As shown in FIG. 26, the orthogonal signal generation unit 110D is provided with an orthogonalization unit 112D instead of the orthogonalization unit 112C and the FILs 113C ₁ and 113C ₂ as compared with the orthogonal signal generation unit 110C. The difference is that FILs 113D ₁ and 113D ₂ are provided.

  As shown in FIG. 27, the orthogonalizing unit 112D is different from the orthogonalizing unit 112C only in that an oscillating unit 210D is provided instead of the oscillating unit 210C.

The oscillator 210D generates a signal having an angular frequency ω _SHM specified by the signal FTC from the controller 520D, and thereby supplies the signal OTV ₁ to be supplied to the multiplier 220 ₁ and the multiplier 220 ₂ . Signal OTV ₂ is generated. In the fourth embodiment, the signal OTV ₁ and the signal OTV ₂ are represented by the following equations (18) and (19).
OTV ₁ (t) = B ₀ · cos (ω _SHM · t) (18)
OTV ₂ (t) = B ₀ · sin (ω _SHM · t) (19)
Here, B ₀ is a constant.

Returning to FIG. 26, each of the FIL113D _j (j = 1, 2) is configured as a digital filter such as an infinite impulse response filter (IIR). The FIL 113D _j receives the signal OSD _j from the orthogonalizing unit 112D and the signal FTC from the control unit 520D. The FIL 113D _j selectively passes one of the components of the angular frequency (ω _SHM −ω _C ) in the signal OSD _j , that is, the signal component corresponding to the pilot signal PS, to the phase calculation unit 120D as the signal PSD _j . Output toward.

Note that a fixed phase shift Δθ associated with the filter delay may occur through the FIL 113D _j . In the fourth embodiment, it is assumed that such a phase shift Δθ occurs. This phase shift Δθ is determined by the configuration of the FIL 113D _j and is determined at the design stage of the FIL 113D _j .

As a result, the signals PSD ₁ and PSD ₂ output from the orthogonal signal generation unit 110D are expressed by the following equations (20) and (21).
PSD ₁ (t) = − A (t) · sin [ω _SHM t− (ω _C t + φ ₀ ) −Δθ] (20)
PSD ₂ (t) = A (t) · cos [ω _SHM t− (ω _C t + φ ₀ ) −Δθ] (21)

Returning to FIG. 25, the phase calculation unit 120D generates the pilot signal PS based on the signals PSD ₁ and PSD ₂ from the quadrature signal generation unit 110D and the angular frequency ω _SHM specified by the signal FTC from the control unit 520D. Calculate the reflected phase. At the time of such calculation, the phase calculation unit 120D, for example, performs an arctan calculation on the signal PSD ₁ and the signal PSD ₂ and then considers the angular frequency ω _SH and the phase shift Δθ to reflect the phase of the pilot signal PS. θ _C (t) is calculated.

  The phase thus calculated is output as a signal PHA from the phase calculation unit 120D to the reference signal generation unit 130D.

The reference signal generation unit 130D generates reference signals BSD ₁ and BSD ₂ having angular frequencies (ω _SHM −2ω _C ) based on the signal PHA from the phase calculation unit 120D and the signal FTC from the control unit 520D. As shown in FIG. 28, the reference signal generation unit 130D having such a function is different from the above-described reference signal generation unit 130C only in that a phase processing unit 131D is provided instead of the phase processing unit 131C.

The phase processing unit 131D receives the phase θ _C (t) indicated by the signal PHA from the phase calculation unit 120D and the signal FTC from the control unit 520D. And the phase process part 131D calculates phase (theta) _M (t) by following (22) Formula.
θ _M (t) = ω _SHM t−2θ _C (t) = ω _SHM t−2 (ω _C t + φ ₀ ) (22)

That is, in the fourth embodiment, the phase processing unit 131D calculates the phase θ _M (t) that changes at the angular frequency (ω _SHM −2ω _C ). The phase θ _M (t) calculated in this way is output to the signal generators 132 ₁ and 132 ₂ as the phase processing signal MPD.

As a result, in the fourth embodiment, the signal generators 132 ₁ and 132 ₂ are based on the phase processing signal MPD from the phase processing unit 131D and the reference signals BSD ₁ , BSD ₂ is generated.
BSD ₁ (t) = − C ₀ · sin [θ _M (t)]
= −C ₀ · sin [ω _SHM t−2θ _C (t)]
= −C ₀ · sin [ω _SHM t−2 (ω _C t + φ ₀ )] (23)
BSD ₂ (t) = C ₀ · cos [θ _M (t)]
= C ₀ · cos [ω _SHM t−2θ _C (t)]
= C ₀ · cos [ω _SHM t−2 (ω _C t + φ ₀ )] (24)

The reference signals BSD ₁ and BSD ₂ generated in this way are output toward the signal processing unit 140.

Returning to FIG. 25, each of the HPFs 115 _j (j = 1, 2) is configured as a digital filter such as a finite impulse response filter (FIR) having no or very little phase shift (phase delay). Each HPF115 _j, has a filtering characteristic as shown by a chain line in FIG. 29, the intermediate frequency signal IFD signal level is is sufficient, and is employed by the control unit 520D in the case interference signal level is low When the shift angular frequency ω _SH1 is designated to the orthogonalizing unit 112D, the signal component in the frequency band related to the broadcast sound included in the signal OSD _j is passed through almost as it is.

On the other hand, each HPF 115 _j has a shift angular frequency ω _SH2 (<) that is adopted by the control unit 520D when the signal level of the intermediate frequency signal IFD is not sufficient or the interference signal level is also large, as shown in FIG. When ω _SH1 ) is designated for the orthogonalizing unit 112D, a part of the low frequency side signal component SG2M _j corresponding to the subchannel signal (L _M −R _M ) is cut off. Here, components that are blocked by HPF115 _j among signal components SG2M _j, at the time of reproduction to be a part of the high range, the filtering characteristics of HPF115 _j is defined.

As a result, when the shift frequency ω _SHM determined by the control unit 520 changes, the high cut characteristic during FM sound reproduction changes to D. Each filtering characteristic of HPF 115 _j is shown by a one-dot chain line in FIGS. 29 and 30, so that the amount of filtering calculation can be optimized even when the detection signal DCD has a high sample rate. It is like that.

Signals via the HPFs 115 ₁ and 115 ₂ configured as described above are output to the signal processing unit 140 as signals HCC ₁ and HCC ₂ . Therefore, the signal processing unit 140 processes the signals HCC ₁ and HCC ₂ using the signals BSD ₁ and BSD ₂ to generate the signal SCP. As a result, as shown in FIG. 24, signals LDD and RDD are output from the stereo demodulator 550D.

<Operation>
Next, the operation of FM receiving apparatus 500D configured as described above will be described mainly focusing on stereo demodulation processing.

  As a premise, similarly to the case of the third embodiment described above, in the FM receiver 500D, it is assumed that the reception desired station is selected according to the channel selection designation from the operation input unit 570. In FM receiving apparatus 500D, it is assumed that the output volume from speakers 580L and 580R is adjusted in accordance with the designation of the volume from operation input unit 570.

  When broadcast waves from various broadcast stations are received by the antenna 510, the reception result is sent to the front end unit 530 as a reception signal RFS. The front end unit 530 that has received the reception signal RFS extracts a signal corresponding to the desired station from the reception signal RFS, and sends it to the control unit 520D and the detection unit 540 as an intermediate frequency signal IFD of an intermediate frequency (see FIG. 24).

As described above, the control unit 520D that has received the intermediate frequency signal IFD calculates the shift frequency ω _SHM based on the signal level of the intermediate frequency signal IFD and the interference signal level, and uses the stereo demodulation unit 550D (more details) as the signal FTC. Are sent to the quadrature signal generator 110D, the phase calculator 120D, and the reference signal generator 130D (see FIG. 24).

  The detection unit 540 that has received the intermediate frequency signal IFD performs digital detection processing on the intermediate frequency signal IFD by a predetermined method to generate a signal DCD that is a composite signal, as in the third embodiment. The signal DCD generated in this way is sent to the stereo demodulator 550D.

Upon receiving the signal DCD, the stereo demodulation unit 550D supplies the signal DCD to the LPF 551M and the subchannel processing unit 552D. LPF551M which has received the signal DCD, as in the case of the third embodiment, selectively pass a main channel signal is a component of the angular frequency 0~ω _c (L + R), the addition unit 553L and subtraction as a signal MCA Send to part 553R.

On the other hand, in the subchannel processing unit 552D, the orthogonal signal generation unit 110D receives the signal DCD (see FIG. 25). In the orthogonal signal generation unit 110D that receives the supply of the signal DCD, first, as in the case of the third embodiment, the band limiting filter 111 performs a signal in a predetermined frequency band including the angular frequencies ω _{C to} 3ω _C in the signal DCD. The components are selectively passed and sent to the orthogonalization unit 112D as a band limited signal LSI with a limited frequency band (see FIG. 26).

The orthogonalizing unit 112D that has received the band limited signal LSI generates _two signals OSD ₁ and OSD ₂ that are orthogonal to each other based on the signal FTC and the band limited signal LSI received from the control unit 520D, and FILs 113D ₁ and 113D _2. To HPF 115 ₁ and 115 ₂ (see FIG. 26). The HPFs 115 ₁ and 115 ₂ that have received the signals OSD ₁ and OSD ₂ pass signal components in a predetermined frequency band in accordance with the filtering characteristics determined as described above, and the signals HCC ₁ and HCC ₂ are sent to the signal processing unit 140. Send (see FIG. 25).

Subsequently, FIL113D ₁ which has received the signal OSD ₁ is, component selectively passed through the angular frequency of the signal _{OSD 1 (ω SHM -ω C)} , is output to the signal PSD ₁ to phase calculation section 120D. Further, FIL113D ₂ which has received the signal OSD ₂ is the component of the angular frequency (ω _SHM -ω _C) in the signal OSD _2, i.e., selectively pass a signal component corresponding to the pilot signal PS, phase as the signal PSD ₂ It outputs toward calculation part 120D (refer FIG. 26).

Here, the signal PSD ₁ has a waveform represented by the above-described equation (20). On the other hand, the signal PSD ₂ has a waveform represented by the above-described equation (22).

The phase calculation unit 120D that has received the signals PSD ₁ and PSD ₂ operates as described above to calculate the phase θ _C (t) of the pilot signal PS. The phase θ _C (t) calculated in this way is sent to the reference signal generation unit 130D as a signal PHA (see FIG. 25).

The reference signal generator 130D that receives the signal PHA generates the reference signals BSD ₁ and BSD ₂ based on the signal PHA. In generating the reference signals BSD ₁ and BSD ₂ , in the reference signal generation unit 130D, first, the phase processing unit 131D calculates the phase θ _M (t) by the above-described equation (22), and the signal generation unit 132 _1. , 132 ₂ (see FIG. 28). Signal generating unit 132 _1, 132 ₂ having received the phase theta _M a (t) refers to the internal table on the basis of the phase theta _M (t), the above-described (23), represented by equation (24) The reference signals BSD ₁ and BSD ₂ which are signals are generated. The reference signals BSD ₁ and BSD ₂ generated in this way are sent to the signal processing unit 140 (see FIG. 28).

The signal processing unit 140 that receives the reference signals BSD ₁ and BSD ₂ from the reference signal generation unit 130D and the signals HCC ₁ and HCC ₂ from the HPFs 115 ₁ and 115 ₂ uses the reference signals BSD ₁ and BSD _2. The signals HCC ₁ and HCC ₂ are processed to generate a signal SCP. The signal SCP generated in this way is sent to the LPF 551S.

LPF551S which receives the signal SCP from subchannel processor 552D selectively passes the component of the angular frequency 0~ω _c. As a result, the sub-channel signal (LR) selectively passes through the LPF 551S. Thus, the subchannel signal (LR) that has passed through the LPF 551S is output as the signal SCD to the adding unit 553L and the subtracting unit 553R.

  Thereafter, as in the case of the third embodiment, the adder 553L and the subtractor 553R generate the signal RDD related only to the LDD and R channel by the adder 553L and the subtractor 553R, and send it to the audio processor 560. (See FIG. 24).

  The audio processing unit 560 that has received the signals LDD and RDD from the stereo demodulation unit 550D converts each of the signals LDD and RDD into an analog signal in the same manner as in the first embodiment, and then converts the signals from the control unit 520D. The volume and the like are adjusted according to the audio processing command VLC. The results thus obtained are output to the L channel speaker 580L and the R channel speaker 580R as an L channel audio signal LSD and an R channel audio signal RSD (see FIG. 24).

  Upon receiving the audio signal LSD from the audio processing unit 560, the speaker 580L reproduces and outputs L channel audio in accordance with the audio signal LSD. The speaker 580R that has received the audio signal RSD from the audio processing unit 560 reproduces and outputs the R channel audio in accordance with the audio signal RSD.

As described above, in the fourth embodiment, as in the case of the third embodiment, from the detection signal DCD including the pilot signal PS of the angular frequency ω _C , the feedback loop in the PLL method, the reference for phase measurement, The phase reflecting the pilot signal PS is derived without using the base signal. Therefore, it is possible to generate the reference signals BSD ₁ and BSD ₂ for generating the baseband subchannel signal (LR) that is easily and quickly synchronized with the pilot signal PS. The sound contained in the waves can be played faithfully. As a result, it is possible to reproduce the sound included in the FM broadcast wave with good quality.

Further, in the fourth embodiment, since the phase that reflects the pilot signal PS is calculated in consideration of the phase shift Δθ in the FILs 113D ₁ and 113D _2, the phase can be calculated with high accuracy. Therefore, it is possible to generate the reference signals BSD ₁ and BSD ₂ that are accurately synchronized with the pilot signal PS.

  Further, in the fourth embodiment, since orthogonalization is performed while performing frequency conversion in the orthogonalization unit 112D, the amount of calculation can be reduced compared to the case of orthogonalization using Hilbert transform or the like.

  In the fourth embodiment, since the band limited signal LSI obtained by band limiting the signal DCD by the band limiting filter 111 is orthogonalized while performing frequency conversion, noise contamination in the frequency conversion result of the pilot signal PS is reduced. Can be made.

In the fourth embodiment, the high-cut processing is performed using HPF 115 ₁ and 115 ₂ on the signal obtained by performing frequency conversion on the band limited signal LSI obtained by band limiting the signal DCD. Simplification in designing the FILs 113D ₁ and 113D ₂ can be achieved.

[Modification of Embodiment]
The present invention is not limited to the first to fourth embodiments described above, and various modifications can be made.

For example, in the first to fourth embodiments described above, for performing processing sub-channel signal (L _M -R _M), it has been described with reference to the case of obtaining a baseband auxiliary channel signal (L-R). On the other hand, when the broadcast wave further includes an RDS (Radio Data System) signal or the like, the same processing as in the case of the subchannel signal (L _M -R _M ) is performed to convert the RDS signal into the baseband band. It can also be a signal.

In the third and fourth embodiments, the signals having the angular frequencies (ω _SH −ω _C ) and (ω _SHM −ω _C ) are extracted from the signals corresponding to the pilot signal PS after the frequency conversion. However, the signals having the angular frequencies (ω _SH + ω _C ) and (ω _SHM + ω _C ) may be extracted and used.

In the fourth embodiment, the band for performing the high cut is adjusted by changing the shift frequency for performing the frequency conversion on the subchannel signal. On the other hand, the band for performing the high cut may be adjusted by making the pass bands of the FILs 115 ₁ and 115 ₂ variable.

  Moreover, the deformation | transformation to 2nd Embodiment with respect to 1st Embodiment can also be applied to 3rd or 4th Embodiment.

  The control unit, the detection unit, and the stereo demodulation unit in the FM receivers of the first to fourth embodiments are configured by a computer including a CPU (Central Processing Unit) and a DSP (Digital Signal Processor), and the program in the computer By execution, the functions of the FM receivers of the first to fourth embodiments can be realized. These programs may be acquired in the form recorded on a portable recording medium such as a CD-ROM or DVD, or may be acquired in the form of delivery via a network such as the Internet. Good.

It is a block diagram which shows the schematic structure of the FM receiver which concerns on 1st Embodiment of this invention. It is a figure for demonstrating the frequency distribution of the detection signal assumed in 1st Embodiment. It is a block diagram which shows the structure of the stereo demodulation part in the apparatus of FIG. It is a block diagram which shows the structure of the subchannel process part in FIG. It is a block diagram which shows the structure of the orthogonal signal generation part in FIG. It is a block diagram which shows the structure of the orthogonalization part in FIG. FIG. 5 is a block diagram illustrating a configuration of a reference signal generation unit in FIG. 4. It is a block diagram which shows the structure of the signal processing part in the apparatus of FIG. It is a block diagram which shows the structure of the process part of FIG. It is a figure for demonstrating the frequency distribution of the process result signal by the signal processing part in 1st Embodiment. It is a block diagram which shows schematic structure of the FM receiver which concerns on 2nd Embodiment of this invention. It is a block diagram which shows the structure of the subchannel process part in the apparatus of FIG. It is a block diagram which shows the structure of the reference signal production | generation part in FIG. It is a figure for demonstrating the frequency distribution of the processing result signal by the signal processing part in 2nd Embodiment. It is a block diagram which shows schematic structure of the FM receiver which concerns on 3rd Embodiment of this invention. It is a block diagram which shows the structure of the subchannel process part in the apparatus of FIG. It is a block diagram which shows the structure of the orthogonal signal generation part in FIG. It is a figure for demonstrating the characteristic of a band-limiting filter. It is a block diagram which shows the structure of the orthogonalization part in FIG. It is a block diagram which shows the structure of the reference signal production | generation part in FIG. It is a figure for demonstrating the frequency distribution of the detection signal assumed in 3rd Embodiment. It is a figure for demonstrating the frequency distribution of the band limitation signal output from a band limitation filter. It is a figure for showing the frequency conversion result by the orthogonalization part of FIG. It is a block diagram which shows schematic structure of the FM receiver which concerns on 4th Embodiment of this invention. It is a block diagram which shows the structure of the subchannel process part in the apparatus of FIG. It is a block diagram which shows the structure of the orthogonal signal generation part in FIG. It is a block diagram which shows the structure of the orthogonalization part in FIG. FIG. 26 is a block diagram illustrating a configuration of a reference signal generation unit in FIG. 25. It is FIG. (1) for demonstrating the characteristic and effect | action of HPF in FIG. It is FIG. (2) for demonstrating the characteristic and effect | action of HPF in FIG.

Explanation of symbols

110A to 110D: Orthogonal signal generator (orthogonal signal generator)
111 ... Band limiting filter (composite band limiting means)
112A to 112D ... Orthogonalizing unit (orthogonalizing means)
113A ₁ , 113A ₂ ... Filter (filter means)
113C ₁ , 113C ₂ ... Filter (filter means)
113D ₁ , 113D ₂ ... Filter (filter means)
115 ₁ , 115 ₂ ... High-pass filter (orthogonalization band limiting means)
120A to 120D ... Phase calculation unit (phase calculation means)
130A to 130D ... reference signal generation unit (reference signal generation means)
140 ... Signal processing unit (baseband conversion means)
500A-500D ... FM receiver

Claims (9)

Orthogonal signal generation means for generating a first signal and a second signal that are orthogonalized to each other by reflecting the phase of the pilot signal in the composite signal with respect to the composite signal obtained by FM detection of the FM broadcast wave;
Phase calculating means for calculating a phase reflecting the pilot signal based on the first signal and the second signal;
Reference signal generating means for generating a reference signal having a predetermined relationship with the pilot signal based on a calculation result by the phase calculating means;
A signal corresponding to a sub-channel signal that is a signal in a frequency band higher than the frequency of the pilot signal in the composite signal is a base that is a signal in a frequency band lower than the frequency of the pilot signal using the reference signal baseband conversion means for converting the band subchannel signal; equipped with,
The orthogonal signal generating means includes
Orthogonalizing means for orthogonalizing the composite signal to generate a first orthogonalized signal and a second orthogonalized signal orthogonal to each other;
Filter means for selectively passing the first signal included in the first orthogonalized signal and the second signal included in the second orthogonalized signal;
The orthogonal signal generation means performs a frequency shift in addition to the orthogonalization.
An FM receiver characterized by that.   Stereo demodulation means for performing stereo demodulation based on a main channel signal that is a signal in a frequency band lower than the frequency of the pilot signal in the composite signal and the baseband subchannel signal is further provided. The FM receiver according to claim 1.   3. The FM receiver according to claim 1, wherein the reference signal is a signal whose phase is further shifted by a predetermined fixed phase value. 4. The phase calculation means calculates the phase of the pilot signal in consideration of the frequency shift amount due to the frequency shift.
The FM receiver as described in any one of Claims 1-3 characterized by the above-mentioned. 5. The FM according to claim 1 , further comprising a composite band limiting unit that generates a band limited signal that limits a frequency band of the composite signal and supplies the band limited signal to the orthogonalizing unit. Receiver device. The orthogonal signal generating means further comprises orthogonal band limiting means for limiting signal bands in the first orthogonal signal and the second orthogonal signal,
The baseband conversion means converts the signal that has passed through the orthogonalization band limiting means to a signal having a frequency band lower than the frequency of the pilot signal, using the reference signal.
The FM receiver as described in any one of Claims 1-5 characterized by the above-mentioned. An orthogonal signal generation step of generating a first signal and a second signal that are orthogonalized to each other by reflecting the phase of the pilot signal in the composite signal with respect to the composite signal obtained by FM detection of the FM broadcast wave;
A phase calculating step of calculating a phase reflecting the pilot signal based on the first signal and the second signal;
A reference signal generating step for generating a reference signal having a predetermined relationship with the pilot signal based on a calculation result in the phase calculating step;
A signal corresponding to a sub-channel signal that is a signal in a frequency band higher than the frequency of the pilot signal in the composite signal is a base that is a signal in a frequency band lower than the frequency of the pilot signal using the reference signal baseband conversion step of converting the band subchannel signal; equipped with,
The orthogonal signal generation step includes:
An orthogonalization step of orthogonalizing the composite signal to generate a first orthogonalized signal and a second orthogonalized signal orthogonal to each other;
A filtering step of selectively passing the first signal included in the first orthogonalized signal and the second signal included in the second orthogonalized signal;
In the orthogonal signal generation step, in addition to the orthogonalization, a frequency shift is performed.
An FM reception processing method. An FM reception processing program for causing an arithmetic means to execute the FM reception processing method according to claim 7 . 9. A recording medium, wherein the FM reception processing program according to claim 8 is recorded so as to be readable by an arithmetic means.

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