JP4798639B2

JP4798639B2 - Induction motor control device, control method, and vehicle using the control device

Info

Publication number: JP4798639B2
Application number: JP2009062491A
Authority: JP
Inventors: 郁也刀谷
Original assignee: 日本輸送機株式会社
Priority date: 2009-03-16
Filing date: 2009-03-16
Publication date: 2011-10-19
Anticipated expiration: 2029-03-16
Also published as: JP2010220331A

Description

本発明は、誘導電動機の制御装置、制御方法および該制御装置を用いた車両、特に、誘導電動機の低回転時のトルクアップのための高効率な制御装置、制御方法および該制御装置を用いた例えばフォークリフトなどの電気車両に関するものである。 The present invention relates to an induction motor control device, a control method, and a vehicle using the control device, in particular, a high-efficiency control device, a control method, and the control device for increasing torque during low rotation of the induction motor. For example, it relates to an electric vehicle such as a forklift.

従来から、誘導電動機の制御装置として、例えば図４に示すように、誘導電動機に流れる１次電流の２つのベクトル成分のうち、励磁電流指令値およびトルク電流指令値に基づいて、当該誘導電動機の滑り角周波数を演算する滑り周波数制御型ベクトル制御方法を用いた誘導電動機の制御装置が知られている（例えば、非特許文献１参照）。 Conventionally, as a control device for an induction motor, for example, as shown in FIG. 4, based on an excitation current command value and a torque current command value among two vector components of a primary current flowing in the induction motor, A control device for an induction motor using a slip frequency control type vector control method for calculating a slip angular frequency is known (for example, see Non-Patent Document 1).

図４を参照して、この誘導電動機２７の制御装置２０は、外部入力された励磁電流指令値ｉ_１ｄ ^＊およびトルク電流指令値ｉ_１q ^＊に基づいて滑り角周波数ω_ｓを演算するとともに、当該誘導電動機２７に供給される１次電流および当該誘導電動機から検出される角周波数ω_ｒのフィードバックを受けながら、当該誘導電動機２７の駆動を制御するものである。
この制御装置２０において、電流制御器２４は、励磁電流成分ｉ_１ｄおよびトルク電流成分ｉ_１ｑをそれぞれ外部入力された励磁電流指令値ｉ_１ｄ ^＊およびトルク電流指令値ｉ_１q ^＊に基づいて制御して、ｄ軸電圧指令値ｖ_１ｄ ^＊およびｑ軸電圧指令値ｖ_１ｑ ^＊を出力する。ここで、励磁電流成分ｉ_１ｄおよびトルク電流成分ｉ_１ｑは、誘導電動機２７に供給される１次電流が１次電流検出手段２９によって検出され、さらに３相−２相変換器３０（ｄｑ−αβ座標変換器３０ａおよびαβ−３相座標変換器３０ｂ）によって変換されることにより得られる。
滑り角周波数演算手段２１は、励磁電流指令値ｉ_１ｄ ^＊およびトルク電流指令値ｉ_１q ^＊を用いて、誘導電動機２７の滑り角周波数ω_ｓを、次式、

によって演算する。
１次角周波数演算器２２は、滑り角周波数演算手段２１から出力される滑り角周波数ω_ｓと誘導電動機２７から検出される角周波数ω_ｒを加算して、１次角周波数ω_１＝ω_ｓ＋ω_ｒを出力する。
積分器２３は、１次角周波数演算器２２から出力された１次角周波数ω_１を、次式、

によって時間積分して、位相θ_１を出力する。
２相−３相変換器２５（ｄｑ−αβ座標変換器２５ａおよびαβ−３相座標変換器２５ｂ）は、ｄ軸電圧指令値ｖ_１ｄ ^＊およびｑ軸電圧指令値ｖ_１ｑ ^＊を位相θ_１により座標変換して１次電圧指令値ｖ_１ｕ ^＊、ｖ_１ｖ ^＊、ｖ_１ｗ ^＊を出力する。
インバータ２６は、この１次電圧指令値ｖ_１ｕ ^＊、ｖ_１ｖ ^＊、ｖ_１ｗ ^＊に基づいて誘導電動機２７に駆動電力を供給して当該誘導電動機２７を駆動する。 Referring to FIG. 4, control device 20 of induction motor 27 calculates slip angular frequency ω _s based on excitation current command value i _1d ^* and torque current command value i _1q ^* inputted externally, and while receiving feedback of the detected angular frequency omega _r from the primary current and the induction motor is supplied to the induction motor 27, and controls the driving of the induction motor 27.
In the control device 20, the current controller 24 controls the excitation current component i _1d and the torque current component i _{1q based} on the excitation current command value i _1d ^* and the torque current command value i _1q ^* which are externally input, respectively. , D-axis voltage command value v _1d ^* and q-axis voltage command value v _1q ^* are output. Here, as for the excitation current component i _1d and the torque current component i _1q , the primary current supplied to the induction motor 27 is detected by the primary current detection means 29, and the three-phase to two-phase converter 30 (dq-αβ It is obtained by being converted by the coordinate converter 30a and the αβ-3 phase coordinate converter 30b).
The slip angle frequency calculating means 21 uses the excitation current command value i _1d ^* and the torque current command value i _1q ^* to calculate the slip angle frequency ω _s of the induction motor 27 by the following equation:

Calculate by
The primary angular frequency computing unit 22 adds the slip angular frequency ω _s output from the slip angular frequency computing means 21 and the angular frequency ω _r detected from the induction motor 27 to obtain a primary angular frequency ω ₁ = ω _s. and it outputs a + ω _r.
The integrator 23 converts the primary angular frequency ω ₁ output from the primary angular frequency calculator 22 into the following formula:

And integration time by outputting a phase theta _1.
The two-phase to three-phase converter 25 (dq-αβ coordinate converter 25a and αβ-3 phase coordinate converter 25b) converts the d-axis voltage command value v _1d ^* and the q-axis voltage command value v _1q ^{* according} to the phase θ _1. coordinate transformation to the primary voltage command value _{^{_{^{_{v 1u *, v 1v *,}}}}} v and outputs a _{1 w} ^*.
The inverter 26 supplies drive power to the induction motor 27 based on the primary voltage command values v _1u ^* , v _1v ^* , v _1w ^* to drive the induction motor 27.

ここで、この誘導電動機２７のトルクτおよび２次側磁束φ_２は、それぞれ、次式、

および、

により与えられる。（３）式および（４）式より、励磁電流指令値ｉ_１ｄ ^＊が一定の場合には誘導電動機２７のトルクτがトルク電流指令値ｉ_１q ^＊に比例し、当該トルクτが当該トルク電流指令値ｉ_１q ^＊によって定まることから、この制御装置２０では、励磁電流指令値ｉ_１ｄ ^＊を一定値に制限していた。言い換えれば、図５（ａ）に示すように、励磁電流指令値ｉ_１ｄ ^＊が２次側磁束φ_２の磁束飽和する励磁電流指令値（以下、飽和電流制限閾値ｉ_１ｄｔｈ ^＊という）を超える非線形領域の範囲（領域Ｂ）にある場合、実際の２次側磁束φ_２は、（４）式に基づいて計算した２次側磁束φ_２より小さくなり、（４）式の下では正確な２次側磁束φ_２が算出されない。その結果、トルクτも（３）式の下では正確な値が算出されない。したがって、（３）式および（４）式の下で正確な２次側磁束φ_２およびトルクτを算出するために、励磁電流指令値ｉ_１ｄ ^＊を、飽和電流制限閾値ｉ_１ｄｔｈ ^＊以下の線形領域の範囲（領域Ａ）に制限していた。なお、この場合、励磁電流指令値ｉ_１ｄ ^＊および滑り角周波数ω_ｓは、それぞれ図５（ｂ）および（ｃ）に示すように、回転数（回転速度に）応じて制御される。 Here, the torque τ and the secondary magnetic flux φ ₂ of the induction motor 27 are respectively expressed by the following equations:

and,

Given by. From the equations (3) and (4), when the excitation current command value i _1d ^* is constant, the torque τ of the induction motor 27 is proportional to the torque current command value i _1q ^* , and the torque τ is related to the torque current command. Since it is determined by the value i _1q ^* , the control device 20 limits the excitation current command value i _1d ^* to a constant value. In other words, as shown in FIG. 5 (a), the excitation current command value i _1d ^* exceeds the excitation current command value that saturates the magnetic flux of the secondary side magnetic flux φ ₂ (hereinafter referred to as saturation current limit threshold i _1dth ^* ). In the region range (region B), the actual secondary magnetic flux φ ₂ is smaller than the secondary magnetic flux φ ₂ calculated based on the equation (4). next side magnetic flux phi ₂ is not calculated. As a result, an accurate value of the torque τ is not calculated under the expression (3). Therefore, in order to calculate the accurate secondary-side magnetic flux φ ₂ and torque τ under the equations (3) and (4), the excitation current command value i _1d ^* is linearly equal to or less than the saturation current limit threshold i _1dth ^*. It was limited to the area range (area A). In this case, the excitation current command value i _1d ^* and the slip angular frequency ω _s are controlled according to the number of rotations (according to the rotation speed), as shown in FIGS. 5B and 5C, respectively.

この制御装置２０では、励磁電流指令値ｉ_１ｄ ^＊が、飽和電流制限閾値ｉ_１ｄｔｈ ^＊以下の値に制限され、飽和電流制限閾値ｉ_１ｄｔｈ ^＊を超える非線形領域の範囲で使用されていないので、誘導電動機の低回転時における高トルクを実現したい場合に、効率の面で最良の制御が実施されていなかった。 In the control device 20, the exciting current command value _{i 1d} ^* is limited to the saturation current limit threshold _{i 1dth} ^* The following values, because they are not used in the range of non-linear region which exceeds the saturation current limit threshold _{i 1dth} ^*, induced When it is desired to achieve a high torque at a low rotation of the electric motor, the best control in terms of efficiency has not been implemented.

「交流電動機可変速駆動の基礎と応用」、電気学会交流電動機駆動方式の技術分類・用語整理調査専門委員会編、１９９８年１０月２８日、Ｐ．８７の図４．１３“Basics and Applications of AC Motor Variable Speed Drives”, IEEJ Technical Class / Terminology Organizing Survey Committee, October 28, 1998, p. 87 Figure 4.13

本発明は、誘導電動機の低回転時のトルクアップのための高効率な制御装置、制御方法および該制御装置を用いた車両を提供する。 The present invention provides a highly efficient control device, a control method, and a vehicle using the control device for increasing torque during low rotation of an induction motor.

上記課題を解決するため、本発明の誘導電動機の制御装置は、誘導電動機の１次電流の２つのベクトル成分のうち、励磁電流成分ｉ_１ｄおよびトルク電流成分ｉ_１ｑをそれぞれ外部入力された励磁電流指令値ｉ_１ｄ ^＊およびトルク電流指令値ｉ_１q ^＊に基づいて制御して、ｄ軸電圧指令値ｖ_１ｄ ^＊およびｑ軸電圧指令値ｖ_１ｑ ^＊を出力する電流制御器と、前記励磁電流指令値ｉ_１ｄ ^＊または前記トルク電流指令値ｉ_１q ^＊のうちの少なくとも１つを用いて、前記誘導電動機の滑り角周波数ω_ｓを演算する滑り角周波数演算手段と、前記滑り角周波数演算手段から出力される前記滑り角周波数ω_ｓと前記誘導電動機から検出される角周波数ω_ｒを加算して１次角周波数ω_１を出力する１次角周波数演算器と、前記１次角周波数演算器から出力された前記１次角周波数ω_１を積分して位相θ_１を出力する積分器と、前記ｄ軸電圧指令値ｖ_１ｄ ^＊および前記ｑ軸電圧指令値ｖ_１ｑ ^＊を前記位相θ_１により座標変換して１次電圧指令値ｖ_１ｕ ^＊、ｖ_１ｖ ^＊、ｖ_１ｗ ^＊を出力する２相−３相変換器と、前記１次電圧指令値ｖ_１ｕ ^＊、ｖ_１ｖ ^＊、ｖ_１ｗ ^＊に基づいて前記誘導電動機に駆動電力を供給して当該誘導電動機を駆動するインバータと、を備えた誘導電動機の制御装置であって、前記励磁電流指令値ｉ_１ｄ ^＊を、予め設定された飽和電流制限閾値ｉ_１ｄｔｈ ^＊と比較して、第１の励磁電流指令値ｉ_１ｄ１ ^＊と第２の励磁電流指令値ｉ_１ｄ２ ^＊とに分け、前記励磁電流指令値ｉ_１ｄ ^＊が前記飽和電流制限閾値ｉ_１ｄｔｈ ^＊以下である場合には、前記第１の励磁電流指令値ｉ_１ｄ１ ^＊として当該励磁電流指令値ｉ_１ｄ ^＊を、前記第２の励磁電流指令値ｉ_１ｄ２ ^＊として０を出力する一方で、前記励磁電流指令値ｉ_１ｄ ^＊が前記飽和電流制限閾値ｉ_１ｄｔｈ ^＊を超える場合には、前記第１の励磁電流指令値ｉ_１ｄ１ ^＊として前記飽和電流制限閾値ｉ_１ｄｔｈ ^＊を、前記第２の励磁電流指令値ｉ_１ｄ２ ^＊として前記励磁電流指令値ｉ_１ｄ ^＊と前記飽和電流制限閾値ｉ_１ｄｔｈ ^＊との差分値（ｉ_１ｄ ^＊−ｉ_１ｄｔｈ ^＊）を出力する励磁電流指令値制御手段と、を備え、前記滑り角周波数演算手段は、前記第１の励磁電流指令値ｉ_１ｄ１ ^＊および前記トルク電流指令値ｉ_１ｑ ^＊を用いて第１の滑り角周波数ω_ｓ１を演算する第１の滑り角周波数演算手段と、前記第２の励磁電流指令値ｉ_１ｄ２ ^＊を用いて第２の滑り角周波数ω_ｓ２を演算する第２の滑り角周波数演算手段と、前記第１の滑り角周波数演算手段から出力される前記第１の滑り角周波数ω_ｓ１と前記第２の滑り角周波数演算手段から出力される前記第２の滑り角周波数ω_ｓ２を加算して前記滑り角周波数ω_ｓを出力する滑り角周波数加算器と、からなり、前記１次角周波数演算器は、前記滑り角周波数加算器から出力される前記滑り角周波数ω_ｓと前記誘導電動機から検出される前記角周波数ω_ｒとを加算して前記１次角周波数ω_１を出力する。
この発明の誘導電動機の制御装置によれば、励磁電流指令値制御手段を備え、滑り角周波数演算手段を、第１の滑り角周波数演算手段と、第２の滑り角周波数演算手段と、滑り角周波数加算器とから構成する。そして、励磁電流指令値制御手段によって、励磁電流指令値ｉ_１ｄ ^＊が、予め設定された飽和電流制限閾値ｉ_１ｄｔｈ ^＊と比較され、励磁電流指令値ｉ_１ｄ ^＊が飽和電流制限閾値ｉ_１ｄｔｈ ^＊以下である場合には、第１の励磁電流指令値ｉ_１ｄ１ ^＊として当該励磁電流指令値ｉ_１ｄ ^＊が出力されて、第２の励磁電流指令値ｉ_１ｄ２ ^＊として０が出力される。一方で、励磁電流指令値ｉ_１ｄ ^＊が飽和電流制限閾値ｉ_１ｄｔｈ ^＊を超える場合には、第１の励磁電流指令値ｉ_１ｄ１ ^＊として飽和電流制限閾値ｉ_１ｄｔｈ ^＊が、第２の励磁電流指令値ｉ_１ｄ２ ^＊として励磁電流指令値ｉ_１ｄ ^＊と飽和電流制限閾値ｉ_１ｄｔｈ ^＊の差分値（ｉ_１ｄ ^＊−ｉ_１ｄｔｈ ^＊）が出力される。さらに、誘導電動機に必要な滑り角周波数ω_ｓが、第１の励磁電流指令値ｉ_１ｄ１ ^＊、第２の励磁電流指令値ｉ_１ｄ２ ^＊およびトルク電流指令値ｉ_１q ^＊を用いて、演算されるようにしたので、励磁電流指令値ｉ_１ｄ ^＊が飽和電流制限閾値ｉ_１ｄｔｈ ^＊を越える場合においても誘導電動機の制御が行え、誘導電動機の制御の効率化を高めることができ、誘導電動機の低回転時における高トルクを実現できる。 In order to solve the above problems, an induction motor control apparatus according to the present invention includes an excitation current component i _1d and a torque current component i _1q that are externally input from two vector components of a primary current of the induction motor. A current controller that controls the command value i _1d ^* and the torque current command value i _1q ^* to output the d-axis voltage command value v _1d ^* and the q-axis voltage command value v _1q ^* , and the excitation current command value The slip angle frequency calculating means for calculating the slip angle frequency ω _s of the induction motor using at least one of i _1d ^* or the torque current command value i _1q ^* and the slip angle frequency calculating means A primary angular frequency calculator that outputs a primary angular frequency ω ₁ by adding the slip angular frequency ω _s and the angular frequency ω _r detected from the induction motor, and the primary angular frequency calculator An integrator for outputting a phase theta ₁ by integrating the primary angular frequency omega ₁ outputted coordinate the d-axis voltage command value v _1d ^* and the q-axis voltage command value v _1q ^* by the phase theta ₁ converted to the primary voltage command value _{^{_{^{_{v 1u *, v 1v *,}}}}} v 1w * and 2-phase three-phase converter for outputting said primary voltage command value _^v ^1u _{_*,} _v 1v _*, based on _{v 1 w} ^* And an inverter for driving the induction motor by supplying drive power to the induction motor, wherein the excitation current command value i _1d ^* is set to a preset saturation current limit threshold value. Compared to i _1dth ^* , the excitation current command value i _1d2 ^* is divided into a first excitation current command value i _1d1 ^* and a second excitation current command value i _1d2 ^*, and the excitation current command value i _1d ^* is the saturation current limit threshold i _1dth ^*. If the following is true: The exciting current command value _{i 1d} ^* as the current command value _{i 1d1} ^*, the second excitation current command value _{i 1d2} ^* as while outputs 0, the excitation current command value _{i 1d} ^* is the saturation current limit threshold If more than _i 1dth ^*, the first exciting current command value _{i 1d1} ^* the saturation current limit threshold _{i 1dth} ^* as the excitation current command value _{i 1d} as the second excitation current command value _{i 1d2} ^* Excitation current command value control means for outputting a difference value (i _1d ^* −i _1dth ^* ) between ^* and the saturation current limit threshold value i _1dth ^*, and the slip angular frequency calculation means includes the first excitation frequency calculation means. a first slip angular frequency calculation means for calculating a first slip angular frequency omega _s1 using a current command value i _1d1 ^* and the torque current command value i _1q ^*, said second excitation current command value i _d2 ^* and a second slip angular frequency calculation means for calculating a second slip angular frequency omega _s2 with, said first slip angular frequency of the first slip angular frequency omega _s1 outputted from the arithmetic means and the A slip angular frequency adder for adding the second slip angular frequency ω _s2 output from the second slip angular frequency calculating means and outputting the slip angular frequency ω _s, and calculating the primary angular frequency calculation The sum adds the slip angular frequency ω _s output from the slip angular frequency adder and the angular frequency ω _r detected from the induction motor, and outputs the primary angular frequency ω ₁ .
According to the control apparatus for an induction motor of the present invention, the excitation current command value control means is provided, and the slip angle frequency calculation means includes a first slip angle frequency calculation means, a second slip angle frequency calculation means, and a slip angle. And a frequency adder. Then, the excitation current command value control means compares the excitation current command value i _1d ^* with a preset saturation current limit threshold i _1dth ^*, and the excitation current command value i _1d ^* is equal to or less than the saturation current limit threshold i _1dth ^*. If it is the first of the excitation current command value _{i 1d} ^* is output as the excitation current command value _{i 1d1} ^*, the second exciting current command value _{i 1d2} ^* 0 is output. On the other hand, when the excitation current command value i _1d ^* exceeds the saturation current limit threshold value i _1dth ^* , the saturation current limit threshold value i _1dth ^* becomes the second excitation current command value as the first excitation current command value i _1d1 ^*. the value _{i 1d2} ^* as an excitation current command value _{i 1d} ^* and the saturation current limit threshold _{i 1dth} ^* difference value _{^_(i} 1d _{^* -i} ^1dth _*) is outputted. Further, the slip angular frequency ω _s required for the induction motor is calculated using the first excitation current command value i _1d1 ^* , the second excitation current command value i _1d2 ^*, and the torque current command value i _1q ^*. As a result, the induction motor can be controlled even when the excitation current command value i _1d ^* exceeds the saturation current limit threshold i _1dth ^*, and the efficiency of the induction motor control can be improved. High torque at the time can be realized.

また、上記構成において、前記第１の滑り角周波数演算手段は、前記第１の励磁電流指令値ｉ_１ｄ１ ^＊および前記トルク電流指令値ｉ_１ｑ ^＊を用いて第１の滑り角周波数ω_ｓ１を、次式、

によって演算し、前記第２の滑り角周波数演算手段は、前記第２の励磁電流指令値ｉ_１ｄ２ ^＊を用いて第２の滑り角周波数ω_ｓ２を、次式、

によって演算することが好ましい。なお、（６）式は、第２の滑り角周波数ω_ｓ２が第２の励磁電流指令値ｉ_１ｄ２ ^＊を用いて線形近似されることに基づく。
この構成によれば、誘導電動機に必要な滑り角周波数ω_ｓ＝ω_ｓ１＋ω_ｓ２が、第１の励磁電流指令値ｉ_１ｄ１ ^＊、第２の励磁電流指令値ｉ_１ｄ２ ^＊およびトルク電流指令値ｉ_１q ^＊を用いて、容易に演算されるようにしたので、励磁電流指令値ｉ_１ｄ ^＊が飽和電流制限閾値ｉ_１ｄｔｈ ^＊を越える場合においても、誘導電動機の制御を容易に効率よく行える。 Further, in the above configuration, the first slip angular frequency calculating means calculates the first slip angular frequency ω _s1 using the first excitation current command value i _1d1 ^* and the torque current command value i _1q ^* , The following formula,

The second slip angular frequency calculating means calculates the second slip angular frequency ω _s2 using the second excitation current command value i _1d2 ^* by the following formula:

It is preferable to calculate by: The expression (6) is based on the fact that the second slip angular frequency ω _s2 is linearly approximated by using the second excitation current command value i _1d2 ^* .
According to this configuration, the slip angular frequency ω _s = ω _s1 + ω _s2 necessary for the induction motor is determined by the first excitation current command value i _1d1 ^* , the second excitation current command value i _1d2 ^*, and the torque current command value i. _{Since 1q} ^* is used for easy calculation, the induction motor can be controlled easily and efficiently even when the excitation current command value i _1d ^* exceeds the saturation current limit threshold i _1dth ^* .

さらに、上記の誘導電動機の制御装置を、例えばフォークリフトなどの車両に、その走行用車輪を回転駆動するための、または油圧ポンプを駆動するための誘導電動機の制御装置として用いてもよい。 Further, the induction motor control device described above may be used as a control device for an induction motor for driving a traveling wheel or driving a hydraulic pump in a vehicle such as a forklift.

上記課題を解決するため、本発明の誘導電動機の制御方法は、誘導電動機の１次電流の２つのベクトル成分である、外部入力された励磁電流指令値ｉ_１ｄ ^＊およびトルク電流指令値ｉ_１ｑ ^＊に基づいて、前記誘導電動機の滑り角周波数ω_ｓを演算するとともに、前記誘導電動機に供給される１次電流および前記誘導電動機から検出される角周波数ω_ｒのフィードバックを受けながら、前記誘導電動機の駆動を制御する誘導電動機の滑り周波数型ベクトル制御方法であって、前記励磁電流指令値ｉ_１ｄ ^＊を、予め設定された飽和電流制限閾値ｉ_１ｄｔｈ ^＊と比較するステップと、前記励磁電流指令値ｉ_１ｄ ^＊を、第１の励磁電流指令値ｉ_１ｄ１ ^＊と第２の励磁電流指令値ｉ_１ｄ２ ^＊とに分け、前記励磁電流指令値ｉ_１ｄ ^＊が前記飽和電流制限閾値ｉ_１ｄｔｈ ^＊以下である場合には、前記第１の励磁電流指令値ｉ_１ｄ１ ^＊として当該励磁電流指令値ｉ_１ｄ ^＊を、前記第２の励磁電流指令値ｉ_１ｄ２ ^＊として０を出力する一方で、前記励磁電流指令値ｉ_１ｄ ^＊が前記飽和電流制限閾値ｉ_１ｄｔｈ ^＊を超える場合には、前記第１の励磁電流指令値ｉ_１ｄ１ ^＊として前記飽和電流制限閾値ｉ_１ｄｔｈ ^＊を、前記第２の励磁電流指令値ｉ_１ｄ２ ^＊として前記励磁電流指令値ｉ_１ｄ ^＊と前記飽和電流制限閾値ｉ_１ｄｔｈ ^＊との差分値（ｉ_１ｄ ^＊−ｉ_１ｄｔｈ ^＊）を出力するステップと、前記第１の励磁電流指令値ｉ_１ｄ１ ^＊および前記トルク電流指令値ｉ_１ｑ ^＊を用いて第１の滑り角周波数ω_ｓ１を演算するステップと、前記第２の励磁電流指令値ｉ_１ｄ２ ^＊を用いて第２の滑り角周波数ω_ｓ２を演算するステップと、前記第１の滑り角周波数ω_ｓ１と前記第２の滑り角周波数ω_ｓ２を加算して滑り角周波数ω_ｓを出力するステップと、前記滑り角周波数ω_ｓと前記誘導電動機１２から検出される前記角周波数ω_ｒとを加算して１次角周波数ω_１を出力するステップと、前記１次角周波数ω_１を積分して位相θ_１を出力するステップと、前記誘導電動機に供給される前記１次電流から検出される励磁電流成分ｉ_１ｄおよびトルク電流成分ｉ_１ｑをそれぞれ前記励磁電流指令値ｉ_１ｄ ^＊および前記トルク電流指令値ｉ_１q ^＊に基づいて制御して、ｄ軸電圧指令値ｖ_１ｄ ^＊およびｑ軸電圧指令値ｖ_１ｑ ^＊を出力するステップと、前記ｄ軸電圧指令値ｖ_１ｄ ^＊および前記ｑ軸電圧指令値ｖ_１ｑ ^＊を前記位相θ_１により座標変換して１次電圧指令値ｖ_１ｕ ^＊、ｖ_１ｖ ^＊、ｖ_１ｗ ^＊を出力するステップと、前記１次電圧指令値ｖ_１ｕ ^＊、ｖ_１ｖ ^＊、ｖ_１ｗ ^＊に基づいて前記誘導電動機に駆動電力を供給して当該誘導電動機を駆動するステップと、を備える。 In order to solve the above-described problem, an induction motor control method according to the present invention includes an excitation current command value i _1d ^* and a torque current command value i _1q ^* that are externally input two vector components of the primary current of the induction motor ^. And calculating the slip angular frequency ω _s of the induction motor and receiving the feedback of the primary current supplied to the induction motor and the angular frequency ω _r detected from the induction motor. A method of controlling a slip frequency type vector of an induction motor for controlling driving, the step of comparing the excitation current command value i _1d ^* with a preset saturation current limit threshold i _1dth ^*, and the excitation current command value i _1d ^* is divided into a first excitation current command value i _1d1 ^* and a second excitation current command value i _1d2 ^*, and the excitation current command value i _1d ^* is saturated. If it is less than the current limit threshold _{i 1dth} ^*, the first exciting current command value _{i 1d1} ^* as the excitation current command value _{i 1d} ^*, outputs 0 as the second excitation current command value _{i 1d2} ^* while, when said excitation current command value _{i 1d} ^* exceeds the saturation current limit threshold _{i 1dth} ^* is the first exciting current command value _{i 1d1} ^* as the saturation current limit threshold _{i 1dth} ^*, wherein and outputting the second exciting current command value _{i 1d2} ^* as the exciting current command value _{i 1d} ^* and the difference value between the saturation current limit threshold _{i 1dth} ^* a _{^{_{^{(i 1d * -i 1dth *)}}}} , the first The step of calculating the first slip angular frequency ω _s1 using the excitation current command value i _1d1 ^* and the torque current command value i _1q ^* , and the second excitation current command value i _1d2 ^* And outputting the slip angular frequency omega _s by adding a step of computing a second slip angular frequency omega _s2, the first slip angular frequency omega _s1 and the second slip angular frequency omega _s2 using, and outputting a primary angular frequency omega ₁ by adding the said angular frequency omega _r detected from the induction motor 12 and the slip angular frequency omega _s, phase θ by integrating the primary angular frequency omega ₁ ₁ and the exciting current component i _1d and torque current component i _1q detected from the primary current supplied to the induction motor are converted into the exciting current command value i _1d ^* and the torque current command value i, respectively. A step of outputting the d-axis voltage command value v _1d ^* and the q-axis voltage command value v _1q ^* by controlling based on _1q ^* , the d-axis voltage command value v _1d ^* and the q-axis voltage command value v _1q ^* the previous Phase theta ₁ by the coordinate transformation to the primary voltage command value _^v ^1u _{_*,} _v 1v _*, and outputting the _{v 1 w} ^*, the primary voltage command value _^v ^1u _{_*,} _v 1v _*, based on _{v 1 w} ^* Supplying drive power to the induction motor to drive the induction motor.

また、上記構成において、前記第１の滑り角周波数ω_ｓ１を演算するステップにおいて、当該第１の滑り角周波数ω_ｓ１を、前記第１の励磁電流指令値ｉ_１ｄ１ ^＊および前記トルク電流指令値ｉ_１ｑ ^＊を用いて、次式、

によって演算し、前記第２の滑り角周波数ω_ｓ２を演算するステップにおいて、当該第２の滑り角周波数ω_ｓ２を、前記第２の励磁電流指令値ｉ_１ｄ２ ^＊を用いて、次式、

によって演算することが好ましい。 In the above structure, in the step of calculating the first slip angular frequency omega _s1, the first slip angular frequency omega _s1, the first exciting current command value i _1d1 ^* and the torque current command value i _{Using 1q} ^* , the following formula:

Computed by, in the step of calculating the second slip angular frequency omega _s2, the second slip angular frequency omega _s2, using said second excitation current command value i _1d2 ^*, the following equation,

It is preferable to calculate by:

本発明によれば、励磁電流指令値ｉ_１ｄ ^＊が、予め設定された飽和電流制限閾値ｉ_１ｄｔｈ ^＊と比較されて第１の励磁電流指令値ｉ_１ｄ１ ^＊と第２の励磁電流指令値ｉ_１ｄ２ ^＊とに分けられ、誘導電動機に必要な滑り角周波数ω_ｓが、当該第１の励磁電流指令値ｉ_１ｄ１ ^＊、当該第２の励磁電流指令値ｉ_１ｄ２ ^＊および当該トルク電流指令値ｉ_１q ^＊を用いて、演算されるようにした。これにより、励磁電流指令値ｉ_１ｄ ^＊が飽和電流制限閾値ｉ_１ｄｔｈ ^＊を越えた場合にも誘導電動機の制御を容易に行えるように構成したので、誘導電動機の制御の効率化を高めることができ、誘導電動機の低回転時における高トルクを実現できる。 According to the present invention, the excitation current command value i _1d ^* is compared with a preset saturation current limit threshold i _1dth ^* to compare the first excitation current command value i _1d1 ^* and the second excitation current command value i _1d2. ^*, And the slip angular frequency ω _s required for the induction motor is determined based on the first excitation current command value i _1d1 ^* , the second excitation current command value i _1d2 ^*, and the torque current command value i _1q ^*. It was made to calculate using. As a result, the induction motor can be easily controlled even when the excitation current command value i _1d ^* exceeds the saturation current limit threshold i _1dth ^* , so that the efficiency of the control of the induction motor can be increased. Thus, high torque can be realized at the time of low rotation of the induction motor.

本発明の実施例による誘導電動機の制御装置のブロック図である。It is a block diagram of the control apparatus of the induction motor by the Example of this invention. 本発明の実施例による誘導電動機の制御方法を説明するフロー図である。It is a flowchart explaining the control method of the induction motor by the Example of this invention. 本発明の実施例による誘導電動機の制御方法と従来の誘導電動機の制御方法の違いを説明するグラフであり、（ａ）は、２次側磁束−励磁電流指令値特性を示すグラフ、（ｂ）は励磁電流指令値−回転数特性を示すグラフ、（ｃ）は滑り角周波数−回転数特性を示すグラフである。It is a graph explaining the difference between the control method of the induction motor by the Example of this invention, and the control method of the conventional induction motor, (a) is a graph which shows the secondary side magnetic flux-excitation current command value characteristic, (b) Is a graph showing an excitation current command value-rotational speed characteristic, and (c) is a graph showing a slip angle frequency-rotational speed characteristic. 従来の誘導電動機の制御装置のブロック図である。It is a block diagram of the control apparatus of the conventional induction motor. 従来の誘導電動機の制御方法を説明するグラフであり、（ａ）は、２次側磁束−励磁電流指令値特性を示すグラフ、（ｂ）は励磁電流指令値−回転数特性を示すグラフ、（ｃ）は滑り角周波数−回転数特性を示すグラフである。It is a graph explaining the control method of the conventional induction motor, (a) is a graph which shows a secondary side magnetic flux-excitation current command value characteristic, (b) is a graph which shows an excitation current command value-rotational speed characteristic, c) is a graph showing slip angular frequency-rotational speed characteristics.

以下、本発明の好ましい１実施例について図面を参照しながら説明する。 Hereinafter, a preferred embodiment of the present invention will be described with reference to the drawings.

（実施例）
図１は、本発明の実施例による誘導電動機の制御装置のブロック図である。
図１を参照して、この誘導電動機１２の制御装置１は、外部入力された励磁電流指令値ｉ_１ｄ ^＊およびトルク電流指令値ｉ_１q ^＊に基づいて滑り角周波数ω_ｓを演算するとともに、当該誘導電動機１２に供給される１次電流および当該誘導電動機１２から検出される角周波数ω_ｒのフィードバックを受けながら、当該誘導電動機１２の駆動を制御するものである。 (Example)
FIG. 1 is a block diagram of an induction motor control apparatus according to an embodiment of the present invention.
Referring to FIG. 1, control device 1 for induction motor 12 calculates slip angular frequency ω _s based on externally input excitation current command value i _1d ^* and torque current command value i _1q ^*, and while receiving feedback of the angular frequency omega _r detected from the primary current and the induction motor 12 is supplied to the induction motor 12, and controls the driving of the induction motor 12.

この制御装置１において、励磁電流指令値制御手段２は、励磁電流指令値ｉ_１ｄ ^＊を、予め設定された飽和電流制限閾値ｉ_１ｄｔｈ ^＊と比較して、第１の励磁電流指令値ｉ_１ｄ１ ^＊と第２の励磁電流指令値ｉ_１ｄ２ ^＊とに分けるものである。より具体的には、励磁電流指令値制御手段２は、励磁電流指令値ｉ_１ｄ ^＊が飽和電流制限閾値ｉ_１ｄｔｈ ^＊以下である場合には、第１の励磁電流指令値ｉ_１ｄ１ ^＊として当該励磁電流指令値ｉ_１ｄ ^＊を、第２の励磁電流指令値ｉ_１ｄ２ ^＊として０を出力する一方で、励磁電流指令値ｉ_１ｄ ^＊が飽和電流制限閾値ｉ_１ｄｔｈ ^＊を超える場合には、第１の励磁電流指令値ｉ_１ｄ１ ^＊として飽和電流制限閾値ｉ_１ｄｔｈ ^＊を、第２の励磁電流指令値ｉ_１ｄ２ ^＊として励磁電流指令値ｉ_１ｄ ^＊と飽和電流制限閾値ｉ_１ｄｔｈ ^＊との差分値（ｉ_１ｄ ^＊−ｉ_１ｄｔｈ ^＊）を出力する。 In this control device 1, the excitation current command value control means 2 compares the excitation current command value i _1d ^* with a preset saturation current limit threshold i _1dth ^* to determine the first excitation current command value i _1d1 ^*. And the second excitation current command value i _1d2 ^* . More specifically, when the excitation current command value i _1d ^* is equal to or less than the saturation current limit threshold i _1dth ^* , the excitation current command value control means 2 determines the excitation current command value i _1d1 ^* as the first excitation current command value i _1d1 ^*. When the current command value i _1d ^* is output as 0 as the second excitation current command value i _1d2 ^* , while the excitation current command value i _1d ^* exceeds the saturation current limit threshold i _1dth ^* , the first the excitation current command value _{i 1d1} ^* as the saturation current limit threshold _{i 1dth} ^*, the second exciting current command value _{i 1d2} ^* as an excitation current command value _{i 1d} ^* and the saturation current limit threshold _{i 1dth} ^* difference value of _{(i 1d} ^* -I _1dth ^* ) is output.

滑り角周波数演算手段３は、第１の滑り角周波数演算手段４、第２の滑り角周波数演算手段５、および滑り角周波数加算器６からなる。
ここで、第１の滑り角周波数演算手段４は、第１の励磁電流指令値ｉ_１ｄ１ ^＊およびトルク電流指令値ｉ_１ｑ ^＊を用いて第１の滑り角周波数ω_ｓ１を、次式、

によって演算する。
また、第２の滑り角周波数演算手段５は、第２の励磁電流指令値ｉ_１ｄ２ ^＊を用いて第２の滑り角周波数ω_ｓ２を、次式、

によって演算する。なお、（１０）式は、第２の滑り角周波数ω_ｓ２が第２の励磁電流指令値ｉ_１ｄ２ ^＊を用いて線形近似されることに基づく。
滑り角周波数加算器６は、第１の滑り角周波数演算手段４から出力される第１の滑り角周波数ω_ｓ１と第２の滑り角周波数演算手段５から出力される第２の滑り角周波数ω_ｓ２を加算して滑り角周波数ω_ｓ＝ω_ｓ１＋ω_ｓ２を出力する。 The slip angle frequency calculation means 3 includes a first slip angle frequency calculation means 4, a second slip angle frequency calculation means 5, and a slip angle frequency adder 6.
Here, the first slip angular frequency calculating means 4 uses the first excitation current command value i _1d1 ^* and the torque current command value i _1q ^* to calculate the first slip angular frequency ω _s1 by the following formula:

Calculate by
Further, the second slip angular frequency calculating means 5 uses the second excitation current command value i _1d2 ^* to calculate the second slip angular frequency ω _s2 by the following formula:

Calculate by Equation (10) is based on the fact that the second slip angular frequency ω _s2 is linearly approximated using the second excitation current command value i _1d2 ^* .
The slip angle frequency adder 6 includes a first slip angle frequency ω _s1 output from the first slip angle frequency calculation means 4 and a second slip angle frequency ω output from the second slip angle frequency calculation means 5. _The slip angular frequency ω _s = ω _s1 + ω _s2 is output by adding _s2 .

１次角周波数演算器７は、滑り角周波数加算器６から出力される滑り角周波数ω_ｓと誘導電動機から検出される角周波数ω_ｒとを加算して１次角周波数ω_１＝ω_ｓ＋ω_ｒを出力する。 The primary angular frequency calculator 7 adds the slip angular frequency ω _s output from the slip angular frequency adder 6 and the angular frequency ω _r detected from the induction motor to obtain a primary angular frequency ω ₁ = ω _s + ω. Output _r .

積分器８は、１次角周波数演算器７から出力された１次角周波数ω_１を、次式、

によって時間積分して、位相θ_１を出力する。 The integrator 8 converts the primary angular frequency ω ₁ output from the primary angular frequency calculator 7 into the following formula:

And integration time by outputting a phase theta _1.

電流制御器９は、励磁電流成分ｉ_１ｄおよびトルク電流成分ｉ_１ｑを励磁電流指令値ｉ_１ｄ ^＊およびトルク電流指令値ｉ_１q ^＊に基づいて制御して、ｄ軸電圧指令値ｖ_１ｄ ^＊およびｑ軸電圧指令値ｖ_１ｑ ^＊を出力する。ここで、励磁電流成分ｉ_１ｄおよびトルク電流成分ｉ_１ｑは、誘導電動機１２に供給される１次電流が１次電流検出手段１４によって検出され、さらに３相−２相変換器１５（ｄｑ−αβ座標変換器１５ａおよびαβ−３相座標変換器１５ｂ）によって変換されることにより得られる。 The current controller 9 controls the excitation current component i _1d and the torque current component i _{1q based} on the excitation current command value i _1d ^* and the torque current command value i _1q ^* , so that the d-axis voltage command value v _1d ^* and q The shaft voltage command value v _1q ^* is output. Here, as for the exciting current component i _1d and the torque current component i _1q , the primary current supplied to the induction motor 12 is detected by the primary current detecting means 14, and the three-phase to two-phase converter 15 (dq-αβ It is obtained by being converted by the coordinate converter 15a and the αβ-3 phase coordinate converter 15b).

２相−３相変換器１０（ｄｑ−αβ座標変換器１０ａおよびαβ−３相座標変換器１０ｂ）は、ｄ軸電圧指令値ｖ_１ｄ ^＊およびｑ軸電圧指令値ｖ_１ｑ ^＊を位相θ_１により座標変換して１次電圧指令値ｖ_１ｕ ^＊、ｖ_１ｖ ^＊、ｖ_１ｗ ^＊を出力する。 The two-phase to three-phase converter 10 (dq-αβ coordinate converter 10a and αβ-3 phase coordinate converter 10b) converts the d-axis voltage command value v _1d ^* and the q-axis voltage command value v _1q ^{* according} to the phase θ _1. coordinate transformation to the primary voltage command value _{^{_{^{_{v 1u *, v 1v *,}}}}} v and outputs a _{1 w} ^*.

インバータ１１は、この１次電圧指令値ｖ_１ｕ ^＊、ｖ_１ｖ ^＊、ｖ_１ｗ ^＊に基づいて誘導電動機１２に駆動電力を供給して当該誘導電動機１２を駆動する。 The inverter 11 supplies drive power to the induction motor 12 based on the primary voltage command values v _1u ^* , v _1v ^* , v _1w ^* to drive the induction motor 12.

次に、本発明の誘導電動機の制御方法について、以下に簡単に説明する。 Next, the control method of the induction motor of the present invention will be briefly described below.

図２は、本発明の実施例による誘導電動機１２の制御方法を説明するフロー図である。図３は、本発明の実施例による誘導電動機の制御方法と従来の誘導電動機の制御方法の違いを説明するグラフである。
ここで、本発明は、励磁電流指令値ｉ_１ｄ ^＊が図３（ａ）に示す非線形領域の範囲（領域Ｂ）であっても誘導電動機の制御を行うことができる、誘導電動機の制御方法を提供するものである。 FIG. 2 is a flowchart illustrating a method for controlling the induction motor 12 according to the embodiment of the present invention. FIG. 3 is a graph for explaining a difference between a control method for an induction motor according to an embodiment of the present invention and a control method for a conventional induction motor.
Here, the present invention provides a control method for an induction motor that can control the induction motor even when the excitation current command value i _1d ^* is in the non-linear range (region B) shown in FIG. It is to provide.

まず、外部入力される励磁電流指令値ｉ_１ｄ ^＊およびトルク電流指令値ｉ_１q ^＊に基づいて、励磁電流指令値制御手段２によって、励磁電流指令値ｉ_１ｄ ^＊を予め設定された飽和電流制限閾値ｉ_１ｄｔｈ ^＊と比較する（ステップＳ１）。そして、励磁電流指令値制御手段２によって、励磁電流指令値ｉ_１ｄ ^＊を、第１の励磁電流指令値ｉ_１ｄ１ ^＊と第２の励磁電流指令値ｉ_１ｄ２ ^＊とに分け、励磁電流指令値ｉ_１ｄ ^＊が飽和電流制限閾値ｉ_１ｄｔｈ ^＊以下である場合（図３（ａ）に示す領域Ａの場合）には、第１の励磁電流指令値ｉ_１ｄ１ ^＊として当該励磁電流指令値ｉ_１ｄ ^＊を出力し、第２の励磁電流指令値ｉ_１ｄ２ ^＊として０を出力する（ステップＳ２）。一方で、励磁電流指令値ｉ_１ｄ ^＊が飽和電流制限閾値ｉ_１ｄｔｈ ^＊を超える場合（図３（ａ）に示す領域Ｂの場合）には、第１の励磁電流指令値ｉ_１ｄ１ ^＊として飽和電流制限閾値ｉ_１ｄｔｈ ^＊を出力し、第２の励磁電流指令値ｉ_１ｄ２ ^＊として励磁電流指令値ｉ_１ｄ ^＊と前記飽和電流制限閾値ｉ_１ｄｔｈ ^＊との差分値（ｉ_１ｄ ^＊−ｉ_１ｄｔｈ ^＊）を出力する（ステップＳ３）。 First, based on the excitation current command value i _1d ^* and the torque current command value i _1q ^* inputted externally, the excitation current command value control means 2 sets the excitation current command value i _1d ^* to a preset saturation current limit threshold value. Compare with i _1dth ^* (step S1). Then, the excitation current command value control means 2 divides the excitation current command value i _1d ^* into a first excitation current command value i _1d1 ^* and a second excitation current command value i _1d2 ^* , and the excitation current command value i _{When 1d} ^* is equal to or less than the saturation current limit threshold i _1dth ^* (in the case of the region A shown in FIG. 3A), the excitation current command value i _1d ^* is used as the first excitation current command value i _1d1 ^*. And 0 is output as the second excitation current command value i _1d2 ^* (step S2). On the other hand, when the excitation current command value i _1d ^* exceeds the saturation current limit threshold i _1dth ^* (in the case of the region B shown in FIG. 3A), the saturation current is set as the first excitation current command value i _1d1 ^*. outputs limit threshold _{i 1dth} ^*, the second exciting current command value _{i 1d2} ^* as an excitation current command value _{i 1d} ^* and the saturation current limit threshold _{i 1dth} ^* difference value of the _{^_(i} 1d _{^* -i} ^1dth _*) Output (step S3).

次に、滑り角周波数演算手段３の第１の滑り角周波数演算手段４によって、第１の滑り角周波数ω_ｓ１を、第１の励磁電流指令値ｉ_１ｄ１ ^＊およびトルク電流指令値ｉ_１ｑ ^＊を用いて、次式、

によって演算する（ステップＳ４）。
また、滑り角周波数演算手段３の第２の滑り角周波数演算手段５によって、第２の滑り角周波数ω_ｓ２を、第２の励磁電流指令値ｉ_１ｄ２ ^＊を用いて、次式、

によって演算する（ステップＳ５）。
そして、滑り角周波数加算器６によって、第１の滑り角周波数ω_ｓ１と第２の滑り角周波数ω_ｓ２を加算して滑り角周波数ω_ｓ＝ω_ｓ１＋ω_ｓ２を出力する（ステップＳ６）。 Next, the first slip angular frequency calculation means 4 of the slip angular frequency calculation means 3 converts the first slip angular frequency ω _s1 into the first excitation current command value i _1d1 ^* and the torque current command value i _1q ^* . Using the following formula:

(Step S4).
Further, the second slip angular frequency calculation means 5 of the slip angular frequency calculation means 3 uses the second excitation current command value i _1d2 ^* to calculate the second slip angular frequency ω _s2 as follows:

(Step S5).
Then, the slip angular frequency adder 6 adds the first slip angular frequency omega _s1 the second slip angular frequency omega _s2 outputs the slip angular frequency _{_{_{ω s = ω s1 + ω s2}}} ( step S6).

さらに、１次角周波数演算器７によって、滑り角周波数加算器６から出力される滑り角周波数ω_ｓと誘導電動機から検出される角周波数ω_ｒとを加算し、１次角周波数ω_１＝ω_ｓ＋ω_ｒを出力し（ステップＳ７）、積分器８によって、１次角周波数演算器７から出力された１次角周波数ω_１を、次式、

によって時間積分することにより、位相θ_１を出力する（ステップＳ８）。 Further, the primary angular frequency calculator 7 adds the slip angular frequency ω _s output from the slip angular frequency adder 6 and the angular frequency ω _r detected from the induction motor, and the primary angular frequency ω ₁ = ω. _s + ω _r is output (step S7), and the primary angular frequency ω ₁ output from the primary angular frequency calculator 7 by the integrator 8 is _expressed by the following equation:

By integration time by outputting a phase theta ₁ (step S8).

そして、電流制御器９によって、励磁電流成分ｉ_１ｄおよびトルク電流成分ｉ_１ｑをそれぞれ励磁電流指令値ｉ_１ｄ ^＊およびトルク電流指令値ｉ_１q ^＊に基づいて制御し、ｄ軸電圧指令値ｖ_１ｄ ^＊およびｑ軸電圧指令値ｖ_１ｑ ^＊を出力する（ステップＳ９）。そして、２相−３相変換器１０によって、ｄ軸電圧指令値ｖ_１ｄ ^＊およびｑ軸電圧指令値ｖ_１ｑ ^＊を位相θ_１を用いて座標変換して１次電圧指令値ｖ_１ｕ ^＊、ｖ_１ｖ ^＊、ｖ_１ｗ ^＊を出力し（ステップＳ１０）、インバータ１１によって、この１次電圧指令値ｖ_１ｕ ^＊、ｖ_１ｖ ^＊、ｖ_１ｗ ^＊に基づいて誘導電動機１２に駆動電力を供給し、当該誘導電動機１２を駆動する（ステップＳ１１）。なお、このステップＳ９乃至ステップＳ１１において、誘導電動機１２の制御装置１によって、励磁電流成分ｉ_１ｄおよびトルク電流成分ｉ_１ｑと、励磁電流指令値ｉ_１ｄ ^＊およびトルク電流指令値ｉ_１q ^＊との偏差をそれぞれ０にするように、常にフィードバック制御を行って当該誘導電動機１２を駆動する。 The current controller 9 controls the excitation current component i _1d and the torque current component i _{1q based} on the excitation current command value i _1d ^* and the torque current command value i _1q ^* , respectively, and the d-axis voltage command value v _1d ^*. And q-axis voltage command value _v1q ^* is output (step S9). Then, the two-phase to three-phase converter 10 _performs coordinate conversion of the d-axis voltage command value v _1d ^* and the q-axis voltage command value v _1q ^* using the phase θ ₁ and _performs primary voltage command values v _1u ^* , v _{1 v} ^*, _v outputs _{1 w} ^* (step S10), and the inverter 11, the primary voltage command value _{^{_{^{_{v 1u *, v 1v *,}}}}} v drive power supplied to the induction motor 12 on the basis of _{1 w} ^*, the induction The electric motor 12 is driven (step S11). In steps S9 to S11, the controller 1 of the induction motor 12 causes the deviation between the excitation current component i _1d and the torque current component i _1q and the excitation current command value i _1d ^* and the torque current command value i _1q ^*. Thus, the induction motor 12 is driven by always performing feedback control so that each is set to 0.

図３を参照して、従来の誘導電動機２７の制御方法では、点線のグラフに示すように、回転数が磁束飽和点より低い領域（以下、低回転数領域という）において、励磁電流指令値ｉ_１ｄ ^＊の上限値を飽和電流制限閾値ｉ_１ｄｔｈ ^＊に制限し、（１）式を用いて滑り角周波数ω_ｓ＝ω_ｓ１を算出するとともに、（３）式および（４）式を用いて正確なトルクτを算出していた。
一方、実線のグラフに示すように、本発明の誘導電動機１２の制御方法では、低回転数領域において、励磁電流指令値ｉ_１ｄ ^＊を、非線形領域（領域Ｂ）の範囲まで拡げるべくｉ_１ｄ ^＊＝ｉ_１ｄ１ ^＊＋ｉ_１ｄ２ ^＊＝ｉ_１ｄｔｈ ^＊＋ｉ_１ｄ２ ^＊とした。ここで、誘導電動機１２のトルクτ、２次側磁束φ_２および滑り角周波数ω_ｓは、誘導電動機１２のどのような回転数（回転速度）領域においても、それぞれ、次式

、

、および

により、算出される。
しかしながら、非線形領域（領域Ｂ）の範囲にある励磁電流指令値ｉ_１ｄ ^＊を用いた場合においては、上述したように２次側磁束の磁束飽和が生じるため、実際の２次側磁束φ_２が（１６）式に基づいて計算した２次側磁束φ_２より小さくなり、実際のトルクτも（１５）式に基づいて計算したトルクτより小さくなる。
そこで、図３（ｂ）および（ｃ）ならびに（１７）式に示すように、低回転数領域において、誘導電動機１２の滑り角周波数ω_ｓを、従来のω_ｓ１から、励磁電流指令値ｉ_１ｄ ^＊と飽和電流制限閾値ｉ_１ｄｔｈ ^＊の差分値（ｉ_１ｄ２ ^＊＝ｉ_１ｄ ^＊−ｉ_１ｄｔｈ ^＊）に比例した滑り角周波数ω_ｓ２だけアップさせた。そして、（１５）式、（１６）式および（１７）式の関係を満たすように、２次側磁束φ_２およびトルク電流指令値ｉ_１q ^＊の値を調整することにより、励磁電流指令値ｉ_１ｄ ^＊が線形領域（領域Ａ）または非線形領域（領域Ｂ）のうちのいずれの範囲にある場合であっても、正確なトルクτを算出でき、かつ従来よりも高いトルクを得ることができる。 Referring to FIG. 3, in the conventional method for controlling induction motor 27, as shown by the dotted line graph, in a region where the rotational speed is lower than the magnetic flux saturation point (hereinafter referred to as a low rotational speed region), excitation current command value i The upper limit value of _1d ^* is limited to the saturation current limit threshold i _1dth ^* , and the slip angular frequency ω _s = ω _s1 is calculated using the equation (1), and is accurately calculated using the equations (3) and (4). Torque τ was calculated.
On the other hand, as shown by the solid line in the graph, the control method of the induction motor 12 of the present invention, in the low speed range, the excitation current command value i _1d ^* a, i _1d to expand to the extent of non-linear region (region B) ^* _{^{_{^{_{= i 1d1 * + i 1d2 *}}}}} = i 1dth * + i 1d2 * and the. Here, the torque τ, the secondary magnetic flux φ ₂ and the slip angular frequency ω _s of the induction motor 12 are respectively expressed by the following formulas in any rotation speed (rotation speed) region of the induction motor 12.

,

,and

Is calculated by
However, when the excitation current command value i _1d ^* in the non-linear region (region B) is used, the magnetic flux saturation of the secondary magnetic flux occurs as described above, so that the actual secondary magnetic flux φ ₂ is The secondary side magnetic flux φ ₂ calculated based on the equation (16) is smaller, and the actual torque τ is also smaller than the torque τ calculated based on the equation (15).
Therefore, as shown in FIGS. 3B, 3C, and 17, the slip angular frequency ω _s of the induction motor 12 is changed from the conventional ω _s1 to the excitation current command value i _1d in the low rotation speed region. ^* and was only allowed to up the saturation current limit threshold _{i 1dth} ^* of the difference value _{^{_{^{_{(i 1d2 * = i 1d *}}}}} -i 1dth *) slip angular frequency ω _s2 which is proportional to. Then, the excitation current command value i is adjusted by adjusting the secondary magnetic flux φ ₂ and the torque current command value i _1q ^* so as to satisfy the relationships of the equations (15), (16), and (17). _{Even if 1d} ^* is in any of the linear region (region A) and the non-linear region (region B), it is possible to calculate an accurate torque τ and to obtain a higher torque than before.

したがって、上記ステップＳ１乃至Ｓ１１を実行することにより、励磁電流指令値ｉ_１ｄ ^＊が飽和電流制限閾値ｉ_１ｄｔｈ ^＊を越えた範囲にある場合にも誘導電動機１２の制御を容易に行えることがわかる。故に、誘導電動機の制御の効率を高めることができ、誘導電動機の低回転時における高トルクを実現できる。 Therefore, it can be seen that by executing the steps S1 to S11, the induction motor 12 can be easily controlled even when the excitation current command value i _1d ^* is in the range _{exceeding the} saturation current limit threshold i _1dth ^* . Therefore, the control efficiency of the induction motor can be increased, and a high torque can be realized at the time of low rotation of the induction motor.

なお、本発明は上記実施例に限定されるものではなく、その趣旨を逸脱しない限りにおいて上述したもの以外に種々の変更を行うことが可能である。
例えば、本実施例において、励磁電流指令値を飽和電流制限閾値と比較して滑り角周波数を演算するものであれば、その励磁電流指令値制御手段、滑り角周波数演算手段の構成は何ら限定されない。
また、本実施例において、第２の滑り角周波数ω_ｓ２を第２の励磁電流指令値ｉ_１ｄ２ ^＊の線形近似式を用いて演算したが、第２の滑り角周波数ω_ｓ２と第２の励磁電流指令値ｉ_１ｄ２ ^＊の相関関係を考慮して、第２の滑り角周波数ω_ｓ２を第２の励磁電流指令値ｉ_１ｄ２ ^＊の多項式の近似式を用いて演算するようにしてもよい。
さらに、本発明による誘導電動機の制御装置を例えばフォークリフトなどの電気車両に応用することができる。 The present invention is not limited to the above-described embodiments, and various modifications other than those described above can be made without departing from the spirit of the present invention.
For example, in the present embodiment, the configuration of the excitation current command value control means and the slip angle frequency calculation means is not limited as long as the slip current frequency is calculated by comparing the excitation current command value with the saturation current limit threshold. .
In the present embodiment, the second slip angular frequency ω _s2 is calculated using a linear approximation formula of the second excitation current command value i _1d2 ^* , but the second slip angular frequency ω _s2 and the second excitation angular frequency ω _s2 are calculated. In consideration of the correlation of the current command value i _1d2 ^* , the second slip angular frequency ω _s2 may be calculated using an approximate expression of a polynomial of the second excitation current command value i _1d2 ^* .
Furthermore, the induction motor control device according to the present invention can be applied to an electric vehicle such as a forklift.

１制御装置
２励磁電流指令値制御手段
３滑り角周波数演算手段
４第１の滑り角周波数演算手段
５第２の滑り角周波数演算手段
６滑り角周波数加算器
７１次角周波数演算器
８積分器
９電流制御器
１０２相−３相変換器
１０ａｄｑ−αβ座標変換器
１０ｂ αβ−３相座標変換器
１１インバータ（電力変換器）
１２誘導電動機
１３パルスジェネレータ（パルス生成器）
１４１次電流検出手段
１５３相−２相変換器
１５ａｄｑ−αβ座標変換器
１５ｂ αβ−３相座標変換器
２０制御装置
２１滑り角周波数演算手段
２２１次角周波数演算器
２３積分器
２４電流制御器
２５２相−３相変換器
２５ａｄｑ−αβ座標変換器
２５ｂ αβ−３相座標変換器
２６インバータ（電力変換器）
２７誘導電動機
２８パルスジェネレータ（パルス生成器）
２９１次電流検出手段
３０３相−２相変換器
３０ａｄｑ−αβ座標変換器
３０ｂ αβ−３相座標変換器 DESCRIPTION OF SYMBOLS 1 Control apparatus 2 Excitation current command value control means 3 Sliding angular frequency calculating means 4 1st sliding angular frequency calculating means 5 2nd sliding angular frequency calculating means 6 Sliding angular frequency adder 7 Primary angular frequency calculating unit 8 Integrator 9 Current controller 10 2-phase-3 phase converter 10a dq-αβ coordinate converter 10b αβ-3 phase coordinate converter 11 Inverter (power converter)
12 Induction motor 13 Pulse generator (pulse generator)
14 Primary current detector 15 Three-phase to two-phase converter 15a dq-αβ coordinate converter 15b αβ-3 phase coordinate converter 20 Controller 21 Slip angular frequency calculator 22 Primary angular frequency calculator 23 Integrator 24 Current Controller 25 2-phase-3 phase converter 25a dq-αβ coordinate converter 25b αβ-3 phase coordinate converter 26 Inverter (power converter)
27 Induction motor 28 Pulse generator (pulse generator)
29 Primary current detection means 30 3-phase-2 phase converter 30a dq-αβ coordinate converter 30b αβ-3 phase coordinate converter

Claims

Of the two vector components of the primary current of the induction motor, the excitation current component i _1d and the torque current component i _1q are controlled based on the excitation current command value i _1d ^* and the torque current command value i _1q ^* respectively input from the outside. A current controller that outputs the d-axis voltage command value v _1d ^* and the q-axis voltage command value v _1q ^* , and at least one of the excitation current command value i _1d ^* or the torque current command value i _1q ^* One out using a slip angular frequency calculation means for calculating a slip angular frequency omega _s of the induction motor, the angular frequency to be detected from the slip angular frequency omega _s and the induction motor output from the slip angular frequency calculation means and primary angular frequency arithmetic unit for outputting a first-order angular frequency omega ₁ by adding omega _r, the primary angular frequency arithmetic unit by integrating the primary angular frequency omega ₁ outputted from the output phase theta ₁ You Integrators, and the d-axis voltage command value v _1d ^* and the q-axis voltage command value v _1q ^* are coordinate-transformed by the phase θ ₁ to produce primary voltage command values v _1u ^* , v _1v ^* , v _1w ^*. A two-phase to three-phase converter, and an inverter for driving the induction motor by supplying drive power to the induction motor based on the primary voltage command values v _1u ^* , v _1v ^* , v _1w ^* An induction motor control device comprising:
The excitation current command value i _1d ^* is compared with a _preset saturation current limit threshold i _1dth ^* , and is divided into a first excitation current command value i _1d1 ^* and a second excitation current command value i _1d2 ^*. , wherein when exciting current command value _{i 1d} ^* is equal to or less than the saturation current limit threshold _{i 1dth} ^* is the first exciting current command value _{i 1d1} ^* as the excitation current command value _{i 1d} ^*, the second while outputting the excitation current command value _{i 1d2} ^* 0 of, when said excitation current command value _{i 1d} ^* exceeds the saturation current limit threshold _{i 1dth} ^*, the first exciting current command value _{i 1d1} ^* the saturation current limit threshold _{i 1dth} ^*, the second excitation current command value _{i 1d2} ^* as the exciting current command value _{i 1d} ^* and the saturation current limit threshold _{i 1dth} ^* difference value of _{(i 1d} ^* as - Comprising an excitation current command value control means for outputting a _1dth ^*), and
The slip angular frequency calculation means includes
First slip angular frequency calculating means for calculating a first slip angular frequency ω _s1 using the first excitation current command value i _1d1 ^* and the torque current command value i _1q ^* ;
Second slip angular frequency calculating means for calculating a second slip angular frequency ω _s2 using the second excitation current command value i _1d2 ^* ;
The first slip angular frequency ω _s1 output from the first slip angular frequency calculating means and the second slip angular frequency ω _s2 output from the second slip angular frequency calculating means are added together to add the A slip angular frequency adder that outputs a slip angular frequency ω _s ,
The primary angular frequency calculator is
The primary angular frequency ω ₁ is output by adding the sliding angular frequency ω _s output from the sliding angular frequency adder and the angular frequency ω _r detected from the induction motor. Electric motor control device.

The first slip angular frequency calculating means calculates the first slip angular frequency ω _s1 using the first excitation current command value i _1d1 ^* and the torque current command value i _1q ^* by the following formula:

Is calculated by
The second slip angular frequency calculating means calculates the second slip angular frequency ω _s2 using the second excitation current command value i _1d2 ^* by the following formula:

The control apparatus for an induction motor according to claim 1, wherein the calculation is performed by:

A vehicle using the control device for an induction motor according to claim 1 or 2 as a control device for a driving wheel or an induction motor for driving a hydraulic pump.

While calculating the slip angular frequency ω _s of the induction motor based on the excitation current command value i _1d ^* and the torque current command value i _1q ^* which are externally input, which are two vector components of the primary current of the induction motor. A slip frequency type vector control method for an induction motor that controls driving of the induction motor while receiving feedback of a primary current supplied to the induction motor and an angular frequency ω _r detected from the induction motor,
Comparing the excitation current command value i _1d ^* with a preset saturation current limit threshold i _1dth ^* ;
The excitation current command value i _1d ^* is divided into a first excitation current command value i _1d1 ^* and a second excitation current command value i _1d2 ^*, and the excitation current command value i _1d ^* is the saturation current limit threshold i. If _1Dth ^* or less, said first excitation current command value _{i 1d1} ^* as the excitation current command value _{i 1d} ^*, while outputting 0 as the second excitation current command value _{i 1d2} ^*, when said exciting current command value _{i 1d} ^* exceeds the saturation current limit threshold _{i 1dth} ^* is the first exciting current command value _{i 1d1} ^* as the saturation current limit threshold _{i 1dth} ^*, the second excitation and outputting the current command value _{i 1d2} ^* as the exciting current command value _{i 1d} ^* and the saturation current limit threshold _{i 1dth} ^* difference value of the _{^{_{^{(i 1d * -i 1dth *)}}}} ,
Calculating a first slip angular frequency ω _s1 using the first excitation current command value i _1d1 ^* and the torque current command value i _1q ^* ;
Calculating a second slip angular frequency ω _s2 using the second excitation current command value i _1d2 ^* ;
Adding the first slip angular frequency ω _s1 and the second slip angular frequency ω _s2 to output the slip angular frequency ω _s ;
Adding the slip angular frequency ω _s and the angular frequency ω _r detected from the induction motor to output a primary angular frequency ω ₁ ;
And outputting the phase theta ₁ by integrating the primary angular frequency omega _1,
The exciting current component i _1d and the torque current component i _1q detected from the primary current supplied to the induction motor are controlled based on the exciting current command value i _1d ^* and the torque current command value i _1q ^* , respectively. _Outputting a d-axis voltage command value v _1d ^* and a q-axis voltage command value v _1q ^* ;
The d-axis voltage command value _{v 1d} ^* and the q-axis voltage command value _{v 1q} ^* the phase theta ₁ coordinate transformation to the primary voltage command value by _^v ^1u _{_*,} _v 1v _*, and outputting the _{v 1 w} ^* ,
And a step of supplying drive power to the induction motor based on the primary voltage command values v _1u ^* , v _1v ^* , v _1w ^* to drive the induction motor.

In the step of calculating the first slip angular frequency omega _s1, the first slip angular frequency omega _s1, using the first exciting current command value i _1d1 ^* and the torque current command value i _1q ^*, The following formula,

Is calculated by
Wherein in a second step of calculating the slip angular frequency omega _s2, the second slip angular frequency omega _s2, using said second excitation current command value i _1d2 ^*, the following equation,

The control method according to claim 4, wherein the calculation is performed by: