JP4649381B2 - Wraparound canceller - Google Patents

Description

  The present invention relates to a relay station or a relay apparatus in digital broadcasting or digital transmission using OFDM (Orthogonal Frequency Division Multiplexing), and in particular, between transmitting and receiving antennas in an SFN (Single Frequency Network) broadcasting wave relay station. The present invention relates to a sneak canceller for removing a sneak current of radio waves in the network.

  The sneak canceller cancels a sneak component generated by coupling between transmitting and receiving antennas included in the received signal and retransmits only the upper station signal in the SFN broadcast wave relay station that performs retransmission at the same frequency as the frequency of the received signal. It is a device.

  In the SFN broadcast wave relay station, wraparound occurs between a transmission antenna that radiates a radio wave of a transmission signal and a reception antenna that receives a radio wave from an upper station as a desired wave. The wraparound occurs when a part of the radio wave radiated from the transmission antenna (the radio wave of the transmission signal) passes through the wraparound propagation path and is received by the reception antenna that receives the higher-order local wave.

  In order to cancel this sneak component from the received signal, it is necessary to realize the same transmission characteristics as the sneak path and generate a sneak replica signal inside the sneak canceller. The wraparound canceller cancels the wraparound by subtracting the wraparound replica signal generated inside the wraparound canceller from the received signal, and can extract only the signal from the upper station and transmit it as a transmission signal. Such wraparound cancellers include those described in Patent Documents 1 to 6, for example.

  FIG. 1 is a block diagram showing an outline of a general configuration of a wraparound canceller. This wraparound canceller 100 performs frequency conversion, AD conversion, and further subtracts wraparound replicas from an equivalent baseband signal obtained by performing quadrature demodulation on a received signal input via an external receiving antenna and receiving unit (not shown). The subtracting unit 201 and the signal output from the subtracting unit 201 are subjected to filter processing to generate a FFF (Feed-Forward Filter) 202 that equalizes a bandwidth limit of a predetermined bandwidth and multipath distortion of a higher station, and a wraparound replica FBF (Feed-Back Filter) 203, and a filter coefficient control unit 110 that generates a filter coefficient of FFF 202 and a filter coefficient of FBF 203 are provided. The equivalent baseband signal that has been filtered by the FFF 202 is subjected to quadrature modulation, D / A conversion, and frequency conversion, and is output to the outside. The output signal of the wraparound canceller 100 is amplified to a predetermined power by a PA (not shown) and then transmitted from a transmission antenna (not shown).

Under such a configuration, the FBF 203 and the subtraction unit 201 cancel the wraparound, and the FFF 202 performs multipath distortion equalization of the upper station signal and signal band limitation.
To do.

  FIG. 3 is a block diagram showing a first configuration of a conventional wraparound canceller. This conventional wraparound canceller 101 includes a subtractor 201, FFF 202, FBF 203, and filter coefficient controller 110. The filter coefficient control unit 110 is an effective symbol period extraction unit 111 that extracts an effective symbol period of the OFDM signal in the time domain based on the symbol timing of the received OFDM signal that is reproduced and supplied by a synchronous reproduction unit (not shown). FFT (Fast Fourier Transform) means 112 that performs fast Fourier transform on a time-domain OFDM signal and outputs a carrier symbol in the frequency domain, channel response estimation that estimates a channel response (frequency characteristic of a transmission path) from the carrier symbol Means 113, cancellation residual calculation means 114 for calculating and outputting the frequency characteristics of the cancellation residual from the channel response, inverse fast Fourier transform of the cancellation residual response which is a frequency domain signal, and canceling the residual of the time domain signal Impulse of IFFT (Inverse Fast Fourier Transform) means 115 for outputting a response, coefficient cutout means 122 and 123 for cutting out coefficients in a predetermined time range from the impulse response of the cancellation residual output by IFFT means, Multiplication means 116, 119 for multiplying the impulse response of the cancellation residual by a predetermined constant, addition means 117, 120 for generating the filter coefficient by adding the multiplied impulse response of the cancellation residual, and a predetermined filter coefficient There are provided delay means 118 and 121 for delaying the output. The filter coefficient generated by the adding unit 117 is output to the FBF 203, and the filter coefficient generated by the adding unit 120 is output to the FFF 202.

  As described in Patent Document 7, the bandwidth of the FFF 202 illustrated in FIG. 3 is transmitted from the output end of the FFF 202 via a transmission unit, a PA unit, a transmission antenna, a sneak path, a reception antenna, and a reception unit (not shown). It is necessary not to exceed the bandwidth of the wraparound loop that reaches the input end of the FFF 202 via the subtraction unit 201 and the cancel loop that reaches the input end of the FFF 202 via the FBF 203 from the output end of the FFF 202.

  Patent Document 8 discloses a wraparound canceller that equalizes and retransmits the distortion due to multipath of the upper station signal in an environment where a multipath component having a small delay time difference from the main wave exists in the propagation path from the upper station. Are listed. However, in this wraparound canceller, when FFF control that equalizes multipath distortion is performed, the passband of the FFF sometimes exceeds the passband width of the wraparound loop and the cancellation loop. In this case, when the loop gain of the wraparound loop is larger than 1, the noise level outside the OFDM signal band increases in the output signal of the wraparound canceller, and there is a problem that oscillation is caused by the increased noise.

  FIG. 4 is a block diagram showing a second configuration of a conventional wraparound canceller. This conventional wraparound canceller 102 includes a subtractor 201, FFF 202, FBF 203, and filter coefficient controller 210. The filter coefficient control unit 210 includes an effective symbol period extraction unit 211, an FFT unit 212, a channel response estimation unit 213, an LPF unit 214, a division unit 215, a cancellation residual calculation unit 216, an IFFT unit 217, a coefficient extraction unit 221, and a multiplication unit. 218, adding means 219, and delay means 220 are provided. The LPF unit 214 extracts a low frequency component of the channel response estimated by the channel response estimation unit 213. The dividing unit 215 divides the channel response estimated by the channel response estimating unit 213 by the low frequency component of the channel response extracted by the LPF unit 214, and removes the multipath distortion component of the upper station signal from the channel response. .

  Comparing the wraparound canceller 101 shown in FIG. 3 and the wraparound canceller 102 of FIG. 4 is the same in that it includes a subtracting unit 201, an FFF 202, and an FBF 203, but the configuration of the filter coefficient control units 110 and 210, and The wraparound canceller 102 is different in that the filter coefficient of the FFF 202 is not input from the filter coefficient control unit 210. Further, when the filter coefficient control unit 110 of the wraparound canceller 101 and the filter coefficient control unit 210 of the wraparound canceller 102 are compared, the filter coefficient control unit 210 extracts the low-frequency component of the channel response, and the channel response Is divided by the low-frequency component to remove the multipath distortion component of the higher-order station signal from the channel response, and further differs in that no means for generating the filter coefficient of the FFF 202 is provided. To do.

  Patent Document 9 describes a wraparound canceller that corrects an error in the FFT window position with respect to the channel response. According to this wraparound canceller, the LPF means 214 and the division shown in FIG. 4 are used when obtaining the filter coefficient of the FBF in an environment where a multipath component having a small delay time difference from the main wave exists in the transmission path from the upper station. Similar to the means 215, the channel response is divided by its low frequency component to remove the multipath distortion component of the upper station signal. Thereby, since the multipath component having a small delay time difference from the main wave is removed, it is possible to improve the estimation accuracy of the wraparound characteristic.

JP-A-11-355160 JP 2000-341238 A JP 2000-349734 A JP 2001-94528 A JP 2000-295195 A Japanese Patent Laid-Open No. 2001-237749 Japanese Patent Laid-Open No. 2002-77096 JP 2003-174430 A JP 2004-320677 A

  By the way, by combining the above-described Patent Documents 8 and 9, it is possible to assume a wraparound canceller that controls the FFF that equalizes distortion due to multipath of the upper station signal. FIG. 5 is a block diagram showing a configuration of a wraparound canceler assumed from Patent Documents 8 and 9. In FIG. The wraparound canceller 103 includes a subtractor 201, FFF 202, FBF 203, and filter coefficient controller 310. The filter coefficient control unit 310 includes an effective symbol period extraction unit 311, an FFT unit 312, a channel response estimation unit 313, an LPF unit 314, a division unit 315, a cancellation residual calculation unit 316, an IFFT unit 317, coefficient extraction units 324, 325, Multiplication means 318 and 321, addition means 319 and 322, and delay means 320 and 323 are provided.

  Comparing the wraparound canceller 102 shown in FIG. 4 and the wraparound canceller 103 shown in FIG. 5 is the same in that it includes a subtracting unit 201, an FFF 202, and an FBF 203, but the configuration of the filter coefficient control units 210 and 310, and The wraparound canceller 103 is different in that the filter coefficient of the FFF 202 is input from the filter coefficient control unit 310. Further, when the filter coefficient control unit 310 of the wraparound canceller 103 is compared with the filter coefficient control unit 210 of the wraparound canceller 102, the filter coefficient control unit 310 generates a coefficient cutout unit 325 and a multiplication unit 321 for generating the filter coefficient of the FFF 202. , Except that an adding means 322 and a delay means 323 are provided.

  According to the wraparound canceller 103 shown in FIG. 5, the LPF unit 314 extracts the low frequency component of the channel response estimated by the channel response estimation unit 313, and the division unit 315 is estimated by the channel response estimation unit 313. The channel response is divided by the low frequency component of the channel response extracted by the LPF means 314 to remove multipath distortion components from the channel response. As a result, the filter coefficient of the FBF 203 is generated from the channel response from which the multipath distortion component is removed in an environment where there is a multipath component having a small delay time difference from the main wave in the transmission path from the upper station. The estimation accuracy of can be improved. However, since the filter coefficient of the FFF 202 is generated from the channel response from which the distortion component due to multipath having a small delay time difference from the main wave is removed, it is impossible to realize a wraparound canceller that equalizes and retransmits distortion due to multipath. There was a problem.

  Therefore, the present invention has been made to solve such a problem, and its purpose is to combine the transmitting and receiving antennas in an environment in which a multipath component having a small delay time difference from the main wave exists in the transmission path of the upper station signal. It is an object of the present invention to provide a sneak canceller that eliminates the sneak path caused by multipath and equalizes distortion due to multipath, and a relay apparatus that relays a higher-level station signal satisfactorily and stably using the sneak canceller.

  In order to solve the above-described problem, the wraparound canceller according to the present invention generates a wraparound replica for canceling the wraparound by combining in reverse phase with an input OFDM (Orthogonal Frequency Division Multiplexing) signal. In a wraparound canceller comprising at least a filter, a FFF (Feed-Forward Filter) that equalizes multipath distortion of a signal after wraparound cancellation, and a filter coefficient control unit that generates a filter coefficient for controlling the FFF The filter coefficient control unit extracts an effective symbol period extraction unit that extracts a period corresponding to an effective symbol period of the OFDM signal input from the FFF, and the effective symbol period FFT means for converting the OFDM signal input from the inter-phase extraction section into a carrier symbol which is a frequency domain signal by FFT (Fast Fourier Transform), and channel response estimation for estimating the channel response from the carrier symbol input from the FFT means LPF (Low Pass Filter) that extracts a low-frequency component including a multipath component in a delay time range that can be equalized by a main wave from an upper station and an FFF from a channel response input from the channel response estimation unit ) Means, FFT window position correcting means for correcting a phase rotation component due to the shift of the FFT window position included in the low frequency component of the channel response inputted from the LPF means, and a channel inputted from the FFT window position correcting means Multipass distortion equalization for low frequency response components An equalization residual calculation means for converting to a difference (frequency domain signal) and an equalization residual input from the equalization residual calculation means are subjected to IFFT (Inverse Fast Fourier Transform) processing, and an impulse response of the equalization residual IFFT means for converting into IFFT, cutting out means for cutting out a coefficient in a predetermined time range from the impulse response of the equalization residual inputted from the IFFT means, and an impulse response after cutting out inputted from the coefficient cutting out means Multiplication means for multiplying the constants, delay means for holding the filter coefficients of the FFF for unit update time, impulse response of equalization residual after constant multiplication input from the multiplication means, and input from the delay means Addition for generating a new FFF filter coefficient by adding the FFF filter coefficient before the unit update time And means.

  Further, the wraparound canceller according to the present invention generates FBF replica signals for canceling wraparound by synthesizing in reverse phase with the input OFDM signal, multipath distortion of the signal after wraparound cancellation, etc. And a filter coefficient control unit that generates a filter coefficient for controlling the FBF and a filter coefficient for controlling the FFF, the filter coefficient control unit receives an input from the FFF An effective symbol period extraction unit for extracting a period corresponding to an effective symbol period of the OFDM signal to be performed, and an FFT means for converting the OFDM signal input from the effective symbol period extraction unit into a carrier symbol which is a frequency domain signal by FFT And input from the FFT means Channel response estimation means for estimating a channel response from a carrier symbol, and a channel response input from the channel response estimation means, including a multipath component in a delay time range that can be equalized by a main wave from an upper station and FFF Dividing the channel response input from the channel response estimation means by the low frequency component of the channel response input from the LPF means for each subcarrier of the OFDM signal, A division means for removing multipath distortion components of the channel response; a cancellation residual calculation means for calculating a wraparound cancellation residual from the channel response from which the multipath distortion components input from the division means are removed; and the cancellation residual. The wraparound cancellation residual (frequency domain signal) input from the difference calculation means is I IFFT means for performing FT processing and converting it into an impulse response of a wraparound cancellation residual, a coefficient cutout means for cutting out a coefficient in a predetermined time range from the impulse response of the wraparound cancellation residual inputted from the IFFT means, and the coefficient cutout Multiplication means for multiplying the impulse response of the cancellation residual after clipping inputted from the means, a delay means for holding the filter coefficient of the FBF for a unit update time, and a wraparound inputted from the multiplication means An adding means for adding a cancellation residual impulse response and an FBF filter coefficient before the unit update time inputted from the delay means to generate a new FBF filter coefficient, and a channel response inputted from the LPF means Correct the phase rotation component due to the shift of the FFT window position included in the low frequency component FFT window position correcting means, an equalization residual calculating means for converting a low frequency component of the channel response input from the FFT window position correcting means into an equalization residual (frequency domain signal) of multipath distortion, IFFT means that performs an IFFT (Inverse Fast Fourier Transform) process on the equalization residual input from the equalization residual calculation means and converts it into an impulse response of the equalization residual, and the equalization residual input from the IFFT means A unit for cutting out a coefficient in a predetermined time range from the impulse response, a multiplier for multiplying the impulse response inputted from the coefficient cutting unit by a predetermined constant, and updating the filter coefficients of the FFF Delay means for holding time, impulse response of equalization residual after multiplication by constant inputted from the multiplication means and the previous Characterized in that it comprises adding means which adds the filter coefficients of the unit updates time before FFF inputted from the delay means for generating a filter coefficient of the new FFF, the.

  In the wraparound canceller according to the present invention, the filter coefficient control unit further controls the FFF by convolving a predetermined filter coefficient with the filter coefficient of the new FFF generated by the adding unit. Convolution operation means for generating filter coefficients is provided.

  The relay device according to the present invention uses the wraparound canceller.

  As described above, according to the present invention, a wraparound canceller that equalizes distortion due to multipath and retransmits in an environment where a multipath component with a small delay time difference from the main wave exists in the transmission path of the higher-level station signal, and By using this, it is possible to realize a relay device that relays the upper station wave satisfactorily and stably.

The best mode for carrying out the present invention will be described below in detail with reference to the drawings.
FIG. 2 is a block diagram showing the configuration of the wraparound canceller according to the embodiment of the present invention. The wraparound canceller 1 includes a subtracting unit 201, an FFF 202, an FBF 203, and a filter coefficient control unit 10. The filter coefficient control unit 10 includes an effective symbol period extraction unit 11, an FFT unit 12, a channel response estimation unit 13, an LPF unit 14, a division unit 15, a cancellation residual calculation unit 16, IFFT units 17, 23, a coefficient cutout unit 30, 31, multiplication means 18 and 24, addition means 19 and 25, delay means 20 and 26, FFT window position correction means 21, equalization residual calculation means 22, and convolution calculation means 27. The convolution operation means 27 includes a convolution operation stage 28 and a filter coefficient setting stage 29.

  Comparing the conventional wraparound canceller 102 shown in FIG. 4 with the wraparound canceller 1 shown in FIG. 2, the wraparound canceller 1 is such that the FFF 202 inputs filter coefficients from the filter coefficient control unit 10, FFT window position correcting means 21, The difference is that the equalization residual calculation means 22, the IFFT means 23, the multiplication means 24, the addition means 25, the delay means 26 and the convolution calculation means 27 generate the filter coefficients of the FFF 202.

  Further, the wraparound canceller 1 includes a frequency conversion unit, an A / D conversion unit, a quadrature demodulation unit, and a synchronous reproduction unit (not shown) before the subtraction unit 201, and a quadrature modulation unit (not shown) after the FFF 202, A D / A converter and a frequency converter are provided.

  A frequency conversion unit (not shown) frequency-converts an IF signal input from the outside and outputs a second IF signal to the A / D conversion unit. The A / D conversion unit receives the second IF signal frequency-converted by the frequency conversion unit, and A / D converts the second IF signal into a digital IF signal using the sampling clock supplied from the synchronous reproduction unit. The digital IF signal is output to the quadrature demodulator. The quadrature demodulator inputs the digital IF signal output from the A / D converter, and converts the digital IF signal into an equivalent baseband of an I component (In-phase component) and a Q component (Quadrature component). The signal is orthogonally demodulated and the obtained equivalent baseband signal is output to the subtractor 201.

  Note that the A / D converter and the orthogonal demodulator may be configured in reverse order. That is, an orthogonal demodulator is provided after the frequency converter, and an A / D converter is provided after the orthogonal demodulator. In this case, the quadrature demodulator inputs the second IF signal frequency-converted by the frequency converter, performs analog quadrature demodulation on the second IF signal to an equivalent baseband signal of I component and Q component, and the obtained I component signal and Each Q component signal is output to the A / D converter. The A / D conversion unit is composed of two A / D conversion units, which respectively input I-component and Q-component equivalent baseband signals and A / D-convert them into digital I and Q signals. The synchronous reproduction unit receives the equivalent baseband signal from the FFF 202 and reproduces the sampling clock and the carrier frequency.

  The subtracting unit 201 inputs the equivalent baseband signal after quadrature demodulation and the wraparound replica signal from the FBF 203, and subtracts the wraparound replica signal from the equivalent baseband signal and outputs the result. That is, the subtraction unit 201 cancels the wraparound.

  The FFF 202 receives the equivalent baseband signal whose wraparound is canceled from the subtraction unit 201, receives the filter coefficient from the filter coefficient control unit 10, and performs filter processing on the equivalent baseband signal. Here, the equivalent baseband signal (filtered equivalent baseband signal) from the FFF 202 is divided into four, and a quadrature modulation unit (not shown), a synchronous reproduction unit (not shown), an FBF 203, and a filter coefficient control unit provided in the subsequent stage of the FFF 202 10 respectively.

  A quadrature modulation unit (not shown) provided at the subsequent stage of the FFF 202 receives the equivalent baseband signal from the FFF 202, performs quadrature modulation, converts the digital signal into a digital IF signal, and outputs the digital IF signal to the D / A conversion unit. A D / A conversion unit (not shown) receives the digital IF signal from the quadrature modulation unit, performs D / A conversion on the second IF signal, and outputs the second IF signal to the frequency conversion unit. The frequency converter receives the second IF signal from the D / A converter, converts the frequency into an IF signal, and outputs the IF signal to the outside.

  The FBF 203 inputs a second equivalent baseband signal among the equivalent baseband signals distributed in four stages after the FFF 202, inputs a filter coefficient from the filter coefficient control unit 10, and uses the filter coefficient to input the equivalent baseband signal. The band signal is subjected to filter processing, a transmission characteristic of a wraparound loop including a wraparound propagation path is realized, a wraparound replica signal is generated, and output to the subtraction unit 201.

  The filter coefficient control unit 10 inputs the third equivalent baseband signal among the equivalent baseband signals distributed in four stages in the subsequent stage of the FFF 202, generates the filter coefficient of the FFF 202 and the filter coefficient of the FBF 203, and outputs the filter coefficient to each filter. Output. Hereinafter, the configuration of the filter coefficient control unit 10 will be described.

  The effective symbol period extraction means 11 of the filter coefficient control unit 10 receives an equivalent baseband signal from the FFF 202, and corresponds to an effective symbol period in one OFDM transmission symbol period based on the symbol timing input from the synchronous reproduction unit (not shown). The signal is extracted, and the obtained OFDM signal in the time domain of the effective symbol period is output to the FFT means 12. The FFT means 12 receives the OFDM signal in the time domain of the effective symbol period from the effective symbol extraction means 11, and fast Fourier transforms the OFDM signal in the time domain into a carrier symbol that is an OFDM signal in the frequency domain. The symbol is output to channel response estimation means 13.

  The channel response estimation means 13 receives the carrier symbol from the FFT means 12 and calculates the channel response after the wraparound cancellation from the carrier symbol. Details of the channel estimation calculation process will be described later. Here, the channel response output from the channel response estimating means 13 is divided into two, one being supplied to the LPF means 14 and the other being supplied to the dividing means 15.

  The LPF means 14 receives the channel response from the channel response estimation means 13, extracts the low frequency component of the channel response, and outputs it. Here, the low frequency component of the channel response output from the LPF means 14 is divided into two, one being supplied to the dividing means 15 and the other being supplied to the FFT window position correcting means 21.

  The dividing means 15 inputs the channel response from the channel response estimating means 13, receives the low frequency component of the channel response from the LPF means 14, and divides the channel response by the low frequency component for each subcarrier of the OFDM signal. Remove multipath distortion components from the channel response. Then, the channel response from which the multipath distortion component has been removed is output to the cancellation residual calculation means 16. Details of the LPF process and the process of dividing the channel response by its low frequency component will be described later.

  The cancellation residual calculation means 16 inputs the channel response from which the multipath distortion component has been removed from the division means 15, the transmission characteristics of the wraparound loop including the actual wraparound propagation path, the output from the FFF 202 to the input of the subtraction unit 201. A cancellation residual (frequency domain signal) that is a difference between the transmission characteristics and the transmission characteristics of the FBF 203 is calculated and output to the IFFT means 17. Details of the cancellation residual calculation process will be described later.

  The IFFT unit 17 receives the cancellation residual from the cancellation residual calculation unit 16, performs inverse fast Fourier transform on the cancellation residual in the frequency domain into an impulse response of the cancellation residual in the time domain, and outputs the result to the coefficient extraction unit 30. . The coefficient cutout unit 30 inputs an impulse response of the cancellation residual from the IFFT unit, cuts out a coefficient in a predetermined time range, and outputs it to the multiplication unit 18.

  The multiplying unit 18 receives the impulse response of the cancellation residual after clipping from the coefficient clipping unit 30, multiplies the coefficient by a preset value μ, and outputs the result to the adding unit 19. Here, the preset value μ is a value in the range of 0 <μ <1, and the following performance with respect to the fluctuation of the transmission characteristic of the wraparound loop including the wraparound propagation path realized by the FBF 203 and its estimation accuracy It is an adaptive parameter for adjusting the trade-off between the two. When μ is set to a large value, the fluctuation tracking performance of the sneak canceller in a sneak fluctuation environment is improved. On the other hand, when μ is set to a small value, the estimation accuracy of the transmission characteristics of the wraparound loop in an environment with little fluctuation is improved.

  The addition means 19 receives the impulse response of the cancellation residual after multiplication by the constant μ from the multiplication means 18, inputs the filter coefficient before the unit coefficient update time from the delay means 20, and the impulse response of the cancellation residual to this filter coefficient. Are added and output as new filter coefficients. Here, the filter coefficient output from the adding means 19 is divided into two, one being supplied to the FBF 203 and the other being supplied to the delay means 20. The delay means 20 receives one of the filter coefficients from the adding means 19, holds the filter coefficient and delays it until the next coefficient update time (during the unit coefficient update time), and sends it to the adding means 19 after the delay time elapses. Output. Details of the inverse fast Fourier transform by the IFFT unit 17, the multiplication process by the multiplication unit 18, the addition process by the addition unit 19, and the delay process by the delay unit 20 will be described later.

  The FFT window position correcting means 21 receives the low frequency component of the channel response from the LPF means 14 and calculates the effective symbol period and the period extracted by the effective symbol period extracting means 11 included in the low frequency component of the channel response. The phase rotation component due to the gap is corrected, and a channel response including only the characteristics of the transmission path is output.

  The equalization residual calculation means 22 receives the channel response in which the phase rotation due to the FFT window position shift output from the FFT window position correction means 21 is input, and is realized by the equalization processing to be actually performed and the FFF 202. The equalization residual, which is the difference between the current equalization process, is calculated and output to the IFFT means 23. Details of the process of calculating the equalization residual (frequency domain signal) will be described later.

  The IFFT unit 23 receives the equalization residual from the equalization residual calculation unit 22, performs inverse fast Fourier transform on the impulse response of the equalization residual, and outputs the result to the coefficient cutout unit 31. The coefficient cutout unit 31 receives the impulse response of the equalization residual from the IFFT unit, cuts out a coefficient in a predetermined time range, and outputs it to the multiplication unit 24. The multiplying unit 24 receives the impulse response of the equalized residual after clipping from the coefficient clipping unit 31, multiplies the impulse response by a preset value, and outputs the result to the adding unit 25.

  The adding means 25 receives the impulse response of the equalization residual from the multiplication means 24, inputs the filter coefficient before the unit coefficient update time from the delay means 26, and adds the impulse response of the equalization residual to this filter coefficient. And output as a new filter coefficient. Here, the filter coefficients output from the adding means 25 are divided into two, one being supplied to the convolution calculating means 27 and the other being supplied to the delay means 26. The delay means 26 receives one of the two distributed filter coefficients output from the adding means 25, holds the filter coefficient, delays it until the next coefficient update time (during unit coefficient update time), and the delay time After the elapse, the data is output to the adding means 25. Details of the inverse fast Fourier transform by the IFFT means 23, the multiplication processing by the multiplication means 24, the addition processing by the addition means 25, and the delay processing by the delay means 26 will be described later.

  A convolution operation stage 28 of the convolution operation means 27 receives a new filter coefficient from the addition means 25, inputs a filter coefficient preset by the filter coefficient setting stage 29, performs a convolution operation on these filter coefficients, and sends them to the FFF 202. Output. Details of the convolution operation processing will be described later.

  Next, in the case where the wraparound canceller 1 shown in FIG. 2 is applied to ISDB-T (Integrated Services Digital Broadcasting-Terrestrial) terrestrial digital broadcasting, the channel response estimation means 13, LPF means 14, LPF means 14, Division means 15, cancellation residual calculation means 16, FFT window position correction means 21, equalization residual calculation means 22, IFFT means 17, 23, multiplication means 18, 24, addition means 19, 25, delay means 20, 26, The operation of the convolution calculator 27 will be described. However, the following is a description of the principle, and basic portions such as frequency conversion, A / D, D / A, quadrature modulation / demodulation, and a transmission / reception unit are well-known techniques, and thus description thereof is omitted. In addition, it is assumed that synchronous reproduction is realized with sufficient accuracy.

[Channel estimation]
First, details of the operation of the channel response estimation means 13 will be described. As described above, the channel response estimation means 13 receives the carrier symbol and calculates the channel response after the wraparound cancellation from the carrier symbol.

  FIG. 9 shows an SP (scattered pilot) assigned to a specific subcarrier of a specific symbol in the ISDB-T system and DVB-T (Digital Video Broadcasting Terrestrial) system, which are broadcasting systems for digital terrestrial television broadcasting. FIG. In FIG. 9, SP is indicated by a black circle, and other carrier symbols such as data symbols are indicated by white circles. Since the amplitude and phase of the SP are determined in advance, the sneak canceller 1 on the receiving side can generate the same signal.

を満足する。但し、ｍｏｄは剰余を示す。以下、式（1）を満足するｉ，ｋをそれぞれｉ_ｐ，ｋ_ｐとする。 FIG. 7 is a block diagram showing a first configuration of the channel response estimation means 13 shown in FIG. The channel response estimation unit 13-1 includes an SP signal extraction stage 31, a reference SP signal generation stage 32, a division stage 33, and an interpolation stage 34. For a subcarrier to which an SP is assigned, if the symbol number is i and the subcarrier number is k,

Satisfied. However, mod indicates a remainder. Hereinafter, i and k satisfying the expression (1) are assumed to be i _p and k _p , respectively.

  The SP signal extraction stage 31 receives a carrier symbol, extracts an SP from the carrier symbol, and outputs it to the division stage 33. The reference SP signal generation stage 32 generates a reference SP having the same amplitude and phase as the SP at the time of signal generation in the ISDB-T modulator, and outputs it to the division stage 33. The division stage 33 divides the SP extracted by the SP signal extraction stage 31 by the reference SP generated by the reference SP signal generation stage 32 and outputs the result to the interpolation stage 34. The interpolation stage 34 interpolates the channel response generated by the division stage 33.

Here, if the SP extracted in the SP signal extraction stage 31 is Y _{ip, kp} and the reference SP generated in the reference SP signal generation stage 32 is X _{ip, kp} , the symbol number i _p and subcarrier number k _p The channel response F _{ip, kp} is expressed by the following equation.

  The channel response estimation unit 13-1 uses the SP employed in the ISDB-T method as a reference signal for calculating the channel response, but is not limited to this. Another carrier symbol that has a known amplitude and phase and can be generated by the wraparound canceller 1 on the receiving side may be used as the reference signal.

  FIG. 8 is a block diagram showing a second configuration of the channel response estimation means 13 shown in FIG. The channel response estimation means 13-2 includes a channel equalization stage 40, a determination stage 46, and a division stage 47. The channel equalization stage 40 includes an SP signal extraction stage 41, a reference SP signal generation stage 42, a division stage 43, an interpolation stage 44, and a division stage 45. When comparing the channel response estimation means 13-1 shown in FIG. 7 with the channel response estimation means 13-2 shown in FIG. 8, the SP signal extraction stages 31, 41, the reference SP signal generation stages 32, 42, the division stage 33, 43 and the interpolation stages 34 and 44, but the channel response estimation means 13-2 includes a division stage 45, a determination stage 46, and a division stage 47 in addition to the channel response estimation means 13-1. It differs in that it has.

  The channel equalization stage 40 receives a carrier symbol from the FFT means 12, and the division stage 45 outputs a carrier symbol other than SP (hereinafter referred to as a data symbol) of the carrier symbol by the interpolation stage 44. The channel equalization is performed by dividing the channel response F. Channel equalization stage 40 outputs the data symbols after channel equalization to decision stage 46.

ここで、Ｄ_ｋはＩＳＤＢ-Ｔ変調器における信号生成時のサブキャリヤ番号ｋのキャリヤシンボル（以下、送信シンボル）、Ｎ_ｋは雑音を示す。式（３）によれば、Ｚ_ｋは、第２項により信号点Ｄ_ｋを中心に分散することがわかる。 Data symbol Z _k after channel equalization can be expressed as:

Here, D _k is a carrier symbol (hereinafter referred to as a transmission symbol) of subcarrier number k at the time of signal generation in the ISDB-T modulator, and N _k is noise. According to Equation (3), it can be seen that Z _k is distributed around the signal point D _k by the second term.

  Next, the decision stage 46 inputs the data symbol after channel equalization from the channel equalization stage 40, performs threshold judgment, and is a known value having the smallest Euclidean distance in the signal space from the carrier symbol after equalization. The signal point is output to the division stage 47 as the estimated value of the transmission symbol.

ｄｅｃ（ｙ）は、しきい値判定の関数であり、信号空間においてデータシンボルＹに最も近い送信信号を返す。 Here, the estimated value of the transmission symbol can be expressed by Equation (4).

Dec (y) is a threshold determination function and returns the transmission signal closest to the data symbol Y in the signal space.

つまり、除算段４７は、キャリヤシンボルを、式（４）に示す送信シンボルの推定値である基準信号で除算し、その結果のチャネル応答を出力する。 Assuming that there is no determination error in equation (4), the estimated value of the transmission symbol shown in equation (4) is equal to the transmission symbol D _{k at} the frequency of the subcarrier of subcarrier number k. Therefore, the estimated value of the transmission symbol shown in Equation (4) can be used as a reference signal in the same manner as SP, and the channel response can be obtained as shown in Equation (5).

That is, the division stage 47 divides the carrier symbol by the reference signal that is the estimated value of the transmission symbol shown in Equation (4), and outputs the resulting channel response.

[Channel response delay time separation]
Next, details of the operations of the LPF means 14 and the dividing means 15 will be described. As described above, the LPF means 14 extracts the low frequency component of the channel response, and the dividing means 15 divides the channel response by the low frequency component for each subcarrier of the OFDM signal, and multipath from the channel response. The distortion component is removed.

  FIG. 10 is a block diagram showing a first configuration of the LPF means 14 shown in FIG. This LPF means 14-1 inputs the channel response (frequency characteristic), extracts the low frequency component, and outputs it. Here, the low frequency component of the channel response is a low frequency component of the channel response (frequency characteristic) waveform, and corresponds to a response in a certain delay time range including the main wave on the delay profile obtained by IFFT of the frequency characteristic. To do.

  FIG. 11 is a block diagram showing a second configuration of the LPF means 14 shown in FIG. The LPF means 14-2 includes an IFFT stage 51, a window function stage 52, and an FFT stage 53. The IFFT stage 51 receives the channel response from the channel response estimation means 13, performs inverse fast Fourier transform to a delay profile that is a time domain representation of the channel response, and outputs the result to the window function stage 52. The window function stage 52 receives the delay profile from the IFFT stage 51, multiplies the window function, extracts only the components in the delay time range that can be equalized by the FFF 202, and outputs them to the FFT stage 53. The FFT stage 53 inputs the window function multiplication result from the window function stage 52, performs fast Fourier transform again on the channel response which is the frequency domain representation, and outputs the low-frequency component of the channel response obtained thereby.

ここで、式（６）の左辺は、サブキャリヤ番号ｋについて、主波とＦＦＦ２０２によって等化可能なマルチパス成分のみが含まれるチャネル応答を示す。 Such processing by the LPF units 14-1 and 14-2 can be expressed by Expression (6).

Here, the left side of Expression (6) indicates a channel response including only the multipath component that can be equalized by the main wave and the FFF 202 with respect to the subcarrier number k.

ここで、式（７）の左辺は、サブキャリヤ番号ｋについて、主波とＦＢＦ２０３によってそのレプリカを生成可能な遅延時間範囲の回り込み成分のみが含まれるチャネル応答を示す。 Division means 15 divides channel response F _k from channel response estimation means 13 by the left side of equation (6) from LPF means 14. That is, it can be expressed by equation (7).

Here, the left side of Equation (7) indicates a channel response including only the wraparound component in the delay time range in which the replica can be generated by the main wave and the FBF 203 for the subcarrier number k.

  FIG. 12 is a diagram expressing the processing contents of the LPF means 14 and the dividing means 15 by a delay profile which is a time domain expression of the channel response. FIG. 12A shows a delay profile of the channel response input by the LPF unit 14 and the dividing unit 15. (B) shows the delay profile of the low frequency component of the channel response which is the output signal of the LPF means 14. (C) shows a delay profile obtained by removing multipath distortion components from the channel response, which is an output signal of the dividing means 15.

  From FIG. 12, the channel response (a) indicates that the main wave and the signal (b) including only the multipath component in the delay time range that can be equalized by the FFF 202, and the delay time that can generate the replica by the main wave and the FBF 203. It can be seen that the signal (c) including only the sneak component of the range is separated. That is, the purpose of the LPF means 14 and the dividing means 15 is to use the channel response signal shown in (a), the main wave and multipath component signals shown in (b) (signals for generating the filter coefficients of the FFF 202). ) And the signal of the main wave and the sneak component shown in (c) (the signal for generating the filter coefficient of the FBF 203). For details, refer to Patent Document 9 described above.

[Calculation of wraparound cancellation residual, update of filter coefficient of FBF]
Next, details of the operations of the cancellation residual calculation unit 16, the IFFT unit 17, the coefficient extraction unit 30, the multiplication unit 18, the addition unit 19 and the delay unit 20 will be described. As described above, the cancellation residual calculation unit 16 inputs the channel response from which the multipath distortion component has been removed, calculates the cancellation residual of the sneak path, and the IFFT unit 17 cancels the frequency domain cancellation residual. Inverse fast Fourier transform to cancel residual impulse response. The coefficient cutout unit 30 cuts out a coefficient in a predetermined time range from the impulse response of the cancellation residual. The multiplying means 18 multiplies the extracted cancellation residual impulse response by a preset value μ, and the adding means 19 adds the cancellation residual impulse response to the filter coefficient before the unit coefficient update time. The delay means 20 holds the input filter coefficient and delays it until the next coefficient update time (during the unit coefficient update time).

Assuming that the wraparound cancellation residual relating to the subcarrier number k is δW _k , the cancellation residual calculated by the cancellation residual calculation means 16 can be expressed by equation (8).

ここで、ｎは任意の離散時間を示す。 The IFFT means 17 calculates the impulse response δw _(n) of the wraparound cancellation residual when the frequency domain cancellation residual δW _k is converted into the time domain by IFFT processing.

Here, n represents an arbitrary discrete time.

ここでμ_ｂは、雑音抑圧のための適応係数を示し、乗算手段１８により乗算される定数である。尚、フィルタ係数の更新にあたっては、特開２００１−２３７７４９、特開２０００−３４１２４２等の手法を用いることができる。 Assuming that the current time is i, an arbitrary discrete time is n, and the filter coefficient of the FBF 203 is w (i, n), the multiplying means 18, the adding means 19 and the delay means 20 are the filter coefficients w (i, n) of the FBF 203. ) Is updated by equation (10).

Here, μ _b represents an adaptive coefficient for noise suppression, and is a constant multiplied by the multiplying means 18. In updating the filter coefficients, methods such as Japanese Patent Laid-Open Nos. 2001-237749 and 2000-341242 can be used.

[Calculation of equalization residual, update of FFF202 filter coefficient]
Next, details of operations of the equalization residual calculation unit 22, the IFFT unit 23, the multiplication unit 24, the coefficient cutout unit 31, the addition unit 25, and the delay unit 26 will be described. The equalization residual calculation means 22 calculates the equalization residual by inputting the low frequency component of the channel response including only the characteristics of the transmission path of the upper station signal, and the IFFT means 23 calculates the frequency domain equalization residual. The difference is inverse fast Fourier transformed into an impulse response of the equalization residual in the time domain. The coefficient cutout unit 31 cuts out a coefficient in a predetermined time range. The multiplication unit 24 multiplies the impulse response of the equalization residual by a preset value, and the addition unit 25 adds the impulse response of the equalization residual to the filter coefficient before the unit coefficient update time, The delay means 26 holds the filter coefficient and delays it until the next coefficient update time (during the unit coefficient update time).

If the equalization residual for the subcarrier number k is δH _k , the equalization residual calculated by the equalization residual calculation means 22 can be expressed by equation (11).

The IFFT means 23 calculates the impulse response δh (n) of the equalization residual when the frequency domain equalization residual δH _k is converted into the time domain by IFFT processing.

ここでμ_ｆは、雑音抑圧のための適応係数を示し、乗算手段２４により乗算される定数である。 Assuming that the current time is i, an arbitrary discrete time is n, and the filter coefficient of the FFF 202 is h (i, n), the multiplication means 24, the addition means 25, and the delay means 26 express the filter coefficients of the FFF 202 by the equation (13). ) To update.

Here, μ _f represents an adaptive coefficient for noise suppression, and is a constant multiplied by the multiplication unit 24.

[Convolution operation]
Next, details of the operation of the convolution calculator 27 will be described. In the convolution operation means 27, the convolution operation stage 28 receives a new filter coefficient from the addition means 25, receives a filter coefficient preset by the filter coefficient setting stage 29, and performs a convolution operation between these filter coefficients. The purpose of the processing by the convolution calculator 27 is to limit the bandwidth so that the bandwidth of the FFF 202 does not exceed the bandwidth of either the wraparound loop or the cancel loop.

ここで、ｌは非適応狭帯域フィルタｇ（ｎ）のフィルタ長（タップ長）を示す。 The convolution operation means 27 is a non-adaptive filter, and a filter coefficient of a narrowband filter whose bandwidth does not exceed any of the wraparound loop and the cancel loop is determined in advance as g (n), and the expression (13 The time domain convolution operation is performed on the filter coefficient of the FFF 202 generated by the following equation (14).

Here, l indicates the filter length (tap length) of the non-adaptive narrowband filter g (n).

  When the convolution operation means 27 that is such a non-adaptive narrowband filter is connected in cascade to the FFF 202, a processing delay due to the wraparound canceller becomes large. However, by convolving the filter coefficients in the time domain in this way, an increase in device delay due to the wraparound canceller can be prevented, and the bandwidth of the FFF 202 does not exceed any bandwidth of the wraparound loop or the cancellation loop. Can be.

  As described above, according to the wraparound canceller 1 shown in FIG. 2, when the filter coefficient control unit 10 generates the filter coefficient of the FFF 202, the LPF means 14 extracts the low frequency component of the channel response, The FFF 202 generates a channel response including only the characteristics of the multipath transmission path within the delay time range that can be equalized. Thereby, the filter coefficient of the FFF 202 can be equalized by the main wave and the FFF 202 by the FFT window position correcting means 21, the equalization residual calculating means 22, the IFFT means 23, the multiplying means 24, the adding means 25, and the delay means 26. It is generated from the channel response including only the characteristics of the transmission path due to the multipath within the delay time range. Therefore, even in an environment where a multipath component having a small delay time difference from the main wave exists in the propagation path from the upper station, the FFF 202 can equalize distortion due to the multipath.

  Further, according to the wraparound canceller 1 shown in FIG. 2, when the filter coefficient control unit 10 generates the filter coefficient of the FBF 203, the dividing unit 15 divides the channel response by the low frequency component, and depending on the FBF 203, A channel response that does not contain multi-distortion components within the range of delay time that cannot generate replicas is generated. That is, a channel response from which multipath distortion components are removed is generated. Accordingly, the filter coefficient of the FBF 203 is generated from the channel response from which the multipath distortion component is removed by the cancellation residual calculation unit 16, the IFFT unit 17, the multiplication unit 18, the addition unit 19, and the delay unit 20. Become. Therefore, even in an environment where a multipath component with a small delay time difference from the main wave exists in the transmission path from the upper station, the FFF 202 equalizes the distortion due to the multipath, and the FBF 203 and the subtraction unit 201 cancel the wraparound. can do.

  Next, the effect of the wraparound canceller 1 shown in FIG. 2 will be described. FIG. 13 is a diagram illustrating an example of a delay profile. Hereinafter, an example of the operation of the sneak canceller in the propagation environment represented by FIG. 13 is shown in FIG. 3, the sneak canceller 103 assumed from the prior art shown in FIG. 4, and FIG. 2. Each wraparound canceller 1 according to the embodiment of the present invention will be described.

  FIG. 14 is a diagram showing an example of filter coefficients of the FFF 202 and the FBF 203 when the conventional wraparound canceller 101 shown in FIG. 3 is used. In order to simplify the description, the filter coefficient after one update of the filter coefficient is shown below. From FIG. 14, it can be seen that the filter coefficient of the FFF 202 is generated so as to equalize the multipath, but the filter coefficient of the FBF 203 is not generated so as to cancel the wraparound. This is because the filter coefficient of the FBF 203 is generated with a coefficient for canceling a component that is caused by multipath distortion and does not exist in the actual sneak path, so that the sneak replica generated by the FBF 203 is subtracted. This is because they are not canceled out in the part 201. That is, the wraparound is generated by the wraparound canceller 101 itself.

  FIG. 15 is a diagram illustrating an example of filter coefficients of the FFF 202 and the FBF 203 when the wraparound canceller 103 assumed from the related art illustrated in FIG. 5 is used. From FIG. 15, the filter coefficient of the FBF 203 is generated so as to cancel the wraparound, and the characteristic of the wraparound propagation path is correctly estimated. However, the filter coefficient of the FFF 202 is generated so as to equalize the multipath. You can see that it is not. This is because the filter coefficients of the FFF 202 are generated from the channel response after removing the multipath components in the delay time range that can be equalized by the FFF 202, so that the multipath components in the delay time range that can be equalized by the FFF 202 are This is because filter coefficients for equalization are not generated.

  FIG. 16 is a diagram illustrating an example of filter coefficients of the FFF 202 and the FBF 203 when the wraparound canceller 1 according to the embodiment of the present invention illustrated in FIG. 2 is used. From FIG. 16, the filter coefficients of the FFF 202 are correctly generated so as to equalize the multipath, and the filter coefficients of the FBF 203 are correctly generated so as to generate a wraparound replica and cancel the wraparound. Recognize.

[Repeater]
Next, the relay device will be described. FIG. 6 is a block diagram showing a configuration of a relay apparatus using the wraparound canceller 1 shown in FIG. The relay device 9 includes a reception antenna 2, a reception filter 3, a reception unit 4, a sneak canceller 1, a transmission unit 5, a PA (amplification unit) 6, a transmission filter 7, and a transmission antenna 8.

  The relay apparatus 9 of the broadcast wave relay station provided with the wraparound canceller 1 shown in FIG. 2 receives the desired wave transmitted from the upper station by the receiving antenna 2. The reception filter 3 inputs the reception signal output from the reception antenna 2 via a feeder cable, and removes unnecessary signal components outside the frequency band of the desired wave. The reception (conversion) unit 4 receives the output signal of the reception filter 3, performs AGC amplification so that the output level becomes constant, and then performs frequency conversion to generate and output an IF signal. As the center frequency of this IF signal, 37.15 MHz is generally used.

  The wraparound canceller 1 receives an IF signal from the receiving unit 4. The frequency conversion unit of the wraparound canceller 1 frequency-converts the IF signal to generate a second IF signal. As the center frequency of the second IF signal, 512/63 (8.127689) MHz is generally used. The A / D converter receives the second IF signal from the frequency converter, performs A / D conversion, and outputs a digital IF signal. The quadrature demodulation unit receives the digital IF signal from the A / D conversion unit, performs quadrature demodulation processing, converts the digital IF signal into an equivalent baseband signal, and outputs it. The subtracting unit receives the equivalent baseband signal from the quadrature demodulating unit, synthesizes the wraparound replica from the FBF in reverse phase, and outputs the resultant to the FFF. The FFF equalizes the multipath component and outputs it as an equivalent baseband signal that has been subjected to cancellation of wraparound and multipath equalization. The quadrature modulation unit inputs an equivalent baseband signal, performs quadrature modulation processing, converts it to a digital IF signal, and outputs the digital IF signal. The D / A converter receives the digital IF signal from the quadrature modulator, converts it to a second IF signal, and outputs it. The frequency converter receives the second IF signal from the D / A converter, converts it to an IF signal, and outputs the IF signal.

  The transmission (conversion) unit 5 receives the IF signal from the frequency conversion unit of the wraparound canceller 1, converts the frequency to the RF band, amplifies the signal to a certain level, and outputs it. The PA 6 receives an RF band signal from the transmitter 5 and amplifies the power to obtain a desired output transmission signal. The transmission filter 7 receives an RF band transmission signal from the PA 6 and removes unnecessary radiation components outside the band. The transmission signal from which unnecessary components outside the band are removed by the transmission filter 7 is supplied to the transmission antenna 8 via the feeder cable and radiated as radio waves.

It is a block diagram which shows the outline of the general structure of a wraparound canceller. It is a block diagram which shows the structure of the wraparound canceller by embodiment of this invention. It is a block diagram which shows the 1st structure of the conventional wraparound canceller. It is a block diagram which shows the 2nd structure of the conventional wraparound canceller. It is a block diagram which shows the structure of the wraparound canceller assumed from a prior art. It is a block diagram which shows the structure of the relay apparatus using the wraparound canceller of FIG. It is a block diagram which shows the 1st structure of a channel response estimation means. It is a block diagram which shows the 2nd structure of a channel response estimation means. It is a figure which shows arrangement | positioning of SP. It is a block diagram which shows the 1st structure of a LPF means. It is a block diagram which shows the 2nd structure of a LPF means. It is a figure explaining delay time separation of a channel response. It is a figure which shows the example of a delay profile. It is a figure which shows the example of the filter coefficient by the 1st structure of the conventional wraparound canceller. It is a figure which shows the example of the filter coefficient by the wraparound canceller assumed from a prior art. It is a figure which shows the example of the filter coefficient by the wraparound canceller of FIG.

Explanation of symbols

1,100,101,102,103 wraparound canceller 2 receiving antenna 3 receiving filter 4 receiving unit 5 transmitting unit 6 PA
7 Transmission filter 8 Transmission antenna 9 Relay device 10, 110, 204, 210, 310 Filter coefficient control unit 11, 111, 211, 311 Effective symbol period extraction unit 12, 112, 212, 312 FFT unit 13, 113, 213, 313 Channel response estimation means 14, 214, 314 LPF means 15, 215, 315 Division means 16, 114, 216, 316 Cancel residual calculation means 17, 23, 115, 217, 317 IFFT means 18, 24, 116, 119, 218 , 318, 321 Multiplication means 19, 25, 117, 120, 219, 322 Addition means 20, 26, 118, 121, 220, 320, 323 Delay means 21 FFT window position correction means 22 Equalization residual calculation means 27 Convolution calculation Means 28 Convolution Operation Stage 29 Fill Coefficient setting stage 30, 122, 123, 221, 324, 325 coefficient extraction means 31, 41 SP signal extraction stage 32, 42 reference SP signal generation stage 33, 43, 45, 47 division stage 34, 44 interpolation stage 40 channels, etc. Conversion stage 46 Judgment stage 51 IFFT stage 52 Window function stage 53 FFT stage 201 Subtraction unit 202 FBF
203 FFF

Claims (4)

FBF (Feed-Back Filter) for generating a wraparound replica for canceling the wraparound by synthesizing in reverse phase with the input OFDM (Orthogonal Frequency Division Multiplexing) signal, and distortion due to the multipath of the signal after the wraparound cancellation In a wraparound canceller comprising at least a FFF (Feed-Forward Filter) for equalizing and a filter coefficient control unit for generating a filter coefficient for controlling the FFF,
The filter coefficient control unit
An effective symbol period extraction unit that extracts a period corresponding to an effective symbol period of the OFDM signal input from the FFF;
FFT means for converting an OFDM signal input from the effective symbol period extraction unit into a carrier symbol that is a frequency domain signal by FFT (Fast Fourier Transform);
Channel response estimation means for estimating a channel response from a carrier symbol input from the FFT means;
LPF (Low Pass Filter) means for extracting a low frequency component including a multipath component in a delay time range that can be equalized by a main wave from an upper station and FFF from a channel response input from the channel response estimation means; ,
FFT window position correcting means for correcting a phase rotation component caused by a shift of the FFT window position included in the low frequency component of the channel response input from the LPF means;
Equalization residual calculation means for converting a low frequency component of the channel response input from the FFT window position correction means into an equalization residual of multipath distortion;
IFFT means that performs an IFFT (Inverse Fast Fourier Transform) process on the equalization residual input from the equalization residual calculation means, and converts it into an impulse response of the equalization residual;
Clipping means for cutting out a coefficient in a predetermined time range from the impulse response of the equalization residual input from the IFFT means;
Multiplication means for multiplying the impulse response after clipping input from the coefficient clipping means by a predetermined constant;
Delay means for holding the filter coefficient of the FFF for a unit update time;
Adding means for adding the impulse response of the equalization residual input from the multiplication means and the filter coefficient of the FFF before the unit update time input from the delay means to generate a new FFF filter coefficient;
A wraparound canceller characterized by comprising: FBF that generates a wraparound replica for canceling wraparound by combining with an input OFDM signal in reverse phase, FFF that equalizes distortion due to multipath of the signal after wraparound cancellation, and controls the FBF In a wraparound canceller comprising at least a filter coefficient for generating a filter coefficient for controlling the FFF and a filter coefficient control unit for controlling the FFF,
The filter coefficient control unit
An effective symbol period extraction unit that extracts a period corresponding to an effective symbol period of the OFDM signal input from the FFF;
FFT means for converting the OFDM signal input from the effective symbol period extraction unit into a carrier symbol which is a frequency domain signal by FFT;
Channel response estimation means for estimating a channel response from a carrier symbol input from the FFT means;
LPF means for extracting a low frequency component including a multipath component in a delay time range that can be equalized by a main wave from an upper station and FFF from a channel response input from the channel response estimation means;
The multipath distortion component of the channel response is removed by dividing the channel response input from the channel response estimation unit by the low frequency component of the channel response input from the LPF unit for each subcarrier of the OFDM signal. Dividing means;
A cancellation residual calculation means for calculating a wraparound cancellation residual from the channel response from which the multipath distortion component input from the division means is removed;
IFFT means for performing IFFT processing on the wraparound cancellation residual input from the cancellation residual calculation means and converting it into an impulse response of the wraparound cancellation residual;
A cutout unit that cuts out a coefficient in a predetermined time range from the impulse response of the wraparound cancellation residual input from the IFFT unit;
Multiplying means for multiplying the impulse response of the cancellation residual after clipping input from the clipping means by a predetermined constant;
Delay means for holding FBF filter coefficients for a unit update time;
Adding means for adding the impulse response of the wraparound residual input from the multiplication means and the filter coefficient of the FBF before the unit update time input from the delay means to generate a new FBF filter coefficient;
FFT window position correcting means for correcting a phase rotation component caused by a shift of the FFT window position included in the low frequency component of the channel response input from the LPF means;
Equalization residual calculation means for converting a low frequency component of the channel response input from the FFT window position correction means into an equalization residual of multipath distortion;
IFFT means that performs an IFFT (Inverse Fast Fourier Transform) process on the equalization residual input from the equalization residual calculation means, and converts it into an impulse response of the equalization residual;
Clipping means for cutting out a coefficient in a predetermined time range from the impulse response of the equalization residual input from the IFFT means;
Multiplication means for multiplying the impulse response after clipping input from the coefficient clipping means by a predetermined constant;
Delay means for holding the filter coefficient of the FFF for a unit update time;
Adding means for adding the impulse response of the equalization residual input from the multiplication means and the filter coefficient of the FFF before the unit update time input from the delay means to generate a new FFF filter coefficient;
A wraparound canceller characterized by comprising: In the wraparound canceller according to claim 1 or 2,
The filter coefficient control unit further includes a convolution operation unit that convolves a predetermined filter coefficient with the filter coefficient of the new FFF generated by the adding unit, and generates a filter coefficient for controlling the FFF. A wraparound canceller characterized by comprising.   The relay apparatus using the wraparound canceller according to any one of claims 1 to 3.

Images