JP3528819B2 - Switching power supply circuit - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply in various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a switching power supply circuit that employs a switching converter such as a flyback converter or a forward converter is widely known. Since the switching operation waveforms of these switching converters are rectangular waves, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It is known that there is a limit to the improvement of power conversion efficiency. Therefore, the applicant has previously proposed various switching power supply circuits using various resonant converters.
The resonant converter can easily obtain high power conversion efficiency, and the switching operation waveform has a sinusoidal waveform to realize low noise. It also has the advantage that it can be configured with a relatively small number of parts.

The circuit diagram of FIG. 13 shows an example of a switching power supply circuit as a prior art which can be constructed based on the invention previously proposed by the present applicant. As a basic configuration of the power supply circuit shown in this figure, a voltage resonance type converter is provided as a primary side switching converter.

In the power supply circuit shown in this figure, a bridge rectifier circuit Di and a smoothing capacitor Ci generate a rectified and smoothed voltage Ei corresponding to a level of one time the AC input voltage VAC from a commercial AC power supply (AC input voltage VAC). .

As the voltage resonance type converter which receives and connects the rectified and smoothed voltage Ei (DC input voltage), a single-ended system using one stone is adopted. The drive system is a self-excited type. In this case, the switching element Q1 forming the voltage resonance type converter is
A high voltage bipolar transistor (BJT; junction type transistor) is selected. A primary side parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. A series circuit of a clamp diode DD and a resistor RD is connected between the base and the emitter. Here, the parallel resonance capacitor Cr
Forms a primary side parallel resonance circuit together with a leakage inductance L1 obtained in the primary winding N1 of the insulating converter transformer PIT, whereby an operation as a voltage resonance type converter can be obtained. Then, with respect to the base of the switching element Q1, the drive winding NB-resonance capacitor CB-base current limiting resistor RB
Is connected to the self-excited oscillation drive circuit. The switching element Q1 is switching-driven by being supplied with a base current based on an oscillation signal generated by the self-excited oscillation drive circuit. It should be noted that, at the time of start-up, it is started by the start-up current flowing from the line of the rectified and smoothed voltage Ei through the start-up resistor Rs to the base.

The orthogonal control transformer PRT is constructed by winding a control winding Nc such that the winding directions of the drive winding NB and the current detection winding ND are orthogonal to each other. It is provided to control the switching frequency of the primary side voltage resonance type converter as described later. The structure of the orthogonal control transformer PRT will be described later.

The insulating converter transformer PIT is provided for transmitting the switching output of the switching converter obtained on the primary side to the secondary side. This isolation converter transformer PIT has a primary winding N with respect to the EE type core.
1 and the secondary winding N2 are divided and wound, and a gap is formed for the central magnetic leg so that a loose coupling state with a required coupling coefficient can be obtained and a saturation state can be obtained. I try to make it difficult.

The primary winding N1 of the insulating converter transformer PIT has one end connected to the DC input voltage (rectified and smoothed voltage Ei) line via the current detection winding ND, and the other end connected to the collector of the switching element Q1. Has been done.
The switching element Q1 performs switching with respect to the DC input voltage.
The switching output of the switching element Q1 is supplied to the primary winding N1 and the current detection winding ND, and an alternating voltage having a cycle corresponding to the switching frequency is generated.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonant capacitor C2 are used. A parallel resonant circuit is formed. With this parallel resonant circuit,
The alternating voltage induced in the secondary winding N2 becomes a resonance voltage.
That is, the voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the parallel resonance circuit for making the switching operation a voltage resonance type is provided on the primary side, and the parallel resonance circuit for obtaining the voltage resonance operation is provided on the secondary side. In the present specification, the switching converter configured such that the resonance circuits are provided for the primary side and the secondary side as described above is also referred to as a “composite resonance type switching converter”.

Insulation converter transformer PIT in this case
On the secondary side, first, the half-wave rectification circuit is formed by connecting the anode of the rectifying diode DO1 to the end of the secondary winding N2 and connecting the cathode to the positive terminal of the smoothing capacitor CO1. Is forming. Depending on this half-wave rectifier circuit, the secondary side DC output voltage EO1 is obtained across the smoothing capacitor CO1. Further, in this case, a tap is provided on the secondary winding N2, and a half-wave rectifying circuit including a rectifying diode D02 and a smoothing capacitor C02 is formed for the tap output as shown in the figure. Then, depending on this half-wave rectification circuit, the secondary side DC output voltage E lower than the secondary side DC output voltage EO1 is obtained.
O2 is obtained. In addition, specifically, the secondary side DC output voltage EO1 = 135V and the secondary side DC output voltage EO2 = 15V.

These secondary side DC output voltages EO1 and EO2 are
It will be supplied to each required load circuit. The secondary side DC output voltage EO1 is branched and output as a detection voltage of the control circuit 1.

In the control circuit 1, resistors R3-R4 are connected in series between the DC output voltage EO1 and the secondary side ground, and the control terminal of the shunt regulator Q3 is connected to this connection point (division point). . The anode of the shunt regulator Q3 is grounded, and the cathode is connected to the line of the secondary side DC output voltage EO2 via the control winding NC of the orthogonal control transformer PRT. Further, here, the cathode of the shunt regulator Q3 is connected to the capacitor C1.
It is connected via 1 to the connection point of resistors R3 and R4. Further, a series connection circuit of a capacitor C3 and a resistor R5 is connected in parallel with the resistor R4.

The control circuit 1 formed by the above-mentioned connection form functions as an error amplifier having the DC output voltage EO1 as a detection input. That is, the DC output voltage EO1 is applied to the resistor R
The voltage divided by 3 and R4 is input to the control terminal of the shunt regulator Q3 as a control voltage. Therefore, in the shunt regulator Q3, a current having a level corresponding to the DC output voltage E01 is caused to flow as the control current Ic to the control winding NC. That is,
The control current level flowing through the control winding NC is variably controlled. By varying the level of the control current flowing through the control winding Nc, the quadrature control transformer PRT is controlled to vary the inductance LB of the drive winding NB. As a result, the resonance frequency of the resonance circuit composed of the drive winding NB and the resonance capacitor CB in the self-excited oscillation drive circuit changes, and the switching frequency of the switching element Q1 is variably controlled. By varying the switching frequency of the switching element Q1 in this manner, the secondary side DC output voltage is controlled to be constant. That is, the power supply is stabilized. Here, in changing the switching frequency, the period during which the main switching element Q1 is off is kept constant, and then the period during which it is on is variably controlled. That is, the conduction angle control for the ON period is performed and the switching frequency control is performed. In this specification, such composite control is referred to as a "composite control method".

FIG. 14 shows operation waveforms of each part under heavy load as the operation of the main part of the power supply circuit having the configuration shown in FIG. Here, the operation on the primary side is mainly shown. Series resonance circuit (NB,
In CB), a resonance operation is performed by the alternating voltage obtained in the drive winding NB, so that a sinusoidal series resonance current I2 is obtained as shown in FIG. 14 (e). Then, as the series resonance current I2 passes through the base current limiting resistor RB, a base current (driving current) IB flows through the base of the switching element Q1 as shown in FIG. 14 (d). The drive current IB causes the switching element Q1 to perform a switching operation.

At this time, the collector current IQ1 flowing through the collector of the switching element Q1 has the waveform shown in FIG. 14 (b). In addition, at both ends of the parallel connection circuit of the switching element Q1 // parallel resonance capacitor Cr,
As shown in (a), the parallel resonance voltage V1 is generated by the action of this parallel resonance circuit. This parallel resonance voltage V1
As shown in the figure, the period TON in which the switching element Q1 is on is 0 level, and a waveform of a sinusoidal pulse is obtained in the period TOFF in which the switching element Q1 is off, which corresponds to the operation as the voltage resonance type.

Further, as the switching element Q1 performs the switching operation at the above timing, the winding current I1 flowing through the primary winding N1 has an alternating waveform according to the switching cycle as shown in FIG. 14 (c). Become.

Here, in the period TON during which the switching element Q1 is turned on, the series resonance current I2 of FIG.
The region of positive polarity corresponds to the region of forward bias current of the drive current IB in FIG. Also, the same period TON
In the region where the series resonance current I2 is negative, the reverse bias current of the drive current IB is obtained. Then, the region of the reverse bias current of the drive current IB in this period TON becomes the accumulation time (tstg) of the switching element Q1.

A low speed damper diode D having a long reverse recovery time is provided between the base and emitter of the switching element Q1.
A series circuit of D and a resistor RD is connected. In the period TOFF in which the switching element Q1 is off, a negative series resonance current I2 flows through the resistor RD → the clamp diode DD → the base current limiting resistor RB → the resonance capacitor CB → the drive winding NB, which is shown in FIG. (G) Damper current I
The waveform is obtained in the period TOFF as D1. Then, when the period TON is started next, the parallel resonance capacitor Cr
Charging / discharging energy flows through the clamp diode DD → the base of the switching element Q1 → the collector, and this becomes a negative damper current (ID) at the start (turn-on) of the period TON. Then, when this period ends, the damper diode DD becomes the region of the reverse recovery time and steeply rises in the positive polarity direction, and thereafter, as shown in the figure, gradually becomes 0 level at the end of the period TON. You can get a good waveform.

Corresponding to the flow of the drive current IB and the damper current ID1 as described above, the base-emitter voltage VBE of the switching element Q1 becomes negative during the period TOFF as shown in FIG. 14 (f). In the period TON, the waveform has a sharp negative peak in the negative polarity in the damper period at the start of the period, and when this is finished, an offset is given to the 0 level at a constant level of positive polarity. is there. This offset level is determined by the resistance value of the resistor RD, for example.

FIG. 15 shows the constant voltage control characteristics of the power supply circuit of FIG. 13 which operates as described above. As the load current Io of the secondary side DC output voltage EO1 changes in the range of 0 to 1.5 A, the control current Ic changes as shown in the figure. In other words, the control is performed such that the control current level is decreased as the load current increases and becomes a heavy load condition, and the secondary side DC output voltage EO1 decreases. As a result, the switching frequency fs is controlled so as to decrease as the heavy load condition is met. In addition, as a measure to deal with the fluctuation of the AC input voltage VAC, the AC input voltage VAC = 120V and VAC = 90.
Although the case of V is shown, the control current Ic is the AC input voltage VAC under the condition when the AC input voltage VAC = 120V.
= 90V, the switching frequency fs is higher under the condition of AC input voltage VAC = 120V than under the condition of AC input voltage VAC = 90V. This is controlled so that the control current Ic is increased when the level of the AC input voltage VAC is increased and the secondary side DC output voltage EO1 is increased, and accordingly the switching frequency fs is also increased. It is shown that it is controlled so as to be raised.

[0022]

By the way, in the above configuration, the orthogonal control transformer PRT has the resonance current detection winding N
It is a saturable reactor in which D, a drive winding NB, and a control winding NC are wound. FIG. 17 shows the orthogonal control transformer PRT.
Shows the structure of. FIG. 17A is an external perspective view for explaining the entire structure, and FIG. 17B is a cross-sectional perspective view for explaining the winding direction of the winding to be wound. FIG. 17
As shown in (a), the orthogonal control transformer PRT is formed by a three-dimensional core 20 in which two double U-shaped cores 21 and 22 made of ferrite are combined.
One of the double U-shaped cores 21 is shown in FIGS.
4 magnetic legs 21a, 21b, 21
c, 21d. The other double U-shaped core 22 also has four magnetic legs 22a, 22b, 22c, 22 as shown in FIGS. 17 (a) and 17 (b), for example.
It is configured with d. The magnetic legs 21a to 21d, 2 of the two double U-shaped cores 21, 22 are
The three-dimensional core 20 is formed by joining the ends of 2a to 22d.

Each of the magnetic legs 21a to 21d and the magnetic leg 22
For each of the joints a to 22d, see the upper 2
A mylar film of 10 μm is inserted in the set or the two sets in the lower stage so that the gap G = 10 μm. Then, as shown in FIG. 16, the inductance LB of the drive winding NB
The direct current superposition characteristics of the control current Ic = 10 mA to 60 mA
However, the inductance LB changes from 8 μH to 2.5 μH.

Then, as shown in FIG. 17B, for example, the two magnetic legs 22 of the double U-shaped core 22 are provided.
A control winding NC is wound around c and 22d, and a detection winding ND and a drive winding NB are wound around magnetic legs 21c and 21b of the double U-shaped core 21. That is, the orthogonal control transformer PRT is configured as a saturable reactor in which the control winding NC is wound in the direction orthogonal to the detection winding ND and the drive winding NB. This orthogonal control transformer P
As the control winding NC of the RT, for example, 1000 T (turn) is wound with a polyurethane-coated copper wire of 60 μmφ, the detection winding ND is 1 T with a polyurethane-coated copper wire of 0.3 mmφ, and the drive winding NB is 0.3 mmφ. It is wound 3T by a polyurethane-coated copper wire.

In such an orthogonal control transformer PRT, the gap G is as small as about 10 μm as described above in order to reduce the amount of control current flowing through the control winding. However, the precision error of the gap thickness is unavoidable at the time of manufacturing, but this is due to the drive winding N wound around the orthogonal control transformer PRT.
This causes variations in the inductance value of B.
Further, variations in the magnetic permeability of the ferrite core, deviations when the magnetic legs are joined, and the like also cause variations in the inductance value of the drive winding NB. For these reasons, the allowable value of the inductance LB must be such that the inductance value changes by ± 10%. For this reason, variations in the amplification factor hFE of the switching element Q1 and the accumulation time tstg occur, but the constant voltage guarantee range of the composite resonant converter is set against this variation, for example, when the commercial AC power supply is a 100V system, the AC input voltage VAC. = 100V ± 10%, the variable inductance range of the orthogonal control transformer PRT must be designed with a sufficient margin. In other words, it becomes difficult to design the margin for practical use.

Further, the winding specifications of the orthogonal control transformer PRT are as described above, and it is not necessary to wind the control winding NC, the detection winding ND and the drive winding NB in directions orthogonal to each other. Moreover, the winding process becomes very complicated. Further, joining the four magnetic legs of each of the double U-shaped cores 21 and 22 through the Mylar film without displacement also complicates the assembly process. That is, the orthogonal control transformer PRT
Is difficult to manufacture and cost reduction is difficult.

The DC control current Ic flowing through the control winding NC of the orthogonal control transformer PRT is the DC output voltage E02 line (15V) on the secondary side of the insulating converter transformer PIT.
Line), and the supplied power is 0.9 W to 0.
Although it fluctuates within the range of 15 W, this supplied power is reactive power, and power loss during light load increases.

[0028]

In view of the above problems, the present invention is configured as a switching power supply circuit as follows. That is, a switching means having a switching element for switching a DC input voltage, a primary winding and a secondary winding are provided, and the output of the switching means obtained in the primary winding is transmitted to the secondary winding. And a primary side parallel resonance circuit which is formed by the primary winding of the insulation converter transformer and the primary side parallel resonance capacitor, and which is provided so that the operation of the switching means is a voltage resonance type. Further, the drive transformer is provided with the detection winding, the drive winding, and the control winding wound so that a magnetic coupling state can be obtained between the drive winding and the control winding. In addition, a switching drive means is provided which has a series resonance circuit formed by a drive winding and a resonance capacitor, and which drives the switching element by switching based on the output of the series resonance circuit. Further, the insulating converter transformer secondary winding, and a circuit oscillating secondary co formed by connecting to the capacitor vibration secondary co. Further, it is provided with a direct current output voltage generating means configured to obtain a direct current output voltage by inputting an alternating voltage obtained in the secondary side resonance circuit and performing a rectifying operation. A conduction control circuit formed by connecting a control winding and a transistor element as a conduction control element in series is provided on the secondary side of the insulation converter transformer, and the conduction control element in the conduction control element is provided in accordance with the level of the DC output voltage. By variably controlling the current conduction amount, the switching frequency of the switching element is variably controlled, and constant voltage control means for performing constant voltage control of the DC output voltage is provided.

The power supply circuit having the above-mentioned structure has a basic structure as a composite resonance type converter which is provided with a drive transformer to drive the switching element by self-excitation. A conduction control circuit including a conduction control element is provided for constant voltage control. In this conduction control circuit, a level current according to the secondary side DC voltage level is made to flow through the control winding of the drive transformer that is connected in series with the conduction control element. If the amount of current in the control winding is changed in this way, the amount of current that tends to flow through the drive winding that is magnetically coupled to the control winding will change. The amount of current flowing through the formed switching drive means is changed. Thereby, the switching frequency of the switching element is variably controlled. With such a constant voltage control configuration, it becomes possible to omit the orthogonal control transformer used for variable switching frequency control in the case of the self-excited type, for example.

[0030]

FIG. 1 shows the configuration of a power supply circuit as a first embodiment of the present invention. The power supply circuit shown in FIG. 1 has a configuration as a composite resonance type switching converter having a voltage resonance type converter on the primary side and a parallel resonance circuit on the secondary side. In the power supply circuit shown in this figure, first, as a rectifying / smoothing circuit for inputting a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage, a full-wave rectifying circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci. And an AC input voltage V
The rectified smoothed voltage Ei corresponding to a level of AC is generated.

A switching converter for inputting the rectified and smoothed voltage Ei (DC input voltage) to connect and disconnect is 1
A voltage resonance type converter including a stone switching element Q1 and performing a switching operation by a so-called single-ended system is provided. The voltage resonance type converter here has a self-excited configuration, and a high breakdown voltage bipolar transistor (BJT; junction type transistor) is used as the switching element Q1. Switching element Q
The collector of 1 is connected to the positive electrode of the smoothing capacitor Ci via the primary winding N1 of the insulating converter transformer PIT, and the emitter is connected to the primary side ground.

Further, the collector of the switching element Q1
A parallel resonance capacitor Cr is connected in parallel between the emitters. The capacitance of the parallel resonance capacitor Cr and the primary winding N of the insulating converter transformer PIT
The leakage inductance obtained in 1 forms a primary side parallel resonant circuit. Then, according to the switching operation of the switching element Q1, the resonance operation by the parallel resonant circuit is obtained, so that the switching element Q1
The switching operation of is a voltage resonance type.

Further, a clamp diode DD is connected between the base and the emitter of the switching element Q1 in the direction shown. Here, the clamp diode DD
Is connected to the emitter (primary side ground) and the cathode is connected to the base. A slow recovery type diode element is selected as the clamp diode DD in this case.

The drive transformer CDT is provided to drive the switching element Q1 by self-excitation. In this case, the primary side of the drive transformer CDT is the detection winding N
A, the detection winding NA is an insulating converter transformer P
By being connected in series to the primary winding N1 of IT, the switching output of the switching element Q1 transmitted to the primary winding N1 of the insulating converter transformer PIT is detected. The drive winding NB is connected to the secondary side where the alternating voltage obtained in the detection winding NA is induced.
Is wound. The drive winding NB forms a self-excited oscillation drive circuit for switching-driving the switching element Q1. Further, in the case of the present embodiment, the control winding Nc is wound around the primary side of the drive transformer CDT.
Here, the winding direction of each of the windings is such that the drive winding NB and the control winding Nc are in phase with each other and the detection winding NA is in reverse phase with respect to these windings NB and Nc, as shown in the drawing. It is wrapped around.

The drive transformer CDT around which the windings (NA, NB, Nc) are wound is, for example, an H-shaped ferrite core as shown in FIG.
An EI type ferrite core can be used. Figure 5
In this case, the H-shaped ferrite core 100 is formed by winding the detection winding NA, the drive winding NB, and the control winding Nc.

In the case of FIG. 6, the I-type core 102 and the E-type core 103 are combined as shown to form an EI-type core. Then, the split bobbin 10 is attached to the central magnetic leg of the E-shaped core 103.
3 are arranged, and the detection winding NA, the drive winding NB, and the control winding Nc are arranged on the divided bobbin 103, for example, as illustrated.
It is formed by winding respectively. The drive transformer CDT having the structure shown in FIG. 5 or 6 is considerably small and lightweight when compared with, for example, the orthogonal control transformer PRT described in FIG.

In the drive transformer CDT, the detection winding NA and the drive winding NB are arranged on the primary side of the insulating converter transformer PIT. On the other hand, the control winding Nc is arranged on the secondary side of the insulating converter transformer PIT. Therefore, for example, in order to secure the direct current insulation between the primary side and the secondary side, a photo coupler or the like is generally provided. On the other hand, in the present embodiment, by selecting a triple insulated wire for the control winding Nc, the insulation property with respect to the detection winding NA and the drive winding NB is ensured, and thus the photocoupler is formed. A sufficient insulation state is obtained without any interposition.

As shown in the drawing, a series connection circuit of [base current limiting resistor RB-driving winding NB-time constant (for resonance) capacitor CB] is connected to the base of the switching element Q1. This series connection circuit is a self-excited oscillation drive circuit for switching-driving the switching element Q1 by self-excitation.

In this case, the detection winding NA of the drive transformer CDT is the primary winding N1 of the insulating converter transformer PIT.
Are connected in series. Therefore, in the drive winding NB of the drive transformer CDT in the self-excited oscillation drive circuit, an alternating voltage is excited by the switching output voltage transmitted to the primary winding N1. As the self-excited oscillation drive circuit described above, a series resonance circuit is formed by the capacitor CB and the inductance of the drive winding NB.

As described above, the drive winding NB of the self-excited oscillation drive circuit is excited by the detection winding NA to generate an alternating voltage as a drive voltage. The drive voltage causes the series resonance circuit (NB-CB) to oscillate in a self-excited manner to obtain a resonance output. This resonance output is the base current limiting resistance RB
To the base of the switching element Q1,
A base current as a switching drive signal is made to flow. As a result, the switching element Q1 performs the switching operation at the switching frequency determined by the resonance frequency of the series resonance circuit. Then, the switching output obtained at the collector is transmitted to the primary winding N1 of the insulating converter transformer PIT.

Further, the starting resistance Rs in this case is inserted between the line of the rectified and smoothed voltage Ei and the connection point of the drive winding NB and the time constant capacitor CB. For example, when the power supply is started, the rectification smoothed voltage Ei changes the starting resistance R
drive winding NB-base current limiting resistor RB
The base current via the base current of the switching element Q1 flows into the base of the switching element Q1 to start the switching operation.

Further, the drive transformer C of the present embodiment
The winding start end of the control winding Nc wound around the DT is grounded to the secondary side ground, and the winding end thereof is connected to the collector of the conduction control element Q2. The emitter of the conduction control element Q2 is grounded to the secondary side ground. That is, the control winding Nc and the conduction control element Q2 (collector-emitter)
Can be regarded as forming a series connection circuit connected in series on the secondary side of the insulating converter transformer PIT. In this case, the conduction control element Q2 is NP
N-type bipolar transistors have been selected.

The base of the conduction control element Q2 is connected to the collector of a PNP type transistor Q4 provided in the control circuit 1 described later via the resistor R1. A resistor R2 is inserted between the base and emitter of the conduction control element Q2. A base-emitter voltage VBE2 is obtained across the resistor R2.

The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side. This insulating converter transformer PIT is provided with an EE type core in which, for example, two sets of E type cores made of ferrite material are combined so that their magnetic legs face each other, and a split bobbin is attached to the central magnetic leg of the EE type core. Is used to wind the primary winding N1 and the secondary winding N2 separately. A gap is formed with respect to the central magnetic leg. As a result, a loosely coupled state with a required coupling coefficient is obtained. The gap can be formed by making the center magnetic legs of the two sets of E-shaped cores shorter than the two outer magnetic legs. Further, as the coupling coefficient k, for example, a loose coupling state of k≈0.85 is obtained, and accordingly, it is difficult to obtain a saturated state.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In the circuit shown in FIG. 1, the secondary side parallel resonance capacitor C2 is connected to the secondary winding N2.
Are connected in parallel. Therefore, in this case, the parallel resonant circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonant capacitor C2. With this parallel resonant circuit, the secondary winding N2
The alternating voltage induced in the coil and the alternating voltage obtained in the detection winding NA become a resonance voltage. That is, the voltage resonance operation is obtained on the secondary side.

That is, this power supply circuit is provided with a parallel resonance circuit for providing a voltage resonance type switching operation on the primary side and a parallel resonance circuit for obtaining a voltage resonance operation on the secondary side. It is a "resonant switching converter".

For the secondary side of the power supply circuit formed as described above, there is provided a half-wave rectification circuit including a secondary side rectification diode DO1 connected to the secondary winding N2 and a smoothing capacitor CO1. As a result, the secondary side DC output voltage EO1 corresponding to the almost equal level of the alternating voltage induced in the secondary winding N2 is obtained. Further, here, a tap output is provided for the secondary winding N2, and as shown in the drawing, a tap output is provided between the tap output and the secondary side earth and is composed of a secondary side rectifying diode D02 and a smoothing capacitor CO2. By connecting the wave rectification circuit, a low-voltage secondary side DC output voltage EO2 is obtained. In this case, the secondary side DC output voltage EO1 is input to the control circuit 1 as a detection voltage for constant voltage control. In addition, the secondary side DC voltage EO2
Is used as an operating power supply of the control circuit 1.

In the control circuit 1, resistors R3-R4 are connected in series between the DC output voltage EO1 and the secondary side ground, and the control terminal of the shunt regulator Q3 is connected to this connection point (voltage dividing point). . For the resistor R4,
The series connection circuit of the capacitor C3 and the resistor R5 is connected in parallel. The anode of the shunt regulator Q3 is grounded to the secondary side ground, and the cathode is connected to the base of the PNP type transistor Q4 via the resistor R7. The transistor Q4 is provided to amplify the current flowing through the anode of the shunt regulator Q3. Transistor Q
The emitter of 4 is connected to the line of the secondary side DC voltage EO2, and the collector is connected to the base of the conduction control element Q2 described above via the resistor R1. The output of this collector becomes the detection output of the control circuit 1. Also, the base-
A resistor R6 is inserted between the emitters.

The control circuit 1 formed by the above-mentioned connection form functions as an error amplifier having the DC output voltage EO1 as a detection input. That is, the DC output voltage EO1 is applied to the resistor R
By inputting the voltage divided by 3 and R4 to the control terminal of the shunt regulator Q3 as a control voltage, a current having a level corresponding to the DC output voltage EO1 flows through the shunt regulator Q3. Then, this current is amplified by the transistor Q4 and flows to the base of the conduction control element Q2. In this way, the base of the conduction control element Q2 is
A direct current, the level of which is varied according to the secondary side DC voltage EO2, flows as a base current. Depending on this, the level of the collector current IQ2 flowing through the collector of the conduction control element Q2 also changes to the secondary side. It is variable according to the level of the DC voltage EO2. This means that the level (amplitude) of the current flowing through the control winding Nc connected between the collector of the conduction control element Q2 and the secondary side ground is variably controlled. According to this operation, the switching frequency of the switching element Q1 is changed, and thereby the secondary side DC output voltage is stabilized so as to be constant, which will be described later.

2 and 3 are waveform charts showing the operation of the main part of the power supply circuit having the configuration shown in FIG. Figure 2
3 shows the operation under the condition of heavy load of load power Po = 162 W, and FIG. 3 shows the operation under the condition of no load of load power Po = 0 W. In order to obtain the operation shown in these figures, each component is selected as follows. Drive winding NB = 3T (turn), inductance LB = 1
0 μH Control winding Nc = 5T, Inductance Lc = 28 μH Detection winding NA = 1T Capacitor CB = 0.56 μF Base current limiting resistance RB = 0.47Ω In addition, each selection of the above windings (NB, Nc, NA) The values are obtained when the EI type core shown in FIG. 6 is adopted for the drive transformer CDT, and when the H-shaped ferrite core 100 shown in FIG. 5 is used, the number of turns of each winding is slightly larger than this. Become.

In the drive winding NB wound around the drive transformer CDT, the alternating voltage due to the exciting action is generated as described above. Then, the self-excited oscillation drive circuit (CB-NB
-RB) performs a self-excited oscillation operation based on the alternating voltage generated in the drive winding NB. That is, the time constant capacitor C
The series resonant circuit formed by B and the drive winding NB performs resonant operation, and the resonant output is transferred to the base current limiting resistor R
A base current is caused to flow through B to the base of the switching element Q1.

Here, due to the resonance operation of the series resonance circuit (CB-NB), the time constant capacitor CB is connected as shown in FIG.
As shown in (i) and FIG. 3 (i), a sinusoidal resonance voltage V3 corresponding to the switching cycle is generated. And
The resonance output of this series resonance circuit (CB-NB) is as shown in FIGS. 2 (j) and 3 (j).
A resonance current I0 flowing into the base side of the switching element Q1 via the base current limiting resistor RB can be obtained from. Further, this resonance current IO is branched and flows as a base current IB flowing through the base of the switching element Q1 and a damper current ID flowing through the clamp diode DD.

In the case of the present embodiment, the drive winding NB and the control winding Nc forming the series resonance circuit are wound so as to be magnetically tightly coupled in the drive transformer CDT. For this reason, the series resonance circuit (CB-NB) resonates as described above, so that the drive winding NB
An alternating voltage is induced in the control winding Nc by the alternating voltage generated at. Due to the induced voltage in the control winding Nc, the control winding Nc and the control winding Nc
The collector current IQ2 shown in FIG. 2 (g) and FIG. 3 (g) flows in the transistor as the conduction control element Q2 connected in series between the secondary side ground and the ground.

During the period TON in which the switching element is turned on, the reverse recovery time trr of the clamp diode DD first causes the slow recovery type clamp diode DD to conduct during the period t3 to t1.
(E), as shown in FIG. 3 (e), the damper current ID
Flows. The damper current ID during this period t3 to t1 is
It flows through the PN junction of the base → collector of the switching element Q1. Accordingly, as the collector current IQ1 of the switching element Q1 in the period t3 to t1, as shown in FIGS. 2B and 3B, a waveform flowing in the negative polarity direction is obtained. Also, the base current IB is
As shown in FIGS. 2 (c) and 3 (c), the positive polarity rises at time t3 and decreases to 0 level by time t1.

In the subsequent period t1 to t2, the clamp diode DD is turned off. At this time,
Regarding the base current IB (FIGS. 2C and 3C), first, the forward current IB1 due to the positive polarity flows, and thereafter, the base current IB reverses to the negative polarity due to the base stored carrier annihilation time tstg.
A reverse current IB2 flows. The switching element Q1 becomes conductive according to the base current IB, and
As shown in FIGS. 3B and 3B, a positive collector current IQ1 flows through the collector of the switching element Q1.

Then, the base current IB (FIGS. 2 (c) and 3 (c)) becomes zero level when the base accumulated carrier erasing time tstg in which the reverse current IB2 flows is completed, whereby the switching element Q1 becomes Off period T
Move to OFF.

As shown in FIGS. 2D and 3D, the base-emitter voltage VBE1 of the switching element Q1 has a period TOFF (periods t2 to t3) and a period t3 to t1 within the period TON. In the above, since the negative polarity is obtained, the switching element Q1 becomes reverse bias. Then, in the periods t1 to t2 in the period TON, the level to which the certain offset is given with respect to the 0 level is maintained.

As the switching element Q1 performs the switching operation as described above, the resonance voltage V1 obtained across the primary side parallel resonance capacitor Cr is as shown in FIG.
As shown in (a) and FIG. 3 (a), a waveform having a 0 level in the period TON in which the switching element Q1 is on and a sinusoidal pulse in the period TOFF in the off state is obtained. This indicates that the primary side switching converter is a voltage resonance type operation.

The operation of the conduction control element Q2, which is an NPN type bipolar transistor, is as follows. As described above, the conduction control element Q2 and the control winding N
c is connected in series on the secondary side. Then, the control winding Nc is induced by the drive winding NB,
A collector current IQ2 flows from the control winding Nc to the collector of the conduction control element Q2.

Here, the base-emitter voltage VBE2 of the conduction control element Q2 has a period TOFF (periods t2 to t3) and a period subsequent thereto as shown in FIGS. 2 (h) and 3 (h). During the period t3 to t1 within TON, the conduction control element Q2 is in the state of being reverse-biased due to the negative polarity. Accordingly, the resonance current IO in the self-excited oscillation drive circuit (CB-NB-RB) also has a waveform inverted to the negative polarity during the period t2 to t3 to t1. Therefore, the conduction control element Q2 is
It operates as a so-called reverse transistor, and has a period of t2 to t3.
From t1 to t1, a current flows from the emitter to the collector in the conduction control element Q2. As a result, the collector current IQ2 of the conduction control element Q2 has a period of t2 to t as shown in FIGS. 2 (g) and 3 (g).
From 3 to t1, a negative waveform is obtained.

Then, the base-emitter voltage VBE2 of the conduction control element Q2 has a period t1 to t2 within the remaining period TON.
At, a constant level of positive polarity is maintained and positive bias is applied. In response to this, also in the conduction control element Q2 in the periods t1 to t2, the current reversely flows from the collector to the emitter, and the collector current I
As Q2, as shown in FIG. 2 (g) and FIG. 3 (g), it has a waveform inverted to the positive polarity. It should be noted that the collector-emitter voltage V2 of the conduction control element Q2 is as shown in FIG.
As shown in (f), a waveform having a negative polarity in the periods t2 to t3 to t1 and a positive polarity in the periods t1 to t2 is obtained.

Here, it is assumed that the level of the secondary side DC output voltage EO1 rises because the AC input voltage VAC rises or the load power decreases, for example. Then, the control circuit 1 operates so as to increase the collector current of the transistor Q4 which is the detection output. As a result, the base current of the conduction control element Q2 increases and the amplitude of the base-emitter voltage VBE2 also increases.

By controlling the conduction control element Q2 as described above, the collector current IQ2 of the conduction control element Q2 is controlled.
Will be controlled so that the amplitude of is also large. Here, as described above, since the drive winding NB and the control winding Nc are in a tightly coupled state in the drive transformer CDT, the drive winding NB and the control winding Nc are equalized. It can be seen that Nc and Nc are connected in parallel. Therefore, as for the equalization circuit, for example, a self-oscillation circuit (C
B-NB-RB), the control winding N is connected to the connection point between the winding end portion of the drive winding NB and the base current limiting resistor RB.
A circuit in which the winding start ends of c are connected is formed.
In this case, the collector current IQ2 flowing through the control winding Nc,
The relationship between the resonance current I0 that is the output of the self-excited oscillation circuit (CB-NB-RB) is that the current that flows in the drive winding NB with the connection point of the drive winding NB and the base current limiting resistor RB as the branch point. Can be said to be the current obtained by branching.

Therefore, when the amplitude of the collector current IQ2 of the conduction control element Q2 increases and the amount of the collector current IQ2 in the periods t1 to t2 increases, the amount of the resonance current IO decreases accordingly. Thus, the behavior that changes is obtained. The waveform of the base current IB obtained based on the resonance current I0 is, for example, as shown in FIGS.
Although it changes as shown as a transition to (c), the base accumulated carrier disappearance time (tstg) of the switching element Q1 becomes short accordingly. Along with this, the lengths of the periods t1 to t2 within the period TON in which the switching element Q1 is turned on are shortened and varied.

If the periods t1 to t2 in the period TON are shortened, the length of the entire period TON is shortened even if the previous periods t3 to t1 are unchanged. Therefore, the switching frequency of the switching element Q1 is It will be controlled to be higher. This is shown by comparing FIG. 2 and FIG. 3 that the time length of one switching period consisting of the period TON + TOFF becomes shorter as the condition of the light load is shortened. By variably controlling the switching frequency, for example, the resonance impedance of the primary side parallel resonance circuit is changed, and the power transmitted from the primary side to the secondary side of the insulating converter transformer PIT is also changed. It will be variable. As a result, the level of the secondary side DC output voltage is finally variably controlled, and the power source is stabilized. Specifically, the load power Po = 162W
Switching frequency fs =
A constant voltage can be achieved by variably controlling in the range of 80 KHz to 135 KHz.

In the present embodiment, when the switching frequency is variably controlled, the period TOFF during which the switching element Q1 is off is constant, and the period TON during which it is on is variable. That is, also in this case, the constant voltage control operation by the composite control method is obtained.

With the structure of the constant voltage control circuit system according to the present embodiment described above, the orthogonal control transformer PRT shown in FIG. 13 is omitted. As a result, in this embodiment, the orthogonal control transformer PRT is used.
The problem of the variation in the inductance value of the drive winding NB due to the variation in the gap at the time of manufacturing is solved. Therefore, the margin for the range of the AC input voltage VAC can be set to be small, and the circuit design can be facilitated.
Further, the problem of difficulty in the manufacturing process of the orthogonal control transformer PRT is solved. Further, the AC / DC power conversion efficiency can be improved.

Also, unlike the example of FIG. 13, the control power is not supplied to the control winding NC of the orthogonal control transformer PRT to control the switching frequency, so that the reactive power at the light load is reduced and the power is reduced. The loss can be reduced. Further, the drive transformer CDT is, as described with reference to FIG. 5 or FIG. 6, an H-shaped ferrite core or an ultra-compact EI-1.
It can be configured with a type 2 ferrite core.
The size and weight can be greatly reduced as compared with the case where the orthogonal control transformer PRT is provided as shown in the prior art.

Particularly in the case of the present embodiment, the current to flow from the control winding Nc to the conduction control element Q2 is excited by the drive winding NB magnetically tightly coupled to the control winding Nc in the drive transformer CDT. It is designed to be obtained based on the alternating voltage. For example, as a power supply for obtaining a current to be passed through the control winding Nc, a tertiary winding wound around an insulating converter transformer PIT and a rectifying circuit connected to this tertiary winding are formed, and generated by this rectifying circuit. There is a case where a low-voltage DC voltage is used, but in the case of the present embodiment, such a rectifying circuit is unnecessary, and it can contribute to the promotion of reduction in size and weight of the circuit.

Further, although the series connection circuit composed of the control winding Nc and the conduction control element Q2 is connected to the primary side self-excited oscillation drive circuit in terms of equalization as described above, it is actually used. As can be seen from the fact that it is provided between the secondary side ground, the isolation converter transformer PIT
Is a circuit provided on the secondary side of the. Then, as described above, for the control winding Nc, for example, a triple insulated wire is selected to secure direct current insulation with the primary side, and
A state in which it is magnetically coupled to the drive winding NB on the primary side is obtained. As a result, in the present embodiment, by adopting the above-mentioned configuration, it is possible to secure the DC insulation state between the secondary side control circuit system and the primary side switching converter without adding a component element such as a photocoupler. It is what For example, suppose that the control winding Nc and the conduction control element Q2 are
If the primary side is provided with a series connection circuit consisting of, a DC coupling must be provided between the secondary side control circuit system and a photocoupler, for example.

Further, with the structure of the present embodiment, the current flowing through the conduction control element Q2 is very small, and the voltage applied to the conduction control element Q2 is also low.
Therefore, for the bipolar transistor as the conduction control element Q2, a low withstand voltage small capacity product having a withstand voltage of 30 V and a rated current of 0.15 A or less may be selected. For example, the present applicant previously provided an active clamp circuit that clamps a primary side parallel resonance voltage or a secondary side resonance voltage for a composite resonance type switching converter, and stabilizes the power supply by controlling the conduction angle of the active clamp circuit. Although various configurations have been proposed for achieving this, in this case, the switching element (transistor) that forms the active clamp circuit has a high voltage corresponding to the primary side parallel resonance voltage level or the secondary side resonance voltage level. It is necessary to select a pressure resistant product, which is disadvantageous in terms of cost and size. On the other hand, in the present embodiment, a low withstand voltage small capacity product is selected as the bipolar transistor as the conduction control element Q2, so that it is possible to realize cost reduction, size reduction and weight reduction. is there.

Here, FIG. 4 shows the power conversion efficiency characteristics and the control power required for switching frequency control as the characteristics of the power supply circuit according to the present embodiment shown in FIG.
This is shown by comparison with the power supply circuit of FIG. 13 shown as the prior art. As can be seen from this figure, the load power Po =
Control power Pc corresponding to the load fluctuation range of 0 W to 162 W
The power supply circuit shown in FIG.
The power supply circuit shown in FIG.
It is 19 W to 0.06 W, which is significantly lower than that of the power supply circuit shown in FIG. this is,
This is because the amount of current flowing through the conduction control element Q2 in the circuit shown in FIG. 1 is significantly smaller than the amount of control current flowing through the control winding Nc in the circuit shown in FIG. Regarding the power conversion efficiency ηAC-DC, it is shown that the power supply circuit shown in FIG. 1 is higher than the power supply circuit shown in FIG. 13 over the load fluctuation range of the load power Po = 0W to 162W. Has been done. That is, in the power supply circuit shown in FIG. 1, the power conversion efficiency is comprehensively improved as compared with the power supply circuit shown in FIG.

FIG. 7 shows a configuration example of a switching power supply circuit according to the second embodiment of the present invention. In addition,
In this figure, the same parts as those in FIG. Even if the power supply circuit shown in FIG.
Alternatively, the drive transformer CDT having the structure shown in FIG. 6 is provided.

In the power supply circuit shown in this figure, a voltage doubler rectifier circuit is provided to generate the rectified and smoothed voltage Ei which is the input voltage to the primary side switching converter. This voltage doubler rectifier circuit
It is formed by connecting rectifier diodes Di1 and Di2 and smoothing capacitors Ci1 and Ci2 to C as shown in the figure. Then, the AC input voltage VAC is input and the rectified and smoothed voltage Ei having a level corresponding to a level twice the AC input voltage VAC is generated. The configuration in which the voltage doubler rectifier circuit is provided in this way is, for example, a commercial AC power supply AC.
It is suitable when a relatively heavy load condition is met in a 100V system.

On the secondary side of the power supply circuit shown in this figure, the secondary winding N2 of the insulating converter transformer PIT is provided.
After connecting the center tap to the secondary side ground, the two rectifying diodes D0
A full-wave rectifier circuit is formed by connecting 1, D02 and the smoothing capacitor C01. Then, this full-wave rectification circuit obtains the secondary side DC output voltage EO1 as the voltage across the smoothing capacitor CO1. Although the control circuit 1 is shown as one functional block in this drawing, for example, actually, the internal configuration similar to that of the control circuit 1 shown in FIG. 1 may be adopted.

Also in the configuration as the second embodiment, a drive transformer CDT having the structure described with reference to FIGS. 5 and 6, for example, is provided, and a constant voltage control circuit similar to the power supply circuit of FIG. 1 is provided. By adopting the system configuration, it is possible to obtain the same effects as those of the power supply circuit of the first embodiment, including the omission of the orthogonal control transformer PRT.

By the way, in the power supply circuit of each of the above-mentioned embodiments, the rectification circuit system provided on the secondary side is not limited to the configuration shown in each drawing, and for example, FIGS.
It is also possible to adopt the configuration shown in FIG. Figure 8
In the above, in the secondary side parallel resonance circuit (N2 // C2), the secondary side DC output voltage EO1 is connected by connecting the full-wave rectification circuit composed of the bridge rectification circuit DBR and the smoothing capacitor CO1 in the illustrated connection form. The configuration that is intended to be obtained is shown.

Further, in FIG. 9, for the secondary side parallel resonant circuit (N2 // C2), the rectifying diode DO1, the rectifying diode DO2 and the smoothing capacitor CO are provided as shown in the figure.
By connecting A and COB, a voltage doubler rectifier circuit by the full-wave rectification method is formed. In this case, the secondary side DC output voltage EO1 corresponding to twice the alternating voltage level generated in the secondary winding N2 is obtained as the voltage across the series connection circuit of the smoothing capacitors COA-COB.

As for the secondary side structure shown in FIG. 10, the secondary side series resonance capacitor Cs is connected in series to the winding end portion of the secondary winding N2. by this,
On the secondary side of the insulating converter transformer PIT, the leakage inductance of the secondary winding N2 and the capacitance of the secondary side series resonant capacitor Cs form a secondary side series resonant circuit. Therefore, in this case, the primary side parallel resonance circuit (N1, C
r) and a secondary side series resonance circuit (N
2, Cs) constitutes a composite resonance type converter. And, for this secondary side series resonant circuit,
By connecting the bridge rectifier circuit DBR and the smoothing capacitor CO1 as shown in the figure, the secondary side series resonance circuit (N
2, Cs) forms a full-wave rectification circuit for full-wave rectifying the secondary side series resonance voltage generated by the resonance action. And
For both ends of the smoothing capacitor CO1, the secondary side DC output voltage EO1 corresponding to the same level as the alternating voltage level generated in the secondary winding N2 is obtained.

In FIG. 11, the rectifier diode DO1, the rectifier diode DO2, and the smoothing capacitors COA and COB are connected to the secondary side series resonance circuit (N2, Cs) as shown in FIG. It forms a voltage rectifier circuit.

In FIG. 12, two sets of secondary side series resonance capacitors Cs1 and Cs2 are connected to the secondary winding N2 as shown in the figure, and four rectifying diodes DO1, DO2 and D03 are further connected. , D0 4 are connected by the illustrated connection form to form a secondary side rectifier circuit. A quadruple voltage rectifier circuit is formed as the secondary side rectifier circuit configured in this manner. In explaining the operation of this quadruple voltage rectifier circuit, [series resonance capacitor Cs1, rectifier diode D0
The operation of the circuit consisting of 1, D02 and smoothing capacitor COA] will be described. First, during the period when the rectifying diode DO1 is off and the rectifying diode D02 is on,
Due to the leakage inductance of the secondary winding N2 and the series resonance action of the series resonance capacitor Cs1, the rectified current rectified by the rectification diode D02 is applied to the series resonance capacitor Cs.
The operation of charging for 1 is obtained. Then, during the period in which the rectifying diode D02 is turned off and the rectifying diode D01 is turned on to perform the rectifying operation, a series resonance occurs in which the potential of the series resonance capacitor Cs1 is added to the voltage induced in the secondary winding N2. Smoothing capacitor COA
The operation is performed for charging. By performing the rectifying operation as described above, a DC voltage (rectified smoothed voltage) corresponding to almost twice the induced voltage of the secondary winding N2 is obtained across the smoothing capacitor COA. Further, the same operation is performed by a rectifying circuit composed of a set of [series resonance capacitor Cs2, rectifying diodes DO3, DO4, smoothing capacitor COB], and by the same operation, the induced voltage of the secondary winding N2 is almost across the smoothing capacitor COB. A DC voltage corresponding to twice the voltage will be obtained.

As a result of the voltage doubler rectifying operation performed by the rectifying circuits of the respective stages as described above, the induced voltage of the secondary winding N2 is applied across the smoothing capacitor COA and the smoothing capacitor COB connected in series. The secondary side DC output voltage EO1 corresponding to almost four times is obtained.

In each of the above-mentioned embodiments, the single-end type having one set of switching elements is shown. However, a so-called push-pull type self-exciting voltage resonance converter having two sets of switching elements is shown. It is okay to say that. The secondary side may also be provided with a rectifier circuit having a circuit configuration other than that shown in each drawing.

[0084]

As described above, the present invention adopts a self-excited configuration including a drive transformer as a composite resonance type converter. At the same time, the switching frequency control for constant voltage winding is performed by winding a control winding around the drive transformer and magnetically coupling the drive winding, and connecting the control winding and the conduction control element in series (conduction control). Circuit). The conduction of the series connection circuit is controlled to variably control the amount of current conduction in the self-excited oscillation drive circuit (switching drive means) that drives the switching element. Then, with such a configuration, it becomes possible to omit the orthogonal control transformer which has been required for the switching frequency control up to now. This solves the problem of variations in the inductance value due to variations in the gap of the orthogonal control transformer. In particular, the variation of the inductance of the drive transformer in the present invention is about ± 5%. For this reason, it is possible to set a small margin for the range of the AC input voltage, so that the circuit design can be facilitated. Further, the problem of difficulty in the manufacturing process of the orthogonal control transformer can be solved. Since it is not configured to supply the control power to the control winding of the orthogonal control transformer to control the switching frequency, it is possible to reduce the reactive power and the power loss at the light load. That is, A
The C / DC power conversion efficiency will also be improved. Further, since the drive transformer according to the present invention is much smaller than the orthogonal control transformer, it is possible to promote the reduction in size and weight of the power supply circuit.

The conduction control element (bipolar transistor) provided in the conduction control circuit according to the present invention includes:
Since the low voltage and the small level current are applied, the power required for the constant voltage control is also reduced. That is, the reactive power in the power supply circuit is reduced, which also promotes the reduction of the AC / DC power conversion efficiency of the power supply circuit. Further, as the conduction control element, a low withstand voltage, small capacity product is selected, and in this respect as well, reduction in size and weight and cost reduction are promoted.

Further, according to the present invention, the conduction control circuit is provided on the secondary side of the insulating converter transformer by utilizing the fact that the control winding is galvanically isolated from other windings in the drive transformer. However, this makes it possible to omit the photocoupler, for example. Further, since the control winding is magnetically coupled to the drive winding in the drive transformer, the conduction control circuit can operate based on the voltage excited in the control winding by the drive winding. Therefore, it is not necessary to add a circuit component for generating a low voltage DC power supply, for example. That is, also in these respects, it is possible to further promote the reduction in size and weight of the circuit.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a waveform diagram showing an operation (under heavy load) of a main part of the power supply circuit according to the present embodiment.

FIG. 3 is a waveform diagram showing an operation (under no load) of a main part of the power supply circuit according to the present embodiment.

FIG. 4 is a characteristic diagram showing the characteristics of AC-DC power conversion efficiency with respect to load fluctuations and control power for constant voltage in the power supply circuit shown in FIG. 1, as compared with the prior art.

FIG. 5 is a perspective view showing a structural example of a drive transformer having an H-shaped core.

FIG. 6 is a cross-sectional view showing a structural example of a drive transformer having an EI type core.

FIG. 7 is a circuit diagram showing a configuration example of a switching power supply circuit according to a second embodiment of the present invention.

FIG. 8 is a circuit diagram showing another configuration example on the secondary side of the embodiment.

FIG. 9 is a circuit diagram showing another configuration example on the secondary side of the embodiment.

FIG. 10 is a circuit diagram showing another configuration example on the secondary side of the embodiment.

FIG. 11 is a circuit diagram showing another configuration example on the secondary side of the embodiment.

FIG. 12 is a circuit diagram showing another configuration example of the secondary side of the embodiment.

FIG. 13 is a circuit diagram showing a configuration example of a switching power supply circuit as a prior art.

14 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

15 is an explanatory diagram showing constant voltage control characteristics of the power supply circuit shown in FIG.

16 is an explanatory diagram showing a DC superimposition characteristic of the inductance of the drive winding NB in the circuit shown in FIG.

FIG. 17 is a perspective view and a cross-sectional view showing a structural example of an orthogonal control transformer.

[Explanation of symbols]

1 control circuit, Q1 switching element, Q2 conduction control element (bipolar transistor), PIT insulation converter transformer, CDT drive transformer, DD clamp diode, N1 primary winding, N2 secondary winding, Cr
Primary side parallel resonance capacitor, NA detection winding, NB drive winding, Nc control winding, C2 Secondary side parallel resonance capacitor

─────────────────────────────────────────────────── ─── Continuation of front page (58) Fields surveyed (Int.Cl. ⁷ , DB name) H02M 3/00-3/44

Claims (2)

(57) [Claims] 1. A switching means comprising a switching element for switching a DC input voltage, a primary winding and a secondary winding, and the output of the switching means obtained at the primary winding is used as the secondary winding. And a primary side parallel resonance circuit formed by a primary winding of the insulation converter transformer and a primary side parallel resonance capacitor, the primary side parallel resonance circuit provided so as to make the operation of the switching means a voltage resonance type. A drive transformer in which the detection winding, the drive winding, and the control winding are wound so that a magnetic coupling state is obtained between the drive winding and the control winding; It has a series resonance circuit formed by a drive winding and a resonance capacitor, and switches the switching element based on the output of this series resonance circuit. Switching driving means for moving, relative to the insulating converter transformer secondary winding, the secondary
And the secondary side both <br/> oscillation circuit formed by connect the side co oscillation capacitor, the DC output voltage by performing a rectifying operation by entering the alternating voltage obtained at the secondary side both oscillating circuits And a conduction control circuit formed by connecting in series the control winding and a transistor element as a conduction control element on the secondary side of the insulation converter transformer. By variably controlling the amount of current conduction in the conduction control element according to the level of the DC output voltage, the switching frequency of the switching element is variably controlled, and constant voltage control of the DC output voltage is performed. A switching power supply circuit comprising: a constant voltage control means. 2. The switching power supply circuit according to claim 1, wherein a triple insulated wire is used for the control winding.