JP3528816B2 - Switching power supply circuit - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply in various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a switching power supply circuit that employs a switching converter such as a flyback converter or a forward converter is widely known. Since the switching operation waveforms of these switching converters are rectangular waves, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It is known that there is a limit to the improvement of power conversion efficiency. Therefore, the applicant has previously proposed various switching power supply circuits using various resonant converters.
The resonant converter can easily obtain high power conversion efficiency, and the switching operation waveform has a sinusoidal waveform to realize low noise. It also has the advantage that it can be configured with a relatively small number of parts.

The circuit diagram of FIG. 15 shows an example of a switching power supply circuit as a prior art, which can be constructed based on the invention previously proposed by the present applicant. As a basic configuration of the power supply circuit shown in this figure, a voltage resonance type converter is provided as a primary side switching converter.

In the power supply circuit shown in this figure, a bridge rectifier circuit Di and a smoothing capacitor Ci generate a rectified and smoothed voltage Ei corresponding to a level of one time the AC input voltage VAC from a commercial AC power supply (AC input voltage VAC). .

As the voltage resonance type converter which receives and connects the rectified and smoothed voltage Ei (DC input voltage), a single-ended system using one stone is adopted. The drive system is a self-excited type. In this case, the switching element Q1 forming the voltage resonance type converter is
A high voltage bipolar transistor (BJT; junction type transistor) is selected. A primary side parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. A series circuit of a clamp diode DD and a resistor RD is connected between the base and the emitter. Here, the parallel resonance capacitor Cr
Forms a primary side parallel resonance circuit together with a leakage inductance L1 obtained in the primary winding N1 of the insulating converter transformer PIT, whereby an operation as a voltage resonance type converter can be obtained. Then, with respect to the base of the switching element Q1, the drive winding NB-resonance capacitor CB-base current limiting resistor RB
Is connected to the self-excited oscillation drive circuit. The switching element Q1 is switching-driven by being supplied with a base current based on an oscillation signal generated by the self-excited oscillation drive circuit. It is to be noted that, at the time of starting, it is started by the starting current flowing from the line of the rectified and smoothed voltage Ei to the base via the starting resistor RS.

The orthogonal control transformer PRT is constructed by winding a control winding Nc such that the winding directions of the drive winding NB and the current detection winding ND are orthogonal to each other. It is provided to control the switching frequency of the primary side voltage resonance type converter as described later. The structure of the orthogonal control transformer PRT will be described later.

The insulating converter transformer PIT is provided for transmitting the switching output of the switching converter obtained on the primary side to the secondary side. This isolation converter transformer PIT has a primary winding N with respect to the EE type core.
1 and the secondary winding N2 are divided and wound, and a gap is formed for the central magnetic leg so that a loose coupling state with a required coupling coefficient can be obtained and a saturation state can be obtained. I try to make it difficult.

The primary winding N1 of the insulating converter transformer PIT is connected between the line of the DC input voltage (rectified and smoothed voltage Ei) and the collector of the switching element Q1. The switching element Q1 performs switching with respect to the DC input voltage, whereby the switching output of the switching element Q1 is supplied to the primary winding N1, and an alternating voltage having a cycle corresponding to the switching frequency is generated. To do.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary side parallel resonant capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonant capacitor C2 are used. A parallel resonant circuit is formed. With this parallel resonant circuit,
The alternating voltage induced in the secondary winding N2 becomes a resonance voltage.
That is, the voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the parallel resonance circuit for making the switching operation a voltage resonance type is provided on the primary side, and the parallel resonance circuit for obtaining the voltage resonance operation is provided on the secondary side. In the present specification, the switching converter configured such that the resonance circuits are provided for the primary side and the secondary side as described above is also referred to as a “composite resonance type switching converter”.

Insulation converter transformer PIT in this case
On the secondary side, first, the half-wave rectification circuit is formed by connecting the anode of the rectifying diode DO1 to the end of the secondary winding N2 and connecting the cathode to the positive terminal of the smoothing capacitor CO1. Is forming. Depending on this half-wave rectifier circuit, the secondary side DC output voltage EO1 is obtained across the smoothing capacitor CO1. Further, in this case, a tap is provided on the secondary winding N2, and a half-wave rectifying circuit including a rectifying diode D02 and a smoothing capacitor C02 is formed for the tap output as shown in the figure. Then, depending on this half-wave rectification circuit, the secondary side DC output voltage E lower than the secondary side DC output voltage EO1 is obtained.
O2 is obtained. In addition, specifically, the secondary side DC output voltage EO1 = 135V and the secondary side DC output voltage EO2 = 15V.

These secondary side DC output voltages EO1 and EO2 are
It will be supplied to each required load circuit. The secondary side DC output voltage EO1 is branched and output as a detection voltage of the control circuit 1.

In the control circuit 1, resistors R3-R4 are connected in series between the DC output voltage EO1 and the secondary side ground, and the control terminal of the shunt regulator Q3 is connected to this connection point (division point). . The anode of the shunt regulator Q3 is grounded, and the cathode is connected to the line of the secondary side DC output voltage EO2 via the control winding NC of the orthogonal control transformer PRT. Further, here, the cathode of the shunt regulator Q3 is connected to the capacitor C1.
It is connected via 1 to the connection point of resistors R3 and R4. Further, a series connection circuit of a capacitor C3 and a resistor R5 is connected in parallel with the resistor R4.

The control circuit 1 formed by the above-mentioned connection form functions as an error amplifier having the DC output voltage EO1 as a detection input. That is, the DC output voltage EO1 is applied to the resistor R
The voltage divided by 3 and R4 is input to the control terminal of the shunt regulator Q3 as a control voltage. Therefore, in the shunt regulator Q3, a current having a level corresponding to the DC output voltage E01 is caused to flow as the control current Ic to the control winding NC. That is,
The control current level flowing through the control winding NC is variably controlled. By varying the level of the control current flowing through the control winding Nc, the quadrature control transformer PRT is controlled to vary the inductance LB of the drive winding NB. As a result, the resonance frequency of the resonance circuit composed of the drive winding NB and the resonance capacitor CB in the self-excited oscillation drive circuit changes, and the switching frequency of the switching element Q1 is variably controlled. By varying the switching frequency of the switching element Q1 in this manner, the secondary side DC output voltage is controlled to be constant. That is, the power supply is stabilized. Here, in changing the switching frequency, the period during which the main switching element Q1 is off is kept constant, and then the period during which it is on is variably controlled. That is, the conduction angle control for the ON period is performed and the switching frequency control is performed. In this specification, such composite control is referred to as a "composite control method".

FIG. 16 shows operation waveforms of each part under heavy load as the operation of the main part of the power supply circuit having the configuration shown in FIG. Here, the operation on the primary side is mainly shown. Series resonance circuit (N
In B and CB), a resonance operation is performed by the alternating voltage obtained in the drive winding NB, so that a sinusoidal series resonance current I2 is obtained as shown in FIG. 16 (e). Then, as the series resonance current I2 passes through the base current limiting resistor RB, a base current (driving current) IB flows through the base of the switching element Q1 as shown in FIG. 16 (d). The drive current IB causes the switching element Q1 to perform a switching operation.

At this time, the collector current IQ1 flowing through the collector of the switching element Q1 has the waveform shown in FIG. 16 (b). In addition, at both ends of the parallel connection circuit of the [switching element Q1 // parallel resonance capacitor Cr],
As shown in (a), the parallel resonance voltage V1 is generated by the action of this parallel resonance circuit. This parallel resonance voltage V1
As shown in the figure, the period TON in which the switching element Q1 is on is at 0 level, and a sinusoidal pulse waveform is obtained in the period TOFF in which it is off, which corresponds to the operation as the voltage resonance type.

Further, as the switching element Q1 performs the switching operation at the above timing, the winding current I1 flowing through the primary winding N1 has an alternating waveform corresponding to the switching cycle as shown in FIG. 16 (c). Become.

Here, in the period TON during which the switching element Q1 is turned on, the series resonance current I2 of FIG.
The region of positive polarity corresponds to the region of forward bias current of the drive current IB in FIG. Also, the same period TON
In the region where the series resonance current I2 is negative, the reverse bias current of the drive current IB is obtained. Then, the region of the reverse bias current of the drive current IB in this period TON becomes the accumulation time (tstg) of the switching element Q1.

A low speed damper diode D having a long reverse recovery time is provided between the base and emitter of the switching element Q1.
A series circuit of D and a resistor RD is connected. In the period TOFF during which the switching element Q1 is off, the negative series resonance current I2 flows through the resistor RD → the clamp diode DD → the base current limiting resistor RB → the resonance capacitor CB → the drive winding NB, which is shown in FIG. (G) Damper current I
The waveform is obtained in the period TOFF as D1. Then, when the period TON is started next, the parallel resonance capacitor Cr
Charging / discharging energy flows through the clamp diode DD → the base of the switching element Q1 → the collector, and this becomes a negative damper current (ID) at the start (turn-on) of the period TON. Then, when this period ends, the damper diode DD becomes the region of the reverse recovery time and steeply rises in the positive polarity direction, and thereafter, as shown in the figure, gradually becomes 0 level at the end of the period TON. You can get a good waveform.

Corresponding to the flow of the drive current IB and the damper current ID1 as described above, the base-emitter voltage VBE of the switching element Q1 becomes negative during the period TOFF as shown in FIG. 16 (f). In the period TON, the waveform has a sharp negative peak in the negative polarity in the damper period at the start of the period, and when this is finished, an offset is given to the 0 level at a constant level of positive polarity. is there. This offset level is determined by the resistance value of the resistor RD, for example.

FIG. 17 shows the constant voltage control characteristic of the power supply circuit of FIG. 15 which operates as described above. As the load current Io of the secondary side DC output voltage EO1 changes in the range of 0 to 1.5 A, the control current Ic changes as shown in the figure. In other words, the control is performed such that the control current level is decreased as the load current increases and becomes a heavy load condition, and the secondary side DC output voltage EO1 decreases. As a result, the switching frequency fs is controlled so as to decrease as the heavy load condition is met. In addition, as a measure to deal with the fluctuation of the AC input voltage VAC, the AC input voltage VAC = 120V and VAC = 90
Although the case of V is shown, the control current Ic is the AC input voltage VAC under the condition when the AC input voltage VAC = 120V.
= 90V, the switching frequency fs is higher under the condition of AC input voltage VAC = 120V than under the condition of AC input voltage VAC = 90V. This is controlled so that the control current Ic is increased when the level of the AC input voltage VAC is increased and the secondary side DC output voltage EO1 is increased, and accordingly the switching frequency fs is also increased. It is shown that it is controlled so as to be raised.

[0022]

By the way, in the above configuration, the orthogonal control transformer PRT has the resonance current detection winding N
It is a saturable reactor in which D, a drive winding NB, and a control winding NC are wound. FIG. 19 shows the orthogonal control transformer PRT.
Shows the structure of. FIG. 19 (a) is an external perspective view for explaining the overall structure, and FIG. 19 (b) is a sectional perspective view for explaining the winding direction of the winding to be wound. FIG. 19
As shown in (a), the orthogonal control transformer PRT is formed by a three-dimensional core 20 in which two double U-shaped cores 21 and 22 made of ferrite are combined.
One of the double U-shaped cores 21 is shown in FIGS.
4 magnetic legs 21a, 21b, 21
c, 21d. The other double U-shaped core 22 also has four magnetic legs 22a, 22b, 22c, 22 as shown in FIGS. 19 (a) and 19 (b), for example.
It is configured with d. The magnetic legs 21a to 21d, 2 of the two double U-shaped cores 21, 22 are
The three-dimensional core 20 is formed by joining the ends of 2a to 22d.

Each of the magnetic legs 21a to 21d and the magnetic leg 22
For each of the joints a to 22d, see the upper 2
A mylar film of 10 μm is inserted in the set or the two sets in the lower stage so that the gap G = 10 μm. Then, as shown in FIG. 18, the inductance LB of the drive winding NB
The direct current superposition characteristics of the control current Ic = 10 mA to 60 mA
However, the inductance LB changes from 8 μH to 2.5 μH.

Then, as shown in FIG. 19B, for example, two magnetic legs 22 of a double U-shaped core 22 are provided.
A control winding NC is wound around c and 22d, and a detection winding ND and a drive winding NB are wound around magnetic legs 21c and 21b of the double U-shaped core 21. That is, the orthogonal control transformer PRT is configured as a saturable reactor in which the control winding NC is wound in the direction orthogonal to the detection winding ND and the drive winding NB. This orthogonal control transformer P
As the control winding NC of the RT, for example, 1000 T (turn) is wound with a polyurethane-coated copper wire of 60 μmφ, the detection winding ND is 1 T with a polyurethane-coated copper wire of 0.3 mmφ, and the drive winding NB is 0.3 mmφ. It is wound 3T by a polyurethane-coated copper wire.

In such an orthogonal control transformer PRT, the gap G is as small as about 10 μm as described above in order to reduce the amount of control current flowing through the control winding. For this reason, an accuracy error of the gap thickness inevitably occurs at the time of manufacture, but this causes variations in the inductance value of the drive winding NB wound around the orthogonal control transformer PRT. Further, variations in the magnetic permeability of the ferrite core, deviations when the magnetic legs are joined, and the like also cause variations in the inductance value of the drive winding NB. For these reasons, the allowable value of the inductance LB must be such that the inductance value changes by ± 10%. For this reason, variations in the amplification factor hFE of the switching element Q1 and the accumulation time tstg occur, but the constant voltage guarantee range of the composite resonant converter is set against this variation, for example, when the commercial AC power supply is a 100V system, the AC input voltage VAC. = 100V ± 10%,
The variable inductance range of the orthogonal control transformer PRT must be designed with a sufficient margin. In other words, it becomes difficult to design the margin for practical use.

Further, the winding specifications of the orthogonal control transformer PRT are as described above, and it is not necessary to wind the control winding NC, the detection winding ND and the drive winding NB in directions orthogonal to each other. Moreover, the winding process becomes very complicated. Further, joining the four magnetic legs of each of the double U-shaped cores 21 and 22 through the Mylar film without displacement also complicates the assembly process. That is, the orthogonal control transformer PRT
Is difficult to manufacture and cost reduction is difficult.

The DC control current Ic flowing through the control winding NC of the orthogonal control transformer PRT is the DC output voltage E02 line (15V) on the secondary side of the insulating converter transformer PIT.
Line), and the supplied power is 0.9 W to 0.
Although it fluctuates within the range of 15 W, this supplied power is reactive power, and power loss during light load increases.

[0028]

In view of the above problems, the present invention is configured as a switching power supply circuit as follows. In other words, input the DC input voltage and switch
Switching means including a switching element for performing switching
Equipped with. Further, the primary winding and the secondary winding are provided, and the primary winding
The output of the switching means obtained on the wire to the secondary winding
Insulation converter transformer that transmits to
Also the primary winding of the isolated converter transformer and the primary side parallel
Formed by a vibration capacitor,
Primary side parallel resonance provided so that the operation is a voltage resonance type
And a circuit. Also, the primary of the isolated converter transformer
Winding or series with secondary winding of the above isolated converter transformer
Detection winding as a primary winding connected to the
Drive winding in which the drive winding and the tertiary winding are wound
Have a lance. Also, the secondary of the isolated converter transformer
With a secondary resonant capacitor connected to the winding
The secondary side resonance circuit formed is provided. Also with the drive winding
Has a series resonant circuit formed by a resonant capacitor
The switching element based on the output of this series resonant circuit.
Equipped with switching drive means for switching drive the child
It It is also connected in parallel to the tertiary winding,
Series circuit formed by connecting a controller and a conduction control element in series
Equipped with. In addition, the alternating voltage obtained in the secondary side resonance circuit is
Obtain DC output voltage by inputting and rectifying operation
DC output voltage generating means configured in
The current conduction amount of the conduction control element can be changed according to the level.
Constant voltage control for DC output voltage
And a constant voltage control means to be performed.
It was

[0029]

The power supply circuit having the above-mentioned structure has a basic structure as a composite resonance type switching converter having a self-exciting voltage resonance type converter on the primary side. After that, the drive winding and the tertiary winding of the self-excited oscillation drive circuit are magnetically coupled by a drive transformer, and the tertiary winding and the [inductor-conduction control element] series circuit are connected in parallel. Then, by varying the current conduction amount of the series circuit according to the level fluctuation of the secondary side DC output voltage, the oscillation output of the self-excited oscillation drive circuit is changed, whereby the conduction angle and the switching of the switching element are changed. The constant voltage control is realized by the composite control system that controls the frequency at the same time. As a result, in the present invention, when controlling the switching frequency, for example, the orthogonal control transformer can be omitted.

[0031]

FIG. 1 shows the configuration of a power supply circuit as a first embodiment of the present invention. The power supply circuit shown in FIG. 1 has a configuration as a composite resonance type switching converter having a voltage resonance type converter on the primary side and a parallel resonance circuit on the secondary side. In the power supply circuit shown in this figure, first, as a rectifying / smoothing circuit for inputting a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage, a full-wave rectifying circuit including a bridge rectifying circuit Di and a smoothing capacitor Ci. And an AC input voltage V
The rectified smoothed voltage Ei corresponding to a level of AC is generated.

As a switching converter which receives the rectified and smoothed voltage Ei (DC input voltage) and is intermittently connected,
A voltage resonance type converter including a stone switching element Q1 and performing a switching operation by a so-called single-ended system is provided. The voltage resonance type converter here has a self-excited configuration, and a high breakdown voltage bipolar transistor (BJT; junction type transistor) is used as the switching element Q1. Switching element Q
The collector of 1 is connected to the positive electrode of the smoothing capacitor Ci via the primary winding N1 of the insulating converter transformer PIT, and the emitter is connected to the primary side ground.

Further, the collector of the switching element Q1
A parallel resonance capacitor Cr is connected in parallel between the emitters. The capacitance of the parallel resonance capacitor Cr and the primary winding N of the insulating converter transformer PIT
The leakage inductance obtained in 1 forms a primary side parallel resonant circuit. Then, according to the switching operation of the switching element Q1, the resonance operation by the parallel resonant circuit is obtained, so that the switching element Q1
The switching operation of is a voltage resonance type.

A clamp diode DD is connected between the base and emitter of the switching element Q1 in the direction shown in the figure. That is, the anode of the clamp diode DD is connected to the primary side ground (emitter), and the cathode is connected to the base. A low speed recovery type diode element is selected as the clamp diode DD.

Further, the base of the switching element Q1 is connected to the line of the rectified and smoothed voltage Ei via the starting resistor RS. For example, at the time of starting the power source, the starting resistor R1 is connected.
It is designed to start up when the base current obtained via S flows.

The drive transformer CDT is provided to drive the switching element Q1 by self-excitation. In this case, the primary side of the drive transformer CDT is the detection winding N
A, the detection winding NA is an insulating converter transformer P
By being connected in series to the secondary winding N2 of IT, the switching output of the switching element Q1 induced in the secondary winding N2 from the primary winding N1 via the insulating converter transformer PIT is detected. ing. Then, the drive winding NB is wound around the secondary side in which the alternating voltage obtained in the detection winding NA is induced. This drive winding NB
Form a self-excited oscillation drive circuit for switching-driving the switching element Q1. Further, in the case of the present embodiment, the tertiary winding N3 is wound around the primary side of the drive transformer CDT. A series circuit of an inductor LO and a conduction control element Q2 described later is connected to the tertiary winding N3 in parallel.

Here, the drive winding NB and the tertiary winding N3 which are magnetically coupled in the drive transformer CDT are equalized so that the tertiary winding N3 is connected in parallel to the drive winding NB. It can be seen as being present.

The drive transformer CDT around which the above windings are wound is, for example, an H-shaped ferrite magnetic core as shown in FIG. 4 (a), or an EI shown in FIG. 4 (b).
A -12 type ferrite magnetic core can be used. Figure 4
In the case of (a), it is formed by winding the detection winding NA, the drive winding NB, and the tertiary winding N3 on the H-shaped ferrite magnetic core 100. Since the primary side and the secondary side of the CDT are located on the secondary side and the primary side of the insulating converter transformer PIT, in practice, a photocoupler or the like is provided to provide DC insulation. Is required. However, if a triple insulated wire is selected for the drive winding NB, a sufficient insulation state can be obtained without interposing a photocoupler.

In the case of FIG. 4B, the E-shaped core 101 and I
The mold core 102 is arranged as shown. E-shaped core 101 and I
A gap G is formed at the junction of the magnetic legs of the mold core 102. The split bobbin 10 is attached to the central magnetic leg of the E-shaped core 101.
3 is arranged, and the detection winding NA and the drive winding NB are wound around the divided bobbin 103, respectively. A drive transformer CD as shown in FIG. 4 (a) or FIG. 4 (b).
T is, for example, the orthogonal control transformer PR described in FIG.
Compared with the T, the size and weight can be significantly reduced.

As shown in the drawing, a series connection circuit of [drive winding NB-time constant capacitor CB-base current limiting resistor RB] is connected to the base of the switching element Q1. This series connection circuit is a self-excited oscillation drive circuit for switching-driving the switching element Q1 by self-excitation.

In this case, in the drive winding NB of the drive transformer CDT in the self-excited oscillation drive circuit, the detection winding NA is connected in series to the secondary winding N2 of the insulating converter transformer PIT as described above. It is excited by the switching output voltage available on the secondary winding N2. The self-excited oscillation drive circuit includes a capacitor CB and a drive winding NB.
And the inductance of form a series resonance circuit.

The drive winding NB of the self-excited oscillation drive circuit is excited by the detection winding NA to generate an alternating voltage as a drive voltage. The drive voltage causes the series resonance circuit (NB-CB) to oscillate in a self-excited manner to obtain a resonance output. The resonance output passes through the base current limiting resistor RB, so that the base current as a switching drive signal flows through the base of the switching element Q1. As a result, the switching element Q1 performs the switching operation at the switching frequency determined by the resonance frequency of the series resonance circuit. Then, the switching output obtained at the collector is transmitted to the primary winding N1 of the insulating converter transformer PIT.

The insulating converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side. The insulating converter transformer PIT is provided with an EE type core in which, for example, two sets of E type cores made of ferrite material are combined so that their magnetic legs face each other, and a split bobbin is attached to the central magnetic leg of the EE type core. Use primary winding N
1 and the secondary winding N2 are wound separately. A gap is formed with respect to the central magnetic leg. As a result, loose coupling with a required coupling coefficient is obtained. The gap is two sets of E
It can be formed by making each central magnetic leg of the die core shorter than the two outer magnetic legs. Further, as the coupling coefficient k, for example, a loose coupling state of k≈0.85 is obtained, and accordingly, it is difficult to obtain a saturated state.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In the circuit shown in FIG. 1, the detection winding NA is connected in series to the secondary winding N2 as described above. The secondary side parallel resonance capacitor C2 is connected to the series connection of the secondary winding N2 and the detection winding NA.
Are connected in parallel. Therefore, in this case, a parallel resonance circuit is formed by the leakage inductance L2 of the secondary winding N2, the inductance of the detection winding NA, and the capacitance of the secondary side parallel resonance capacitor C2. By this parallel resonance circuit, the alternating voltage induced in the secondary winding N2 and the alternating voltage obtained in the detection winding NA become a resonance voltage. That is, the voltage resonance operation is obtained on the secondary side.

In other words, this power supply circuit is equipped with a parallel resonance circuit for making the switching operation a voltage resonance type on the primary side and a parallel resonance circuit for obtaining the voltage resonance operation on the secondary side. It is a "resonant switching converter".

For the secondary side of the power supply circuit formed as described above, a half of a secondary side rectifying diode DO1 connected to one end of the secondary winding N2 and a smoothing capacitor CO1. A wave rectifier circuit is provided so as to obtain a secondary side DC output voltage EO1 corresponding to a level approximately equal to the alternating voltage induced in the secondary winding N2. Also,
Here, a tap output is provided for the secondary winding N2,
Between the tap output and the secondary side ground, as shown in the figure, the secondary side rectifying diode D02 and the smoothing capacitor CO2
By connecting a half-wave rectification circuit consisting of the following, a low-voltage secondary side DC output voltage EO2 is obtained. In this case, the secondary side DC output voltage EO1 is input to the control circuit 1 as a detection voltage for constant voltage control.

The control circuit 1 functions as an error amplifier which receives the DC output voltage EO1 as a detection input, and outputs a voltage of a level varied according to the level of the DC output voltage EO1 to the MOS-F.
Output to the gate of the conduction control element Q2 as ET. In this case, the control circuit 1 is configured to increase the output voltage level according to the increase in the level of the DC output voltage EO1.

On the secondary side of the insulating converter transformer PIT, the inductor LO is connected to the tertiary winding N3 wound around the drive transformer CDT, and the drain of the conduction control element Q2 is connected to the inductor LO. In addition to the connection, the source is connected to one end side (primary side ground) of the tertiary winding N3. That is, the series connection circuit including the inductor LO and the conduction control element Q2 is connected in parallel to the tertiary winding N3. A diode DD2 for forming a reverse current path is connected in parallel with the conduction control element Q2. A body diode incorporated in the conduction control element Q2 may be selected as the diode DD2. This circuit part is for the parallel circuit consisting of the tertiary winding N3 // inductor LO,
It can be regarded as being formed so as to interpose the conduction control element Q2. In this embodiment, a constant voltage control circuit system including the control circuit 1, a conduction control element Q2, a tertiary winding N3, and a parallel circuit including an inductor LO is configured. The constant voltage control circuit system operates so as to variably control the switching frequency of the switching element Q1, thereby achieving a constant voltage. This operation will be described later.

2 and 3 are waveform charts showing the operation of the main part of the power supply circuit having the configuration shown in FIG. In FIG. 2, when the AC input voltage VAC = 100V, the load power Po
= 150 W, the operation under the heavy load condition is shown, and FIG. 3 shows the operation under the light load condition with the AC input voltage VAC = 100 V and the load power Po = 25 W. In order to obtain the operation shown in these figures, each component is selected as follows. Drive winding NB = 10 μH, tertiary winding N3 = 7T (turn) Detection winding NA = 1T Capacitor CB = 0.56 μF Inductor LO = 4.7 μH

In the drive winding NB wound around the drive transformer CDT, an alternating voltage VNB is generated as shown in FIGS. 2 (f) and 3 (f), and FIG. 2 (d), Figure 3
The drive winding current INB shown in (d) flows. Then, the self-oscillation drive circuit (NB-CB-RB) is operated by the alternating output obtained in the drive winding NB, so that the switching element Q1 is switched to the switching element Q1 shown in FIGS. 2 (c) and 3 (c). The base current IB flows as shown. The base current IB becomes zero level when the base accumulated carrier extinction time tstg is completed, whereby the switching element Q
A transition is made to a period TOFF in which 1 is turned off.

When the period TON during which the switching element is turned on is reached, the clamp diode DD becomes conductive due to the effect of the reverse recovery time trr of the clamp diode DD. At this time, assuming that the current flowing through the clamp diode DD is the current ID, the waveform thereof has waveforms corresponding to the cycle of the base current IB as shown in FIGS. 2 (e) and 3 (e). Then, by making the clamp diode DD conductive,
The drive winding current INB passes through the clamp diode DD,
Further, current flows through the PN junction of Q1 base → Q1 collector. As a result, as the collector current IQ1 shown in FIGS. 2B and 3B, first, a waveform flowing in the negative direction is obtained, and the base current IB at this time is also obtained.
A positive waveform corresponding to the collector current IQ1 is obtained.
Then, after that, the clamp diode DD is turned off and the switching element Q1 becomes conductive,
Via the collector → emitter of the switching element Q1,
The collector current IQ1 that increases in the positive polarity flows.

When the switching element Q1 performs the switching operation as described above, the resonance voltage V1 obtained across the primary side parallel resonance capacitor Cr is as shown in FIG.
As shown in (a) and FIG. 3 (a), a waveform having a 0 level in the period TON in which the switching element Q1 is on and a sinusoidal pulse in the period TOFF in the off state is obtained. This indicates that the primary side switching converter is a voltage resonance type operation. Further, the collector current IQ1 becomes 0 level during the period TOFF as shown in FIGS. 2 (b) and 3 (b).

The resonance voltage generated on the primary side is excited in the secondary winding N2 on the secondary side of the insulating converter transformer PIT. As a result, the secondary side parallel resonance current I2 is applied to the secondary winding N2.
2 will flow, and the waveform of this current I2 is shown in FIG.
As shown in (g) and FIG. 3 (g), the waveform flows as an alternating waveform corresponding to the switching cycle.

The tertiary winding N3 is excited by the alternating voltage obtained in the detection winding NA, so that FIG. 2 (h) and FIG. 3 (h)
As shown as the third winding voltage VN3, a positive polarity waveform is obtained in the period TON, a positive polarity peak is obtained at the beginning of the period TOFF, and an alternating waveform that inverts to the negative polarity in the remaining period of the period TOFF is obtained. can get. Then, the drain current IQ2 flows through the conduction control element Q2 whose current conduction amount is controlled by the control circuit 1 as shown in FIGS. 2 (i) and 3 (i).

Here, as described in FIG. 1, the tertiary winding N3 and the inductor LO wound around the drive transformer CDT.
Can be regarded as being connected in parallel via the conduction control element Q2. Then, by varying the current conduction amount of the conduction control element Q2, the impedance of the inductor LO connected in series with the conduction control element Q2 is varied.

In the power supply circuit shown in FIG. 1, the operation as the waveform shown in FIGS. 2 and 3 can be obtained. As can be seen by comparing FIGS. 2 and 3, for example, under a light load condition. When the secondary side DC output voltage rises due to, for example, the switching frequency is increased. That is, switching frequency control is performed, and a constant voltage is achieved by this. The operation of changing the switching frequency in this way is obtained as follows.

For example, when the AC input voltage VAC increases or the load power decreases, the secondary side DC output EO1
Suppose that the level of has risen. Then, as described with reference to FIG. 1, in the control circuit 1, the current conduction amount (drain current IQ2) flowing in the conduction control element Q2, which is a MOS-FET, is increased in accordance with the rise of the secondary side DC output voltage E01. become. When the drain current IQ2 increases, the current flowing through the inductor LO also increases and the impedance is reduced. Here, as described with reference to FIG. 1, since the tertiary winding N3 and the series circuit of the conduction control element Q2-inductor LO are in parallel, the current conduction amount (drain current IQ2) shunted from LO to Q2 is Depending on the increase, the tertiary winding N
The amount of current flowing through 3 will decrease. This lowers the alternating voltage level generated in the tertiary winding N3, so that the drive winding N on the secondary side of the live transformer CDT is reduced.
The alternating voltage excited by B also decreases.

Along with this, the current level of the drive winding current INB decreases, so that the self-excited oscillation drive circuit (NB-CB-RB).
As a result, the level of the base current IB passed through the base of the switching element Q1 also decreases. Therefore, the period in which the base current IB has the positive polarity becomes shorter, which means that the period in which the switching element Q1 is turned on becomes shorter, which increases the switching frequency. Therefore, as a result, as the secondary side DC output voltage E01 increases, the switching frequency is variably controlled so as to increase. By variably controlling the switching frequency, for example, the resonance impedance of the primary side parallel resonance circuit is changed, and the power transmitted from the primary side to the secondary side of the insulating converter transformer PIT is also changed. It will be. As a result, finally, the level of the secondary side DC output voltage E01 is also variably controlled, and the power supply is stabilized.

Further, as can be seen from the above operation,
In this power supply circuit according to the present embodiment, the switching element Q is used to variably control the switching frequency.
The period TOFF in which 1 is off is kept constant, and the period TON in which it is on is variable. That is, the control is performed so that the TON period in which the switching element Q1 is on becomes longer when the load is heavy, and the TON period becomes shorter when the load is light. Therefore, in the present embodiment, the "combined control method" is adopted in which the conduction angle control for the ON period is performed and the switching frequency control is executed.

The configuration of the constant voltage control circuit according to the present embodiment has been described above. With this circuit configuration, FIG.
The orthogonal control transformer PRT shown in (3) will be omitted. As a result, in the present embodiment, the problem of the variation in the inductance value of the drive winding NB due to the variation in the gap at the time of manufacturing the orthogonal control transformer PRT is solved. Therefore, the margin for the range of the AC input voltage VAC can be set to be small, and the circuit design can be facilitated. Further, the problem of difficulty in the manufacturing process of the orthogonal control transformer PRT is solved. AC
/ DC power conversion efficiency can be improved.

Further, unlike the example of FIG. 15, the control current is not supplied to the control winding NC of the orthogonal control transformer PRT to control the switching frequency, so that the power loss can be reduced, and particularly the control can be performed. The effect becomes greater when the current level increases and the load is light. Further, the drive transformer CDT is an ultra-compact EI-, as described in FIG.
It can be constituted by a 12-type ferrite magnetic core or an H-shaped ferrite magnetic core, and can be made much smaller and lighter than in the case of providing the orthogonal control transformer PRT as shown in the prior art of FIG.

Further, the MOS-FET (Q2) may be a low withstand voltage small capacity product having a withstand voltage of 30 V and a rated current of 1 A or less. For example, the present applicant previously provided an active clamp circuit that clamps a primary side parallel resonance voltage or a secondary side resonance voltage for a composite resonance type switching converter, and stabilizes the power supply by controlling the conduction angle of the active clamp circuit. Various configurations have been proposed for achieving this, but in this case, switching elements such as MOS-FETs that form an active clamp circuit may be changed depending on the primary side parallel resonance voltage level or the secondary side resonance voltage level. It is necessary to select a high withstand voltage product, which is disadvantageous in terms of cost and size. On the other hand, in the present embodiment, since a low withstand voltage small capacity product is selected for the MOS-FET (Q2), it is possible to realize cost reduction and size reduction and weight reduction accordingly.

Further, in the present embodiment, the switching element is driven by using the alternating voltage excited in the drive winding NB by the detection winding NA. Here, since the detection winding NA is an inductance forming a secondary side parallel resonance circuit, a resonance waveform is obtained in the detection winding NA, and therefore the drive winding NB excited by the resonance waveform is obtained.
As a result, a resonance waveform can be obtained. As a result, as the base current IB flowing through the switching element Q1, as shown in the waveform diagrams of FIGS. 2 (c) and 3 (c), the backward base current IB2 is the reverse base current IB2 rather than the forward base current IB1. , The peak value becomes large and the switching element Q
Since the fall time of 1 is reduced, the switching loss when the switching element Q1 is off is reduced accordingly. Therefore, the heat generation of the switching element Q1 can be further reduced.

FIG. 5 shows a configuration example of a power supply circuit according to the second embodiment of the present invention. In this figure, the same parts as those in FIG. In the power supply circuit shown in this figure, the detection winding NA wound around the drive transformer CDT is connected in series to the primary winding N1 of the insulating converter transformer PIT. In this case, one end of the detection winding NA is connected to the primary winding N1 and the other end is connected to the collector of the switching element Q1. Depending on such a connection form, the detection winding NA forms a primary side parallel resonance circuit together with the leakage inductance of the primary winding N1 of the insulating converter transformer PIT. Then, even as the alternating voltage obtained at the detection winding NA, the waveform as the parallel resonance voltage is obtained.

The switching operation and the constant voltage control operation in the power supply circuit having such a configuration are shown in FIGS.
This is the same as the case described above, and the same effect as that of the power supply circuit shown in FIG. 1 can be obtained.

On the secondary side of the power supply circuit shown in FIG. 5, a full-wave rectifying circuit is formed by the rectifying diodes DO1 and DO2 connected to both ends of the secondary winding N2 and the smoothing capacitor C01. The full-wave rectifier circuit is used to obtain the secondary side DC output voltage EO1. Also,
As in the case of FIG. 1, the secondary side DC output voltage EO2 is obtained by a half-wave rectification circuit formed by a rectification diode DO2 connected to the tap output of the secondary winding N2 and a smoothing capacitor CO2. There is.

FIG. 6 shows a configuration example of a switching power supply circuit according to the third embodiment. In this figure, the same parts as those in FIGS. 1 and 5 are designated by the same reference numerals and the description thereof will be omitted. The basic circuit configuration of the power supply circuit shown in this figure is the same as that of the power supply circuit according to the first embodiment shown in FIG. However, in the circuit shown in FIG.
A power saving capacitor CD and a power saving resistor RD are newly added. The connection form is as follows.
And the power-saving capacitor CD are connected in parallel to form a parallel circuit, and this parallel circuit (RD-CD) is connected to the connection point between the anode of the clamp diode DD and the primary side ground. In other words, this parallel circuit (RD-CD)
It is provided so as to be inserted in the reverse current path via the clamp diode DD. The operation of this circuit configuration will be described below with reference to FIGS. 7 and 8.

7 and 8 are waveform diagrams showing the operation of the main part of the power supply circuit according to the present embodiment. Here, as can be seen by comparing FIG. 2 and FIG. 3 described in the first embodiment, regarding the waveforms of the respective portions shown in FIG. 7 and FIG. Are similar. This is because the basic configuration of the power supply circuit of the present embodiment is the first
Since the power supply circuit is similar to that of the embodiment, the basic operation is also similar. Therefore, in FIGS. 7 and 8, the description of the operation described in FIGS. 2 and 3 will be omitted, and the operation of the newly added portion will be described. In order to obtain the operation shown in these figures, each component is selected as follows. Drive winding NB = 10 μH, tertiary winding N3 = 7T (turn) Detection winding NA = 1T Capacitor CB = 0.56 μF Inductor LO = 4.7 μH Power saving resistor RD = 22Ω Power saving capacitor CD = 0.1 μF

First, in the primary side self-excited oscillation drive circuit,
The period TOFF in which the switching element Q1 is off overlaps with the period DON in which the clamp diode DD is on. In this period DON, the negative drive winding current INB shown in FIGS. 7D and 8D is generated. , Drive winding NB → power saving capacitor CD → clamp diode DD → base current limiting resistor RB
→ Flows through the time constant capacitor CB. This is because the current ID (FIG. 7) obtained by the clamp diode DD in the period TOFF within the period DON in FIGS.
(E) and FIG. 8 (e)) waveforms in the positive polarity direction and current ICD obtained by the capacitor CD (FIGS. 7 (g) and 8 (g))
It is also shown by the correspondence between the waveforms of and.

Next, as shown in the figure, when the switching element Q1 is turned on, the beginning of the period TON is the end of the period Don, and also in this period, FIG. 7 (e) and FIG.
The drive winding current INB continues to flow through the power saving capacitor CD and the clamp diode DD, as shown by the current ID in (e) and the current ICD in FIGS. 7 (g) and 8 (g). . The current flowing through the clamp diode DD in this period is a damper current, and the current is as shown in FIGS.
The switching element Q1 is used as the base current IB shown in (c).
Flows into the base from the base through the PN junction to the collector.

Here, the current INB flows through the power-saving capacitor CD as described above, so that the capacitor CD is charged. And
During the period TON, when the clamp diode DD is turned off to the period DOFF, the capacitor CD discharges the charge, and the discharged current is the power saving resistor R.
It flows to D and is consumed here. As shown in FIGS. 7 and 8, this operation is performed when the period DOFF is reached.
(E), current ID of FIG. 8 (e), FIG. 7 (g), FIG.
It is shown that the waveform of the current ICD in (g) is at 0 level, which means that the current discharged by the capacitor CD is flowing not in the clamp diode DD direction but in the resistance RD direction. It is a thing. At the same time, the fact that the waveforms of the current IRD in FIGS. 7 (h) and 8 (h) are inverted in the negative direction also indicates that the current discharged by the capacitor CD is flowing in the resistor RD. Has been done.

As described above, in the circuit configuration of this embodiment including the parallel circuit of the power saving resistor RD and the power saving capacitor CD, the drive winding current INB as the alternating output of the drive winding NB is saved in the period DON. The power capacitor CD is charged and the power saving resistor RD is started to be discharged in the period DOFF. That is, although the on / off timings of the clamp diode DD and the switching element Q1 are not completely coincident with each other, the power saving capacitor CD is charged and turned off at the timing when the switching element Q1 is turned on. Power saving capacitor C
The operation of discharging the charged electric charge of D to the power saving resistor RD is obtained. Then, as a result of an experiment using this circuit configuration, it was found that the base current IB of the switching element Q1 increased.
As the base current IB increases, the collector saturation voltage (VCE (SAT)) of the switching element Q1 decreases, which results in conduction loss during the period TON during which the switching element Q1 is on and the Q1 turn-off time. The switching loss will be reduced. As a result, the power supply circuit according to the present embodiment can reduce the input power by 21.5W.

As described above, in the power supply circuit of this embodiment, the power saving resistor RD is added to the primary side self-excited oscillation drive circuit.
And a power-saving capacitor CD in parallel circuit, the clamp diode DD is turned off during the period TON when the switching element Q1 is turned on.
By discharging the negative current charged in the power saving capacitor CD and consuming it by the power saving resistor RD, Q1
The base current IB of is increased. Thus, the conduction loss in the period TON and the switching loss at the time of Q1 turn-off are reduced, and the input power of the power supply circuit is reduced.

Next, FIG. 9 shows a structural example of a switching power supply circuit as a fourth embodiment. In this figure, the same parts as those in FIG. 1, FIG. 5 and FIG. The power supply circuit shown in this figure has the same basic configuration as the circuit shown in FIG. 5 as the second embodiment. That is, the detection winding NA wound around the drive transformer CDT is the insulating converter transformer P.
It is connected in series with the primary winding N1 of IT. Then, in the circuit of the clamp diode DD and the power saving resistor RD, the power saving capacitor CD is connected in parallel to the power saving resistor RD. Also, with the configuration of this embodiment, the same effect as that of each of the embodiments described above can be obtained.

By the way, in the power supply circuits of FIGS. 5 and 9 in which the detection winding NA is provided on the primary side among the above-described embodiments, the rectification circuit system provided on the secondary side is shown in each drawing. Without being limited to the configuration, for example, FIG.
It may be possible to adopt the configuration shown in FIG. In FIG. 10, the secondary side parallel resonant circuit (N2 // C
2), a full-wave rectification circuit composed of a bridge rectification circuit DBR and a smoothing capacitor CO1 is connected in the illustrated connection form to obtain a secondary side DC output voltage EO1.

Further, in FIG. 11, for the secondary side parallel resonance circuit (N2 // C2), the rectifying diode DO1, the rectifying diode DO2 and the smoothing capacitor C are provided as shown in the figure.
By connecting OA and COB, a voltage doubler rectifier circuit based on the full-wave rectification method is formed. In this case, the secondary side DC output voltage EO1 corresponding to twice the alternating voltage level generated in the secondary winding N2 is obtained as the voltage across the series connection circuit of the smoothing capacitors COA-COB.

Further, as the configuration of the secondary side shown in FIG. 12, the secondary side series resonance capacitor Cs is connected in series to the winding end portion of the secondary winding N2. by this,
On the secondary side of the insulating converter transformer PIT, the leakage inductance of the secondary winding N2 and the capacitance of the secondary side series resonant capacitor Cs form a secondary side series resonant circuit. Therefore, in this case, the primary side parallel resonance circuit (N1, C
r) and a secondary side series resonance circuit (N
2, Cs) constitutes a composite resonance type converter. And, for this secondary side series resonant circuit,
By connecting the bridge rectification circuit DBR and the smoothing capacitor CO1 as shown in FIG. 10, full-wave rectification of the secondary side series resonance voltage generated by the resonance action of the secondary side series resonance circuit (N2, Cs) is performed. A wave rectifier circuit is formed. The secondary winding N is connected to both ends of the smoothing capacitor CO1.
Therefore, the secondary side DC output voltage EO1 corresponding to the equal voltage of the alternating voltage level generated in 2 is obtained.

In FIG. 13, the rectifying diode DO1, the rectifying diode DO2, and the smoothing capacitors COA and COB are connected to the secondary side series resonant circuit (N2, Cs) as shown in FIG. It forms a voltage rectifier circuit.

In FIG. 14, two sets of secondary side series resonance capacitors Cs1 and Cs2 are connected to the secondary winding N2 as shown in the figure, and four rectifying diodes DO1, DO2 and D03 are further connected. , D0 4 are connected by the illustrated connection form to form a secondary side rectifier circuit. A quadruple voltage rectifier circuit is formed as the secondary side rectifier circuit configured in this manner. In explaining the operation of this quadruple voltage rectifier circuit, [series resonance capacitor Cs1, rectifier diode D0
1, the operation of the circuit composed of D02 and the smoothing capacitor COA] will be described. First, during the period when the rectifying diode DO1 is off and the rectifying diode D02 is on,
Due to the leakage inductance of the secondary winding N2 and the series resonance action of the series resonance capacitor Cs1, the rectified current rectified by the rectification diode D02 is applied to the series resonance capacitor Cs.
The operation of charging for 1 is obtained. Then, during the period in which the rectifying diode D02 is turned off and the rectifying diode D01 is turned on to perform the rectifying operation, a series resonance occurs in which the potential of the series resonance capacitor Cs1 is added to the voltage induced in the secondary winding N2. Smoothing capacitor COA
The operation is performed for charging. By performing the rectifying operation as described above, a DC voltage (rectified smoothed voltage) corresponding to almost twice the induced voltage of the secondary winding N2 is obtained across the smoothing capacitor COA. Further, the same operation is performed by a rectifying circuit composed of a set of [series resonance capacitor Cs2, rectifying diodes DO3, DO4, smoothing capacitor COB], and by the same operation, the induced voltage of the secondary winding N2 is almost across the smoothing capacitor COB. A DC voltage corresponding to twice the voltage will be obtained.

As a result of the voltage doubler rectifying operation performed by the rectifying circuits of the respective stages as described above, the induced voltage of the secondary winding N2 is applied across the smoothing capacitor COA and the smoothing capacitor COB connected in series. The secondary side DC output voltage EO1 corresponding to almost four times is obtained.

Although the embodiments have been described above, the present invention is not limited to the configurations shown in the drawings as the above embodiments. For example, in each of the above-described embodiments, the case of the single-end system including one set of switching elements is shown, but the so-called push-pull system including two sets of switching elements
It may be a self-excited voltage resonance converter. The secondary side may also be provided with a rectifier circuit having a circuit configuration other than that shown in each drawing.

[0082]

As described above, the present invention has a basic structure of a composite resonance type switching converter having a self-exciting voltage resonance type converter on the primary side. Then, for constant voltage control, the drive winding and the tertiary winding of the self-excited oscillation drive circuit are magnetically coupled by a drive transformer, and then the tertiary winding and the series circuit of the inductor and the conduction control element are connected in parallel. Form a circuit to connect. Then, by varying the amount of current conduction in the series circuit according to the level fluctuation of the secondary side DC output voltage, the voltage generated in the tertiary winding is dropped, and the oscillation frequency of the self-excited oscillation drive circuit is reduced. (Switching frequency) is variably controlled. With such a configuration, constant voltage control by a composite control method that simultaneously changes the conduction angle and the switching frequency of the switching element can be realized, which is required for the switching frequency control. It is possible to omit the orthogonal control transformer. Therefore, in the present invention, the problem of the variation in the inductance value due to the variation in the gap of the orthogonal control transformer is solved, and the variation in the inductance of the drive transformer in the present invention is about ± 5%. For this reason, it is possible to set a small margin for the range of the AC input voltage, so that the circuit design can be facilitated. Further, the problem of difficulty in the manufacturing process of the orthogonal control transformer is solved. Further, the AC / DC power conversion efficiency can be improved.

Since the control current is not supplied to the control winding of the quadrature control transformer to control the switching frequency, the power loss is reduced, and particularly, the effect of the control current supply is increased at a light load. growing. Further, a small and lightweight drive transformer can be selected, and a low withstand voltage small capacity MOS-FET can be used as the auxiliary switching element, which is preferable in terms of design and cost.

Further, according to the present invention, since the detection winding is included in the primary side parallel resonance circuit or the secondary side resonance circuit, the alternating voltage excited in the drive winding by the detection winding. The switching drive signal obtained by using the signal corresponds to a sine waveform. Therefore, as for the base current flowing through the switching element, the reverse base current has a larger peak value than the forward base current and the fall time of the switching element is shorter, so that the switching loss when the switching element is off is reduced. Will be reduced. In addition, there is an advantage that the switching element generates less heat.

Furthermore, if a parallel circuit of a power-saving capacitor and a resistor is connected in series to the diode provided in the primary side self-excited oscillation drive circuit, conduction loss and turn-off in the ON period of the switching element will occur. It becomes possible to reduce the switching loss at the time and to reduce the input power of the power supply circuit.

In the power supply circuit according to the embodiment of the present invention, since the triple insulated wire is selected for the drive winding of the drive transformer, it is sufficient to provide a photocoupler or the like to prevent direct current insulation. A good insulation state can be obtained. Therefore, the number of parts can be reduced, and the cost of parts can be reduced.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a waveform diagram showing an operation (at a heavy load) of a main part of the power supply circuit according to the first embodiment.

FIG. 3 is a waveform diagram showing an operation (at a light load) of a main part of the power supply circuit according to the first embodiment.

4A and 4B are a perspective view and a sectional view showing a structural example of a drive transformer.

FIG. 5 is a circuit diagram showing a configuration example of a switching power supply circuit according to a second embodiment of the present invention.

FIG. 6 is a circuit diagram showing a configuration example of a switching power supply circuit according to a third embodiment of the present invention.

FIG. 7 is a waveform chart showing an operation (under heavy load) of a main part of the power supply circuit according to the third embodiment.

FIG. 8 is a waveform chart showing an operation (at a light load) of a main part of the power supply circuit according to the third embodiment.

FIG. 9 is a circuit diagram showing a configuration example of a switching power supply circuit according to a fourth embodiment of the present invention.

FIG. 10 is a circuit diagram showing another configuration example on the secondary side of the embodiment.

FIG. 11 is a circuit diagram showing another configuration example on the secondary side of the embodiment.

FIG. 12 is a circuit diagram showing another configuration example of the secondary side of the embodiment.

FIG. 13 is a circuit diagram showing another configuration example on the secondary side of the embodiment.

FIG. 14 is a circuit diagram showing another configuration example of the secondary side of the embodiment.

FIG. 15 is a circuit diagram showing a configuration example of a switching power supply circuit as a prior art.

16 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

17 is an explanatory diagram showing constant voltage control characteristics of the power supply circuit shown in FIG.

18 is an explanatory diagram showing a DC superimposition characteristic of the inductance of the drive winding NB in the circuit shown in FIG.

FIG. 19 is a perspective view and a cross-sectional view showing a structural example of an orthogonal control transformer.

[Explanation of symbols]

1 Control circuit, Di-fridge rectifier circuit, RS start resistor, RB base current limiting resistor, D01, D02 Secondary side rectifier diode, DD, DD2 clamp diode, C
i, C01, C02 smoothing capacitor, CB time constant capacitor, Q1 switching element, Q2 conduction control element (transistor element), PIT insulation converter transformer, CDT drive transformer, N1 primary winding, N2
Secondary winding, N3 tertiary winding, Cr primary side parallel resonance capacitor, C2 secondary side parallel resonance capacitor, NA detection winding,
NB drive winding, LO inductor, CD power saving capacitor, RD power saving resistor

─────────────────────────────────────────────────── ─── Continuation of front page (58) Fields surveyed (Int.Cl. ⁷ , DB name) H02M 3/00-3/44

Claims (3)

(57) [Claims] 1. Switching is performed by inputting a DC input voltage.
A switching means having a switching element for performing, a primary winding and a secondary winding are provided, and the primary winding is obtained.
The output of the switching means is transmitted to the secondary winding.
An insulating converter transformer for transmitting, and at least a primary winding of the insulating converter transformer
It is formed by the primary side parallel resonance capacitor and
It is provided so that the operation of the
Primary side parallel resonance circuit and the primary winding of the insulation converter transformer or the insulation coil
Primary winding connected in series with secondary winding of inverter transformer
A detection winding as a wire and a drive winding as a secondary winding,
Connected to the drive transformer on which the tertiary winding is wound, and the secondary winding of the isolated converter transformer.
A secondary-side resonance capacitor
Vibration circuit, a series formed by the drive winding and the resonance capacitor
Having a resonant circuit, based on the output of this series resonant circuit
A switch for switching driving the above switching element
Driving means and the tertiary winding, which are connected in parallel to the inductor and the inductor.
Rectification by inputting the alternating voltage obtained in the secondary side resonance circuit and the series circuit formed by connecting the communication control elements in series
It is configured to obtain a DC output voltage by performing an operation
DC output voltage generating means and the conduction control element according to the level of the DC output voltage
By changing the current conduction amount of
Constant voltage control hand to make constant voltage control about
A switching power supply circuit comprising: a stage . 2. The switching driving means includes a parallel circuit of a capacitor and a resistor, and the parallel circuit charges the capacitor by the output of the series resonance circuit in correspondence with a period in which the switching element is off. Is done
2. The switching power supply according to claim 1 , wherein the switching power supply is connected so as to obtain an operation of discharging the electric charge charged in the capacitor to the resistor according to a period in which the switching element is turned on. circuit. 3. The switching power supply circuit according to claim 1 , wherein a triple insulated wire is used for the drive winding.