JP2019017125A - Device and method for driving induction motor - Google Patents

Abstract

An initial speed including zero speed in vector control without a speed sensor of an induction motor is estimated in a short time and with a low torque shock. In an induction motor driven by vector control without a speed sensor, when an AC voltage is applied to the induction motor, the current vector obtained from the current detector according to the rotor frequency of the induction motor draws a circular locus. Thus, by estimating the rotor frequency based on the phase difference between the voltage vector and the current vector obtained from the AC voltage, the initial speed including zero speed can be estimated at high speed and with low torque shock. [Selection] Figure 1

Description

  The present invention relates to a drive device and a drive method for an induction motor, and in particular, the drive device and the drive method include a control device and a control method that can reduce vehicle body vibration when the railway vehicle is stopped or restarted from coasting. It is characterized by a point.

  In the vector control without the speed sensor of the induction motor, the rotational frequency of the rotor is controlled based on the AC voltage command value and the detected current value applied to the induction motor by the control software recorded in the microcomputer provided in the power converter (drive device) such as an inverter. An algorithm for estimating (hereinafter referred to as “rotor frequency”) is provided to estimate the rotor frequency. Therefore, when restarting from a stop or free-run state in which the induction motor is not driven by the drive device, it is necessary to perform initial speed estimation that instantaneously estimates the rotor frequency immediately after the start of energization. In this initial speed estimation, in addition to reliably estimating the rotor frequency without failure, it is required not to generate a torque shock of the induction motor during the estimation period.

In the vector control of an induction motor, physical quantities such as voltage, current, and magnetic flux are defined in a rotating coordinate system in which an excitation (magnetic flux) axis that rotates with a magnetic pole of the induction motor is defined as a d axis and an axis orthogonal to the d axis is defined as a q axis. handle. Here, the q-axis is an axis for operating the torque current. Expression (1) is an expression representing the motor torque τ. In general vector control, an arbitrary torque τ is obtained by freely adjusting the d-axis secondary magnetic flux Φ _2d and the q-axis current I _q after controlling the q-axis secondary magnetic flux Φ _2q to zero. .
Here, P _m is the number of pole pairs of the induction motor, M is a mutual inductance, L ₂ is a secondary self-inductance, and I _d is a d-axis current.

However, in speed sensorless vector control, if the estimated rotor frequency has an error with respect to the actual rotor frequency, a q-axis secondary magnetic flux Φ _2q that is normally controlled to zero is generated. When the q-axis secondary magnetic flux Φ _2q is generated, the term of (Φ _2q · I _d ) in the equation (1) does not become zero, and an unintended torque shock occurs. This q-axis secondary magnetic flux Φ _2q is defined by equation (2).
Here, ω _s is a slip frequency, T ₂ is a second-order time constant, and s is a differential operator.

In other words, the error of the rotor frequency estimation value is an error of the slip frequency ω _s . When this slip frequency ω _s error occurs, the term (I _q · M−ω _s · Φ _2d · T ₂ ) in equation (2) does not become zero, and the residual of this term is the time constant T ₂ . A q-axis secondary magnetic flux Φ _2q is generated by the primary delay. That is, to reduce the initial speed estimation time corresponding to the time when the residual of (I _q · M−ω _s · Φ _2d · T ₂ ) is as short as possible, the q-axis secondary magnetic flux Φ _2q Occurrence is prevented, leading to a reduction in torque shock during initial speed estimation.

  For these reasons, improved techniques have been proposed for the purpose of speeding up the estimation time and reducing torque shock in the initial speed estimation of the speed sensorless vector control of the induction motor.

  In Patent Document 1, in a control device that detects pulsation superimposed on a current waveform by velocity electromotive force, a pulsation signal superimposed on a current feedback value on a fixed coordinate is extracted, and the rotational coordinate having a predetermined coordinate rotation frequency is extracted. A technique for obtaining a free-running rotation frequency and a rotation direction by converting the pulsation signal and detecting the frequency of the pulsation signal on the rotation coordinate is disclosed.

In Patent Document 2, in order to accurately identify the rotation speed of the motor in the free-run state of speed sensorless vector control, the motor control unit has a frequency sweep unit that performs frequency sweep when performing a speed search, When the polarity of the detected q-axis current _Iq is reversed during the frequency sweep, if the value of the detected d-axis current _Id is equal to or less than a predetermined determination threshold, the motor rotation speed (actual speed) and the command rotation speed (sweep) (Frequency) is determined to match.

JP 2004-350459 A JP 2013-106461 A

  The inventor of the present application diligently studied that initial speed estimation in speed sensorless vector control is performed in a short time and with low torque shock, and as a result, the following knowledge was obtained.

  According to the technique described in Patent Document 1, since the pulsation itself does not occur when the rotor frequency is around 0 Hz, and the zero speed cannot be estimated in principle, it is necessary to switch the estimation method according to the speed (exception processing). There is a problem that the control system becomes complicated.

  Further, according to the technique described in Patent Document 2, it is necessary to sweep (sweep) the frequency when performing the speed search from the highest frequency, and there is a problem that the initial speed estimation time becomes long.

  An object of the present invention is to estimate an initial speed including zero speed in vector control without a speed sensor of an induction motor in a short time and with a low torque shock.

  The present invention uses the property that the current vector obtained from the current detector according to the rotor frequency of the induction motor draws a circular locus when an AC voltage is applied to the induction motor, and the voltage vector and current obtained from the AC voltage are used. The rotor frequency is estimated based on the phase difference of the vector.

  According to the present invention, in an induction motor driven by vector control without a speed sensor, an initial speed including a zero speed can be estimated at a high speed and with a low torque shock.

FIG. 1 is a functional block diagram of an induction motor driving apparatus according to Embodiment 1 of the present invention. FIG. 2 is a diagram showing a T-type equivalent circuit of the induction motor. FIG. 3 is a diagram showing the locus of the current vector when the rotor frequency changes when the voltage vector is applied. FIG. 4 is a diagram illustrating the principle of speed estimation in the first embodiment. FIG. 5 is a diagram illustrating an example of a circular locus of the current vector when the voltage vector and the primary frequency are changed. FIG. 6 is a diagram illustrating the effect of the initial speed estimation according to the first embodiment. FIG. 7 is a functional block diagram of an induction motor driving apparatus according to Embodiment 2 of the present invention. FIG. 8 is a functional block diagram of an induction motor driving apparatus according to Embodiment 3 of the present invention. FIG. 9 is a functional block diagram of an induction motor driving apparatus according to Embodiment 4 of the present invention. FIG. 10 is a diagram for explaining the accuracy of speed estimation according to the present invention. FIG. 11 is a functional block diagram of an induction motor driving apparatus according to Embodiment 5 of the present invention. FIG. 12 is a functional block diagram of an induction motor drive device according to Embodiment 6 of the present invention. FIG. 13 is a diagram illustrating a railway vehicle having a drive device for an induction motor according to Embodiment 7 of the present invention.

  Hereinafter, Examples 1 to 7 will be described in detail with reference to the drawings as modes for carrying out the present invention. In each embodiment, the same reference numerals indicate the same constituent elements or constituent elements having similar functions. In addition, the structure demonstrated below is only an Example to the last, and it is not the meaning which intends that the embodiment which concerns on this invention is limited to the following specific aspects.

  FIG. 1 is a diagram illustrating functional blocks of a drive device for an induction motor according to Embodiment 1 of the present invention. The configuration when the induction motor is restarted from the stop or free run state by vector control without speed sensor is shown as a functional block.

  The drive device 1 shown in FIG. 1 is mounted with a control program for vector control of the induction motor 3 connected as a load without a speed sensor. However, in FIG. 1, only the minimum configuration necessary for calculating the initial speed estimation is shown, and a power converter composed of a driving transistor such as an IGBT (Insulated Gate Bipolar Transistor) and a power device such as a diode ( The main circuit) and other control configurations are not shown.

The drive device 1 drives the induction motor 3 by switching control modes between initial speed estimation and normal operation. When the rotor of the induction motor 3 is restarted from a stopped or free-run state and energized, the initial speed estimation command generator 31 is a d-axis that is an excitation axis (flux axis) on rotational coordinates in vector control. D-axis voltage command V _d ^* , q-axis voltage command V _q ^* orthogonal to d-axis, and primary frequency command f ₁ ^* are output. After the initial speed estimation is completed, the control mode is switched by the control mode switching unit 22 in order to output the command value of the vector control unit 30 responsible for control during power running or regenerative operation.

The coordinate converter 20 uses the phase angle θ obtained by integrating the d-axis voltage command V _d ^* , the q-axis voltage command V _q ^*, and the primary frequency command f ₁ ^* by the integrator 21 to obtain a rotation coordinate from the dq coordinate. Three-phase conversion is performed, and command values V _u ^* , V _v ^*, and V _w ^* of a three-phase AC voltage waveform are generated.

The voltage output device 2 converts the command values V _u ^* , V _v ^*, and V _w ^* of the three-phase AC voltage waveform into a PWM (Pulse Width Modulation) signal, and then a drive circuit and a main circuit (not shown, but a voltage output device) via included) _2, is applied to the induction motor 3 and V u, _{V v,} and _{V w} as 3-phase AC voltage.

  The induction motor 3 generates a rotational torque by the interaction between the rotating magnetic field generated on the stator side by the application of the three-phase AC voltage and the induced current flowing in the rotor conductor when the rotor slips.

The current detector 4 is configured by a Hall CT (Current Transformer) or the like, and detects U-phase, V-phase, and W-phase three-phase currents I _u , I _v, and I _w that flow through the induction motor 3 along with their waveforms. However, it is not always necessary to detect all three phases of current with the current detector 4, and any two phases are detected, and the remaining one phase is calculated by assuming that the three-phase current is in an equilibrium state. Good.

The coordinate converter 5 converts the three-phase currents I _u , I _v, and I _w of the induction motor detected by the current detector 4 into dq coordinates of a rotating coordinate system used for vector control, and a vector as I _d and I _q It outputs to the control part 30 and the phase difference calculating part 6.

  The vector control unit 30 is configured so that, during power running or regenerative operation, a current controller (not shown) provided in the vector control unit 30 matches the current command value based on the detected Id and Iq. The output voltages Vd * and Vq * are adjusted.

The phase difference calculation unit 6 calculates a phase difference θ _VI between the voltage vector command (V _d ^* and V _q ^* ) to be applied to the induction motor 3 and the detected and converted current vector (I _d and I _q ). It outputs to the speed estimation calculation part 7.

The initial speed estimation calculation unit 7 takes in the calculated phase difference θ _VI and calculates the rotor frequency estimation value f _r ^ based on the phase difference θ _VI .

  The switching of the control mode at the time of restart will be described. The estimation of the rotor frequency by the initial speed estimation calculation unit 7 is performed when the rotor is stopped or restarted from the free-run state.

  At the initial speed estimation, the control mode switch 22 applies a substantially constant voltage from the initial speed estimation command generator 31 to the induction motor 3. That is, the initial speed estimation is performed when another controller such as a current controller (not shown) provided in the vector controller 30 is stopped.

  When the estimation of the rotor frequency is completed by the initial speed estimation calculation unit 7, the estimated value of the rotor frequency is delivered to the vector controller 30. Inside the vector controller 30, the rotor frequency estimation value is set as an initial value of the speed estimation system. Thereafter, each control module (not shown) such as a current controller (not shown) and a speed estimation system provided in the vector controller 30 is promptly operated. Switch to output voltage command. After switching, the torque is increased by increasing the d-axis current and the q-axis current to shift to the normal control mode.

  Although the configuration of the driving device 1 has been described above, the configuration obtained by removing the voltage output device 2 and the current detector 4 from the driving device 1 constitutes a control device, and the same applies to the following Examples 2 to 7. It is.

Next, a method for calculating the rotor frequency estimated value f _r ^ from the phase difference θ _VI , which is the main point of the first embodiment, will be described.

  In the first embodiment, the rotor frequency is estimated by applying the property that the impedance of the induction motor 3 changes according to the slip frequency (difference between the primary frequency and the rotor frequency).

FIG. 2 is a diagram showing a T-type equivalent circuit of the induction motor. When the phase voltage vector V ₁ is defined as Equation (3) and the phase current vector I ₁ is defined as Equation (4) and the impedances of the primary side and the secondary side are combined, the combined impedance Z of the induction motor 3 is expressed by the equation ( 8).

Where r ₁ is the primary resistance, r ₂ is the secondary resistance, x ₁ is the primary leakage reactance, x ₂ is the secondary leakage reactance, Z ₁ is the primary side impedance, Z _2S is the secondary side impedance, and Y ₀ is the excitation. admittance, _{X m} is the excitation reactance and slip slip, to. However, slip slip is, _{f 1} the primary frequency, the rotor frequency as _{f r,} is defined by the following equation (9).

Finally, the current vector I ₁ of the induction motor 3 can be expressed by the following equation (10).
That is, as shown in Expression (10), the current vector I ₁ is uniquely determined if the equivalent circuit constant, voltage vector, primary frequency, and slip (rotor frequency) are determined.

However, the current vector I ₁ shown in the equation (10) cannot represent a transient state and represents a steady state to the last. In the case of an induction motor for a railway vehicle, the electrical time constant until the steady state is reached is generally as short as about 5 to 20 ms, and the time until the steady state is obtained does not matter so much.

  In addition, the equivalent circuit shown in FIG. 2 simply ignores the iron loss resistance, and the d-axis current slightly changes. However, in the model and voltage application conditions where the influence of this iron loss resistance cannot be ignored, It is necessary to consider appropriately.

FIG. 3 is a diagram showing the trajectory of the current vector. The voltage vector is constant (V _d ^* = 0 [V], V _q ^* = 20 [V]), and the primary frequency f ₁ ^* is constant (f ₁ ^* = 20). as [Hz]), a trajectory when changing the rotor frequency _{f r} from -10Hz to 200 Hz. In FIG. 3, the voltage vector is displayed as a vector by adding “→” to the beginning of “V”.

As a result of studies by the inventors, it has been clarified that when the AC voltage is constant, the current vector draws a circular locus according to the rotor frequency. Starting from f _r = −10 [Hz], a counterclockwise circular locus is exhibited until the rotor frequency increases and reaches f _r = 200 [Hz], and f _r = 20 [Hz], that is, the slip is 0. In this case, I _q = 0 and I _d are minimum values.

Using the maximum values I _{d_max} and I _{q_max} and the minimum values I _{d_min} and I _{q_min} of the current vectors of the d axis and the q axis in FIG. 3, the center coordinates Cx and Cy of this circular locus are expressed by the following equations (11), respectively. And can be defined as shown in equation (12).

Current correction values I _d ′ and I _q moved to the origin of the dq coordinate by subtracting Cx and Cy from I _d and I _q in FIG. 3 (the following equations (13) and (14)), respectively. 'Can be used to obtain a circular locus of the current vector.

FIG. 4 is a diagram illustrating a circular locus of current vectors of the current correction values I _d ′ and I _q ′. As is apparent from FIG. 4, the rotor frequency _fr can be uniquely defined by using the phase difference θ _VI between the voltage vector and the current vector corrected as described above. That is, in the first embodiment, the initial speed is estimated using the above principle. In FIG. 4, like the voltage vector, the current vector is also displayed as a vector by adding “→” to the beginning of “I”.

  Many common initial speed estimation methods converge the speed estimation error to zero by PI control (proportional / integral control) or the like so that the estimated rotor frequency value matches the actual rotor frequency. On the other hand, in the first embodiment, it is possible to instantaneously estimate the rotor frequency by applying a predetermined AC voltage and calculating the phase difference from the detected current vector.

FIG. 5 is a diagram illustrating two examples of circular trajectories of the current vector when the voltage and the primary frequency are changed. As shown, the voltage vectors, also changes the relationship between theta _VI and f _r by the primary frequency and a circuit constant, although the opening position of the size and the circle of circular locus changes, the nature itself draws a circular locus change Absent. That is, even when the equivalent circuit constant of the induction motor and the input voltage condition are changed, the calculation formulas (3) to (14) are built in the control software, so that sequential calculation is possible.

  In the first embodiment, the rotor frequency is uniquely estimated based on the circular locus of the current vector in the steady state calculated by the equations (3) to (14). Therefore, it is desirable that the applied voltage be applied with a constant amplitude and frequency without changing transiently to obtain a steady-state current vector.

  Further, in the estimation of the rotor frequency according to the first embodiment, the output voltage cannot be operated during the period in which the applied voltage is constant, and other control systems (such as a current controller) cannot be operated. Therefore, it is a preferred form to use when starting energization when the rotor is stopped or restarted from a free-run state.

In the first embodiment, for the purpose of simplification, the speed is estimated only from the phase difference θ _VI. However, I _d ′ and I _q ′ corrected by the origin movement are not used, and I _d shown in FIG. 3 is used. It is also possible to estimate the initial speed with a configuration for speed estimation based on the magnitude of I _q and the phase difference with respect to the voltage vector.

  In the first embodiment, since the rotor frequency can be estimated without using the speed electromotive force generated in proportion to the product of the magnetic flux and the rotor frequency, the zero speed (stop state) is set as described in Patent Document 1. There are no problems that cannot be estimated. That is, it is possible to perform estimation for all speed ranges by a single method without providing exceptional processing for the speed ranges, and there is no problem that the control system becomes complicated.

  Furthermore, as described above, when a predetermined alternating voltage is applied and the current vector is in a steady state, the initial speed can be instantaneously estimated. Therefore, as described in Patent Document 2, the primary frequency is set for the entire speed range. There is no need to sweep. That is, it is possible to realize a faster restart than in the prior art, and to realize a low torque shock for the reason described in the equations (1) and (2).

FIG. 6 is a diagram illustrating an example of a result of initial speed estimation according to the first embodiment. FIG. 6 shows the result of initial speed estimation under the condition of the rotor frequency f _r = 0 [Hz] for a high-efficiency induction motor having a 1-hour rated slip of 1.0% or less. As an AC voltage for estimating the initial speed at the time of 0.1 [sec], the primary frequency f ₁ ^* = 100 [Hz] (upper left diagram in FIG. 6), Vd = 0 [V], and Vq = 100 [V ] (Right middle diagram in FIG. 6) is applied to start estimation (“▼” shown in FIG. 6 is the estimation start time). As a result, as shown in the upper left diagram of FIG. 6, 0.2 [sec] at the time, the rotor both frequency estimate _{f r} ^ and the rotor frequency _{f r} is matched, duration 100 [ms] from the estimated start The initial speed estimation can be completed within a short time.

Focusing on the magnetic flux (lower right diagram in FIG. 6), although vibration is generated due to interference between the dq axes, the amplitude is as small as about 0.01 [wb]. In the general initial speed estimation method, the premise is that the d-axis secondary magnetic flux Φ _2d is used. However, in the method of the first embodiment, a region where the slip is sufficiently large (magnetic flux is hardly generated). (Condition)), since the initial speed can be estimated, the estimation can be shortened. In addition, since the Φ _2d term itself represented by the equation (2) is reduced, the torque shock can be significantly reduced by a synergistic effect with a shorter time. As shown in the middle left diagram of FIG. 6, τ _% indicating the ratio of torque shock to the rated torque is as small as 1 [%] or less.

Note that the magnitude and primary frequency of the voltage vector applied at the time of initial speed estimation take into consideration the voltage output accuracy of the power output device 2 and the current detection accuracy of the current detector 4 due to the influence of dead time and the like, and the phase difference θ _VI It is determined as appropriate so that no error occurs.

  In addition, regarding the equivalent circuit constant used for the initial speed estimation calculation unit 7, the inductance is set to a constant value using a value measured in advance. The resistance may be a constant value, but since it has temperature dependence, the resistance value estimated based on the fluctuation of the current control response accompanying the change in the current deviation amount and motor electric time constant in the normal control mode may be used. Good.

  In the first embodiment, the initial speed is estimated by applying the property that the impedance of the induction motor 3 changes according to the slip frequency that is the difference between the frequency of the primary voltage applied to the induction motor 3 and the rotor frequency. Is. This focuses on the property of the induction motor 3 that generates torque by causing an induction current to flow on the secondary side, and thus can be applied to any linear induction motor or the like in which a conductor is disposed on the secondary side. However, it is difficult to apply to a permanent magnet synchronous motor or the like in which an impedance change according to the slip frequency is difficult to see, and is an initial speed estimation method unique to an induction motor.

  As described above, according to the first embodiment, it is possible to instantaneously estimate the rotor frequency by detecting the phase difference between the voltage vector and the detected current vector when an AC voltage is applied to the induction motor 3. As a result, it is possible to provide a drive device for an induction motor that can be restarted at a high speed and with a low torque shock.

  FIG. 7 is a functional block diagram of the induction motor driving apparatus according to the second embodiment of the present invention. The functional block shows the configuration when the induction motor is restarted from the stop or free run state by vector control without speed sensor.

  The purpose of the second embodiment is to make the initial speed estimation more accurate than the first embodiment. Therefore, the current detected by the current detector 4 is different from the first embodiment in that the current detected by the current detector 4 is input to the phase difference calculation unit 6 through the DC detection filter 15.

Since a highly efficient induction motor has a small resistance value, vibration due to interference between dq axes is likely to occur. That is, as the resistance value decreases with higher efficiency, vibration components superimposed on the detection currents I _d and I _q shown in the upper right diagram of FIG. 6 tend to appear. When current vibration due to interference between the dq axes is superimposed on I _d and I _q on the rotation coordinates, the phase difference θ _VI defined in the first embodiment vibrates, which leads to vibration of the rotor frequency estimation value. This causes a decrease in accuracy of speed estimation.

In the second embodiment, as shown in FIG. 7, the currents I _d0 and I _q0 output from the coordinate converter 5 are passed through the DC detection filter 15 and input to the phase difference calculation unit 6 as I _d and I _q . . As a result, it is possible to obtain a DC value from which the vibration component superimposed on the current on the dq axis is removed, and even for a highly efficient induction motor, the influence of interference between the dq axes is reduced and the initial speed is increased. It is possible to estimate with accuracy.

  The DC detection filter 15 used in the second embodiment is configured by a low-pass filter, a band-pass filter, a notch filter, or the like, but the type of filter is limited as long as it can detect a substantially DC component on the dq axis. It is not a thing.

  As described above, according to the second embodiment, the detected current value of the induction motor passes through the DC detection filter 15 to remove the vibration component and is detected as the DC component. It is possible to provide a drive device for an induction motor that can be restarted with higher accuracy by a low torque shock.

  FIG. 8 is a diagram showing functional blocks of the induction motor driving apparatus according to Embodiment 3 of the present invention. A configuration when the induction motor is restarted by speed sensorless vector control from the stop or free-run state is shown by functional blocks.

  The third embodiment aims to reduce the calculation load of the microcomputer at the time of initial speed estimation than the second embodiment. Therefore, the initial speed estimation calculation unit 7 is different from the second embodiment in that the initial speed estimation calculation unit 7 includes a speed estimation map 8 that is derived using Expression (3) to Expression (14). Further, the speed estimation map 8 may be applied to the first embodiment (that is, the configuration obtained by removing the DC detection filter 15 from the configuration of the second embodiment).

In the initial speed estimation method of the third embodiment, if the voltage vector, the primary frequency, and the equivalent circuit constant are determined in advance, the relationship between the phase difference θ _VI and the rotor frequency estimated value f _r ^ can be mapped in advance. Is possible. The principle of estimation of the initial speed is the same as that of the first embodiment. However, by providing the driving apparatus 1 with the speed estimation map 8, it is possible to more easily implement it as software. Furthermore, since the sequential calculation using the above equations (3) to (14) becomes unnecessary, the calculation load of the microcomputer can be reduced.

As described above, according to the third embodiment, the relationship between the phase difference θ _VI between the voltage vector and the detected current vector and the rotor frequency f _r ^ when an AC voltage is applied to the induction motor 3 is mapped and incorporated in the control software. By doing so, it is possible to provide a drive device for an induction motor that can be restarted at a high speed and with a low torque shock and that further reduces the computational load of the microcomputer.

  FIG. 9 is a diagram illustrating functional blocks of the induction motor drive device according to the fourth embodiment of the present invention. The functional block shows the configuration when the induction motor is restarted from the stop or free run state by vector control without speed sensor.

  The purpose of the fourth embodiment is to make the initial velocity estimation more accurate than the second embodiment. Therefore, the second embodiment is different from the second embodiment in that a series of processing from voltage application to the induction motor to initial speed estimation is repeated at least twice by changing the applied voltage vector and the primary frequency command value. In addition, the processing method according to the fourth embodiment may be applied to the first or third embodiment.

FIG. 10 shows the current vector when the rotor frequency changes when the primary frequency command value f ₁ ^* of the voltage vector (V _d ^* = 0 [V], V _q ^* = 20 [V]) is 20 Hz. It is a figure which shows a circular locus. When the primary frequency command value f ₁ ^* is 20 Hz, if the difference from the actual rotor frequency greatly deviates, such as fr = 200 [Hz] or fr is a negative value, the speed with respect to the fluctuation amount of the phase difference The amount of fluctuation of the estimated value also increases, the resolution of speed estimation decreases (“Low-Resolution Region” shown in FIG. 10), and is easily affected by errors in current detection and voltage output. That is, in order to improve the robustness against these errors, the primary frequency command value f ₁ ^* is close to the rotor frequency so as to belong to at least the “High-Resolution Region” shown in FIG. It is desirable to set by value.

In the fourth embodiment, the rotor frequency estimated value f _r ^ estimated by the initial speed estimation calculating unit 7 is input to the estimated value holding unit 9, and the rotor frequency estimated value f is determined according to the timing of the trigger signal output from the trigger signal output unit 10. Hold _r ^. The held rotor frequency estimation value f _r ^ is fed back to the initial speed estimation command generator 31. The initial speed estimation command generator 31 generates a new primary frequency command value f ₁ ^* based on this feedback value. Then, a series of processes from application of AC voltage to initial speed estimation is performed again.

If this new primary frequency command value f ₁ ^* is reset in a direction closer to the actual rotor frequency than the previous value, an effect of improving the speed estimation accuracy can be obtained. The new voltage command values (V _d ^* and V _q ^* ) are changed based on the new primary frequency command value f ₁ ^* so that an excessive current does not flow. For example, the voltage and the primary frequency The V / F ratio, which is the ratio of the above, is changed to be constant.

  The timing at which the trigger signal is output from the trigger signal output unit 10 does not necessarily have to wait until the estimated speed value reaches a steady state, and may be a point in time when the approximate speed can be estimated in about 30 ms. The number of times this initial speed estimation is repeated is not limited to two, and may be any number as long as it is two or more.

  As described above, according to the fourth embodiment, the initial speed estimation is performed with higher accuracy than in the second embodiment by changing the command value of the primary frequency of the applied AC voltage waveform and repeating the speed estimation at least twice. It is possible to provide a drive device for an induction motor capable of achieving the above.

  FIG. 11 is a diagram illustrating functional blocks of the induction motor drive device according to the fifth embodiment of the present invention. The functional block shows the configuration when the induction motor is restarted from the stop or free run state by vector control without speed sensor.

The purpose of the fifth embodiment is to make the initial speed estimation more accurate than the fourth embodiment. For this purpose, the rotor frequency estimation value f _r ^ obtained in the first to fourth embodiments is input to the convergence calculation unit 13 and the rotor frequency estimation value f _r ^ is calculated to converge to the actual rotor frequency. Different from 1 to Example 4.

The approximate speed calculation unit 12 shown in FIG. 11 includes a phase difference calculation unit 6 and an initial speed estimation calculation unit 7 according to the first embodiment, or a phase difference calculation unit 6 according to the second to fourth embodiments, an initial speed estimation calculation unit 7 and a direct current. The rotor frequency f _r0 ^ that is configured by the detection filter 15 (shown in FIG. 11 shows the configuration of the second embodiment) and is estimated by each method described in the first to fourth embodiments is output to the convergence calculation unit 13.

  Based on the current detection value from the current detector 4 via the coordinate converter 5, the convergence calculation unit 13 changes the primary frequency command by PI control so that the q-axis current (slip frequency) becomes zero and induction. Any configuration may be used as long as the speed estimation error is converged to zero by feedback control including at least one integrator, such as a method of performing PI control by an observer method based on the voltage equation of the motor.

  Even if the values estimated by the respective methods shown in the first to fourth embodiments include a speed estimation error of about several Hz, the convergence calculation unit 13 finally calculates the convergence, thereby achieving higher accuracy. Calculation is possible, and robustness against the influence of disturbance can be improved. When PI control is performed from an initial value at which the rotor frequency estimation value greatly deviates from the actual rotor frequency, it takes an enormous amount of time to complete convergence. However, if the deviation is about several Hz, it can be instantaneously converged.

  As described above, according to the fifth embodiment, it is possible to provide a drive device for an induction motor capable of performing initial speed estimation with higher accuracy than that of the fourth embodiment by causing the convergence calculation unit 13 to perform convergence calculation of the speed estimation error.

FIG. 12 is a diagram illustrating functional blocks of the induction motor driving apparatus according to the sixth embodiment of the present invention. The functional block shows the configuration when the induction motor is restarted from the stop or free run state by vector control without speed sensor. In FIG. 12, the phase difference calculation unit 6, the initial speed estimation calculation unit 7, and the DC detection filter 15 according to the second embodiment are shown as the configuration for calculating the rotor frequency estimation value f _r ^. Alternatively, the configurations of the third to fifth embodiments may be used.

  The purpose of the sixth embodiment is to further reduce the initial speed estimation and reduce the torque shock. For this purpose, the speed information used in the drive system of a railway vehicle or the like is taken into the induction motor drive device 1 and is set as the initial value of the initial speed estimation command generator 31, which is different from the first to fifth embodiments. . Here, the speed information used in a drive system such as a railway vehicle is information related to the rotation speed (rotor rotation speed) of the induction motor.

When the speed information V _car used in the drive system of a railway vehicle or the like is taken into the induction motor drive device 1, the speed information V _car is converted into the rotor frequency estimated value f _{rin i} by the electrical angular frequency conversion unit 14 to _obtain the initial speed. This is input to the estimation command generator 31 and reflected in the primary frequency command value f ₁ ^* to be output. The speed information V _car does not necessarily have to be accurate with respect to the rotation speed (rotor frequency) of the induction motor, and may be an approximate value such as a relay signal of a speed stage used in a railway vehicle. By applying the primary frequency command based on the speed information input from the outside of the induction motor drive device to the configuration shown in the first to fifth embodiments, the initial speed can be estimated at a higher speed and with a lower torque shock. . The sixth embodiment can be applied to a system configuration in which speed information used in a drive system such as a railway vehicle can be taken into the drive device 1 for an induction motor.

  In the sixth embodiment, since the estimated initial value used for the initial speed estimation command generator 31 can be made closer to the actual speed, the initial speed estimation can be realized at a higher speed and with a lower torque shock than in the fifth embodiment.

  FIG. 13 is a schematic configuration diagram showing a part of a railway vehicle having an induction motor drive device according to the present invention as a seventh embodiment.

  In FIG. 13, the drive device 1 mounted on the railway vehicle receives supply of electric power from the overhead line 101 via the current collector, outputs alternating current power and supplies it to the induction motor 3, so that electric energy is converted into mechanical torque. Converted. The induction motor 3 is connected to an axle of the railway vehicle via a reduction gear, and the railway vehicle travels by a tangential force generated between the wheel 103 connected to the axle and the rail 102.

  By applying the induction motor driving device 1 according to each of the embodiments described above as the driving device 1 mounted on the railway vehicle, the initial speed estimation time can be shortened and the torque shock can be greatly reduced. Thereby, even when the railway vehicle is stopped or restarted from coasting, the vehicle body vibration of the railway vehicle can be reduced and the riding comfort of passengers can be further improved.

DESCRIPTION OF SYMBOLS 1 ... Induction motor drive device, 2 ... Voltage output device, 3 ... Induction motor, 4 ... Current detector,
5 ... Coordinate converter, 6 ... Phase difference calculation unit, 7 ... Initial speed estimation calculation unit, 8 ... Speed estimation map,
9 ... Estimated value hold unit, 10 ... Trigger signal output unit, 12 ... Estimated speed calculation unit,
13 ... Convergence calculation unit, 14 ... Electrical angular frequency calculation unit, 15 ... DC detection filter,
20 ... Coordinate converter, 21 ... Integrator, 22 ... Control mode switch, 30 ... Vector control unit,
31 ... Command generator for initial speed estimation, 101 ... Overhead wire, 102 ... Rail, 103 ... Wheel

Claims (20)

A voltage output device that outputs an alternating voltage applied to the induction motor;
A current detector for detecting a current flowing in the induction motor;
A drive device for an induction motor comprising a control device for controlling the output of the voltage output device,
From the circular locus drawn by the current vector determined from the detected current from the current detector according to the rotational frequency of the rotor of the induction motor, the control device can generate the alternating current when the rotor is stopped or started from a free-run state. An induction motor driving apparatus, wherein a rotational frequency of the rotor is estimated based on a phase difference between a voltage vector obtained from a voltage and the current vector. It is a drive device of the induction motor according to claim 1,
An induction motor drive device, wherein the control device estimates a rotation frequency of the rotor when the rotor is stopped or started from a free-run state. It is a drive device of the induction motor according to claim 1,
An induction motor drive device, wherein the control device estimates a rotational frequency of the rotor when a substantially constant voltage is applied to the induction motor. It is a drive device of the induction motor according to claim 1,
An induction motor drive device, wherein the control device estimates a rotational frequency of the rotor when a current controller that performs feedback control so that the value of the detected current becomes a predetermined current value is stopped. It is a drive device of the induction motor of any one of Claims 1-4, Comprising:
The control device includes a coordinate converter that converts the AC voltage and the detected current into the voltage vector and the current vector, respectively, based on an orthogonal biaxial rotating coordinate system, and a phase difference between the voltage vector and the current vector. An induction motor drive device comprising: an arithmetic unit that calculates and estimates a rotational frequency of the rotor based on the phase difference from the circular locus. A drive device for an induction motor according to claim 5,
The drive device for an induction motor, wherein the control device includes a filter that removes a vibration component of the current vector and outputs a substantially DC component before the current vector is input to the arithmetic unit. The induction motor driving device according to claim 5 or 6,
The drive unit for an induction motor, wherein the calculation unit estimates a rotation frequency of the rotor using a speed estimation map in which a correlation between the phase difference and the rotation frequency of the rotor is defined in advance. The induction motor driving device according to any one of claims 5 to 7,
The arithmetic unit repeatedly performs a series of processes from application of the AC voltage to the induction motor to estimation of the rotational frequency of the rotor at least twice by changing a primary frequency command of the AC voltage. An induction motor drive device characterized by the above. It is a drive device of the induction motor according to claim 8,
When changing the primary frequency command of the alternating voltage, the voltage command of the alternating voltage is also changed to make the ratio of the alternating voltage and the primary frequency of the alternating voltage constant. Drive device. The induction motor drive device according to any one of claims 5 to 9,
The control device further includes a convergence calculation unit that converges the speed estimation deviation to zero based on the detected current using at least one integrator provided for the rotational frequency of the rotor estimated by the calculation unit. An induction motor drive device characterized by that. The induction motor driving device according to any one of claims 5 to 10,
A drive device for an induction motor, wherein a primary frequency command for the AC voltage is set based on speed information related to a rotation frequency of the rotor used in a system including the drive device. A railway vehicle equipped with an induction motor for driving wheels,
The railway vehicle according to any one of claims 1 to 11, wherein the induction motor drive device is the induction motor drive device according to any one of claims 1 to 11. A method of driving an induction motor to which an alternating voltage output from a voltage output device is applied,
From the circular locus drawn by the current vector flowing from the current flowing through the induction motor in accordance with the rotation frequency of the rotor of the induction motor, the rotor of the rotor is based on the phase difference between the voltage vector obtained from the AC voltage and the current vector. A method for driving an induction motor, wherein the voltage output device is controlled by estimating a rotation frequency. A method for driving an induction motor according to claim 13,
A method of driving an induction motor, wherein the rotational frequency of the rotor is estimated when the rotor is stopped or started from a free-run state. A method for driving an induction motor according to claim 13,
A method for driving an induction motor, wherein the rotational frequency of the rotor is estimated when a substantially constant voltage is applied to the induction motor. A method for driving an induction motor according to claim 13,
A method for driving an induction motor, comprising: estimating a rotation frequency of the rotor when feedback for controlling a current value flowing through the induction motor to a predetermined current value is stopped. A method for driving an induction motor according to any one of claims 13 to 16,
Each of the voltage vector and the current vector is obtained by conversion based on a rotating coordinate system of two orthogonal axes. A method for driving an induction motor according to any one of claims 13 to 17,
A series of processing from application of the AC voltage to the induction motor to estimation of the rotational frequency of the rotor is repeatedly performed by changing a primary frequency command of the AC voltage at least twice or more. Driving method of the electric motor. A method for driving an induction motor according to claim 18,
When changing the primary frequency command of the alternating voltage, the voltage command of the alternating voltage is also changed to make the ratio of the alternating voltage and the primary frequency of the alternating voltage constant. Driving method. A method for driving an induction motor according to any one of claims 13 to 19,
A method for driving an induction motor, wherein a speed estimation deviation is converged to zero based on a current flowing through the induction motor using PI control with respect to the estimated rotational frequency of the rotor.