JP2016082790A - Electric motor control device, electric motor control system - Google Patents

Abstract

PROBLEM TO BE SOLVED: To prevent efficiency of a motor from being deteriorated, by suppressing fluctuation of output torque relative to load torque even while maintaining controllability of a rotational speed.SOLUTION: In a load angle command generation part 4A, proportional control is performed on a deviation ωerr between a rotational speed ωe^ of a motor 2 and a rotational speed command value ω0to obtain an arithmetic value φc. A differential value sφcof the arithmetic value φcis defined as a speed correction amount ωcorr. A sum of the rotational speed ωe^ and the speed correction amount ωcorr is adopted as a control axis rotational speed ω0 of a rotational coordinate system. An integral value of the control axis rotational speed ω0 is defined as a rotation angle θ0 of the rotational coordinate system with respect to a stationary coordinate system. On the basis of a command value [λ] regarding primary magnetic flux, the control axis rotational speed ω0 and the rotational angle θ0, a voltage command generation part 70 generates voltage command values v, vand vand gives them to a voltage supply source.SELECTED DRAWING: Figure 3

Description

  The present invention relates to a motor control device and a motor control system, for example, a motor control device that applies an AC voltage to a synchronous motor.

  Patent Documents 1 and 2 describe a method for controlling a synchronous motor. The synchronous motor has an armature having windings and a field.

  Patent Document 1 introduces a technique for controlling the current flowing through the armature (hereinafter referred to as “armature current”) to be constant.

  In Patent Document 2, the primary magnetic flux of the synchronous motor is controlled. More specifically, the γ-axis and δ-axis that are orthogonal to each other are adopted as control axes, and the γ-axis component of the primary magnetic flux is controlled to zero. At this time, the control frequency of the inverter driving the synchronous motor is performed by feeding back the γ-axis component of the armature current.

In Patent Document 3, a δc-γc rotating coordinate system for performing primary magnetic flux control is adopted, and a speed command value ω ^* is corrected by a correction amount based on the γc-axis current, and the δc-γc rotating coordinate system is corrected. The rotational speed (hereinafter referred to as “control shaft rotational speed”) is calculated. As the correction amount, similar to Patent Document 2, the product of the harmonic component obtained by removing the DC component from the γc-axis current and the gain is employed.

  As is well known (see, for example, Patent Document 4), the output torque depends on the sine value of the load angle and the sine value of twice the load angle. The output torque depends on the γc axis component of the armature current (hereinafter referred to as “γc axis current”).

Japanese Patent No. 2551132 Japanese Patent No. 3672761 Japanese Patent No. 5494760 JP 2002-34280 A Japanese Patent No. 3551919

  When the primary magnetic flux control is performed on the motor, the rotational speed increases transiently when the load torque decreases. This means that the phase angle (load angle) between the δc-γc rotating coordinate system and the dq rotating coordinate system increases. At this time, if a large load angle is obtained, the γc-axis current is also large.

  In the technique described in Patent Document 3, the rotational speed of the δc-γc rotating coordinate system can be reduced and the load angle can be reduced based on the decrease in the γc-axis current. Conversely, when the load torque decreases, the γc-axis current takes a large value, and the rotational speed of the δc-γc rotational coordinate system increases. Thus, the control reflecting the magnitude of the load angle prevents the motor from stepping out.

  However, in such control, control based on the value of the load angle is performed. That is, after the γc-axis current has actually obtained a large value, the step-out is prevented based on the value. In such control, the γc-axis current fluctuates, and this fluctuation leads to pulsation of the armature current, which in turn degrades the efficiency of the motor.

  Therefore, an object of the present invention is to provide an electric motor control device that suppresses fluctuations of the output torque with respect to the load torque while maintaining the controllability of the rotation speed, and does not deteriorate the efficiency of the electric motor.

The motor control device (3) according to the present invention applies an AC voltage to a synchronous motor (2) having an armature (21) and a field (23) that generates a field magnetic flux ([Λ0]). And a control device for controlling the voltage supply source (1) for supplying an alternating current (i _u , i _v , i _w ) to the armature.

The first aspect is that the rotational speed (ωe ^) at the electrical angle of the synchronous motor and the command value (ω0 ^* ) of the control shaft rotational speed (ω0), which is the rotational speed of the rotational coordinates (δc−γc). A first subtractor (41) for obtaining a deviation (ωerr); a load angle which is a command value of a control load angle (φc) which is a phase of the rotational coordinate with respect to the field magnetic flux by performing at least proportional control on the deviation A load angle command generation unit (4A, 4B) having a gain unit (42) for obtaining a command (φc ^* ); and the load angle command and rotation information (ωe ^, θe) of the field magnetic flux. A rotation information acquisition unit (5A to 5D) for obtaining a rotation speed (θ0) of the control shaft rotation speed and the rotation coordinate; an electric machine generated by the alternating current using the control shaft rotation speed and the rotation angle; Between the magnetic flux of the child reaction and the field magnetic flux. Which is the formation primary flux ([lambda]) is controlled in the rotating coordinate, the command value serving voltage command of the AC voltage ^{(Vu *, Vv *, Vw} *) comprises a voltage-command generating unit for obtaining the a and (70).

For example, the rotation information acquisition unit (5A) includes a differentiator (51) for obtaining a speed correction amount (ωcorr) as a differential value of the load angle command (φc ^* ), and the rotation of the field magnetic flux ([Λ0]). An adder (52) for adding the rotational speed (ωe ^) of the field magnetic flux as information and the speed correction amount to output the control shaft rotational speed (ω0); and the control shaft rotational speed (ω0); And an integrator (53) for integrating and outputting the rotation angle (θ0).

For example, the rotation information acquisition unit (5B) adds the load angle command (φc ^* ) and the phase (θe) of the field magnetic flux that is the rotation information of the field magnetic flux ([Λ0]). An adder (54) that outputs a rotation angle (θ0); and a differentiator (55) that differentiates the rotation angle and outputs the control shaft rotation speed (ω0).

For example, the rotation information acquisition unit (5C) outputs a deviation (φerr) between the load angle command (φc ^* ) and an estimated value (φc ^) of the control load angle (φc) (56 ) And; a proportional-integral control unit (57) that performs proportional-integral control on the output of the second subtractor to obtain a speed correction amount (ωcorr); and the rotation information of the field magnetic flux ([Λ0]). An adder (52) for adding the rotational speed (ωe ^) of the field magnetic flux and the speed correction amount to output the control shaft rotational speed (ω0); and integrating the control shaft rotational speed (ω0) And an integrator (53) for outputting the rotation angle (θ0).

For example, the rotation information acquisition unit (5D) outputs a deviation (φerr) between the load angle command (φc ^* ) and the estimated value (φc ^) of the control load angle (φc) (56 ) ;; proportional-integral control on the output of the second subtractor to obtain an angle correction amount (θcorr); 58; the load angle command (φc ^* ); and the angle correction An adder (59) for adding the amount and the phase (θe) of the field magnetic flux ([Λ0]) as the rotation information of the field magnetic flux to output the rotation angle (θ0); And a differentiator (55) that outputs the control shaft rotational speed (ω0).

  A second aspect of the motor control device according to the present invention is the first aspect, in which the load angle command generation unit (4A, 4B) is configured to operate the synchronous motor (2 And a low-pass filter (49) that removes the pulsation component caused by the periodic load torque pulsation and outputs the pulsation component to the first subtracter (41).

  A third aspect of the motor control device according to the present invention is the first aspect thereof, wherein the load angle command generation unit (4A, 4B) is configured to reduce the deviation (ωerr) of the synchronous motor (2). A low-pass filter (49) for removing pulsation components caused by periodic load torque pulsations is further provided.

A fourth aspect of the motor control device according to the present invention is the first aspect, in which the load angle command generation unit (4A, 4B) is configured to output the synchronous motor of the load angle command (φc ^* ) (φc ^* ). Further provided is a low-pass filter (49) for removing a pulsation component caused by the periodic load torque pulsation of 2).

In the first to fourth aspects of the motor control device according to the present invention, preferably, the alternating current (i _u , i _v , i _w ) is _{converted into a} rotation current (i _δc ) as a current in the rotation coordinates based on the rotation angle. , I _γc ) is further provided.

The voltage command generator (70) is based on the primary magnetic flux command ([λ ^* ]) that is the command value of the primary magnetic flux ([λ]), the control shaft rotational speed (ω0), and the rotational current. An original voltage command calculation unit (71) for _{obtaining an} original voltage command (V _δc ^* , V _γc ^* ) as a voltage command value in the rotation coordinates; and the original voltage command based on the rotation angle (θ0) A coordinate conversion unit (72) that converts the voltage command (V _u ^* , V _v ^* , V _w ^* ).

In the first aspect of the motor control device according to the present invention, preferably, a rotation speed acquisition unit (61) for obtaining the rotation speed (ωe ^) based on the alternating current (i _u , i _v , i _w ) is further provided. Prepare.

  The motor control system according to the present invention includes the motor control device (3) of the motor control device according to the present invention and the voltage supply source (1).

  According to the first aspect of the electric motor control device of the present invention, while maintaining the controllability of the rotational speed, the fluctuation of the output torque with respect to the load torque is suppressed, so that the efficiency of the electric motor does not deteriorate.

  According to the 2nd-4th aspect of the electric motor control apparatus concerning this invention, control is stabilized.

It is a vector diagram which shows an example of a rotation coordinate and magnetic flux typically. It is a block diagram which shows an example of a notional structure of the control apparatus of an electric motor, and its peripheral device. It is a block diagram which shows a conceptual example of a structure of a control part. It is a graph which shows the waveform of load torque, output torque, and armature current. It is a block diagram which shows the other conceptual example of a structure of a control part. It is a block diagram which illustrates the aspect which inserts a low-pass filter. It is a graph which shows the waveform of the load torque at the time of employ | adopting a low-pass filter, an output torque, and an armature current. It is a block diagram which illustrates the aspect which inserts a low-pass filter. It is a block diagram which illustrates the aspect which inserts a low-pass filter. It is a block diagram which shows the structure of a rotation information acquisition part. It is a block diagram which shows the other structure of a rotation information acquisition part.

  Before going into the detailed description of the embodiment, the premise of the present invention will be described.

<1. Premise>
FIG. 1 shows a synchronous motor (hereinafter, simply referred to as “motor”. As a special type of synchronous motor, there is a motor that does not have a field such as a switched reluctance motor. It is a vector diagram showing the relationship between the gap magnetic flux [λ] (symbol [] represents a vector quantity; the same applies hereinafter) and the field magnetic flux [Λ0] in the electric motor. For example, when the electric motor has a permanent magnet, the field magnetic flux [Λ0] is generated by the permanent magnet, and when the electric motor has a field winding, current flows in the field winding. Generated by flowing.

  A dq rotating coordinate system is introduced as a rotating coordinate system synchronized with the rotation of the electric motor. Here, the d-axis is set in phase with the field magnetic flux [Λ0], and the q-axis is phase-shifted with respect to the d-axis in a direction (hereinafter simply referred to as “rotation direction”) that is desired to be rotated by the motor control. Advance 90 degrees. Therefore, the rotation speed ωe of the dq rotation coordinate system can be regarded as the rotation speed of the electric motor. For convenience of later explanation, the phase of the d axis with respect to the fixed axis α is introduced as the field magnetic flux phase θe.

  Further, a δ-γ rotating coordinate system and a δc-γc rotating coordinate system are introduced as rotating coordinate systems. The phase advances at a phase angle φ in the direction of rotation of the motor with respect to the δ axis with respect to the d axis and the γ axis with respect to the q axis, respectively. The phase advances at a phase angle φc toward the rotational direction of the motor with respect to the δc axis with respect to the d axis and the γc axis with respect to the q axis, respectively.

  For example, in a motor control method known as “primary magnetic flux control”, the δ axis is set in phase with the gap magnetic flux [λ]. In this case, the phase angle φ is grasped as a load angle (phase angle between the field magnetic flux [Λ0] and the gap magnetic flux [λ]).

  Hereinafter, for convenience of explanation, the phase angle φ of the δ axis with respect to the d axis is referred to as an actual load angle φ, and the phase angle φc of the δc axis with respect to the d axis is referred to as a control load angle φc. For convenience of later explanation, the phase of the δc axis with respect to the fixed axis α is introduced as the rotation angle θ0.

As is well known, the gap magnetic flux [λ] is a voltage and current supplied to an electric motor (more specifically, an armature winding included in an armature included in the electric motor) and device constants (for example, inductance, armature) of the electric motor. It is determined by the resistance component of the winding, the field magnetic flux ([Λ0])) and the rotation speed of the motor. Therefore, in the above-described “primary magnetic flux control”, the control device that controls the electric motor performs control so that the air gap magnetic flux [λ] becomes equal to the command value [λ ^* ] of the air gap magnetic flux [λ]. The γ-axis component of the command value [λ ^* ] is zero.

  When the δc-γc rotational coordinate system is employed in such control, the rotation of the motor can be appropriately controlled because the control load angle φc matches the actual load angle φ.

<2. Configuration of power conversion device>
FIG. 2 is a block diagram showing the motor control device 3 and its peripheral devices that perform the control of the present embodiment based on the above premise.

  The electric motor 2 is a three-phase electric motor, and includes an armature 21 and a rotor as a field magnet 23. As technical common sense, the armature 21 has an armature winding 22, and the rotor rotates relative to the armature 21. The case where the field magnet 23 includes a magnet that generates the above-described field magnetic flux [Λ0] will be described. The electric motor 2 drives a compressor, for example.

The voltage supply source 1 includes, for example, a voltage control type inverter and its control unit, and a three-phase voltage command value [v _x ^* ] = [v _u ^* v _v ^* v _w ^* ] ^t (“ t ″ indicates transposition of the matrix. The same applies hereinafter), and three-phase voltages v _u , v _v , and v _w are applied to the motor 2. As a result, a three-phase alternating current (hereinafter also referred to as “three-phase current”) [i _x ] = [i _{u iv} i _w ] ^t flows in the motor 2. However, components included in the voltage command value [v _x ^* ] and the three-phase current [i _x ] are described, for example, in the order of the U-phase component, the V-phase component, and the W-phase component.

The electric motor control device 3 is a device that controls the air gap magnetic flux [λ] and the rotation speed with respect to the electric motor 2. The air gap magnetic flux [λ] is also referred to as a primary magnetic flux, and the magnetic flux of the armature reaction generated by the field magnetic flux [Λ0] and the armature current flowing through the armature 21 (this is also the three-phase current [i _x ]). Is a synthesis of Hereinafter, the gap magnetic flux [λ] is also referred to as a primary magnetic flux [λ].

  It can be understood that the electric motor control device 3 constitutes an electric motor control system together with the voltage supply source 1.

  FIG. 3 is a block diagram showing a conceptual example of the internal configuration of the motor control device 3. The electric motor control device 3 includes a load angle command generation unit 4A, a rotation information acquisition unit 5A, an original voltage command calculation unit 71, and coordinate conversion units 72 and 73.

The coordinate conversion unit 73 converts the three-phase current [i _x ] detected by the current detection unit 8 (see also FIG. 2) into a current in the _δc-γc rotating coordinate system (hereinafter also referred to as “rotating current”) [i _δγc ]. = [I _δc i _γc ] Convert to ^t . Coordinate conversion section 72, (hereinafter referred to as "original voltage command") a voltage command value in .delta.c-[gamma] c rotating coordinate system ^{_{[v δγc *] = [v}} δc * v γc *] Voltage command value ^{t _[v x} *] Convert to For these conversions, the rotation angle θ0 of the δc-γc rotation coordinate system with respect to the fixed coordinate system (for example, the UVW fixed coordinate system) of the electric motor 2 is used. Since these coordinate transformations are realized by a known technique, the details thereof are omitted here.

The original voltage command calculation unit 71 generates an original voltage command [v _δγc ^* ] based on the primary magnetic flux command value [λ ^* ] and the control shaft rotational speed ω0. The primary magnetic flux command value [λ ^* ] is a command value for the primary magnetic flux [λ], and the γc-axis component λγ ^* is set to zero in the primary magnetic flux control. Thereby, the primary magnetic flux [λ] is controlled along the δc axis.

Any known method may be adopted as a method for generating such an original voltage command [v _δγc ^* ]. Since this method is described in detail in, for example, Patent Document 3, the description thereof is omitted here.

Note that one set of the original voltage command calculation unit 71 and the coordinate conversion unit 72 generates a voltage command value [v _x ^* ] for the AC voltage of the electric motor 2 based on the primary magnetic flux command value [λ ^* ] and the rotation angle θ0. It can be grasped as a voltage command generation unit 70 to be generated.

The load angle command generation unit 4A receives the rotation speed command value ω0 ^* and the rotation speed ωe ^, and calculates the control shaft rotation speed ω0. These are all electrical angles.

The rotation speed command value ω0 ^* is a command value for the control shaft rotation speed ω0, and is given from the outside of the load angle command generation unit 4A. The rotational speed ωe ^ is an estimated value of the rotational speed ωe and is acquired by the rotational speed acquisition unit 61. For example, the rotational speed acquisition unit 61 receives the rotational current [i _δγc ] and calculates (estimates) the rotational speed ωe ^ by a known technique. The rotational speed acquisition unit 61 may be an arbitrary rotational speed sensor. In this case, the rotational speed ωe ^ is used as a detection value.

The load angle command generation unit 4A first calculates a load angle command φc ^* based on the rotation speed command value ω0 ^* and the rotation speed ωe ^. The load angle command φc ^* is calculated by performing proportional control on the deviation ωerr between the rotational speed command value ω0 ^* and the rotational speed ωe ^, for example, as shown in the following equation (1). Kp is a proportional gain, and is stored in advance in the load angle command generation unit 4A, for example.

φc ^* = Kp · ωerr = Kp · (ω0 ^* −ωe ^) (1).

The load angle command φc ^* can be adopted as a command value for the control load angle φc. This is because if the rotational speed command value ω0 ^* is different from the rotational speed ωe ^, the deviation of the control load angle φc varies with time.

  Referring also to FIG. 1, the control shaft rotational speed ω0 is the sum of the rotational speed ωe and the differential value of the control load angle φc, as shown by the following equation (2). However, a differential operator s indicating time differentiation was introduced.

  ω0 = ωe + sφc (2).

  Therefore, the control shaft rotational speed ω0 is calculated by the following equation (3) with reference to equations (1) and (2).

ω0 = ωe ^ + s {Kp · (ω0 ^* −ωe ^)} (3).

  For example, the load angle command generation unit 4A includes a subtractor 41 and a gain unit 42 in order to perform a calculation based on Expression (3).

The subtractor 41 subtracts the rotational speed ωe ^ from the rotational speed command value ω0 ^* to obtain a deviation ωerr. The gain unit 42 multiplies the deviation ωerr by the gain Kp, and obtains a load angle command φc ^* (= Kp · (ω0 ^* −ωe ^)) as the calculation result.

The rotation information acquisition unit 5A receives the load angle command φc ^* and outputs information about the rotation of the δc-γc rotation coordinate system, here, the control shaft rotation speed ω0 that is the rotation speed and the rotation angle θ0. . The rotation angle θ0 is obtained by integrating the control shaft rotation speed ω0. Therefore, in the illustration of FIG. 3, the rotation information acquisition unit 5A has an integrator 53 that integrates the control shaft rotation speed ω0.

The rotation information acquisition unit 5 </ b> A further includes a differentiator 51 and an adder 52. Differentiator 51 differentiates the load angle command .phi.c ^*, differential operator because the differential value Esufaishi ^* serving speed correction amount ^{ωcorr (= s (ω0 * -ωe} ^)) to obtain the (gain Kp is not dependent on time s and the calculation order can be interchanged). The adder 52 adds the speed correction amount ωcorr and the rotational speed ωe ^, and outputs this as the control shaft rotational speed ω0.

A gain portion 42 and the differentiator 51 can be grasped at least arithmetic unit for determining the speed correction amount ωcorr as by performing proportional control obtained calculation value .phi.c ^* differential value Esufaishi ^*, as against the deviation Omegaerr.

The load angle command generation unit 4A may perform PI control (proportional integration control) when calculating the calculated value φc ^* . Specifically, the calculated value φc ^* is calculated by the following equation (4).

φc ^* = {Kp + Ki (1 / s)} · (ω0 ^* −ωe ^) (4).

  Such a configuration can be realized by providing a PI control unit instead of the gain unit 42 of FIG.

Using the control shaft rotational speed ω 0 obtained as described above, the original voltage command calculation unit 71 obtains the original voltage command [v _δγc ^* ]. In addition, using the rotation angle θ0 obtained as described above, the coordinate conversion unit 72 converts the original voltage command [v _δγc ^* ] into a voltage command value [v _x ^* ]. Based on the voltage command value [v _x ^* ], the voltage supply source 1 applies the three-phase voltages v _u , v _v , v _w to the electric motor 2, and the electric motor 2 rotates at the rotational speed ωe ^. In this way, the rotational speed ωe ^ is controlled so as to approach the rotational speed command value ω0 ^* .

  The deviation ωerr indicates the amount of change in the control load angle φc between control timings. Therefore, correcting the control shaft rotational speed ω0 with the speed correction amount ωcorr based on the differential value of the deviation ωerr corrects the fluctuation of the rotational speed ωe ^ resulting from the fluctuation of the load torque with the differential value of the control load angle φc. It will be. In other words, the speed correction amount ωcorr can be grasped as a command value of the differential value sφc of the control load angle φc.

In the conventional technique, as described above, control that reflects the value of the control load angle φc itself (more specifically, the control load angle φc that depends on the value of the γc-axis current i _γc depending on the value of the control load angle φc) is performed. Is going. On the other hand, in the technique of the present embodiment, the control shaft rotational speed ω0 is corrected by the load angle command φc ^* which is the r command value of the control load angle φc. Therefore, the fluctuation of the armature current is suppressed as compared with the control based on the fluctuation of the armature current as in the prior art.

  FIG. 4 shows the variation of the output torque and the armature current in the mechanical angle of the electric motor 2 for each of the case where the conventional technique is adopted and the case of the present embodiment when the load torque exhibits the same periodic fluctuation. It is a graph shown by one rotation (for two electrical angles). In FIG. 4, the broken lines indicate the waveforms of the output torque and the armature current when the conventional technique is employed, and the solid lines indicate the waveforms of the respective embodiments when the present embodiment is employed.

  As is clear from FIG. 4, it can be seen that the variation of the armature current is smaller when the present embodiment is adopted.

  In this way, a technique is provided in which fluctuation of the output torque with respect to the load torque is suppressed while maintaining the controllability of the rotation speed, and thus the efficiency of the electric motor is not deteriorated.

  FIG. 5 is a block diagram illustrating another conceptual example of the configuration of the control unit. The configuration shown in FIG. 3 is different from the configuration shown in FIG. 3 in that the load angle command generation unit 4A is a load angle command generation unit 4B, the rotation information acquisition unit 5A is a rotation information acquisition unit 5B, and the rotation speed acquisition unit 61 is a field magnet. The magnetic flux phase acquisition unit 62 is provided with a replaced configuration.

Similar to the load angle command generation unit 4A, the load angle command generation unit 4B includes a subtractor 41 for obtaining the deviation ωerr, a gain unit 42 for obtaining the load angle command φc ^* from the deviation ωerr, and a differentiator 45. ing. The differentiator 45 receives the field magnetic flux phase θe from the field magnetic flux phase acquisition unit 62 and differentiates it to obtain the rotational speed ωe ^.

The field magnetic flux phase acquisition unit 62 receives the rotational current [i _δγc ] and calculates the field magnetic flux phase θe by a known technique (see, for example, Patent Document 5). The field magnetic flux phase acquisition unit 62 may be an arbitrary rotational position sensor. In this case, the field magnetic flux phase θe is used as a detection value.

The rotation information acquisition unit 5B includes an adder 54 and a differentiator 55. The adder 54 adds the load angle command φc ^* and the field magnetic flux phase θe, and outputs the rotation angle θ0. This is a technically reasonable addition because Equation (5) is established in FIG. 1 and the load angle command φc ^* is treated as a command value for the control load angle φc.

  θ0 = φc + θe (5).

  The control shaft rotation speed ω0 necessary for the calculation of the original voltage command calculation unit 71 is generated by the differentiator 55 differentiating the rotation angle θ0.

  Since the differentiation and integration described in this case are both related to time, even when the rotation information acquisition unit 5B is used, the control shaft rotation speed ω0 and the rotation are the same as when the rotation information acquisition unit 5A is used. Obviously, the angle θ0 can be determined.

The rotation information acquisition units 5A and 5B generate rotation information of the δc-γc rotation coordinate system, that is, the control shaft rotation speed ω0 and the rotation angle θ0. At this time, the rotation information acquisition unit 5A inputs the rotation speed ωe ^ from the rotation speed acquisition unit 61, the rotation information acquisition unit 5B inputs the field magnetic flux phase θe from the field magnetic flux phase acquisition unit 62, and acquires the rotation information. The parts 5A and 5B also input a load angle command φc ^* . Therefore, the rotation information acquisition units 5A and 5B use the rotation information of the field magnetic flux ([Λ0]) and the load angle command φc ^* to generate the rotation information of the δc-γc rotation coordinate system. . Thus, controlling the control load angle φc directly by the load angle command φc ^* contributes to at least the potential effect of the present case, or at least potentially contributes.

  In the load angle command generation units 4A and 4B, it is also preferable to calculate the speed correction amount ωcorr by removing the pulsation component caused by the periodic load torque pulsation of the electric motor 2. By removing the pulsation component caused by the periodic load torque pulsation, the pulsation component of the armature current is removed, and as a result, the efficiency of the motor is improved.

  6, 8, and 9 are block diagrams showing various modes for inserting a low-pass filter (LPF) 49, taking the configuration of FIG. 3 as an example.

  In FIG. 6, the low-pass filter 49 removes the pulsation component due to the periodic load torque pulsation of the electric motor 2 at the rotational speed ωe ^, and outputs it to the subtractor 41.

  FIG. 7 shows the variation of the output torque and the armature current for each of the case of adopting the conventional technique and the case of the present embodiment when the load torque exhibits the same periodic variation, as in FIG. It is a graph shown by one rotation for the mechanical angle of the electric motor 2 (for two electrical angles). Similar to FIG. 4, the broken line indicates the waveform of the output torque and the armature current when the conventional technique is employed, and the solid line indicates the waveform of the output torque and the armature current when the present embodiment is employed. However, FIG. 7 is a graph when the low-pass filter 49 is employed, and the graph is drawn on the same scale as that shown in FIG.

  As is apparent from the comparison between FIG. 7 and FIG. 4, it can be seen that the use of the low-pass filter 49 reduces fluctuations in both the output torque and the armature current.

In FIG. 8, the low-pass filter 49 removes the pulsation component due to the cyclic load torque pulsation of the electric motor 2 with the deviation ωerr. In FIG. 9, the low-pass filter 49 removes the pulsation component caused by the periodic load torque pulsation of the electric motor 2 of the load angle command φc ^* and outputs it to the differentiator 43. It is obvious that the low-pass filter 49 can be inserted in the configuration of FIG.

The rotation information acquisition units 5A and 5B can also generate rotation information of the δc-γc rotation coordinate system by feeding back the estimated value φc ^ of the control load angle φc. Specifically, PI control is performed based on the deviation between the estimated value φc ^ and the load angle command φc ^* .

The estimated value φc ^ can be obtained from the original voltage command [v _δγc ^* ] and the rotation current [i _δγc ] by a known technique (see, for example, Patent Document 3).

  FIG. 10 is a block diagram showing a configuration of a rotation information acquisition unit 5C that can be used in place of the rotation information acquisition unit 5A (see FIG. 3). The rotation information acquisition unit 5C has a configuration in which the differentiator 51 of the rotation information acquisition unit 5A is replaced with a subtractor 56 and a proportional integration control unit 57.

The subtractor 56 outputs a deviation φerr between the estimated value φc ^ and the load angle command φc ^* . The proportional-integral control unit 57 performs proportional-integral control on the deviation φerr to obtain the speed correction amount ωcorr.

  The functions of the adder 52 and the integrator 53 are the same as those of the rotation information acquisition unit 5A.

  FIG. 11 is a block diagram showing a configuration of a rotation information acquisition unit 5D that can be used in place of the rotation information acquisition unit 5B (see FIG. 5). The rotation information acquisition unit 5D has a configuration in which the adder 54 of the rotation information acquisition unit 5B is replaced with a subtractor 56, a proportional integration control unit 58, and an adder 59.

The subtractor 56 outputs the deviation φerr in the same manner as the rotation information acquisition unit 5C. The proportional integral control unit 58 performs proportional integral control on the deviation φerr to obtain an angle correction amount θcorr. The adder 59 adds the field magnetic flux phase θe, the load angle command φc ^*, and the angle correction amount θcorr to output the rotation angle θ0. The function of the differentiator 55 is the same as that of the rotation information acquisition unit 5C.

Since the rotation information acquisition units 5C and 5D obtain the rotation information of the field magnetic flux ([Λ0]) based on the deviation φerr between the estimated value φc ^ and the load angle command φc ^* , the followability of the rotation angle θ0 is improved. Can do.

  As pulsation of the load torque, there are those that exhibit one cycle during one rotation and those that exhibit two cycles. As an example of the former, a case where a one-cylinder compressor or a scroll compressor is employed as the load of the electric motor 2 can be cited. An example of the latter is a case where a two-cylinder compressor is employed as the load of the electric motor 2.

  The various aspects described above can be appropriately combined as long as the functions of each other are not impaired.

  The above block diagram is schematic, and each unit may be configured by hardware, or may be configured by a microcomputer (including a storage device) whose function is realized by software. Various procedures executed by each unit or various means or various functions implemented may be realized by hardware.

  The microcomputer executes each processing step (in other words, a procedure) described in the program. The storage device is composed of one or more of various storage devices such as a ROM (Read Only Memory), a RAM (Random Access Memory), a rewritable nonvolatile memory (EPROM (Erasable Programmable ROM), etc.), and a hard disk device, for example. Is possible. The storage device stores various information, data, and the like, stores a program executed by the microcomputer, and provides a work area for executing the program. It can be understood that the microcomputer functions as various means corresponding to each processing step described in the program, or can realize that various functions corresponding to each processing step are realized.

DESCRIPTION OF SYMBOLS 1 Voltage supply source 2 Synchronous motor 21 Armature 22 Armature winding 23 Field 3 Motor controller 4A, 4B Load angle command generation part 41,56 Subtractor 42 Gain part 43,47,51,55 Differentiator 44,52 , 54, 59 Adder 49 Low-pass filter 5A, 5B, 5C, 5D Rotation information acquisition unit 53 Integrator 57, 58 Proportional integration control unit 70 Voltage command generation unit 71 Original voltage command calculation unit 72, 73 Coordinate conversion unit φc Control load corner φc ^* load angle command _{i u,} i v, i _w alternating current _{^{V u *, V v *,}} V w * voltage instruction _V δc ^*, V γc ^* original voltage command θe field magnetic flux phase θ0 rotation angle θcorr angle correction Amount ωcorr speed correction amount φerr, ωerr deviation ωe ^ (motor) rotational speed ω0 ^* command value ω0 control shaft rotational speed [Λ0] field magnetic flux

Claims (12)

An AC voltage is applied to a synchronous motor (2) having an armature (21) and a field (23) that generates a field magnetic flux ([Λ0]), and an AC current (i _u , i _v , i _w ) a control device for controlling the voltage supply source (1) flowing through the armature,
First, a deviation (ωerr) between the rotational speed (ωe ^) at the electrical angle of the synchronous motor and the command value (ω0 ^* ) of the control shaft rotational speed (ω0), which is the rotational speed of the rotational coordinate (δc−γc), is obtained. A subtractor (41);
A gain unit (42) that performs at least proportional control on the deviation and obtains a load angle command (φc ^* ) that is a command value of a control load angle (φc) that is a phase of the rotation coordinate with respect to the field magnetic flux;
A load angle command generator (4A, 4B) having
Using the load angle command and the rotation information (ωe ^, θe) of the field magnetic flux, rotation information acquisition units (5A to 5D) for obtaining the control shaft rotation speed and the rotation angle (θ0) of the rotation coordinates. When;
Using the control shaft rotation speed and the rotation angle, a primary magnetic flux ([λ]), which is a combination of the magnetic flux of the armature reaction generated by the alternating current and the field magnetic flux, is controlled in the rotational coordinates, and A motor control device (3), comprising: a voltage command generation unit (70) for obtaining a voltage command (V _u ^* , V _v ^* , V _w ^* ) as a command value of an AC voltage. The rotation information acquisition unit (5A)
A differentiator (51) for obtaining a speed correction amount (ωcorr) as a differential value of the load angle command (φc ^* );
An adder (52) that adds the rotational speed (ωe ^) of the field magnetic flux as the rotational information of the field magnetic flux ([Λ0]) and the speed correction amount to output the control shaft rotational speed (ω0). )When;
The motor control device (3) according to claim 1, further comprising an integrator (53) that integrates the control shaft rotation speed (ω0) and outputs the rotation angle (θ0). The rotation information acquisition unit (5B)
An adder that outputs the rotation angle (θ0) by adding the load angle command (φc ^* ) and the phase (θe) of the field magnetic flux as the rotation information of the field magnetic flux ([Λ0]). 54) and;
The motor control device (3) according to claim 1, further comprising a differentiator (55) for differentiating the rotation angle and outputting the control shaft rotation speed (ω0). The rotation information acquisition unit (5C)
A second subtractor (56) for outputting a deviation (φerr) between the load angle command (φc ^* ) and an estimated value (φc ^) of the control load angle (φc);
A proportional-integral control unit (57) that performs proportional-integral control on the output of the second subtractor to obtain a speed correction amount (ωcorr);
An adder (52) that adds the rotational speed (ωe ^) of the field magnetic flux as the rotational information of the field magnetic flux ([Λ0]) and the speed correction amount to output the control shaft rotational speed (ω0). )When;
The motor control device (3) according to claim 1, further comprising an integrator (53) that integrates the control shaft rotation speed (ω0) and outputs the rotation angle (θ0). The rotation information acquisition unit (5D)
A second subtractor (56) for outputting a deviation (φerr) between the load angle command (φc ^* ) and an estimated value (φc ^) of the control load angle (φc);
A proportional-integral control unit (58) that performs proportional-integral control on the output of the second subtractor to obtain an angle correction amount (θcorr);
The rotation angle (θ0) is obtained by adding the load angle command (φc ^* ), the angle correction amount, and the phase (θe) of the field magnetic flux ([Λ0]) as the rotation information of the field magnetic flux. An adder (59) for outputting
The motor control device (3) according to claim 1, further comprising a differentiator (55) for differentiating the rotation angle and outputting the control shaft rotation speed (ω0). The load angle command generator (4A, 4B)
A low-pass filter (49) for removing the pulsation component of the rotational speed (ωe ^) due to the pulsation of the periodic load torque of the synchronous motor (2) and outputting it to the first subtractor (41)
The electric motor control device (3) according to any one of claims 1 to 5, further comprising: The load angle command generator (4A, 4B)
A low-pass filter (49) for removing a pulsation component caused by a periodic load torque pulsation of the synchronous motor (2) of the deviation (ωerr)
The electric motor control device (3) according to any one of claims 1 to 5, further comprising: The load angle command generator (4A, 4B)
A low-pass filter (49) for removing a pulsation component caused by a pulsation of a periodic load torque of the synchronous motor (2) of the load angle command (φc ^* )
The electric motor control device (3) according to any one of claims 1 to 5, further comprising: A coordinate conversion unit (73) that converts the alternating current (i _u , i _v , i _w ) into a rotation current (i _δc , i _γc ) as a current in the rotation coordinates based on the rotation angle.
Further comprising
The voltage command generator (70)
Based on the primary magnetic flux command ([λ ^* ]), which is the command value of the primary magnetic flux ([λ]), the control shaft rotational speed (ω0), and the rotational current, the command value of the voltage at the rotational coordinate is obtained. An original voltage command calculation unit (71) for _{obtaining an} original voltage command (V _δc ^* , V _γc ^* );
The coordinate converter (72) which converts the original voltage command into the voltage commands (V _u ^* , V _v ^* , V _w ^* ) based on the rotation angle (θ0). The motor control device (3) according to any one of the above. A rotational speed acquisition unit (61) for determining the rotational speed (ωe ^) based on the alternating current (i _u , i _v , i _w )
The motor control device (3) according to claim 2 or 4, further comprising: Field magnetic flux phase acquisition unit (62) for obtaining the phase (θe) of the field magnetic flux based on the alternating current (i _u , i _v , i _w ).
The motor control device (3) according to claim 3 or 5, further comprising: The motor control device (3) according to any one of claims 1 to 11,
An electric motor control system comprising the voltage supply source (1).

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