JP2011250286A - Operational transconductance amplifier and filter circuit using the same - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide an OTA which exhibits superior linear performance even in an expanded input voltage range and excels in the accuracy of transconductance values and a filter circuit using the OTA.SOLUTION: The OTA is comprised of an I-V converter and current control circuits 1 and 2 which amplify the output current of the I-V converter at an amplification rate proportional to the resistance value of an internal resistive element. The current control circuits 1 and 2 are composed of a drain to which an input current is fed, a MOS transistor 10 having a gate to which a first control voltage is supplied, a MOS transistor 13 having a drain to which an output current is output, a MOS transistor 11 having a gate to which a second control voltage is supplied, a non-inverted input terminal connected to the drain of the MOS transistor 10, an output terminal connected to the gate of the MOS transistor 13, and a differential amplifier 12 having inverted input terminals connected to the source of the MOS transistor 13 and the drain of the MOS transistor 11.

Description

  The present invention can operate at high speed by combining an operational transconductance amplifier (Operational Transconductance Amplifier: generally referred to as OTA, and hereinafter referred to as OTA), an OTA and a capacitive element (C), and an input voltage. The present invention relates to a filter circuit (hereinafter referred to as an OTA-C filter) using an operational transconductance amplifier having a wide range and excellent linear performance.

  Filter circuits (hereinafter also simply referred to as filters) have been proposed in a wide variety of configurations, and each is used properly depending on the purpose and specifications. For example, an active filter includes an operational amplifier, a resistor, and a capacitor. However, a desired frequency characteristic cannot be obtained because it is affected by variations in characteristics of resistors and capacitors formed on the chip and temperature fluctuations. Digital filters and SCFs (Switched-Capacitor Filters) are not suitable for high-speed operation because they perform sampling operations. An OTA-C filter (also referred to as a Gm-C filter) is suitable for high-speed operation, but has a problem that it is not suitable for large signal processing. This OTA-C filter is composed of an OTA and a capacity, and many types of OTA have been proposed.

  FIG. 10 is a diagram for explaining a typical OTA. The illustrated OTA includes a pair of input MOS transistors including MOS transistors 112 and 113, a pair of load MOS transistors including MOS transistors 110 and 111, and a MOS transistor 114 having a current source function. By supplying the input voltage Vin to the gate terminal pair 115 and 116 of the input transistor, the output current Iout is output from the output terminal pair 117 and 118. The relationship between the input voltage Vin and the output current Iout can be described as in Expression (1) (Non-Patent Document 1).

Iout = (K · Io) ^0.5 Vin {1- (K · Vin ² ) / (4 · Io)} ^0.5 (1)
Here, K in the equation (1) is a parameter representing the performance of the MOS transistor as shown in the equation (2), and Io is a current flowing through the MOS transistor.
K = μ · Cox · (W / L) (2)
Μ, Cox, W, and L shown in the equation (2) are the mobility of the MOS transistor, the gate capacitance per unit area, the channel width, and the channel length, respectively. Further, the current Io can be expressed as in equation (3) using K and the overdrive voltage Vov.
Io = K · Vov ² (3)
Here, the overdrive voltage Vov indicates a value obtained by subtracting the threshold voltage Vth from the gate-source voltage Vgs of the MOS transistor as shown in the equation (4).
Vov = Vgs−Vth (4)

Formula (1) described above can be expressed as Formula (5) using Formula (3) and Formula (4).
Iout = (K · Io) ^0.5 Vin {1-Vin ² / (4 · Vov ² )} ^0.5 (5)
Here, when the input voltage Vin is sufficiently smaller than the overdrive voltage Vov, Expression (5) can be expressed as Expression (6).
Iout = gm · Vin Equation (6)
In equation (6), gm is a proportionality constant as represented by equation (7), is a transconductance value of the OTA circuit of FIG. 10, and is also a transconductance value of the input MOS transistors 112 and 113.
gm = (K · Io) ^0.5 (7)

  That is, the output current Iout is proportional to the input voltage Vin, and an OTA-C filter can be realized by combining the OTA and the capacitive element (C) shown in FIG. However, when the input signal Vin is large, for example, if Vin becomes so large that it cannot be ignored with respect to the value of Vov, the linear relationship between the output current Iout and the input voltage Vin does not hold, as can be seen from equation (5). In such a case, there is a problem that a desired frequency characteristic cannot be obtained with the OTA-C filter. Usually, since the value of Vov is about 0.3 to 0.7 V, the input voltage range in which linearity can be maintained becomes very small.

FIG. 11 shows an OTA that can maintain linear performance over a wide input voltage. The OTA shown in the figure is a circuit called Caprio Quad and is described in, for example, Patent Document 1 and Non-Patent Document 2. The circuit shown in FIG. 11 is a signal for controlling the common mode level to a pair of input MOS transistors composed of MOS transistors 132 and 133, a pair of MOS transistors 134 and 135, a pair of current sources 136 and 137, and a gate terminal 142 thereof. Is provided with a pair of MOS transistors 130 and 131 and a resistance element 143.
Input terminals 138 and 139 for inputting signals to the gates of the MOS transistors 130 and 131, output terminals 140 and 141, connection nodes 144 and 145 for connecting the gates and drains of the MOS transistors 134 and 135, and both sides of the resistance element 143 The terminals 146 and 147 are shown in the figure.

Next, the behavior of the output current Iout output from the output terminals 140 and 141 with respect to the input voltage Vin of the circuit shown in FIG. 11 will be described.
Using the reference numerals shown in FIG. 11, voltages V146 and V147 of both terminals 146 and 147 of the resistor 143 can be expressed as follows.
V147 = V138−Vgs132−Vgs135 (8)
V146 = V139−Vgs133−Vgs134 (9)

V138 and V139 in the equations (8) and (9) indicate terminal voltages of the input terminals 138 and 139, respectively. Vgs132, Vgs135, Vgs133, and Vgs134 are the gate-source voltages of the MOS transistors 132, 135, 133, and 134, respectively. It is assumed that the sizes of the MOS transistors 132 to 135 have the same value. That is, when the same current flows through the MOS transistors 132 to 135, the gate-source voltages of the MOS transistors 132 to 135 are the same. Since the same current always flows through the MOS transistors 132 and 134,
Vgs132 = Vgs134 (10)
Holds.

Similarly, since the same current always flows through the MOS transistors 133 and 135,
Vgs133 = Vgs135 (11)
Holds. When the voltage applied to both ends of the resistance element 143 is calculated using the input voltage Vin = V138−V139 and Expressions (8) to (11), the voltage is always equal to the input voltage Vin as shown in Expression (12). .
V147−V146 = Vin Equation (12)
Therefore, the current I143 flowing through the resistance element 143 is as shown in Expression (13).
I143 = Vin / R (13)

R in the equation (13) is the resistance value of the resistor 143. Since the current flowing through the MOS transistors 130 and 131 is equal to the current flowing through the current sources 136 and 137, the output current Iout is a value obtained by dividing the input voltage Vin by the resistance value R as shown in Expression (14).
Iout = I140−I141 = I143 = Vin / R (14)
Thus, since the output current Iout is exactly proportional to the value of the input voltage Vin, even when the input voltage is wide, the output current Iout has a remarkably excellent linear performance. Since the OTA in FIG. 11 converts the input voltage Vin into the output current Iout, it can also be called a V-I converter. The name of the OTA or VI converter is generally used according to its function, but the basic function is the same.

  FIG. 12 is a circuit diagram for explaining a conventional OTA-C filter. In FIG. 12, reference numerals 41 to 44 denote OTAs whose Gm values are Gm1 to Gm4, respectively. 45 and 46 are capacitive elements having capacitance values C1 and C2, 49 and 50 are input terminal pairs to which input signals are inputted as differentials, and 51 and 52 are output as low-pass filter (LPF) output signals as differentials. The output terminal pairs 53 and 54 are output terminal pairs from which bandpass filter (BPF) output signals are output.

The OTA-C filter shown in FIG. 12 is generally known as a biquadratic OTA-C filter, and has a transfer function H representing the filter frequency characteristic at the terminal pair 51 and 52 from which a low-pass filter output signal is output. (S) can be represented by Formula (40).
H (s) = (Gm1 · Gm2) / (C1 · C2) /
[s ² + s (Gm 4 / C 1) + [Gm ² · Gm 3 / (C 1 · C 2)]] Formula (15)
In this way, a filter having an arbitrary characteristic can be realized by Gm1 to Gm4 and the capacitors C1 and C2.

JP, 2008-92266, A paragraphs 0017-0018

IEEE Transactions on Circuits and Systems, R.R.Torrance, 32, No. 11, page 1097. 1985. Paper title "High-Frequency CMOS Continuous-Time Filters" R. Caprio, IEE Electron. Lett, Vol. 9, No. 6, pp. 147-148, 1973. Paper title "Precision differential Voltage-Current Convertor"

As already described, the OTA shown in FIG. 10 has a problem in terms of linear performance. On the other hand, the OTA shown in FIG. 11 has excellent linear performance, but the transconductance value of the OTA in FIG. 11 depends on the resistance value R of the resistance element 143. As the resistance element 143, either a resistance element formed on the IC chip or an external resistance element can be selected.
In general, a resistance element formed on an IC chip is a resistance element (hereinafter also referred to as an internal resistance element) formed on an IC chip using a poly resistor or a diffused resistor. Since the value has manufacturing variation and temperature dependence, there is a fluctuation of about 20 to 40%. For this reason, an accurate transconductance value cannot be realized using the OTA as shown in FIG.

  In addition, when an external resistance element is used in order to increase the accuracy of the transconductance value, the problem of variation in resistance value can be avoided, but the number of external resistance elements required for the number of OTAs provided on the chip is required. Therefore, there are problems that the number of IC pins is increased and the mounting cost is increased. Furthermore, there is at least about 3 to 5 pF of parasitic capacitance such as pads or the like generated by bonding to attach the external resistance element to the terminal of the circuit of the OTA body, or high speed performance is exhibited. There is a problem that can not be.

Further, in the case of an OTA-C filter using OTA as shown in FIG. 12, any kind of OTA including the generally known OTA shown in FIGS. 10 and 11 may be used. It can be used for OTA of C filter. However, when the OTA of FIG. 10 is used, if an internal resistance element provided on an IC chip with poor linear performance is used as the resistance element, the frequency characteristic varies by the variation of the resistance value of the resistance element.
In addition, when the OTA of FIG. 11 is used, the linear performance is good, and when an external resistance element is used in the OTA of FIG. 11, the frequency characteristic accuracy is good, but external resistors corresponding to the number of OTAs are required. . For this reason, there are problems that the number of pins is increased and the mounting cost is increased. Further, since the parasitic capacitance of the bonding pad and the mounting substrate is at least about 3 to 5 pF, there is a problem that a high-speed filter cannot be realized.

  In addition, in the case of an OTA-C filter using OTA as shown in FIG. 12, any kind of OTA including the generally known OTA shown in FIGS. 10 and 11 can be used as the OTA of the OTA-C filter shown in FIG. Can be used. For example, when the OTA of FIG. 10 is used, there is a problem that the linear performance is poor, and when the OTA of FIG. 11 is used, the linear performance is good, but a built-in resistor is used as the resistance of the OTA of FIG. Then, the frequency characteristics change by the amount of change in the resistance value. Further, when using an external resistor, the frequency characteristic accuracy is good. However, since external resistors are required by the number of OTAs, there are problems that the number of pins is increased and the mounting cost is increased. Furthermore, since the parasitic capacitance of the bonding pad and the mounting substrate is at least about 3 to 5 pF, there is a problem that a high-speed filter cannot be realized.

  The present invention has been made in view of the above points, and has higher transconductance value accuracy than conventional OTA circuits, and has excellent linear performance even when the input voltage range is widened.・ The purpose is to provide an amplifier. A filter circuit using an operational transconductance amplifier that can operate at high speed using this operational transconductance amplifier, has a wide input voltage range, excellent linear performance, and excellent frequency characteristic accuracy. The purpose is to provide.

  In order to solve the above-described problems of the present invention, an operational transconductance amplifier according to the present invention includes an input terminal (for example, the input terminals 138 and 139 illustrated in FIG. 1) and an output terminal (for example, the terminals illustrated in FIG. 1). 140, 141) and a first internal resistance element (for example, the resistance element 143 shown in FIG. 1) formed on at least one IC chip, and the amplification factor of the resistance value of the first internal resistance element A voltage-current converter (for example, the IV converter 8 shown in FIG. 1) that is the reciprocal number and an output current output from the output terminal of the voltage-current converter are input, and the first internal resistance element A current amplifier (for example, the current control circuits 1 and 2 shown in FIG. 1) that amplifies at a gain proportional to the resistance value, the current amplifier including a drain terminal to which an input current is input, a first control voltage Is A first MOS transistor having a gate terminal (for example, the MOS transistor 10 shown in FIG. 2), a second MOS transistor having a drain terminal for outputting an output current (for example, the MOS transistor 13 shown in FIG. 2), A third MOS transistor having a gate terminal to which a second control voltage is supplied (for example, the MOS transistor 11 shown in FIG. 2), a non-inverting input terminal connected to the drain terminal of the first MOS transistor, and the second MOS transistor A first differential amplifier (for example, the differential shown in FIG. 2) having an output terminal connected to the gate terminal, and an inverting input terminal connected to the source terminal of the second MOS transistor and the drain terminal of the third MOS transistor. An amplifier 12), the first control voltage and the first Control voltage, characterized in that it is a voltage value realizing the amplification factor of the current amplifier.

The operational transconductance amplifier according to the present invention is the above-described invention, wherein the first MOS transistor and the third MOS transistor have a drain-source voltage sufficiently smaller than a gate-source voltage, It is preferable to operate in the linear region.
Further, the operational transconductance amplifier according to the present invention is the first control voltage generation circuit (for example, the voltage source 16 shown in FIG. 2 or the circuit shown in FIG. 3) for generating the first control voltage in the above-described invention. 28) and a second control voltage generation circuit (for example, the voltage source 17 shown in FIG. 2 and the circuit 38 shown in FIG. 3) for generating the second control voltage, and the first control voltage generation circuit Comprises a second internal resistance element (for example, the resistance element 23 shown in FIG. 3) manufactured by the same process as the first internal resistance element included in the voltage-current converter. One end is connected to a first voltage source (eg, Vss shown in FIG. 3), and the other end is a non-inverting input terminal (eg, shown in FIG. 3) of a second differential amplifier (eg, differential amplifier 22 shown in FIG. 3). Connected to the non-inverting input terminal 26) At the same time, it is connected to the drain terminal of a fourth MOS transistor (for example, the MOS transistor 21 shown in FIG. 3), and the inverting input terminal of the second differential amplifier is connected to the third reference voltage (for example, the voltage source 27 shown in FIG. 3). And the output of the second differential amplifier is connected to the gate of the fourth MOS transistor and to a terminal for outputting the first control voltage, and the second control voltage generating circuit (for example, FIG. 3 includes a reference resistance element (eg, the resistance element 33 shown in FIG. 3) externally attached to the IC chip, and one end of the reference resistance element is connected to a second voltage source (eg, FIG. 3). Vss), the other end is connected to a non-inverting input terminal of a third differential amplifier (eg, the differential amplifier 32 shown in FIG. 3), and a fifth MOS transistor (eg, shown in FIG. 3). The OS transistor 31) is connected to the drain terminal, the inverting input terminal of the third differential amplifier is connected to a fourth reference voltage (for example, the voltage source 37 shown in FIG. 3), and the output of the third differential amplifier is The second MOS transistor is preferably connected to the gate of the fourth MOS transistor and to a terminal for outputting the second control voltage.

  In the operational transconductance amplifier according to the present invention, in the above-described invention, the voltage-current converter is configured such that the first input signal is input to the gate terminal, and the first output signal of the voltage-current converter is An eleventh MOS transistor (for example, the MOS transistor 132 shown in FIG. 1) whose first output terminal is connected to the drain terminal and a second input signal is input to the gate terminal, and the second input of the voltage-current converter is A twelfth MOS transistor (for example, the MOS transistor 133 shown in FIG. 1) whose second output terminal from which an output signal is output is connected to the drain terminal, and a drain terminal are the source terminal of the eleventh MOS transistor and the gate of the fourteenth MOS transistor. And the gate terminal is connected to the source terminal of the twelfth MOS transistor and the first terminal. A thirteenth MOS transistor connected to the drain of the MOS transistor (for example, the MOS transistor 134 shown in FIG. 1), a drain terminal connected to the source terminal of the twelfth MOS transistor and the gate terminal of the thirteenth MOS transistor, and a gate terminal connected to the first MOS transistor A 14th MOS transistor (for example, the MOS transistor 135 shown in FIG. 1) connected to the source terminal of the 11MOS transistor and the drain of the 13th MOS transistor, and the source terminal of the 13th MOS transistor is a first current source (for example, FIG. 1). Connected to one terminal of an eleventh internal resistance element (for example, the resistance element 143 shown in FIG. 1) formed on the IC chip and connected to the source terminal of the fourteenth MOS transistor. But second Current sources (e.g., current source 137 shown in FIG. 1) the other are preferably connected to the terminals of the first 11 internal resistance elements as well as connect to.

  According to the operational transconductance amplifier of the present invention, in the above-described invention, the voltage-current converter has an eleventh input signal input to the gate terminal, and the eleventh output signal of the voltage-current converter is A twenty-first MOS transistor (for example, MOS transistor 132 shown in FIG. 4) whose eleventh output terminal is connected to the drain terminal and a twelfth input signal is inputted to the gate terminal, and the twelfth input of the voltage-current converter is inputted. The 22nd MOS transistor (for example, the MOS transistor 133 shown in FIG. 4) in which the 12th output terminal from which the output signal is output is connected to the drain terminal, and the drain terminal are the source terminal of the 21st MOS transistor and the gate of the 24th MOS transistor. And the gate terminal is the source of the 22nd MOS transistor. A 23rd MOS transistor (for example, MOS transistor 134 shown in FIG. 4) connected to the child and the drain of the 24th MOS transistor, and a drain terminal connected to the source terminal of the 22nd MOS transistor and the gate terminal of the 23rd MOS transistor; Includes a 24th MOS transistor (for example, MOS transistor 135 shown in FIG. 4) connected to the source terminal of the 21st MOS transistor and the drain of the 23rd MOS transistor, and the source terminal of the 23rd MOS transistor is formed on the IC chip. Connected to one terminal of the twenty-first internal resistance element (for example, the resistance element 148 shown in FIG. 4), and the source terminal of the 24th MOS transistor is formed on the IC chip. It is connected to one terminal of the resistor 149) shown in desirable.

  In the operational transconductance amplifier according to the present invention, the voltage-current converter is configured such that the first input signal is input to the non-inverting input terminal and the inverting input terminal is the 31st MOS transistor (for example, FIG. 5 is connected to the source terminal of the MOS transistor 62 shown in FIG. 5 and the 31st current source (eg, current source 64 shown in FIG. 5), and the 31st internal resistance element (eg shown in FIG. 5) formed on the IC chip. A resistance element 68) connected to one terminal and an output terminal connected to the gate terminal of the 31st MOS transistor (for example, the differential amplifier 66 shown in FIG. 5), and a non-inverting input terminal The second input signal is input, and the inverting input terminal is the source terminal of the 32nd MOS transistor (for example, the MOS transistor 63 shown in FIG. 5). And the thirty-second current source (for example, the current source 65 shown in FIG. 5), the other terminal of the thirty-first internal resistance element, and the output terminal connected to the gate terminal of the thirty-second MOS transistor. And a 31st output terminal from which the first output signal of the voltage-current converter is output is connected to the drain terminal of the 31st MOS transistor, and a differential amplifier (for example, the differential amplifier 67 shown in FIG. 5). The 32nd output terminal from which the second output signal of the voltage-current converter is output is preferably connected to the drain terminal of the 32nd MOS transistor.

  In the operational transconductance amplifier according to the present invention, the voltage-current converter is configured such that the first input signal is input to the non-inverting input terminal and the inverting input terminal is the 41st MOS transistor (for example, FIG. 6 is connected to the source terminal of the MOS transistor 62) and one terminal of the 41st internal resistance element (for example, internal resistance element 73 shown in FIG. 6) formed on the IC chip, and the output terminal is connected to the 41st MOS transistor. A 41st differential amplifier (for example, the differential amplifier 66 shown in FIG. 6) connected to the gate terminal of the transistor, a second input signal is input to the non-inverting input terminal, and a 42nd MOS transistor (for example, FIG. 6) is input to the inverting input terminal. The 42nd internal resistance element (for example, formed on the source terminal of the MOS transistor 63) and the IC chip shown in FIG. A forty-second differential amplifier (for example, differential amplifier 67 shown in FIG. 6) connected to one terminal of the internal resistance element 74) shown in FIG. 6 and having an output terminal connected to the gate terminal of the forty-second MOS transistor; A first output signal of the voltage-current converter is connected to a drain terminal of the 41th MOS transistor, and a second output signal of the voltage-current converter is output. The 42 output terminal is preferably connected to the drain terminal of the 42nd MOS transistor.

  In the operational transconductance amplifier according to the present invention, in the above-described invention, the voltage-current converter has a first input signal input to the gate terminal and a 51st current source (for example, as shown in FIG. 7) to the drain terminal. And the source terminal is connected to the source terminal of the 52nd MOS transistor (eg, MOS transistor 82 shown in FIG. 7) and the 52nd current source (eg, current source 84 shown in FIG. 7). A 51st MOS transistor (for example, MOS transistor 93 shown in FIG. 7) connected to one terminal of a 51st internal resistance element (for example, resistance element 88 shown in FIG. 7) formed on the IC chip, Connected to the drain terminal of the 51MOS transistor, and the output terminal connected to the gate of the 52nd MOS transistor. 51 inverter (for example, the inverter 86 shown in FIG. 7), the second input signal is input to the gate terminal, the 53rd current source (for example, the MOS transistor 96 shown in FIG. 7) is connected to the drain terminal, and the source terminal is connected. It is connected to the source terminal of the 54th MOS transistor (for example, the MOS transistor 83 shown in FIG. 7) and the 54th current source (for example, the current source 85 shown in FIG. 7) and to the other terminal of the 51st internal resistance element. A 53rd MOS transistor (for example, MOS transistor 94 shown in FIG. 7) and a 52nd inverter (for example, FIG. 7) having an input terminal connected to the drain terminal of the 53MOS transistor and an output terminal connected to the gate terminal of the 54th MOS transistor. And a first output signal of the voltage-current converter is The 51st output terminal is connected to the drain terminal of the 52nd MOS transistor, and the 52nd output terminal from which the second output signal of the voltage-current converter is output is connected to the drain terminal of the 54th MOS transistor. It is desirable.

In the operational transconductance amplifier according to the present invention, in the above-described invention, the voltage-current converter is configured such that the first input signal is input to the gate terminal, and the 61st current source (for example, FIG. The current source 95 shown in FIG. 8 is connected to the source terminal of the 62nd MOS transistor (eg MOS transistor 82 shown in FIG. 8) and the 62nd internal resistance element (eg shown in FIG. 8) formed on the IC chip. A 61st MOS transistor (for example, the MOS transistor 93 shown in FIG. 8) connected to one terminal of the internal resistance element 97), an input terminal is connected to the drain terminal of the 61MOS transistor, and an output terminal is the gate of the 62nd MOS transistor. A 61st inverter (for example, the inverter 86 shown in FIG. 8) connected to the The second input signal is input to the first terminal, the 62nd current source (for example, the current source 96 shown in FIG. 8) is connected to the drain terminal, and the 64th MOS transistor (for example, the MOS transistor 83 shown in FIG. 8) is connected to the source terminal. A 63th MOS transistor (for example, the MOS transistor 94 shown in FIG. 8) connected to one terminal of the source terminal and the 63rd internal resistance element (for example, the internal resistance element 98 shown in FIG. 8) formed on the IC chip; A 62nd inverter (for example, the inverter 87 shown in FIG. 8) whose input terminal is connected to the drain terminal of the 63th MOS transistor and whose output terminal is connected to the gate of the 64th MOS transistor. The 61st output terminal of the voltage-current converter from which the first output signal is output is connected to the 62nd MOS transistor. It is connected to the drain terminal of static, the voltage - current converter of the second output signal that is the 62 output terminal to be output is connected to the drain terminal of the first 64MOS transistor desirable.
Further, a filter circuit using the operational transconductance amplifier of the present invention includes any one of the operational transconductance amplifier described above and a capacitive element (for example, the capacitive element 45 shown in FIG. 46).

  According to the present invention, since the transconductance value is directly affected by the manufacturing fluctuation and the temperature fluctuation, the OTA is used in combination with the current amplifier that eliminates the influence of the manufacturing fluctuation and the temperature fluctuation of the transconductance value. It is possible to realize an OTA having excellent linearity with a wide input voltage range that is not affected by manufacturing fluctuations and temperature fluctuations. Further, by using this OTA in combination with a capacitor, it is possible to provide an OTA-C filter having high frequency characteristic accuracy, high speed and excellent linearity.

It is a circuit diagram for demonstrating OTA of Embodiment 1 of this invention. It is a circuit diagram for demonstrating the current control circuit used for Embodiment 1 of this invention. FIG. 3 is a control voltage generation circuit diagram used in the current control circuit of FIG. 2. FIG. 5 is an OTA circuit diagram of the present invention for explaining a second embodiment. FIG. 5 is an OTA circuit diagram of the present invention for explaining a third embodiment. It is an OTA circuit diagram of the present invention for explaining Embodiment 3-2. FIG. 6 is an OTA circuit diagram of the present invention for explaining a fourth embodiment. It is an OTA circuit diagram of the present invention for explaining embodiment 4-2. It is a circuit diagram for demonstrating the OTA-C filter of Embodiment 5 of this invention. It is a conventional OTA circuit diagram. It is a conventional OTA circuit diagram, and is a voltage-current converter used in the first embodiment. It is a circuit diagram for demonstrating the conventional OTA-C filter.

Hereinafter, Embodiments 1 to 5 of the present invention will be described with reference to the drawings.
(Embodiment 1)
The OTA of the first embodiment will be described with reference to FIG. FIG. 1 includes a voltage-current converter (hereinafter referred to as a V-I converter) 8 surrounded by a dotted line, current control circuits 1 and 2 having a current amplification factor A, and MOS transistors 3 and 4. ing. The current control circuits 1 and 2 are exactly the same circuit. The VI converter 8 surrounded by a dotted line corresponds to a portion obtained by removing the MOS transistors 130 and 131 from the conventional OTA shown in FIG.
The VI converter 8 in FIG. 1 can also be referred to as OTA, but in order to distinguish it from the OTA in which the VI converter 8 and the current control circuits 1 and 2 are combined, in the following description, in FIG. The VI converter 8 is not referred to as OTA but is also referred to as an OTA main body.

  Next, the operation of the OTA in FIG. 1 will be described. Since the VI converter 8 has already been described, it is omitted to some extent. Also, among the elements and terminals shown in FIG. 1, the same elements and terminals as those shown in FIG. In the OTA of FIG. 1, a voltage Vin is input as an input signal to terminals 138 and 139 that are input terminals of the VI converter 8. The input voltage signal Vin is converted into a current signal I1. The current signal I1 is input to the current control circuits 1 and 2 from the terminals 140 and 141, multiplied by A by the current control circuits 1 and 2, and then output from the output terminals 5 and 6 of the OTA as the output current I2.

The MOS transistors 3 and 4 are for controlling the common mode level of the voltage at the output terminals 5 and 6, and the common mode control signal is supplied from the gate terminal 7 of the MOS transistors 3 and 4. As described with reference to FIG. 11, the output current of the V-I converter 8 is affected by manufacturing variations and temperature variations of the resistance element 143 used for current conversion in the V-I converter 8. The output current I1 from the V-I converter 8 is input to the current control circuits 1 and 2 and controlled so that the current value becomes A times. The output current I2 is output from which the influence of fluctuation due to the resistance element 143 is eliminated.
The conversion coefficient of the VI converter in FIG. 1 is the reciprocal of the resistance value R of the resistance element formed on the chip, as shown in the equation (14). For this reason, the relationship between the input voltage Vin and the output current I2 output from the output terminals 5 and 6 in FIG. 1 is expressed by equation (16) using equation (14).
I2 = I1.A = (1 / R) .A.Vin (16)

Next, the current control circuits 1 and 2 will be described with reference to FIG. As described above, since the current control circuit has the same configuration, only one of the current control circuits 1 and 2 is illustrated in FIG.
The current control circuits 1 and 2 shown in FIG. 2 have voltage sources 16 and 17 for supplying control voltages VC1 and VC2 to the MOS transistors 10, 11, and 13, the differential amplifier 12, and the gate terminals of the MOS transistors 10 and 11, respectively. Consists of The value (Vgs−Vth) obtained by subtracting the threshold voltage Vth from the gate-source voltage Vgs of the MOS transistors 10 and 11 is higher than the drain-source voltage Vds. In this case, the drain / source current Ids characteristics with respect to Vgs of the MOS transistors 10 and 11 can be expressed by Expression (17), which is well known as a linear operation region.

Ids = 2K · (Vgs−Vth−0.5Vds) · Vds Equation (17)
In the formula (17), K is represented by the above-described formula (2). In the formula (2), W, L, μ, and Cox are the channel width, channel length, mobility, and gate capacitance per unit area of the MOS transistors 10 and 11, respectively. Vth is the threshold voltage of the MOS transistors 10 and 11. According to Expression (17), when (Vgs−Vth) is sufficiently larger than Vds, Expression (17) can be approximated as Expression (18). At this time, Ids is proportional to Vds, and it can be seen that it operates in the linear region.
Ids = 2K · (Vgs−Vth) · Vds Equation (18)
In other words, when (Vgs−Vth) is sufficiently larger than Vds, the resistance values of the MOS transistors 10 and 11 are the same as the resistance value R of the resistance element as shown in the following equation (19).
R = 1 / {2K · (Vgs−Vth)} (19)
As can be seen from equation (19), the resistance value R can be controlled by the voltage Vgs supplied to the gate of the MOS transistor.

Next, the operation of the current control circuit 1 shown in FIG. 2 will be described.
The input current I14 is supplied from the terminal 14, passes through the MOS transistor 10, and flows to the positive power supply terminal 19. The output current I15 flows from the terminal 15 through the MOS transistor 13 and the MOS transistor 11 to the positive power supply terminal 19.
The resistance values of the MOS transistors 10 and 11 operating in the linear region are R10 and R11, respectively. In this case, the voltages at the terminals 14 and 18 are Vdd−I14 · R10 and Vdd−I15 · R11, respectively, from Ohm's law. Here, Vdd is a positive power supply voltage.

If the voltage at the terminal 14 is lower than that at the terminal 18, the output of the differential amplifier 12 becomes low, and as a result, the voltage at the terminal 18 becomes low. Conversely, when the voltage at the terminal 14 is higher than that at the terminal 18, the output of the differential amplifier 12 becomes high, and as a result, the voltage at the terminal 18 becomes high. In this way, when the gain of the differential amplifier 12 is sufficiently high, the voltages at the terminals 14 and 18 are equal. Such a relationship can be expressed by Expression (20).
I14 · R10 = I15 · R11 (20)

When Expression (20) is rewritten as Expression (21), the ratio of the output current to the input current, that is, the current amplification factor A becomes (R10 / R11) as shown in Expression (22).
I2 = (R10 / R11) · I1 Formula (21)
A = (R10 / R11) ... Formula (22)
Here, when the voltage VC2 of the voltage source 17 shown in FIG. 2 is changed, the resistance value R11 of the MOS transistor 11 is changed by the equation (19). For this reason, the current amplification factor A given by the equation (22) also changes as the resistance value R11 changes. That is, in FIG. 2, by controlling the voltage of the voltage source 17, the circuit shown in FIG. 2 can be used as a current control circuit.

Similarly, the current amplification factor A can be changed by changing the voltage of the voltage source 16. Thus, in the current control circuit of the OTA of the present invention, by controlling either one or both of the voltage source 16 and the voltage source 17, 1 / R of which appears in the conversion coefficient of the VI converter 8 of FIG. By compensating for manufacturing fluctuations and temperature fluctuations, the current at the output terminal 5 of the OTA in FIG. 1 is not affected by manufacturing fluctuations and temperature fluctuations.
Next, a specific circuit of the voltage sources 16 and 17 is shown in FIG. Since the voltage sources 16 and 17 supply a voltage to the gate terminals of the MOS transistors 10 and 11 in FIG. 2 to control the current amplification factor, they are hereinafter also referred to as control voltage generation circuits. The control voltage generation circuit includes a circuit 28 corresponding to the voltage source 16 for generating the voltage VC1 and a circuit 38 corresponding to the voltage source 17 for generating the voltage VC2.

  The circuit 28 includes a MOS transistor 21, a resistance element 23, a differential amplifier 22, and a voltage source 27 that supplies a voltage Va to the inverting input terminal 25. The resistance element 23 is a resistance element formed on the IC, and is the same type of resistance element as the resistance element 143 of the VI converter 8 of FIG. Here, the “same type of resistor” is a resistor formed in the same manufacturing process, and is formed in the same manufacturing process, so that there is no manufacturing variation or temperature variation between the resistance elements.

The resistance value of the resistance element 23 is R1. Since the resistance element 143 in FIG. 1 and the resistance element 23 in FIG. 3 are the same type of resistance element, even if the respective values fluctuate due to manufacturing variations and temperature fluctuations, the resistance ratio R1 / R is always constant. is there.
The circuit 38 includes a MOS transistor 31, a resistance element 33, a differential amplifier 32, and a voltage source 37 that supplies a voltage Vb to a terminal 35. The resistance element 33 is a reference resistance element, and is externally attached to the IC, for example. Therefore, unlike the resistance element 23, the resistance value R2 is always a predetermined resistance value without being affected by manufacturing variations and temperature variations. Therefore, the resistance element 33 is realized by externally attaching to the IC.

Next, the operation of the voltage sources 16 and 17 shown in FIG. 3 will be described. Since the circuit 28 and the circuit 38 are similar, the circuit 28 will be described in detail first.
When the voltage V26 supplied to the non-inverting input terminal 26 of the differential amplifier 22 is lower than the voltage V25 supplied to the inverting input terminal 25, the output voltage VC1 of the differential amplifier 22 becomes low. This output voltage VC1 is supplied to the gate terminal of the MOS transistor 21, and the MOS transistor 21 and the resistance element 23 constitute an inverter. At this time, the voltage at the non-inverting input terminal 26 corresponding to the output of the inverter is inverted and increased.

  Conversely, when the voltage V26 supplied to the non-inverting input terminal 26 of the differential amplifier 22 is higher than the voltage V25 supplied to the inverting input terminal 25, the output voltage VC1 of the differential amplifier 22 becomes high. This output voltage VC1 is supplied to the gate terminal of the MOS transistor 21, and the MOS transistor 21 and the resistance element 23 constitute an inverter. At this time, the voltage at the non-inverting input terminal 26 corresponding to the output of the inverter is inverted and lowered. Thus, since the differential amplifier 22, the MOS transistor 21, and the resistance element 23 form a negative feedback circuit, if the gain of the differential amplifier 22 is sufficiently high, the voltage V26 of the non-inverting input terminal 26 and the inverting input terminal The voltage Va of 25 is equal.

Here, if the gate-source voltage of the MOS transistor 21 is sufficiently large to be in a linear region, the MOS transistor 21 operates as a resistance element having a resistance value R21. On the other hand, the MOS transistor 21 and the resistance element 23 are inserted between the positive power supply terminal 30 to which the voltage Vdd is supplied and the negative power supply terminal 29 to which the voltage Vss is supplied. Equation (23) holds from the law.
(Vdd-Va) / R21 = (Va-Vss) R1 (23)
From equation (23), the resistance value R21 can be expressed by equation (24).
R21 = R1. (Vdd-Va) / (Va-Vss) (24)

The voltage VC1 generated in the circuit 28 is supplied to the gate of the MOS transistor 21 and also to the gate of the MOS transistor 10 in FIG. Here, in order to simplify the description, it is assumed that the sizes of the MOS transistors 10 and 21 are the same. At this time, the resistance values R10 and R21 of the MOS transistors 10 and 21 are the same because the gate voltages of the MOS transistors 10 and 21 are the same.
Similarly, for the circuit 38, the resistance value R31 of the MOS transistor is given by equation (25).
R31 = R2 · (Vdd−Vb) / (Vb−Vss) Equation (25)

The voltage VC2 generated by the circuit 38 is supplied to the gate of the MOS transistor 31 and also to the gate of the MOS transistor 11 in FIG. Here, in order to simplify the description, it is assumed that the sizes of the MOS transistors 11 and 31 are the same. At this time, the resistance values R11 and R31 of the MOS transistors 11 and 31 have the same value because the gate voltages of the MOS transistors 11 and 31 are the same.
Therefore, the current amplification factor A of the current control circuits 1 and 2 of FIG. 2 shown by the equation (22) is expressed by the equation (26) from the equations (24) and (25).
A = (R10 / R11) = (R21 / R31)
= (R1 / R2). (Vdd-Va) (Vb-Vss) / (Va-Vss)
/ (Vdd-Vb) ... Formula (26)

From equation (26), if Va and Vb are constant, terms other than (R1 / R2) are always constant. Therefore, the output current I2 with respect to the input voltage Vin of the OTA circuit of FIG. 1 can be expressed by the equation (27) from the equations (16) and (26).
I2 = Vin. (R1 / R2). (1 / R) .B Equation (27)
However, B in Formula (27) is a constant represented by Formula (28).
B = (Vdd-Va) (Vb-Vss) / (Va-Vss) / (Vdd-Vb) (28)
Furthermore, Expression (27) and Expression (28) can be described as Expression (29) and Expression (30).
I2 = Vin · D (29)
D = (R1 / R2) · (1 / R) · B Formula (30)

  That is, the circuit of FIG. 1 can be said to be an OTA circuit whose transconductance value is D. In the equations (29) and (30), R1 and R are resistance values of the resistance elements formed on the IC, so that the ratio is always constant, and R2 is affected by manufacturing variations and temperature variations. This is the resistance value of the reference resistance element that does not occur. For this reason, the transconductance value D of the OTA in FIG. 1 is always constant regardless of manufacturing variations and temperature variations.

  The OTA of FIG. 1 has excellent linearity performance over a wide input voltage range, which is a feature of the OTA of FIG. 11, and has no influence of manufacturing fluctuations and temperature fluctuations by adding current control circuits 1 and 2. The characteristic is that it always has a constant transconductance value. Further, according to the first embodiment, as shown later, even when a plurality of OTAs are used, the control voltage generation circuit can be shared, so that only one external resistance element is required. Further, since this external resistance element is formed in a separate control voltage generation circuit separated from the OTA body, the bonding pad accompanying the attachment of the resistance element and the parasitic capacitance of the mounting substrate adversely affect the high-speed performance of the OTA. Will not affect. When an OTA-C filter is designed using such an OTA, it has excellent linearity performance over a wide input voltage range by adding only one external resistance element while maintaining high-speed performance. A filter having excellent characteristic accuracy can be provided.

(Embodiment 2)
The OTA of Embodiment 2 will be described with reference to FIG. In FIG. 4, the same components as those shown in FIG. 1 used in the description of the first embodiment are denoted by the same reference numerals, and the description thereof is partially omitted.
FIG. 4 includes a VI converter 480 surrounded by a dotted line, current control circuits 1 and 2 having a current amplification factor A, and MOS transistors 3 and 4. The difference between FIG. 4 and FIG. 1 is that, in the VI converter 480 surrounded by a dotted line, the resistance element 143 of FIG. 1 is removed, and a resistance having a resistance value R is used instead of the OTA current sources 136 and 137. The elements 148 and 149 are provided, and the other parts are all the same.

Next, the operation of the OTA in FIG. 4 will be described. The behavior of the output current I40 as a difference current output from the output terminals 140 and 141 of the VI converter 480 to the current control circuits 1 and 2 with respect to the input voltage Vin input from the input terminals 138 and 139 of the circuit of FIG. Will be described. The relationship between the terminal voltages V138 and V139 of the input terminals 138 and 139, the terminal voltages V146 and V147 of the terminals 146 and 147, and the input voltage Vin is the same as in FIG.
V146−V147 = V138−V139 = Vin Formula (31)

Further, the currents I148 and I149 flowing through the resistance element 148 and the resistance element 149 are expressed by the equations (32) and (33).
I148 = V146 / R Formula (32)
I149 = V147 / R Formula (33)
R is a resistance value of the resistance elements 148 and 149. Since the output current I40 as the difference current between the terminals 140 and 141 is also the difference current between the resistance elements 148 and 149, the input voltage Vin as shown in the expression (34) from the expressions (31), (32) and (33). Is divided by the resistance value R.
I40 = I146−I147 = (V146−V147) / R = Vin / R Equation (34)

  Thus, since the output current I40 is proportional to the value of the input voltage Vin, even when the input voltage is wide, it has a remarkably excellent linear performance. However, similarly to the VI converter 8 of FIG. 1, the resistance values of the resistance elements 148 and 149 expressed by the equation (34) are affected by manufacturing variations and temperature variations. As in the case of FIG. 1 of the first embodiment, the output currents appearing at the output terminals 5 and 6 of the OTA by the action of the current control circuits 1 and 2 are the same as the expressions (29) and (30). In other words, the OTA of the second embodiment has an excellent linearity performance over a wide input voltage range, which is a feature of the OTA of FIG. 11, and the addition of the current control circuits 1 and 2 has the effect of manufacturing variations and temperature variations. The characteristic is that it always has a constant transconductance value.

  Even when a plurality of OTAs are used, the control voltage generation circuit can be shared, so that only one external resistance element is required. When an OTA-C filter is designed using such an OTA, it has excellent linearity performance over a wide input voltage range by adding only one external resistance element while maintaining high-speed performance. A filter having excellent characteristic accuracy can be provided.

(Embodiment 3)
The OTA of the third embodiment will be described with reference to FIG. In FIG. 5, the same components as those shown in FIG. 1 of the first embodiment are denoted by the same reference numerals, and the description thereof is partially omitted. FIG. 5 includes a VI converter 580 surrounded by a dotted line, current control circuits 1 and 2 having a current amplification factor A, and MOS transistors 3 and 4. The difference between FIG. 5 and FIG. 1 is the VI converter 580 surrounded by a dotted line. That is, the VI converter 580 of the third embodiment includes differential amplifiers 66 and 67, current sources 64 and 65, MOS transistors 62 and 63, and a resistance element 68.

  Next, the operation of the OTA in FIG. 5 will be described. The behavior of the output current I40 as the difference current output from the output terminals 60 and 61 of the VI converter 580 to the current control circuits 1 and 2 with respect to the input voltage Vin input from the input terminals 69 and 70 of the circuit of FIG. Will be described. The differential amplifier 66, the current source 64, and the MOS transistor 62 form a negative feedback circuit that passes from the output of the differential amplifier 66 through the MOS transistor 62 to the inverting input terminal of the differential amplifier 66.

In such a case, if the gain of the differential amplifier 66 is sufficiently high, the voltage V71 of the inverting input terminal 71 (common to the source terminal of the MOS transistor 62) of the differential amplifier 66 becomes equal to the voltage V69 of the non-inverting input terminal 69. . Similarly, the voltage V72 at the inverting input terminal 72 of the differential amplifier 67 is equal to the voltage V70 at the non-inverting input terminal 70. The relationship between the terminal voltages V69 and V70 of the input terminals 69 and 70 and the terminal voltages V71 and V72 of the terminals 71 and 72 and the input voltage Vin is expressed by the expression (35) in the same manner as in FIG.
V71−V72 = V69−V70 = Vin Equation (35)
Therefore, the current I68 flowing through the resistance element 68 is as shown in Expression (36).
I68 = Vin / R ... Formula (36)

R is the resistance value of the resistance element 68. The output current I40 that is the difference current between the terminals 60 and 61 is the difference current between the MOS transistors 62 and 63, that is, the current that flows through the resistance element 68. Therefore, from the expressions (35) and (36), the expression (37) As shown, the input voltage Vin is divided by the resistance value R.
I40 = I68 = Vin / R ... Formula (37)
As described above, since the output current I40 is surely proportional to the value of the input voltage Vin, the OTA of the third embodiment has remarkably excellent linear performance even when the input voltage is wide. However, since the resistance value of the resistance element 68 expressed by the equation (37) is formed on the IC as in the case of the IV converter 8 of the first embodiment, it is affected by manufacturing variations and temperature variations.

  As in the case of the OTA of FIG. 1 described in the first embodiment, the output current appearing at the output terminals 5 and 6 of the OTA due to the action of the current control circuits 1 and 2 is the same as the expressions (29) and (30). Become. That is, the OTA of the third embodiment has excellent linearity performance over a wide input voltage range, which is a feature of the VI converter 580 of FIG. It is characterized by having a constant transconductance value that is not affected by temperature fluctuations.

  Even when a plurality of OTAs are used, the control voltage generation circuit can be shared, so that only one external resistance element is required. When an OTA-C filter is designed using such an OTA, it has excellent linearity performance over a wide input voltage range by adding only one external resistance element while maintaining high-speed performance. A filter having excellent characteristic accuracy can be provided.

Embodiment 3-2
The OTA of the embodiment 3-2 will be described with reference to FIG. In FIG. 6, the same components as those shown in FIG. 5 used in the description of the third embodiment are denoted by the same reference numerals, and the description thereof is partially omitted.
FIG. 6 includes a VI converter 780 surrounded by a dotted line, current control circuits 1 and 2 having a current amplification factor A, and MOS transistors 3 and 4. The difference between FIG. 6 and FIG. 5 is that in the VI converter 780 surrounded by a dotted line, the resistance element 68 of FIG. 5 is removed, and a resistance having a resistance value R is used instead of the current sources 64 and 65 of OTA. The elements 73 and 74 are provided, and the other parts are all the same.

Next, the operation of the OTA in FIG. 6 will be described. The behavior of the output current I40 as the difference current output from the output terminals 60 and 61 of the VI converter 780 to the current control circuits 1 and 2 with respect to the input voltage Vin input from the input terminals 69 and 70 of the circuit of FIG. Will be described. The relationship between the terminal voltages V69 and V70 of the input terminals 69 and 70, the terminal voltages V71 and V72 of the terminals 71 and 72, and the input voltage Vin is the same as in FIG.
V71−V72 = V69−V70 = Vin Formula (51)

Further, the currents I73 and I74 flowing through the resistance element 73 and the resistance element 74 are expressed by the equations (52) and (53).
I73 = V71 / R Formula (52)
I74 = V72 / R Formula (53)
R is the resistance value of the resistance elements 73 and 74. Since the output current I40 as the difference current between the terminals 60 and 61 is also the difference current between the resistance elements 73 and 74, the input voltage is expressed by the equation (54) from the equations (51), (52), and (53). A value obtained by dividing Vin by the resistance value R.
I40 = I73-I74 = (V71-V72) / R = Vin / R (54)

  Thus, since the output current I40 is proportional to the value of the input voltage Vin, even when the input voltage is wide, it has a remarkably excellent linear performance. However, similarly to the VI converter 8 of FIG. 1, the resistance values of the resistance elements 73 and 74 expressed in the equations (52) and (53) are affected by manufacturing fluctuations and temperature fluctuations. As in the case of FIG. 1 of the first embodiment, the output currents appearing at the output terminals 5 and 6 of the OTA by the action of the current control circuits 1 and 2 are the same as the expressions (29) and (30). That is, the OTA of the embodiment 3-2 has excellent linearity performance over a wide input voltage range, which is a feature of the VI converter 780 of FIG. It is characterized by having a constant transconductance value that is not affected by manufacturing fluctuations and temperature fluctuations.

  Even when a plurality of OTAs are used, the control voltage generation circuit can be shared, so that only one external resistance element is required. When an OTA-C filter is designed using such an OTA, it has excellent linearity performance over a wide input voltage range by adding only one external resistance element while maintaining high-speed performance. A filter having excellent characteristic accuracy can be provided.

(Embodiment 4)
The OTA of the fourth embodiment will be described with reference to FIG. In FIG. 7, the same components as those shown in FIG. 1 used for the description of the first embodiment are denoted by the same reference numerals, and the description thereof is partially omitted. FIG. 7 includes a V-I converter 680, current control circuits 1 and 2 having a current amplification factor A, and MOS transistors 3 and 4. The difference between FIG. 7 and FIG. 1 is in the VI converter 680 surrounded by a dotted line. That is, the VI converter 680 includes inverters 86 and 87, current sources 84, 85, 95, and 96, MOS transistors 93, 94, 82, and 83, and a resistance element 88.

Next, the operation of the OTA in FIG. 7 will be described. The behavior of the output current I40 as the difference current output from the output terminals 80 and 81 of the VI converter 680 to the current control circuits 1 and 2 with respect to the input voltage Vin input from the input terminals 89 and 90 of the circuit of FIG. Since is described in detail in Japanese Patent Application Laid-Open No. 2005-057535, only the results are described below. According to this, the relationship between the terminal voltages V89 and V90 of the input terminals 89 and 90, the terminal voltages V91 and V92 of the terminals 91 and 92, and the input voltage Vin is the same as that of the MOS transistors 93 and 94. Since it is supplied from the sources 95 and 96, it is expressed as shown in Expression (38) as in FIG.
V91-V92 = V89-V90 = Vin Formula (38)

Therefore, the current I88 flowing through the resistance element 88 is as shown in Expression (39).
I88 = Vin / R ... Formula (39)
R is the resistance value of the resistance element 88. Since the output current I40 as the difference current between the terminals 80 and 81 is the difference current between the MOS transistors 82 and 83, that is, the current flowing through the resistance element 88, the expression (40) is obtained from the expressions (38) and (39). As shown, the input voltage Vin is divided by the resistance value R.
I40 = I88 = Vin / R (40)
As described above, since the output current I40 is surely proportional to the value of the input voltage Vin, the OTA of the fourth embodiment has a remarkably excellent linear performance even when the input voltage is wide. However, similarly to the VI converter 8 of FIG. 1, the resistance value of the resistance element 88 expressed by the equation (40) is formed on the IC, and thus is affected by manufacturing variations and temperature variations.

As in the case of FIG. 1 of the first embodiment, the output currents appearing at the output terminals 5 and 6 of the OTA by the action of the current control circuits 1 and 2 are the same as the expressions (29) and (30). That is, the OTA of Embodiment 4 has excellent linearity performance over a wide input voltage range, which is a feature of the VI converter 680 in FIG. It is characterized by having a constant transconductance value that is not affected by temperature fluctuations.
Even when a plurality of OTAs are used, the control voltage generating circuit can be shared, so that only one external resistance element is required. When an OTA-C filter is designed using such an OTA, it has excellent linearity performance over a wide input voltage range by adding only one external resistance element while maintaining high-speed performance. A filter having excellent characteristic accuracy can be provided.

(Embodiment 4-2)
The OTA of the embodiment 4-2 will be described with reference to FIG. In FIG. 8, the same components as those shown in FIG. 7 used in the description of the fourth embodiment are denoted by the same reference numerals, and the description thereof is partially omitted.
FIG. 8 includes a VI converter 880 surrounded by a dotted line, current control circuits 1 and 2 having a current amplification factor A, and MOS transistors 3 and 4. The difference between FIG. 8 and FIG. 7 is that in the VI converter 880 surrounded by a dotted line, the resistance element 88 of FIG. 7 is removed, and a resistance having a resistance value R is used instead of the current sources 84 and 85 of OTA. The elements 97 and 98 are provided, and the other parts are all the same.

Next, the operation of the OTA in FIG. 8 will be described. The behavior of the output current I40 as the difference current output from the output terminals 80 and 81 of the VI converter 880 to the current control circuits 1 and 2 with respect to the input voltage Vin input from the input terminals 89 and 90 of the circuit of FIG. Will be described. The relationship between the terminal voltages V89 and V90 of the input terminals 89 and 90, the terminal voltages V91 and V92 of the terminals 91 and 92, and the input voltage Vin is the same as in FIG.
V91-V92 = V89-V90 = Vin Formula (61)

Further, the currents I97 and I98 flowing through the resistance element 97 and the resistance element 98 are expressed by the equations (62) and (63).
I97 = V91 / R Formula (62)
I98 = V92 / R Formula (63)
R is the resistance value of the resistance elements 97 and 98. Since the output current I40 as the difference current between the terminals 80 and 81 is also the difference current between the resistance elements 97 and 98, the input voltage as shown in the expression (64) from the expressions (61), (62), and (63). A value obtained by dividing Vin by the resistance value R.
I40 = I97-I98 = (V91-V92) / R = Vin / R (64)

  Thus, since the output current I40 is proportional to the value of the input voltage Vin, even when the input voltage is wide, it has a remarkably excellent linear performance. However, similarly to the VI converter 8 of FIG. 1, the resistance values of the resistance elements 97 and 98 expressed by the equations (62) and (63) are affected by manufacturing variations and temperature variations. As in the case of FIG. 1 of the first embodiment, the output currents appearing at the output terminals 5 and 6 of the OTA by the action of the current control circuits 1 and 2 are the same as the expressions (29) and (30). That is, the OTA of the embodiment 4-2 has an excellent linearity performance over a wide input voltage range, which is a feature of the VI converter 880 in FIG. It is characterized by having a constant transconductance value that is not affected by manufacturing fluctuations and temperature fluctuations.

  Even when a plurality of OTAs are used, the control voltage generation circuit can be shared, so that only one external resistance element is required. When an OTA-C filter is designed using such an OTA, it has excellent linearity performance over a wide input voltage range by adding only one external resistance element while maintaining high-speed performance. A filter having excellent characteristic accuracy can be provided.

(Embodiment 5)
Next, an embodiment of the OTA-C filter using the OTA shown in the first to fourth embodiments will be described with reference to FIG. In FIG. 9, the same components as those shown in FIG. 12 are denoted by the same reference numerals, and the description thereof is partially omitted. 9, 41 to 44 are OTAs having Gm values of Gm1 to Gm4, 45 and 46 are capacitive elements having capacitance values of C1 and C2, and 49 and 50 are input terminal pairs to which input signals are inputted as differentials. , 51 and 52 are differential output terminal pairs that output a low-pass filter (LPF) output signal as differentials, 53 and 54 are output terminal pairs that output a band-pass filter (BPF) output signal, and 47 is a current control circuit. 3 is a control voltage generation circuit shown in FIG. 3 for supplying a control signal, and the same circuit as that shown in FIG. 3 can be used. Control voltages VC1 and VC2 are commonly supplied to OTAs 41-44.

In FIG. 9, the circuit 48 excluding the control voltage generation circuit 47 is generally known as a biquadratic OTA-C filter as described in FIG. 12, and a terminal pair 51 for outputting a low-pass filter signal. The transfer function H (s) representing the filter frequency characteristic at 52 can be expressed by the same equation (41) as the equation (15).
H (s) = (Gm1 · Gm2) / (C1 · C2) /
[s ² + s (Gm4 / C1) + [Gm2 · Gm3 / (C1 · C2)]] (41)

  As described above, the filter of FIG. 9 can realize a filter having an arbitrary characteristic using Gm1 to Gm4 and capacitors C1 and C2 as in the filter of FIG. As the OTA of the OTA-C filter shown in FIG. 9, the OTAs shown in FIGS. 1 and 4 to 8 can be used. For example, in the OTA-C filter of FIG. 9, the OTA shown in FIG. 1 can be used for Gm1 to Gm4. As the current control circuits 1 and 2 in the OTA circuit, the circuit shown in FIG. 2 can be used, and the control signals from the voltage sources 16 and 17 used in FIG. 2 are the same as those in FIG. Supplied from the control voltage generation circuit 47.

  Since the OTA-C filter of Embodiment 5 uses the OTA of FIG. 1 as Gm1 to Gm4, it has a performance with excellent linearity over a wide input voltage range. The four OTAs are supplied with a common signal from the control voltage generation circuit 47 to the current control circuit. That is, the OTA-C filter of FIG. 9 only needs to have one external resistance element no matter how the filter order increases. In the fifth embodiment, in addition to the high-speed performance inherently possessed by the OTA-C filter, a filter having excellent linearity over a wide input voltage range and excellent frequency characteristic accuracy can be provided. .

Furthermore, since this external resistance element is provided in a voltage generation circuit that is a separate circuit from the OTA main body, the bonding pad for providing the external resistance element and the parasitic capacitance of the mounting substrate are High speed performance is not adversely affected. Moreover, even if it uses any OTA shown in FIG. 1 and FIGS. 4-8 as OTA, the same outstanding effect can be provided.
Moreover, although Embodiment 5 demonstrated using the biquadratic filter like FIG. 9 as embodiment of an OTA-C filter, Embodiment 5 is not limited to such a structure. For example, the OTA-C filter of the fifth embodiment is various types of OTA-C filters using the OTA of the first to fourth, third, second, and fourth embodiments, such as a leapfrog filter and a state variable filter. Can also be configured.

  Since the OTA of the present invention has excellent linear performance, it can be suitably applied to high-speed and high-precision OTA-C filter design.

1, 2, Current control circuit 3, 4, 10, 11, 13, 21, 31, 62, 63, 82, 83, 93, 94, 132, 133, 134, 135 MOS transistor input terminal 138, 139
140, 141 Output terminal 8, 480, 580, 680 V-I converter 12, 22, 32, 66, 67 Differential amplifier 16, 17, 27, 37 Voltage source 23, 33, 68, 73, 74, 88, 97, 98, 143, 148, 149 Resistance element 28, 38, 48 Circuit 47 Control voltage generation circuit 64, 65, 84, 85, 136, 137 Current source

Claims (10)

A voltage-current converter comprising: an input terminal; an output terminal; and a first internal resistance element formed on at least one IC chip, wherein the amplification factor is the reciprocal of the resistance value of the first internal resistance element; ,
A current amplifier that inputs an output current output from an output terminal of the voltage-current converter and amplifies the output current with an amplification factor proportional to a resistance value of the first internal resistance element;
The current amplifier is
A first MOS transistor having a drain terminal to which an input current is input and a gate terminal to which a first control voltage is supplied;
A second MOS transistor having a drain terminal from which an output current is output;
A third MOS transistor having a gate terminal to which a second control voltage is supplied;
A non-inverting input terminal connected to the drain terminal of the first MOS transistor, an output terminal connected to the gate terminal of the second MOS transistor, a source terminal of the second MOS transistor, and a drain terminal of the third MOS transistor. And a first differential amplifier having an inverting input terminal.
The operational transconductance amplifier, wherein the first control voltage and the second control voltage are voltages having values that realize an amplification factor of the current amplifier.   2. The operation according to claim 1, wherein the first MOS transistor and the third MOS transistor have a drain-source voltage sufficiently smaller than a gate-source voltage and operate in a linear region. National transconductance amplifier. A first control voltage generating circuit for generating the first control voltage; and a second control voltage generating circuit for generating the second control voltage;
The first control voltage generation circuit includes:
A second internal resistance element manufactured by the same process as the first internal resistance element included in the voltage-current converter, wherein one end of the second internal resistance element is connected to the first voltage source, and the other end is The second differential amplifier is connected to the non-inverting input terminal of the second differential amplifier, connected to the drain terminal of the fourth MOS transistor, and the inverting input terminal of the second differential amplifier is connected to a third reference voltage. Is connected to the gate of the fourth MOS transistor and to a terminal for outputting the first control voltage,
The second control voltage generation circuit includes:
A reference resistance element externally attached to the IC chip, one end of the reference resistance element connected to the second voltage source, the other end connected to the non-inverting input terminal of the third differential amplifier, and a fifth MOS transistor; The drain terminal of the third differential amplifier, the inverting input terminal of the third differential amplifier is connected to a fourth reference voltage, and the output of the third differential amplifier is connected to the gate of the fourth MOS transistor and the second control. The operational transconductance amplifier according to claim 1, wherein the operational transconductance amplifier is connected to a terminal for outputting a voltage. The voltage-current converter is
An eleventh MOS transistor in which a first input signal is input to a gate terminal, and a first output terminal from which a first output signal of the voltage-current converter is output is connected to a drain terminal;
A twelfth MOS transistor in which a second input signal is input to the gate terminal, and a second output terminal from which the second output signal of the voltage-current converter is output is connected to the drain terminal;
A thirteenth MOS transistor having a drain terminal connected to the source terminal of the eleventh MOS transistor and the gate terminal of the fourteenth MOS transistor, and a gate terminal connected to the source terminal of the twelfth MOS transistor and the drain of the fourteenth MOS transistor;
A fourteenth MOS transistor having a drain terminal connected to a source terminal of the twelfth MOS transistor and a gate terminal of the thirteenth MOS transistor, and a gate terminal connected to a source terminal of the eleventh MOS transistor and a drain of the thirteenth MOS transistor;
A source terminal of the thirteenth MOS transistor is connected to a first current source and to one terminal of an eleventh internal resistance element formed on the IC chip;
4. The operation according to claim 1, wherein a source terminal of the fourteenth MOS transistor is connected to a second current source and to the other terminal of the eleventh internal resistance element. 5. National transconductance amplifier. The voltage-current converter is
An eleventh input signal input to the gate terminal, an eleventh output terminal from which the eleventh output signal of the voltage-current converter is output, a twenty-first MOS transistor connected to the drain terminal;
A twelfth MOS transistor having a twelfth input signal input to the gate terminal and a twelfth output terminal from which the twelfth output signal of the voltage-current converter is output connected to the drain terminal;
A 23rd MOS transistor having a drain terminal connected to a source terminal of the 21st MOS transistor and a gate terminal of the 24th MOS transistor, and a gate terminal connected to a source terminal of the 22nd MOS transistor and a drain of the 24th MOS transistor;
A drain terminal connected to a source terminal of the 22nd MOS transistor and a gate terminal of the 23rd MOS transistor, and a 24th MOS transistor connected to a source terminal of the 21st MOS transistor and a drain of the 23rd MOS transistor;
A source terminal of the 23rd MOS transistor is connected to one terminal of a 21st internal resistance element formed on the IC chip;
The operational terminal according to any one of claims 1 to 3, wherein a source terminal of the 24th MOS transistor is connected to one terminal of a 22nd internal resistance element formed on the IC chip. Transconductance amplifier. The voltage-current converter is
The first input signal is input to the non-inverting input terminal, the inverting input terminal is connected to the source terminal of the 31st MOS transistor and the 31st current source, and is connected to one terminal of the 31st internal resistance element formed on the IC chip. A thirty-first differential amplifier having an output terminal connected to the gate terminal of the thirty-first MOS transistor;
The second input signal is input to the non-inverting input terminal, the inverting input terminal is connected to the source terminal of the 32nd MOS transistor and the 32nd current source, and is connected to the other terminal of the 31st internal resistance element, and the output terminal is A thirty-second differential amplifier connected to the gate terminal of the thirty-second MOS transistor,
A thirty-first output terminal from which the first output signal of the voltage-current converter is output is connected to a drain terminal of the thirty-first MOS transistor;
4. The operation according to claim 1, wherein a 32nd output terminal from which the second output signal of the voltage-current converter is output is connected to a drain terminal of the 32nd MOS transistor. 5. National transconductance amplifier. The voltage-current converter is
The first input signal is inputted to the non-inverting input terminal, the inverting input terminal is connected to the source terminal of the 41st MOS transistor and one terminal of the 41st internal resistance element formed on the IC chip, and the output terminal is connected to the 41st MOS. A forty-first differential amplifier connected to the gate terminal of the transistor;
The second input signal is input to the non-inverting input terminal, the inverting input terminal is connected to the source terminal of the 42nd MOS transistor and one terminal of the 42nd internal resistance element formed on the IC chip, and the output terminal is connected to the 42nd MOS. A forty-second differential amplifier connected to the gate terminal of the transistor;
A 41st output terminal from which the first output signal of the voltage-current converter is output is connected to a drain terminal of the 41st MOS transistor;
4. The operation according to claim 1, wherein a forty second output terminal from which the second output signal of the voltage-current converter is output is connected to a drain terminal of the forty-second MOS transistor. 5. National transconductance amplifier. The voltage-current converter is
The first input signal is input to the gate terminal, the 51st current source is connected to the drain terminal, the source terminal is connected to the source terminal of the 52nd MOS transistor and the 52nd current source, and the 51st internal formed on the IC chip. A 51st MOS transistor connected to one terminal of the resistance element;
A 51st inverter having an input terminal connected to the drain terminal of the 51th MOS transistor and an output terminal connected to the gate of the 52nd MOS transistor;
The second input signal is input to the gate terminal, the 53rd current source is connected to the drain terminal, the source terminal is connected to the source terminal of the 54th MOS transistor and the 54th current source, and the other terminal of the 51st internal resistance element. A 53rd MOS transistor connected to
A 52nd inverter having an input terminal connected to the drain terminal of the 53MOS transistor and an output terminal connected to the gate of the 54th MOS transistor;
A 51st output terminal from which a first output signal of the voltage-current converter is output is connected to a drain terminal of the 52nd MOS transistor;
4. The operation according to claim 1, wherein a 52nd output terminal from which the second output signal of the voltage-current converter is output is connected to a drain terminal of the 54th MOS transistor. 5. National transconductance amplifier. The voltage-current converter is
The first input signal is input to the gate terminal, the 61st current source is connected to the drain terminal, the source terminal is connected to the source terminal of the 62nd MOS transistor and one terminal of the 62nd internal resistance element formed on the IC chip. 61st MOS transistor to
A 61st inverter having an input terminal connected to the drain terminal of the 61th MOS transistor and an output terminal connected to the gate of the 62nd MOS transistor;
The second input signal is input to the gate terminal, the 62nd current source is connected to the drain terminal, the source terminal is connected to the source terminal of the 64th MOS transistor and one terminal of the 63rd internal resistance element formed on the IC chip. A 63rd MOS transistor,
A 62nd inverter having an input terminal connected to the drain terminal of the 63th MOS transistor and an output terminal connected to the gate of the 64th MOS transistor;
A 61st output terminal from which the first output signal of the voltage-current converter is output is connected to a drain terminal of the 62nd MOS transistor;
4. The operation according to claim 1, wherein a 62nd output terminal from which the second output signal of the voltage-current converter is output is connected to a drain terminal of the 64th MOS transistor. 5. National transconductance amplifier.   A filter circuit using an operational transconductance amplifier, comprising: the operational transconductance amplifier according to claim 1; and a capacitive element.

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