JP2004147040A - Antenna assembly - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide an antenna assembly for which the beam width of the E surface element pattern of an excitation dipole antenna is widened. <P>SOLUTION: The antenna assembly is provided with the excitation dipole antenna 2 equipped separated from a conductor plate 1 by roughly 1/4 wavelength at a prescribed frequency, which resonates at the prescribed frequency, and a plurality of non-excitation monopole antennas 3 erected on the conductor plate 1 and provided with a length shorter than 1/4 wavelength at the prescribed frequency. The excitation dipole antenna 2 is provided with a feeding point C2 on a center part and is arranged roughly parallelly to the conductor plate 1. The respective non-excitation monopole antennas 3 are separately arranged along the extension direction of the excitation dipole antenna 2, provided with a feeding terminal C3 provided on the side connected with the conductor plate 1 and a reactance element 4 loaded on the feeding terminal C3, and provided roughly vertically on the conductor plate 1 such that an end part on the opposite side of the feeding terminal is turned to the side of the excitation dipole antenna 2. <P>COPYRIGHT: (C)2004,JPO

Description

【００４１】
ただし、式（１）において、Ｄ_１は非励振ダイポール７ａ（＃１）および７ｂ（＃３）からの（２素子アレーとしての）再放射パターン、Ｄ_２は励振ダイポール６ａ（＃２）、６ｂ（＃４）からの（同じく、２素子アレーとしての）放射パターンを示している。
また、式（１）において、ｃはＥ面ビーム幅を規定する正面方向からの振幅低下レベルである。
【００４２】
電流励振係数比Ｉ_１／Ｉ_２を複素平面上に展開したとき、上記式（１）から、Ｅ面ビーム幅を規定する領域は、円形を示し、この円形の外側領域として、所望Ｅ面ビーム幅を満たす電流励振係数比が与えられる。
この例を図１０の説明図に示す。
図１０において、「ビーム幅１」を規定する円形４１は破線で示され、「ビーム幅２」を規定する円形４２は一点鎖線で示され、「ビーム幅３」を規定する円形４３は、実線で示されている。
また、リアクタンス素子８ａ、８ｂのリアクタンス値Ｘを変化させた場合の電流励振係数比の軌跡４４は、点線で示されている。
【００４３】
ここで、「各ビーム幅１、２、３」の大小関係は、「ビーム幅１」＜「ビーム幅２」＜「ビーム幅３」を満たしており、ビーム幅が広がるほど、円形は小さくなりながら、その中心点は実軸（横軸）上を移動する。
図１０の場合、円形４３の外側領域に存在する電流励振係数比の軌跡４４の範囲は、ビーム幅として狭い円形４１の外側領域に存在する軌跡４４の範囲に比べて、極めて小さくなっている。
このことは、ビーム幅を拡大すればするほど、この条件を満たす電流励振係数比は少なくなることを意味している。
【００４４】
ここで注意すべき点は、ある所望ビーム幅に対して上記式（１）で表される条件を満たす電流励振係数比を求める際、所望ビーム幅より狭いビーム幅においては、常に条件を満足している必要があるということである。
すなわち、図１０から明らかなように、「ビーム幅３」を満足する電流励振係数比の範囲は、「ビーム幅１」および「ビーム幅２」においても常に成立している。
以上のことから、条件を表す式（１）を電流励振係数比Ｉ_１／Ｉ_２の複素平面に展開した場合、求める電流励振係数比は、所望ビーム幅を規定する円形領域の外側で表すことができ、このことは、図１０のように、図形的に容易に解釈することができる。
【００４５】
実施の形態６．
なお、上記実施の形態５では、電流励振係数比に対する複素平面上での図形的解釈を適用して、ビーム幅拡大を実現するリアクタンス値Ｘの条件について詳細に言及しなかったが、以下のように、ビーム幅拡大を実現するためのリアクタンス値Ｘの条件を設定してもよい。
以下、図１１を参照しながら、この発明の実施の形態６によるリアクタンス値Ｘの条件について説明する。ここでは、前述の電流励振係数比に対する複素平面上での図形的解釈を適用して、ビーム幅拡大を実現するリアクタンス値Ｘの条件について設定する。
【００４６】
図１１はこの発明の実施の形態６によるビーム幅拡大条件を示す説明図であり、電流励振係数比に対する複素平面を示している。
図１１において、破線で示す半径Ｒ_０の円５１は、ビーム幅を規定する領域であり、実線で示す半径ｒ_０の円５２は、リアクタンス値Ｘを変化させた場合に電流励振係数比の描く軌跡である。なお、流励振係数比の軌跡５２が図１１のように円を描くことは、数学的に証明されている。
ここで、図１１のように、規定領域を示す円５１と電流励振係数比の軌跡５２とが交点ｑ_１、ｑ_２を有し、円５１の外側領域に軌跡５２が存在する場合には、所望ビーム幅を満たす電流励振係数比が存在することになる。すなわち、この場合、リアクタンス素子４（図１参照）のリアクタンス値Ｘを決定することができる。
【００４７】
また、ここでは、図９のように、図１に示すアンテナ装置を等価的にモデル化し、各素子に番号＃１〜＃４を付して、上記条件について説明する。
前述した通り、リアクタンス素子８ａ、８ｂの変化に対して、非励振ダイポール７ａ、７ｂの励振ダイポール６ａ、６ｂに対する電流励振係数比は、複素平面上で円軌跡を描くことが数学的に証明される（ここでは、説明を省略する）。
また、前述の式（１）から励振ダイポールのＥ面ビーム幅を規定する領域も円形となる（図１１内の円５１参照）。
所望ビーム幅を満足する電流励振係数比が存在する条件は、上記２つの円が交点を有し、Ｅ面ビーム幅規定領域円よりも外側領域に電流励振係数比の軌跡があることである。
【００４８】
まず、図１１において、規定領域の円５１の中心をＰ_０、電流励振係数比の軌跡５２の中心をｐ_０とし、上記２つの円５１、５２の交点をｑ_１、ｑ_２とする。また、規定領域の中心点Ｐ_０と軌跡の中心点ｐ_０とを結ぶ直線を線分Ａとし、軌跡の中心点ｐ_０と一方の交点ｑ_１とを結ぶ直線を線分Ｂとする。
次に、線分Ａと実軸（横軸）との成す角を
【数７】

とし、線分Ａと線分Ｂとの成す角を
【数８】

とする。
また、図９内の非励振ダイポール７ａ（非励振モノポールアンテナ：＃１）と励振ダイポール６ａ（＃２）との間の相互インピーダンスＺ_１２を、位相ξを用いて、
【数９】

のように、極座標で表示する。
同様に、非励振ダイポール（＃１）の自己インピーダンスＺ_１１と、非励振ダイポール７ａ（＃１）と非励振ダイポール７ｂ（＃３）との間の相互インピーダンスＺ_１３との差分を、位相ηを用いて、
【数１０】

のように、極座標で表示する。
なお、自己インピーダンスＺ_１１、相互インピーダンスＺ_１２、Ｚ_１３は、周知の起電力法などによって解析的に求めることができる。
【００４９】
この場合、非励振ダイポール７ａ、７ｂに装荷されるリアクタンス素子８ａ、８ｂのリアクタンス値Ｘに関して所望ビーム幅を得るための適用範囲は、次式で表すことができる。
【数１１】

ただし、上式において、「γ＜」、「γ＞」は、「γ±」として与えられる値のうちの、それぞれ、小さい方の値、大きい方の値を示している。
このように、電流励振係数比に対する複素平面上に展開した軌跡５２を用いて図形的に解釈することにより、所望ビーム幅を得るリアクタンス値Ｘの最適値を得ることができる。
【００５０】
実施の形態７．
なお、上記実施の形態１〜６では、導体地板１の上方に配置された単一の励振ダイポールアンテナ２と、導体地板１上に立設された一対の非励振モノポールアンテナ３とを備えたアンテナ装置について説明したが、励振ダイポールアンテナ２および非励振モノポールアンテナ３からなるアンテナ素子を複数配列したアレーアンテナにより構成されたアンテナ装置であってもよい。
以下、図１２を参照しながら、前述のアンテナ素子を複数配列したアレーアンテナとしたこの発明の実施の形態７について説明する。
【００５１】
図１２はこの発明の実施の形態７によるアレーアンテナの一例を概略的に示す構成図であり、前述（図１参照）と同様のものについては、前述と同一符号を付して詳述を省略する。
この場合、励振ダイポールアンテナ２の延長方向に沿って、励振ダイポールアンテナ２と一対の非励振モノポールアンテナ３とからなるアンテナ素子６０を単位として、複数のアンテナ素子６０が多数配列されている。
図１２のように、アレー状態で構成されたアンテナ装置においても、前述と同様に、リアクタンス素子４のリアクタンス値を調整することにより、アレー素子パターンのビーム幅を拡大させることができ、広角へのビーム走査時に利得低下を防ぐことができる。
すなわち、励振ダイポールアンテナ２および非励振モノポールアンテナ３を複数配列したダイポールアレーアンテナにおけるアレー素子パターンのビーム幅を広げることができる。
【００５２】
【発明の効果】
以上のように、この発明によれば、導体地板と、導体地板から所定周波数でほぼ１／４波長だけ離れた位置に設けられて、所定周波数で共振する励振ダイポールアンテナと、導体地板上に立設されて、所定周波数でほぼ１／４波長よりも短い長さを有する複数個の非励振モノポールアンテナとを備え、励振ダイポールアンテナは、中央部に給電点を有し且つ導体地板に対してほぼ平行に配置され、各非励振モノポールアンテナは、励振ダイポールアンテナの延長方向に沿って離間配置されるとともに、導体地板との接続側に設けられた給電端子と、給電端子に装荷されたリアクタンス素子とを有し、給電端子とは反対側の端部が励振ダイポールアンテナ側を向くように、導体地板上にほぼ垂直に設けられたので、励振ダイポールアンテナのＥ面素子パターンのビーム幅を広げたアンテナ装置が得られる効果がある。
【図面の簡単な説明】
【図１】この発明の実施の形態１を示す構成図である。
【図２】この発明の実施の形態１を等価的にモデル化した場合を示す構成図である。
【図３】この発明の実施の形態１による各非励振ダイポールからの再放射パターンを示す説明図である。
【図４】図３内の再放射パターンを合成して示す説明図である。
【図５】この発明の実施の形態１による合成パターンを模式的に示す説明図である。
【図６】この発明の実施の形態１による励振ダイポールアンテナのＥ面放射パターンの一例を示す説明図である。
【図７】この発明の実施の形態３の要部を示す構成図である。
【図８】この発明の実施の形態４の要部を示す構成図である。
【図９】この発明の実施の形態５を説明するための等価的にモデル化した構成図である。
【図１０】この発明の実施の形態５によるビーム幅拡大条件を示す説明図である。
【図１１】この発明の実施の形態６によるビーム幅拡大条件を示す説明図である。
【図１２】この発明の実施の形態７によるアレーアンテナの一例を概略的に示す構成図である。
【符号の説明】
１、２０　導体地板、２　励振ダイポールアンテナ、３、２１、３１　非励振モノポールアンテナ、４、８ａ、８ｂ、２２　リアクタンス素子、５　導体地板の仮想位置、６ａ　励振ダイポール、６ｂ　イメージ励振ダイポール、７ａ、７ｂ　非励振ダイポール、９ａ　励振ダイポール上の電流、９ｂ　イメージ励振ダイポール上の電流、１０ａ、１０ｂ　非励振ダイポール上の誘起電流、２３　電子スイッチ（リアクタンス値設定手段）、３２　スタブ、５１　規定領域（第１の円軌跡）、５２　流励振係数比の軌跡（第２の円軌跡）、６０　アンテナ素子、Ａ、Ｂ　線分、Ｃ２、Ｃ６ａ、Ｃ６ｂ　給電点、Ｃ３、Ｃ２１、Ｃ３１　給電端子、Ｃ７ａ、Ｃ７ｂ　給電部、Ｐ_０、ｐ_０　中心点、ｑ１、ｑ２　交点。[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to a structure for adjusting a beam width of an element pattern, particularly an E-plane beam width, and a means for expanding the beam width in an excitation dipole antenna. Also, this means can be applied to enlarge the beam width of an array element pattern in a dipole array antenna having a small element interval.
[0002]
[Prior art]
In general, when performing wide-angle beam scanning in an array antenna, it is necessary to arrange the element antennas with a small distance in order to suppress the generation of grating lobes that cause a decrease in gain. Therefore, the coupling amount between the element antennas may increase, and the radiation pattern (array element pattern) of the element antenna may be disturbed, and the antenna gain at the time of beam scanning may decrease. Therefore, it is necessary to control the beam width to increase the beam width in the array element pattern.
[0003]
In a conventional antenna device (for example, see Non-Patent Document 1), a radiation pattern is obtained based on, for example, a radiation pattern analysis method of a half-wave excitation dipole antenna element alone.
In this analysis method, correspondence between the radiation pattern and the actually measured value is established. This analysis model uses a conductor ground plane (infinity) and an excitation dipole antenna when a single excitation dipole antenna element is arranged above a conductor ground plane at a height of １ ／ wavelength of the resonance frequency. Here, when the excitation dipole antenna element alone operates normally, an E-plane radiation pattern is generated, and its 3 [dB] beam width is about 76 degrees.
[0004]
In an array antenna having a plurality of excitation dipole antennas, beam scanning within an E-plane cut plane is considered.
In order to facilitate the explanation, in consideration of an ideal case where there is no coupling between elements, the array element pattern of this array antenna is the same as the element single radiation pattern. The gain at the time of beam scanning depends on the radiation pattern of the element alone. Therefore, for example, in a wide-angle direction more than 38 degrees from the front of the antenna, the gain is reduced by 3 [dB] or more, and beam scanning in the E-plane wide-angle direction involves a large gain reduction. Therefore, it is required to increase the beam width in the array element pattern.
[0005]
Actually, under the influence of the coupling between the elements, the array element pattern is deformed from the element single radiation pattern, and the beam width also changes. However, expanding the beam width of the array element pattern is considered to be an application of expanding the beam width of the element pattern in the element antenna alone, and it is important to increase the beam width in the radiation pattern of the element alone. It is. Therefore, it is important to increase the beam width of the E-plane element pattern in the excitation dipole antenna alone.
[0006]
As an example of a conventional base station antenna for mobile communication, for example, there is a type in which a bent reflector is combined with an excitation dipole antenna.
In this case, the bent reflector (corner reflector) forms a sector beam by adjusting dimensions such as a bending angle and a reflector length of the corner reflector 71. The excitation dipole antenna is provided with a non-excitation element, and the beam width is adjusted by adjusting the length of the non-excitation element, the interval between the excitation dipole antenna, and the like.
However, in any case, although it is suitable for fine adjustment of the beam width, it is impossible to greatly increase the beam width.
[0007]
[Non-patent document 1]
Roger F. Harrington, "Time-harmonic electromagnetic fields", McGraw-Hill, p. 81-85
[0008]
[Problems to be solved by the invention]
As described above, the conventional antenna device uses an array antenna having an excitation dipole antenna having a 3 [dB] beam width of about 76 degrees in the E-plane radiation pattern as an element antenna, and a wide-angle beam in the E-plane cut plane. In the case of scanning, there is a problem that a gain is reduced.
In addition, when a sector beam is formed by a mobile communication base station antenna, there is a problem that it is necessary to adjust a reflector dimension in order to adjust a beam width.
[0009]
SUMMARY OF THE INVENTION The present invention has been made to solve the above-described problems, and can increase the beam width of an E-plane element pattern (or an array element pattern in a dipole array antenna) of a single element of an excitation dipole antenna. It is intended to obtain an antenna device.
[0010]
[Means for Solving the Problems]
An antenna device according to the present invention is provided with a conductive ground plane, an excitation dipole antenna that is provided at a position separated from the conductive ground plane by approximately 1/4 wavelength at a predetermined frequency, and resonates at a predetermined frequency, and is erected on the conductive ground plane. A plurality of non-excited monopole antennas having a length shorter than substantially 1/4 wavelength at a predetermined frequency, wherein the excited dipole antenna has a feed point at the center and is substantially parallel to the conductive ground plane. The non-excited monopole antennas are arranged so as to be spaced apart from each other along the extension direction of the excited dipole antenna, and a feed terminal provided on the connection side with the conductor ground plane and a reactance element loaded on the feed terminal. The power supply terminal is provided substantially vertically on the conductor ground plane such that the end opposite to the power supply terminal faces the excitation dipole antenna side.
[0011]
BEST MODE FOR CARRYING OUT THE INVENTION
Embodiment 1 FIG.
Hereinafter, Embodiment 1 of the present invention will be described in detail with reference to the drawings.
FIG. 1 is a configuration diagram schematically showing Embodiment 1 of the present invention.
First, the expansion of the E-plane beam width in the excitation dipole antenna alone will be described. Here, in order to simplify the description, an example in which the number of non-excitation monopole antennas installed below the excitation dipole antenna is two will be described.
In FIG. 1, an excitation dipole antenna 2 is disposed above a conductor ground plane 1, and a plurality (two) of non-excitation monopole antennas 3 are erected on the conductor ground plane 1.
[0012]
The excitation dipole antenna 2 has a predetermined frequency f ₁ At a predetermined frequency f ₁ At a position separated from the conductive ground plane 1 by approximately 1/4 wavelength. The excitation dipole antenna 2 has a feed point C2 in the center and is arranged substantially parallel to the conductor ground plane 1.
Each non-excitation monopole antenna 3 has a predetermined frequency f ₁ And has a length shorter than approximately ほ ぼ wavelength, and is spaced apart along the extension direction of the excitation dipole antenna 2. The non-excitation monopole antenna 3 has a power supply terminal C3 provided on the connection side with the conductor ground plane 1 and a load impedance, that is, a reactance element 4 loaded on each power supply terminal C3. It is provided substantially vertically on the conductor ground plane 1 such that the opposite end faces the excitation dipole antenna 2 side.
Here, in order to simplify the description, a case is shown in which the non-excited monopole antenna is disposed at a symmetrical position with respect to the feeding point C2 of the excited dipole antenna 2.
[0013]
Next, the operation according to the first embodiment of the present invention shown in FIG. 1 will be described with reference to FIGS.
FIG. 2 is a configuration diagram showing a case where the first embodiment of the present invention is modeled. The area of the conductive ground plane 1 in FIG. 1 is regarded as infinite, and the reflected wave from the conductive ground plane 1 is a virtual image. This figure shows a state modeled using the concept replaced with an excitation dipole.
[0014]
2, an excitation dipole 6a corresponding to the excitation dipole antenna 2 is disposed above the virtual position 5 of the conductor ground plane 1 (see FIG. 1). In addition, below the virtual position 5, an image excitation dipole (hereinafter, also simply referred to as "excitation dipole") 6b, which is an image element similar to the excitation dipole 6a, is regarded as equivalently arranged. The image excitation dipole 6b has a line-symmetric relationship with the excitation dipole 6a with respect to a virtual position 5 indicated by a broken line in place of the conductor ground plane 1.
[0015]
First, referring to FIG. 2, a mechanism of generating induced currents 10a and 10b in the non-excited dipoles 7a and 7b induced by the currents 9a and 9b flowing through the feeding points C6a and C6b of the excited dipoles 6a and 6b will be described. .
At the virtual position 5, a first non-excitation dipole 7a and a second non-excitation dipole 7b corresponding to the non-excitation monopole antenna 3 (see FIG. 1) when an image is considered are provided.
[0016]
Each of the non-excited dipoles 7a and 7b is set to have a dipole length that is twice as long as the above-described non-excited monopole antenna 3, and the reactance elements 8a and 7 8b has a symmetrical shape.
The currents 9a and 9b at the feeding points C6a and C6b of the excitation dipoles 6a and 6b flow in the directions indicated by the arrows. Further, the induced currents 10a and 10b in the feeding portions C7a and C7b of the non-excitation dipoles 7a and 7b flow in the directions of the arrows.
[0017]
First, when the current 9a at the feeding point C6a flows in the direction of the arrow when the excitation dipole 6a in FIG. 2 operates as a normal dipole, the current 9b flowing through the feeding point C6b of the image excitation dipole 6b becomes the current 9a. The direction is opposite to the direction.
The radiation pattern of the excitation dipole antenna 2 with the conductor ground plane 1 described above (see FIG. 1) can be expressed as a radiation pattern of a two-element array composed of the excitation dipoles 6a and 6b in FIG.
The currents 10a and 10b induced by the coupled waves from the excitation dipoles 6a and 6b on the non-excitation dipoles 7a and 7b flow in the directions of the arrows. The directions of the induced currents 10a and 10b have opposite phases.
[0018]
The radiation from the induced currents 10a and 10b (hereinafter referred to as "re-radiation") affects the E-plane radiation pattern (polarization of the θ-direction component in the plane on the paper) of the excitation dipole 9a.
Next, the mechanism of the induced currents 10a and 10b affecting the E-plane radiation pattern will be described with reference to FIGS.
FIG. 3 is an explanatory diagram showing re-radiation patterns 12a and 12b from the respective non-excited dipoles 7a and 7b.
In FIG. 3, the directions of the currents 11a and 11b contributing to the re-radiation of the respective non-excited dipoles 7a and 7b are opposite in phase to the directions of the induced currents 10a and 10b described above (see FIG. 2).
When considering the re-emission from the induced currents 10a and 10b (see FIG. 2) on the non-excited dipoles 7a and 7b, the phases of the induced currents 10a and 10b can be considered as opposite phases. , 11b are shown as in FIG.
[0019]
Since these currents 11a and 11b are in an anti-phase relationship with each other, the re-radiation patterns 12a and 12b from the respective non-excited dipoles 7a and 7b are respectively shown in FIG.
The phases of the re-radiation patterns 12a and 12b are opposite to each other with respect to the non-excited dipoles 7a and 7b.
Therefore, the non-excited dipole two-element array radiation pattern in the case where two elements of the non-excited dipoles 7a and 7b are used as an array antenna is synthesized as shown in a pattern 13 in FIG. Become.
The radiation pattern 13 in FIG. 4 corresponds to a combination of the re-radiation patterns 12a and 12b in FIG.
[0020]
FIG. 5 is an explanatory diagram schematically showing a combined pattern, and shows a pattern in which radiation from the excitation dipoles 6a and 6b and re-radiation from the non-excitation dipoles 7a and 7b are combined.
In FIG. 5, a pattern 13 is a re-radiation pattern (radiation pattern of a non-excited dipole two-element array) described above (see FIG. 4).
The pattern 14 is a radiation pattern when the two elements of the excitation dipoles 6a and 6b are used as an array antenna (two-element array radiation pattern of the excitation dipole), and corresponds to a combination of the radiation patterns of the excitation dipoles 6a and 6b. I do.
[0021]
Therefore, the final combined radiation pattern obtained from the antenna configuration described above (see FIG. 1) can be regarded as the radiation pattern of the excited dipoles 6a and 6b in consideration of re-radiation from the non-excited dipoles 7a and 7b. 5 can be represented by a combination of the non-excited dipole two-element array radiation pattern 13 and the excited dipole two-element array radiation pattern 14.
At this time, the relative phases of the radiation patterns 13 and 14 can be adjusted by variably setting the reactance values of the reactance element 4 (see FIG. 1) or the reactance elements 8a and 8b (see FIG. 2).
[0022]
The radiation pattern 13 by the non-excited dipoles 7a and 7b has an amplitude peak in the direction of θ = 0 degrees, and has an amplitude “0” in the direction of θ = 90 degrees.
On the other hand, the radiation pattern 14 by the excitation dipoles 6a and 6b has an amplitude peak in the direction of θ = 90 degrees, and has an amplitude “0” in the direction of θ = 0 degrees.
Therefore, as shown in FIG. 5, for example, when the relative phases of the two patterns 13 and 14 are the same, the patterns 13 and 14 are synthesized so as to complement the directions of the amplitude “0” of each other, so that the final pattern is finally obtained. The beam width of the combined pattern (pattern 13 + pattern 14) is increased.
[0023]
Here, how to give the reactance values of the reactance elements 4, 8a, 8b will be described.
First, the length of the non-excitation monopole antenna 3 (or the non-excitation dipoles 7a and 7b) is set to be short in consideration of the antenna structure, cost, and the like.
For example, in FIG. 1 (FIG. 2), the excitation dipole antenna 2 (excitation dipoles 6a and 6b) is moved from the conductor ground plane 1 (virtual position 5) to a predetermined frequency f. ₁ Are located at positions separated by approximately 1/4 wavelength.
Here, the length of the non-excitation monopole antennas (non-excitation dipoles 7a and 7b) provided below the excitation dipole antenna 2 (excitation dipole 6a) is equal to the predetermined frequency f. ₁ Is shorter than the 波長 wavelength (that is, shorter than the resonance length), so that the induced currents 10a and 10b (see FIG. 2) due to the coupling hardly occur in this state.
[0024]
Therefore, the reactance elements 4 (8a, 8b) are provided to compensate for the lack of resonance length of the non-excitation monopole antenna 3 (non-excitation dipoles 7a, 7b). Therefore, the reactance elements 4 (8a, 8b) have an inductance component, and the reactance value is a value resulting from the length of the non-excited monopole antenna 3.
For example, assuming that the length of the non-excitation monopole antenna 3 (1/2 of the non-excitation dipoles 7a and 7b) is shorter than the resonance length and that the reactance elements 4 (8a and 8b) are not loaded, The amount of re-radiation from the monopole antenna 3 (the non-excited dipoles 7a and 7b) is small, and it is impossible to expect an increase in the beam width as described above.
[0025]
FIG. 6 is an explanatory diagram showing an example of the E-plane radiation pattern of the excitation dipole antenna 2 (the excitation dipole 6a).
In FIG. 6, a solid line indicates a relative output characteristic when the reactance element 4 (8a, 8b) having an appropriate reactance value is loaded on the non-excitation monopole antenna 3 (non-excitation dipoles 7a, 7b), and a broken line indicates the reactance. The relative output characteristics when the element 4 (8a, 8b) is not provided are shown.
As is clear from FIG. 6, it can be confirmed that the beam width is expanded by the loading of the reactance elements 4 (8a, 8b) on the non-excitation monopole antenna 3 (non-excitation dipoles 7a, 7b).
[0026]
As shown in FIG. 1, according to the first embodiment of the present invention, the predetermined frequency f ₁ A plurality of non-excited monopole antennas 3 having a length less than the resonance length (1/4 wavelength) are arranged substantially perpendicular to the conductive ground plane 1 below the excited dipole antenna 2 operated by By loading a reactance element 4 having an optimum reactance value on the feeding terminal C3 of the antenna 3, the influence of the re-radiation from the induced currents 10a and 10b distributed on the non-excited monopole antenna 3 (see FIG. 2). Accordingly, the beam width of the E-plane element pattern of the element of the excitation dipole antenna 2 alone can be increased.
The reactance element 4 can be realized by a lumped constant element or the like.
[0027]
Further, in FIG. 1, the non-excited monopole antenna 3 is regularly arranged in a symmetrical position with respect to the excited dipole antenna 2, but this arrangement structure increases the beam width while maintaining the symmetry of the radiation pattern. This is because it is intended to do so.
However, achieving this object is not an essential requirement. For example, when it is necessary to change the radiation pattern shape according to the application, the arrangement position of the non-excited monopole antenna 3 is variable as a parameter during radiation pattern control. Can be set.
At this time, the interval between the non-excited monopole antennas 3 can also be a parameter, but if focusing on increasing the amplitude level in the wide angle direction of the angle θ in the composite pattern, the interval between the non-excited monopole antennas 3 is For example, in FIG. ₁ Is set to approximately 波長 wavelength.
Further, the length of the non-excited monopole antenna 3 is also a parameter. For example, the shorter the length of the non-excited monopole antenna 3, the larger the reactance value.
[0028]
Embodiment 2 FIG.
In the first embodiment, the reactance value of the reactance element 4 is set to the same value for all the non-excitation monopole antennas 3. However, the reactance value of each reactance element 4 is set to the same value for each non-excitation monopole antenna 3. May be variably set to different values for each element.
Hereinafter, a second embodiment of the present invention in which the reactance value is set to be different for each element of the non-excitation monopole antenna 3 will be described with reference to FIG.
[0029]
In this case, by changing the value of the reactance element 4 for each element of the non-excitation monopole antenna 3, the shape of the re-radiation pattern from the non-excitation monopole antenna 3 and the excitation phase can be arbitrarily adjusted. 2, a pattern shape having high gain in any direction can be obtained.
Conversely, an amplitude "0 (null)" point can be formed in a certain arbitrary direction.
[0030]
Embodiment 3 FIG.
In the first and second embodiments, the means for variably setting the reactance value of the reactance element 4 has not been particularly described. However, a reactance value setting means for electronically switching the reactance value may be provided. .
Hereinafter, a third embodiment of the present invention provided with electronic reactance value setting means will be described with reference to FIG.
FIG. 7 is a configuration diagram showing a main part of the third embodiment of the present invention, in which only a single non-excitation monopole antenna is shown to avoid complexity, and the excitation dipole antenna 2 is not shown. Is omitted.
[0031]
In FIG. 7, a non-exciting monopole antenna 21 is provided upright on a conductive ground plane 20, and a reactance element 22 whose reactance value is variably set is loaded on a feed terminal C 21 of the non-exciting monopole antenna 21. An electronic switch 23 functioning as a reactance value setting unit is connected to the reactance element 22.
The electronically controllable electronic switch 23 has a reactance value (jX ₁ , JX ₂ , JX ₃ ) Includes an arrangement of a plurality of elements different from each other, and one of the plurality of elements is connected to the reactance element 22.
[0032]
Next, the operation of the electronic switch 23 according to the third embodiment of the present invention shown in FIG. 7 will be described. Note that the operation of expanding the beam width of the E-plane radiation pattern of the excitation dipole antenna 2 (see FIG. 1) is the same as that of the first embodiment, and a description thereof will be omitted.
In this case, the reactance value of the reactance element 22 loaded on the non-excitation monopole antenna 21 can be electronically variably set by the electronic switch 23.
[0033]
If a reactance element having a constant reactance value is used, the frequency band that satisfies the effect of expanding the beam width is limited. However, by using the electronic switch 23 as shown in FIG. Accordingly, the reactance value can be variably set, and the effect of expanding the beam width over a wide band can be obtained.
Also, by adjusting the reactance value, the re-radiation pattern from the non-excited monopole antenna 21 can be adjusted, so that the beam width can be further increased and the radiation pattern shape can be changed. That is, a desired radiation pattern shape can be obtained by variably setting the reactance value by the electronic switch 23. The reactance element 22 can be realized by using a lumped constant element or the like.
[0034]
Embodiment 4 FIG.
In the third embodiment, the reactance element 21 is realized by a lumped constant element or the like. However, the reactance element may be realized by using a stub.
Hereinafter, a fourth embodiment of the present invention using a stub as a reactance element will be described with reference to the configuration diagram of FIG.
8, the same components as those described above (see FIG. 1) are denoted by the same reference numerals as those described above, and the detailed description will be omitted. FIG. 8A is a configuration diagram showing a main part according to Embodiment 4 of the present invention. 8B, similarly to the above (see FIG. 2), the non-excited monopole antenna 31 is equivalently shown as a non-excited dipole 33 by image theory when the conductive ground plane 1 is regarded as an infinite ground plane. FIG.
[0035]
In FIG. 8A, the reactance element loaded on the feeding terminal C31 of the non-excitation monopole antenna 31 is configured by a stub 32 in which the non-excitation monopole antenna 31 meanders near the conductive ground plane 1.
In FIG. 8B, the excitation dipole 33 has an equivalent relation to the non-excitation monopole antenna 31, and it is assumed that the excitation dipole 33 is equivalently loaded with the meandering stub 32. Can be.
The stub 32 has a function as an inductance and operates as a reactance element.
[0036]
Next, an operation according to the fourth embodiment of the present invention shown in FIG. 8 will be described. The operation for expanding the beam width is the same as that described above, and the description is omitted here.
In this case, since the stub 32 is provided as a reactance element, the operation of the stub 32 will be described. That is, by adjusting the length of the stub 32, the reactance value can be changed. Therefore, the length of the stub 32 is a parameter for expanding the beam width, and has the same operation and effect as described above.
[0037]
Embodiment 5 FIG.
In the fourth embodiment, the condition for realizing the beam width expansion with respect to the ratio of the current excitation coefficient between the excited dipole antenna 2 and the non-excited monopole antenna 3 is not particularly described. May be set.
Hereinafter, a fifth embodiment of the present invention in which the beam width expansion condition is set will be described with reference to FIGS. 9 and 10 together with FIG.
[0038]
FIG. 9 is a configuration diagram for explaining the fifth embodiment of the present invention, and shows a case where equivalent modeling is performed using image theory, as in the above (see FIG. 2).
Here, the current excitation coefficients I of the excitation dipoles 6a and 6b (excitation dipole antennas) and the non-excitation dipoles 7a and 7b (non-excitation monopole antennas) ₁ ~ I ₄ The condition for expanding the beam width with respect to the ratio of will be described.
In addition, as shown in FIG. 1, a model in which two non-excitation monopole antennas 3 loaded with reactance elements 4 are arranged at symmetric positions below the excitation dipole antenna 2 is handled.
[0039]
In FIG. 9, the elements of the non-excitation dipoles 7a and 7b and the excitation dipoles 6a and 6b are numbered by # 1 to # 4.
That is, element numbers # 1 and # 3 correspond to the non-excitation dipoles 7a and 7b loaded with the reactance elements 8a and 8b, and element numbers # 2 and # 4 correspond to the excitation dipoles 6a and 6b.
The excitation coefficient of the induced current 10a in the non-excited dipole 7a (# 1) is represented by I ₁ , The excitation coefficient of the current 9a in the excitation dipole 6a (# 2) is represented by I ₂ , The current excitation coefficient ratio I to the desired beam width ₁ / I ₂ Is represented by the following equation (1).
[0040]
(Equation 6)

[0041]
However, in equation (1), D ₁ Are the re-radiation patterns (as a two-element array) from the parasitic dipoles 7a (# 1) and 7b (# 3), D ₂ Indicates radiation patterns (also as a two-element array) from the excitation dipoles 6a (# 2) and 6b (# 4).
In the equation (1), c is a level of amplitude reduction from the front which defines the E-plane beam width.
[0042]
Current excitation coefficient ratio I ₁ / I ₂ Is expanded on a complex plane, from the above equation (1), the region defining the E-plane beam width indicates a circle, and the current excitation coefficient ratio satisfying the desired E-plane beam width is given as the outer region of the circle. Can be
This example is shown in the explanatory diagram of FIG.
In FIG. 10, a circle 41 defining “beam width 1” is indicated by a broken line, a circle 42 defining “beam width 2” is indicated by a dashed line, and a circle 43 defining “beam width 3” is a solid line. Indicated by
The locus 44 of the current excitation coefficient ratio when the reactance value X of the reactance elements 8a and 8b is changed is indicated by a dotted line.
[0043]
Here, the magnitude relation of “each beam width 1, 2, 3” satisfies “beam width 1” <“beam width 2” <“beam width 3”, and the larger the beam width, the smaller the circle. However, the center point moves on the real axis (horizontal axis).
In the case of FIG. 10, the range of the locus 44 of the current excitation coefficient ratio existing in the outer region of the circle 43 is extremely smaller than the range of the locus 44 existing in the outer region of the circle 41 having a narrow beam width.
This means that the larger the beam width, the smaller the current excitation coefficient ratio that satisfies this condition.
[0044]
It should be noted here that when calculating the current excitation coefficient ratio that satisfies the condition represented by the above formula (1) for a certain desired beam width, the condition is always satisfied for a beam width smaller than the desired beam width. It is necessary to be.
That is, as is clear from FIG. 10, the range of the current excitation coefficient ratio that satisfies “beam width 3” always holds for “beam width 1” and “beam width 2”.
From the above, the equation (1) representing the condition is expressed by the current excitation coefficient ratio I ₁ / I ₂ , The current excitation coefficient ratio to be obtained can be expressed outside the circular region defining the desired beam width, which can be easily interpreted graphically as shown in FIG. it can.
[0045]
Embodiment 6 FIG.
In the fifth embodiment, the condition of the reactance value X for realizing the beam width expansion by applying the graphical interpretation on the complex plane with respect to the current excitation coefficient ratio is not described in detail. In addition, the condition of the reactance value X for realizing the beam width expansion may be set.
Hereinafter, the condition of the reactance value X according to the sixth embodiment of the present invention will be described with reference to FIG. Here, the condition of the reactance value X for realizing the beam width expansion is set by applying the above-described graphical interpretation on the complex plane with respect to the current excitation coefficient ratio.
[0046]
FIG. 11 is an explanatory diagram showing a beam width expanding condition according to the sixth embodiment of the present invention, and shows a complex plane with respect to a current excitation coefficient ratio.
In FIG. 11, a radius R indicated by a broken line ₀ Circle 51 is an area defining the beam width, and has a radius r indicated by a solid line. ₀ Is a locus drawn by the current excitation coefficient ratio when the reactance value X is changed. It is mathematically proved that the trajectory 52 of the flow excitation coefficient ratio draws a circle as shown in FIG.
Here, as shown in FIG. 11, a circle 51 indicating the defined area and a locus 52 of the current excitation coefficient ratio intersect at q ₁ , Q ₂ When the locus 52 exists in the area outside the circle 51, the current excitation coefficient ratio that satisfies the desired beam width exists. That is, in this case, the reactance value X of the reactance element 4 (see FIG. 1) can be determined.
[0047]
In addition, here, as shown in FIG. 9, the antenna device shown in FIG. 1 is equivalently modeled, and the above-described conditions are described by assigning numbers # 1 to # 4 to the respective elements.
As described above, it is mathematically proved that the current excitation coefficient ratio of the non-exciting dipoles 7a and 7b to the exciting dipoles 6a and 6b draws a circular locus on a complex plane with respect to the change of the reactance elements 8a and 8b. (The description is omitted here).
In addition, the area defining the E-plane beam width of the excitation dipole is also circular from the above-described equation (1) (see circle 51 in FIG. 11).
The condition that the current excitation coefficient ratio that satisfies the desired beam width exists is that the two circles have an intersection, and the locus of the current excitation coefficient ratio is located outside the E-plane beam width defining region circle.
[0048]
First, in FIG. 11, the center of the circle 51 in the specified area is defined as P ₀ , The center of the locus 52 of the current excitation coefficient ratio is p ₀ And the intersection of the two circles 51 and 52 is q ₁ , Q ₂ And Also, the center point P of the specified area ₀ And the center point p of the trajectory ₀ Is the line segment A, and the center point p of the locus ₀ And one intersection q ₁ Is a line segment B.
Next, the angle between the line segment A and the real axis (horizontal axis) is
(Equation 7)

And the angle between line segment A and line segment B is
(Equation 8)

And
The mutual impedance Z between the non-excited dipole 7a (non-excited monopole antenna: # 1) and the excited dipole 6a (# 2) in FIG. ₁₂ Using the phase ξ
(Equation 9)

Displayed in polar coordinates, as in
Similarly, the self-impedance Z of the non-excited dipole (# 1) ₁₁ And the mutual impedance Z between the non-excited dipole 7a (# 1) and the non-excited dipole 7b (# 3) _Thirteen Is calculated using the phase η.
(Equation 10)

Displayed in polar coordinates, as in
Note that the self impedance Z ₁₁ , Mutual impedance Z ₁₂ , Z _Thirteen Can be analytically obtained by a well-known electromotive force method or the like.
[0049]
In this case, the applicable range for obtaining a desired beam width with respect to the reactance value X of the reactance elements 8a and 8b loaded on the non-excitation dipoles 7a and 7b can be expressed by the following equation.
[Equation 11]

However, in the above equation, “γ <” and “γ>” indicate a smaller value and a larger value of the values given as “γ ±”, respectively.
As described above, by performing the graphic interpretation using the trajectory 52 developed on the complex plane with respect to the current excitation coefficient ratio, the optimum value of the reactance value X for obtaining the desired beam width can be obtained.
[0050]
Embodiment 7 FIG.
In the first to sixth embodiments, the single excitation dipole antenna 2 disposed above the conductor ground plane 1 and the pair of non-excitation monopole antennas 3 erected on the conductor ground plane 1 are provided. Although the antenna device has been described, the antenna device may be configured by an array antenna in which a plurality of antenna elements including the excited dipole antenna 2 and the non-excited monopole antenna 3 are arranged.
Hereinafter, a seventh embodiment of the present invention, which is an array antenna in which a plurality of the above-described antenna elements are arranged, will be described with reference to FIG.
[0051]
FIG. 12 is a configuration diagram schematically showing an example of an array antenna according to a seventh embodiment of the present invention. Components similar to those described above (see FIG. 1) are denoted by the same reference numerals as those described above, and detailed description thereof will be omitted. I do.
In this case, a plurality of antenna elements 60 are arranged along the extension direction of the excitation dipole antenna 2 with the antenna element 60 including the excitation dipole antenna 2 and the pair of non-excitation monopole antennas 3 as a unit.
As shown in FIG. 12, in the antenna device configured in the array state as well, the beam width of the array element pattern can be expanded by adjusting the reactance value of the reactance element 4 in the same manner as described above. It is possible to prevent a decrease in gain during beam scanning.
That is, the beam width of the array element pattern in a dipole array antenna in which a plurality of the excited dipole antennas 2 and the non-excited monopole antennas 3 are arranged can be increased.
[0052]
【The invention's effect】
As described above, according to the present invention, the conductive ground plane, the excitation dipole antenna that is provided at a position separated from the conductive ground plane by approximately 1/4 wavelength at a predetermined frequency and resonates at the predetermined frequency, and standing on the conductive ground plane. A plurality of non-excited monopole antennas having a length shorter than substantially 1/4 wavelength at a predetermined frequency, wherein the excited dipole antenna has a feed point in the center and The non-excited monopole antennas are arranged substantially parallel to each other, and are spaced apart from each other along the extension direction of the excited dipole antenna.A feed terminal provided on the connection side with the conductive ground plane and a reactance loaded on the feed terminal Element, and is provided almost vertically on the conductor ground plane such that the end opposite to the feed terminal faces the excitation dipole antenna side. The effect of the antenna device widening the beam width of the element pattern.
[Brief description of the drawings]
FIG. 1 is a configuration diagram showing a first embodiment of the present invention.
FIG. 2 is a configuration diagram showing a case where the first embodiment of the present invention is equivalently modeled.
FIG. 3 is an explanatory diagram showing a re-radiation pattern from each passive dipole according to the first embodiment of the present invention.
FIG. 4 is an explanatory diagram showing a combination of re-radiation patterns in FIG. 3;
FIG. 5 is an explanatory diagram schematically showing a combined pattern according to the first embodiment of the present invention.
FIG. 6 is an explanatory diagram showing an example of an E-plane radiation pattern of the excitation dipole antenna according to the first embodiment of the present invention.
FIG. 7 is a configuration diagram showing a main part of a third embodiment of the present invention.
FIG. 8 is a configuration diagram showing a main part of a fourth embodiment of the present invention.
FIG. 9 is an equivalently modeled configuration diagram for explaining Embodiment 5 of the present invention.
FIG. 10 is an explanatory diagram showing a beam width expansion condition according to a fifth embodiment of the present invention.
FIG. 11 is an explanatory diagram showing a beam width expanding condition according to a sixth embodiment of the present invention.
FIG. 12 is a configuration diagram schematically showing an example of an array antenna according to a seventh embodiment of the present invention.
[Explanation of symbols]
1, 20 conductor ground plane, 2 excitation dipole antenna, 3, 21, 31 non-excitation monopole antenna, 4, 8a, 8b, 22 reactance element, 5 virtual position of conductor ground plane, 6a excitation dipole, 6b image excitation dipole, 7a, 7b Non-excited dipole, 9a Current on excited dipole, 9b Current on image-excited dipole, 10a, 10b Induced current on non-excited dipole, 23 electronic switch (reactance value setting means), 32 stub, 51 defined area (first area) ), 52 trajectory of the excitation coefficient ratio (second circular trajectory), 60 antenna elements, A, B line segments, C2, C6a, C6b feeding points, C3, C21, C31 feeding terminals, C7a, C7b feeding Part, P ₀ , P ₀ Center point, q1, q2 intersection.

Claims (7)

A conductive ground plane,
An excitation dipole antenna that is provided at a position separated from the conductive ground plane by approximately a quarter wavelength at a predetermined frequency and resonates at the predetermined frequency;
A plurality of non-excited monopole antennas that are erected on the conductive ground plane and have a length shorter than substantially １ ／ wavelength at the predetermined frequency;
The excitation dipole antenna has a feed point in the center and is disposed substantially parallel to the conductive ground plane,
Each of the non-excited monopole antennas,
While being spaced apart along the extension direction of the excitation dipole antenna,
A power supply terminal provided on the connection side with the conductive ground plane,
A reactance element loaded on the power supply terminal,
An antenna device, wherein the antenna device is provided substantially vertically on the conductive ground plane such that an end opposite to the power supply terminal faces the excitation dipole antenna. The antenna device according to claim 1, wherein a reactance value of the reactance element is set to be different for each of the non-excited monopole antennas to be loaded. 3. The antenna device according to claim 1, further comprising a reactance value setting unit for electronically variably setting a reactance value of the reactance element. 2. The antenna device according to claim 1, wherein the reactance element is configured by a stub in which the non-excitation monopole antenna meanders near the conductive ground plane. 3. The non-excitation monopole antenna includes first and second non-excitation monopole antennas disposed at symmetrical positions within an E-plane cut plane with respect to a feed point of the excitation dipole antenna,
When the current excitation coefficient of the first non-excitation monopole antenna is I ₁ and the current excitation coefficient of the excitation dipole antenna is I ₂ , the current excitation coefficient ratio I ₁ / I ₂ displayed on the complex plane is The antenna apparatus according to claim 1, wherein the antenna apparatus is set outside a region where a first circular locus is drawn on the complex plane to define an E-plane beam width. The current excitation coefficient ratio I ₁ / I ₂ draws a second circular locus on the complex plane with respect to the change in the reactance value,
When the second circular locus has an intersection with the first circular locus, an angle between a line segment A connecting the center points of the first and second circular locus and a real axis of the complex plane is defined as

age,
An angle formed by a line segment B connecting the center of the second circular locus and the intersection of the first and second circular locuses with the line segment A

age,
And, the mutual impedance Z ₁₂ between the first parasitic monopole antenna and the excitation dipole antenna, with a phase xi],

Expressed in polar coordinates as
Wherein a first of the parasitic monopole antenna self impedances Z _11, a difference between the mutual impedance Z ₁₃ between the first and second parasitic monopole antenna, with a phase eta,

When expressed in polar coordinates like
The applicable range for obtaining a desired beam width with respect to the reactance value X of the reactance element is expressed by the following equation.

The antenna device according to claim 5, wherein the antenna device is provided so as to satisfy the following. The antenna device according to any one of claims 1 to 6, comprising an array antenna in which a plurality of antenna elements including the excitation dipole antenna and the non-excitation monopole antenna are arranged.

Images