JP2002051555A - Switching power circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To make constant high direct-current voltage outputted from a flyback transformer. SOLUTION: Negative resonance voltage V2 outputted from the secondary side of an insulating converter transformer PIT constituting a compositely resonant switching power circuit is inputted to the primary side of a flyback transformer FBT through a series resonance capacitor C3. Further, the inductance LR of an orthogonal control transformer PRT-2 whose controlled winding NR is series-connected with the primary winding N4 of the flyback transformer FBT is varied and controlled through a control circuit 2, so that the high direct- current voltage EHV outputted from a high voltage generating circuit 4 can be made constant.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit suitable for use in, for example, a large-sized color television receiver having a high resolution, a projector device, and the like.

[0002]

2. Description of the Related Art Cathode ray tubes (CRTs)
For example, among the video equipment provided with a high-definition television receiver, a high-definition television receiver, a high-definition television broadcast (HDTV;
Definition Television (hereinafter referred to as “HD television”), projector devices, and the like, have a horizontal synchronization frequency twice as high as that of a normal television receiver (in order to achieve high resolution). 3
1.5 KHz). For example, a high voltage supplied to an anode electrode of a cathode ray tube (hereinafter, referred to as “CRT”) is set to 30 KV or more, and a beam current is set to 2 mA or more.

In the above-described high-resolution television receiver, when a flyback pulse voltage generated during a horizontal retrace period is used to generate a high voltage to be supplied to an anode electrode of a CRT, a flyback is generated. Since the pulse width of the pulse voltage is narrower than that of a normal television receiver, for example, when the beam current of the CRT becomes zero,
A problem occurs in that the high pressure supplied to the anode of the CRT fluctuates.

Therefore, as a power supply circuit of a television receiver as described above, a high voltage regulator circuit is inserted before the flyback transformer, and the input voltage level input to the primary side of the flyback transformer is changed. In some cases, the voltage is variably controlled in accordance with the high-voltage level output from the secondary side to make the high-voltage level constant.

FIG. 11 is a diagram showing a configuration of a conventional power supply circuit provided in a high-resolution television receiver or the like as described above. In FIG. 11, a switching power supply 10 performs switching of an input DC voltage, converts the input DC voltage to a predetermined DC voltage level, and outputs the DC voltage.
Although it is a DC converter, it is constituted by, for example, a switching power supply circuit of a current resonance type (not shown).

In the preceding stage of the switching power supply 10, a bridge rectifier circuit Di of full-wave rectification and a smoothing capacitor Ci are provided.
The DC voltage Ei is obtained by rectifying and smoothing a commercial AC power supply (AC input voltage VAC) by the rectification / smoothing circuit, and the DC voltage Ei is input to the switching power supply 10. . As a result, the switching power supply 10 outputs at least the DC output voltage EO1 at a predetermined voltage level. This DC output voltage EO1 is a drive voltage for driving the horizontal deflection circuit of the television receiver.

On the secondary side of the switching power supply 10, a smoothing capacitor CO1 and a high-voltage regulator circuit 20 indicated by a broken line are provided. The smoothing capacitor CO1 is
The output of the switching power supply 10 is smoothed and the DC output voltage E
Get O1. The high voltage regulator circuit 20 includes a PWM (Puls
e Width Modulation) This is a step-down type chopper circuit of a control system, and is composed of, for example, a switching element Q11 composed of a MOS-FET, a flywheel diode D11, a choke coil CH, and a smoothing capacitor CO11. In this case, the gate of the switching element Q11 is connected to the drive circuit 14 of the switching drive unit 30, which will be described later, and its drain is connected to the secondary output terminal (DC output voltage EO1) of the switching power supply 10. The source of the high voltage generating circuit 40 is connected via a choke coil CH.
Is connected to the winding start end of the primary winding N11 of the flyback transformer FBT provided in the first embodiment. The flywheel diode D11 is connected between the source of the switching element Q11 and the secondary side ground in the direction shown in the figure, and the smoothing capacitor CO11 is connected to the connection line between the choke coil CH and the primary winding N11 of the flyback transformer FBT. Connected. The high-voltage regulator circuit 20 having such a configuration steps down the input DC output voltage EO1 to a predetermined level DC output voltage EO11 according to the switching operation of the switching element Q11 and outputs it.

[0008] The flyback transformer FBT boosts the input on the primary side and transmits it to the secondary side. The primary winding N11 provided on the primary side of the flyback transformer FBT is
The winding start end is connected to the positive electrode of the smoothing capacitor CO11 of the high-voltage regulator circuit 20, and the winding end is connected to the collector of the switching element Q12. The switching element Q12 is, for example, an IGBT (Insulated Gate B
The base is connected to the drive circuit 15 of the switching drive unit 30. A damper diode D12 and a parallel resonance capacitor Cr11 are connected in parallel between the collector and the emitter.

That is, in the power supply circuit shown in FIG. 11, a primary winding N11 of a flyback transformer FBT, a switching element Q12, a damper diode D12, A voltage resonance type converter composed of a parallel resonance capacitor Cr11 is connected.

[0010] Switching drive unit 3 shown by a broken line
0 of the horizontal synchronization signal fH (3
1.5KHz). The oscillation circuit 12 performs an oscillation operation based on the synchronization signal from the synchronization circuit 11, and outputs an oscillation signal generated by the oscillation operation to the PW
It outputs to the M circuit 13 and the drive circuit 15. The PWM circuit 13 outputs a pulse width modulated signal to the drive circuit 14 based on an oscillation signal from the oscillation circuit 12 and a control signal from the control circuit 1 described later. Drive circuit 14 converts the modulated signal from PWM circuit 13 into a drive voltage and outputs the drive voltage to the gate of switching element Q11. The drive circuit 15 receives an oscillation signal from the oscillation circuit 12, and outputs a drive current obtained by converting the oscillation signal to the switching element Q12.

Therefore, the switching element Q11 has a PWM
A switching operation is performed based on the modulation signal generated by the circuit 13, and the switching element Q12
Is performing the switching operation based on the oscillation signal generated in.

On the secondary side of the flyback transformer FBT, five sets of boost windings NVH11, NVH12, NVH13, NHV14,
NHV15 is divided and wound. In this case, the winding (winding direction) of each of the step-up windings NHV11 to NHV15 with respect to the primary winding N11 is wound so as to have the opposite polarity. The anodes of the high-voltage rectifier diodes DHV11, DHV12, DHV13, DHV14, and DHV15 are connected to the winding ends of the boosting windings NHV11 to NHV15. And the high voltage rectifier diode DHV11
Are connected to the positive terminal of the smoothing capacitor COHV, and the cathodes of the high-voltage rectifier diodes DHV12 to DHV15 are connected to the winding start ends of the step-up windings NHV11 to NHV14, respectively. That is, on the secondary side of the flyback transformer FBT shown in this figure, a so-called multisingle-type half-wave rectification circuit in which five sets of half-wave rectification circuits are connected in series is formed.

Therefore, on the secondary side of the flyback transformer FBT, these five sets of half-wave rectifiers are connected to the boost winding NVH11.
To rectify the current induced in NHV15 to smoothing capacitor C
The operation of charging the HV is performed, and a DC voltage having a level corresponding to five times the induced voltage induced in each of the boost windings NVH11 to NHV15 is obtained at both ends of the smoothing capacitor COHV. And this smoothing capacitor COHV
Is supplied as an anode voltage to, for example, an anode electrode of a CRT. In addition,
In this case, an induced voltage boosted to 6 KV is obtained in each of the boost windings NHV11 to NHV15, and the DC high voltage EHV is 30 V.
A high voltage of KV is obtained.

A smoothing capacitor COFV is inserted between the cathode of the high-voltage rectifier diode DHV14 and the secondary side ground, and the DC output voltage EFV obtained at both ends of the smoothing capacitor COFV is used as a focus voltage.
T is supplied to, for example, a fourth grid (focus electrode).

Further, a series circuit in which resistors R1 and R2 are connected in series is provided between the cathode of the high-voltage rectifier diode DHV11 and the winding start end (secondary ground) of the boost winding NHV15. The voltage divided by the series circuit is input to the control circuit 1 as a detection voltage of the DC high voltage EHV.

The control circuit 1 uses the control current or control voltage corresponding to the level change of the DC high voltage EHV output from the secondary side of the high voltage generation circuit 40 as a control signal as a PWM signal.
3 is output. As described above, the switching drive unit 30 supplies the drive voltage changed according to the voltage level of the secondary-side DC high voltage EHV output from the high voltage generation circuit 40 to the switching element Q11 of the high voltage regulator circuit 20. By controlling the on / off period of the switching element Q11, the high voltage regulator circuit 20 outputs a DC output voltage EO11 corresponding to the voltage level of the DC high voltage EHV.
Is output. By using this DC output voltage EO11 as the input voltage of the voltage resonance type converter formed on the primary side of the flyback transformer FBT, the DC high voltage E01 output from the secondary side of the flyback transformer FBT is obtained.
It is trying to stabilize HV.

Here, for example, 30
When the power supply circuit shown in FIG. 11 is actually configured so that a DC high voltage EHV of KV can be obtained, the primary winding N11 of the flyback transformer FBT = 70T (turn), and each of the boost windings NHV11 to NHV15 = 462T, parallel resonance capacitor C
r11 = 0.01 μF, smoothing capacitor COHV = 2000
PF, smoothing capacitor COFV = 220PF, resistance R1 = 1
GΩ is selected.

FIG. 12 shows operation waveforms of the power supply circuit shown in FIG. 12A to 12D, for example, when the AC input voltage VAC is 100 V and the high-voltage load of the high-voltage generation circuit 40 is the maximum load power Pomax = 60 W (IHV = 2 m
Operation waveforms under the condition A) are shown in FIGS.
FIG. 2H shows an operation waveform under the condition that the high-voltage load of the high-voltage generation circuit 40 is set to the minimum load power Pomin = 0 W (IHV = 0 mA).

When the high voltage load of the high voltage generating circuit 40 is set to the maximum load power, the ON / OFF period TON1 / TOFF1 of the switching element Q11 is controlled to 28.5 μs / 3.3 μs, and the drain-source of the switching element Q11 is controlled. Voltage V
FIG. 11 is shown as in FIG. The drain current I11 flowing through the switching element Q11 is as shown in FIG.
It is shown as (b). At this time, the DC output voltage EO11 generated in the smoothing capacitor CO11 of the high-voltage regulator circuit 20 is, for example, 121 V, as shown in FIG.

On the other hand, the ON / OFF period TON2 / TOFF2 of the switching element Q12 is controlled to 26.25 μs / 5.5 μs, and the switching element Q12 has a collector current I12 having a waveform as shown in FIG. Flows. And
In the period TOFF2 in which the switching element Q12 is turned off, both ends of the parallel resonance capacitor Cr11
A flyback pulse voltage (resonance pulse voltage) as shown in FIG.

On the other hand, when the high voltage load of the high voltage generation circuit 40 is set to the minimum load power (no load), the ON / OFF period TON1 / TOFF1 of the switching element Q11 is set to 26 μs.
/5.8 μs, and the drain-source voltage V11 of the switching element Q11 is as shown in FIG. The drain current I11 flowing through the switching element Q11 is shown as in FIG. At this time, the DC output voltage EO11 generated in the smoothing capacitor CO11 of the high-voltage regulator circuit 20 has a voltage level of, for example, 111 V as shown in FIG.
DC output voltage EO11 at maximum load power shown in (c)
(121 V). That is, from FIG. 1, the high-voltage regulator circuit 20 changes the on / off period of the switching element Q11 due to the high-voltage load of the high-voltage generation circuit 40, and the voltage level of the DC output voltage EO11 output accordingly. It can be seen that has changed.

On the other hand, the ON / OFF period TON2 / TOFF2 of the switching element Q12 is kept at 26.25 μs / 5.5 μs, and the flyback pulse voltage generated at both ends of the parallel resonance capacitor Cr11 is as shown in FIG. The collector current I12 flowing through the switching element Q12 is as shown in FIG. 12 (h).

FIG. 13A shows how the DC high voltage EHV and the DC output voltages EO1 and EO11 change with respect to the change in the beam current IHV output from the high voltage generating circuit 40 of the power supply circuit shown in FIG. . From FIG. 13A, the beam current IHV output from the high voltage generation circuit 40 is 0 mA to
When the voltage fluctuates in the range of 2 mA, only the voltage level of the DC output voltage EO11 which is the output of the high-voltage regulator circuit 20 fluctuates according to the change in the beam current IHV, and the voltage level of the DC output voltage EO1 and the DC high voltage EHV It can be seen that the voltage level is kept at a substantially constant level.

[0024]

In the power supply circuit shown in FIG. 11, since the switching power supply 10, the high-voltage regulator circuit 20, and the high-voltage generation circuit 40 perform power conversion, respectively, There is a disadvantage that the conversion efficiency is reduced. For example, the AC / DC conversion efficiency (ηAC-DC) of the switching power supply 10 is about 90%, and the voltage conversion efficiency (ηDC-DC) of the high-voltage regulator circuit 20 is about 95%. In the high-voltage generating circuit 40, power loss in the resistor R1 for obtaining the detection voltage corresponding to the DC high voltage EHV, power loss in the focus circuit for obtaining the focus voltage EFV,
Power loss in high voltage rectifier diodes DHV11 to DHV15,
The power conversion efficiency characteristic of the high-voltage generation circuit 40 is represented by, for example, FIG.
Is 60 W (EHV = 30 KV, IHV = 2 mA), the power conversion efficiency (ηDC−DC) is about 82%. Therefore,
The overall power conversion efficiency η of the power supply circuit shown in FIG. 11 at 60 W is about 70.1%, for example, 60%.
In order to obtain the high-voltage load power PHV of W, about 85.6 W of AC input power Pin is required, so that a power loss of about 25.6 W occurs in the power supply circuit shown in FIG.

Further, the power supply circuit shown in FIG. 11 requires the provision of the high-voltage regulator circuit 20.
There are drawbacks such as an increase in the number of parts, a complicated circuit configuration, and an increase in the area for mounting the parts. Further, since two sets of relatively expensive switching elements and two sets of diode elements are required, there is also a drawback that component material costs are significantly increased.

Since the switching operation of the high-voltage regulator circuit 20 has a rectangular waveform as shown in FIGS. 12A and 12D, switching noise is generated in the switching element Q11. When the illustrated power supply circuit is provided in an actual television receiver, it is necessary to take some measures to suppress noise.

[0027]

Therefore, a switching power supply circuit according to the present invention is configured as follows in consideration of the above-mentioned problems. In other words, a switching unit formed with a switching element for intermittently outputting the input DC input voltage, and a primary-side parallel resonance circuit that makes the operation of the switching unit a voltage resonance type are formed. A primary side parallel resonance capacitor is provided, and is provided for transmitting an output of the primary side to the secondary side. A primary side winding is wound on the primary side, and at least a first secondary side is wound on the secondary side. A secondary winding having a side winding part and a second secondary winding part formed so as to be wound up with respect to the first secondary winding is wound, An insulated converter transformer that can obtain the required degree of coupling, which is loosely coupled to the side winding and the secondary winding, and a secondary parallel resonance capacitor connected in parallel to the secondary winding Secondary side parallel resonance circuit formed by connecting Equipped with a. A DC output voltage formed by including a secondary parallel resonance circuit and configured to obtain a DC output voltage by performing a half-wave rectification operation on a voltage in a positive period of the alternating voltage obtained from the secondary winding. The output voltage generation means and the switching frequency of the switching element are variably controlled according to the DC output voltage level, and the switching element is switched by changing the ON period after the OFF period in the switching cycle is fixed. A first constant voltage control unit configured to perform constant voltage control by being driven. Further, by transmitting the resonance voltage input to the primary side to the secondary side, a flyback transformer that inverts the resonance voltage from the secondary side to obtain a boosted voltage, and a flyback transformer In order to make the primary side operation a resonance operation, a resonance voltage formed from a series resonance capacitor connected in series to at least the primary side winding of the flyback transformer and obtained from the secondary side parallel resonance circuit is substantially sinusoidal. By performing a rectification operation on a series resonance circuit that generates a wavy resonance voltage on the primary side of the flyback transformer and a boosted voltage obtained on the secondary side of the flyback transformer, the DC voltage corresponding to almost equal to the boosted voltage is obtained. DC high voltage generating means configured to obtain a voltage, and according to the DC high voltage level,
A second constant voltage control means for performing constant voltage control of DC high voltage is provided.

That is, according to the present invention, a resonance voltage obtained from a secondary parallel resonance circuit provided on a secondary side of an insulating converter transformer constituting a switching power supply circuit of a complex resonance type is converted into a series resonance circuit. As a result, after generating a substantially sinusoidal resonance voltage on the primary side of the flyback transformer, a predetermined boosted voltage is obtained from the secondary side of the flyback transformer. The second high-voltage control means makes the high-voltage DC output from the high-voltage DC generation means constant, thereby suppressing fluctuations in the high-voltage DC caused by high-voltage load fluctuations.

[0029]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to an embodiment of the present invention. The power supply circuit shown in this figure has a configuration as a complex resonance type switching converter including a voltage resonance type converter on the primary side and a parallel resonance circuit on the secondary side. In the power supply circuit shown in FIG. 1, first, a full-wave rectifier comprising a bridge rectifier circuit Di and a smoothing capacitor Ci as a rectifying and smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage. A smoothing circuit is provided to generate a rectified smoothed voltage (DC input voltage) Ei corresponding to a level that is one time higher than the AC input voltage VAC.

The switching converter which inputs and outputs the DC input voltage Ei intermittently has a single switching element Q
1 and a voltage resonance type converter that performs a switching operation in a so-called single-ended manner by a self-excited system. In this case, the switching element Q1
A high breakdown voltage bipolar transistor (BJT; junction transistor) is used.

The base of the switching element Q1 is connected to the positive electrode of the smoothing capacitor Ci via the current limiting resistor RB and the starting resistor RS, and the emitter is grounded to the primary side ground. Further, a series resonance circuit for self-excited oscillation driving, which is composed of a series connection circuit of a drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB, is connected between the base of the switching element Q1 and the primary side ground. The clamp diode DD1 inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the switching element Q1 is turned off. Switching element Q
The first collector is connected to one end of a primary winding N1 formed on the primary side of the isolated converter transformer PIT,
Its emitter is grounded.

A parallel resonance capacitor Cr1 is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr1 is
The primary parallel resonance circuit of the voltage resonance type converter is formed by its own capacitance and the leakage inductance L1 of the primary winding N1. Although not described in detail here, when the switching element Q1 is turned off, the voltage V1 across the resonance capacitor Cr1 due to the operation of the primary side parallel resonance circuit actually has a sinusoidal pulse waveform. Thus, a voltage resonance type operation is obtained.

The orthogonal control transformer PRT-1 is a saturable reactor in which a resonance current detection winding ND, a drive winding NB, and a control winding NC1 are wound. The orthogonal control transformer PRT-1 is provided for driving the switching element Q1 and for controlling the constant voltage. Although the illustration of the structure of the orthogonal control transformer PRT-1 is omitted,
A three-dimensional core is formed by joining the ends of the magnetic legs of two double U-shaped cores having four magnetic legs. Then, the resonance current detecting winding ND and the driving winding NB are wound in the same winding direction with respect to two predetermined magnetic legs of the three-dimensional core.
And the control winding NC1 is wound in a direction orthogonal to the resonance current detection winding ND and the drive winding NB.

In this case, the orthogonal control transformer PRT-1
Is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1, so that the switching output of the switching element Q1 is connected to the resonance current through the primary winding N1. It is transmitted to the detection winding ND. In the orthogonal control transformer PRT-1, the switching output obtained in the resonance current detection winding ND is induced in the drive winding NB via the transformer coupling, so that the drive winding NB has an alternating voltage as a drive voltage. Voltage is generated. This drive voltage is applied to a series resonance circuit (N
B, CB) is output to the base of the switching element Q1 as a drive current via the base current limiting resistor RB. As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit.

Insulation converter transformer (Power Isolation
Transformer) PIT transmits the switching output of switching element Q1 to the secondary side. As shown in FIG. 9, the structure of the insulating converter transformer PIT includes an EE-type core in which E-type cores CR1 and CR2 made of, for example, a ferrite material are combined so that their magnetic legs face each other.
The primary winding N1 and the secondary winding N2 are wound around the center magnetic leg of the EE type core using the split bobbin B, respectively. A gap G is formed for the center magnetic leg as shown in the figure. Thereby, loose coupling with a required coupling coefficient is obtained. The gap G is formed by connecting the central magnetic legs of the E-shaped cores CR1 and CR2 to 2
It can be formed by making it shorter than the outer magnetic leg of the book. The coupling coefficient k is set to a loosely coupled state of, for example, k ≒ 0.85, so that a saturated state is hardly obtained.

By the way, the insulation converter transformer PIT
Of the secondary winding includes a primary winding N1 and a secondary winding N2.
Of the primary winding N1 and the secondary winding N1 by the polarity (winding direction) of the rectifier diode D0 and the polarity of the alternating voltage excited in the secondary winding.
2 with respect to the mutual inductance M with the inductance L2, the operation mode of + M (polarization mode; forward operation) and the operation mode of -M (depolarization mode;
Flyback operation). For example, FIG.
Mutual inductance is + M when equivalent to the circuit shown in FIG. 10A, and −M when equivalent to the circuit shown in FIG. 10B. In addition, in the power supply circuit shown in FIG.
The polarity of the primary winding N1 and the secondary windings N2, N3 of T is + M
During the operation mode of the rectifier diode DO
1. The charging operation of the smoothing capacitors CO1 and CO3 is performed via DO3.

The winding start end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1 as shown in FIG. 1, and the winding end is connected in series with the resonance current detection winding ND. Connected to the positive electrode of the smoothing capacitor Ci. On the secondary side, a secondary winding N2 serving as a first secondary winding, and a second secondary side formed by winding up a winding end portion of the secondary winding N2. A tertiary winding N3 is provided as a winding. And
A secondary side parallel resonance capacitor C2 is connected to a secondary side winding (N2 + N3) including the secondary winding N2 and the tertiary winding N3.
Are connected in parallel.

In this case, the winding start end of the secondary winding N2 is connected to the secondary side ground, and the winding end is connected to the rectifier diode D.
Connected to the anode of O1. A half-wave rectifying / smoothing circuit composed of the rectifying diode DO1 and the smoothing capacitor CO1 obtains a DC output voltage EO1 (for example, 135 V) for horizontal deflection of 110V to 140V.

A tap is provided at a required position of the secondary winding N2, and a rectifier diode DO2
The anode is connected. Then, a half-wave rectifying and smoothing circuit including the rectifying diode DO2 and the smoothing capacitor CO2 provides a DC output voltage EO2 for a signal system circuit.
(15V).

Further, on the secondary side of the insulated converter transformer PIT, the ending of the tertiary winding N3 wound around the secondary winding N2 is connected to the anode of the rectifying diode DO3, so that the rectifying diode DO3 is connected to the rectifying diode DO3. The DC output voltage EO3 (200 V) for the video output circuit is obtained by the half-wave rectifier circuit including the smoothing capacitor CO3. In the power supply circuit shown in FIG. 1, the negative side of the smoothing capacitor CO3 is connected to the smoothing capacitor CO1. By connecting to the positive electrode side,
The DC output voltage EO3 for the video output circuit is obtained from both ends of the series connection circuit of the smoothing capacitors CO1 to CO3. That is, in the power supply circuit shown in FIG. 1, in order to obtain the DC output voltage EO3 for the video output circuit, the DC output voltage EO1 generated across the smoothing capacitor CO1 is added to the smoothing capacitor C01.
The DC output voltage generated at both ends of O3 is accumulated, that is, the DC output voltage EO1 obtained from the secondary winding N2 and the DC output voltage obtained from the tertiary winding N3 are superimposed to obtain the DC output voltage EO3. ing. For this reason, the configuration of the rectifying and smoothing circuit including the tertiary winding N3, the rectifying diode DO3 and the smoothing capacitor CO3 includes a DC output voltage EO3.
(200 V), the DC output voltage EO1 (110 V to 14
0 V) is subtracted and a DC output voltage of 90 V to 60 V can be obtained. Although not shown, a DC output voltage (± 15 V) for the vertical deflection circuit and a DC output voltage (6.3 V) for the heater may be obtained from the secondary side of the insulating converter transformer PIT. It is possible.

In this case, the secondary side parallel resonance capacitor C2 is connected in parallel to the secondary side winding (N2 + N3) composed of the secondary winding N2 and the tertiary winding N3, so that the secondary side is connected. Leakage inductance of side winding (N2 + N3) (L2 +
L3) and the capacitance of the secondary parallel resonance capacitor C2 form a secondary parallel resonance circuit. The alternating voltage induced in the secondary winding N2 and the tertiary winding N3 becomes a resonance voltage, and the isolated converter transformer PIT , A voltage resonance operation is obtained on the secondary side.

That is, in the power supply circuit shown in FIG. 1, the primary side is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and the secondary side is provided with a parallel resonance circuit for obtaining a voltage resonance operation. Be provided. In the present specification, a switching converter having a configuration in which a resonance circuit is provided for the primary side and the secondary side and operates as described above is described below.
It is also referred to as a “composite resonance type switching converter”. The configuration as such a composite resonance type switching converter is realized by forming a gap G with respect to the insulated converter transformer PIT and performing loose coupling by a required coupling coefficient as described with reference to FIG. ,
This is further realized by obtaining a state that is unlikely to be saturated. For example, when the gap G is not provided for the insulating converter transformer PIT, there is a high possibility that the insulating converter transformer PIT becomes saturated during flyback operation and the operation becomes abnormal. It is difficult to hope that this is done properly.

The above-described DC output voltage EO1 is also branched and input to the control circuit 1. The control circuit 1 is constituted by, for example, an error amplifier or the like, and changes the orthogonal control transformer PRT-1 according to a change in the DC output voltage level EO1 output from the secondary side of the insulating converter transformer PIT.
, The inductance LB of the drive winding NB wound around the orthogonal control transformer PRT-1 is variably controlled by varying the level of the control current (DC current) supplied to the control winding NC1.
As a result, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB of the drive winding NB changes, and the switching frequency of the switching element Q1 is varied. Become. This operation stabilizes the DC output voltage output from the secondary side of the insulating converter transformer PIT. Note that the DC output voltage EO3 may be branched and input to the control circuit 1 to make the DC output voltage constant.

When the orthogonal control transformer PRT-1 for variably controlling the inductance LB of the drive winding NB is provided as in the power supply circuit shown in FIG. 1, the switching element Q1 is turned off to change the switching frequency. Is kept constant, and the ON period TO is kept constant.
N is variably controlled. In other words, in the power supply circuit shown in FIG. 1, as the constant voltage control operation, the switching frequency is variably controlled to perform the resonance impedance control on the switching output, and at the same time, the conduction angle control (PWM) of the switching element Q1 in the switching cycle. Control).
This complex control operation is realized by a set of control circuit systems. In the present specification, such complex control is also referred to as “complex control method”.

Further, in the power supply circuit shown in FIG. 1, the series resonance capacitor C3, the primary winding N4 of the flyback transformer FBT and the orthogonal control transformer PRT-2 are connected in parallel with the secondary parallel resonance capacitor C2. A series resonance circuit including a series connection of control windings (main windings) NR is provided. That is, in the power supply circuit shown in FIG. 1, the secondary winding (N2 + N3) of the insulated converter transformer PIT, the series resonance capacitor C3, the primary winding N4, and the controlled winding are connected to the secondary parallel resonance capacitor C2. The series resonance circuits composed of the line NR are connected in parallel.

In the power supply circuit having such a configuration, the isolated converter transformer PIT operates as a complex resonance type switching converter, so that a resonance pulse voltage is generated across the secondary side parallel resonance capacitor C2. Then, the DC output voltage EO3 is obtained from the positive resonance pulse voltage generated during the positive period in which the insulating converter transformer PIT performs the forward operation, and the resonance pulse voltage generated in the secondary parallel resonance capacitor C2 is converted into the series resonance capacitor C3. The signal is input to the primary winding N4 of the flyback transformer FBT through the input terminal. In this case, a series resonance circuit composed of the series resonance capacitor C3, the primary winding N4, and the controlled winding NR is formed on the primary side of the flyback transformer FBT. The resonance pulse voltage V2 generated at both ends is
The capacitance of the DC resonance capacitor C3, the inductance of the primary winding N4 of the flyback transformer FBT, and the controlled winding N of the orthogonal control transformer PRT-2.
By the series resonance operation by the inductance LR of R, the current I4 flowing through the primary winding N4 of the flyback transformer FBT and the voltage V4 across the primary winding N4 both have a substantially sinusoidal resonance waveform.

The orthogonal control transformer PRT-2 is a saturable reactor around which the controlled winding NR and the control winding NC2 are wound, and a constant DC high voltage EHV output from a high voltage generating circuit 4 described later. Provided for voltage control. As shown in FIG. 4A, the structure of the orthogonal control transformer PRT-2 is such that two double U-shaped cores 50a and 50b having four magnetic legs are joined to each other at the ends of the magnetic legs. Then, a three-dimensional core (ferrite core) 50 is formed.
A controlled winding NR is wound around predetermined two magnetic legs of the three-dimensional core 50, and a control winding NC2 is wound in a direction orthogonal to the controlled winding NR. It is configured by mounting. In this case, the U-shaped cores 50a, 50
A gap G is provided at a portion where b faces.

Such an orthogonal control transformer PRT-2
As shown in FIG. 4B, according to the level of the control current (DC current) IC2 flowing through the control winding NC2, the inductance LR of the controlled winding NR is, for example, 50 μH to 10 μH.
It changes in the range of μH. In other words, the orthogonal control transformer PRT-2 is provided with a primary winding N4 of the flyback transformer FBT in order to make the DC high voltage EHV constant.
Variably controls the inductance of the controlled winding NR connected in series with the flyback transformer FB.
The inductance control of the series resonance circuit formed on the primary side of T is performed.

High-voltage generating circuit 4 enclosed by a dashed line
Is configured by a flyback transformer FBT and a high-voltage rectifier circuit, and boosts the resonance voltage V4 generated in the primary winding N4 of the flyback transformer FBT, for example, by C
A high voltage corresponding to the anode voltage level of RT is generated. For this reason, on the secondary side of the flyback transformer FBT, four to five sets of step-up windings NHV are divided and wound by slit winding or interlayer winding. In this case, the primary winding N4 and the step-up winding NHV are wound so as to be tightly coupled, and furthermore, are wound so that their polarities (winding directions) are opposite. Therefore, the flyback transformer F
On the secondary side of the BT, the negative resonance voltage of the resonance voltage V4 generated in the primary winding N4 is inverted, and the winding ratio (NHV / N4) between the boost winding NHV and the primary winding N4 is determined. The boosted voltage is obtained.

Here, FIG. 3 shows a flyback transformer FB.
A cross-sectional view of T is shown, and the structure of the flyback transformer FBT will be described with reference to FIG. The flyback transformer FBT shown in this figure has, for example, two U-shaped cores C.
By combining the magnetic legs of R1 and CR2 so as to face each other, a square core CR30 is formed. A gap G is provided at a portion where the end of the U-shaped core CR1 and the end of the U-shaped core CR2 face each other. Further, as shown in the figure, by attaching a low-voltage winding bobbin LB and a high-voltage winding bobbin HB to one of the magnetic legs of the square core CR30, the low-voltage winding bobbin LB and the high-voltage winding bobbin HB are attached. The primary winding N4 and the boost winding NHV are respectively divided and wound. In this case, the primary winding N4 is wound around the low-voltage winding bobbin LB, and a plurality of boosting windings NHV are wound around the high-voltage winding bobbin HB by inserting and winding up the interlayer film F. become.

In the power supply circuit shown in FIG. 1, the resonance voltage V4 input to the primary winding N4 of the flyback transformer FBT is obtained from the secondary side of the insulating converter transformer PIT. The frequency corresponds to the switching frequency of the switching element Q1, and is, for example, in the range of about several tens of kHz to about 200 kHz. For this reason, in the present embodiment, a litz wire is used for the primary winding N4 of the flyback transformer FBT to prevent eddy current from being generated in the primary winding N4.

In this embodiment, the case where the boost winding NHV of the flyback transformer FBT is wound by interlayer winding is shown, but the winding of the boost winding NVH is limited to the interlayer winding. Although not shown, for example, the high-pressure bobbin HB is divided into a plurality of
It is also possible to wind the step-up windings NHV around each divided region, that is, to wind them by so-called divided winding. That is,
The structure of the flyback transformer FBT may be such that a plurality of boost windings NHV wound around the high-voltage bobbin HB are wound in an insulated state.

In the power supply circuit shown in FIG. 1, on the secondary side of the flyback transformer FBT, five sets of boost windings NVH1, NHH are provided.
V2, NHV3, NHV4, and NHV5 are wound independently of each other. High-voltage rectifier diodes DHV1, DHV2, and DHV are provided at the winding ends of the respective boost windings NVH1 to NVH5.
3, the anode sides of DHV4 and DHV5 are connected. The cathode of the high-voltage rectifier diode DHV1 is connected to the positive terminal of the smoothing capacitor COHV, and the remaining cathodes of the high-voltage rectifier diodes DHV2 to DHV5 are connected to the boost winding N
Connected to the winding start ends of HV1 to NHV4.

That is, on the secondary side of the flyback transformer FBT, [the boost winding NHV1, the high-voltage rectifier diode DHV
1], [boost winding NHV2, high voltage rectifier diode DHV2],
[Step-up winding NHV3, high-voltage rectifier diode DHV3], [Step-up winding NHV4, high-voltage rectifier diode DHV4], [Step-up winding N
HV5, high-voltage rectifier diode DHV5], that is, a so-called multisingler type half-wave rectifier circuit in which five sets of half-wave rectifier circuits are connected in series.

Therefore, on the secondary side of the flyback transformer FBT, five sets of half-wave rectifier circuits
~ NHV5 is rectified to smooth the capacitor CO
An operation of charging the HV is performed, and a DC high voltage (anode voltage) EHV having a level corresponding to about five times the induced voltage induced in each of the boost windings NVH1 to NHV5 is provided across the smoothing capacitor COHV. Is obtained.

Further, a capacitor COFV is inserted between the cathode of the high voltage rectifier diode DHV4 and the secondary side ground, and a DC output voltage of a voltage level lower than the anode voltage EHV obtained at both ends of the capacitor COFV is obtained. Focus voltage) EFV is obtained.

Further, a series circuit composed of a resistor R1 and a resistor R2 is connected between the cathode of the high voltage rectifier diode DHV1 and the secondary side ground, and the voltage divided by the resistors R1 and R2 is used as a control circuit. 2 is input. The control circuit 2 is also formed of, for example, an error amplifier and the like, and controls a control current (DC current) level flowing through the control winding NC2 of the orthogonal control transformer PRT-2 according to a level change of the voltage obtained from the DC high voltage EHV. By varying IC2, the inductance LR of the controlled winding NR is variably controlled. And
By varying the current I4 flowing through the primary winding N4 of the flyback transformer FBT, the anode voltage EHV and the focus voltage EFV can be made constant.

When the power supply circuit shown in FIG. 1 is actually constructed, the secondary winding N2 of the insulated converter transformer PIT =
45T, tertiary winding N3 = 22T, secondary parallel resonance capacitor C2 = 2700PF, series resonance capacitor C3 = 39
00PF, primary winding N of flyback transformer FBT
4 = 30T, boost windings NHV1 to NHV5 = 460T, smoothing capacitor COHV = 2000PF, smoothing capacitor COFV =
220 PF and a resistance R1 = 1 GΩ are selected.

FIG. 2 shows operation waveforms of the power supply circuit shown in FIG. 2A to 2F, for example, when the AC input voltage VAC is 100 V and the high-voltage load power of the high-voltage generation circuit 4 is the maximum load power Pomax = 60 W (IHV = 2 mA)
FIG. 2 (g) to FIG.
(L) shows an operation waveform under the condition that the high-voltage load power of the high-voltage generation circuit 4 is the minimum load power Pomin = 0W (IHV = 0mA), for example.

When the high-voltage load of the high-voltage generation circuit 4 is set to the maximum load power, the switching frequency of the switching element Q1 is controlled to, for example, 90.9 kHz, and the actual on / off period TON / OFF of the switching element Q1 is controlled. TOFF
Is 7.4 μs / 3.6 μs. The resonance voltage V1 generated at both ends of the parallel resonance capacitor Cr1 by the on / off operation of the switching element Q1 is shown in FIG.
As shown in (a), in the period TOFF during which the switching element Q1 is turned off, a sine-wave-like waveform is obtained, indicating that the operation of the switching converter is of the voltage resonance type. At this time, a collector current IC as shown in FIG. 2B flows through the switching element Q1. For example, when the switching element Q1 is turned on, a damper current (negative direction) flows to the primary winding N1 via the clamp diode DD1 and the base-collector of the switching element Q1, and a damper period (for example, 1.4 μs) in which the damper current flows. Is completed, the collector current IC rapidly rises in level from the negative level to the positive level.

By such an operation, the secondary winding (N2 + N3) of the insulated converter transformer PIT is connected to FIG.
As shown in FIG. 2C, the resonance current I2 flows as shown in FIG. 2D, and the voltage V2 generated in the secondary-side parallel resonance capacitor C2, as shown in FIG.
Due to the operation of the rectifier diodes DO1, DO3, the voltage level becomes a positive voltage level of 200 V, and during a period TOFF during which the switching element Q1 is turned off, a negative resonance pulse voltage whose peak voltage level is 500 Vp Become. The resonance pulse voltage generated in the secondary side resonance capacitor C2 is supplied to the flyback transformer FB.
The primary winding N4 of T and the controlled winding NR of the orthogonal control transformer PRT-2 are input to the primary winding N4 via a series resonance capacitor C3 which forms a series resonance circuit, so that the primary winding N4 becomes primary. Resonant voltage V4 generated at both ends of the side winding N4
As shown in FIG. 2 (e), the resonance voltage waveform has a peak voltage level of 400 Vp, and the resonance current I4 flowing through the primary winding N4 is as shown in FIG. 2 (f). The resonance current waveform has a peak value of 2 Ap.

On the other hand, when the high voltage load of the high voltage generation circuit 4 is set to the minimum load power (no load), the switching element Q1
Is controlled to be, for example, 100 kHz, and the actual ON / OFF period TON / TOFF of the switching element Q1 is 6.4 μs / 3.6 μs. In this case, the resonance voltage V1 generated at both ends of the parallel resonance capacitor Cr1 by the on / off operation of the switching element Q1, as shown in FIG. 2 (g), has a sinusoidal waveform during the period TOFF when the switching element Q1 is off. Is obtained. At this time, a collector current IC as shown in FIG. 2 (h) flows through the switching element Q1. In this case, the period of the damper current flowing when the switching element Q1 is turned on is set to about 2 μs. It is longer than the damper period (1.4 μs) at the time of the maximum load power shown in (b).

By such an operation, the secondary winding (N2 + N3) of the insulated converter transformer PIT is connected to FIG.
As shown in FIG. 2 (j), the resonance current I2 flows, and the voltage V2 generated across the secondary-side parallel resonance capacitor C2 is as shown in FIG.
As shown in (i). The negative resonance pulse generated in the secondary resonance capacitor C2 is input to the primary winding N4 of the flyback transformer FBT via the series resonance capacitor C3, and the waveform of the voltage V4 across the primary winding N4 is 2 (k), and the current waveform of the current I4 flowing through the primary winding N4 has a resonance waveform as shown in FIG. 2 (l).

FIGS. 2A to 2F and FIG. 2G
2 (l), the power supply circuit shown in FIG. 1 shows that when the high-voltage load power of the high-voltage generation circuit 4 fluctuates from the maximum load power Pomax to the minimum load power Pomin, the switching element Q1 Switching frequency changes from 90.9 kHz to 100 kHz.
In other words, the power supply circuit shown in FIG. 1 variably controls the switching frequency of the primary-side switching converter in accordance with the change in the high-voltage load of the high-voltage generation circuit 4. The inductance LR of the controlled winding NR of the orthogonal control transformer PRT-2 is variably controlled to make the anode voltage EHV and the focus voltage output from the high voltage generation circuit 4 constant.

FIG. 5 shows the power conversion efficiency characteristics of the high voltage generating circuit 4 configured as described above. For example, when the high effective load power PHV is 60 W (EHV = 30 KV, IHV = 2 m)
The power conversion efficiency (ηDC-DC) at the time A) is about 92%.
Therefore, as compared with the power supply circuit shown in FIG. 11, the power conversion efficiency (ηDC−DC) can be improved by about 10%, and for example, the power loss in the high voltage generation circuit 4 when obtaining the high voltage load power PHV of 60 W Can be reduced by about 8 W. Accordingly, the overall power conversion efficiency η is about 8
Since it is improved to 8.3% and the AC input power Pin is about 68 W, it is possible to reduce the AC input power Pin by about 17.6 W as compared with the power supply circuit shown in FIG.

In the conventional power supply circuit shown in FIG. 11, the switching power supply 10 and the high-voltage regulator circuit 20
However, the power supply circuit of the present embodiment shown in FIG. 1 does not require a high-voltage regulator circuit, and the switching power supply can supply the peak power. Therefore, the supply capacity is large. Therefore, for example, when a white peak window screen is displayed on a CRT screen displaying a black screen as shown in FIG. 6, the conventional power supply circuit shown in FIG. While the image is curved, the power supply circuit of the present embodiment shown in FIG. 1 can eliminate the curved image of the white peak window screen as shown by the solid line.

Further, since it is not necessary to provide the high voltage regulator circuit 20 provided in the conventional power supply circuit shown in FIG. 11, the number of parts can be reduced. Therefore, the circuit can be simplified and the mounting area of the components can be reduced, so that the switching power supply can be reduced in size. Further, since one set of a relatively expensive switching element and diode element may be used, component material costs can be reduced.

Further, since the operation waveforms of the respective parts are all resonance waveforms, it is possible to suppress the switching noise generated due to the switching operation and to suppress the switching noise required in the conventional power supply circuit. There is also an advantage that no countermeasure is required.

Further, the power supply circuit of the present invention is not limited to the circuit configuration shown in FIG. FIG. 7 is a diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention. The same parts as those of the power supply circuit shown in FIG. In the power supply circuit shown in FIG. 7, the voltage resonance type converter provided on the primary side employs a separately-excited configuration, and is provided with, for example, a single MOS-FET switching element Q2. The drain of the switching element Q2 is connected to the insulation converter transformer PIT.
Is connected to the positive electrode of the smoothing capacitor Ci through the primary winding N1, and the source is connected to the primary side ground. Also,
Here, the parallel resonance capacitor Cr2 is connected in parallel between the drain and the source. Further, a clamp diode DD2 is connected in parallel between the drain and the source.

The switching element Q 2 is driven by the oscillation / drive circuit 3 so that the switching operation described above with reference to FIG. 1 is obtained. That is, the control circuit 1 supplies the oscillation / drive circuit 3 with a current or voltage at a level that fluctuates according to the fluctuation of the DC output voltage EO1. In the oscillation / drive circuit 3, a switching drive whose cycle is varied according to the output level from the control circuit 1 so that the DC output voltage output from the secondary side of the insulating converter transformer PIT is stabilized. A signal (voltage) is output to the gate of the switching element Q2. As a result, the switching frequency of the switching element Q2 is varied. At this time, as described in FIG. 1, the period during which the switching element Q2 is off is constant, and the period during which the switching element Q2 is on is varied. The switching drive signal generated as much as possible is output. The starting resistor RS is provided to supply a starting current obtained in the rectifying / smoothing line to the oscillation / drive circuit 3 when the commercial AC power is turned on.

On the secondary side of the insulating converter transformer PIT formed as described above, the tertiary winding N5 (third secondary winding) is wound around the winding end of the secondary winding N3. The ending of the tertiary winding N5 is connected to the starting end of the primary winding N4 of the flyback transformer FBT via the series resonance capacitor C3. N4, a series resonance circuit composed of the controlled winding NR of the orthogonal control transformer PRT-2 is formed.

In this case, the same effects as those of the power supply circuit of FIG. 1 described above can be obtained, and the flyback transformer FB can be further connected from the secondary side of the isolated converter transformer PIT.
Since the voltage level of the resonance voltage V4 input to the primary winding N4 of T can be increased, the boosted voltage level boosted by the flyback transformer FBT is suppressed lower than that of the power supply circuit shown in FIG. It becomes possible. Therefore, comparing the power supply circuit shown in FIG. 7 with the power supply circuit shown in FIG. 1, the power supply circuit shown in FIG. 7 reduces the number of windings of each boost winding NHV1 to NHV5 of the flyback transformer FBT. The size of the flyback transformer FBT can be further reduced.

Further, the secondary side circuit configuration of the power supply circuit according to the present embodiment described above is conceivable. FIG. 8 is a diagram showing another configuration example of the secondary circuit of the power supply circuit according to the present embodiment described above. The primary-side circuit configuration of the power supply circuit shown in FIG. 8 may be either the self-excited type voltage resonance converter shown in FIG. 1 or the separately-excited type voltage resonance converter shown in FIG. Also, the same parts as those in FIG.

The insulated converter transformer P shown in FIG.
On the secondary side of the IT, the tertiary winding N5 is formed by winding up the winding end of the tertiary winding N3. In this case, the winding end of the tertiary winding N5 is directly connected to the flyback transformer FBT. It is connected to the winding start end of the primary winding N4. The difference is that a series resonance capacitor C3 is connected in series to the winding end of the primary winding N4, and an active clamp circuit 5 is connected in parallel to the series resonance capacitor C3. .

In this case, the active clamp circuit 5
Auxiliary switching element Q3, clamp capacitor CCL,
It is formed with a clamp diode DD3. Note that, for example, a so-called body diode is selected as the clamp diode DD2. In addition, the auxiliary switching element Q3
Is provided with a drive winding Ng, a capacitor Cg, and a resistor Rg.

In this case, a clamp diode DD3 is provided between the drain and source of the auxiliary switching element Q3.
Are connected in parallel. Here, the clamp diode D
The anode of D3 is connected to the source and the cathode is connected to the drain. The drain of the auxiliary switching element Q3 is connected to one terminal of the clamp capacitor CCL, and the other terminal is connected to the connection point between the winding termination of the primary winding N4 of the flyback transformer FBT and the series resonance capacitor C3. Connected to The source of the auxiliary switching element Q3 is grounded with respect to the secondary ground. That is, the active clamp circuit 5 of the present embodiment is configured by connecting the clamp capacitor CCL in series to the parallel connection circuit of the auxiliary switching element Q3 // clamp diode DD3. The circuit thus formed is connected in parallel to the series resonance capacitor C3.

As a driving circuit system for the auxiliary switching element Q3, a series connection circuit of a resistor Rg, a capacitor Cg, and a driving winding Ng is connected to the gate of the auxiliary switching element Q3, as shown in the figure. This series connection circuit forms a self-excited oscillation driving circuit for the auxiliary switching element Q3. Here, the drive winding Ng is formed so as to wind up the winding start end side of the secondary winding N2 in the insulating converter transformer PIT, and the number of turns in this case is, for example, 1T (turn). As a result, a voltage excited by the alternating voltage obtained in the secondary winding N2 is generated in the driving winding Ng. In practice, if the number of turns of the drive winding Ng is 1T, the operation is guaranteed, but the present invention is not limited to this. further,
The gate of the auxiliary switching element Q3 is also connected to the control circuit 2, and the flyback transformer F
A control voltage corresponding to the level change of the DC high voltage EHV output from the secondary side of the BT is input.

In the case of such a configuration, the control circuit 2 controls the conduction angle (PWM control) of the auxiliary switching element Q3 of the active clamp circuit 5, thereby equivalently varying the capacitance of the series resonance capacitor C3. Therefore, the series resonance frequency of the series resonance circuit including the primary winding N4 and the series resonance capacitor C3 can be controlled. This makes it possible to control the resonance voltage level input to the primary winding N4 of the flyback transformer FBT according to the voltage level of the DC high voltage EHV output from the secondary side of the flyback transformer FBT. In this case as well, it is possible to realize a constant DC high voltage EHV output from the high voltage generation circuit 4.

In this case as well, a tertiary winding N5, which is a third secondary winding, is wound around the terminal end of the tertiary winding N3.
The winding end of this tertiary winding N5 is connected to a flyback transformer F
By connecting to the primary winding N4 of the BT, the primary winding N4
Since the resonance voltage level inputted to the flyback transformer FBT is increased, the boosted level of the boosted voltage in the flyback transformer FBT can be suppressed lower than that of the power supply circuit shown in FIG. , The number of windings of each of the boost windings NHV1 to NHV5 can be reduced. Further, in the power supply circuit shown in FIG. 8, the tertiary winding N5 is wound around the winding end of the tertiary winding N3. It is also possible to configure such that the terminal portion is connected to the primary winding N4 of the flyback transformer FBT.

In the present embodiment, the orthogonal control transformer PRT is used as a control transformer for performing constant voltage control under a configuration having a self-excited resonance converter for the primary side. Instead of the orthogonal control transformer PRT, an oblique control transformer previously proposed by the present applicant can be employed. Although the illustration of the structure of the oblique control transformer is omitted here, as in the case of the orthogonal control transformer, for example, 4
A three-dimensional core is formed by combining two sets of double U-shaped cores having two magnetic legs. Then, the control winding NC and the driving winding NB are wound around the three-dimensional core. At this time, the winding directions of the control winding and the driving winding obliquely intersect. To be. Specifically, one of the control winding NC and the drive winding NB is
Of the four magnetic legs, winding is performed on two magnetic legs adjacent to each other, and the other winding is wound on two magnetic legs that are considered to be in a diagonal positional relationship. It is to be wound.
When such an oblique control transformer is provided, even when the AC current flowing through the drive winding changes from a negative current level to a positive current level, the operation tendency that the inductance of the drive winding increases. Is obtained. This allows
The current level in the negative direction for turning off the switching element increases, and the accumulation time of the switching element is shortened.Accordingly, the fall time at the time of turning off the switching element is also shortened, and the switching element is turned off. The power loss can be further reduced.

[0081]

As described above, the switching power supply circuit of the present invention comprises a secondary parallel resonance circuit provided on the secondary side of an insulating converter transformer constituting a switching power supply circuit of a complex resonance type. The obtained resonance voltage is generated on the primary side of the flyback transformer as a substantially sinusoidal resonance voltage by the series resonance circuit, and a predetermined boosted voltage is obtained from the secondary side of the flyback transformer,
The second constant voltage control means is configured to make the DC high voltage output from the DC high voltage generation means constant.

Therefore, if the switching power supply circuit of the present invention is applied to a video device such as a large color television receiver or a projector device having a high resolution, for example,
Since the power conversion efficiency of the DC high-voltage generating means can be improved, the overall power conversion efficiency can be improved, and the power loss in the switching power supply circuit can be greatly reduced. Become.

Further, the switching power supply circuit of the present invention
Since the peak power supply capability can be increased as compared with the conventional power supply circuit, it is possible to eliminate image bending or the like when a white peak window screen is displayed on a CRT screen, for example.

Further, since the circuit configuration can be simplified as compared with the conventional power supply circuit, the cost of parts can be significantly reduced, and the switching power supply circuit can be downsized. Furthermore, since the operation waveforms of the respective units are all resonance waveforms, there is an advantage that noise generated due to the switching operation is suppressed and a measure for suppressing the noise is not required.

A third secondary winding is wound around the second secondary winding of the insulated converter transformer, and a series resonance circuit is connected from the third secondary winding. If so, the resonance voltage level obtained on the secondary side of the insulating converter transformer can be increased, so that the resonance voltage level generated on the primary side of the flyback transformer can be increased. Thereby, the number of windings of the flyback transformer can be reduced, and the flyback transformer can be downsized.

[Brief description of the drawings]

FIG. 1 is a diagram showing a configuration of a power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

FIG. 3 is a sectional view showing a configuration of a flyback transformer provided in the power supply circuit shown in FIG. 1;

FIG. 4 is a sectional view showing a configuration of an orthogonal control transformer PRT-2 provided in the power supply circuit shown in FIG. 1, and a diagram showing its inductance characteristics.

FIG. 5 is a diagram showing a voltage conversion efficiency characteristic of the voltage resonance converter according to the present embodiment.

FIG. 6 is an explanatory diagram of image curving that occurs when a white peak window screen is displayed on a CRT screen.

FIG. 7 is a diagram illustrating a configuration of a power supply circuit according to a second embodiment of the present invention;

FIG. 8 is a diagram showing another circuit configuration of the secondary circuit applicable to the power supply circuit according to the present embodiment.

FIG. 9 is a cross-sectional view showing the structure of the insulating converter transformer.

FIG. 10 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 11 is a diagram showing a configuration of a power supply circuit provided in a conventional high-resolution television receiver.

12 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

13A and 13B are diagrams illustrating a state of a change in each unit with respect to a change in a beam current of the power supply circuit illustrated in FIG. 11, and a diagram illustrating a voltage conversion efficiency characteristic of a high voltage generation circuit.

[Explanation of symbols]

1 2 control circuit, 3 oscillation / drive circuit, 4 high voltage generation circuit, 5 active clamp circuit, AC commercial AC power supply, Ci smoothing capacitor, Cr1 Cr2 primary parallel resonance capacitor, C2 secondary parallel resonance capacitor,
C3 series resonance capacitor, CB resonance capacitor, CCL
Clamp capacitor, COHV COFV smoothing capacitor,
Di bridge rectifier circuit, DD1 to DD3 clamp diode, DHV1 to DHV5 High voltage rectifier diode, DO1 to DO3
Rectifier diode, FBT flyback transformer, NHV
1 to NHV5 boost winding, N1 N4 primary winding, N2 secondary winding (secondary winding), N3 N5 tertiary winding (secondary winding), NB Ng drive winding, NC1 NC2 control winding Line, NR
Controlled winding, PIT insulation converter transformer, PR
T-1 PRT-2 orthogonal control transformer, Q1 Q2
Switching element, Q3 auxiliary switching element, R1
R2 Rg RS RB Resistance

 ──────────────────────────────────────────────────続 き Continued on the front page F term (reference) 5C068 AA06 CA04 CA07 CB04 CC03 KA02 5H730 AA14 AA15 AS01 AS04 AS15 BB23 BB43 BB52 BB57 BB72 CC01 DD02 DD04 DD23 EE02 EE07 EE45 EE73 FD01 FD41 FG03 FG07 ZZ16

Claims (5)

[Claims] 1. A switching means formed with a switching element for intermittently outputting an input DC input voltage, and a primary-side parallel resonance circuit having a voltage resonance type operation of the switching means. A primary side parallel resonance capacitor provided in such a manner as to be provided for transmitting the output of the primary side to the secondary side, a primary side winding is wound on the primary side, and at least the A secondary winding having a first secondary winding and a second secondary winding formed to wind up with respect to the first secondary winding. In addition, an insulation converter transformer capable of obtaining a required degree of coupling that is loosely coupled between the primary winding and the secondary winding, and a secondary winding with respect to the secondary winding. Side parallel resonant capacitors are connected in parallel. A secondary parallel resonance circuit, and the secondary parallel resonance circuit, the DC output is obtained by performing a half-wave rectifying operation on a voltage in a positive period of an alternating voltage obtained from the secondary winding. A DC output voltage generating means configured to obtain a voltage, and variably controlling a switching frequency of the switching element according to the DC output voltage level, while keeping an OFF period in a switching cycle constant, A first constant voltage control means for performing constant voltage control by switching the switching element so as to change the period, and transmitting a resonance voltage input to the primary side to the secondary side. By doing so, the flyback transformer configured to obtain a boosted voltage by inverting the resonance voltage from the secondary side to obtain a boosted voltage, and the primary side operation of the flyback transformer co-oscillating In order to achieve this, at least the resonance voltage formed from a serial resonance capacitor connected in series to the primary winding of the flyback transformer and obtained from the secondary parallel resonance circuit is converted into a substantially sinusoidal resonance. By performing a rectification operation on a boosted voltage obtained on the secondary side of the flyback transformer and a series resonance circuit generated on the primary side of the flyback transformer as a voltage, a DC voltage corresponding to almost equal to the boosted voltage is obtained. DC high voltage generation means configured to obtain a voltage, and second constant voltage control means configured to perform constant voltage control of the DC high voltage according to the DC high voltage level. A switching power supply circuit. 2. The controlled constant winding connected in series to a primary winding of the flyback transformer forming the series resonance circuit, wherein:
A control transformer comprising a control winding for controlling the inductance of the controlled winding; and variably controlling the inductance of the controlled winding in accordance with the DC high voltage level, thereby setting the DC high voltage constant. The switching power supply circuit according to claim 1, wherein voltage control is performed. 3. The active clamp means comprising a series connection circuit of at least a clamp capacitor and an auxiliary switching element, wherein the series connection circuit is connected in parallel to the series resonance capacitor. 2. The switching device according to claim 1, wherein a constant voltage control of the DC high voltage is performed by controlling a conduction angle of the auxiliary switching element according to the DC high voltage level. Power circuit. 4. An insulating converter transformer, wherein a third secondary winding is wound around a second secondary winding of the insulation converter transformer, and the second secondary winding is wound with respect to the third secondary winding. The switching power supply circuit according to claim 1, wherein the series resonance circuit is connected to the switching power supply circuit. 5. The DC high voltage generating means includes: a first DC high voltage having a predetermined high voltage level;
2. The switching power supply circuit according to claim 1, wherein the switching power supply circuit is configured to be able to output a second DC high voltage having a voltage level lower than the DC high voltage.