JP2002034244A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To make a DC high voltage outputted by a flyback transformer constant. SOLUTION: A negative resonance voltage V2 outputted by a secondary side of an insulated converter transformer PIT comprising a composite resonance type switching power supply circuit is inputted to a primary side of a flyback transformer FBT via a series resonance capacitor C3. After an approximately sinusoidal resonance voltage V4 generated in the primary winding N4 of the flyback transformer FBT is stepped up, a DC high voltage EHV with a predetermined high voltage level is obtained. With such a constitution, in order to obtain, for instance, the DC high voltage EHV necessary for the horizontal deflection of a television receiver, the mediation of a horizontal deflection circuit system is avoided, so that a power loss can be reduced and the DC high voltage can be kept constant even if a power for a high voltage load fluctuates.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit suitable as, for example, a power supply for a television receiver.

[0002]

2. Description of the Related Art For example, in a color television receiver provided with a cathode ray tube (CRT: Cathode-ray Tube) for displaying an image, an electron beam emitted from an electron gun inside the CRT is horizontally (horizontally) directed. The power supply circuit mainly includes a horizontal deflection system circuit that deflects the current and a switching power supply of a soft switching type using a current resonance type converter.

FIG. 15 is a diagram showing a configuration of a horizontal deflection system circuit provided in a television receiver and its peripheral circuits. The switching power supply 10 shown in FIG.
The DC-DC converter performs switching of the input DC voltage, converts the DC voltage into a DC voltage of a predetermined voltage level, and outputs the DC voltage. The DC-DC converter is configured by a current resonance type switching power supply circuit as described later.

In the preceding stage of the switching power supply 10, a full-wave rectification type bridge rectifier circuit Di and a smoothing capacitor Ci are provided.
And a rectifying / smoothing circuit is provided. The rectifying / smoothing circuit rectifies and smoothes a commercial AC power supply (AC input voltage VAC) to obtain a DC voltage Ei, and the DC voltage Ei is input to the switching power supply 10. I have. The switching power supply 10 outputs a DC output voltage EO (EO1, EO2, EO4, EO5) converted to a predetermined voltage level.
In this case, for example, the DC output voltage EO1 is a drive voltage for driving the horizontal deflection circuit of the television receiver,
O2 is a drive voltage for driving the signal circuit, a DC output voltage EO
4, EO5 are driving voltages for driving the audio output circuit. As the actual voltage levels of the DC output voltages EO1, EO2, EO4, EO5, for example, the DC output voltage EO1 is 135V, the DC output voltage EO2 is 15V, Output voltage EO4, EO5 is ± 2
0V.

The horizontal output circuit 20 generates a horizontal deflection current IDY for scanning the electron beam emitted from the electron gun of the CRT in the horizontal direction, and generates a high voltage in a high voltage generation circuit 40 described later. It is configured to generate a flyback pulse.

For this reason, a horizontal synchronizing signal fH (15.73) of a video signal is supplied from a horizontal drive circuit (not shown) to the base of the horizontal output transistor Q11 of the horizontal output circuit 20.
45 kHz). The collector is a flyback transformer FB to be described later.
T is connected to a secondary output terminal (secondary DC output voltage EO1) of the switching power supply 10 via a primary winding N11 of T.
Its emitter is grounded. A damper diode D is provided between the collector and the emitter of the horizontal output transistor Q11.
11, horizontal retrace capacitor Cr11, and [horizontal deflection yoke H. DY, a horizontal straight line correction coil HLC, and an S-shaped correction capacitor CS1] are connected in parallel.

The horizontal output circuit 20 having such a configuration is
Then, the capacitance of the horizontal retrace capacitor Cr11,
A voltage resonance type converter is formed by the leakage inductance component of the primary winding N11 of the flyback transformer FBT. The horizontal output transistor Q11 performs a switching operation by a pulse voltage input from a horizontal drive circuit (not shown), so that the horizontal deflection yoke H.H. A horizontal deflection current IDY having a sawtooth waveform flows through DY, and during a period in which the horizontal output transistor Q11 is turned off,
Horizontal deflection yoke H. Due to the resonance operation of the inductance LDY of DY and the capacitance of the horizontal retrace capacitor Cr11, and the action of the damper diode D11, a relatively high voltage pulse voltage (flyback pulse voltage) is applied across the horizontal retrace capacitor Cr11. V11 occurs. Although the detailed description is omitted, the horizontal linear correction coil HL
The C and S-shaped correction capacitor CS1 corrects, for example, the horizontal deflection current IDY to correct the distortion of the image displayed on the screen of the CRT.

A high-voltage generating circuit 40 shown by a dashed line.
Is a flyback transformer FBT (Fly Back Tra
nsformer) and a high-voltage rectifying / smoothing circuit, and boosts the flyback pulse voltage V11 generated by the horizontal output circuit 20 to generate a high voltage corresponding to the anode voltage level of the CRT.

A primary winding N11 is wound on the primary side of the flyback transformer FBT, and five sets of boost windings NVH1, NVH2, NVH3, NHV4, NHV are wound on the secondary side thereof.
HV5 is divided and wound by slit winding or interlayer winding. In addition, flyback transformer FBT
On the primary side, tertiary windings N12 and N13 are wound in a tightly coupled state with the primary winding N11. In this case, the polarity (winding direction) of each of the boost windings NHV1 to NHV5 with respect to the primary winding N11.
Are wound so as to have opposite polarities, and the tertiary windings N12, N
13 is wound so as to have the same polarity as the primary winding N11.

The winding start end of the primary winding N11 is connected to a secondary output terminal (DC output voltage EO1) of the switching power supply 10, and the winding end is connected to the collector of the horizontal output transistor Q11. Further, high voltage rectifier diodes DHV1, DHV2, DHV
The anodes of 3, DHV4 and DHV5 are connected. And
The cathode of the high-voltage rectifier diode DHV1 is connected to the positive terminal of the smoothing capacitor COHV, and the cathodes of the high-voltage rectifier diodes DHV2 to DHV5 are connected to the respective boost windings N.
Connected to the winding start ends of HV1 to NHV4.

That is, on the secondary side of the flyback transformer FBT, [the boost winding NHV1, the high-voltage rectifier diode DHV
1], [boost winding NHV2, high voltage rectifier diode DHV2],
[Step-up winding NHV3, high-voltage rectifier diode DHV3], [Step-up winding NHV4, high-voltage rectifier diode DHV4], [Step-up winding N
HV5, high-voltage rectifier diode DHV5] are connected in series to form a so-called multisingler type half-wave rectifier circuit.

Accordingly, on the secondary side of the flyback transformer FBT, these five sets of half-wave rectifier circuits rectify the current induced in the boost windings NVH1 to NHV5 and charge the smoothing capacitor COHV. Is performed, the boosting windings NHV1 to NHV are connected to both ends of the smoothing capacitor COHV.
The DC high voltage EHV at a level corresponding to five times the induced voltage induced at 5 is obtained. Then, the DC high voltage EHV obtained at both ends of the smoothing capacitor COHV is used, for example, as an anode voltage of a CRT. An induced voltage boosted to 6 KV is obtained in each of the boost windings NHV1 to NHV5.
A KV anode voltage is obtained.

The primary winding N11 of the flyback transformer FBT is provided with a tap, and a positive pulse voltage obtained from the tap is rectified by a half-wave rectifying / smoothing circuit including a rectifying diode DO3 and a smoothing capacitor CO3. By performing the smoothing, a DC output voltage EO3 is obtained from both ends of the smoothing capacitor CO3. The DC output voltage EO3 has a voltage level of, for example, 200 V, and is supplied to a cathode electrode of a cathode ray tube via a video signal amplifier (not shown).

The negative pulse voltage obtained from the tertiary winding N12 wound on the primary side of the flyback transformer FBT is supplied to a rectifying / smoothing circuit comprising a rectifying diode DO6 and a smoothing capacitor CO6, and a rectifying diode DO7 to a smoothing circuit. DC output voltages EO6 and EO7 are obtained from both ends of the smoothing capacitors CO6 and CO7, respectively, by performing rectification and smoothing by a rectifying and smoothing circuit including a capacitor CO7. The voltage level of the DC output voltages EO6 and EO7 is +
15 V and -15 V, which are used as drive voltages for a vertical deflection circuit (not shown). Further, the negative pulse voltage obtained from the tertiary winding N13 is rectified and smoothed by a rectifying / smoothing circuit including a rectifying diode DO8 and a smoothing capacitor CO8, so that a DC output voltage EO8 is obtained from both ends of the smoothing capacitor CO8. . The DC output voltage EO8 is, for example, 6.3 V, and is used as a heater voltage for the CRT.

FIG. 16 shows a configuration of a switching converter circuit which is currently frequently used as the switching power supply 10 shown in FIG. The switching converter circuit shown in FIG. 16 employs a configuration including a current resonance type switching converter in which two switching elements (bipolar transistors) Q21 and Q22 are half-bridge-coupled. The DC input voltage Ei is input.

The gates of the switching elements Q21 and Q22 are connected to the oscillation / drive circuit 21. The drain of the switching element Q21 is connected to the line of the DC input voltage Ei, and the source is connected to the series resonance capacitor Ci.
1. The insulation converter transformer PIT is grounded to the primary side ground via the primary side winding N21. Clamp diodes DD11 and DD12 are connected in parallel between the drain and source of each of the switching elements Q21 and Q22. Here, for the switching elements Q21 and Q22,
As shown, the series resonance capacitor C1 is connected in series to the primary winding N21 of the insulated converter transformer PIT, so that the capacitance of the series resonance capacitor C1 and the insulation converter transformer PI including the primary winding N21 are connected.
The leakage inductance component of T forms a primary side series resonance circuit for making the operation of the switching converter a current resonance type.

The switching elements Q21 and Q22 are driven by the oscillation / drive circuit 21 so that a switching operation is obtained. Therefore, the control circuit 1 supplies a current or voltage of a level corresponding to the fluctuation of the DC output voltage EO1 to the oscillation / drive circuit 21. Then, in the oscillation / drive circuit 21, the switching drive signal (voltage) whose cycle is varied according to the output level from the control circuit 1 is switched to the switching elements Q21 and Q21 so that the DC output voltage EO1 is stabilized. Output is made to the gate of Q22. As a result, the switching frequency of the switching elements Q21 and Q22 is varied, and the DC output voltage EO1 is stabilized. Starting resistance RS
Is provided to supply a starting current obtained in the rectifying / smoothing line to the oscillation / drive circuit 21 when the commercial AC power is turned on.

The insulating converter transformer PIT transmits the switching output of the switching elements Q21 and Q22 to the secondary side. Although a detailed description is omitted here, a loose coupling state can be obtained by forming a gap with respect to the core in the insulating converter transformer PIT. On the secondary side of the insulation converter transformer PIT,
As shown, three sets of secondary windings N22, N23, N24 are wound independently of each other. Then, two rectifier diodes DO11 and DO12 and a smoothing capacitor CO1 are connected to the secondary winding N22 in a connection form as shown in the drawing, so that the DC output voltage EO1 (1
35V) and two rectifier diodes DO21, DO22 and a smoothing capacitor CO2 are connected to the secondary winding N23 in the same connection form so that the DC output voltage EO2 (15V) of the signal system circuit is obtained. ing. Further, two sets of rectifying / smoothing circuits including a rectifying / smoothing circuit including rectifying diodes DO41 and DO42 and a smoothing capacitor CO4 and a rectifying / smoothing circuit including rectifying diodes DO51 and DO52 and a smoothing capacitor CO5 are illustrated in the secondary winding N24. By connecting in this manner, the DC output voltages EO4 (+20 V) and EO5 (-20 V) for the audio output circuit are obtained.

The operation waveforms of each part of the circuit shown in FIG. 15 are as shown in FIG. In the circuit shown in FIG. 15, a pulse voltage synchronized with the horizontal synchronizing signal fH (15.7345 kHz) of the video signal is input to the base of the horizontal output transistor Q11. , Horizontal synchronization signal f
H, the ON period (horizontal scanning period) Tt of the horizontal output transistor Q11 is 51.5 μs, and the OFF period (horizontal retrace period) Tr is 12 μs. That is, the period TH (63.5 μS) obtained by adding the horizontal scanning period Tt and the horizontal scanning period Tr corresponds to the cycle of the horizontal synchronization signal fH.

In this case, due to the switching operation of the horizontal output transistor Q11, a primary current I11 having a waveform as shown in FIG. . A horizontal deflection current IDY having a waveform as shown in FIG. Also, a rectified current I3 having a waveform as shown in FIG. 17E flows through the rectifier diode DO3 via a tap provided on the primary winding N11.

At this time, the voltage V11 across the horizontal retrace capacitor Cr11 connected between the collector and the emitter of the horizontal output transistor Q11 is, as shown in FIG.
During the period Tt when the horizontal output transistor Q11 is on, 0
Level, and during the period Tr during which the horizontal output transistor Q11 is turned off, the horizontal deflection yoke H.H. The flyback pulse voltage V11 of, for example, about 1200 Vp is generated by the resonance operation of the inductance component LDY of DY and the capacitance of the horizontal retrace capacitor Cr11. Therefore, in the high-voltage generating circuit 40 shown in FIG. 15, the flyback transformer FBT is driven by the flyback pulse voltage V11 as described above.
The positive pulse voltage applied to the primary side is boosted to obtain various DC output voltages having a predetermined voltage level from the secondary side boosting windings NHV1 to NHV5 and the tertiary windings N12 and N13. I have.

Further, as shown in FIG. 17D, a pulse voltage V3 of, for example, about 200 Vp is generated at both ends of the smoothing capacitor CO3 during the period Tr in which the horizontal output transistor Q11 is turned off. DC output voltage EO3 is obtained by rectifying and smoothing V3 with a rectifying diode DO3 and a smoothing capacitor CO3.

[0023]

The circuit shown in FIG. 15 is used for converting the DC voltage EO1 input from the switching power supply 10 into the DC high voltage EHV in the flyback transformer FBT of the high voltage generating circuit 40. Since the power conversion efficiency is about 85%, for example, when the high-voltage load power is 60 W, a loss power of about 9 W is generated.

In the high-voltage generation circuit 40, the positive side pulse voltage (flyback pulse voltage) input to the primary winding N11 of the flyback transformer FBT induces the boosting windings NHV1 to NVV5 on the secondary side. The peak value of the applied current is half-wave rectified to obtain a high DC voltage EHV. However, in this case, the conduction angle of the high-voltage rectifier diodes DHV1 to DHV5 is narrow, and equivalently, the power supply impedance is high. Therefore, there is a disadvantage that the voltage level of the DC high voltage EHV is easily affected by the fluctuation of the high-voltage load. .

For example, when the circuit shown in FIG. 15 is applied to a television receiver having a CRT screen size of 34 inches or more, the CRT screen brightness is set to the brightest state (highlight) by, for example, It is necessary to supply a beam current IHV of 2 mA or more to the anode electrode of the CRT. Therefore, for example, when the voltage level of the DC high voltage EHV supplied to the anode electrode is 30 KV, 60 W (30 KV × 2 mA) is required as the high-voltage load power applied to the high-voltage generation circuit 40 in the highlight state. Therefore, the high voltage load power supplied from the high voltage generation circuit 40 to the anode electrode of the CRT is at least 0 W (IHV = 0 m
A) It is conceivable to vary from 60 W (IHV = 2 mA).

In this case, assuming that a beam current IHV of 2 mA flows to the anode electrode of the CRT and the voltage level of the DC high voltage EHV is 30 KV when the high voltage load power of the high voltage generating circuit 40 is 60 W, a high voltage is generated. When the high-voltage load power of the circuit 40 becomes the no-load state (0 W), for example, the voltage level of the DC high voltage EHV rises to 32.5 KV. That is, when the circuit shown in FIG. 15 is applied to an actual television receiver or the like, the voltage fluctuation width ΔEHV of the DC high voltage EHV in the actual use range (0 W to 60 W) of the high-voltage load power is about 2.5 KV become. This is because the high-voltage load power applied to the high-voltage generation circuit 40 fluctuates and the high-voltage rectifier diodes DHV1 to DHV1 to
This is due to the voltage drop in DHV5 and the like.

When the voltage level of the DC high voltage EHV fluctuates in this way, for example, when the current value of the horizontal deflection current IDY is constant, the horizontal amplitude of the electron beam output from the CRT changes. For this reason, in an actual television receiver, a zooming correction circuit or the like for correcting the current value of the horizontal deflection current IDY is provided in a horizontal output circuit so that the horizontal amplitude of the electron beam does not change due to the fluctuation of the DC high voltage EHV. 20 had to be provided.

Further, due to the structure of the flyback transformer FBT, the induced voltage induced in each of the boost windings NHV1 to NHV5 becomes a negative level due to, for example, the leakage inductance of the booster windings NVH1 to NVH5 on the secondary side. Causes ringing (vibration). When this ringing component is superimposed on the primary current I11 flowing through the primary side of the flyback transformer FBT shown in FIG. 17B, there is a drawback that raster ringing, curtain stripes, etc. occur on the left end of the screen of the CRT. Was. For this reason, in an actual television receiver, it was necessary to take some measures to prevent such raster ringing and curtain stripes.

[0029]

Therefore, a switching power supply circuit according to the present invention is configured as follows in consideration of the above-mentioned problems. In other words, a switching unit formed with a switching element for intermittently outputting the input DC input voltage, and a primary-side parallel resonance circuit that makes the operation of the switching unit a voltage resonance type are formed. A primary side parallel resonance capacitor is provided, and is provided for transmitting an output of the primary side to the secondary side. A primary side winding is wound on the primary side, and at least a first secondary side is wound on the secondary side. A secondary winding having a side winding part and a second secondary winding part formed so as to be wound up with respect to the first secondary winding is wound, An insulated converter transformer that can obtain the required degree of coupling, which is loosely coupled to the side winding and the secondary winding, and a secondary parallel resonance capacitor connected in parallel to the secondary winding Secondary side parallel resonance circuit formed by connecting , A first DC output voltage is obtained by performing a half-wave rectification operation on a voltage in the positive period of the alternating voltage obtained from the first secondary winding, which is formed including a secondary parallel resonance circuit. The DC output voltage obtained by performing a half-wave rectification operation on the voltage of the alternating voltage obtained from the second secondary winding in a positive period with respect to the first DC output voltage is stacked on the second DC output voltage. DC output voltage generation means configured to obtain the DC output voltage of the first DC output voltage, the switching frequency of the switching element is variably controlled according to the level of the first DC output voltage, and the OFF period in the switching cycle is kept constant. After that, a constant voltage control means is provided which performs constant voltage control by switchingly driving the switching element so as to vary the ON period. And, by transmitting the resonance voltage input to the primary side to the secondary side, a step-up transformer configured to obtain a step-up voltage obtained by boosting the resonance voltage from the secondary side,
In order to make the primary side operation of the step-up transformer a resonance operation, it is inserted between the secondary side winding of the insulating converter transformer and the primary side winding of the step-up transformer, and is obtained from the secondary side winding of the insulating converter transformer. A DC high voltage of a predetermined high voltage level is obtained by performing a rectifying operation on a series resonance capacitor having a negative resonance voltage obtained as a substantially sinusoidal resonance voltage and a boost voltage obtained on the secondary side of the boost transformer. And a DC high voltage generating means configured as described above.

That is, according to the present invention, the negative resonance voltage output from the secondary side of the insulated converter transformer constituting the switching power supply circuit of the complex resonance type is converted into the voltage of the step-up transformer via the series resonance capacitor. By inputting to the primary side, it is possible to prevent the horizontal deflection circuit system from intervening to obtain a DC high voltage required for performing horizontal deflection of the television receiver, for example. At this time, the resonance voltage waveform input to the primary side of the step-up transformer can be made substantially sinusoidal.

[0031]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to an embodiment of the present invention. The power supply circuit shown in this figure has a configuration as a complex resonance type switching converter including a voltage resonance type converter on the primary side and a parallel resonance circuit on the secondary side. In the power supply circuit shown in FIG. 1, first, a full-wave rectifier comprising a bridge rectifier circuit Di and a smoothing capacitor Ci as a rectifying and smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) to obtain a DC input voltage. A smoothing circuit is provided to generate a rectified smoothed voltage (DC input voltage) Ei corresponding to a level that is one time higher than the AC input voltage VAC.

The switching converter which inputs and outputs the DC input voltage Ei is provided with a switching device Q1 having a single element and a voltage resonance type converter which performs a switching operation in a so-called single-end system by a self-excited system. . In this case, the switching element Q1
Uses a high breakdown voltage bipolar transistor (BJT; junction type transistor).

The base of the switching element Q1 is connected to the positive electrode of the smoothing capacitor Ci via the current limiting resistor RB and the starting resistor RS, and the emitter is grounded to the primary side ground. Further, a series resonance circuit for self-excited oscillation driving, which is composed of a series connection circuit of a drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB, is connected between the base of the switching element Q1 and the primary side ground. The clamp diode DD1 inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the switching element Q1 is turned off. Switching element Q
The first collector is connected to one end of a primary winding N1 formed on the primary side of the isolated converter transformer PIT,
Its emitter is grounded.

A parallel resonance capacitor Cr1 is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr1 is
The primary parallel resonance circuit of the voltage resonance type converter is formed by its own capacitance and the leakage inductance L1 of the primary winding N1. Although not described in detail here, when the switching element Q1 is turned off, the voltage V1 across the resonance capacitor Cr1 due to the operation of the primary side parallel resonance circuit actually has a sinusoidal pulse waveform. Thus, a voltage resonance type operation is obtained.

The orthogonal control transformer PRT is a saturable reactor on which a resonance current detection winding ND, a driving winding NB, and a control winding NC are wound. This orthogonal control transformer P
RT drives switching element Q1 and is provided for constant voltage control. This orthogonal control transformer P
As a structure of the RT, although not shown, a three-dimensional core is formed by joining ends of two magnetic legs of two double U-shaped cores having four magnetic legs. Then, a resonance current detection winding ND and a drive winding NB are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is connected to the resonance current detection winding. It is configured to be wound in a direction orthogonal to the winding ND and the driving winding NB.

In this case, the resonance current detecting winding ND of the orthogonal control transformer PRT is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1, so that the switching output of the switching element Q1 is changed. , Through the primary winding N1 to the resonance current detection winding ND. In the orthogonal control transformer PRT, the switching output obtained in the resonance current detection winding ND is induced in the drive winding NB via the transformer coupling, so that an alternating voltage as a drive voltage is applied to the drive winding NB. appear. This drive voltage is
Series resonant circuit (NB, CB) forming self-excited oscillation drive circuit
Is output as a drive current to the base of the switching element Q1 via the base current limiting resistor RB. As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit.

An insulation converter transformer (Power Isolation)
Transformer) PIT transmits the switching output of switching element Q1 to the secondary side. As shown in FIG. 13, the structure of the insulating converter transformer PIT includes an EE-type core in which E-type cores CR1 and CR2 made of, for example, a ferrite material are combined so that their magnetic legs face each other. Split bobbin B for central magnetic leg
The primary winding N1 and the secondary winding N2 are wound in a state of being divided respectively. A gap G is formed for the center magnetic leg as shown in the figure. Thereby, loose coupling with a required coupling coefficient is obtained. The gap G is formed by connecting the central magnetic legs of the E-shaped cores CR1 and CR2 to 2
It can be formed by making it shorter than the outer magnetic leg of the book. The coupling coefficient k is set to a loosely coupled state of, for example, k ≒ 0.85, so that a saturated state is hardly obtained.

By the way, the insulation converter transformer PIT
Of the secondary winding includes a primary winding N1 and a secondary winding N2.
Of the primary winding N1 and the secondary winding N1 by the polarity (winding direction) of the rectifier diode D0 and the polarity of the alternating voltage excited in the secondary winding.
2 with respect to the mutual inductance M with the inductance L2, the operation mode of + M (polarization mode; forward operation) and the operation mode of -M (depolarization mode;
Flyback operation). For example, FIG.
Mutual inductance is + M when equivalent to the circuit shown in FIG. 14A, and −M when equivalent to the circuit shown in FIG. 14B. In addition, in the power supply circuit shown in FIG.
The polarity of the primary winding N1 and the secondary windings N2, N3 of T is + M
During the operation mode of the rectifier diode DO
1. The charging operation of the smoothing capacitors CO1 and CO3 is performed via DO3.

The winding start end of the primary winding N1 of the insulated converter transformer PIT is connected to the collector of the switching element Q1 as shown in FIG. 1, and the winding end is connected in series with the resonance current detection winding ND. Connected to the positive electrode of the smoothing capacitor Ci. On the secondary side, a secondary winding N2 serving as a first secondary winding, and a second secondary side formed by winding up a winding end portion of the secondary winding N2. A tertiary winding N3 is provided as a winding. And
A secondary side parallel resonance capacitor C2 is connected to a secondary side winding (N2 + N3) including the secondary winding N2 and the tertiary winding N3.
Are connected in parallel.

In this case, the winding start end of the secondary winding N2 is connected to the secondary side ground, and the winding end is connected to the rectifier diode D2.
Connected to the anode of O1. The half-wave rectifying and smoothing circuit including the rectifying diode DO1 and the smoothing capacitor CO1 obtains a DC output voltage EO1 (for example, 135 V) for horizontal deflection of 100 V to 140 V.

A tap is provided at a required position of the secondary winding N2, and a rectifier diode DO2
The anode is connected. Then, a half-wave rectifying and smoothing circuit including the rectifying diode DO2 and the smoothing capacitor CO2 provides a DC output voltage EO2 for a signal system circuit.
(15V).

Further, on the secondary side of the insulated converter transformer PIT, the winding end of the tertiary winding N3 wound around the secondary winding N2 is connected to the anode of the rectifying diode DO3, whereby A DC output voltage EO3 (200 V) for a video output circuit is obtained by a half-wave rectifier circuit composed of DO3 and a smoothing capacitor CO3. In the power supply circuit shown in FIG. 1, the negative side of the smoothing capacitor CO3 is connected to a smoothing capacitor. By connecting to the positive electrode side of CO1, a DC output voltage EO3 for the video output circuit is obtained from both ends of the series connection circuit of the smoothing capacitors CO1-CO3. That is, in the power supply circuit shown in FIG. 1, in order to obtain the DC output voltage EO3 for the video output circuit, the DC output voltage EO1 generated across the smoothing capacitor CO1 is applied to the DC output voltage generated across the smoothing capacitor CO3. Pile up,
That is, the DC output voltage EO1 obtained from the secondary winding N2,
The DC output voltage EO3 is obtained by superimposing the DC output voltage obtained from the tertiary winding N3. For this reason,
The configuration of the rectifying / smoothing circuit including the tertiary winding N3, the rectifying diode DO3, and the smoothing capacitor CO3 includes a DC output voltage EO3 (200 V), a DC output voltage EO1 (110 V
１４０140 V) minus 90 V to 60 V DC output voltage.

In this case, the secondary side parallel resonance capacitor C2 is connected in parallel to the secondary side winding (N2 + N3) composed of the secondary winding N2 and the tertiary winding N3, so that the secondary side is connected. Leakage inductance of side winding (N2 + N3) (L2 +
L3) and the capacitance of the secondary parallel resonance capacitor C2 form a secondary parallel resonance circuit. The alternating voltage induced in the secondary winding N2 and the tertiary winding N3 becomes a resonance voltage, and the isolated converter transformer PIT , A voltage resonance operation is obtained on the secondary side.

That is, in the power supply circuit shown in FIG. 1, a primary side is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and a secondary side is provided with a parallel resonance circuit for obtaining a voltage resonance operation. Be provided. In the present specification, a switching converter having a configuration in which a resonance circuit is provided for the primary side and the secondary side and operates as described above is described below.
It is also referred to as a “composite resonance type switching converter”. Note that the configuration as such a composite resonance type switching converter is as described above with reference to FIG.
This is realized by forming a gap G with respect to the insulating converter transformer PIT and performing loose coupling by a required coupling coefficient, thereby obtaining a state in which a saturation state is more unlikely to be obtained. For example, isolation converter transformer PI
If no gap G is provided for T, there is a high possibility that the insulation converter transformer PIT will be in a saturated state during flyback operation and the operation will be abnormal, and the rectification operation on the secondary side will be performed properly. It's hard to want.

The above-described DC output voltage EO1 is also branched and input to the control circuit 1. The control circuit 1 varies the level of the control current (DC current) flowing through the control winding NC of the orthogonal control transformer PRT in accordance with the change in the DC output voltage level EO1 on the secondary side.
The inductance LB of the drive winding NB wound around T is variably controlled. As a result, the resonance condition of the series resonance circuit in the self-excited oscillation driving circuit for the switching element Q1 formed including the inductance LB of the driving winding NB changes, so that the switching frequency of the switching element Q1 is changed. The operation becomes variable, and this operation stabilizes the DC output voltage output from the secondary side of the insulating converter transformer PIT. Note that the DC output voltage EO3 may be branched and input to the control circuit 1 to make the DC output voltage constant.

When an orthogonal control transformer PRT for variably controlling the inductance LB of the drive winding NB is provided as in the power supply circuit shown in FIG. 1, the switching element Q1 is turned off to change the switching frequency. After the period TOFF is fixed, the ON period TON is variably controlled. In other words, in the power supply circuit shown in FIG. 1, as the constant voltage control operation, the switching frequency is variably controlled to perform the resonance impedance control on the switching output, and at the same time, the conduction of the switching element Q1 in the switching cycle. It can be seen that the angle control (PWM control) is also performed. This complex control operation is realized by a set of control circuit systems. In the present specification, such complex control is also referred to as “complex control method”.

Further, in the power supply circuit shown in FIG. 1, the winding termination of the tertiary winding N3 is connected to the winding of the primary winding N4 of the flyback transformer FBT provided in the high-voltage generating circuit 4 via the series resonance capacitor C3. It is connected to the start end. That is, the power supply circuit shown in FIG.
Secondary winding (N2 + N3) and series resonance capacitor C3
And the series connection circuit of the primary winding N4 of the flyback transformer FBT are connected in parallel.

In the power supply circuit having such a configuration, the isolated converter transformer PIT operates as a complex resonance type switching converter, so that a resonance pulse voltage is generated across the secondary side parallel resonance capacitor C2. Then, the DC output voltage E03 is obtained from the positive resonance pulse voltage generated during the positive period when the insulating converter transformer PIT performs the forward operation, and the negative resonance pulse voltage generated during the negative period during the flyback operation is changed to the series resonance. Flyback transformer FB via capacitor C3
T is inputted to the primary winding N4. In this case, a current resonance circuit is formed on the primary side of the flyback transformer FBT by a series connection circuit of the capacitance of the DC resonance capacitor C3 and the inductance of the primary winding N4. A substantially sinusoidal resonance current I4 flows through the side winding N4, and the voltage V4 across the primary winding N4 also becomes a substantially sinusoidal resonance voltage.

High-voltage generating circuit 4 enclosed by a dashed line
Is configured by a flyback transformer FBT and a high-voltage rectifier circuit, and boosts the resonance voltage V4 generated in the primary winding N4 of the flyback transformer FBT, for example, by C
A high voltage corresponding to the anode voltage level of RT is generated. For this reason, on the secondary side of the flyback transformer FBT, four to five sets of step-up windings NHV are divided and wound by slit winding or interlayer winding. In this case, the primary winding N4 and the step-up winding NHV are wound so as to be tightly coupled, and furthermore, are wound so that their polarities (winding directions) are opposite. Therefore, the flyback transformer F
On the secondary side of the BT, the negative resonance voltage of the resonance voltage V4 generated in the primary winding N4 is inverted, and the turn ratio (NHV / N4) between the boost winding NHV and the primary winding N4. As a result, a boosted voltage is obtained.

Here, FIG. 3 shows a flyback transformer FB.
A cross-sectional view of T is shown, and the structure of the flyback transformer FBT will be described with reference to FIG. The flyback transformer FBT shown in this figure has, for example, two U-shaped cores C.
The square core CR30 is formed by combining the magnetic legs of R1 and CR2 so as to face each other. A gap G is provided at a portion where the end of the U-shaped core CR1 and the end of the U-shaped core CR2 face each other. Then, as shown in the figure, by attaching the low-voltage winding bobbin LB and the high-voltage winding bobbin HB to one of the magnetic legs of the square core CR30, the low-voltage winding bobbin LB and the high-voltage winding bobbin HB are attached. The primary winding N4 and the boost winding NHV are respectively divided and wound. In this case, the primary winding N4 is wound around the low-voltage winding bobbin LB, and a plurality of boosting windings NHV are wound around the high-voltage winding bobbin HB by inserting and winding up the interlayer film F. become.

In the power supply circuit shown in FIG. 1, the resonance voltage V4 input to the primary winding N4 of the flyback transformer FBT is obtained from the secondary side of the insulating converter transformer PIT. The frequency corresponds to the switching frequency of the switching element Q1, and is, for example, in the range of about several tens of kHz to about 200 kHz. For this reason, in the present embodiment, a litz wire is used for the primary winding N4 of the flyback transformer FBT to prevent eddy current from being generated in the primary winding N4.

In this embodiment, the case where the boost winding NHV of the flyback transformer FBT is wound by interlayer winding is shown, but the winding of the boost winding NHV is limited to the interlayer winding. Instead, for example, the high-pressure bobbin HB may be divided into a plurality of regions, and the winding may be wound by a so-called divided winding in which the boosting winding NHV is wound around each divided region. That is, as the structure of the flyback transformer FBT, a plurality of step-up windings NHV wound around the high-voltage bobbin HB may be wound in a state in which they are insulated.

In the power supply circuit shown in FIG. 1, on the secondary side of the flyback transformer FBT, five sets of boost windings NVH1, NHH are provided.
V2, NHV3, NHV4, and NHV5 are wound independently of each other. High-voltage rectifier diodes DHV1, DHV2, and DHV are provided at the winding ends of the respective boost windings NVH1 to NVH5.
3, the anode sides of DHV4 and DHV5 are connected. The cathode of the high-voltage rectifier diode DHV1 is connected to the positive terminal of the smoothing capacitor COHV, and the remaining cathodes of the high-voltage rectifier diodes DHV2 to DHV5 are connected to the boost winding N
Connected to the winding start ends of HV1 to NHV4.

That is, on the secondary side of the flyback transformer FBT, [the boost winding NHV1, the high-voltage rectifier diode DHV
1], [boost winding NHV2, high voltage rectifier diode DHV2],
[Step-up winding NHV3, high-voltage rectifier diode DHV3], [Step-up winding NHV4, high-voltage rectifier diode DHV4], [Step-up winding N
HV5, high-voltage rectifier diode DHV5], that is, a so-called multisingler type half-wave rectifier circuit in which five sets of half-wave rectifier circuits are connected in series.

Therefore, on the secondary side of the flyback transformer FBT, five sets of half-wave rectifier circuits
~ NHV5 is rectified to smooth the capacitor CO
An operation of charging the HV is performed, and a high DC voltage E of a level corresponding to about five times the induced voltage induced in each of the boost windings NVH1 to NHV5 is provided across the smoothing capacitor COHV.
HV will be obtained. Then, the DC high voltage EHV obtained at both ends of the high voltage converter COHV is, for example, CR
It is used as the anode voltage of T.

A capacitor COFV is inserted between the cathode of the high-voltage rectifier diode DHV4 and the secondary-side ground. EFV
Is output as a focus voltage to, for example, a fourth grid (focus electrode) of the CRT.

Here, for example, 30K
When the power supply circuit shown in FIG. 1 is actually constructed so as to obtain a DC high voltage EHV of V, the secondary winding N2 of the insulating converter transformer PIT = 45T and the tertiary winding N3 = 22
T, secondary parallel resonance capacitor C2 = 2700 PF, series resonance capacitor C3 = 3900 PF, primary winding N4 of the flyback transformer FBT = 30T, boost winding NHV1
ＮNHV5 = 460T is selected.

FIG. 2 shows operation waveforms of the power supply circuit shown in FIG. 2A to 2F, for example, when the AC input voltage VAC is 100 V and the high-voltage load power of the high-voltage generation circuit 4 is the maximum load power Pomax = 60 W (IHV = 2 mA)
FIG. 2 (g) to FIG.
(L) shows an operation waveform under the condition that the high-voltage load power of the high-voltage generation circuit 4 is the minimum load power Pomin = 0W (IHV = 0mA), for example.

When the high-voltage load power of the high-voltage generating circuit 4 is set to the maximum load power, the switching frequency of the switching element Q1 is controlled to, for example, 90.9 kHz, and the actual on / off period TON of the switching element Q1 is turned on.
/ TOFF is 7.4 μs / 3.6 μs. As shown in FIG. 2A, the resonance voltage V1 generated at both ends of the parallel resonance capacitor Cr1 by the on / off operation of the switching element Q1 is a sinusoidal pulse during the period TOFF when the switching element Q1 is off. Is obtained, which indicates that the operation of the switching converter is of the voltage resonance type. At this time, a collector current IC as shown in FIG. 2B flows through the switching element Q1. For example, when the switching element Q1 is turned on, a damper current (negative direction) flows to the primary winding N1 via the clamp diode DD1 and the base-collector of the switching element Q1, and a damper period (for example, 1.4 μs) in which the damper current flows. Is completed, the collector current IC rapidly rises in level from the negative level to the positive level.

By such an operation, the secondary winding (N 2 + N 3) of the insulating converter transformer PIT is connected to the circuit shown in FIG.
As shown in FIG. 2C, the resonance current I2 flows as shown in FIG. 2D, and the voltage V2 generated in the secondary-side parallel resonance capacitor C2, as shown in FIG.
Due to the operation of the rectifier diodes DO1, DO3, the voltage level becomes a positive voltage level of 200 V, and during a period TOFF during which the switching element Q1 is turned off, a negative resonance pulse voltage whose peak voltage level is 500 Vp Become. The negative resonance pulse voltage generated in the secondary resonance capacitor C2 is applied to the primary winding N4 via the series resonance capacitor C3 forming a series resonance circuit together with the primary winding N4 of the flyback transformer FBT. 2E, the resonance voltage V4 generated at both ends of the primary winding N4 becomes a resonance voltage waveform having a peak voltage level of 400 Vp, as shown in FIG. The resonance current I4 flowing through the winding N4 is shown in FIG.
As shown in the figure, the resonance current waveform has a peak value of 2 Ap.

On the other hand, when the high-voltage load power of the high-voltage generation circuit 4 is set to the minimum load power (no load), the switching frequency of the switching element Q1 is controlled to, for example, 100 kHz, and the actual switching element Q1 is turned on. /
6.4 μs / 3.6 μs as the OFF period TON / TOFF
Becomes In this case, the resonance voltage V1 generated across the parallel resonance capacitor Cr1 by the on / off operation of the switching element Q1, as shown in FIG.
In the period TOFF during which the switching element Q1 is turned off, a sine-wave-like waveform is obtained. At this time, a collector current IC as shown in FIG. 2H flows through the switching element Q1, but in this case, the switching element Q1
The period of the damper current flowing at the time of turn-on is about 2 μs
The damper period is longer than the damper period (1.4 μs) at the time of the maximum load power shown in FIG.

By such an operation, the secondary winding (N2 + N3) of the insulating converter transformer PIT is connected to the circuit shown in FIG.
As shown in FIG. 2 (j), the resonance current I2 flows, and the voltage V2 generated across the secondary-side parallel resonance capacitor C2 is as shown in FIG.
As shown in (i). The negative resonance pulse generated in the secondary resonance capacitor C2 is input to the primary winding N4 of the flyback transformer FBT via the series resonance capacitor C3, and the waveform of the voltage V4 across the primary winding N4 is 2 (k), and the current waveform of the current I4 flowing through the primary winding N4 has a resonance waveform as shown in FIG. 2 (l).

FIGS. 2A to 2F and FIG. 2G
2 (l), the power supply circuit shown in FIG. 1 is configured such that the high-voltage load power of the high-voltage generation circuit 4 is changed from the maximum load power Pomax to the minimum load power P.
Omin, the switching frequency of the switching element Q1 changes from 90.9 kHz to 100 kHz. That is, the power supply circuit shown in FIG.
, The switching frequency of the primary-side switching converter is variably controlled in accordance with the fluctuation of the high-voltage load. This means that the alternating voltage cycle of the flyback transformer FBT (the switching frequency of the rectifier diode in the high-voltage generating circuit 4) is varied depending on the switching frequency of the primary-side switching converter.

A comparison between the power supply circuit having such a configuration shown in FIG. 1 and the conventional circuit shown in FIG. 15 shows that the conventional circuit shown in FIG. The back pulse voltage V11 is stepped up to obtain the DC high voltage EHV from the high voltage generation circuit 40. On the other hand, in the circuit shown in FIG. 1, the resonance pulse voltage V2 generated at both ends of the secondary side parallel resonance capacitor C2 is obtained.
Is directly input to the series high-voltage generation circuit 4 via the series resonance capacitor C3, so that the DC output voltage EO1 of the switching power supply 10 is changed like the conventional circuit shown in FIG.
Horizontal output circuit 20 for converting a signal into a flyback pulse voltage
, The DC high voltage EHV is obtained in the high voltage generation circuit 4. Also, a composite resonance type switching is provided in which a primary resonance side of the insulating converter transformer PIT is provided with a parallel resonance circuit for making the switching operation a voltage resonance type, and a secondary side is provided with a parallel resonance circuit for obtaining the voltage resonance operation. It is composed of a converter.

In this case, the conventional circuit shown in FIG.
While the voltage conversion efficiency ηDC-DC is about 85%,
In the power supply circuit shown in FIG. 1, the switching converter has a complex resonance type, and its voltage conversion efficiency ηDC-DC is about 95%.
Therefore, the power supply circuit shown in FIG. 1 can reduce power loss.

In the conventional circuit shown in FIG.
The current-resonant converter constituting the switching power supply 10 has an AC-DC power conversion efficiency ηAC-DC of about 90%, whereas the power supply circuit shown in FIG.
Since the DC power conversion efficiency ηAC-DC is about 92%, comparing the power supply circuit shown in FIG. 1 with the conventional circuit shown in FIG. 15, the power supply circuit shown in FIG. The power can be reduced, and energy can be saved.

Further, in the power supply circuit shown in FIG.
Since a series resonance circuit is provided to make the primary side operation of the flyback transformer FBT a current resonance type,
The resonance voltage V4 input to the primary winding N4 of the flyback transformer FBT has a resonance voltage waveform as shown in FIGS. 2 (e) and 2 (k), and the flyback transformer FB
The induced voltage induced in the T step-up windings NHV1 to NHV5 also has a resonance voltage waveform. Therefore, when such an induced voltage is rectified by the high-voltage rectifier diodes DHV1 to DHV5, the conduction angle of each of the high-voltage rectifier diodes DHV1 to DHV5 increases,
Equivalently, the power supply impedance can be lowered,
Even when the high-voltage load power varies from 0 W to 60 W, it is possible to suppress the variation in the voltage level of the DC high voltage EHV.

FIG. 7 is a diagram showing the relationship between the DC high voltage output from the high voltage generation circuit and the high voltage load power when the power supply circuit shown in FIG. 1 is applied to an actual television receiver. As shown in FIG. 7, for example, if the voltage level of the DC high voltage EHV when the high voltage load power is 60 W is 30 KV, the voltage level of the DC high voltage EHV when the high voltage load power is 0 W is It becomes about 31.5 KV.
That is, in the power supply circuit shown in FIG.
Even when the voltage fluctuates from W to 0 W, the voltage fluctuation width ΔEHV of the DC high voltage EHV is 1.5 KV, and the fluctuation width ΔEHV is reduced as compared with the conventional circuit (ΔEHV = 2.5 KV) shown in FIG. It becomes possible.

Therefore, when the DC high voltage EHV is supplied from the power supply circuit according to the present embodiment shown in FIG. 1 to the anode electrode of the CRT, the fluctuation in the horizontal amplitude of the electron beam output from the CRT is reduced. Since the luminance can be suppressed, when the luminance of the cathode ray tube surface is low, a screen missing or the like does not occur, and it is not necessary to provide a zooming correction circuit or the like in the horizontal output circuit of the television receiver.

The switching frequency of the flyback transformer FBT corresponds to the switching frequency of the switching element Q1, and is not synchronized with, for example, the cycle of the horizontal synchronizing signal fH of the video signal. As a result, due to the leakage magnetic flux and leakage inductance from the flyback transformer FBT, the flyback transformer FBT
Ringing does not occur in the induced voltages of the step-up windings NHV1 to NHV5. Therefore, for example, raster ringing does not occur on the screen of the CRT, and even if ringing occurs, the horizontal deflection current IDY is formed independently of the horizontal deflection circuit of the high-voltage generation circuit 4.
Without the ringing current component being superimposed on
There is also an advantage that raster ringing and curtain stripes can be prevented from being generated on the screen of the CRT.

Further, in the conventional circuit shown in FIG. 15, 73T (turn) is required as the number of turns of the primary winding N4 of the flyback transformer FBT, whereas
In the power supply circuit shown in FIG. 1, since the number of turns of the primary winding N4 can be reduced to 30T and, for example, the operating frequency under no load increases to about 100 KHz,
Ferrite core of flyback transformer FBT (core)
Can be reduced to about 1/2 of the conventional one.

Further, as shown in FIG. 16, the switching power supply 10 provided in the conventional circuit shown in FIG. 15 includes a DC output voltage EO1 for the horizontal deflection circuit output from the secondary side of the insulating converter transformer PIT. Each of the DC output voltage EO2 for the signal system circuit and the DC output voltages EO4 and EO5 for the audio output circuit is obtained by a double-wave rectification method using a center tap. For this reason, the number of pins of the winding bobbin for winding the secondary winding of the insulating converter transformer PIT is required to be nine, but in the power supply circuit shown in FIG. 1, the secondary side of the insulating converter transformer PIT is required. Since the number of pins of the winding bobbin for winding the winding is only four, there is an advantage that the structure of the insulating converter transformer PIT can be simplified accordingly.

Further, the power supply circuit of the present invention is not limited to the circuit configuration shown in FIG. FIG. 4 is a diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention. The same parts as those of the power supply circuit shown in FIG. The power supply circuit shown in FIG. 4 is provided with a parallel resonance circuit on the primary side of the insulated converter transformer PIT for making the switching operation a voltage resonance type, and on the secondary side, similarly to the power supply circuit shown in FIG. The composite resonance type switching converter includes a parallel resonance circuit for obtaining a voltage resonance operation. However, FIG.
The power supply circuit shown in FIG. 4 includes a high-voltage generation circuit 4 having a flyback transformer FBT to obtain a high DC voltage EHV, whereas the power supply circuit shown in FIG. A different point is that a high-voltage generating circuit 5 is provided.

In this case, the insulation converter transformer PIT
Of the tertiary winding N3 is connected to the winding start end of the primary winding N4 of the step-up transformer HVT via a series resonance capacitor C3, and a sinusoidal resonance current is applied to the primary winding N4 of the step-up transformer HVT. By flowing, the voltage V4 across the primary winding N4 has a sinusoidal resonance voltage waveform.

High-voltage generating circuit 5 enclosed by a dashed line
Is configured by a step-up transformer HVT and a high-voltage rectifier circuit, and boosts the resonance voltage V4 generated in the primary winding N4 of the step-up transformer HVT to generate a high voltage corresponding to, for example, an anode voltage level of a CRT. To be.
Therefore, for example, the secondary side of the step-up transformer HVT
A pair of boost windings NHV11 and NHV12 are divided and wound by slit winding or interlayer winding. Also in this case, the primary winding N4 and the boost windings NHV11 and NHV12 are tightly coupled and wound so that their polarities (winding directions) are opposite to each other. A boosted voltage is obtained by the turn ratio (NHV / N4) between each boost winding NHV and the primary winding N4.

FIG. 6 is a sectional view of the step-up transformer HVT. The structure of the step-up transformer HVT will be described with reference to FIG. The step-up transformer HVT shown in FIG.
In the same manner as the flyback transformer FBT shown in FIG. 3, for example, the square core
Are formed, and these two U-shaped cores CR1, CR2 are formed.
A gap G is provided at a portion where the ends of the. Then, as shown in the figure, by attaching the low-voltage winding bobbin LB and the high-voltage winding bobbin HB to one of the magnetic legs of the square core CR30, the low-voltage winding bobbin LB and the high-voltage winding bobbin HB are attached. , Each primary winding N
Four and two sets of step-up windings NHV are divided and wound. The primary winding N is connected to the low-voltage winding bobbin LB.
4 are wound around the high-voltage winding bobbin HB, and a plurality of step-up windings NHV are wound by interlayer winding in which the interlayer film F is inserted and wound. Also in this case, a litz wire is used for the primary winding N4 of the step-up transformer HVT.

In the power supply circuit shown in FIG.
On the secondary side of the VT, for example, two sets of boost windings NVH11, NVH
12 is wound in a divided state. For example, the winding end of the boost winding NVH11 is connected to a high-voltage capacitor CHV11 made of a film capacitor or a ceramic capacitor.
Of the high-voltage rectifier diode DHV11. The cathode of the high voltage rectifier diode DHV11 is connected to the other end of the high voltage capacitor CHV11.

In such a connection form, as a result, a double voltage half-wave rectifier circuit composed of [boost winding NHV11, high-voltage rectifier diode DHV11, and high-voltage capacitor CHV11] is formed. The voltage doubler rectification operation of such a voltage doubler half-wave rectifier circuit is as follows. First, during the period when the high-voltage rectifier diode DHV13 is off,
An operation of charging the high-voltage capacitor CHV11 with the rectified current rectified by the high-voltage rectifier diode DHV11 is obtained.
Then, during the period in which the high-voltage rectifier diode DHV13 is turned on, an operation is obtained in which the voltage across the high-voltage capacitor CHV11 is added to the induced voltage induced in the boost winding NHV11. As a result, a voltage corresponding to almost twice the induced voltage induced in the boost winding NHV11 can be obtained in the voltage doubler rectifier circuit including the boost winding NHV11, the high voltage rectifier diode DHV11, and the high voltage capacitor CHV11. .

In the power supply circuit shown in FIG. 4, the above-mentioned [boost winding NHV11,
A voltage doubler rectifier circuit composed of a high voltage rectifier diode DHV11 and a high voltage capacitor CHV11] and two sets of voltage doubler rectifier circuits composed of a [boost winding NHV12, a high voltage rectifier diode DHV12 and a high voltage capacitor CHV12] are connected via a high voltage rectifier diode DHV14. These two sets of voltage doubler rectifiers perform a rectifying operation, so that a high voltage rectifier diode D
The smoothing capacitor COHV is charged via the HV13 and DHV14 with a voltage substantially four times the induced voltage induced in each of the boost windings NVH11 and NHV12. Thereby, the boost winding NVH1 is connected to both ends of the smoothing capacitor COHV.
1, a high level DC voltage EHV corresponding to four times the voltage level obtained at each of the NHV12 is obtained, and the DC high voltage EHV obtained at both ends of the smoothing capacitor COHV is used as the anode voltage of the CRT. It is.

In this case, a smoothing capacitor COFV is connected between the cathode of the high-voltage rectifier diode DHV14 and the secondary side ground as shown in FIG. Is a focus voltage and is output to, for example, a fourth grid (focus electrode) of the CRT.

Here, when the circuit is actually constructed so that a DC high voltage EHV of, for example, 30 KV can be obtained from the high voltage generating circuit 5 of the power supply circuit shown in FIG. N2 = 45T, tertiary winding N3 =
22T, secondary side parallel resonance capacitor C2 = 2200P
F, series resonance capacitor C3 = 3900PF, primary winding N4 of boost transformer HVT = 28T, each boost winding NHV1
1, NHV12 = 530T, smoothing capacitors COHV, COFV =
1000 PF is selected.

FIG. 5 shows operation waveforms of the power supply circuit shown in FIG. 5 (a) to 5 (l) are the same as the operation waveforms shown in FIG.
To (f), for example, when the AC input voltage VAC is 100 V and the high-voltage load power of the high-voltage generation circuit 5 is the maximum load power Pomax =
Operation waveforms under the condition of 60 W (IHV = 2 mA) are shown in FIGS. 5 (g) to 5 (l). The high-voltage load power of the high-voltage generation circuit 5 is the minimum load power Pomin = 0W (IHV = 0m).
The operation waveform under the condition A) is shown.

In this case, when the high-voltage load power of the high-voltage generating circuit 5 is set to the maximum load power, the switching frequency of the switching element Q1 is controlled to be about 104 kHz, and the actual on / off period of the switching element Q1 is controlled. TO
N / TOFF is 6 μs / 3.6 μs. The resonance voltage V1 generated across the parallel resonance capacitor Cr1 due to the on / off operation of the switching element Q1 is shown in FIG.
As shown in (a), a waveform that is a sinusoidal pulse is obtained during the period TOFF when the switching element Q1 is off. At this time, the switching element Q1 is
The collector current IC flows as shown in FIG.

By such an operation, the secondary winding (N2 + N3) of the insulated converter transformer PIT is connected to FIG.
As shown in FIG. 5D, a resonance current I2 flows, and a voltage V2 generated across the secondary-side parallel resonance capacitor C2 is as shown in FIG.
As shown in (c), during the period in which the switching element Q1 is turned on, the rectifier diodes DO1, DO3 operate to have a positive voltage level at which the voltage level is 200 V, and the switching element Q1 is turned off in the period TOFF. Becomes a negative resonance pulse voltage having a peak voltage level of 500 Vp. Then, a negative resonance pulse generated at both ends of the secondary resonance capacitor C2 is input to the primary winding N4 of the step-up transformer HVT via the series resonance capacitor C3 forming a series resonance circuit. , The resonance voltage V4 generated at both ends of the primary winding N4 is
As shown in FIG. 5E, the peak voltage level is 40
The resonance voltage waveform becomes 0 Vp, and the resonance current I4 flowing through the primary winding N4 has a resonance current waveform having a peak value of 2 Ap as shown in FIG.

On the other hand, when the high-voltage load power in the high-voltage generation circuit 5 is set to the minimum load power, the switching frequency of the switching element Q1 is controlled to, for example, 116 kHz, and the actual on / off period of the switching element Q1 is controlled. TON / TOFF is 5 μs / 3.6 μs. In this case as well, a resonance voltage V1 as shown in FIG. 5 (g) is generated at both ends of the parallel resonance capacitor Cr1 by the on / off operation of the switching element Q1, and FIG.
A collector current IC as shown in FIG.

By such an operation, the secondary winding (N2 + N3) of the insulated converter transformer PIT is connected to FIG.
As shown in FIG. 5 (j), the resonance current I2 flows, and the voltage V2 generated across the secondary-side parallel resonance capacitor C2 is as shown in FIG.
As shown in (i). The negative resonance pulse generated in the secondary resonance capacitor C2 is input to the primary winding N4 of the flyback transformer FBT via the series resonance capacitor C3, and the waveform of the voltage V4 across the primary winding N4 is 5 (k), and the current waveform of the current I4 flowing through the primary winding N4 has a resonance waveform as shown in FIG. 2 (l).

FIG. 5A to FIG. 5F and FIG.
As can be understood by comparing the operation waveforms shown in FIGS. 5A to 5L, in the power supply circuit shown in FIG. 4, the high-voltage load power of the high-voltage generation circuit 5 is changed from the maximum load power Pomax to the minimum load power P.
Omin, the switching frequency of the switching element Q1 changes from 104 kHz to 116 kHz. In this case, too, the switching frequency of the primary-side switching converter is variably controlled according to the change in the high-voltage load of the high-voltage generation circuit 5. . That is, the power supply circuit shown in FIG. 4 operates similarly to the power supply circuit shown in FIG.

Therefore, even in the case where the power supply circuit is configured as shown in FIG. 4, similarly to the power supply circuit shown in FIG. 1, the DC voltage is generated in the high-voltage generation circuit 5 without interposing the horizontal output circuit of the television receiver. Since a high voltage EHV can be obtained, compared with the case where each DC output voltage is output from the horizontal deflection circuit as in the conventional circuit shown in FIG.
The power loss is small, and the DC input power can be reduced. Accordingly, the AC input power input to the power supply circuit can be reduced, and energy saving can be achieved.

Further, also in this case, the step-up transformer HV
The voltage waveform of the voltage V4 input to the primary winding N4 of T is
As shown in FIG. 5 (e) and FIG. 5 (k), since the resonance voltage waveform is obtained, the boost windings NHV11,
The induced voltage induced in the NHV12 also has a resonance waveform, and the conduction angle of each of the high-voltage rectifier diodes DHV13 and DHV14 is widened. Therefore, for example, as shown in FIG.
If the voltage level of the DC high voltage EHV is 30 KV, the DC high voltage EHV when the high-voltage load power is 0 W
The voltage level of HV is about 31 KV. That is, in the power supply circuit shown in FIG. 4, even when the high-voltage load power fluctuates from 60 W to 0 W, the voltage fluctuation width ΔEHV of the DC high voltage EHV is 1
KV, as compared with the conventional circuit shown in FIG.
The variation width ΔEHV can be reduced to half or less.

Accordingly, if the power supply circuit shown in FIG. 4 is applied to, for example, a television receiver and a high DC voltage EHV is supplied to the anode electrode of the CRT, the brightness of the surface of the CRT becomes low. In such a case, there is no occurrence of screen missing or the like, and it is not necessary to provide a zooming correction circuit or the like in the horizontal output circuit of the television receiver unlike a conventional circuit.

The switching frequency of the step-up transformer HVT also corresponds to the switching frequency of the switching element Q1, and is not synchronized with, for example, the cycle of the horizontal synchronizing signal fH of the video signal. Due to the leakage inductance, it is possible to suppress ringing generated in the induced voltage induced from the boost windings NHV11 and NHV12, and it is possible to prevent raster ringing and curtain stripes from being generated on the screen of the CRT.

Further, in the power supply circuit shown in FIG. 4, the number of turns of the primary winding N4 of the step-up transformer HVT is
The number of windings of the step-up windings NHV11 and NHV12 can be reduced from the conventional 2650T (530T × 5) to the 1060T (53T).
0T × 2). In this case, the operating frequency of the step-up transformer HVT is also 100 KH.
z or more, it is possible to reduce the cross-sectional area of the ferrite core of the step-up transformer HVT to about 1/2 of the conventional one.

Next, FIG. 9 shows a configuration of a power supply circuit according to a third embodiment of the present invention. In the power supply circuit shown in FIG. 9, the voltage resonance type converter provided on the primary side employs a separately-excited configuration, and is provided with a switching element Q2 composed of, for example, one MOS-FET. The drain of the switching element Q2 is connected to the positive electrode of the smoothing capacitor Ci via the primary winding N1 of the insulating converter transformer PIT, and the source is connected to the primary side ground. Here, the parallel resonance capacitor Cr2 is connected in parallel between the drain and the source. Further, a clamp diode DD2 is connected in parallel between the drain and the source.

The switching element Q2 is driven by the oscillation / drive circuit 2 so that the switching operation described above with reference to FIG. 1 is obtained. That is, the control circuit 1 supplies a current or voltage of a level fluctuated according to the fluctuation of the DC output voltage EO1 to the oscillation / drive circuit 2. In the oscillation / drive circuit 2, a switching drive whose cycle is varied according to the output level from the control circuit 1 so that the DC output voltage output from the secondary side of the insulating converter transformer PIT is stabilized. A signal (voltage) is output to the gate of the switching element Q2. As a result, the switching frequency of the switching element Q2 is varied. At this time, as described in FIG. 1, the period during which the switching element Q2 is off is constant, and the period during which the switching element Q2 is on is varied. The generated switching drive signal is output. The starting resistor RS is provided to supply a starting current obtained in the rectifying / smoothing line to the oscillation / drive circuit 2 when the commercial AC power is turned on.

On the secondary side of the insulating converter transformer PIT formed as described above, the tertiary winding N5 (third secondary winding) is wound around the winding end of the secondary winding N3. The ending of the tertiary winding N5 is connected to the starting end of the primary winding N4 of the flyback transformer FBT via a series resonance capacitor C3.

In this case as well, the same effects as those of the power supply circuit of FIG. 1 described above can be obtained, and the flyback transformer FB is further connected from the secondary side of the isolated converter transformer PIT.
Since the voltage level of the resonance voltage V4 input to the primary winding N4 of T can be increased, the boosted voltage level boosted by the flyback transformer FBT is suppressed lower than that of the power supply circuit shown in FIG. It becomes possible. Therefore, in the power supply circuit shown in FIG. 9, the boost windings NVH1 to NHH provided on the secondary side of the flyback transformer FBT
There is an advantage that the number of windings of V5 can be reduced, and the flyback transformer FBT can be further reduced in size.

Next, FIG. 10 shows the configuration of a power supply circuit according to a fourth embodiment of the present invention. The same parts as those in FIG. 9 are denoted by the same reference numerals, and detailed description is omitted. This figure 1
The power supply circuit denoted by reference numeral 0 is provided with a high voltage generation circuit 5 instead of the high voltage generation circuit 4 provided in the power supply circuit shown in FIG. That is, the flyback transformer FB
Instead of the high voltage generation circuit 4 configured by T, a high voltage generation circuit 5 configured by a step-up transformer HVT is provided.

In this case as well, the same effects as those of the power supply circuit of FIG. 4 described above can be expected, and the resonance input from the secondary side of the insulating converter transformer PIT to the primary winding N4 of the step-up transformer HVT. Since the voltage level V4 can be increased, the boosted voltage level boosted by the boost transformer HVT can be suppressed lower than that of the power supply circuit shown in FIG. Therefore, also in this case, the boost winding NHV11 provided on the secondary side of the boost transformer HVT,
The number of turns of the NHV12 can be reduced, and the step-up transformer H
The VT can be further reduced in size.

Further, other secondary circuit configurations of the power supply circuit according to the present embodiment described above are conceivable. FIG. 11 is a diagram illustrating another configuration example of the secondary circuit of the power supply circuit according to the present embodiment described above. The primary circuit configuration of the power supply circuit shown in FIG. 11 may be either the self-excited voltage resonance converter shown in FIG. 1 or the separately excited voltage resonance converter shown in FIG. Also, the same parts as those in FIG.

On the secondary side of the insulating converter transformer PIT shown in FIG. 11, a tertiary winding N3 is formed so as to wind up around the winding end of the secondary winding N2, as in FIG. In this case, a tap is provided in the secondary winding N2, and the output obtained from this tap is rectified and smoothed by the rectifying diode DO6 and the smoothing capacitor CO6, so that, for example, the DC output voltage EO2 for the signal system circuit is reduced. And a DC output voltage EO6 (+15 V) for the vertical deflection circuit.

In this case, the winding end of the secondary winding N2 is connected to the winding end of the tertiary winding (fourth secondary winding) N6. The output obtained from N6 is rectified and smoothed by a rectifying diode DO7 and a smoothing capacitor CO7, so that a DC output voltage E for a vertical deflection circuit is obtained.
O7 (-15V) is obtained.

Further, in the power supply circuit shown in FIG.
An independent tertiary winding (fifth secondary winding) N7 is connected to the secondary winding of the insulation converter transformer PIT by a primary winding N1.
Are wound in the same polarity direction. In this case, the winding start end of the tertiary winding N7 is grounded to the secondary earth, and the winding end is connected to the anode of the rectifier diode DO8.
After the output obtained from the tertiary winding N7 is rectified and smoothed by the rectifier diode DO8 and the smoothing capacitor CO8, the rectified smoothed output is made constant by a three-terminal regulator Q3 and the like, and the smoothing capacitor CO9 is charged. For example, a DC output voltage EO8 (6.3 V) for the heater is obtained from both ends of the capacitor CO9.

Therefore, when the secondary side of the power supply circuit according to the present embodiment is configured in this manner, in the conventional circuit shown in FIG. 15, the DC output voltage EO3 for the video output circuit is
At the same time, the DC output voltages EO6, E0 for the vertical deflection circuit, which are obtained from the primary side of the flyback transformer FBT,
O7 (± 15V) / 7W, DC output voltage EO8 for heater
(6.3V) / 4W etc. are also isolated converter transformers PIT
Can be obtained from the secondary side.

As a result, for example, when the conventional circuit shown in FIG. 15 is applied to a 34-inch television receiver, the voltage conversion efficiency ηDC-DC is about 85%. DC output voltage EO3 (200V) / 10W for video output circuit, DC output voltage EO6, EO7 (± 15V) / 7W for vertical deflection circuit, DC output voltage EO8 (6.3V) / 4W for heater Therefore, a DC input power of 21W ÷ 0.85 = 24.7W was required. On the other hand, the switching converter provided in the power supply circuit shown in FIG. 11 is of a complex resonance type, and its voltage conversion efficiency ηDC-DC is improved to about 95%. Therefore, as shown in FIG. DC output voltages EO3, EO6, EO from the secondary side of the transformer PIT
7, EO8, 21W２１0.95 =
22.1W DC input power is sufficient, and about 2.6
It is also possible to reduce the power loss of W. As a result, for example, as in the power supply circuit shown in FIG. 11, the DC output voltages EO1, E0 are output from the secondary side of the insulating converter transformer PIT.
The power loss in the case of obtaining O3, EO6, EO7, EO8 and the DC output voltages EO1, EO3, EO6, EO6, EO6 from the flyback transformer FBT as in the conventional circuit shown in FIG.
Comparing the power loss when obtaining EO7 and EO8, FIG.
The power supply circuit shown in (1) can reduce the power loss of about 14 W.

Further, in the conventional circuit shown in FIG. 15, the AC-DC power conversion efficiency ηAC-DC of the current resonance type converter constituting the switching power supply 10 is about 90%,
When the maximum load power (Pomax) obtained by integrating the DC output voltages EO1, EO2, EO4, and EO5 is 200W, the AC input power is 222.2W. However, in the power supply circuit shown in FIG. Since the power conversion efficiency ηAC-DC is improved, the AC input power can be reduced by about 18W.

FIG. 12 is a diagram showing still another configuration example of the secondary circuit of the power supply circuit according to the present embodiment. The same parts as those in FIG. 11 are denoted by the same reference numerals, and detailed description will be omitted. The power supply circuit shown in FIG.
As in the power supply circuit shown in FIG.
From the secondary side of T, for example, a DC output voltage E for a signal system circuit
O2 and DC output voltages EO6, EO7 (± 1
5 V) and a DC output voltage EO8 (6.3 V) for the heater. However, instead of the high voltage generating circuit 4, a high voltage generating circuit 6 constituted by a step-up transformer HVT is provided. In this case, in the high-voltage generation circuit 6, the boost winding NHV11 of the boost transformer HVT is used.
Is connected to the connection point between the anode of the high-voltage rectifier diode DHV21 and the cathode of the high-voltage rectifier diode DHV22, and its winding start is connected to the connection point between the negative electrode of the smoothing capacitor COHV1 and the positive electrode of the smoothing capacitor COHV2. You.
The positive electrode of the smoothing capacitor COHV1 is connected to the cathode of the high-voltage rectifier diode DHV21, and the smoothing capacitor COHV2
Is connected to the anode of the high-voltage rectifier diode DHV22.

In such a connection form, as a result, the output from the boost winding NHV11 causes the smoothing capacitor COHV1
And the charging operation for the smoothing capacitor COHV2 are performed, so that a DC voltage corresponding to twice the alternating voltage obtained in the boost winding NHV11 is applied across the smoothing capacitor COHV1 and the smoothing capacitor COHV2 connected in series. An output voltage is obtained. The high voltage generating circuit 6 includes [boost winding NHV11, high voltage rectifier diodes DHV21, DHV22, smoothing capacitors COHV1, COHV2], [boost winding NHV12, high voltage rectifier diodes DHV23, DHV24, smoothing capacitors COFV1, COFV1.
OFV2] are provided.
Smoothing capacitors COHV1 and COHV of each voltage rectifying and smoothing circuit
2, COFV1 and COFV2 are connected in series. As a result, from both ends of the smoothing capacitors COHV1-COHV2-COF1-COFV2, the DC voltage EHV corresponding to the level approximately four times the induced voltage induced in the two sets of boost windings NVH11 and NVH12.
(Anode voltage) and COFV1−
From both ends of COFV2, a DC voltage EFV (focus voltage) corresponding to a level almost twice the induced voltage induced in the boost winding NHV12 is obtained. When the secondary side of the power supply circuit according to the present embodiment is configured in this manner, the same effects as those of the power supply circuit shown in FIG. 11 can be obtained.

In the present embodiment, in the high-voltage generating circuit, the output voltage of a level corresponding to substantially equal to or twice the induced voltage induced in the flyback transformer FBT or the boost winding NHV of the boost transformer HVT. In this case, a rectifier circuit is connected in series to obtain a DC high voltage of a predetermined level. Naturally, it is also possible to obtain a DC high voltage of a predetermined level by connecting in series a rectifier circuit for obtaining an output voltage of a level corresponding to three times.

In the present embodiment, the orthogonal control transformer PRT is used as a control transformer for performing constant voltage control under a configuration having a self-excited resonance converter for the primary side. Instead of the orthogonal control transformer PRT, an oblique control transformer previously proposed by the present applicant can be employed. Although the illustration of the structure of the oblique control transformer is omitted here, as in the case of the orthogonal control transformer, for example, 4
A three-dimensional core is formed by combining two sets of double U-shaped cores having two magnetic legs. Then, the control winding NC and the driving winding NB are wound around the three-dimensional core. At this time, the winding directions of the control winding and the driving winding obliquely intersect. To be. Specifically, one of the control winding NC and the drive winding NB is
Of the four magnetic legs, winding is performed on two magnetic legs adjacent to each other, and the other winding is wound on two magnetic legs that are considered to be in a diagonal positional relationship. It is to be wound.
When such an oblique control transformer is provided, even when the AC current flowing through the drive winding changes from a negative current level to a positive current level, the operation tendency that the inductance of the drive winding increases. Is obtained. This allows
The current level in the negative direction for turning off the switching element increases, and the accumulation time of the switching element is shortened.Accordingly, the fall time at the time of turning off the switching element is also shortened, and the switching element is turned off. The power loss can be further reduced.

[0110]

As described above, the switching power supply circuit of the present invention uses a series resonance capacitor to convert a negative resonance voltage obtained from the secondary side of the insulating converter transformer constituting the switching power supply circuit as a complex resonance type. The voltage is input to the primary side of the step-up transformer via the power supply. Then, after boosting the substantially sinusoidal resonance voltage input to the step-up transformer via the series resonance capacitor, the DC high voltage generating means obtains a DC high voltage at a predetermined high voltage level. Therefore, if the switching power supply circuit of the present invention is applied to a television receiver, for example, when obtaining a high DC voltage to be supplied to the anode of a cathode ray tube,
There is no need to convert the secondary side DC output voltage to a flyback pulse voltage in the horizontal deflection circuit, and the configuration without the horizontal deflection circuit can be used. Can be improved.

Since the resonance voltage input to the primary side of the step-up transformer via the series resonance capacitor is substantially sinusoidal, the DC high voltage output from the DC high voltage generating means is Is fluctuated, the voltage fluctuation width can be reduced as compared with the conventional case. Therefore, if the present invention is applied to, for example, a high voltage supply unit of a television receiver, it becomes possible to suppress, for example, horizontal amplitude fluctuation of an electron beam output from a cathode ray tube.

The primary side of the step-up transformer is separated from the horizontal deflection circuit of the television receiver, and the switching frequency of the step-up transformer corresponds to the switching frequency of the switching element provided in the power supply circuit. Therefore, ringing does not occur in the induced voltage of the boost winding due to leakage magnetic flux or leakage inductance from the boost transformer. Thus, even when the present invention is applied to a television receiver, there is an advantage that, for example, raster ringing and curtain stripes do not occur on the screen of a CRT.

Further, a third secondary winding is formed so as to be wound around a winding end of the second secondary winding of the insulating converter transformer, and the third secondary winding is connected in series from the third secondary winding. If the resonance voltage is input to the primary side of the step-up transformer via the resonance capacitor, the number of windings of the step-up transformer can be reduced, so that the size of the step-up transformer can be further reduced. Furthermore, a fourth secondary winding side is formed so as to be wound around the winding start end of the first secondary winding of the insulating converter transformer. The power loss can be reduced by independently winding the fifth secondary winding side to obtain the third DC output voltage and the fourth DC output voltage from the secondary side of the insulating converter transformer. As a result, the AC input power can be reduced, and energy can be saved.

[Brief description of the drawings]

FIG. 1 is a diagram showing a configuration of a power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

FIG. 3 is a sectional view showing a configuration of a flyback transformer provided in the power supply circuit shown in FIG. 1;

FIG. 4 is a diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention.

FIG. 5 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIG.

6 is a cross-sectional view illustrating a configuration of a boosting transformer provided in the power supply circuit illustrated in FIG.

FIG. 7 is a diagram showing a relationship between a high DC voltage output from a high voltage generating circuit of the power supply circuit shown in FIG. 1 and a high voltage load power.

8 is a diagram showing a relationship between a DC high voltage output from a high voltage generation circuit of the power supply circuit shown in FIG. 4 and a high voltage load power.

FIG. 9 is a diagram illustrating a configuration of a power supply circuit according to a third embodiment of the present invention.

FIG. 10 is a diagram showing a configuration of a power supply circuit according to a fourth embodiment of the present invention.

FIG. 11 is a diagram showing another circuit configuration of a secondary circuit applicable to the power supply circuit according to the present embodiment.

FIG. 12 is a diagram showing still another circuit configuration of a secondary circuit applicable to the power supply circuit according to the present embodiment.

FIG. 13 is a cross-sectional view showing the structure of the insulating converter transformer.

FIG. 14 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 15 is a diagram showing a configuration of a conventional television power supply circuit.

16 is a circuit diagram showing a configuration of a switching power supply circuit provided in the television power supply circuit shown in FIG.

17 is a waveform chart showing an operation of a main part of the television power supply circuit shown in FIG.

[Explanation of symbols]

1 control circuit, 4 5 6 high voltage generation circuit, AC commercial AC power supply, Ci smoothing capacitor, Cr1 Cr2 primary side parallel resonance capacitor, C2 secondary side parallel resonance capacitor, C3 series resonance capacitor, CB resonance capacitor,
CO1 to CO9 COHV COHV1 COHV2 COFV Smoothing capacitor, CHV11 CHV12 High voltage capacitor, Di bridge rectifier circuit, DD1 DD2 Clamp diode, DHV
1 ~ DHV5 DHV11 ~ DHV14 DHV21 ~ DHV24 High voltage rectifier diode, DO1 ~ DO8 Rectifier diode, EO1 ~ EO8
DC output voltage, EHV DC high voltage, FBT flyback transformer, HVT step-up transformer, NHV1 to NHV5 N
HV11 NHV12 Boost winding, N1 N4 Primary winding, N2
Secondary winding (secondary winding), N3 N5 to N7 tertiary winding (secondary winding), NB drive winding, NC control winding, PI
T isolation converter transformer, PRT orthogonal control transformer, Q1 Q2 switching element, Q3 three-terminal regulator, RS starting resistance, RB base current limiting resistance

──────────────────────────────────────────────────の Continued on the front page (51) Int.Cl. ⁷ Identification symbol FI Theme coat ゛ (Reference) H04N 3/18 H04N 3/18 Z 5/63 5/63 A F term (Reference) 5C026 EA02 EA04 5C068 AA06 CB01 CC07 CC08 CC09 5H006 AA01 BB04 CA01 CA07 CA12 CA13 CB03 CC02 DA04 DC05 HA09 5H730 AA01 AA16 AS01 AS04 BB26 BB43 BB52 BB57 BB66 BB67 BB72 BB74 BB76 BB86 BB94 CC01 DD02 DD22 EE02 FF01 Z16Z04

Claims (8)

[Claims] 1. A switching means formed with a switching element for intermittently outputting an input DC input voltage, and a primary-side parallel resonance circuit having a voltage resonance type operation of the switching means. A primary side parallel resonance capacitor provided in such a manner as to be provided for transmitting the output of the primary side to the secondary side, a primary side winding is wound on the primary side, and at least the A secondary winding having a first secondary winding and a second secondary winding formed to wind up with respect to the first secondary winding. In addition, an insulation converter transformer capable of obtaining a required degree of coupling that is loosely coupled between the primary winding and the secondary winding, and a secondary winding with respect to the secondary winding. Side parallel resonant capacitors are connected in parallel. And a half-wave rectifying operation on a voltage in a positive period of an alternating voltage obtained from the first secondary winding, the secondary parallel resonance circuit being formed including the secondary parallel resonance circuit. , A first DC output voltage, and a DC output voltage obtained by performing a half-wave rectification operation on a voltage in a positive period of the alternating voltage obtained from the second secondary winding, by the first DC output voltage. DC output voltage generating means configured to obtain a second DC output voltage by accumulating the DC output voltage; and varying a switching frequency of the switching element according to a level of the first DC output voltage. A constant voltage control that performs constant voltage control by switching and driving the switching element while changing the ON period while controlling the OFF period within the switching cycle. Means, a boosting transformer configured to obtain a boosted voltage obtained by boosting the resonance voltage from the secondary side by transmitting a resonance voltage input to the primary side to the secondary side, and a primary side operation of the boosting transformer. In order to make the resonance operation, a negative winding obtained from the secondary winding of the insulating converter transformer is inserted between the secondary winding of the insulating converter transformer and the primary winding of the step-up transformer. A predetermined high voltage level is obtained by performing a rectification operation on a series resonance capacitor input to the primary side of the step-up transformer as a resonance voltage having a substantially sinusoidal resonance voltage and a step-up voltage obtained on the secondary side of the step-up transformer. And a DC high voltage generator configured to obtain a DC high voltage. 2. The switching power supply circuit according to claim 1, wherein said step-up transformer is a flyback transformer. 3. The step-up high-voltage generating means includes: a plurality of step-up windings each independently wound on a secondary side of the step-up transformer; and the step-up winding obtained by each of the step-up windings. A multiple rectification circuit formed by connecting in series a plurality of rectification circuits provided so as to perform a half-wave rectification operation on a voltage and to obtain an output voltage of a level corresponding to substantially equal to the boosted voltage. The switching power supply circuit according to claim 1, comprising: 4. The step-up high-voltage generating means includes: a plurality of step-up windings each independently wound on a secondary side of the step-up transformer; and the step-up winding obtained by each of the step-up windings. By performing a rectifying operation on a voltage, a multiple voltage rectifier circuit formed by connecting in series a plurality of double voltage rectifier circuits provided so as to obtain an output voltage of a level corresponding to approximately twice the boosted voltage. The switching power supply circuit according to claim 1, comprising: a rectifier circuit. 5. The DC high voltage generating means outputs a first DC high voltage having a predetermined high voltage level and a second DC high voltage having a voltage level lower than the first DC high voltage. The switching power supply circuit according to claim 1, wherein the switching power supply circuit is configured to be capable of being operated. 6. A third secondary winding is wound around a winding end portion of a second secondary winding of the insulating converter transformer, and the third secondary winding is formed. The switching power supply circuit according to claim 1, wherein a primary winding of the step-up transformer is connected between the winding and a secondary ground via the series resonance capacitor. 7. A fourth secondary winding is formed so as to wind up a fourth secondary winding with respect to a winding start end of a first secondary winding of the insulating converter transformer. The switching power supply circuit according to claim 1, further comprising a third DC output voltage generating unit configured to obtain a third DC output voltage by performing a rectification operation on an alternating voltage obtained on the line. . 8. A fifth secondary winding is independently wound around a secondary side of the insulating converter transformer, and a fourth secondary winding is obtained from an alternating voltage obtained in the fifth secondary winding. 2. The switching power supply circuit according to claim 1, further comprising: a fourth DC output voltage generating unit configured to obtain a DC output voltage of the switching power supply.