JP2001128451A - Switching power circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To eliminate abnormal operation of a switching element under intermediate load conditions. SOLUTION: A secondary resonance circuit is formed by combining a secondary parallel resonance circuit with a secondary series resonance circuit with respect to a secondary winding N2 of an insulated converter transformer PIT as a secondary resonance circuit. As a result, the contact-voltage control of secondary DC output voltage is a combined control controlling both switching frequency and the conducting angle of switching current passing through a switching element Q1, thereby restraining increase in period, during which the switching element Q1 is turned off.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a circuit employing a switching converter of a type such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular switching operation waveform, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It has been found that there is a limit in improving the power conversion efficiency. Therefore, the present applicant has previously proposed various switching power supply circuits using various resonance type converters.
The resonance type converter can easily obtain high power conversion efficiency and realize low noise because the switching operation waveform is sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

FIG. 6 is a circuit diagram showing an example of a prior art switching power supply circuit which can be constructed based on the invention proposed by the present applicant. This power supply circuit is provided with a voltage resonance type converter having a single switching element Q1 and performing a switching operation by a self-excited method in a so-called single-ended system.

In the power supply circuit shown in FIG. 1, a bridge rectification circuit Di is used as a rectification and smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) and obtaining a DC input voltage.
And a full-wave rectifier circuit composed of a smoothing capacitor Ci, and generates a rectified smoothed voltage Ei corresponding to a level that is one time the AC input voltage VAC. Further, in the rectifying / smoothing circuit, an inrush current limiting resistor Ri is inserted in the rectifying current path to suppress, for example, an inrush current flowing into the smoothing capacitor Ci when the power is turned on.

A voltage resonance type switching converter provided in this power supply circuit has a single switching element Q1.
It has a self-excited configuration with In this case, a high withstand voltage bipolar transistor (BJT: junction type transistor) is employed as the switching element Q1.

The base of the switching element Q1 is connected to a smoothing capacitor Ci (rectified smoothing voltage E) via a starting resistor RS.
It is connected to the positive electrode side of i) so that the base current at the time of startup is obtained from the rectification smoothing line. Further, a detection drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB are provided between the base of the switching element Q1 and the primary side ground.
A series resonance circuit for driving self-excited oscillation is connected. A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the switching element Q1 is turned off. , The collector of switching element Q1 is connected to one end of primary winding N1 of insulating converter transformer PIT, and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr
Forms a primary side parallel resonance circuit of the voltage resonance type converter with its own capacitance and a leakage inductance L1 on the primary winding N1 side of the insulating converter transformer PIT described later. Although the detailed description is omitted here, when the switching element Q1 is turned off, the voltage VCP across the resonance capacitor Cr actually becomes a sinusoidal pulse waveform due to the operation of the parallel resonance circuit due to the operation of the voltage resonance type operation. Is obtained.

The orthogonal control transformer PRT shown in FIG.
Is a saturable reactor on which a resonance current detecting winding ND, a driving winding NB, and a control winding NC are wound. The orthogonal transformer PRT drives the switching element Q1 and is provided for constant voltage control. Although the illustration is omitted as the structure of the orthogonal control transformer PRT,
A three-dimensional core is formed by joining the ends of the two magnetic legs of two double U-shaped cores having four magnetic legs. Then, the resonance current detection winding ND and the drive winding N are applied to the predetermined two magnetic legs of the three-dimensional core in the same winding direction.
B, and a control winding NC is wound in a direction perpendicular to the resonance current detecting winding ND and the driving winding NB.

In this case, the resonance current detection winding ND of the orthogonal control transformer PRT is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1 of the insulating converter transformer PIT, so that the switching element Q1 Is transmitted to the resonance current detection winding ND via the primary winding N1. In the orthogonal control transformer PRT, the switching output obtained in the resonance current detection winding ND is excited by the drive winding NB through the transformer coupling, so that the alternating voltage as the drive voltage is applied to the drive winding NB. appear. This drive voltage is output as a drive current from the series resonance circuit (NB, CB) forming the self-excited oscillation drive circuit via the base current limiting resistor RB to the base of the switching element Q1. As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit.

The insulated converter transformer PIT transmits the switching output of the switching element Q1 to the secondary side. As shown in FIG. 15, the insulated converter transformer PIT has E-shaped cores CR1, CR2 made of, for example, ferrite material.
An EE-type core is provided so that the magnetic legs of the EE-type core are opposed to each other. Each is wound in a divided state. A gap G is formed for the center magnetic leg as shown in the figure. As a result, loose coupling with a required coupling coefficient can be obtained. Gap G is E-type core CR
1 and CR2 can be formed by making the center magnetic leg shorter than the two outer magnetic legs. The coupling coefficient k is set to a loosely coupled state of, for example, k ≒ 0.85, so that a saturated state is hardly obtained.

One end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1 as shown in FIG. 6, and the other end is connected to the smoothing capacitor through a series connection of the resonance current detecting winding ND. It is connected to the positive electrode of Ci (rectified smoothed voltage Ei).

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, for the secondary winding N2,
By connecting the secondary parallel resonance capacitor C2 in parallel, a parallel resonance circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2. The alternating voltage excited by the secondary winding N2 by this parallel resonance circuit becomes a resonance voltage. That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, a primary side is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and a secondary side is provided with a parallel resonance circuit for obtaining a voltage resonance operation. In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

With respect to the secondary parallel resonance circuit formed as described above, a rectifying / smoothing circuit including a bridge rectifying circuit DBR and a smoothing capacitor CO1 is provided for the secondary winding N2. The secondary side output voltage EO1 is obtained. That is, in this configuration, a full-wave rectification operation is obtained by the bridge rectification circuit DBR on the secondary side. In this case, the bridge rectifier circuit DBR receives the resonance voltage supplied from the secondary-side parallel resonance circuit, thereby forming the secondary winding N2.
To generate a DC output voltage EO1 corresponding to a level substantially equal to the alternating voltage excited at the same time. Note that this DC output voltage EO1
Is also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage and an operation power supply of the control circuit 1.

As the secondary side of the insulating converter transformer PIT of the power supply circuit shown in FIG. 6, a circuit configuration as shown in FIG. 7 can be adopted based on the proposal of the present applicant. Insulated converter transformer PIT shown in this figure
Is connected to a secondary parallel resonance capacitor C2 in parallel with the secondary winding N2. And the secondary winding N
2 is provided with a center tap, and rectifying diodes DO1, DO2 and a smoothing capacitor CO1 are connected as shown in the figure to form a full-wave rectifier circuit. A DC output voltage EO1 corresponding to a substantially equal-magnification level is generated.

Incidentally, the insulation converter transformer PIT
, The inductance L1 of the primary winding N1 and the inductance of the secondary winding N2 depend on the relationship between the polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the connection of the rectifier diode DO (DO1, DO2). The mutual inductance M with L2 may be + M or -M. For example, when the connection configuration shown in FIG. 16A is employed, the mutual inductance is + M, and when the connection configuration shown in FIG. 16B is employed, the mutual inductance is -M. When this is made to correspond to the operation on the secondary side shown in FIGS. 6 and 7, for example, in the power supply circuit shown in FIG. 6, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, the bridge rectifier circuit The operation in which the rectified current I3 flows through the DBR can be regarded as a + M operation mode (forward mode). Conversely, when the alternating voltage obtained in the secondary winding N2 has a negative polarity, the rectified current I4 flows through the bridge rectifier diode DBR. Is -M
Operation mode (flyback method). For example, in the power supply circuit shown in FIG. 7, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, an operation in which a rectified current flows through the rectifier diode DO1 can be regarded as a + M operation mode (forward mode). Conversely, the secondary winding N2
Rectifier diode D when the alternating voltage obtained at
It can be seen that the rectified current flowing in O2 is in the -M operation mode (flyback method). That is, in the power supply circuits shown in FIGS. 6 and 7, each time the alternating voltage obtained on the secondary winding N2 becomes positive / negative, the mutual inductance becomes + M
/ -M mode.

In such a configuration, power is supplied to the load side increased by the action of the primary side parallel resonance circuit and the secondary side parallel resonance circuit, and the power supplied to the load side is also increased. The rate of increase in load power is also improved. This is because, as described above with reference to FIG. 15, the gap G is formed in the insulating converter transformer PIT and loose coupling is performed with a required coupling coefficient, thereby obtaining a state that is more difficult to be saturated. It is realized. For example, when the gap G is not provided for the insulating converter transformer PIT, there is a high possibility that the insulating converter transformer PIT becomes saturated during flyback operation and the operation becomes abnormal. It is difficult to hope that this is done properly.

In the control circuit 1, the level of the control current (DC current) flowing through the control winding NC is changed according to the change in the DC output voltage level (EO1) on the secondary side, so that the orthogonal control transformer PRT is controlled. The inductance LB of the wound drive winding NB is variably controlled. Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB of the drive winding NB changes. This changes the switching frequency of the switching element Q1, thereby stabilizing the DC output voltage on the secondary side.

In the power supply circuit shown in FIG. 6, when an orthogonal control transformer PRT for variably controlling the inductance LB of the drive winding NB is provided, the period during which the switching element Q1 is turned off for changing the switching frequency is used. TOFF is kept constant, and the ON period TON is variably controlled. In other words, in this power supply circuit, as a constant voltage control operation, an operation is performed to variably control the switching frequency, thereby performing resonance impedance control on the switching output, and at the same time, controlling the conduction angle of the switching element in the switching cycle (PWM). Control). This complex control operation is realized by a set of control circuit systems. In the present specification, such complex control is also referred to as “complex control method”.

Here, for example, the input AC input voltage V
AC fluctuation is 85V to 144V, load power Po that can be supported
The power supply circuit corresponding to the input / output condition of 200 W to 0 W (no load) is connected to the orthogonal control transformer P shown in FIG.
In the case of using the self-excited oscillation type switching frequency control method by RT, for example, the isolation converter transformer P
The number of turns of the IT ferrite core EE-35, the primary winding N1 and the secondary winding N2 are each 43 T (turn), the gap G = 1 mm, and the primary side resonance capacitor Cr = 6800.
It has been experimentally confirmed that the optimum operation is obtained when the pF and the secondary parallel resonance capacitor C2 are set to 0.01 .mu.F.

FIG. 8 shows an example of an operation waveform obtained when the input AC voltage VAC is set to 100 V in the power supply circuit having the configuration shown in FIG. 6 in which the values of the respective components are set as described above. In such a circuit, the switching operation of the switching element Q1 by a series resonance circuit (NB, CB) as a self-excited oscillation driving circuit causes the switching element Q1 // parallel resonance capacitor Cr to have both ends of a parallel connection circuit. 8 by the action of the parallel resonance circuit.
A primary side parallel resonance voltage Vcp as shown in FIG. As shown in the figure, the parallel resonance voltage Vcp is at the 0 level during the period when the switching element Q1 is on,
In the OFF period TOFF, a waveform of a sine wave pulse is obtained, which corresponds to the operation as the voltage resonance type.

The switching output is transmitted to the secondary side of the insulated converter transformer PIT by the on / off operation of the switching element Q1. In this case, in the bridge rectifier circuit DBR provided on the secondary side of the insulation converter transformer PIT, when the insulation converter transformer PIT is in the + M operation mode (forward system), the insulation converter transformer PIT operates as a forward converter. A rectified current I3 having a waveform as shown in FIG. Conversely, the insulation converter transformer PIT
In the M operation mode (flyback method), the isolated converter transformer PIT operates as a flyback converter, and the rectified current I4 having a waveform as shown in FIG.
Will flow.

FIG. 9 is a diagram showing the constant voltage control characteristics when the load fluctuates in the power supply circuit having the configuration shown in FIG. 6 in which the values of the respective components are set as described above. In this case, the input AC voltage VAC is set to 100V. As shown in this figure, in the power supply circuit shown in FIG. 6, as the constant voltage control of the DC output voltage EO1 output from the secondary side, as the load power Po increases, the switching element Q1 The switching frequency fs is controlled so as to decrease with a substantially constant gradient, and at the same time, the period TON during which the switching element Q1 is turned on is controlled so as to increase with a substantially constant gradient. In addition,
The period TOFF during which the switching element Q1 is turned off is substantially constant, and is not shown. That is, FIG.
In the power supply circuit shown in (1), as a constant voltage control operation, the switching frequency is variably controlled to perform resonance impedance control on the switching output, and at the same time, control the ON period (conduction angle) of the switching element. Has been adopted.

FIG. 10 is a diagram showing another example of the configuration of a switching power supply circuit as a prior art which can be configured based on the invention previously proposed by the present applicant. The same parts as those in FIG. 6 are denoted by the same reference numerals, and description thereof will be omitted.

The insulation converter transformer PI shown in FIG.
The secondary winding N2 provided on the secondary side of T is wound with a different number of turns from the secondary winding N2 of the power supply circuit shown in FIG. 6 as described later. One end of the secondary winding N2 is connected to the secondary side ground, and the other end is a series resonance capacitor C2.
s is connected to the connection point of the anode of the rectifier diode DO1 and the cathode of the rectifier diode DO2 via the series connection of s. The cathode of the rectifier diode DO1 is connected to the positive electrode of the smoothing capacitor CO1, and the anode of the rectifier diode DO2 is connected to the secondary side ground. Smoothing capacitor C
The negative side of O1 is connected to the secondary side ground.

In such a connection form, the secondary winding N2
, Series resonance capacitor Cs, rectifier diodes DO1, DO
2, a voltage doubler half-wave rectifier circuit composed of a set of smoothing capacitors CO1]. Here, the series resonance capacitor Cs forms a series resonance circuit corresponding to the on / off operation of the rectifier diodes DO1, DO2 by its own capacitance and the leakage inductance component L2 of the secondary winding N2. Also, the primary side parallel resonance circuit (N1, Cr)
Let fo1 denote the parallel resonance frequency of fo1 and fo2 denote the series resonance frequency of the series resonance circuit on the secondary side.
The capacitance of the secondary-side series resonance capacitor Cs is selected so as to be 2. That is, the power supply circuit shown in this figure is also provided with a parallel resonance circuit on the primary side to make the switching operation a voltage resonance type, and a series resonance circuit on the secondary side for obtaining a doubled voltage half-wave rectification operation. This is configured as a composite resonance type switching converter.

[Second winding N2, series resonance capacitor Cs, rectifier diodes DO1, DO2, smoothing capacitor CO]
In the voltage doubler rectifying operation by the set [1], when a switching output is obtained in the primary winding N1 by the switching operation on the primary side, the switching output is excited by the secondary winding N2. The voltage doubler rectifier circuit performs the rectification operation by inputting the obtained alternating voltage to the secondary winding N2. In this case, first, during the period when the rectifier diode DO1 is turned off and the rectifier diode DO2 is turned on, the rectifier diode DO1 operates in the depolarization mode in which the polarity of the primary winding N1 and the secondary winding N2 is -M. The rectified current rectified by the rectifier diode DO2 is converted into the series resonance capacitor C2 by the series resonance effect of the leakage inductance of the secondary winding N2 and the series resonance capacitor Cs.
The operation of charging s is obtained. During the period in which the rectifier diode DO2 is turned off and the rectifier diode DO1 is turned on and the rectification operation is performed, the polarity of the primary winding N1 and the secondary winding N2 becomes + M, and the secondary winding is turned on. The operation is such that the smoothing capacitor CO1 is charged in a state where series resonance occurs in which the potential of the series resonance capacitor Cs is added to the voltage induced on the line N2.
As described above, the isolated converter transformer PIT alternately repeats the polarity polarity mode and the polarity reduction mode, so that the DC output voltage corresponding to almost twice the level of the induced voltage of the secondary winding N2 is applied to the smoothing capacitor CO1. (Rectified smoothed voltage) is obtained. That is, on the secondary side shown in FIG.
A double voltage half-wave rectifier circuit that performs a so-called double voltage half-wave rectification operation is provided.

Here, for example, the input AC input voltage V
85V to 144V AC fluctuation, Possible load power Po
When the power supply circuit corresponding to the input / output condition of 200 W to 0 W is constituted by the power supply circuit having the voltage doubler half-wave rectifier circuit shown in FIG. 10, for example, the winding of the primary winding N1 of the insulation converter transformer PIT may be used. The number of wires is 39T, the number of turns of the secondary winding N2 is 23T, and the primary side resonance capacitor Cr = 4.
700 pF, secondary side series resonance capacitor Cs = 0.1μ
It has been confirmed by experiments that the optimum operating condition is obtained when F is set.

FIG. 12 shows a power supply circuit in which the secondary side of the insulated converter transformer PIT shown in FIG. 10 is constituted by a voltage doubler half-wave rectifier circuit and the values of the respective components are optimally set. It is an example of an operation waveform obtained when VAC is set to 100V. Even in such a circuit, a series resonance circuit (NB
, CB), the switching element Q1 performs a switching operation, so that both ends of the parallel connection circuit of the switching element Q1 // parallel resonance capacitor Cr are actuated by the parallel resonance circuit to form a primary element as shown in FIG. Side parallel resonance voltage Vcp is obtained. This parallel resonance voltage Vc
As shown in the figure, p is a level during which the switching element Q1 is turned on, TON is 0 level, and a sine-wave shaped pulse is obtained during the off time TOFF, which corresponds to the operation as a voltage resonance type. Become. The currents I3 and I4 flowing through the rectifier diodes DO1 and DO2 are as shown in FIG.
As shown in FIG. 12B and FIG. 12C, the waveform of the series resonance current flowing through the secondary winding N2 alternately and continuously flows.

FIG. 13 is a diagram showing a constant voltage control characteristic when the load is varied in the power supply circuit shown in FIG. 10 in which the values of the respective components are set by the above design. In this case, the input AC voltage VAC is 100
V, and the level of the secondary side DC output voltage EO1 is 135V. As shown in FIG. 13, in the power supply circuit shown in FIG. 10, even when the load power Po fluctuates, the switching frequency fs hardly changes, and the DC output voltage output from the secondary side The constant voltage control for making EO1 a constant voltage is performed during a period T during which the switching element Q1 is turned on.
This is realized by controlling the ON and OFF periods TOFF. That is, the power supply circuit shown in FIG. 10 does not perform variable control of the switching frequency fs, and does not employ the complex control method as described above.

The secondary side of the insulated converter transformer PIT of the power supply circuit shown in FIG. 10 may have a circuit configuration as shown in FIG. 11 based on a proposal from the present applicant. Insulated converter transformer P shown in this figure
The secondary winding N2 of the IT is also wound with a different number of turns from the secondary winding N2 of the power supply circuit shown in FIG. 10 as described later. One end of the secondary winding N2 is connected to the connection point between the anode of the rectifier diode DO1 and the cathode of the rectifier diode DO2 via a series connection of the series resonance capacitor Cs1, and the series resonance capacitor Cs1.
2 is connected to the connection point between the anode of the rectifier diode DO3 and the cathode of the rectifier diode DO4 via the series connection of the two. On the other hand, the other end of the secondary winding N2 is connected to a smoothing capacitor C2.
It is connected to the connection point between the negative electrode of O10 and the positive electrode of the smoothing capacitor CO11. The connection point between the negative electrode of the smoothing capacitor CO10 and the positive electrode of the smoothing capacitor CO11 is
The anode of the rectifier diode DO2 and the cathode of the rectifier diode DO3 are connected. The smoothing capacitor CO10 and the smoothing capacitor CO11 are connected in series with the negative electrode of the smoothing capacitor CO10 and the positive electrode of the smoothing capacitor CO11, and then the positive electrode of the smoothing capacitor CO10 is connected to the rectifying diode DO1.
And the negative electrode of CO11 is connected to the secondary side ground.

In such a connection form, as a result,
[Series resonant capacitor Cs1, rectifier diodes DO1, D
A first voltage doubler rectifying circuit composed of a set of [O2, smoothing capacitor CO10] and a second voltage doubler rectifying circuit composed of a set of [series resonant capacitor Cs2, rectifying diodes DO3, DO4, smoothing capacitor CO11] are formed. , The outputs of these first and second voltage doubler rectifier circuits (smoothing capacitors CO10, C10
O11) will be provided connected in series. As a whole of the rectifier circuit combining the first and second voltage doubler rectifier circuits, a secondary winding N is connected between both ends of a series-connected smoothing capacitor C010 and a smoothing capacitor C011.
2, a secondary output voltage corresponding to four times the obtained alternating voltage is obtained. That is, as a whole rectifier circuit combining the first and second voltage doubler rectifier circuits, a quadruple voltage full wave rectifier circuit is formed. The rectifying operation of the quadruple voltage full-wave rectifier circuit will be described later.

The series resonance capacitor Cs1 has its own capacitance and the leakage inductance component L of the secondary winding N2.
2 form a series resonance circuit corresponding to the on / off operation of the rectifier diodes DO1, DO2 in the first voltage doubler rectifier circuit. Similarly, the series resonance capacitor Cs2
Forms a series resonance circuit corresponding to the on / off operation of the rectifier diodes DO3 and DO4 in the second voltage doubler rectifier circuit by using its own capacitance and the leakage inductance component L2 of the secondary winding N2.

As for the resonance frequencies of these series resonance circuits, the parallel resonance frequency of the primary side parallel resonance circuit (N1, Cr) is fo1, and the secondary side series resonance circuit (N2, Cr) is
Assuming that the series resonance frequency of Cs1) is fo2 and the series resonance frequency of the same secondary-side series resonance circuit (N2, Cs2) is fo3, the secondary-side series resonance capacitors Cs1,. The capacitance of Cs2 is selected.

Next, the operation of the above-described quadruple voltage full-wave rectifier circuit will be described. When a switching output is obtained on the primary winding N1 by the switching operation on the primary side, the switching output is excited by the secondary winding N2. The quadruple voltage rectifier circuit performs a rectification operation by inputting the obtained alternating voltage to the secondary winding N2, and comprises a [series resonance capacitor Cs1, rectifier diodes DO1, DO2, and a smoothing capacitor CO10]. The operation of the first voltage doubler rectifier circuit is described below. First, the rectifier diode DO1 is turned off,
During the period when the rectifier diode DO2 is on, the primary winding N1 and the secondary winding N2 operate in the depolarization mode in which the polarity is -M, and the leakage inductance of the secondary winding N2 and the series resonance capacitor Cs1 , The operation of charging the series resonant capacitor Cs1 with the rectified current rectified by the rectifier diode DO2 is obtained. During the period in which the rectifier diode DO2 is turned off and the rectifier diode DO1 is turned on and the rectification operation is performed, the polarity of the primary winding N1 and the secondary winding N2 becomes + M, and the secondary winding is turned on. The smoothing capacitor CO10 is charged in a state where series resonance occurs in which the potential of the series resonance capacitor Cs1 is added to the voltage induced on the line N2.

As described above, the polarity mode (+ M;
The rectifying operation is performed using both the forward operation) and the depolarization mode (-M; flyback operation), so that the smoothing capacitor CO10 has the secondary winding N
2, a DC voltage (rectified smoothed voltage) corresponding to almost twice the induced voltage is obtained. Also, [series resonance capacitor Cs2
, Rectifier diodes DO3, DO4, smoothing capacitor CO11
] In the second voltage doubler rectifier circuit having the same set, the two ends of the smoothing capacitor CO11
A DC voltage corresponding to almost twice the induced voltage of the secondary winding N2 is obtained.

The voltage doubler rectification operation is performed by each of the first and second voltage doubler rectifier circuits as described above. As a result, a secondary capacitor is connected across both ends of the series-connected smoothing capacitor CO10 and smoothing capacitor CO11. A secondary side DC output voltage EO1 corresponding to almost four times the induced voltage of the winding N2 is obtained.

Here, for example, the input AC input voltage V
85V to 144V AC fluctuation, Possible load power Po
When the power supply circuit corresponding to the input / output condition of 200 W to 0 W is constituted by the power supply circuit provided with the quadruple voltage full-wave rectifier circuit shown in FIG. 11, for example, the primary winding N1 of the insulation converter transformer PIT = 46T , Secondary winding N2 =
It has been experimentally confirmed that optimum operation can be obtained when 14T, the primary side resonance capacitor Cr = 3900 pF, and the secondary side series resonance capacitors Cs1, Cs2 = 0.1 μF.

[0039]

By the way, in the power supply circuit shown in FIG. 6, when the rectifier diodes constituting the bridge rectifier circuit DBR are turned on, the currents I3 and I4 flowing through the rectifier diodes of the bridge rectifier circuit DBR are turned on. In
8 (b) and 8 (c), based on the leakage inductance component L2 of the secondary winding N2 of the insulating converter transformer PIT and the junction capacitance (several pF) of each rectifier diode constituting the bridge rectifier circuit DBR. ), A high-frequency ringing current (oscillating current) is superimposed. Such a high-frequency oscillating current is generated by power supply noise (EMI; Elec) from four sets of rectifier diodes constituting the bridge rectifier circuit DBR.
tromagnetic interference). Therefore, when actually configuring the power supply circuit shown in FIG. 6, EMI measures must be taken by adding a ferrite bead inductor or a ceramic capacitor to the secondary side of the insulating converter transformer PIT. Increase.

For example, in the power supply circuit shown in FIG.
A secondary side DC output voltage is obtained by providing a bridge rectifier circuit DBR on the secondary side. That is,
This is to generate a DC output voltage EO1 corresponding to a level substantially equal to the alternating voltage excited in the secondary winding N2. Therefore, in this case, it is necessary that the number of turns of the primary winding N1 and the number of turns of the secondary winding N2 are substantially the same (for example, 43T). For this reason, for example, the number of turns of the secondary winding N2 becomes appropriate, so that the isolated converter transformer PIT in which a litz wire is wound as the primary winding N1 and the secondary winding N2.
It was difficult to reduce the size of the divided bobbin B (see FIG. 15), and it was not possible to reduce the size and weight of the insulated converter transformer PIT.

On the other hand, in the power supply circuit shown in FIG. 10, the resonance currents I3 and I4 flowing through the rectifier diodes DO1 and DO2 of the voltage doubler rectifier circuit provided on the secondary side are different from those shown in FIG. As shown in FIG. 12C, high-frequency ringing noise is not superimposed when the rectifier diodes DO1 and DO2 are turned on.

In the power supply circuit shown in FIG. 11, the operation waveforms are not shown, but the series resonance currents alternately and continuously flow through the rectifier diodes DO1 to DO4 constituting the quadruple voltage rectifier circuit. Rectifier diode D
High-frequency ringing noise is not superimposed on the resonance current flowing through O1 to DO4.

However, in the power supply circuits shown in FIGS. 10 and 11, as shown in FIG.
In the range of the intermediate load state of the range of W to 120 W, the switching element Q1 operates abnormally as described later. FIG. 14 is a waveform diagram showing operation waveforms of the power supply circuit shown in FIG. 10 in an intermediate load state. Also in this case, the series resonance circuit (N
The primary side parallel resonance voltage Vcp as shown in FIG. 14A is obtained by performing the switching operation of the switching element Q1 by B, CB). In this case, the switching element Q1 is turned off. In the period T1 immediately before the end of the period TOFF, the collector current Icp flows for a short time to the collector of the switching element Q1 as shown in FIG. The waveform of the secondary side resonance current I2 flowing through the secondary winding N2 of the insulating converter transformer PIT has a waveform as shown in FIG.

In this case, as shown in FIGS. 14A and 14B, in the period T1 immediately before the end of the OFF period TOFF of the switching element Q1, the switching element Q1 is in the conductive state, and the switching is performed. Primary side resonance voltage Vcp supplied between the collector and the emitter of the element Q1
(ZVS), which is a so-called resonance-type basic operation that performs a switching operation when is at a zero level.
tching) operation.

Such an operation occurs in the power supply circuit shown in FIG. 10 because the period TOFF during which the switching element Q1 is turned off increases as the load power Po decreases. In the period T1 in which such an abnormal operation occurs, the switching operation is performed with the switching element Q1 having a certain voltage level and a current level, so that the power loss in the switching element Q1 increases. For this reason, it is necessary to enlarge the heat radiating plate for suppressing the heat generation of the switching element Q1.

[0046]

SUMMARY OF THE INVENTION In view of the above-mentioned problems, the present invention prevents a high-frequency ringing current from being superimposed on a secondary-side resonance current flowing through a rectifier diode provided on a secondary side. It is an object of the present invention to provide a switching power supply circuit in which the switching element operates in a ZVS operation even in an intermediate load state.

Therefore, a switching power supply circuit according to the present invention includes a switching element having a switching element for intermittently outputting an input DC input voltage, and an insulating converter for transmitting an output of the switching means to a secondary side. By connecting a transformer, a primary-side voltage resonance circuit inserted so as to make the operation of the switching means a voltage resonance type, and a secondary-side parallel resonance capacitor connected in parallel to the secondary winding of the insulating converter transformer. The secondary parallel resonance circuit formed is combined with a secondary series resonance circuit formed by connecting a secondary series resonance capacitor in series to the secondary winding of the insulating converter transformer. A secondary side resonance circuit. Then, by inputting the alternating voltage obtained in the secondary winding of the insulating converter transformer and performing the rectification operation, the secondary voltage having a level corresponding to 2n times (where n is a natural number of 1 or more) the level of the alternating voltage is obtained. DC output voltage generating means configured to obtain a side DC output voltage, and a constant voltage configured to perform constant voltage control by varying a switching frequency of a switching element according to a secondary DC output voltage level. And a control unit.

According to the above arrangement, the secondary winding is provided by combining the secondary series resonance circuit and the secondary parallel resonance circuit with respect to the secondary winding of the insulating converter transformer. By the resonance operation of the side parallel resonance circuit, the secondary side resonance current flowing through the secondary winding N2 of the insulating converter transformer can be made substantially sinusoidal. This allows
Since the conduction angle of the resonance current flowing through the rectifier diode provided on the secondary side becomes substantially equal, the high-frequency ringing current is not superimposed on the resonance current flowing through the rectifier diode. In addition, the constant voltage control of the secondary side DC output voltage is a combined control that controls the switching frequency and the conduction angle of the switching current flowing through the switching element, so that the switching element is turned off even when the load fluctuates. Can be suppressed, and the switching element can perform the ZVS operation even in the intermediate load state.

[0049]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to an embodiment of the present invention. The power supply circuit shown in this figure is provided with a self-excited voltage resonance type switching converter constituted by one switching element (bipolar transistor), similarly to the power supply circuit described above. In this figure, the same parts as those in FIGS. 6 and 10 are denoted by the same reference numerals, and description thereof will be omitted.

In the power supply circuit of the present embodiment shown in FIG. 1, a voltage doubler half-wave rectifier circuit is provided on the secondary side of the insulating converter transformer PIT, similarly to the power supply circuit shown in FIG. Is a secondary parallel resonance capacitor C2,
The difference from the power supply circuit shown in FIG. 10 is that the secondary side series capacitor Cs is connected in combination. That is,
A secondary-side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2 of the insulating converter transformer PIT, and one end of the secondary winding N2 and a rectifier diode DO1 are connected.
A secondary side series resonance capacitor Cs is inserted in series between the anode of the rectifier diode DO2 and the connection point of the cathode of the rectifier diode DO2.

According to the above configuration, on the secondary side of the power supply circuit of the present embodiment, the voltage resonance circuit is formed by the capacitance of the secondary parallel resonance capacitor C2 and the leakage inductance L2 of the secondary winding N2. Is formed, and a current resonance circuit is formed by the capacitance of the secondary side series resonance capacitor Cs and the leakage inductance L2 of the secondary winding N2. In other words, on the secondary side,
This embodiment employs a configuration in which a voltage resonance circuit and a current resonance circuit are combined in a complex manner, and the secondary winding N2 is commonly provided as an inductance. In this specification, a configuration in which the voltage resonance circuit and the current resonance circuit are combined on the secondary side in this manner is also referred to as a “secondary-side voltage / current resonance circuit”.

FIG. 2 is a diagram showing an example of the operation waveform of each part of the power supply circuit of the present embodiment as described above. In the power supply circuit of this embodiment, the switching operation of the switching element Q1 by the series resonance circuit (NB, CB) as a self-excited oscillation driving circuit allows the parallel connection of the switching element Q1 // parallel resonance capacitor Cr. A primary parallel resonance voltage Vcp as shown in FIG. 2A is obtained at both ends of the connection circuit by the action of the parallel resonance circuit. As shown in the figure, the parallel resonance voltage Vcp has a waveform that becomes a sine-wave pulse during the period TOFF when the switching element Q1 is on and the period TON when the switching element Q1 is off. Also, the collector of the switching element Q1 is shown in FIG.
A collector current ICP having a waveform as shown in FIG.

The waveform of the secondary-side resonance current I2 flowing through the secondary winding N2 of the insulating converter transformer PIT is such that the secondary-side series resonance capacitor Cs and the secondary-side parallel resonance capacitor C2 are connected in combination. Thus, the parallel resonance capacitor C2 and the series resonance capacitor C2
s and the voltage / current resonance operation by the leakage inductance L2 of the secondary winding N2.
As shown in (c), it is substantially sinusoidal. In this case, the resonance current I3 flowing through the rectifier diodes DO1, DO2
, I4 have substantially the same conduction angle, and the resonance current I3
, I4 are as shown in FIGS. 2 (d) and 2 (e).

FIG. 10 shows a comparison between the power supply circuit of this embodiment and the power supply circuit shown in FIG. 10 in which the secondary parallel resonance capacitor C2 is not inserted into the secondary winding N2. In the power supply circuit shown, the period TOFF in which the switching element Q1 is turned off is expanded as the load power Po becomes lighter load. For example, in an intermediate load state, immediately before the end of the period TOFF in which the switching element Q1 is turned off. In the period T1, the switching element Q1 is turned on and the collector current ICP flows as shown by the broken lines in FIGS. 14 (a) and 14 (b). On the other hand, in the power supply circuit according to the present embodiment, as shown in FIGS. 2A and 2B, even during the intermediate load state, the period TOFF during which the switching element Q1 is turned off, as described later, is almost zero. Since the current does not expand, the collector current ICP does not flow through the switching element Q1 during the period TOFF. Thereby, abnormal operation in the intermediate load state is prevented, and stable ZV
S operation is performed. That is, stable ZVS operation is realized in the entire load range that can be supported. Also, since the abnormal operation in the intermediate load state is prevented, the power loss caused by the abnormal operation is also eliminated,
Since the power conversion efficiency in the intermediate load state can be improved and the heat generated by the switching element Q1 is reduced, it is not necessary to enlarge the heat sink attached to the switching element Q1.

Further, in the power supply circuit of the present embodiment, FIG.
(D), as shown in FIG.
Since the conduction angles of the resonance currents I3 and I4 flowing through O1 and DO2 are substantially equal, for example, in the power supply circuit shown in FIG. 6, when the four rectification diodes constituting the bridge rectification circuit DBR are turned on. The generated high-frequency ringing current is not superimposed on the resonance currents I3 and I4. Thus, in the power supply circuit of the present embodiment, EMI is hardly radiated from the rectifier diodes DO1 and DO2, and therefore, for example, in the power supply circuit shown in FIG.
As a measure against EMI, it is possible to eliminate the ferrite bead inductor and the ceramic capacitor that actually need to be provided, and accordingly, the number of parts can be reduced.

According to experiments, for example, the number of windings of the primary winding N1 of the insulating converter transformer PIT is 43T, the number of windings of the secondary winding N2 is 23T, and the primary side resonance capacitor Cr = 3.
300 pF, secondary side series resonance capacitor Cs = 0.06
8 .mu.F, secondary side parallel resonance capacitor C2 = 6800 pF
In this case, it has been confirmed that the power supply circuit of the present embodiment operates optimally.

Here, FIG. 3 is a diagram showing the constant voltage control characteristics when the load is varied in the power supply circuit of the present embodiment in which the values of the respective components are set by the above design. . In this case, the input AC voltage VAC is also 10
0V and the secondary side DC output voltage EO1 is 135V. As shown in FIG. 3, in the power supply circuit of the present embodiment, as the constant voltage control of the DC output voltage EO1 output from the secondary side, as the load power Po increases, the switching frequency fs increases. It is controlled so as to be lower and the period TON during which the switching element Q1 is on is controlled to be longer. That is, it can be seen that the composite control method is used as the constant voltage control operation.

Accordingly, in the power supply circuit of the present embodiment, the change in the period TOFF during which the switching element Q1 is turned off due to the change in the load is, for example, the period TOFF during which the switching element Q1 of the power supply circuit is turned off shown in FIG. Change (Fig. 1
3), and the change amount is small. This also indicates that even when the power supply circuit according to the present embodiment has an intermediate load state, the period TOFF during which the switching element Q1 is off does not increase. Therefore, the power supply circuit of the present embodiment does not have an abnormal operation in which the switching operation deviates from the ZVS operation in the intermediate load state, as in the power supply circuit shown in FIG.

In the power supply circuit of the present embodiment, the switching element Q1 can be controlled by the complex control method, so that the maximum load power PoMAX that can be supported is 200.
In addition to an increase from W to 220 W, it is possible to increase the controllable operation range.

Further, the power supply circuit of the present embodiment
Since the secondary side is provided with a voltage doubler rectifier circuit for obtaining a DC output voltage corresponding to almost twice the level of the excitation voltage of the secondary winding N2, for example, the secondary winding shown in FIG. The number of turns of the secondary winding N2 can be reduced as compared with a power supply circuit that outputs a DC output voltage corresponding to a level equal to the excitation voltage of the line N2. It is also possible to reduce the weight.

The secondary side of the power supply circuit according to the present embodiment is not limited to the voltage doubler half-wave rectifier circuit shown in FIG. Therefore, a configuration on the secondary side is shown in FIGS. 4 and 5 as a modified example of the power supply circuit of the present embodiment. In FIGS. 4 and 5, the configuration on the primary side is the same as the configuration in FIG. 1 and 6 are denoted by the same reference numerals and description thereof is omitted.

FIG. 4 shows a configuration on the secondary side as a first modification. On the secondary side shown in this figure, a series resonance capacitor Cs, a rectifier diode DO1,
By connecting DO2 and the smoothing capacitors CO10 and CO11, a so-called voltage doubler full-wave rectifier circuit is formed.
As a connection form of the above elements, one end of the secondary winding N2 of the insulating converter transformer PIT is connected to a series resonance capacitor C.
s is connected to the connection point of the anode of the rectifier diode DO1 and the cathode of the rectifier diode DO2 via the series connection of s. The cathode of the rectifier diode DO1 is connected to the positive electrode of the smoothing capacitor CO10, and the anode of the rectifier diode DO2 is connected to the secondary ground. On the other hand, the other end of the secondary winding N2 of the insulating converter transformer PIT is connected to a connection point of the series connection of the negative electrode of the smoothing capacitor CO10 and the smoothing capacitor CO11, and the negative electrode side of the smoothing capacitor CO11 is connected to the secondary side ground. Connected. The point that the secondary parallel resonance capacitor C2 is provided in parallel with the secondary winding N2 is the same as the case of FIG.

The rectification operation of the voltage doubler full-wave rectifier circuit formed by the above-described connection form is as follows. When the switching output on the primary winding N1 is obtained by the switching operation on the primary side, the switching output is excited by the secondary winding N2, and the alternating voltage obtained on the secondary winding N2 is input to perform the rectification operation. Do. In this case, first, during a period in which the rectifier diode DO1 is turned off and the rectifier diode DO2 is turned on, the rectifier diode DO1 operates in the reduced polarity mode in which the polarity of the primary winding N1 and the secondary winding N2 is -M, and the rectifier operates. The rectified current rectified by the diode DO2 flows through the series resonance capacitor Cs → the secondary winding N2 → the smoothing capacitor CO11, whereby the operation of charging the smoothing capacitor CO11 is obtained. During the period in which the rectifier diode DO2 is turned off and the rectifier diode DO1 is turned on and the rectification operation is performed, the polarity of the primary winding N1 and the secondary winding N2 becomes + M, and the secondary winding is turned on. The current induced in the line N2 flows through the series resonance capacitor Cs → the rectifier diode DO1 → the smoothing capacitor CO10, so that the smoothing capacitor CO1
This is an operation in which charging is performed for 0. In this manner, the full-wave rectification operation using both the additive polarity mode (+ M; forward operation) and the depolarization mode (-M; flyback operation) is performed, so that the smoothing connected in series is performed. The secondary winding N is connected to both ends of the capacitors CO10 and CO11.
A DC voltage (rectified smoothed voltage) EO1 corresponding to almost twice the induced voltage of 2 is obtained.

When the secondary side of the power supply circuit according to the present embodiment has such a configuration, for example, the number of secondary windings N2 of the insulating converter transformer PIT is 20T, and the secondary side series resonance capacitor Cs = By setting 0.1 .mu.F and the secondary side parallel resonance capacitor C2 = 0.01 .mu.F, the optimum operation was obtained, and the same effect as that of the power supply circuit shown in FIG. Further, when the secondary side is constituted by a voltage doubler full-wave rectifier circuit as shown in FIG. 4, compared with the power supply circuit shown in FIG. 1 in which the secondary side is a voltage doubler half-wave rectifier circuit. Also, since the number of turns of the secondary winding N2 can be reduced from 23T to 20T, the size and weight of the divided bobbin B can be further reduced.

FIG. 5 shows a configuration on the secondary side as a second modification. On the secondary side shown in this figure, a quadruple voltage composed of series resonance capacitors Cs1, Cs2, rectifier diodes DO1, DO2, and smoothing capacitors CO10, CO11 is applied to the secondary winding N2 of the insulating converter transformer PIT. A wave rectifier circuit is provided, and a secondary side parallel resonance capacitor C2 is provided for the secondary winding N2. Note that the rectifying operation of the quadruple voltage full-wave rectifier circuit has been described above with reference to FIG.

In the case where the secondary side of the power supply circuit according to the present embodiment has such a configuration, for example, the number of windings of the secondary winding N2 of the insulating converter transformer PIT is 14T, and the secondary side series resonance capacitor Cs1, By setting Cs2 = 0.1 .mu.F and the secondary side parallel resonance capacitor C2 = 0.022 .mu.F, an optimum operation was obtained, and the same effect as that of the power supply circuit shown in FIG. 1 was obtained. In this case, the insulation converter transformer PI
Since the number of turns of the secondary winding N2 of T can be reduced to 14T, the size and weight of the divided bobbin B can be further reduced.

In the present embodiment, the excitation voltage applied to the secondary winding N2 of the insulating converter transformer PIT is approximately 2 to the secondary side of the switching power supply circuit.
A voltage doubler half-wave rectifier, a voltage doubler full-wave rectifier, and a secondary winding N that output a DC output voltage EO1 corresponding to the double level.
2, a quadruple voltage full-wave rectifier circuit that outputs a DC output voltage EO1 corresponding to a level approximately four times higher than the excitation voltage to be excited is provided as an example. The present invention is not limited to this. The present invention provides a secondary output voltage EO1 having a level corresponding to 2n times (n is a natural number of 1 or more) the excitation voltage level excited in the secondary winding N2. What is necessary is just to provide a rectifier circuit configured to obtain the same. Therefore, for example, a rectifier circuit for generating a DC output voltage EO1 corresponding to a level approximately six times, eight times... (2n) times the excitation voltage excited by the secondary winding N2 may be provided. Things.

The power supply circuit of the present invention may be appropriately modified in accordance with actual use conditions, in addition to the configuration shown in FIGS. 1, 4 and 5. For example, in each of the above embodiments, a configuration of switching drive by a self-excited system is adopted, but the present invention is also applicable to a configuration in which a switching element is driven by a separately excited system. Further, as the switching element, other component elements other than the bipolar transistor and the MOS-FET may be employed.

[0069]

As described above, according to the present invention, a secondary side resonance circuit formed by combining a secondary side series resonance circuit and a secondary side parallel resonance circuit with respect to the secondary winding of an insulating converter transformer is provided. It is formed. In this case, first, the secondary-side resonance current flowing through the secondary winding of the insulating converter transformer becomes substantially sinusoidal due to the resonance operation of the secondary-side parallel resonance circuit. The conduction angle of the current flowing through the rectifier diode is also substantially equal. Accordingly, EMI generated from the rectifier diode can be suppressed without superimposing a high-frequency ringing current on the current flowing through the rectifier diode. Therefore, there is no need to provide a component for EMI countermeasures on the secondary side, and the circuit scale can be reduced accordingly.

Further, since the constant voltage control of the secondary side DC output voltage is a combined control for controlling the switching frequency and the conduction angle of the switching current flowing through the switching element, the switching element is turned off even when the load fluctuates. The expansion of the period can be suppressed, so that the switching element does not deviate from the ZVS operation even in the intermediate load state. As a result, the abnormal operation that has occurred in the intermediate load state is resolved, and as a result, a stable ZVS operation can be realized in the entire region within the load fluctuation range that can be handled. Also, by obtaining the ZVS operation,
Since the power loss in the switching element is also reduced, the power conversion efficiency can be improved, and there is no need to increase the size of the heat sink attached to the switching element.

Furthermore, the constant voltage control of the secondary DC output voltage output from the DC output voltage generating means is a composite control for controlling the switching frequency and the conduction angle of the switching current flowing through the switching element. There is also an advantage that the load power can be increased and the controllable range can be expanded.

The DC output voltage generating means includes, for example, a double voltage half-wave rectification operation, a double voltage full-wave rectification operation for generating a DC output voltage corresponding to almost twice the alternating voltage obtained in the secondary winding, and the like. Alternatively, if the configuration is such that a quadruple voltage full-wave rectification operation for generating a DC output voltage corresponding to almost four times the alternating voltage is performed, the number of windings of the insulating converter transformer can be reduced. It is possible to reduce the size and weight of the transformer.

[Brief description of the drawings]

FIG. 1 is a circuit diagram illustrating a configuration example of a power supply circuit according to an embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part of the power supply circuit of the present embodiment.

FIG. 3 is a diagram showing constant voltage control characteristics when the load of the power supply circuit of the present embodiment fluctuates.

FIG. 4 is a circuit diagram showing a secondary-side configuration example of a power supply circuit as a first modified example.

FIG. 5 is a circuit diagram showing a secondary-side configuration example of a power supply circuit as a second modification.

FIG. 6 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 7 is a circuit diagram showing another configuration of the secondary side of the power supply circuit according to the prior art.

8 is a waveform chart showing an operation of a main part of the power supply circuit according to the prior art shown in FIG.

9 is a diagram showing constant voltage control characteristics when the load of the power supply circuit as the prior art shown in FIG. 6 fluctuates.

FIG. 10 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 11 is a circuit diagram showing another configuration of the secondary side of the power supply circuit according to the prior art.

12 is a waveform chart showing a secondary-side operation of the power supply circuit shown in FIG. 10 as the prior art.

FIG. 13 is a diagram showing constant voltage control characteristics when the load of the power supply circuit as the prior art shown in FIG. 10 fluctuates.

14 is a waveform chart showing a primary-side operation of the power supply circuit shown in FIG. 10 as the prior art.

FIG. 15 is a cross-sectional view illustrating a configuration of an insulating converter transformer.

FIG. 16 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

[Explanation of symbols]

1 control circuit, Ci smoothing capacitor, Q1 switching element, PIT isolation converter transformer, PRT
Quadrature control (drive) transformer, Cr primary parallel resonance capacitor, C2 secondary parallel resonance capacitor, Cs
Cs1 Cs2 Secondary side series resonance capacitor, NC
Control winding, NB drive winding, ND detection winding, CB resonance capacitor, DO1 DO2 DO3 DO4 rectifier diode, CO1 CO2 CO1 CO10 CO11 smoothing capacitor

Claims (4)

[Claims] A switching means for intermittently outputting an input DC input voltage; an insulating converter transformer for transmitting an output of the switching means to a secondary side; A primary-side voltage resonance circuit inserted so as to be a resonance type, and a secondary-side parallel resonance formed by connecting a secondary-side parallel resonance capacitor in parallel to the secondary winding of the insulating converter transformer. A secondary-side resonance circuit formed by combining a circuit and a secondary-side series resonance circuit formed by connecting a secondary-side series resonance capacitor in series to the secondary winding of the insulating converter transformer; and By inputting the alternating voltage obtained in the secondary winding of the insulating converter transformer and performing rectification, 2n times the level of the alternating voltage (where n is 1) DC output voltage generating means configured to obtain a secondary-side DC output voltage at a level corresponding to the above natural number), and changing a switching frequency of the switching element according to the secondary-side DC output voltage level. And a constant voltage control means for performing constant voltage control. 2. The DC output voltage generating means includes two sets of rectifier diodes and one set of smoothing capacitors, and is formed by inserting the secondary side series resonance capacitor into a rectification current path. 2. The switching power supply circuit according to claim 1, wherein a double voltage half-wave rectifying operation is performed to generate a secondary side DC output voltage having a level corresponding to twice the alternating voltage level. 3. The DC output voltage generation means includes two sets of rectifier diodes and two sets of smoothing capacitors, and is formed by inserting the secondary side series resonance capacitor into a rectification current path. 2. The switching power supply circuit according to claim 1, wherein a doubled voltage full-wave rectifying operation is performed to generate a secondary side DC output voltage having a level corresponding to twice the alternating voltage level. 4. The DC output voltage generating means includes two sets of rectifier diodes and two sets of smoothing capacitors, and inserts two sets of the secondary-side series resonance capacitors into a rectified current path. 2. A quadruple voltage full-wave rectifying operation for generating a secondary side DC output voltage having a level corresponding to four times the alternating voltage level by being formed. Switching power supply circuit.