JP2001103751A - Switching power circuit and isolation converter transformer - Google Patents

Abstract

PROBLEM TO BE SOLVED: To suppress electric power loss in the secondary side. SOLUTION: A tertiary winding N3 provided on the secondary side of an isolation converter transformer PIT is wound to obtain lose coupling between a primary winding N1 provided on the primary side of the isolation converter transformer PIT and a secondary winding N2 provided on the secondary side, thus it is possible to reduce the peak current of secondary current I4 outputted from a second half-wave rectifier circuit 3 with the tertiary winding N3 of the isolation converter transformer PIT.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an insulated converter transformer and a switching power supply circuit provided as an electric power source for various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a circuit employing a switching converter of a type such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular switching operation waveform, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It has been found that there is a limit in improving the power conversion efficiency. Therefore, the present applicant has previously proposed various switching power supply circuits using various resonance type converters.
The resonance type converter can easily obtain high power conversion efficiency and realize low noise because the switching operation waveform is sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

FIG. 7 is a circuit diagram showing an example of a switching power supply circuit that can be configured based on the invention proposed by the present applicant. This power supply circuit is provided with a voltage resonance type converter having a single switching element Q1 and performing a switching operation by a self-excited method in a so-called single-ended system.

In the power supply circuit shown in FIG. 1, a bridge rectification circuit Di is used as a rectification and smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) and obtaining a DC input voltage.
And a full-wave rectifier circuit composed of a smoothing capacitor Ci. The full-wave rectifier circuit generates a rectified smoothed voltage Ei corresponding to, for example, a level one time higher than the AC input voltage VAC. Further, in the rectifying / smoothing circuit, an inrush current limiting resistor Ri is inserted in the rectifying current path to suppress, for example, an inrush current flowing into the smoothing capacitor Ci when the power is turned on.

A voltage resonance type switching converter provided in this power supply circuit has a single switching element Q1.
It has a self-excited configuration with In this case, a high withstand voltage bipolar transistor (BJT: junction type transistor) is employed as the switching element Q1.

The base of the switching element Q1 is connected to a smoothing capacitor Ci (rectified smoothing voltage E) via a starting resistor RS.
It is connected to the positive electrode side of i) so that the base current at the time of starting can be obtained from the rectification smoothing line. A drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB are provided between the base of the switching element Q1 and the primary side ground.
A series resonance circuit for driving self-excited oscillation is connected. A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the switching element Q1 is turned off. , The collector of switching element Q1 is connected to one end of primary winding N1 of insulating converter transformer PIT, and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr
Forms a primary parallel resonance circuit of a voltage resonance type converter by its own capacitance and a leakage inductance L1 on the primary winding N1 side of an insulating converter transformer PIT described later. Although the detailed description is omitted here, when the switching element Q1 is turned off, the voltage Vcp across the resonance capacitor Cr actually becomes a sinusoidal pulse waveform due to the action of the parallel resonance circuit due to the operation of the voltage resonance type. Is obtained.

The orthogonal control transformer PRT shown in FIG.
Is a saturable reactor on which a detection winding ND, a drive winding NB, and a control winding NC are wound. This orthogonal transformer PRT drives the switching element Q1 and
Provided for constant voltage control. As the structure of the orthogonal control transformer PRT, although not shown, a three-dimensional core is formed by joining the ends of the two magnetic legs of two double U-shaped cores having four magnetic legs. I do. Then, a detection winding ND and a drive winding NB are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is further connected to the detection windings ND and ND. It is constructed by being wound in a direction orthogonal to the drive winding NB.

In this case, the detection winding ND of the orthogonal control transformer PRT is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1 of the insulating converter transformer PIT, so that the switching of the switching element Q1 is performed. The output is transmitted to the detection winding ND via the primary winding N1. In the orthogonal control transformer PRT, the detection winding N
When the switching output obtained at D is excited by the drive winding NB via the transformer coupling, an alternating voltage as a drive voltage is generated in the drive winding NB. This drive voltage is applied to a series resonance circuit (N
B, CB) is output as a drive current to the base of the switching element Q1 via the base current limiting resistor RB. As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit (NB, CB).

The insulating converter transformer PIT has a structure of an insulating converter transformer PIT for transmitting the switching output of the switching element Q1 to the secondary side, as shown in FIG.
1. An EE-type core in which CR2 and CR2 are combined so that their magnetic legs face each other is provided. A primary winding N1 and a secondary winding The wires N2 (and N2A) are wound separately. A gap G is formed for the center magnetic leg as shown in the figure. As a result, loose coupling with a required coupling coefficient can be obtained. Gap G
Can be formed by making the central magnetic legs of the E-shaped cores CR1 and CR2 shorter than the two outer magnetic legs. Also,
As the coupling coefficient k, for example, a loosely coupled state of k ≒ 0.85 is obtained, so that a saturated state is hardly obtained.

Here, the primary winding N1 and the secondary winding N2 (and N2A) wound on the split bobbin B of the insulating converter transformer PIT will be described with reference to FIGS. FIG. 10 is a diagram schematically showing how to wind the primary winding N1 and the secondary winding N2 (and N2A) wound around the divided bobbin B. The split bobbin B has a primary winding N
Regions for winding the primary winding N2 and the secondary winding N2 (and N2A) are separately provided. Here, the winding width of the primary winding N1 wound around the divided bobbin B is shown as a bobbin inner winding width K1 shown in the drawing, and the winding width of the secondary winding N2 (and N2A) wound around the divided bobbin B is shown. The winding width is shown as a bobbin inner winding width K2.

In this case, the primary winding N1 is wound in a predetermined direction from a predetermined winding start position N1S. Then, when the primary winding N1 is wound to the end of the bobbin inner winding width K1, the primary winding N1 that has been wound so far is placed on the primary winding N1 that has been wound so far.
It is made to wind in the opposite direction. The primary winding N1 is wound in such a winding manner, and the primary winding N1 is wound.
Is wound by a predetermined number of turns.
At the winding end position N1E. For example, in FIG. 10, the winding start position N1S of the primary winding N1 is the center side (lower side) of the divided bobbin B and the position farthest from the secondary winding N2 (left side position).
The left side of the outer side (upper surface side) of the divided bobbin B, which is the winding end position N1E, while alternately changing the winding direction from the winding start position N1S, such as rightward → leftward → rightward →. It is schematically shown that the primary winding N1 is wound up to the position.

The winding of the secondary winding N2 (and N2A) is also performed in the same manner as in the case of the primary winding N1, by changing the winding direction alternately within the bobbin winding width K2 from the predetermined winding start position N2S. It will be wound by the number of turns. However, since the secondary winding N2 in this case is configured so that a part thereof becomes the secondary winding N2A as described later, the secondary winding N2 is once drawn out as the intermediate tap CT. After that, the secondary winding N2A is wound. The number of turns of the secondary winding N2A in this case is, for example, the secondary winding N2A.
The output level of 2A is set to 1/1 of the output level of secondary winding N2.
Assuming that it is about 0, the number of turns of the secondary winding N2 is about 1/6 to
This is 1/10, which is 5 turns or less. For example, FIG.
, The secondary winding N2 started to be wound from the winding start position N1S is drawn out as an intermediate tap CT at the winding end position N2E, and the winding end position N2A is changed from the winding start position N2AS to the secondary winding N2A.
It is schematically shown that winding is performed up to E.

Incidentally, both ends of the primary winding N1 and the secondary winding N2 (and N2A) drawn from the winding start positions N1S and N2S and the winding end positions N1E and N2AE, and the intermediate tap CT
Are soldered to predetermined pin terminals P, P,... Provided on the upper surface of the divided bobbin B, for example.

The primary winding N1 wound around the bobbin winding width K1 of the divided bobbin B and the secondary winding N2 (and N2A) wound around the bobbin winding width K2 are as shown in FIG. In, for example, aligned without providing a gap,
That is, the adjacent primary windings N1 are wound so as to be in contact with each other.

One end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1, and the other end of the primary winding N1 is connected to the positive electrode of the smoothing capacitor Ci through a series connection of the detection winding ND as shown in the figure. Rectified smoothing voltage E
i).

On the secondary side of the insulation converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, for the secondary winding N2,
When the secondary parallel resonance capacitor C2 is connected in parallel, a parallel resonance circuit is formed by the leakage inductance L2 (and L2A) of the secondary winding N2 (and N2A) and the capacitance of the secondary parallel resonance capacitor C2. Is done. With this parallel resonance circuit, the secondary winding N2 (and N2
The alternating voltage excited in 2A) becomes a resonance voltage. That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for making the switching operation a voltage resonance type, and the secondary side is provided with a parallel resonance circuit for obtaining a half-wave rectification operation (voltage resonance operation). A resonance circuit is provided. In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

For the secondary-side parallel resonance circuit formed as described above, an intermediate tap is provided for the secondary winding N2 so that the rectifier diodes DO1, DO2 and the smoothing capacitors CO1, CO2 are connected. By connecting as shown in the figure, a first half-wave rectifier circuit 2 composed of a set of [rectifier diode DO1, smoothing capacitor CO1] and a second half-wave rectifier circuit composed of a set of [rectifier diode DO2, smoothing capacitor CO2]. Wave rectification circuit 3
Is provided. The first half-wave rectifier circuit 2 receives the resonance voltage supplied from the secondary side parallel resonance circuit and receives a DC output voltage E.
O1 is generated, and the second half-wave rectifier circuit 3 similarly receives the resonance voltage supplied from the secondary side parallel resonance circuit and generates the DC output voltage EO2. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

Incidentally, the insulation converter transformer PIT
, The inductance L of the primary winding N1 depends on the relationship between the polarity (winding direction) of the primary winding N1, the secondary windings N2 and N2A and the connection of the rectifier diode DO (DO1, DO2).
1 and the mutual inductance M between the inductances L2 and L2A of the secondary windings N2 and N2A may be + M or -M. For example, when the connection configuration shown in FIG. 9A is employed, the mutual inductance is + M, and when the connection configuration shown in FIG. 9B is employed, the mutual inductance is -M. If this is made to correspond to the operation on the secondary side shown in FIG. 7, for example, the first half-wave rectifier circuit 2
For example, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, a rectified current flows through the rectifier diode DO1 to perform + M operation mode (forward mode).
When the alternating voltage obtained in FIG. 2 has a negative polarity, the rectifying diode DO is turned off, and no rectifying current flows. That is, in this power supply circuit, the primary winding N1 and the secondary winding N
The rectifying operation is performed in a mode where the mutual inductance with 2 is + M.

In such a configuration, the electric power is supplied to the load side increased by the action of the primary side parallel resonance circuit and the secondary side parallel resonance circuit, and the electric power supplied to the load side is also increased, thereby increasing the maximum load. The rate of increase in power also increases. this is,
As described above with reference to FIG. 8, the gap G is formed in the insulating converter transformer PIT so as to be loosely coupled with a required coupling coefficient, thereby achieving a state in which a saturation state is further reduced. Things. For example,
If no gap G is provided for the insulated converter transformer PIT, there is a high possibility that the insulated converter transformer PIT will be in a saturated state during flyback operation and the operation will be abnormal. It is difficult to want to be done.

In the control circuit 1, the level of the control current (DC current) flowing through the control winding NC is varied in accordance with the change in the secondary DC output voltage level EO1, so that the control current is wound around the orthogonal control transformer PRT. The inductance LB of the driving winding NB is variably controlled. Thus, the switching element Q including the inductance LB of the driving winding NB is formed.
The resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for 1 changes. This, as explained in FIG.
The operation of varying the switching frequency of the switching element Q1 is performed.
Stabilizes O1.

Here, in the case where an orthogonal control transformer PRT having a variable control structure is provided for the inductance LB of the drive winding NB as shown in FIG. 7, the switching element Q1 is turned off to change the switching frequency. The period TOFF is kept constant, and the ON period TON is variably controlled. In other words, in this power supply circuit, as a constant voltage control operation, an operation is performed to variably control the switching frequency, thereby performing resonance impedance control on the switching output, and at the same time, controlling the conduction angle of the switching element in the switching cycle (PWM). Control). This complex control operation is realized by a set of control circuit systems.

That is, in such a power supply circuit, the secondary-side DC output voltage EO1 output from the first half-wave rectifier circuit 2
Is supplied to the control circuit 1 as a detection voltage, and the resonance voltage level obtained from the secondary winding N2 of the insulating converter transformer PIT is variably controlled, so that the secondary side DC output voltage EO1 is made constant. However, the secondary DC output voltage EO2 output from the second half-wave rectifier circuit 3 is a constant voltage of the secondary DC output voltage EO2 supplied as an operating voltage to the control circuit 1. It is not planned.

FIG. 12 is a diagram showing an example of an operation waveform of each part of the power supply circuit shown in FIG. 7, and mainly shows an output waveform on the secondary side. The output waveform shown in FIG. 12 is obtained by, for example, applying the maximum load power PoMAX =
It is assumed that the optimum drive condition is obtained at 217 W, and the AC input voltage VAC is 100 V
The isolated converter transformer PIT has, for example, a secondary winding in order to obtain 135 V as a secondary-side DC output voltage EO1 that has been converted into a constant voltage and 15 V as a secondary-side DC output voltage EO2 that has not been converted into a constant voltage. A winding of 38 turns is applied as the line N2, and a winding of 5 turns is applied as the secondary winding N2A.

In such a circuit, the switching operation of the switching element Q1 by a series resonance circuit (NB, CB) as a self-excited oscillation driving circuit allows the switching element Q1 // parallel resonance capacitor Cr to be connected in parallel. At both ends, the action of the parallel resonance circuit
A primary side parallel resonance voltage Vcp as shown in FIG. As shown in the figure, the parallel resonance voltage Vcp is at the 0 level during the period when the switching element Q1 is on,
In the OFF period TOFF, a waveform of a sine wave pulse is obtained, which corresponds to the operation as the voltage resonance type.

The switching output is transmitted to the secondary side of the insulated converter transformer PIT by the on / off operation of the switching element Q1.
A secondary resonance current I2 having a waveform as shown in FIG. 12 (b) flows through the connection end between the secondary winding N2 and the rectifier diode DO1. 12 (c)
A secondary resonance voltage V2 as shown in FIG. In this case, the rectifier diode DO1 becomes conductive at a timing when the secondary resonance voltage V2 shown in FIG. 12C becomes higher than the level of the secondary DC output voltage EO1. A secondary side rectified current I3 having a waveform as shown in FIG.

On the other hand, at both ends of the secondary winding N2A,
A secondary resonance voltage V3 having a waveform as shown in FIG. In this case, the waveform of the secondary side resonance voltage V3 is shown in FIG.
The waveform is similar to the waveform of the secondary side resonance voltage V2 shown in (c), and the similarity ratio is determined by the number of turns of the entire secondary winding N2 (38 turns) and the number of turns of the secondary winding N2A. (5 turns)
, For example, the secondary side resonance voltage V
2 multiplied by 5/38. Also in this case, the rectifier diode DO2 shown in FIG. 7 becomes conductive at the timing when the secondary resonance voltage V3 shown in FIG. 12E becomes higher than the level of the secondary DC output voltage EO2. The secondary side rectified current I4 flows during the period (6 .mu.S) shown in FIG.
5 Ap.

[0029]

The above-mentioned FIG.
The operation waveform of the secondary side resonance voltage V2 shown in FIG.
The operation waveform of the secondary side resonance voltage V3 shown in FIG.
Although having a similar waveform, the secondary-side rectified current I3 flowing through the rectifier diode DO1 shown in FIG.
The waveform is different from the secondary side rectified current I4 flowing through the rectifier diode DO2 shown in (f). This is attributed to the fact that the secondary winding N2A wound around the divided bobbin B of the insulating converter transformer PIT is loosely coupled to the primary winding N1 and the secondary winding N2. You. This is because the number of turns of the secondary winding N2A is 5 turns or less as described above. For example, if the secondary winding N2A is wound around the bobbin inner winding width K2 of the divided bobbin B by aligned winding, for example, As shown in FIG. 11, the secondary winding N2A has a bobbin inner winding width K2.
For example, it is caused by being wound around only a part such as the right side.

As described above, the secondary winding N2A is connected to the secondary winding N2.
Is wound unevenly with respect to the winding width K2 in the bobbin of the divided bobbin B to be wound, for example, the load power supplied from the secondary side DC output voltage EO2 that is not constant voltage is 10 W As described above, the output current output from the secondary winding N2A is such that the + M operation mode (forward converter operation) is superimposed on the -M operation mode (flyback converter operation). As a result, the secondary-side rectified current I3 is superimposed on the secondary-side rectified current I4, and the secondary-side rectified current I4 has a peak in the latter half of the conduction angle of the rectifier diode DO2, as shown in FIG. The waveform has a value. When the peak current value of the secondary side rectified current I4 flowing through the rectifier diode DO2 increases, the heat generation of the rectifier diode DO2 increases, so that the power loss in the rectifier diode DO2 increases and the temperature increases due to the heat generation. Reliability will be impaired.

FIG. 13 is a diagram showing the relationship between the secondary DC output voltages EO1 and EO2 and the load current IL2 supplied from the second half-wave rectifier circuit 3 in the power supply circuit shown in FIG. is there. In FIG. 13, a straight line indicated by a white circle “○” indicates a secondary-side DC output when the power supply circuit shown in FIG. 7 is configured to have optimal driving conditions at the maximum load power PoMAX = 217 W. The relationship between the voltage EO2 and the load current IL2 is shown. Also, white triangle “△”
When the power supply circuit shown in FIG. 7 is configured to have the optimum driving condition at the maximum load power PoMAX = 82 W, the secondary side DC output voltage EO2
And the load current IL2.

As can be seen from FIG. 13, when the load current IL2 changes from 0A to 1.0A when the power supply circuit is configured to support the maximum load power POMAX = 217W, the secondary side DC output voltage EO2 The voltage fluctuation level ΔEO2 is about 6.
6V. Similarly, the power supply circuit is connected to the maximum load power POM.
AX = 82W, the load current IL2
Changes from 0A to 1.0A, the voltage fluctuation level ΔEO2 of the secondary side DC output voltage EO2 becomes about 6.4V. That is, in any case, the secondary-side DC output voltage level EO2 greatly fluctuates with the fluctuation of the load current IL2 output from the second half-wave rectifier circuit 3, so that the cross regulation deteriorates. become.

For this reason, for example, a local regulator of 12 V is connected to the output of the second half-wave rectifier circuit 3 so that a stable constant voltage is output from the local regulator by increasing the load current IL2. Accordingly, even when the voltage level of the secondary side DC output voltage EO2 decreases, it is necessary to keep the secondary side DC output voltage EO2 at or above a predetermined voltage level. However, in this case, conversely, when the level of the load current IL2 decreases, the voltage level of the secondary side DC output voltage EO2 increases, and the power loss in the local regulator increases as the voltage level increases.

Also, when the maximum load power PO of the load to which the secondary side DC output voltage EO1 output from the first half-wave rectifier circuit 2 is changed, the second half-wave rectifier circuit 3 The level of the secondary side DC output voltage EO2 output from the output fluctuates. For example, as shown in FIG. 13, the maximum load power Po = 217 of the load to which the secondary side DC output voltage EO1 is supplied.
The relationship between the voltage level of the secondary side DC output voltage EO2 and the load current IL2 is different between the case of W and the case of the maximum load power Po = 82W. This means that the level of the secondary-side DC output voltage EO2 also fluctuates due to the fluctuation of the load connected to the secondary-side DC output voltage EO1 that has been converted into a constant voltage. Cross regulation was also deteriorated by the level fluctuation of the DC output voltage EO2.

[0035]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. That is, a rectifying / smoothing means for receiving a commercial AC power supply, generating a rectified smoothed voltage and outputting the rectified smoothed voltage as a DC input voltage, winding a primary winding on a primary side, and a secondary winding on a secondary side as a secondary winding. 1 and a second secondary winding having a predetermined number of turns or less, and a required coupling coefficient that is loosely coupled between the primary winding and the first secondary winding is obtained. And an insulating converter transformer provided for transmitting an output on the primary side to the secondary side. A switching unit that includes a switching element and is configured to intermittently output a DC input voltage to output to a primary winding of the insulating converter transformer; and a leakage inductance component including at least a primary winding of the insulating converter transformer; A primary side resonance circuit formed by the capacitance of the side resonance capacitor and making the operation of the switching means a resonance type.
Also, by connecting a secondary resonance capacitor to at least one of the secondary windings of the insulating converter transformer, a leakage inductance component of the winding to which the secondary resonance capacitor is connected, A secondary resonance circuit forming a resonance circuit by the capacitance of the side resonance capacitor; and obtaining a first secondary DC output voltage from an alternating voltage excited in the first secondary winding, DC output voltage generating means configured to obtain a second secondary DC output voltage from an alternating voltage excited by the secondary winding; and a switching element based on the level of the first secondary DC output voltage. Constant voltage control means for performing constant voltage control of the first secondary-side DC output voltage by varying the switching frequency of the first DC voltage. Then, the second secondary winding of the insulating converter transformer is wound so as to obtain a state of being tightly coupled to the primary winding and the first secondary winding.

The insulating converter transformer of the present invention includes a primary winding wound on the primary side, a first secondary winding wound on the secondary side, and a second winding having a predetermined number of turns or less. An EE type in which a required coupling coefficient for loose coupling is obtained for the primary winding and the first secondary winding by forming a gap between the secondary winding and the center magnetic leg. And a second secondary winding wound around the primary winding and the first secondary winding so as to obtain a tightly coupled state.

According to the above configuration, the second secondary winding provided on the secondary side of the insulating converter transformer is replaced with the primary winding provided on the primary side of the insulating converter transformer and the first secondary winding provided on the secondary side. The secondary winding of the second converter is wound so as to be in a tightly coupled state, so that the secondary current generated by the flyback converter operation of the insulating converter transformer is suppressed in the second secondary winding. I have. Further, the second secondary winding provided on the secondary side is tightly coupled with the primary winding provided on the primary side and the first secondary winding provided on the secondary side. By configuring the switching power supply circuit using the wound insulating converter transformer, it is possible to suppress the peak current value of the secondary-side current excited by the second secondary winding.

[0038]

FIG. 1 is a circuit diagram showing a configuration of a power supply circuit according to an embodiment of the present invention. In this figure, the same parts as those in FIG. 7 are denoted by the same reference numerals, and the description is omitted. Although the details of the power supply circuit of the present embodiment shown in FIG.
A secondary winding N2 having a secondary side of IT as a first secondary winding.
And an insulation converter transformer PIT which is wound independently of the secondary winding N2 and is constituted by a tertiary winding N3 serving as a second secondary winding. When a power supply circuit is constituted by such an insulating converter transformer PIT of the present embodiment, the connection between the tertiary winding N3, the primary winding N1 and the secondary winding N2 is provided in the power supply circuit shown in FIG. The secondary winding N2A of the insulated converter transformer PIT and the primary winding N1 and the secondary winding N2 can be more tightly coupled.

Thus, in the present embodiment, the secondary current obtained based on the current excited in the tertiary winding N3 of the isolated converter transformer PIT is, for example, the load power of the second half-wave rectifier circuit 3. Even when the power is set to 10 W or more, without being affected by the -M operation mode (flyback converter operation) of the isolated converter transformer PIT,
It is excited by the M operation mode (forward converter operation). As a result, the secondary side rectified current I4 flowing through the rectifier diode DO2 constituting the second half-wave rectifier circuit 3 can be reduced in its peak current value as described later.

Here, the secondary winding N2 and the tertiary winding N3 wound around the split bobbin B of the insulating converter transformer PIT according to the present embodiment will be described with reference to FIGS. FIG. 2 is a schematic diagram for explaining how to wind a winding wound around the divided bobbin B of the insulating converter transformer PIT provided in the power supply circuit shown in FIG. The same parts as those in FIG.
The description is omitted. In this case, the secondary winding N2 is wound in the same manner as in FIG. 10 from the winding start position N2S, but the secondary winding N2 is formed independently of the tertiary winding N3. Therefore, the secondary winding N2 is wound by a predetermined number of turns from the winding start position N2S to the winding end position N2E.

A tertiary winding N3 formed independently of the secondary winding N2 is wound on the secondary winding N2 wound around the divided bobbin B. In this case, the tertiary winding N
3 is wound around the entire bobbin inner winding width K2 of the divided bobbin B from the winding start position N3S to the winding end position N3E shown in FIG. In other words, in the present embodiment, the tertiary winding N3 wound on the secondary winding N2 is not wound by the aligned winding wound in such a manner that the windings shown in FIG. And the tertiary winding N as shown in FIG.
The winding is performed by providing a uniform pitch interval CP between the three.

As described above, in the present embodiment, the secondary winding N
2 and a tertiary winding N3 formed independently of the secondary winding N2.
At the same time, a uniform pitch interval is provided so that the tertiary winding N3 is wound around the entire bobbin winding width K2 of the divided bobbin B. According to such a winding method, for example, in the insulated converter transformer PIT having the structure described with reference to FIGS. 10 and 11, the secondary winding N2A is loosely coupled to the primary winding N1 and the secondary winding N2. On the other hand, in the present embodiment, the tertiary winding N3 can obtain a tightly coupled state with the primary winding N1 and the secondary winding N2.

FIG. 4 is a diagram showing an example of an operation waveform of each part of the power supply circuit of the present embodiment provided with the insulating converter transformer PIT shown in FIG. 2, and mainly shows an output waveform on the secondary side. Have been. The output waveform shown in FIG.
The power supply circuit previously shown in FIG.
Assuming that the optimum drive conditions are obtained at 7 W, 100 V is input as the AC input voltage VAC, and 135 V is set as the constant-side secondary DC output voltage EO1 as in FIG. In order to obtain 15 V as the unconverted secondary-side DC output voltage EO2, the insulation converter transformer PIT has a secondary winding N2 of 33V.
It is assumed that a five-turn winding is provided as the tertiary winding N3.

In this case, the switching element Q1 performs a switching operation, and the collector of the switching element Q1 is connected to the collector of the switching element Q1 by the action of the parallel resonance circuit shown in FIG.
A primary parallel resonance voltage Vcp as shown in FIG. Then, by the on / off operation of the switching element Q1,
The switching output is transmitted to the secondary side of the insulated converter transformer PIT, and the connection end between the secondary winding N2 of the insulated converter transformer PIT and the anode of the rectifier diode DO1, as in the waveform shown in FIG. A secondary resonance current I2 having a waveform as shown in FIG. 4B is obtained, and a secondary resonance voltage V2 as shown in FIG. 4C is generated at both ends of the secondary winding N2.

In this case, the rectifier diode DO1 is connected to the rectifier diode DO1 shown in FIG.
Since the secondary resonance voltage V2 shown in (c) becomes conductive at a timing when it becomes higher than the level of the secondary DC output voltage EO1, the rectifier diode DO1 has a second waveform having a waveform as shown in FIG. The secondary side rectified current I3 flows. Also, the secondary-side resonance voltage V3 generated at both ends of the tertiary winding N3
Has a waveform as shown in FIG.
Is substantially similar to the secondary resonance voltage V3 generated at both ends of the secondary winding N2A, and is similar to the waveform of the secondary resonance voltage V2.

On the other hand, the secondary side rectified current I4 flowing through the rectifier diode DO2 of the second half-wave rectifier circuit 3 connected to the tertiary winding N3 of the insulating converter transformer PIT is as shown in FIG. FIG. 12 (f)
The output period is reduced from 6 .mu.S to 5 .mu.S, for example, and the peak level is reduced from 4.5 Ap to 2.5 Ap, for example, as compared with the secondary side rectified current I4 shown in FIG.

This is because, as described above with reference to FIG.
Is formed on the divided bobbin B, a tertiary winding N3 independent of the secondary winding N2 is formed, and the tertiary winding N3 is formed.
Are wound at equal pitch intervals so that the tertiary winding N3 is not biasedly wound on the secondary winding N2, so that the tertiary winding N3 is tightly coupled to the primary winding N1 and the secondary winding N2. 4F, the secondary side rectified current I4 shown in FIG. 4 (f) is obtained by the + M operation mode (forward converter operation) with the influence of the -M operation mode of the isolated converter transformer PIT suppressed. Becomes As a result, the current generated in the tertiary winding N3 by the flyback operation is suppressed, and the rectifier diode D2 of the secondary side rectified current I4 generated by the current by the flyback operation is reduced.
For example, the level of the peak current value in the latter half of the conduction angle of O2 can be reduced from 4.5 Ap to 2.5 Ap.

Therefore, if a power supply circuit is constituted by such an insulated converter transformer PIT, the peak current value of the secondary side rectified current I4 flowing through the rectifier diode DO2 can be suppressed, so that the power loss in the rectifier diode DO2 can be reduced. And the temperature rise due to the heat generated by the rectifier diode DO2 can be suppressed, and the reliability of the rectifier diode DO2 can be improved.

In this embodiment, the tertiary winding N3 is formed on the secondary side of the insulating converter transformer PIT independently of the secondary winding N2.
In order to form the secondary winding N2A, as in the case of the insulated converter transformer PIT shown in FIG. 7, the step of drawing out the secondary winding N2 as an intermediate tap CT and tying it to the pin terminal P is unnecessary. Therefore, there is an advantage that the manufacturing process is easy.

In the power supply circuit of the present embodiment, the second
For example, it is also possible to suppress the voltage fluctuation of the secondary-side DC output voltage EO2 output from the half-wave rectifier circuit 3 which is not constant voltage. For example, as shown in FIG. 5, when the power supply circuit is configured to support the maximum load power PoMAX = 217 W, when the load current IL2 output from the second half-wave rectifier circuit 3 changes from 0A to 1.0A, The voltage fluctuation level ΔEO2 of the secondary side DC output voltage EO2 is about 1.3V. Similarly, when the power supply circuit is configured to correspond to the maximum load power PoMAX = 82 W, when the load current IL2 output from the second half-wave rectifier circuit 3 changes from 0A to 1.0A, the secondary side The voltage fluctuation level ΔEO2 of the DC output voltage EO2 is about 2.2.

That is, as is apparent from comparison with FIG. 13, when the power supply circuit is configured to support the load power = 217 W, the secondary-side DC output voltage output from the second half-wave rectifier circuit 3 The voltage fluctuation level ΔE of EO2 is about 5.3V,
When the power supply circuit has a configuration corresponding to load power = 82 W, the voltage fluctuation level ΔE of the secondary side DC output voltage EO2 output from the second half-wave rectifier circuit 3 is about 4.2 V. Also in the circuit, the fluctuation of the secondary side DC output voltage level EO2 due to the fluctuation of the load current IL2 output from the second half-wave rectifier circuit 3 is suppressed, and it is possible to prevent the cross regulation from deteriorating.

As a result, even when a local regulator of, for example, 12 V is connected to the output of the second half-wave rectifier circuit 3, the secondary side DC output voltage EO2 accompanying the fluctuation of the load current IL2.
Therefore, power loss due to heat generated by the local regulator can be reduced.
This also makes it possible to reduce the size of the heat sink provided in the local regulator.

Also, as shown in FIG. 5, the maximum load power (217 W and 82 W) of the load connected to the first half-wave rectifier circuit 2 of the power supply circuit of the present embodiment is different. Since the generated voltage level difference is also reduced, the level fluctuation of the secondary side DC output voltage EO2 is caused by the fluctuation of the load connected to the constant side secondary DC output voltage EO1. It becomes possible to reduce.

FIG. 6 is a schematic diagram for explaining a method of winding an insulating converter transformer PIT of another embodiment. 2 and 10 are denoted by the same reference numerals,
The description is omitted. The insulation converter transformer PIT shown in FIG. 6A has a configuration in which the winding positions of the secondary winding N2 and the tertiary winding N3 of the insulation converter transformer PIT shown in FIG. 2 are interchanged. That is, as shown in the figure, the tertiary winding N3 is wound on the center side (lower side) of the divided bobbin B, and the secondary winding N2 is wound on the tertiary winding N3. Also in this case, the tertiary winding N3 is wound so that each pitch interval CP is equal.

When the power supply circuit is provided with the insulating converter transformer PIT having such a configuration, for example, the voltage level of the secondary side DC output voltage EO2 can be increased, and the secondary side DC output voltage EO2 can be increased. Even if the fluctuation of the load current IL2 supplied from the output voltage EO2 becomes large, it is possible to suppress the level fluctuation of the secondary DC output voltage EO2. For example, depending on the device, a 24V regulator for audio output may be connected to the secondary-side DC output voltage EO2, but this audio output regulator has a relatively wide load current IL2 of 0A to 2A. fluctuate. The insulation converter transformer PIT having the above-described configuration includes:
This is suitable for connecting a circuit having a large load current fluctuation, such as an audio output regulator.

The configuration of the secondary side of the insulation converter transformer PIT shown in FIG. 6B is similar to that of the insulation converter transformer PIT shown in FIG. 9 with respect to the secondary winding N2 of the insulation converter transformer PIT. An intermediate tap CT is provided. In this case, the secondary winding N2A is wound on the secondary winding N2 wound around the divided bobbin B at equal pitch intervals. That is, the secondary winding N
2A is wound without bias over the entire bobbin inner winding width K2 of the divided bobbin B.

The power supply circuit having the insulating converter transformer PIT having such a structure is suitable for a light load in which the load connected to the second half-wave rectifier circuit 3 is 10 W or less, for example. In this case, the operation waveform of the secondary-side rectified current I4 flowing through the rectifier diode DO2 of the second half-wave rectifier circuit 3 is the same as the operation waveform shown in FIG. 4C, and the voltage of the secondary-side DC output voltage EO2 The fluctuation value ΔEO2 can be reduced.

In the above-described embodiment, a case where a so-called single-ended voltage resonance type converter having one switching element on the primary side is provided as an example of the composite resonance type switching converter has been described. The present invention can be applied to a so-called push-pull system in which two switching elements are alternately switched.

In the above embodiment, a self-excited voltage resonance type converter is provided on the primary side.
According to the present invention, for example, instead of the self-excited oscillation driving circuit,
It is also possible to adopt a configuration in which an oscillation / drive circuit by C (integrated circuit) is provided, and the switching element of the voltage resonance type converter is driven by the oscillation / drive circuit. In addition, when adopting such a separately excited configuration,
The orthogonal control transformer PRT is omitted.

In the case of employing a separately excited configuration as described above, a single bipolar transistor (BJT)
Instead of the switching element Q, it is possible to employ a Darlington circuit in which two bipolar transistors (BJT) are connected in Darlington, for example.
Further, instead of the switching element Q as a single bipolar transistor (BJT), a MOS-FET (M
It is possible to use an OS type field effect transistor; metal oxide semiconductor, or IGBT (insulated gate bipolar transistor), or SIT (static induction thyristor), and switch any of these Darlington circuits or each of the above elements. When used as an element, it is possible to achieve higher efficiency. In the case where these elements are used as switching elements, although not shown here, the configuration of the drive circuit is changed so as to conform to the characteristics of elements to be actually used instead of the switching element Q. If a MOS-FET is used as a switching element, a configuration in which voltage driving is performed by a separately-excited system may be employed.

[0061]

As described above, in the insulating converter transformer of the present invention, the second secondary winding (tertiary winding) provided on the secondary side is replaced by the primary winding provided on the primary side of the insulating converter transformer. By winding the wire and the first secondary winding provided on the secondary side in a tightly coupled state, the second secondary winding is generated by the flyback converter operation of the insulating converter transformer. It is possible to suppress the secondary current that occurs. Therefore, if a switching power supply circuit having such an isolated converter transformer is configured,
Since the secondary current generated in the second secondary winding by the flyback converter operation can be suppressed, the second
It is possible to suppress the peak current value of the secondary current generated in the secondary winding. As a result, the power loss of the rectifier diode constituting the DC output voltage generator can be reduced, and the rectifier diode can be prevented from generating heat and its reliability can be improved.

Further, a second secondary winding provided on the secondary side of the insulating converter transformer is formed independently of the first secondary winding, and this second secondary winding is formed on a divided bobbin. If the first secondary winding to be wound is wound at an upper or lower position at an equal interval, the direct current output voltage generation means can output, for example, 1
Even when a heavy load of 0 W or more is connected, a stable secondary-side DC output voltage can be supplied.

A second secondary winding provided on the secondary side of the insulating converter transformer is provided with a first secondary winding and an intermediate tap, and the first secondary winding is wound around a divided bobbin. By winding the secondary winding at an equal interval above the secondary winding, it is possible to output a stable secondary DC output voltage to the DC output voltage generating means when, for example, a light load of 10 W or less is connected. Will be possible.

[Brief description of the drawings]

FIG. 1 is a circuit diagram illustrating a configuration example of a power supply circuit according to an embodiment of the present invention.

FIG. 2 is a schematic diagram for explaining a winding direction of an insulating converter transformer.

FIG. 3 is a diagram showing a state of a tertiary winding N3 wound around a divided bobbin B;

FIG. 4 is a waveform chart showing an operation of a main part of the power supply circuit of the present embodiment.

FIG. 5 is a diagram showing a relationship between a secondary-side DC output voltage output from a secondary side of the insulating converter transformer and a load current.

FIG. 6 is a schematic diagram for explaining a winding direction of an insulating converter transformer according to another embodiment.

FIG. 7 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 8 is a cross-sectional view showing the structure of the insulating converter transformer.

FIG. 9 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 10 is a schematic diagram for explaining a winding direction of an insulating converter transformer provided in the power supply circuit shown in FIG. 7;

FIG. 11 is a diagram showing a state of a secondary winding N2A wound around a divided bobbin B.

FIG. 12 is a waveform diagram showing an operation of a main part of a power supply circuit according to the prior art.

FIG. 13 is a diagram showing a relationship between a secondary-side DC output voltage output from a secondary side of an insulating converter transformer provided in a power supply circuit as a prior art and a load current.

[Explanation of symbols]

REFERENCE SIGNS LIST 1 control circuit, 2 first half-wave rectifier circuit, 3 second half-wave rectifier circuit, Ci smoothing capacitor, Q1 switching element, PIT isolation converter transformer, PRT orthogonal type control (drive) transformer, Cr primary side parallel resonant capacitor , C2 secondary side parallel resonance capacitor, N1 primary winding, N2 N2A secondary winding, N3 tertiary winding, NC control winding, NB drive winding, ND detection winding, CB resonance capacitor, DD clamp diode, DO1 DO2 Rectifier diode, CO1 CO2 Smoothing capacitor

Claims (6)

[Claims] 1. A rectifying / smoothing means for receiving a commercial AC power supply to generate a rectified / smoothed voltage and outputting the rectified / smoothed voltage as a DC input voltage, a primary winding on a primary side, and a secondary winding on a secondary side. At least a first secondary winding and a second secondary winding having a predetermined number of turns or less are wound, and the primary winding and the first secondary winding are loosely coupled. A gap is formed with respect to the core so as to obtain the coupling coefficient, and an insulating converter transformer is provided for transmitting the output of the primary side to the secondary side, and a switching element, and the DC input voltage is intermittently provided. Switching means configured to output to the primary winding of the insulated converter transformer, at least a leakage inductance component including the primary winding of the insulated converter transformer, and a capacity of the primary side resonance capacitor. A primary-side resonance circuit formed by the capacitance and the resonance means for making the operation of the switching means a resonance type; and at least one of a secondary winding of the insulating converter transformer.
By connecting a secondary resonance capacitor to one of the windings, a resonance circuit is formed by the leakage inductance component of the winding to which the secondary resonance capacitor is connected and the capacitance of the secondary resonance capacitor. A secondary-side resonance circuit, and a first secondary-side DC output voltage is obtained from the alternating voltage excited by the first secondary winding, and the first secondary-side DC output voltage is obtained from the alternating voltage excited by the second secondary winding. DC output voltage generating means configured to obtain a second secondary DC output voltage; and varying a switching frequency of the switching element based on a level of the first secondary DC output voltage. Constant voltage control means for performing constant voltage control of the first secondary-side DC output voltage, wherein the second secondary winding of the insulating converter transformer comprises:
A switching power supply circuit wound so as to obtain a state in which the primary winding and the first secondary winding are tightly coupled to each other. 2. The second secondary winding of the insulating converter transformer is wound independently of the first secondary winding, and then wound on a primary side and a secondary side. For the divided bobbin formed so that the region to be turned is divided,
2. The switching power supply circuit according to claim 1, wherein the secondary winding is wound at an upper or lower position of a winding portion with an equal winding interval. 3. 3. A second secondary winding of the insulating converter transformer, one end of which is connected to the first secondary winding via a tap. With respect to the divided bobbin formed so that the region where the winding is wound is divided between the first and second windings, the divided winding bobbin has a uniform winding interval at a position above the winding portion of the first secondary winding. The switching power supply circuit according to claim 1, wherein the switching power supply circuit is wound. 4. A primary winding wound on a primary side, a first secondary winding wound on a secondary side, a second secondary winding having a predetermined number of turns or less, a central magnetic leg. The primary winding and the first secondary winding are provided with an EE-type core capable of obtaining a required coupling coefficient for loose coupling. An insulating converter transformer, wherein the second secondary winding is wound so as to obtain a tightly coupled state with respect to the primary winding and the first secondary winding. 5. The second secondary winding of the insulating converter transformer is wound independently of the first secondary winding and then wound on a primary side and a secondary side. For the divided bobbin formed so that the region to be turned is divided,
5. The insulated converter transformer according to claim 4, wherein the secondary winding is wound at an upper or lower position of a winding portion of the secondary winding with an equal winding interval. 6. A second secondary winding of the insulating converter transformer, one end of which is connected to the first secondary winding via a tap. With respect to the divided bobbin formed so that the region where the winding is wound is divided between the first and second windings, the divided winding bobbin has a uniform winding interval at a position above the winding portion of the first secondary winding. 5. The insulation converter transformer according to claim 4, wherein the insulation converter transformer is wound.