FR2689704A1 - Zero voltage, zero current, resonant converter - Google Patents

Abstract

The resonant tank transformer assy is placed in series with either the output or the input of any multi ended transformer used in DC to DC conversion. It has one winding used for the resonant tank which is connected to a resonant capacitor and at least two additional windings, one for each side of the transformers windings if on the primary and at least two windings for each output if placed in the secondary with each of the two windings being placed in series with each respective secondary windings ground. The additional output windings being correctly ratio to achieve balanced reflected voltages. In series with a primary winding is a control switch; in series with the other primary winding is a control switch. The switches are operated alternately, by the control circuit and with a predetermined delay or dead time between the opening of one switch and closing of the other.

Description

 This invention relates to direct current to direct current converters and more particularly to self-oscillating direct current to direct current converters.

There are many converters which swap one DC voltage quantity over another. Conventional converters such as direct converters and indirect converters are well described in the prior art. Many texts like that of George Chryssis "High-Frequency Switching Power Supplies: Theory and
Design ", McGraw Hill Book Co, can explain how converters like these work.

 Conventional converters all use forced switching means to control the voltage and current during the regulation and power transfer process. This means of forced regulation switching causes two types of difficulty. The first is that of losses associated with forced switching. As there is a finite time associated with the closing and opening of a switch element, the current flowing through the switch and the voltage across the switch overlap during changes of state. switch causing a loss called switching loss. The faster the switching, the lower the switching loss. The second type of difficulty is the noise generated by the forced switching process. This noise is mainly caused by the rate of change of the voltage dV / dT on the high voltage switch. This change in voltage, coupled with the parasitic capacity of the primary - secondary windings of the transformer causes a common mode current to flow through this path.

To help control this noise on opening, a straight-line shock absorber is typically used. This way of doing things requires that the surface of the mounting board is sufficient and also leads to energy losses and expenditure of money. In addition, this approach provides assistance with opening only and the difficulty of the noise generated still remains at closing. Finally, the question of yield associated with the classical technique is the most serious difficulty. The current technique operates with a yield of approximately 73%, which is the cause of significant losses which must be absorbed by the power supply unit, which decreases the density of the power supply.

 Two types of the new generation of the classical technique, visible in FIGS. 1A and 1B, tackle the difficulties associated with the prior art described above. The first type is the resonant power supply in series with zero current switching. Such a power supply reduces the noise created by switching to a point very close to zero current (magnetizing current) and eliminates about half of the switching losses without the need for straight load formatting circuits. However, there is still the loss associated with the switching voltage when the switch closes. During the change of state of the switch, there may exist at the terminals of the latter a significant voltage which essentially charges the parasitic capacity of the switch and which finally discharges when the switch is closed. A typical field effect transistor with a capacity of the drain relative to the source of 120 pf, operating at 100khz, with a voltage of 700 volts at its terminals can have a parasitic loss of approximately 2.98 watts. . The rate of change of the current is as low as the current (dI / dT) is sinusoidal. The main problem with the type of converter visible in FIG. 1A lies in the relation of the effective current with respect to the line. The peak currents flowing in the resonance switches, transformers and output diodes are at their lowest value during operation at low line level when the duty cycle is maximum. The yield at this time can be in the low values of eighty. However, when the input voltage is high, the effective current flowing in the supply elements rises rapidly, causing the effective current to increase by as much as 1.7. This ultimately causes the efficiency to drop to the low values of the seventy at which the initial forced switching power supply operates. There are means to overcome this phenomenon, but these would require the addition of a new converter in front of the self-oscillating circuits for the stabilization of the input voltage and, thus, the optimization of the efficiency. The additional cost involved and the additional surface area required on the assembly panel make this approach acceptable only in high power areas. The operation of this type of converter is described, for example, in US Patent No. 4,415,959 in the name of Vinciarelli.

A more recent solution to this problem of effective currents is visible in FIG. 1B and described, for example, in Japanese patent No. 1503925 in the name of
Matsushita. In this type of self-oscillating converter, the self-oscillating circuit is a combination of resonant circuits both in series and in parallel. With this type of approach, the offset of the frequency of the power supplied to the regulation area is greatly reduced.

In the Vinciarelli type converter, the frequency offset can be greater than 10: 1 at all conditions and even more when operation without load is necessary. In the serial / parallel converter, the frequency offset is a function of the ratio between the inductance in parallel and the inductance in series of the plug circuit. You can take advantage of practical frequency shifts of only 2: 1. As the energy to be transferred to the output load is a function not only of the voltage in the resonant circuit (1 / 2CV2F) but also a function of the phase relationship between the two resonant circuits, the effective current flowing in the switches , the transformer and the diodes change little with the corresponding changes in the finish line. This indeed stabilizes the yield in relation to the characteristics of the supply line. In the series-parallel converters the losses of the resonant circuit tend to rise with those of the line but the effective currents in the switch, the transformer and the magnetic parts tend to remain the same. One of the main drawbacks of the Matsushita solution is the inability to adjust the operating voltage of the plug circuit.

In all resonant converters the Q factor of the oscillating circuit is of the greatest importance for the general efficiency of the power converter. A primary means of controlling losses at any given output power in the resonant type design is to draw a low operating current from the plug circuit. In the converters of the prior art (Vinciarelli: American patent no. 4,415,959 Japanese patent no. 1,503,925) the voltage of the plug circuit in operation is not adjustable independently of the ratios of the operating voltages of the converter. This forces the designer to adjust other equally important parameters such as the leakage inductance, the value of the resonance capacity and the density of the operating flow. The result of this is greatly reduced design efficiency due to a lower Q factor as well as a larger capacity of the plug circuit and a potentially more difficult magnetic element from the manufacturing point of view. In addition, as the ratio of the magnetization inductance of the plug circuit to the voltage of the latter cannot be manipulated in conventional converters, the minimum frequency offset is difficult to optimize. In the converter of the prior art according to Archer patent no. 4,774,649, there is a new self-oscillating converter which is built on an integrated magnetic element. Thanks to this approach, a certain control of the operating voltage of the plug circuit is possible; however, this control is obtained at the expense of other variables and the integrated magnetic transformer tends to have a low Q factor for this reason.

 Finally, the remaining problem associated with both the Vinciarelli and Matsushita converters is the inability to operate at both zero current and zero voltage simultaneously. As previously indicated, a controlled rate of change of voltage (dV / dT) is desirable to achieve quiet operation from the standpoint of protection against interference and noise. To further complicate this requirement, it is desirable to achieve this without the use of external components since their number and size have a direct influence on the size of the finished feed.

 In the converter to be described, a primary switching takes place at a current very close to zero at the same time as a substantially sinusoidal shape is preserved during the switching intervals.

 The converter described below achieves zero voltage operation of the primary switch, thus having a controlled rate of voltage changes across the transformer primary without the addition of external parts and facilitating the elimination of parasitic losses from switching associated with the interruption of the high frequency voltage.

 The converter provides the ability to transform the operating voltage of the plug circuit into any desired voltage by using a transformer and plug circuit assembly.

 Thus, the converter has a high operating efficiency, a high safety and a reduced cost and noise. The converter in service is placed in series with either the output or the input of any transformer with several terminals used in the conversion of direct current into direct current. It has a winding used for the plug circuit which is connected to a resonance capacitor and at least two additional windings, one for each side of the transformer windings if it is the primary and at least two windings for each output in the case of the secondary with each of the two windings connected in series with each respective secondary winding grounded. The additional output windings are in the correct ratio to provide balanced reflected voltages.

 In a secondary embodiment of the stopper circuit with reference, the following would be typical of the embodiment.

The primary winding of a transformer with symmetrical windings would have opposite phases connected to a DC supply voltage, the corresponding secondary windings would have their outputs correctly in phase connected to a corresponding rectifier and their earth terminals would be connected to a winding. respective of the resonant transformer assembly with each respective winding of the resonant assembly out of phase with the others. The corresponding outputs of the resonant assembly would be joined together and connected to the ground of the smoothing capacitor of the output with the other side of the smoothing capacitor connected to the output of the rectifiers.

The flow of current through the primary is established in pulses with the opening time of the two switches held constant and the closing time controlled by means of an error amplifier responding to changes in the output load. The transformer leakage field is granted to resonant transformers by a resonant capacitor with the leakage inductance being scaled by means of the corresponding secondary ratio of the transformers to the winding of the plug circuit. This resonant frequency is fixed at about twice the frequency of the parallel plug circuit which is regulated by the magnetization inductance of the plug circuit winding. The currents flowing in the primary and secondary windings are therefore forced to take a sinusoidal shape in response to the common sinusoidal flow flowing in the whole of the plug circuit transformer.

 The converter optimizes the Q factor of the plug circuit by selecting the material of the core, the density of the operating flow and the frequency.

The use of the capacity of the plug circuit through the adjustment of the ratio of the turns between the load windings and of the plug circuit minimizes the shift of the frequency from minimum to maximum for regulation by selecting the permeability of the core and the ratio of the winding of the plug circuit to the load winding.

This is made possible by the use of a transformer for the parallel resonant plug circuit.

 According to the present invention there is provided a resonant power supply converter for changing the magnitude of a DC voltage, in a power operating range, comprising: a main transformer having a core, a primary winding to produce alternating flows flowing in opposite directions through the core, a control switch for producing alternating pulses of current flow through the primary winding for the purpose of obtaining this alternating flow flow, this switch control and this primary winding having a predetermined parasitic capacitance with a predetermined delay time between the end of a given pulse and the start of the next pulse, this transformer comprising a pair of secondary windings arranged to be alternately conductive in phase with alternating flow circulation, being inherent in this transformer main a magnetizing inductor such that the magnetizing current is large enough to charge in a controlled manner this parasitic capacitance, the secondary circuit comprising said secondary windings and having internally rectifying means for rectifying the currents in these secondary windings, an output circuit having output terminals and an output capacitor, a circuit for applying the rectified current from the rectifier to the output terminals as a load current, a plug circuit comprising a plug circuit transformer having a load current winding disposed in the converter to respond to the load current, a plug circuit winding magnetically connected to said load current winding and a plug circuit capacitor mounted in parallel with this plug circuit winding.

Direct current to direct current converters according to the invention will now be described by way of example with reference to the appended schematic drawings in which
FIGS. 1A and 1B show, respectively, two kinds of circuits of the prior art,
FIG. 2 is an exemplary embodiment of the present invention with the plug circuit located in the secondary of the main transformer,
FIGS. 3 and 4 show waveforms existing in the circuit of FIG. 2 at relatively heavy and relatively light load conditions, respectively,
FIG. 5 is an alternative embodiment of the invention with the plug circuit located in the primary of the main transformer,
- Figure 6 is a control circuit for the operation of switches 28, 30, 28 ', 30' of Figures 2 and 5 for the creation of waveforms shown in Figures 3 and 4. Details of the operation of the The circuit of Figure 6 are exposed in a Motorola publication entitled Semi-conductor Technical Data Prototype Information, High Performance Resonant Mode Controller, PC 34067.

 FIG. 2 shows a main transformer 10 having primary windings 12 and 14, a core 16 and secondary windings 18 and 20. The windings 12 and 14 have equal numbers of turns, are wound in opposition as indicated by conventional points, so that the primary currents (Ip1 and Ip2) produce opposite flows in the core 16. The windings 12 and 14 are produced in such a way that there is a minimum parasitic capacitance and a minimum leakage inductance between the windings.

 Electric power is supplied from a continuous source 22 connected in bridge with capacitors 24 and 26 which serve to eliminate the ripples and which also provide a stable source of energy for the converter.

The windings 12 and 14 are mounted in parallel at the terminals at the input voltage 22. In series with the winding 12 there is a control switch 28, in series with the winding 14 there is a control switch 30 The switches 28 and 30 are operated alternately by the control circuit visible in FIG. 6 and with a predetermined delay or dead time between the opening of one switch and the closing of the other. Switches 28 and 30 are type switches
MOSFET, metal, oxide, semiconductor, having respectively intrinsic diodes 32 and 34 which restore the excursion of the frequency when the opposite switch opens.

 The delay time is achieved, among other things, by the choice of a predetermined parasitic capacitance in the control switches 28, 30, 32, 34. The main transformer 10 is designed so that the magnetization current is large enough to charge the in a controlled manner, the parasitic capacity of the primary windings 12, 14. For example, the magnetization inductance can be established by the existence of a suitable air gap in the core 16.

 The secondary windings 18 and 20 are magnetically connected by the core 16 to the primary windings 12 and 14 and these secondary windings are wound to be in phase with the windings 12 and 14 as indicated by the dots. The inductors 36 and 38 are not discrete circuit elements but they represent the leakage field of the windings 12, 14, 18, 20 reflected in the secondary.

 The output of the secondary is rectified by any suitable means represented here for example by a diode 40 in series with the winding 18 and a diode 42 in series with the winding 20. The output of the diodes 40, 42 is connected to the terminal outlet 44, as seen at 46.

 Interposed in the secondary between the windings 18, 20 and the output terminals 44, 48 is a plug circuit 50 comprising a resonant plug transformer 51 having a core 52 and associated windings 54, 56, 58.

The winding 54 is in series with the secondary winding 18; the winding 56 is in series with the winding 20. The two outputs are joined together with the terminal 48. The capacitor 60 between the output terminals 44/48 smoothes the output signal. Resistor 62 represents the load at the output of the converter. In this way the windings 54 and 56 are sensitive to the load current of the converter.

The third winding 58 constitutes a plug winding at the terminals of which is a plug circuit capacitor 64.

 The resonant plug transformer 51 has two basic functions. First, it controls the shape of the current through the main transformer 10 through the common flow of the winding 68 of the resonant plug circuit and associated windings through which all the current flowing in the main transformer 10 must pass. Second, it allows the designer to minimize losses in the 58/64 elements of the resonant plug circuit as well as controlling frequency offset through the ability to adjust the operating voltage of the plug circuit using the ratio of the turns of the transformer as well as fixing the ratio of the magnetization inductance of the plug circuit to the reflected leakage inductance of the main transformer 10.

 Figure 5 is similar to Figure 2 with the plug circuit 50 moved from the secondary of the main transformer to the primary as it is visible at 50 '. As with circuit 50, the plug circuit 50 'is located to be sensitive to the load current through 62. The correspondence between the constituent parts of FIG. 5 and of FIG. 2 are indicated by the symbol' applied to the numerical references on Figure 5.

 The winding 54 'which corresponds to the winding 54 of Figure 2 is in series with the primary winding 12', the winding 56 'is in series with the winding 14'.

Like the core 52 of Figure 2, the core 52 'of Figure 5 is common to the three windings 54', 56 ', 58'.

 The waveforms of Figures 3 and 4 refer to the circuit of Figure 5 substantially without change. In the primary windings of Figure 5 there is a weak magnetization current flowing through the windings 54 ', 56', which is the cause of negligible distortion.

 The operation of the apparatus of FIG. 2 is as follows. Switch 28 is closed for the start of a cycle. I1 is assumed that the converter is in the permanent operating state (that is to say not at start-up), so the instant of closing of this switch corresponds to an appropriate phase relationship of the resonant plug transformer 51. The current rises in the winding 12 and in the corresponding switch 28 in a sinusoidal manner due to the influence of the flux of the plug transformer and of the circuit created by the leakage inductance of the output circuit 36 and of the reflected magnetization inductance of the resonant plug circuit. From circuit 50, a plug circuit resonating in series is thus formed until the voltage of the latter is sufficiently reversed and the output diode 40 opens. At this moment the primary switch 28 continues to remain closed but the current follows a downward ramp and cuts the magnetization current of the transformer 10. When the control circuit of FIG. 6 opens the switches 28/30, the two switches remain open while the voltage at the terminals of the switch 28 and of the switch 30 reverses, until their respective diodes 32/34 become conductive. At this time the respective voltages are at their zero point and the switch 30 acting alternately can be closed. The secondary winding 20 acting alternately now conducts through its resonant path in series 38/51 and the currents of the previous cycle are repeated in the opposite direction. This time report can be kept until the load changes. When the load is increased, the control circuit of Figure 6 decreases the closing time of the two switches, which effectively increases the operating frequency of the converter. However, the delay or dead time between the switches remains the same since the voltage recovery time changes only slightly due to the magnitude of the opening current. The opening current is always the magnetization current because the load current decreases to the value of the magnetization current during each cycle if it is assumed that the power supply is operating within its limits dump. The remaining magnetization current, flowing through the parasitic capacitor of the switches and windings, drives the voltage to the corresponding point of recovery in a period determined by the value of the capacity and the magnitude of the current. This drive current changes with frequency since the closing time changes with frequency but this change is not sufficient to produce any significant change in the dV / dT ratio. The actual change in a practical realization would be about 2: 1.

In this way, the converter operates in a zero voltage mode, which avoids the noise and loss problems associated with the closing and opening of the switches with a voltage across their terminals. This loss can be estimated approximately when the voltage at the terminals of the switch is known as well as the importance of the parasitic capacitance as well as the operating frequency. In a typical high frequency converter controlled by a field effect transistor operating on opening at a high voltage (300 v), this loss would be approximately 2 watts, which would require a heat sink and the increase in l importance of the power supply as well as an increase in the value of the input filter. As the control circuit raises the frequency in response to an increased demand for power, the duration of series resonance increases and the duration of the current flowing through the corresponding reflected leakage field and the secondary diode increases. The opposite is true when the demand for power decreases and the corresponding operating frequency decreases.

Figure 3 shows the relationship of the converter waveforms to a medium to relatively high state of charge. Waves A and B are corresponding commands from the control circuit of FIG. 6 to the respective switches 28 and 30. The waveforms C and D are the corresponding currents I1 and Ip2 flowing in the switches during their conduction cycle. As can be seen from these waveforms, the initial current (load current) at 100 is sinusoidal in nature ending at 102 at the magnetization current 104 of the primary of the transformer (linear part). E and
F show the corresponding voltages Vp1 and Vp2 at the terminals of switches 30 and 28 respectively, showing the interval 106 of interruption at zero voltage (forms A and B). The waveform G shows the voltage V t of the resonant plug circuit 58/64 and the current It flowing in the resonance capacitor 64, for the two parts in series and in parallel with the operation of the converter. The time t1 is the time during which the equivalent circuit is a plug circuit resonant in series with the reflected leakage inductance 36 in series with the reflected resonance capacity fixing the operating frequency of the plug circuit.

As this path passes through the diode 40, the reflected current flows through the diode 40 to the load 62.

The time t2 is the time during which the output diodes 40 and 42 are open and that there is no reflected leakage field in series with the plug circuit 50. The operating frequency at this point is the frequency of the circuit plug in parallel comprising the winding 58 of the plug magnetization circuit of the inductor in parallel with the resonance capacitor 64. The times t3 and t4 correspond to the identical waveforms seen in a mirror following the action of the opposite switch 30. Figure 4 shows the functional waveforms of the converter at a lower state of charge. The relationships between times are the same as in Figure 3.

Claims (8)

RESELL ICAT IONS  1. Oscillating power converter for changing the magnitude of a DC voltage, in a power operating range, comprising  a main transformer (10) having a core (16) and primary windings (12, 14) for producing alternating flows flowing in opposite directions through the core (16),  control switches (28, 30) for creating alternating pulses of current flowing through the primary windings (12, 14) to generate said alternating flows,  these control switches (28, 30) and said primary windings (12, 14) having a predetermined parasitic capacitance with a predetermined delay time between the completion of a given pulse and the triggering of the next pulse,  the transformer (10) comprising a pair of secondary windings (18, 20) arranged to conduct alternately in phase with the current of the alternating flows,  being inherent in the main transformer (10) a magnetization inductor such that the magnetization current is large enough to charge said parasitic capacity in a controlled manner,  secondary circuits comprising the secondary windings (18, 20) and having internally rectifying means (40, 42) for rectifying the currents in the secondary windings,  an output circuit having output terminals (44, 48) and an output capacitor (60),  a circuit for applying the rectified current from the rectifiers to said output terminals (44, 48) in the form of a load current,  a plug circuit (50) comprising a plug transformer (51) having a load current winding arranged in the converter to be sensitive to the load current,  a plug circuit winding (58) magnetically linked to the load current windings (54, 56),  a plug circuit capacitor (64) in parallel with the plug circuit winding (58).  2. Converter according to claim 1 wherein the control switches (28, 30) are sensitive to the magnitude of a load applied to the output terminals (44, 48) to control the triggering and the duration of each current pulse.  3. Converter according to claim 1 wherein the magnitude of the magnetization inductance is established by the existence of an air gap in the core (16).  4. Converter according to claim 3 characterized in that the control switches (28, 30) are sensitive to the magnitude of a load applied to the output terminals (44, 48) to control the triggering and the duration of each pulse of current.  5. Converter according to any one of claims 1 to 4 characterized in that the primary winding comprises a pair of primary windings (12, 14) connected in parallel, the control switches comprise a pair of switches (28, 30 ) in series respectively with the primary windings.  6. Converter according to any one of the preceding claims, characterized in that the plug circuit (50 ') is arranged in series with the primary windings (12', 14 ').  7. Oscillating power converter for changing the magnitude of a DC voltage, in a power operating range, including  a main transformer (10) having a core (16), primary windings (12, 14) to create alternating flows flowing in opposite directions through the core (16),  control switches (28, 30) for creating alternating pulses of current flowing through the primary windings (12, 14) for creating said alternating flows,  these control switches (28, 30) and the primary windings (12, 14) having a predetermined parasitic capacitance with a predetermined delay time between the completion of a given pulse and the triggering of the next pulse,  this transformer (10) having a pair of secondary windings (18, 20) arranged to conduct alternately in phase with said alternating flows,  being inherent in this main transformer (10) a magnetization inductor the magnitude of which is established by the existence of an air gap in the core (16) such that the magnetization current is large enough to charge said parasitic capacity in a controlled manner,  a secondary circuit comprising secondary windings (18, 20) and having internally a pair of diodes (40, 42) for respectively rectifying the current in the secondary windings (18, 20),  a plug circuit (50) comprising a transformer (51) having a first winding (54) in series with one of the diodes (40) and a secondary winding (56) in series with the other diode (42), and a plug circuit winding (58),  a plug circuit capacitor (64) in parallel with the plug circuit winding (58),  an output circuit having a pair of output terminals (44, 48) and an output capacitor (60) between these terminals, and  a circuit for applying to the output terminals the rectified current coming from the diodes.  8. Converter according to claim 7 characterized in that the control switches (28, 30) are sensitive to the magnitude of a load applied to the output terminals (44, 48) to control the triggering and the duration of each pulse of current.