CN117240089B

CN117240089B - Control circuit and control method of bidirectional four-transistor Buck-Boost converter

Info

Publication number: CN117240089B
Application number: CN202311171635.3A
Authority: CN
Inventors: 肖凌萱; 阮新波; 刘飞
Original assignee: Nanjing University of Aeronautics and Astronautics
Current assignee: Nanjing University of Aeronautics and Astronautics
Priority date: 2023-09-12
Filing date: 2023-09-12
Publication date: 2025-02-14
Anticipated expiration: 2043-09-12
Also published as: CN117240089A

Abstract

The control circuit and the control method of the bidirectional four-tube Buck-Boost converter are characterized in that the converter consists of four switching tubes and a filter inductor, two ends of the filter inductor are respectively connected with the middle point of a bridge arm of a switching tube Q ₁、Q₂ and the middle point of a bridge arm of a Q ₃、Q₄, Q ₁ is conducted complementarily with Q ₂, Q ₃ is conducted complementarily with Q ₄, and the converter can realize bidirectional flow of energy. In order to improve the working efficiency of the converter, the four switching tubes realize ZVS and simultaneously minimize the effective current value, and a quadrilateral current control mode is adopted for the four-tube Buck-Boost converter. The invention provides a control circuit and a control method for realizing stable operation and smooth switching of the converter in a constant voltage mode and a constant current mode and realizing natural commutation and smooth transition of the inductance current, so that the inductance current can realize smooth commutation of the current in a single switching period, and meanwhile, the realization condition that a switching tube does not lose a soft switch after a commutation signal is triggered can be ensured, and the running efficiency and the reliability of the converter are improved.

Description

Control circuit and control method of bidirectional four-tube Buck-Boost converter

Technical Field

The invention belongs to the technical field of power converters, and particularly relates to a control circuit and a control method of a bidirectional four-tube Buck-Boost converter.

Background

In order to alleviate energy crisis and environmental pollution and realize sustainable development of human society, renewable energy sources represented by solar energy and wind energy are gradually replacing traditional fossil energy sources. However, the power generated by renewable energy power generation systems may fluctuate significantly, subject to weather conditions. To solve this problem, energy storage devices such as a storage battery and a supercapacitor can be introduced. The bidirectional DC-DC converter is used for controlling the power flow between the energy storage device and the direct current bus, so that the electric energy can smoothly flow to a load end and the stability of the bus voltage is ensured. The bidirectional DC-DC converter can be divided into two main types of isolation type and non-isolation type according to different application requirements. Because the terminal voltage of the energy storage device has a certain variation range in the charge and discharge process, if the bus voltage is within the variation range, the bidirectional converter can have higher conversion efficiency in the variation range of the voltage of the energy storage device. To meet this design requirement, a bi-directional converter with buck-boost function is selected. The topological structure of the four-tube Buck-Boost converter (Four-Switch Buck-Boost, FSBB) has the advantages of positive output voltage, low voltage stress of a switching tube and the like, so that the four-tube Buck-Boost converter is widely applied.

The working modes of the bidirectional converter are various working modes including constant-current charging, constant-voltage charging, constant-current power supply and bus constant-voltage power supply according to the change of renewable energy power generation and load requirements. The renewable energy power generation system is greatly influenced by weather environment, and in order to maintain the stability of bus voltage, the bidirectional converter needs to frequently switch working modes according to external conditions. Therefore, the natural commutation and smooth transition of the inductor current in the bidirectional FSBB converter are beneficial to improving the stability and reliability of the converter in operation. Finding a simple and feasible control method is a problem to be solved by researchers in the field.

Disclosure of Invention

The invention fully considers the problem that the bidirectional converter is required to perform frequent reversing work because the renewable energy power generation system is greatly influenced by weather environment, and provides a control circuit and a control method of the bidirectional four-pipe Buck-Boost converter. For the bidirectional converter applied to the energy storage system, the natural commutation, smooth transition and mode switching of the inductance current in the bidirectional FSBB converter are effectively realized, the stability of the bus voltage can be maintained, and meanwhile, the stability and the reliability of the converter in operation are improved, so that the converter has better dynamic performance.

In order to achieve the above purpose, the present invention adopts the following technical scheme:

The control circuit of the bidirectional four-tube Buck-Boost converter comprises four switching tubes Q ₁、Q₂、Q₃、Q₄ and a filter inductor L _c, wherein two ends of the filter inductor L _c are respectively connected with the midpoint of a bridge arm of a switching tube Q ₁、Q₂ and the midpoint of a bridge arm of a switching tube Q ₃、Q₄, the switching tube Q ₁ is complementarily conducted with the Q ₂, the switching tube Q ₃ is complementarily conducted with the Q ₄, one end of the converter is a bus voltage V _bus, and the other end of the converter is a battery voltage V _bat, and the control circuit is characterized by comprising:

The driving signal generating module A is used for generating driving signals of the switching tubes Q ₁ and Q ₂ when the converter operates in the forward direction and generating driving signals of the switching tubes Q ₃ and Q ₄ when the converter operates in the reverse direction;

The driving signal generating module B is used for generating driving signals of the switching tubes Q ₃ and Q ₄ when the converter operates in the forward direction and generating driving signals of the switching tubes Q ₁ and Q ₂ when the converter operates in the reverse direction;

The phase shift duty cycle circuit generating module is used for generating a phase difference between the on time of the switching tube Q ₁ and the on time of the switching tube Q ₃ when the converter runs in the forward direction and runs in the reverse direction;

the natural reversing module is used for generating reversing signals for controlling the converter to switch the running state;

The current flow direction smooth transition module is used for realizing the smooth transition of the inductance current i _Lc of the filter inductance L _c after receiving the reversing signal of the natural reversing module.

In order to optimize the technical scheme, the specific measures adopted further comprise:

further, the driving signal generating module a includes a current regulator, a voltage regulator, a comparator 1 and an RS flip-flop 1, wherein the outputs of the current regulator and the voltage regulator are respectively connected through diodes D ₁ and D ₂, and the final output v _r is the smaller value of the two outputs;

When the converter operates in the forward direction, the input of the current regulator is a current sampling value flowing into a battery, the input of the voltage regulator is a battery voltage sampling value, the voltage reference of the voltage regulator is the maximum battery voltage, when the battery voltage V _bat is less than the maximum battery voltage, the voltage regulator keeps positive saturated output, the current regulator works, the battery charges in a constant current mode, the diode D ₁ is conducted, the diode D ₂ is cut off, when the battery voltage V _bat reaches the maximum battery voltage, the voltage regulator works, the battery voltage V _bat is kept unchanged, the current regulator is saturated output, the diode D ₂ is conducted, the diode D ₁ is cut off, the final output V _r and the sawtooth wave V _saw are sent to the RS trigger 1, and the output of the RS trigger 1 and the clock signal CLK1 are sent to the RS trigger 1, so that driving signals of Q ₁ and Q ₂ are obtained;

When the converter runs reversely, the input of the current regulator is a current sampling value flowing into a bus, the input of the voltage regulator is a bus voltage sampling value, the voltage reference of the voltage regulator is rated voltage, when the bus voltage V _bus is smaller than the rated voltage, the voltage regulator keeps positive saturated output, the current regulator works, the battery discharges, the diode D ₁ is conducted, the diode D ₂ is cut off, when the bus voltage V _bus is the same as the rated voltage, the voltage regulator works, the bus voltage V _bus is kept unchanged, the current regulator is saturated output, the diode D ₂ is conducted, the diode D ₁ is cut off, the final output V _r and the sawtooth wave V _saw are sent to the RS trigger 1, and the output of the RS trigger 1 and the clock signal CLK1 are sent to the RS trigger 1, so that driving signals of Q ₃ and Q ₄ are obtained.

Further, the driving signal generating module B includes a comparator 2 and an RS flip-flop 2, the positive input end of the comparator 2 is an inductance current sampling value, the negative input end is-I _ZVS,-I_ZVS, which is a negative current reference required for ensuring that the switching tube Q ₁、Q₄ realizes the soft switching, and I _ZVS is a negative current reference required for ensuring that the switching tube Q ₂、Q₃ realizes the soft switching, and after the output v _comp of the comparator 2 and the clock signal CLK1 perform an or operation, the output v _comp and the clock signal CLK2 generated by the phase-shifting duty cycle circuit generating module are sent to the RS flip-flop 2;

When the inductor current I _Lc drops to-I _ZVS, the output v _comp of the comparator 2 becomes high potential, so that the RS trigger 2 is reset, thereby turning off Q ₃ and turning on Q ₄;

When the converter is operated in reverse, the clock signal CLK2 is used to turn on the switching transistor Q ₁, and when the inductor current I _Lc rises to I _ZVs, the output v _comp of the comparator 2 becomes high, resetting the RS flip-flop 2, thereby turning off Q ₁ and turning on Q ₂.

Further, the phase shift duty ratio circuit generating module generates a phase difference between the on time of the switching tube Q ₁ and the on time of the switching tube Q ₃ according to the phase shift duty ratio between the two bridge arms:

during forward operation of the converter, the approximation D _{θ_appr}(V_bat,i_bat) of the phase difference between the Q ₁ and Q ₃ on times is:

D_{θ_appr}(V_bat,i_bat)＝a₁V_bat+b₁i_bat+V_c1

When the inverter is operating in reverse, the approximation D' _θappr(V_bat,i_bus of the phase difference between the on times of Q ₃ and Q ₁) is:

D′_{θ_appr}(V_bat,i_bus)＝a₂V_bat+b₂i_bus+V_c2

Wherein a ₁、b₁、a₂、b₂ represents a linear coefficient, and V _c1 and V _c2 are constant terms;

according to the phase difference between the on timings of the switching transistors Q ₁ and Q ₃, the clock signal CLK2 is generated and input to the driving signal generating module B.

Further, the natural commutation module includes a comparator 4, a comparator 5 and an RS trigger 3, where a forward input end of the comparator 4 is a lowest reference value V _{bus_L}, a reverse input end is a bus voltage sampling value V _{bus_s}, a forward input end of the comparator 5 is a bus voltage sampling value V _{bus_s}, and a reverse input end is a highest reference value V _{bus_H};

When the bus voltage V _bus drops below V _{bus_L}, the output of the comparator 4 is high, after the output of the comparator 4 and the clock signal CLK1 are subjected to OR operation, the generated signal V _flip1 is input into the RS trigger 3, when the rising edge of the clock signal CLK1 arrives, V _flip1 is set high, the RS trigger 3 is reset, the q ₁ signal output by the RS trigger 3 is set low, the q ₂ signal is set high, the converter works reversely, and the generated reversing signal q ₂ inputs current to the smooth transition module;

When the bus voltage V _bus rises to be higher than V _{bus_H}, the output of the comparator 5 is high, after the output of the comparator 5 and the clock signal CLK1 are subjected to OR operation, the generated signal V _flip2 is input into the RS flip-flop 3, when the rising edge of the clock signal CLK1 comes, V _flip2 is set high, the RS flip-flop 3 is reset, the q ₁ signal output by the RS flip-flop 3 is set high, the q ₂ signal is set low, the converter works forward, and the generated commutation signal q ₂ inputs current to the smooth transition module.

Further, the current flow direction smooth transition module enables a reversing signal q ₂ generated by the natural reversing module to generate a narrow pulse signal on both the rising edge and the falling edge of the jump, a q _{2_d} signal is generated after the two signals are subjected to OR operation and used as a driving signal to enable a triode arranged at a reference position of the current regulator to be conducted, the reference value of the current regulator is pulled down to 0 after the triode is conducted, the triode is cut off after the narrow pulse signal disappears, the reference voltage of the current regulator is gradually increased from 0, the reference voltage of the current regulator is set to 0 at the moment of reversing under the action of the q _{2_d} signal, and then the reference voltage of the current regulator is gradually increased, so that the inductance current i _Lc is reversely and gradually increased from zero.

The invention also provides a control method of the bidirectional four-tube Buck-Boost converter based on the control circuit, which is characterized by comprising the following steps:

When the converter works in the forward direction, the battery voltage V _bat and the inflow battery current i _bat are sampled, and the phase shift angle D _θ of the phase difference between the switching tube Q ₁、Q₃ and the switching time is approximately calculated;

When the converter works reversely, the battery voltage V _bat and the inflow bus current i _bus are sampled, and the phase shift angle D' _θ of the phase difference between the switching tube Q ₃、Q₁ and the switching time is approximately calculated;

When the converter works in the forward direction, the inductance current I _Lc is sampled and compared with a negative current reference I _zvs required by ensuring that the switching tube Q ₁、Q₄ realizes soft switching, and when I _Lc linearly drops to-I _zvs, the switching tube Q ₃ is turned off to obtain the duty ratio 1-D _y2 of the switching tube Q ₃;

When the converter works reversely, negative inductance current-I _Lc is sampled and compared with negative current reference I _zvs required by ensuring that the switching tube Q ₃、Q₂ realizes soft switching, and when I _Lc rises linearly to I _zvs, the switching tube Q ₁ is turned off to obtain the duty ratio 1-D' _y2 of the switching tube Q ₁.

Further, a lowest reference value and a highest reference value of the bus voltage V _bus are set, when the bus voltage V _bus is lower than the lowest reference value, the converter is operated reversely, the battery supplies power to the bus end to enable the bus voltage V _bus to rise, when the power generated by the power generation system is greater than the load demand, the bus voltage V _bus rises, the energy storage device stores redundant energy, when the bus voltage V _bus is higher than the highest reference value, the converter is operated positively, the battery stores redundant energy, and the bus voltage V _bus falls to the rated value.

The control circuit and the control method have the beneficial effects that the control circuit and the control method for realizing natural commutation, smooth transition and mode switching of the bidirectional four-tube Buck-Boost converter are provided, and the current flow direction and the working mode of the converter are determined by detecting the voltage of the bus, so that the stability of the bus voltage is maintained. Meanwhile, by means of the current loop reference soft start method, smooth transition during current commutation is achieved, large current overshoot is avoided, and accordingly reliability of the converter is improved.

Drawings

Fig. 1 is a circuit configuration diagram of a four-tube Buck-Boost converter.

Fig. 2 is a current waveform diagram of the load change and current switching, taking V _bus＜V_bat as an example.

Fig. 3a is a circuit diagram of the drive signals of Q ₁ and Q ₂ in forward operation and the drive signal generation modules of Q ₃ and Q ₄ in reverse operation of the inverter.

Fig. 3b is a circuit diagram of the drive signals of Q ₃ and Q ₄ in forward operation and the drive signal generation modules of Q ₁ and Q ₂ in reverse operation of the inverter.

Fig. 3c is a circuit diagram of the phase-shift duty cycle circuit generating module.

Fig. 3d is a circuit diagram of the natural commutation module.

Fig. 3e is a circuit diagram of a current flow to a smooth transition module.

Fig. 4a is a simulated waveform diagram of the forward operation of the converter with a battery voltage of 75V.

Fig. 4b is a simulated waveform diagram of the converter operating in the forward direction and battery voltage of 100V.

Fig. 4c is a simulated waveform diagram of the converter operating in the forward direction and the battery voltage at 125V.

Fig. 5a is a simulated waveform diagram of the inverter operating in reverse and battery voltage of 75V.

Fig. 5b is a simulated waveform diagram of the inverter operating in reverse and battery voltage of 100V.

Fig. 5c is a simulated waveform diagram of the inverter operating in reverse and the battery voltage at 125V.

Fig. 6 is a dynamic waveform diagram of a bi-directional converter switching from constant voltage charging to maximum current constant current discharging of a battery.

Fig. 7 is a dynamic waveform diagram of a bi-directional converter switching from maximum power constant current charging to maximum current constant current discharging of a battery.

Detailed Description

The invention will now be described in further detail with reference to the accompanying drawings.

Fig. 1 is a circuit structure diagram of a four-tube Buck-Boost converter, mainly comprising four switching tubes and a filter inductor, wherein the switching tubes Q ₁ and Q ₂ are complementarily conducted to form a bridge arm unit, the switching tubes Q ₃ and Q ₄ are complementarily conducted to form a bridge arm unit, and the filter inductor L _c is respectively connected with the midpoints of the two bridge arms for storing and transmitting energy. One end of the converter is a bus voltage V _bus, and the other end is a battery voltage V _bat. The current flowing into the bus-terminal is defined as i _bus, and the current flowing into the battery terminal is defined as i _bat.

In order to increase the operating efficiency of the converter, it is desirable to minimize the inductor current effective value while achieving all switching tubes ZVS (zero voltage switching). Fig. 2 illustrates the operating waveforms of the FSBB converter during load changes and current commutation, using V _bus＜V_bat as an example. When the converter is in forward operation, the on time of Q ₁ is defined as the starting time of each switching cycle, and when the inductance current is linearly increased to-I _ZVS, Q ₃ is immediately turned off, so that Q ₄ is ensured to realize ZVS and meanwhile current pulsation is reduced as much as possible. The commutation signal is triggered at the time t=4t _s, T _s is a switching period, and the driving signals of the two bridge arms are exchanged, so that the starting time of each switching period is Q ₃ on time, and the phase shift duty ratio is the phase difference between the Q ₃ and Q ₁ on time. Immediately after the commutation signal is triggered, Q ₁ should be turned off when the inductor current rises linearly to I _ZVS to ensure that Q ₂ achieves ZVS while minimizing current ripple.

FSBB the converter has three control amounts in forward operation, namely duty cycle D _y2 of duty cycle D _y1、Q₄ of Q ₁, And a phase shift duty ratio D _θ corresponding to the phase difference between the turn-on moments of Q ₁ and Q ₃, wherein when the converter is operated reversely, the driving signals of the two bridge arms are exchanged, and the driving signals are controlled according to three control amounts, namely, the duty ratio of Q ₃ is D '_y1、Q₂, the duty ratio is D' _y2, And the phase shift duty cycle corresponding to the phase difference between the on times of Q ₃ and Q ₁ is D' _θ. The duty ratio is used for controlling the switching action, and the phase shift duty ratio is used for controlling the phase difference of the switching-on time. The soft switching of the switching tubes Q ₁ and Q ₄ is realized, the inductor current needs to be ensured to be over negative before the switching tube is turned on, the junction capacitance of the switching tube is discharged to zero to enable the junction capacitance to be reversely conducted by the diode to be naturally conducted, and the magnitude of negative current needed to be realized is defined as-I _ZVS. The soft switching of the switching tubes Q ₂ and Q ₃ is realized, the inductor current is required to be ensured to be positive before the switching tube is turned on, the junction capacitance of the switching tube is discharged to zero to enable the junction capacitance to be inverted, the diode is naturally conducted, and the size of positive current required to be realized is defined as I _ZVS.

In one embodiment, the invention provides a control circuit of a bidirectional four-pipe Buck-Boost converter, which is exemplified by an analog control circuit, and schematic diagrams of the control circuit are shown in fig. 3a to 3e, and the control circuit mainly comprises five components, namely a driving signal generating module of Q ₁ and Q ₂ in forward operation of the converter, a driving signal generating module of Q ₃ and Q ₄ in reverse operation of the converter, a driving signal generating module of Q ₃ and Q ₄ in forward operation of the converter, a driving signal generating module of Q ₁ and Q ₂ in reverse operation of the converter, a phase-shifting duty ratio circuit generating module, a natural commutation module and a current flow smooth transition module.

The partial parameters in the figure are I _{bat_s} as a current sampling value flowing into the storage battery, I _{bat_s} as a current sampling value flowing into a bus terminal, I _os as an output value (determined according to current flow direction) of I _{bat_s} and I _{bat_s} after selecting a switch through an S1, I _{o_ref} as a reference value of a current regulator, q _{2_d} as a reference soft start of the current regulator when a commutation signal arrives due to the q ₂ signal passing through a current flow smoothing transition module, V _{bat_s} as a battery voltage sampling value, V _{bus_s} as a bus voltage sampling value, V _os as an output value (determined according to current flow direction) of V _{bat_s} and V _{bus_s} after selecting a switch through an S2, V _{o_ref} as a reference value of the voltage regulator, C _ref as a capacitance value for soft start of the current regulator, and I _{Lc_s} as an inductance current sampling value.

1. Drive signals of Q ₁ and Q ₂ in forward operation of the inverter and drive signal generation modules of Q ₃ and Q ₄ in reverse operation (drive signal generation module A)

As shown in fig. 3a, when a battery is selected as the energy storage device, the battery voltage fluctuates with different respective states of charge of the battery. When the converter works in the forward direction, the storage battery is charged, and the charging process is that constant current charging is carried out until the maximum voltage of the battery is reached, and then constant voltage charging is carried out. The outputs of the current regulator and the voltage regulator are connected by diodes D ₁ and D ₂, respectively, the final output v _r being the smaller of the two. The voltage reference of the voltage regulator is the voltage sampling value corresponding to the maximum battery voltage, so when the battery voltage is not the maximum value, the voltage regulator keeps positive saturated output, the current regulator works, the storage battery is charged with constant current, and at the moment, the diode D ₁ is turned on, and the diode D ₂ is turned off. When the voltage of the battery reaches the maximum value, the voltage regulator works to maintain the voltage at two ends of the storage battery unchanged, at the moment, the charging current i _bat is smaller than the set value, the current regulator outputs positive saturation, the diode D ₂ is turned on, and the diode D ₁ is turned off. The final output v _r and the sawtooth v _saw are fed to comparator 1, the output of which is fed to RS flip-flop 1, resulting in the drive signals of Q ₁ and Q ₂. Note that v _saw is synchronized with the clock signal CLK 1.

When the bus voltage is the same as the rated voltage, the converter works in a constant voltage mode, the voltage regulator keeps positive saturated output, the current regulator works, the storage battery discharges, the diode D ₁ is conducted, the diode D ₂ is cut off, the voltage regulator works in a constant voltage mode, the voltage at two ends of the bus is kept unchanged, the charging current i _bat is smaller than a set value, the current regulator keeps positive saturated output, the diode D ₂ is conducted, and the diode D ₁ is cut off. The operation of the bi-directional converter is different depending on the external environmental conditions.

2. Drive signals of Q ₃ and Q ₄ in forward operation of the inverter and drive signal generation modules of Q ₁ and Q ₂ in reverse operation (drive signal generation module B)

As shown in fig. 3b, when the converter is operating in the forward direction, the clock signal CLK2 (the generating circuit thereof see the phase-shifted duty cycle circuit generating module) is used to turn on Q ₃. When the inductor current I _Lc drops to-I _ZVS, the output v _comp of the comparator 2 goes high, resetting the RS flip-flop 2, turning off Q ₃ and turning on Q ₄. In order to prevent I _Lc from falling to-I _ZVS before Q ₂ turns off, CLK1 and v _comp are intentionally added to perform or to forcibly turn off Q ₃, so as to ensure that the switching timing of the converter is unchanged.

When the inverter is operating in reverse, -I _Lc is compared to-I _ZVS and CLK2 is used to turn on Q ₁. When the inductor current I _Lc rises to I _ZVS, the output v _comp of the comparator 2 goes high, resetting the RS flip-flop 2, turning off Q ₁ and turning on Q ₂. In order to prevent I _Lc from rising to I _ZVS before Q ₄ turns off, CLK1 and v _comp are intentionally added to perform or to forcibly turn off Q ₁, so as to ensure that the switching timing of the converter is unchanged.

3. Phase shift duty cycle circuit generating module

After the duty ratio of each switching tube is determined, the phase shift duty ratio between the two bridge arms is also determined. The theoretical calculation of the phase shift duty cycle D _θ, i.e. the phase difference between the on moments of Q ₁ and Q ₃, is in fact a function of the battery voltage V _bat and the battery charging current i _ba when the bi-directional converter is operating in the forward direction, and the theoretical calculation of the phase shift duty cycle D' _θ, i.e. the phase difference between the on moments of Q ₃ and Q ₁, is in fact a function of the battery voltage V _bat and the charging current i _bus flowing into the bus bar when the bi-directional converter is operating in the reverse direction. The curved surface of the phase shift duty ratio is approximated by a plane, thereby realizing the aim of simplifying control. The simplified post-phase shift duty cycle is a linear combination of voltage and current sample values. When the converter works in the forward direction, the phase difference approximate value between the turn-on moments of the Q ₁ and the Q ₃ is as follows:

D_{θ_appr}(V_bat,i_bat)＝a₁V_bat+b₁i_bat+V_c1

When the inverter is operating in reverse, the phase difference approximation between the on times of Q ₃ and Q ₁ can be expressed as:

D′_{θ_appr}(V_bat,i_bus)＝a₂V_bat+b₂i_bus+V_c2

Where a ₁、b₁、a₂、b₂ denotes a linear coefficient, and V _c1 and V _c2 are constant terms. D _{θ_appr}(V_bat,i_bat) and D' _{θ_appr}(V_bat,i_bus) are obtained by a proportional-plus-add circuit when the converter operates in forward and reverse directions, respectively, from the module shown in fig. 3 c.

4. Natural reversing module

In order to maintain the stability of the bus voltage, when the electric energy generated by the renewable energy power generation system is greater than the load demand, the bus voltage starts to rise, and the energy storage equipment starts to store redundant energy at the moment. In fig. 3d, when the output q ₁ of the RS flip-flop 3 is high, the switch S ₁～S₈ is turned on with the up signal, the inverter operates in the forward direction, and when q ₁ is low, the inverter operates in the reverse direction. The positive input end of the comparator 4 is the lowest reference value V _{bus_L}, the negative input end is the busbar voltage sampling value, the positive input end of the comparator 5 is the busbar voltage sampling value, and the negative input end is the highest reference value V _{bus_H}. When the bus voltage V _bus drops below V _{bus_L}, the comparator 4 outputs a high potential. The starting time of the next switching period is waited, namely, when the rising edge of the CLK1 signal arrives, V _flip1 is set high, the trigger is reset, the q ₁ signal is set low, the q ₂ signal is set high, the converter works reversely, and when the bus voltage V _bus rises to be higher than V _{bus_H}, the comparator 5 outputs high potential. When the starting time of the next switching period arrives, v _flip2 is set high, the trigger is set, the q ₁ signal is set high, the q ₂ signal is set low, and the converter works in the forward direction.

5. Current flow direction smooth transition module

If smooth commutation of the bi-directional converter can be achieved, the reliability of the converter during commutation can be improved. From the above analysis, q ₂ is set high when the inverter is switched from forward to reverse operation, and q ₂ is set low when the inverter is switched from reverse to forward operation. Thus, the q ₂ signal is toggled whenever the commutation condition is triggered. The invention makes q ₂ signal generate narrow pulse signal on rising edge and falling edge of jump through logic circuit shown in fig. 3e, q _{2_d} is signal after OR operation of both, and is used as drive to make triode at reference position of current regulator conduct. After the transistor is turned on, the capacitor C _ref discharges and the reference value is pulled down to 0. After the narrow pulse signal disappears, the transistor turns off, the reference V _{i_ref} charges the capacitor C _ref again, and the reference voltage gradually rises from 0. Under the action of q _{2_d}, the reference voltage of the current regulator is set to 0 at the moment of commutation, and then gradually rises, so that the inductance current reversely starts to gradually rise from zero, thereby realizing smooth transition of current and ensuring that the switching tube does not lose soft switching before and after the commutation signal is triggered.

In another embodiment, the invention provides a control method of a bidirectional four-pipe Buck-Boost converter, which is realized based on the control circuit. In order to improve the working efficiency of the converter, the four switching tubes realize ZVS and simultaneously minimize the effective current value, and the control method adopts a quadrilateral current control mode for the four-tube Buck-Boost converter.

According to the difference of current flow direction and working mode, the battery voltage V _bat, the bus voltage V _bus, the inflow battery current i _bat and the inflow bus current i _bus are sampled and compared with different reference values respectively in different working modes, and the duty ratio D _y1 of the Q ₁ in forward working or the duty ratio D' _y1 of the Q3 in reverse working are regulated and controlled through a current regulator and a voltage regulator, so that current stability and voltage stability are realized. Meanwhile, the two regulators are connected through the diode to realize the switching from the constant current mode to the constant voltage mode.

When the converter operates in the forward direction, the phase shift angle D _θ of the phase difference between the battery voltage V _bat and the inflow battery current i _bat is approximately calculated, and when the converter operates in the reverse direction, the phase shift angle D' _θ of the phase difference between the battery voltage V _bat and the inflow bus current i _bus is approximately calculated.

The converter is operated in the forward direction to sample the inductance current I _Lc and compare with the negative current reference I _zvs required by ensuring the switching tube Q ₁、Q₄ to realize soft switching, the switching tube Q ₃ is turned off when the I _Lc is linearly reduced to-I _zvs to obtain the duty ratio 1-D _y2 of the switching tube Q ₃, the converter is operated in the reverse direction to sample the negative inductance current-I _Lc and compare with the negative current reference I _zvs required by ensuring the switching tube Q ₃、Q₂ to realize soft switching, and the switching tube Q ₁ is turned off when the I _Lc is linearly increased to I _zvs to obtain the duty ratio 1-D' _y2 of the switching tube Q ₁.

In order to realize natural commutation, smooth transition and mode switching of the inductive current, specific measures are as follows:

And natural commutation, namely, when the electric energy generated by the power generation system is insufficient to meet the load demand, the bus voltage drops, and the energy storage device releases energy to supply energy to the load. In order to realize the natural commutation of the inductor current, a lower limit value and an upper limit value of the bus voltage are required to be set. When the voltage of the bus is lower than the lower limit value, the signal is enabled to be high, the converter works reversely, the storage battery supplies energy to the bus end to enable the voltage of the bus to rise, and when the electric energy generated by the power generation system is greater than the load demand, the voltage of the bus rises, and at the moment, the energy storage equipment stores redundant energy. When the bus voltage is higher than the upper limit value, the enable signal is low, the converter works forward, the storage battery stores redundant energy, and the bus voltage drops to the rated value. Thereby realizing the natural commutation of the inductive current.

Smooth transition, namely, if the smooth reversing of the bidirectional converter can be realized, the reliability of the converter during reversing can be improved. Since there is a moment in the commutation process when the current is 0, the regulator will be saturated in output without any measures, which will cause a reverse current overshoot of the inductor current. Therefore, the invention adds a triode at the reference position of the current regulator, and uses the commutation enabling signal to generate a pulse signal to conduct the triode at the moment of triggering the commutation condition, so that the reference value and the output of the regulator are reduced to 0. After a short period of time is maintained, the reference voltage is gradually increased, and the current is increased from zero at the commutation moment, so that the smooth transition of the inductance current is realized, and the switching tube is ensured not to lose the soft switch before and after the commutation signal is triggered.

Mode switching, namely, the output of the current regulator and the output of the voltage regulator are respectively connected through a diode, and the final output is the smaller value of the current regulator and the voltage regulator. The voltage reference of the voltage regulator is a voltage sampling value corresponding to rated voltage. When the voltage is less than the rated value, the voltage regulator keeps positive saturated output, the current regulator works, and the converter works in a constant current mode. When the voltage reaches a set value, the voltage regulator works, the converter works in a constant voltage mode, and the current regulator outputs positive saturation. The mode switching of the bidirectional converter is adopted to adapt to different external environment requirements.

To further illustrate the advantages of the control method proposed by the present invention, a simulation example of the present invention is given below.

The simulation circuit was built with Saber simulation software according to the 1.25kW four-tube Buck-Boost converter parameters given in table 1. Fig. 4a-4c show simulation waveforms of the four-tube Buck-Boost converter during forward operation under rated power, wherein fig. 4a is a simulation waveform diagram of the battery voltage under 75V full load, fig. 4b is a simulation waveform diagram of the battery voltage under 100V full load, and fig. 4c is a simulation waveform diagram of the battery voltage under 125V full load, and it can be seen that all switching tubes can realize ZVS and the inductor current pulsation is small. Fig. 5a to 5c show simulation waveforms of the four-tube Buck-Boost converter when the four-tube Buck-Boost converter works reversely under rated power, wherein fig. 5a is a simulation waveform diagram of the battery voltage under 75V full load, fig. 5b is a simulation waveform diagram of the battery voltage under 100V full load, and fig. 5c is a simulation waveform diagram of the battery voltage under 125V full load. Fig. 6 is a dynamic waveform of the bi-directional converter from constant voltage charging of the battery to maximum current constant current discharging. Fig. 7 is a dynamic waveform of the bi-directional converter from maximum power constant current charging to maximum current constant current discharging of the storage battery. It can be seen that on the basis of adopting the control mode, the natural commutation of the converter is realized, the bidirectional converter meets the requirements of different working modes in the commutation process, and meanwhile, the inductor current realizes smooth transition, so that the bidirectional converter has a faster dynamic response speed.

Table 1 major parameters of bidirectional converter

Parameters (parameters)	Numerical value	Parameters (parameters)	Numerical value
				Bus voltage V _bus	100V	Switching frequency f _s	1MHz
Storage battery voltage V _bat	75~125V	Inductance L _c	0.8μH
				Maximum charge/discharge current	10A	Maximum output power	1.25kW

The above is only a preferred embodiment of the present invention, and the protection scope of the present invention is not limited to the above examples, and all technical solutions belonging to the concept of the present invention belong to the protection scope of the present invention. It should be noted that modifications and adaptations to the invention without departing from the principles thereof are intended to be within the scope of the invention as set forth in the following claims.

Claims

1. The control circuit of the bidirectional four-tube Buck-Boost converter comprises four switching tubes Q ₁、Q₂、Q₃、Q₄ and a filter inductor L _c, wherein two ends of the filter inductor L _c are respectively connected with the midpoint of a bridge arm of a switching tube Q ₁、Q₂ and the midpoint of a bridge arm of a switching tube Q ₃、Q₄, the switching tube Q ₁ is complementarily conducted with the Q ₂, the switching tube Q ₃ is complementarily conducted with the Q ₄, one end of the converter is a bus voltage V _bus, and the other end of the converter is a battery voltage V _bat, and the control circuit is characterized by comprising:

The current flow direction smooth transition module is used for realizing the smooth transition of the inductance current i _Lc of the filter inductance L _c after receiving the reversing signal of the natural reversing module;

The driving signal generating module A comprises a current regulator, a voltage regulator, a comparator 1 and an RS trigger 1, wherein the outputs of the current regulator and the voltage regulator are respectively connected through a diode D ₁ and a diode D ₂, and the final output v _r is the smaller value of the two outputs;

2. The control circuit of the bidirectional four-tube Buck-Boost converter of claim 1, wherein the driving signal generating module B comprises a comparator 2 and an RS trigger 2, wherein the positive input end of the comparator 2 is an inductance current sampling value, the reverse input end is a negative current reference required by-I _ZVS,-I_ZVS for ensuring the soft switching of the switching tube Q ₁、Q₄, and I _ZVS is a negative current reference required by ensuring the soft switching of the switching tube Q ₂、Q₃, and after the output v _comp of the comparator 2 and the clock signal CLK1 are subjected to OR operation, the clock signal CLK2 generated by the phase-shifting duty cycle circuit generating module is fed into the RS trigger 2;

3. The control circuit of the bidirectional four-tube Buck-Boost converter of claim 1, wherein the phase shift duty cycle circuit generating module generates a phase difference between on times of the switching tubes Q ₁ and Q ₃ according to a phase shift duty cycle between two bridge arms:

D_{θ_appr}(V_bat,i_bat)＝a₁V_bat+b₁i_bat+V_c1

when the inverter is operating in reverse, the approximation D' _{θ_appr}(V_bat,i_bus of the phase difference between the on times of Q ₃ and Q ₁) is:

D'_{θ_appr}(V_bat,i_bus)＝a₂V_bat+b₂i_bus+V_c2

4. The control circuit of the bidirectional four-tube Buck-Boost converter of claim 1, wherein the natural commutation module comprises a comparator 4, a comparator 5 and an RS trigger 3, the forward input end of the comparator 4 is the lowest reference value V _{bus_L}, the reverse input end is a bus voltage sampling value V _{bus_s}, the forward input end of the comparator 5 is a bus voltage sampling value V _{bus_s}, and the reverse input end is the highest reference value V _{bus_H};

5. The control circuit of the bidirectional four-transistor Buck-Boost converter of claim 1, wherein the current flow direction smoothing transition module enables a reversing signal q ₂ generated by the natural reversing module to generate a narrow pulse signal at the rising edge and the falling edge of a jump, the narrow pulse signal q ₂ and the narrow pulse signal are subjected to OR operation to generate a q _{2_d} signal, a triode arranged at a reference position of the current regulator is conducted as a driving signal, the reference value of the current regulator is pulled down to 0 after the triode is conducted, the triode is cut off after the narrow pulse signal disappears, the reference voltage of the current regulator is gradually increased from 0, and under the action of the q _{2_d} signal, the reference voltage of the current regulator is set to 0 at the reversing moment and then gradually increased, so that an inductance current i _Lc reversely and gradually increases from zero.

6. A control method of a bi-directional four-pipe Buck-Boost converter based on a control circuit according to any one of claims 1-5, characterized in that the control method comprises:

7. The method of controlling a bi-directional four-pipe Buck-Boost converter of claim 6, wherein a minimum reference value and a maximum reference value of a bus voltage V _bus are set, when the bus voltage V _bus is lower than the minimum reference value, the converter is operated in a reverse direction, the battery supplies power to the bus terminal to raise the bus voltage V _bus, when the power generated by the power generation system is greater than a load demand, the bus voltage V _bus is raised, the energy storage device stores excessive energy, when the bus voltage V _bus is higher than the maximum reference value, the converter is operated in a forward direction, the battery stores excessive energy, and the bus voltage V _bus is lowered to a rated value.