CN112510976B - Active clamp flyback converter, controller and control method thereof - Google Patents
Active clamp flyback converter, controller and control method thereof Download PDFInfo
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- CN112510976B CN112510976B CN202011524409.5A CN202011524409A CN112510976B CN 112510976 B CN112510976 B CN 112510976B CN 202011524409 A CN202011524409 A CN 202011524409A CN 112510976 B CN112510976 B CN 112510976B
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- 238000004804 winding Methods 0.000 claims description 16
- 238000003079 width control Methods 0.000 claims description 9
- 238000002955 isolation Methods 0.000 claims description 5
- 238000011084 recovery Methods 0.000 abstract description 7
- 238000005265 energy consumption Methods 0.000 abstract description 2
- 238000010586 diagram Methods 0.000 description 20
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
- H02M1/083—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the ignition at the zero crossing of the voltage or the current
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of DC power input into DC power output
- H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
- H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
- H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
- H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33507—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
- H02M3/33523—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters with galvanic isolation between input and output of both the power stage and the feedback loop
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Abstract
The invention provides a controller of an active clamp flyback converter and a control method thereof, in the flyback converter, the controller realizes the switching between a double-pulse non-complementary mode and a leading edge non-complementary mode or the switching between the double-pulse non-complementary mode and the flyback mode after detecting the comparison of a feedback voltage and a set mode switching threshold voltage, the performance of a power supply system is better, and the requirement of European Union six-level energy consumption is met; compared with the back-edge non-complementary mode, the double-pulse non-complementary mode has the advantages that after the main switch tube is turned off, the resonance of the leakage inductance and the clamping capacitor is not realized through the body diode of the clamping switch tube any more, but is realized through the clamping switch tube, and therefore the problem of reverse recovery of the diode of the clamping switch tube body is solved. The conduction time of the first pulse signal of the double-pulse non-complementary is controllable, the conduction time is 1/4 of the resonance period of the clamping capacitor and the leakage inductor, the reverse recovery problem is avoided, the leakage inductor is reversely charged after the resonant current passes zero, and the EMI problem of the converter is solved.
Description
Technical Field
The invention relates to the field of switching power supply design, in particular to an active clamp flyback converter, a controller and a control method of the active clamp flyback converter.
Background
The flyback converter is widely applied to medium and small power switching power supplies due to the advantages of low cost, simple topology and the like. In the actual working process, the energy of the primary side of the flyback converter cannot be completely transmitted to the secondary side due to the existence of the leakage inductance, and the resonance between the leakage inductance energy of the primary side and the MOS tube junction capacitor causes the drain electrode of the main switching tube to generate a high-frequency voltage peak. In the product design process, in order to reduce the voltage stress of the switching tube, it is a common practice to add a suitable snubber circuit, and the common snubber circuit includes an RCD snubber circuit, an LCD snubber circuit, and an active clamp circuit. The active clamping circuit is additionally provided with an additional clamping switch tube and a larger clamping capacitor, so that leakage inductance energy can be stored in the clamping capacitor, and the energy is recycled to the input end of the converter. In addition, due to the electric inertia of the leakage inductance, the active clamping circuit extracts the charges on a termination capacitor at the drain end of the main switching tube through reverse exciting current after the recovery process of the leakage inductance energy is finished, so that the drain voltage of the main switching tube is reduced to zero, zero voltage switching-on (ZVS) of the main switching tube is realized, the switching-on loss of the main switching tube is reduced, and the power density of a product is further improved.
Referring to fig. 1, fig. 1 is a circuit diagram of a typical active-clamp flyback converter, and an active-clamp flyback converter 100 includes: leakage inductance LK, excitation inductance LM, clamping capacitor C _ C, main switch tube S1, clamping switch tube S2, excitation inductance LM current sampling resistor RCS, transformer primary winding NP, transformer secondary winding NS, rectifier diode DR, converter output capacitor COUTA controller 120 (i.e., the main control chip for the converter), and an isolated feedback circuit 130. The controller realizes the work of the active clamping flyback converter by sampling the output voltage of the converterAnd performing mode control.
However, at present, for the control of the working mode of the active clamp flyback converter, it cannot be ensured that the active clamp flyback converter operates at high efficiency within the full-voltage full-load range, and simultaneously, light-load efficiency and no-load power consumption are considered.
Therefore, how to ensure that the active clamp flyback converter operates at high efficiency within a full-voltage full-load range, and simultaneously give consideration to light-load efficiency and no-load power consumption, and how to control the first driving signal conduction time of the clamp switching tube when the active clamp flyback converter works in a double-pulse non-complementary mode is a technical problem to be solved urgently in the industry.
Disclosure of Invention
In view of the defects in the prior art, the technical problem to be solved by the present invention is to provide an active clamp flyback converter, a controller and a control method thereof, so as to solve the problems of low efficiency, large no-load power consumption and control of the on-time of the first driving signal of the clamp switching tube in the existing control mode under light load.
In order to solve the above technical problem, the present invention provides a control method for an active clamp flyback converter, including: by detecting the comparison between the feedback voltage and the set threshold voltage, when the feedback voltage is greater than the threshold voltage Vth1, operating the active clamping flyback converter to work in a double-pulse non-complementary mode; when the feedback voltage is less than the threshold voltage Vth2, the active clamp flyback converter is operated in a front edge non-complementary mode or a flyback mode, and when the feedback voltage is between the threshold voltage Vth2 and the threshold voltage Vth1, the operation mode of the current period is kept consistent with the operation mode of the previous period, wherein the threshold voltage Vth1 is greater than the threshold voltage Vth 2.
In one embodiment, when the active clamp flyback converter operates in the double-pulse non-complementary mode, the second turn-on time of the clamp switching tube is when the drain-source resonant voltage of the main switching tube is at a peak time, and simultaneously, the drain-source resonant voltage of the clamp switching tube is at a trough time.
In an embodiment, the second turn-on time of the clamping switch tube may be any peak of the drain-source resonant voltage of the main switch tube, that is, the second turn-on time of the clamping switch tube may be the first peak of the drain-source resonant voltage of the main switch tube, or may be any one of the second to nth peaks, where N is a positive integer.
In one embodiment, the switching between the double-pulse non-complementary mode and the leading-edge non-complementary mode has a transition process, in the transition process, the conduction time of the first driving signal transmitted to the clamping switch tube is kept unchanged, and the conduction time of the second driving signal transmitted to the clamping switch tube is gradually reduced in the transition process until the last driving signal is not carried out; the switching between the leading edge non-complementary mode and the double-pulse non-complementary mode has another transition process, in the transition process, the conduction time of the first conduction driving signal transmitted to the clamping switch tube is unchanged, and the conduction time of the second conduction driving signal transmitted to the clamping switch tube is gradually increased from a certain minimum value to the main switch tube to just realize zero-voltage switching-on.
The present invention also provides an active clamp flyback converter, comprising: a transformer, a primary circuit, a secondary circuit, an isolation feedback circuit and a controller,
the primary winding of the transformer is connected with the primary circuit, and the secondary winding of the transformer is connected with the secondary circuit;
the primary side circuit is provided with a leakage inductor, a clamping capacitor, a main switching tube and a clamping switching tube, wherein the drain electrode of the clamping switch is connected with one end of the leakage inductor through the clamping capacitor, and the source electrode of the clamping switch is connected with the drain electrode of the main switching tube; the other end of the leakage inductance is connected with the primary winding;
the isolation feedback circuit is connected with the output end of the secondary side circuit and used for detecting the feedback voltage of the output end;
the controller is connected with the main switching tube and the clamping switching tube respectively and is used for controlling the working mode of the active clamping flyback converter according to the feedback voltage, and specifically:
when the feedback voltage is greater than the threshold voltage Vth1, controlling the active clamp flyback converter to work in a double-pulse non-complementary mode; when the feedback voltage is less than the threshold voltage Vth2, the active clamp flyback converter is controlled to work in a front edge non-complementary mode or a flyback mode, and when the feedback voltage is between the threshold voltage Vth2 and the threshold voltage Vth1, the working mode of the current period is kept consistent with the working mode of the previous period, wherein the threshold voltage Vth1 is greater than the threshold voltage Vth 2.
In one embodiment, the active clamp flyback converter is further provided with a resistor connected in parallel with the clamp capacitor, and the resistor is used for consuming energy on the clamp capacitor when the active clamp flyback converter works in a flyback mode.
The present invention also provides a controller for an active clamp flyback converter, comprising:
the mode judging module is used for operating the converter to work in a double-pulse non-complementary mode or a leading edge non-complementary mode according to the load;
the frequency control module is used for controlling the working frequency of the converter according to the load size;
the pulse width control module is used for controlling the conduction time of a main switching tube of the converter according to the load;
the time sequence control module is respectively connected with the mode judgment module, the frequency control module and the pulse width control module and is used for generating a driving signal according to control signals output by the mode judgment module, the frequency control module and the pulse width control module;
and the driving module is used for converting the driving signal and inputting the converted driving signal to the main switching tube and the clamping switching tube.
In one embodiment, the load magnitude is achieved by sensing a feedback voltage or by sensing a load current magnitude.
The invention also provides a control method of the active clamping flyback converter, the active clamping flyback converter is provided with a primary side circuit, a secondary side circuit and a transformer which is connected with the primary side circuit and the secondary side circuit, the primary side circuit is provided with a main switch tube, a clamping capacitor and a leakage inductance, and the control method is characterized by comprising the following steps:
detecting the size of the load;
when the size of the load is larger than a first threshold value, driving signals transmitted to the main switching tube and the clamping switching tube are controlled, so that the active clamping flyback converter works in a double-pulse non-complementary mode, and when the active clamping flyback converter works in the double-pulse non-complementary mode, the clamping switching tube receives the driving signals twice in each working period, wherein the conducting time of the first driving signal received by the clamping switching tube is 1/4 of the resonance period of the clamping capacitor and the leakage inductor;
when the size of the load is smaller than a second threshold value, driving signals transmitted to the main switching tube and the clamping switching tube are controlled, so that the active clamping flyback converter works in a front edge non-complementary mode or a flyback mode;
when the size of the load is between a second threshold and a first threshold, keeping the working mode of the current period consistent with the working mode of the last period, wherein the first threshold is larger than the second threshold.
In one embodiment, when the size of the load is smaller than the third threshold or zero, the active clamp flyback converter is controlled to operate in the burst mode, and when the active clamp flyback converter operates in the burst mode, the active clamp flyback converter is controlled to operate for a period of time and then rest for a period of time.
The invention also provides a control method of the active clamping flyback converter, the active clamping flyback converter is provided with a primary side circuit, a secondary side circuit and a transformer for connecting the primary side circuit and the secondary side circuit, the primary side circuit is provided with a main switching tube, a clamping capacitor and a leakage inductance, and the control method comprises the following steps:
detecting the size of the load;
controlling driving signals transmitted to a main switching tube and a clamping switching tube according to the load size, so that the active clamping flyback converter is switched between a double-pulse non-complementary mode and a leading-edge non-complementary mode;
when the active clamp flyback converter is switched from a double-pulse non-complementary mode to a leading-edge non-complementary mode, a first driving signal transmitted to a clamp switch tube is controlled within a certain set time, so that the conduction time of the first driving signal is kept unchanged, and meanwhile, the conduction time of a second driving signal transmitted to the clamp switch tube is adjusted, so that the conduction time of the second driving signal is gradually reduced until the second driving signal is reduced to zero;
when the active clamp flyback converter is switched from a leading edge non-complementary mode to a double-pulse non-complementary mode, a first driving signal transmitted to the clamp switch tube is controlled within a certain set time, so that the conduction time of the first driving signal is kept unchanged, and meanwhile, the conduction time of a second driving signal is adjusted, so that the conduction time of the second driving signal is gradually increased until a main switch tube just realizes zero-voltage switching.
Compared with the prior art, the invention has the beneficial effects that:
1. the converter works in a leading edge non-complementary mode when the converter is under light load and no load, and works in a double-pulse non-complementary mode when the converter is over light load, so that the performance of a power supply system is better, and the requirement of EU six-level energy consumption can be met;
2. compared with the conventional back-edge non-complementary mode, the double-pulse non-complementary mode has the advantages that a front-edge driving signal (namely a first driving signal) is added, so that after the main switch tube is turned off, the resonance of leakage inductance and the clamping capacitor is not realized through the body diode of the clamping switch tube any more, but is realized through the clamping switch tube, and the problem of reverse recovery of the body diode of the clamping switch tube is solved.
3. The conduction time of the first pulse signal in the double-pulse non-complementary mode is controllable, the conduction time is 1/4 of the resonant period of the clamping capacitor and the leakage inductor, the reverse recovery problem is avoided, and the leakage inductor is reversely charged after the resonant current crosses zero.
Drawings
Fig. 1 is a circuit schematic block diagram of an active clamp flyback converter;
fig. 2 is a schematic diagram of an active clamp flyback converter according to an embodiment of the present invention;
FIG. 3 is a diagram illustrating the relationship between the FB voltage and the frequency according to an embodiment of the present invention;
fig. 4 is a waveform diagram of an ideal case where the active clamp flyback converter operates in the leading edge non-complementary mode according to the embodiment of the present invention;
fig. 5 is a waveform diagram of an ideal case where the active clamp flyback converter operates in the double-pulse non-complementary mode according to the embodiment of the present invention;
FIG. 6 is a schematic diagram illustrating a mode switching threshold return difference according to an embodiment of the present invention;
FIG. 7 is a waveform of a transition from a double-pulse non-complementary mode to a leading-edge non-complementary mode according to an embodiment of the present invention;
FIG. 8 is a schematic diagram of a first driving signal in a double-pulse non-complementary mode according to an embodiment of the present invention;
fig. 9 is a schematic diagram of an active clamp flyback converter according to a second embodiment of the present invention;
fig. 10 is an ideal operating waveform diagram of a second active clamp flyback converter according to an embodiment of the present invention in a flyback mode of operation;
FIG. 11 is a diagram illustrating the relationship between the FB voltage and the frequency in the second mode switching according to the embodiment of the present invention;
FIG. 12 is a diagram illustrating a second mode switching threshold return difference according to an embodiment of the present invention;
FIG. 13 is a waveform diagram of a second transition process according to the embodiment of the present invention.
Detailed Description
First embodiment
Referring to fig. 2, fig. 2 is a schematic diagram of an active-clamp flyback converter according to a first embodiment of the present invention, and an active-clamp flyback converter 700 (hereinafter, referred to as a converter) is used for converting an input voltage into an output voltage to be provided to a load, and includes: a transformer, a primary circuit, a secondary circuit, a controller 710, and an isolated feedback circuit 720.
The primary winding NP of the transformer is connected with the primary circuit, and the secondary winding NS of the transformer is connected with the secondary circuit.
The primary circuit consists of a leakage inductor LK, an excitation inductor LM, a clamping capacitor C _ C, a main switch tube S1, a clamping switch tube S2, an excitation inductor LM and a current sampling resistor RCS. In this embodiment, the main switch tube S1 and the clamp switch tube S2 are MOS tubes, respectively, a drain of the clamp switch S2 is connected to one end of the leakage inductor LK through a clamp capacitor C _ C, and a source of the clamp switch S2 is connected to a drain of the main switch tube S1; the other end of the leakage inductance LK is connected with the synonym end of the primary winding NP.
The secondary circuit comprises a rectifier diode DR and an output capacitor COUTThe anode of the rectifier diode DR is connected with the homonymous end of the secondary winding NS; cathode of rectifier diode DR and output capacitor COUTOne end of the output capacitor is connected with the output end of the transformerOUTAnd the other end of the secondary winding is connected with the synonym end of the secondary winding NS.
The isolation feedback circuit 720 is used for detecting the magnitude of the load, and in this embodiment, the detection of the magnitude of the load is realized by detecting the feedback signal FB voltage (i.e., the feedback voltage, hereinafter referred to as the FB voltage), and in other embodiments, the detection of the magnitude of the load is realized by detecting the magnitude of the load current.
The controller 710 (i.e., the main control chip of the converter 700) is connected to the gates of the main switch transistor S1 and the clamp switch transistor S2, respectively.
The key module circuits of the controller 710 include a mode determination module 711, a frequency control module 712, a pulse width control module 713, a timing control module 714, and a driving module 715.
Please refer to fig. 2 and fig. 3, wherein fig. 3 is a diagram of a relationship between a mode switching FB voltage and a frequency. The mode determination module 711 compares the detected FB voltage with a set mode switching threshold voltage Vth, and determines that the converter needs to operate in a double-pulse non-complementary mode and output a corresponding mode control signal when the FB voltage is greater than the threshold voltage Vth; and when the FB voltage is smaller than the threshold voltage Vth, judging that the converter needs to work in a leading edge non-complementary mode or a flyback mode and outputting a corresponding mode control signal.
The frequency control module 712 implements control of the operating frequency of the converter 700 (PFM control) by detecting the FB voltage, which is higher, the higher the frequency of the converter 700; when the FB voltage decreases to a certain extent, the frequency is locked and no longer follows the FB voltage variations.
The pulse width control module 713 samples the FB voltage and a peak signal Vcs generated by a primary side sampling resistor RCS, and turns off a main switching tube S1 when the primary side peak signal Vcs is greater than the FB voltage, so as to control the primary side peak current.
The timing control module 714 generates driving signals DRV _1 and DRV _2 according to the control signals output from the mode determination module, the frequency control module, and the pulse width control module.
The driving module 715 is configured to convert the driving signal DRV _1 of the main switching transistor S1 and the driving signal DRV _2 of the clamp switching transistor S2 into the driving signal GS _1 of the main switching transistor S1 and the driving signal GS _2 of the clamp switching transistor S2, so as to implement control of the operating mode, the operating frequency, and the primary side peak current of the converter 700.
The converter 700 of the present invention operates in the leading edge non-complementary mode during light load and no load operation of the converter 700 and operates in the double pulse non-complementary mode above light load. That is, when the FB voltage is detected to be less than the threshold voltage Vth, the load is considered to be a light load, and when the FB voltage is detected to be greater than the threshold voltage Vth, the load is considered to be a light load or more (or a heavy load).
Referring to fig. 4, fig. 4 shows key waveforms of the leading edge non-complementary mode. Wherein, the representation S1 is the driving signal waveform of the main switch tube S1, S2 is the driving voltage waveform of the clamp switch tube S2, and VDS1 is the drain-source voltage waveform of the main switch tube S1; ILM represents the current waveform through the excitation inductance LM.
Stage one (t)0～t1): this phase is the dead time, at t0At the moment, the driving signal of the main switch tube S1 is switched from high level to low level, the primary side excitation current charges the output junction capacitor of the main switch tube S1, the leakage inductor LK and the clamping capacitor C _ C charge the clamping capacitor C _ C through the body diode resonant current of the clamping switch tube S2, and when the voltage on the junction capacitor of the main switch tube S1 rises to Vin+nVoutWhen the voltage across the drain and source of the clamp switch tube S2 drops to zero, t1At which time the transformer begins to transfer energy to the secondary side.
Stage two (t)1～t2): at t1At the moment, the voltage at the two ends of the clamping switch tube S2 is reduced to zero, the clamping switch tube S2 realizes zero voltage switching-on, the leakage inductor LK and the clamping capacitor C _ C resonate through the clamping switch tube S2, the resonant current continuously charges the clamping capacitor C _ C, the energy stored in the leakage inductor LK is transferred to the clamping capacitor C _ C to be stored, and at the moment, the transformer still transfers the energy to the secondary side.
Stage three (t)2～t3): at t2At this time, the clamp switching tube S2 is turned off, the excitation current does not drop to zero, and energy continues to be transmitted to the secondary side until the excitation current becomes zero.
Stage four (t)3～t4): at t3At the moment, the exciting current is zero, the primary side does not transfer energy to the secondary side any more, the voltage at two ends of the primary winding NP of the transformer is zero, and at the moment, the leakage inductance LK and the exciting inductance LM of the transformer and the output junction capacitor of the main switching tube S1 resonate until t4At the moment, the main switch tube S1 is turned on, and the next cycle is entered.
When the converter 700 is in a low-load operation in the leading-edge non-complementary mode, the clamp capacitor C _ C is charged by the converter 700 in each period, and when the energy on the clamp capacitor C _ C is charged to a certain extent, the energy on the clamp capacitor C _ C is released and recovered through the clamp switch tube S2 in a certain period, so that the condition that a large resistor is connected to the clamp capacitor C _ C in parallel to consume the energy on the clamp capacitor C _ C is avoided, and the low-load efficiency is improved. When the circuit works under light load, the switching frequency of the main switching tube S1 and the clamping switching tube S2 is between 20KHz and 35KHz, so that the driving loss and the switching loss are reduced, and audible noise is not generated.
Referring to fig. 5, fig. 5 shows waveforms of the converter 700 operating in the double-pulse non-complementary mode. Wherein, the representation S1 is the driving signal waveform of the main switch tube S1, S2 is the driving voltage waveform of the clamp switch tube S2, and VDS1 is the drain-source voltage waveform of the main switch tube S1; ILM represents the current waveform through the excitation inductor LM; VDS1 represents the drain-source voltage waveform of clamp switch S2.
The working principle of the double-pulse non-complementary mode is as follows:
stage one (t)0～t1): this phase is the dead time, at t0At the moment, the driving signal of the main switch tube S1 is switched from high level to low level, the primary side excitation current charges the output junction capacitor of the main switch tube S1, the leakage inductor LK and the clamping capacitor C _ C charge the clamping capacitor C _ C through the body diode resonant current of the clamping switch tube S2, and when the voltage on the junction capacitor of the main switch tube S1 rises to Vin+nVoutWhile clampingThe voltage across the drain and source of the switch tube S2 drops to zero, t1At which time the transformer begins to transfer energy to the secondary side.
Stage two (t)1～t2): at t1At the moment, the voltage at the two ends of the clamping switch tube S2 is reduced to zero, the clamping switch tube S2 realizes zero voltage switching-on, the leakage inductor LK and the clamping capacitor C _ C resonate through the clamping switch tube S2, the resonant current continuously charges the clamping capacitor C _ C, the energy stored in the leakage inductor LK is transferred to the clamping capacitor C _ C to be stored, and at the moment, the transformer still transfers the energy to the secondary side.
Stage three (t)2～t3): at t2At the moment, the clamping switch tube S2 is closed, the exciting current does not drop to zero, and energy is continuously transmitted to the secondary side until t3The exciting current is reduced to zero at the moment.
Stage four (t)3～t4): at t3At the moment, the exciting current is zero, the primary side does not transfer energy to the secondary side any more, the voltage at two ends of the primary winding NP of the transformer is zero, and at the moment, the leakage inductance LK and the exciting inductance LM of the transformer and the output junction capacitor of the main switching tube S1 resonate until t4The time clamp switching tube S2 turns on again.
Stage five (t)4～t5): since the clamp switch tube S2 is turned on again, the clamp capacitor C _ C starts to discharge and reversely excites the exciting inductor, thereby generating a negative current at t5The clamp switch S2 is again turned off at this point.
Stage six (t)5～t6)：t5When the time goes into the dead time, as the clamping switch tube S2 is turned off, the reverse current extracts the charges on the capacitor of the S1 junction of the main switch tube, thereby creating a condition for realizing ZVS turn-on of the main switch tube S1 in the next period, and at t6The voltage at the two ends of the drain and the source of the main switch tube S1 is reduced to zero, and the main switch tube S1 is turned on to realize ZVS turning-on.
Referring to fig. 3 and 5, it can be seen from fig. 3 that when the load is gradually increased from no load, and the inverter 700 operates in the leading-edge non-complementary mode, the switching frequency is kept constant within a certain load range, when the load continues to increase, the switching frequency is correspondingly increased, and the FB voltage is also increased with the increase of the load, and when the load increases so that the feedback voltage FB is greater than the mode switching threshold voltage Vth, the operating mode of the inverter 700 is switched from the leading-edge non-complementary mode to the double-pulse non-complementary mode. As the load continues to increase, the FB voltage also increases, the switching frequency increases, when the clamp switching tube S2 is at the first peak of the Vds1 resonant waveform of the main switching tube S1 during the second conduction, the load increases thereafter, so that the switching frequency decreases, because the load increases, the required peak current becomes larger, and the excitation time becomes longer, at this time, the clamp switching tube S2 has already been turned on at the first peak of the resonant voltage, and the switching frequency cannot be raised by changing the turn-on time of the clamp switching tube S2, so when the load increases to a certain extent, the switching frequency decreases with the increase of the load. The working principle of load reduction is similar and is not described in detail herein.
Referring to fig. 6, fig. 6 is a schematic diagram of mode switching threshold back-off, in the present embodiment, there is back-off of the threshold voltage Vth of the mode switching reference (the threshold voltage Vth1 and the threshold voltage Vth2 are respectively, and the threshold voltage Vth1 is greater than the threshold voltage Vth2), so as to prevent the two modes from switching back and forth, that is, the actual FB voltage point for switching from the double-pulse mode to the leading-edge non-complementary mode and the leading-edge non-complementary mode to the double-pulse mode is different, and the back-off value can be freely set.
Referring to fig. 7, fig. 7 is a waveform diagram illustrating a transition process of the dual-pulse non-complementary mode to the leading edge non-complementary mode, when the FB voltage gradually decreases from being greater than the threshold voltage Vth2 of the mode switching to being less than the threshold voltage Vth2 of the mode switching, the converter 700 enters the transition process of the mode switching. The transition process is as follows: the on-time of the first driving signal of the double-pulse non-complementary mode is not changed and the on-time of the second driving signal is gradually reduced within a certain set time until the driving is reduced to be absent, at which time the transition process is completed and the converter 700 operates in the leading-edge non-complementary mode. When the FB voltage increases from less than the mode switching threshold voltage Vth1 to greater than the mode switching voltage Vth1, the converter 700 enters a transition of mode switching as follows: after the leading-edge non-complementary mode driving is kept unchanged and changed into the double-pulse non-complementary mode within the set time, the on-time of the second-time driving signal is gradually increased until the main switching tube S1 just achieves zero-voltage switching-on.
Referring to fig. 5 and 8, wherein fig. 8 is a schematic diagram of a first driving signal in a double-pulse non-complementary mode, Vds1 represents a drain-source voltage waveform of the main switch tube S1; GS-2 sends the first driving signal to the clamping switch tube S1; IlK denotes the resonant current on the leakage inductance LK.
After the main switch tube S1 is turned off, the clamp switch tube S2 is turned on for the first time after a dead time, if the conduction time of the clamp switch tube S2 is too short, taking the conduction time of the clamp switch tube S2 as a time period t0-t1 as an example, only the resonant current of the time period t0-t1 charges the clamp capacitor C _ C through the clamp switch tube S2, and the leakage inductance energy of the time period t1-t2 still has a part of energy to charge the clamp capacitor C _ C through the body diode of the clamp switch tube S2, and at this time, because the body diode flows current, the reverse recovery causes a negative current in the time period t2-t3, and finally affects the efficiency; if the on-time of the clamp switch tube S2 is too long, from t0-t3, after the energy of the leakage inductance LK is completely transferred to the clamp capacitor C _ C at time t2, since the clamp switch tube S2 is still on, at this time, the clamp capacitor C _ C is discharged through the clamp switch tube S2 to charge the leakage inductance LK, which causes voltage fluctuation at both drain and source terminals of the switch tube, and affects subsequent QR (quasi-resonant) conduction of the clamp switch tube S2. The clamp switch S2 will not have the body diode reverse recovery problem, but will have the same effect if turned on for too long.
If the conduction time of the clamp switch S2 is exactly at the point where the resonant current crosses zero, i.e., during the time period t0-t2, then there is no negative current. Therefore, the on-time control of the first pulse signal in the double-pulse non-complementary mode is important, and it is more appropriate to derive from the test waveform that the on-time of the clamp switch transistor S2 is equal to the resonance period 1/4 of the clamp capacitor C _ C and the leakage inductor LK.
Second embodiment
Referring to fig. 9, fig. 9 is a circuit diagram of an active-clamp flyback converter according to a second embodiment of the present invention, which is compared with the first embodiment, in the embodiment, a resistor Rbleed is connected in parallel to two ends of the clamping capacitor C _ C, since the clamping switch tube S2 is not driven in the light load, that is, in the light no-load mode, the converter 700 operates in the flyback mode, however, no matter what mode the leakage inductor LK operates in, the energy of the leakage inductor LK is always present, so that the energy of the leakage inductor LK charges the clamping capacitor C _ C through the body diode of the clamping switch tube S2, if the resistor Rbleed is not connected in parallel to the clamp capacitor C, the energy on the clamp capacitor C can accumulate, when the leakage inductance LK is accumulated to a certain degree, the energy of the leakage inductance LK cannot be charged, which may cause the converter 700 to operate abnormally, therefore, it is necessary to connect a resistor across the clamp capacitor C _ C to dissipate the energy in the clamp capacitor C _ C.
Referring to fig. 10, fig. 10 shows waveforms of converter 700 operating in the flyback mode. When the converter works in a flyback mode, the clamping switch tube S2 is always in an off state.
Referring to fig. 11, fig. 11 is a graph showing the relationship between the switching frequency and the feedback voltage of the dual-pulse non-complementary mode and the flyback mode. As can be seen from fig. 11, when the load is slowly increased from no load, and the converter 700 is operated in the flyback mode, the switching frequency is kept constant within a certain load range, and when the load is continuously increased, the switching frequency is correspondingly increased, and the FB voltage is also increased with the increase of the load, and when the load is increased, so that the voltage FB is greater than the mode switching threshold voltage Vth, the operation mode of the converter 700 is switched from the flyback mode to the double-pulse non-complementary mode.
As the load continues to increase, the FB voltage also increases, the switching frequency increases, when the clamping switch tube S2 is at the first peak of the Vds1 resonance waveform of the main switch tube S1 at the second turn-on time, the load increases thereafter, so that the switching frequency decreases, because the load increases, the required peak current becomes larger, and the excitation time becomes longer, at this time, the clamping switch tube S2 is already turned on at the first peak of the resonance voltage, and the switching frequency cannot be raised by changing the turn-on time of the clamping switch tube S2, so when the load increases to a certain extent, the switching frequency decreases with the increase of the load. The working principle of the load reduction is substantially the same as that of the first embodiment, and is not described herein again.
Referring to fig. 12, it should be noted that there is a back difference between the mode switching threshold voltage Vth (i.e. the threshold voltage Vth1 and the threshold voltage Vth2, and the threshold voltage Vth1 is greater than the threshold voltage Vth2), which prevents the switching between the two modes, that is, the actual FB voltage point for switching from the double-pulse non-complementary mode to the flyback mode is different from the actual FB voltage point for switching from the double-pulse non-complementary mode to the double-pulse non-complementary mode, and the back difference value can be set freely.
Referring to fig. 12 and fig. 13, in which fig. 13 is a waveform diagram of a transition process of the dual-pulse non-complementary mode and flyback mode switching, when the FB voltage gradually decreases from being greater than the threshold voltage Vth2 of the mode switching to being less than the threshold voltage Vth2 of the mode switching, the converter 700 enters the transition process of the mode switching. The transition process is as follows: in a set time, the on-time of the first driving signal and the on-time of the second driving signal in the double-pulse non-complementary mode are gradually reduced at the same time until the driving is not performed, at this time, the transition process is completed, and the converter 700 works in the flyback mode.
When the FB voltage increases from less than the mode switching threshold voltage Vth1 to greater than the mode switching voltage Vth1, the converter 700 enters a transition of mode switching as follows: and in a set time, the conduction time of the first driving signal and the conduction time of the second driving signal in the double-pulse non-complementary mode are simultaneously and gradually increased, wherein the conduction time of the first driving signal in the double-pulse non-complementary mode is kept unchanged when the conduction time of the first driving signal in the double-pulse non-complementary mode is increased to 1/4 of the resonance period of the clamping capacitor C _ C and the leakage inductor LK, and the conduction time of the second driving signal is continuously increased until the main switch tube S1 just achieves zero voltage switching-on.
Claims (10)
1. A control method of an active clamp flyback converter is characterized in that: comparing the detected feedback voltage with a set threshold voltage, and operating the active clamping flyback converter to work in a double-pulse non-complementary mode when the feedback voltage is greater than the threshold voltage Vth 1; when the feedback voltage is less than the threshold voltage Vth2, operating the active clamp flyback converter to work in a front edge non-complementary mode or a flyback mode, and when the feedback voltage is between the threshold voltage Vth2 and the threshold voltage Vth1, keeping the working mode of the current period consistent with the working mode of the previous period, wherein the threshold voltage Vth1 is greater than the threshold voltage Vth 2;
the switching between the double-pulse non-complementary mode and the leading-edge non-complementary mode has a transition process, in the transition process, the conduction time of a first driving signal transmitted to the clamping switch tube is kept unchanged, and the conduction time of a second driving signal transmitted to the clamping switch tube is gradually reduced in the transition process until the conduction time is not achieved finally; the switching between the leading edge non-complementary mode and the double-pulse non-complementary mode has another transition process, in the transition process, the conduction time of the first conduction driving signal transmitted to the clamping switch tube is unchanged, and the conduction time of the second conduction driving signal transmitted to the clamping switch tube is gradually increased from a certain minimum value to the main switch tube to just realize zero-voltage switching-on.
2. The control method according to claim 1, characterized in that: when the active clamping flyback converter works in a double-pulse non-complementary mode, the second switching-on moment of the clamping switch tube is the moment when the drain-source resonant voltage of the main switch tube is at a wave crest, and simultaneously, the moment when the drain-source resonant voltage of the clamping switch tube is at a wave trough.
3. The control method according to claim 2, characterized in that: and the second conduction moment of the clamping switch tube is the moment when any peak of the drain-source resonant voltage of the main switch tube appears.
4. An active-clamp flyback converter, comprising: a transformer, a primary circuit, a secondary circuit, an isolation feedback circuit and a controller,
the primary winding of the transformer is connected with the primary circuit, and the secondary winding of the transformer is connected with the secondary circuit;
the primary side circuit is provided with a leakage inductor, a clamping capacitor, a main switching tube and a clamping switching tube, wherein the drain electrode of the clamping switch is connected with one end of the leakage inductor through the clamping capacitor, and the source electrode of the clamping switch is connected with the drain electrode of the main switching tube; the other end of the leakage inductance is connected with the primary winding;
the isolation feedback circuit is connected with the output end of the secondary side circuit and used for detecting the feedback voltage of the output end;
the controller is respectively connected with the main switch tube and the clamping switch tube and is used for controlling the working mode of the active clamping flyback converter according to the feedback voltage, and specifically:
when the feedback voltage is greater than a threshold voltage Vth1, controlling the active clamp flyback converter to work in a double-pulse non-complementary mode; when the feedback voltage is less than a threshold voltage Vth2, controlling the active clamp flyback converter to work in a front edge non-complementary mode or a flyback mode, and when the feedback voltage is between a threshold voltage Vth2 and a threshold voltage Vth1, keeping the working mode of the current period consistent with the working mode of the previous period, wherein the threshold voltage Vth1 is greater than the threshold voltage Vth 2;
the switching between the double-pulse non-complementary mode and the leading-edge non-complementary mode has a transition process, in the transition process, the conduction time of a first driving signal transmitted to the clamping switch tube is kept unchanged, and the conduction time of a second driving signal transmitted to the clamping switch tube is gradually reduced in the transition process until the conduction time is not achieved finally; and in the transition process, the conduction time of the first conducted driving signal transmitted to the clamping switch tube is unchanged, and the conduction time of the second conducted driving signal transmitted to the clamping switch tube is gradually increased from a certain minimum value to the main switch tube to just realize zero-voltage switching-on.
5. The active-clamp flyback converter of claim 4, wherein: and the resistor is connected with the clamping capacitor in parallel and used for consuming energy on the clamping capacitor when the resistor works in a flyback mode.
6. A controller for an active clamp flyback converter, comprising:
the mode judging module is used for operating the converter to work in a double-pulse non-complementary mode or a leading edge non-complementary mode according to the load; specifically, when the size of the load is larger than a first threshold value, the active clamp flyback converter is controlled to work in a double-pulse non-complementary mode; when the size of the load is smaller than a second threshold value, controlling the active clamp flyback converter to work in a front edge non-complementary mode, wherein the first threshold value is larger than the second threshold value;
the frequency control module is used for controlling the working frequency of the converter according to the load size;
the pulse width control module is used for controlling the conduction time of a main switching tube of the converter according to the load;
the time sequence control module is respectively connected with the mode judgment module, the frequency control module and the pulse width control module and is used for generating a driving signal according to control signals output by the mode judgment module, the frequency control module and the pulse width control module;
the driving module is used for converting the driving signal and inputting the converted driving signal to the main switching tube and the clamping switching tube;
the switching between the double-pulse non-complementary mode and the leading-edge non-complementary mode has a transition process, in the transition process, the conduction time of a first driving signal transmitted to the clamping switch tube is kept unchanged, and the conduction time of a second driving signal transmitted to the clamping switch tube is gradually reduced in the transition process until the conduction time is not achieved finally; and in the transition process, the conduction time of the first conducted driving signal transmitted to the clamping switch tube is unchanged, and the conduction time of the second conducted driving signal transmitted to the clamping switch tube is gradually increased from a certain minimum value to the main switch tube to just realize zero-voltage switching-on.
7. The controller of claim 6, wherein: the load size is realized by detecting the feedback voltage or the load current.
8. A control method of an active clamp flyback converter is provided, the active clamp flyback converter is provided with a primary side circuit, a secondary side circuit and a transformer connected with the primary side circuit and the secondary side circuit, the primary side circuit is provided with a main switch tube, a clamp capacitor and a leakage inductance, and the control method is characterized by comprising the following steps:
detecting the size of the load;
when the size of the load is larger than a first threshold value, driving signals transmitted to the main switching tube and the clamping switching tube are controlled, so that the active clamping flyback converter works in a double-pulse non-complementary mode, and when the active clamping flyback converter works in the double-pulse non-complementary mode, the clamping switching tube receives the driving signals twice in each working period, wherein the conducting time of the first driving signal received by the clamping switching tube is 1/4 of the resonance period of the clamping capacitor and the leakage inductor;
when the size of the load is smaller than a second threshold value, driving signals transmitted to the main switching tube and the clamping switching tube are controlled, so that the active clamping flyback converter works in a front edge non-complementary mode or a flyback mode;
when the size of the load is between a second threshold and a first threshold, keeping the working mode of the current period consistent with the working mode of the last period, wherein the first threshold is larger than the second threshold.
9. The control method according to claim 8, characterized in that: when the size of the load is smaller than a third threshold value or zero, the active clamp flyback converter is controlled to work in a burst mode, and when the active clamp flyback converter works in the burst mode, the active clamp flyback converter works for a period of time and then has a rest for a period of time.
10. A control method of an active clamp flyback converter is provided, the active clamp flyback converter is provided with a primary side circuit, a secondary side circuit and a transformer connected with the primary side circuit and the secondary side circuit, the primary side circuit is provided with a main switch tube, a clamp capacitor and a leakage inductance, and the control method is characterized by comprising the following steps:
detecting the size of the load;
according to the load, driving signals transmitted to a main switching tube and a clamping switching tube are controlled, so that the active clamping flyback converter is switched between a double-pulse non-complementary mode and a leading-edge non-complementary mode;
when the active clamp flyback converter is switched from a double-pulse non-complementary mode to a leading-edge non-complementary mode, a first driving signal transmitted to a clamp switch tube is controlled within a certain set time, so that the conduction time of the first driving signal is kept unchanged, and meanwhile, the conduction time of a second driving signal transmitted to the clamp switch tube is adjusted, so that the conduction time of the second driving signal is gradually reduced until the second driving signal is reduced to zero;
when the active clamp flyback converter is switched from a leading edge non-complementary mode to a double-pulse non-complementary mode, a first driving signal transmitted to the clamp switch tube is controlled within a certain set time, so that the conduction time of the first driving signal is kept unchanged, and meanwhile, the conduction time of a second driving signal is adjusted, so that the conduction time of the second driving signal is gradually increased until a main switch tube just realizes zero-voltage switching.
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| CN114039487B (en) * | 2021-07-26 | 2024-06-25 | 杰华特微电子股份有限公司 | Asymmetric half-bridge flyback converter and control method thereof |
| CN113824328B (en) * | 2021-08-17 | 2024-06-18 | 广州金升阳科技有限公司 | Flyback converter |
| CN114189160B (en) * | 2021-12-07 | 2023-12-12 | 南京微盟电子有限公司 | Light-load conduction control method and circuit for clamping tube in active clamping flyback topology |
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| CN103795260B (en) * | 2014-01-21 | 2016-01-20 | 广州金升阳科技有限公司 | A kind of incomplementarity flyback active clamp converter |
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