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CN110943641A - Pulse width modulation method of current type three-phase high-frequency link matrix inverter - Google Patents

Pulse width modulation method of current type three-phase high-frequency link matrix inverter Download PDF

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CN110943641A
CN110943641A CN201911155589.1A CN201911155589A CN110943641A CN 110943641 A CN110943641 A CN 110943641A CN 201911155589 A CN201911155589 A CN 201911155589A CN 110943641 A CN110943641 A CN 110943641A
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controllable switch
switch tube
frequency
current
collector
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CN110943641B (en
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闫朝阳
代会文
赵丁选
张祝新
奚子伟
段浩天
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Yanshan University
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Yanshan University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/42—Conversion of DC power input into AC power output without possibility of reversal
    • H02M7/44—Conversion of DC power input into AC power output without possibility of reversal by static converters
    • H02M7/48—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/539—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency
    • H02M7/5395—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency by pulse-width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/42—Conversion of DC power input into AC power output without possibility of reversal
    • H02M7/44—Conversion of DC power input into AC power output without possibility of reversal by static converters
    • H02M7/48—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53875—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with analogue control of three-phase output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Inverter Devices (AREA)
  • Ac-Ac Conversion (AREA)

Abstract

本发明公开一种电流型三相高频链矩阵式逆变器的脉宽调制方法。所述电流型三相高频链矩阵式逆变器拓扑由全桥电流型高频逆变器、高频变压器T、三相矩阵式变换器、输出CL型滤波器、负载依次连接构成；其前级变压器T原边的全桥电流型高频逆变器采用一对具有重叠导通时间的占空比大于50％的高频方波驱动控制，可控开关管S1、S4和可控开关管S2、S3交替导通；后级变压器T副边的三相矩阵式变换器可以等效分解成两组普通三相电流型逆变器进行控制。采用本发明提供的脉宽调制方法，可以实现拓扑中后级矩阵变换器所有可控开关管ZCS，减小可控开关管的开关损耗，提高变换器的效率。本发明还具有高效、控制方法简单且易于实现、电路稳定性高的优点。

Figure 201911155589

The invention discloses a pulse width modulation method of a current-type three-phase high-frequency chain-matrix inverter. The current-type three-phase high-frequency chain-matrix inverter topology is composed of a full-bridge current-type high-frequency inverter, a high-frequency transformer T, a three-phase matrix converter, an output CL-type filter, and a load connected in sequence; The full-bridge current-mode high-frequency inverter on the primary side of the transformer T adopts a pair of high-frequency square-wave drive controls with overlapping on-time duty ratios greater than 50%, controllable switch tubes S 1 , S 4 and controllable switches The tubes S 2 and S 3 are turned on alternately; the three-phase matrix converter on the secondary side of the latter stage transformer T can be equivalently decomposed into two groups of ordinary three-phase current-mode inverters for control. By using the pulse width modulation method provided by the invention, all the controllable switch tubes ZCS of the rear-stage matrix converter in the topology can be realized, the switching loss of the controllable switch tubes can be reduced, and the efficiency of the converter can be improved. The invention also has the advantages of high efficiency, simple and easy control method, and high circuit stability.

Figure 201911155589

Description

Pulse width modulation method of current type three-phase high-frequency link matrix inverter
Technical Field
The invention relates to the technical field of topology and modulation of power electronic power converters, in particular to a pulse width modulation method of a current type three-phase high-frequency link matrix inverter.
Background
An inverter is a topological device that converts direct current electrical energy into alternating current electrical energy. The high-frequency chain inverter adopts a high-frequency transformer to replace a power frequency transformer, and overcomes the defects of large volume, high noise, high cost and the like of the traditional transformer. The conversion process of the high-frequency link matrix inverter has three power characteristics of DC (direct current)/HFAC (high frequency alternating current)/LFAC (low frequency alternating current). It is known that a DC/AC (i.e., direct current/alternating current) inversion link occurs in such an inverter, and the DC/AC inversion link is located at the primary side of the transformer, and an AC/AC (i.e., alternating current/alternating current) conversion link also occurs, and the DC/AC inversion link is also called a matrix converter link and is located at the secondary side of the transformer. Compared with the traditional converter, the matrix converter has no intermediate energy storage link, adopts a bidirectional switch, can realize bidirectional flow of energy, has compact structure, small volume and high efficiency, and can independently control the amplitude and frequency of output voltage.
The method mainly comprises the following safety commutation strategies that ① can realize soft switching by adding active clamping to inhibit voltage overshoot, but the introduced clamping circuit increases cost, and the added controllable power tube also makes control more complicated, the ② unipolar and bipolar phase shift control strategies realize natural commutation of inductive current by means of commutation overlapping of the matrix converter, and realize ZVS (zero voltage Switch) of the power tube, but the problems that commutation overlapping time is not easy to control and the like, ③ introduces a series resonant circuit in a front-stage inverter to realize soft commutation of the power tube, at the moment, switching of the power tube is required to occur at a zero current moment, and control output energy needs to judge the working state of the resonant circuit, so that the control mode is complicated.
Therefore, although the above strategy can realize safe commutation, modulation and control of the inverter are more complicated, and system reliability is reduced, so that popularization and application of the converter are affected.
Disclosure of Invention
The invention aims to provide a pulse width modulation method of a current type three-phase high-frequency chain matrix inverter, which aims to solve the problems of complex modulation process and low reliability of a safety commutation strategy of the conventional high-frequency chain matrix inverter.
In order to achieve the purpose, the invention provides the following scheme:
a pulse width modulation method of a current type three-phase high-frequency chain matrix inverter comprises a full-bridge current type high-frequency inverter, a high-frequency transformer T, a three-phase matrix converter, an output CL type filter and a load which are connected in sequence;
the full-bridge current type high-frequency inverter comprises a direct-current voltage source E, an energy storage inductor L and a controllable switching tube S1Controllable switch tube S2Controllable switch tube S3And a controllable switching tube S4(ii) a The three-phase matrix converter comprises a controllable switch tube S1aControllable switch tube S4bControllable switch tube S4aControllable switch tube S1bControllable switch tube S3aControllable switch tube S6bControllable switch tube S6aControllable switch tube S3bControllable switch tube S5aControllable switch tube S2bControllable switch tube S2aAnd a controllable switching tube S5b(ii) a The output CL type filter comprises a first capacitor Cf1A second capacitor Cf2A third capacitor Cf3A first inductor Lf1A second inductor Lf2And a third inductance Lf3(ii) a The load comprises a first load R1A second load R2And a third load R3；
The positive electrode of the direct-current voltage source E and the energy storage batteryOne end of the inductor L is connected; the other end of the energy storage inductor L is respectively connected with the controllable switch tube S1Collector electrode of, the controllable switch tube S3The collector electrodes are connected; the negative electrode of the direct current voltage source E is respectively connected with the controllable switch tube S2Emitter of, the controllable switching tube S4The emitting electrodes are connected; the controllable switch tube S1The emitting electrode of the high-frequency transformer T is respectively connected with one end of the primary side of the high-frequency transformer T and the controllable switch tube S2The collector electrodes are connected; the controllable switch tube S3The emitting electrode of the high-frequency transformer T is respectively connected with the other end of the primary side of the high-frequency transformer T and the controllable switch tube S4The collector electrodes are connected;
one end of the secondary side of the high-frequency transformer T is respectively connected with the controllable switch tube S1aCollector electrode of, the controllable switch tube S3aCollector electrode of, the controllable switch tube S5aThe collector electrodes are connected; the other end of the secondary side of the transformer T is respectively connected with the controllable switch tube S1bCollector electrode of, the controllable switch tube S3bCollector electrode of, the controllable switch tube S5bThe collector electrodes are connected; the controllable switch tube S1aAnd the controllable switch tube S4bIs connected with the emitting electrode of the controllable switch tube S3aAnd the controllable switch tube S6bIs connected with the emitting electrode of the controllable switch tube S5aAnd the controllable switch tube S2bThe emitting electrodes are connected; the controllable switch tube S1bAnd the controllable switch tube S4aIs connected with the emitting electrode of the controllable switch tube S3bAnd the controllable switch tube S6aIs connected with the emitting electrode of the controllable switch tube S5bAnd the controllable switch tube S2aThe emitting electrodes are connected; the controllable switch tube S4aCollector and the controllable switch tube S4bThe collector electrodes are connected; the controllable switch tube S6aCollector and the controllable switch tube S6bThe collector electrodes are connected; the controllable switch tube S2aCollector and the controllable switch tube S2bThe collector electrodes are connected;
said canControl switch tube S4aCollector and the controllable switch tube S4bRespectively connected with the first capacitor Cf1One terminal of, the first inductance Lf1One end of the two ends are connected; the controllable switch tube S6aCollector and the controllable switch tube S6bRespectively connected with the second capacitor Cf2One terminal of the second inductor Lf2One end of the two ends are connected; the controllable switch tube S2aCollector and the controllable switch tube S2bAre connected with the third capacitor C respectivelyf3One terminal of, the third inductance Lf3One end of the two ends are connected; the first capacitor Cf1Is connected to the second capacitor Cf2The other end of the third capacitor Cf3The other ends of the two are connected;
the first inductor Lf1And the other end of the first load R1One end of the two ends are connected; the second inductor Lf2And the other end of the second load R2One end of the two ends are connected; the third inductor Lf3And the other end of the third load R3One end of the two ends are connected; the first load R1Respectively with the second load R2The other end of (b), the third load R3The other ends of the two are connected;
the pulse width modulation method comprises the following steps:
a full-bridge current type high-frequency inverter on the primary side of a high-frequency transformer T converts direct current into high-frequency square wave current under the drive of high-frequency square waves;
the three-phase matrix converter converts the high-frequency square wave current output by the secondary side of the high-frequency transformer T into low-frequency pulsating current;
and the output CL filter converts the low-frequency pulsating current into three-phase power-frequency sinusoidal current to supply power to a load.
Optionally, the full-bridge current type high-frequency inverter converts the direct current into the high-frequency square wave current under the driving of the high-frequency square wave, and specifically includes:
driving the said with a pair of high frequency square waves with overlapping on-time and duty ratio greater than 50%The controllable switch tube S is connected to the full-bridge current type high-frequency inverter on the primary side of the high-frequency transformer T1The controllable switch tube S4The first bridge arm and the controllable switch tube S2The controllable switch tube S3And the second bridge arm is alternatively conducted, and the direct current output by the direct current voltage source E is converted into high-frequency square wave current.
Optionally, the controllable switch tube S1The controllable switch tube S4The first bridge arm and the controllable switch tube S2The controllable switch tube S3The overlapping conduction angle of the second bridge arm is theta, and the value range of the overlapping conduction angle is more than 0 degree and less than theta and less than 180 degrees.
Optionally, during a switching period TsWithin time, the controllable switch tube S1The controllable switch tube S4The first bridge arm and the controllable switch tube S2The controllable switch tube S3The common conduction time of the second bridge arm is Tcom＝Ts*2θ/360°。
Optionally, the three-phase matrix converter converts the high-frequency square wave current output by the high-frequency transformer T into a low-frequency pulsating current, and specifically includes:
equivalently decomposing the three-phase matrix converter into two groups of common current type three-phase inverters which are respectively a positive group inverter and a negative group inverter; the positive group inverter comprises a controllable switch tube S1aControllable switch tube S2aControllable switch tube S3aControllable switch tube S4aControllable switch tube S5aAnd a controllable switching tube S6a(ii) a The negative group inverter comprises a controllable switch tube S1bControllable switch tube S2bControllable switch tube S3bControllable switch tube S4bControllable switch tube S5bAnd a controllable switching tube S6b；
Each group of current type three-phase inverters adopts a 120-degree modulation method of a common current type three-phase inverter, and driving signals of the positive group of inverters and the negative group of inverters are subjected to AND logic synthesis, so that when the positive group of inverters work, all controllable switching tubes of the negative group of inverters are turned off; when the negative group inverter works, all the controllable switching tubes of the positive group inverter are switched off, and the high-frequency square wave current output by the high-frequency transformer T is converted into low-frequency pulsating current.
According to the specific embodiment provided by the invention, the invention discloses the following technical effects:
the invention provides a pulse width modulation method of a current type three-phase high-frequency chain matrix inverter, wherein the topology of the current type three-phase high-frequency chain matrix inverter is formed by sequentially connecting a full-bridge current type high-frequency inverter, a high-frequency transformer T, a three-phase matrix converter, an output CL type filter and a load; the full-bridge current type high-frequency inverter of the primary side of the pre-stage transformer T adopts a pair of high-frequency square waves with overlapping conduction time and duty ratio more than 50 percent for driving control, and the controllable switching tube S1Controllable switch tube S4And a controllable switching tube S2Controllable switch tube S3Conducting alternately; the three-phase matrix converter on the T secondary side of the post-stage transformer can be equivalently decomposed into two groups of common three-phase current type inverters for control. By adopting the pulse width modulation method, all controllable switching tubes ZCS (Zero Current Switch) of the post-stage matrix converter in the topology can be realized, the switching loss of the controllable switching tubes is reduced, and the efficiency of the converter is improved. The invention also has the advantages of high efficiency, simple control method, easy realization, high circuit stability and the like.
Drawings
In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the drawings required to be used in the embodiments will be briefly described below, and it is obvious that the drawings in the following description are only some embodiments of the present invention, and for those skilled in the art, other drawings can be obtained according to the drawings provided by the present invention without any creative effort.
Fig. 1 is a circuit topology diagram of a current type three-phase high-frequency link matrix inverter provided by the invention.
Fig. 2 is a schematic waveform diagram of the operating state of the inverter in a high frequency period according to the present invention.
Fig. 3 is a schematic diagram of the three-phase matrix converter on the secondary side of the post-stage transformer T provided by the present invention being equivalently decomposed into two sets of current-mode three-phase inverters.
Fig. 4 is a schematic diagram illustrating a pulse width modulation method of the current-type three-phase high-frequency link matrix inverter according to the present invention.
Fig. 5 is a mode circuit diagram of the current-type three-phase high-frequency link matrix inverter provided by the present invention within one high-frequency period; fig. 5(a) is a mode circuit diagram of the current-type three-phase high-frequency link matrix inverter provided by the present invention in an operating mode 1; fig. 5(b) is a mode circuit diagram of the current-type three-phase high-frequency link matrix inverter provided in the present invention in the operating mode 2; fig. 5(c) is a mode circuit diagram of the current-type three-phase high-frequency link matrix inverter provided in the present invention in the operating mode 3; fig. 5(d) is a mode circuit diagram of the current-type three-phase high-frequency link matrix inverter according to the present invention in the operating mode 4.
Detailed Description
The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
The invention aims to provide a pulse width modulation method of a current type three-phase high-frequency link matrix inverter with less power conversion grade and simple modulation.
In order to make the aforementioned objects, features and advantages of the present invention comprehensible, embodiments accompanied with figures are described in further detail below.
Fig. 1 is a circuit topology diagram of a current type three-phase high-frequency link matrix inverter provided by the invention. As shown in fig. 1, the current-type three-phase high-frequency chain matrix inverter topology according to the present invention is formed by sequentially connecting a full-bridge current-type high-frequency inverter, a high-frequency transformer T, a three-phase matrix converter, an output CL type filter, and a load.
Wherein, the full-bridge current type high-frequency inverter comprises a DC voltage source E, an energy storage inductor L and a controllable switch tube S1Controllable switch tube S2Controllable switch tube S3Controllable switch tube S4And (4) forming.
The three-phase matrix converter is composed of a controllable switch tube S1aControllable switch tube S4bControllable switch tube S4aControllable switch tube S1bControllable switch tube S3aControllable switch tube S6bControllable switch tube S6aControllable switch tube S3bControllable switch tube S5aControllable switch tube S2bControllable switch tube S2aControllable switch tube S5bAnd (4) forming.
The output CL type filter is composed of a first capacitor Cf1A second capacitor Cf2A third capacitor Cf3A first inductor Lf1A second inductor Lf2A third inductor Lf3And (4) forming.
The load is composed of a first load R1A second load R2A third load R3And (4) forming.
As shown in fig. 1, the positive electrode of the dc voltage source E is connected to one end of the energy storage inductor L, and the other end of the energy storage inductor L is connected to the controllable switch tube S respectively1Collector electrode, controllable switch tube S3The negative pole of the direct current voltage source E is respectively connected with a controllable switch tube S2Emitter and controllable switch tube S4Are connected.
Controllable switch tube S1The emitting electrode of the high-frequency transformer T is respectively connected with one end of the primary side of the high-frequency transformer T and the controllable switch tube S2The collector electrodes are connected; controllable switch tube S3The emitting electrode of the high-frequency transformer T is respectively connected with the other end of the primary side of the high-frequency transformer T and the controllable switch tube S4Is connected to the collector of the collector.
One end of the secondary side of the high-frequency transformer T is respectively connected with a controllable switch tube S1aCollector electrode, controllable switch tube S3aCollector electrode, controllable switch tube S5aCollector electrode ofThe other end of the secondary side of the transformer T is respectively connected with a controllable switch tube S1bCollector electrode, controllable switch tube S3bCollector electrode, controllable switch tube S5bIs connected to the collector of the collector. Controllable switch tube S1aEmitter and controllable switch tube S4bIs connected with a controllable switch tube S3aEmitter and controllable switch tube S6bIs connected with a controllable switch tube S5aEmitter and controllable switch tube S2bThe emitting electrodes are connected; controllable switch tube S1bEmitter and controllable switch tube S4aIs connected with a controllable switch tube S3bEmitter and controllable switch tube S6aIs connected with a controllable switch tube S5bEmitter and controllable switch tube S2aAre connected.
Controllable switch tube S4aCollector and controllable switch tube S4bAre respectively connected with a first capacitor Cf1One end of (1), the first inductance Lf1Is connected to one end of a first inductor Lf1Another end of (1) and a load R1Are connected at one end to a load R1Respectively with the other end of the load R2Another end of (1), load R3And the other end of the two are connected. A first capacitor Cf1The other end of the first capacitor is respectively connected with a second capacitor Cf2Another terminal of (1), a third capacitance Cf3And the other end of the two are connected.
Controllable switch tube S6aCollector and controllable switch tube S6bAre respectively connected with a second capacitor Cf2One terminal of (1), a second inductance Lf2Is connected to one end of a second inductor Lf2Another end of (1) and a load R2Are connected at one end to a load R2Respectively with the other end of the load R1Another end of (1), load R3And the other end of the two are connected. Second capacitor Cf2The other end of the first capacitor C is connected with the first capacitor Cf1Another terminal of (1), a third capacitance Cf3And the other end of the two are connected.
Controllable switch tube S2aCollector and controllable switch tube S2bAre connected with a third capacitor C respectivelyf3One terminal of (1), third inductance Lf3Is connected to one end of a third inductor Lf3Another end of (1) and a load R3Are connected at one end to a load R3Respectively with the other end of the load R1Another end of (1), load R2And the other end of the two are connected. Third capacitor Cf3The other end of the first capacitor C is connected with the first capacitor Cf1Another terminal of (1), a second capacitor Cf2And the other end of the two are connected.
Based on the topological structure of the current type three-phase high-frequency chain matrix inverter, the invention provides a pulse width modulation method of the current type three-phase high-frequency chain matrix inverter, which comprises the following steps:
the full-bridge current type high-frequency inverter on the primary side of the pre-stage transformer T is driven by a pair of high-frequency square waves with overlapping conduction time and duty ratio of more than 50 percent, and the controllable switching tube S1Controllable switch tube S4The first bridge arm and the controllable switch tube S2Controllable switch tube S3The second bridge arm is alternatively conducted. Because of the existence of the preceding stage energy storage inductor L, the direct current side current pulsation is small and is approximate to direct current. The direct current is converted into a high-frequency square wave current through a full-bridge current type high-frequency inverter.
The three-phase matrix converter on the secondary side of the T of the post-stage transformer is equivalently decomposed into two groups of common current type three-phase inverters (a positive group inverter and a negative group inverter) for control, and the controllable switch tube S is controlled according to current type modulation logic1aControllable switch tube S6aControllable switch tube S1bControllable switch tube S6bAnd controlling to convert the high-frequency square wave current output by the transformer T into low-frequency pulsating current.
Further, controllable switch tube S of full-bridge current type high-frequency inverter on primary side of preceding stage transformer T1And S4、S2And S3The overlapping conduction angle is theta, and the value range of theta is more than 0 degree and less than 180 degrees. In a switching period TsWithin time, the switch tube S can be controlled1And S4、S2And S3The common on-time of (c) is: t iscom＝Ts*2θ/360°。
Further, three-phase matrix type transformer of T secondary side of post-stage transformerThe converter is equivalently decomposed into two groups of common current type three-phase inverters, the controllable switch tubes of the three-phase matrix converter are decomposed into a positive group and a negative group by adopting a 120-degree modulation mode, and the positive group of controllable switch tubes S is defined1a～S6aControllable switching tube S of sum-negative group1b～S6bWhen the positive group of controllable switch tubes work, the negative group of controllable switch tubes are all turned off, and when the negative group of controllable switch tubes work, the positive group of controllable switch tubes are all turned off.
The working process of the current type three-phase high-frequency chain matrix inverter is as follows:
a current type full-bridge high-frequency inverter of a preceding stage high-frequency transformer T primary side adopts a pair of high-frequency square wave drive control methods with overlapping conduction time and duty ratio of more than 50 percent to enable a controllable switching tube S of the preceding stage inverter1And S4、S2And S3Alternately conducting and controllable switch tube S1And S4、S2And S3The bridge arm has an overlapping conduction angle theta. The direct current is modulated and converted into high-frequency alternating square wave current with a dead zone with the width of theta through a front-stage high-frequency inverter.
And the three-phase matrix converter on the T secondary side of the rear-stage high-frequency transformer is equivalently decomposed into two groups of common current type three-phase inverters. Wherein, the controllable switch tube S1a-S6aForm a positive inverter with a controllable switching tube S1b-S6bForming a negative group inverter. Each group of inverters adopts a 120-degree modulation method of a common current type three-phase inverter, and when the positive group of inverters work, the controllable switching tubes of the negative group of inverters are all turned off through AND logic synthesis. When the negative group inverter works, the controllable switching tubes of the positive group inverter are all turned off. The high-frequency alternating current output by the transformer is converted into three-phase power frequency pulsating current through the three-phase matrix converter, and the three-phase power frequency pulsating current is converted into three-phase power frequency sinusoidal current through the output CL filter to supply power to a load.
Fig. 2 is a schematic waveform diagram of the operating state of the inverter in a high frequency period according to the present invention. Controllable switch tube S of post-level matrix converter1a、S2a、S1b、S2bThe working time is for illustration. In FIG. 2VS1&S4、VS2&S3Controllable switch tube S of full-bridge current type high-frequency inverter respectively serving as primary side of preceding-stage transformer T1&S4、S2&S3The drive signal of (1). VS1a、VS2a、VS1b、VS2bControllable switch tube S in three-phase matrix converter with secondary side of post-stage transformer T1a、S2a、S1b、S2bThe drive signal of (1). i.e. isThe current waveform of the secondary side of the transformer T. As can be seen from fig. 2, the preceding stage power switch tube driving signal is a pair of high frequency square wave signals with overlapping on-time and duty ratio greater than 50%; meanwhile, in a high-frequency period of the topology, when the output current of the secondary side of the transformer T is zero, the rear-stage controllable switch tube acts, and overvoltage peak caused by cutting off the leakage inductance current flow path of the transformer can be avoided.
Fig. 3 is a schematic circuit equivalent decomposition diagram of the three-phase matrix converter of the post-stage transformer T according to the present invention. The pulse width modulation method enables the three-phase matrix converter to be equivalently decomposed into two common three-phase current type inverters which are respectively a positive group inverter and a negative group inverter. When the input current of the transformer is up-positive and down-negative, the controllable switch tube S of the positive group inverter1a、S2a、S3a、S4a、S5a、S6aControllable switch tube S of working negative group inverter1b、S2b、S3b、S4b、S5b、S6bIn an off state; controllable switch tube S of negative group inverter when input current signal of transformer is up-negative-up-right-time1b、S2b、S3b、S4b、S5b、S6bControllable switch tube S of working positive group inverter1a、S2a、S3a、S4a、S5a、S6aIn an off state.
Fig. 4 is a schematic diagram of the pulse width modulation method of the current-type three-phase high-frequency link matrix inverter according to the present invention. The pulse width modulation strategy adopted by the pulse width modulation method of the invention is realized by an AND gate, u in figure 4sIs the voltage waveform of the secondary side of the transformer T, T is time, VSPWM1And VSPWM4120 degree modulated wave of A phase 50Hz, VpAnd VnIs a pair of complementary square waves with the same frequency as the high-frequency inversion of the preceding stage and the duty ratio of 50 percent, VS1a、VS4b、VS4a、VS1bControllable switch tube S in three-phase matrix converter with secondary side of post-stage transformer T1a、S4b、S4a、S1bThe drive signal of (1). As can be seen from fig. 4, only one switching tube of the rear-stage controllable switching tube is turned on (the amplitude is 0 off, and the amplitude is 1 on) at any time in the same bridge arm.
Fig. 5 is a mode circuit diagram of the current-type three-phase high-frequency link matrix inverter provided by the invention in one high-frequency period. Fig. 5(a) to (d) correspond to the following operation modes 1 to 4, respectively. For the sake of analysis, it is assumed that the circuit operates in an ideal state, i.e. without taking into account the influence of any distribution and spurious parameters of the system on the circuit. According to the working principle, the current type three-phase high-frequency chain matrix inverter has 4 working modes in a high-frequency period, and the specific mode analysis is as follows:
(1) working mode 1[ t ]1-t2]: as shown in fig. 2 and 5(a), t1Before the moment, the preceding stage switching tube S1、S4Off, S2、S3Opening; rear stage switch tube S1b、S2bAnd the power is in an on state, and the rest are all turned off. t is t2After the moment, the preceding switching tube S2、S3Off, S1、S4Opening; rear stage switch tube S1a、S2aAnd the power is in an on state, and the rest are all turned off.
At t1Four switching tubes S at the moment and before1～S4Simultaneously, a DC voltage source E is connected to the switch tube S1、S2And a switching tube S3、S4Two loops are formed for follow current; transformer T primary side leakage inductance current and switch tube S2A DC voltage source E, an energy storage inductor L and a switch tube S3Forming a circulation path to discharge to zero rapidly; transformer T secondary side leakage current is because of not having primary side energy to transmitThe discharge is gradually and rapidly reduced to zero. Then, the rear-stage inverter is changed from negative group work to positive group work, and the switching tube realizes zero current switching-on and switching-off. At this stage, the first capacitor Cf1From a charged state to a discharged state and provides energy to the load. Second capacitor Cf2Is always in a charging state. Third capacitor Cf3From a discharged state to a charged state.
(2) Working mode 2[ t ]2-t3]: at this stage, the pre-switch S is in the position shown in FIGS. 2 and 5(b)1、S4Opening, S2、S3And (6) turning off. Rear stage switch tube S1a、S2aIs in an on state. The front stage is in normal working state, the primary side of the transformer T inputs the current with positive upper part and negative lower part, the secondary side induces and outputs the current with positive lower part and negative upper part, and the current passes through S1aLoad, S2aForming a flow path. At this stage, the first capacitor Cf1In a charging state. Second capacitor Cf2In a charging state. Third capacitor Cf3Is in a discharge state.
(3) Working mode 3[ t ]3-t4]: as shown in FIGS. 2 and 5(c), t3Before the moment, the preceding stage switching tube S2、S3Off, S1、S4And (4) opening. Rear stage switch tube S1a、S2aAnd the power is in an on state, and the rest are all turned off. t is t4After the moment, the preceding switching tube S1、S4Off, S2、S3And (4) opening. Rear stage switch tube S1b、S2bAnd the power is in an on state, and the rest are all turned off.
At t3At the moment, the four switching tubes of the preceding stage are simultaneously switched on, and the power supply is respectively connected with the switching tubes S1、S2And a switching tube S3、S4Two loops are formed for follow current. Transformer T primary side leakage inductance current and switch tube S4A DC voltage source E, an energy storage inductor L and a switch tube S1The circulating path is formed to discharge to zero rapidly. The leakage current of the secondary side of the transformer T is quickly discharged to zero due to no primary side energy transfer. Then, the back stage is changed from positive group work to negative group work, and the switching tube realizes zero current switching on and off. At this stage, the first powerContainer Cf1From a charged state to a discharged state and provides energy to the load. Second capacitor Cf2Is always in a charging state. Third capacitor Cf3From a discharged state to a charged state.
(4) Working mode 4[ t ]4-t5]: at this stage, the pre-switch S is in the position shown in FIGS. 2 and 5(d)2、S3Opening, S1、S4And (6) turning off. Rear stage switch tube S1b、S2bIs in an on state. The front stage is in normal working state, the primary side of the transformer T inputs the current with positive lower part and negative upper part, the secondary side induces and outputs the current with positive upper part and negative lower part, and the current passes through S1bLoad, S2bForming a flow path. At this stage, the first capacitor Cf1In a charging state. Second capacitor Cf2In a charging state. Third capacitor Cf3Is in a discharge state.
It can be seen from the above working processes that the Current type three-phase high-frequency link matrix inverter adopting the pulse width modulation method of the present invention can realize Zero Current Switch (ZCS) of the switching tube at the rear stage in the topology, thereby greatly reducing the overvoltage peak caused by the leakage inductance of the transformer and the voltage oscillation caused by the parasitic capacitance of the controllable switching tube.
Compared with the prior art, the pulse width modulation method of the current type three-phase high-frequency link matrix inverter provided by the invention has the following advantages:
(1) the existing three-phase Boost type high-frequency alternating current link grid-connected inverter mostly adopts a Modulation mode of Space Vector Pulse Width Modulation (SVPWM), and the method is complex. The pulse width modulation method provided by the invention has the advantages of simple modulation process, safety and reliability.
(2) The front stage of the current type three-phase high-frequency chain matrix inverter is driven by high-frequency square waves with overlapping conduction time and duty ratio of more than 50%, so that a controllable switch tube S of the front stage inverter1And S4、S2And S3The bridge arm has the time of overlapping conduction angle, which not only accords with the characteristic that the equivalent current source can not be broken, but also ensures that the high-frequency transformer of the rear-stage three-phase matrix converter has no transmission during the commutation periodAnd current is output to realize a rear-stage controllable switching tube ZCS, so that the phenomenon of very high voltage peak generated at two ends of the switching tube due to interruption of a circulation path of leakage inductance current of the transformer is reduced, the switching loss of the controllable switching tube is reduced, and the reliability and efficiency of the circuit are improved.
(3) Aiming at the current type three-phase high-frequency chain matrix inverter, no relevant research on modulation design through logic synthesis exists at present, and the later-stage three-phase matrix converter adopts AND logic to drive and synthesize, so that the modulation principle is simple and is easy to realize.
The above description is only for the preferred embodiment of the present invention, but the scope of the present invention is not limited thereto, and any changes or substitutions that can be easily conceived by those skilled in the art within the technical scope of the present invention disclosed herein should be covered within the scope of the present invention. Therefore, the protection scope of the present invention shall be subject to the protection scope of the claims.
The principles and embodiments of the present invention have been described herein using specific examples, which are presented solely to aid in the understanding of the apparatus and its core concepts; meanwhile, for a person skilled in the art, according to the idea of the present invention, the specific embodiments and the application range may be changed. In view of the above, the present disclosure should not be construed as limiting the invention.

Claims (5)

1.一种电流型三相高频链矩阵式逆变器的脉宽调制方法，其特征在于，所述脉宽调制方法应用于一种电流型三相高频链矩阵式逆变器；所述电流型三相高频链矩阵式逆变器包括依次连接的全桥电流型高频逆变器、高频变压器T、三相矩阵式变换器、输出CL型滤波器以及负载；1. A pulse width modulation method for a current-type three-phase high-frequency chain-matrix inverter, wherein the pulse-width modulation method is applied to a current-type three-phase high-frequency chain-matrix inverter; The phase high-frequency chain-matrix inverter includes a full-bridge current-type high-frequency inverter, a high-frequency transformer T, a three-phase matrix converter, an output CL filter and a load, which are connected in sequence; 所述全桥电流型高频逆变器包括直流电压源E、储能电感L、可控开关管S1、可控开关管S2、可控开关管S3以及可控开关管S4；所述三相矩阵式变换器包括可控开关管S1a、可控开关管S4b、可控开关管S4a、可控开关管S1b、可控开关管S3a、可控开关管S6b、可控开关管S6a、可控开关管S3b、可控开关管S5a、可控开关管S2b、可控开关管S2a以及可控开关管S5b；所述输出CL型滤波器包括第一电容Cf1、第二电容Cf2、第三电容Cf3、第一电感Lf1、第二电感Lf2以及第三电感Lf3；所述负载包括第一负载R1、第二负载R2以及第三负载R3；The full-bridge current-mode high-frequency inverter includes a DC voltage source E, an energy storage inductor L, a controllable switch tube S 1 , a controllable switch tube S 2 , a controllable switch tube S 3 and a controllable switch tube S 4 ; The three-phase matrix converter includes a controllable switch tube S 1a , a controllable switch tube S 4b , a controllable switch tube S 4a , a controllable switch tube S 1b , a controllable switch tube S 3a , and a controllable switch tube S 6b , controllable switch tube S6a, controllable switch tube S3b, controllable switch tube S5a, controllable switch tube S2b, controllable switch tube S2a and controllable switch tube S5b ; the output CL filter Including a first capacitor C f1 , a second capacitor C f2 , a third capacitor C f3 , a first inductance L f1 , a second inductance L f2 and a third inductance L f3 ; the load includes a first load R 1 , a second load R 2 and a third load R 3 ; 所述直流电压源E的正极与所述储能电感L的一端相连；所述储能电感L的另一端分别与所述可控开关管S1的集电极、所述可控开关管S3的集电极相连；所述直流电压源E的负极分别与所述可控开关管S2的发射极、所述可控开关管S4的发射极相连；所述可控开关管S1的发射极分别与所述高频变压器T原边的一端、所述可控开关管S2的集电极相连；所述可控开关管S3的发射极分别与所述高频变压器T原边的另一端、所述可控开关管S4的集电极相连；The positive pole of the DC voltage source E is connected to one end of the energy storage inductance L; the other end of the energy storage inductance L is respectively connected to the collector of the controllable switch tube S1 and the controllable switch tube S3 The collector of said DC voltage source E is connected to the emitter of said controllable switch tube S2 and the emitter of said controllable switch tube S4 respectively ; the emitter of said controllable switch tube S1 The poles are respectively connected with one end of the primary side of the high-frequency transformer T and the collector of the controllable switch tube S2 ; the emitter of the controllable switch tube S3 is respectively connected with the other side of the primary side of the high-frequency transformer T. One end is connected to the collector of the controllable switch tube S4 ; 所述高频变压器T副边的一端分别与所述可控开关管S1a的集电极、所述可控开关管S3a的集电极、所述可控开关管S5a的集电极相连；所述变压器T副边的另一端分别与所述可控开关管S1b的集电极、所述可控开关管S3b的集电极、所述可控开关管S5b的集电极相连；所述可控开关管S1a的发射极与所述可控开关管S4b的发射极相连，所述可控开关管S3a的发射极与所述可控开关管S6b的发射极相连，所述可控开关管S5a的发射极与所述可控开关管S2b的发射极相连；所述可控开关管S1b的发射极与所述可控开关管S4a的发射极相连，所述可控开关管S3b的发射极与所述可控开关管S6a的发射极相连，所述可控开关管S5b的发射极与所述可控开关管S2a的发射极相连；所述可控开关管S4a的集电极与所述可控开关管S4b的集电极相连；所述可控开关管S6a的集电极与所述可控开关管S6b的集电极相连；所述可控开关管S2a的集电极与所述可控开关管S2b的集电极相连。One end of the secondary side of the high-frequency transformer T is respectively connected to the collector of the controllable switch tube S1a , the collector of the controllable switch tube S3a , and the collector of the controllable switch tube S5a ; The other end of the secondary side of the transformer T is respectively connected with the collector of the controllable switch tube S1b , the collector of the controllable switch tube S3b, and the collector of the controllable switch tube S5b ; The emitter of the controllable switch S1a is connected to the emitter of the controllable switch S4b , the emitter of the controllable switch S3a is connected to the emitter of the controllable switch S6b , the The emitter of the controllable switch S5a is connected to the emitter of the controllable switch S2b ; the emitter of the controllable switch S1b is connected to the emitter of the controllable switch S4a . The emitter of the controllable switch S3b is connected to the emitter of the controllable switch S6a, and the emitter of the controllable switch S5b is connected to the emitter of the controllable switch S2a ; The collector of the controllable switch S4a is connected to the collector of the controllable switch S4b ; the collector of the controllable switch S6a is connected to the collector of the controllable switch S6b ; The collector of the controllable switch S2a is connected to the collector of the controllable switch S2b . 所述可控开关管S4a的集电极与所述可控开关管S4b的集电极相连后分别与所述第一电容Cf1的一端、所述第一电感Lf1的一端相连；所述可控开关管S6a的集电极与所述可控开关管S6b的集电极相连后分别与所述第二电容Cf2的一端、所述第二电感Lf2的一端相连；所述可控开关管S2a的集电极与所述可控开关管S2b的集电极相连后分别与所述第三电容Cf3的一端、所述第三电感Lf3的一端相连；所述第一电容Cf1的另一端分别与所述第二电容Cf2的另一端、所述第三电容Cf3的另一端相连；The collector of the controllable switch tube S 4a is connected to the collector of the controllable switch tube S 4b and then connected to one end of the first capacitor C f1 and one end of the first inductance L f1 respectively; the The collector of the controllable switch tube S 6a is connected to the collector of the controllable switch tube S 6b and then connected to one end of the second capacitor C f2 and one end of the second inductance L f2 respectively; The collector of the switch tube S 2a is connected to the collector of the controllable switch tube S 2b and is respectively connected to one end of the third capacitor C f3 and one end of the third inductor L f3 ; the first capacitor C The other end of f1 is respectively connected to the other end of the second capacitor C f2 and the other end of the third capacitor C f3 ; 所述第一电感Lf1的另一端与所述第一负载R1的一端相连；所述第二电感Lf2的另一端与所述第二负载R2的一端相连；所述第三电感Lf3的另一端与所述第三负载R3的一端相连；所述第一负载R1的另一端分别与所述第二负载R2的另一端、所述第三负载R3的另一端相连；The other end of the first inductor L f1 is connected to one end of the first load R 1 ; the other end of the second inductor L f2 is connected to one end of the second load R 2 ; the third inductor L The other end of f3 is connected to one end of the third load R3 ; the other end of the first load R1 is connected to the other end of the second load R2 and the other end of the third load R3 respectively ; 所述脉宽调制方法包括：The pulse width modulation method includes: 高频变压器T原边的全桥电流型高频逆变器在高频方波驱动下将直流电流变换为高频方波电流；The full-bridge current-type high-frequency inverter on the primary side of the high-frequency transformer T converts the DC current into a high-frequency square-wave current under the drive of a high-frequency square wave; 三相矩阵式变换器将所述高频变压器T副边输出的所述高频方波电流转换成低频脉动电流；The three-phase matrix converter converts the high-frequency square wave current output by the secondary side of the high-frequency transformer T into a low-frequency pulsating current; 输出CL滤波器将所述低频脉动电流转换为三相工频正弦电流给负载供电。The output CL filter converts the low-frequency pulsating current into a three-phase power-frequency sinusoidal current to supply power to the load. 2.根据权利要求1所述的脉宽调制方法，其特征在于，所述全桥电流型高频逆变器在高频方波驱动下将直流电流变换为高频方波电流，具体包括：2. The pulse width modulation method according to claim 1, wherein the full-bridge current-type high-frequency inverter converts the direct current into a high-frequency square-wave current under the high-frequency square-wave drive, specifically comprising: 采用一对具有重叠导通时间的占空比大于50％的高频方波驱动所述高频变压器T原边的所述全桥电流型高频逆变器，令所述可控开关管S1、所述可控开关管S4所在的第一桥臂和所述可控开关管S2、所述可控开关管S3所在的第二桥臂交替导通，将直流电压源E输出的直流电流变换为高频方波电流。A pair of high-frequency square waves with overlapping on-time duty ratios greater than 50% are used to drive the full-bridge current-mode high-frequency inverter on the primary side of the high-frequency transformer T, so that the controllable switch transistor S is 1. The first bridge arm where the controllable switch transistor S4 is located, the second bridge arm where the controllable switch transistor S2 and the controllable switch transistor S3 are located are alternately turned on , and the DC voltage source E is output The DC current is converted into a high frequency square wave current. 3.根据权利要求2所述的脉宽调制方法，其特征在于，所述可控开关管S1、所述可控开关管S4所在的第一桥臂和所述可控开关管S2、所述可控开关管S3所在的第二桥臂的重叠导通角度为θ，其取值范围为0°＜θ＜180°。3 . The pulse width modulation method according to claim 2 , wherein the controllable switch tube S 1 , the first bridge arm where the controllable switch tube S 4 is located, and the controllable switch tube S 2 . . The overlapping conduction angle of the second bridge arm where the controllable switch tube S3 is located is θ, and its value range is 0°<θ<180°. 4.根据权利要求3所述的脉宽调制方法，其特征在于，在一个开关周期Ts时间内，所述可控开关管S1、所述可控开关管S4所在的第一桥臂和所述可控开关管S2、所述可控开关管S3所在的第二桥臂的共同导通时间为Tcom＝Ts*2θ/360°。4 . The pulse width modulation method according to claim 3 , wherein, within one switching period T s , the first bridge arm where the controllable switch tube S 1 and the controllable switch tube S 4 are located. 5 . The common conduction time with the controllable switch tube S 2 and the second bridge arm where the controllable switch tube S 3 is located is T com =T s *2θ/360°. 5.根据权利要求1所述的脉宽调制方法，其特征在于，所述三相矩阵式变换器将高频变压器T输出的所述高频方波电流转换成低频脉动电流，具体包括：5. The pulse width modulation method according to claim 1, wherein the three-phase matrix converter converts the high-frequency square wave current output by the high-frequency transformer T into a low-frequency pulsating current, specifically comprising: 将所述三相矩阵式变换器等效分解为两组普通的电流型三相逆变器，分别为正组逆变器和负组逆变器；所述正组逆变器包括可控开关管S1a、可控开关管S2a、可控开关管S3a、可控开关管S4a、可控开关管S5a以及可控开关管S6a；所述负组逆变器包括可控开关管S1b、可控开关管S2b、可控开关管S3b、可控开关管S4b、可控开关管S5b以及可控开关管S6b；The three-phase matrix converter is equivalently decomposed into two groups of ordinary current-mode three-phase inverters, which are respectively positive group inverters and negative group inverters; the positive group inverters include controllable switches tube S 1a , controllable switch tube S 2a , controllable switch tube S 3a , controllable switch tube S 4a , controllable switch tube S 5a and controllable switch tube S 6a ; the negative group inverter includes a controllable switch tube tube S 1b , controllable switch tube S 2b , controllable switch tube S 3b , controllable switch tube S 4b , controllable switch tube S 5b and controllable switch tube S 6b ; 每组所述电流型三相逆变器采用普通电流型三相逆变器120°调制方法，所述正组逆变器和所述负组逆变器的驱动信号通过与逻辑合成，使所述正组逆变器工作时，所述负组逆变器的可控开关管全部关断；所述负组逆变器工作时，所述正组逆变器的可控开关管全部关断，将所述高频变压器T输出的所述高频方波电流转换成低频脉动电流。Each group of the current-mode three-phase inverters adopts the common current-mode three-phase inverter 120° modulation method, and the driving signals of the positive group inverters and the negative group inverters are synthesized by AND logic, so that all When the positive group inverters are working, all the controllable switching tubes of the negative group inverters are turned off; when the negative group inverters are working, all the controllable switching tubes of the positive group inverters are turned off , converting the high-frequency square wave current output by the high-frequency transformer T into a low-frequency pulsating current.
CN201911155589.1A 2019-11-22 2019-11-22 A pulse width modulation method of a current-mode three-phase high-frequency chain-matrix inverter Active CN110943641B (en)

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