CN109039171A - A kind of high-speed permanent-magnet brushless DC motor control method based on variable turn-on cycle - Google Patents

Abstract

The invention discloses a high-speed permanent magnet brushless DC motor control method based on a variable conduction period. The method realizes the stepless speed regulation of the system by adjusting the conduction period of the control signal, and obtains the required conduction through the double closed-loop control of speed and current. According to the principle of minimum current ripple, the conduction period is divided into one pulse or two symmetrical pulses or two asymmetrical pulses, and then applied to each power tube according to the three-phase six-state conduction law. By dynamically adjusting the optimal commutation angle, the invention can make the high-speed operating current waveform of the motor system completely symmetrical and minimize the pulsation, effectively improve the stator current waveform of the high-speed permanent magnet brushless DC motor, and reduce the eddy current loss of the rotor.

Description

A control method for high-speed permanent magnet brushless DC motor based on variable conduction period

technical field

The invention belongs to the technical field of motor control, and in particular relates to a high-speed permanent magnet brushless DC motor control method based on a variable conduction period.

Background technique

The permanent magnet brushless DC motor is mainly composed of three parts: the motor body, the rotor position sensor and the power inverter circuit. It is a typical mechatronic motor. The structure of the motor body is similar to that of ordinary synchronous motors. The stator generally adopts three-phase windings, and the rotor adopts permanent magnet excitation; the power inverter circuit inverts the DC power supply into AC power according to the conduction sequence output by the rotor position sensor, which is used to power the motor stator. The three-phase winding supplies power in order to interact with the permanent magnetic field of the rotor to generate continuous electromagnetic torque; the power inverter circuit generally adopts a three-phase bridge structure (also called a three-phase bridge inverter), consisting of six power switch tubes , The rotor position sensor detects the position of the permanent magnet rotor, outputs a three-phase position signal, and forms the trigger signal of the power inverter circuit after logic processing and power amplification. At present, Hall integrated circuits are mostly used.

The permanent magnet brushless DC motor has the characteristics of small size, high power density, high efficiency, and simple rotor structure. Wide range of applications, but common brushless motor control strategies are not suitable for high-speed operation. The permanent magnet brushless DC motor generally adopts the three-phase six-state conduction mode, and the speed control generally adopts the PWM (pulse width modulation) mode of adjusting the duty cycle, through the six commutation drive signals of the three-phase power inverter bridge Superimpose high-frequency PWM waves to adjust the voltage applied to the motor windings. When the motor speed is low, the PWM chopping frequency is much higher than the motor operating frequency, which will not cause distortion of the winding phase current waveform. However, when the motor operating frequency and PWM frequency When approached, the winding phase current will be severely distorted. For example, when the running speed of the motor reaches 120,000 revolutions per minute, the three-phase six-state 120° conduction control method is adopted, and the commutation frequency of the motor is 12kHz (take a motor with 1 pair of poles as an example), which is comparable to the commonly used chopping frequency range (10～20kHz) is very close, and the number of chopping in a commutation cycle is only about one. Under this condition, PWM chopping speed regulation will cause serious distortion of the motor phase current waveform, which not only restricts the maximum speed of the motor, but also It will cause problems such as noise and temperature rise. In order to solve this problem, PAM (Pulse Amplitude Modulation) speed regulation method can usually be used, that is, a DC chopper (Buck circuit) is added in the front stage of the three-phase power inverter bridge, but the circuit topology of PAM speed regulation method is complex , the hardware circuit is increased, and when driving a high-power permanent magnet brushless motor, the DC reactor in the Buck circuit will generate additional losses, which will affect the efficiency of the entire drive system.

In addition, under the conventional three-phase six-state conduction mode of the permanent magnet brushless DC motor, the ideal optimal commutation angle is to conduct at 30° after the back EMF zero crossing point, and to turn off at 150°. The conduction period is Fixed 120°. However, during ultra-high-speed operation, the inductance of the motor winding has a great influence. In the ideal commutation mode, the motor current will lag behind the back EMF, resulting in a decrease in the equivalent power factor of the motor, and an increase in the effective value and peak value of the current. The loss and the stress of the inverter increase. Since the current lag angle changes with the change of the motor speed and load, it cannot be solved by fixed advance commutation. With the increase of the commutation lead angle, the electromagnetic torque ripple of the motor will will increase.

Contents of the invention

In view of the above, the present invention provides a high-speed permanent magnet brushless DC motor control method based on variable conduction period, by adjusting the conduction period of the control signal to realize stepless speed regulation of the system, which can make the high-speed operation current waveform of the motor system completely symmetrical And the pulsation is minimal.

A high-speed permanent magnet brushless DC motor control method based on a variable conduction period, comprising the following steps:

(1) Use the Hall sensor to collect and capture the rotor discrete position angle of the motor and calculate the motor speed and the rotor electrical angular velocity, and use the current sensor to collect and obtain the stator current signals of the two phases of the motor BC;

(2) After filtering the motor BC two-phase stator current signal and AD sampling, the DC bus current of the motor is reconstructed;

(3) Calculate the continuous position angle of the rotor of the motor according to the electrical angular velocity of the rotor and the discrete position angle of the rotor;

(4) Calculate the conduction angle of the motor through the double closed-loop control of current and speed;

(5) Determine the switch state of the motor power inverter at the next moment through the corresponding conduction logic according to the continuous position angle and conduction angle of the rotor, and apply switch control accordingly.

Further, the discrete position angle of the rotor in the step (1) is the initial position angle when the rotor enters the current sector.

Further, the specific criteria for reconstructing the DC bus current of the motor in the step (2) are as follows:

If the switch state corresponding to the sector where the motor rotor is currently located is that the upper bridge arm of the power inverter phase A and the lower bridge arm of the B phase are turned on, the DC bus current of the motor is -i _b ;

If the switch state corresponding to the sector where the motor rotor is currently located is that the upper bridge arm of the power inverter phase A and the lower bridge arm of the phase C of the power inverter are turned on, the DC bus current of the motor is _-ic ;

If the switch state corresponding to the sector where the motor rotor is currently located is that the upper bridge arm of the power inverter phase B and the lower bridge arm of the phase A of the power inverter are turned on, then the DC bus current of the motor is i _b ;

If the switch state corresponding to the sector where the motor rotor is currently located is that the upper bridge arm of the B-phase and the lower bridge arm of the C-phase of the power inverter are turned on, the DC bus current of the motor is i _b ;

If the switch state corresponding to the sector where the motor rotor is currently located is that the upper bridge arm of phase C and the lower bridge arm of phase A of the power inverter are turned on, then the DC bus current of the motor is i _c ;

If the switch state corresponding to the sector where the motor rotor is currently located is that the upper bridge arm of the C-phase and the lower bridge arm of the B-phase of the power inverter are turned on, the DC bus current of the motor is i _c ;

Among them: i _b and i _c are the instantaneous values of the stator currents of the two phases of the current motor BC respectively.

Further, in the step (3), the rotor continuous position angle of the motor is calculated by the following formula:

Among them: θ(t) is the continuous position angle of the rotor of the motor at time t, ω is the electrical angular velocity of the rotor of the motor in the previous electrical cycle, θ _i is the discrete position angle of the rotor of the motor, t represents the current moment, and t ₁ represents that the rotor enters the current position The initial moment of the sector.

Further, the specific implementation process of the step (4) is as follows: firstly, the current motor speed is subtracted from the given reference speed, and a current reference value is generated after performing PI adjustment on the speed error obtained by the subtraction; then, the The current reference value is subtracted from the reconstructed DC bus current of the motor, and then the conduction angle of the motor is generated after PI adjustment is performed on the subtracted current error.

Further, the conduction logic in the step (5) is as follows:

When 30°≤θ≤150°-θ ₁ and 150°+(x ₁ -1)θ ₁ ≤θ≤150°+(x ₁ +x ₂ -1)θ ₁ , the upper arm of phase A of the power inverter open;

When 210°≤θ≤330°-θ ₁ and 330°+(x ₁ -1)θ ₁ ≤θ≤330°+(x ₁ +x ₂ -1)θ ₁ , the lower arm of the phase A of the power inverter open;

When 330°≤θ≤360° and 0°≤θ≤90°-θ ₁ and 90°+(x ₁ -1)θ ₁ ≤θ≤90°+(x ₁ +x ₂ -1)θ ₁ , power The lower bridge arm of phase B of the inverter is turned on;

When 150°≤θ≤270°-θ ₁ and 270°+(x ₁ -1)θ ₁ ≤θ≤270°+(x ₁ +x ₂ -1)θ ₁ , the upper bridge arm of phase B of the power inverter open;

When 270°≤θ≤390°-θ ₁ and (x ₁ -1)θ ₁ +30°≤θ≤(x ₁ +x ₂ -1)θ ₁ +30°, the C-phase upper arm of the power inverter open;

When 90°≤θ≤210°-θ ₁ and 210°+(x ₁ -1)θ ₁ ≤θ≤210°+(x ₁ +x ₂ -1)θ ₁ , the C-phase lower arm of the power inverter open;

Among them: θ is the rotor continuous position angle of the motor, θ ₁ is the conduction angle of the motor, x ₁ and x ₂ are the conduction coefficients (related to the motor parameters, these two conduction coefficients should be adjusted appropriately for different motors to minimize the stator current ripple).

The motor control method of the present invention realizes the stepless speed regulation of the system by adjusting the conduction period of the control signal, obtains the required conduction period through the double closed-loop control of speed and current, and divides the conduction period into one pulse or two according to the principle of minimum current ripple. A symmetrical pulse or two asymmetrical pulses are applied to each power tube according to the three-phase six-state conduction law. By dynamically adjusting the optimal commutation angle, the invention can make the high-speed operating current waveform of the motor system completely symmetrical and minimize the pulsation, effectively improve the stator current waveform of the high-speed permanent magnet brushless DC motor, and reduce the eddy current loss of the rotor.

Description of drawings

Figure 1 is a system block diagram of the traditional PWM control method.

FIG. 2 is a system block diagram of the present invention based on a new control mode of variable conduction period.

Figure 3(a) is the motor winding current waveform diagram under the traditional PWM control mode.

Fig. 3(b) is a waveform diagram of the motor winding current under the new control method based on the variable conduction period of the present invention.

FIG. 4 is a structural block diagram of the DSP-based variable conduction period motor control system of the present invention.

Fig. 5(a) is a schematic diagram of the principle of the single-pulse control method based on the variable conduction period.

Fig. 5(b) is a schematic diagram of the principle of the symmetrical double-pulse control method based on the variable conduction period.

Fig. 5(c) is a schematic diagram of the principle of an asymmetrical double-pulse control method based on a variable conduction period.

FIG. 6 is a schematic diagram of the conduction logic of six power transistors under the asymmetrical double pulse control mode.

Detailed ways

In order to describe the present invention more specifically, the technical solutions of the present invention will be described in detail below in conjunction with the accompanying drawings and specific embodiments.

Figure 1 is a block diagram of a traditional permanent magnet brushless DC motor speed and current double-closed-loop PWM control system. The control strategy adopts a three-phase six-state 120° conduction mode, and each state conducts two power switch tubes. The conduction of each power tube The pass cycle is 120° electrical angle, and the Hall position sensor of the motor provides the position information required for state switching. According to the principles of electromechanics, in order to realize the speed regulation of the brushless DC motor, the voltage applied to both ends of the motor winding must be changed. The traditional control method is to generate a PWM wave with a fixed chopping cycle and an adjustable duty cycle and the 120° conduction control logic signal of the power tube. The PWM duty cycle is determined by the output of the double closed-loop regulator. By changing the PWM The duty cycle changes the actual conduction time of each power tube to realize stepless speed regulation.

The system block diagram of the new control method based on the variable conduction period of the present invention is shown in Figure 2. The regulation of the voltage at both ends of the motor winding is realized by directly changing the conduction time of each power tube. The difference between the given speed and the actual feedback speed The difference passes through the output of the speed PI regulator as the given value of the current loop, and the difference between it and the feedback value of the motor current passes through the current PI regulator to output the conduction period required by each power tube. According to the principle of minimum current ripple, the conduction period The on-cycle is divided into one pulse or two symmetrical pulses or two asymmetrical pulses, and then applied to each power tube according to the three-phase six-state commutation law, and then the commutation angle is dynamically adjusted. At this time, the duration of each operating state is different. Even if the electrical angle is 60°, the conduction period of each power tube is no longer fixed at 120° electrical angle, but stabilized at a variable angle according to the operating conditions.

The traditional PWM control method has a good control effect when the motor speed is low and the operating fundamental frequency is much lower than the chopping frequency, but as the motor speed increases, especially when the operating fundamental frequency is close to the chopping frequency , the number of PWMs in the 120° conduction period of each power tube is only a limited number and is no longer an integer multiple, resulting in serious distortion of the winding current waveform. Figure 3(a) shows the motor speed under the traditional PWM control mode The winding current waveform at 120000r/min.

Based on the traditional three-phase bridge inverter, the present invention does not increase the hardware circuit, and controls the system speed by adjusting the conduction period of the control signal and the optimal commutation angle, so that the winding current waveform of the motor system is completely symmetrical when the motor system is running at high speed And the pulsation is the smallest. Figure 3(b) shows the winding current waveform when the motor speed is 120000r/min under the new control mode of the present invention.

It can be seen from the comparison that at the same rotational speed of 120000r/min, the winding current waveform of the new control mode of the present invention is completely symmetrical, which effectively reduces the distortion of the phase current, thereby effectively reducing the rotor eddy current loss of the high-speed brushless DC motor.

The control object of this embodiment is a permanent magnet brushless DC motor, and the system includes a DC power supply, a power drive board (three-phase inverter) and a control board (the control chip used in this example is TMS320F28035DSP of the Piccolo series of TI Company), in which the DC The power supply converts the direct current into alternating current through the three-phase inverter and then supplies the three-phase winding of the motor stator. The control board is responsible for processing the collected data (motor rotor position, stator current, speed and other information) and sending corresponding control signals to the three-phase inverter The driver of the inverter, and the power driver board is used to drive the motor to rotate.

As shown in Figure 4, the control system of the permanent magnet brushless DC motor based on the variable conduction period of the present invention includes a Hall sensor, a Hall signal capture unit, a current sensor, a current filter circuit, a current sampling unit, a speed calculation unit, The rotor continuous position calculation unit, double closed-loop adjustment unit and conduction logic unit, Hall signal capture unit, current sampling unit, speed calculation unit, rotor continuous position calculation unit and conduction logic unit are all realized by DSP software programming, among which:

Hall sensor The three Hall sensors installed on the motor are used to collect six discrete positions of the motor rotor; the DSP captures the pulse signals of the three Hall sensors (a total of six Hall states) and uses the Hall signal capture unit The internal counter records the time interval Δt between the two rising edges of a Hall signal; the position of the six Hall state jumps corresponds to the six discrete positions of the motor rotor θ _i (i=1,2,3, 4,5,6), the time interval recorded by the counter is the time corresponding to one electrical cycle of the rotor rotation.

The speed calculation unit uses the time interval Δt obtained by the capture unit, that is, the rotor electrical angular velocity and rotor speed can be calculated by the following formula:

Among them: w is the electrical angular velocity of the rotor, in rad/s; Δt is the duration of the rotor in the last electrical cycle, in s; n is the rotor speed, in r/min; p is the number of pole pairs of the motor.

Since this embodiment needs to control the stator winding to be turned on or off at a suitable rotor position (no longer the rotor position where the Hall signal jumps) according to the conduction angle of the double closed-loop output, so as to achieve the goal of minimizing the stator current ripple , so the continuous position of the motor rotor must be obtained; the continuous rotor position calculation unit in this example uses the six discrete rotor positions collected by the Hall sensor to obtain the continuous position of the rotor through linear interpolation, and the rotor position is obtained by integrating the electrical angular velocity of the rotor. The formula is as follows:

Where: ω is the electrical angular velocity of the motor rotor calculated in the previous electrical cycle, θi is the initial position angle when the rotor enters sector _{i, t i is} the moment when the rotor enters sector i, and t is the moment when the rotor is in the current sector.

The current sensor used in this example is the ACS712 series product of Allegro Company, and the current sensor with a range of 5A is selected according to the rated current of the motor, and the stator current is measured by two current sensors connected in series at the B and C phases of the motor stator winding.

The current sampling unit uses the AD module in the DSP to sample the two-phase current values of the stator winding B and C of the motor and convert them into digital quantities for internal calculation of the DSP.

The current reconstruction unit uses the stator two-phase winding current value obtained by AD sampling to reconstruct the bus current value for the feedback of the current loop. The principle of reconstruction is as follows:

①When the DSP sends out a control signal to turn on the upper bridge power tube of phase A and the lower bridge power tube of phase B (that is, the stator winding is turned on A+B-), at this time the bus current i=-i _b (i _b is the stator Winding B-phase current value).

②When the DSP sends a control signal to turn on the power tube of the upper bridge power tube of phase A and the lower bridge power tube of phase _C (that is, the stator winding is turned on A+C-), at this time the bus current i= _-ic (ic is the stator Winding phase C current value).

③When the DSP sends a control signal to turn on the B-phase upper bridge power tube and the A-phase lower bridge power tube (that is, the stator winding is turned on B+A-), or when the DSP sends a control signal to make the B-phase upper bridge power tube and the When the C-phase lower bridge power tube is turned on (that is, the stator winding is turned on B+C-), the bus current i=i _b (i _b is the B-phase current value of the stator winding).

④When the DSP sends a control signal to turn on the C-phase upper bridge power tube and the A-phase lower bridge power tube (that is, the stator winding is turned on C+A-), or when the DSP sends a control signal to make the C-phase upper bridge power tube and the When the B-phase lower bridge power tube is turned on (that is, the stator winding is turned on C+B-), at this time the bus current i=ic ( _ic is the current value of the _C -phase of the stator winding).

The double-closed-loop adjustment unit realizes the stepless speed regulation of the motor through the double-closed-loop control of the speed and current. The output of the double-closed-loop control is the conduction angle θ ₁ to be controlled: first, subtract the current motor speed from the given reference speed , to generate a current reference value after performing PI adjustment on the rotational speed error obtained by subtraction; then, subtract the reconstructed DC bus current of the motor from the current reference value, and then perform PI adjustment on the current error obtained by subtraction to generate the motor’s conduction angle θ ₁ .

The control strategy of the present invention includes three implementation modes of single pulse, symmetrical double pulse and asymmetrical double pulse, and realizes stepless speed regulation of the system by adjusting the conduction period of the control signal. Among them, the single-pulse control method realizes system speed regulation by changing the conduction period of the control signal, the pulse width θ of the control signal in Figure 5(a), and the control signal pulse with a width of θ can be within 30° ~150° flat top range to obtain the best commutation angle; in the symmetrical double-pulse control mode, the control signal pulse with a width of θ can be divided into two pulses with equal width (width is f1(θ)), as shown in Figure 5 As shown in (b), and the position of the two pulses can move within the range of 30°~150° flat top of the back EMF waveform e, the interval f2(θ) between the two pulses is also variable to obtain the best commutation angle ; In the asymmetric double-pulse control mode, the control signal pulse with a width of θ can be divided into two pulses with unequal widths (the widths are f1(θ) and f3(θ) respectively), as shown in Figure 5(c), and the two The position of a pulse can move within the range of 30°-150° flat top of the back EMF waveform e, and the interval f2(θ) between two pulses is also variable to obtain the best commutation angle.

In this embodiment, the conduction logic unit determines the switch state of the motor power inverter at the next moment through the corresponding conduction logic according to the continuous position angle of the rotor and the conduction angle, and applies switch control accordingly; the six power tubes of the inverter The conduction logic is shown in Figure 6, where the abscissa is the continuous position θ of the motor rotor, and the switching of the six power transistors is subject to the conduction angle θ ₁ of the double closed-loop output in the control system in Figure 4, where x1 and x2 are the conduction coefficients , is related to the motor parameters. For different motors, these two coefficients should be properly adjusted to minimize the stator current ripple.

The specific implementation steps of this example are as follows:

(1) Motor start: use the 6 discrete position signals of the rotor collected by the Hall sensor to obtain the continuous position of the rotor through linear interpolation, and give a fixed conduction period to make the motor open-loop.

(2) Adding the current loop: given the current reference value, the AD module of the DSP samples the B-phase and C-phase currents of the motor stator windings, and performs current reconstruction through software programming to obtain the motor bus current value for current feedback. The current loop The output is the conduction period to be adjusted; in the area of the back EMF flat-top wave, adjust the two pulses and the width between the two pulses, adjust the proportional integral parameter of the current regulator, so that the actual current of the motor stator is within the current reference value Nearby fluctuations, to achieve closed-loop control of the motor current.

(3) Join the speed loop: given the speed reference value, the value obtained through the above speed calculation formula is used as the speed feedback, the output of the speed loop is used as the given value of the current loop, adjust the proportional integral parameter of the speed loop, and properly adjust the current loop The proportional-integral parameters of the motor make the motor run stably at a given speed and realize the double-closed-loop control of motor speed and current.

The above description of the embodiments is for those of ordinary skill in the art to understand and apply the present invention. It is obvious that those skilled in the art can easily make various modifications to the above-mentioned embodiments, and apply the general principles described here to other embodiments without creative efforts. Therefore, the present invention is not limited to the above embodiments, and improvements and modifications made by those skilled in the art according to the disclosure of the present invention should fall within the protection scope of the present invention.

Claims (6)

1. A high-speed permanent magnet brushless DC motor control method based on variable conduction period, comprising the steps: (1) Use the Hall sensor to collect and capture the rotor discrete position angle of the motor and calculate the motor speed and the rotor electrical angular velocity, and use the current sensor to collect and obtain the stator current signals of the two phases of the motor BC; (2) After filtering the motor BC two-phase stator current signal and AD sampling, the DC bus current of the motor is reconstructed; (3) Calculate the continuous position angle of the rotor of the motor according to the electrical angular velocity of the rotor and the discrete position angle of the rotor; (4) Calculate the conduction angle of the motor through the double closed-loop control of current and speed; (5) Determine the switch state of the motor power inverter at the next moment through the corresponding conduction logic according to the continuous position angle and conduction angle of the rotor, and apply switch control accordingly. 2. The high-speed permanent magnet brushless DC motor control method according to claim 1, characterized in that: the rotor discrete position angle in the step (1) is the initial position angle when the rotor enters the current sector. 3. the high-speed permanent-magnet brushless DC motor control method according to claim 1, is characterized in that: the concrete standard of reconstruction motor DC bus current in the described step (2) is as follows: If the switch state corresponding to the sector where the motor rotor is currently located is that the upper bridge arm of the power inverter phase A and the lower bridge arm of the B phase are turned on, the DC bus current of the motor is -i _b ; If the switch state corresponding to the sector where the motor rotor is currently located is that the upper bridge arm of the power inverter phase A and the lower bridge arm of the phase C of the power inverter are turned on, the DC bus current of the motor is _-ic ; If the switch state corresponding to the sector where the motor rotor is currently located is that the upper bridge arm of the power inverter phase B and the lower bridge arm of the phase A of the power inverter are turned on, then the DC bus current of the motor is i _b ; If the switch state corresponding to the sector where the motor rotor is currently located is that the upper bridge arm of the B-phase and the lower bridge arm of the C-phase of the power inverter are turned on, the DC bus current of the motor is i _b ; If the switch state corresponding to the sector where the motor rotor is currently located is that the upper bridge arm of phase C and the lower bridge arm of phase A of the power inverter are turned on, then the DC bus current of the motor is i _c ; If the switch state corresponding to the sector where the motor rotor is currently located is that the upper bridge arm of the C-phase and the lower bridge arm of the B-phase of the power inverter are turned on, the DC bus current of the motor is i _c ; Among them: i _b and i _c are the instantaneous values of the stator currents of the two phases of the current motor BC respectively. 4. the high-speed permanent magnet brushless DC motor control method according to claim 1, is characterized in that: in described step (3), calculate the rotor continuous position angle of motor by following formula: Among them: θ(t) is the continuous position angle of the rotor of the motor at time t, ω is the electrical angular velocity of the rotor of the motor in the previous electrical cycle, θ _i is the discrete position angle of the rotor of the motor, t represents the current moment, and t ₁ represents that the rotor enters the current position The initial moment of the sector. 5. the high-speed permanent-magnet brushless DC motor control method according to claim 1 is characterized in that: the concrete realization process of described step (4) is: first, make given reference rotating speed subtract current motor rotating speed, to The speed error obtained by subtraction is PI-adjusted to generate a current reference value; then, the current reference value is subtracted from the reconstructed DC bus current of the motor, and then the current error obtained by subtraction is PI-adjusted to generate the conduction of the motor horn. 6. The high-speed permanent magnet brushless DC motor control method according to claim 1, characterized in that: the conduction logic in the step (5) is as follows: When 30°≤θ≤150°-θ ₁ and 150°+(x ₁ -1)θ ₁ ≤θ≤150°+(x ₁ +x ₂ -1)θ ₁ , the upper arm of phase A of the power inverter open; When 210°≤θ≤330°-θ ₁ and 330°+(x ₁ -1)θ ₁ ≤θ≤330°+(x ₁ +x ₂ -1)θ ₁ , the lower arm of the phase A of the power inverter open; When 330°≤θ≤360° and 0°≤θ≤90°-θ ₁ and 90°+(x ₁ -1)θ ₁ ≤θ≤90°+(x ₁ +x ₂ -1)θ ₁ , power The lower bridge arm of phase B of the inverter is turned on; When 150°≤θ≤270°-θ ₁ and 270°+(x ₁ -1)θ ₁ ≤θ≤270°+(x ₁ +x ₂ -1)θ ₁ , the upper bridge arm of phase B of the power inverter open; When 270°≤θ≤390°-θ ₁ and (x ₁ -1)θ ₁ +30°≤θ≤(x ₁ +x ₂ -1)θ ₁ +30°, the C-phase upper arm of the power inverter open; When 90°≤θ≤210°-θ ₁ and 210°+(x ₁ -1)θ ₁ ≤θ≤210°+(x ₁ +x ₂ -1)θ ₁ , the C-phase lower arm of the power inverter open; Among them: θ is the rotor continuous position angle of the motor, θ ₁ is the conduction angle of the motor, x ₁ and x ₂ are conduction coefficients.