CN101005475A

CN101005475A - Method and system for synchronizing time and frequency in orthogonal frequency division multiplex communication

Info

Publication number: CN101005475A
Application number: CN 200610157618
Authority: CN
Inventors: 刘田; 唐友喜; 邵士海; 王吉滨; 李云岗
Original assignee: Huawei Technologies Co Ltd; University of Electronic Science and Technology of China
Current assignee: Huawei Technologies Co Ltd; University of Electronic Science and Technology of China
Priority date: 2006-12-14
Filing date: 2006-12-14
Publication date: 2007-07-25

Abstract

The invention relates to a method and system for synchronizing time and frequency in Orthogonal Frequency Division Multiplexing communication. Wherein, at the transmitting end, the carrier-modulated original information is serial-parallel-converted into branch data streams with the same number of antennas, and mapped together with the frequency-domain pilot sequence to the corresponding data position of the predetermined frequency-domain pilot pattern, Generate the data information of each antenna and perform OFDM modulation and radio frequency transmission on it; at the receiving end, the time domain output sequence is generated by the radio frequency reception of each antenna, and is correlated with the reference sequence to construct the synchronization objective function between each pair of antennas and realize the time and frequency synchronization. The method is particularly suitable for multiple-input multiple-output systems. The invention provides a method and system for OFDM time and frequency synchronization, which effectively solves the problems in the prior art that the system overhead is large, the effective data bearer is much occupied, it is not suitable for MIMO systems, and the requirements for channel characteristics are strict.

Description

Method and system for time and frequency synchronization in OFDM communication

技术领域technical field

本发明涉及多路复用通信，更确切地说，涉及一种正交频分复用(OFDM)通信中时间和频率同步的方法及系统。The present invention relates to multiplexing communication, more specifically, relates to a method and system for synchronizing time and frequency in Orthogonal Frequency Division Multiplexing (OFDM) communication.

背景技术Background technique

MIMO(Multi-Input Multi-Output，多输入多输出)和OFDM(OrthogonalFrequency Division Multiplexing，正交频分复用)技术为现代通信技术的两项重要内容。OFDM技术作为一种高效的多载波传输技术，其优点在于具备高的频谱利用率和抗频率选择性衰落能力强。MIMO技术则能够提供比SISO(Single-Input Single-Output，单输入单输出)技术更大的信道容量，是高速数据传输的一种潜在方法。结合MIMO和OFDM的MIMO-OFDM方法可以实现在宽带无线通信系统中提供高数据传输速率业务的目的。由于OFDM系统对时间和频率同步的要求高，特别是对频率同步的要求非常严格，较小的频偏就会导致较大的性能损失，所以时间和频率同步在MIMO-OFDM中至关重要。MIMO (Multi-Input Multi-Output) and OFDM (Orthogonal Frequency Division Multiplexing) technologies are two important contents of modern communication technology. As an efficient multi-carrier transmission technology, OFDM technology has the advantages of high spectrum utilization rate and strong ability to resist frequency selective fading. MIMO technology can provide a larger channel capacity than SISO (Single-Input Single-Output) technology, and is a potential method for high-speed data transmission. The MIMO-OFDM method combining MIMO and OFDM can realize the purpose of providing high data transmission rate services in broadband wireless communication systems. Since the OFDM system has high requirements on time and frequency synchronization, especially the very strict requirements on frequency synchronization, a small frequency offset will lead to a large performance loss, so time and frequency synchronization is very important in MIMO-OFDM.

OFDM时间和频率同步方法分为数据辅助的同步方法和盲同步(即无需数据辅助的同步)方法，本发明涉及数据辅助的同步方法。数据辅助的同步方法需要在信号中插入时域训练序列或频域导频，因此与盲同步方法相比降低了数据传输效率，但数据辅助方法的估计精度高且计算复杂度较低。OFDM time and frequency synchronization methods are divided into data-assisted synchronization methods and blind synchronization (that is, synchronization without data assistance) methods. The present invention relates to data-assisted synchronization methods. Data-assisted synchronization methods need to insert time-domain training sequences or frequency-domain pilots into the signal, thus reducing data transmission efficiency compared with blind synchronization methods, but data-assisted methods have high estimation accuracy and low computational complexity.

目前已有的OFDM数据辅助的同步方法包括利用一个或多个OFDM符号作训练序列进行时间和频率同步，以及利用插入导频的OFDM符号进行频率同步。其中用插入导频的OFDM符号进行频率同步为在相邻OFDM符号的相同子载波位置上插入同样的导频序列，在精确的时间同步已完成情况下，利用相同导频序列的差分相关来估计载波频偏。Currently existing OFDM data-assisted synchronization methods include using one or more OFDM symbols as training sequences to perform time and frequency synchronization, and using pilot-inserted OFDM symbols to perform frequency synchronization. The OFDM symbols with inserted pilots are used for frequency synchronization to insert the same pilot sequence at the same subcarrier position of adjacent OFDM symbols. When the precise time synchronization has been completed, the differential correlation of the same pilot sequence is used to estimate Carrier frequency offset.

设定训练序列的位置和内容可以被用来获得更好的同步性能，如一种“三明治”方式在多个数据符号两端放置训练序列，由于训练序列间隔较远，利用其做频率同步时精度较高；而多个CAZAC(Constant Amplitude Zero Auto-Correlation，恒模自相关)序列的重复排列则可以被用作训练序列的内容来进行频率同步。Setting the position and content of the training sequence can be used to obtain better synchronization performance. For example, a "sandwich" method is used to place the training sequence at both ends of multiple data symbols. Since the training sequence is far apart, the accuracy of frequency synchronization can be improved by using it. Higher; while the repeated arrangement of multiple CAZAC (Constant Amplitude Zero Auto-Correlation, constant modulus autocorrelation) sequences can be used as the content of the training sequence for frequency synchronization.

目前对于MIMO-OFDM时间和频率的同步的研究还处于起步阶段，主要通过以下两种方式进行：一是各天线发送多个训练序列的方式，此方式建立在信号从所有发射天线到所有接收天线时间延迟相同，频率偏移也相同的假设基础之上。通过对接收信号序列进行DFT(Discrete Fourier Transform，离散傅立叶变换)后与本地时域序列的DFT形式作相关完成频率粗同步，通过接收信号作延迟相关取模实现时间粗同步，作延迟相关取相位实现频率精同步，时间精同步通过接收信号与本地序列作相关完成；二是各天线发送时域上错开的训练序列的方式，此方式的训练序列仍由多个OFDM符号构成，时间粗同步通过接收信号作延迟相关取模并累加实现，时间精同步通过接收信号与本地序列作相关取模实现，频率同步利用接收信号延迟作相关所得的相位信息完成。At present, the research on the synchronization of MIMO-OFDM time and frequency is still in its infancy, and it is mainly carried out in the following two ways: one is the way that each antenna sends multiple training sequences, which is based on the fact that the signal is transmitted from all transmitting antennas to all receiving antennas It is based on the assumption that the time delay is the same and the frequency offset is the same. After performing DFT (Discrete Fourier Transform) on the received signal sequence and correlating with the DFT form of the local time domain sequence to complete the frequency coarse synchronization, the received signal is used for delay correlation modulo to achieve time coarse synchronization, and for delay correlation phase acquisition Realize frequency fine synchronization, and time fine synchronization is completed by correlating the received signal with the local sequence; the second is the way that each antenna sends a training sequence staggered in the time domain. The training sequence of this method is still composed of multiple OFDM symbols, and the time coarse synchronization is passed The received signal is implemented by delay correlation modulo acquisition and accumulation, the time fine synchronization is achieved by correlation modulo acquisition between the received signal and the local sequence, and the frequency synchronization is completed by using the phase information obtained by the delay correlation of the received signal.

这里要说明的是，在OFDM时间和频率同步方法中，所述本地序列和所述训练序列对应相同的数学实体，本地序列和训练序列在时域和频域也分别有相同的表达形式。在本领域一般技术文献中，所述数学实体在发送端被称为训练序列；在接收端则被称为本地序列，以区分于接收信号序列。本公开中亦采用所述表示方式。It should be noted here that in the OFDM time and frequency synchronization method, the local sequence and the training sequence correspond to the same mathematical entity, and the local sequence and the training sequence also have the same expression form in the time domain and the frequency domain respectively. In the general technical literature in this field, the mathematical entity is called a training sequence at the sending end; it is called a local sequence at the receiving end to distinguish it from the received signal sequence. This representation is also used in the present disclosure.

现有技术的两种时间和频率同步方法如下：The two methods of time and frequency synchronization in the prior art are as follows:

第一种是叠加PN(Pseudo Noise伪噪声)序列的方法。用于同步的训练序列的构成如图1所示，其中N为OFDM子载波数，PN序列的周期为k，L为N/K的整数部分。从图中可见，最后一个PN序列可能不完整。多个PN序列重复放置的目的在于利用PN序列的周期相关特性在接收端将接收信号与本地序列进行相关处理，因此可以选择具有良好自相关特性的PN序列，如M序列或Gold序列。The first is the method of superimposing PN (Pseudo Noise) sequences. The composition of the training sequence used for synchronization is shown in Figure 1, where N is the number of OFDM subcarriers, the period of the PN sequence is k, and L is the integer part of N/K. It can be seen from the figure that the last PN sequence may not be complete. The purpose of repeated placement of multiple PN sequences is to use the periodic correlation characteristics of PN sequences to correlate the received signal with the local sequence at the receiving end, so PN sequences with good autocorrelation characteristics, such as M sequences or Gold sequences, can be selected.

在接收端将接收序列与本地序列作相关，得到如下相关函数：Correlate the received sequence with the local sequence at the receiving end to obtain the following correlation function:

$γ γ ((k k,, a a)) = = {Σ Σ}_{l l = = 00}^{L L - - P P - - 11} [[(({Σ Σ}_{m m = = 00}^{k k - - 11} {c c}^{* *} ((k k - - m m - - lK k - - a a)) r r ((k k - - m m - - lK k))))$

$\cdot &Center Dot; (({Σ Σ}_{m m = = 00}^{K K - - 11} {c c}^{* *} ((k k - - m m - - ((l l + + P P)) K K - - a a)) r r ((k k - - m m - - ((l l + + P P)) K K)) {))}^{* *}]] - - - - - - ((11))$

公式(1)中c(k)是由重复PN序列构成的训练序列，r(k)为接收信号经A/D(模/数)变换后得到的序列。P为正整数，用于调整频率偏移估计范围和精度。取时间同步的目标函数为|γ(k，a)|。在接收端采用一个长为KL的滑动窗口(Sliding Window)，将滑动窗内的数据按公式(1)进行处理，a表示滑动窗口相对于精确同步点的距离，则a＝0表示时间同步完成。例如：设各参数值为N＝128，K＝15，P＝1，则训练序列内包含完整的PN序列共8个。当时间延迟为128个采样间隔时，随着滑动窗口的移动，当本地序列c(k)对齐2个PN序列时即有小的尖峰输出，当本地序列c(k)对齐8个PN序列时有最大的尖峰输出，而当PN序列与本地序列不对齐时输出值很小，这样很容易找到时间同步点。这种时间同步方法可以一次完成粗时间同步和精时间同步，捕获概率大，虚警概率和漏报概率小。In formula (1), c(k) is a training sequence composed of repeated PN sequences, and r(k) is a sequence obtained after A/D (analog/digital) conversion of the received signal. P is a positive integer and is used to adjust the frequency offset estimation range and precision. The objective function of time synchronization is |γ(k, a)|. A sliding window (Sliding Window) with a length of KL is used at the receiving end, and the data in the sliding window is processed according to formula (1). a indicates the distance of the sliding window from the precise synchronization point, and a=0 indicates that the time synchronization is completed . For example, assuming that each parameter value is N=128, K=15, and P=1, then the training sequence contains a total of 8 complete PN sequences. When the time delay is 128 sampling intervals, with the movement of the sliding window, there is a small spike output when the local sequence c(k) is aligned with 2 PN sequences, and when the local sequence c(k) is aligned with 8 PN sequences There is the largest spike output, and the output value is small when the PN sequence is not aligned with the local sequence, so it is easy to find the time synchronization point. This time synchronization method can complete coarse time synchronization and fine time synchronization at one time, has high capture probability, and small false alarm probability and false negative probability.

在实际传输中，将训练序列以一定的功率比叠加到数据上，如图2所示，其中训练序列与数据功率比为ρ∶(1-ρ)。根据直接序列扩频的工作原理，在对PN序列作相关时，可抑制一部分数据对PN序列的干扰。In actual transmission, the training sequence is superimposed on the data with a certain power ratio, as shown in Figure 2, where the power ratio of the training sequence to the data is ρ:(1-ρ). According to the working principle of direct sequence spread spectrum, when the PN sequence is correlated, the interference of a part of data to the PN sequence can be suppressed.

频率同步方法是在取得时间同步后，对时间同步的目标函数|γ(k，a)|归一化并取相位，得The frequency synchronization method is to normalize the objective function |γ(k, a)| of time synchronization and take the phase after obtaining time synchronization, and obtain

$\overset{^^}{ϵ ϵ} = = \frac{M m}{KP KP} \cdot \cdot arg arg ((γ γ ((k k,, 00)) / / {LK LK}^{22} {ρσ ρσ}_{s the s}^{22})) - - - - - - ((22))$

其中σ_s ²为接收到的有用信号的功率，γ(k，0)表示时间同步已准确实现，即a＝0。Where σ _s ² is the power of the received useful signal, and γ(k, 0) indicates that the time synchronization has been accurately realized, that is, a=0.

所述叠加PN序列的方法主要缺点是：PN序列的功率限定了误码率的最小值，通常传输信号要求尽可能低的误码率，而时间和频率同步要求PN序列有一定功率，二者相矛盾；PN序列叠加在随机的数据上，只能用来同步，要实现信号的正确解调还需要另外的开销用做信道估计等；所述方法最初是针对SISO系统的，若直接应用于MIMO-OFDM系统，每根天线都需要叠加训练序列，在总发射功率一定的客观条件下，数据功率占总功率的比例会进一步降低，导致数据的检测将受到PN序列更严重的干扰。The main disadvantage of the method of superimposing the PN sequence is: the power of the PN sequence limits the minimum value of the bit error rate, usually the transmission signal requires a bit error rate as low as possible, and time and frequency synchronization require the PN sequence to have a certain power, the two Contradictory; the PN sequence is superimposed on the random data and can only be used for synchronization. To achieve the correct demodulation of the signal, additional overhead is required for channel estimation, etc.; the method is originally aimed at the SISO system. If it is directly applied to In the MIMO-OFDM system, each antenna needs to superimpose the training sequence. Under the objective condition of a certain total transmission power, the ratio of data power to the total power will be further reduced, resulting in more serious interference of data detection by the PN sequence.

现有技术的第二种方法是导频辅助OFDM系统时间同步的方法，利用已知的用于信道估计的频域导频符号并结合循环前缀，完成时间同步。具体过程如下：The second method in the prior art is a method of pilot-assisted OFDM system time synchronization, which uses known frequency-domain pilot symbols for channel estimation combined with a cyclic prefix to complete time synchronization. The specific process is as follows:

设一个OFDM符号包括N个子载波，在其中N_p个子载波中插入已知导频符号，用γ表示导频子载波的位置序号集合，则传输信号可看作是数据和导频两部分的和。第一部分是N-N_p个数据子载波，其对应的时域信号为：Assuming that an OFDM symbol includes N subcarriers, and inserting known pilot symbols into _Np subcarriers, and using γ to represent the set of position numbers of pilot subcarriers, the transmission signal can be regarded as the sum of two parts of data and pilot . The first part is NN _p data subcarriers, and the corresponding time domain signal is:

$s the s ((n no)) = = \frac{11}{\sqrt{N N}} \underset{k k &Element; &Element; {{00,, KN KN - - 11}} \ \ γ γ}{Σ Σ} {x x}_{k k} {e e}^{} j j 22 πkn πkn / / N N - - - - - - ((33))$

其中，{0，K N-1}\γ表示集合{0，K N-1}与集合γ的差集，x_k是第k个子载波上传输的数据符号，x_k可以用任意一种已知的星座映射的方式进行映射，并且具有平均期望功率 $E {{| x_{k} |}^{2}} = σ_{x}^{2}$ 。第二部分包括N_p个导频子载波，对应时域信号为Among them, {0, K N-1}\γ represents the difference between the set {0, K N-1} and the set γ, x _k is the data symbol transmitted on the kth subcarrier, and x _k can be used by any known constellation mapping and has an average expected power $E. {{| x_{k} |}^{2}} = σ_{x}^{2}$ . The second part includes N _p pilot subcarriers, and the corresponding time domain signal is

$m m ((n no)) = = \frac{11}{\sqrt{N N}} \underset{k k &Element; &Element; γ γ}{Σ Σ} {p p}_{k k} {e e}^{} j j 22 πkn πkn / / N N - - - - - - ((44))$

其中p_k是第k个子载波上发射的导频符号，其期望功率 $E {{| p_{k} |}^{2}} = σ_{x}^{2}$ ，在AWGN(Additive White Gaussian Noise，加性白高斯噪声)信道下接收信号可以表示为where p _k is the pilot symbol transmitted on the kth subcarrier with expected power $E. {{| p_{k} |}^{2}} = σ_{x}^{2}$ , the received signal under the AWGN (Additive White Gaussian Noise) channel can be expressed as

r(n)＝s(n-θ)+m(n-θ)+w(n) (5)r(n)=s(n-θ)+m(n-θ)+w(n) (5)

其中θ表示未知的整数倍的归一化时间延迟，w(n)是加性复白高斯噪声，w(n)方差为σ_w ²。接收信号r(n)的统计特性进行如下：Where θ represents the normalized time delay of an unknown integer multiple, w(n) is additive complex white Gaussian noise, and the variance of w(n) is σ _w ² . The statistical properties of the received signal r(n) proceed as follows:

对于具有大量数据子载波的OFDM系统(N_p＝N)，时域信号s(n)可近似为离散时间高斯过程，具有方差ασ_x ²，α＝(N-N_p)/N；而导频信号m(n)是确定信号。故接收信号r(n)是高斯过程，具有时变均值m(n)，方差ασ_x ²+σ_w ²。采用ML算法，最大似然条件满足时，时延估计

的计算公式为：For an OFDM system with a large number of data subcarriers (N _p =N), the time-domain signal s(n) can be approximated as a discrete-time Gaussian process with variance ασ _x ² , α=(NN _p )/N; while the pilot signal m(n) is a definite signal. Therefore, the received signal r(n) is a Gaussian process with time-varying mean value m(n) and variance ασ _x ² +σ _w ² . Using the ML algorithm, when the maximum likelihood condition is satisfied, the time delay estimation

The calculation formula is:

${\overset{^^}{θ θ}}_{ML ML} = = arg arg \underset{θ θ}{max max} {{Λ Λ ((θ θ))}} - - - - - - ((66))$

其中，in,

Λ(θ)＝ρΛ_cp(θ)+(1-ρ)Λ_p(θ) (7)Λ(θ)= _ρΛcp (θ)+(1-ρ) _Λp (θ) (7)

$ρ ρ = = \frac{{ασ ασ}_{x x}^{22}}{{ασ ασ}_{x x}^{22} + + {σ σ}_{w w}^{22}} - - - - - - ((88))$

${Λ Λ}_{cp cp} ((θ θ)) = = Re Re {{{Σ Σ}_{n no = = θ θ}^{θ θ + + L L - - 11} {r r}^{* *} ((n no)) r r ((n no + + N N))}}$

$- - \frac{ρ ρ}{22} {Σ Σ}_{n no = = θ θ}^{θ θ + + L L - - 11} {| | r r ((n no)) | |}^{22} + + {| | r r ((n no + + N N)) | |}^{22} - - - - - - ((99))$

${Λ Λ}_{p p} ((θ θ)) = = ((11 + + ρ ρ)) Re Re {{\underset{n no}{Σ Σ} {r r}^{* *} ((n no)) m m ((n no - - θ θ))}}$

$- - ρRe ρRe {{{Σ Σ}_{n no = = θ θ}^{θ θ + + L L - - 11} {((r r ((n no)) + + r r ((n no + + N N))))}^{* *} m m ((n no - - θ θ))}} - - - - - - ((1010))$

公式(5)到公式(10)所述的ML算法适合于没有频偏的情况，当存在频偏时，接收信号可以表示为The ML algorithm described in formula (5) to formula (10) is suitable for the case of no frequency offset. When there is frequency offset, the received signal can be expressed as

r(n)＝(s(n-θ)+m(n-θ))e^j2πεn/N+w(n) (11)r(n)=(s(n-θ)+m(n-θ))e ^j2πεn/N +w(n) (11)

其中，ε是归一化频偏，将ε表示成两部分的和：ε＝ε_I+ε_f，其中，ε_I表示整数部分频偏，ε_f表示小数部分频偏。Wherein, ε is the normalized frequency offset, and ε is expressed as a sum of two parts: ε=ε _I +ε _f , wherein, ε _I indicates the frequency offset of the integer part, and ε _f indicates the frequency offset of the fractional part.

由于频偏或信道相位的变化会引起公式(7)存在一定的随机性，导致估计结果

方差增大，而且，上述ML算法在计算ρ时需要知道接收端SNR(信噪比)，这点往往不可实现。所以针对这两点，又提出了鲁棒的时间同步算法：Due to the change of frequency offset or channel phase, there will be some randomness in the formula (7), resulting in the estimation result

The variance increases, and the above ML algorithm needs to know the SNR (signal-to-noise ratio) of the receiving end when calculating ρ, which is often not achievable. Therefore, for these two points, a robust time synchronization algorithm is proposed:

${\overset{^^}{θ θ}}_{ML ML} = = arg arg \underset{θ θ}{max max} {{{ρ ρ}^{% %} {Λ Λ}_{cp cp}^{% %} ((θ θ)) + + ((11 - - {ρ ρ}^{% %})) {Λ Λ}_{p p}^{% %} ((θ θ))}} - - - - - - ((1212))$

其中，in,

${ρ ρ}^{% %} = = \frac{α α {S S}^{22} NR NR}{α α {S S}^{22} NR NR + + 11} - - - - - - ((1313))$

${Λ Λ}_{cp cp}^{% %} ((θ θ)) = = | | {Σ Σ}_{n no = = θ θ}^{θ θ + + L L - - 11} {r r}^{* *} ((n no)) r r ((n no + + N N)) | |$

$- - \frac{{ρ ρ}^{% %}}{22} {Σ Σ}_{n no = = θ θ}^{θ θ + + L L - - 11} {| | r r ((n no)) | |}^{22} + + {| | r r + + ((n no + + N N)) | |}^{22} - - - - - - ((1414))$

${Λ Λ}_{p p}^{% %} ((θ θ)) = = ((11 + + {ρ ρ}^{% %})) | | \underset{n no}{Σ Σ} {r r}^{* *} ((n no)) m m ((n no - - θ θ)) | |$

$- - {ρ ρ}^{% %} | | {Σ Σ}_{n no = = θ θ}^{θ θ + + L L - - 11} {((r r ((n no)) + + r r ((n no + + N N))))}^{* *} m m ((n no - - θ θ)) | | - - - - - - ((1515))$

其中S²NR是假设的SNR值，并且固定不变。Wherein S ² NR is the assumed SNR value, and it is fixed.

可见，所述导频辅助OFDM系统时间同步的方法有如下缺点：所采用的ML算法是基于AWGN信道导出，用于多径衰落信道时，接收序列r(k)的自相关特性会呈现复杂的变化，导致似然函数的随机变化，算法性能明显下降；需要对接收信号的信噪比进行估计，估计误差会导致同步精度下降；虽然可同时完成时间同步和信道估计，但频率同步尚需要另外完成，并未在实质上减少系统的开销；在更复杂的MIMO信道环境更难以使用。It can be seen that the method for time synchronization of the pilot-assisted OFDM system has the following disadvantages: the adopted ML algorithm is derived based on the AWGN channel, and when used in a multipath fading channel, the autocorrelation characteristics of the received sequence r(k) will present complex changes, resulting in random changes in the likelihood function, and the performance of the algorithm is significantly reduced; the signal-to-noise ratio of the received signal needs to be estimated, and the estimation error will lead to a decrease in synchronization accuracy; although time synchronization and channel estimation can be completed at the same time, frequency synchronization still requires additional Complete, without substantially reducing system overhead; it is more difficult to use in a more complex MIMO channel environment.

发明内容Contents of the invention

本发明实施例所要解决的技术问题是提供一种承载有效数据效率高，同步过程消耗小的正交频分复用通信中时间和频率同步的方法及系统。The technical problem to be solved by the embodiments of the present invention is to provide a method and system for synchronizing time and frequency in OFDM communication with high efficiency of carrying effective data and low consumption of the synchronization process.

为解决上述技术问题，通过如下技术方案实现：In order to solve the above-mentioned technical problems, it is realized through the following technical solutions:

本发明实施例提供一种正交频分复用通信中时间和频率同步的方法，其包括：An embodiment of the present invention provides a method for time and frequency synchronization in OFDM communication, which includes:

将载波调制后的原始信息串并转换为n_T条支路数据流，n_T为发射天线数量；The original information after carrier modulation is serially converted into n _T branch data streams, where n _T is the number of transmitting antennas;

构造发射天线的频域导频序列；Construct the frequency domain pilot sequence of the transmitting antenna;

将发射天线的支路数据流与频域导频序列映射到预定频域导频图案的相应数据位置，生成数据信息；Mapping the branch data stream of the transmitting antenna and the frequency domain pilot sequence to the corresponding data position of the predetermined frequency domain pilot pattern to generate data information;

将数据信息进行OFDM调制后由发射天线发射出去；After the data information is modulated by OFDM, it is transmitted by the transmitting antenna;

接收天线接收电磁波，产生时域输出序列；The receiving antenna receives electromagnetic waves and generates time-domain output sequences;

根据时域输出序列与参考序列的相关性，构造天线间的同步目标函数；According to the correlation between the time-domain output sequence and the reference sequence, the synchronization objective function between the antennas is constructed;

利用同步目标函数的峰值信息估计时间同步点和频偏。The time synchronization point and frequency offset are estimated by using the peak information of the synchronization objective function.

根据所述的时间同步点和频偏信息完成时间和频率的同步。Time and frequency synchronization is completed according to the time synchronization point and frequency offset information.

本发明实施例还提供一种可实现时间和频率同步的正交频分复用通信系统，其发射端包括：用于将原始输入信息进行编码、并根据频域导频图案生成各天线的数据信息的多输入多输出编码器，至少一个用于对各天线的数据信息进行OFDM调制以生成OFDM符号的OFDM调制器，以及，将所述OFDM符号进行射频处理并通过发射天线发射出去的射频发射部分；The embodiment of the present invention also provides an OFDM communication system that can realize time and frequency synchronization, and its transmitting end includes: encoding the original input information and generating data for each antenna according to the frequency domain pilot pattern A multiple-input multiple-output encoder for information, at least one OFDM modulator for performing OFDM modulation on the data information of each antenna to generate OFDM symbols, and a radio frequency transmitter for performing radio frequency processing on the OFDM symbols and transmitting them through the transmitting antennas part;

其接收端包括：接收射频信号并进行射频处理以产生接收信号的时域输出序列的射频接收部分，根据所述时域输出序列与本地序列的相关性、构造各对天线间以时间点为自变量的同步目标函数、并使用同步目标函数的峰值信息估计时间同步点和频偏的时间和频率同步装置，至少一个结合所述估计出的时间同步点和频偏信息对所述接收信号的时域输出序列进行OFDM解调、以得到还原的各天线数据信息的OFDM解调器，以及，对还原的各天线的数据信息先利用频域导频进行导引辅助信道估计、再进行空时频解码以得到恢复的原始的输入信息的多输入多输出解码器。Its receiving end includes: receiving the radio frequency signal and performing radio frequency processing to generate the radio frequency receiving part of the time domain output sequence of the received signal, according to the correlation between the time domain output sequence and the local sequence, construct the time point between each pair of antenna A variable synchronization objective function, and a time and frequency synchronization device for estimating a time synchronization point and a frequency offset using the peak information of the synchronization objective function, at least one time synchronization point and frequency offset information combined with the estimated time synchronization point and frequency offset information for the received signal The OFDM demodulator that performs OFDM demodulation on the output sequence in the domain domain to obtain the restored data information of each antenna, and uses the frequency domain pilot to conduct pilot auxiliary channel estimation for the restored data information of each antenna, and then performs space-time-frequency MIMO decoder for decoding to recover the original input information.

实施本发明的实施例所提供的正交频分复用通信中时间和频率同步的方法及系统，可以获得如下有益效果：通过设计特殊的各发射天线的导频序列，使系统能够以更低的开销实现时间和频率同步，各天线只需要发送一个在频域插入导频的OFDM符号，不需要另外的训练序列，即可完成同步任务。Implementing the method and system for time and frequency synchronization in OFDM communication provided by the embodiments of the present invention can obtain the following beneficial effects: by designing special pilot sequences for each transmitting antenna, the system can be used at a lower The cost of time and frequency synchronization is realized, and each antenna only needs to send an OFDM symbol with a pilot inserted in the frequency domain, without additional training sequences, to complete the synchronization task.

附图说明Description of drawings

图1是OFDM同步训练序列原理框图。Fig. 1 is a functional block diagram of OFDM synchronous training sequence.

图2是OFDM同步训练序列信号叠加方式示意图。Fig. 2 is a schematic diagram of a superposition manner of OFDM synchronous training sequence signals.

图3是本发明一个优选实施例中的时间和频率同步方法的流程示意图。Fig. 3 is a schematic flowchart of a time and frequency synchronization method in a preferred embodiment of the present invention.

图4是本发明一个优选实施例中的可实现时间和频率同步的OFDM通信系统的原理框图。Fig. 4 is a functional block diagram of an OFDM communication system capable of realizing time and frequency synchronization in a preferred embodiment of the present invention.

图5是本发明一个优选实施例中各天线联合设计的频域导频图案实施例的示意图。Fig. 5 is a schematic diagram of an embodiment of a frequency-domain pilot pattern jointly designed by each antenna in a preferred embodiment of the present invention.

图6是本发明一个优选实施例中含循环前缀的时域导频序列的示意图。Fig. 6 is a schematic diagram of a time-domain pilot sequence with a cyclic prefix in a preferred embodiment of the present invention.

图7是本发明一个优选实施例中MIMO编码器的原理框图。Fig. 7 is a functional block diagram of a MIMO encoder in a preferred embodiment of the present invention.

图8是本发明一个优选实施例中OFDM调制器的原理框图。Fig. 8 is a functional block diagram of an OFDM modulator in a preferred embodiment of the present invention.

图9是本发明一个优选实施例中射频发射机的原理框图。Fig. 9 is a functional block diagram of a radio frequency transmitter in a preferred embodiment of the present invention.

图10是本发明一个优选实施例中时间和频率同步模块的同步算法模块示意图。Fig. 10 is a schematic diagram of a synchronization algorithm module of the time and frequency synchronization module in a preferred embodiment of the present invention.

图11是本发明一个优选实施例中射频接收机的原理框图。Fig. 11 is a functional block diagram of a radio frequency receiver in a preferred embodiment of the present invention.

图12是本发明一个优选实施例中OFDM解调器的原理框图。Fig. 12 is a functional block diagram of an OFDM demodulator in a preferred embodiment of the present invention.

具体实施方式Detailed ways

本发明实施例提供的OFDM通信中发射信号与接收信号之间时间和频率同步的方法可以用于MIMO系统。所述方法是包括在发射端的发送数据构造和在接收端利用特定结构的发送数据进行时间和频率同步在内的联合过程。其思想是通信过程使用导频，通过设计发射端各天线的频域联合导频图案，使其时域序列具有特定的相关性，然后利用所述时域相关性进行时间和频率同步。The method for time and frequency synchronization between the transmitted signal and the received signal in OFDM communication provided by the embodiment of the present invention can be used in a MIMO system. The method is a joint process including the construction of the transmitted data at the transmitting end and the time and frequency synchronization at the receiving end using the specially structured transmitted data. The idea is to use pilots in the communication process. By designing the frequency-domain joint pilot patterns of each antenna at the transmitting end, the time-domain sequences have specific correlations, and then use the time-domain correlations to synchronize time and frequency.

下面结合附图对本发明实施例提供的MIMO-OFDM时间和频率同步的方法与系统进行详细描述。The MIMO-OFDM time and frequency synchronization method and system provided by the embodiments of the present invention will be described in detail below with reference to the accompanying drawings.

本发明一个优选实施例中，MIMO-OFDM时间和频率同步方法的流程示意图如图3所示，图中步骤301到步骤304为发射端步骤，步骤305到步骤307为接收端步骤。In a preferred embodiment of the present invention, a schematic flowchart of the MIMO-OFDM time and frequency synchronization method is shown in Figure 3, in which steps 301 to 304 are steps at the transmitting end, and steps 305 to 307 are steps at the receiving end.

首先，在步骤301，将载波调制后的原始信息串并转换为n_T条支路数据流，其中n_T为发射天线数量；然后，步骤302构造各发射天线的频域导频序列；之后，在步骤303，将步骤301产生的各发射天线的支路数据流与步骤302构造的频域导频序列共同映射到预定频域导频图案的相应数据位置，生成各发射天线的数据信息；在发射端的最后，步骤304，对各天线的数据信息进行OFDM调制，经射频发射出去。First, in step 301, the original information after carrier modulation is serially converted into n _T branch data streams, where n _T is the number of transmitting antennas; then, step 302 constructs the frequency domain pilot sequence of each transmitting antenna; after that, In step 303, the branch data streams of each transmitting antenna generated in step 301 and the frequency domain pilot sequence constructed in step 302 are jointly mapped to the corresponding data positions of the predetermined frequency domain pilot pattern to generate data information of each transmitting antenna; At the end of the transmitting end, step 304, perform OFDM modulation on the data information of each antenna, and transmit it through radio frequency.

在信号接收端，则首先在步骤305，由各接收天线接收电磁波，产生时域输出序列；然后，在步骤306，根据时域输出序列与本地序列的相关性，构造各对天线间以时间点为自变量的同步目标函数；之后，在步骤307利用同步目标函数的峰值信息估计时间同步点和频偏。At the signal receiving end, firstly, in step 305, each receiving antenna receives electromagnetic waves to generate a time-domain output sequence; then, in step 306, according to the correlation between the time-domain output sequence and the local sequence, a time point is the synchronization objective function of the independent variable; then, in step 307, the time synchronization point and the frequency offset are estimated by using the peak information of the synchronization objective function.

这里要说明的是，步骤301和步骤302并没有承接关系，二者的顺序可以是任意的，即可以首先执行任何一个，也可以二者同时分别进行。It should be noted here that step 301 and step 302 have no succession relationship, and the order of the two can be arbitrary, that is, either one can be performed first, or the two can be performed separately at the same time.

本发明实施例的OFDM通信时间和频率同步方法将通过应用于OFDM通信系统得以具体实施。本发明一个优选实施例中，使用MIMO-OFDM通信时间和频率同步方法的系统原理框图如图4所示。所述MIMO-OFDM通信系统包括空时频调制部分401、射频发射部分406、射频接收部分413、时间频率同步装置420和空时频解调部分421。The OFDM communication time and frequency synchronization method in the embodiment of the present invention will be implemented by being applied to an OFDM communication system. In a preferred embodiment of the present invention, the functional block diagram of the system using the MIMO-OFDM communication time and frequency synchronization method is shown in FIG. 4 . The MIMO-OFDM communication system includes a space-time-frequency modulation part 401 , a radio frequency transmitting part 406 , a radio frequency receiving part 413 , a time-frequency synchronization device 420 and a space-time-frequency demodulation part 421 .

其中空时频调制部分401、射频发射部分406位于发射端。空时频调制部分401进一步包括：多输入多输出编码器402，用于将原始输入信息进行编码，根据频域导频图案生成各天线的数据信息，MIMO编码器402的更具体的内部结构将结合图6在下文详细介绍；多个OFDM调制器(403，404，405)，针对各天线的数据信息进行OFDM调制，生成OFDM符号，这里OFDM调制可以通过IDFT(Inverse Discrete Fourier Transform，逆离散傅立叶变换)实现；射频发射部分，包括多组射频处理一(407，408，409)和发射天线(410，411，412)的组合，每组组合(如射频处理一407和发射天线410的组合)对应一个OFDM调制器，由射频处理一将OFDM符号进行射频处理并通过发射天线发射出去。The space-time-frequency modulation part 401 and the radio frequency transmitting part 406 are located at the transmitting end. The space-time-frequency modulation part 401 further includes: a multiple-input multiple-output encoder 402, which is used to encode the original input information and generate data information of each antenna according to the frequency domain pilot pattern. The more specific internal structure of the MIMO encoder 402 will In conjunction with Fig. 6, it will be described in detail below; a plurality of OFDM modulators (403, 404, 405) perform OFDM modulation for the data information of each antenna to generate OFDM symbols, where OFDM modulation can pass IDFT (Inverse Discrete Fourier Transform, Inverse Discrete Fourier Transform) transformation) realization; radio frequency transmission part, including the combination of multiple groups of radio frequency processing one (407,408,409) and transmitting antenna (410,411,412), each combination (such as the combination of radio frequency processing one 407 and transmitting antenna 410) Corresponding to an OFDM modulator, the radio frequency processing-processes the OFDM symbols through the radio frequency and transmits them through the transmitting antenna.

射频接收部分413、时间和频率同步装置420和空时频解调部分421位于接收端。射频接收部分413包括多组接收天线(430，431，432)、接收本振(417，418，419)和射频处理二(414，415，416)的组合，每组组合对应一个OFDM解调器，每组组合通过接收天线接收射频信号，并根据接收本振的频率信息由射频处理二进行射频处理，产生接收信号的时域输出序列。时间和频率同步装置420的作用是进行时间和频率同步，根据时域输出序列与本地序列的相关性，构造各对天线间以时间点为自变量的同步目标函数，使用同步目标函数的峰值信息估计时间同步点和频偏，关于其具体方法将在后文结合图10进行详细阐述。时间和频率同步装置420所估计出的时间同步点和频偏信息被传送到接收本振(417，418，419)和OFDM解调器(422，423，424)。OFDM解调器结合所述估计出的时间同步点和频偏信息对所述接收信号的时域输出序列进行OFDM解调，得到还原的各天线数据信息，其中主要用到的是时间同步点的信息。多输入多输出解码器425对还原的各天线的数据信息先利用其中的频域导频进行导引辅助信道估计，再进行空时频解码，得到恢复的原始的输入信息。The radio frequency receiving part 413, the time and frequency synchronization device 420 and the space-time-frequency demodulation part 421 are located at the receiving end. The radio frequency receiving part 413 includes the combination of multiple groups of receiving antennas (430, 431, 432), receiving local oscillators (417, 418, 419) and radio frequency processing two (414, 415, 416), and each group of combinations corresponds to an OFDM demodulator , each combination receives the radio frequency signal through the receiving antenna, and performs radio frequency processing by radio frequency processing 2 according to the frequency information of the received local oscillator to generate a time domain output sequence of the received signal. The function of the time and frequency synchronization device 420 is to perform time and frequency synchronization. According to the correlation between the time domain output sequence and the local sequence, construct a synchronization objective function between each pair of antennas with the time point as an argument, and use the peak information of the synchronization objective function The specific method of estimating the time synchronization point and the frequency offset will be described in detail later in conjunction with FIG. 10 . The time synchronization point and frequency offset information estimated by the time and frequency synchronization device 420 are transmitted to the receiving local oscillators (417, 418, 419) and OFDM demodulators (422, 423, 424). The OFDM demodulator performs OFDM demodulation on the time-domain output sequence of the received signal in combination with the estimated time synchronization point and frequency offset information to obtain the restored data information of each antenna, which mainly uses the time synchronization point information. The multiple-input multiple-output decoder 425 uses the frequency-domain pilots in the restored data information of each antenna to guide and assist channel estimation, and then performs space-time-frequency decoding to obtain the restored original input information.

下面对图4中部分装置的工作过程及原理进行进一步的详细说明。The working process and principle of some devices in Fig. 4 will be further described in detail below.

在发射端，MIMO编码器402的作用是根据频域导频图案生成各天线的数据信息。图5是本发明一个优选实施例中各天线联合设计的频域导频图案实施例的示意图，图中横轴为各根天线，纵轴对应各个子载波。从图5中可以看出，各天线均采用导频辅助OFDM(Pilot-Assisted OFDM)发射结构，选择等间隔排列的子载波位置放置频域导频(该子载波称为导频子载波)，各天线导频子载波的位置互不相同，数据信息加载于含有导频子载波的频段以外的位置。若OFDM的子载波数为N，其中导频子载波为N_p，则每根天线的数据位置为(N-n_TN_p)个子载波。每根天线导频子载波间的间隔为D_f，显然D_f应至少大于总的天线数量。At the transmitting end, the function of the MIMO encoder 402 is to generate data information of each antenna according to the frequency domain pilot pattern. Fig. 5 is a schematic diagram of an embodiment of a frequency-domain pilot pattern jointly designed by each antenna in a preferred embodiment of the present invention, the horizontal axis in the figure represents each antenna, and the vertical axis corresponds to each subcarrier. It can be seen from Figure 5 that each antenna adopts a pilot-assisted OFDM (Pilot-Assisted OFDM) transmission structure, and selects subcarriers arranged at equal intervals to place frequency-domain pilots (the subcarriers are called pilot subcarriers). The positions of the pilot sub-carriers of each antenna are different from each other, and the data information is loaded in a position other than the frequency band containing the pilot sub-carriers. If the number of subcarriers of OFDM is N, and the pilot subcarrier is N _p , then the data position of each antenna is (Nn _T N _p ) subcarriers. The interval between the pilot subcarriers of each antenna is D _f , obviously D _f should be at least greater than the total number of antennas.

下面分别介绍频域导频图案中导频和数据的构成及映射方式。首先介绍导频序列的构造。第i根发射天线，i＝1，2，K n_T，发射的一个OFDM符号包含的导频序列构造方法如下：选择多段相关性较好的序列作为基础序列；将选出的多段基础序列重复排列，且每段基础序列乘以相应系数构成时域导频序列；对时域导频序列进行离散傅立叶变换得到频域导频序列。The composition and mapping manner of the pilot and data in the frequency domain pilot pattern are respectively introduced below. First introduce the structure of the pilot sequence. The i-th transmitting antenna, i=1, 2, K n _T , the method of constructing the pilot sequence contained in an OFDM symbol transmitted is as follows: select multiple sequences with better correlation as the basic sequence; repeat the selected multiple basic sequences Arrangement, and each segment of the basic sequence is multiplied by a corresponding coefficient to form a time-domain pilot sequence; the time-domain pilot sequence is subjected to discrete Fourier transform to obtain a frequency-domain pilot sequence.

采用此种构造方法的目的是使频域导频满足预定导频图案且具有良好的相关性，下面对频域导频序列的构造原理和具体内容加以阐述。The purpose of adopting this construction method is to make the frequency domain pilot meet the predetermined pilot pattern and have good correlation. The construction principle and specific content of the frequency domain pilot sequence will be described below.

先从频域出发，设DFT的点数为N，各天线的导频子载波间隔为M，要求M可以整除N。为了使用FFT(Fast Fourier Transform，快速傅立叶变换)方法计算DFT，一般要求DFT点数N为2ⁿ，故M整除N的条件容易满足。第i根天线的频域导频X_i可以表示为冲激串的形式Starting from the frequency domain, set the number of DFT points to be N, and the pilot subcarrier interval of each antenna to be M, and M is required to be divisible by N. In order to use the FFT (Fast Fourier Transform, Fast Fourier Transform) method to calculate DFT, the number of DFT points N is generally required to be 2 ⁿ , so the condition that M divides N by M is easy to satisfy. The frequency-domain pilot X _i of the i-th antenna can be expressed in the form of a burst

${X x}_{i i} ((k k)) = = {Σ Σ}_{r r = = 00}^{\frac{N N}{M m} - - 11} {α α}_{r r}^{i i} δ δ ((k k - - Mr Mr. - - {p p}_{i i})),, k k = = 00,, K K,, N N - - 11;; i i = = 11,, K K,, {n no}_{T T} - - - - - - ((1616))$

其中，p_i表示第i根发射天线的第一个导频子载波的位置相对于k＝0的移位，p_i＝0，1，K M-1，a_r ⁱ表示第i根发射天线的第r个导频符号。不同天线的导频要错开放置，故p_i必须不同，因此这种导频配置方法要求n_T≤M。Among them, p _i represents the shift of the position of the first pilot subcarrier of the i-th transmitting antenna relative to k=0, p _i =0, 1, K M-1, a _r ⁱ represents the i-th transmitting antenna The rth pilot symbol of . The pilots of different antennas must be staggered, so p _i must be different, so this pilot configuration method requires n _T ≤ M.

X_i(k)经过IDFT后对应的N点时域序列为：The time-domain sequence of N points corresponding to _Xi (k) after IDFT is:

${x x}_{i i} ((n no)) = = {e e}^{j j \frac{22 π π}{N N} {p p}_{i i} n no} {Σ Σ}_{r r = = 00}^{\frac{N N}{M m} - - 11} {α α}_{r r}^{i i} {e e}^{j j \frac{22 π π}{N N} Mnr Mnr} - - - - - - ((1717))$

可见时域序列具有如下准周期性的特点：对于p_i＝0，时域序列明显以N/M为周期；对于p_i＝1，K，M-1，时域序列无周期性，但整个序列可视作将序列的前N/M点重复M次，每一段分别乘上It can be seen that the time-domain sequence has the following quasi-periodic characteristics: for p _i =0, the time-domain sequence obviously takes N/M as the period; for p _i =1, K, M-1, the time-domain sequence has no periodicity, but the entire The sequence can be regarded as repeating the first N/M points of the sequence M times, and each segment is multiplied by

$α α = = [[11,, {e e}^{j j \frac{22 π π}{M m} {p p}_{i i}},, {e e}^{j j \frac{22 π π}{M m} ((22 {p p}_{i i}))},, K K,, {e e}^{j j \frac{22 π π}{M m} ((((M m - - 11)) {p p}_{i i}))}]] - - - - - - ((1818))$

当p_i＝0，结论也成立，此时公式(18)为α＝[1，...，1]。When p _i =0, the conclusion also holds, and formula (18) is α=[1, . . . , 1].

频域导频序列经过IDFT后对应的N点序列称为时域导频序列。二者是同一数学实体序列的频域和时域表示。The N-point sequence corresponding to the IDFT of the frequency domain pilot sequence is called the time domain pilot sequence. Both are frequency and time domain representations of the same sequence of mathematical entities.

根据频域导频序列的时域特性，可以直接在时域产生N点导频序列，这样可以选择相关特性较好的PN序列以提高时间和频率同步的性能。直接在时域生成导频序列的步骤为：According to the time-domain characteristics of the frequency-domain pilot sequence, the N-point pilot sequence can be directly generated in the time domain, so that the PN sequence with better correlation characteristics can be selected to improve the performance of time and frequency synchronization. The steps to generate the pilot sequence directly in the time domain are:

各发射天线时域序列发生器生成长为K＝N/M点的PN序列c_i，i＝1，K，n_T。即基础序列，要求各天线的PN序列c_i之间互相关特性良好，。将c_i重复M次，得到M段相同的PN序列，每段分别乘上相应α_t ⁱ，t＝0，K，M-1，从而构成了天线i的N点时域导频序列x_i(n)。Each transmit antenna time-domain sequence generator generates a PN sequence c _i , i=1, K, n _T , whose length is K=N/M points. That is, the basic sequence requires good cross-correlation characteristics between the PN sequences _ci of each antenna. Repeat c _i for M times to obtain M segments of the same PN sequence, each segment is multiplied by the corresponding α _t ⁱ , t=0, K, M-1, thus forming the N-point time-domain pilot sequence x _i of antenna i (n).

通常，作为分隔和设计上的需要，在导频序列之前要加上循环前缀，图6是本发明一个优选实施例中含循环前缀的时域导频序列的示意图。图中c_qα_t ^q表示第q根发射天线的时域导频序列的第t段。其中α_t ⁱ定义为Usually, as a separation and design requirement, a cyclic prefix is added before the pilot sequence. FIG. 6 is a schematic diagram of a time-domain pilot sequence with a cyclic prefix in a preferred embodiment of the present invention. In the figure, c _q α _t ^q represents the tth section of the time-domain pilot sequence of the qth root transmitting antenna. where α _t ⁱ is defined as

${α α}_{t t}^{i i} = = {e e}^{j j \frac{22 πt πt}{M m} {p p}_{i i}},, t t = = 00,, K K,, M m - - 11,, i i = = 11,, K K,, {n no}_{T T} - - - - - - ((1919))$

将这些参数带入公式(17)构成x_i(n)，显见其DFT形式X_i(k)为等间隔分布的冲激串形式，用作导频时不同天线是X_i(k)互相错开的。Putting these parameters into formula (17) to form x _i (n), it is obvious that its DFT form Xi ₍ k) is in the form of impulse trains distributed at equal intervals. When used as a pilot, different antennas are staggered by _Xi (k) of.

接下来说明频域导频图案中数据的映射方式：对某一根天线，导频子载波数N_p＝N/M，所有天线的导频共占据n_TN_p个子载波，于是各天线的数据符号都映射到剩余的N-n_TN_p＝N(1-n_T/M)个数据子载波中。这样导频与数据在频域也是不重叠的。Next, the mapping method of data in the frequency domain pilot pattern is explained: for a certain antenna, the number of pilot subcarriers N _p =N/M, and the pilots of all antennas occupy n _T N _p subcarriers in total, so the All data symbols are mapped to the remaining Nn _T N _p =N(1-n _T /M) data subcarriers. In this way, the pilot frequency and the data do not overlap in the frequency domain.

下面结合导频序列生成和预定导频图案的设置以及数据映射方法，对图4中MIMO编码器402的结构进行详细说明。MIMO编码器402的原理框图如图7所示，其包括调制器726，串并转换装置一727和导频映射部分。The structure of the MIMO encoder 402 in FIG. 4 will be described in detail below in combination with pilot sequence generation, setting of a predetermined pilot pattern, and a data mapping method. The functional block diagram of the MIMO encoder 402 is shown in FIG. 7 , which includes a modulator 726 , a serial-to-parallel conversion device 1 727 and a pilot mapping part.

其中调制器726对原始输入信息进行载波调制，如BPSK(Binary PhaseShift Keying，二进制相移键控)调制，产生调制后的数据流；串并转换装置一727将载波调制后的数据流串并转换为n_T条支路数据流，n_T为发射天线数量；导频映射部分构造各发射天线的频域导频序列，将各发射天线的支路数据流与频域导频序列共同映射到预定频域导频图案的相应数据位置，生成各发射天线的数据信息。导频映射部分具体工作方式为将每条支路数据流经串并转换装置728，729再次串并转换为N-n_TN_p条，并通过映射装置732，733映射到频域导频图案的相应数据位置，显然数据将占据子载波的N-n_TN_p个数据位置；导引序列生成装置730，731产生对应每根天线的时域导频序列，并经过DFT装置734，735变换为频域导频序列，每根天线的频域导频将占据N_p个数据位置。要说明的是，导引序列生成装置可以包括适当的电路、逻辑或者处理器，其通过适当的电路或逻辑连接或执行适当的程序来产生时域导频序列。得到频域导频序列之后，将频域导频序列与映射数据通过合成装置736，737进行合成，生成各发射天线的数据信息。Wherein modulator 726 carries out carrier modulation to original input information, such as BPSK (Binary PhaseShift Keying, binary phase shift keying) modulation, produces the data stream after modulation; is n _T branch data streams, n _T is the number of transmitting antennas; the pilot mapping part constructs the frequency domain pilot sequence of each transmitting antenna, and maps the branch data stream and frequency domain pilot sequence of each transmitting antenna to a predetermined The corresponding data positions of the frequency-domain pilot patterns are used to generate the data information of each transmitting antenna. The specific working mode of the pilot mapping part is to convert each branch data stream into Nn _T N _p again through the serial-to-parallel conversion device 728, 729, and map to the corresponding frequency-domain pilot pattern through the mapping device 732, 733 data position, obviously the data will occupy Nn _T N _p data positions of the subcarrier; the pilot sequence generating means 730, 731 generate the time domain pilot sequence corresponding to each antenna, and transform it into the frequency domain pilot sequence through the DFT means 734, 735 frequency sequence, the frequency domain pilot of each antenna will occupy N _p data positions. It should be noted that the pilot sequence generation device may include appropriate circuits, logic or processors, which generate time-domain pilot sequences through appropriate circuit or logic connections or execute appropriate programs. After the frequency-domain pilot sequence is obtained, the frequency-domain pilot sequence and the mapping data are synthesized by combining means 736 and 737 to generate data information of each transmitting antenna.

要说明的是，在图6中，作为示例，仅标示了两根天线对应的各个装置，但此种标示不应作为对本发明的限制。如图6中两根天线之间的省略号部分所示，发射天线的数量n_T可以是任意自然数。显然单根发射天线作为特例也将落入本发明的保护范围。It should be noted that, in FIG. 6 , as an example, only devices corresponding to two antennas are marked, but such marking should not be regarded as a limitation to the present invention. As shown by the ellipsis between the two antennas in FIG. 6 , the number n _T of transmitting antennas may be any natural number. Apparently, a single transmitting antenna as a special case also falls within the protection scope of the present invention.

生成各发射天线的数据信息之后，多个OFDM调制器(403，404，405)对各天线的数据信息进行OFDM调制，生成OFDM符号，这里OFDM调制可以通过IDFT实现。完成数据映射后，一种优选方式为添加前缀生成OFDM符号。第i根发射天线发射的第l个OFDM符号可看作导频和数据两部分之和，添加N_g点循环前缀后的发射信号可表示为After the data information of each transmitting antenna is generated, multiple OFDM modulators (403, 404, 405) perform OFDM modulation on the data information of each antenna to generate OFDM symbols, where the OFDM modulation can be realized by IDFT. After the data mapping is completed, a preferred way is to generate OFDM symbols by adding a prefix. The l-th OFDM symbol transmitted by the i-th transmit antenna can be regarded as the sum of the two parts of the pilot and data, and the transmitted signal after adding the N _g -point cyclic prefix can be expressed as

${s the s}_{i i}^{l l} ((n no)) = = {d d}_{i i}^{l l} ((n no)) + + {x x}_{i i}^{l l} ((n no)),, n no = = 00,, K K,, N N + + {N N}_{g g} - - 11,, i i = = K K,, {n no}_{T T} - - - - - - ((2020))$

其中d_l ⁱ(n)和x_l ⁱ(n)分别表示第i根发射天线的第l个OFDM符号所包含数据序列和导频序列的时域第n个采样值。Among them, d _l ⁱ (n) and x _l ⁱ (n) respectively denote the nth sampling value of the data sequence and the pilot sequence contained in the lth OFDM symbol of the ith transmitting antenna in the time domain.

OFDM调制器的原理框图如图8所示。由于OFDM调制可以通过IDFT实现，因此OFDM调制器包括一个IDFT装置和一个添加循环前缀的装置。The functional block diagram of the OFDM modulator is shown in Figure 8. Since OFDM modulation can be realized by IDFT, the OFDM modulator includes an IDFT device and a device for adding a cyclic prefix.

调制生成的OFDM符号被送至射频发射部分，发射部分包括多组射频处理一(407，408，409)和发射天线(410，411，412)的组合，每组组合(如射频处理一407和发射天线410的组合)对应一个OFDM调制器，由射频处理一将OFDM符号进行射频处理并通过发射天线发射出去。The OFDM symbols generated by modulation are sent to the radio frequency transmission part, and the transmission part includes a combination of multiple groups of radio frequency processing one (407, 408, 409) and transmitting antennas (410, 411, 412), and each group of combinations (such as radio frequency processing one 407 and The combination of transmitting antennas 410) corresponds to an OFDM modulator, and radio frequency processing-the OFDM symbols are subjected to radio frequency processing and transmitted through the transmitting antennas.

其中，射频处理一为射频发射机，图9是本发明一个优选实施例中射频发射机原理框图。其包括发射本振、滤波器一和放大器一。输入信号和发射本振进行合成，并经过滤波器一进行滤波，再经过放大器一进行放大产生发射信号。Wherein, the radio frequency processing one is a radio frequency transmitter, and FIG. 9 is a functional block diagram of the radio frequency transmitter in a preferred embodiment of the present invention. It includes transmit local oscillator, filter one and amplifier one. The input signal and the transmitting local oscillator are synthesized, filtered by the filter 1, and then amplified by the amplifier 1 to generate the transmitting signal.

发射信号要在空间传播一段距离之后才能达到接收端。在信号发射端和接收端之间是复杂的信道空间，对信道空间进行建模是利用接收信号与本地序列共同构成同步函数的基础。例如，发射天线i到接收天线j之间的信道可以为准静态(quasi-static)频率选择性非相干散射瑞利衰落信道，即信道在每一个OFDM符号开始时随机变化，但在一个OFDM符号时间内保持不变。设信道存在L条不同时延的信道冲激响应，则采用抽头延迟线模型，信道冲激响应可以表示为：The transmitted signal has to travel a certain distance in space before reaching the receiving end. There is a complex channel space between the signal transmitter and the receiver. Modeling the channel space is the basis of using the received signal and the local sequence to form a synchronization function. For example, the channel between transmitting antenna i and receiving antenna j can be a quasi-static frequency-selective non-coherent scattering Rayleigh fading channel, that is, the channel changes randomly at the beginning of each OFDM symbol, but in an OFDM symbol time remains unchanged. Assuming that there are L channel impulse responses with different delays in the channel, the tapped delay line model is adopted, and the channel impulse response can be expressed as:

${h h}_{j j,, i i}^{l l} ((τ τ)) = = {Σ Σ}_{m m = = 00}^{L L - - 11} {h h}_{j j,, i i}^{% %} ((m m)) δ δ ((τ τ - - {τ τ}_{m m})) - - - - - - ((21 twenty one))$

其中，h_j，i ^l(τ)代表第l个OFDM符号时间内，发射天线i到接收天线j之间多径信道的冲激响应，L为信道多径的数目，

代表第l个OFDM符号时间内，发射天线i到接收天线j之间多径信道第m条径的等效低通冲激响应，t_m表示第m条径的时间延迟。Among them, h _{j, i} ^l (τ) represents the impulse response of the multipath channel between the transmitting antenna i and the receiving antenna j within the l-th OFDM symbol time, L is the number of channel multipath,

Represents the equivalent low-pass impulse response of the m-th path of the multi-path channel between the transmitting antenna i and the receiving antenna j within the l-th OFDM symbol time, and t _m represents the time delay of the m-th path.

发送信号可以表示成一个(n_T(N+N_g))×1的列向量S_l The transmitted signal can be expressed as a (n _T (N+N _g ))×1 column vector S _l

${S S}_{l l} = = {(({(\begin{matrix} {s the s}_{11}^{l l} ((00)) \\ M m \\ {S S}_{{n no}_{T T}}^{l l} \end{matrix})}^{T T},, {(\begin{matrix} {s the s}_{11}^{l l} ((11)) \\ M m \\ {s the s}_{{n no}_{T T}}^{l l} ((11)) \end{matrix})}^{T T},, K K,, {(\begin{matrix} {s the s}_{11}^{l l} ((N N + + {N N}_{g g} - - 11)) \\ M m \\ {s the s}_{{n no}_{T T}}^{l l} ((N N + + {N N}_{g g} - - 11)) \end{matrix})}^{T T}))}^{T T} - - - - - - ((22 twenty two))$

其中，(g)^T表示转置运算，s_i ^l(n)表示第l个OFDM符号时间第i根发射天线发送OFDM符号的时域第n个采样值。Among them, (g) ^T represents the transpose operation, and s _i ^l (n) represents the nth sampling value in the time domain of the OFDM symbol transmitted by the i-th transmitting antenna at the l-th OFDM symbol time.

公式(21)所示的信道冲激响应模型可以表示成时变FIR(Finite ImpulseResponse，有限长冲激响应)滤波器的形式，于是n_T条发射天线n_R条接收天线的多输入多输出信道模型可以表示为(n_R(N+N_g+N_L-1))×(n_T(N+N_g))的矩阵H_l，其中N_L表示第L条径的延时采样点数，The channel impulse response model shown in formula (21) can be expressed in the form of a time-varying FIR (Finite Impulse Response, finite impulse response) filter, so the multiple-input multiple-output channel of n _T transmitting antennas and n _R receiving antennas The model can be expressed as a matrix H _l of (n _R (N+N _g +N _L -1))×(n _T (N+N _g )), where N _L represents the number of delayed sampling points of the Lth path,

${H h}_{l l} = = (\begin{matrix} H h ((00)) & 00 & L L & 00 \\ H h ((11)) & H h ((00)) & L L & M m \\ M m & H h ((11)) & L L & 00 \\ H h (({N N}_{L L} - - 11)) & M m & L L & H h ((00)) \\ 00 & H h (({N N}_{L L} - - 11)) & L L & H h ((11)) \\ M m & M m & L L & M m \\ 00 & 00 & L L & H h (({N N}_{L L} - - 11)) \end{matrix}) - - - - - - ((23 twenty three))$

其中，in,

$H h ((m m)) = = (\begin{matrix} {h h}_{1,1 1,1}^{l l} ((m m)) & K K & {h h}_{{11,, n no}_{T T}}^{l l} ((m m)) \\ M m & O o & M m \\ {h h}_{{n no}_{R R},, 11}^{l l} ((m m)) & L L & {h h}_{{n no}_{R R},, {n no}_{T T}}^{l l} ((m m)) \end{matrix}),, m m = = 00,, . . . . . .,, {N N}_{L L} - - 11 - - - - - - ((24 twenty four))$

考虑加性复高斯白噪声，在时间和频率完全同步的情况下接收信号R_l可以表示为(n_R(N+N_g+N_L-1))×1的列向量Considering additive complex Gaussian white noise, the received signal R _l can be expressed as a column vector of (n _R (N+N _g +N _L -1))×1 when the time and frequency are fully synchronized

R_l＝H_lS_l+W_l (25)R _l ＝H _l S _l +W _l (25)

其中，W_l的元素是零均值加性复高斯白噪声采样，方差为σ_w ²，可表示成Among them, the elements of W _l are zero-mean additive complex Gaussian white noise samples with variance σ _w ² , which can be expressed as

${W W}_{l l} = = {(({(\begin{matrix} {w w}_{11}^{l l} ((00)) \\ M m \\ {w w}_{{n no}_{R R}}^{l l} ((00)) \end{matrix}),,}^{T T} {(\begin{matrix} {w w}_{11}^{l l} ((11)) \\ M m \\ {w w}_{{n no}_{R R}}^{l l} ((11)) \end{matrix})}^{T T},, K K,, {(\begin{matrix} {w w}_{11}^{l l} ((N N + + {N N}_{g g} + + {N N}_{L L} - - 22)) \\ M m \\ {w w}_{{n no}_{R R}}^{l l} ((N N + + {N N}_{g g} + + {N N}_{L L} - - 22)) \end{matrix})}^{T T}))}^{T T} - - - - - - ((2626))$

信号经信道传播后，由发送端到达接收端。在接收端，首先由射频接收部分413接收电磁波，产生时域输出序列。射频接收部分413包括多组接收天线(430，431，432)、接收本振(417，418，419)和射频处理二(414，415，416)的组合，每组组合对应一个OFDM解调器，每组组合通过接收天线接收射频信号，并根据接收本振的频率信息由射频处理二进行射频处理，产生接收信号的时域输出序列。After the signal propagates through the channel, it reaches the receiving end from the sending end. At the receiving end, the radio frequency receiving part 413 first receives the electromagnetic wave, and generates a time-domain output sequence. The radio frequency receiving part 413 includes the combination of multiple groups of receiving antennas (430, 431, 432), receiving local oscillators (417, 418, 419) and radio frequency processing two (414, 415, 416), and each group of combinations corresponds to an OFDM demodulator , each combination receives the radio frequency signal through the receiving antenna, and performs radio frequency processing by radio frequency processing 2 according to the frequency information of the received local oscillator to generate a time domain output sequence of the received signal.

射频处理二为射频接收机。图11是本发明一个优选实施例中射频接收机的原理框图。输入信号首先经过放大器放大，之后通过滤波器二进行滤波，滤波后的信号与来自接收本振的信号结合，将合成的信号再次经过滤波器的滤波，得到射频接收机的输出。The second radio frequency processing is the radio frequency receiver. Fig. 11 is a functional block diagram of a radio frequency receiver in a preferred embodiment of the present invention. The input signal is first amplified by the amplifier, and then filtered by the second filter. The filtered signal is combined with the signal from the receiving local oscillator, and the synthesized signal is filtered by the filter again to obtain the output of the RF receiver.

接着，由时间和频率同步装置420进行时间和频率同步，根据时域输出序列与参考序列的相关性，构造各对天线间以时间点为自变量的同步目标函数，使用同步目标函数的峰值信息估计时间同步点和频偏。下面对时间和频率同步装置420的工作原理加以具体阐述。Next, time and frequency synchronization is performed by the time and frequency synchronization device 420, and according to the correlation between the time domain output sequence and the reference sequence, a synchronization objective function with the time point as an argument between each pair of antennas is constructed, and the peak information of the synchronization objective function is used Estimate time synchronization point and frequency offset. The working principle of the time and frequency synchronization device 420 will be described in detail below.

当时间和频率完全同步时，由公式(22)到公式(26)可得，第l个符号时间第j根接收天线的接收信号可表示成When the time and frequency are fully synchronized, from formula (22) to formula (26), the received signal of the jth receiving antenna at the lth symbol time can be expressed as

${r r}_{j j}^{l l} ((n no)) = = {Σ Σ}_{i i = = 11}^{{n no}_{T T}} {Σ Σ}_{m m = = 00}^{L L - - 11} {h h}_{j j,, i i}^{l l} ((m m)) {s the s}_{i i}^{l l} ((n no - - m m)) + + {w w}_{j j}^{l l} ((n no)),, n no = = 00,, K K,, N N + + {N N}_{g g} - - 11 - - - - - - ((2727))$

由于l与n同为时间变量，有 $r_{j}^{l} (n) = r_{j} (n + l (N + N_{g}))$ 。为表述方便，令n′＝n+l(N+N_g)，于是n′是实际的OFDM符号时域采样点的序号：n′＝0，1，K。Since l and n are both time variables, we have $r_{j}^{l} (no) = r_{j} (no + l (N + N_{g}))$ . For the convenience of expression, let n'=n+l(N+N _g ), then n' is the serial number of the actual OFDM symbol time domain sampling point: n'=0, 1, K.

考虑到时间延迟和频率偏移，令第j根接收天线对第i根发射天线的接收延时为θ_j，i，设射频接收机归一化频偏为ε_j，i，i＝1，...，n_T，j＝1，...，n_R。第j根接收天线的接收信号可表示成Considering the time delay and frequency offset, let the receiving delay of the j-th receiving antenna to the i-th transmitting antenna be θ _j,i , and set the normalized frequency offset of the RF receiver as ε _j,i , i=1, ..., n _T , j=1, ..., n _R . The received signal of the jth receiving antenna can be expressed as

${r r}_{j j} (({n no}^{' '})) = = {Σ Σ}_{i i = = 11}^{{n no}_{T T}} {Σ Σ}_{m m = = 00}^{{N N}_{L L} - - 11} {h h}_{j j,, i i}^{l l} ((m m)) {s the s}_{i i} (({n no}^{' '} - - m m - - {θ θ}_{j j,, i i})) {e e}^{j j 22 {πϵ πϵ}_{j j,, i i} {n no}^{' '} / / N N} + + {w w}_{j j} (({n no}^{' '}))$

其中，g表示向下取整。Among them, g represents rounding down.

下面以第j根接收天线与第q根发射天线间同步为例对同步的具体算法进行说明，其它各对天线间方法与之类似。由公式(28)可得The specific algorithm of synchronization will be described below by taking the synchronization between the jth receiving antenna and the qth transmitting antenna as an example, and the methods between other pairs of antennas are similar. From formula (28) can get

${r r}_{j j} (({n no}^{' '})) = = {Σ Σ}_{m m = = 00}^{{N N}_{L L} - - 11} {h h}_{j j,, q q}^{l l} ((m m)) {s the s}_{q q} (({n no}^{' '} - - m m - - {θ θ}_{j j,, q q})) {e e}^{j j 22 {πϵ πϵ}_{j j,, q q} {n no}^{' '} / / N N}$

$+ + {Σ Σ}_{\underset{i i &NotEqual; &NotEqual; q q}{i i = = 11}}^{{n no}_{T T}} {Σ Σ}_{m m = = 00}^{{N N}_{L L} - - 11} {h h}_{j j,, i i}^{l l} ((m m)) {s the s}_{i i} (({n no}^{' '} - - m m - - {θ θ}_{j j,, i i})) {e e}^{j j 22 {πϵ πϵ}_{j j,, i i} {n no}^{' '} / / N N} + + {w w}_{j j} (({n no}^{' '}))$

$= = {h h}_{j j,, q q}^{l l} (({m m}_{max max})) {s the s}_{q q} (({n no}^{' '} - - {m m}_{max max} - - {θ θ}_{j j,, q q})) {e e}^{j j 22 {πϵ πϵ}_{j j,, q q} {n no}^{' '} / / N N} - - - - - - ((2929))$

$+ + {Σ Σ}_{\underset{m m &NotEqual; &NotEqual; {m m}_{max max}}{m m = = 00}}^{{N N}_{L L} - - 11} {h h}_{j j,, q q}^{l l} ((m m)) {s the s}_{q q} (({n no}^{' '} - - m m - - {θ θ}_{j j,, q q})) {e e}^{j j 22 {πϵ πϵ}_{j j,, q q} {n no}^{' '} / / N N}$

$+ + {Σ Σ}_{\underset{i i &NotEqual; &NotEqual; q q}{i i = = 11}}^{{n no}_{T T}} {Σ Σ}_{m m = = 00}^{{N N}_{L L}} {h h}_{j j,, i i}^{l l} ((m m)) {s the s}_{i i} (({n no}^{' '} - - m m - - {θ θ}_{j j,, i i})) {e e}^{j j 22 {πϵ πϵ}_{j j,, i i} {n no}^{' '} / / N N} + + {w w}_{j j} (({n no}^{' '}))$

其中，m_max是第l个OFDM符号时间发射天线q和接收天线j之间多径信道最强径时延。最强径定义为 $m_{\max} = \arg \max_{m} {E [{| h_{j, q}^{l} (m) |}^{2}]}$ ，E[g]表示求数学期望。将公式(20)代入公式(29)，r_j(n′)可以进一步表示为Among them, m _max is the strongest multipath channel delay between transmitting antenna q and receiving antenna j at the lth OFDM symbol time. The strongest diameter is defined as $m_{\max} = \arg \max_{m} {E. [{| h_{j, q}^{l} (m) |}^{2}]}$ , E[g] means seeking mathematical expectation. Substituting formula (20) into formula (29), r _j (n′) can be further expressed as

${r r}_{j j} (({n no}^{' '})) = = {h h}_{j j,, q q}^{l l} (({m m}_{max max})) {x x}_{q q} (({n no}^{' '} - - {m m}_{max max} - - {θ θ}_{j j,, q q})) {e e}^{j j 22 {πϵ πϵ}_{j j,, q q} {n no}^{' '} / / N N} - - - - - - ((3030))$

$+ + {I I}_{Data Data} + + {I I}_{ISI ISI} + + {I I}_{OtherAntenna OtherAntenna} + + {w w}_{j j} (({n no}^{' '}))$

其中，in,

${I I}_{Data Data} = = {h h}_{j j,, q q}^{l l} (({m m}_{max max})) {d d}_{q q} (({n no}^{' '} - - {m m}_{max max} - - {θ θ}_{j j,, q q})) {e e}^{j j 22 {πϵ πϵ}_{j j,, q q} {n no}^{' '} / / N N}$

${I I}_{ISI ISI} = = {Σ Σ}_{\underset{m m &NotEqual; &NotEqual; {m m}_{max max}}{m m = = 00}}^{{N N}_{L L} - - 11} {h h}_{j j,, q q}^{l l} ((m m)) {s the s}_{q q} (({n no}^{' '} - - m m - - {θ θ}_{j j,, q q})) {e e}^{j j 22 {πϵ πϵ}_{j j,, q q} {n no}^{' '} / / N N} - - - - - - ((3131))$

${I I}_{OtherAntenna OtherAntenna} = = {Σ Σ}_{\underset{i i &NotEqual; &NotEqual; q q}{i i = = 00}}^{{n no}_{T T}} {Σ Σ}_{m m = = 00}^{{N N}_{L L} - - 11} {h h}_{j j,, i i}^{l l} ((m m)) {s the s}_{i i} (({n no}^{' '} - - m m - - {θ θ}_{j j,, i i})) {e e}^{j j 22 {πϵ πϵ}_{j j,, q q} {n no}^{' '} / / N N}$

I_Data表示第q根发射天线最强径的数据对导频信号的干扰，I_ISI表示第q根发射天线其它各径对最强径信号的干扰。I_{Other Antenna}表示其它发射天线发射信号对第q根发射天线的干扰。令 $w_{j}^{%} (n^{'}) = I_{Data} + I_{ISI} + I_{OtherAntenna} + w_{j} (n^{'})$ ，则r_j(n′)可以表示为I _Data represents the interference of the data on the strongest path of the qth transmitting antenna to the pilot signal, and _IISI represents the interference of other paths of the qth transmitting antenna on the signal of the strongest path. I _{Other Antenna} indicates the interference of signals transmitted by other transmitting antennas to the qth transmitting antenna. make $w_{j}^{%} ({no}^{'}) = I_{Data} + I_{ISI} + I_{OtherAntenna} + w_{j} ({no}^{'})$ , then r _j (n′) can be expressed as

${r r}_{j j} (({n no}^{' '})) = = {h h}_{j j,, q q}^{l l} (({m m}_{max max})) {x x}_{q q} (({n no}^{' '} - - {m m}_{max max} - - {θ θ}_{j j,, q q})) {e e}^{j j 22 {πϵ πϵ}_{j j,, q q} {n no}^{' '} / / N N} + + {w w}_{j j}^{% %} (({n no}^{' '})) - - - - - - ((3232))$

根据前文结合图6对导频序列的描述，x_q ^l(n)除去循环前缀外的部分具有准周期性质，c_qα_t ^q表示第q根天线的时域导频序列的第t段，t＝0，K，M-1。其中c_q是第q根发射天线的PN序列，长度为N/M。令K＝N/M，于是x_q ^l(n)可以表示成According to the previous description of the pilot sequence in conjunction with Figure 6, the part of x _q ^l (n) except the cyclic prefix has quasi-periodic properties, c _q α _t ^q represents the t-th segment of the time-domain pilot sequence of the qth antenna, t=0, K, M-1. Among them, c _q is the PN sequence of the qth transmitting antenna, and the length is N/M. Let K=N/M, then x _q ^l (n) can be expressed as

对第q根发射天线设定一个长为N的滑动窗口，随时间变量n′不断变化构造：A sliding window of length N is set for the qth transmitting antenna, and the structure is continuously changed with the time variable n′:

${λ λ}_{j j,, q q} (({n no}^{' '})) = = {Σ Σ}_{t t = = 00}^{M m - - 11} [[(({Σ Σ}_{k k = = 00}^{K K - - 11} {(({α α}_{t t}^{q q}))}^{* *} {c c}_{q q}^{* *} ((k k)) {r r}_{j j} (({n no}^{' '} + + k k + + tK tK)))) - - - - - - ((3434))$

$\cdot &Center Dot; {(({Σ Σ}_{k k = = 00}^{K K - - 11} {(({α α}_{t t + + 11}^{q q}))}^{* *} {c c}_{q q}^{* *} ((k k)) {r r}_{j j} (({n no}^{' '} + + k k + + ((t t + + 11)) K K))))}^{* *}]]$

令η_j，q(n′)＝|λ_j，q(n′)|，则η_j，q(n′)即为同步目标函数。Let η _{j, q} (n')=|λ _{j, q} (n')|, then η _{j, q} (n') is the synchronization objective function.

有了同步目标函数，即可以利用同步目标函数的峰值信息估计时间同步点和频偏。一种最直接判定同步目标函数相关峰的方法是设定门限，当目标函数值超过门限时判为同步，对应采样点为时间同步点。With the synchronization objective function, the time synchronization point and frequency offset can be estimated by using the peak information of the synchronization objective function. One of the most direct ways to determine the correlation peak of the synchronization objective function is to set a threshold. When the objective function value exceeds the threshold, it is judged as synchronization, and the corresponding sampling point is the time synchronization point.

门限用自适应门限方式产生，其方法是将同步目标函数当前输出值与前N₁点均值的T_h倍比较，当大于等于时判为时间同步，判定条件如下式：The threshold is generated by an adaptive threshold method. The method is to compare the current output value of the synchronization objective function with T _h times the average value of the previous _N1 points. When it is greater than or equal to, it is judged as time synchronization. The judgment condition is as follows:

$| | {λ λ}_{j j,, q q} (({n no}^{' '})) | | &GreaterEqual; &Greater Equal; {T T}_{h h} \frac{{Σ Σ}_{i i = = 11}^{{N N}_{11}} | | {λ λ}_{j j,, q q} (({n no}^{' '} - - i i)) | |}{{N N}_{11}} - - - - - - ((3535))$

当时间同步精确完成时，目标函数η_j，q(n′)应出现最大峰值。对第一个OFDM符号来说，这个峰值应该在n′＝m_max+θ_j，q+N_g时出现。由公式(32)公式(34)可得，When the time synchronization is done precisely, the objective function η _j,q (n′) should have the largest peak. For the first OFDM symbol, this peak should appear when n' = m _max + θ _{j, q} + N _g . From formula (32) and formula (34), we can get,

此时at this time

${λ λ}_{j j,, q q} (({m m}_{max max} + + {θ θ}_{j j,, q q} + + {N N}_{g g}))$

$= = {Σ Σ}_{t t = = 00}^{M m - - 11} [[(({Σ Σ}_{k k = = 00}^{K K - - 11} {(({α α}_{t t}^{q q}))}^{* *} {c c}_{q q}^{* *} ((k k)) {r r}_{j j} (({m m}_{max max} + + {θ θ}_{j j,, q q} + + {N N}_{g g} + + k k + + tK tK))))$

$. . {(({Σ Σ}_{k k = = 00}^{K K - - 11} {(({α α}_{t t + + 11}^{q q}))}^{* *} {c c}_{q q}^{* *} ((k k)) {r r}_{j j} (({m m}_{max max} + + {θ θ}_{j j,, q q} + + {N N}_{g g} + + k k + + ((t t + + 11)) K K))))}^{* *}]]$

$= = {Σ Σ}_{t t = = 00}^{M m - - 11} [[(({Σ Σ}_{k k = = 00}^{K K - - 11} {(({α α}_{t t}^{q q}))}^{* *} {c c}_{q q}^{* *} ((k k)) (({h h}_{j j,, q q}^{l l} (({m m}_{max max})) {x x}_{q q} (({N N}_{g g} + + k k + + tK tK)) {e e}^{j j 22 {πϵ πϵ}_{j j,, q q} (({m m}_{max max} + + {θ θ}_{j j,, q q} + + {N N}_{g g} + + tK tK)) / / N N} - - - - - - ((3636))$

$+ + {w w}_{j j}^{% %} (({m m}_{max max} + + {θ θ}_{j j,, q q} + + {N N}_{g g} + + k k + + tK tK))))))$

$. . (({Σ Σ}_{k k = = 00}^{K K - - 11} {(({α α}_{t t + + 11}^{q q}))}^{* *} {c c}_{q q}^{* *} ((k k)) (({h h}_{j j,, q q}^{l l} (({m m}_{max max})) {x x}_{q q} (({N N}_{g g} + + k k + + ((t t + + 11)) K K)) {e e}^{j j 22 {πϵ πϵ}_{j j,, q q} (({m m}_{max max} + + {θ θ}_{j j,, q q} + + {N N}_{g g} + + k k + + ((t t + + 11)) K K)) / / N N}$

$+ + {w w}_{j j}^{% %} (({m m}_{max max} + + {θ θ}_{j j,, q q} + + {N N}_{g g} + + k k + + ((t t + + 11)) K K)))) {))}^{* *}]]$

由于 $x_{q} (n^{'}) = x_{q}^{l} (n^{'} - l (N + N_{g})),$

利用公式(33)，上式可以化简为because

x_{q} ({no}^{'}) = x_{q}^{l} ({no}^{'} - l (N + N_{g})),

Using formula (33), the above formula can be simplified as

${λ λ}_{j j,, q q} (({m m}_{max max} + + {θ θ}_{j j,, q q} + + {N N}_{g g})) = = {| | {h h}_{j j,, q q}^{l l} (({m m}_{max max})) | |}^{22} {Me Me}^{- - j j 22 {πϵ πϵ}_{j j,, q q} K K / / N N} {| | \frac{sin sin (({πϵ πϵ}_{j j,, q q} K K / / N N))}{sin sin (({πϵ πϵ}_{j j,, q q} / / N N))} | |}^{22} + + {w w}_{j j}^{' '} - - - - - - ((3737))$

其中，w_j′是所有加性的无关项之和。可见公式(37)在精确同步时有峰值输出，根据扩频原理，公式(37)的第一项取模后远远大于第二项的模，故忽略干扰项，于是在时间精确同步时刻，同步目标函数值为：where w _j ′ is the sum of all additive irrelevant terms. It can be seen that formula (37) has a peak output during precise synchronization. According to the principle of spread spectrum, the modulus of the first term of formula (37) is far greater than the modulus of the second term, so the interference term is ignored, so at the time of precise synchronization, The synchronization objective function value is:

${η η}_{j j,, q q} (({m m}_{max max} + + {θ θ}_{j j,, q q} + + {N N}_{g g})) = = | | {λ λ}_{j j,, q q} (({m m}_{max max} + + {θ θ}_{j j,, q q} + + {N N}_{g g})) | |$

$= = {| | {h h}_{j j,, q q}^{l l} (({m m}_{max max})) | |}^{22} M m {| | \frac{sin sin (({πϵ πϵ}_{j j,, q q} K K / / N N))}{sin sin (({πϵ πϵ}_{j j,, q q} / / N N))} | |}^{22} - - - - - - ((3838))$

$\approx \approx {| | {h h}_{j j,, q q}^{l l} (({m m}_{max max})) | |}^{22} {K K}^{22} M m {| | sin sin c c ((\frac{{πϵ πϵ}_{j j,, q q} K K}{N N})) | |}^{22}$

可见，目标函数与归一化频偏ε和最强径衰落有关。例如，若N＝1024，K＝128，则目标函数随ε变化的第一个零点在ε＝8取得。新一代移动通信中，载频为通常为3~3.5GHz，若载频为3.5GHz，设接收机晶振的稳定系数为10ppm(百万分之一)，OFDM子载波间隔为20KHz，则ε＝1.75。此时只要最强径不发生深衰落，时间同步目标函数峰值仍然很高，满足时间同步的要求。It can be seen that the objective function is related to the normalized frequency offset ε and the strongest path fading. For example, if N=1024, K=128, then the first zero point of the objective function changing with ε is obtained at ε=8. In the new generation of mobile communication, the carrier frequency is usually 3~3.5GHz. If the carrier frequency is 3.5GHz, the stability factor of the receiver crystal oscillator is set to 10ppm (one millionth), and the OFDM subcarrier interval is 20KHz, then ε= 1.75. At this time, as long as the strongest path does not undergo deep fading, the peak value of the time synchronization objective function is still high, which meets the requirements of time synchronization.

由于接收信号具有准周期性，由公式(32)，公式(33)和公式(34)可见目标函数η_j，q(n′)有多个峰值，这要求门限设定需相当精确，否则单纯的门限判定法就容易同步到旁瓣。为了获得精确的同步点，需要对该方法加以改进，设定一个搜索窗，改进后的方法步骤如下：首先找到满足公式(35)的n₁′点；在区间

中搜索η_j，q(n′)的最大值点n₂′；点n₂′即对应同步点(同步到最强径)。Due to the quasi-periodicity of the received signal, it can be seen from the formula (32), formula (33) and formula (34) that the objective function η _{j, q} (n′) has multiple peaks, which requires the threshold setting to be quite accurate, otherwise the simple The threshold judgment method is easy to synchronize to the side lobe. In order to obtain an accurate synchronization point, this method needs to be improved, and a search window is set. The steps of the improved method are as follows: firstly, find n ₁ ′ points satisfying the formula (35);

Search for the maximum value point n ₂ ′ of η _j,q (n′); point n ₂ ′ corresponds to the synchronization point (synchronization to the strongest path).

这种改进后的方法实现了对最强径的同步，但最强径未必是第一径，虽然最强径在通常情况下是第一径。由于循环前缀的作用，对OFDM的解调并不需要严格同步到第一径。但若不能同步到第一径，最强径前的多径能量不但不能利用，还会对信号解调产生干扰。为此，本发明一个优选实施例中提供了下述进行第一径同步的方法：先找到最强径；从最强径前N₂点开始向后搜索；设定判定以熬煎为目标函数比其前N₃点的均值的N₄倍大；找到满足所述判定条件的点即为第一径，否则最强径就是第一径。This improved method realizes the synchronization of the strongest path, but the strongest path is not necessarily the first path, although the strongest path is usually the first path. Due to the role of the cyclic prefix, OFDM demodulation does not need to be strictly synchronized to the first path. However, if it cannot be synchronized to the first path, the multipath energy before the strongest path will not only be unusable, but will also interfere with signal demodulation. For this reason, in a preferred embodiment of the present invention, the following method for synchronizing the first path is provided: find the strongest path first; start searching backward from the N ₂ points before the strongest path; The average value of the previous N ₃ points is N ₄ times larger; finding the point that meets the above judgment conditions is the first path, otherwise the strongest path is the first path.

这样做是因为即使最强径前面还有多径分量，但其增益对应的目标函数值不满足所述判定条件，说明其增益远小于最强径，可以忽略。This is because even if there are multipath components in front of the strongest path, the objective function value corresponding to its gain does not meet the above judgment conditions, indicating that its gain is much smaller than the strongest path and can be ignored.

假设各天线时间同步已完成。由(37)可见，ε_j，q可以用如下方法进行估计：It is assumed that the time synchronization of each antenna has been completed. It can be seen from (37) that ε _{j, q} can be estimated by the following method:

${\overset{^^}{ϵ ϵ}}_{j j,, q q} = = \frac{N N}{22 πK πK} \cdot \cdot arg arg (({λ λ}_{j j,, q q} (({m m}_{max max} + + {θ θ}_{j j,, q q} + + {N N}_{g g})))) - - - - - - ((3939))$

其中arg(g)是求相角运算。Where arg(g) is the phase angle operation.

根据上述时间和频率同步方法，本发明一个优选实施例中，时间和频率同步装置420的一种同步算法实施例的模块示意图如图10所示。其中模块1038，模块1039，模块1040，模块1041用于构造公式(34)中的λ_j，q(n′)，模块1042对λ_j，q(n′)取绝对值，得到同步目标函数|λ_i(k)|。之后，模块1043用于根据已知的|λ_i(k)|计算判决门限值，之后比较模块根据公式(35)进行判断，找到同步目标函数的峰值，初步确定同步点。当初步确定同步点以后，进一步通过模块1045寻找天线间第一径。同时模块1046对同步目标函数的峰值取相角，得到频偏估计值。According to the above time and frequency synchronization method, in a preferred embodiment of the present invention, a block diagram of a synchronization algorithm embodiment of the time and frequency synchronization device 420 is shown in FIG. 10 . Wherein module 1038, module 1039, module 1040, module 1041 are used for constructing λ _{j in the formula (34), q} (n '), module 1042 is taken absolute value to λ _{j, q} (n '), obtains synchronous objective function | λ _i (k)|. After that, the module 1043 is used to calculate the judgment threshold value according to the known |λ _i (k)|, and then the comparison module judges according to formula (35), finds the peak value of the synchronization objective function, and preliminarily determines the synchronization point. After the synchronization point is preliminarily determined, the first path between the antennas is further searched through module 1045 . At the same time, module 1046 takes a phase angle for the peak value of the synchronization objective function to obtain an estimated value of frequency offset.

采用本发明实施例的同步方法可以同时实现整数部分频偏估计和小数部分频偏估计。频偏估计范围是 $| \hat{ϵ} | \leq \frac{N}{2 K} .$ 设N＝1024，K＝128，则 $| \hat{ϵ} | \leq 4$ 。设载频为3.5GHz，接收机晶振的稳定系数为10ppm(百万分之一)，OFDM子载波间隔为20KHz，则ε＝1.75＜4，说明本发明的频率同步可以满足新一代移动通信的频率同步要求。The frequency offset estimation of the integer part and the frequency offset estimation of the fractional part can be realized simultaneously by adopting the synchronization method of the embodiment of the present invention. The frequency offset estimation range is $| \hat{ϵ} | \leq \frac{N}{2 K} .$ Suppose N=1024, K=128, then $| \hat{ϵ} | \leq 4$ . If the carrier frequency is 3.5GHz, the stability factor of the receiver crystal oscillator is 10ppm (one part per million), and the OFDM subcarrier spacing is 20KHz, then ε=1.75＜4, illustrating that the frequency synchronization of the present invention can meet the requirements of the new generation of mobile communications Frequency synchronization requirements.

在时间和频率同步过程中，可以利用发射和接收分集的方法来提高频率同步算法的精度。对于各对收发天线之间的频偏都相同的MIMO系统，有In the process of time and frequency synchronization, the method of transmitting and receiving diversity can be used to improve the accuracy of the frequency synchronization algorithm. For a MIMO system with the same frequency offset between each pair of transmit and receive antennas, we have

$\overset{^^}{ϵ ϵ} = = - - \frac{N N}{22 πK πK} \cdot &Center Dot; arg arg (({Σ Σ}_{q q = = 11}^{{n no}_{T T}} {Σ Σ}_{j j = = 11}^{{n no}_{R R}} {λ λ}_{j j,, q q} (({m m}_{max max} + + {θ θ}_{j j,, q q} + + {N N}_{g g})))) - - - - - - ((4040))$

经过第一次频率同步后，由于干扰的影响还会存在残余频偏，这可以通过二次频率同步来实现。假设第一次频率同步能够完全抵消整数部分频偏，在接收端将其补偿后，残余频偏可以通过利用较远的数据再做一次频偏估计来实现。将公式(34)改写成After the first frequency synchronization, there will still be residual frequency offset due to the influence of interference, which can be realized through the second frequency synchronization. Assuming that the first frequency synchronization can completely offset the frequency offset of the integer part, after the receiver compensates it, the residual frequency offset can be realized by using the farther data to do another frequency offset estimation. Rewrite formula (34) as

${λ λ}_{j j,, q q} (({n no}^{' '})) = = {Σ Σ}_{t t = = 00}^{M m - - P P - - 11} [[(({Σ Σ}_{k k = = 00}^{K K - - 11} {(({α α}_{t t}^{q q}))}^{* *} {c c}_{q q}^{* *} ((k k)) {r r}_{j j} (({n no}^{' '} + + k k + + tK tK))))$

$. . {(({Σ Σ}_{k k = = 00}^{K K - - 11} {(({α α}_{t t + + 11}^{q q}))}^{* *} {c c}_{q q}^{* *} ((k k)) {r r}_{j j} (({n no}^{' '} + + k k + + ((t t + + P P)) K K))))}^{* *}]] - - - - - - ((4141))$

用与第一次频偏估计类似的方法可以导出

的计算式：Using a method similar to the first frequency offset estimation, it can be derived

The calculation formula:

${\overset{^^}{ϵ ϵ}}_{j j,, q q} = = - - \frac{N N}{22 πPK πPK} \cdot \cdot arg arg (({λ λ}_{j j,, q q} (({m m}_{max max} + + {θ θ}_{j j,, q q} + + {N N}_{g g})))) - - - - - - ((4242))$

其中，P用于调整精度。P越大精度越高，但频偏估计范围也越小。于是

可表示成两部分的和：

{\hat{ϵ}}_{j, q} = {\hat{ϵ}}_{j, q}^{1} + {\hat{ϵ}}_{j, q}^{2},

和

分别表示第一次和第二次频偏估计的结果。Among them, P is used to adjust the precision. The larger P is, the higher the accuracy is, but the frequency offset estimation range is also smaller. then

Can be expressed as a sum of two parts:

{\hat{ϵ}}_{j, q} = {\hat{ϵ}}_{j, q}^{1} + {\hat{ϵ}}_{j, q}^{2},

and

Denote the results of the first and second frequency offset estimations, respectively.

上述目标函数的构造方法是针对各对收发天线分别构造目标函数，其中每根发射天线的时间同步都是基于对其它天线数据进行抑制实现的。当发射天线数较大时，干扰增大，性能会有所下降，由公式(38)知，可以通过增大K使目标函数峰值更大。还通过使用如下的分集增益重新构造同步函数来提高同步精度。下面具体说明所述特定条件。The method for constructing the above objective function is to construct the objective function for each pair of transmitting and receiving antennas respectively, wherein the time synchronization of each transmitting antenna is realized based on suppressing the data of other antennas. When the number of transmitting antennas is large, the interference will increase and the performance will decrease. According to formula (38), the peak value of the objective function can be made larger by increasing K. Synchronization accuracy is also improved by reconstructing the synchronization function using the diversity gain as follows. The specific conditions are specifically described below.

例如，当同一根发射天线到各接收天线的延迟相同时，可以利用接收分集，此时第q根发射天线的同步目标函数为：For example, when the delay from the same transmit antenna to each receive antenna is the same, receive diversity can be utilized. At this time, the synchronization objective function of the qth transmit antenna is:

${η η}_{q q} (({n no}^{' '})) = = | | {Σ Σ}_{j j = = 11}^{{n no}_{R R}} {λ λ}_{j j,, q q} (({n no}^{' '})) | | - - - - - - ((4343))$

同理，当各发射天线到第j根接收天线的延迟相同时，可以利用发射分集，此时第j根接收天线的同步目标函数为：Similarly, when the delays from each transmit antenna to the j-th receive antenna are the same, transmit diversity can be used. At this time, the synchronization objective function of the j-th receive antenna is:

${η η}_{j j} (({n no}^{' '})) = = | | {Σ Σ}_{q q = = 11}^{{n no}_{T T}} {λ λ}_{j j,, q q} (({n no}^{' '})) | | - - - - - - ((4444))$

当各发射天线到各接收天线时延相同，并且各对收发天线之间的频偏都相同时，可以进一步利用发射分集，同步目标函数为：When the time delay from each transmit antenna to each receive antenna is the same, and the frequency offset between each pair of transmit and receive antennas is the same, transmit diversity can be further utilized, and the synchronization objective function is:

$η η (({n no}^{' '})) = = | | {Σ Σ}_{q q = = 11}^{{n no}_{T T}} {Σ Σ}_{j j = = 11}^{{n no}_{R R}} {λ λ}_{j j,, q q} (({n no}^{' '})) | | - - - - - - ((4545))$

特别地，对于各对收发天线之间频偏都相同的MIMO系统，若果满足N为偶数，频率同步有另一种方法：In particular, for a MIMO system with the same frequency offset between each pair of transmitting and receiving antennas, if N is an even number, there is another method for frequency synchronization:

由公式(18)可见N点时域导频序列具有如下性质：当p_i为奇数时，时域导频序列的前N/2点等于后N/2点乘以-1；当p_i为偶数时，时域导频序列的前N/2点与后N/2点相同。It can be seen from formula (18) that the N-point time-domain pilot sequence has the following properties: when p _i is an odd number, the first N/2 points of the time-domain pilot sequence are equal to the rear N/2 points multiplied by -1; when p _i is When the number is even, the first N/2 points of the time-domain pilot sequence are the same as the last N/2 points.

发射天线数是受到导频子载波间隔M限制的，理论上发射天线数n_T≤M，当n_T＝M时不再有数据子载波，为了不降低频谱利用率，要求至少满足n_T≤M/2。The number of transmitting antennas is limited by the pilot subcarrier spacing M. In theory, the number of transmitting antennas n _T ≤ M, when n _T = M, there are no data subcarriers. In order not to reduce the spectrum utilization, it is required to meet at least n _T ≤ M/2.

于是预定导频图案中发射天线的第一个导频子载波位置相对于OFDM子载波原点的位移，即第i根天线的导频子载波位置p_i的集合，i＝0，...，n_T-1，可以在如下两个集合中任选其一：Then the displacement of the first pilot subcarrier position of the transmitting antenna relative to the origin of the OFDM subcarrier in the predetermined pilot pattern, that is, the set of pilot subcarrier positions p _i of the i-th antenna, i=0,..., n _T -1, you can choose one of the following two sets:

p_i∈{0，2，...，M-2}或p_i∈{1，3，...，M-1} (46)p _i ∈ {0, 2, ..., M-2} or p _i ∈ {1, 3, ..., M-1} (46)

即：p_i必须全为奇数或全为偶数。That is: p _i must be all odd numbers or all even numbers.

设时间同步点为n₁′，假设各天线时间同步已完成，则频偏计算式为：Assuming that the time synchronization point is n ₁ ′, assuming that the time synchronization of each antenna has been completed, the frequency offset calculation formula is:

$\overset{^^}{ϵ ϵ} = = \frac{- - N N}{22 πKM πKM / / 22} arg arg (({Σ Σ}_{t t = = 00}^{M m / / 22 - - 11} {Σ Σ}_{k k = = 00}^{K K - - 11} {((- - 11))}^{f f} r r (({n no}_{11}^{' '} + + k k + + tK tK)) {r r}^{* *} (({n no}_{11}^{' '} + + k k + + ((t t + + M m / / 22)) K K)))) - - - - - - ((4747))$

其中，

in,

公式(46)可以进一步写成Equation (46) can be further written as

$\overset{^^}{ϵ ϵ} = = - - \frac{11}{π π} arg arg (({Σ Σ}_{k k = = 00}^{N N / / 22 - - 11} {((- - 11))}^{f f} r r (({n no' '}_{11} + + k k)) {r r}^{* *} (({n no' '}_{11} + + k k + + N N / / 22)))) - - - - - - ((4848))$

该方法完全利用了分集增益，发射天线数越多同步越精确，频偏估计范围是 $| \hat{ϵ} | \leq 1 .$ This method fully utilizes the diversity gain, the more the number of transmit antennas, the more accurate the synchronization, and the frequency offset estimation range is $| \hat{ϵ} | \leq 1 .$

得到时间同步点和频偏信息后，时间和频率同步装置420所估计出的时间同步点和频偏信息被传送到接收本振(417，418，419)和OFDM解调器(422，423，424)。OFDM解调器结合所述估计出的时间同步点和频偏信息对所述接收信号的时域输出序列进行OFDM解调，得到还原的各天线数据信息。之后，多输入多输出解码器425对还原的各天线的数据信息先利用其中的频域导频进行导引辅助信道估计，再进行空时频解码，得到恢复的原始的输入信息。After obtaining the time synchronization point and frequency offset information, the time synchronization point and frequency offset information estimated by the time and frequency synchronization device 420 are transmitted to the receiving local oscillator (417, 418, 419) and OFDM demodulator (422, 423, 424). The OFDM demodulator performs OFDM demodulation on the time-domain output sequence of the received signal in combination with the estimated time synchronization point and frequency offset information, to obtain restored data information of each antenna. After that, the MIMO decoder 425 uses the frequency-domain pilots in the restored data information of each antenna to guide and assist channel estimation, and then performs space-time-frequency decoding to obtain the restored original input information.

本发明一个优选实施例中，OFDM解调器的原理框图如图12所示。与图8中的OFDM调制器相对应，图12中的OFDM解调器包括一个去掉循环前缀模块和一个DFT模块。输入信号首先被去掉循环前缀然后进行DFT，产生输出信号。In a preferred embodiment of the present invention, the functional block diagram of the OFDM demodulator is shown in FIG. 12 . Corresponding to the OFDM modulator in FIG. 8 , the OFDM demodulator in FIG. 12 includes a cyclic prefix removal module and a DFT module. The input signal is first removed from the cyclic prefix and then subjected to DFT to generate the output signal.

由前述实施例可以看出，本发明中，通过设计特殊的各发射天线的导频序列，使系统能够以更低的开销实现时间和频率同步，各天线只需要发送一个在频域插入导频的OFDM符号，不需要另外的训练序列，即可在完成同步任务同时，并完成MIMO信道估计和承载数据业务的传输。所述时间和频率同步的方法适用于复杂的实际的MIMO系统，各发射天线到各接收天线时延可以不同，并且各对收发天线之间的频偏也可以不同，在频偏或时延相同时还可以利用天线的分集增益进一步提高同步性能。与传统的只能捕获到最强径的同步方法相比，本发明实施例提供了捕获第一径的方法。同时，本发明与现有方法相比，还具有适用于多径信道和无需对信噪比进行估计的优点。It can be seen from the foregoing embodiments that in the present invention, by designing special pilot sequences for each transmitting antenna, the system can realize time and frequency synchronization with lower overhead, and each antenna only needs to send a pilot sequence inserted in the frequency domain OFDM symbols, without additional training sequence, it can complete the synchronization task and complete the MIMO channel estimation and the transmission of the bearer data service at the same time. The method for time and frequency synchronization is applicable to complex actual MIMO systems. The time delays from each transmitting antenna to each receiving antenna can be different, and the frequency offset between each pair of receiving and transmitting antennas can also be different. At the same time, the diversity gain of the antenna can be used to further improve the synchronization performance. Compared with the traditional synchronous method that can only capture the strongest path, the embodiment of the present invention provides a method for capturing the first path. At the same time, compared with the existing method, the present invention also has the advantages of being applicable to multi-path channels and not needing to estimate the signal-to-noise ratio.

要说明的是，在结合图4对本发明的具体实施方式进行说明的过程中，作为示例，各个环节的射频处理一、射频处理二、OFDM调制器、OFDM解调器、接收本振等装置只表现了一定的个数(例如只标示了2个发射器602，603)，但是此种标示并不能作为对本发明范围的限制。本发明的范围将包括所有满足本发明条件的发射天线和接收天线数量的组合。特别地，单输入天线或单输出天线的情况作为本发明的一个特例，本发明的各个方面涉及的方法和系统完全适用于所述情况，因此也将落入本发明的保护范围之内。It should be noted that, in the process of describing the specific implementation of the present invention in conjunction with FIG. 4, as an example, the radio frequency processing one, radio frequency processing two, OFDM modulator, OFDM demodulator, receiving local oscillator and other devices in each link are only Certain numbers are shown (for example only 2 emitters 602, 603 are labeled), but such labeling is not to be taken as limiting the scope of the invention. The scope of the present invention shall include all combinations of the number of transmitting antennas and receiving antennas satisfying the conditions of the present invention. In particular, the case of single-input antenna or single-output antenna is a special case of the present invention, and the methods and systems involved in various aspects of the present invention are fully applicable to the case, and therefore also fall within the protection scope of the present invention.

Claims

1. A method for time and frequency synchronization in orthogonal frequency division multiplexing communications, characterized by:

original information after carrier modulation is serial-parallel converted into n_TStrip tributary data stream, n_TIs the number of transmit antennas;

constructing a frequency domain pilot frequency sequence of a transmitting antenna;

mapping the branch data stream of the transmitting antenna and the frequency domain pilot frequency sequence to the corresponding data position of a preset frequency domain pilot frequency pattern to generate data information;

the data information is transmitted by a transmitting antenna after OFDM modulation;

the receiving antenna receives electromagnetic waves and generates a time domain output sequence;

constructing a synchronous target function between the antennas according to the correlation between the time domain output sequence and the reference sequence;

estimating a time synchronization point and a frequency offset by using peak value information of a synchronization objective function;

and completing the time and frequency synchronization according to the time synchronization point and the frequency offset information.

2. The method of claim 1, wherein the step of constructing the frequency-domain pilot sequences for the transmit antennas comprises:

selecting a plurality of sequences as a basic sequence;

repeatedly arranging the basic sequences, and multiplying each section of basic sequence by a corresponding coefficient to form a time domain pilot frequency sequence;

and performing discrete Fourier transform on the time domain pilot sequence to obtain a frequency domain pilot sequence.

3. The method according to claim 1, wherein in the predetermined frequency domain pilot pattern, the frequency domain pilot of each antenna is located at the position of equally spaced subcarriers, and the subcarriers are called pilot subcarriers; the positions of the pilot frequency sub-carriers of the antennas are different from each other; the data information is located at a position outside the frequency band containing the pilot subcarriers.

4. The method of claim 1, wherein the step of estimating the time synchronization point and the frequency offset using the peak information of the synchronization objective function comprises:

judging a correlation peak of a synchronization target function by using a self-adaptive threshold method, and estimating a time synchronization point according to corresponding peak value information; performing first frequency offset estimation by taking a phase angle from a peak value of a synchronous target function by using the obtained time synchronization point; and performing second frequency offset estimation by using data farther away from the current moment than the first frequency offset estimation by adopting the same method as the first frequency offset estimation.

5. The method of claim 1, wherein the synchronization objective function is constructed using receive diversity if the time delay from the same transmit antenna to each receive antenna is the same; if the time delay from each transmitting antenna to the same receiving antenna is the same, then using transmit diversity; if the time delay and frequency offset are the same between each pair of transmit and receive antennas, full diversity, including receive diversity and transmit diversity, is used.

6. The method of claim 3, wherein when the frequency offsets between the transceiving antennas of each pair are the same and the length of the time domain pilot sequence is even, the predetermined frequency domain pilot pattern is designed such that the displacements of the first pilot subcarrier positions of the respective transmitting antennas relative to the origin of the OFDM subcarriers are all odd numbers or all even numbers; and a synchronization objective function is constructed with full diversity gain.

7. The method of claim 4, wherein the time synchronization point is estimated by determining a sampling point where a peak of the synchronization objective function is located as the time synchronization point.

8. The method of claim 4, wherein the step of estimating a time synchronization point comprises: and setting a search window in a specific range near a sampling point where a peak point of the synchronous target function is located, and searching a maximum point of the synchronous target function in the search window, wherein the maximum point corresponds to a synchronous point which is synchronized to the strongest path between the antennas.

9. The method of claim 8, wherein the specific range is a length of each half of the time domain pilot sequence around the sampling point where the peak of the synchronous objective function is located.

10. The method of claim 8, wherein after finding the strongest path synchronization point between the antennas, further comprising the step of finding a first path between the antennas by:

comparing the synchronization target function with the former N₃N of the mean of the points₄Multiplying the size as a first diameter judgment condition; n before the sampling point corresponding to the strongest path₂The point starts to search backwards until the sampling point corresponding to the strongest path is self; if there is a sampling point satisfying the first path determination condition, the sampling point corresponds to the first path, otherwise the strongest path is the first path, where N is₂、N₃、N₄Are the corresponding coefficients.

11. The method of claim 1, wherein the step of constructing the frequency-domain pilot sequences of the transmitting antennas further comprises: a cyclic prefix is added before the pilot sequence.

12. An orthogonal frequency division multiplexing communication system capable of realizing time and frequency synchronization, characterized in that,

its transmitting end includes:

a multiple-input multiple-output encoder for encoding the original input information and generating data information for each antenna from the frequency domain pilot pattern,

at least one OFDM modulator for OFDM modulating data information for each antenna to generate OFDM symbols,

and a radio frequency transmitting part for performing radio frequency processing on the OFDM symbol and transmitting the OFDM symbol through a transmitting antenna;

its receiving end includes:

a radio frequency receiving portion that receives a radio frequency signal and performs radio frequency processing to generate a time domain output sequence of the received signal,

a time and frequency synchronization device which constructs a synchronization objective function with a time point as an independent variable among each pair of antennas according to the correlation between the time domain output sequence and the local sequence and estimates a time synchronization point and a frequency offset by using peak information of the synchronization objective function,

at least one OFDM demodulator for OFDM demodulating the time domain output sequence of the received signal by combining the estimated time synchronization point and the frequency offset information to obtain the restored data information of each antenna,

and a multi-input multi-output decoder for performing pilot-assisted channel estimation on the restored data information of each antenna by using the frequency domain pilot frequency and then performing space-time-frequency decoding to obtain the restored original input information.

13. The orthogonal frequency division multiplexing communication system capable of achieving time and frequency synchronization according to claim 12, wherein the multiple-input multiple-output encoder comprises:

a modulation module for carrier modulating the original input information to produce a modulated data stream,

original information after carrier modulation is serial-parallel converted into n_TSerial-to-parallel conversion module for a strip-tributary data stream, n of which_TIn order to be able to transmit the number of antennas,

and the pilot mapping module part is used for constructing frequency domain pilot sequences of all the transmitting antennas and mapping the branch data streams of all the transmitting antennas and the frequency domain pilot sequences to corresponding data positions of a preset frequency domain pilot pattern together so as to generate data information of all the transmitting antennas.

14. The orthogonal frequency division multiplexing communication system that can achieve time and frequency synchronization according to claim 12,

each OFDM modulator comprises: an IDFT device for performing inverse discrete Fourier transform on data information of an antenna, and a cyclic prefix adding device for adding a cyclic prefix before a pilot sequence output by the IDFT device;

each OFDM demodulator comprises: a cyclic prefix removing means for removing the cyclic prefix, and a DFT means for performing discrete fourier transform.

15. The orthogonal frequency division multiplexing communication system capable of achieving time and frequency synchronization according to claim 12, wherein the time and frequency synchronization means comprises:

a correlation module for performing correlation processing on the signal inputted from the radio frequency receiving section,

a delay module for delaying an output of the correlation module,

a conjugate extracting module for extracting conjugate from the output of the correlation module,

a multiplication module for performing multiplication operation on the outputs of the delay module and the conjugate taking module,

a summation module for summing the sequence values output by the multiplication module,

a modulus module for performing modulus processing on the output of the summation module,

a threshold module for calculating a threshold according to the output of the modulus module,

a comparison module for comparing the output of the modulus taking module and the threshold module;

a time synchronization module for finding a first path according to the output of the comparison module and further obtaining a time synchronization point;

and the frequency offset estimation module is used for obtaining a phase angle finger according to the output of the comparison module so as to obtain a frequency offset estimation value.