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Microwave Passive Components, 2nd Edition
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Microwave Passive Components, 2nd Edition

A special issue of Micromachines (ISSN 2072-666X). This special issue belongs to the section "E：Engineering and Technology".

Deadline for manuscript submissions: 20 February 2025 | Viewed by 6799

Special Issue Editors

Dr. Guoxiang Shu

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Guest Editor
College of Electronics and Information Engineering, Shenzhen University, Shenzhen 518060, China
Interests: vacuum electronic devices; millimeter-wave/THz passive devices; dielectric microwave measurement
Special Issues, Collections and Topics in MDPI journals
Prof. Dr. Guo Liu

E-Mail Website
Guest Editor
School of Electronic Science and Engineering, University of Electronic Science and Technology of China, Chengdu 611731, China
Interests: vacuum electronic devices; passive pulse compressor; microwave biosensor; dielectric microwave measurement
Special Issues, Collections and Topics in MDPI journals
Dr. Dian Zhang

E-Mail Website
Guest Editor
College of Advanced Interdisciplinary Studies, National University of Defense Technology, Changsha 410073, China
Interests: high power microwave devices; microwave mode converters; millimeter wave sources
Dr. Luqi Zhang

E-Mail Website
Guest Editor
Institute of Applied Electronics, China Academy of Engineering Physics, Mianyang 621900, China
Interests: vacuum electronic devices; microwave passive components; terahertz transmission line; terahertz micromachining

Special Issue Information

Dear Colleagues,

Today, micro-, millimeter and terahertz wave devices and systems have been widely used in various aspects of life, such as the 5G/6G communication, vehicle imaging radar, medical, bio-science, security, etc. As is well-known, microwave passive components play an important role in the design and application of these devices and systems. Especially with the development of advanced machining technologies, such as the micro-electro-mechanical system (MEMS), 3D printing, and micro-/nano-machining, the machining accuracy and ability of the passive components have been improved greatly. In addition, in the past two decades, novel concepts and mechanisms have been continually introduced or proposed from other fields, including the meta-material, vortex electromagnetic wave, and spoof surface plasmon. This has made the microwave passive devices/components enter a new stage controlled by information coding. That means the performance of microwave passive devices still has great potential in the future, which may contribute to the miniaturization and integration of RF circuits and devices. Prof. Guo Liu has organized a Special Issue entitled “Microwave Passive Components”, which received a strong response within the field. Therefore, this Special Issue, entitled “Microwave Passive Components, 2nd Edition”, was created, devoted to continue exploring for research papers, short communications, and review articles focusing on the theory, modeling, simulation, measurement and applications of passive components, circuits, devices and systems in the microwave, millimeter-wave and terahertz-wave bands.

We look forward to receiving your contributions to this Special Issue.

Dr. Guoxiang Shu
Dr. Guo Liu
Dr. Dian Zhang
Dr. Luqi Zhang
Guest Editors

Manuscript Submission Information

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Submitted manuscripts should not have been published previously, nor be under consideration for publication elsewhere (except conference proceedings papers). All manuscripts are thoroughly refereed through a single-blind peer-review process. A guide for authors and other relevant information for submission of manuscripts is available on the Instructions for Authors page. Micromachines is an international peer-reviewed open access monthly journal published by MDPI.

Please visit the Instructions for Authors page before submitting a manuscript. The Article Processing Charge (APC) for publication in this open access journal is 2600 CHF (Swiss Francs). Submitted papers should be well formatted and use good English. Authors may use MDPI's English editing service prior to publication or during author revisions.

Keywords

  • theory, modeling, fabrication, measurement and applications
  • microwave, millimeter and terahertz wave passive component/devices applied in the communication, radar and some other systems
  • passive component/devices in antenna, filters, biosensors, vacuum electronic devices, pulse compressor, particle accelerator, etc.
  • other work related to microwave devices

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Related Special Issue

Published Papers (6 papers)

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Research

11 pages, 2762 KiB  
Article
Two CMOS Wilkinson Power Dividers Using High Slow-Wave and Low-Loss Transmission Lines
by Chatrpol Pakasiri, Wei-Sen Teng and Sen Wang
Micromachines 2024, 15(8), 1009; https://doi.org/10.3390/mi15081009 - 5 Aug 2024
Viewed by 629
Abstract
This work presents two Wilkinson power dividers (WPDs) using multi-layer pseudo coplanar waveguide (PCPW) structures. The PCPW-based WPDs were designed, implemented, and verified in a standard 180 nm CMOS process. The proposed PCPW features high slow-wave and low-loss performances compared to other common [...] Read more.
This work presents two Wilkinson power dividers (WPDs) using multi-layer pseudo coplanar waveguide (PCPW) structures. The PCPW-based WPDs were designed, implemented, and verified in a standard 180 nm CMOS process. The proposed PCPW features high slow-wave and low-loss performances compared to other common transmission lines. The two WPDs are based on the same PCPW structure parameters in terms of line width, spacing, and used metal layers. One WPD was realized in a straight PCPW-based layout, and the other WPD was realized in a meandered PCPW-based layout. Both the two WPDs worked up to V-band frequencies, as expected, which also demonstrates that the PCPW guiding structure is less susceptible to the effects of meanderings on the propagation constant and characteristic impedance. The meandered design shows that the measured insertion losses were about 5.1 dB, and its return losses were better than 17.5 dB at 60 GHz. In addition, its isolation, amplitude imbalance, and phase imbalance were 18.5 dB, 0.03 dB, and 0.4°, respectively. The core area was merely 0.2 mm × 0.23 mm, or 1.8 × 10−3λo2. Full article
(This article belongs to the Special Issue Microwave Passive Components, 2nd Edition)
►▼ Show Figures

Figure 1

Figure 1
<p>A Conventional WPD topology using two 70.7-Ω quarter-wavelength transmission lines and one 100-Ω isolation resistor.</p> Full article ">Figure 2
<p>Cross-section of a standard 0.18 μm CMOS process.</p> Full article ">Figure 3
<p>Cross-section view of (<b>a</b>) the thin-film microstrip line, (<b>b</b>) coplanar waveguide, and (<b>c</b>) conductor-back coplanar waveguide.</p> Full article ">Figure 4
<p>(<b>a</b>) Cross-sectional view and (<b>b</b>) 3D structure of the PCPW.</p> Full article ">Figure 5
<p>(<b>a</b>) Simulated <span class="html-italic">Z<sub>c</sub></span> of the PCPW structure. Simulated (<b>b</b>) slow-wave factor (SWF) and (<b>c</b>) attenuation results of 70.7-Ω CBCPW, CPW, TFMSL, and PCPW structures.</p> Full article ">Figure 6
<p>Chip photo of the two proposed WPDs. (<b>a</b>) Straight PCPW-based layout and (<b>b</b>) meandered PCPW-based layout.</p> Full article ">Figure 7
<p>Simulated and measured results: (<b>a</b>) insertion and return losses, (<b>b</b>) isolation, and (<b>c</b>) phase and amplitude imbalances of the straight PCPW-based WPD.</p> Full article ">Figure 7 Cont.
<p>Simulated and measured results: (<b>a</b>) insertion and return losses, (<b>b</b>) isolation, and (<b>c</b>) phase and amplitude imbalances of the straight PCPW-based WPD.</p> Full article ">Figure 8
<p>Simulated and measured results: (<b>a</b>) insertion and return losses, (<b>b</b>) isolation, and (<b>c</b>) phase and amplitude imbalances of the meandered PCPW-based WPD.</p> Full article ">Figure 8 Cont.
<p>Simulated and measured results: (<b>a</b>) insertion and return losses, (<b>b</b>) isolation, and (<b>c</b>) phase and amplitude imbalances of the meandered PCPW-based WPD.</p> Full article ">
18 pages, 6136 KiB  
Article
An Electronically Reconfigurable Highly Selective Stop-Band Ultra-Wideband Antenna Applying Electromagnetic Bandgaps and Positive-Intrinsic-Negative Diodes
by Anees Abbas, Niamat Hussain, Md. Abu Sufian, Wahaj Abbas Awan, Jaemin Lee and Nam Kim
Micromachines 2024, 15(5), 638; https://doi.org/10.3390/mi15050638 - 9 May 2024
Viewed by 771
Abstract
In this article, an ultra-wideband (UWB) antenna featuring two reconfigurable quasi-perfect stop bands at WLAN (5.25–5.75 GHz) and lower 5G (3.4–3.8 GHz) utilizing electromagnetic bandgaps (EBGs) and positive-intrinsic-negative (P-I-N) diodes is proposed. A pair of EBG structures are applied to generate sharp notch [...] Read more.
In this article, an ultra-wideband (UWB) antenna featuring two reconfigurable quasi-perfect stop bands at WLAN (5.25–5.75 GHz) and lower 5G (3.4–3.8 GHz) utilizing electromagnetic bandgaps (EBGs) and positive-intrinsic-negative (P-I-N) diodes is proposed. A pair of EBG structures are applied to generate sharp notch bands in the targeted frequency spectrum. Each EBG creates a traditional notch, while two regular notches are combined to make a quasi-perfect, sharp, notch band. Four P-I-N diodes are engraved into the EBG structures to enable notch band reconfigurability. By switching the operational condition of the four diodes, the UWB antenna can dynamically adjust its notching characteristics to enhance its adaptability to various communication standards and applications. The antenna can be reconfigured as a UWB (3–11.6 GHz) without any notch band, a UWB with a single sharp notch (either at WLAN or 5G), or a UWB with two quasi-perfect notch bands. Moreover, the antenna’s notch bands can also be switched from a traditional notch to a quasi-perfect notch and vice versa. To confirm the validity of the simulated outcomes, the proposed reconfigurable UWB antenna is fabricated and measured. The experimental findings are aligned closely with simulation results, and the antenna offers notch band reconfigurability. The antenna shows a consistently favorable radiation pattern and gain. The dimension of the presented antenna is 20 × 27 × 1.52 mm3 (0.45 λc × 0.33 λc × 0.025 λc, where λc is the wavelength in free space). Full article
(This article belongs to the Special Issue Microwave Passive Components, 2nd Edition)
►▼ Show Figures

Figure 1

Figure 1
<p>Representation of regular notch bands and quasi-perfect notch bands in a UWB antenna.</p> Full article ">Figure 2
<p>Equivalent circuit of EBG structure.</p> Full article ">Figure 3
<p>(<b>a</b>) The unit cell of EBG designed for the WLAN band and (<b>b</b>) the reflection phase of the EBG at the WLAN band.</p> Full article ">Figure 4
<p>The schematic diagram shows the EBG structure connected with the patch.</p> Full article ">Figure 5
<p>The configuration of sharp dual notch UWB antenna.</p> Full article ">Figure 6
<p>The |S<sub>11</sub>| results of the sharp dual quasi-perfect notch-band antenna.</p> Full article ">Figure 7
<p>Equivalent circuit of a P-I-N diode: (<b>a</b>) the DC circuit diagram of P-I-N diode, (<b>b</b>) forward-biased, (<b>c</b>) reverse-biased.</p> Full article ">Figure 8
<p>The geometry of the proposed dual-notch-band UWB antenna.</p> Full article ">Figure 9
<p>Reflection coefficient of the antenna (<b>a</b>) without any stop band and (<b>b</b>) coefficient with WLAN-restricted frequency range.</p> Full article ">Figure 10
<p>The reflection coefficient of the antenna (<b>a</b>) with a lower 5G rejection band. (<b>b</b>) Notch bands at lower 5G GHz and WLAN bands.</p> Full article ">Figure 11
<p>The reflection antenna when (<b>a</b>) diodes 1 and 3 are OFF and (<b>b</b>) diodes 2 and 4 are OFF.</p> Full article ">Figure 12
<p>Notch bandwidth flexibility: (<b>a</b>) single conventional notch band at WLAN band by setting three diodes to OFF and diode 4 to ON, and (<b>b</b>) a sharp notch band and a conventional notch band when three diodes are in ON state.</p> Full article ">Figure 13
<p>The fabricated reconfigurable UWB antenna: (<b>a</b>) frontside, (<b>b</b>) backside, (<b>c</b>) antenna with controller circuitry.</p> Full article ">Figure 14
<p>Radiation measurement setup of fabricated UWB antenna.</p> Full article ">Figure 15
<p>The overall efficiency of the proposed reconfigurable antenna.</p> Full article ">Figure 16
<p>Simulated versus measured gains of the proposed antenna.</p> Full article ">Figure 17
<p>Analyzed simulated radiation behavior in the E- and-H-planes at frequencies 4.5, 6.5, 8.5, and 11.5 GHz.</p> Full article ">Figure 18
<p>Simulated and observed results with (<b>a</b>) a sharp notch at 5G sub-6 GHz band, (<b>b</b>) a single sharp notch at WLAN band, and (<b>c</b>) dual rejected bands.</p> Full article ">Figure 19
<p>Simulated and observed results of an antenna: (<b>a</b>) single conventional notch band at 5G sub-6 GHz when P-I-N diode 1 is forward-biased, (<b>b</b>) dual conventional notch bands at WLAN band and 5G sub-6 GHz when P-I-N diodes 1 and 3 are forward-biased, and (<b>c</b>) a sharp notch band and a conventional notch band when all three diodes are forward-biased.</p> Full article ">
12 pages, 34515 KiB  
Article
A Compact Broadband Power Combiner for High-Power, Continuous-Wave Applications
by Zihan Yang, Qiang Zhang, Kelin Zhou, Lishan Zhao and Jun Zhang
Micromachines 2024, 15(2), 207; https://doi.org/10.3390/mi15020207 - 30 Jan 2024
Viewed by 1207
Abstract
A compact broadband combiner with a high power capacity and a low insertion loss, which is especially useful for solid-state power sources where multi-way power synthesis is needed, was designed and experimentally investigated. The combiner could combine the microwave signals of sixteen terminals [...] Read more.
A compact broadband combiner with a high power capacity and a low insertion loss, which is especially useful for solid-state power sources where multi-way power synthesis is needed, was designed and experimentally investigated. The combiner could combine the microwave signals of sixteen terminals into a single one and was based on a radial-line waveguide whose circumferential symmetry benefited the amplitude and phase consistency of the combiner. Simulation and experimental results showed that the prototype device, designed for S-band applications, exhibited a reflection coefficient S1,1 < −20 dB in the range of 2.06–2.93 GHz, which corresponds to a relative bandwidth of approximately 34.6%. At 2.45 GHz, the phase imbalance was ±4.5° and the 16-way transmission coefficient was concentrated around −12.0~−12.3 dB. The insertion loss of the device at ambient and elevated temperatures was simulated and experimentally verified, which is of importance for the design of similar high-power microwave combiners. High-power tests proved that even without enforced wind or liquid cooling, the device can handle continuous power (CW) of at least 3.9 kW, which can be much enhanced by taking regular cooling measures. The combined features of the designed combiner suggest promising applications for power synthesis in high-power, solid-state RF sources. Full article
(This article belongs to the Special Issue Microwave Passive Components, 2nd Edition)
►▼ Show Figures

Figure 1

Figure 1
<p>Illustration of the of the proposed power combiner.</p> Full article ">Figure 2
<p>Profile view of the power combiner: (<b>a</b>) overall sectional structure and (<b>b</b>) transition structure of the power combiner. The green area represents the medium within the joint, while the gray area represents the metal. (<b>a</b>) displays the parameters for the radial line waveguide and coaxial waveguide in the proposed combiner and the transition structure connecting them. (<b>b</b>) provides a more detailed view of this transition structure and its parameters.</p> Full article ">Figure 3
<p>Structure of the input side: (<b>a</b>) N-type RF coaxial connector and (<b>b</b>) profile view. (The joint’s medium is represented in green, the internal channel of the combiner (also the electromagnetic wave transmission area) in blue, and the metal region in gray. (<b>a</b>) shows an input connector model, while (<b>b</b>) is a cross-section of all input connectors within the combiner).</p> Full article ">Figure 4
<p>Profile view of the input side. This diagram shows the transitional internal space channel at one of the combiner’s inputs and how it connects to the radial line waveguide. Other input ports on the combiner are evenly distributed around the outside of the waveguide in this particular model.</p> Full article ">Figure 5
<p>Structure of the model converter: (<b>a</b>) perspective view and (<b>b</b>) profile view. The proposed mode converter is shown in (<b>a</b>), with the blue section representing the internal spatial channel for electromagnetic wave transmission and the gray area describing the metal. (<b>b</b>) shows a cross-sectional view of the mode converter, including all structural parameters.</p> Full article ">Figure 6
<p>Results of <span class="html-italic">S</span>-parameter simulation of the power combiner. The figure shows the frequency on the horizontal axis and the <span class="html-italic">S</span>-parameter simulation result of the combiner on the vertical axis. For the 1:16 power divider/combiner simulation, the ideal single-way transmission coefficient <span class="html-italic">S</span><sub>n,1</sub> is approximately −12.04 dB, calculated as 10 log (1/16). During simulation, when Port1’s reflection coefficient <span class="html-italic">S</span><sub>11</sub> is better than −20 dB, transmission coefficient <span class="html-italic">S</span><sub>21</sub>~<span class="html-italic">S</span><sub>17,1</sub> will be around −12.04 dB due to the combiner’s excellent symmetry, while good power distribution/synthesis can be realized.</p> Full article ">Figure 7
<p>Cold tests: (<b>a</b>) the block diagram and (<b>b</b>) a snapshot of the experimental setup. The output port of the combiner is connected to a mode converter (WR340~N-type RF coaxial) and then to port 1 of the network analyzer. Fifteen output ports of the combiner are connected to the standard 50 Ω matched load, and the remaining port <span class="html-italic">X</span> to be tested is connected to the network analyzer port 2. The transmission coefficients and phases of the 16 input ports are measured sequentially in the experiment.</p> Full article ">Figure 8
<p>Results of <span class="html-italic">S</span>-parameter measurements: (<b>a</b>) <span class="html-italic">S</span>-parameter measurement results within a wide frequency band range and (<b>b</b>) transmission coefficients measured. It is seen from (<b>a</b>) that the reflection <span class="html-italic">S</span><sub>11</sub> is less than −20 dB in the range of 2.065–2.93 GHz. (<b>b</b>) shows the measured transmission coefficient <span class="html-italic">S<sub>n</sub></span><sub>,1</sub> from port 2~17 to port 1. The transmission coefficients of the 16 ports are distributed within a narrow range around −12 dB.</p> Full article ">Figure 9
<p>Phase measured (the figure shows the measured transmission phase from port 2~17 to port 1).</p> Full article ">Figure 10
<p>Block diagram of the experimental setup. A high-power CW RF source is connected to combiner 1 and combiner 2. The former here is used as a power divider, to divide the input microwave signal into 16 ways, which are subsequently connected to the inputs of combiner 2 through coaxial cables. Power meters 1 and 2 are attached to the coupling Port A of coupler 1 and Port B of coupler 2, respectively, to measure the coupled input and output power (the microwave energy is relatively low and falls within the power measurement range of the power meter). The load is employed to absorb the high-power microwave energy output from the straight-through end of coupler 2, thereby preventing any potential damage to other devices. In addition, a temperature sensor is connected to the combiner’s outer surface to record the device’s temperature.</p> Full article ">Figure 11
<p>The insertion loss and P-point temperature of the combiner vs. time, for a power of 2700 W in the power combiner. The horizontal axis represents the time of power injection to the combiner. The left vertical axis indicates the insertion loss of the device, and the right vertical axis is the temperature at point P indicated in the right panel.</p> Full article ">Figure 12
<p>Relationship between insertion loss and temperature of the device (this graph shows how the insertion loss of the combiner changes with the temperature; the horizontal axis represents the temperature, while the vertical axis represents the insertion loss).</p> Full article ">Figure 13
<p>Simulated steady-state temperature distribution of the combiner subjected to 3000 W CW injection at 2.45 GHz. In the simulation process, for the purpose of enhancing the stringency of simulation conditions, the convective heat transfer coefficient between the metal and air was adjusted to 5 W/(m<sup>2</sup>·K), and the thermal conductivity of the 6061 aluminum alloy was configured to 155 W/(m<sup>2</sup>·K). The displayed results were obtained with electromagnetic and thermal multiphysics field simulation. Since in experiments of high-power operation, it was observed that the inner conductor of the input port connector exhibited the highest temperature, while the outer surface of the waveguide showed the lowest temperature, which is consistent with the simulated results above, one can believe that by measuring the temperature at point P, it is possible to estimate the highest temperature point of the combiner.</p> Full article ">Figure 14
<p>The P-point temperature of the combiner vs. input power. The stabilized temperatures at point P of the combiner under various power conditions are shown in the figure. It is seen that the stabilized temperature increases monotonically with respect to the input power.</p> Full article ">
13 pages, 16250 KiB  
Article
Broadband Continuous Transverse Stub (CTS) Array Antenna for High-Power Applications
by Yunfei Sun, Kelin Zhou, Juntao He, Zihan Yang, Chengwei Yuan and Qiang Zhang
Micromachines 2023, 14(11), 2127; https://doi.org/10.3390/mi14112127 - 20 Nov 2023
Viewed by 1276
Abstract
A continuous transverse stub (CTS) array antenna with broad bandwidth and high-power handling capacity is proposed in this paper. The technologies of multi-step impedance matching and T-shaped electromagnetic band-gap (EBG) loading are utilized, which improved the antenna operating frequency bandwidth. An H-plane lens [...] Read more.
A continuous transverse stub (CTS) array antenna with broad bandwidth and high-power handling capacity is proposed in this paper. The technologies of multi-step impedance matching and T-shaped electromagnetic band-gap (EBG) loading are utilized, which improved the antenna operating frequency bandwidth. An H-plane lens horn is used to feed the CTS array. As a result, a good bandwidth capability of more than 32% is achieved, with a gain variation less than 3.0 dB. The measured sidelobe level (SLL) is below −18 dB in the entire frequency range. Moreover, the power handling capacity of the antenna is more than 80 MW and can reach the GW level after arraying, which indicates that this antenna has application potential in the high-power microwave (HPM) field. Full article
(This article belongs to the Special Issue Microwave Passive Components, 2nd Edition)
►▼ Show Figures

Figure 1

Figure 1
<p>Three-dimensional structure of non-uniform four-step CTS antenna.</p> Full article ">Figure 2
<p>The structure of non-uniform four-step CTS element.</p> Full article ">Figure 3
<p>Coupling coefficient of the three structures.</p> Full article ">Figure 4
<p>Data fitting and simulation comparison.</p> Full article ">Figure 5
<p><span class="html-italic">S</span> parameters.</p> Full article ">Figure 6
<p>Simulated gain and sidelobe.</p> Full article ">Figure 7
<p>Antenna 3D radiation pattern at 10 GHz.</p> Full article ">Figure 8
<p>Electric field inside the antenna: (<b>a</b>) without T-shaped EBG, (<b>b</b>) with T-shaped EBG.</p> Full article ">Figure 9
<p>Simulated results of the feed horn. (<b>a</b>) <span class="html-italic">S</span> parameters of the horn. (<b>b</b>) Phase distribution of electric field on flare surface. (<b>c</b>) Electric field distribution.</p> Full article ">Figure 10
<p>Measurement of S<sub>11</sub>.</p> Full article ">Figure 11
<p>Reflection coefficient for antenna simulation and measurement.</p> Full article ">Figure 12
<p>Measurement environment of the antenna pattern.</p> Full article ">Figure 13
<p>Measured and simulated radiation pattern of the antenna at 8.4 GHz. (<b>a</b>) Co-polarization radiation pattern of E-plane. (<b>b</b>) Cross-polarization radiation pattern of E-plane. (<b>c</b>) Co-polarization radiation pattern of H-plane. (<b>d</b>) Cross-polarization radiation pattern of H-plane.</p> Full article ">Figure 14
<p>Measured and simulated radiation pattern of the antenna at 10 GHz. (<b>a</b>) Co-polarization radiation pattern of E-plane. (<b>b</b>) Cross-polarization radiation pattern of E-plane. (<b>c</b>) Co-polarization radiation pattern of H-plane. (<b>d</b>) Cross-polarization radiation pattern of H-plane.</p> Full article ">Figure 15
<p>Measured radiation pattern of the antenna at 8.4–11.6 GHz.</p> Full article ">Figure 16
<p>The structure of the coupler.</p> Full article ">Figure 17
<p><span class="html-italic">S</span> parameters of the coupler.</p> Full article ">Figure 18
<p>Schematic diagram of HPM injection experiment.</p> Full article ">Figure 19
<p>Typical waveforms received by an oscilloscope.</p> Full article ">Figure 20
<p>The feed pillbox box.</p> Full article ">Figure 21
<p>The feed network.</p> Full article ">
18 pages, 6109 KiB  
Article
Accurate Microwave Circuit Co-Simulation Method Based on Simplified Equivalent Circuit Modeling
by Sanghyun Kim, Won-Sang Yoon, Jongsik Lim and Sang-Min Han
Micromachines 2023, 14(10), 1847; https://doi.org/10.3390/mi14101847 - 27 Sep 2023
Cited by 1 | Viewed by 1238
Abstract
A new co-simulation method is proposed for active devices and electromagnetic resonant circuits at microwave frequency range. For the measured and extracted device parameters, three steps of equivalent circuit models are processed of the general, simplified, and EM RLC models. To overcome the [...] Read more.
A new co-simulation method is proposed for active devices and electromagnetic resonant circuits at microwave frequency range. For the measured and extracted device parameters, three steps of equivalent circuit models are processed of the general, simplified, and EM RLC models. To overcome the limited lumped element simulation in an electromagnetic simulator, the simplified equivalent circuit model is established by mathematical computation. The co-simulation procedures are described and experimentally verified for commercial diodes. The application circuit is designed and implemented using the proposed co-simulation method. The experimental results verify that design using the proposed co-simulated method presented excellent agreement for a wideband frequency range of 0–4 GHz, compared with that using a conventional design method. The proposed co-simulation method can be applied to any commercial EM simulation tools without active model error. Full article
(This article belongs to the Special Issue Microwave Passive Components, 2nd Edition)
►▼ Show Figures

Figure 1

Figure 1
<p>Co-simulation procedures using the simplified equivalent circuit (EC) model.</p> Full article ">Figure 2
<p>Customized calibration kits of an open, a short, a load, and through circuits for de-embedded calibration.</p> Full article ">Figure 3
<p>Varactor diode equivalent circuit models. (<b>a</b>) General equivalent circuit model and chip photograph. (<b>b</b>) Simplified equivalent circuit model. (<b>c</b>) EM RLC model with transmission lines in EM simulators.</p> Full article ">Figure 4
<p>Comparison of varactor diode parameters of the measured extraction, general equivalent circuit models, and simplified equivalent circuit models for control voltages of (<b>a</b>) V<sub>R</sub> = 0 V, (<b>b</b>) V<sub>R</sub> = −1 V, (<b>c</b>) V<sub>R</sub> = −2 V (<b>d</b>) V<sub>R</sub> = −3 V, and (<b>e</b>) V<sub>R</sub> = −4 V.</p> Full article ">Figure 4 Cont.
<p>Comparison of varactor diode parameters of the measured extraction, general equivalent circuit models, and simplified equivalent circuit models for control voltages of (<b>a</b>) V<sub>R</sub> = 0 V, (<b>b</b>) V<sub>R</sub> = −1 V, (<b>c</b>) V<sub>R</sub> = −2 V (<b>d</b>) V<sub>R</sub> = −3 V, and (<b>e</b>) V<sub>R</sub> = −4 V.</p> Full article ">Figure 4 Cont.
<p>Comparison of varactor diode parameters of the measured extraction, general equivalent circuit models, and simplified equivalent circuit models for control voltages of (<b>a</b>) V<sub>R</sub> = 0 V, (<b>b</b>) V<sub>R</sub> = −1 V, (<b>c</b>) V<sub>R</sub> = −2 V (<b>d</b>) V<sub>R</sub> = −3 V, and (<b>e</b>) V<sub>R</sub> = −4 V.</p> Full article ">Figure 5
<p>EM RLC modeling. (<b>a</b>) Configuration of EM RLC model. (<b>b</b>) Reference simulation setup for the extracted parameter modification.</p> Full article ">Figure 6
<p>Comparison of varactor diode parameters of the measured extraction and the EM RLC model with transmission lines for control voltages of (<b>a</b>) V<sub>R</sub> = 0 V, (<b>b</b>) V<sub>R</sub> = −1 V, (<b>c</b>) V<sub>R</sub> = −2 V, (<b>d</b>) V<sub>R</sub> = −3 V, and (<b>e</b>) V<sub>R</sub> = −4 V.</p> Full article ">Figure 6 Cont.
<p>Comparison of varactor diode parameters of the measured extraction and the EM RLC model with transmission lines for control voltages of (<b>a</b>) V<sub>R</sub> = 0 V, (<b>b</b>) V<sub>R</sub> = −1 V, (<b>c</b>) V<sub>R</sub> = −2 V, (<b>d</b>) V<sub>R</sub> = −3 V, and (<b>e</b>) V<sub>R</sub> = −4 V.</p> Full article ">Figure 7
<p>Schottky diode equivalent circuit model. (<b>a</b>) Vendor equivalent circuit model. (<b>b</b>) Proposed equivalent circuit model.</p> Full article ">Figure 8
<p>Comparison of measured and modeled impedances for variable RF input power.</p> Full article ">Figure 9
<p>Lowpass filter design using the proposed co-simulation method. (<b>a</b>) Design layout. (<b>b</b>) Measurement setup.</p> Full article ">Figure 9 Cont.
<p>Lowpass filter design using the proposed co-simulation method. (<b>a</b>) Design layout. (<b>b</b>) Measurement setup.</p> Full article ">Figure 10
<p>S11 comparison of simulated designs with the ideal capacitors and proposed varactor diode models, and measured results for the control voltages of (<b>a</b>) V<sub>R</sub> = 0 V, (<b>b</b>) V<sub>R</sub> = −1 V, (<b>c</b>) V<sub>R</sub> = −2 V, (<b>d</b>) V<sub>R</sub> = −3 V, and (<b>e</b>) V<sub>R</sub> = −4 V.</p> Full article ">Figure 10 Cont.
<p>S11 comparison of simulated designs with the ideal capacitors and proposed varactor diode models, and measured results for the control voltages of (<b>a</b>) V<sub>R</sub> = 0 V, (<b>b</b>) V<sub>R</sub> = −1 V, (<b>c</b>) V<sub>R</sub> = −2 V, (<b>d</b>) V<sub>R</sub> = −3 V, and (<b>e</b>) V<sub>R</sub> = −4 V.</p> Full article ">Figure 10 Cont.
<p>S11 comparison of simulated designs with the ideal capacitors and proposed varactor diode models, and measured results for the control voltages of (<b>a</b>) V<sub>R</sub> = 0 V, (<b>b</b>) V<sub>R</sub> = −1 V, (<b>c</b>) V<sub>R</sub> = −2 V, (<b>d</b>) V<sub>R</sub> = −3 V, and (<b>e</b>) V<sub>R</sub> = −4 V.</p> Full article ">Figure 11
<p>S21 comparison of simulated designs with the ideal capacitors and proposed varactor diode models, and measured results for the control voltages of (<b>a</b>) V<sub>R</sub> = 0 V, (<b>b</b>) V<sub>R</sub> = −1 V, (<b>c</b>) V<sub>R</sub> = −2 V, (<b>d</b>) V<sub>R</sub> = −3 V, and (<b>e</b>) V<sub>R</sub> = −4 V.</p> Full article ">Figure 11 Cont.
<p>S21 comparison of simulated designs with the ideal capacitors and proposed varactor diode models, and measured results for the control voltages of (<b>a</b>) V<sub>R</sub> = 0 V, (<b>b</b>) V<sub>R</sub> = −1 V, (<b>c</b>) V<sub>R</sub> = −2 V, (<b>d</b>) V<sub>R</sub> = −3 V, and (<b>e</b>) V<sub>R</sub> = −4 V.</p> Full article ">Figure 11 Cont.
<p>S21 comparison of simulated designs with the ideal capacitors and proposed varactor diode models, and measured results for the control voltages of (<b>a</b>) V<sub>R</sub> = 0 V, (<b>b</b>) V<sub>R</sub> = −1 V, (<b>c</b>) V<sub>R</sub> = −2 V, (<b>d</b>) V<sub>R</sub> = −3 V, and (<b>e</b>) V<sub>R</sub> = −4 V.</p> Full article ">
14 pages, 3358 KiB  
Article
Design of Multiplexers for IoT-Based Applications Using Stub-Loaded Coupled-Line Resonators
by Muhammad Idrees, Sohail Khalid, Muhammad Abdul Rehman, Syed Sajid Ullah, Saddam Hussain and Jawaid Iqbal
Micromachines 2023, 14(10), 1821; https://doi.org/10.3390/mi14101821 - 23 Sep 2023
Cited by 1 | Viewed by 1014
Abstract
This paper presents the design of microstrip-based multiplexers using stub-loaded coupled-line resonators. The proposed multiplexers consist of a diplexer and a triplexer, meticulously engineered to operate at specific frequency bands relevant to IoT systems: 2.55 GHz, 3.94 GHz, and 5.75 GHz. To enhance [...] Read more.
This paper presents the design of microstrip-based multiplexers using stub-loaded coupled-line resonators. The proposed multiplexers consist of a diplexer and a triplexer, meticulously engineered to operate at specific frequency bands relevant to IoT systems: 2.55 GHz, 3.94 GHz, and 5.75 GHz. To enhance isolation and selectivity between the two passband regions, the diplexer incorporates five transmission poles (TPs) within its design. Similarly, the triplexer filter employs seven transmission poles to attain the desired performance across all three passbands. A comprehensive comparison was conducted against previously reported designs, considering crucial parameters such as size, insertion loss, return loss, and isolation between the two frequency bands. The fabrication of the diplexer and triplexer was carried out on a compact Rogers Duroid 5880 substrate. The experimental results demonstrate an exceptional performance, with the diplexer exhibiting a low insertion loss of 0.3 dB at 2.55 GHz and 0.4 dB at 3.94 GHz. The triplexer exhibits an insertion loss of 0.3 dB at 2.55 GHz, 0.37 dB at 3.94 GHz, and 0.2 dB at 5.75 GHz. The measured performance of the fabricated diplexer and triplexer aligns well with the simulated results, validating their effectiveness in meeting the desired specifications. Full article
(This article belongs to the Special Issue Microwave Passive Components, 2nd Edition)
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Figure 1
<p>Layout model of proposed diplexer filter.</p> Full article ">Figure 2
<p>Schematic of <math display="inline"><semantics> <mrow> <mi>B</mi> <mi>P</mi> <msub> <mi>F</mi> <mn>1</mn> </msub> </mrow> </semantics></math> with its equivalent circuit model.</p> Full article ">Figure 3
<p>Schematic of <math display="inline"><semantics> <mrow> <mi>B</mi> <mi>P</mi> <msub> <mi>F</mi> <mn>2</mn> </msub> </mrow> </semantics></math> with its equivalent circuit model.</p> Full article ">Figure 4
<p>Frequency response with a variable electrical length. (<b>a</b>) By increasing the electrical length; (<b>b</b>) By decrising the electrical length.</p> Full article ">Figure 5
<p>Frequency response with a variable width of coupled lines. (<b>a</b>) By Varing <math display="inline"><semantics> <msub> <mi>W</mi> <mn>1</mn> </msub> </semantics></math>; (<b>b</b>) By Varing <math display="inline"><semantics> <msub> <mi>W</mi> <mn>2</mn> </msub> </semantics></math>.</p> Full article ">Figure 6
<p>Schematic of <math display="inline"><semantics> <mrow> <mi>B</mi> <mi>P</mi> <msub> <mi>F</mi> <mn>3</mn> </msub> </mrow> </semantics></math> with its equivalent circuit model.</p> Full article ">Figure 7
<p>Frequency response with a variable electrical length. (<b>a</b>) By increasing the Electrical Length (<b>b</b>) By Decrising the Electrical Length.</p> Full article ">Figure 8
<p>Layout model of proposed triplexer filter.</p> Full article ">Figure 9
<p>S-parameter response of proposed diplexer.</p> Full article ">Figure 10
<p>Fabricated prototype of proposed diplexer.</p> Full article ">Figure 11
<p>S-parameter response of proposed triplexer.</p> Full article ">Figure 12
<p>Fabricated prototype of proposed triplexer.</p> Full article ">Figure 13
<p>Frequency response with a variable electrical length. (<b>a</b>) <math display="inline"><semantics> <mrow> <mi>B</mi> <mi>P</mi> <msub> <mi>F</mi> <mn>1</mn> </msub> </mrow> </semantics></math>; (<b>b</b>) <math display="inline"><semantics> <mrow> <mi>B</mi> <mi>P</mi> <msub> <mi>F</mi> <mn>2</mn> </msub> </mrow> </semantics></math>; (<b>c</b>) <math display="inline"><semantics> <mrow> <mi>B</mi> <mi>P</mi> <msub> <mi>F</mi> <mn>3</mn> </msub> </mrow> </semantics></math>.</p> Full article ">
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