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Walter Fischer
Digital Video and Audio
Broadcasting Technology
A Practical Engineering Guide
Third Edition
123
Dipl. Ing. (FH) Walter Fischer
Rohde & Schwarz GmbH & Co. KG
Geschäftsbereich Meßtechnik
Mühldorfstr. 15
81671 München
Germany
Walter.Fischer@Rohde-Schwarz.com
Translator :
Horst von Renouard
36 Chester Road
DA 15 8S6 Sidcup, Kent
United Kingdom
ISSN 1860-4862
ISBN 978-3-642-11611-7
e-ISBN 978-3-642-11612-4
DOI 10.1007/978-3-642-11612-4
Springer Heidelberg Dordrecht London New York
Library of Congress Control Number: 2010923775
Originally published with the title Digital Television
c Springer-Verlag Berlin Heidelberg 2003, 2007, 2010
This work is subject to copyright. All rights are reserved, whether the whole or part of the material is
concerned, specifically the rights of translation, reprinting, reuse of illustrations, recitation, broadcasting,
reproduction on microfilm or in any other way, and storage in data banks. Duplication of this publication
or parts thereof is permitted only under the provisions of the German Copyright Law of September 9,
1965, in its current version, and permission for use must always be obtained from Springer. Violations
are liable to prosecution under the German Copyright Law.
The use of general descriptive names, registered names, trademarks, etc. in this publication does not
imply, even in the absence of a specific statement, that such names are exempt from the relevant protective
laws and regulations and therefore free for general use.
Cover design: WMXDesign GmbH, Heidelberg
Printed on acid-free paper
Springer is part of Springer Science+Business Media (www.springer.com)
Preface
It is not so long ago that the second English edition of this book appeared.
In many countries, the switch-over from analog to digital television has
now been completed, especially in terrestrial television. Everyone is talking about high-definition TV - HDTV - which is supported by virtually
every TV display on the market today. The reason that the HDTV supply
chain is not as yet gapless is not to be found in the technology but only in
the lack of available HD source material. However, this is expected to
change from 2010 onward. The number of studios which are being reequipped with the new technology is ever increasing. Suitable compression
standards for HDTV have been around for years and there is now also sufficient bandwidth available with the second generation of DVB transmission standards.
In comparison with digital television, digital audio broadcasting - DAB
- still has its problems. Although DAB is on the air in many countries, it is
still largely unknown to the general public, Good old FM radio is still the
Number One audio transmission medium. It will be interesting to find out
what will happen in this respect in the next few years. Will DVB also
"gobble up" DAB?
What is now the third edition of the English version has been updated
further to match the facts of present conditions. This book contains all the
modern source encoding standards for digital television and digital audio
broadcasting. Nevertheless, one or the other areas will still seem to be
slightly inadequate to the liking of some readers. As the author, may aim is
not simply to copy standards - which I may even have misunderstood in
one point or the other - as the standards themselves are publicly available
and more or less comprehensible in their form although their interpretation
sometimes presents problems. Rather, I am mainly concerned with passing
on the knowledge I have actually acquired myself.
The fact that this book has now been published in four languages and
thus has found unexpectedly wide circulation all over the world is my
greatest compensation for all the work done since 2001. The depth of technical knowledge presented in many chapters would have been impossible
to achieve without the numerous discussions with my colleagues from the
Broadcasting Division at Rohde&Schwarz. Many more impulses also
VI
Preface
came from many readers and participants in seminars, and my special
thanks are due to the team of my publishers, Springer Verlag and to my
translator, collaboration with whom had been outstanding.
Moosburg an der Isar, near Munich, August 2009
Walter Fischer
Preface to the Second English Edition
A few years have passed since the first English edition of this book appeared in the bookshops. Digital television has become a fact of life in
many countries, conveyed to the viewer either by satellite, by cable or terrestrially through the rooftop antenna, and there are now also first indications of a fourth distribution path through IPTV, television by Internet, not
to forget Mobile TV which is being mentioned more and more frequently
in advertising. All these are reasons why it became necessary to update and
expand much of the book. But there are also some new chapters such as
DAB, Data Broadcasting, Mobile TV in the form of DVB-H, and T-DMB,
DRM, etc. Sections on modern source encoding methods such as MPEG-4
have also been amended, incorporating many suggestions from readers and
participants in seminars.
My previous publications "Digital Television - A Practical Guide for
Engineers" and "Digitale Fernsehtechnik in Theorie und Praxis" have
found warm acceptance by a wide circle of readers and both works have
also been used as welcome support material in numerous seminars.
My lectureship in the subject of "Television Engineering" at the Munich
University of Applied Science, which I am carrying on in the spirit of Prof.
Mäusl's lectures on the subject, is also providing me with new impulses in
the ways in which knowledge of the subject can be imparted and in the selection of contents, whilst at the same time enriching my own experience.
Since the last edition, many new findings and experiences have been
gathered by myself in many seminars throughout the world but also by me
personally participating when the DVB-T networks were being switched
on in Bavaria. Some of these findings and experiences will be found again
in this book.
Many thanks to my publishers, Springer Verlag, especially to Dr.
Merkle and Mrs. Jantzen, and to Horst von Renouard, the translator of this
book, and to my colleagues from Rohde&Schwarz, for their excellent collaboration in producing the finished book.
Moosburg an der Isar, near Munich, August 2007
Walter Fischer
Preface to the First English Edition
The world of television engineering has long fascinated me and from the
day I wrote my diploma paper on "The Generation of Test Lines" at the
Fachhochschule München (Munich University or Applied Sciences) under
Prof. Rudolf Mäusl in 1983, it has never released its grip on me. My research for this paper led to contacts with Rohde&Schwarz who where subsequently to become my employers. I worked there as development engineer until 1999, always in video test engineering but within various fields
of products and activities. For many years, this activity involved analog
video testing and there mainly video insertion test signals (VITS), but from
the mid-nineties onward, the focus shifted more and more to MPEG-2 and
digital video broadcasting (DVB) and then quite generally to the field of
digital television. Naturally, as a consequence of my work as a development engineer I also became intensively engaged in the field of firmware
and software development and my involvement with the programming
language C and C++ led me into the domain of software training where I
was increasingly active in-house from the early nineties onward. I have
lost count of the number of seminars and of the participants in these seminars who succeeded in implanting in me a joy in this type of "work".
Whatever the cause, it was in the course of these, perhaps forty seminars
that I discovered my love for instructing and in 1999 I chose this to be my
main occupation. Since March 1999, I have been active as instructor in the
field of television engineering, main subject "digital television", in the
Rohde&Schwarz Training Center. Since then, I have travelled more than
500,000 km by air all over the world, from Stockholm to Sydney, to provide instruction about the new field of digital television and especially
about test engineering and transmitter technology.
A key event in my professional life has been a seminar request from
Australia in July 1999 which resulted in, thus far, 7 trips to Australia with
a total stay of about half a year, more than 50 seminar days and almost 400
participants. From this has sprung a love for this far-distant, wonderful
continent which, I am sure, will be apparent between the lines throughout
this book. One of the main suggestions to write this book as a résumé of
my seminars came from the circle of participants in Australia. These trips
gave rise to significant impulses and I have gained a large amount of prac-
X
Preface to the First English Edition
tical experience during my seminars "Down Under" and during the construction of their DVB-T network, which proved to be invaluable in the
creation of this book. I owe special thanks to my colleague, Simon Haynes
from Rohde&Schwarz Australia, who provided me with the closest support for the seminars and with helpful suggestions for this book. We often
talked about publishing the contents of the seminars but I had underestimated the effort involved. The original documentation for the seminars did
not easily lend itself to being directly for the book. Virtually all the texts
had be completely revised, but now I had plenty to occupy me during the
100 days or so of travelling a year, even at night, an important factor with
all the boredom of being absent from home.
My readers will be people who have a practical interest in the new subject of "Digital Television", engineers and technicians who want to or have
to familiarize themselves with this new field and the book, therefore, contains only a minimum ballast of mathematics although, by the nature of
thins, there have to be some.
In the meantime, I have been able to extend my seminar travels to other
countries as, for example, Greenland, and to gather numerous impressions
there, too. However, although it is very nice to see the world as a result of
one's professional activities, it is not easy for one's family or for oneself,
for that matter. For this reason, I would like to take this opportunity to express special thanks to those who had to stay at home for whom I was then
not available. To some extent this also applies to the time when this book
was written. In particular, I thank my daughter Christine for her help in
writing the manuscript.
I would like to thank Horst von Renouard from London for his successful translation. As chance would have it, he, too, had spent many years in
Australia and also comes from the field of television engineering. He thus
was able to empathize with what I was trying to express and to convey this
in his translation. And while I am on the subject of translation, my gratitude is due also to the Rohde&Schwarz Translation Department who also
contributed some chapters which were required in advance for seminar
purposes.
To my former patron, Prof. Rudolf Mäusl, who initiated me into the
world of television engineering as no-one else could have done, my heartfelt thanks four our many conversations and for all his helpful suggestions.
His lectures at the Fachhochschule and his way of imparting knowledge
have always been of guiding influence on me and, I hope, have also been a
positive influence on how this book has turned out. His many publications
and books are models in their field and can only be recommended.
Preface to the First English Edition
XI
Many thanks also to my publishers, Springer Verlag, to Dr. Merkle,
Mrs. Jantzen and Mrs. Maas for their active support, and for the opportunity to have this book published by this renowned publishing house.
And many thanks for the many discussions and suggestions by the participants in my seminars throughout the world, in Australia, Austria, Canada, the Czech Republic, France, Germany, Greenland, Latvia, Mexico,
the Netherlands, Portugal, Spain, Sweden, Switzerland, Turkey, the United
States and all the other countries in which I have been or from which participants have come to Munich or elsewhere in order to join me in finding
out about the complex subject of digital television.
To the present day, there have been worldwide seminars on the subject
of analog and digital television for just on 300 days, with about 2000 participants from all corners of the world. These international seminars present a rich personal experience and I am filled with gratitude at the many
contacts made, some are still ongoing via email.
Moosburg an der Isar, near Munich, June 2003
Walter Fischer
Foreword
Without a doubt, this book can definitely be called a reference work and is
a true “Engineering Guide to Digital Television”. Walter Fischer is an outstandingly knowledgeable author and an expert in his chosen field. I have
known him since the beginning of the eighties when he attended my lectures at the Fachhochschule München (Munich University of Applied Sciences). He attracted attention even then with his excellent knowledge and
with the way he tackled new and complex problems. After he had concluded his studies, continuing contacts with my erstwhile employer Rohde
& Schwarz then provided him with the opportunity to give free rein to his
talent in their Department of Television Test Engineering.
In 1988 the Fernseh- und Kinotechnische Gesellschaft (Television and
Cinematographic Association) awarded him their Rudolf Urtel Price for
independently developing a test method for determining the parameters of
a video channel by means of the Fast Fourier Transform (FFT).
After a long period of developing test instruments for analog and digital
television signals and equipped with the extensive knowledge in digital
television practice gained from this, he finally realized his long-standing
ambition to change over into the field of teaching. For some years now he
has been active for the Rohde&Schwarz Training Center and is passing on
this knowledge in seminars all over the world. I may add that I, too, have
been able to benefit from Walter Fischer’s expertise in my own relatively
brief volume on digital television.
I wish Walter Fischer continuing success, particularly with regard to a
good acceptance of this reference work throughout the world.
Aschheim near Munich, February 2003
Professor Rudolf Mäusl
Note from the Translator
When I was first asked to translate Walter Fischer's book on digital television a few years ago, digital television was a relatively new concept for me
who had 'grown up' in the age of valves and analog black-and-white, and
later color, television. I had been a television technician with the Australian Broadcasting Corporation, the ABC, and later, in 1969, operated the
TV scan converter at the Honeysuckle Creek tracking station near Canberra in Australia that brought live television of the Apollo 11 moon landing beamed down from the moon over a distance of more than 350,000
kilometers to a world audience ("TELE-vision", seeing at a distance, indeed!).
That was over 40 years ago, narrow-band (to save transmission power)
analog black-and-white slow scan television (SSTV) at 10 frames per second that was recorded on a large-diameter magnetic disk from where each
frame was read out 5 times to produce the 60 frames-a-second NTSCstandard broadcast picture which was relayed to the world (not "done in a
shed", either, as some people want you to believe today - that SSTV signal
came into our receivers from the moon as verified by our ranging pulse
which indicated that the source of the signal was, indeed, some 250,000
miles distant, and by the vidicon camera mounted on the receiving antenna
dish which showed us that the antenna was pointing at the moon).
After the Apollo and Skylab days I left engineering to pander to my
second love - languages, combining the two to become a technical translator. But television has long been in my blood, and when Walter Fischer
asked me to translate his supremely accomplished reference work on digital television, I felt singularly honoured and, I admit, flattered. In the end,
its success has justified our combined efforts and, I feel sure, will continue
to do so.
London, December 2009
Horst E. von Renouard
Table of Contents
1 Introduction............................................................................................1
2 Analog Television....................................................................................7
2.1 Scanning an Original Black/White Picture.....................................10
2.2 Horizontal and Vertical Synchronization Pulses ............................12
2.3 Adding the Color Information ........................................................14
2.4 Transmission Methods....................................................................17
2.5 Distortion and Interference .............................................................18
2.6 Signals in the Vertical Blanking Interval........................................20
2.7 Measurements on Analog Video Signals........................................24
2.8 Analog and Digital TV in a Broadband Cable Network.................29
3 The MPEG Data Stream ......................................................................31
3.1 The Packetized Elementary Stream (PES)......................................35
3.2 The MPEG-2 Transport Stream Packet ..........................................38
3.3 Information for the Receiver...........................................................42
3.3.1 Synchronizing to the Transport Stream ...................................43
3.3.2 Reading out the Current Program Structure ............................44
3.3.3 Accessing a Program ...............................................................46
3.3.4 Accessing Scrambled Programs ..............................................46
3.3.5 Program Synchronization (PCR, DTS, PTS)...........................48
3.3.6 Additional Information in the Transport Stream .....................51
3.3.7 Non-Private and Private Sections and Tables..........................52
3.3.8 The Service Information according to DVB (SI).....................60
3.4 The PSIP according to the ATSC ...................................................73
3.5 ARIB Tables according to ISDB-T ................................................76
3.6 DTMB (China) Tables....................................................................78
3.7 Other Important Details of the MPEG-2 Transport Stream............78
3.7.1 The Transport Priority .............................................................78
3.7.2 The Transport Scrambling Control Bits ..................................79
3.7.3 The Adaptation Field Control Bits ..........................................79
3.7.4 The Continuity Counter ...........................................................80
XVIII
Table of Contents
4 Digital Video Signal According to ITU-BT.R.601 (CCIR 601).........81
5 High Definition Television – HDTV ....................................................87
6 Transforms to and from the Frequency Domain ...............................93
6.1 The Fourier Transform ...................................................................95
6.2 The Discrete Fourier Transform (DFT) ..........................................97
6.3 The Fast Fourier Transform (FFT) .................................................99
6.4 Implementation and Practical Applications of DFT and FFT.......101
6.5 The Discrete Cosine Transform (DCT) ........................................101
6.6 Time Domain Signals and their Transforms in the Frequency
Domain ...............................................................................................105
6.7 Systematic Errors in DFT or FFT, and How to Prevent them ......107
6.8 Window Functions........................................................................110
7 MPEG-2 Video Coding.......................................................................111
7.1 Video Compression ......................................................................111
7.1.1 Reducing the Quantization from 10 Bits to 8 ........................115
7.1.2 Omitting the Horizontal and Vertical Blanking Intervals......115
7.1.3 Reduction in Vertical Color Resolution (4:2:0).....................116
7.1.4 Further Data Reduction Steps................................................117
7.1.5 Differential Pulse Code Modulation of Moving Pictures ......118
7.1.6 Discrete Cosine Transform Followed by Quantization .........123
7.1.7 Zig-Zag Scanning with Run-Length Coding
of Zero Sequences ..........................................................................131
7.1.8 Huffman Coding ....................................................................131
7.2 Summary.......................................................................................132
7.3 Structure of the Video Elementary Stream ...................................136
7.4 More Recent Video Compression Methods..................................137
7.5 MPEG-4 Advanced Video Coding ...............................................138
8 Compression of Audio Signals to MPEG and Dolby Digital...........147
8.1 Digital Audio Source Signal .........................................................147
8.2 History of Audio Coding ..............................................................148
8.3 Psychoacoustic Model of the Human Ear.....................................151
8.4 Basic Principles of Audio Coding ................................................155
8.5 Subband Coding in Accordance with MPEG Layer I, II ..............157
8.6 Transform Coding for MPEG Layer III and Dolby Digital..........159
8.7 Multichannel Sound......................................................................161
8.8 New Developments - MPEG-4 .....................................................162
9 Teletext, Subtitles and VPS for DVB ................................................165
Table of Contents
XIX
9.1 Teletext and Subtitles ...................................................................165
9.2 Video Program System .................................................................169
9.3 WSS – Wide Screen Signalling ....................................................171
9.4 Practical examples ........................................................................173
10 A Comparison of Digital Video Standards .....................................175
10.1 MPEG-1 and MPEG-2, VCD and DVD, M-JPEG and
MiniDV/DV........................................................................................175
10.2 MPEG-3, MPEG-4, MPEG-7 and MPEG-21 .............................178
10.3 Physical Interfaces for Digital Video Signals.............................182
10.3.1 Parallel and Serial CCIR 601...............................................182
10.3.2 Synchronous Parallel Transport Stream Interface ...............184
10.3.3 Asynchronous Serial Transport Stream Interface (TS ASI) 185
10.3.4 SMPTE 310 Interface .........................................................186
10.3.5 DVI Interface......................................................................186
10.3.6 HDMI Interface ..................................................................187
10.3.7 HD-SDI Interface ...............................................................188
10.3.8 Gigabit Ethernet Interface as Transport Stream Distributor188
11 Measurements on the MPEG-2 Transport Stream........................189
11.1 Loss of Synchronisation (TS_sync_loss)....................................191
11.2 Errored Sync Bytes (sync_byte_error) .......................................192
11.3 Missing or Errored Program Association Table (PAT) ..............192
11.4 Missing or Errored Program Map Table (PMT) (PMT_error) ...193
11.5 The PID_Error ............................................................................194
11.6 The Continuity_Count_Error......................................................195
11.7 The Transport_Error (Priority 2) ................................................196
11.8 The Cyclic Redundancy Check Error .........................................197
11.9 The Program Clock Reference Error ..........................................197
11.10 The Presentation Time Stamp Error (PTS_Error) ....................199
11.11 The Conditional Access Table Error (CAT_Error) ..................200
11.12 Service Information Repetition Rate Error ...............................201
11.13 Monitoring the NIT, SDT, EIT, RST and TDT/TOT Tables....202
11.14 Undefined PIDs (Unreferenced_PID).......................................202
11.15 Errors in the Transmission of Additional Service Information 203
11.16 Faulty tables NIT_other_error, SDT_other_error,
EIT_other_error ..................................................................................204
11.17 Monitoring an ATSC-Compliant MPEG-2 Transport Stream..204
12 Picture Quality Analysis of Digital TV Signals ..............................207
12.1 Methods for Measuring Video Quality.......................................209
12.1.1 Subjective Picture Quality Analysis ....................................210
XX
Table of Contents
12.1.2 Double Stimulus Continual Quality Scale Method DSCQS 211
12.1.3 Single Stimulus Continual Quality Evaluation Method
SSCQE............................................................................................211
12.2 Objective Picture Quality Analysis.............................................211
12.3 Summary and Outlook...............................................................217
13 Basic Principles of Digital Modulation ...........................................219
13.1 Introduction.................................................................................219
13.2 Mixer...........................................................................................221
13.3 Amplitude Modulator .................................................................223
13.4 IQ Modulator ..............................................................................225
13.5 The IQ Demodulator...................................................................233
13.6 Use of the Hilbert transform in IQ modulation...........................238
13.7 Practical Applications of the Hilbert Transform.........................240
13.8 Channel Coding/Forward Error Correction ................................242
13.9 A Comparison to Analog Modulation Methods.........................246
13.9.1 Amplitude Modulation .......................................................247
13.9.2 Variants of Amplitude Modulation.....................................251
13.9.3 Frequency Modulation........................................................252
13.9.4 Phase Modulation ...............................................................255
13.10 Band Limiting of Modulated Carrier Signals ..........................256
13.11 Summary..................................................................................259
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2 261
14.1 The DVB-S System Parameters..................................................264
14.2 The DVB-S Modulator ...............................................................266
14.3 Convolutional Coding.................................................................272
14.4 Signal Processing in the Satellite................................................277
14.5 The DVB-S Receiver ..................................................................278
14.6 Influences Affecting the Satellite Transmission Link ................281
14.7 DVB-S2 ......................................................................................285
15 DVB-S/S2 Measuring Technology ...................................................293
15.1 Introduction.................................................................................293
15.2 Measuring Bit Error Ratios.........................................................293
15.3 Measurements on DVB-S Signals using a Spectrum Analyzer ..296
15.3.1 Approximate Determination of the Noise Power N.............298
15.3.2 C/N, S/N and Eb/N0 .............................................................299
15.3.3 Finding the EB/N0 Ratio .......................................................300
15.4 Modulation Error Ratio (MER) ..................................................301
15.5 Measuring the Shoulder Attenuation ..........................................302
15.6 DVB-S/S2 Receiver Test ............................................................303
Table of Contents
XXI
16 Broadband Cable Transmission According to DVB-C .................305
16.1 The DVB-C Standard .................................................................306
16.2 The DVB-C Modulator...............................................................308
16.3 The DVB-C Receiver .................................................................309
16.4 Interference Effects on the DVB-C Transmission Link .............310
17 Broadband Cable Transmission According to ITU-T J83B (US) 315
17.1 J83B Transmission Parameters...................................................317
17.2 J83B Baseband Input Signals .....................................................318
17.3 Forward Error Correction ...........................................................318
17.4 Calculation of the Net Data Rate ...............................................320
17.5 Roll-off Filtering.........................................................................321
17.6 Fall-off-the-Cliff .........................................................................321
18 Measuring Digital TV Signals in the Broadband Cable................325
18.1 DVB-C/J83A, B, C Test Receivers with Constellation Analysis326
18.2 Detecting Interference Effects Using Constellation Analysis.....330
18.2.1 Additive White Gaussian Noise (AWGN)...........................330
18.2.2 Phase Jitter...........................................................................333
18.2.3 Sinusoidal Interferer ............................................................334
18.2.4 Effects of the I/Q Modulator................................................334
18.2.5 Modulation Error Ratio (MER) ...........................................337
18.2.6 Error Vector Magnitude (EVM) ..........................................339
18.3 Measuring the Bit Error Ratio (BER) .........................................339
18.4 Using a Spectrum Analyzer for Measuring DVB-C Signals ......340
18.5 Measuring the Shoulder Attenuation ..........................................342
18.6 Measuring the Ripple or Tilt in the Channel ..............................343
18.7 DVB-C/J83A,B,C Receiver Test ................................................343
19 Coded Orthogonal Frequency Division Multiplex (COFDM) ......345
19.1 Why Multi-Carrier? ....................................................................347
19.2 What is COFDM? .......................................................................350
19.3 Generating the COFDM Symbols...............................................354
19.4 Supplementary Signals in the COFDM Spectrum......................364
19.5 Hierarchical Modulation .............................................................366
19.6 Summary.....................................................................................367
20 Terrestrial Transmission of Digital Television Signals (DVB-T) 369
20.1 The DVB-T Standard..................................................................371
20.2 The DVB-T Carriers ...................................................................373
20.3 Hierarchical Modulation .............................................................379
XXII
Table of Contents
20.4 DVB-T System Parameters of the 8/7/6 MHz Channel..............381
20.5 The DVB-T Modulator and Transmitter.....................................390
20.6 The DVB-T Receiver..................................................................393
20.7 Interference on the DVB-T Transmission Link and its Effects ..398
20.8 DVB-T Single-Frequency Networks (SFN) ...............................406
20.9 Minimum Receiver Input Level Required with DVB-T.............414
21 Measuring DVB-T Signals ...............................................................421
21.1 Measuring the Bit Error Ratio ....................................................423
21.2 Measuring DVB-T Signals Using a Spectrum Analyzer ...........425
21.3 Constellation Analysis of DVB-T Signals..................................429
21.3.1 Additive White Gaussian Noise (AWGN)...........................430
21.3.2 Phase Jitter...........................................................................430
21.3.3 Interference Sources ............................................................430
21.3.4 Echoes, Multipath Reception...............................................431
21.3.5 Doppler Effect .....................................................................431
21.3.6 I/Q Errors of the Modulator.................................................431
21.3.7 Cause and Effect of I/Q Errors in DVB-T ...........................435
21.4 Measuring the Crest Factor.........................................................444
21.5 Measuring the Amplitude, Phase and Group Delay Response ...444
21.6 Measuring the Impulse Response ...............................................445
21.7 Measuring the Shoulder Attenuation ..........................................446
22 DVB-H/DVB-SH - Digital Video Broadcasting for Handhelds ....451
22.1 Introduction.................................................................................451
22.2 Convergence between Mobile Radio and Broadcasting .............453
22.3 Essential Parameters of DVB-H .................................................454
22.4 DSM-CC Sections ......................................................................455
22.5 Multiprotocol Encapsulation.......................................................457
22.6 DVB-H Standard ........................................................................458
22.7 Summary.....................................................................................462
22.8 DVB-SH ....................................................................................464
23 Digital Terrestrial TV to North American ATSC Standard.........467
23.1 The 8VSB Modulator .................................................................472
23.2 8VSB Gross Data Rate and Net Data Rate .................................480
23.3 The ATSC Receiver....................................................................482
23.4 Causes of Interference on the ATSC Transmission Path............482
23.5 ATSC-M/H Mobile DTV............................................................483
23.5.1 Compatibility with the Existing Frame Structure ................483
23.5.2 MPEG-4 Video and Audio Streaming.................................485
23.5.3 ATSC M/H Multiplexer.......................................................487
Table of Contents
XXIII
23.5.4 ATSC M/H Modulator.........................................................490
23.5.5 Forming Single Frequency Networks..................................490
23.5.6 Summary..............................................................................491
23.6 Closed Captioning ......................................................................492
23.7 Analog Switch-off ......................................................................493
24 ATSC/8VSB Measurements.............................................................495
24.1 Bit Error Ratio (BER) Measurement ..........................................495
24.2 8VSB Measurements Using a Spectrum Analyzer .....................496
24.3 Constellation Analysis on 8VSB Signals....................................497
24.4 Measuring Amplitude Response and Group Delay Response ....500
25 Digital Terrestrial Television according to ISDB-T.......................503
25.1 Introduction.................................................................................503
25.2 ISDB-T Concept .........................................................................504
25.1 Forming Layers...........................................................................506
25.2 Baseband Encoding ....................................................................507
25.3 Changes in the Transport Stream Structure ................................507
25.4 Channel Tables ...........................................................................509
25.5 Performance of ISDB-T..............................................................510
25.6 Other ISDB Standards ................................................................511
25.7 ISDB-T measurements................................................................512
25.8 Summary.....................................................................................513
26 Digital Audio Broadcasting - DAB ..................................................515
26.1 Comparing DAB and DVB.........................................................517
26.2 An Overview of DAB.................................................................520
26.3 The Physical Layer of DAB........................................................525
26.4 DAB – Forward Error Correction...............................................537
26.5 DAB Modulator and Transmitter................................................542
26.6 DAB Data Structure....................................................................546
26.7 DAB Single-Frequency Networks ..............................................551
26.8 DAB Data Broadcasting .............................................................553
26.9 DAB+..........................................................................................554
26.10 DAB Measuring Technology....................................................554
26.10.1 Testing DAB Receivers ....................................................555
26.10.2 Measuring the DAB Signal...............................................555
27 DVB Data Services: MHP and SSU ................................................559
27.1 Data Broadcasting in DVB .........................................................560
27.2 Object Carousels.........................................................................561
27.3 The Multimedia Home Platform MHP .......................................563
XXIV
Table of Contents
27.4 System Software Update SSU ....................................................565
28 T-DMB ...............................................................................................567
29 IPTV – Television over the Internet................................................569
29.1 DVB-IP .......................................................................................571
29.2 IP Interface Replaces TS-ASI.....................................................572
29.3 Summary.....................................................................................573
30 DRM – Digital Radio Mondiale .......................................................575
30.1 Audio source encoding ..............................................................579
30.2 Forward Error Correction ...........................................................579
30.3 Modulation Method ....................................................................580
30.4 Frame structure ...........................................................................581
30.5 Interference on the transmission link..........................................582
30.6 DRM data rates ...........................................................................583
30.7 DRM transmitting stations and DRM receivers..........................584
30.8 DRM+........................................................................................585
31 Digital Terrestrial TV Networks in Practice ..................................587
31.1 The DVB-T SFNs Southern and Eastern Bavaria.......................587
31.2 Playout Center and Feed Networks ............................................590
31.3 Technical Configuration of the Transmitter Sites.......................591
31.3.1 Mount Wendelstein Transmitter ..........................................592
31.3.2 Olympic Tower Transmitter, Munich..................................606
31.3.3 Brotjacklriegel Transmitter..................................................609
31.4 Measurements in DVB-T Single-Frequency Networks..............612
31.4.1 Test Parameters....................................................................612
31.4.2 Practical Examples ..............................................................620
31.4.3 Response of DVB-T Receivers............................................627
31.4.4 Receiver Test and Simulation of Receiving Conditions in
Single-Frequency Networks ...........................................................628
31.5 Network Planning .......................................................................632
31.6 Filling the Gaps in the Coverage ................................................632
31.7 Fall-off-the-Cliff .........................................................................635
31.8 Summary.....................................................................................636
32 DTMB ................................................................................................637
32.1 DMB-T, or now DTMB..............................................................637
32.2 Some more Details......................................................................638
33 Return Channel Techniques ............................................................641
Table of Contents
XXV
34 Display Technologies ........................................................................643
34.1 Previous Converter Systems - the Nipkow Disk.........................645
34.2 The Cathode Ray Tube (CRT)....................................................647
34.3 The Plasma Screen......................................................................651
34.4 The Liquid Crystal Display Screen.............................................652
34.5 Digital Light Processing Systems ...............................................654
34.6 Organic Light-Emitting Diodes ..................................................654
34.7 Effects on Image Reproduction ..................................................654
34.8 Compensation Methods ..............................................................657
34.9 Test Methods ..............................................................................657
34.10 Current State of the Technology...............................................658
35 The New Generation of DVB Standards.........................................661
35.1 Overview of the DVB Standards ................................................662
35.2 Characteristics of the Old and the New Standards......................663
35.3 Capabilities and Aims of the New DVB Standards ....................664
36 Baseband Signals for DVB-x2..........................................................667
36.1 Input Signal Formats...................................................................667
36.1.1 MPEG-2 Transport Streams - TS ........................................668
36.1.2 Generic Fixed Packetized Streams - GFPS .........................669
36.1.3 Generic Continuous Streams - GCS ....................................669
36.1.4 Generic Encapsulated Streams - GSE..................................670
36.2 Signal Processing and Conditioning in the Modulator Input
Section ................................................................................................670
36.2.1 Single Input Stream .............................................................671
36.2.2 Multiple Input Streams ........................................................674
36.3 Standard-related Special Features...............................................677
36.3.1 DVB-S2 ...............................................................................677
36.3.2 DVB-T2...............................................................................678
36.3.3 DVB-C2...............................................................................684
37 DVB-T2..............................................................................................687
37.1 Introduction.................................................................................687
37.2 Theoretical Maximum Channel Capacity ...................................688
37.3 DVB-T2 - Overview ...................................................................690
37.4 Baseband Interface......................................................................690
37.5 Forward Error Correction ...........................................................691
37.6 COFDM Parameters ...................................................................695
37.6.1 Normal Carrier Mode ..........................................................697
37.6.2 Extended Carrier Mode........................................................699
37.7 Modulation Patterns....................................................................700
XXVI
Table of Contents
37.7.1 Normal Constellation Diagrams ..........................................701
37.7.2 Definition of 'Cell' ...............................................................701
37.7.3 Rotated Q-delayed Constellation Diagrams ........................702
37.8 Frame Structure ..........................................................................704
37.8.1 P1 Symbol............................................................................706
37.8.2 P2 Symbols ..........................................................................707
37.8.3 Symbol, Frame, Superframe ................................................709
37.9 Block Diagram............................................................................709
37.10 Interleavers ...............................................................................710
37.10.1 Types of Interleaver...........................................................711
37.10.2 DVB-T2 Time Interleaver Configuration..........................712
37.11 Pilots .........................................................................................715
37.12 Sub-Slicing ...............................................................................717
37.13 Time-Frequency-Slicing (TFS).................................................718
37.14 PAPR Reduction.......................................................................719
37.15 SISO/MISO Multi-Antenna Systems........................................720
37.15.1 MISO according to Alamouti ............................................721
37.15.2 Modified Alamouti in DVB-T2 .........................................722
37.16 Future Extension Frames ..........................................................726
37.17 Auxiliary Data Streams.............................................................726
37.18 DVB-T2-MI..............................................................................726
37.19 SFNs in DVB-T2 ......................................................................726
37.20 Transmitter Identification Information in DVB-T2 ............. .... 728
37.21 Capacity ....................................................................................728
37.22 Outlook .....................................................................................729
38 DVB-C2 – the New DVB Broadband Cable Standard ..................731
38.1 Introduction.................................................................................731
38.2 Theoretical Maximum Channel Capacity ...................................733
38.3 DVB-C2 – An Overview ............................................................734
38.4 Baseband Interface......................................................................735
38.5 Forward Error Correction ...........................................................735
38.6 COFDM Parameters ...................................................................735
38.7 Modulation Pattern .....................................................................737
38.8 Definition of a Cell .....................................................................739
38.9 Interleavers .................................................................................739
38.10 Variable Coding and Modulation (VCM).................................740
38.11 Frame Structure ........................................................................740
38.12 Channel Bundling and Slice Building ......................................741
38.13 Preamble Symbols ....................................................................742
38.14 Pilots in DVB-C2......................................................................745
38.15 PAPR ........................................................................................746
38.16 Block Diagram..........................................................................746
Table of Contents
XXVII
38.17 Levels in Broadband Cables .....................................................746
38.18 Capacity ....................................................................................748
38.19 Outlook .....................................................................................748
39 DVB-x2 Measuring Techniques.......................................................749
39.1 DVB-S2 ......................................................................................749
39.2 DVB-T2 ......................................................................................750
39.3 DVB-C2......................................................................................752
39.4 Summary.....................................................................................753
40 CMMB – Chinese Multimedia Mobile Broadcasting ....................755
41 Other Transmission Standards........................................................757
41.1 MediaFLO...................................................................................757
41.2 IBOC - HD Radio .......................................................................758
41.3 FMextra.......................................................................................759
41.4 Effects of the Digital Dividend on Cable and Terrestrial TV
Networks.............................................................................................759
41.4.1 Anatomy of the Mobile Radio Signals ................................760
41.4.2 Terrestrial TV Networks and Mobile Radio........................761
41.4.3 Broadband Cable TV Networks and Mobile Radio.............761
41.4.4 Electromagnetic Field Immunity Standard for Sound and
Television Broadcast Receivers......................................................763
41.4.5 Summary..............................................................................764
42 Digital Television throughout the World - an Overview ..............765
Bibliography...........................................................................................769
Definition of Terms................................................................................777
TV Channel Tables ................................................................................791
Europe, Terrestrial and Cable .............................................................791
Australia, Terrestrial ...........................................................................794
North America, Terrestrial..................................................................795
North America, Cable.........................................................................797
Europe, Satellite..................................................................................801
Index........................................................................................................803
1 Introduction
For many decades, television and data transmission have followed parallel
paths which, however, were completely independent of one another. Although television sets were used as first home computer monitors back in
the eighties of the century now past, this was the only interaction between
the two fields. Today, however, it is becoming more and more difficult to
distinguish between the two media of TV and computers which are converging increasingly in this age of multimedia. There are now excellent
TV cards for PCs so that the PC can easily become another TV set. On the
other side, teletext was introduced back in the eighties to provide an early
medium for supplementary digital information in analog TV. For young
people, this type of information is such a natural part of viewing, e.g. as
electronic program guide, as if there had been teletext from the beginnings
of television.
And now we are living in the age of digital TV, since 1995 in fact, and
the distinction between data and television has virtually disappeared.
When one is able to follow the developments in this field throughout the
world like the author has on numerous seminar trips, one will encounter
more and more applications where either both television and data services
are found jointly in one data signal, or the services are even just pure data
services, e.g. fast Internet access via channels which were actually provided for digital TV. The common factor leading to this fusion is the high
data rate. Today’s generation is hungry for information and used to getting
it in large quantity and variety. Talking to telecommunication specialists
about data rates, one hears time and again how envious they are of the data
rates used in digital TV. Thus GSM, for example, works with data rates of
9600 bit/sec and UMTS uses a maximum of 2 Mbit/sec under optimum
conditions, e.g. for Internet accesses. An ISDN basic access telephone
channel has two times 64 kbit/sec. By comparison, the data rate of an uncompressed digital Standard Definition TV signal is already 270 Mbit/sec
and High Definition TV begins at about 1.5 Gbit/s and extends into the 3
Gigabit range. One would be fully justified to call television a broadband
technology, not only from the point of view of digital TV but even in analog TV where the channels have always been very wide. An analog or
digital terrestrial TV channel has a width of 6, 7 or 8 MHz and the chanW. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_1, © Springer-Verlag Berlin Heidelberg 2010
2
1 Introduction
nels broadcast via satellite are even 36 MHz wide. It is not surprising that
a new boom is being experienced especially in broadband TV cable, which
is being used as a medium for high-speed home Internet access in the
Mbit/sec range, uplinking via cable modems.
The foundation stone for analog television was laid by Paul Nipkow
back in 1883 when he developed what is now known as the Nipkow disc.
He had the idea of transmitting a picture by disecting it into lines. The first
real analog TV transmissions per se took place in the thirties but, held back
by World War II, analog television didn’t have its proper start until the fifties, in black and white at first. The television set acquired color towards
the end of the sixties and from then on, this technology has been basically
only refined, both in the studio and in the home. There have been no further changes in the principles of the technology. Analog TV transmissions
are often so perfect, at least in quality if not in content, that it is difficult to
interest many people in buying a receiver for digital TV.
In the eighties, an attempt was made to depart from traditional analog
TV by way of D2MAC. For various reasons, this did not succeed and
D2MAC vanished from view again. In Europe, the PAL system was given
a slight boost by the introduction of PALplus but this, too, did not achieve
much success in the TV set market, either. At the same time, various approaches were tried, mainly in Japan and in the US, to achieve success
with the transmission of HDTV, but these also failed to gain the universal
popular appeal hoped for.
In the studio, digital television signals have been used since the beginning of the nineties as uncompressed digital TV signals conforming to
“CCIR 601”. These data signals have a date rate of 270 Mbit/sec and are
highly suitable for distribution and processing in the studio, and are very
popular today. But they are not at all suitable for broadcasting and transmission to the end user. The channel capacities available via cable, terrestrial channels and satellite would not be even nearly adequate enough for
these signals. In the case of HDTV signals, the data rate is about 1.5
Gbit/sec uncompressed. Without compression, these signals could not be
broadcast.
The key event in the field of digital television can be considered to be
the establishment of the JPEG standard. JPEG stands for Joint Photographic Experts Group, a group of experts specializing in still frame compression. It was here that the discrete cosine transform (DCT) was used for
the first time for compressing still frames towards the end of the eighties.
Today, JPEG is a commonly used standard in the data field and is being
used very successfully in the field of digital photography. Digital cameras
are experiencing quite a boom and are becoming better and better so that
this medium has replaced traditional photography in many areas.
1 Introduction
3
The DCT also became the basic algorithm for MPEG, the Motion Picture Experts Group, which developed the MPEG-1 standard by 1993 and
the MPEG-2 standard by 1995. The aim of MPEG-1 was to achieve the reproduction of full-motion pictures at data rates of up to 1.44 Mbit/sec, using the CD as a data medium. The aim for MPEG-2 was higher and
MPEG-2, finally, was to become the baseband signal for digital television
world-wide. Initially, only Standard Definition Television (SDTV) was
provided for in MPEG-2, but High Definition Television (HDTV) was also
implemented which was apparently originally intended for MPEG-3.
However, there is no MPEG-3 (nor does it have anything to do with MP3
files, either). In MPEG-2, both the MPEG data structure was described
(ISO/IEC 13818-1) and a method for full-motion picture compression
(ISO/IEC 13818-2) and for audio compression (ISO/IEC 13818-3) defined.
These methods are now used throughout the world. MPEG-2 allows the
digital TV signals of originally 270 Mbit/sec to be compressed to about 2
to 7 Mbit/sec. The uncompressed data rate of a stereo audio signal of about
1.5 Mbit/sec, too, can be reduced to about 100 to 400 kbit/sec, typically to
192 kbit/s. As a result of these high compression factors it is now possible
even to combine a number of programs to form one data signal which can
then be accommodated in what was originally an e.g. 8-MHz-wide analog
TV channel.
In the meantime, there is MPEG-4, MPEG-7 and MPEG-21.
At the beginning of the nineties, Digital Video Broadcasting (DVB) was
then created as a European project. In the course of this project, several
transmission methods were developed: DVB-S, DVB-C and DVB-T. The
satellite transmission method DVB-S has been in use since about 1995.
Using the QPSK method of modulation and with channel bandwidths of
about 33…36 MHz, a gross data rate of 38 Mbit/sec is possible with satellite transmission. With approximately 6 Mbit/sec per program, up to 6, 8 or
even 10 programs can now be transmitted in one channel depending on
data rate and content and when mainly audio programs are broadcast, more
than 20 programs are often found in one channel. In the case of DVB-C,
transmitted via coaxial cable, the 64QAM modulation also provides a data
rate of 38 Mbit/sec at a bandwidth of only 8 MHz. Current HFC (hybrid
fibre coax) networks now allow data rates of more than 50 Mbit/s per
channel. DVB-C, too, has been in use since about 1995. The digital terrestrial TV system DVB-T started in 1998 in Great Britain in 2K mode and is
now available nationwide. This terrestrial path to broadcasting digital TV
signals is being used more and more, spreading from the UK, Scandinavia
and Spain all the way to Australia. DVB-T provides for data rates of between 5 to 31 Mbit/sec and the data rate actually used is normally about 22
to 22 Mbit/sec if a DVB-T network has been designed for roof antenna re-
4
1 Introduction
ception, or about 13 to 15 Mbit/sec for portable indoor use. Germany was
changing, region by region, from analog terrestrial TV to DVB-T. This
change-over was completed in Germany at the end of 2008.
In North America, other methods are in use. Instead of DVB-C, a very
similar system which conforms to ITU-J83B is used for cable transmission. Terrestrial transmission makes use of the ATSC method where ATSC
stands for Advanced Television System Committee. In Japan, too, other
transmission methods are used, such as ITU-J83C for cable transmission,
again very similar to DVB-C (which corresponds to ITU-J83A), and the
ISDB-T standard for terrestrial transmission. Yet another terrestrial transmission system is being developed in China. The common factor for all
these methods is the MPEG-2 baseband signal.
In 1999, another application was given the green light, namely the digital versatile disc, or DVD. The video DVD also uses an MPEG-2 data
stream with MPEG video and MPEG or Dolby Digital audio.
In the meantime, the range of digital television has been extended to
mobile reception with the development of standards for use with mobile
telephones, designated as DVB-H (Digital Video Broadcasting for Handhelds) and T-DMB (Terrestrial Digital Multimedia Broadcasting) and
CMMB.
This book deals with all present-day TV compression and transmission
methods, i.e. MPEG, DVB, ATSC, ISDB-T and DTMB. The video DVD
is also discussed to some extent. The discussion is focused on dealing with
these subjects in as practical a way as possible. Although mathematical
formulations are used, they are in most cases only utilized to supplement
the text. The mathematical ballast will be kept to a minimum for the practical field engineer. This has nothing to do with any possible aversion the
author may have against mathematics. Quite on the contrary. In the course
of many seminars involving thousands of participants throughout the
world, forms of presentation were developed which have contributed to a
better and easier understanding of these in some cases highly complex subjects. The book also contains chapters dealing with basic concepts such as
digital modulation or transformations into the frequency domain, some of
which can be skipped by a reader if he so desires. Experience has shown,
however, that it is better to read these chapters, too, before starting with
the actual subject of digital television. A major emphasis is placed on the
measuring techniques used on these various digital TV signals. Necessary
and appropriate measuring techniques are discussed in detail and practical
examples and hints are provided.
1 Introduction
5
Table 1.1. Digital video and audio broadcasting standards and methods
Method/Standards
JPEG
Motion JPEG
MPEG-1
MPEG-2
MPEG-4
DVB
DVB-S
DVB-C
DVB-T
DVB-H
MMDS
ITU-T J83A
ITU-T J83B
ITU-T J83C
ATSC
ISDB-T
DTMB
DAB
DRM
T-DMB
DVB-SH
DVB-S2
DVB-T2
DVB-C2
CMMB
Application
Still picture compression, photography
MiniDV, digital home video cameras
Video on CD
Baseband signal for digital television,
DVD Video
New codecs for video and audio compression
Digital Video Broadcasting
Digital Video Broadcasting via satellite
Digital Video Broadcasting via broadband cable
Digital Terrestrial Video Broadcasting
Digital Video Broadcasting for Heldhelds; mobile TV standard
Multipoint Microwave Distribution System, local terrestrial multipoint transmission of digital television to supplement broadband cable
ITU equivalent of DVB-C
US cable standard
Japanese cable standard
Standard for digital terrestrial television
(US, Canada)
Japanese standard for digital terrestrial
television
Chinese standard for digital terrestrial
television
Digital Audio Broadcasting; standard
for digital terrestrial audio broadcasting
Digital Radio Mondiale; standard for
digital terrestrial audio broadcasting
Terrestrial Digital Multimedia Broadcasting; mobile TV standard
DVB for handheld mobile terminals,
satellite and terrestrial, hybrid standard
for satellite and terrestrial broadcasting
Second Generation DVB for Satellite
Second Generation DVB Terrestrial
Second Generation DVB for Cable
Chinese
Mobile
Multimedia
Broadcasting
Note: Many of the terms listed in the table are protected by copyright ©
6
1 Introduction
As far as possible, practical findings and experiences have been incorporated time and again in the individual chapters. In some cases it will be
possible to recognize one or the other experience of the author on his travels. Particularly extensive practical insights were gained especially far
away from Europe in Australia during the introductory phase of DVB-T
and were written down in this book. But it is not intended to be a travel
guide to Australia or the world even though it would be very interesting to
tell about these areas and many beautiful locations where digital television
is being newly introduced. The content of this book is structured in such a
way that it starts with the analog TV baseband signal and then continues
with a discussion of the MPEG-2 data stream, digital video, digital audio
and the compression methods. After an excursion into the digital modulation methods, all the transmission methods like DVB-S, DVB-C, ITUJ83ABC, DVB-T, ATSC and ISDB-T are discussed in detail. Interspersed
between these are found the chapters on the relevant measuring technique.
As well, transmission methods based on Digital Audio Broadcasting
(DAB) are being discussed. The book deals more intensively with the subject of "digital audio broadcasting" with the DRM (Digital Radio
Mondiale) audio transmission standard and the possibilities of transmitting
digital audio also by DVB. Since it is no longer possible to separate the
subjects of television, audio broadcasting, in any case, the title of the book
has been changed to "Digital Video and Audio Broadcasting Technology,
A Practical Engineering Guide" from its second edition. Television still
claims the greater part, however. Part of the reason for this is that digital
audio broadcasting is still having problems with its practical implementation. It will be interesting to see what will be the end result of the competition between DAB and DVB-T2.
The methods and standards relating to the subject of “digital television
and digital audio broadcasting” and discussed in this book are listed in Table 1.1. In the meantime, new standards such as DVB-SH, DVB-T2 and
DVB-C2 have appeared which are also described as far as possible even if
there is only little practical experience available as yet. It should also not
be forgotten that digital television and digital audio can also be transmitted
via the Internet and IPTV is also mentioned briefly. Digital SDTV is now a
reality when it was only being introduced at the time the first English edition of this book appeared. It is now only necessary for HDTV to become
reality, and that not only by the complete range of technology now available but also by widely provided contents, i.e. available programs.
Bibliography: [ISO13818-1], [ISO13818-2], [ISO13818-3], [ETS300421],
[ETS300429],
[ETS300744],
[A53],
[ITU205],
[ETS300401],
[ETS101980]
2 Analog Television
Throughout the world, there are only two major analog television standards, the 625-line system with a 50 Hz frame rate and the 525-line system
with a 60 Hz frame rate. The composite color video-and-blanking signal
(CVBS, CCVS) of these systems is transmitted in the following color
transmission standards:
•
•
•
PAL (Phase Alternating Line)
NTSC (National Television System Committee)
SECAM (Séquentiel Couleur a Mémoire)
Vertical
blanking
ABC
Horizontal
blanking
Visible
lines
Visible
horizontal
part
Fig. 2.1. Dividing a frame into lines
PAL, NTSC and SECAM color transmission is possible in 625-line systems and in 525-line systems. However, not all the possible combinations
have actually been implemented. The video signal with its composite coding is then modulated onto a carrier, the vision carrier, mostly with negative-going amplitude modulation. It is only in Std. L (France) that positive-
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_2, © Springer-Verlag Berlin Heidelberg 2010
8
2 Analog Television
going modulation (sync inside) is used. The first and second sound subcarrier is usually an FM-modulated subcarrier but an amplitude-modulated
sound subcarrier is also used (Standard L, France). In Northern Europe, the
second sound subcarrier is a digitally modulated NICAM subcarrier. Although the differences between the methods applied in the various countries are only minor, together they result in a multiplicity of standards
which are mutually incompatible. The analog television standards are
numbered through alphabetically from A to Z and essentially describe the
channel frequencies and bandwidths in VHF bands I and III (47 ... 68
MHz, 174 ... 230 MHz) and UHF bands IV and V (470 ... 862MHz); An
example is Standard B, G Germany: B =7 MHz VHF, G = 8 MHz UHF.
In the television camera, each field is dissected into a line structure of
625 or 525 lines. Because of the finite beam flyback time in the television
receiver, however, a vertical and horizontal blanking interval became necessary and as a result, not all lines are visible but form part of the vertical
blanking interval. In a line, too, only a certain part is actually visible. In
the 625-line system, 50 lines are blanked out and the number of visible lines is 575. In the 525-line system, between 38 and 42 lines fall into the area of the vertical blanking interval.
To reduce the flickering effect, each frame is divided into two fields
combining the even-numbered lines and odd-numbered lines in each case.
The fields are transmitted alternately and together they result in a field repetition rate of twice the frame rate. The beginning of a line is marked by
the horizontal sync pulse, a pulse which is below the zero volt level in the
video signal and has a magnitude of -300 mV.. All the timing in the video
signal is referred to the front edge of the sync pulse and there exactly to the
50% point. 10 s after the sync pulse falling edge, the active image area in
the line begins in the 625-line system. The active image area itself has a
length of 52 s.
In the matrix in the television camera, the luminance (luminous density)
signal (Y signal or black/white signal) is first obtained and converted into
a signal having a voltage range from 0 Volt (corresponding to black level)
to 700 mV (100% white). The matrix in the television camera also produces the color difference signals from the Red, Green and Blue outputs. It
was decided to use color difference signals because, on the one hand, the
luminance has to be transmitted separately for reasons of compatibility
with black/white television and, on the other hand, color transmission had
to conserve bandwidth as effectively as possible. Due to the reduced color
resolution of the human eye, it was possible to reduce the bandwidth of the
color information. In fact, the color bandwidth is reduced quite significantly compared with the luminance bandwidth: The luminance bandwidth
2 Analog Television
9
is between 4.2 MHz (PAL M), 5 MHz (PAL B/G) and 6 MHz (PAL D/K,
L) whereas the chrominance bandwidth is only 1.3 MHz in most cases.
Fig. 2.2. Analog composite video signal (PAL)
Fig. 2.3. Vector diagram of a composite PAL video signal
10
2 Analog Television
In the studio, the color difference signals U=B-Y and V=R-Y are still
used directly. For transmission purposes, however, the color difference
signals U and V are vector modulated (IQ modulated) onto a color subcarrier in PAL and NTSC. In SECAM, the color information is transmitted
frequency-modulated. The common feature of PAL, SECAM and NTSC is
that the color information is modulated onto a color subcarrier of a higher
frequency which is placed at the upper end of the video frequency band
and is simply added to the luminance signal. The frequency of the color
subcarrier was selected such that it causes as little interference to the luminance channel as possible. It is frequently impossible, however, to avoid
crosstalk between luminance and chrominance and conversely, e.g. if a
newsreader is wearing a pinstriped suit. The colored effects which are then
visible on the pinstriped pattern are the result of this crosstalk (cross-color
or cross-luminance effects).
Vision terminals can have the following video interfaces:
•
•
•
•
CVBS, CCVS 75 Ohms 1 VPP (video signal with composite
coding)
RGB components (SCART, Peritel)
Y/C (separate luminance and chrominance to avoid cross color
or cross luminance effects)
In the case of digital television, it is advisable to use an RGB (SCART)
connection or a Y/C connection for the cabling between the receiver and
the TV monitor in order to achieve optimum picture quality.
In digital television only frames are transmitted, no fields. It is only at
the very end of the transmission link that fields are regenerated in the set
top box or in the decoder of the IDTV receiver. The original source material, too, is provided in interlaced format which must be taken into account
in the compression (field coding).
2.1 Scanning an Original Black/White Picture
At the beginning of the age of television, the pictures were only in “black
and white”. The circuit technology available in the 1950s consisted of tube
circuits which were relatively large and susceptible to faults and consumed
a lot of power. The television technician was still a real repairman and, in
the case of a fault, visited his customers carrying his box of vacuum tubes.
2.1 Scanning an Original Black/White Picture
11
Let us look at how such a black/white signal, the “luminance signal”, is
produced. Using the letter “A” as an example, its image is filmed by a TV
camera which scans it line by line (see Fig. 2.4.). In the early days, this
was done by a tube camera in which a light-sensitive layer, onto which the
image was projected by optics, was scanned line by line by an electron
beam deflected by horizontal and vertical magnetic fields.
1
2
3
4
5
6
7
8
9
10
11
12
700mV = white
Line 3
0mV = black
Fig. 2.4. Scanning an original black/white picture
Hblanking
700mV = white
0mV = black
H-sync
-300mV
Visible
part
Fig. 2.5. Inserting the horizontal sync pulse
Today, CCD (charge coupled device) chips are universally used in the
cameras and the principle of the deflected electron beam is now only pre-
12
2 Analog Television
served in TV receivers; and even there the technology is changing to LCD
and plasma screens.
The result of scanning the original is the luminance signal where 0 mV
corresponds to 100% black and 700 mV is 100% white. The original picture is scanned line by line from top to bottom, resulting in 625 or 525 active
lines depending on the TV standard used. However, not all lines are visible. Because of the finite beam flyback time, a vertical blanking interval
of up to 50 lines had to be inserted. In the line itself, too, only a certain
part represents visible picture content, the reason being the finite flyback
time from the right-hand to the left-hand edge of the line which results in
the horizontal blanking interval. Fig. 2.4. shows the original to be scanned
and Fig. 2.5. shows the associated video signal.
2.5 lines
Vertical sync puls
1 line
Begin of line 1 of 1st field
Center of line 313 of 2nd field
Fig. 2.6. Vertical synchronization pulse
2.2 Horizontal and Vertical Synchronization Pulses
However, it is also necessary to mark the top edge and the bottom edge of
the image in some way, in addition to the left-hand and right-hand edges.
This is done by means of the horizontal and vertical synchronization pulses. Both types of pulses were created at the beginning of the television
age so as to be easily recognizable and distinguishable by the receiver and
are located in the blacker than black region below zero volts.
The horizontal sync pulse (Fig. 2.5.) marks the beginning of a line. The
beginning is considered to be the 50% value of the front edge of the sync
pulse (nominally -150 mV). All the timing within a line is referred to this
time. By definition, the active line, which has a length of 52 s, begins
2.2 Horizontal and Vertical Synchronization Pulses
13
10 s after the sync pulse front edge. The sync pulse itself is 4.7 s long
and stays at -300 mV during this time.
At the beginning of television, the capabilities of the restricted processing techniques of the time which, nevertheless, were quite remarkable, had
to be sufficient. This is also reflected in the nature of the sync pulses. The
horizontal sync pulse (H sync) was designed as a relatively short pulse
(appr. 5 s) whereas the vertical sync pulse (V sync) has a length of 2.5
lines (appr. 160 s). In a 625-line system, the length of a line including H
sync is 64 s. The V sync pulse can, therefore, be easily distinguished
from H sync. The V sync pulse (Fig. 2.6.) is also in the blacker than black
region below zero volts and marks the beginning of a frame or field,
respectively.
Beginn of
field
Begin
1. 1st
H a lbbild
EndEof
2nd
n de
2. field
H a lbbild
0,3 V
S yn c.s ign a l
0V
622
623
624
625
End
field
E nof
de1st
1. H
a lbbild
1
2
3
4
5
6
Zeilen n u m m er
Beginnof2.2nd
field
Begin
H a lbbild
0,3 V
S yn c.s ign a l
0V
310
311 312
313
314
315
316
317 318
Zeilen n u m m er
Fig 2.7. Vertical synchronization pulses with pre- and post-equalizing pulses in
the 625 line-system
As already mentioned, a frame, which has a frame rate of 25 Hz = 25
frames per second in a 625-line system, is subdivided into 2 fields. This
makes it possible to cheat the eye, rendering flickering effects largely invisible. One field is made up of the odd-numbered lines and the other one is
made up of the even-numbered lines. They are transmitted alternatingly,
resulting in a field rate of 50 Hz in a 625-line system. A frame (beginning
of the first field) begins when the V sync pulse goes to the -300 mV level
for 2.5 lines at the precise beginning of a line. The second field begins
14
2 Analog Television
when the, V sync pulse drops to the -300 mV level for 2.5 lines at the center of line 313.
The first and second field are transmitted interlaced with one another,
thus reducing the flickering effect. Because of the limitations of the pulse
technology at the beginnings of television, a 2.5-line-long V sync pulse
would have caused the line oscillator to lose lock. For this reason, additional pre- and post-equalizing pulses were gated in which contribute to the
current appearance of the V sync pulse (Fig. 2.7.). Today’s signal processing technology renders these unnecessary.
Red
Green
Blue
Y
U
I
+
Matrix
V
CVBS
+
Q
90
PAL color subcarrier 4.43 MHz
Fig. 2.8. Block diagram of a PAL modulator
2.3 Adding the Color Information
At the beginning of the television age, black/white rendition was adequate
because the human eye has its highest resolution and sensitivity in the area
of brightness differences and the brain receives its most important information from these. There are many more black/white receptors than color
receptors in the retina. But just as in the cinema, television managed the
transition from black/white to color because its viewers desired it. Today
this is called innovation. When color was added in the sixties, knowledge
about the anatomy of the human eye was taken into consideration. With
only about 1.3 MHz, color (chrominance) was allowed much less resolution, i.e. bandwidth, than brightness (luminance) which is transmitted with
about 5 MHz. At the same time, chrominance is embedded compatibly into
2.3 Adding the Color Information
15
the luminance signal so that a black/white receiver was undisturbed but a
color receiver was able to reproduce both color and black/white correctly.
If a receiver falls short of these ideals, so-called cross-luminance and
cross-color effects are produced.
In all three systems, PAL, SECAM and NTSC, the Red, Green and Blue
color components are first acquired in three separate pickup systems (initially tube cameras, now CCD chips) and then supplied to a matrix where
the luminance signal is formed as the sum of R + G + B, and the chrominance signal. The chrominance signal consists of two signals, the color difference signals Blue minus luminance and Red minus luminance. However, the luminance signal and the chrominance signal formed must be
matrixed, i.e. calculated, provided correctly with the appropriate weighting
factors according to the eye’s sensitivity, using the following formula
Y = 0.3 • R + 0.59 • G + 0.ll • B;
U = 0.49 • (B-Y);
V = 0.88 • (R-Y);
The luminance signal Y can be used directly for reproduction by a
black/white receiver. The two chrominance signals are also transmitted
and are used by the color receiver. From Y, U and V it is possible to recover R, G and B. The color information is then available in correspondingly
reduced bandwidth, and the luminance information in greater bandwidth
(“paintbox principle”).
To embed the color information into a CVBS (composite video, blanking and sync) signal intended initially for black/white receivers, a method
had to be found which has the fewest possible adverse effects on a
black/white receiver, i.e. keeps it free of color information, and at the same
time contains all that is necessary for a color receiver.
Two basic methods were chosen, namely embedding the information
either by analog amplitude/phase modulation (IQ modulation) as in PAL
or NTSC, or by frequency modulation as in SECAM. In PAL and NTSC,
the color difference signals are supplied to an IQ modulator with a reduced
bandwidth compared to the luminance signal (Fig. 2.8.) The IQ modulator
generates a chrominance signal as amplitude/phase modulated color subcarrier, the amplitude of which carries the color saturation and the phase of
which carries the hue. An oscilloscope would only show, therefore, if there
is color, and how much, but would not identify the hue. This would require
a vectorscope which supplies information on both.
In PAL and in NTSC, the color information is modulated onto a color
subcarrier which lies within the frequency band of the luminance signal
16
2 Analog Television
but is spectrally intermeshed with the latter in such a way that it is not visible in the luminance channel. This is achieved by the appropriate choice
of color subcarrier frequency. In PAL (Europe), the color subcarrier frequency was chosen by using the following formula
fSC = 283.75 • fH + 25 Hz = 4.43351875 MHz;
Fig. 2.9. Oscillogram of a CVBS, CCVS (composite color video and sync) signal
In SECAM, the frequency modulated color difference signals are alternately modulated onto two different color subcarriers from line to line. The
SECAM process is currently only used in France and in French-speaking
countries in North Africa, and also in Greece. Countries of the previous
Eastern Block changed from SECAM to PAL in the nineties.
Compared with NTSC, PAL has a great advantage due to its insensitivity to phase distortion because its phase changes from line to line. The color
cannot be changed by phase distortion on the transmission path, therefore.
NTSC is used in analog television, mainly in North America, where it is
sometimes ridiculed as “Never Twice the Same Color” because of the color distortions.
The composite PAL, NTSC or SECAM video signal (Fig. 2.9.) is generated by mixing the black/white signal, the sync information and the chrominance signal and is now called a CCVS (Composite Color, Video and
2.4 Transmission Methods
17
Sync) signal. Fig. 2.9. shows the CCVS signal of a color bar signal. The
color burst can be seen clearly. It is used for conveying the reference phase
of the color subcarrier to the receiver so that its color oscillator can lock to
it.
CVBS
AM
Audio 1
FM 1
Audio 2
FM 2
VSB
f
+
VSB = vestigialside band filter
Fig. 2.10. Principle of a TV modulator for analog terrestrial TV and analog TV
broadband cable
2.4 Transmission Methods
Analog television is disseminated over three transmission paths which are:
terrestrial transmission paths, via satellite and by broadband cable. The
priority given to any particular transmission path depends greatly on the
countries and regions concerned. In Germany, the traditional analog “antenna TV” has currently only a minor status with fewer than 10%, this
term being used mainly by the viewers themselves whereas the actual
technical term is “terrestrial TV”. The reason for this is the good coverage
by satellite and cable, and more programs. This will change when DVB-T
is introduced as has already become apparent in some regions.
Transmission of analog television via terrestrial and satellite paths will
shrivel away into insignificance within a few years. Whether this will also
be true of broadband cable cannot yet be predicted.
In the terrestrial transmission of analog TV signals, and that by cable,
the modulation method used is amplitude modulation, in most cases with
18
2 Analog Television
negative modulation. Positive modulation is only used in the French Standard L.
The sound subcarriers are frequency modulated in most cases. To save
bandwidth, the vision carrier is VSB-AM (vestigial sideband amplitude
modulation) modulated, i.e. a part of the spectrum is suppressed by bandpass filtering. The principle is shown in Fig. 2.10. and 2.11. Because of the
nonlinearities and the low signal/noise ratio on the transmission link, frequency modulation is used in satellite transmission.
Since these analog transmission paths are losing more and more in
significance, they will not be discussed in greater detail in this book and
the reader is referred to the appropriate literature, instead.
Power =
sync peak power
Mod.
CVBS
Vision
carrier
10% Residual
picture carrier
Fig. 2.11. Vision modulator
2.5 Distortion and Interference
Over the entire transmission link, an analog video signal is subjected to influences which have a direct effect on its quality and are immediately visible in most cases. These distortions and interferences can be roughly
grouped in the following categories:
•
•
•
Linear distortion (amplitude and phase distortion)
Non-linear distortion
Noise
2.5 Distortion and Interference
•
•
19
Interference
Intermodulation
Linear distortion is caused by passive electronic components. The amplitude or group delay is no longer constant over a certain frequency range
which is 0 ... 5 MHz in the case of video. Parts of the relevant frequency
range are distorted to a greater or lesser extent, depending on the characteristic of the transmission link involved. As a result, certain signal components of the video signal are rounded. The worst effect is rounding of the
sync pulses which leads to synchronization problems in the TV receiver
such as, e.g. horizontal "pulling" or "rolling" of the picture from top to bottom. These terms have been known since the early days of television.
Changing of heads from field to field produces similar effects at the top
edge of the picture with some older videorecorders, the picture is "pulling".
These effects have become relatively rare thanks to modern receiver
technology and relatively good transmission techniques. In the active picture area, linear distortion manifests itself either as lack of definition, ringing, optical distortion or displacement of the color picture with respect to
the luminance picture.
Nonlinear distortion can be grouped into
•
•
•
Static nonlinearity
Differential gain
Differential phase
With non-linear distortion, neither the gray steps nor the color subcarrier
are reproduced correctly in amplitude and phase. Non-linear distortion is
caused by active components (transmitter tubes, transistors) in the transmission link. However, they become only visible ultimately when many
processes are added together since the human eye is very tolerant in this
respect. Putting it another way: "Although this isn’t the right gray step,
who is to know?". And in color television this effect is less prominent, in
any case, because of the way in which color is transmitted, particularly
with PAL.
One of the most visible effects is the influence of noise-like disturbances. These are simply produced by superimposition of the ever-present
gaussian noise, the level of which is only a question of its separation from
the useful signal level. I.e., if the signal level is too low, noise becomes visible. The level of thermal noise can be determined in a simple way via the
Boltzmann constant, the bandwidth of the useful channel and the normal
20
2 Analog Television
ambient temperature and is thus almost a fixed constant. Noise is immediately visible in the analog video signal which is the great difference compared with digital television.
Intermodulation products and interference are also very obvious in the
video signal and have a very disturbing effect, forming moiré patterns in
the picture. These effects are the result of heterodyning of the video signal
with an interfering product either from an adjacent channel or interferers
entering the useful spectrum directly from the environment. This type of
interference is the one most visible and thus also causes the greatest
disturbance in the overall impression of the picture. It is also most apparent
in cable television because of its multichannel nature.
2.6 Signals in the Vertical Blanking Interval
Since the middle of the seventies, the vertical blanking interval, which was
originally used for the vertical flyback, is no longer only "empty" or
"black". At first, so-called VITS (vertical insertion test signals), or test lines, were inserted there which could be used for assessing the quality of
the analog video signal. In addition, teletext and the data line can be found
there. Test lines were and are used for monitoring the transmission quality
of a TV transmission link or section virtually on-line without having to
isolate the link. These test lines contain test signals which can be used for
identifying the causes of faults.
Fig. 2.12. CCIR 17 and 330 test lines
Test line "CCIR 17" (now ITU 17, on the left in Fig. 2.12.) begins with
the so-called white pulse (bar) and is used as technical voltage reference
2.6 Signals in the Vertical Blanking Interval
21
for 100% white. Its nominal amplitude is 700 mV. The "roof" of the white
pulse is 10 µs long and should be flat and without overshoots. This is followed by the 2T pulse which is a so-called cos2 pulse with a halfamplitude period of 2T 2 • 100 ns = 200 ns. The main components of its
spectrum extend to the end of the luminance channel of 5 MHz. It reacts
very sensitively to amplitude response and group delay distortion from 0 ...
5 MHz and can thus be used for assessing linear distortion both visually
and by measurement. The next pulse is a 20T pulse, a cos2 pulse with superimposed color subcarrier and with a half-amplitude period of 20T = 20
• 100 ns = 2 s. It clearly shows linear distortion of the color channel with
respect to the luminance channel.
Linear distortion of the color channel with respect to the luminance
channel is
•
•
Differential gain of the color channel with respect to the luminance channel
Luminance-chrominance delay caused by group delay
Non-linear distortion can be easily identified by means of the 5-step
gray scale. All five steps must have identical height. If they do not have
equal height due to nonlinearities, this is called static nonlinearity (luminance nonlinearity). In test line 330, the gray scale is replaced by a staircase on which the color subcarrier is superimposed. This can be used for
identifying non-linear effects on the color cubcarrier such as differential
amplitude and phase. The color bursts superimposed on the staircase
should all be ideally of the same amplitude and must not have a phase discontinuity at the transition points of the steps.
Teletext is well known by now (Fig. 2.13. and 2.14.). It is a data service
offered in analog television. The data rate is about 6.9 Mbit/s, but only in
the area of the lines really used in the vertical blanking interval. In actual
fact, the data rate is much lower. In each teletext line, 40 useful characters
are transmitted. A teletext page consists of 40 characters times 24 lines. If
the entire vertical blanking interval were to be used, just short of one teletext page could be transmitted per field. Teletext is transmitted in NRZ
(non-return-to-zero) code. A teletext line begins with the 16-bit-long runin, a sequence of 10101010… for synchronizing the phase of the teletext
decoder in the receiver. This is followed by the framing code. This hexadecimal number 0xE4 marks the beginning of the active teletext. After the
magazine and line number, the 40 characters of a line of the teletext are
transmitted. One teletext page consists of 24 text lines.
22
2 Analog Television
Fig. 2.13. Teletext line
40 character
24
lines
Fig. 2.14. Teletext page
The most important teletext parameters are as follows:
•
•
•
•
•
Non-return-to-zero code
Data rate: 444 • 15625 kbit/s = 6.9375 Mbit/s
Error protection: Even parity
Characters per line: 40
Lines per teletext page: 24
The data line (e.g. line 16 and corresponding line in the second field,
Fig. 2.15.) is used for transmitting control information, signaling and,
2.6 Signals in the Vertical Blanking Interval
23
among other things, the VPS (video program system) data for controlling
video recorders. In detail, the data line is used for transmitting the following data:
•
•
•
•
•
•
•
•
•
•
•
•
Byte 1: Run-in 10101010
Byte 2: Start code 01011101
Byte 3: Source ID
Byte 4: Serial ASCII text transmission (source)
Byte 5: Mono/stereo/dual sound
Byte 6: Video content ID
Byte 7: Serial ASCII text transmission
Byte 8: Remote control (routing)
Byte 9: Remote control (routing)
Byte 10: Remote control
Byte 11 to 14: Video program system (VPS)
Byte 15: Reserved
Fig. 2.15. Data line (mostly line 16 in the vertical blanking interval)
The VPS bytes contain the following information:
•
•
•
•
•
•
Day (5 bits)
Month (4 bits)
Hour (5 bits)
Minute (6 bits) = virtual starting time of the program
Country ID (4 bits)
Program source 11) (6 bits)
24
2 Analog Television
The transmission parameters of the data line are:
•
•
•
•
•
Line: 16/329
Code: Return-to-zero code
Data rate: 2.5 Mbit/s
Level: 500 mV
Data: 15 bytes per line
According to DVB, these signals from the vertical blanking interval are
partially regenerated in the receiver to retain compatibility with analog
television. The test line signals, however, are no longer provided.
2.7 Measurements on Analog Video Signals
Analog video signals have been measured since the beginning of the TV
age, initially with simple oscilloscopes and vectorscopes and later with ever more elaborate video analyzers, the latest models of which were digital
(Fig. 2.22.). These video measurements are intended to identify the distortions in the analog video signal. The following test parameters are determined with the aid of test lines:
•
•
•
•
•
•
•
•
•
•
•
•
•
White bar amplitude
Sync amplitude
Burst amplitude
Tilt on the white bar
2T pulse amplitude
2T K factor
Luminance-chrominance amplitude on the 20T pulse
Luminance-chrominance delay on the 20T pulse
Static nonlinearity on the grayscale
Differential gain on the grayscale with color subcarrier
Differential phase on the grayscale with color subcarrier
Weighted and unweighted luminance signal/noise ratio
Hum
In addition, an analog TV test receiver also provides information on:
•
Vision carrier level
2.7 Measurements on Analog Video Signals
•
•
•
•
•
25
Sound carrier level
Deviation of the sound carriers
Frequencies of vision and sound carriers
Residual picture carrier
ICPM (Incidential phase modulation)
17µs
A
White bar
amplitude
= A – B;
B
37µs
H sync (50% falling edge)
B=black
level
Fig. 2.16. Measuring the white bar amplitude
Fig. 2.17. Sync pulse and burst
The most important parameter to be measured on an analog TV signal is
the white bar amplitude which is measured as shown in Fig. 2.16. In the
worst case, the white bar can also be quite rounded due to linear distortions, as indicated in the figure. The sync amplitude (s. Fig. 2.17.) is used as
26
2 Analog Television
voltage reference in the terminals and is of special importance for this reason. The sync amplitude is nominally 300 mV below black. The 50% value of the falling edge of the sync pulse is considered to be the timing reference in the analog video signal. The burst (s. Fig. 2.17.) is used as voltage
and phase reference for the color subcarrier. Its amplitude is 300 mVPP. In
practice, amplitude distortions of the burst have little influence on the picture quality.
Linear distortion leads to tilt on the white bar (Fig. 2.18.). This is also an
important test parameter. To measure it, the white bar is sampled at the beginning and at the end and the difference is calculated which is then related
to the white pulse amplitude.
The 2T pulse reacts sensitively to linear distortion in the entire transmission channel of relevance. Fig. 2.19. shows the undistorted 2T pulse on the
left. It has been used as test signal for identifying linear distortion since the
seventies. A 2T pulse altered by linear distortion is also shown on the right
in Fig. 2.19. If the distortion of the 2T pulse is symmetric, it is caused by
amplitude response errors. If the 2T pulse appears to be unsymmetric, then
group delay errors are involved (non-linear phase response).
13µs
A
B
21µs
(B-A)
Tilt = ----------- * 100%
bar amplitude
H sync (50% falling edge)
Fig. 2.18. Tilt on the white bar
The 20T pulse (Fig. 2.20., center) was created especially for measurements in the color channel. It reacts immediately to differences between
luminance and chrominance. Special attention must be paid to the bottom
of the 20T pulse. It should be straight, without any type of indentation. In
2.7 Measurements on Analog Video Signals
27
the ideal case, the 20T pulse, like the 2T pulse, should have the same magnitude as the white pulse (700 mV nominal).
Fig. 2.19. Undistorted (left) and distorted (right) 2T pulse
Fig. 2.20. Linearly distorted white pulse, 2T pulse and 20T pulse
Nonlinearities distort the video signal in dependence on modulation.
This can be shown best on staircase signals. To this end, the gray scale and
the staircase with color subcarrier were introduced as test signal, the steps
simply being of different size in the presence of nonlinearitics. Noise and
intermodulation can be verified best in a black line (Fig. 2.21.). In most cases, line 22 was kept free of information for this purpose but this is not necessarily so any longer, either, since it carries teletext in most cases. To
measure these effects, it is only necessary to look for an empty line suitable for this purpose among the 625 or 525 lines and this differs from
program to program.
28
2 Analog Television
Luminance noise measurement
in a “black line“
Fig. 2.21. Luminance noise measurement in a ”black line“
Fig. 2.22. Analog video test and measurement equipment: video test signal generator and video analyzer (Rohde&Schwarz SAF and VSA)
In digital television, test line testing now only makes sense for assessing
the beginning (studio equipment) and the end (receiver) of the transmis-
2.8 Analog and Digital TV in a Broadband Cable Network
29
sion link. In between - on the actual transmission link - nothing happens
that can be verified by this means. The corresponding measurements on
the digital transmission links will be described in detail in the respective
chapters.
Fig. 2.23. TV Analyzer Rohde&Schwarz ETL for analog and digital TV measurements; ETL offers spectrum analyzer functionality and analog and digital TV
measurements
2.8 Analog and Digital TV in a Broadband Cable Network
There is still both analog and digital TV and FM radio in a broadband cable network. That means analog TV and analog TV measurements are still
a topic today.
Fig. 2.23. and 2.24. shows a current example of a mix of analog FM
audio and analog and digital TV channels in a broadband cable network
(year 2008, Germany and Austria).
30
2 Analog Television
Fig. 2.24. Analog and digital broadband cable channels (example Munich, Germany, 0 … 800 MHz)
Fig. 2.25. Analog and digital broadband cable channels (example Klagenfurt, Austria, 0 … 1000 MHz)
Bibliography: [MÄUSL3], [MÄUSL5], [VSA], [FISCHER6], [ETL]
3 The MPEG Data Stream
The abbreviation MPEG, first of all, stands for Moving Pictures Experts
Group, that is to say MPEG deals mainly with the digital transmission of
moving pictures. However, the data signal defined in the MPEG-2 Standard can also generally carry data which have nothing at all to do with
video and audio and could be Internet data, for example. And indeed,
throughout the world there are MPEG applications in which it would be
futile to look for video and audio signals. Thus, in Wollongong, about 70
km south of Sydney in Australia, an Australian pay TV provider is operating a pure data broadcasting service using MPEG-2 data signals via
MMDS (Microwave Multipoint Distribution System). “Austar” are here
providing their customers with fast Internet links in the Mbit/s range.
MPEG = Moving Pictures Expert Group
MPEG-7
Metadata,
XML based
ISO/IEC15938
“Multimedia
Content
Part2: video
Part2: video
Part2: video
ISO/IEC11172-2 ISO/IEC13818-2 ISO/IEC14496-2 Description
Interface“
Part3: audio
Part3: audio
Part3: audio
ISO/IEC11172-3 ISO/IEC13818-3 (AAC)
ISO/IEC14496-3
MPEG-1
Part1: systems
ISO/IEC11172-1
“PES layer“
MPEG-4
MPEG-2
Part1: systems
Part1: systems
ISO/IEC13818-1 ISO/IEC14496
“Transportation“
MPEG-21
additional
“tools“
ISO/IEC21000
Part6: DSM-CC Part10: video
ISO/IEC13818-6 (AVC, H.264)
ISO/14496-10
Part7: AAC
ISO/IEC13818-7
Fig. 3.1. MPEG standards
As in the MPEG Standard itself, first the general structure of the MPEG
data signal will be described in complete isolation from video and audio.
An understanding of the data signal structure is also of greater importance
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_3, © Springer-Verlag Berlin Heidelberg 2010
32
3 The MPEG Data Stream
in practice than a detailed understanding of the video and audio coding
which will be discussed later.
Luminance
sampling frequency
13.5 MHz
5.75 MHz
8/10 Bit
Matrix
Y
R
G
B
A
Cb
A
Cr
A
D
Y
8/10 Bit
Cb
D
8/10 Bit
Cr
D
270 Mbit/s
ITU-BT.R 601
„CCIR601“
2.75 MHz
6.75 MHz
Chrominance
sampling frequency
16 Bit
Right
A
up to 768 kbit/s
D
15 ... 20 kHz
32/44.1/48 kHz
bandwidth
Sampling frequency
= approx.
1.5 Mbit/s
16 Bit
Left
A
D
up to 768 kbit/s
15 ... 20 kHz
32/44.1/48 kHz
bandwidth
Sampling frequency
Fig. 3.2. Video and audio data signals
All the same, the description of the data signal structure will begin with
the uncompressed video and audio signals. An SDTV (Standard Definition
Television) signal without data reduction has a data rate of 270 Mbit/s and
a digital stereo audio signal in CD quality has a data rate of about 1.5
Mbit/s (Fig. 3.2.).
The video signals are compressed to about 1 Mbit/s in MPEG-1 and to
about 2 - 7 Mbit/s in MPEG-2. The video data rate can be constant or variable (statistical multiplex). The audio signals have a data rate of about 100
- 400 kbit/s (mostly 192 kbit/s) after compression (to be discussed in a
3 The MPEG Data Stream
33
separate chapter) but the audio data rate is always constant and a multiple
of 8 kbit/s. The compression itself will be dealt with in a separate chapter.
The compressed video and audio signals in MPEG are called “elementary
streams”, ES in brief. There are thus video streams, audio streams and,
quite generally, data streams, the latter containing any type of compressed
or uncompressed data. Immediately after having been compressed (i.e. encoded), all the elementary streams are divided into variable-length packets,
both in MPEG-1 and in MPEG-2 (Fig. 3.3.).
var. length
up to 64 kbyte
at MPEG-1
PES packet
Video PES
PES header
Audio PES
Data PES
Fig. 3.3. MPEG Elementary Streams
Since it is possible to have sometimes more and sometimes less compression depending on the instantaneous video and audio content, variablelength containers are needed in the data signal. These containers carry one
or more compressed frames in the case of the video signal and one or more
compressed audio signal segments in the case of the audio signal. These
elementary streams (Fig. 3.3.) thus divided into packets are called “packetized elementary streams”, or simply PES for short. Each PES packet
usually has a size of up to 64 kbytes. It consists of a relatively short header
and of a payload. The header contains inter alia a 16-bit-long length indicator for the maximum packet length of 64 kbytes. The payload part contains either the compressed video and audio streams or a pure data stream.
According to the MPEG Standard, however, the video packets can also be
longer than 64 kbytes in some cases. The length indicator is then set to
34
3 The MPEG Data Stream
zero and the MPEG decoder has to use other mechanisms for finding the
end of the packet.
max. 64 kbyte + 6
max. 64 kbyte
payload
6 byte
header
Optional
PES
header
PES header
3 Byte start
code prefix
00 00 01
Stream ID
PES packet
length
24
8
16
max. 64 kByte + 6
max. 64 kbyte
payload
6 Byte
header
Optional
PES
header
PES Header
„10“
PES
scrambling
control
2
11 flags
PES header
data length
12
8
2
ESCR
33
33
42
Stuffing
bytes „FF“
bit
Optional
PES
header
PES header
DTS
Optional
fields depending
on flags
max. 64 kbyte + 6
max. 64 kbyte
payload
6 byte
header
PTS
bit
ES
rate
22
DSM
trick
mode
8
Additional
copy
info
8
Previous
PES
CRC
16
Optional fields inside of optional PES header
Fig. 3.4. The PES packet
PES
extension
bit
3.1 The Packetized Elementary Stream (PES)
35
3.1 The Packetized Elementary Stream (PES)
All elementary streams in MPEG are first packetized in variable-length
packets called PES packets. The packets, which primarily have a length of
64 kbytes, begin with a PES header of 6 bytes minimum length. The first 3
bytes of this header represent the “start code prefix”, the content of which
is always 00 00 01 and which is used for identifying the start of a PES
packet. The byte following the start code is the “stream ID” which describes the type of elementary stream following in the payload. It indicates
whether it is, e.g. a video stream, an audio stream or a data stream which
follows. After that there are two “packet length” bytes which are used to
address the up to 64 kbytes of payload. If both of these bytes are set to
zero, a PES packet having a length which may exceed these 64 kbytes can
be expected. The MPEG decoder then has to use other arrangements to
find the PES packet limits, e.g. the start code.
After these 6 bytes of PES header, an “optional PES header” is transmitted which is an optional extension of the PES header and is adapted to the
requirements of the elementary stream currently being transmitted. It is
controlled by 11 flags in a total of 12 bits in this optional PES header.
These flags show which components are actually present in the “optional
fields” in the optional PES header and which are not. The total length of
the PES header is shown in the “PES header data length” field. The optional fields in the optional header contain, among other things, the “Presentation Time Stamps” (PTS) and the “decoding time stamps” (DTS)
which are important for synchronizing video and audio. At the end of the
optional PES header there may also be stuffing bytes. Following the complete PES header, the actual payload of the elementary stream is transmitted which can usually be up to 64 kbytes long or even longer in special
cases, plus the optional header.
In MPEG-1, video PES packets are simply multiplexed with PES packets and stored on a data medium (Fig. 3.5.). The maximum data rate is
about 1.5 Mbit/s for video and audio and the data stream only includes a
video stream and an audio stream.
This “Packetized Elementary Stream” (PES) with its relatively long
packet structures is not, however, suitable for transmission and especially
not for broadcasting a number of programs in one multiplexed data signal.
In MPEG-2, on the other hand, the objective has been to assemble up to
6, 10 or even 20 independent TV or radio programs to form one common
multiplexed MPEG-2 data signal. This data signal is then transmitted via
satellite, cable or terrestrial transmission links. To this end, the long PES
packets are additionally divided into smaller packets of constant-length.
36
3 The MPEG Data Stream
From the PES packets, 184-byte-long pieces are taken and to these another
4-byte-long header is added (Fig. 3.6.), making up 188-byte-long packets
called “transport stream packets” which are then multiplexed.
... Video PES Audio
PES
V
A
V
V
...
Multiplexed video and audio PES packets
Application:
MPEG-1 Video CD
MPEG-2 SVCD
MPEG-2 Video DVD
Fig. 3.5. Multiplexed PES packets
PES
header
PES
header
Packetized elementary stream
Transport stream
Payload
unit start
indicator = 1
4 byte
TS header
184 byte
payload
Payload
unit start
indicator = 1
Fig. 3.6. Forming MPEG-2 transport stream packets
To do this, first the transport stream packets of one program are multiplexed together. A program can consist of one or more video and audio
signals and an extreme example of this is a Formula 1 transmission with a
3.1 The Packetized Elementary Stream (PES)
37
number of camera angles (track, spectators, car, helicopter) and presented
in different languages. All the multiplexed data streams of all the programs
are then multiplexed again and combined to form a complete data stream
which is called an “MPEG-2 transport stream” (TS for short).
Program 1
Encoder
Audio1
Video2
Program 2
Encoder
Audio2
Video3
Encoder
Program 3
PID=0x200
MPEG-2 multiplexer
Video1
PID=0x300
MPEG-2 TS
PID=0x100
PID = Packet identifier
Audio3
Fig. 3.7. Multiplexed MPEG-2 transport stream packets
An MPEG-2 transport stream contains the 188-byte-long transport
stream packets of all programs with all their video, audio and data signals.
Depending on the data rates, packets of one or the other elementary
streams will occur more or less frequently in the MPEG-2 transport
stream. For each program there is one MPEG encoder which encodes all
elementary streams, generates a PES structure and then packetizes these
PES packets into transport stream packets. The data rate for each program
is usually approx. 2 - 7 Mbit/s but the aggregate data rate for video, audio
and data can be constant or vary in accordance with the program content at
the time. This is then called “statistical multiplex”. The transport streams
of all the programs are then combined in a multiplexed MPEG-2 data
stream to form one overall transport stream (Fig. 3.7.) which can then have
a data rate of up to about 40 Mbit/s. There are often up to 6, 8 or 10 or
even 20 programs in one transport stream. The data rates can vary during
the transmission but the overall data rate has to remain constant. A program can contain video and audio, only audio (audio broadcast) or only
data, and the structure is thus flexible and can also change during the transmission. To be able to determine the current structure of the transport
38
3 The MPEG Data Stream
stream during the decoding, the transport stream also carries lists describing the structure, so-called “tables”.
3.2 The MPEG-2 Transport Stream Packet
The MPEG-2 transport stream consists of packets having a constant length
(Fig. 3.8.). This length is always 188 bytes, with 4 bytes of header and 184
bytes of payload. The payload contains the video, audio or general data.
The header contains numerous items of importance to the transmission of
the packets. The first header byte is the “sync byte”. It always has a value
of 47hex (0x47 in C/C++ syntax) and is spaced a constant 188 bytes apart in
the transport stream. It is quite possible, and certainly not illegal, for there
to be a byte having the value 0x47 somewhere else in the packet.
188 byte
4 Byte
TS Header
184 byte
payload
13 bit packet identifier = PID
1 bit transport error indicator
1 byte sync byte = 0x47
Fig. 3.8. MPEG-2 transport stream packet
The sync byte is used for synchronizing the packet to the transport
stream and it is its value plus the constant spacing which is being used for
synchronization. According to MPEG, synchronization at the decoder occurs after five transport stream packets have been received. Another important component of the transport stream is the 13 bit-long “packet identifier” or PID for short. The PID describes the current content of the payload
part of this packet. The hexadecimal 13 bit number in combination with
tables also included in the transport stream show which elementary stream
or content this is.
3.2 The MPEG-2 Transport Stream Packet
Transmission
link
MPEG-2
TS
DVB /
ATSC
mod.
RS
(DVB)
DVB /
ATSC
demod.
39
MPEG-2
TS
RS
(ATSC)
204 or 208 byte
(DVB)
4 byte
header
184 byte
payload
(ATSC)
16 or 20 byte
RS FEC
188 byte
Sync byte 0x47
1 bit transport error indicator
184 byte
payload
4 byte
header
188 byte
Fig. 3.9. Reed-Solomon FEC
The bit immediately following the sync bit is the “transport error indicator” bit (Fig. 3.8.). With this bit, transport stream packets are flagged as errored after their transmission. It is set by demodulators at the end of the
transmission link if e.g. too many errors have occurred and there had been
no further possibility to correct these by means of error correction mechanisms used during the transmission. In DVB (Digital Video Broadcasting),
e.g., the primary error protection used is always the Reed Solomon error
correction code (Fig. 3.9.). In one of the first stages of the (DVB-S, DVBC or DVB-T) modulator, 16 bytes of error protection are added to the initially 188 bytes of the packet. These 16 bytes of error protection are a special checksum which can be used for repairing up to 8 errors per packet at
the receiving end. If, however, there are more than 8 errors in a packet,
there is no further possibility for correcting the errors, the error protection
has failed and the packet is flagged as errored by the transport error indica-
40
3 The MPEG Data Stream
tor. This packet must now no longer be decoded by the MPEG decoder
which, instead, has to mask the error which, in most cases, can be seen as a
type of “blocking“ in the picture.
It may be necessary occasionally to transmit more than 4 bytes of header
per transport stream packet. The header is extended into the payload field
in this case. The payload part becomes correspondingly shorter but the total packet length remains a constant 188 bytes. This extended header is
called an “adaptation field” (Fig. 3.10.). The other contents of the header
and of the adaptation field will be discussed later. “Adaptation control
bits” in the 4 byte-long header show if there is an adaptation field or not.
188 byte
4 byte
header
184 byte
payload
Optional
adaptation
field
Header
Adaption
Discontinuity ...
field
indicator
length
8
5 flags
1
5
Optional
fields
dep. on
flags
...
PCR
42
...
bit
Fig. 3.10. Adaptation field
The structure and especially the length of a transport stream packet are
very similar to a type of data transmission known from telephony and
LAN technology, namely the “asynchronous transfer mode” or ATM in
short. Today, ATM is used both in long-haul networks for telephony and
Internet calls and for interconnecting computers in a LAN network in
buildings. ATM also has a packet structure. The length of one ATM cell is
53 bytes containing 5 bytes of header and 48 bytes of payload. Right at the
beginning of MPEG-2 it was considered to transmit MPEG-2 data signals
via ATM links. Hence the length of an MPEG-2 transport stream packet.
Taking into consideration one special byte in the payload part of an ATM
cell, this leaves 47 bytes of payload data. It is then possible to transmit 188
bytes of useful information by means of 4 ATM cells, corresponding exactly to the length of one MPEG-2 transport stream packet. And indeed,
MPEG-2 transmissions over ATM links are nowadays a fact of life. Ex-
3.2 The MPEG-2 Transport Stream Packet
41
amples of this are found, e.g. in Austria where all national studios of the
Austrian broadcasting institution ORF (Österreichischer Rundfunk) are
linked via an ATM network (called LNET). In Germany, too, MPEG
streams are exchanged over ATM links.
ATM = Asynchronous Transfer Mode
53 byte
5 byte
header
48 byte
payload
(48 - 1) * 4 = 188 byte
= MPEG-2 transportstrom
stream packet length
188 byte MPEG-2 TS packet
47 byte
payload
5 byte header
47 byte
payload
47 byte
payload
47 byte
payload
4 ATM cells
1 byte spec. information
Fig. 3.11. ATM cell
When MPEG signals are transmitted via ATM links, various transmission modes called ATM Adaptation Layers can be applied at the ATM
level. The mode shown in Fig. 3.11. corresponds to ATM Adaptation
Layer 1 without FEC (i.e. AAL1 without FEC (forward error correction)).
ATM Adaptation Layer 1 with FEC (AAL1 with FEC) or ATM Adaptation Layer 5 (AAL5) are also possible. The most suitable layer appears to
be AAL1 with FEC since the contents are error-protected during the ATM
transmission in this case.
The fact that the MPEG-2 transport stream is a completely asynchronous data signal is of particularly decisive significance. There is no way of
knowing what information will follow in the next time slot (= transport
stream packet). This can only be determined by means of the PID of the
transport stream packet. The actual payload data rates in the payload can
fluctuate; there may be stuffing to supplement the missing 184 bytes. This
asynchronism has great advantages with regard to future flexibility, mak-
42
3 The MPEG Data Stream
ing it possible to implement any new method without much adaptation.
But there are also disadvantages: the receiver must always be monitoring
and thus uses more power; unequal error protection as, e.g., in DAB (digital audio broadcasting) cannot be applied and different contents can not be
protected to a greater or lesser degree as required.
Transport stream
synchronization
Sync byte
0x47
Read-out of
TS content
Program Specific
Information PAT, PMT
Accessing a
program
Packet identification
PID
Descrambling,
if required
Conditional Access
Table CAT
Program
synchronization
Program Clock Ref.
PCR, PTS, DTS
Decoding
additional data
Service Information
SI
Fig. 3.12. Information for the receiver
3.3 Information for the Receiver
In the following paragraphs, the components of the transport stream which
are necessary for the receiver will be considered. Necessary components
means in this case: What does the receiver, i.e. the MPEG decoder, need
for extracting from the large number of transport stream packets with the
most varied contents exactly those which are needed for decoding the desired program? In addition, the decoder must be able to synchronize correctly to this program. The MPEG-2 transport stream is a completely asynchronous signal and its contents occur in a purely random fashion or on
demand in the individual time slots. There is no absolute rule which can be
used for determining what information will be contained in the next transport stream packet. The decoder and every element on the transmission
link must lock to the packet structure. The PID (packet identifier) can be
3.3 Information for the Receiver
43
used for finding out what is actually being transmitted in the respective
element. On the one hand, this asynchronism has advantages because of
the total flexibility provided but there are also disadvantages with regard to
power saving. Every single transport stream packet must first be analysed
in the receiver.
PID = 0x00
Pointer to
PMT2
Pointer to
PMT1
PID4
PID3
PID2
Payload of
TS packet
PID1
TS
header
...
PAT =
Program
Association
Table
Pointer to
PMT4
Pointer to
PMT3
1 PID entry per program
PID2
Payload of
TS packet
PID1
TS
header
...
PMT =
Program
Map
Table
PID from PAT
Pointer to
video ES
Pointer to
audio ES
1 PID entry per elementary stream
Fig. 3.13. PAT and PMT
3.3.1 Synchronizing to the Transport Stream
When the MPEG-2 decoder input is connected to an MPEG-2 transport
stream, it must first lock to the transport stream, i.e. to the packet structure.
The decoder, therefore, looks for the sync bytes in the transport stream.
These always have the value of 0x47 and always appear at the beginning
of a transport stream packet. They are thus present at constant intervals of
188 bytes. These two factors together, the constant value of 0x47 and the
constant spacing of 188 bytes, are used for the synchronization. If a byte
44
3 The MPEG Data Stream
having a value of 0x47 appears, the decoder will examine the positions of
n times 188 bytes before and after this byte in the transport stream for the
presence of another sync byte. If there is, then this is a sync byte. If not,
then this is simply some code word which has accidentally assumed this
value. It is inevitable that the code word of 0x47 will also occur in the continuous transport stream. Synchronization will occur after 5 transport
stream packets and the decoder will lose lock after a loss of 3 packets (as
quoted in the MPEG-2 Standard).
3.3.2 Reading out the Current Program Structure
The number and the structure of the programs transmitted in the transport
stream is flexible and open. The transport stream can contain one program
with one video and audio elementary stream, or there can be 20 programs
or more, some with only audio, some with video and audio and some with
video and a number of audio signals which are being broadcast. It is, therefore, necessary to include certain lists in the transport stream which describe the instantaneous structure of the transport stream.
These lists provide the so-called “program specific information”, or PSI
in short (Fig. 3.13.). They are tables which are occasionally transmitted in
the payload part. The first table is the “Program Association Table” (PAT).
This table occurs precisely once per transport stream but is repeated every
0.5 sec.. This table shows how many programs there are in this transport
stream. Transport stream packets containing this table have the value zero
as packet identifier (PID) and can thus be easily identified. In the payload
part of the program association table, a list of special PIDs is transmitted.
There is exactly one PID per program in the program association table
(Fig. 3.13.).
These PIDs are pointers, as it were, to other information describing each
individual program in more detail. They point to other tables, the so-called
“Program Map Tables” (PMT). The program map tables, in turn, are special transport stream packets with a special payload part and special PID.
The PIDs of the PMTs are transmitted in the PAT. If it is intended to receive, e.g. program No.3, PID no. 3 is selected in the list of all PIDs in the
payload part in the program association table (PAT). If this is, e.g. 0x1FF3,
the decoder looks for transport stream packets having PID = 0x1FF3 in
their header. These packets are then the program map table for program no.
3 in the transport stream. The program map table, in turn, contains PIDs
which are the PIDs for all elementary streams contained in this program
(video, audio, data).
3.3 Information for the Receiver
45
Since there can be a number of video and audio streams - as for instance
in a Formula-1 broadcast in various languages - the viewer must select the
elementary streams to be decoded. Ultimately he will select exactly 2 PIDs
- one for the video stream and one for the audio stream, resulting e.g. in
the two hexadecimal numbers PID1 = 0x100 and PID2 = 0x110. PID1 is
then e.g. the PID for the video stream to be decoded and PID2 is the PID
for the audio stream to be decoded. From now on, the MPEG-2 decoder
will only be interested in these transport stream packets, collect them, i.e.
demultiplex them and assemble them again to form the PES packets. It is
precisely these PES packets which are supplied to the video and audio decoder in order to generate another video-and-audio signal.
The composition of the transport stream can change during the transmission, e.g. local programs can only be transmitted within certain windows.
A set-top box decoder, e.g. for DVB-S signals must, therefore, continuously monitor in the background the instantaneous structure of the transport stream, read out the PAT and PMTs and adapt to new situations. The
header of a table contains a so-called version management for this purpose
which signals to the receiver whether something has changed in the structure. It is regrettable that this does still not hold true for all DVB receivers.
A receiver often recognizes a change in the program structure only after a
new program search has been started. In many regions in Germany, socalled “regional window programs” are inserted into the public service
broadcast programs at certain times of the day. These are implemented by
a so-called “dynamic PMT”, i.e. the contents of the PMT are altered and
signal changes in the PIDs of the elementary streams.
Video PID = 0x100
MPEG-2 TS
Audio PID = 0x200
Fig. 3.14. Accessing a program via video and audio PIDs
46
3 The MPEG Data Stream
3.3.3 Accessing a Program
After the PIDs of all elementary streams contained in the transport stream
have become known from the information contained in the PAT and the
PMTs and the user has committed himself to a program, a video and audio
stream, precisely two PIDs are now defined (Fig. 3.14.): the PID for the
video signal to be decoded and the PID for the audio signal to be decoded.
The MPEG-2 decoder, on instruction by the user of the set-top box, will
now only be interested in these packets. Assuming then that the video PID
is 0x100 and the audio PID is 0x110: in the following demultiplexing
process all TS packets with 0x100 will be assembled into video PES packets and supplied to the video decoder. The same applies to the 0x110 audio
packets which are collected together and reassembled to form PES packets
which are supplied to the audio decoder. If the elementary streams are not
scrambled, they can now also be decoded directly.
Key codes
ECM
Entitlement Control Messages
EMM Entitlement
Management Messages
Allocation rights
CAT
(PID=1)
PID
PID
Fig. 3.15. The Conditional Access Table
3.3.4 Accessing Scrambled Programs
However, the elementary streams are transmitted scrambled. All or some
of the elementary streams are transmitted protected by an electronic code
in the case of pay TV or for licencing reasons involving local restrictions
on reception. The elementary streams are scrambled (Fig. 3.17.) by various
methods (Viaccess, Betacrypt, Irdeto, Conax, Nagravision etc.) and cannot
3.3 Information for the Receiver
47
MPEG-2 TS
Demultiplexer
be received without additional hardware and authorization. This additional
hardware must be supplied with the appropriate descrambling and authorization data from the transport stream. For this purpose, a special table is
transmitted in the transport stream, the “conditional access table” (CAT)
(Fig. 3.15.).
The CAT supplies the PIDs for other data packets in the transport
stream in which this descrambling information is transmitted. This additional descrambling information is called ECM (entitlement control message) and EMM (entitlement management message). The ECMs are used
for transmitting the scrambling codes and the EMMs are used for user administration. The important factor is that only the elementary streams
themselves may be scrambled, and no transport stream headers (or tables,
either). Neither is it permitted to scramble the transport stream header or
the adaptation field.
Video decoder
Audio decoder
Video
Audio
Common Interface (CI)
MPEG-2 decoder
Descrambler
Smart
card
(user data)
Fig. 3.16. Descrambling in the DVB receiver
The descrambling itself is done outside the MPEG decoder in additional
hardware related to the descrambling method, which can be plugged into a
so-called “common interface” (CI) in the set-top box. The transport stream
is looped through this hardware before being processed further in the
MPEG decoder. The information from the ECMs and EMMs and the
user’s personal code from the smart card then enable the streams to be descrambled.
3 The MPEG Data Stream
Exor
48
S
?
S
S
...
S
S
Exor
Pseudo Random Binary Sequency (PRBS)
Descrambled data
Scrambled data
Fig. 3.17. Scrambling and descrambling by PRBS generator in the CA system and
the receiver
3.3.5 Program Synchronization (PCR, DTS, PTS)
Once the PIDs for video and audio have been determined and any scrambled programs have been descrambled and the streams have been demultiplexed, video and audio PES packets are generated again. These are then
supplied to the video and audio decoder. The actual decoding, however,
requires a few more synchronization steps. The first step consists of linking the receiver clock to the transmitter clock. As indicated initially, the
luminance signal is sampled at 13.5 MHz and the two chrominance signals
are sampled at 6.75 MHz. 27 MHz is a multiple of these sampling frequencies, which is why this frequency is used as reference, or basic, frequency for all processing steps in the MPEG encoding at the transmitter
end. A 27 MHz oscillator in the MPEG encoder feeds the “system time
clock” (STC). The STC is essentially a 42 bit counter which is clocked by
this same 27 MHz clock and starts again at zero after an overflow. The
LSB positions do not go up to FFF but only to 300. Approximately every
26.5 hours the counter restarts at zero. At the receiving end, another system time clock (STC) must be provided, i.e. another 27 MHz oscillator
connected to a 42 bit counter is needed. However, the frequency of this
27 MHz oscillator must be in complete synchronism with the transmitting
end, and the 42 bit counter must also count in complete synchronism.
3.3 Information for the Receiver
Video
Audio
MPEG-2
encoder
PCR
every
~40 ms
PCR
42 Bit
PCR
MPEG-2 TS
Counter
MPEG-2
decoder
Video
42 Bit
Audio
49
Counter
Load
Copy
27 MHz
STC
+
27 MHz
STC
Numerically
controlled
oscillator
(NCO)
Fig. 3.18. Program Clock Reference
To accomplish this, reference information is transmitted in the MPEG
data stream (Fig. 3.18.). In MPEG-2, these are the “Program Clock Reference” (PCR) values which are nothing else than an up-to-date copy of the
STC counter fed into the transport stream at a certain time. The data
stream thus carries an accurate internal “clock time”. All coding and decoding processes are controlled by this clock time. To do this, the receiver,
i.e. the MPEG decoder, must read out the “clock time”, namely the PCR
values, and compare them with its own internal system clock, that is to say
its own 42 bit counter.
If the received PCR values are locked to the system clock in the decoder, the 27 MHz clock at the receiving end matches the transmitting end.
If there is a deviation, a controlled variable for a PLL can be generated
from the magnitude of the deviation, i.e. the oscillator at the receiving end
can be corrected. In parallel, the 42 bit count is always reset to the received
PCR value, a basic requirement for system initialization and in the event of
a program change.
The PCR values must be present in sufficient numbers, that is to say
with a maximum spacing, and relatively accurately, that is to say free of
jitter. According to MPEG, the maximum spacing per program is 40 ms
between individual PCR values. The PCR jitter must be less than ± 500 ns.
PCR problems manifest themselves in the first instance in that instead of a
color picture, a black/white picture is displayed. PCR jitter problems can
occur during the remultiplexing of a transport stream, among other things.
The reason is that e.g., the order of the transport stream packets is changed
50
3 The MPEG Data Stream
without the PCR information continued in them also being changed. There
is frequently a PCR jitter of up to ± 30 µs even though only ± 500 ns is allowed. This can be handled by many set-top boxes but not by all. The
PCR information is transmitted in the adaptation field of a transport stream
packet belonging to the corresponding program. The precise information
about the type of TS packets in which this is done can be found in the corresponding program map table (PMT). The PMT contains the so-called
PCR_PID which, however, corresponds to the video PID of the respective
program in most cases. After program clock synchronization has been
achieved, the video and audio coding steps are then executed in lock with
the system time clock (STC).
PTS of video PES
Video PES
PES
header
Video lip sync to audio
Audio PES
PTS of audio PES
DTS of video PES
Video PES
PES
header
Fig. 3.19. PTS and DTS
However, another problem now presents itself. Video and audio must be
decoded and reproduced with lip synchronization. In order to be able to
achieve “lip sync”, i.e. synchronization between video and audio, additional timing information is keyed into the headers of the video and audio
PESs. This timing information is derived from the system time clock
(STC, 42 bits). Using the 33 most significant bits (MSB) of the STC, these
3.3 Information for the Receiver
51
values are entered into the video and audio PES headers at maximum intervals of 700 ms and are called “presentation time stamps” (PTS)
As will be seen later in the section on video coding, the order in which
the compressed picture information is transmitted will differ from the order in which it is recorded. The frame sequence is now scrambled in conformity with certain coding rules, a necessary measure in order to save
memory space in the decoder. To recover the original sequence, additional
time stamps must be keyed into the video stream. These are called “decoding time stamps” (DTS) and are also transmitted in the PES header.
An MPEG-2 decoder in a set-top box is then able to decode the video
and audio streams of a program, resulting again in video and audio signals,
either in analog form or in digital form.
CRC
SECTION #0
T ID
SECTION #1
CRC
SECTION #2
CRC
T ID
T ID
Table
CRC
T ID
max. 4 kbyte
SECTION #3
SECTION #n
CRC
T ID
…
Fig. 3.20. Sections and tables
3.3.6 Additional Information in the Transport Stream
(SI/PSI/PSIP)
According to MPEG, the information transmitted in the transport stream is
fairly hardware-oriented, only relating to the absolute minimum requirements, as it were. However, this does not make the operation of a set-top
box particularly user-friendly. For example, it makes sense, and is necessary, to transmit program names for identification purposes. It is also desirable to simplify the search for adjacent physical transmission channels.
It is also necessary to transmit electronic program guides (EPG) and time
and date information. In this respect, both the European DVB Project
52
3 The MPEG Data Stream
group and the US ATSC Project group have defined additional information
for the transmission of digital video and audio programs which is intended
to simplify the operation of set-top boxes and make it much more userfriendly.
3.3.7 Non-Private and Private Sections and Tables
To cope with any extensions, the MPEG Group has incorporated an “open
door” in the MPEG-2 Standard. In addition to the “program specific information” (PSI), the “program map table” (PMT) and the “conditional access table” (CAT), it created the possibility to incorporate so-called “private sections and private tables” (Fig. 3.20.) in the transport stream. The
group has defined mechanisms which specify what a section or table has to
look like, what its structure has to be and by what rules it is to be linked
into the transport stream.
According to MPEG-2 Systems (ISO/IEC 13818-1), the following was
specified for each type of table:
•
•
•
A table is transmitted in the payload part of one or more transport stream packets with a special PID which is reserved for
only this table (DVB) or some types of tables (ATSC).
Each table begins with a table ID which is a special byte which
identifies only this table alone. The table ID is the first payload
byte of a table.
Each table is subdivided into sections which are allowed to have
a maximum size of 4 bytes. Each section of a table is terminated with a 32-bit-long CRC checksum over the entire section.
The “Program Specific Information” (PSI) has exactly the same structure. The PAT has a PID of zero and begins with a table ID of zero. The
PMT has the PIDs defined in the PAT as PID and has a table ID of 2. The
CAT has a PID and a table ID of one in each case. The PSI can be composed of one or more transport stream packets for PAT, PMT and CAT
depending on content.
Apart from the PSI tables PAT, PMT and CAT mentioned above, another table, the so-called “network information table” (NIT) was provided
in principle but not standardized in detail. It was actually implemented as
part of the DVB (Digital Video Broadcasting) project.
All tables are implemented through the mechanism of sections. There
are non-private and private sections (Fig. 3.21.). Non-private sections are
defined in the original MPEG-2 Systems Standard. All others are corre-
3.3 Information for the Receiver
53
spondingly private. The non-private sections include the PSI tables and the
private ones include the SI sections of DVB and the MPEG-2 DSM-CC
(Digital Storage Media Command and Control) sections which are used for
data broadcasting. The header of a table contains administration of the version number of a table and information about the number of sections of
which a table is made up. A receiver must first of all scan through the
header of these sections before it can evaluate the rest of the sections and
tables. Naturally, all sections must be broken down from an original
maximum length of 4 kbytes to maximally 148 bytes payload length of an
MPEG-2 transport stream packet before they are transmitted.
MPEG-2 Section
Non-private
-Defined in ISO/IEC13818-1
-MPEG-2 Program
Specific Information
(PSI Tables)
Private
-Not defined in ISO/IEC13818-1
-Used for MPEG-2 section
structure
-DVB Service Information
(SI Tables)
-ISO/IEC13818-6 DSM-CC
A table is = 1 … N sections of same type
(max. 1024 byte / 4096 byte per section)
Fig. 3.21. Sections and tables according to MPEG-2
In the case of PSI/SI, the limit of the section length has been lowered to
1 kbyte in almost all tables, the only exception being the EIT (Event Information Table) which is used for transmitting the electronic program
guide (EPG). The sections of the EIT can assume the maximum length of 4
kbytes because they carry a large amount of information as in the case of a
week-long EPG.
If a section begins in a transport stream packet (Fig. 3.22.), the payload
unit start indicator of its header is set to “1”. The TS header is then followed immediately by the pointer which points (in number of bytes) to the
actual beginning of the section. In most cases (and always in the case of
54
3 The MPEG Data Stream
PSI/SI), this pointer is set to zero which means that the section begins immediately after the pointer.
Pointer to begin of a section payload,
mostly set to 0x00
188 Byte
4 byte
header
184 byte
payload
Section payload
Header
Sync
byte
Transport
error
indicator
Payload
unit start
indicator
Transport
priority
PID
8
1
1
1
13
Transport
scrambling
control
2
Adaptation Continuity
field
counter
control
2
4
bit
Payload unit start indicator: set to “1“
Fig. 3.22. Beginning of a section in an MPEG-2 transport stream packet
table_id
section_syntax_indicator
private_indicator
reserved
section_length
if (section_syntax_indicator == 0)
table_body1() /* short table */
else
table_body2() /* long table */
if (section_syntax_indicator == 1)
CRC
Fig. 3.23. Structure of a section
8 Bit
1
1
2
12
32 Bit
3.3 Information for the Receiver
55
If the pointer has a value which differs from zero, remainders of the
preceding section can still be found in this transport stream packet. This is
utilized for saving TS packets, an example being MPE (multi-protocol encapsulation) over DSM-CC sections in the case of IP over MPEG-2 (see
DVB-H).
The structure of sections always follow the same plan (Fig. 3.23., Fig.
3.24.). A section begins with the table_ID, a byte which signals the type of
table. The section_syntax_indicator bit indicates whether this is a short
type of section (bit = 0) or a long one (bit = 1). If it is a long section, this is
then followed by an extended header which contains, among other things,
the version management of the section and its length and the number of the
last section. The version number indicates if the content of the section has
changed (e.g. in case of a dynamic PMT or if the program structure has
changed). A long section is always concluded with a 32-bit-long CRC
checksum over the entire section.
table_body1()
{
for (i=0;i<N;i++)
data_byte
}
table_body2()
{
table_id_extension
reserved
version_number
current_next_indicator
section_number
last_section_number
for (i=0;i<N;i++)
data_byte
}
Fig. 3.24. Structure of the section payload
8 Bit
16 Bit
2
5
1
8
8
8 Bit
56
3 The MPEG Data Stream
The detailed structure of a PAT and PMT can now also be understood
more easily. A PAT (Fig. 3.25, Fig. 3.26) begins with the table_ID = 0x00.
Its type is that of a non-private long table, i.e. the version management follows in the header. Since the information about the program structure to be
transmitted is very short, a single section is virtually always sufficient
(last_section_no = 0) and it also fits inside a transport packet. In the program loop, the program number and the associated program map ID are
listed for each program. Program No. Zero is a special exception, it informs about the PID of the later NIT (program information table). The
PAT is then concluded with the CRC checksum. There is one PAT per
transport stream but it is repeated every 0.5 sec. In the header of the table,
an unambiguous number, the transport stream_ID, is allocated to the transport stream via which it can be addressed in a network (e.g. a satellite network with many transport streams). The PAT does not contain any text information.
PAT
PID=0x00;
Table_ID=0x00;
Section length
Program
Loop
(Length
calculated
from
section
length)
Transport_stream_ID;
Version Management;
for i=0
i<N
repetition time:
25ms …500ms
i++
program_number; 16 Bit
reserved; 3 Bit
if (program_number ==0)
network_PID=0x10; 13 Bit
else
program_map_PID; 13 Bit
PID of NIT
PID of PMT
CRC_32
Fig. 3.25. Detailed structure of the PAT
The program map table (PMT) begins with the table_ID = 0x02. The
PID is signalled via the PAT and is in the range of 0x20 ... 0x1FFE. The
PMT is also a so-called non-private tale with version management and
concluding CRC checksum. The header of the PMT carries the program_no, already familiar from the PAT. The program_no in PAT and
PMT must match, i.e. be equal.
3.3 Information for the Receiver
Table ID
Table
header/
version
management
0 = „not private“
Transport Stream ID
Program Loop
Program Loop
CRC_32
Fig. 3.26. Details of the Program Association Table (practical example)
57
58
3 The MPEG Data Stream
The header of the PMT is followed by the program_info_loop into
which various descriptors can be inserted as required which describe program components in more detail. It does not have to be utilized, however.
The actual program components like video, audio or teletext are identified
via the stream loop which contains the entries for the respective stream
type and the PID of the elementary stream.
PMT
PID=
0x20…0x1FFE;
Table_ID=0x02;
Section_length
Program_number;
Version Management;
Repetition time:
25ms …500ms
PCR_PID
Program_info_length
for i=0
i<N
i++
Descriptor();
for i=0
Stream
Loop
(Length
calculated
from
section
length)
i<N
Program info
i++
Stream_type;
Reserved;
Elementary_PID;
Reserved;
ES_info_length
for i=0
i<N
i++
Descriptor();
ES Info
(Length from
ES_info_length)
CRC_32
Fig. 3.27. Detailed structure of the Program Map Table
It is possible to include a number of descriptors for each program component in the ES_info_loop. There is one PMT for each program and it is
sent out every 0.5 sec. There is no text information in the PMT, either.
3.3 Information for the Receiver
59
Fig. 3.28. shows an actual example of the structure of a Program Map
Table, which is quite short in this case. It will be discussed in more detail
as representative of many other tables following. The example, recorded
with an MPEG-2 analyzer, shows that the PMT begins with the table ID
0x02, a byte which clearly identifies it as such.
Table_ID
Table
header/
version
management
„0“ = not
private
Program no.
PCR_PID
Video PID
Stream
loop
Audio PID
Fig. 3.28. Details of the Program Map Table (practical example)
The section syntax indicator bit is set to “1” and tells one that this is a
long table with version management. The subsequent bit is set to “0” and
identifies this table as a so/called non/private MPEG table. The section
length says how long this current section of this table happens to be,
namely 23 bytes long in this case. The field of the table_ID extension contains the program number; there must also be a corresponding entry in the
PAT. The version number and the current/next indicator signal a change in
the program map table. This information must be continuously checked by
a receiver which must respond to a change in the program structure (dynamic PMT) if necessary. The section number tells what section this happens to be and the Last Section No informs about the number of the last
section of a table. It is set to zero in this case, i.e. the table consists of only
one section.
The PCR_PID (program check reference – packet identifier) provides
the PID on which the PCR value is broadcast. This is the video PID in
most cases.
60
3 The MPEG Data Stream
There should now be a program_info loop but there is none in this example, a fact which is signalled by the length indicator “program_info_length = 0.
However, the stream loop provides information about the video and audio PID. The stream type (see Table 3.1.) shows the type of payload,
namely MPEG-2 video and MPEG-2 audio in this case.
Table 3.1. Stream types of the Program Map Table
Value
0x00
0x01
0x02
0x03
0x04
0x05
0x06
0x07
0x08
0x09
0x0A
0x0B
0x0C
0x0D
0x0E
0x0F-0x7F
0x80-0xFF
Description
ITU-T/ISO/IEC reserved
ISO/IEC 11172 MPEG-1 video
ITU-T H.262 / ISO/IEC13818-2 MPEG-2 video
ISO/IEC 11172 MPEG-1 audio
ISO/IEC 13818-3 MPEG-2 audio
ITU-T H222.0 / ISO/IEC 13818-1 private sections
ITU-T H.222.0 / ISO/IEC 13818-1 PES packets containing private data
ISO/IEC 13522 MHEG
ITU-T H.222.0 /ISO/IEC 13818-1 annex A DSM-CC
ITU-T H.222.1
ISO/IEC 13818-6 DSM-CC type A
ISO/IEC 13818-6 DSM-CC type B
ISO/IEC 13818-6 DSM-CC type C
ISO/IEC 13818-6 DSM-CC type D
ISO/IEC 13818-1 auxiliary
ITU-T H.222.0 / ISO/IEC 13818-1 reserved
User private
3.3.8 The Service Information according to DVB (SI)
Taking advantage of the “private section” and “private table” features, the
European DVB Group has introduced numerous additional tables intended
to simplify the operation of the set-top boxes or quite generally of the
DVB receivers. Called “service information” (SI), they are defined in
ETSI Standard ETS300468.
They are the following tables (Fig. 3.29.): the “network information table” (NIT), the “service descriptor table” (SDT), the “bouquet association
table” (BAT), the “event information table” (EIT), the “running status table” (RST), the “time&date table” (TDT), the “time offset table” (TOT)
3.3 Information for the Receiver
61
and, finally, the “stuffing table” (ST). These eight tables will now be
described in more detail.
PAT Program Association Table
PMT‘s Program Map Table
CAT Conditional Access Table
(NIT) Network Information Table
Private Sections / Tables
MPEG-2 PSI
Program Specific
Information
NIT
SDT
BAT
EIT
RST
TDT
TOT
ST
DVB SI
Service
Information
Network Information Table
Service Descriptor Table
Bouquet Association Table
Event Information Table
Running Status Table
Time&Date Table
Time Offset Table
Stuffing Table
Fig. 3.29. MPEG-2 PSI and DVB SI
NIT
Network Information Table
(PID=0x10, Table_ID=0x40/0x41)
Information about
physical network
(satellite, cable, terrestrial)
Netzwork provider name,
transmission parameter
(RF, QAM, FEC)
Fig. 3.30. Network Information Table (NIT)
The “network information table” (NIT) (Fig. 3.30., Fig. 3.31., Fig.
3.32.) describes all physical parameters of a DVB transmission channel. It
contains, e.g. the received frequency and the type of transmission (satellite,
cable, terrestrial) and also all the technical data of the transmission, i.e. error protection, type of modulation etc.. This table has the purpose of optimizing the channel scan as much as possible. A set-top box is able to store
all the parameters of a physical channel when scanning during setup, and it
62
3 The MPEG Data Stream
is possible, e.g. to broadcast information about all available physical channels within a network (e.g. satellite, cable), making it possible to do away
with the actual physical search for channels.
The NIT contains the following information:
•
•
•
•
•
Transmission path (satellite, cable, terrestrial)
Received frequency
Type of modulation
Error protection
Transmission parameters
NIT
PID=0x10;
Table_ID=0x40/0x41;
Section_length
Network_ID;
Version management;
Repetition time:
25ms …10s
Network_descriptors_length
Network
descriptors
loop
for i=0
i<N
i++
Descriptor();
Transport_stream_loop_length
for i=0
Transport
stream
loop
i<N
i++
Transport_stream_ID;
Original_network_ID;
Reserved;
Transport_descriptors_length
for i=0
i<N
Descriptor();
i++
Transport
descriptors
loop
CRC_32
Fig. 3.31. Structure of the Network Information Table (NIT)
3.3 Information for the Receiver
63
The important factor in relation to the NIT is that many receivers, i.e.
set-top boxes, may behave in a “peculiar” manner if the transmission parameters in the NIT do not match the actual transmission. If, e.g. the
transmit frequency given in the NIT does not correspond to the actual received frequency, many receivers, without any indication of reasons, may
simply refuse to reproduce any picture or sound.
Table
header/
version
management
Table_ID
Network_ID
Network
descriptor
loop
Transport_stream_ID
Terrestrial
delivery
descriptor
Transport
stream
loop
Transport_stream_ID
Terrestrial
delivery
descriptor
Transport_stream_ID
Terrestrial
delivery
descriptor
Fig. 3.32. Practical example of a Network Information Table (NIT)
64
3 The MPEG Data Stream
SDT
Service Descriptor Table
(PID=0x11, Table_ID=0x42/0x46)
Information about all
services (= programs)
in a transport stream
Service provider name
service names = program
names
Fig. 3.33. Service Descriptor Table (SDT)
SDT
PID=0x11;
Table_ID=0x42/0x46;
Section_length
Transport_stream_ID;
Version Management;
Repetition time:
25ms …2s
Original_network_ID
for i=0
Service
loop
i<N
i++
Service_ID;
Reserved;
EIT_schedule_flag;
EIT_present_following_flag;
Running_status;
Free_CA_mode;
Descriptors_loop_length
for i=0
i<N
i++
Descriptor();
CRC_32
Fig. 3.34. Structure of the Service Descriptor Table (SDT)
Descriptors
loop
3.3 Information for the Receiver
65
The “service descriptor table” (SDT) contains more detailed descriptions of the programs carried in the transport stream, the “services”.
Among other things, these are the program titles such as, e.g. “CNN”,
“CBS”, “Eurosport”, “ARD”, “ZDF”, “BBC”, “ITN” etc.. That is to say, in
parallel with the program PIDs entered in the PAT, the SDT now contains
textual information for the user. This is intended to facilitate the operation
of the receiving device by providing lists of text.
Table_ID
Transport_ID
Service_ID
Service
loop
Service_name
Service_ID
Fig. 3.35. Practical Example of an SDT
BAT
Bouquet Association Table
(PID=0x11, Table_ID=0x4A)
Combining some
services to one
bouquet
Fig. 3.36. Bouquet Association Table
66
3 The MPEG Data Stream
BAT
PID=0x11;
Table_ID=0x4A;
Section_length
Bouquet_ID;
Version Management;
Repetition time:
25ms …10s
Bouquet_descriptors_length
Bouquet
descriptors
loop
for i=0
i<N
i++
Descriptor();
Transport_stream_loop_length
for i=0
Transport
stream
loop
i<N
i++
Transport_stream_ID;
Original_network_ID;
Reserved;
Transport_descriptors_length
for i=0
i<N
i++
Descriptor();
Transport
descriptors
loop
CRC_32
Fig. 3.37. Structure of a Bouquet Association Table (BAT)
A close relative of the service descriptor table is the “bouquet association table” (BAT). SDT and BAT have the same PID and differ only in the
table ID. Whereas the SDT describes the program structure of one physical
channel, a BAT describes the program structure of several physical channels or of a large number of physical channels.
The BAT is thus nothing else than a multi-channel program table. It
provides an overview of all services contained in a group. Program providers can make use of e.g. an entire bouquet of physical channels if a single
channel is insufficient for transmitting the complete range of programs
3.3 Information for the Receiver
67
provided. An example of this is the pay TV provider “Premiere”. A handful of satellite or cable DVB channels are combined here to form a bouquet
of this provider’s channels. The associated BAT is transmitted in all individual channels and links this bouquet together.
In fact, however, a bouquet association table is found very rarely in a
transport stream. The broadcasters ARD and ZDF in Germany, and Premiere, are broadcasting a BAT for their respective bouquet and sometimes
a BAT can be found in networks of cable network providers.
But frequently, the BAT doesn’t exist at all, as already mentioned.
When it does exist, it tells by way of so-called linkage descriptors which
service of a particular service ID can be found in which transport streams.
Many providers are also transmitting an “electronic program guide”
(EPG) which has its own table in DVB, the so-called “event information
table”, or EIT for short (Fig. 3.38. and 3.39.). It contains the planned starting and stopping times of all broadcasts of, e.g. one day or one week. The
structure which is possible here is very flexible and also allows any
amount of additional information to be transmitted. Unfortunately it is true
that this feature is not supported by all set-top boxes, or only inadequately
so.
EIT
Event Information Table
(PID=0x12, Table_ID=0x4E..0x6F)
Electronical Program
Guide
(EPG)
Fig. 3.38. EIT
Frequently, however, there are variations and delays in the planned
starting and stopping times of broadcasts. To be able to start and stop, e.g.
a video recorder at a given time, the relevant control information is transmitted in the “Running Status Table” (RST). The RST can thus be compared to the VPS (video program system) signal in the data line of an analog TV signal. The RST is currently not being used in practice, or, at least,
68
3 The MPEG Data Stream
has not been found by the author in a transport stream anywhere in the
world, excepting “synthetic” transport streams. Instead, the data line containing the VPS has been adapted within DVB for controlling video recorders and similar recording media.
EIT
PID=0x12;
Table_ID=0x4E…0x6F;
Section_length
Service_ID;
Version Management;
Repetition time:
25ms …2s
Transport_ID;
Original_network_ID;
Segment_last_section_no.
for i=0
Event
loop
i<N
i++
Event_ID;
Start_time;
Duration;
Running_status;
Free_CA_mode;
Descriptors_loop_length
for i=0
i<N
i++
Descriptor();
Event
descriptors
loop
CRC_32
Fig. 3.39. Structure of the Event Information Table (EIT)
The operation of the set-top box also requires the transmission of the
current clock time and the current date. This is done in two stages. In the
“Time&Date Table” (TDT) (Fig. 3.42. and 3.43.), Greenwich Mean Time
(GMT or UTC), i.e. the current clock time on the Zero-Degree meridian
without any daylight saving time shift is transmitted. The respective applicable time offsets can then be broadcast in a “time offset table” (TOT)
(Fig. 3.42. and 3.43.) for the various time zones. It depends on the software
of the set-top box how the information contained in the TDT and TOT is
3.3 Information for the Receiver
69
evaluated, and to what extent. Complete support for this broadcast time information would require the set-top box to be informed of its current location and in a country having a number of time zones such as Australia, especially, more attention should be paid to this point.
It may sometimes be necessary to cancel certain information, especially
tables in the transport stream. After a DVB-S signal has been received in a
CATV head station, it can quite easily happen that, e.g. the NIT must be
exchanged or overwritten or that individual programs must be rendered
unusable for relaying. This can be done by means of the “stuffing table”
(ST) (Fig. 3.44.) which enables information in the transport stream to be
overwritten.
Service_ID
Transport_stream_ID
Event_ID
Start Time &
Duration
Fig. 3.40. Event Information Table (practical example)
RST
Running Status Table
(PID=0x13,Table ID=0x71)
Current status of a
event
Fig. 3.41. Running Status Table (RST)
70
3 The MPEG Data Stream
RST
PID=0x13;
Table_ID=0x71;
Section_length
Event
loop
Repetition time:
25ms …infinite
Table Header
for i=0
i<N
i++
Transport_stream_ID;
Original_network_ID;
Service_ID;
Event_ID;
Reserved;
Running_status;
Fig. 3.42. Structure of the Running Status Table (RST)
TDT/TOT
Time and Date Table,
Time Offset Table
(PID=0x14, Table ID =0x70, 0x74)
Current time and date
(UTC/GMT)
and
local time offset
Fig. 3.43. Time and Date Table (TDT) and Time Offset Table (TOT)
The PIDs and the table IDs for the service information have been permanently allocated within DVB in Table 3.2.
The PSI/SI tables are linked to one another via the most varied identifiers (Fig. 3.45.). These are both PIDs and special, table-dependent identifiers. In the PAT, the PMT_PIDs are chained together by way of the
prog_no. To each prog_no, a PMT_PID is allocated which refers to a
transport stream packet with the corresponding PMT of this associated
program. The prog_no can then also be found in the header of the respective PMT. Prog_no = 0 is allocated to the NIT where the PID of the NIT
can be found.
3.3 Information for the Receiver
71
Fig. 3.44. Example of a Time and Date Table (TDT and Time Offset Table (TOT)
ST
Stuffing Table
(Table ID=0x72)
Cancellation
of sections
and tables in a
distribution network
e.g. at cable headends
Fig. 3.45. Stuffing Table (ST)
72
3 The MPEG Data Stream
Table 3.2. PIDs and table IDs of the PSI/SI tables
Table
PID
Table_ID
PAT
0x0000
0x00
PMT
0x0020...0x1FFE
0x02
CAT
0x0001
0x01
NIT
0x0010
0x40...0x41
BAT
0x0011
0x4A
SDT
0x0011
0x42, 0x46
EIT
0x0012
0x4E...0x6F
RST
0x0013
0x71
TDT
0x0014
0x70
TOT
0x0014
0x73
ST
0x0010...0x0014
0x72
PMT
Prog_no
other TS
NIT
TS_ID
TS_ID
Descriptor
TS_ID
Descriptor
PAT
SDT/BAT
TS_ID
Service_ID
Descriptor()
Prog_no
PMT_PID
Prog_no
PMT_PID
PMT
Prog_no
Prog_no
PMT_PID
Service_ID
Descriptor()
Service_ID
Descriptor()
SDT/BAT
Service_ID
Descriptor()
EIT
Service_ID
Descriptor()
Service_ID
Event_ID
Event_ID
Event_ID
Service_ID
Descriptor()
Fig. 3.46. Links between the PSI/SI tables
RST
Event_ID
3.4 The PSIP according to the ATSC
73
In the NIT, the physical parameters of all transport streams of a network
are described via their TS_IDs. A TS_ID corresponds to the current transport stream; precisely this TS_ID can be found in the header of the PAT at
the position of the Table ID extension.
The services (= programs) contained in this transport stream are listed in
the service descriptor table via the service IDs. The service IDs must correspond to the prog_no in the PAT and in the PMTs.
This is continued in the EIT: there is an EIT for every service. In the
header of the EIT, the table_ID_extension corresponds to the service_ID of
the associated program. In the EIT, the events are associated with these by
way of event_IDs. If there are associated RSTs, then these are chained to
the respective RST via these event_IDs.
Table 3.3. Repetition rates of the PSI/SI tables according to MPEG/DVB
PSI/SI table
PAT
CAT
PMT
NIT
SDT
BAT
EIT
RST
TDT
TOT
Max. interval
(complete table)
0.5 s
0.5 s
0.5 s
10 s
2s
10 s
2s
30 s
30 s
Min. interval
(single sections)
25 ms
25 ms
25 ms
25 ms
25 ms
25 ms
25 ms
25 ms
25 ms
25 ms
The repetition rates of the PSI/SI tables are regulated through MPEG-2
Systems [ISO&IEC 13818/1] and DVB/SI [ETS 300468] (Table 3.3)
3.4 The PSIP according to the ATSC
In the US, a separate standard was specified for digital terrestrial and cable
TV. This is the ATSC standard, where ATSC stands for Advanced Television System Committee. During the work on the ATSC standard, the decision was made to use the MPEG-2 transport stream with MPEG-2 video
and AC-3 Dolby Digital audio as the baseband signal. The type of modulation used is 8 or 16VSB. In addition, it was recognized that other tables
going beyond PSI are needed. Like the SI tables in DVB, ATSC, therefore,
74
3 The MPEG Data Stream
has the PSIP tables, listed below and described in more detail in the text
which follows.
PAT Program Association Table
PMTs Program Map Table
CAT Conditional Access Table
Private Tables
MPEG-2 PSI
Program Specific
Information
MGT Master Guide Table
EIT Event Information Table
ETT Extended Text Table
STT System Time Table
RTT Rating Region Table
CVCT Cable Virtual Channel Table
TVCT Terrestrial Virtual Channel Table
ATSC PSIP
Program and
System
Information
Protocol
Fig. 3.47. ATSC PSIP tables
EIT1
MGT
3 hours
EPG
per EIT
PID=0x1FFB
ETT
PID
PID
EIT2
PID
4 EIT‘s
Fig. 3.48. Referencing the PSIP in the MGT
PSIP stands for “program and system information protocol” and is nothing else than another way of representing similar information to that given
in the previous section on DVB SI. In ATSC, the following tables are
used: the Master Guide Table (MGT) (Fig. 3.47.), the Event Information
Table (EIT), the Extended Text Table (ETT), the System Time Table
3.4 The PSIP according to the ATSC
75
(STT), the Rating Region Table (RRT), and the Cable Virtual Channel Table (CVCT) or the Terrestrial Virtual Channel Table (TVCT).
According to ATSC, the PSI tables defined in MPEG-2 and provided in
the MPEG Standard are used for accessing the video and audio streams,
i.e. the transport stream carries one PAT and several PMTs. The conditional access information is also referenced via a CAT.
The actual ATSC tables are implemented as “private tables”. The Master Guide Table, the main table, so to say, contains the PIDs for some of
these ATSC tables. The Master Guide Table can be recognized by the
packet ID = 0x1FFB and the table ID = 0xC7. The transport stream must
contain at least four Event Information Tables (EIT-0, EIT-1, EIT-2, EIT3) and the PIDs for these EITs are found in the Master Guide Table. Up to
128 further Event Information Tables are possible but are optional. An EIT
contains a 3-hour section of an electronic program guide (EPG). Together
with the 4 mandatory EITs, it is thus possible to cover a period of 12
hours. Furthermore, Extended Text Tables can be optionally accessed
through the MGT. Each existing Extended Text Table (ETT) is allocated
to one EIT. Thus, e.g. ETT-0 contains extended text information for EIT0. It is possible to have up to a total of 128 ETTs.
Table 3.4. PSIP tables
Table
Program Association Table (PAT)
Program Map Table (PMT)
Conditional Access Table (CAT)
Master Guide Table (MGT)
Terrestrial Virtual Channel Table (TVCT)
Cable Virtual Channel Table (CVCT)
Rating Region Table (RRT)
Event Information Table (EIT)
Extended Text Table (ETT)
System Time Table (STT)
PID
0x0
über PAT
0x1
0x1FFB
0x1FFB
0x1FFB
0x1FFB
über PAT
über PAT
0x1FFB
Table ID
0x0
0x2
0x1
0xC7
0xC8
0xC9
0xCA
0xCB
0xCC
0xCD
In the Virtual Channel Table, which can be present either as Terrestrial
Virtual Channel Table (TVCT) or as Cable Virtual Channel Table (CVCT)
depending on the transmission path, identification information for the virtual channels, i.e. programs, contained in a multiplexed transport stream
are transmitted. The VCT contains, among other things, the program
names. The VCT is thus comparable to the SDT table in DVB:
In the System Time Table (STT), all the necessary time information is
transmitted. The STT can be recognized by the packet ID = 0x1FFB and
the table ID = 0xCD. In the STT, the GPS (Global Positioning System)
76
3 The MPEG Data Stream
time and the time difference between GPS time and UTC (Universal Time
Coordinated (= GMT)) is transmitted. The Rating Region Table (RRT) can
be used for restricting the size of the audience in terms of age or region. In
addition to the information about region (e.g. a Federal State in the US),
information relating to the minimum age set for the program currently being broadcast is also included. Using the RRT, a type of parental lock can
thus be implemented in the set-top box. The RRT is recognized by the
packet ID = 0x1FFB and the table ID = 0xCA.
The PIDs and Table IDs of the PSIP tables are listed in Table 3.4.
3.5 ARIB Tables according to ISDB-T
Like DVB (Digital Video Broadcasting) and ATSC (Advanced Television
Systems Committee), Japan, too, has defined its own tables in its ISDB-T
(Integrated Services Digital Broadcasting – Terrestrial) standard. These are
called ARIB (Association of Radio Industries and Business) tables according to ARIB Std. B.10.
According to the ARIB standard, the following tables are proposed:
Table 3.5. ARIB tables
Type
PAT
PMT
CAT
NIT
SDT
BAT
EIT
RST
TDT
TOT
LIT
ERT
ITT
PCAT
ST
BIT
NBIT
Name
Program Association Table
Program Map Table
Conditional Access Table
Network Information Table
Service Description Table
Bouquet Association Table
Event Information Table
Running Status Table
Time&Date Table
Time Offset Table
Local Event
Information Table
Event Relation Table
Index Transmission Table
Partial Content
Announcement Table
Stuffing Table
Broadcaster
Information Table
Network Board
Information Table
Note
ISO/IEC 13818-1 MPEG-2
ISO/IEC 13818-1 MPEG-2
ISO/IEC 13818-1 MPEG-2
like DVB-SI, ETS 300468
like DVB-SI, ETS 300468
like DVB-SI, ETS 300468
like DVB-SI, ETS 300468
like DVB-SI, ETS 300468
like DVB-SI, ETS 300468
like DVB-SI, ETS 300468
like DVB-SI, ETS 300468
3.5 ARIB Tables according to ISDB-T
LDT
and others
ECM
EMM
DCT
DLT
SIT
SDTT
DSMCC
77
Linked Description Table
Entitlement Control Message
Entitlement
Management Message
Download Control Table
Download Table
Selection Information Table
Software Download
Trigger Table
Digital Storage Media
Command & Control
Table 3.6. PID’s and table IDs of the ARIB tables
Table
PAT
CAT
PMT
DSM-CC
NIT
SDT
BAT
EIT
TDT
RST
ST
TOT
DIT
SIT
ECM
EMM
DCT
DLT
PCAT
SDTT
BIT
NBIT
LDT
LIT
PID
0x0000
0x0001
über PAT
über PMT
0x0010
0x0011
0x0011
0x0012
0x0014
0x0013
all except 0x0000,
0x0001, 0x0014
0x0014
0x001E
0x001F
via PMT
via CAT
0x0017
via DCT
0x0022
0x0023
0x0024
0x0025
0x0025
via PMT or 0x0020
Table ID
0x00
0x01
0x02
0x3A...0x3E
0x40, 0x41
0x42, 0x46
0x4A
0x4E...0x6F
0x70
0x71
0x72
0x73
0x7E
0x7F
0x82...0x83
0x84...0x85
0xC0
0xC1
0xC2
0xC3
0xC4
0xC5, 0xC6
0xC7
0xD0
The BAT, PMT and CAT tables fully correspond to the MPEG-2 PSI.
Similarly, the NIT, SDT, BAT, EIT, RST, TDT. TOT and ST tables have
78
3 The MPEG Data Stream
exactly the same structure as in DVB SI and also have the same functionality. The ARIB Standard thus also makes reference to ETSI 300468.
3.6 DTMB (China) Tables
China, too, have their own digital terrestrial television standard named
DTMB – Digital Terrestrial Multimedia Broadcasting. It can be assumed
that there is also an independent or modified or copied table of comparable
significance to DVB-SI but there have been no publications regarding
what modifications, if any, were made.
3.7 Other Important Details of the MPEG-2 Transport
Stream
In the section below, other details of the MPEG-2 transport stream will be
discussed in more detail.
Apart from the sync bytes (synchronization to the transport stream) already mentioned, the transport stream error indicator and the packet identifier (PID), the transport stream header also contains:
•
•
•
•
•
the Payload Unit Start Indicator,
the Transport Priority,
the Transport Scrambling Control,
the Adaptation Field Control, and
the Continuity Counter.
The Payload Unit Start Indicator is a bit which marks the start of a payload. If this bit is set, it means that a new payload is starting in this transport stream packet: this transport stream packet contains either the start of
a video or audio PES packet plus PES header, or the beginning of a table
plus table ID as the first byte in the payload part of the transport stream
packet.
3.7.1 The Transport Priority
This bit indicates that this transport stream packet has a higher priority
than other TS packets with the same PID.
3.7 Other Important Details of the MPEG-2 Transport Stream
79
3.7.2 The Transport Scrambling Control Bits
The two Transport Scrambling Control Bits show whether the payload part
of a TS packet is scrambled or not. If both bits are set to zero, this means
that the payload section is transmitted unscrambled. If one of the two bits
is not zero, the payload is transmitted scrambled. A Conditional Access
Table is then needed to descramble the payload.
188 byte
4 byte
header
184 byte
payload
Optional
adaptation
field
header
Sync Transport Payload Transport
unit start priority
byte error
indicator indicator
8
1
1
1
Transport Adaptation Continuity
PID scrambling field
counter
control
control
13
2
2
4
bit
188 byte
4 byte
header
Header
Adaption
Discontinuity ...
field
indicator
length
8
1
184 byte
payload
Optional
adaptation
field
5 flags
5
Optional
fields
...
depending
on flags
PCR
42
...
bit
Fig. 3.49. Other details in the MPEG-2 transport stream
3.7.3 The Adaptation Field Control Bits
These two bits indicate whether there is an extended header, i.e. an adaptation field, or not. If both bits are set to zero, there is no adaptation field. If
there is an adaptation field, the payload part is shortened and the header
becomes longer but the total packet length remains a constant 188 bytes.
80
3 The MPEG Data Stream
3.7.4 The Continuity Counter
Each transport stream packet with the same PID carries its own 4-bit
counter. This is the continuity counter which continuously counts from 0 15 from TS packet to TS packet and then begins again from 0. The continuity counter makes it possible to recognize missing TS packets and to
identify an errored data stream (counter discontinuity). It is possible, and
permissible, to have a discontinuity with a program change which is then
indicated by the Discontinuity Indicator in the adaptation field.
Bibliography: [ISO 13818/1], [ETS 300468], [A53], [REIMERS],
[SIGMUNT], [DVG], [DVDM], [GRUNWALD], [FISCHER],
[FISCHER4], [DVM], [ARIB]
4 Digital Video Signal According to ITU-BT.R.601
(CCIR 601)
Uncompressed digital video signals have been used for some time in television studios. Based on the original CCIR Standard CCIR 601, designated
as IBU-BT.R601 today, this data signal is obtained as follows:
To start with, the video camera supplies the analog Red, Green and Blue
(R, G, B) signals. These signals are matrixed in the camera to form luminance (Y) and chrominance (color difference CB and CR) signals.
Luminance
sampling frequency
13.5 MHz
5.75 MHz
8/10 Bit
R
G
B
Matrix
Y
A
Cb
A
Cr
A
D
Y
8/10 Bit
Cb
D
8/10 Bit
D
Cr
270 Mbit/s
ITU-BT.R 601
„CCIR601“
2.75 MHz
6.75 MHz
chrominance
sampling frequency
Fig. 4.1. Digitization of luminance and chrominance
These signals are produced by simple addition or subtraction of R =
Red, G = Green, B = Blue:
Y = (0.30 • R) + (0.59 • G) + (0.11 • B);
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_4, © Springer-Verlag Berlin Heidelberg 2010
82
4 Digital Video Signal According to ITU-BT.R.601 (CCIR 601)
CB = 0.56 • (B-Y);
CR = 0.71 • (R-Y) ;
The luminance bandwidth is then limited to 5.75 MHz using a low-pass
filter. The two color difference signals are limited to 2.75 MHz, i.e. the
color resolution is clearly reduced compared with the brightness resolution. This principle is familiar from children’s books where the impression
of sharpness is simply conveyed by printed black lines. In analog television (NTSC, PAL, SECAM), too, the color resolution is reduced to about
1.3 MHz. The low-pass filtered Y, CB and CR signals are then sampled and
digitized by means of analog/digital converters. The A/D converter in the
luminance branch operates at a sampling frequency of 13.5 MHz and the
two CB and CR color difference signals are sampled at 6.75 MHz each.
13.5 MHz luminance sampling frequency
Y
CB
SAV = Start of
active video
EAV = End of
active video
6.75 MHz chrominance sampling frequency
Blanking
SAV
......
EAV
SAV
Cb
Y
Cr
Y
Cb
Y
EAV
CR
Active video
Fig. 4.2. Sampling of the components in accordance with ITU-BT.R601
This meets the requirements of the sampling theorem: There are no
more signal components above half the sampling frequency. The three A/D
converters can all have a resolution of 8 or 10 bits. With a resolution of 10
bits, this will result in a gross data rate of 270 Mbit/s which is suitable for
distribution in the studio but much too high for TV transmission via existing channels (terrestrial, satellite or cable). The samples of all three A/D
converters are multiplexed in the following order: CB Y CR Y CB Y ... In
this digital video signal (Fig. 4.1.), the luminance value thus alternates
4 Digital Video Signal According to ITU-BT.R.601 (CCIR 601)
83
00000000(00) = 0/0
00000000(00) = 0/0
1 F V H P3 P2 P1 P0 00
.....
11111111(11) = 255/1023
with a CB value or a CR value and there are twice as many Y values as
there are CB or CR values. This is called a 4:2:2 resolution, compared with
the resolution immediately after the matrixing, which was the same for all
components, namely 4:4:4.
1
2
3
4
.....
TRS = Timing
Reference Sequence
4 code words
(SAV or EAV
= Start of active
video or end
of active video)
Fig. 4.3. SAV and EAV code words in the ITU-BT.R601 signal
This digital signal can be present in parallel form at a 25 pin sub-D connector or serially at a 75 Ohm BNC socket. The serial interface is called
SDI which stands for serial digital interface and has become the most
widely used interface because a conventional 75-Ohm BNC cable can be
used.
Within the data stream, the start and the end of the active video signal is
marked by special code words called SAV (start of active video) and EAV
(end of active video), naturally enough (Fig. 4.2.). Between EAV and
SAV, there is the horizontal blanking interval which does not contain any
information related to the video signal, i.e. the digital signal does not contain the sync pulse. In the horizontal blanking interval, supplementary information can be transmitted such as, e.g. audio signals (embedded audio).
The SAV and EAV code words (Fig. 4.3.) consist of four 8 or 10 bit
code words each. SAV and EAV begins with one code word in which all
bits are set to one, followed by two words in which all bits are set to zero.
The fourth code word contains information about the respective field or the
vertical blanking interval, respectively. This fourth code word is used for
detecting the start of a frame, field and active picture area in the vertical
84
4 Digital Video Signal According to ITU-BT.R.601 (CCIR 601)
direction. The most significant bit of the fourth code word is always 1. The
next bit (bit 8 in a 10 bit transmission or bit 6 in an 8 bit transmission)
flags the field; if this bit is set to zero, it is a line of the first field and if it is
set to one, it is a line of the second field. The next bit (bit 7 in a 10 bit
transmission or bit 5 in an 8 bit transmission) flags the active video area in
the vertical direction. If this bit is set to zero, then this is the visible active
video area and if not, it is the vertical blanking interval. Bit 6 (10 bit) or bit
4 (8 bit) provides information about whether the present code word is an
SAV or an EAV. It is SAV if this bit is set to zero and EAV if it is not.
Bits 5...2 (10 bit) or 3...0 (8 bit) are used for error protection of the SAV
and EAV code words. Code word 4 of the timing reference sequence
(TRS) contains the following information:
•
•
•
•
F = Field (0 = 1st field, 1 = 2nd field)
V = Vertical blanking (1 = vertical blanking interval active)
H = SAV/EAV identification (0 = SAV, 1 = EAV)
P0, P1, P2, P3 = Protection bits (Hamming code)
Neither the luminance signal (Y) nor the color difference signals (CB,
CR) use the full dynamic range available for them. There is a prohibited
range which is reserved as headroom, on the one hand, and, on the other
hand, allows SAV and EAV to be easily identified. A Y signal ranges between 16 and 64 decimal (8 bits) or 240 and 960 decimal (10 bits).
255/1023
700 mV
235/940
Y
255/1023
350 mV
CB/CR
0 mV
16/64
0
Fig. 4.4. Level diagram
240/960
0 mV
128/512
-350 mV
16/64
0
4 Digital Video Signal According to ITU-BT.R.601 (CCIR 601)
85
The dynamic range of CB and CR is 16 to 240 decimal (8 bits) or 64 to
960 decimal (10 bits). The area outside this range is used as headroom and
for sync identification purposes.
This video signal conforming to ITU-BT.R601, which is normally
available as an SDI (Serial Digital Interface) signal, forms the input signal
to an MPEG encoder.
Bibliography: [ITU601], [MÄUSL4], [GRUNWALD]
5 High Definition Television – HDTV
The Standard Definition Television – SDTV – introduced in the 50s is still
virtually the main standard for analog and digital television in all countries
throughout the world. However, as in the field of computers, modern TV
cameras and terminal devices such as plasma screens and LCD receivers
provide for much higher pixel resolution.
In computer monitors, the resolutions are:
•
•
•
•
•
•
•
VGA 640 x 480 (4:3)
SVGA 800 x 600 (4:3)
XGA 1024 x 768 (4:3)
SXGA 1280 x 1024 (5:4)
UXGA 1600 x 1200 (4:3)
HDTV 1920 x 1080 (16:9)
QXGA 2048 x 1536 (4:3)
pixels, together with the respective aspect ratios (width x height).
Since the 1990s, there have been efforts in some countries to switch
from the standard resolution SDTV to high resolution HDTV (High Definition Television). The first attempts were made in Japan with MUSE
(Multiple Sub-Nyquist Sampling Encoding), developed by the broadcaster
NHK (Nippon Hoso Kyokai). In Europe, too, HDTV was on the agenda at
the beginning of the 1990s as HD-MAC (High-Definition Multiplexed
Analog Components) but never entered the market. In the US, it was decided in the mid 90s to introduce HDTV as part of the ATSC (Advanced
Television System Committee) effort, and in Australia it was decided to
transmit HDTV as part of digital terrestrial television when the DVB-T
standard was adopted. Europe, too, is now beginning to introduce HDTV.
HDTV is currently implemented by MPEG-2 coding both in Japan, in the
US and in Australia.
Europe, too, is now beginning to introduce HDTV. Since 2006, the
channels Premiere (now Sky), Pro7 and Sat1 had also been on the air in
HD. Pro7 and Sat1 are currently suspending their HD transmissions until
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_5, © Springer-Verlag Berlin Heidelberg 2010
88
5 High Definition Television – HDTV
2010 when the public broadcasters in Germany will begin to transmit in
HD.
The usual field rate in a 625 line TV system is 50 Hz and in a 525 line
system it is 60 Hz. This is related to the power line frequency used in the
original countries. The aspect ratio for HDTV will normally be 16:9 which
is also becoming the norm for SDTV.
Initially, HDTV was to be based on twice the number of lines and twice
the number of pixels per line. This would result
•
•
in 1250 lines total with 1152 active lines and 1440 active pixels
in a 625 line system
and 1050 lines total with 960 active lines and 1440 active pixels
in a 525 line system.
However, the resolution used for ATSC and HDTV in the US is 1280 x
720 pixels at 60 Hz. In Australia it is usually 1920/1440 x 1080 pixels at
50 Hz. The resolution of the European HDTV satellite channel EURO1080
is 1080 active lines x 1920 pixels at a field rate of 50 Hz.
625
or
525
lines
SDTV
4:3/16:9
576 1250
or
or
480 1125
aktive lines
lines
720 pixel
HDTV
16:9
1080
(720)
aktive
lines
1920
(1280)
pixel
Fig. 5.1. SDTV and HDTV resolution
Once HDTV is introduced throughout Europe, MPEG-2 coding will be
replaced by MPEG-4 Part 10, H.264, which will be more effective by a
factor of 2 to 3. In this chapter, however, only the uncompressed digital
baseband signal for HDTV as defined by the ITU-R BT.709 and ITU-R
BT.1120 standards will be described.
The ITU has generally decided on a total number of 1125 lines in the 50
Hz and 60 Hz system, with 1080 active lines (Fig. 5.4.) and 1920 pixels
per line both in the 50 Hz and the 60 Hz system. An active image of 1080
lines x 1920 pixels is called the Common Image Format (CIF). The sam-
5 High Definition Television – HDTV
89
pling rate of the luminance signal is 74.25 MHz (Fig. 5.2.). The Y:CB:CR
format is 4:2:2. The sampling rate of the color difference signals is 0.5 x
74.25 MHz = 37.125 MHz. ITU-R BT.709 provided a sampling rate of 72
MHz for the luminance and 36 MHz for the chrominance. To avoid aliasing, the luminance signal bandwidth is limited to 30 MHz and that of the
chrominance signals to 15 MHz by low-pass filtering them before they are
sampled.
In the 1125/60 system (Fig. 5.2.), and with a 10 bit resolution, this results in a gross physical data rate of:
Y:
74,25 x 10 Mbit/s = 742.5 Mbit/s
CB: 0.5 x 74,25 x 10 Mbit/s = 371.25 Mbit/s
CR: 0.5 x 74,25 x 10 Mbit/s = 371.25 Mbit/s
---------------1.485 Gbit/s
gross data rate (1125/60)
Because of the slightly lower sampling rates in the 1250/50 system (Fig.
5.2.), the gross data rate, with 10 bit resolution, is then:
Y:
72 x 10 Mbit/s = 720 Mbit/s
CB: 0.5 x 72 x 10 Mbit/s = 360 Mbit/s
CR: 0.5 x 7,2 x 10 Mbit/s = 360 Mbit/s
---------------1.44 Gbit/s
gross data rate (1250/50)
Both interlaced and progressive scanning are provided for. Plasma and
LCD screens support progressive scanning due to the technology used, and
interlaced scanning can lead to unattractive artefacts. With 50/60 progressively scanned frames the sampling rates are doubled to 148.5 and 144
MHz, respectively, for the luminance signal and 74.25 and 72 MHz, respectively, for the chrominance signals. The gross data rates are then doubled to 2.97 Gbit/s and 2.88 Gbit/s, respectively.
The structure of the uncompressed digital HDTV data signal is similar
to ITU-R BT.601. A parallel and a serial interface (HD-SDI) are defined.
Since, apart from a few exceptions, the large-scale introduction of
HDTV is still pending, however, the actual technical parameters are still
subject to modification.
The European receiver manufacturers have defined the logos "HD
Ready" and "Full HD" in order to describe characteristics of a display or
90
5 High Definition Television – HDTV
projector. "HD Ready" defines a display or projector which offers the following features:
•
•
•
•
•
•
•
at least minimum 720 lines physical resolution
aspect ratio of 16:9
supports a resolution of 1280 x 720 at 50 Hz or 60 Hz frame rate,
progressive
supports a resolution of 1980 x 1080 at 50 Hz or 60 Hz frame rate,
interlaced
analog Y Pb Pr interface
digital DVI or HDMI interface
HDCP encryption in the digital interfaces
Sync
+700mV
+300mV
74.25 MHz (1125/60)
72 MHz (1250/50)
(x 2 @ 50/60p 1:1)
30 MHz
(x 2 @ 50/60p)
Y
0mV
A
-300mV
8/10 Bit
D
Y
1.44 Gbit/s,
1.485 Gbit/s
+350mV
CB
-350mV
Blanking
8/10 Bit
D
15 MHz
(x 2 bei 50/60p)
CR
aktive
video
A
15 MHz
(x 2 @ 50/60p)
CB
8/10 Bit
A
D
gross data rate
2.88 Gbit/s,
2.97 Gbit/s
(@ 50/60p)
CR
37.125 MHz (1125/60)
36 MHz (1250/50)
(x 2 @ 50/60p 1:1)
Fig. 5.2. Sampling of an HDTV signal according to ITU-R.BT709
DVI stands for Digital Visual Interface and is already known from PCs
where it will replace the usual VGA interface. DVI allows a data rate of
1.65 Gbit/s. HDMI stands for High Definition Multimedia Interface and
supports a data rate of up to 5 Gbit/s, transporting both picture and sound.
HDCP means High Bandwidth Digital Content Protection and protects
5 High Definition Television – HDTV
91
digital HD material in the DVI and HDMI interface against illegal recording, a requirement set by the film industry.
In contrast to "HD Ready", "Full HD" provides the full physical resolution of 1920 x 1080 pixels.
Bibliography: [MÄUSL6], [ITU709], [ITU1120]
6 Transforms to and from the Frequency Domain
1.0
0.5
2nd harmonic
harmonic
2nd
|U|(f)
1stharmonic
harmonic
1st
In this chapter, principles of transforms to and from the frequency domain
are discussed. Although it describes methods which are used quite generally throughout the field of electrical communication, a thorough knowledge of these principles is of great importance to understanding the subsequent chapters on video encoding, audio encoding and Orthogonal
Frequency Division Multiplex (OFDM), i.e. DVB-T and DAB. Experts, of
course, can simply skip this chapter.
0.5
0.2
0.1
0
1
2
3
4
f
DC
fundamental
Fundamentalwave
wave
t
u(t) = 0.5 + 1.0sin(t+0.2)+0.5sin(2t)+0.2sin(3t-1)+0.1sin(4t-1.5);
Fig. 6.1. Fourier Analysis of a periodic time domain signal
Signals are normally represented as signal variation with time. An oscilloscope, for example, shows an electrical signal, a voltage, in the time domain. Voltmeters provide only a few parameters of these electrical signals,
e.g. the DC component and the RMS value. These two parameters can also
be calculated from the voltage variation by using a modern digital oscilloscope. A spectrum analyzer shows the signal in the frequency domain. It is
possible to think of any time domain signal as being composed of an infi-
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_6, © Springer-Verlag Berlin Heidelberg 2010
94
6 Transforms to and from the Frequency Domain
nite number of sinusoidal signals of a certain amplitude, phase and frequency.
The time domain signal is obtained by adding together all the sinusoidal
signals at every point in time, i.e. the original signal is obtained from the
superposition. A spectrum analyzer, however, only shows us the information about the amplitude or power of these sinusoidal part-signals, the
harmonics.
A periodic time domain signal can be resolved into its harmonics
mathematically by means of Fourier Analysis (Fig. 6.1.). This signal,
which can have any shape, can be thought of as being composed of the
fundamental wave which has the same period length as the signal itself,
and of the harmonics which are simply multiples of the fundamental. In
addition, each time domain signal also has a certain DC component. This
direct voltage corresponds to a zero frequency. Non-periodic signals can
also be represented in the frequency domain. but non-periodic signals do
not have a line spectrum but a continuous spectrum. Thus, the spectral
band contains spectral lines not only at certain points but at any number of
points.
+∞
H ( f ) = ³ h(t )e − j 2πft dt; Fourier Transform (FT)
−∞
∞
h(t ) =
³ H ( f )e
j 2πft
df ; Inverse Fourier Transform (IFT)
−∞
Time domain
Frequency domain
h(t)
FT
Re(f)
IFT
H(f)
time
f
Im(f)
f
Fig. 6.2. Fourier Transform
6.1 The Fourier Transform
95
6.1 The Fourier Transform
The spectrum of any time domain signal can be obtained mathematically
by means of the so-called Fourier Transform (Fig. 6.2.). This is an integral
transform in which the time domain signal has to be observed from minus
infinity to plus infinity. Such a Fourier Transform can thus only be solved
correctly if the time domain signal can be described in unambiguous terms
mathematically. The Fourier Transform then calculates the variation of the
real components and the variation of the imaginary components versus
frequency from the time domain signal. It is possible to assemble any sinusoidal signal of any amplitude, phase and frequency from a cosinusoidal
signal component of this frequency with a special amplitude and from a sinusoidal signal component of this frequency and special amplitude. The
real component accurately describes the amplitude of the cosinusoidal
component and the imaginary component accurately describes the amplitude of the sinusoidal component.
In the vector diagram (Fig. 6.3.), the vector of a sinusoidal quantity is
obtained by the vectorial addition of the real and imaginary parts, i.e. of
the sine and cosine components. The Fourier Transform thus provides the
information about the real part, i.e. the cosine component, and the imaginary part, i.e. the sine component, at any point in the spectrum in infinitely
fine resolution. The Fourier Transform is possible forwards and backwards
and is referred to as Fourier Transform (FT) and Inverse Fourier Transform (IFT), respectively.
The Fourier transform turns a real time domain signal into a complex
spectrum which is composed of real parts and imaginary parts as described. The spectrum consists of positive and negative frequencies and the
negative frequency range does not provide any additional information
about the time domain signal in question. The real part is mirrorsymmetrical with respect to the zero frequency and Re(-f) = Re(f) holds
true whereas the imaginary part is point-to-point-symmetrical and Im(-f) =
-Im(f) holds true. The Inverse Fourier Transform supplies a single real
time domain signal again from the complex spectrum. The Fourier Analysis, i.e. the analysis of the harmonics, is nothing else than a special case of
a Fourier Transform where the Fourier Transform is simply applied to a
periodic signal and the integral can then be replaced by a summation formula. The signal can be unambiguously described since it is periodic. The
information over one period is sufficient.
By applying Pythagoras’s theorem or the arc tangent, respectively, amplitude and phase information can be obtained from the real and imaginary
96
6 Transforms to and from the Frequency Domain
parts if required (Fig. 6.4.). The group delay characteristic is obtained by
differentiating the phase variation with frequency.
u(t)
u(t)
sin
Q
Im
AA=vector
= vectorlength
length
A
A
im
tt
ϕ
f=1/T;
f=1/T
re
ϕ
TT
Re
I
cos
Euler formula:
A e (2 ft + ) = re cos (2 ft) + j im(2 ft);
u(t)=A sin(2 t/T + );
Fig. 6.3. Vector diagram of a sinusoidal signal
Time domain
Frequency domain
A(f)
u(t)
f
time
(f)
A( f ) = (Re( f ) 2 + Im( f ) 2 );
ϕ ( f ) = arctan(
Im( f )
);
Re( f )
f
Fig. 6.4. Amplitude and phase characteristic
6.2 The Discrete Fourier Transform (DFT)
97
6.2 The Discrete Fourier Transform (DFT)
Signals of a quite general format cannot be described mathematically;
there are no periodicities and they would have to be observed for an infinite period of time which is impossible in practice. There is thus no possible mathematical or numerical approach for calculating its spectrum. One
solution which approximately supplies the frequency band is the Discrete
Fourier Transform (DFT). Using, e.g. an analog/digital converter, the signal is sampled at discrete points in the time domain at intervals Δt and observed only within a limited time window at N points (Fig. 6.5.).
Time domain
Frequency domain
Re(f)
u(t)
N points
DFT
ts
f
N points
time
Im(f)
f=fs/N
T
N points
f
fs = 1/ts
Fig. 6.5. Discrete Fourier Transform (DFT)
Instead of an integral from minus infinity to plus infinity, only a
summation formula has to be solved then and this can even be done purely
numerically by means of digital signal processing. The discrete Fourier
transform results in N points for the real part(f) and N points for the
imaginary part(f) in the spectral band.
The Discrete Fourier Transform (DFT) and the Inverse Discrete Fourier
Transform (IDFT) are obtained through the following mathematical relations:
98
6 Transforms to and from the Frequency Domain
N −1
H n = ¦ hk e
− j 2 πk
n
N
N −1
= − ¦ hk cos(2πk
k =0
hk =
1
N
k =0
N −1
¦H
n
e
n
j 2πk
N
N −1
n
n
) − j ¦ hk sin(2πk );
N
N
k =0
;
n =0
The frequency band thus no longer has an infinitely fine resolution and
is described only at discrete frequency interpolation points. The band extends from DC to half the sampling frequency and then continues symmetrically or point-to-point-symmetrically up to the sampling frequency. The
real-time graph is symmetrical up to half the sampling frequency and the
imaginary part is point-to-point-symmetrical. The frequency resolution is a
function of the number of points in the window of observation and on the
sampling frequency.
The following applies:
Δf =
1
fs
; Δt = ;
N
fs
The Discrete Fourier Transform (DFT), in reality, actually corresponds
to a Fourier analysis within the observed time window of the band-limited
signal. It is thus assumed that the signal in the observed time window continues periodically. This assumption results in “uncertainties” in the analysis so that the Discrete Fourier Transform can only supply approximate information about the actual frequency band. ‘Approximate’ in as much as
the areas preceding and following the time window are not taken into consideration and the signal window is sharply truncated. However, the DFT
can be solved by simple mathematical and numerical means and it functions both forwards and in the reverse direction in the time domain (Inverse Discrete Fourier Transform – IDFT, Fig. 6.6.). The result of
performing a DFT on a real time domain signal interval is a discrete complex spectrum (real and imaginary parts). The IDFT transforms the
complex spectrum back into a real time domain signal again. In reality,
however, the section of time domain signal cut out and transformed into
the frequency domain has been converted into a periodic signal.
Once a rectangular time domain signal segment has been windowed, the
spectrum corresponds to a convolution of a sin(x)/x function with the
original spectrum of the signal. This produces different effects which in a
spectrum analysis done by means of the DFT disturb and affect the measurement result to a greater or lesser extent. In test applications, therefore,
the choice would be not to select a rectangular window function but, e.g.
cos2 function which would cut out a smoother window and lead to fewer
6.3 The Fast Fourier Transform (FFT)
99
disturbances in the frequency domain. Various types of window function
are used, e.g. rectangular windows, Hanning windows, Hamming windows, Blackman windows etc.. Windowing means that the signal segment
is first cut out to a rectangular shape and then multiplied by the window
function.
Time domain
Frequency domain
Re(f)
u(t)
IDFT
N points
N
points
f
time
Im(f)
f=fs/N
T
Periodic signal
N points
f
fs = 1/ts
Fig. 6.6. IDFT
6.3 The Fast Fourier Transform (FFT)
The Discrete Fourier Transform is a simple but fairly time-consuming algorithm. However, if the number of points N within the window of observation is restricted to N=2x, i.e. a power of two (Cooley, Tukey, 1965), a
more complex, but less time-consuming algorithm, the Fast Fourier Transform (FFT), can be used. This algorithm itself provides exactly the same
result as a DFT but is much faster and is restricted to N=2x points (2, 4, 8,
16, 32, 64, ...,256, …,1024, 2048, …,8192, ...). The Fast Fourier Transform can also be inverted (Inverse Fast Fourier Transform - IFFT).
The FFT algorithm makes use of methods of linear algebra. The samples
are presorted in co-called bit reversal and then processed by means of butterfly operations. These operations are implemented as machine codes in
signal processors and special FFT chips.
100
6 Transforms to and from the Frequency Domain
The number of multiplications given below shows the time gained by
the FFT compared with the DFT:
Number of multiplications needed:
N•N
N • log(2N)
DFT:
FFT;
The FFT has long been used in the field of acoustics (surveying concert
halls and churches) and in geology (searching for minerals, ores and oil).
However, the analyses were performed off-line with fast computers, using
a Dirac impulse to excite the medium to be examined (hall, rocks) and then
recording the impulse response of the medium under investigation. A
Dirac impulse is a very short and very strong impulse, an example of an
acoustical Dirac impulse being a pistol shot and a geological Dirac impulse
being the explosion of a blasting charge.
Back in 1988, a 256 point FFT still consumed minutes of PC time. Today, an 8192 point FFT (8k FFT) takes less than one millisecond of computing time! This opens the door for new and interesting applications such
as video and audio compression or Orthogonal Frequency Division Multiplex (OFDM). FFT has also been used increasingly for spectrum analysis
in analog video testing and for detecting the amplitude and group delay response of video transmission links since the late 1980s. In modern storage
oscilloscopes, too, this interesting test function is frequently found today
and makes it possible to perform a low-cost spectrum analysis, especially
also in audio test engineering.
Frequency domain
Time domain
u(t)
Re(f)
re(t)
ts
N points
Re(f)
N points
FFT/DFT
f
time
Im(f)
T
0
im(t)
f=fs/N
N points
IFFT/IDFT
Im(f)
f
fs = 1/ts
Fig. 6.7. Implementation and practical applications of DFT and FFT
6.5 The Discrete Cosine Transform (DCT)
101
6.4 Implementation and Practical Applications of DFT and
FFT
The Fourier Transform, the Discrete Fourier Transform and the Fast Fourier Transform are all defined through the field of complex numbers. This
means that both the time domain signal and the frequency domain signal
have real and imaginary parts. Typical time domain signals are, however,
always purely real, i.e. the imaginary part is zero at every point in time.
The imaginary part must, therefore, be set to zero before the Fourier transform or its numerical variations DFT and FFT are performed.
When DFT or FFT and IDFT or IFFT are performed in practice two input signals are required (Fig. 6.7.). The input signals are implemented as
real-part and imaginary-part tables and correspond to the sampled time or
frequency domain. As the N samples of a typical time domain signal are
always real, the corresponding imaginary part must be set to zero for each
of the N points. This means that the imaginary-part table for the time domain must be filled with zeros. When the inverse transform is performed,
the imaginary part of the time domain signal must again be zero assuming
that the frequency range for the real part is about half the sampling frequency and the frequency range for the imaginary part is point-to-point
symmetric about half the sampling frequency. If these symmetries are not
present in the frequency domain, a complex time domain signal is output,
i.e. the signal also has imaginary components in the time domain.
6.5 The Discrete Cosine Transform (DCT)
The Discrete Cosine Transform (DCT), and thus also the fast Fourier
Transform which is a special case of the DCT, is a cosine-sine transform
as can be seen from its formula; it is an attempt to assemble a time-domain
signal segment by the superposition of many different cosine and sine signals of different frequency and amplitude. A similar result can also be
achieved by using only cosine signals or only sine signals.
They are then called Discrete Cosine Transform (DCT) (Fig. 6.8.) or
Discrete Sine Transform (DST) (Fig. 6.9.). Compared with the DFT, the
sum of single signals required remains the same but twice as many cosine
or sine signals are required. In addition, half-integral multiples of the fundamental are needed as well as integral multiples. The Discrete Cosine
Transform (Fig. 6.8.), especially, has become quite important for audio and
video compression.
102
6 Transforms to and from the Frequency Domain
The formulas of the Discrete Cosine Transform (DCT) and the Discrete
Sine Transform (DST) are:
N −1
Fk = ¦ f z cos(
z =0
1
N −1
πzk
2 ); F =
f z sin(
);
¦
k
N
N
z =0
πk ( z + )
cos(0)
cos(0.5x)
cos(x)
cos(1.5x)
cos(2x)
Fig. 6.8. Discrete Cosine Transform (DCT)
sin(0.5x)
sin(x)
sin(1.5x)
sin(2x)
sin(2.5x)
Fig. 6.9. Discrete Sine Transform (DST)
The DCT supplies, in the time domain, the amplitudes of the cosine signals from which the time interval analyzed can be assembled. The zero coefficient corresponds to the DC component of the signal segment. All the
6.5 The Discrete Cosine Transform (DCT)
103
other coefficients first describe the low-frequency components, then the
medium-frequency and then the higher-frequency components of the signal or, respectively, the amplitudes of the cosine functions from which the
time-domain signal segment can be generated by adding them together.
The response of the DCT is relatively gentle at the edges of the signal
segment cut out and will lead to lesser discontinuities if a signal is transformed and retransformed segment by segment. This may well be the reason why the DCT has attained such great importance in the field of compression.
250
200
150
100
50
x
0
1
2
3
4
5
6
7
250
200
150
110
100
50
50
20
10
0
1
2
3
10
4
3
5
20
6
8
7
n
y
250
200
150
100
50
0
-50
x
y(x)=110+10cos(0.5x)+50cos(x)+20cos(1.5x)
+10cos(2x)+3cos(2.5x)+20cos(3x)+8cos(3.5x);
Fig. 6.10. DCT and IDCT
104
6 Transforms to and from the Frequency Domain
The DCT is the algorithm at the core of the JPEG and MPEG image
compression (digital photography and video) in which an image is transformed two-dimensionally block by block into the frequency domain and
compressed block by block. It is of particular importance that the block
edges cannot be recognized in the image after its decompression (no discontinuities at the edges).
The discrete cosine transform does not supply the coefficients in the
frequency domain in pairs, i.e. separated according to real and imaginary
parts and does not provide any information about the phase, only about the
amplitude. Neither does the amplitude characteristic correspond directly to
the result of the DFT. But this type of frequency transform is adequate for
many applications and is also possible in both directions (Inverse Discrete
Cosine Transform - IDCT) (Fig. 6.10.).
In principle, of course, there is also a Discrete Sine Transform (Fig. 6.9)
where it is attempted to duplicate a time domain signal by the superposition of pure sinusoidal signals.
U(f)
u(t)
t
f
T
1/T
Fig. 6.11. Fourier Transform of a single squarewave pulse
U(f)
u(t)
t
T
f
1/T
Fig. 6.12. Fourier Transform of a periodic squarewave pulse
6.6 Time Domain Signals and their Transforms in the Frequency Domain
105
6.6 Time Domain Signals and their Transforms in the
Frequency Domain
In the following paragraphs, some important time-domain signals and their
transforms in the frequency domain will be discussed. The purpose of
these observations is to get some feel for the results of the Fast Fourier
Transform.
Let us begin with a periodic squarewave signal (Fig. 6.12.): since it is a
periodic signal, it has discrete lines in the frequency spectrum; all discrete
spectral lines of the squarewave signal are located at integral multiples of
the fundamental frequency of the squarewave signal. Most of the energy
will be found in the fundamental wave itself. If there is a DC component, it
will result in a spectral line at zero frequency (Fig. 6.14.). The envelope of
the spectral lines of the fundamental and the harmonics is the sin(x)/x
function.
U(f)
u(t)
t
f
Fig. 6.13. Fourier Transform of a Dirac impulse
U(f)
u(t)
t
f
Fig. 6.14. Fourier Transform of a pure direct voltage (DC)
If then the duration of the period T of the squarewave signal is allowed
to tend towards infinity, the discrete spectral lines move closer and closer
together until a continuous spectrum of a single pulse is obtained (Fig.
6.11.).
106
6 Transforms to and from the Frequency Domain
The spectrum of a single squarewave pulse is a sin(x)/x function. If then
the pulse width T is allowed to become narrower and narrower and to tend
towards zero, all zero points of the sin(x)/x function will tend towards infinity. In the time domain, this provides an infinitely short pulse, a socalled Dirac impulse, the Fourier Transform of which is a straight line; i.e.
the energy is distributed uniformly from zero frequency to infinity (Fig.
6.13.). Conversely, a single Dirac needle at f=0 in the frequency domain
corresponds to a direct voltage (DC) in the time domain.
U(f)
u(t)
-T
t
T 2T
f
-1/T 1/T
2/T
Fig. 6.15. Fourier Transform of a sequence of Dirac impulses
U(f)
u(t)
t
-1/T
1/T
f
T
Fig. 6.16. Fourier Transform of a sinusoidal signal
A sequence of Dirac impulses spaced apart at intervals T from one another again results in a discrete spectrum of Dirac needles spaced apart by
1/T (Fig. 6.15.). The Dirac impulse train is of importance when considering a sampled signal. Sampling an analog signal has the consequence that
this signal is convoluted with a sequence of Dirac impulses.
To conclude, a purely sinusoidal signal will be considered. Its Fourier
transform is a Dirac needle at the frequency of the sinewave fs and –fs
(Fig. 6.16.).
6.7 Systematic Errors in DFT or FFT, and How to Prevent them
107
6.7 Systematic Errors in DFT or FFT, and How to Prevent
them
To obtain the precise result of the Fourier Transform, a time-domain signal
would have to be observed for an infinitely long period of time. In the case
of the Discrete Fourier Transform, however, a signal segment is only observed for a finite period of time and transformed. The result of the DFT or
FFT, respectively, will thus always differ from that of the Fourier Transform. It has been seen that, in principle, this analyzed time segment is converted into periodic signals in the DFT, i.e., the result of the DFT must be
considered to be the Fourier Transform of this converted time segment.
u(t)
u(t)
N points
time
N points
time
T
T
Fig. 6.17. Conversion of a signal segment into periodic signals by the DFT or
FFT, resp.
u(t)
u‘(t)
T2
T2
t
T1
t
T1
Fig. 6.18. Windowing (T1, T2) a sinusoidal signal
It is clear that, naturally, the result of the transform depends greatly on
the type and position of the “cutting-out” process, the so-called windowing. This can be visualized best by performing the DFT on a sinusoidal
108
6 Transforms to and from the Frequency Domain
signal. If exactly one sample is taken from the sinusoidal signal so that it
has a length of a multiple n=1, 2, 3 etc of the period, the result of the DFT
will exactly match that of the Fourier transform because converting this
time segment into periodic signals will again produce a signal which is exactly sinusoidal.
|U(f)|
f
Fig. 6.19. Picket fence effect
|U(f)|
f
Fig. 6.20. Dispersal of the energy to main and side lobes
If, however, the length of the window (Fig. 6.18.) cut out differs from
the length of the period, the result of the transform will differ more or less
from the expected value depending on the number of cycles included. A
sample of less than one fundamental wave will have the worst effect. A
Dirac needle will become a wider “lobe”, in some cases with “sidelobes”.
The amplitude of the main lobe will correspond more or less to the expected value. Leaving the period of observation constant and varying the
frequency of the signal, the amplitude of the spectral line will fluctuate and
will correspond to the expected value whenever there is exactly one multi-
6.7 Systematic Errors in DFT or FFT, and How to Prevent them
109
ple of the period within the window of observation; in between that it will
become smaller and assume the exact value time and again. This is called
the “picket fence” effect (Fig. 6.19.).
The fluctuation in the amplitude of the spectral line is caused by a dispersal of the energy due to a widening of the main lobe and by the appearance of sidelobes (Fig. 6.20.).
In addition, aliasing products may appear if the measurement signal is
not properly band-limited; moreover quantization noise becomes visible
and will limit the dynamic range.
These systematic errors can be prevented or suppressed by programming an observation time of corresponding length, by good suppression of
aliasing products and by using A/D converters having a correspondingly
high resolution. In the next section, “windowing” will be discussed as a
further aid in suppressing DFT system errors.
u(t)
u(t)
original
signal
Original signal
k(t)
k(t)
window
function
Window
function
(e.g. Hanning)
(e.g.
Hanning)
u'(t) = u(t)*k(t);
u`(t)
= k(t)•u(t);
Windowed
windowed
time signal timedomain signal
Fig. 6.21. Multiplying a signal by a window function
110
6 Transforms to and from the Frequency Domain
6.8 Window Functions
In the last section it was shown that windows with abrupt or “hard” edge
transitions produced spurious effects, so-called leakage, as picket fence effect and sidelobes. The main lobe is dispersed depending on whether an integral multiple of the period has been sampled or not.
These leakage effects can be reduced by using soft windowing, i.e. a
window function with soft edges, instead of a rectangular window with
hard rectangular edges.
Fig. 6.21. shows that in windowing, the original signal is weighted, i.e.
multiplied by the window function k(t). The signal is cut out softly towards
the edge. The window function shown is the Hanning window function - a
simple cosine squared window which is the most commonly used window.
The sidelobes are attenuated more and the picket fence effect is reduced.
There are a number of windows used in practice, examples of which are:
•
•
•
•
•
•
•
•
Rectangular window
Hanning window
Hamming window
Triangular window
Tukey window
Kaiser Bessel window
Gaussian window
Blackman window
Depending on the window selected, the main lobes are widened to a
greater or lesser extent, the sidelobes are attenuated more or less, and the
picket fence effect is reduced to a greater or lesser extent. Rectangular
windowing means maximum or no cutting out, the Hanning cosine squared
window was shown in the Figure. Regarding the other windows, reference
is also made to the relevant literature references and to the article by
[HARRIS].
Bibliography: [COOLEY], [PRESS], [BRIGHAM],
[FISCHER], [GIROD], [KUEPF], [BRONSTEIN]
[HARRIS],
7 MPEG-2 Video Coding
Digital SDTV (Standard Definition Television) video signals (uncompressed) have a data rate of 270 Mbit/s. This data rate is much too high for
broadcasting purposes, which is why they are subjected to a compression
process before being processed for transmission. The 270 Mbit/s must be
compressed to about 2...7 Mbit/s - a very high compression factor which,
however, is possible due to the use of a variety of redundancy and irrelevance reduction mechanisms. The data rate of an uncompressed HDTV
signal is even higher than 1 Gbit/s and MPEG-2 coded HDTV signals have
a data rate of about 15 … 20 Mbit/s.
Lens
Iris
Retina
Pupil
Anterior
chamber
Optic nerve
Vitreous chamber
Rods
(B / W
receptors)
Cones
(color
receptors)
Fig. 7.1. Anatomics of the human eye
7.1 Video Compression
To compress data, it is possible to remove redundant or irrelevant information from the data stream. Redundant means superfluous, irrelevant means
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_7, © Springer-Verlag Berlin Heidelberg 2010
112
7 MPEG-2 Video Coding
unnecessary. Superfluous information is information which exists several
times in the data stream, or information which has no information content,
or simply information which can be easily and losslessly recovered by
mathematical processes at the receiving end. Redundancy reduction can be
achieved, e.g. by variable-length coding. Instead of transmitting ten zeroes,
the information ‘ten times zero’ can be sent by means of a special code
which is much shorter.
Limit angle of
perceptibility of
structures
of the
human
eye:
~1.5 minutes of angle
Fig. 7.2. Limit angle of perceptibility of structures of the human eye
The alphabet of the Morse code, too, uses a type of redundancy reduction. Letters which are used frequently are represented by short code sequences whereas letters which are used less frequently are represented by
longer code sequences. In information technology, this type of coding is
called Huffman coding or variable length coding.
Wavelength [nm]
Blue cyan green yellow red
Fig. 7.3. Luminance sensitivity of the human eye
Irrelevant information is the type which cannot be perceived by the human senses. In case of the video signal, they are the components which the
7.1 Video Compression
113
eye does not register due to its anatomy. The human eye (Fig. 7.1.) has far
fewer color receptors than detection cells for brightness information. For
this reason, the “sharpness in the color” can be reduced which means a reduction in the bandwidth of the color information. The receptors for
black/white are called rods and the color receptors are cones, both of
which are located on the retina of the eye, behind the lens. The lens focuses the image sharply onto the retina. The rods have their main function
in night vision and are much more sensitive and present in much greater
numbers. The limit angle of the perceptibility of structures is a function of
the number of rods in the human eye and is about 1.5 minutes of angle
(Fig. 7.2.). There are red-, green- and blue-sensitive cones, the sensitivity
to green being much more pronounced than that for blue and red and that
for red, in turn, being greater than that for blue (Fig. 7.3.). This also finds
its expression in the matrixing formula for forming the luminance signal:
Y = 0.30 • R + 0.59 • G + 0.11 • B
Fig. 7.4. Perception test for coarse and fine image structures
It is also known that we cannot discern fine structures in a picture, e.g.
thin lines, as well as coarse structures. This can be illustrated well by perception tests (Fig. 7.4). If, e.g., one varies the size of a spot and its brightness against a background, the color of which can also be varied, it can be
demonstrated that at some point the human eye can no longer see a small
spot which differs only slightly from the background. This is precisely the
main point of attack for data reduction methods like JPEG and MPEG,
where coarse structures are transmitted with much greater accuracy, i.e.
with many more bits, than fine structures, performing, in fact, an irrelevance reduction, a so-called perception coding. However, irrelevance re-
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7 MPEG-2 Video Coding
duction is always associated with an irretrievable loss of information
which is why the only method considered in data processing is redundancy
reduction as, e.g. in the well known ZIP files.
Data reduction
Redundancy reduction
(no loss of information)
reversible
Irrelevance reduction
(loss of information)
non-reversible
Fig. 7.5. Data reduction
In MPEG, the following steps are carried out in order to achieve a data
reduction factor of up to 130:
•
•
•
•
•
•
•
8 bits resolution instead of 10 bits (irrelevance reduction)
Omitting the horizontal and vertical blanking interval (redundancy reduction)
Reducing the color resolution also in the vertical direction
(4:2:0) (irrelevance reduction)
Differential pulse code modulation (DPCM) of moving pictures
(redundancy reduction)
Discrete cosine transform (DCT) followed by quantization (irrelevance reduction)
Zig-zag scanning with variable-length coding (redundancy reduction)
Huffman coding (redundancy reduction)
Let us begin again with the analog video signal from a television camera. The red, green and blue (RGB) output signals are matrixed to become
Y, CB and CR signals. After that, the bandwidth of these signals is limited
and they are analog/digital converted. According to ITU-BT.R601, this
provides a data signal with a data rate of 270 Mbit/s. The color resolution
is reduced in comparison with the brightness resolution, making the number of brightness samples twice that of the CB and CR values and resulting
in a 4:2:2 signal; there is thus already an irrelevance reduction in ITU-
7.1 Video Compression
115
BT.R601. It is this 270 Mbit/s signal which must be compressed to about
2...7 (15) Mbit/s in the MPEG video coding process.
7.1.1 Reducing the Quantization from 10 Bits to 8
In analog television, the rule of thumb was that when a video signal has a
signal/noise ratio, referred to white level and weighted, of more than 48
dB, the noise component is just below the threshold of perception of the
human eye. Given the appropriate drive to the A/D converter, the quantization noise from the 8 bit resolution is already well below this threshold so
that a 10 bit resolution in Y, CB and CR is unnecessary outside the studio.
In the studio, 10 bit resolution is better because post-processing is easier
and gives better results. Reducing the data rate from 10 bits to 8 bits compared with ITU BT.R601 means a reduction in the data rate of 20 % ((108)/10 = 2/10 = 20 %), but this is an irrelevance reduction and the original
signal cannot be recovered in the decoding at the receiving end. According
to the rule of thumb that S/N [dB] = 6•N, the quantization noise level has
now risen by 12 dB.
625
lines
Horzontal blankig
Vertical blanking
Visible, active picture
575
visible lines
Active line
Fig. 7.6. Horizontal and vertical blanking
7.1.2 Omitting the Horizontal and Vertical Blanking Intervals
The horizontal and vertical blanking intervals of a digital video signal according to ITU BT.R601 (Fig. 7.6.) do not contain any relevant information, not even teletext. These areas can contain supplementary data such as
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sound signals but these must be transmitted coded separately according to
MPEG. The horizontal and vertical blanking intervals are, therefore, left
out completely in MPEG. The horizontal and vertical blanking intervals
and all signals in them can be regenerated again without problems at the
receiving end.
A PAL signal has 625 lines, only 575 of which are visible. The difference of 50 lines, divided by 625, is 8 % which is the saving in data rate
achieved when the vertical blanking is omitted. The length of one line is
64 µs but the active video area is only 52 µs which, divided by 64,
amounts to a further saving of 19 % in the data rate. Since there is some
overlap in the two savings, the total result of this redundancy reduction is
about 25 %.
7.1.3 Reduction in Vertical Color Resolution (4:2:0)
The two color difference signals CB and CR are sampled at half the data
rate compared with the luminance signal Y. In addition, the bandwidth of
CB and CR is also reduced to 2.75 MHz in comparison with the luminance
bandwidth of 5.75 MHz - a 4:2:2 signal (Fig. 7.7.). However, the color
resolution of this 4:2:2 signal is only reduced in the horizontal direction.
The vertical color resolution corresponds to the full resolution resulting
from the number of lines in a television frame.
Column m
Y
Cb
Column m + 1
Y
Column m
Cb
Line n
Cr
Y
Cb
Y
Cb
Column m + 2
Y
Y
Line n
Y
Y
Line n + 1
Cr
Cr
Cb
Cb
Line n + 1
Cr
Y
Column m + 1
Cr
Y
Cr
Fig. 7.7. 4:4:4 and 4:2:2 resolution
However, the human eye cannot distinguish between horizontal and vertical as far as color resolution is concerned. It is possible, therefore, to also
reduce the color resolution to one half in the vertical direction without perceptible effect. MPEG-2 does this usually in one of the first steps and the
signal then becomes a 4:2:0 signal (Fig.7.8.). Four Y pixels are now in
7.1 Video Compression
117
each case associated with only one CB value and one CR value each. This
type of irrelevance reduction results in another saving of exactly 25 % data
rate.
Column m
Column m + 1
Y
Column m + 2
Y
Y
Line n
Y
Y
Line n + 1
Cb
Cr
Y
Fig. 7.8. 4:2:0 resolution
Horizontal blankig
Vertical blanking
625
lines
Visible, active picture
720x576 pixel Y
360x576 (288@4:2:0)
pixel Cb, Cr
25 frames/s
576
visible lines
Active line
Fig. 7.9. Physical parameters of a SDTV signal
7.1.4 Further Data Reduction Steps
The data reduction carried out up to now has produced the following result: Beginning with an original data rate of 270 Mbit/s, this ITU BT.R601
signal has now been compressed to 124.5 Mbit/s, i.e. to less than half its
original rate, by applying the following steps:
•
•
•
•
ITU BT.R601
8 bits instead of 10 (-20%)
Hor. and vert. blanking (appr. -25%)
4:2:0 (-25%)
= 270 Mbit/s
= 216 Mbit/s
= 166 Mbit/s
= 124.5 Mbit/s
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7 MPEG-2 Video Coding
There is, however, still a large gap between the 124.5 Mbit/s now
achieved and the required 2...6 Mbit/s with its upper limit of 15 Mbit/s,
and this gap needs to be closed by means of further steps which are much
more complex.
255
0
time
ts
Fig. 7.10. Pulse Code Modulation
255
0
time
ts
Fig. 7.11. Differential Pulse Code Modulation
7.1.5 Differential Pulse Code Modulation of Moving Pictures
Adjoining moving pictures differ only very slightly from each other. They
contain stationary areas which won’t change at all from frame to frame;
there are areas which only change their position and there are objects
which are newly added. If each frame were to be transmitted completely
every time, some of the information transmitted would always be the same,
resulting in a very high data rate. The obvious conclusion is to differentiate
7.1 Video Compression
119
between these types of picture areas and to transmit only the difference,
i.e. the delta value, from one frame to the next. This particular method of
redundancy reduction, which is based on a method which has been known
for a long time, is called differential pulse code modulation (DPCM).
255
0
time
ts
Fig. 7.12. Differential Pulse Code Modulation with reference values
Fig. 7.13. Dividing a picture into blocks and macroblocks
What then is differential pulse code modulation? If a continuous analog
signal is sampled and digitized, discrete values, i.e. values which are no
longer continuous, are obtained at equidistant time intervals (Fig.7.10.).
These values can be represented as pulses spaced apart at equidistant intervals, which corresponds to a pulse code modulation. The height of each
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7 MPEG-2 Video Coding
pulse carries information in discrete, non-continuous form about the current state of the sampled signal at precisely this point in time.
In reality, the differences between adjacent samples, i.e. the PCM values, are not very large because of the previous band-limiting. If only the
difference between adjacent samples is transmitted, transmission capacity
can be saved and the required data rate is reduced. This type of pulse code
modulation is a relatively old idea and is now called differential pulse code
modulation (Fig. 7.11.).
The problem with the usual DPCM is, however, that after a switch-on or
after transmission errors it takes a very long time until the demodulated
time domain signal again matches the original signal to some extent. This
problem can be eliminated though by employing the small trick of transmitting at regular intervals firstly complete samples, then a few differences
followed again by a complete sample etc. (Fig.7.12.) This very closely
approaches the differential pulse code modulation method used in the
MPEG-1/-2 image compression.
GOP
I
P
= Motion vector
P
P
= Block
P
I
I = Intra frame encoded picture
P = “Predicted" forward encoded picture
GOP = Group of Pictures
Fig. 7.14. Forward predicted delta frames
Before a frame is examined for stationary and moving components, it is
first divided into numerous square blocks of 16x16 luminance pixels and
8x8 CB and CR pixels each (Fig.7.13.). Due to the 4:2:0 pattern, 8x8 CB
pixels and 8x8 CR pixels are in each case overlaid on one layer of 16x16
luminance pixels each. This arrangement is now called a macroblock
(Fig.7.25.). One single frame is composed of a large number of macro-
7.1 Video Compression
121
blocks and the horizontal and vertical number of pixels is selected to be
such that it is divisible by 16 and also by 8 (Y: 720 x 576 pixels). At certain intervals complete reference frames, so-called I (intracoded) frames,
formed without forming the difference, are then repeatedly transmitted and
interspersed between them the delta frames (interframes).
Forming the difference is done at macroblock level, i.e. the respective
macroblock of a following frame is always compared with the macroblock
of the preceding frame. Put more precisely, this macroblock is first examined to see whether it has shifted in any direction due to movement in the
picture, has not shifted at all or whether the picture information in this
macroblock is completely new. If there is a simple displacement, only a
so-called motion vector is transmitted. In addition to the motion vector, it
is also possible to transmit the difference, if any, with respect to the preceding macroblock. If the macroblock has neither shifted nor changed in
any way, nothing needs to be transmitted at all. If no correlation with an
adjoining preceding macroblock can be found, the macroblock is completely recoded. Such pictures produced by simple forward prediction are
called P (predicted) pictures (Fig.7.14.).
GOP
I
B
Forward
encoding
B
P
B
Backward
encoding
I
I = Intra frame encoded picture
P = “predicted" forward encoded picture
B = "bidirectional" encoded picture
GOP = Group of Pictures
Fig. 7.15. Bidirectionally predicted delta frames
Apart from unidirectionally forward predicted frames there are also
bidirectionally, i.e. forward and backward, predicted delta frames, socalled B pictures. The reason for this is the much lower data rate in the B
pictures compared with the P pictures or even I pictures, which becomes
122
7 MPEG-2 Video Coding
possible as a result of this. The arrangement of frames occurring between
two I pictures, i.e. complete pictures, is called a group of pictures (GOP)
(Fig.7.14.).
The motion estimation for obtaining the motion vectors proceeds as follows: Starting with a delta frame to be encoded, the system looks in the
preceding frame (forward prediction P) and possibly also in the subsequent
frame (bidirectional prediction B) for suitable macroblock information in
the environment of the macroblock to be encoded. This is done by using
the principle of block matching within a certain search area around the
macroblock.
Matching window
Frame N-1,
motion vector
forward
Frame N,
B encoded
macro block
Frame N+1,
motion vector
backward
Fig. 7.16. Motion vectors
If a matching block is found in front, and also behind in the case of bidirectional coding, the motion vectors are determined forward and backward and transmitted. In addition, any additional block delta which may be
necessary can also be transmitted, both forward and backward. However,
the block delta is coded separately by DCT with quantization, described in
the next chapter, a method which saves a particularly large amount of storage space.
A group of pictures (GOP) then consists of a particular number and a
particular structure of B pictures and P pictures arranged between two I
pictures. A GOP usually has a length of about 12 frames and corresponds
to the order of I, B, B, P, B, B, P, .... The B pictures are thus embedded between I and P pictures. Before it is possible to decode a B picture at the re-
7.1 Video Compression
123
ceiving end, however, it is absolutely necessary to have the information of
the preceding I and P pictures and that of the following I or P picture in
each case. But according to MPEG, the GOP structure can be variable. So
that not too much storage space needs to be reserved at the receiving end,
the GOP structure must be altered during the transmission so that the respective backward prediction information is already available before the
actual B pictures. For this reason, the frames are transmitted in an order
which no longer corresponds to the original order.
B5
I10
B4
P6
B2
B1
P3
B-1
B-2
I0
Fig. 7.17. Order of picture transmission
Instead of the order I0, B1, B2, P3, B4, B5, P6, B7, B8, P9, the pictures are
now transmitted in the following order: I0, B-2, B-1, P3, B1, B2, P6, B4, B5,
P9, etc. (Fig. 7.17.). That is to say, the P or I pictures following the B pictures are now available at the receiving end before the corresponding B
pictures are received and must be decoded. The storage space to be reserved at the receiving end is now calculable and limited. To be able to restore the original order, the frame numbers must be transmitted coded in
some way. For this purpose, the DTS (decoding time stamp) values contained in the PES header are used, among other things (see Section 3, The
MPEG Data Stream).
7.1.6 Discrete Cosine Transform Followed by Quantization
A very successful method for still-frame compression has been in use since
the end of the eighties: the JPEG method, which today is also being used
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7 MPEG-2 Video Coding
for digital cameras and produces excellent picture quality. JPEG stands for
Joint Photographic Experts Group, i.e. a group of experts in the coding of
still frames. The basic algorithm used in JPEG is the Discrete Cosine
Transform (Fig.7.18.), or DCT in brief. This DCT also forms the central
algorithm for the MPEG video coding method.
y
x
0
1
2
3
fz =
5
π k (z + )
z =0
N
2
N
k
7
IDCT
1
2 );
π k (z + )
N −1
¦F
6
1
2 );
N −1
Fk = ¦ f z cos(
DCT
4
cos(
N
k =0
250
200
150
110
100
50
50
20
10
0
1
2
3
10
3
4
5
20
6
8
7
n
Fig. 7.18. One-dimensional Discrete Cosine Transform
The human eye perceives fine structures in a picture differently from
coarse structures. In analog video test engineering it was already known
that low-frequency picture disturbances, i.e. picture disturbances which
correspond to coarse image structures or interfere with these are perceived
more readily than high-frequency disturbances, i.e. those corresponding to
fine image structures or interfering with these.
7.1 Video Compression
125
250
200
150
110
100
50
50
20
10
0
1
2
3
10
3
4
5
20
6
8
7
n
Q
50
40
Q
32
30
20
16
10
8
4
1
0
1
1
1
1
2
3
4
5
6
7
n
250
200
150
110
100
50
50
20
10
0
1
2
3
3
0
1
0
4
5
6
7
n
Fig. 7.19. Quantization of the DCT coefficients
For this reason, the signal/noise ratio had been measured weighted, i.e.
referred to the sensitivity of the eye, even at the beginning of video testing.
It was possible to allow for much more noise in the direction of higherfrequency image structures than with coarse, low-frequency image components. This knowledge is utilized in JPEG and in MPEG. Low-frequency,
coarse image components are coded with finer quantization and fine image
components are coded with coarser quantization in order to save data rate.
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7 MPEG-2 Video Coding
But how to separate coarse components from medium and fine image
components? It is done by means of transform coding (Fig.7.18.).
Firstly, a transition is made from the time domain of the video signal
into the frequency domain. The Discrete Cosine Transform is a special
case of the Discrete Fourier Transform or the Fast Fourier Transform, respectively.
y
y‘
y(x)=110+10cos(0.5x)+50cos(x)+20cos(1.5x)+10cos(2x)+3cos(2.5x)+20cos(3x)+8cos(3.5x);
y‘(x)=110+10cos(0.5x)+50cos(x)+20cos(1.5x)+12cos(2x)+0cos(2.5x)+16cos(3x)+0cos(3.5x);
Fig. 7.20. Original curve (y) and quantized Curve (y’)
These transforms are dealt with in a separate Section (5) of this book.
To begin with a simple example: Using the DCT, 8 samples in a video
line are transformed into the frequency domain (Fig.7.18.). This again provides 8 values which, however, no longer correspond to video voltage values in the time domain but to 8 power values in the frequency domain,
graded into DC, low- and medium- to high-frequency components within
these 8 transformed video voltage values. The first value (DC coefficient)
in the frequency domain corresponds to the energy of the video component
with the lowest frequency in this section up to medium- or higherfrequency signal components. The information in one video signal section
has now been processed in such a way that an irrelevance reduction can be
performed which corresponds to the sensitivity characteristic of the human
eye.
In a first step in this process, these coefficients are quantized in the frequency domain, i.e. each coefficient is divided by a certain quantization
factor (Fig.7.19.). The higher the value of the quantization factor, the
coarser the quantization. In the case of coarse image structures, the quanti-
7.1 Video Compression
127
zation must be changed only a little or not at all and in the case of fine image structures, the quantization is reduced more, meaning that the quantization factors increase in the direction of finer image structures. Due to the
quantization, many values which have become zero are obtained as the
fineness of the image structure increases, that is to say in the direction of
higher frequency coefficients.
f(x,y)
Fig. 7.21. 8 x 8 pixel block
55
70
92
111
116
108
94
83
70
81
94
103
101
87
70
58
16
81
42
56
62
62
57
52
34
36
38
39
37
33
29
26
67
63
57
52
50
50
51
53
46
40
33
28
32
42
55
64
46
32
10
-9
-16
-13
-3
5
24
10
-9
-23
-24
-13
5
18
Fig. 7.22. Subtracting 128
These values can then be coded in a special space-saving way. However, the characteristic recovered by decoding at the receiving end after the
quantization then no longer corresponds perfectly to the original curve
(Fig.7.20.) and exhibits quantization errors.
In practice, however, the coding in JPEG and MPEG is two-dimensional
transform coding. For this, the picture is divided into 8 x 8 pixel blocks
(Fig.7.13.). Each 8 x 8 pixel block (Fig.7.23.) is then transformed into the
frequency domain by means of the two-dimensional Discrete Cosine
Transform. Before that is done, the value 128 is first subtracted from all
pixel values in order to obtain signed values (Fig.7.22.).
128
7 MPEG-2 Video Coding
Frequency domain
Time domain
DCT
IDCT
8 x 8 pixel block
8 x 8 DCT coefficients
DC coefficient
F(v,u) = DCT(f(x,u);
Fig. 7.23. Two-dimensional DCT
The result (Fig.7.23.) of the two-dimensional Discrete Cosine Transform of an 8 x 8 pixel array is another 8 x 8 pixel array, but now in the frequency domain. The first coefficient of the first row is the DC coefficient
which corresponds to the DC component of the entire block. The second
coefficient corresponds to the energy of the coarsest image structures in
the horizontal direction and the last coefficient of the first row corresponds
to the energy of the finest image structures in the horizontal direction. The
first column of the 8 x 8 pixel block contains from top to bottom the energies of the coarsest image structures down to the finest image structures in
the vertical direction. The coefficients of the coarse to fine image structures in the diagonal direction can be found diagonally.
The next step is the quantization (Fig.7.24.). All coefficients are divided
by suitable quantization factors. The MPEG standard defines quantization
tables but these can be exchanged by any encoder which can replace them
with its own tables. These are then made known to the decoder by being
transmitted to it. The quantization usually results in a great number of values which have now become zero. After the quantization, the matrix is also
7.1 Video Compression
129
relatively symmetric to the diagonal axis from top left to bottom right. The
matrix is, therefore, read out in a zig-zag scanning process which then provides a large number of adjacent zeroes. These can then be variable-length
coded in the next step, resulting in a very large data reduction. The quantization is the only ‘adjusting screw’ for controlling the data rate of the
video elementary stream.
8
16
19
22
22
26
26
27
16
16
22
22
26
27
27
29
19
22
26
26
27
29
29
35
22
24
27
27
29
32
34
38
26
27
29
29
32
35
38
46
27
29
34
34
35
40
46
56
29
34
34
37
40
48
56
69
34
37
38
40
48
58
69
83
Q(v,u)
scale_factor = 2 ;
QF(v,u) = F(v,u) / Q(v,u) / scale_factor ;
Fig. 7.24. Quantization after the DCT
With 4:2:0, four 8 x 8 Y pixel blocks and one 8 x 8 CB and 8 x 8 CR
pixel block each are combined to form one macroblock (Fig.7.25.). The
quantization for Y, CB and CR can be changed by means of a special quantizer scale factor from macroblock to macroblock. This factor alters all
quantization factors either of the standard MPEG tables or of the quantization tables provided by the encoder, by a simple multiplication by a certain
factor. The complete quantization table can only be exchanged at sequence
level at certain times, as will be seen later.
This transform coding followed by quantization must be performed for
the Y pixel plane and for the CB and CR planes.
130
7 MPEG-2 Video Coding
Y0
Y1
Cb
Y2
Cr
Y3
Fig. 7.25. Macroblock Structure with 4:2:0
173
6
0
0
-1
0
2
0
-2
0
0
0
0
0
0
-1
0
0
0
0
0
0
0
0
0
0
0
0
0
-1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
QFS(v,u)
Fig. 7.26. Zig-zag scanning
In the case of I frames, all macroblocks are coded in the manner described above. In the case of P and B frames, however, the pixel differences are transform coded from macroblock in one frame to macroblock in
7.1 Video Compression
131
another frame. I.e., first the macroblock of the preceding frame may have
to be shifted to a suitable position with the aid of the motion vector of the
macroblock and then the difference with respect to the macroblock at this
position is calculated. Using the DCT, these 8 x 8 difference values are
then transformed into the frequency domain and then quantized. The same
also applies to the backward prediction of B pictures.
7.1.7 Zig-Zag Scanning with Run-Length Coding of Zero
Sequences
After the zig-zag scanning (Fig.7.26.) of the quantized DCT coefficients, a
large number of adjacent zeroes is obtained. Instead of these many zeroes,
only their number is then simply transmitted by using run-length coding
(RLC) (Fig.7.27.), transmitting, e.g. the information 10 times 0 instead of
0, 0, 0, ...0. This type of redundancy reduction, in conjunction with DCT
and quantization, provides the main gain in the data compression.
173
6
0
0
-1
0
2
0
-2
0
0
0
0
0
0
-1
0
0
0
0
0
0
0
0
0
0
0
0
0
-1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
RLC
173, 6, 2*0, -1, 1*0, 2, 1*0, -2, 6*0, -1, 13*0, -1, 34*0
Fig. 7.27. Run-length coding (RLC)
7.1.8 Huffman Coding
Codes occurring frequently in the RLC-coded data stream are also subjected to Huffman coding (Fig.7.28.), i.e., the code words are suitably recoded, resulting in further redundancy reduction. In this type of coding, the
codes used frequently are recoded into particularly short codes as in Morse
code.
132
7 MPEG-2 Video Coding
7.2 Summary
By using a few methods of redundancy reduction and irrelevance reduction, it has been possible to reduce the data rate of a standard definition
television signal with an initial data rate of 270 Mbit/s in the 4:2:2 format
according to ITU BT.R601 to about 2...6 Mbit/s with an upper limit of 15
Mbit/s. The heart of this compression method can be considered to be a
differential pulse code modulation with motion compensation in combination with DCT transform coding. MPEG-2 signals intended for distribution
to homes have their color resolution reduced both in the horizontal and in
the vertical direction. This is then the 4:2:0 format. For studio-studio contribution, MPEG also provides the 4:2:2 format and the data rate is naturally somewhat higher.
Codes of constant length
173, 6, 2*0, -1, 1*0, 2, 1*0, -2, 6*0, -1, 13*0, -1, 34*0
DC
coefficient
Huffman code table
Codes of variable length
Fig. 7.28. Huffman coding (variable length coding - VLC)
Standard Definition (4:2:0) is called Main Profile@Main Level
(Fig.7.29.) and Standard Definition (4:2:2) is called High Profile@Main
Level. However, the MPEG standard also implements High Definition
Television, both as a 4:2:0 signal (Main Profile@High Level) and as a
4:2:2 signal (High Profile@High Level). At over 800 Mbit/s, the initial
data rate of an HDTV signal is clearly higher than that of an SDTV signal
but the compression processes in HDTV, SDTV and 4:2:2 and 4:2:0 are
the same as described before. The resultant signals only differ in their different quality and, naturally, their data rate.
7.2 Summary
133
Levels
Max. Bit
Max. No. of
Pixels x Lines Rate
Mbit/s
x Fields
1440
x1080
x30
1440
x1152
x25
60
(80)
High1440
MP@
H14L
720
x480
x30
720
x576
x25
15
(20)
Main
352
x240
x30
352
x288
x25
4
Low
SSP
@H14
L
SNRP
@ML
MP@
LL
SNRP
@LL
Main + SNR
Scalability
Main
SNR
scalable
MP@
ML
4:2:0, no
Scalability
4:2:0, no
bidirectional
prediction
Simple
SP@
ML
HP@
HL
HP@
H14L
HP@
ML
Profiles
Coding Tools,
Functionality
MP@
HL
High
High
Total
Functionality
(incl. 4:2:2)
80
(100)
Spatial
scalable
1920
x1152
x25
Main +
resolution
Scalability
1920
x1080
x30
Fig. 7.29. MPEG-2 profiles and levels
control
Y
Cb
Cr
redund.
reduct.
-
DPCM
predict.
V
memory
+
DCT
quant.
inv.
DCT
inv.
quant.
Zig
Zag
RLC
VLC
MUX
data
buffer
video
ES
motion
predict.
Fig. 7.30. MPEG-2 encoder
The quality of a 6 Mbit/s SDTV signal in 4:2:0 format approximately
corresponds to the quality of a conventional analog TV signal. In practice,
however, there are data rates ranging from 2 to 7 Mbit/s which, naturally,
determines the picture quality. Correspondingly high data rates are needed,
especially for sports broadcasts.
134
7 MPEG-2 Video Coding
The data rate of the elementary video stream can be constant or can vary
depending on the current picture content. The data rate is controlled by
changing the quantization factors in dependence on the level of the output
buffer of the MPEG encoder (Fig.7.30.).
The macroblocks of an I, P or B picture can be coded in various ways.
The most numerous variants occur especially in the case of the B picture
where a macroblock can be coded in the following ways:
•
•
•
•
Intraframe coded (completely new)
Forward coded
Forward and backward coded
Skipped (not coded at all)
Frame encoding
DCT1 DCT2
DCT3 DCT4
Field encoding
DCT1 DCT2
DCT3 DCT4
Fig. 7.31. Frame/field coding of macroblocks
The type of coding is decided by the encoder (Fig.7.30.) with reference
to the current picture content and the available channel capacity (data rate).
In contrast to analog television, no fields are transmitted but only
frames. The fields are then recreated at the receiving end by reading out
the frame buffer in a particular way.
7.2 Summary
135
There is, however, a special type of DCT coding which results in better
image quality for the interlaced scanning method. This involves frame and
field coding of macroblocks (Fig.7.31.). In this method, macroblocks are
first recoded line by line before being subjected to DCT coding.
Macroblock
Slice
Video ES:
sequence
GOP
picture
slice
macroblock
block
Block
Frame
Fig. 7.32. Block, macroblock, slice and frame
Quantizer
scale factor
Quantizer matrix
extension
Pointer to
1st frame
of PES
packet
Sequence GOP
Picture
Slice
Sequence Picture header
Slice header
header
GOP header
Packetized Elementary Stream
DTS
PTS
Macroblock
PES
header
Length indicator
PES
header
Transport stream
Payload unit
start indicator = 1
4 byte
184 byte
TS header payload
Fig. 7.33. Structure of MPEG-2 video elementary stream
Macroblock
136
7 MPEG-2 Video Coding
7.3 Structure of the Video Elementary Stream
The smallest unit of the video stream is a block consisting of 8 x 8 pixels.
Each block is subjected to a separate Discrete Cosine Transform (DCT)
during the encoding. In the case of a 4:2:0 profile, four luminance blocks
and one CB block and one CR block in each case together form one macroblock. Each macroblock can exhibit a different amount of quantization,
i.e. be compressed to a greater or lesser extent. To this end, the video encoder can select different scaling factors by which each DCT coefficient is
additionally divided. These quantizer scaling factors are the actual “set
screws” for the data rate of the video PES stream. The quantization table
itself cannot be exchanged from macroblock to macroblock. Each macroblock can be either frame encoded or field encoded. This is decided by the
encoder on the basis of necessity and opportunity. One necessity for field
encoding arises from the existence of motion components between the first
and second field and an opportunity is presented by the available data rate.
Together, a certain number of macroblocks in a row form a slice
(Fig.7.32.). Each slice starts with a header which is used for resynchronization, e.g. in the case of bit errors. At the level of the video stream, error
concealment mainly takes place at slice level, i.e. in the case of bit errors,
the MPEG decoders copy the slice of the preceding frame into the current
frame. The MPEG decoder can resynchronize itself again with the beginning of a new slice. The shorter the slices, the lower the interference
caused by bit errors.
Many slices together will then form a frame (picture). A frame, too,
starts with a header, the picture header. There are different types of frames,
called I (intraframe) frame, P (predicted) frame and B (bidirectionally predicted) frame. Because of the bidirectional differential coding, the order of
the frames does not correspond to the original order and the headers and
especially the PES headers, therefore, carry a time stamp so that the original order can be restored (DTS).
Together, a certain number of frames corresponding to a coding pattern
of the I, P and B frame coding predetermined by the encoder, form a group
of pictures (GOP). Each GOP has a GOP header. In broadcasting, relatively short GOPs are used which, as a rule, have a length of about 12
frames, i.e about half a second. The MPEG decoder can only lock to the
signal and begin to reproduce pictures when it receives the start of a GOP,
i.e. the first I frame. Longer GOPs can be chosen for mass storage devices
such as the DVD since it is easy to position their read head on the first I
frame.
7.4 More Recent Video Compression Methods
137
One or more GOPs produce a sequence, each of which also starts with a
header. At the sequence header level, it is possible to change essential
video parameters such as the quantization table. If an MPEG encoder uses
its own table which differs from the standard, this is where it will be found
or, respectively, where the decoder is informed of this.
The structure of the video PES stream (Fig. 7.33.) described above is
embedded wholly or partially in the video PES packets. The manner of this
embedment and the length of a PES packet are determined by the video
encoder. On mass storage devices such as the DVD, the PES packets are
additionally inserted in so-called packs. PES packets and packs also start
with a header.
Video Production DVCPRO
Home Video MiniDV
Motion JPEG
1985
1988
1991
1993
JPEG
ITU-T H.120
H.261
H.262
DCT
1995
=
ISO/IEC
MPEG-1
Part 2
MPEG-2
Part 2
2002
H.263
H.264
=
MPEG-4
Part 2
=
MPEG-4
Part 10
AVC
1992
1994
1998
2003
ISO/IEC
11172-2
ISO/IEC
13818-2
ISO/IEC
14496-2
ISO/IEC
14496-10
Windows Media 9
Fig. 7.34. History of the development of video coding
7.4 More Recent Video Compression Methods
Time has not stood still. Today, more modern, more advanced compression methods such as MPEG-4 Part 10 Advanced Video Coding (AVC)
(H.264) or Windows Media 9 (= VC-1) are already available. With data
rates which are lower by a factor of 2 to 3, a better image quality than with
MPEG-2 can be achieved in many cases. Although the basic principle of
138
7 MPEG-2 Video Coding
video coding has not changed, the difference lies in the details. Thus, variable transform block sizes are used e.g. in H.264. Fig.7.34. shows the history of the development of video coding. As has already been mentioned
several times, establishment of the JPEG standard was also a milestone of
sorts for motion picture coding. DCT was used for the first time in JPEG
and was only replaced by a similar transform, an integer transform, in
MPEG-4 Part 10 (= H.264). Video coding was developed as part of the
ITU-T H.xxx standards and then incorporated in the series of MPEG video
coding methods as MPEG-1, MPEG-2 and MPEG-4. MPEG-2 Part 2
Video corresponds to H-262, MPEG-4 Part 2 Video to H.263 and MPEG4 Part 10 AVC (Advanced Video Coding), finally, to ITU-T H.264.
7.5 MPEG-4 Advanced Video Coding
Compared with MPEG-2, the much improved MPEG-4 Part 10 AVC
(H.264) video codec enables the data rates to be decreased by 30 to 50%.
This means that an SDTV signal can now be compressed to approx. 1.5 ...
3 Mbit/s compared with a data rate of 2 ... 7 Mbit/s, the original uncompressed data rate having been 270 Mbit/s. Using MPEG-4, an HDTV signal can be shrunk to about 10 Mbit/s from its original 1.5 Gbit/s. MPEG-2
would have required about 20 Mbit/s for this.
MPEG-4 Part 10 Advanced Video Coding (H.264) is distinguished by
the following features:
•
•
•
•
•
•
•
•
•
•
•
Formats 4:2:0, 4:2:2 and 4:4:4 are supported
Up to 16 reference frames maximum
Improved motion compensation (1/4 pixels accuracy)
Switching P (SP) and Switching I (SI) frames
Higher accuracy due to 16 bit implementation
Flexible macroblock structure (16x16, 16x8, 8x16, 8x4, 4x8,
4x4)
52 selectable sets of quantization tables
Integer or Hadamard transform instead of a DCT (block size
4x4 or 2x2 pixels, resp.)
In-loop deblocking filter (eliminates blocking artefacts)
Flexible slice structure (better bit error performance)
Entropy encoding; variable length coding (VLC) and context
adaptive binary arithmetic coding (CABAC)
The details are as follows:
7.5 MPEG-4 Advanced Video Coding
139
In MPEG-2 video coding, a 4:2:0 format macroblock consists of 4 luminance blocks of 8x8 pixels and one CB and CR block each of 8x8 pixels.
MPEG-4 provides much more flexibility in this respect. Here, a macroblock has a size of either 16x16, 16x8, 8x16, 8x4, 4x8 or 4x4 pixels in the
luminance layer. The block itself comprises either 4x4 or 2x2 pixels
whereas it was always fixed at 8x8 pixels in MPEG-2 and MPEG-1.
The accuracy of the motion compensation is now 1/4 pixel instead of
1/2 pixel in MPEG-2. In the MPEG-2 interframe coding it was only possible to use one reference in each direction. In MPEG-4, it is possible to
form several reference frames which enables the data rate to be reduced
considerably.
In MPEG-2, a slice was always a multiple of macroblocks in the horizontal direction whereas MPEG-4 provides for a flexible macroblock allocation in a slice.
But it is mainly in the field of transform coding that MPEG-4 shows
great changes.
In principle, MPEG-2 transform coding by means of the DCT is actually
performed by a matrix multiplication in the encoder which is then inverted
in the decoder. For this purpose, a lookup table is stored in the hardware.
The formula for the two-dimensional DCT is:
F(u, v) =
N −1
2
C(u)C(v) ¦
N
x= 0
N−1
¦ f (x,y)cos
y =0
1
C(u), C(v) = ® 2
¯ 1
(2x + 1)uπ
(2y + 1)vπ
cos
2N
2N
for u,v = 0
otherwise
Fig. 7.35. Definition of Discrete Cosine Transform (DCT)
It can be split into matrix multiplications based on a matrix of cosine
values:
140
7 MPEG-2 Video Coding
Mab[]
b
a
M ab [ ] = cos(
( 2a + 1)bπ
);
16
cos(0)
=1
cos(0)
=1
cos(0)
=1
cos(0)
=1
cos(0)
=1
cos(0)
=1
cos(0)
=1
cos(0)
=1
cos( /16)
=0.9808
cos(3 /16)
=0.8315
cos(5 /16)
=0.5556
cos(7 /16)
=0.1951
cos(9 /16)
=-0.1951
cos(11 /16)
=-0.5556
cos(13 /16)
=-0.8315
cos(15 /16)
=-0.9808
cos( /8)
=0.9239
cos(3 /8)
=0.3827
cos(5 /8)
=-0.3827
cos(7 /8)
=-0.9239
cos(9 /8)
=-0.9239
cos(11 /8)
=-0.3827
cos(13 /8)
=0.3827
cos(15 /8)
=0.9239
cos(3 /16)
=0.8315
cos(9 /16)
=-0.1950
cos(15 /16)
=-0.9808
cos(21 /16)
=-0.5556
cos(27 /16)
=0.5556
cos(33 /16)
=0.9808
cos(39 /16)
=0.1951
cos(45 /16)
=-0.8315
cos( /4)
=0.7071
cos(3 /4)
=-0.7071
cos(5 /4)
=-0.7071
cos(7 /4)
=0.7071
cos(9 /4)
=0.7071
cos(11 /4)
=-0.7071
cos(13 /4)
=-0.7071
cos(15 /4)
=0.7071
cos(5 /16)
=0.5556
cos(15 /16)
=-0.9807
cos(25 /16)
=0.1951
cos(35 /16)
=0.8315
cos(45 /16)
=-0.8315
cos(55 /16)
=-0.1951
cos(65 /16)
=0.9808
cos(75 /16)
=-0.5556
cos(3 /8)
=0.3827
cos(9 /8)
=-0.9239
cos(15 /8)
=0.9239
cos(21 /8)
-0.3827
cos(27 /8)
=-0.3827
cos(33 /8)
=0.9239
cos(39 /8)
=-0.9239
cos(45 /8)
=0.3827
cos(7 /16)
=0.1951
cos(21 /16)
=-0.5556
cos(35 /16)
=0.8315
cos(49 /16)
=-0.9808
cos(63 /16)
=0.9808
cos(77 /16)
=-0.8315
cos(91 /16)
=0.5556
cos(105 /16)
=-0.1951
Fig. 7.36. Cosine matrix lookup table
The Discrete Cosine Transform can be represented and executed as a
matrix multiplication in both directions:
F[] = C • f[] • Mab[] • Mab[]T;
where Mab[]T is the transposed matrix of Mab[], i.e. columns and rows
have been swapped. This makes it possible to simultaneously perform both
a horizontal and a vertical transformation, i.e. two-dimensional transformation. Linking it to matrix C makes the matrix Mab into a so-called orthonormal matrix which is of great practical significance for implementing
the transformation process. An orthogonal matrix is a matrix in which the
inverted matrix corresponds to the transposed matrix. The following thus
applies in the case of an orthogonal matrix:
MT = M-1;
An orthogonal matrix has the additional property that the vectors of the
matrix all have the same length. The matrix of cosine values becomes an
orthonormal matrix if the first row is multiplied by 1/¥2 which is achieved
by multiplying it by matrix C. Reversing the transformation process requires an inverted matrix.
7.5 MPEG-4 Advanced Video Coding
141
Naturally, inverting the multiplication
M1 = M 2 • M3
is not
M2 = M1/M3-1
but is defined by
M2 = M1 • M3-1;
i.e. by the multiplication by the transposed matrix.
In principle, a matrix multiplication is defined as follows:
ª n
A ⋅ B = « ¦ a ij ⋅ b jk
¬ j =1
º
»;
¼
ªa11 , a12 º ªb11 , b12 º ªa11b11 + a12b21 , a11b21 + a12b22 º
«a , a » ⋅ «b , b » = «a b + a b , a b + a b »;
22 22 ¼
¬ 21 22 ¼ ¬ 21 22 ¼ ¬ 21 11 22 21 21 12
Fig. 7.37. Definition of a matrix multiplication
Apart from the Discrete Cosine Transform (DCT), other transformation
processes are also conceivable for compressing frames and can be represented as matrix multiplications, these being the
•
•
•
•
•
•
Karhunen Loeve Transform (1948/1960)
Haar's Transform (1910)
Walsh-Hadamard Transform (1923)
Slant Transform (Enomoto, Shibata, 1971)
Discrete Cosine Transform (DCT, Ahmet, Natarajan, Rao,
1974)
Short Wavelet Transform
A great advantage of the DCT is the great energy concentration (Fig.
7.38.) to a very few values in the spectral domain, and the avoidance of
Gibbs' phenomenon which would lead to overshoots in the inverse trans-
142
7 MPEG-2 Video Coding
formation and thus to clearly visible blocking. Gibbs' phenomenon (Fig.
7.39.), known from DCT, is based on the sinusoidal component of this
transformation process.
DCT performs energy
concentration; information
can now be stored in some
values; many others become
zeroes
Fig. 7.38. Energy concentration of the DCT
Gibbs‘ phenomenon
using Fourier
synthesis for
a rectangular
signal
Reason: sinusoidal component
of the Fourier Transform;
DCT does not show this effect
Fig. 7.39. Gibbs’ phenomenon
Since the cosine matrix of the DCT has now been converted into 1/¥2
by the conversion of the first row which consisted of all ones, and has thus
7.5 MPEG-4 Advanced Video Coding
143
become orthonormal, implementation of the transform and its inverse is
quite simple.
The transform and its inverse can now be represented as follows:
F = f • Mab • MabT;
f = F • MabT • Mab;
During the quantization, the results of the transform and its inverse were
additionally influenced by a scalar multiplication
F = f • Mab • MabT • Q;
f = F • MabT • Mab • Q`;
If only ones are entered in Q and Q', nothing changes. However, the
quantization of the DCT coefficients is reduced towards higher frequencies
via Q.
In various transformation methods, only other matrices Mab are used, in
principle, i.e. "basic functions" from which it is attempted to represent the
original functions are others. In the case of the DCT, these are cosine patterns.
In MPEG-4, these basic patterns, or the coefficients of the matrix Mab,
respectively, are replaced by others. In the case of MPEG-4, this is called
an integer matrix multiplication or also Hadamard transformation. The
transformation matrices used in MPEG-4 AVC are the following:
1 1 1 1
2 1 -1 -2
T= 1 -1 -1 1
1 -2 2 -1
1 1 1 1
1 1 -1 -1
H=
1 -1 -1 1
1 -1 1 -1
C=
T = integer transform for luma and chroma samples
H = Hadamard transform for luma DC coefficients
C = Hadamard transform for chroma DC coeffients
Fig. 7.40. Transformation matrices in MPEG-4 AVC
1 1
1 -1
144
7 MPEG-2 Video Coding
The matrices used in MPEG-4 AVC have a size of only 4x4 or 2x2 pixels, respectively. In the case of luminance, the transformation is performed
in two steps. In the first step, the original 4x4 pixel blocks are transformed
into the spectral domain by means of the matrix T. Following this, the
DCT coefficients of 16 blocks are again transformed by means of the Hadamard matrix H so that they can be compressed further (Fig. 7.41.)
Example:
16x16 luminance
macro block
DC coefficients
Hadamard
Transform
Fig. 7.41. Hadamard transform of the DC coefficients in MPEG-4 AVC
In MPEG-2, it is either the matrices specified in the Standard which are
used, or they are specified by the encoder and modified and in each case
transmitted to the receiver in the sequence header at the beginning of a sequence. In addition, in MPEG-2, each coefficient is divided by the quantizer scale factor which ultimately determines the actual data rate. MPEG-4
uses a set of 52 quantization matrices.
MPEG-4 also uses a deblocking filter (Fig. 7.42.) which is intended to
additionally suppress the visibility of blocking artefacts. This is also aided
by the smaller block size and the variable macroblock and slice size.
Reference
frames buffer
Uncompressed
video
De-blocking
filter
Motion
estimation
Inverse
transform
Transform
Fig. 7.42. Deblocking filter in MPEG-4
Dequantization
Quantization
Entropy
coding
Compressed
video
7.5 MPEG-4 Advanced Video Coding
145
Like MPEG-2, MPEG-4 also has profiles and levels. SDTV (standard
definition TV) largely corresponds to Main Profile @ Level 3 (MP@L3).
HDTV (high definition TV) is then Main Profile @ Level 4 (MP@L3).
Table 7.1. MPEG-4 AVC profiles
Coding tools
Baseline profile
I, P slices
CAVLC
Error resilience
SP and SI slices
B slices
Interlaced coding
CABAC
x
x
x
Extended
profile
x
x
x
x
x
x
Main profile
x
x
x
x
x
Table 7.2. MPEG-4 levels
Level number Typical picture size
Max. frame
rate for typ.
picture size
Max. compressed
bit rate
1
1.1
15
10
30
15
30
30
30/25
15
30/25
30
60
60p/30i
64 kbit/s
192 kbit/s
384 kbit/s
768 kbit/s
2 Mbit/s
4 Mbit/s
4 Mbit/s
10 Mbit/s
14 Mbit/s
20 Mbit/s
20 Mbit/s
Max. number
of reference
frames of typ.
picture size
4
3
9
6
6
6
6
5
5
5
4
4
60p/30i
50 Mbit/s
4
60p
72
120
30
50 Mbit/s
135 Mbit/s
240 Mbit/s
4
5
5
1.2
1.3
2
2.1
2.2
3
3.1
3.2
4
4.1
4.2
5
5.1
QCIF
320 x 240
QCIF
CIF
CIF
CIF
HHR
SD
SD
1280 x 720p
1280 x 720p
HD 720p,
1080i
HD 720p,
1080i
1920 x 1080p
2k x 1k
2k x 1k
4k x 2k
MPEG-4 Part 10 AVC allows an image compression which is more effective by at least 30% up to 50%, with better image quality. The SDTV
data rate after compression is now less than 3 Mbit/s and the HDTV data
146
7 MPEG-2 Video Coding
rate is less than 10 Mbit/s. MPEG-4 AVC also makes it possible to use
clearly less than 1 Mbit/s for mobile TV with SDTV quality.
MPEG-4 AVC is used today for HDTV in DVB-S2 and mobile TV as
part of DVB-H and T-DMB. MPEG-4 AVC can be incorporated without
problems in the MPEG-2 transport stream. On the contrary, there have
been no attempts at changing anything in the transport layer. The lip synch
mechanisms are the same, too, and have their origin in the MPEG-1 PES
layer.
Bibliography: [ISO13818-2], [TEICHNER], [GRUNWALD], [NELSON],
[MAEUSL4], [REIMERS], [H.264], [ISO14496-10]
8 Compression of Audio Signals to MPEG and
Dolby Digital
8.1 Digital Audio Source Signal
The human ear has a dynamic range of about 140 dB and a hearing bandwidth of up to 20 kHz. High-quality audio signals must, therefore, match
these characteristics. Before the analog audio signals are sampled and digitized, they have to be band-limited by means of a low-pass filter. Then
analog-to-digital conversion is performed at a sampling rate of 32 kHz,
44.1 kHz or 48 kHz (and now also at 96 kHz), and with a resolution of at
least 16 bits. The 44.1 kHz sampling rate corresponds to that of audio CDs,
48/96 kHz are studio quality. While the 32 kHz sampling frequency is still
provided for in the MPEG standard, it is in fact obsolete. A sampling rate
of 48 kHz at 16 bit resolution yields a data rate of 786 kbit/s per channel,
which means approx. 1.5 Mbit/s for a stereo signal (Fig. 8.1.).
16 bit
Right
A
15...20 kHz
Bandwidth
Left
...768 kbit/s
32/44.1/48 kHz
Audio sampling
frequency
16 bit
A
15...20 kHz
Bandwidth
D
D
~1.5 Mbit/s
Compression
...768 kbit/s
100...400 kbit/s
32/44.1/48 kHz
Audio sampling
frequency
Fig. 8.1. Digital audio source signal
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_8, © Springer-Verlag Berlin Heidelberg 2010
148
8 Compression of Audio Signals to MPEG and Dolby Digital
The objective of audio compression is to reduce the 1.5 Mbit/s data rate
to between about 100 kbit/s and 400 kbit/s. MP3 audio files, which are
very widely used today, often have a data rate as low as 32 kbit/s. Similarly as with video compression, this is achieved by way of redundancy reduction and irrelevance reduction. In redundancy reduction, superfluous
information is simply omitted; there is no loss of information. By contrast,
in irrelevance reduction information is eliminated that cannot be perceived
at the receiving end, in this case the human ear. All audio compression
methods are based on a psychoacoustic model, i.e. they make use of the
"imperfection" of the human ear to remove irrelevant information from the
audio signal. The human ear is not capable of perceiving sound events
close to strong sound pulses in frequency or in time. This means that, to
the ear, certain sound events will mask other sound events of lower amplitude.
8.2 History of Audio Coding
In the year 1988, the MASCAM method was developed at the Institut für
Rundfunktechnik (IRT) in Munich in preparation for the digital audio
broadcasting (DAB) system. From MASCAM, the MUSICAM (masking
pattern universal subband integrated coding and multiplexing) method was
developed in 1989 in cooperation with CCETT, Philips and Matsushita.
MUSICAM-coded audio signals are used in DAB. MASCAM and
MUSICAM are both based on subband coding. The audio signal is split
into a large number of subbands, each of which is subjected to irrelevance
reduction to a greater or lesser degree.
At the same time as the subband coding method was developed, the
Fraunhofer Gesellschaft together with Thomson devised the ASPEC
(Adaptive Spectral Perceptual Entropy Coding) method, which is based on
transform coding. The audio signal is transformed from the time to the frequency domain using DCT (Discrete Cosine Transform), and then irrelevant signal components are removed.
Both the subband-coding MUSICAM and the transform-coding ASPEC
method were included in the MPEG-1 audio compression method, which
was established in 1991 (ISO/IEC 11172-3 standard). MPEG-1 audio
comprises three possible layers: II essentially use MUSICAM coding, and
layer III principally uses ASPEC coding. MP3 audio files are coded to
MPEG-1 layer III. MP3 is often mistaken for MPEG-3. MPEG-3 was
originally aimed at implementing HDTV (high definition television), but
HDTV was already integrated in the MPEG-2 standard, so MPEG-3 was
8.2 History of Audio Coding
149
skipped and abandoned altogether. Therefore the MPEG-3 standard does
not exist.
In MPEG-2 audio, the three layers of MPEG-1 audio were taken over,
and layer II was extended to form layer II MC (multichannel). The
ISO/IEC 13818-3 MPEG-2 audio standard was adopted in 1994.
MASCAM
IRT Munich, 1988
Subband
coding
ASPEC
Fraunhofer Gesellschaft,
Thomson
MUSICAM,
IRT, CCETT, Philips,
Matsushita,
1989
Data rates layer I, II, III:
I: 32...384kbit/s
II: 32...448kbit/s
III: 32...192kbit/s
Transform
coding (DCT)
ISO/IEC 11172-3 MPEG1 Audio, 1990/91
Layer I, low complex encoder, low compression
Layer II, medium complex encoder
Layer III high complex encoder, high compression,
subband & transform coding (...mp3)
ISO/IEC 13818-3 MPEG2 Audio, 1994
Layer I, II, III (same as MPEG1)
Layer II MC (Multichannel Audio up to 5.1)
Fig. 8.2. Development of MPEG audio [DAMBACHER]
Simultaneously with MPEG audio, the Dolby digital audio standard
(also known as AC-3 audio) was developed by Dolby Labs in the USA.
This standard was laid down in 1990 and first presented to the public in the
movie "Star Trek VI" shown in December 1991. Nowadays, many movies
employ the Dolby digital technique. In the USA, digital terrestrial TV
broadcasts to ATSC use AC-3 audio coding exclusively. Some other countries too (e.g. Australia) will introduce AC-3 audio in addition to MPEG
audio. The use of both AC-3 audio and MPEG audio is meaningful, if only
because of the fact that this does away with the recoding of movies. As
from the point of quality, there is practically no difference between MPEG
audio and Dolby digital. Modern MPEG decoder chips, therefore, support
both methods. DVD video discs too may use Dolby digital AC-3 audio in
addition to PCM audio and MPEG audio. Below is a short overview of the
development of Dolby digital:
150
8 Compression of Audio Signals to MPEG and Dolby Digital
•
•
•
1990 Dolby digital AC-3 audio
1991 First AC-3 audio coded movie show
Dec. 1991 "Star Trek VI" coded in AC-3 audio
Today:
•
•
AC-3 audio is used as standard in many movies, in ATSC and,
in addition to MPEG audio, in MPEG-2 transport streams all
over the world, and on DVDs.
Dolby AC-3 audio transform coding based on Modified Discrete Cosine Transform (MDCT); 5.1 audio channels (left, center, right, left surround, right surround, subwoofer), 128 kbit/s
per channel.
MPEG, too, has come up with new audio coding methods:
•
•
MPEG-2 AAC ISO/IEC 13818-7
AAC = Advanced Audio Coding
MPEG-4 ISO/IEC 14496-3:
AAC and AAC Plus
Ossicles
(malleus, incus, Inner
ear
stapes)
Semicircular
canals (organ of
balance)
Cochlea
(organ of Corti)
Outer
ear
Auditory
nerves
Eardrum
Middle
ear
Fig. 8.3. Anatomy of the human ear
Eustachian
tube
8.3 Psychoacoustic Model of the Human Ear
151
8.3 Psychoacoustic Model of the Human Ear
In the following section, the process of audio compression will be discussed. Redundancy reduction (lossless) and irrelevance reduction (lossy)
lower the data rate of the original audio signal by about 90 %. Irrelevance
reduction relies on the psychoacoustic model of the human ear, which essentially goes back to Professor Zwicker, former holder of a professorship
for electroacoustics at the Technical University of Munich. This type of
reduction is based on what is referred to as perceptual coding. This means
that audio components which are not perceived by the human ear are not
transmitted.
Let us first have a look at the anatomy of the human ear (Fig. 8.3., 8.4.).
The ear consists of three main parts: the outer ear, the middle ear, and the
inner ear. The outer ear performs the functions of impedance matching,
sound transmission over air, and acts as a filter with a slight resonance
step-up in the region of 3 kHz. It is in the same region, i.e. from 3 kHz to
4 kHz, that the human ear exhibits its maximum sensitivity. The eardrum
or tympanic membrane converts sound waves to mechanical vibrations,
which are transmitted via the malleus, incus and stapes to a membranous
window leading to the sensory inner ear. The air pressure must be the
same, ahead of and behind the eardrum. This is ensured by a tube connecting the region behind the eardrum with the pharynx; the tube is called the
Eustachian tube. Everyone knows the problem of pressure building up in
the ear when climbing large heights. By swallowing, the mucous membrane in the Eustachian tube provides for pressure compensation.
In the inner ear we find the organ of balance, which is made up of several liquid-filled arches, and the cochlea. The cochlea is the actual hearing
organ (organ of Corti) by which sound is directly perceived. If the cochlea
were to be uncoiled, the sensors for the high frequencies would be found at
its entrance, then the sensors for the medium frequencies, and at the end of
the cochlea would be the sensors for the low frequencies.
The cochlea consists of a spiral canal in which lies a smaller membranous spiral passage that becomes wider from the front to the rear. On the
inner membrane rest the frequency-selective sound-collecting sensors from
which the auditory nerves extend to the brain. The auditory nerves transport electrical signals with an amplitude of approx. 100 mVpp. The repetition rate of the electrical pulses is in the order of 1 kHz. The information
contained in this rate is the volume of a tone at a given frequency. The
louder the tone, the higher the repetition rate. Each frequency sensor communicates with the brain via a separate neural line. The frequency selectiv-
152
8 Compression of Audio Signals to MPEG and Dolby Digital
ity of the sensors is highest at low frequencies and decreases towards
higher frequencies.
Ossicles (malleus, incus, stapes)
Eardrum
Inner ear
Membrane
Receptors for
low frequencies
Outer
ear
Middle
ear
Receptors
for high frequencies
Auditory
nerves
Eustachian tube
Outer ear
= mechanical impedance converter
high ........middle...............low frequencies
Filter
Filter characteristics
of outer ear and
ear tube and eardrum
(e.g. resonance at ~3 kHz)
Frequency receptors
inside cochlea
Auditory
nerve signals
Fig. 8.4. Technical model of the human ear
In the following section, we want to investigate those characteristics of
the human ear that are of interest for audio coding. To begin with, the sensitivity of the ear is to a great extent dependent on frequency. Sound signals below 20 Hz and above 20 kHz are practically not audible. The
maximum sensitivity of the ear is in the range around 3 kHz to 4 kHz; outside this range the sensitivity decreases towards higher or lower frequencies. Sounds with a level below a certain threshold (referred to as threshold
of audibility) are not perceived by the human ear. The threshold of audibil-
8.3 Psychoacoustic Model of the Human Ear
153
ity is frequency-dependent. Any components of audio signals whose level
is below the audibility threshold need not be transmitted; they are irrelevant for the human ear. Fig. 8.5. illustrates the general relationship of audibility threshold versus frequency.
The next characteristic of the human ear that is of significance for audio
coding is a characteristic known as masking. For example, a sinusoidal
carrier at 1 kHz with constant amplitude is applied to the ear of a test person, and the region around 1 kHz is investigated by applying other sinusoidal carriers, the frequency and amplitude of which is varied. It will be
found that the other test signals are not audible below a certain frequencydependent level threshold around 1 kHz. This is known as the masking
threshold (Fig. 8.6.). The shape of the masking threshold depends on the
frequency of the masking signal. The higher the frequency of the masking
signal, the wider the masked range.
L [dB]
60
40
20
0
2
4
6
8
10
12
14 f [kHz]
Fig. 8.5. Threshold of audibility
This characteristic of the ear is known as masking in the frequency domain (Fig. 8.6.). The relevant factor for audio coding is the fact that audio
components below a defined masking threshold need not be transmitted.
However, masking not only occurs in the frequency domain but also in
the time domain (Fig. 8.7.). A strong pulse in the time domain masks
sound signals before and after the pulse, provided the levels of these signals are below a certain threshold. This effect, and in particular premasking, is difficult to imagine but very well explicable. It is due to the finite
154
8 Compression of Audio Signals to MPEG and Dolby Digital
time resolution of the human ear in conjunction with the way signals are
transported to the brain via the auditory nerves.
Masking tone (1kHz)
L [dB]
60
40
Masking threshold
20
0
2
4
6
8
10
12
14 f [kHz]
0
2
4
6
8
10
12
14 f [kHz]
L [dB]
60
40
20
Fig. 8.6. Masking in the frequency domain
L[dB] Premasking
50
40
30
20
10
0
Postmasking
Masking
tone
100
200
300
Fig. 8.7. Masking in the time domain
400 t [ms]
8.4 Basic Principles of Audio Coding
155
The audio compression methods known so far use masking only in the
frequency domain, the techniques employed being very similar in all cases.
Full AD range
sinusoidal signal
N bit resolution
A
LP
D
Quantization noise: S/N[dB] = 6 * N
Fig. 8.8. Quantization noise
Frequency
subbands
uncompressed
audio in
Filter
process
Time: fine
frequency: coarse
Irrelevance
reduction
Subbandquantizer
Redundancy
reduction
Data
coding
Compressed
audio out
Spectrum
analysis
Time: coarse
frequency: fine
Psychoacoustic
model
Fig. 8.9. Principle of audio coding based on perceptual coding
8.4 Basic Principles of Audio Coding
Prior to discussing the principle of irrelevance reduction for audio signals,
quantization noise will be examined briefly. If an analog-to-digital converter is driven to full modulation with a sinusoidal signal, an S/N ratio of
approx. 6 • N dB (rule of thumb) is obtained for a resolution of N bits due
156
8 Compression of Audio Signals to MPEG and Dolby Digital
to quantization noise (Fig. 8.8.). This means that approx. 48 dB are obtained for 8 bit resolution and 96 dB for 16 bit resolution. Audio signals
are usually sampled with 16 bits or more. 16 bit resolution, however, still
does not match the dynamic range of the human ear, which is about
140 dB.
Let us now discuss the basic principle of audio coding (Fig. 8.9.). The
digital audio source signal is split into two branches in the coder, filtered
and taken to a frequency analyzer. The frequency analyzer performs spectrum analysis by means of a Fast Fourier transform (FFT) and determines
the components of the audio signal with low time resolution and high frequency resolution.
Based on the knowledge of the psychoacoustic model (masking effect),
irrelevant frequency components of the current signal can be identified.
Simultaneously with spectrum analysis, the audio signal undergoes filtering by which it is split into many subbands. It may happen that a complete subband is masked by signals of other subbands, i.e. the signal level
in this subband is below the masking threshold. If this is the case, the subband in question need not be transmitted; the information carried in this
band is completely irrelevant to the human ear. The filtering process by
which the audio signal is spread to subbands must use very high time resolution so that no information in the time domain will be lost. In contrast for
the frequency domain, coarse resolution will do. As far as irrelevance reduction is concerned, there is another possibility. Sometimes, signals in a
subband are above the masking threshold, but only by a slight margin. In
such cases, quantization in the subband concerned is reduced to the extent
that quantization noise in this band is below the masking threshold and is
therefore not audible.
Likewise, signals below the threshold of audibility need not be transmitted. Here, too, coarser or finer quantization can be selected depending on
the different audibility thresholds of the subbands so that the resulting
quantization noise always remains below the threshold. Lower bit resolution is possible especially at higher frequencies.
The decision of whether a subband is to be suppressed completely, or if
coarser or finer quantization is to be applied is made in the "psychoacoustic model" block, which is fed with the information from the spectrum
analysis block. Quantization is suppressed or controlled by means of the
subband quantizer. It may be followed by redundancy reduction, which is
effected by a special data coding. After these processes are completed, the
compressed audio signal is available.
Perceptual coding may be implemented in various ways. There is pure
subband coding and transform coding, and there are mixed forms which
are referred to as hybrid coding.
8.5 Subband Coding in Accordance with MPEG Layer I, II
157
8.5 Subband Coding in Accordance with MPEG Layer I, II
First the method of subband coding will be discussed. In accordance with
MPEG layer I and II (Fig. 8.10.), the audio signal is passed through a filter
bank of 32 filters that split the signal into frequency subbands of 750 Hz.
For each subband there is a separate quantizer controlled by an FFT block
and a psychoacoustic model. The quantizer either completely suppresses
the subband in question or reduces the number of quantization steps. In the
case of layer II coding, FFT is carried out every 24 milliseconds on 1024
samples. This means that the information fed to the psychoacoustic model
changes every 24 milliseconds. During the 24 ms intervals, the subbands
are subjected to irrelevance reduction in accordance with the information
received from the psychoaoustic model block. In other words, the signal is
treated as if its composition had not altered for 24 ms.
Audio in
BP
Q
BP
Q
compressed
audio out
Frequency
subbands
BP
Bandpass
filter
512 point FFT
@MPEG Layer I,
1024 point FFT
@ Layer II;
every 24ms
FFT
Q
Quantizer
Psycho
acoustic
model
Example:
MPEG Layer I, II
Fig. 8.10. Subband coding using 32 bandpass filters in MPEG-1 and MPEG-2
layer I, II
Because of the different audibility thresholds, bit allocation and thus
quantization is different for the different subbands. Quantization must be
finest at low frequencies; it may be reduced towards higher frequencies.
Fig. 8.11. illustrates the principle of irrelevance reduction in audio
transmission by means of two examples. In one subband, there is a signal
at about 5 kHz with a level above the masking threshold. In the case of this
158
8 Compression of Audio Signals to MPEG and Dolby Digital
subband, only the number of quantization steps can be reduced. In another
subband, we find a signal at about 10 kHz with a level below the masking
threshold. This means that this subband is fully masked by signals of
neighbouring subbands and can therefore be suppressed completely.
Signal level in subband below
masking threshold:
subband completely suppressed
MPEG layer I, II
32 subbands, each 750 Hz wide
L [dB]
60
40
20
0
2
4
6
Spektrum calculated
by means of FFT:
thresholds calculated after
FFT: quantizer controlled by
psychoacoustic model
8
10
12
14 16
18
20 22
24 f [kHz]
Signal level in subband above masking threshold:
quantization noise adjusted to below threshold
Fig. 8.11. Irrelevance reduction utilizing masking effects
Subband filter &
quantizer 0
12
12
12
Samples Samples Samples
Subband filter &
quantizer 1
12
12
12
Samples Samples samples
Subband filter &
quantizer 2
12
12
12
Samples Samples Samples
Subband filter &
quantizer 31
12
12
12
Samples Samples Samples
Layer I
frame
Fig. 8.12. MPEG-2 layer I, II data structure
Layer II
frame
8.6 Transform Coding for MPEG Layer III and Dolby Digital
159
In irrelevance reduction, subbands are also evaluated as to whether they
contain harmonics of signals belonging to a lower subband, i.e. whether
the masked signals are tonal (harmonic) or non-tonal components. Only
non-tonal, masked signals may be completely suppressed.
In MPEG coding, a certain number of samples are always combined into
frames. A layer I frame is formed with 12 samples for each subband. A
layer II frame is formed with 3 x 12 samples for each subband (Fig. 8.12.).
For each 12-sample block, the highest sample is determined. This sample is used as a scaling factor which is applied to all 12 samples of the
block to provide for redundancy reduction (Fig. 8.13.).
Highest value is used
for scale factor determination
for a block of samples
Block of
samples
Fig. 8.13. Redundancy reduction to MPEG-2 layer I, II
8.6 Transform Coding for MPEG Layer III and Dolby
Digital
Transform coding, in contrast to subband coding, uses no filter bank for
subband filtering; the splitting of audio information in the frequency domain is effected by a variation of the Discrete Fourier Transform. Using a
Discrete Cosine Transform (DFT) or Modified Discrete Cosine Transform
(MDFT), the audio signal is processed to give 256 or 512 spectral power
values. At the same time, in the same way as with subband coding, A Fast
Fourier Transform (FFT) is carried out with relatively high resolution in
the frequency domain. Controlled by the psychoacoustic model created
160
8 Compression of Audio Signals to MPEG and Dolby Digital
from the FFT output data, the power values of the audio signal obtained
through MDFT are subjected to coarser or finer quantization or are suppressed completely. The advantage of this method over subband coding is
that it offers higher frequency resolution for the process of irrelevance reduction. This type of coding is used, for example, in Dolby Digital AC-3
Audio (Fig. 8.14.) (AC-3 stands for audio coding 3).
Audio in
(M)DCT
Quantizer
Modified Discrete
Cosine
Transform
Compressed
audio out
Example:
Dolby Digital
AC-3
Psychoacoustic
model
FFT
Fig. 8.14. Transform coding
Audio in
Subband
filter
FFT
(M)DCT
Quantizer
Psychoacoustic
model
Fig. 8.15. Hybrid subband and transform coding
Compressed
audio out
Example:
MPEG layer III
8.7 Multichannel Sound
161
There is also mixed subband coding and transform coding. which is
known as hybrid coding. For example, in MPEG layer III coding, subband
filtering is performed prior to (M)DCT (Fig. 8.15.). This means that first a
coarse splitting into subbands takes place, then (M)DCT is applied to each
subband to obtain a finer resolution. After (M)DCT, data are subjected to
irrelevance reduction controlled by the psychoacoustic model, which in
turn is fed with information from the Fast Fourier Transform. Audio data
coded to MPEG layer III are commonly referred to as MP3 audio files and
are used all over the world today.
8.7 Multichannel Sound
In multichannel audio coding, irrelevances between the channels can be
determined and omitted in transmission. This means that the channels are
investigated for correlated components that do not contribute to the spatial
hearing impression. This procedure is employed, for example, in MPEG
layer II MC and Dolby digital 5.1 surround. In 5.1 audio, the following
channels are transmitted: left, center, right, left surround, right surround
and a low-frequency enhancement (LFE) channel for a subwoofer.
Fig. 8.16. shows the loudspeaker configuration for 5.1 multichannel audio.
Subwoofer
Left
Center
Left surround
Right
Right surround
Fig. 8.16. Multichannel audio
The more detailed structure of these audio coding methods is not relevant in terms of practical applications and will not be further discussed
162
8 Compression of Audio Signals to MPEG and Dolby Digital
here. For more information, consult the relevant literature and the standards.
8.8 New Developments - MPEG-4
Time has not stood still in the field of audio compression, either. MPEG-4
Advanced Audio Coding - MPEG-4 AAC - is a standard which contains a
number of newly developed audio codecs. Fig. 8.17. shows once again the
complete development history of audio compression including MPEG-4
AAC. MPEG-4 AAC = ISO/IEC 14493-3, i.e. MPEG-4 Part 3 includes
both the previous MPEG-2 AAC codec and various new MPEG-4 audio
codecs up to the MPEG-4 HE (High Efficiency) AAC, which is equivalent
to the AAC+ developed by Coding Technologies in Nuremberg. AAC+ allows broadcasting-type audio quality at data rates of 64 kbit/s, i.e. 1/3 of
the data rate in comparison with MPEG-1 Layer II.
1988
1989
DCC „PASC“
MASCAM
IRT
Philips (=layer I)
Subband
coding
Layer
I,II
MUSICAM
ASPEC
Fraunhofer
Gesellschaft
Transform
coding
using
512 MDCT
1992
Layer III
(„MP3“)
1990
Dolby
Digital
Audio
AC-3
MPEG-1
Audio
ISO/IEC
11172-3
1994
Multichannel
extensions,
low
sampling rates
MPEG-2
Audio
ISO/IEC
13818-3
1997
Fraunhofer,
Dolby,
Sony, AT&T
MPEG-2
AAC
ISO/IEC
13818-7
1999
Fraunhofer,
Dolby,
Sony, AT&T
MPEG-4
AAC
ISO/IEC
14496-3
1024 MDCT
DCC=Digital Compact Cassette
HE AAC
AAC=Advanced Audio Coding
IRT=Institut für Rundfunktechnik
2003
MDCT=Modified Discrete Cosine Transform
Coding
MASCAM=Masking Pattern Adapted Subband Coding
Technologies
and Multiplexing
MUSICAM=Masking Pattern Universal Subband Integrated
Coding and Multiplexing
ASPEC=Adaptive Perceptual Spectral Entropy Coding
PASC=Precision Adaptive Sub-band Coding
HE AAC = High Efficiency AAC
Fig. 8.17. History of the development of audio coding
The magic word of the latest audio coding method is "Spectral Band
Replication" (SBR), i.e. the effective transmission and recovery of higherfrequency audio components. This audio compression method is used by
all mobile TV standards. DAB+ and DRM also use the latest MPEG-4
8.8 New Developments - MPEG-4
163
AAC algorithms. Fig. 8.18. once again shows the audio structure of all
MPEG standards. The audio coding method is always described in Part 3
of the respective MPEG standard. DVB uses currently MPEG-1 Layer II
Audio with a data rate of 192 kbit/s in most cases. The MPEG-2 extensions do not provide any advantages in DVB. Multichannel capability is
implemented via Dolby Digital Audio transmitted in parallel. And DVB
does not need the lower sampling rates provided by MPEG-2 Layer II Audio. Audio players mostly supported MPEG-1 Layer III Audio, MPEG-2
Layer III Audio or MPEG-4 AAC. These devices, called MP3 players in
most cases, are known to everyone by now and are also replacing almost
every other audio recording and replaying device.
MPEG-1
Part 3
MPEG-2
Part 3
MPEG-4
Part 3
Layer I
(Philips, DCC, PASC)
Layer II
(DAB, MUSICAM)
Layer III
(ASPEC, Fraunhofer, MP3)
Layer I
(includes MPEG-2 AAC)
AAC LC
AAC LTP
AAC scalable
Twin VQ
CELP
HVXC
TTSI
BSAC
HE AAC = AAC+
Layer II
Layer III
(all layers:
low sampling
rates and
multichannel 3/2
+LFE)
MPEG-2
Part 7
AAC
Fig. 8.18. Audio codec structure in MPEG
Bibliography: [ISO13818-1], [DAMBACHER], [DAVIDSON],
[THIELE], [TODD], [ZWICKER]
9 Teletext, Subtitles and VPS for DVB
In analog television, teletext, subtitles and VPS (Video Program System
for VCR control) have been much used supplementary services for many
years. Apart from being able to create completely new, comparable services in DVB, standards were set up for enabling these familiar services to
be incorporated compatibly in MPEG-2 data streams conforming to DVB.
The approach is for the DVB receiver to insert these services back into the
vertical blanking interval at the composite CVBS video output. This does
not affect any parallel DVB data services such as EPG (Electronic Program Guide) or MHP (Multimedia Home Platform).
40 characters
24
lines
Fig. 9.1. Teletext page
9.1 Teletext and Subtitles
In analog TV, teletext (Fig. 9.1.) is inserted with roll-off filtering as an
NRZ (non-return-to-zero) coded supplementary signal into the vertical
blanking interval. In DVB, by contrast, a teletext elementary stream is
simply multiplexed directly into the MPEG-2 transport stream. The
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_9, © Springer-Verlag Berlin Heidelberg 2010
166
9 Teletext, Subtitles and VPS for DVB
teletext data are processed to give magazines and lines, i.e. the same structure as in British teletext, and combined to form a packetized elementary
stream. A teletext page according to the British or EBU standard is composed of 24 lines of 40 characters each. The data of each line are transmitted in a teletext line in the vertical blanking interval.
16 bit run-in (1010101010101010)
8 bit framing code (0xE4)
8 + 8 bit magazine + row number
40 * 8 bit payload
-------------------------------------360 bit = 45 byte per line
Non-return-to zero code
(NRZ code)
Data rate: 6.9375 Mbit/s
Level: 462 mVPP
Fig. 9.2. Analog TV teletext in the vertical blanking interval
PES header
3 byte start
code prefix
0x00 0x00 0x01
24
Optional
PES
header
Stream ID
„0xBD“
PES paketlength
8
16
Fig. 9.3. PES Packet with teletext
39 Byte
6 byte
header
N * 184 byte
(N * 184) - 6
payload
Data
ID
1 byte
...
Data units
containing
teletext
Bit
PTS,
presentation
time stamp for
program
synchronization
9.1 Teletext and Subtitles
167
The analog TV teletext line shown in Fig. 9.2. begins with the 16-bitlong run-in (1010 sequence) followed by the 1-byte-long framing code
with a value of 0xE4. This marks the beginning of the active teletext. It is
followed by the magazine and line number of 1 byte each. After that, 40
payload characters consisting of 7 bits payload and 1 (even) parity bit are
transmitted. The total amount of data per line is 360 bits (= 45 bytes) and
the data rate is 6.9375 Mbit/s. In DVB teletext (ETS 300472), the teletext
data are inserted into the PES packets after the framing code (Fig. 9.3.).
The 6-byte PES header starts with a 3-byte start code (0x00 0x00 0x01)
which is followed by the stream ID 0xBD which corresponds to a "Private_Stream_1". Next comes a 16-bit (= 2-byte) length indicator which in
the case of teletext is always set so that the total PES length corresponds
to an integral multiple of 184 bytes.
44 byte
TTXT data fields
Structure
according to
EBU teletext
(= 1 TTXT line)
Data unit
ID
Data unit
length
“0x2C“
reserved
8
8
2
Field
parity
Line
offset
1
5
Framing Magazine
code
& packet
„0xE4“
address
8
16
40 byte (characters)
teletext data
320
Bit
“0x02“=EBU teletext without subtitling
“0x03“=EBU teletext with subtitling
Fig. 9.4. Teletext data block in a PES packet
Then comes a 39 byte optional PES header so that the overall PES
header length for teletext is 45 bytes. The actual teletext information is divided into blocks of 44 bytes. The last 43 bytes are identical to the structure of a teletext line of an EBU TTXT after the run-in-code. These bytes
include the magazine and line information as well as the actual 40 bytes of
teletext characters per line. A teletext page consists of 24 lines of 40 characters and the coding is identical to that of EBU or British teletext.
The teletext, processed to form long PES packets, is divided into short
transport stream packets comprising the 184 byte payload and a 4 byte
168
9 Teletext, Subtitles and VPS for DVB
transport stream header, and multiplexed into the transport stream for
transmission the same as video and audio data.
The packet identifiers (PIDs) of transport stream packets containing teletext are included as PIDs for private streams in the program map table
(PMT) of the respective program (Fig. 9.5.).
TTXT
PID
Fig. 9.5. Entry of a teletext service in a Program Map Table (PMT)
TS header
PES header
TTXT0
TTXT1
TTXT2
Fig. 9.6. Transport stream packet with teletext content
These PIDs can then be used for accessing transport stream packets containing teletext. A transport stream packet containing a PES header can be
9.2 Video Program System
169
recognized by its payload start indicator bit being set. The payload unit of
this packet contains the 45 byte PES header and the first teletext packets.
The further teletext packets follow in the next transport stream packets having the same PID. The length of a teletext PES packet is adjusted so that
an integral number of many transport stream packets will yield one
complete PES packet. After a teletext PES packet has been completely
transmitted, it will be retransmitted or a new packet sent if there are any
changes to the teletext. Immediately preceding the teletext data in the PES
packet, the field parity and the line offset indicate the field and line in
which the teletext data are to be inserted back into the composite video signal by the DVB receiver.
Line 16/329
Return to zero code (RZ code)
Data rate: 2.5 Mbit/s
Level: 500 mVPP
15 byte per line
Fig. 9.7. Analog TV data line in the vertical blanking interval
9.2 Video Program System
VPS, the video program system for controlling video recorders, has long
been known and used in public-service TV broadcasting, especially in
Europe. It can be used for controlling recording in video recorders via the
data line, mostly line 16 in the first field. In the data line (Fig. 9.7.), 15
bytes are transmitted in RZ (return-to-zero) coding, including the VPS information. According to ETSI ETS 301775, bytes 3 to 15 of the data line
are simply inserted into the payload part of a PES packet, similarly to
DVB teletext (Fig. 9.8. and 9.9.). The data unit ID is set to 0xC3 in this
170
9 Teletext, Subtitles and VPS for DVB
Data unit
ID
Data unit
length
case, corresponding to the VBI (vertical blanking interval) according to
DVB. As in DVB teletext, this is followed by the Data Unit Length, the
field ID and the line number in the field.
reserved
8
8
2
Field
parity
Line
offset
1
5
13 byte VBI data
(from data line byte 3…15)
104
“0xC3“=VBI data
Fig. 9.8. PES packet with VBI data
PES
data field
PES header
3 byte start
code prefix
0x00 0x00 0x01
24
Optional
PES
header
Stream ID
“0xBD“
PES packet
length
8
16
39 byte
6 byte
header
Bit
Data
ID
1 byte
VBI data
field
PTS,
presentation
time stamp for
program
synchronization
Fig. 9.9. VBI data field
The data line (Fig. 9.7.) contains the following information:
•
•
•
•
•
•
Byte 1:
Byte 2:
Byte 3:
Byte 4:
Byte 5:
Byte 6:
Run-in 10101010
Start code 01011101
Source ID
Serial ASCII text transmission (source)
Monaural/stereo/binaural
Video content ID
Bit
9.3 WSS – Wide Screen Signalling
•
•
•
•
•
•
171
Byte 7: Serial ASCII text transmission
Byte 8: Remote control (routing)
Byte 9: Remote control (routing)
Byte 10: Remote control
Byte 11 to 14: Video Program System
Byte 15: Spare
4 bytes of VPS data (bytes 11 to 14):
•
•
•
•
•
•
Day (5 bits)
Month (4 bits)
Hour (5 bits)
Minute (6 bits)
Country (4 bits)
Program source ID (6 bits)
Fig. 9.10. WSS signal in line 23 of an analog TV signal
9.3 WSS – Wide Screen Signalling
Since PALplus so-called wide screen signalling (WSS) can be provided in
video line 23. WSS informs the monitor or display about the current display format 4:3 or 16:9. For this purpose a digital control signal can be inserted into the first part of line 23 (Fig. 9.10.). A WSS signal can also be
172
9 Teletext, Subtitles and VPS for DVB
tunnelled via DVB in private PES packets. The principle is the same as for
teletext and VPS. For technical reasons, this signal is visible on some TV
screens at the top the left-hand side of the picture (Fig. 9.11.).
Fig. 9.11. Visible WSS signal on a TV screen (top, left-hand side)
Fig. 9.12. Declaration of teletext in a PMT
9.4 Practical examples
173
9.4 Practical examples
In this chapter, the screen shots from a MPEG analyzer show some practical examples of teletext, VBI and WSS tunnelling in a DVB compliant
MPEG-2 transport stream (Fig. 9.12., 9.13., 9.14., 9.15.).
Fig. 9.13. Declaration of VBI and WSS in a PMT [DVM]
Fig. 9.14. Teletext transmission in a private PES packet [DVM]
174
9 Teletext, Subtitles and VPS for DVB
Fig. 9.15. Teletext, VBI and WSS data blocks in a PES packet [DVM]
Bibliography: [ETS 300472], [ETS 301775], [DVM]
10 A Comparison of Digital Video Standards
10.1 MPEG-1 and MPEG-2, VCD and DVD, M-JPEG and
MiniDV/DV
In 1992, MPEG-1 was created as the first standard for encoding moving
pictures accompanied by sound. The aim was to achieve a picture quality
close to that of VHS at CD data rates (< 1.5 Mbit/s). MPEG-1 was provided only for applications on storage media (CD, hard disk) and not for
transmission (broadcasting) and its data structures correspond to this objective. The audio and video coding of MPEG-1 is quite close to that of
MPEG-2 and all the fundamental algorithms and methods are already in
place. There are both I, P and B frames, i.e. forward and backward prediction, and naturally there are the DCT-based irrelevance reduction methods
already found in JPEG. The picture resolution, however, is limited to about
half the VGA resolution (352 x 288). Neither is there any necessity for
field encoding (interlaced scanning method). In MPEG-1, there is only the
so-called Program Stream (PS) which is composed of multiplexed packetized elementary stream (PES) packets of audio and video. The variablelength (64 kbytes max) audio and video PES packets are simply assembled
interleaved in accordance with the present data rate to form a data stream.
This data stream is not processed any further since it is only intended to be
stored on storage media and not used for transmission. A certain number of
audio and video PES packets are combined to form a so-called pack which
consists of a header and the payload just like the PES packets themselves.
A pack is often based on the size of a physical data sector of the storage
medium.
In MPEG-2, the coding methods were developed further in the direction
of higher resolution and better quality. In addition, transmission was also
considered, in addition to the storage of such data. The MPEG-2 transport
stream is the transportation layer, providing much smaller packet structures and more extensive multiplexing mechanisms. In MPEG-1, there is
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_10, © Springer-Verlag Berlin Heidelberg 2010
176
10 A Comparison of Digital Video Standards
only one program (only one movie), whereas MPEG-2 can accommodate a
multiplexed data stream with up to 20 programs and more.
In addition to Standard Definition TV (SDTV), MPEG-2 also supports
High Definition TV (HDTV). MPEG-2 is used throughout the world as
digital baseband signal in broadcasting.
A Video CD (VCD) contains an MPEG-1 coded data signal as a program stream, i.e. there is one program consisting of multiplexed PES
packets. The total data rate is about 1.5 Mbit/s. Many pirate copies of movies are available as Video CD and can be downloaded from the Internet or
bought on the Asian market.
A Super Video CD (SVCD) carries an MPEG-2 data signal coded with
2.4 Mbit/s, also as a program stream with multiplexed PES packets. A Super Video CD approximately corresponds to VHS type quality, sometimes
even better.
On a DVD (Digital Versatile Disk - NOT ‘Digital Video Disk’), the data
material is MPEG-2 coded with data rates of up to 10.5 Mbit/s and exhibits
a much better picture quality than that recorded on VHS tape. A DVD also
carries a multiplexed PES data stream. Subtitles and much else besides are
also possible.
The DVD is intended for a variety of applications including video, audio
and data. In contrast to the CD (approx. 700 Mbytes), the data volume on a
DVD is up to 17 Gbytes and it is possible to have 1, 2 or 4 layers with 4.7
Gbytes each per layer (see table below).
Table 10.1. DVD types
Type
Sides
DVD 5
DVD 9
DVD 10
DVD 18
1
1
2
2
Layers /
side
1
2
1
2
Data
[Gbytes]
4.7
8.5
9.4
17.1
x
CD-ROM
7
13
14
25
Technical data of the Video DVD:
•
•
Storage capacity: 4.7 to 17.1 Gbytes
MPEG-2 Video with variable data rate, 9.8 Mbit/s video max.
Audio:
•
Linear PCM (LPCM) with 48 kHz or 96 kHz sampling frequency at 16, 20 or 24 bits resolution
10.1 MPEG-1 and MPEG-2, VCD and DVD, M-JPEG and MiniDV/DV
•
•
177
MPEG Audio (MUSICAM) mono, stereo, 6-channel sound
(5.1), 8-channel sound (7.1)
Dolby Digital (AC3) mono, stereo, 6-channel sound (5.1)
Table 10.2. Digital video standards
Standard
Video
coding
Resolution
MPEG-1
MPEG-1
352 x 288
192 x 144
384 x 288
MPEG-2
MPEG-2
720 x 576
(SDTV, 25
frames per
second)
different
resolutions
up to HDTV
up to 15
MPEG-4
MPEG-4
Part 2 and
Part 10
(H.264)
MPEG-1
MPEG-2
352 x 288
480 x 576
1.150
2.4
1.4112
2.624
MPEG-2
720 x 576
10.5
MJPEG
variant
MJPEG
variant
720 x 576
up to 9.8,
variable
25
720 x 576
25/50
Video CD
Super
VCD
Video
DVD
MiniDV
DVPro
Video
data rate
[Mbit/s]
0.150 (1.150)
- 3.0
Total data
rate
[Mbit/s]
max.
approx.
3.5
(1.4112)
basically
open,
from the
interfaces
up to
270
approx.
30
approx.
30/55
Apart from MPEG, there are also proprietary methods based on JPEG
all of which have that in common that the video material is only DCT
coded and not interframe coded. Both DV and MiniDV are such methods.
MiniDV has become widely used in the home video camera field and has
revolutionized this field with respect to the picture quality. The data rate is
3.6 Mbyte/s total or 25 Mbit/s video data rate. The picture size is 720 x
576 pixels, the same as in MPEG-2, with 25 frames per second. MiniDV
can be edited at any point since it virtually only consists of frames comparable to I frames. DVCPro is the big brother to MiniDV. DVCPro is a studio standard and supports video data rates of 25 and 50 Mbit/s. The 25
178
10 A Comparison of Digital Video Standards
Mbit/s data rates corresponds to the MiniDV format. DVCPro and MiniDV
are special variants of Motion JPEG. In contrast to MPEG, no quantizer
tables are transmitted, neither are quantizer scale factors varied from macroblock to macroblock. Instead, a set of quantizing tables is provided locally, from which the coder selects the most suitable one from macroblock
to macroblock. MiniDV and DVPro exhibit a very good picture quality at
relatively high data rates and lend themselves easily to postprocessing.
Home editing software for the PC is now available at a cost of around 100
Euros and provides functions available only to professionals a few years
ago. Apart from the actual editing, which is now free of losses and is easy
to handle, the software also allows video material to be coded in MPEG-1,
MPEG-2, VCD, SVCD and video DVD.
Table 10.2. shows the most important technical data of the methods discussed.
STOP
START
Moving picture
PAUSE
Synthetic
person
Text
Fig. 10.1. MPEG-4 example
10.2 MPEG-3, MPEG-4, MPEG-7 and MPEG-21
In the previous chapters, MPEG-2 and MPEG-1 have been discussed in
detail. However, the Moving Picture Experts Group also considered, and is
still working on, other standards such as MPEG-4, MPEG-7 and MPEG21. There was also MPEG-3 but this only had a temporary existence as an
approach to HDTV and has now been completely absorbed into the
MPEG-2 standard. MPEG-4 is a standard for multimedia applications with
interactive components and has been in existence since the end of 1999.
This involves not only video and audio but also applications which can be
composed of a number of different objects. Its structure is object oriented
similar to the programming language C++ and an MPEG-4 application can
thus be composed of, e.g., the following audiovisual objects (Fig. 10.1.):
10.2 MPEG-3, MPEG-4, MPEG-7 and MPEG-21
•
•
•
•
•
•
179
A fixed colored background which may be patterned
An MPEG-4 coded moving picture in a fixed frame
A synthetic figure which moves three-dimensionally in synchronism with the video, e.g. a synthetic person which “mimes”
the sound in gestures (deaf-and-dumb alphabet)
Stop, start, pause, forward and rewind buttons (interactive elements)
Accompanying text
MPEG-coded audio signal
The development of MPEG-4 was continued with respect to the video
and audio coding, adopting and refining the methods known from MPEG-1
and MPEG-2 instead of looking for a wholly new approach. The only new
thing is that MPEG-4 can also cope with synthetic visual and audiovisual
elements such as synthetic sound. MPEG-4 objects can be present as PES
stream both within an MPEG-2 transport stream and as an MPEG-4 file.
MPEG-4 can also be transmitted as a program stream in IP packets.
MPEG-4 applications can be typically employed in
•
•
•
The Internet
Interactive multimedia applications on the PC
New video compression applications with greater compression
factors as required, e.g. for HDTV
MPEG-4 was made into a standard in 1999. At the beginning of the new
millenium, a further new video compression standard H.264 was developed and standardized. Compared with MPEG-2, this method is more effective by a factor of 2 to 3 and thus allows data rates which are lower by a
factor of 2 to 3, often even with improved picture quality. The relevant
standard is ITU-T H.264. H.264 has also been incorporated in the group of
MPEG-4 standards as MPEG-4 Part 10.
The most important standard documents covered by the heading MPEG4 are:
•
•
•
•
MPEG-4 Part 1 – Systems, ISO/IEC 14496-1
MPEG-4 Part 2 – Video Encoding, ISO/IEC 14496-2
MPEG-4 Part 3 – Audio Encoding, ISO/IEC 14996-3
MPEG-4 Part 10 – H.264 Advanced Video Coding. ISO/IEC
14496-10
180
10 A Comparison of Digital Video Standards
Video production DVCPRO
Home video MiniDV
Motion JPEG
1985
1988
1991
1993
JPEG
ITU-T H.120
H.261
H.262
1995
2002
H.263
H.264
DCT
ISO/IEC
MPEG-1
Part 2
MPEG-2
Part 2
MPEG-4
Part 2
MPEG-4
Part 10
AVC
1992
1994
1998
2003
ISO/IEC
11172-2
ISO/IEC
13818-2
ISO/IEC
14496-2
ISO/IEC
14496-10
Windows Media 9
= VC1
Fig. 10.2. History of the development of video encoding
MPEG-4 Part 10 – Advanced Video Coding (AVC) is scheduled for
HDTV applications in Europe as part of the DVB project. Whereas HDTV
requires data rates of about 15 Mbit/s for the video signal with MPEG-2,
these are about 9 Mbit/s or even lower when they are encoded as MPEG-4
AVC signals. In H.264/MPEG-4 Part 10 AVC, the block size is not a constant 8 x 8 pixels but variable within certain limits. Up to 9 motion vectors
are possible and measures for masking blocking have been implemented.
This would be a suitable point at which to pause and to explore some of
the history of the development of video encoding (Fig. 10.2.). A key event
is considered to be the establishment of the JPEG (Joint Photographic Experts Group) Standard in 1985. This was the first time the DCT (Discrete
Cosine Transform) was used for compressing still pictures. Today, JPEG is
a common standard used mainly in digital photography. From JPEG, Motion JPEG applications such as DVCPro for studio applications and
MiniDV for home video applications evolved in one line of development.
The advantage of Motion JPEG lies mainly in the fact that the video material can be edited unrestrictedly at every frame and the picture quality is
extremely good. A further line of development formed through video telephony and video conferencing via the ITU-T standards H.120, H.261 etc.
ITU-T H.261 merged into the MPEG-1 video standard ISO/IEC 11172.2
and H.262 became the MPEG-2 video standard ISO/IEC 13818-2. H-263
formed the basis for MPEG-4 Part 2 video encoding ISO/IEC 14496-2.
10.2 MPEG-3, MPEG-4, MPEG-7 and MPEG-21
181
And finally, H.264 was developed, also known as MPEG-4 Part 10 AVC
(Advanced Video Coding), or as ISO/IEC 14496-10. In parallel to this,
there is also Microsoft Windows Media 9 (now also called VC-1) which
probably arose as a result of Microsoft’s collaboration in MPEG-4 Part 2
and Part 10. Fig. 10.2. shows a rough overview of the history of the development of moving picture encoding.
MPEG-7, in contrast and as a supplement to MPEG-2 and -4, deals exclusively with program-associated data, the so-called meta-data, as a complement to MPEG-2 and MPEG-4. The aim is to transmit background information for a program on air, a type of electronic program guide, with
the aid of XML- and HTML-based data structures together with the program, e.g. in an MPEG-2 transport stream. MPEG-7 has been a standard
since 2001 but has yet to make its debut in practice, at least for the end
user.
MPEG-21 was to be transformed into a full standard by 2003 and was
intended to contain tools and methods to supplement all other MPEG standards (including end-user-to-end-user applications, e.g. via the Internet). It
is not clear what has become of it.
Broadcasting, multimedia and the Internet are converging more and
more. In broadcasting, however, much higher point-to-multipoint data
rates are possible in the downstream than will ever be possible over the
Internet.
Table 10.3. MPEG standards
Standard
MPEG-1
MPEG-2
MPEG-3
MPEG-4
MPEG-7
MPEG-21
Description
Moving pictures and sound,
approx. in VHS quality with
CD data rate (< 1.5 Mbit/s)
Digital television
(SDTV+HDTV)
Existed only temporarily
(no relation to MP3)
Multimedia, interactive
Program-associated
supplementary data
(Meta-data)
Supplementary tools
and methods
Status
Standard since 1992
Standard since 1993
not applicable
Standard since 1999
Standard since 2001
Current
status
???
182
10 A Comparison of Digital Video Standards
10.3 Physical Interfaces for Digital Video Signals
Analog SDTV (Standard Definition Television) signals have a bandwidth
of appr. 4.2 to 6 MHz and are transmitted over 75 Ohm coaxial lines.
These cables, which in most cases are green-jacketed, are fitted with BNC
connectors both in professional applications and in high-end consumer applications. If they are terminated in exactly 75 Ohms, analog video signals
have an amplitude of 1 VPP. The first interfaces for digital TV signals were
designed as parallel interfaces, using the 25-pin Cannon connector known
from the PC printer interface. Because of its noise immunity, transmission
was conducted as low-voltage differential signaling via twisted pair lines.
Today, however, 75 Ohm technology is again being used in most cases.
Digital video signals are transmitted as a serial data signal with a data
rate of 270 Mbit/s via 75 Ohm coax cable fitted with the familiar, rugged
BNC connectors, making no distinction between uncompressed video signals according to the ITU 601 standard and MPEG-2 transport streams.
The distribution paths in the studio are the same, the cables are the same,
amplifiers and cable equalizers are also the same. Engineers often talk of
SDI or of TS-ASI. The physical interface is the same in both cases, only
the content differs. SDI stands for Serial Digital Interface, meaning the serial digital uncompressed video signal of the 601 standard with a data rate
of 270 Mbit/s. TS-ASI stands for Transport Stream Asynchronous Serial
Interface, meaning the MPEG-2 transport stream on a serial interface, the
transport stream having a data rate which is distinctly lower than the data
rate on this serial transmission link. The data rate of the transport stream is
asynchronous to the constant data rate of 270 Mbit/s on the TS-ASI interface. If, e.g., the transport stream has a data rate of 38 Mbit/s, stuffing information is used to fill up the data rate of 270 Mbit/s. The reason for
working with a constant 270 Mbit/s is clear: In the studio, it is desirable to
have uniform distribution paths for 601 signals and the MPEG-2 transport
streams.
10.3.1 Parallel and Serial CCIR 601
Uncompressed SDTV video signals have a data rate of 270 Mbit/s. They
are distributed either as parallel signals via twisted pair lines or serially via
75 Ohm coax cables. The parallel interface is the familiar 25 pin Cannon
socket also known as a PC printer interface. The signals are LVDS (low
voltage differential signaling) signals which means that ECL levels and not
TTL levels are used as voltage levels (800 mVPP). In addition, for each
data bit, the inverted data bit is also transmitted in order to keep the noise
10.3 Physical Interfaces for Digital Video Signals
183
level as low as possible over twisted lines. Table 10.4 shows the pin allocation at the 25 pin parallel interface. The compatible allocation of the parallel MPEG-2 transport stream interface is also entered. In most cases,
however, only the serial ITU 601 interface is used today. It is also called
the Serial Digital Interface (SDI) and uses a 75 Ohm BNC socket with a
voltage level of 800 mVPP. In contrast to the parallel interface, the signals
can be distributed over relatively long distances if cable equalizers are
used.
Table 10.4. Parallel CCIR601 and TS interface
Pin
Signal
Pin
Signal
1
Clock
14
Inverted clock
2
System ground
15
System ground
3
601 data bit 9
(MSB)
TS data bit 7
(MSB)
601 data bit 8
TS data bit 6
601 data bit 7
TS data bit 5
601 data bit 6
TS data bit 4
601 data bit 5
TS data bit 3
601 data bit 4
TS data bit 2
601 data bit 3
TS data bit 1
601 data bit 2
TS data bit 0
601 data bit 1
TS data valid
601 data bit 0
TS packet sync
Case ground
16
601 inverted data bit 9
(MSB)
inverted TS data bit 7
(MSB)
inverted 601 data bit 8
inverted TS data bit 6
inverted 601 data bit 7
inverted TS data bit 5
inverted 601 data bit 6
inverted TS data bit 4
inverted 601 data bit 5
inverted TS data bit 3
inverted 601 data bit 4
inverted TS data bit 2
inverted 601 data bit 3
inverted TS data bit 1
inverted 601 data bit 2
inverted TS data bit 0
inverted 601 daten bit 1
inverted TS data valid
inverted 601 data bit 0
inverted TS packet sync
4
5
6
7
8
9
10
11
12
13
17
18
19
20
21
22
23
24
25
184
10 A Comparison of Digital Video Standards
10.3.2 Synchronous Parallel Transport Stream Interface (TS
Parallel)
The parallel MPEG-2 transport stream interface is designed to be fully
compatible with the parallel ITU 601 interface. The signals are also LVDS
signals, i.e. signals with ECL level which are transmitted with balanced
levels on twisted pairs. The connector is also a 25 pin Cannon connector
with pin allocations which are compatible with the ITU 601 interface. The
pin allocations of the data signal, which is only 8 bits wide in contrast to
the ITU 601 signal, can be found in the table in the previous section.
Fig. 10.3. Parallel TS connector
The data stream transmitted via the transport stream interface (Fig.
10.3., 10.4. and 10.5.) is always synchronous with the MPEG-2 transport
stream to be transmitted, i.e. if the transport stream has a data rate of, e.g.,
38 Mbit/s, the data rate will be 38 Mbit/s here, too. The transport stream
remains unchanged.
Clock
Data [0..7]
187 sync
1
2
186
187 sync
1
DVALID
PSYNC
Fig. 10.4. Transmission Format with 188-Byte Packets [DVG]
However, the transport stream interface can be operated with 188 bytelong packets or 204 or 208 byte-long MPEG-2 transport stream packets.
The 204 or 208 byte-long packets are due to the Reed Solomon error protection of DVB or ATSC signals on the transmission link. At the transport
10.3 Physical Interfaces for Digital Video Signals
185
stream interface, however, any data going beyond 188 bytes are only
dummy bytes and their content can be ignored. Many devices can be configured to these various packet lengths or have the capability of handling
all formats.
Clock
Data [0..7]
16d sync
1
187
1d
16d sync
1
DVALID
PSYNC
Fig. 10.5. Transmission format with 188 byte packets and 16 dummy bytes (=204
Bytes) [DVG]
Fig. 10.6. TS-ASI
10.3.3 Asynchronous Serial Transport Stream Interface (TS
ASI)
The asynchronous serial transport stream interface (Fig. 10.6.) is an interface with a constant data rate of 270 Mbit/s. Data bytes (8 bits each) will
be transmitted via this interface at a maximum rate of 270 Mbit/s, i.e. the
data rate on this interface is not synchronous with the actual MPEG-2
transport stream but always a constant 270 Mbit/s. The advantage of this
is, however, that the same distribution system system can be used as with
the SDI. Each byte is supplemented by 2 additional bits in accordance with
a standardized table. On the one hand, this identifies the data bytes
(dummy bytes) which are irrelevant but necessary for filling up the data
rate of 270 Mbit/s and, on the other hand, it prevents the occurrence of a
DC component in the serial signal.
186
10 A Comparison of Digital Video Standards
The connector uses a BNC socket with an impedance of 75 Ohms. The
level is 800 mVPP (+/-10%).
The TS ASI interface can be operated in two modes: Burst mode, in
which the TS packets remain unchanged in themselves and dummy packets are inserted to attain the data rate of 270 Mbit/s, and single byte mode,
in which dummy bytes are inserted to provide “stuffing” to the output data
rate of 270 Mbit/s.
10bit
K28.5
TSB
K28.5
K28.5
K28.5
TSB
K28.5
188 * 10 bit
n * K28.5
TSP (188 * 8 Bit)
n * K28.5
Special character K28.5
0011111010
1100000101
TSB = Transport stream byte
TSP = Transport stream packet
Fig. 10.7. TS-ASI in Single-Byte Mode (top) and in Burst Mode (bottom)
10.3.4 SMPTE 310 Interface
The SMPTE 310 interface defined for ATSC is a special version of the TS
ASI interface. This interface is fixed at the data rate of precisely 19.39
Mbit/s of the ATSC standard. The connector used is a BNC connector.
10.3.5 DVI Interface
DVI stands for Digital Visual Interface (Fig. 10.8.) and is an interface
which will replace the VGA interface as display interface in the PC domain. It exists as integrated model (DVI-I) and in digital form (DVI-D).
The DVI-I interface additionally contains the analog VGA components,
i.e. using a passive adapter plug it can also be converted into a VGA interface. The DVI-D interface only has the digital monitor signals. The data
rate of the DVI interface is 1.65 Gbit/s. DVI is a possible interface of an
HD Ready monitor or beamer (data projector).
10.3 Physical Interfaces for Digital Video Signals
187
Table 10.5. The DVI interface
Pin
01
02
03
04
05
06
07
08
09
10
11
12
13
14
15
16
17
18
19
20
21
22
23
24
Signal
TMDS data 2TMDS data 2+
Ground for TDMS data 2, 4
TMDS data 4TMDS data 4+
DDC clock
DDC data
Analog V sync
TMDS data 1TMDS data 1+
Ground for TMDS data 1, 3
TMDS data 3TMDS data 3+
+5V
Ground for +5V
Hotplug detect
TMDS data 0TMDS data 0+
Ground for TMDS data 0, 5
TMDS data 5TMDS data 5+
Ground for TMDS clock
TMDS clock+
TMDS clock-
C1
C2
C3
C4
Analog red
Analog green
Analog blue
Analog ground
TDMS - Transmission Minimized Digital Signalling
DDC - Display Data Channel
10.3.6 HDMI Interface
HDMI (High Definition Multimedia Interface) is also an interface (Fig.
10.9.) which can be found on an HD Ready or Full HD monitor. Apart
from the video data, it also contains the audio signals and supports a data
rate of up to 5 Gbit/s. If the data rates are suitable, DVI video can be converted to HDMI video and conversely. This can be done purely passively
by using adapter plugs.
188
10 A Comparison of Digital Video Standards
Fig. 10.8. DVI interface (DVI-D on the left, DVI-I on the right)
Fig. 10.9. HDMI interface
10.3.7 HD-SDI Interface
HD-SDI (High Definition Serial Digital Interface) is the big brother of the
SDI interface. It is used for distributing uncompressed HD video with 10
bits resolution at a data rate of 1.485 Gbit/s or supplying it to an HD
MPEG encoder. This interface can be provided as a BNC interface with an
impedance of 75 ohms and a level of 800 mV or as an optical interface.
10.3.8 Gigabit Ethernet Interface as Transport Stream
Distributor
There are more and more applications in which TS ASI is replaced by a
Gigabit Ethernet interface as MPEG-2 transport stream interface. It is assumed that this interface will replace the TS ASI interface completely in
the medium term. In this arrangement, several transport streams can also
be distributed over one interface. The addressing is then carried out via a
"socket", known from the PC world, which is composed of the port number and the IP address.
Bibliography: [GRUNWALD], [DVG], DVMD], [DVQ], [FISCHER4],
[ITU601], [REIMERS], [TAYLOR], [MPEG4]
11 Measurements on the MPEG-2 Transport
Stream
With the introduction of digital television, neither the hopes of the users
nor the fears of the test equipment makers were confirmed: there is still a
large need for test instruments for digital television, but of a different type.
Where it was mainly video analyzers for evaluating the test lines of an analog baseband signal in analog television, it is mainly MPEG-2 test decoders which are being used in digital TV. Throughout the world, taking
measurements directly at the transport stream has become the most important digital TV test technology with regard to turnover and demand. Thus,
e.g., every MPEG-2 transport stream to be broadcast is analyzed and monitored by means of a MPEG-2 test decoder at almost every transmitter site
of DVB-T networks in some countries.
The input interface of an MPEG-2 test decoder is either a parallel 25-pin
MPEG-2 interface or a serial TS-ASI BNC connector, or both at the same
time.
The MPEG-2 analyzer consists of the essential circuit blocks of
MPEG-2 decoder, MPEG-2 analyzer - usually a signal processor - and a
control computer which acquires all the results, displays them on the display and performs and manages all operating and control operations. A test
decoder is capable of decoding all the video and audio signals contained in
the transport stream and of performing numerous analyses and measurements on the data structure. The MPEG-2 transport stream analysis is a
special type of logic analysis.
The Measurement Group in the DVB Project has defined numerous
measurements on the MPEG-2 transport stream within its Measurement
Guidelines ETR 290. These measurements will be described in more detail
in the following chapters. According to ETR 290, the errors to be detected
by means of these measurements were graded into three levels of priority:
Priority 1, 2, 3.
MPEG-2 transport stream errors:
•
Priority 1 - no decodability
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_11, © Springer-Verlag Berlin Heidelberg 2010
190
11 Measurements on the MPEG-2 Transport Stream
•
•
Priority 2 - partially no decodability
Priority 3 - errors in the supplementary information/SI
If there is a Priority 1 error, there is often no chance to lock to the transport stream or even to decode a program. Priority 2, in contrast, means that
there is partially no possibility of reproducing a program faultlessly. The
presence of a category 3 error, on the other hand, only indicates errors in
the broadcasting of the DVB service information. The effects are then dependent on how the set-top box used reacts.
Apart from the category 3 errors, all measurements can also be applied
with the American ATSC standard where comparable analyses can be made on the PSIP tables.
The following measurements on the MPEG-2 transport stream are defined in the DVB Measurement Guidelines ETR 290:
Table 11.1. MPEG-2 measurement to ETR290 / TR101290 [ETR290]
Measurement
TS_sync_loss
Sync_byte_error
PAT_error
PMT_error
Continuity_count_error
PID_error
Transport_error
CRC_error
PCR_error
PCR_accuracy_error
PTS_error
CAT_error
SI_repetition_error
NIT_error
SDT_error
EIT_error
RST_error
TDT_error
Undefined_PID
Priority
1
1
1
1
1
1
2
2
2
2
2
2
3
3
3
3
3
3
3
11.1 Loss of Synchronisation (TS_sync_loss)
191
11.1 Loss of Synchronisation (TS_sync_loss)
The MPEG-2 transport stream consists of 188-byte-long data packets
composed of 4 bytes of header and 184 bytes of payload. The first byte of
the header is the synchronisation or sync byte which always has the value
0x47 and occurs at constant intervals of 188 bytes. In special cases a spacing of 204 or 208 bytes is also possible, namely when the data frame with
Reed Solomon error protection according to DVB or ATSC is similar. The
additional 16 or 20 bytes are then dummy bytes and can be simply ignored.
At any rate, there is no useful information present since it is not the Reed
Solomon coder and decoder which represent the first or, respectively, last
element of the transmission link but the energy dispersal unit, and thus any
Reed Solomon error protection bytes present would not fit in with the actual transport stream packet. According to DVB, synchronism is achieved
after 5 successive sync bytes have been received at correct intervals and
with the correct content. When 3 successive sync bytes or transport stream
packets have been lost, the MPEG-2 decoder or the corresponding transmission device drops lock again.
188 byte
4 byte
header
Optional
adaptation
field
Header
Sync Transport Payload
unit start
byte error
indicator indicator
8
1
184 byte
payload
1
Transport
priority
1
Transport
PID scrambling
control
13
2
Adaptation Continuity
field
counter
control
2
4
Bit
Fig. 11.1. TS_sync_loss
The state of loss of transport stream synchronization, which may occur
either because of severe interference or simply because of a break in a line,
is called “TS_sync_loss” (Fig. 11.1.).
“TS_sync_loss” occurs when
•
the content of the sync bytes of at least 3 successful transport
stream packets is not equal to 0x47.
192
11 Measurements on the MPEG-2 Transport Stream
The conditions of synchronization (acquisition of lock, loss of lock) can
be adjusted in test decoders.
PID
PAT
PAT
PID=0
PID=0
TableID
ID=0
=0
Table
Program
Program
Association
Association
Table
Table
PID
PMT
PMT
PID=(...PAT)
PID=(...PAT)
TableID
ID=2
=2
Table
Program
Program
Map
Map
Table
Table
PMT
PMT
PID=(...PAT)
PID=(...PAT)
TableID
ID=2
=2
Table
Program
Program
Map
Map
Table
Table
PID
Video ES
Audio ES
PID
Fig. 11.2. PAT and PMT errors
11.2 Errored Sync Bytes (sync_byte_error)
As explained in the previous chapter, the state of synchronism with the
transport stream is considered to be the reception of at least 5 correct sync
bytes. Loss of synchronism occurs after the loss of 3 correctly received
sync bytes. However, incorrect sync bytes may occur occasionally here
and there in the transport stream due to problems in the transmission link.
This state, caused in most cases by too many bit errors, is called
“sync_byte_error” (Fig. 11.2.).
A “sync_byte_error” occurs when
•
the content of a sync byte in the transport stream header is not
equal to 0x47.
11.3 Missing or Errored Program Association Table (PAT)
(PAT_error)
The program structure, i.e. the composition of the MPEG-2 transport
stream is variable or, in other words, open. For this reason, lists for de-
11.4 Missing or Errored Program Map Table (PMT) (PMT_error)
193
scribing the current transport stream composition are transmitted in special
TS packets in the transport stream. The most important one of these is the
Program Association Table (PAT) which is always transmitted in transport
stream packets with PID=0 and Table ID=0. If this table is missing or is errored, identification, and thus decoding, of the programs becomes impossible. In the PAD, the PIDs of all Program Map Tables (PMTs) of all programs are transmitted. The PAT contains pointer information to many
PMTs. A set top box will find all necessary basic information in the PAT.
A PAT which is missing, is transmitted scrambled, is errorred or is
transmitted not frequently enough will lead to an error message “PAT error”. The PAT should be transmitted free of errors and unscrambled every
500 ms at a maximum.
A PAT error occurs when
•
•
•
•
the PAT is missing,
the repetition rate is greater than 500 ms,
the PAT is scrambled,
the table ID is not zero.
Details in the PAT are not checked at that time.
11.4 Missing or Errored Program Map Table (PMT)
(PMT_error)
For each program, a Program Map Table (PMT) is transmitted at maximum intervals of 500 ms. The PIDs of the MAPs are listed in the PAT.
The PMT contains the respective PIDs of all elementary streams belonging
to this program. If a PMT referred to in the PAT is missing, there is no
way for the set top box or decoder to find the elementary streams and to
demultiplex and decode them. A PMT listed in the PAT and is missing, errored or scrambled will lead to the error message “PMT_error”.
A “PMT error” occurs when
•
•
•
•
a PMT listed in the PAT is missing,
a section of the PMT is not repeated after 500 ms at the latest,
a PMT is scrambled,
the table ID is not 2.
Details in the PMT are not checked.
194
11 Measurements on the MPEG-2 Transport Stream
Like any other table, the PMTs can also be divided into sections. Each
section begins with the table_ID=2 and with a PID, specified in the PAT,
of between 0x0010 and 0x1FFE according to MPEG-2 and between
0x0020 and 0x1FFE according to DVB. PID 0x1FFF is intended for the
zero packets.
PMT
PMT
PID=(...PAT)
PID=(...PAT)
TableID
ID=2
=2
Table
Program
Program
Map
Map
Table
Table
PID
Video ES
Audio ES
PID
188 byte
4 byte
header
Optional
adaptation
field
Header
Sync Transport Payload
unit start
byte error
indicator indicator
8
1
184 byte
payload
1
Transport
priority
1
Transport
PID scrambling
control
13
2
Adaptation Continuity
field
counter
control
2
4
Bit
Fig. 11.3. PID_error
11.5 The PID_Error
The PIDs of all elementary streams of a program are contained in the associated program map table (PMT). The PIDs are pointers to the elementary
streams: they are used for the addressed access to the corresponding packets of the elementary stream to be decoded. If a PID is listed in some PMT
but this is not contained in any packet in the transport stream, there is no
way for the MPEG-2 decoder to access the corresponding elementary
stream since this is now not contained in the transport stream or has been
multiplexed with the wrong PID information. This is what one might call a
“classical PID_error”. The time limit for the expected repetition rate of
transport packets having a particular PID must be set in dependence on application during the measuring. This is usually of the order of magnitude of
half a second but is a user-definable quantity, in any case.
11.6 The Continuity_Count_Error
195
A “PID_error” (Fig. 11.3.) occurs when
•
transport stream packets with a PID referred to in a PMT are not
contained in the transport stream or
if their repetition rate exceeds a user-definable limit which is
usually of the order of magnitude of 500 ms.
•
188 byte
4 byte
header
Optional
adaptation
field
Header
Sync Transport Payload
unit start
byte error
indicator indicator
8
1
184 byte
payload
1
Transport
priority
1
Transport
PID scrambling
control
13
2
Adaptation Continuity
field
counter
control
2
4
Bit
Fig. 11.4. Continuity_count_error
11.6 The Continuity_Count_Error
Each MPEG-2 transport stream packet contains in the 4-byte-long header a
4-bit counter which continuously counts from 0 to 15 and then begins at
zero again after an overflow (modulo 16 counter). However, each transport
stream packet of each PID has its own continuity counter, i.e. packets with
a PID=100, e.g., have a different counter, as do packets with a PID=200. It
is the purpose of this counter to enable one to recognize missing or repeated transport stream packets of the same PID in order to draw attention
to any multiplexer problems.
Such problems can also arise as a result of errored remultiplexing or
sporadically due to bit errors on the transmission link. Although MPEG-2
allows discontinuities in the transport stream, they must be indicated in the
adaptation field, e.g. after a switch-over (discontinuity indicator=1). In the
case of zero packets (PID=0x1FF), on the other hand, discontinuities are
allowed and is not checked, therefore.
196
11 Measurements on the MPEG-2 Transport Stream
A continuity_error (Fig. 11.4.) occurs when
•
the same TS packet is transmitted twice without a discontinuity
being indicated, or
if a packet is missing (count incremented by 2) without a
discontinuity being indicated, or
the sequence of packets is completely wrong.
•
•
Note: The way in which an MPEG-2 decoder reacts to a continuity
counter error when the packet sequence is, in fact, correct depends on the
decoder and the decoder chip used in it.
188 byte
4 byte
header
184 byte
payload
Optional
adaptation
field
Header
Sync Transport Payload
unit start
byte error
indicator indicator
8
1
Transport
priority
1
1
Transport
PID scrambling
control
13
Adaptation Continuity
field
counter
control
2
2
4
Bit
Fig. 11.5. Transport_error
11.7 The Transport_Error (Priority 2)
Every MPEG-2 transport stream packet contains a bit called Transport Error Indicator which follows directly after the sync byte. This bit flags any
errored transport stream packets at the receiving end. During the transmission, bit errors may occur due to various types of influences. If error protection (at least Reed Solomon in DVB and ATSC) is no longer able to repair all errors in a packet, this bit is set. This packet can no longer be
utilized by the MPEG-2 decoder and must be discarded.
A transport_error (Fig. 11.5.) occurs when
•
the transport error indicator bit in the TS header is set to 1.
11.9 The Program Clock Reference Error (PCR_Error, PCR_Accuracy)
197
1 byte
table ID
Payload of table
Transport stream
Payload
unit start
indicator = 1
Special
PID‘s
32 bit
CRC
checksum
Fig. 11.6. CRC_error
11.8 The Cyclic Redundancy Check Error
During the transmission, all tables in the MPEG-2 transport stream,
whether they are PSI tables or other private tables according to DVB (SI
tables) or according to ATSC (PSIP tables), are protected by a CRC checksum. It is 32 bits long and is transmitted at the end of each sector. Each
sector, which can be composed of many transport stream packets, is thus
additionally protected. A CRC error has occurred if these checksums do
not match the content of the actual section of the respective table. The
MPEG-2 decoder must then discard this table content and wait for this section to be repeated. The cause of a CRC error is in most cases interference
on the transmission link. If a set top box or decoder were to evaluate such
errored table sections it could become “confused”.
A CRC_error (Fig. 11.6.) occurs when
•
a table (PAT, PMT, CAT, NIT,...) in a section has a wrong
checksum which doesn’t match its content.
11.9 The Program Clock Reference Error (PCR_Error,
PCR_Accuracy)
All coding processes at the MPEG-2 encoder end are derived from a
27 MHz clock reference. This 27 MHz clock oscillator is coupled to a
198
11 Measurements on the MPEG-2 Transport Stream
42 bit-long counter which provides the System Time Clock (STC). For
each program, a separate system time clock (STC) is used. To be able to
link the MPEG-2 decoder to this clock, copies of the current program system time are transmitted about every 40 ms per program in the adaptation
field. The PMT of the respective program carries information about the TS
packets in which this clock time can be found.
The STC reference values are called Program Clock Reference (PCR).
They are nothing else than a 42 bit copy of the 42 bit counter. The MPEG2 decoder links itself to these PCR values via a PLL and derives its own
system clock from them.
PCR
Header
Optional
adaptation
field
Fig. 11.7. PCR value
If the repetition rate of the PCR values is too slow, it may be due to the
fact that the PLL of the receiver has problems in locking to it. MPEG-2
specifies that the maximum interval between two PCR values must not exceed a period of 40 ms. According to the DVB Measurement Guidelines, a
PCR_error has occurred if this time is exceeded.
The timing of the PCR values with respect to one another should also be
relatively accurate, i.e. there should not be any jitter. Jitter may occur, for
example, if the PCR values are not corrected, or are corrected inaccurately,
during remultiplexing.
If the PCR jitter exceeds ±500 ns, a PCR_accuracy_error has occurred.
PCR jitter frequently extends into the ±30 µs range which can be handled
by many set top boxes, but not by all. The first indication that the PCR jitter is too great is a black/white picture instead of a colour picture. The actual effect, however, depends on how the set top box is wired to the TV receiver. An RGB connection (e.g. via a SCART A/V cable) is certainly less
critical than a composite video cable connection.
A PCR_error occurs when
•
the difference between two successive PCR values of a program
is greater than 100 ms and no discontinuity is indicated in the
adaptation field, or
11.10 The Presentation Time Stamp Error (PTS_Error)
•
199
the time interval between two packets with PCR values of a program is more than 40 ms.
A PCR_accuracy_error occurs when
•
the deviation between the PCR values is greater than ±500 ns
(PCR jitter).
11.10 The Presentation Time Stamp Error (PTS_Error)
The Presentation Time Stamps (PTS) transmitted in the PES headers contain a 33 bit-long timing information item about the precise presentation
time. These values are transmitted both in the elementary video streams
and in the elementary audio streams and are used, e.g. for lip synchronisation between video and audio. The PTS values are derived from the system
time clock (STC) which has a total width of 42 bits but only the 33 MSBs
are used in this case. The spacing between two PTS values must not be
greater than 700 ms to avoid a PTS error.
PTS = presentation time stamp
PES payload
PES
header
Fig. 11.8. PTS value in the PES header
A PTS_error occurs when
•
the spacing between two PTS values of a program is greater
than 700 ms.
200
11 Measurements on the MPEG-2 Transport Stream
Although real PTS errors occur only rarely, a perceptible lack of lip
sync between video and audio happens quite frequently. In practice, the
causes of this are difficult to detect and identify during a broadcast and can
be attributable both to older MPEG-2 chips and to faulty MPEG-2 decoders. The direct measurement of lip synchronism would be an important test
parameter.
11.11 The Conditional Access Table Error (CAT_Error)
An MPEG-2 transport stream packet can contain scrambled data but only
the payload part must be scrambled and never the header or the adaptation
field. A scrambled payload part is flagged by two special bits in the TS
header, the Transport Scrambling Control bits. If both bits are set to zero,
there is no scrambling. If one of the two is not zero, the payload part is
scrambled and a Conditional Access Table (CAT) is needed to descramble
it. If this is missing or only rarely there, a CAT_error occurs. The CAT has
a 1 as PID and also a 1 as table ID. Apart from the EIT in the case of the
transmission of a program guide, all DVB tables must be unscrambled.
188 byte
4 byte
header
Optional
adaptation
field
Header
Sync Transport Payload
unit start
byte error
indicator indicator
8
1
1
Value
(binary)
00
184 byte
payload
Transport
priority
1
CAT
CAT
Transport
PID scrambling
control
13
2
Adaptation Continuity
field
counter
control
2
Description
No encrypted data contained in
the packet
01, 10, 11 Defined by user
4
Bit
PID=1
PID=1
TableID
ID=1
=1
Table
Conditional
Conditional
Access
Access
Table
Table
Fig. 11.9. CAT_error
A CAT_error (Fig. 11.9.) occurs when
•
a scrambled TS packet has been found but no CAT is being
transmitted,
11.12 Service Information Repetition Rate Error (SI_Repetition_Error)
•
201
a CAT has been found by means of PID=1 but the table ID is
not equal to 1.
11.12 Service Information Repetition Rate Error
(SI_Repetition_Error)
All the MPEG-2 and DVB tables (PSI/SI) must be regularly repeated at
minimum and maximum intervals. The repetition rates depend on the respective type of table.
The minimum time interval of the table repetition rate (s. Table 11.2.) is
normally about 25 ms and the maximum is between 500 ms and 30 s or
even infinity.
Table 11.2. PSI/SI table repetition time
Service
information
PAT
CAT
PMT
NIT
SDT
BAT
EIT
RST
TDT
TOT
Max. interval
(complete table)
0.5 s
0.5 s
0.5 s
10 s
2s
10 s
2s
30 s
30 s
Min. intervall
(single sections)
25 ms
25 ms
25 ms
25 ms
25 ms
25 ms
25 ms
25 ms
25 ms
25 ms
A SI_repetition_error occurs when
•
•
the time interval between SI tables is too great,
the time interval between SI tables is too small.
The limit values depend on the tables.
Since not every transport stream contains all the types of tables, the test
decoder must be capable of activating or deactivating the limit values.
202
11 Measurements on the MPEG-2 Transport Stream
11.13 Monitoring the NIT, SDT, EIT, RST and TDT/TOT
Tables
In addition to the PSI tables of the MPEG-2 standard, the DVB Group has
specified the NIT, SDT/BAT, EIT, RST and TDT/TOT SI tables.
The DVB Measurement Group recognized that these tables needed to be
monitored for presence, repetition rate and correct identifiability. This does
not include checking the consistency, i.e. the content of the tables. A SI table is identified by means of the PID and its table ID. This is because there
are some tables which have the same PID and can thus only be recognized
from the table ID (SDT/BAT and TDT/TOT).
Table 11.3. SI tables
Service
Information
NIT
SDT
BAT
EIT
RST
TDT
TOT
ST
PID [hex]
0x0010
0x0011
0x0011
0x0012
0x0013
0x0014
0x0014
0x0010
0x0013
Table_id [hex]
0x40, 0x41, 0x42
0x42, 0x46
0x4A
0x4E to 0x4F,
0x50 to 0x6F
0x71
0x70
0x73
to 0x72
Max.
[sec]
10
2
10
2
Interval
30
30
-
A NIT_error, SDT_error, EIT_error, RST_error or TDT_error occurs
when
•
•
a corresponding packet is contained in the TS but has the wrong
table index,
the time interval between two sections of these SI tables is too
great or too small.
11.14 Undefined PIDs (Unreferenced_PID)
All PIDs contained in the transport steam are conveyed to the MPEG-2 decoder via the PAT and the PMTs. There are also the PSI/SI tables. However, it is perfectly possible that the transport stream contains TS packets
11.15 Errors in the Transmission of Additional Service Information
203
whose PID is not indicated by this mechanism, the so-called unreferenced
PIDs. According to DVB, an unreferenced PID may be contained there
only for half a second during a program change.
An unreferenced_PID (Fig. 11.10.) occurs when
•
a packet having an unknown PID is contained in the transport
stream and is not referenced within a PMT after half a second at
the latest.
188 byte
4 byte
header
Optional
adaptation
field
Header
Sync Transport Payload
unit start
byte error
indicator indicator
8
1
184 byte
payload
1
Transport
priority
1
Transport
PID scrambling
control
13
2
Adaptation Continuity
field
counter
control
2
4
Bit
Fig. 11.10. Unreferenced PID
11.15 Errors in the Transmission of Additional Service
Information
Apart from the usual information, additional service information
(SI_other_error) can be transmitted for other channels, according to DVB.
These are the NIT_other, SDT_other and EIT_other tables.
The SI_other tables can be recognized from the PIDs and table_IDs in
Table 10.4. also lists the time limits.
An SI_other_error occurs when
•
•
the time interval between SI_other tables is too great,
the time interval between SI_other tables is too small.
204
11 Measurements on the MPEG-2 Transport Stream
Table 11.4. SI Other
Service
information
NIT_OTHER
SDT_OTHER
EIT_OTHER
Table_ID
0x41
0x46
0x4F, 0x60 to
0x6F
Max. interval
(complete table)
10 s
2s
2s
Min. interval
(single sections)
25 ms
25 ms
25 ms
11.16 Faulty tables NIT_other_error, SDT_other_error,
EIT_other_error
In addition to monitoring the 3 SI_other tables overall, they can also be
monitored individually:
A NIT_other_error, SDT_other_error, EIT_other_error occurs when
•
the time interval between sections of these tables is too great.
11.17 Monitoring an ATSC-Compliant MPEG-2 Transport
Stream
According to the DVB Measurement Guidelines, the following measurements can be made unchanged on an ATSC-compliant MPEG-2 transport
stream:
•
•
•
•
•
•
•
•
•
•
•
•
TS_sync_error
Sync_byte_error
PAT_error
Continuity_count_error
PMT_error
PID_error
Transport_error
CRC_error
PCR_error
PCR_accuracy_error
PTS_error
CAT_error
11.17 Monitoring an ATSC-Compliant MPEG-2 Transport Stream
205
It is only necessary to adapt all Priority 3 measurements to the PSIP tables.
Fig. 11.11. MPEG-2 analyzer, Rohde&Schwarz, on the left: DVM400, on the
right: DVM100
Bibliography: [TR100290], [DVMD], [DVM]
12 Picture Quality Analysis of Digital TV Signals
The picture quality of digital TV signals is subject to quite different effects
and influences than that of analog TV signals. Whereas noise effects in
analog TV signals manifest themselves directly as ‘snow’ in the picture,
they initially only produce an increase in the channel bit error rate in digital television. Due to the error protection included in the signal, however,
most of the bit errors can be repaired up to a certain limit and are thus not
noticeable in the picture or the sound. If the transmission path for digital
television is too noisy, the transmission breaks down abruptly (‘brick wall’
effect, also called ‘fall off the cliff’). Neither does linear or nonlinear
distortion have any direct effect on the picture and sound quality in digital
television but in the extreme case it, too, leads to a total transmission
breakdown. Digital TV does not require VITS (vertical insertion test signal) lines for detecting linear and nonlinear distortion or black-level lines
for measuring noise, neither are they provided there and would not produce any test results concerning the transmission link if they were. Nevertheless, the picture quality can still be good, bad or indifferent but it now
needs to be classified differently and detected by different means. There
are mainly two sources which can disturb the video transmission and
which can cause interference effects of quite a different type:
•
•
the MPEG-2 encoder or sometimes also the multiplexer, and
the transmission link from the modulator to the receiver.
The MPEG-2 encoder has a direct effect on the picture quality due to
the more or less severe compression imposed by it. The transmission link
introduces interference effects resulting in channel bit errors which manifest themselves as large-area blocking effects, as frozen picture areas or
frames or as a total loss of transmission. If the compression of the
MPEG-2 encoder is too great, it causes blocks of unsharp picture areas. All
these effects are simply called blocking. This section explains how the effects caused by the MPEG-2 video coding are produced and analyzed.
All video compression algorithms work in blocks, i.e. the image is in
most cases initially divided into blocks of 8 x 8 pixels. Each of these
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_12, © Springer-Verlag Berlin Heidelberg 2010
208
12 Picture Quality Analysis of Digital TV Signals
blocks is individually compressed to a greater or lesser extent, independently of the other blocks. In the case of MPEG-2, the image is additionally
divided into 16 x 16 pixels called macroblocks which form the basis for
the interframe coding. If the compression is excessive, the block boundaries become visible and blocking occurs. There are discontinuities between
blocks in the luminance and chrominance signals and these are perceptible.
With a predetermined compression, the amount of blocking in an image
also depends on the picture material, among other things. Some source images can be compressed without problems and almost without errors at a
low data rate whereas other material produces strong blocking effects
when compressed. Simple moving picture sources for moving-picture
compression are, for example, scenes with little movement and little detail.
Animated cartoons, but also classical celluloid films, can be compressed
without loss of quality with relatively few problems. The reason for this is,
among other things, that there is no movement between the first and second fields. In addition, the image structures are relatively coarse in animated cartoons. The most critical sources are sports programs, and this, in
turn, depends on the type of sport. By their nature, Formula I programs
will be more difficult to compress without interference than programs involving the thinker’s sport of chess. In addition, however, the actual picture quality depends on the MPEG-2 encoder and the algorithms used
there. In recent years, the picture quality has clearly improved in this department. Fig. 12.1. shows an example of blocking.
Apart from the blocking, the excessively compressed image also shows
the DCT structures, i.e. patterned interference suddenly occurs in the picture.
The decisive factor is that it is always the MPEG-2 encoder which is responsible for such interference effects. Although it is difficult to measure
good or bad picture quality caused by compression processes, it can be
done. Of course, picture quality will never be 100% measurement - there is
always some subjectivity involved. Even so-called objective video quality
analyzers are calibrated by test persons using subjective tests. At least, this
applies to analyzers which do not use a reference signal for quality assessment, but in practice, there is no reference signal with which the compressed video signal could be compared. The requirement that it should be
possible to use reference signals is unrealistic, at least with regard to
transmission testing.
The basis for all video quality analyzers throughout the world - and
there are not many - is the ITU-R BT.500 standard. This standard describes methods for subjective video quality analysis where a group of test
persons analyses video sequences for their picture quality.
12.1 Methods for Measuring Video Quality
209
Fig. 12.1. Blocking effects with excessive compression
12.1 Methods for Measuring Video Quality
The Video Quality Experts Group (VQEG) in the ITU has defined methods for assessing picture quality which have then been incorporated in the
ITU-R BT.500 standard.
In principle, these are two subjective methods for picture quality assessment by test persons, namely:
•
•
the DSCQS (Double Stimulus Continual Quality Scale) method
and
the SSCQE (Single Stimulus Continual Quality Evaluation)
method.
The two methods basically differ only in that one method makes use of
a reference video signal and the other one does not have a reference signal.
The basis is always a subjective picture quality analysis by a group of test
persons who assess an image sequence in accordance with a particular pro-
210
12 Picture Quality Analysis of Digital TV Signals
cedure. It is then attempted to reproduce these subjective methods by
means of objective methods in a test instrument by performing picture
analyses on the macroblocks and using adaptation algorithms.
12.1.1 Subjective Picture Quality Analysis
In subjective picture quality analysis, a group of test persons assesses an
image sequence (SSCQE - Single Stimulus Continual Quality Scale
Method, Fig. 12.2.) or compares an image sequence after compression
with the original (DSCQS - Double Stimulus Continual Quality Scale
Method) and issues marks on a quality scale from 0 (Bad) to 100 (Excellent) by means of a sliding control. The positions of the sliding controls are
detected by a computer connected to them which continually determines
(e.g. every 0.5 sec) a mean value of all the marks issued by the test persons.
Fig. 12.2. Subjective Picture Quality Analysis
A video sequence will then provide a picture quality value versus time,
i.e. a quality profile of the video sequence.
12.2 Objective Picture Quality Analysis
211
12.1.2 Double Stimulus Continual Quality Scale Method DSCQS
In the Double Stimulus Continual Quality Scale method according to
ITU-R BT.500, a group of test persons compares an edited or processed
video sequence with the original video sequence. The result obtained is a
comparative quality profile of the edited or processed video sequence, i.e.
a picture quality value from 0 (Bad) to 100 (Excellent) versus time.
The DSCQS method always requires a reference signal, on the one
hand, but, on the other hand, the purely objective analysis can then be performed very simply by forming the difference. In practice, however, a reference signal is frequently no longer provided. Transmission link measurements cannot be performed using this method. There are test
instruments on the market which reproduce this method (Tektronix).
12.1.3 Single Stimulus Continual Quality Evaluation Method
SSCQE
Since the Single Stimulus Continual Quality Evaluation method SSCQE
deliberately dispenses with a reference signal, this method can be used
much more widely in practice. In this method, a group of test persons only
assesses the processed video sequence and issues marks from 0 (Bad) to
100 (Excellent) which also provides a video quality profile versus time.
Fig. 12.3. Objective picture quality analysis using a test instrument [DVQ]
12.2 Objective Picture Quality Analysis
In the following sections, an objective test method for assessing the picture
quality analysis in accordance with the Single Stimulus Continual Quality
Scale Evaluation method is described. A digital picture analyzer operating
in accordance with this method may provide by Fig. 12.3.
212
12 Picture Quality Analysis of Digital TV Signals
Since the DCT-related artefacts of a compressed video signal are always
associated with blocking, an SSCQE type digital picture analyzer will attempt to verify the existence of this blocking in the picture. To be able to
do this, the macroblocks and blocks must be analyzed in detail.
0
§ AD(i = 0) ·
¸
¨
¨ AD(i = 1) ¸
⋅
¸
¨
¨
AD(i) = AD(i = 8) ¸
¸
¨
⋅
¸
¨
¸
¨
⋅
¸
¨
© AD(i = 15)¹
8
Raster position i
Block borders
Macroblock borders
Fig. 12.4. Pixel difference at the block and macroblock boundaries
In a test procedure developed by the Technical University of Braunschweig (Germany) and Rohde&Schwarz, the differences between adjoining pixels within a macroblock are formed. Pixel difference means that
simply the amplitude values of adjacent pixels of the Y signal within a
macroblock, and also separately those of the Cb and Cr signals are subtracted. For each macroblock, 16 pixel differences are then obtained per
line, e.g. for the Y signal. Then all 16 lines are analyzed. The same is also
done vertically which also provides 16 pixel differences per column for the
macroblock of the Y signal. This analysis is performed for all columns
within the macroblock. The pixel differences at the block borders are of
special significance here and will be particularly large in the case of blocking.
The pixel differences of all macroblocks within a line are then combined
by adding them together in such a way that 16 individual values are obtained per line (Fig. 12.4.). The 16 pixel difference values of the individual
lines are then also added together within a frame, resulting in 16 values per
frame as pixel difference values. This, finally. provides information about
the mean pixel difference 0 ... 15 in the horizontal and vertical direction
within all macroblocks. The same process is repeated for Cb and Cr, i.e.
the color difference signals.
Considering then the pixel differences of a video sequence with good
picture quality and one with poor picture quality, it can be seen quite
12.2 Objective Picture Quality Analysis
213
clearly what this objective test method for assessing the picture quality
amounts to:
16
AD(i=0)
AD(i=1)
AD(i=2)
AD(i=3)
AD(i=4)
AD(i=5)
AD(i=6)
AD(i=7)
AD(i=8)
AD(i=9)
AD(i=10)
AD(i=11)
AD(i=12)
AD(i=13)
AD(i=14)
AD(i=15)
14
Average Difference
12
10
8
6
4
2
0
0
2
4
6
8
10
12 14 16
Frame number
18
20
22
24
26
16
AD(i=0)
AD(i=1)
AD(i=2)
AD(i=3)
AD(i=4)
AD(i=5)
AD(i=6)
AD(i=7)
AD(i=8)
AD(i=9)
AD(i=10)
AD(i=11)
AD(i=12)
AD(i=13)
AD(i=14)
AD(i=15)
14
Average Difference
12
10
8
6
4
2
0
0
2
4
6
8
10
12 14 16
Frame number
18
20
22
24
26
Fig. 12.5. Averaged macroblock pixel differences in a video sequence with good
picture quality (top, flower garden/original, 6 Mbit/s) and with poorer picture
quality (bottom, flower garden/MPEG-2, 2 Mbit/s)
Fig. 12.5. shows clearly that the pixel amplitude differences in the
“good” video sequence are very close to each other for all 16 pixel differences within the macroblocks. In the present example, they are all at about
10 ... 12.
In a video sequence with “poor” quality (bottom display) with blocking,
it can be seen that the macroblock borders exhibit greater jumps, i.e the
pixel differences are greater there.
It can be seen clearly that pixel differences No. 0 and No. 8, in the bottom display, are obviously greater than the remaining difference values.
214
12 Picture Quality Analysis of Digital TV Signals
No. 0 corresponds to the macroblock border and No. 8 corresponds to the
block boundary within a macroblock.
Clearly, this simple analysis of the pixel amplitude differences makes it
possible to verify the existence of blocking (Fig. 12.6.).
16
AD(i=0)
AD(i=1)
AD(i=2)
AD(i=3)
AD(i=4)
AD(i=5)
AD(i=6)
AD(i=7)
AD(i=8)
AD(i=9)
AD(i=10)
AD(i=11)
AD(i=12)
AD(i=13)
AD(i=14)
AD(i=15)
14
Average Difference
12
DVQL-U
10
8
6
spatial activity SA
4
2
0
0
2
4
6
8
10
12 14 16
Frame number
18
20
22
24
26
Fig. 12.6. Determining digital video quality level unweighted (DVQL-U) and spatial activity (SA) from the macroblock pixel differences
The basic test value of a Digital Video Quality Analyzer by Rohde&Schwarz for calculating the picture quality of DCT coded video
sequences is the picture quality test value DVQL-U (digital video quality
level - unweighted). DVQL-U is used as absolute value for the existence of
blocking type interference patterns within an original frame. In contrast to
DVQL-W (digital video quality level - weighted), DVQL-U is a direct
measure of these blocking types of interference. Depending on the original
frame, however, the test value is not always correlated with the impression
of quality of a subjective observation.
To bring the objective picture quality test value closer to the subjectively perceived picture quality, other quantities in the moving picture
must also be taken into consideration. These are
•
•
the spatial activity (SA), and
the temporal activity (TA).
12.2 Objective Picture Quality Analysis
215
This is because both spatial and temporal activity can render blocking
structures invisible, i.e. they can mask them. These artifacts in the picture
are then simply not seen by the human eye.
The spatial activity is a measure of the existence of fine structures in the
picture. A picture rich in detail, i.e. one with many fine structures, exhibits
high spatial activity. An unstructured monochrome picture, on the other
hand, would correspond to a spatial activity of zero. The maximum theoretically achievable spatial activity would occur if a white pixel always alternates with a black pixel both horizontally and vertically in a frame (fine
grid pattern).
Fig. 12.7. Low (left) and high (right) spatial activity
Time
Time
Fig. 12.8. Low (left) and high (right) temporal activity
In addition to the spatial activity in the picture (Fig. 12.7.), the temporal
activity (TA) must be taken into consideration (Fig. 12.8.). The temporal
activity is an aggregate measure of the change (movement) in successive
frames. The maximum temporal activity which could be achieved in theory
would be if all pixels were to change from black to white or conversely in
successive frames. Accordingly, a temporal activity of 0 corresponds to a
sequence of frames without movement.
216
12 Picture Quality Analysis of Digital TV Signals
The two parameters SA and TA must be included when calculating the
weighted video quality from the unweighted DVQL (blocking level).
In a first process, the unweighted digital quality level DVQL-U for all
Y, Cb and Cr signals and the spatial activity SA and temporal activity TA
are determined in the above-mentioned digital video quality analyzer.
TA
Process 1
MPEG-2 TS
16
AD(i=0)
AD(i=1)
AD(i=2)
AD(i=3)
AD(i=4)
AD(i=5)
AD(i=6)
AD(i=7)
AD(i=8)
AD(i=9)
AD(i=10)
AD(i=11)
AD(i=12)
AD(i=13)
AD(i=14)
AD(i=15)
14
Average Difference
12
10
8
6
4
2
SA
DVQL-U
0
0
2
4
6
8
10
12 14 16
Frame number
18
20
22
24
26
Y, Cb, Cr
Process 2
“weighting“
adaptation to
subjective
method
DVQL-W
Fig. 12.9. Using the digital video quality analyzer [DVQ]
Weighting is then performed in a second process which thus takes into
account subjective factors. The display of the digital video quality analyzer
shows both the digital video quality level - weighted or unweighted - and
the spatial and temporal activity. The analyzer is also able to detect decoding problems, in addition to the video quality: These problems include
•
•
•
picture freeze (TA = 0),
picture loss (TA = 0, SA = 0), and
sound loss.
Digital video quality analyzers are mainly used close to the MPEG-2
encoding stages since the transmission has no further effect on the video
quality itself. Naturally, such an analyzer will also detect the decoding
problems caused by bit errors produced by the transmission link. Since in
many cases the network operator is not the program provider, as well, digital video quality analyzers are often also found at the network termination
end so that objective measurement parameters are available as a basis for
any discussion between network operator and program provider. Digital
video quality analyzers are also of great importance in the testing of
MPEG-2 encoders.
12.3 Summary and Outlook
217
12.3 Summary and Outlook
Artefacts caused by the MPEG encoding process lead to a picture quality
which is more or less good. This is apparent as
•
•
•
blocking (visible block transitions)
blurring (lack of high frequencies)
"mosquito" noise (visible DCT structures).
In the case of the more recent image coding methods such as MPEG-4
AVC, however, other approaches must be used in picture quality assessment. E.g., deblocking filters are used here in an attempt to keep blocking
as invisible as possible. The only objective information is provided by an
analysis of the quantization performed by the encoder. Such measurements
are supported by the MPEG analyzer DVM [DVM].
However, the influence of the distribution and transmission paths in the
digital TV network remains the same. It leads to bit errors which, in turn,
can lead to a "fall-off-the-cliff" at some time (Fig. 12.11.), either it works
or it no longer works at some time. In a transition case, slice structures become apparent as can be seen in Fig. 12.11.
Fig. 12.10. Digital video quality analyzer, Rohde&Schwarz [DVQ]
218
12 Picture Quality Analysis of Digital TV Signals
Fig. 12.11. Picture disturbances caused by interference during the transmission;
slice-like structures are recognizable which are caused here by severe rain during
satellite reception.
Bibliography: [ITU500], [DVQ], [DVM]
13 Basic Principles of Digital Modulation
To begin with, this chapter quite generally creates a basis for an approach
to the digital modulation methods. Following this chapter, it would also be
possible to continue e.g. in the field of mobile radio technology (GSM,
IS95 or UMTS) as the basic knowledge discussed here applies to the field
of communication technology and its applications overall. However, its
prime intent is to create the foundation for the subsequent chapters on
DVB-S, DVB-C, OFDM/COFDM, DVB-T, ATSC and ISDB-T. Experts,
of course can simply skip this chapter.
13.1 Introduction
Analog transmission of information has long been effected by means of
amplitude modulation (AM) and frequency modulation (FM). The information to be transmitted is impressed on the carrier by varying either its
amplitude or frequency or phase, this process being referred to as modulation.
To transmit data signals, i.e. digital signals, amplitude or frequency shift
keying was used in the early times of data transmission. To transmit a data
stream of e.g. 10 Mbit/s by means of simple amplitude shift keying (ASK),
a bandwidth of at least 10 MHz is required if a non-return-to-zero code
(NRZ) is used. According to the Nyquist theorem, a bandwidth corresponding to at least half the data rate is required for the NRZ baseband
signal. Using ASK produces two sidebands and that gives a RF signal with
a bandwidth which is equal to the data rate of the baseband signal. The
bandwidth actually required is even larger because of the signal filtering
necessary to suppress adjacent-channel interference.
An analog telephone channel is about 3 kHz wide. Initially, a data rate
of 1200 bit/s could be achieved for this channel. Today, 56 kbit/s is no
problem any more. We are used to our fax and modem links operating at
such data rates. This quantum leap ahead was possible only through the
use of modern digital modulation methods known as IQ modulation. IQ
modulation is basically a form of amplitude modulation.
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_13, © Springer-Verlag Berlin Heidelberg 2010
220
13 Basic Principles of Digital Modulation
We know the following modulation methods:
•
•
•
•
•
•
•
Amplitude modulation
Frequency modulation
Phase modulation
Amplitude shift keying (ASK)
Frequency shift keying (FSK)
Phase shift keying (PSK)
Amplitude and phase shift keying (QAM)
What we want is to reduce the bandwidth for data signal transmission.
This is possible only by using modern digital modulation methods. Our
aim is to cut the required bandwidth by several factors relative to the data
rate of the signal transmitted.
u(t)
u(t)
sin
Q
Im
AA=vector
= vectorlength
length
A
A
im
tt
ϕ
TT
ϕ
re
f=1/T;
f=1/T
Re
I
cos
Euler formula:
A e (2 ft + ) = re cos (2 ft) + j im(2 ft);
u(t)=A sin(2 t/T + );
Fig. 13.1. Vector representation of a sinusoidal signal
It is obvious that this will not go without disadvantages, i.e. susceptibility to noise and interference will increase. In the following, the digital
modulation methods will be discussed.
Before entering into this subject, we should like to point out that in electrical engineering it is customary to represent sinusoidal quantities by
means of vectors (s. Fig. 13.1.). Each sinusoidal quantity can be unambiguously described by its amplitude and zero-phase angle. Moreover, the
frequency must be known. In the vector representation, the rotating vector
13.2 Mixer
221
at the time t = 0 is shown. The vector is then at the zero-phase angle and its
length corresponds to the amplitude of the sinusoidal quantity.
Fig. 13.1. represents a sine signal in the time domain and in the form of
a vector. The rotating vector, whose length corresponds to the amplitude,
is shown at zero-phase angle ϕ. The sine signal is obtained by projecting
the rotating vector on the vertical axis (Im) and recording the position of
the vector tip versus time. The corresponding cosine signal is obtained by
projecting the rotating vector on the horizontal axis (Re).
The vector can be split into its real part and its imaginary part, terms
which are derived from the theory of complex numbers in mathematics.
The real part corresponds to the projection onto the horizontal axis and is
calculated from Re = A • cos . The imaginary part corresponds to the
projection onto the vertical axis and can be calculated from Im = A • sin .
The length of the vector is related to the real part and the imaginary part
via Pythagoras' theorem
A = Re 2 + Im 2 ; .
The real part can also be imagined to be the amplitude of a cosine signal
and the imaginary part as the amplitude of a sine signal.
Any desired sine or cosine signal can be obtained by the superposition
of a sine and a cosine signal of the same frequency and the desired amplitudes.
The real part is also called the I, or in-phase, component and the imaginary part is called the Q, or quadrature, component, where in-phase stands
for 00 phase angle to a reference carrier and quadrature stands for 900
phase angle. The terms real part, imaginary part, cosine and sine component and I and Q component will appear time and again in the sections to
follow.
13.2 Mixer
We will see that the mixer is one of the most important electronic components that make up an IQ modulator. A mixer is basically a multiplier. The
modulation signal is usually converted to the IF by means of a carrier signal. As a result, two sidebands about the carrier are obtained. This type of
modulation is known as double-sideband amplitude modulation with suppressed carrier. The mixer shown in Fig. 13.2. is basically a double switch
driven by the carrier. It reverses the polarity of the modulation signal at the
carrier frequency.
In the case of a purely sinusoidal modulation signal, two spectral lines
are obtained – one above and one below the carrier frequency – each
222
13 Basic Principles of Digital Modulation
spaced from the carrier at an offset of the modulation frequency. Moreover, sub-harmonics at an offset of the carrier frequency are produced. The
latter have to be suppressed by means of a lowpass filter.
Lowpass
filter
low pass filter
us(t)
fs
umix(t)
fs
fif
carrier
uif(t)
ucarrier(t)
f
if
2f
if
umix(t)
us(t)
Modulated carrier
Modulation signal fs
carrier
uif(t)
ucarrier(t)
Fig.13.2. Mixer and mixing process. Amplitude modulation with suppressed carrier
LO
RF
IF
Fig. 13.3. Block diagram of a double-balanced mixer
Fig. 13.3. is a block diagram of a modern analog double-balanced
mixer. The polarity of the modulation signal is switched by 4 PIN diodes.
The carrier signal (LO = local oscillator) is coupled in via an RF transformer, and the modulation product is coupled out via an RF transformer.
The modulation signal is fed DC-coupled.
13.3 Amplitude Modulator
223
Mixers are today often implemented in the form of purely digital multipliers which, except for quantization noise and rounding errors, have an
ideal behaviour.
If a direct voltage is applied as modulation signal, the carrier itself appears at the output of the mixer. Superimposing a sinusoidal signal on the
DC leads to normal amplitude modulation with unsuppressed carrier. (Fig.
13.4.).
Frequency domain
AM
mod.
Modulation signal us(t)
Time domain
Carrier lo(t)
Fig. 13.4. “Normal“ amplitude modulation with unsuppressed carrier
13.3 Amplitude Modulator
In amplitude modulation, the information is contained in the amplitude of
the carrier. The modulation signal changes (modulates) the carrier amplitude. This is effected by means of an AM modulator.
Fig. 13.4. illustrates "normal" AM modulation, in which the carrier is
not suppressed. A sinusoidal modulation signal varies the carrier amplitude
and so is impressed on the carrier as an envelope. In the example in Fig.
13.4., both the carrier and the modulation signal are sinusoidal signals.
Looking at the spectrum, we not only find a spectral line at the carrier fre-
224
13 Basic Principles of Digital Modulation
quency but also two sidebands spaced from the carrier at an offset of the
modulation frequency. For example, if a 1 MHz carrier is amplitudemodulated with a sinusoidal 1 kHz signal, a modulation spectrum with the
carrier signal at 1 MHz and two sideband signals at 1 kHz above and below the carrier will be obtained. The bandwidth is 2 kHz in this case.
Frequency domain
Mixer
Modulation signal us(t)
Time domain
Carrier lo(t)
Fig. 13.5. Amplitude modulation with suppressed carrier
As mentioned above, the carrier is suppressed by the mixer. If a mixer is
used for amplitude modulation and the modulation signal itself has no DC
component, no spectral line at the carrier frequency will be found in the
modulation spectrum. There are only the two sidebands. Fig. 13.5. shows
amplitude modulation effected by means of a double-balanced mixer. In
the modulation spectrum, we find not only the two sidebands but also harmonic sidebands about multiples of the carrier frequency. The latter have
to be suppressed by lowpass filters. Fig. 13.5. also shows a typical amplitude-modulated signal in the time domain with suppressed carrier. The
bandwidth is the same as with "normal" amplitude modulation, i.e. with
unsuppressed carrier.
13.4 IQ Modulator
225
13.4 IQ Modulator
In colour television, quadrature modulation or IQ modulation has been
used for a long time for the transmission of colour information. With PAL
or NTSC colour subcarriers, the chrominance information is contained in
the phase of the subcarrier and the colour saturation, or colour intensity, in
the amplitude of the subcarrier. The colour subcarrier is superimposed on
the luminance signal.
The modulated colour subcarrier is generated by means of an IQ modulator or quadrature modulator, where "I" stands for in-phase and "Q" for
quadrature phase.
I
i(t)
data(t)
Mapper
+
iqmod(t)
q(t)
Q
90°
Carrier lo(t)
Fig. 13.6. IQ modulator
An IQ modulator (see Fig. 13.6.) has an I path and a Q path. The I path
incorporates a mixer which is driven with 0° carrier phase. The mixer in
the Q path is driven with 90° carrier phase. This means that I stands for 0°
and Q for 90° carrier phase. I and Q are orthogonal to each other. In the
vector diagram, the I axis coincides with the real axis and the Q axis with
the imaginary axis.
PAL or NTSC modulators, too, incorporate an IQ modulator. For digital
modulation, a mapper is connected ahead of the IQ modulator. The mapper
is fed with the data stream data(t) to be transmitted; the output signals i(t)
and q(t) of the mapper are the modulation signals for the I and the Q
mixer. i(t) and q(t) are no longer data signals but signed voltages.
If i(t)=0, the I mixer produces no output signal, if q(t)=0, the Q mixer
produces no signal. If i(t) is at 1 V, for example, the I mixer will output a
226
13 Basic Principles of Digital Modulation
carrier signal with constant amplitude and 0° carrier phase. If q(t), on the
other hand, is at 1 V, the Q mixer will output a carrier signal with constant
amplitude and 90° carrier phase (s. Fig. 13.10.).
The I and Q modulation products are combined in an adder.
Q
I
I
i(t)
+
iqmod(t)
q(t)=0
Q
90°
Carrier lo(t)
Fig. 13.7. IQ modulator, I path only
The product iqmod(t) is, therefore, the sum of the output signals of the
I mixer and the Q mixer. If the Q mixer supplies no output signal, iqmod(t)
corresponds to the output signal of the I path and vice versa.
Since the output signals of the I and the Q path are sine and cosine signals of the same frequency (carrier frequency) and differ only in amplitude, a sinusoidal output signal iqmod(t) of variable amplitude and phase is
obtained through the superposition of the sinusoidal I output signal and the
cosinusoidal Q output signal. Therefore, with the aid of control signals i(t)
and q(t), we can vary the amplitude and phase of iqmod(t).
With the IQ modulator, we can generate pure amplitude modulation,
pure phase modulation, or combined amplitude and phase modulation. A
sinusoidal modulator output signal can thus be controlled in amplitude and
phase.
The following applies to the amplitude and phase of iqmod(t):
A=
( Ai )2 + ( Aq )2 ;
13.4 IQ Modulator
ϕ = arctan(
227
Aq
);
Ai
where Ai is the amplitude of the I path and Aq the amplitude of the
Q path.
From the incoming data stream data(t), the mapper generates the two
modulation signals i(t) and q(t). We will see later that bit groups are combined to create certain patterns for i(t) and q(t), i.e. for the modulation signals of the I and the Q path.
Let us first look at the I path only (Fig. 13.7.). The Q path is driven with
q(t)=0, i.e. it delivers no output signal and so does not contribute to
iqmod(t). We now apply alternately +1 V and -1 V to the I path modulation input, so that i(t)=+1 V or i(t)= -1 V. Looking at the output signal
iqmod(t), we see that the carrier lo(t) is present and switched only in phase
between 0° and 180°. By varying the amplitude of i(t), we can vary the
amplitude of iqmod(t).
For the vector diagram this means that the vector changes between 0°
and 180° and varies in length but always remains on the I axis as long as
only i(t) is present and being varied (Fig. 13.7).
2µs
1
1 1
0
1
0 0 0
0
Non Return to Zero Code (NRZ)
Example: 1 Mbit/s
after rolloff filtering:
bandwidth >= 1/2µs = 500 kHz
Fig. 13.8. NRZ code
This is a suitable point for discussing fundamentals relating to bandwidth conditions in the baseband and at RF. In the extreme case, the
bandwidth of a data signal with an NRZ (non-return to zero) code (Fig.
13.8.) at a data rate of 1 Mbit/s can be cut (filtered) to such an extent that
500 kHz bandwidth are just sufficient to ensure reliable decoding. With us-
228
13 Basic Principles of Digital Modulation
ing much mathematics, this can be explained quite simply by the fact that
01 alternations represent the highest frequency; i.e. the period has a length
of 2 bits and is thus 2 s in the case of a data rate of 1 Mbit/s. The reciprocal of 2 s is then 500 kHz and the minimum baseband bandwidth for
transmitting an NRZ code is then:
fbaseband_NRZ [Hz]
0.5 • data rateNRZ [bits/s];
If such a filtered NRZ code (Fig. 13.8.) is then supplied, e.g. without
DC to a mixer as in the I path of this IQ modulator, two sidebands having
each the bandwidth of the input baseband signal (Fig. 13.9.) are produced
at RF. The minimum bandwidth required at RF is thus:
fRF_NRZ [Hz]
data rateNRZ [bits/s];
BPSK
1
µs
1
1 1
0
I
RF
bandwidth
>=1 MHz
1
0 0 0
Q
0
Example: NRZ 1 Mbit/s
LO
600 MHz
600
MHz
Symbol rateBPSK = 1/symbol durationBPSK =
1/bit durationBPSK = 1/1µs = 1 MSymbols/s;
1 MSymbols/s Î RF bandwidth >= 1 MHz
Fig. 13.9. BPSK modulation
In this type of modulation, therefore, the ratio between data rate and
minimum bandwidth required at RF is 1:1. This type of modulation is
called binary phase shift keying, or biphase shift keying, BPSK. With
BPSK, a data rate of 1 Mbit/s requires a minimum bandwidth of 1 MHz at
RF level. The duration of one stable state of the carrier is called a symbol
13.4 IQ Modulator
229
and in BPSK, a symbol has exactly the same duration as one bit. The reciprocal of the symbol duration is called the symbol rate.
Symbol rate = 1/symbol duration;
With a data rate of 1 Mbit/s with BPSK (Fig. 13.9.), the symbol rate is 1
MSymbol/s. The minimum bandwidth required always corresponds to the
symbol rate, i.e. 1 MSymbol/s requires a minimum bandwidth of 1 MHz.
Now let us assume that i(t) is zero and there is only a q(t) output signal.
We now switch q(t) between +1 V and -1 V. iqmod(t) corresponds to the
output signal of the Q mixer; there is no contribution from the I path.
Again, a sine signal is obtained for iqmod(t), but with phase 90° or 270°.
By varying the amplitude of q(t), the amplitude of iqmod(t) can be varied.
For the vector diagram this means that the vector changes between 90° and
180° and varies in length along the Q axis (imaginary axis).
Q
I
I
i(t)=0
+
iqmod(t)
q(t)
Q
90°
Carrier lo(t)
Fig. 13.10. IQ modulator, Q path active only
Next, we want to vary both i(t) and q(t) between +1 V and -1 V. In this
case, the modulation products of the I path and the Q path are added up, so
we can switch the carrier between 45°, 135°, 225° and 315°. This is referred to as quadrature phase shift keying or QPSK. Allowing any voltages
230
13 Basic Principles of Digital Modulation
for i(t) and q(t), any desired amplitude and phase can be generated for
iqmod(t).
The data stream data(t) is converted to the two modulation signals i(t)
for the I path and q(t) for the Q path by means of a mapper. This is shown
in Fig. 13.13. for QPSK modulation. The mapping table is the rule according to which the data stream data(t) is converted to modulation signals i(t)
and q(t). In the case of QPSK, two bits (corresponding to bit 0 and bit 1 in
the mapping table) are combined to form a dibit. For dibit combination 10,
for example, the mapper outputs the signals i(t)= -1 V and q(t)= -1 V according to the mapping table shown here.
Q
QPSK
I
I
+
iqmod(t)
q(t)
Q
90°
Carrier lo(t)
Fig. 13.11. IQ Modulator, I and Q Paths active and identical Amplitudes (QPSK)
The bit combination 11 yields i(t)=+1 V and q(t)= -1 V in this example.
The allocation of bits to modulation signals, defining how the bit stream is
to be read and converted by the mapper, is merely a matter of definition. It
is important that the modulator and the demodulator, i.e. the mapper and
the demapper, use the same mapping rules. Fig. 13.12. also shows that in
this case the data rate is halved after the mapper. QPSK can transmit
two bits per state. Two bits each are combined to form a dibit that determines the state of the mapper output signals i(t) and q(t). Therefore, in this
case, i(t) and q(t) have half the data rate of data(t). i(t) and q(t) in turn
modulate the carrier signal and, in the case of QPSK, switch it only in
phase. There are four possible constellations for iqmod(t): 45°, 135°, 225°
and 315°. The information is contained in the phase of the carrier. Now
13.4 IQ Modulator
231
that we can switch the carrier phase at half the data rate relative to the input rate, the required channel bandwidth is reduced by a factor of 2. The
time the carrier or vector dwells on a specific phase (dwell time = symbol
duration) is referred to as symbol (Fig. 13.12. and 13.14.). The reciprocal
of the symbol duration is the symbol rate. The required bandwidth corresponds to the symbol rate. Compared with simple bit transmission, the
available bandwidth capacity is now boosted by a factor of 2.
Time
data(t)
0
1
1
0
1
1
0
0
0
1
Mapping table
Bit 1
Bit 0
I
Q
0
0
+1
+1
0
1
-1
+1
1
0
-1
-1
1
1
+1
-1
Q
01
i(t)
-1
-1
+1
+1
-1
q(t)
+1
-1
-1
+1
+1
00
QPSK
I
11
10
Vector
I
Symbol
duration
i(t)
data(t)
Mapper
+
iqmod(t)
q(t)
Q
90°
Carrier lo(t)
Fig. 13.12. Mapping with QPSK modulation
In practice, higher-order modulation methods are used besides QPSK.
Fig. 13.11. shows 16QAM produced by varying the amplitude and phase.
The information is in the amplitude, or magnitude, and in the phase. In the
case of 16QAM (= 16 quadrature amplitude modulation), four bits are
combined in the mapper; one carrier constellation can, therefore, carry four
bits, and there are 16 possible carrier constellations. The data rate after the
mapper, or the symbol rate, is a fourth of the input data rate. This means
that the required channel bandwidth has been reduced by a factor of four.
In vector diagrams for IQ modulation, it is common practice to represent
only the end point of the vector. A vector diagram in which all possible
vector constellations are entered is referred to as a constellation diagram.
232
13 Basic Principles of Digital Modulation
Fig 13.13. shows constellation diagrams of real QPSK, 16QAM and
64QAM signals, i.e. impaired by noise. The decision thresholds of the demapper are shown too.
The number of bits transmitted per symbol is the logarithm to the base
of 2 of the constellation.
QPSK = 4QAM
2 bit / symbol
16QAM
4 bit / symbol
64QAM
6 bit / symbol
Fig. 13.13. Constellation diagrams of QPSK, 16QAM and 64QAM
Time
data(t)
0
1
1
0
1
1
0
0
0
1
i(t)
-1
-1
+1
+1
-1
q(t)
+1
-1
-1
+1
+1
01
Q
00
QPSK
Vector
I
Symbol
duration
iqmod(t)
QPSK
symbol
Fig. 13.14. QPSK
10
11
13.5 The IQ Demodulator
233
Fig. 13.14. shows the original data stream data(t), the resulting constellations of the carrier vector, and the switched, or keyed, carrier signal
iqmod(t) in the time domain. Each switching status is referred to as a symbol. The duration of a switching status is called symbol duration. The reciprocal of the symbol duration is the symbol rate.
I
Demapper
iqmod(t)
data(t)
Q
90°
Symbol clock
Carrier and
clock
recovery
Fig. 13.15. IQ demodulator
13.5 The IQ Demodulator
In this section, IQ demodulation will be discussed briefly (s. Fig. 13.15.).
The digitally modulated signal iqmod(t) is fed to the I mixer, which is
driven with 0° carrier phase, and to the Q demodulator, which is driven
with 90° carrier phase. At the same time, the carrier and the symbol clock
are recovered in a signal processing block. To recover the carrier, the input
signal iqmod(t) is squared twice. So, a spectral line at the fourfold carrier
frequency can be isolated by means of a bandpass filter. A clock generator
is locked to this frequency by means of a PLL. Moreover, the symbol
clock has to be recovered, i.e. the point in the middle of the symbol has to
be determined. Some modulation methods allow carrier recovery only with
an uncertainty of multiples of 90°.
234
13 Basic Principles of Digital Modulation
By IQ mixing, the baseband signals i(t) and q(t) are retrieved. The carrier harmonics superimposed on these signals have to be eliminated by
means of a lowpass filter before the signals are applied to the demapper.
The demapper simply reverses the mapping procedure, i.e. it samples
the baseband signals i(t) and q(t) at the middle of the symbol and so recovers the data stream data(t).
Fig. 13.16. illustrates the processes of IQ modulation and demodulation
in the time domain and in the form of constellation diagrams for the QPSK
method. The signal in the first line represents the input data stream data(t).
The second and the third line show the signals i(t) and q(t) at the modulation end. The fourth and the fifth line are the voltage characteristics after
the I and the Q mixer of the modulator, the sixth line the characteristic of
iqmod(t). The phase steps between the symbols are clearly visible. The
amplitude does not change (QPSK). In the last line, the corresponding constellation diagrams are shown. Lines 7 and 8 show the digitally recovered
signals i(t) and q(t) at the demodulation end. It can be seen that, in addition
to the baseband signals, the traces contain the carrier at double the frequency. The latter has to be eliminated both in the I and the Q path by
means of a lowpass filter prior to demapping. In the case of analog mixing,
harmonics would be superimposed in addition which would also be suppressed by the lowpass filters.
data(t)
i(t)
q(t)
imod(t)
qmod(t)
iqmod(t)
idemod(t)
qdemod(t)
vector(t)
Fig. 13.16. IQ modulation and demodulation (mapping table different to examples
before)
13.5 The IQ Demodulator
235
Very frequently, however, demodulation is performed using the fs/4
method, which requires a less complex IQ demodulator. The modulated
signal iqmod(t) is passed through an anti-aliasing lowpass filter and then
sampled by means of an A/D converter which operates at the fourfold IF of
the modulated signal iqmod(t). Therefore, if the carrier of iqmod(t) is at fIF,
the sampling frequency is 4 • fIF. This means that a complete carrier cycle
is sampled four times (see Fig. 13.8.). Provided the A/D converter clock is
fully synchronous with the carrier clock, the rotating carrier vector is sampled exactly at the instants shown in Fig. 13.17. The symbol clock is recovered in a carrier and clock recovery block as described above.
I
i(t)
iqmod(t)
A
Delay
+/-1
D
q(t)
fS=4*fIF
-/+1
fS/2
Fig. 13.17. IQ demodulation using fS/4 method
Q
fIF
I
Fig. 13.18. fS/4 demodulation
Q
FIR
interpolation
236
13 Basic Principles of Digital Modulation
After the A/D converter, a switch splits the data stream into two streams
of half the data rate. For example, the odd samples are taken to the I path
and the even samples to the Q path. This means that only every second
sample is taken to the I path or the Q path, respectively, thus halving the
data rate in both paths. The multipliers in the two paths only reverse the
sign, i.e. they multiply the samples alternately by +1 and -1.
Principle of the fs/4 method:
If the A/D converter operates exactly at the fourfold carrier frequency
(IF) and the A/D converter clock and the carrier clock are fully synchronized, the samples correspond alternately to an I and a Q value. This can
be seen from Fig. 13.18. Each second sample in the I and the Q path has a
negative sign and so has to be multiplied by -1.
Fig. 13.19. IQ demodulation according to the fS/4 method
The baseband signals i(t) and q(t) are thus recovered in a very simple
way. Since the signals i(t) and q(t) have to settle after each symbol change
(change of switching status), and settling is delayed by half a clock cycle
by the switch after the A/D converter, the signals have to be pulled back
into synchronism with the aid of digital filters.
To this effect a signal, for example q(t), is interpolated, so retrieving the
sample between two values. This is done with the aid of an FIR filter (finite impulse response filter, digital filter). Each digital filter has a basic delay, however, which has to be compensated by introducing a corresponding delay in the other path, i.e. the I path in this case, by means of a delay
line. After the FIR filter and the delay line, the sampled and clocksynchronous signals i(t) and q(t) are available and can be applied to the
demapper.
13.5 The IQ Demodulator
237
As already mentioned, the less complex fs/4 method is frequently used
in practice. In the case of OFDM-(Orthogonal Frequency Division Multiplex)-modulated signals, this circuit is provided directly ahead of the FFT
signal processing block. The fs/4 demodulation method is supported by
many modern digital circuits.
Re(f)
Re(f)
f
f
Im(f)
Im(f)
f
f
Sine
Cosine
Fig. 13.20. Fourier Transform of a cosine and a sine
Re(f)
f
Im(f)
f
Fig. 13.21. Fourier Transform of a general real time domain signal
238
13 Basic Principles of Digital Modulation
13.6 Use of the Hilbert transform in IQ modulation
In this section we will discuss the Hilbert Transform, which plays a major
role in some digital modulation methods such as OFDM or 8VSB (c.f.
ATSC, the U.S. version of digital terrestrial TV).
Let us start with sine and cosine signals. At the time t=0, the sine signal
has the value 0, the cosine signal the value 1. The sine signal is shifted 90°
relative to the cosine signal, i.e. it leads the cosine signal by 90°. We will
see later that the sine signal is the Hilbert Transform of the cosine signal.
Based on the sine and cosine functions, we can arrive at some important
definitions: the cosine function is an even function, i.e. it is symmetrical
about t=0, so that cos(x) = cos(-x) applies.
H( )
j
-j
Fig. 13.22. Fourier Transform of the Hilbert Transform
The sine function, on the other hand, is an odd function, i.e. it is halfturn symmetrical about t=0, so that sin(x) = -sin(-x) applies. The spectrum
of the cosine, i.e. its Fourier Transform, is purely real and symmetrical
about f=0. The imaginary component is zero (s. Fig. 13.20.).
The spectrum of the sine, i.e. its Fourier transform, is purely imaginary
and half-turn symmetrical (Fig. 13.20.). The real component is zero. The
above facts are important for understanding the Hilbert Transform. For all
real time-domain signals, the spectrum of all real components versus f
(Re(f)) is symmetrical about f=0, and the spectrum of all imaginary components versus f (Im(f)) is half-turn symmetrical about f=0 (s. Fig. 13.21.).
Any real time-domain signal can be represented as a Fourier series – the
superposition of the cosinusoidal and sinusoidal harmonics of the signal.
The cosine functions are even and the sine functions odd. Therefore, the
13.6 Use of the Hilbert transform in IQ modulation
239
characteristics previously stated for a single cosine function or a single
sine function also generally apply to a sum of cosine functions or a sum of
sine functions. Let us now discuss the Hilbert Transform itself. Fig. 13.22.
shows the transfer function of a Hilbert transformer. A Hilbert transformer
is a signal processing block with special characteristics. Its main purpose is
to phase shift a sine signal by 90°. This means that a cosine is converted to
a sine and a sine to a minus cosine. The amplitude remains invariant under
the Hilbert Transform. These characteristics apply to any type of sinusoidal signal, i.e. of any frequency, amplitude, or phase. Hence, they also apply to all the harmonics of any type of time-domain signal. This is due to
the transfer function of the Hilbert transformer which is shown in
Fig. 13.22. – essentially it only makes use of the symmetry characteristics
of even and odd time-domain signals referred to above.
Examining the transfer function of the Hilbert transformer, we find:
•
•
•
•
All negative frequencies are multiplied by j, all positive frequencies by -j. j is the positive, imaginary square root of -1
The rule j • j = -1 applies
Real spectral components, therefore, become imaginary and
imaginary components become real
Multiplication by j or -j may invert the negative or positive part
of the spectrum
Applying the Hilbert Transform to a cosine signal, the following is obtained: A cosine has a purely real spectrum symmetrical about zero. If the
negative half of the spectrum is multiplied by j, a purely positive imaginary spectrum is obtained for all negative frequencies. If the positive half
of the spectrum is multiplied by -j, a purely negative imaginary spectrum is
obtained for all frequencies above zero. The spectrum of a sine is obtained.
This applies analogously to the Hilbert Transform of a sine signal:
By multiplying the positive imaginary negative sine spectrum by j, the
latter becomes negative real (j • j = -1). By multiplying the negative
imaginary positive sine spectrum by -j, the latter becomes purely positive
real (-j • -j = -(√-1 • √-1) = 1). The spectrum of a minus cosine is obtained.
The cosine-to-sine and sine-to-minus-cosine mapping by the Hilbert
Transform also applies to all the harmonics of any type of time-domain
signal.
Summarizing, the Hilbert Transform shifts the phases of all harmonics
of any type of time-domain signal by 90°, i.e. it acts as a 90° phase shifter
for all harmonics.
240
13 Basic Principles of Digital Modulation
13.7 Practical Applications of the Hilbert Transform
Often, a sideband or parts of a sideband have to be suppressed during
modulation. With single-sideband modulation (SSB modulation), for example, the upper or lower sideband has to be suppressed, which can be
done in a variety of ways. For example, simple lowpass filtering can be
used or, as is common practice in analog TV, vestigial sideband filtering.
Hard lowpass filtering has the disadvantage that significant group delay
distortion is produced. The latter method in any case is technically complex. For a long time, however, an alternative to single-sideband modulation has been available, this alternative being known as the phase method.
A single-sideband modulator using the phase method operates as follows:
the IQ modulator is fed with a modulation signal which is applied unmodified to the I path and to the Q path with 90° phase shift. A phase shift of
plus or minus 90° in the Q path results in suppression of the upper or the
lower sideband, respectively.
Re(f)
f
Im(f)
f
I
u(t)
+
HT
ssb(t)
Q
90º
Fig. 13.23. Practical application of the Hilbert Transform to suppressing a sideband in SSB modulation
13.7 Practical Applications of the Hilbert Transform
241
It is difficult to implement an ideal 90° phase shifter for all harmonics of
a baseband signal as an analog circuit. Digital implementation is no problem – thanks to the Hilbert Transform. A Hilbert transformer is a 90°
phase shifter for all components of a real time-domain signal.
Fig. 13.23. shows the suppression of a sideband by means of an IQ
modulator and a Hilbert transformer. A real baseband signal is directly fed
to the I path of an IQ modulator and to the Q path via a Hilbert transformer. The continuous lines at f=0 represent the spectrum of the baseband
signal, the dashed lines at f=0 the spectrum of the Hilbert Transform of the
baseband signal.
It can be seen clearly that, under the Hilbert Transform, the half-turn
symmetrical imaginary component becomes a mirror symmetrical real
component and the mirror symmetrical real component becomes a halfturn symmetrical imaginary component at the baseband.
If the unmodified baseband signal is then fed into the I path and the Hilbert Transform of the baseband signal into the imaginary path, spectra
about the IQ modulator carrier like those shown in Fig. 13.22. are obtained. It can be seen that in this case the lower sideband is suppressed.
Information
Source coding
Channel coding
e.g. compression
forward error
correction (FEC)
Modulation
Interferences
Transmission
link
Demodulation
Bit errors
Channel decoding
Source decoding
Information
Fig. 13.24. Information transmission
242
13 Basic Principles of Digital Modulation
13.8 Channel Coding/Forward Error Correction
In addition to the most suitable modulation method, the most appropriate
error protection, i.e. channel coding, is selected from among the characteristics of the respective transmission channel. The current aim is to approach the Shannon-Limit as closely as possible. This section discusses
commonly used error protection mechanisms and creates the foundations
for the transmission methods in digital television.
Channel coding
Block
codes
Data
Convolutional
codes
(1955: Elias,
1967: Andrew Viterbi)
Code
+
Cyclical
group code
using
group/field
theory
of linear
algebra
+
Hamming
(1950)
Reed-Solomon
(1963)
o ut1
Block code+
interleaver+
block code
in
+
BCH
(1960)
General
block code
LDPC
(1963: Gallager)
Concatenated
codes
(1966: David Forney)
o ut2
Block code+
interleaver+
Convol.
code
Turbo
codes
(1993:
concat.
convol.
codes)
Fig. 13.25. Channel coding
Before information is transmitted, source encoding (Fig. 13.24.) is used
for changing it into a form in which it can be transmitted in as little space
as possible. This simply means that it is compressed as well as is possible
and tolerable. After that, error protection (Fig. 13.24.) is added before the
data are sent on their journey. This corresponds to channel coding. The error-protected data are then digitally modulated onto a sinusoidal carrier after which the information is sent on its way, subjected to interference such
as noise, linear and nonlinear distortion, discrete and wide-band interferers, intermodulation, multipath propagation etc. Due to the varying degree
of signal quality at the receiving end (Fig. 13.24.), this causes bit errors after its demodulation back to a data stream. Using the error protection
added in the transmitter (FEC - Forward Error Correction), errors can then
be corrected to a certain extent in the channel decoder. The bit error ratio is
reduced back to a tolerable amount, or to zero. The information is then
13.8 Channel Coding/Forward Error Correction
243
processed in such a way that it can be presented. I.e. the data are decompressed, if necessary, which corresponds to source decoding..
The “toolbox“ (Fig. 13.25.) which can be used for providing error protection is not as large as one might assume. The essential basics were
largely created back in 1950 – 1970. Essentially, there are block codes and
convolutional codes. Block codes (Fig. 13.27.) are based on principles of
linear algebra and simply protect a block of data with an error protection
block. From the data to be transmitted, a type of checksum is basically calculated which can be used to find out if errors have crept in during the
transmission or not, and where the errors, if any, are located. Some block
codes also allow a certain number of errors to be repaired. Convolutional
codes (Fig. 13.28.) delay and randomize the data stream with itself and
thus introduce a certain “intelligence“ into the data stream to be transmitted. The counterpart to the convolutional coder is the Viterbi-decoder developed by Andrew Viterbi in 1967.
Scrambler
Coder 1
in
Time
interleaver
Example: DVB-S, DVB-T:
scrambler = energy dispersal,
coder 1 = Reed-Solomon,
time interleaver = Forney interleaver
coder 2 = convolutional coder
Fig. 13.26. Concatenated Forward Error Correction
m
k
Data
l
Code
m=k+l
Algorithm
(linear algebra)
Example: DVB Reed-Solomon code:
k = 188 byte, l = 16 byte, m = 204 byte
Fig. 13.27. Block code
Coder 2
out
244
13 Basic Principles of Digital Modulation
Before the data are supplied to the error protection section (Fig. 13.26.),
however, they are first scrambled in order to bring movement into the data
stream, to break up any adjoining long strings of zeroes or ones into more
or less random data streams. This is done by mixing and EXOR-operations
on a pseudo random binary sequence (PRBS). At the receiving end, the
data stream now encrypted must be recovered by synchronous descrambling. The scrambling is followed by the first FEC. The data stream is then
distributed in time by means of time interleaving. This is necessary so that
during the deinterleaving at the receiving end, burst errors can be broken
up into individual errors. This can be followed by a second FEC.
EXOR
+
out1
+
code rate =
data rate in /
data rate out;
in
+
out2
EXOR
Shift registers
Example: GSM, UMTS, DVB inner coder
Fig. 13.28. Convolutional coding
There is also concatenated error protection (Fig. 13.25. and 13.26.)
(David Forney, 1966). It is possible to concatenate both block codes with
block codes and block codes with convolutional codes or also convolutional codes with convolutional codes. Concatenated convolutional codes
are called turbo codes. They only made their appearance in the 90’s.
It depends on the choice of modulation method and of the error protection how closely the Shannon-Limit is approached. Shannon determined
the theoretical limit of the data rate in a distorted channel of a certain bandwidth. The precise formula for this is:
13.8 Channel Coding/Forward Error Correction
C = B ⋅ log 2 (1 +
245
S
);
N
If the signal/noise ratio is more than 10 dB, the following formula can
also be used:
C[ Bit / s ] ≈
1
⋅ B[ Hz ] ⋅ SNR[dB ];
3
Channel capacity
C[bit/s/Hz]
C[bit/s]=channel capacity
B[Hz]=channel bandwidth
S/N=signal to noise ratio
S/N>>1:
C = B ⋅ log 2 (1 +
S
);
N
1
C ≈ ⋅ B ⋅ SNR;
3
SNR[dB ] = 10 ⋅ log(
Claude Elwood Shannon, USA 1948
The Bell System Technical Journal
„A Mathematical Theory of Communication“
S
);
N
SNR[dB]
Fig. 13.29. Channel capacity
Depending on the properties of the transmission channel, a certain
amount of data can be transmitted within a shorter or longer time period.
The available channel bandwidth determines the maximum possible symbol rate. The signal/noise ratio present in the channel then determines the
modulation method to be selected, in combination with the appropriate error protection. These relationships are illustrated by Prof. Küpfmüller’s socalled information cube (Fig. 13.30.).
The FEC actually used in the transmission process will be discussed in
the relevant chapter.
246
13 Basic Principles of Digital Modulation
“Information cube“
[Prof. Küpfmüller]
Data
set
S/N [dB]
Channel
bandwidth
[Hz]
san on
Tr issi [s]
m e
tim
1
Data _ volume[bit ] ≈ B[ Hz ] ⋅ t[ s ] ⋅ SNR[dB];
3
Fig. 13.30. Information cube
13.9 A Comparison to Analog Modulation Methods
In the age of digital transmission methods, it still pays to look at the traditional analog modulation methods which have been part of our lives for
more than 100 years, from the beginning of carrier keying in Morse code,
to AM radio and then FM radio which, due to its quality, is still giving
digital audio broadcasting a run for its money. Adjacently to a frequencymodulated carrier, however, digitally modulated signals are now also being
transmitted (e.g. IBOC, HD radio) which is why it makes sense to acquaint
oneself again with the analog modulation methods. In this section, therefore, the special features of the traditional modulation methods of
•
•
•
amplitude modulation (AM),
frequency modulation (FM), and
phase modulation (PM)
will be presented. The relevant experience is also quite applicable to the
digital modulation methods. In addition, this chapter is intended to see to it
that this traditional knowledge is not completely lost. In the second half of
the 19th century, budding communication engineers were confronted with
the question of how to convey messages by wire and wirelessly from one
13.9 A Comparison to Analog Modulation Methods
247
place to another. The first variant of message transmission by wire was telephony and the telegram. In telephony, the voice was converted by a carbon microphone into amplitude variations of an electrical voltage and
transmitted as a pure baseband signal via two-wire lines. In the case of the
telegram, a direct voltage was keyed on and off, making it possible, with
the aid of the Morse alphabet, to transmit text messages from point A to
point B. The Morse alphabet was thus already virtually a type of source
encoding, working with redundancy reduction. Short codes were used for
letters occurring frequently in the language and long codes were used for
letters occurring less frequently. After the discovery of and research into
electromagnetic waves, it was then a matter of applying them in the wireless transmission of messages. Baseband signals (voice, various texts)
were then impressed on a sinusoidal carrier of a particular frequency and
this carrier then transmitted the information from a transmitting antenna to
one or more receiving antennas. The necessity of selecting a suitable
transmitting frequency, i.e. carrier frequency, within the correct range of
frequencies or wavelengths and modulating it with the information to be
sent out arises firstly from the mere fact that electromagnetic waves will
emanate from the transmitting antenna only when the wavelength λ = c/f
reaches the order of magnitude of the antenna dimensions. Depending on
the frequency range selected, these messages could then be transmitted
over a greater or lesser distance. But above all else, it was possible to select different carrier frequencies and thus to send out many different messages simultaneously, a principle which applies to the present day. In contrast to the past, however, the main problem today is that very many people
wish to send out great amounts of information at the same time, with the
resultant problem of a lack of frequencies and thus of having to control the
availability of frequencies. At the beginning of communication technology
it didn't matter whether an entire band was occupied or only a part of it,
differently from today where the frequency resource is scarce and must be
well managed. There are separate international organisations especially established for this purpose which deal with this problem.
13.9.1 Amplitude Modulation
In amplitude modulation, the information to be transmitted is impressed on
the amplitude of a sinusoidal carrier (Fig. 13.31.). This type of modulation
can be considered simply as multiplying a modulation signal by a carrier
signal. If the carrier signal is multiplied by the value zero, the result is also
zero. If the carrier signal is multiplied by a particular, informationdependent value, a particular carrier amplitude is obtained. The simplest
248
13 Basic Principles of Digital Modulation
variant of amplitude modulation is amplitude shift keying (ASK). In the
original Morse-type transmission, a carrier was simply switched on and
off. The original characters could be decoded again from the duration of
the on- and off-periods. However, we will now consider the case of modulating a sinusoidal or cosinusoidal carrier with a a sinusoidal or cosinusoidal modulation signal. Using the cosine instead of the sine for representing
the situation results in simpler addition theorems which can be used for
explaining the physics. The modulation signal is described as:
u signal (t ) = U signal ⋅ cos(2πf signal t ) = U signal ⋅ cos(ωsignal t );
The carrier signal is described as:
ucarrier (t ) = U carrier ⋅ cos(2πf carrier t ) = U carrier ⋅ cos(ωcarrier t );
Amin
Amax
Fig. 13.31. Amplitude Modulation
If a direct voltage component of UDC is added to the modulation signal
usignal(t) and then multiplied by the carrier signal by means of a mixer (multiplier) multiplied, the following is obtained
u (t ) = (u signal (t ) + U DC ) ⋅ ucarrier (t ) =
(U signal ⋅ cos(ωsignal t ) + U DC ) ⋅ U carrier ⋅ cos(ωcarrier t );
Applying the addition theorems of mathematics /geometry
13.9 A Comparison to Analog Modulation Methods
cos(α ) ⋅ cos( β ) =
249
1
1
cos(α − β ) + cos(α + β );
2
2
the result is then:
u (t ) = U DC ⋅ ucarrier (t ) + u signal (t ) ⋅ ucarrier (t );
or
u (t ) = U DC ⋅ U carrier cos(ωcarrier t )
1
+ U signalU carrier cos((ωcarrier − ω signal )t )
2
1
+ U signalU carrier cos((ωcarrier + ωsignal )t );
2
A(f)
fS
Baseband
fT-fs fT
LSB
fT+fs
USB
f
Fig. 13.32. Spectrum of the amplitude modulation (AM) with baseband signal,
lower sideband (LSB) and upper sideband (USB)
Reinterpreting this, a carrier component is now produced at the center of
the band, plus two sidebands - one lower than the carrier by the modulation frequency and one higher than the carrier by the modulation frequency. Setting the DC component to zero results in an amplitude modulation with suppressed carrier. Depending on the DC component added to
the modulation component, a greater or lesser carrier component is pro-
250
13 Basic Principles of Digital Modulation
duced. It can be seen that two sidebands of the respective highest modulation frequency are created (Fig. 13.32.) and that, as a result, the minimum
bandwidth required at the RF domain must be greater than/equal to twice
the highest modulation frequency:
bRFAM ≥ 2 ⋅ f signal ;
The amount of amplitude modulation of a sinusoidal or cosinusoidal
carrier is determined by the modulation factor m (Fig. 13.31.). If Amax corresponds to the maximum amplitude of the modulated carrier and Amin
corresponds to the minimum amplitude of the modulated carrier signal, the
modulation factor m is defined as:
m=
Amax − Amin
;
Amax + Amin
S/N
fsignal
Fig. 13.33. LF signal/noise ratio in AM as a function of the baseband signal frequency
In amplitude modulation, the information to be transmitted is located in
the amplitude of the modulated carrier. Nonlinearities in the transmission
channel have a direct effect as amplitude distortions in the demodulated
baseband signal. which means that AM systems have to be very linear.
However, real transmission systems also exhibit interfering effects in the
form of noise. In the case of AM, a baseband signal with superimposed
additive white Gaussian noise (AWGN) is obtained after the demodulation. The noise power density is here constant and independent of the fre-
13.9 A Comparison to Analog Modulation Methods
251
quency of the modulation signal (Fig. 13.33.). The resultant baseband signal/noise ratio (Fig. 13.34.) directly corresponds linearly to the RF signal/noise ratio or may possibly be shifted slightly in parallel due to negative demodulation characteristics.
13.9.2 Variants of Amplitude Modulation
In amplitude modulation, various variants have become successful in practice which are:
•
•
•
•
•
traditional AM with unsuppressed carrier and both sidebands,
AM with suppressed carrier and both sidebands,
single-sideband modulation with unsuppressed carrier,
single-sideband modulation with suppressed carrier,
and vestigial-sideband modulation.
S/NLF
S/NRF
Fig. 13.34. LF signal/noise ratio of AM as a function of the RF signal/noise ratio
In single-sideband modulation, either the upper or the lower sideband is
completely suppressed whereas in vestigial-sideband modulation one sideband is only partially suppressed. The practical use is simply the saving in
bandwidth since the information is present completely both in the upper
sideband and in the lower sideband. The relevant sideband was formerly
suppressed completely or partially by analog filtering and later by applying
a Hilbert transformer or 90-degree phase shifter, and an IQ modulator. In
vestigial sideband modulation (VSB), the analog filters could be made less
severe at the transmitting and receiving end.
252
13 Basic Principles of Digital Modulation
13.9.3 Frequency Modulation
In frequency modulation (Fig. 13.35.), the information to be transmitted is
impressed on the frequency of the carrier, i.e. the frequency of the carrier
changes to a certain extent in dependence on the information to be transmitted. The simplest variant of frequency modulation is frequency shift
keying (FSK). The principle of frequency modulation can be traced back
to Edwin Howard Armstrong (1933) who also invented the superheterodyne receiver. The aim had been to become more insensitive to atmospheric interference. Today, frequency modulation is of great significance
mainly in the field of VHF sound broadcasting. Frequency modulation
(FM) is quite tolerant of nonlinearities and much more insensitive to noiselike influences.
A(f)
fC
fC
f
Fig. 13.35. Spectrum of frequency modulation
This is why FM transmitters are mostly operating in class C mode, i.e.
the amplifiers themselves are highly nonlinear, but this also means that
they are much more efficient. I.e., FM is mainly used where corresponding
channel requirements are set (Low SNR, nonlinearities). In analog TV
transmission via satellite, travelling tube amplifiers (TWAs), which are
quite nonlinear, are used both in the earth terminal and in the satellite.
Moreover, the SNR is about 10 dB due to the long distance of about 36000
km between satellite and Earth.
The frequency modulation can be expressed mathematically by:
u (t ) = U carrier ⋅ cos(2πf (t ) ⋅ t );
13.9 A Comparison to Analog Modulation Methods
253
i.e. the frequency f(t) is a function of time and is influenced by the modulation factor. This involves two parameters, namely
•
•
frequency deviation fcarrier and
maximum modulation frequency fsignal max
In frequency modulation, the modulation index is
M =
Δf carrier
;
f signal max
From Carson's formula (J.R. Carson, 1922), the approximate minimum FM
bandwidth required at the RF level can be specified as:
B10% = 2(Δf carrier + f signal max );
bzw.
B1% = 2(Δf carrier + 2 ⋅ f signal max );
S/NLF
m
FM
AM
S/NRF
FM threshold
Fig. 13.36. LF S/N in frequency modulation as a function of the RF S/N
[MAUESL1]
254
13 Basic Principles of Digital Modulation
All signal components in the channel are here below 10% or 1%,
respectively. The spectral lines produced can be determined by Bessel
functions. Limiting the bandwidth causes nonlinear distortions in the
demodulated signal. The spreading of the bandwidth in the channel results
in a gain in the LF SNR with respect to the amplitude modulation above
the so-called FM threshold (Fig. 13.36.). This gain can be expressed as:
FM gain _ SN _ LF = 10 ⋅ log(3 ⋅ M 3 )dB;
This FM gain is only present above the FM threshold which is a SNR in
the RF domain of about:
FM threshold ≈ 7...10dB + 10 ⋅ log(2 ⋅ ( M + 1))dB;
S/N
fsignal
Fig. 13.37. LF signal-to-noise ratio of frequency modulation as a function of the
baseband signal frequency
The FM threshold itself is defined as disproportionally high drop-off
compared with the FM gain at the 1-dB point. Below the FM threshold,
spike-like noise signals occur due to phase discontinuities of the carrier. In
the case of the wideband FM normally used in analog television by satellite, small white splashes are then produced in the picture. The LF noise
occurring with frequency modulation is called "delta noise" (Fig. 13.37.),
i.e. the noise power density is not constant but increases with increasing
LF bandwidth. To counteract this, preemphasis is applied at the transmitting end, i.e. higher frequencies are emphasized more. At the receiving
13.9 A Comparison to Analog Modulation Methods
255
end, deemphasis is then applied, decreasing the amplitude of the higher
frequencies again in accordance with the preemphasis characteristic so that
a linear frequency response is obtained again.
13.9.4 Phase Modulation
Phase modulation is closely related to frequency modulation. In phase
modulation, the information to be transmitted is impressed on the phase of
the carrier:
u (t ) = U carrier ⋅ cos(2πft + ϕ (t ));
Both frequency modulation and phase modulation are known collectively as angle modulation. Like frequency modulation, phase modulation
is insensitive to nonlinearities. Technically, phase modulation is used
mainly in frequency modulation with preemphasis. To be able to distinguish between frequency modulation and phase modulation, the following
relationships must be considered: In frequency modulation, the frequency
deviation fcarrier is proportional to the amplitude of the modulating signal
Usignal:
Δf carrier _ FM ~ U signal ;
The frequency deviation is not dependent on the modulating signal, i.e. is
not a function of the latter:
Δf carrier _ FM ≠ f ( f signal );
The phase deviation
carrier in frequency modulation corresponds to the
modulation index and is inversely proportional to the frequency of the
modulating signal fsignal:
Δϕcarrier _ FM ~
1
f signal
;
In phase modulation, the phase deviation
amplitude of the modulating signal Usignal:
carrier
is proportional to the
256
13 Basic Principles of Digital Modulation
Δϕcarrier _ PM ~ U signal ;
The frequency deviation in phase modulation is dependent on the maximum signal frequency and proportional to the signal frequency of the
modulating signal:
Δf carrier _ PM ~ f signal ;
The phase deviation in phase modulation is not dependent on the maximum frequency of the modulating signal, i.e. is not a function of the latter:
Δϕ carrier _ PM ≠ f ( f signal );
I.e., frequency modulation can be distinguished physically from phase
modulation only when the frequency of the modulating signal is changing;
in frequency modulation, the frequency deviation does not change then
whereas in phase modulation the frequency deviation of the carrier
changes in dependence on the signal frequency of the modulating signal.
FM and PM can also be distinguished in the LF signal to noise ratio: in
FM, the LF SNR increases with increasing signal frequency (delta noise)
whereas in PM the LF SNR is not a function of the signal frequency.
13.10 Band Limiting of Modulated Carrier Signals
Modulated carrier signals must only occupy their designated channel; they
must not interfere with adjacent or even more remote channels. This applies to any type of modulation, whether amplitude, phase or frequency
modulation or whether analog or digital. To this end, measures are taken
both on the baseband side and on the RF side. The baseband signal itself
must already be band limited. On the RF side, too, precautions for protecting the adjacent channels are taken in most cases by using SAW filters at
the intermediate frequency level. In addition, harmonic traps and channeldependent mask filters are used directly in the RF path.
In the case of digital modulation, in particular, baseband filtering will
still be discussed briefly at this point because this is a matter of concern in
every digital TV or sound broadcasting transmission standard using singlecarrier modulation. It can thus be dealt with here centrally in advance. If
sinusoidal carriers are keyed by a square wave as is the case with digital
13.10 Band Limiting of Modulated Carrier Signals
257
modulation (amplitude and phase shift keying), this results in a multiplication of a square wave by a sinusoidal signal in the time domain and a convolution with the Fourier transform of the square wave signal with the
Fourier transform of the sinusoidal signal in the frequency domain. If a
single square-wave pulse from minus infinity to plus infinity were present,
a continuous sin(x)/x-shaped spectrum with zeroes corresponding to the
inverse of the square-wave pulse duration would be obtained (Fig. 13.38.).
A(f)
f=1/ t;
t
f
f
Fig. 13.38. Spectrum of a single rectangular pulse
A(f)
T
f=1/ t;
t
t
t
f
f
Fig. 13.39. Line spectrum of a symmetric sequence of rectangular pulses
A sequence of rectangular, or square-wave, pulses results in a line spectrum molded to the sin(x)/x function. The spectral lines occur with the
258
13 Basic Principles of Digital Modulation
spacing of the period T. If the period corresponds to exactly twice the
width of the of the square-wave pulse, the line spectrum with the maximum frequency possible is obtained, providing spectral components
spaced apart by 1/(2• t) (Fig. 13.39.). If the period has a longer duration,
the spectral lines move towards lower frequencies. But there will always
be a periodic line spectrum of multiples of the fundamental which corresponds to the inverse of the period of the sequence of square wave pulses.
If the periods are fluctuating and the square wave pulse duration is constant, the sin(x)/x form is more or less "modulated out", resulting in practice in a sin(x)/x-shaped overall spectrum in the case of digital modulation
with a data signal with good energy dispersal. However, only the fundamental is needed for demodulation, i.e. all harmonics can be suppressed.
This is done maximally in rectangular form (Fig. 13.40.).
A(f)
f
f
Fig. 13.40. Rectangular suppression of the harmonics (linear representation)
The control signals at the IQ modulator input i(t) and q(t) are initially
square wave signals and meet the above-mentioned conditions. They must
be band-limited before they are supplied to the IQ modulator. In the case
of digital modulation at the baseband level, this band limiting is carried out
by special "well-mannered" low-pass filters. These are constructed in most
cases, and always digitally today, as root cosine square filters with a special roll-off characteristic. The roll-off factor r describes here where the filtering starts from in relation to the Nyquist bandwidth. The filter curve is
symmetric and has its center at the so-called Nyquist point. The same
matched filtering is performed again in the receiver, resulting in the total
13.11 Summary
259
filter curve. The filter characteristic is designed to result in minimum overshoot in the demodulated signals i(t) and q(t). Fig.13.41. shows the resultant RF spectrum. The dashed curve corresponds to the spectrum after the
modulator and the continuous curve corresponds to the spectrum after additional filtering (matched filter) in the demodulator. In the case of GSM,
Gaussian filtering takes the place of the cosine filtering. In DVB-S, DVBS2 and DVB-C, however, the cosine square or root cosine square filtering
shown in Fig. 13.41, is used. On the baseband side, the spectrum to the
f
f
cos 2
cos 2
cos 2
cos 2
r= f/BN
f
f
2BN = BS
f
left of the vertical axis can be imagined to consist of negative frequencies;
the vertical axis corresponds to the band center of the channel on the RF
side.
Fig. 13.41. Roll-off filtering of digitally modulated signals
13.11 Summary
In this chapter, many basic principles underlying the video and audio
transmission standards have been repeated or recreated. A basic understanding of single carrier modulation (SC modulation) is also the prerequisite for an understanding of multicarrier modulation (MC modulation).
Whilst single carrier modulation is the subject of many transmission stan-
260
13 Basic Principles of Digital Modulation
dards, others employ multicarrier modulation, depending on the characteristics and requirements of the transmission channel.
Bibliography: [MAEUSL1], [BRIGHAM], [KAMMEYER],
[LOCHMANN], [GIROD], [KUEPF], [REIMERS], [STEINBUCH]
14 Transmitting Digital Television Signals by
Satellite - DVB-S/S2
Today, analog television signals are widely received by satellite since this
type of installation has become extremely simple and inexpensive. Thus, in
Europe, a simple satellite receiving system complete with dish, LNB and
receiver is available for less than 100 Euros and there are no follow-up expenses. In the meantime, analog satellite reception in Europe is being replaced more and more by DVB-S - digital video broadcasting by satellite and also DVB-S2 since 2005. In this chapter, the method of transmitting
MPEG-2 source encoded TV signals via satellite is described.
Centrifugal force:
F1 = mSat · ω2 · r ;
ω
Earth
r
F1
Satellite
mSat = mass of satellite;
ω= 2 · π / T = angular speed;
π= 3.141592654 = circular constant;
T = 1 day = 24 · 60 · 60 s = 86400 s ;
Fig. 14.1. Centrifugal force of a geostationary satellite
Every communication satellite is located geostationary (Figs. 14.1.,
14.2. and 14.3.) above the equator in an orbit of about 36000 km above the
Earth’s surface. This means that these satellites are positioned in such a
way that they move around the Earth at the same speed as that with which
the Earth itself is rotating, i.e. once per day. There is precisely only one
orbital position, at a constant distance of about 36,000 km from the Earth’s
surface, where this can be achieved, the only point at which the centrifugal
force of the satellite and the gravitational attraction of the Earth cancel
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_14, © Springer-Verlag Berlin Heidelberg 2010
262
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
each other. However, the various satellites can be positioned at various degrees of longitude, that is to say angular positions above the Earth’s surface. For example, ASTRA is positioned at 19.2o East. It is due to this position of the satellite above the equator that all satellite receiving antennas
point to the South in the Northern hemisphere, and to the North in the Southern hemisphere.
Centripetal force:
F2 = γ · mEearth · mSat / r2 ;
mErde = mass of Earth ;
F2
Earth
r
Satellite
γ = constant of graphitation = 6.67 · 10-11 m3/kg s2 ;
Fig. 14.2. Centripetal force acting on a geostationary satellite
Balance condition:
centrifugal force = centripetal force:
F1 = F2 ;
F2
F1
Satellite
mSat · ω2 · r = γ · mEarth · mSat / r2 ;
r = (γ · mEarth / ω2)1/3 ;
r = 42220 km ;
d = r – rErde = 42220 km – 6370 km = 35850 km ;
Fig. 14.3. Equilibrium condition
The orbital data of a geostationary satellite can be calculated on the basis of the following relationships: The satellite is moving at a speed of one
day per orbit around the Earth. This results in the following centrifugal
force: The satellite is attracted by the Earth with a particular gravitational
force of attraction due to its orbital height: The two forces, centrifugal
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
263
force and centripetal force, must be in equilibrium. From this, it is possible
to determine the orbit of a geostationary satellite (Fig. 14.1. to 14.3.).
Compared with the orbit of a space shuttle, which is about 400 km above the earth's surface, geostationary satellites are far distant from the Earth,
about one tenth of the way to the moon. Geostationary satellites launched,
e.g. by the space shuttle or by similar carrier systems, must first be pushed
up into this distant orbit by firing auxiliary rockets (apogee motors). From
there, they will never pass back into the earth's atmosphere. On the
contrary, shortly before their fuel reserves for path corrections are used up
they must be pushed out into the so-called "satellite cemetery" orbit which
is even farther away. Only satellites close to the earth in a non-stationary
orbit can be "collected" again. As a comparison - the orbital time of nearearth satellites which, in principle, also include the International Space
Station ISS or the space shuttle, is about 90 minutes per orbit at about
27000 km/h.
10
00
11
01
Fig. 14.4. Modulation parameters in DVB-S (QPSK, Gray coded)
But now let us return to DVB-S. In principle, the same satellite systems
can be used for transmitting both analog TV signals and digital TV signals.
However, in Europe, the digital signals are located in a different frequency
band while the previous satellite frequency bands are still occupied with
analog television. Hundreds of programs in Europe can be received both as
analog signals and as digital signals via satellite and a lot of these are completely free to air.
In the following sections, the techniques for transmitting digital television via satellite will be described. This chapter also forms the basis for
understanding digital terrestrial television (DVB-T). Both systems make
264
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
use of the same error protection mechanisms but in DVB-T, a much more
elaborate modulation method is used.
The DVB-S transmission method is defined in the ETSI Standard ETS
300421 “Digital Broadcasting Systems for Television, Sound and Data
Services; Framing Structure, Channel Coding and Modulation for 11/12
GHz Satellite Services” and was adopted in 1994.
14.1 The DVB-S System Parameters
The modulation method selected for DVB-S was quadrature phase shift
keying (QPSK). For some time, the use of 8PSK modulation instead of
QPSK has also been considered in order to increase the data rate. In principle, satellite transmission requires a modulation method which is relatively resistent to noise and, at the same time, is capable of handling severe
nonlinearities. Due to the long distance of 36000 km between the satellite
and the receiving antenna, satellite transmission is subject to severe noise
interference caused by the free-space attenuation of about 205 dB. The active element in a satellite transponder is a traveling wave tube amplifier
(TWA) which exhibits severe nonlinearities in its modulation characteristic. These nonlinearities cannot be compensated for since this would be associated with a decrease in energy efficiency. During daylight, the solar
cells provide power both to the electronics of the satellite and to the batteries. During the night, the energy for the electronics comes exclusively
from the backup batteries. If there are large amounts of nonlinearity, therefore, there must not be any information content in the amplitude of a
modulation signal.
Both in QPSK and in 8PSK, the information content is in the phase
alone. In the satellite transmission of analog TV, too, frequency modulation was used instead of amplitude modulation for this reason.
A satellite channel of a direct broadcasting satellite usually has a width
of 26 to 36 MHz (e.g. 33 MHz in ASTRA 1F, 36 MHz in EUTELSAT Hot
Bird 2), the uplink is in the 14 ... 19 GHz band and the downlink is 11 ...
13 GHz. It is then necessary to select a symbol rate which produces a spectrum which is narrower than the transponder bandwidth. The symbol rate
selected is, therefore, often 27.5 MS/s. As QPSK allows the transmission
of 2 bits per symbol, a gross data rate of 55 Mbit/s is obtained.
gross_data_rate = 2 bits/symbol • 27.5 Megasymbols/s = 55 Mbit/s;
14.1 The DVB-S System Parameters
265
However, the MPEG-2 transport stream now to be sent to the satellite as
QPSK-modulated signal must first be provided with error protection before
being fed into the actual modulator. In DVB-S, two error protection
mechanisms are used, namely a Reed-Solomon block code which is coupled with convolutional (trellis) coding. In the case of the Reed-Solomon
error protection, already known from the audio CD, the data are assembled
into packets of a certain length and these are provided with a special
checksum of a particular length. This checksum allows not only errors to
be detected but also a certain number of errors to be corrected. The number
of errors which can be corrected is a direct function of the length of the
checksum. In Reed-Solomon, the number of repairable errors always corresponds to exactly one half of the error protection bytes (checksum).
x 204/188
Input data rate
Inv. sync.
FEC1/
outer
coder
x2
FEC2/
inner
coder
x (1.5-code_rate)
= Output data rate
[2.17...(1.63)...1.36]
Basisband
interf.
Sync
invers.
Energy
disp.
ReedSolom.
enc.
Conv.
interleaver
Conv.
coder
Puncturing
I
TS in
Coded
data
Q
Synchronization
Code rate
1/2...(3/4)...7/8
same as DVB-C
Fig. 14.5. Forward error correction (FEC) in DVB-S and DVB-T. DVB-S modulator Part 1
It is possible then to always consider exactly one transport stream packet as one data block and to protect this block with Reed Solomon error
protection. An MPEG-2 transport stream packet has a length of 188 bytes.
In DVB-S, it is expanded by 16 bytes Reed Solomon forward error correction to form a data packet of 204 bytes length. This is called RS (204,188)
coding. At the receiving end, up to 8 errors can be corrected in this 204byte-long packet. The position of this/these error/s is not specified. If there
are more than 8 errors in a packet, this will still be reliably detected but it
is no longer possible to correct these errors. The transport stream packet is
then flagged as errored by means of the transport error indicator in the
transport stream header. This packet must then be discarded by the
266
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
MPEG-2 decoder. The Reed Solomon forward error correction reduces the
data rate:
net_data_rate Reed-Solomon = gross_data_rate • 188/204
= 55 Mbit/s • 188/204 =
= 50.69 Mbit/s;
However, simple error protection would not be sufficient for satellite
transmission which is why further error protection in the form of convolutional coding is inserted after the Reed Solomon forward error correction.
This further expands the data stream. This expansion is made controllable
by means of a parameter, the code rate. The code rate describes the ratio
between the input data rate and the output data rate of this second error
correction block:
code _ rate =
input _ data _ rate
;
output _ data _ rate
In DVB-S, the code rate can be selected within the range of 1/2, 3/4,
2/3,...7/8. If the code rate is 1/2, the data stream is expanded by a factor of
2. The error protection is now maximum and the net data rate has dropped
to a minimum. A code rate of 7/8 provides only a minimum overhead but
also only a minimum of error protection. The available net data rate is then
at a maximum. A good compromise is usually a code rate of 3/4. The code
rate can then be used to control the error protection and thus, as a reciprocal of this, also the net data rate.
The net data rate in DVB-S with a code rate of 3/4, after convolutional
coding, is then:
net_data_rate DVB-S 3/4 = code_rate • net_data_rate Reed-Solomon
= 3/4 • 50.69 Mbit/s
= 38.01 Mbit/s;
14.2 The DVB-S Modulator
The following description deals with all component parts of a DVB-S
modulator in detail. Since this part of the circuit is also found in a DVB-T
modulator, it is recommended to read this section also in conjunction with
the latter.
14.2 The DVB-S Modulator
267
The first stage of a DVB-S modulator (Fig. 14.5.) is the baseband interface. This is where the signal is synchronized with the MPEG-2 transport
stream. This MPEG-2 transport stream consists of packets with a constant
length of 188 bytes, consisting of 4 bytes header and 184 bytes payload.,
the header beginning with a sync byte. This has a constant value of 0x47
and follows at constant intervals of 188 bytes. In the baseband interface,
the signal is synchronized to this sync byte structure. Synchronization occurs within about 5 packets and all clock signals are derived from this.
Mapper
cos 2
X
+
FIR
X
LP
BP
X
LO2
90
LO1
Fig. 14.6. DVB-S modulator, part 2
2
3
4
7
0
1
0x47
0x47
0x47
0x47
0x47
0x47
0x47
0xB8
1
0xB8
0
0x47
Sync
byte
2
MPEG-2
TS
packet
Fig. 14.7. Sync byte inversion
In the next block, the energy dispersal unit, every eighth sync byte is
first inverted. I.e., 0x47 then becomes 0xB8 by bit inversion. The other 7
sync bytes between these remain unchanged. Using this sync byte inversion, additional timing stamps are then inserted into the data signal which
are certain long-time stamps over 8 packets, compared with the transport
268
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
stream structure. These time stamps are needed for resetting processes in
the energy dispersal block at the transmitting and receiving end. This, in
turn means that both the modulator or transmitter and the demodulator or
receiver receives this sequence of eight packets of the sync byte inversion
transparently in the transport stream and uses them to control certain processing steps. It may happen that relatively long sequences of zeroes or ones
occur purely accidentally in a data signal. However, these are unwanted
since they do not contain any clock information or cause discrete spectral
lines over a particular period. To eliminate them, virtually every digital
transmission method applies energy dispersal before the actual modulation.
Initializing bits
1
0
0
1
0
1
0
1
0
0
0
0
0
0
0
1 2
3
4
5
6
7
8
9 10 11 12 13 14 15
=1
&
Enable/disable
randomizing
=1
MPEG-2 data In
Randomized
data out /
transparent
sync out
Fig. 14.8. Energy dispersal stage (randomizer)
To achieve energy dispersal, a pseudo random binary sequence (PRBS)
(Fig. 14.8.) is first generated which, however, is restarted time and again in
a defined way. In DVB-S, the starting and resetting takes place whenever
a sync byte is inverted.
The data stream is then mixed with the pseudo random binary sequence
(PRBS) by means of an Exclusive OR operation which breaks up long sequences of ones or zeroes. If this energy-dispersed data stream is mixed
again with the same pseudo random binary sequence at the receiving end,
the dispersal is cancelled again.
The receiving end, therefore, contains the identical circuit, consisting of
a 15-stage shift register with feedback which is loaded in a defined way
with a start word whenever an inverted sync byte occurs. This means that
the two shift registers at the transmitting end and at the receiving end are
operating completely synchronously and are synchronized by the sequence
of 8 packets of the sync byte inversion block. This synchronization only
becomes possible because the sync bytes and the inverted sync bytes are
14.2 The DVB-S Modulator
269
passed through completely transparently and are not mixed with the pseudo random bit sequence.
Transmission
link
MPEG-2
TS
RS
DVB
mod.
MPEG-2
TS
DVB
demod.
RS
204 byte
4 byte
header
184 byte
payload
16 byte
RS FEC
188 byte
Fig. 14.9. Reed-Solomon coding
The next stage contains the outer coder (Fig. 14.5. and 14.9.), the ReedSolomon forward error correction. At this point, 16 bytes of error protection are added to the data packets which are still 188 bytes long but are
now energy-dispersed. The packets now have a length of 204 bytes which
makes it possible to correct up to 8 errors at the receiving end. If there are
more errors, the error protection fails and the packet is flagged as errored
by the demodulator by the transport error indicator in the transport stream
header being set to ‘one’.
Frequently, however, burst errors occur during a transmission. If this results in more than 8 errors in a packet protected by Reed-Solomon coding,
the block error protection will fail. The data are, therefore, interleaved, i.e.
distributed over a certain period of time in a further operating step.
Any burst errors present are then broken up in the de-interleaving (Fig.
14.10.) at the receiving end and are distributed over a number of transport
stream packets. It is then easier to correct these burst errors, which have
now become single errors, and no additional data overhead is required.
In DVB-S, the interleaving is done in a so-called Forney interleaver
(Fig. 14.11.) which is composed of two rotating switches and a number of
shift registers. This ensures that the data are scrambled, and thus distrib-
270
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
uted, as “unsystematically” as possible. Maximum interleaving is over 11
transport stream packets. The sync bytes and inverted sync bytes always
precisely follow a particular path. This means that the speed of rotation of
the switches corresponds to an exact multiple of the packet length and interleaver and de-interleaver are synchronous with the MPEG-2 transport
stream.
Burst error
Single
error
1 2 3 4 5 6
De-interleaving in the
receiver
4
2
5
1
3
6
Fig. 14.10. De-interleaving
The next stage of the modulator is the convolutional coder (trellis
coder). This stage represents the second, so-called inner error protection.
The convolutional coder has a relatively simple structure but understanding it is not quite as simple.
The convolutional coder consists of a 6-stage shift register and two signal paths in which the input signal is mixed with the content of the shift
register at certain tapping points. The input data stream is split into 3 data
streams. The data first run into the shift register where they influence the
upper and lower data stream of the convolutional coder by an Exclusive
OR operation lasting 6 clock cycles. This disperses the information of one
bit over 6 bits. At specific points both in the upper data branch and in the
lower data branch there are EXOR gates which mix the data streams with
the contents of the shift register. This provides two data streams at the output of the convolutional coder, each of which exhibits the same data rate as
the input signal. In addition, the data stream was only provided with a particular memory extending over 6 clock cycles. The total output data rate is
14.2 The DVB-S Modulator
271
then twice as high as the input data rate which corresponds to a code rate =
1/2. An overhead of 100% has now been added to the data signal.
I=12;
M=204/I=204/12=17
Sync path
Interleaver
De-interleaver
M
I
paths
M
1 step
per byte
2M
2M
8 bit
8 bit
8 bit
3M
3M
(I-2)M
(I-2)M
(I-1)M
Sync path
(I-1)M
Max. delay = M(I-1)I=
2244 byte=11 TS packets
n
= Shift register for 8 bit, n stages
Fig. 14.11. Forney interleaver and de-interleaver
Convolutional coder
rBit
+
+
+
out1
+
Puncturing
Shiftregister
register
Shift
rBit
+
Exor
+
+
rout>rBit
+
out2
rBit
Code rate =1/2, ... 7/8
Fig. 14.12. Convolutional coder in DVB-S and DVB-T
272
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
14.3 Convolutional Coding
Each convolutional coder (Fig. 14.12.) consists of stages with more or less
delay and with memory which, in practice, are implemented by using shift
registers. In DVB-S, and also in DVB-T, it was decided to use a six-stage
shift register with 5 taps each in the upper and lower signal path. The timedelayed bit streams taken from these taps are Exclusive-ORed with the undelayed bit stream and thus result in two output data streams, subjected to
a so-called convolution, each with the same data rate as the input data rate.
A convolution occurs whenever a signal “manipulates” itself, delayed in
time.
[]
[]
+
[]
[]
in[ ]
o[ ]
n[ ]
out1[ ]
+
o[ ]
n[ ]
[]
[]
+
out2[ ]
Fig. 14.13. Sample 2-stage convolutional coder
A digital filter (FIR) also performs a convolution. It would take too
much time to analyse the convolutional coder used in DVB-S and DVB-T
directly since, due to its six stages, it has a memory of 26 = 64. Reducing it,
therefore, to a sample encoder having only two stages we only need to
look at 22 = 4 states. The shift register can assume the internal states 00,
01, 10 and 11. To test the behaviour of the circuit arrangement it is then
necessary to feed a zero and a one into the shift register for each of these 4
states and then to analyse the resulting state and also to calculate the output
signals due to the Exclusive OR operations. If, e.g., a zero is fed into the
shift register which has a current content of 00, the resultant new content
will also be 00 since one zero is shifted out and at the same time a new
zero is shifted in. In the upper signal path, the two EXOR operations produce an overall result of 0 at the output. The same applies to the lower
signal path.
If a one is fed into the shift register with contents 00, the new state will
be 10 and a one is obtained as output signal in the upper signal path as well
as in the lower signal path. The other three states can be worked out in the
14.3 Convolutional Coding
273
same way by feeding in a one and a zero in each case. The results are
shown in Fig. 14.14. The total result of the analysis can be illustrated more
clearly in a state diagram (Fig. 14.15.) where the four internal states of the
shift register are entered in circles.
[0]
[0]
+
[1]
[0]
[0]
in[0]
o[0]
n[0]
in[1]
out2[0]
+
out1[1]
in[1]
[0]
+
out1[1]
[1]
o[0]
n[0]
in[1]
[0]
+
out1[0]
[1]
o[1]
n[0]
in[1]
[1]
+
o[1]
n[1]
out1[1]
[1]
o[1]
n[1]
[1]
[1]
out2[1]
out2[0]
+
[1]
o[1]
n[1]
[0]
[1]
[0]
+
[1]
[1]
in[0]
o[1]
n[0]
+
[1]
+
o[0]
n[1]
[1]
out2[1]
out1[0]
[1]
[1]
+
out2[1]
+
[0]
o[1]
n[0]
[0]
[0]
[1]
+
[1]
[0]
in[0]
o[0]
n[1]
+
[0]
+
o[1]
n[1]
[1]
out2[0]
out1[0]
+
[0]
[0]
+
out2[1]
[0]
+
[1]
o[0]
n[1]
[0]
[0]
+
[0]
o[1]
n[0]
o[0]
n[0]
[1]
[1]
in[0]
o[0]
n[1]
[1]
+
[1]
+
[0]
[0]
[0]
out1[1]
+
[0]
o[0]
n[0]
[0]
[1]
+
out1[0]
+
+
out2[0]
Fig. 14.14. States of the sample convolutional coder (o = old state, n = new state)
274
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
The least significant bit is entered on the right and the most significant
bit on the left which means that the shift register arrangement has to be
imagined upended. The arrows between these circles mark the possible
state transitions. The numbers next to the circles describe the respective
stimulus bit and the output bits of the arrangement, respectively. It can be
seen clearly that not all transitions between the individual states are possible. Thus, it is impossible, for instance, to pass directly from 00 to 11
without first passing, e.g. through the 01 state.
1/10
11
1/01
01
0/01
0/10
10
1/00
1/11
0/11
00
x/yy
x = input data
y = output data
(out1, out2)
zz
internal
state of
Shift register,
LSB right,
MSB left
0/00
Fig. 14.15. State diagram of the sample convolutional coder
Plotting the permitted state transitions against time results in a so-called
trellis diagram. Within the trellis diagram, it is only possible to move along
certain paths or branches and not all paths through the trellis are possible.
In many country regions, certain plants (fruit trees, wine) are planted to
grow along trellises on a wall. They are thus forced to grow in an orderly
way in accordance with a particular pattern by being fixed at certain points
on the wall. However, it happens sometimes that such a trellis point breaks
off due to bad weather, and the trellis is then in disarray. The existing pattern makes it possible, however, to find out where the branch must have
been and it can thus be fixed again. The same happens with our data
streams after the transmission where the convolutionally encoded data
streams can be forced out of the trellis due to bit errors caused, e.g. by noise. But since the history of the data streams, i.e. their course through the
trellis diagram is known, bit errors can be corrected on the basis of greatest
14.3 Convolutional Coding
275
probability by reconstructing the paths. This is precisely the principle of
operation of the so-called Viterbi decoder, named after its inventor. The
Viterbi decoder is virtually the counterpart of the convolutional decoder
and there is, therefore, no convolutional decoder. The Viterbi decoder is
also much more complex than the convolutional coder.
t0
00
01
0/00
1/11
t1
t2
0/00
t3
0/00
1/11
0/10
0/11
1/00
10
1/01
11
States
of
shift
register
time
Fig. 14.16. Trellis diagram
After the convolutional coding, the data stream is now inflated by a factor of 2. For example, 10 Mbit/s have now become 20 Mbit/s but the two
output data streams together now carry 100% overhead, i.e. error protection. On the other hand, this correspondingly lowers the net data rate available. This overhead, and thus also the error protection, can be controlled in
the puncturing unit (Fig. 14.17.), e.g. the data rate can be lowered again by
selectively omitting bits. The omitting, i.e. the puncturing, is done in accordance with an arrangement called the puncturing pattern, which is
known to the transmitter and the receiver.
This makes it possible to vary the code rate between 1/2 and 7/8. 1/2
means no puncturing, i.e. maximum error protection, and 7/8 means minimum error protection and a maximum net data rate follows correspondingly. At the receiving end, punctured bits are filled up with ‘Don’t Care’
bits and are treated like errors in the Viterbi decoder and thus reconstructed. Up to here the processing stages of DVB-S and DVB-T are 100%
alike. In the case of DVB-T, the two data streams are combined to form a
common data stream by alternately accessing the upper and lower punctured data stream. In DVB-S, the upper data stream and the lower data
stream in each case run directly into the mapper where the two data
276
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
streams are converted into the corresponding constellation of the QPSK
modulation.
X1
X1
Y1
Y1
X1 X2 X3 X4
X1 Y2 Y3
Y1 Y2 Y3 Y4
Y1 X3 Y4
X1 X2 X3
X1 Y2
Y1 Y2 Y3
Y1 X3
X1 X2 X3 X4 X5
X1 Y2 Y4
Y1 Y2 Y3 Y4 Y5
Y1 X3 Y5
X1 X2 X3 X4 X5 X6 X7
X1 Y2 Y4 Y6
Y1 Y2 Y3 Y4 Y5 Y6 Y7
Y1 Y3 X5 X7
1/2
2/3
3/4
5/6
7/8
Fig. 14.17. Puncturing in DVB-S
f
f
cos 2
cos 2
cos 2
cos 2
r= f/BN
f
f
2BN = BS
f
Fig. 14.18. Roll-off filtering
The mapping is followed by digital filtering so that the spectrum “rolls
off” gently towards the adjacent channels. This limits the bandwidth of the
signal and at the same time optimizes the eye pattern of the data signal. In
14.4 Signal Processing in the Satellite
277
DVB-S, the roll-off filtering is carried out with a roll-off factor of r = 0.35.
The signal rolls off with a root cosine squared shape within the frequency
band. The cosine squared shape of the spectrum actually required is only
produced by combining the transmitter output filter with the receiver filter
because both filters exhibit root cosine squared roll-off filtering. The rolloff factor describes the slope of the roll-off filtering and is defined as r =
Δf/fN. After the roll-off filtering, the signal is QPSK modulated in the IQ
modulator, upconverted to the actual satellite RF and then, after power
amplification, fed to the satellite antenna. It is then uplinked to the satellite
in the 14...17 GHz band.
14.4 Signal Processing in the Satellite
The geostationary direct broadcasting satellites located permanently above
the equator in an orbit of about 36000 km above the Earth’s surface receive the DVB-S signal coming from the uplink station and limit it first
with a band-pass filter. Since the uplink distance of more than 36000 km
results in a free-space loss of over 200 dB and, as a result, the useful signal
is correspondingly attenuated, the uplink antenna and the receiving antenna on the satellite must exhibit corresponding gains. In the satellite, the
DVB-S signal is converted to the downlink frequency in the 11...14 GHz
band and then amplified by means of a TWA (Travelling Wave tube Amplifier). These amplifiers are highly nonlinear and, in practice, can also not
be corrected due to the power budget in the satellite. During the day, the
satellite is supplied with energy by solar cells and this energy is stored in
batteries. During the night, the satellite is then supplied only from its batteries.
Before the signal is sent back to Earth, it is first filtered again in order to
suppress out-of-band components. The transmitting antenna of the satellite
has a certain pattern so that optimum coverage is obtained in the receiving
area to be covered on the ground. This results in a so-called footprint
within which the programs can be received. Because of the high free space
loss of about 200 dB due to the downlink distance of more than 36000 km,
the satellite transmitting antenna must exhibit a correspondingly high gain.
The transmitting power is about 60 … 80 W. The signal processing unit
for a satellite channel is called a transponder. Uplink and downlink are polarized, i.e. there are horizontally and vertically polarized channels. Polarization is used in order to be able to increase the number of channels.
278
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
14.5 The DVB-S Receiver
After the DVB-S signal coming from the satellite has again travelled along
its path of 36000 km and, therefore, been attenuated correspondingly by
200 dB and its power has been reduced further by atmospheric conditions
such as rain or snow, it arrives at the satellite receiving antenna and is focussed at the focal point of the dish. This is the precise point at which the
low noise block (LNB) is mounted. The LNB contains a waveguide with a
detector each for the horizontal and vertical polarization. Depending on
which plane of polarization has been selected it is either the signal from
the horizontal detector or that from the vertical detector which is switched
through. The plane of polarization is selected by selection of the amplitude
of the supply voltage to the LNB (14/18V). The received signal is then
amplified in a low-noise gallium arsenide amplifier and is then downconverted to the first satellite IF in the 900...2100 MHz band.
LNB
Dish
1st sat IF (950...2050 MHz)
DC 14/18V
22 kHz
DVB-S receiver
Fig. 14.19. Satellite receiver with LNB and receiver
Modern “universal” LNBs (suitable for receiving digital TV) contain
two local oscillators which output a carrier at 9.6 GHz and at 10.6 GHz
and the received signal is down-converted by being mixed either with the
9.75 GHz or with the 10.6 GHz depending on whether the received channel is in the upper or lower satellite frequency band. DVB-S channels are
usually in the upper band and the 10.6 GHz oscillator is then used.
The phrase “suitable for receiving digital TV” only refers to the presence of a 10.6 GHz oscillator and is thus misleading. The LNB is switched
between 9.75 and 10.6 GHz by means of a 22 kHz switching voltage
which is superimposed on the LNB supply voltage or not. The LNB is
supplied via the coaxial cable which distributes the satellite intermediate
frequency in the 900...2100 MHz band now output. During installation
14.5 The DVB-S Receiver
279
work, care should be taken, therefore, to deactivate the satellite receiver
since otherwise a possible short circuit could damage the voltage supply
for the LNB.
V
H
10.7...12.75 GHz
LNB noise figure: 0.6 ... 1 dB,
gain: appr. 50 dB
H
950...
2050 MHz
X
BP
V
9.75
GHz
LP
1st
Sat IF
10.6
GHz
Fig. 14.20. Outdoor unit - LNB
Carrier &
clock
recovery
1st
sat
IF
from
LNB
A
ZF1
ZF2
Covol.
deinterleaver
A
D
D
cos 2
Matched
filter
ReedSolomon
decoder
Demapper
Energy
dispersal
removal
Fig. 14.21. DVB-S receiver (without MPEG-2 decoder)
Viterbi
decoder
Base
band
interface
MPEG-2
TS
280
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
In the DVB-S receiver, the so-called DVB-S set-top box or decoder, the
signal undergoes a second down-conversion to a second satellite IF. This
down-conversion is performed with the aid of an IQ mixer which is fed by
an oscillator controlled by the carrier recovery circuit. After the IQ conversion, analog I and Q signals are again available. The I and Q signals are
then A/D converted and supplied to a matched filter in which the same root
cosine squared filtering process as at the transmitting end takes place with
a roll-off factor of 0.35. Together with the transmitter filter, this then results in the actual cosine squared roll-off filtering of the DVB-S signal.
The filtering process must be matched with respect to the roll-off factor at
the transmitting end and at the receiving end.
After the matched filter, the carrier and clock recovery circuit and the
demapper tap off their input signals. The demapper again generates a data
stream from which the first errors are removed in the Viterbi decoder. The
Viterbi decoder is the counterpart of the convolutional coder. The Viterbi
decoder must have knowledge of the code rate currently used. The decoder
must be informed of this code rate (1/2...2/3...7/8) by operator intervention.
The Viterbi decoder is followed by the convolutional de-interleaving
where any burst errors are broken up into individual errors. The bit errors
still present then are corrected in the Reed Solomon decoder. The transport
stream packets, which had an original length of 188 bytes, had been provided with 16 bytes error protection at the transmitting end. These can be
used at the receiving end for correcting up to 8 errors in the packet which
now has a length of 204 bytes. Burst errors, i.e. multiple errors in a packet,
should have been broken up by the preceding deinterleaving process.
However, if an error-protected TS packet with a length of 204 bytes contains more than 8 errors, the error protection will fail. The Transport Error
Indicator in the transport stream header is then set to “1” to flag this packet
as errored. The packet length is now 188 bytes again. TS packets flagged
as errored must not be used by the MPEG-2 decoder and error concealment must be applied.
After the Reed Solomon decoding, the energy dispersal is removed and
the inversion of the sync bytes is cancelled. During this process, the energy
dispersal unit is synchronized by this sequence of 8 packets of the sync
byte inversion. At the output of the following baseband interface, the
MPEG-2 transport stream is available again and is then supplied to an
MPEG-2 decoder.
Today, the entire DVB-S decoder after the A/D converters is located on
one chip which, in turn, is usually integrated in the satellite tuner. I.e., the
tuner, which is controlled via the I2C bus, has an F connector input for the
signal from the LNB and a parallel transport stream output.
14.6 Influences Affecting the Satellite Transmission Link
Nonlinearity
noise
IQ errors
DVB-S
mod.
281
Additive
with Gaussian
noise (AWGN)
DVB-S
rec.
Fig. 14.22. Influences affecting the satellite transmission link
14.6 Influences Affecting the Satellite Transmission Link
This section deals with the influences to be expected on the satellite transmission link (Fig. 14.22.) and it will be seen that these influences are
mainly restricted to noise. However, let us first begin with the modulator.
This can be assumed to be ideal up to the IQ modulator. The IQ modulator
can exhibit different gains in the I and Q branches, a phase error in the 90o
phase shifter and a lack of carrier suppression. There can also be noise effects and phase jitter coming from this circuit section. These problems can
be ignored, however, because of the rugged nature of the QPSK modulation and will normally never reach an order of magnitude which will noticeably affect the signal quality. In the satellite, the travelling wave tube
generates severe nonlinearities but these do not play a part, in practice. In
the region of the uplink and the downlink, however, where the DVB-S signal is attenuated severely by more than 200 dB due to the distance of
36000 km each way travelled by the signal, strong noise effects are experienced. It is these noise effects, the additive white gaussian noise (AWGN)
becoming superimposed on the signal, which form the only influence to be
discussed.
In the part following, the satellite downlink will be analysed by way of
an example with respect to the signal attenuation and the resultant noise effects.
The minimum carrier/noise ratios (C/N) necessary and the channel bit
error rate needed are known and predetermined from forward error correction (FEC, Reed-Solomon and convolutional coding) (Fig. 14.23.).
282
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
To gain an idea about the C/N to be expected, the levels on the satellite
downlink will now be considered.
from
satellite
MPEG-2
transport stream
DVB-Sfront
end
Viterbi
decoder
RS
decoder
BER<3•10-2 BER<2•10-4
CR
1/2
2/3
3/4
5/6
7/8
C/N
>4.1dB
>5.8dB
>6.8dB
>7.8dB
>8.4dB
MPEG-2
Decoder
BER<1•10-11
(QEF) =
1 error/hour
BER
Fig. 14.23. Minimum carrier/noise ratios necessary at the receiving end and bit error ratios
1E+00
1E-01
1E-02
1E-03
1E-04
1E-05
1E-06
1E-07
1E-08
1E-09
1E-10
1E-11
1E-12
1E-13
0
5
10
15
C/N[dB]
Fig. 14.24. Channel bit error ratio in DVB-S as a function of C/N
20
14.6 Influences Affecting the Satellite Transmission Link
283
A geostationary satellite is “parked” in an orbit of 35800 km above the
equator. This is the only orbit in which it can travel around the Earth synchronously. At 45o latitude, the distance from the Earth’s surface is then
d = Earth’s radius • sin(45°) + 35800 km = 6378 km • sin(45°)
+ 35800 km = 37938 km;
Transmitted power (e.g. Astra 1F):
Assumed transponder output power: 82 W =
19 dBW
Gain of the transmitting antenna
33 dB
Satellite EIRP (equivalent isotropic radiated power) 52 dBW
Free space attenuation:
Satellite-Earth distance = 37,938 km
Transmitting frequency = 12.1 GHz
Loss constant
Free space attenuation
91.6 dB
21.7 dB
92.4 dB
205.7 dB
Received power:
Satellite EIRP
Free space attenuation
Clear sky attenuation
Receiver directional error
Polarisation error
52.0 dBW
205.7 dB
0.3 dB
0.5 dB
0.2 dB
Received power at the antenna
Antenna gain
Received power
-154.7 dBW
37 dB
-117.7 dBW
Noise power at the receiver:
Boltzmann’s constant
Bandwidth = 33 MHz
Temperature 20 oC = 273K+20K = 293K
Noise figure of the LNB
Noise power
-228.6 dBW/K/Hz
74.4 dB
24.7 dB
1.0 dB
-128.5 dBW
Carrier/noise ratio C/N:
Received power C
Noise power N
C/N
-117.7 dBW
-128.5 dBW
10.8 dB
284
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
Thus, a C/N of about 10 dB can be expected in the example. Actual C/N
values can be expected between 9 … 12 dB.
The following equations form the basis for the C/N calculation:
Free space attenuation:
L[dB] = 92.4 + 20•log(f/GHz) + 20•log(d/km);
f = transmission frequency in GHz;
d = Transmitter-receiver distance in km;
Antenna gain of a parabolic antenna:
G[dB] = 20 + 20•log(D/m) + 20•log(f/GHz);
D = antenna diameter in m;
f = transmission frequency in GHz;
Noise power at the receiver input:
N[dBW] = -228.6 + 10•log(b/Hz) + 10•log((T/0C +273)) + F;
B = bandwidth in Hz;
T = temperature in 0C;
F = noise figure of the receiver in dB.
Fig. 14.23. shows the minimum C/N ratios as a function of the code rate
used. In addition, the pre-Viterbi, post-Viterbi (= pre-Reed-Solomon) and
post-Reed-Solomon bit error rates are plotted. A frequently used code rate
is 3/4. With a mimum C/N ratio of 6.8 dB, this results in a pre-Viterbi
channel bit error rate of 3-2. The post-Viterbi bit error rate is then 2-4 which
correponds to the limit at which the subsequent Reed-Solomon decoder
still delivers an output bit error rate of 1-11 or better. This approximately
corresponds to one error per day and is defined as quasi error-free (QEF).
At the same time, these conditions also almost correspond to the “fall off
the cliff” (or “brickwall effect”). Slightly more noise and the transmission
breaks down abruptly.
In the calculated example of the C/N to be expected on the satellite
transmission link, there is, therefore, still a margin of about 3 dB available
with a code rate of 3/4. The precise relationship between the channel bit error rate, i.e. the pre-Viterbi bit error rate, and the C/N ratio is shown in Fig.
14.24.
14.7 DVB-S2
285
14.7 DVB-S2
DVB-S was adopted in 1994, using QPSK as a modulation method and a
concatenated error protection system of Reed-Solomon FEC and convolution coding. In 1997, the DVB DSNG standard [ETS301210] was laid
down which was created for reporting purposes (DSNG = Digital Satellite
News Gathering). Live signals are transmitted by satellite, e.g. from outside broadcast vans at big public events to the studios. DVB DSNG already uses 8PSK and 16QAM. In 2003, new methods were defined, both
for direct broadcasting and for professional applications, as "DVB-S2" (s.
Fig. 14.25.) in ETSI document [ETS302307] .
Both QPSK, 8PSK (uniform and non-uniform) and 16APSK (16 amplitude phase shift keying) were provided as modulation methods, the latter
only being used in the professional field (DSNG). The error protection
used is completely new, e.g. LDPC (low density parity check). The standard is quite open for broadcasting, interactive services and DSNG.
MPEG-2
TS
Single
or
multiple
input
streams
(MPEG-2 TS
or generic)
Opt.
DVB-S
FEC
Input
Mode
interadaptation
face
Physical layer
signalling,
pilot insertion,
scrambling
CRC-8 encoding,
baseband signalling
merging
Physical
layer
framing
BCH,
LDPC,
bit interl.
Stream
adaptation
FEC
encoding
Padding,
baseband
scrambling
Code rate
Rolloff
filter
0.20, 0.25,
0.35
IQ
modulator
QPSK,
8PSK,
16APSK,
32APSK,
Hierarch. mod.
Mapper
Upconversion,
amplification,
uplink
Fig. 14.25. Block diagram of a DVB-S2 modulator
Data streams not conforming to the MPEG-2 transport stream can also
be transmitted and it is possible to transmit either one or a number of transport streams. This also applies to generic data streams which can also be
divided into packets. Fig. 14.25. shows the block diagram of a DVB-S2
286
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
modulator. At the input interface, the data stream or streams appear in the
form of an MPEG-2 transport stream or of generic data streams. Following
the mode and stream adaptation blocks, the data are fed to the FEC encoding block.
Q
10
00
I
11
Code rates:
1/4, 1/3, 2/5;
1/2; 3/5, 2/3,
3/4, 4/5, 5/6,
8/9, 9/10
01
Fig. 14.26. Gray-coded QPSK, absolute mapping (like DVB-S)
Q
100
110
000
010
001
Code rates:
I 3/5, 2/3, 3/4,
5/6, 8/9, 9/10
011
101
111
Fig. 14.27. Gray-coded 8PSK
In the downstream mapper, QPSK (Fig. 14.26.), 8PSK (Fig. 14.27.),
16APSK (Fig. 14.28.) or 32APSK (Fig. 14.29.) is then mapped. This is always absolute mapping, i.e. non-differential. Hierarchical modulation is a
special case. It is virtually backward compatible with the DVB-S standard,
making it possible to transmit a DVB-S stream and an additional DVB-S2
stream. In the hierarchical modulation mode (Fig. 14.30.), the constellation
can be interpreted in two different ways.
14.7 DVB-S2
287
The quadrant can be interpreted as a constellation point, gaining 2 bits
for the high priority path conforming to DVB-S. It is also possible, however, to look for the two discrete points in the quadrant, decoding a further
bit for the low priority path in the process. In this case, 3 bits per symbol
are transmitted. There is also hierarchical modulation in DVB-T. After the
mapping, the signal passes through the physical layer framing and roll-off
filtering stages and is then converted into the modulation signal proper in
the IQ modulator. The roll-off factor is 0.20, 0.25 or 0.35.
Q
1010
1000
0010
0000
0110
1110
1100
0100
0111
1111
1101
0101
0011
Code rates:
I 2/3, 3/4, 4/5,
5/6, 8/9, 9/10
0001
1011
1001
Fig. 14.28. 16APSK
Q
01101
11101
01001
01100
11001
00101
00001
00100
11100
00000
10100
10101
10110
10111
01000
10001
Code rates:
I ¾, 4/5, 5/6,
8/9, 9/10
10000 11000
11110
10010
01110
10011
00010
00110
00111
11111
01111
00011
01010
11011
01011
Fig. 14.29. 32 APSK
11010
288
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
The error protection (Fig. 14.31.) consists of a BCH (Bose-ChaudhuriHocquenghem) coder and an LDPC (low density parity check) encoder
followed by the bit interleaver. Possible code rates are 1/4 ... 9/10 and are
shown in the figures for the respective constellation diagrams (QPSK ...
32APSK).
Compared with DVB-S, the minimum C/N ratio necessary in DVB-S2
is much more dependent on the modulation method and can also be varied
by the code rate.
Q
10
quadrant
LP bit
1
1
00
quadrant
0
0
I
HP bit
0
0
11
quadrant
1
1
01
quadrant
Fig. 14.30. Hierarchical QPSK modulation
Table 14.1 shows the minimum C/N ratios from the DVB-S2 Standard
[ETS302307].
Table 14.1. Minimum C/N ratio necessary in DVB-S and DVB-S2
Mod.
CR
=1/3
-1.2
dB
-
CR
=2/5
0
dB
-
CR
=1/2
1
dB
-
16APSK -
-
-
-
CR
=3/5
2.2
dB
5.5
dB
-
32APSK -
-
-
-
-
QPSK
8PSK
CR
=1/4
-2.4
dB
-
CR
=2/3
3.1
dB
6.6
dB
9
dB
-
CR
=3/4
4
dB
7.9
dB
10.2
dB
12.6
dB
CR
=4/5
4.6
dB
-
CR
=5/6
5.2
dB
9.4
dB
11
11.6
dB
dB
13.6 14.3
dB
dB
CR
=8/9
6.2
dB
10.6
dB
12.9
dB
15.7
dB
CR
=9/10
6.5
dB
11
dB
13.1
dB
16.1
dB
Unlike DVB-S, DVB-S2 has a frame structure. There is an FEC frame
and a physical layer frame. An FEC frame firstly contains the data to be
transmitted which are either data which have an MPEG-2 transport stream
structure or data which are quite independent of this, so-called generic
data.
In front of this data field there an 80 bit long baseband header. The data
block with the baseband header is then padded to a length dependent on
14.7 DVB-S2
289
the selected code rate of the error protection and then provided with the
BCH code plus the LDPC code. Depending on the mode, an FEC frame
then has a length of 64800 or 16200 bits.
Code rates:
¼, 1/3, 2/5, ½, 3/5, 2/3,
¾, 4/5, 5/6, 8/9, 9/10
Baseband
scrambler
= part of
stream
adaptation
block
BCH
encoder
LDPC
encoder
Bit
interleaver
FEC encoding block
BCH=Bose-Chaudhuri-Hocquenghem
LDPC=low density parity check code
Fig. 14.31. DVB-S2 FEC block
80 bits
Baseband
header
Data from MPEG-2 TS or generic data
Data field
tBCH=8,10,12
DFL
Padding
kBCH
Outer FEC: BCH coding
16*tBCH bits
BCH
kLDPC = code rate * FEC frame
Inner FEC: LDPC coding
Code
rate
LDPC
64800 or 16200 bits FEC frame
Fig. 14.32. FEC frame in DVB-S2
The FEC frame is then divided into a physical layer frame composed of
n slots. The physical layer frame starts with the one-slot-long physical
290
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
layer header in which the carrier is /2-shift BPSK modulated. This is followed by slot 1 ... slot 16.
Slot 17 may be a pilot block if pilots are transmitted (optional). This is
followed by another 16 time slots with data and then, after slot 32, possibly
another pilot block etc.
Table 14.2. Coding parameters in DVB-S2
LDPC
code rate
¼
1/3
2/5
½
3/5
2/3
¾
4/5
5/6
8/9
9/10
¼
1/3
2/5
½
3/5
2/3
¾
4/5
5/6
8/9
9/10
kBCH
kLDPC
tBCH
FEC frame
16008
21408
25728
32208
38688
43040
48408
51648
53840
57472
58192
3072
5232
6312
7032
9552
10632
11712
12432
13152
14232
NA
16200
21600
25920
32400
38880
43200
48600
51840
54000
57600
58320
3240
5400
6480
7200
9720
10800
11880
12600
13320
14400
NA
12
12
12
12
12
10
12
12
10
8
8
12
12
12
12
12
12
12
12
12
12
NA
64800
64800
64800
64800
64800
64800
64800
64800
64800
64800
64800
16200
16200
16200
16200
16200
16200
16200
16200
16200
16200
16200
Table 14.3. Sample data rates in DVB-S and DVB-S2 with a symbol rate of 27.5
MS/s
Standard
Modulation
CR
Pilots
DVB-S
DVB-S2
DVB-S2
DVB-S2
DVB-S2
DVB-S2
DVB-S2
QPSK
QPSK
QPSK
QPSK
QPSK
8PSK
8PSK
¾
9/10
9/10
8/9
8/9
9/10
9/10
-On
Off
On
Off
On
Off
Net data rate
[Mbit/s]
38.01
48.016345
49.186827
47.421429
48.577408
72.005046
73.678193
14.7 DVB-S2
291
FEC frame
n slots
PL
header
Slot 1 Slot 2
…
Slot 1 Slot 2
… Slot 16
Slot n
90
symbols
per
slot
Pilot
block
Slot n
36
unmodulated
pilot
symbols
after time slot 16, 32, …
1 slot
PL header
/2-shift BPSK
Fig. 14.33. Physical layer frame in DVB-S2
90 symbols
PL header
Net_data_rate = symbol_rate / (FEC_frame/q + 90 +
ceil( (FEC_frame/q/90/16 – 1) ) * 36) * 36 pilot symbols
(FEC_frame * code_rate – (16 * tBCH) – 80);
no. of BCH
polynomes
80 bit DF header
FEC_frame = 64800 or 16200 bit;
q=2,3,4,5 bit/symbol; (QPSK, 8PSK, 16APSK, 32APSK)
ceil(A) rounds A to the nearest integer greater than or equal to A
code_rate = ¼ …9/10;
tBCH = 8,10 or 12;
Fig. 14.34. Formula for calculating the net data rate in DVB-S2
A slot has a length of 90 symbols. A pilot block has a length of 36 symbols. Table 14.1. shows the coding parameters of the FEC frame. The data
rates in DVB-S2 can be calculated by using the formula shown in Fig.
292
14 Transmitting Digital Television Signals by Satellite - DVB-S/S2
14.33. In practice (symbol rate of 27.5 MS/s), they are about 49 Mbit/s.
Examples of data rates are listed in Table 14.3.
The error protection used in DVB-S2 allows the efficiency to be increased enormously (by approx. 30%), approaching the Shannon limit
much more closely. This also requires much greater computing capacity
but this can be provided by today's technology. The error protection applied in DVB-S2 is now also used in the Chinese terrestrial digital TV
standard DTMB, and in the new DVB standards DVB-T2 and DVB-C2.
The other new DVB-SH standard for mobile TV, although largely derived
from DVB-S2 (and DVB-T), uses turbo coding for its error protection.
DVB-S2 is mainly intended for HDTV - High Definition Television.
Since 2005, some HD programs have been broadcast by this means in
Europe, among them Premiere (now Sky), Sat1 and Pro7 in Germany. Sat1
and Pro7 have stopped transmitting HD for the time being, at least until
2010. By then, it will also be possible to receive the public broadcasting
stations in HDTV format via satellite. The content will be MPEG-4 AVC
coded with a data rate of about 10 Mbit/s per HD program.
Bibliography: [ETS300421], [MÄUSL3], [MÄUSL4], [REIMERS],
[GRUNWALD], [FISCHER3], [EN301210], [ETS302307]
15 DVB-S/S2 Measuring Technology
15.1 Introduction
The satellite transmission of digital TV signals has now been discussed in
detail. The following sections will deal with DVB-S/S2 measuring technology. Satellite transmission is relatively rugged and, in principle, only
subject to noise effects (approx. 205 dB free space attenuation), and possible irradiation due to microwave links. There is also noise interference at
the first satellite IF due to cordless telephones (DECT).
The essential tests parameters on a DVB-S signal are:
•
•
•
•
•
•
Signal level
Bit error ratio
C/N (carrier/noise ratio)
Eb/N0
Modulation error ratio (MER)
Shoulder attenuation
The following are required for measurements on DVB-S signals:
• A modern spectrum analyzer (e.g. Rohde&Schwarz FSP, FSU)
• A professional DVB-S receiver with BER measurement or an antenna test instrument (e.g. Kathrein MSK33) or an MPEG analyzer
with corresponding RF interface (Rohde&Schwarz DVM)
• A DVB-S/S2 test transmitter for measurements on set-top boxes and
IDTV receivers (e.g. Rohde&Schwarz SFQ, SFU, SFL, SFE)
15.2 Measuring Bit Error Ratios
Due to the inner and outer error protection, there are three different bit error ratios in DVB-S:
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_15, © Springer-Verlag Berlin Heidelberg 2010
294
15 DVB-S/S2 Measuring Technology
•
•
•
Pre-Viterbi bit error ratio
Pre-Reed-Solomon bit error ratio
Post-Reed-Solomon bit error ratio
The most interesting bit error ratio providing the most information about
the transmission link is the pre- Viterbi bit error ratio. It can be measured
by reapplying the data stream after the Viterbi decoder to a convolutional
coder with the same configuration as that of the transmitter. If then the
data stream before the Viterbi decoder is compared with that after the convolutional coder (Fig. 15.1.) (taking into consideration the delay of the coder), the two are identical if there are no errors. A comparator for the I
branch and for the Q branch then determines the differences, and thus the
bit errors.
The bit errors counted are then related to the number of bits transmitted
in the corresponding period, resulting in the bit error ratio
BER = bit errors / transmitted bits;
The range of the pre-Viterbi bit error ratio is between 1•10-4 to 1•10-2.
This means that every ten-thousandth to hundredth bit is errored.
I
Viterbi
decoder
Data
Q
Convolutional
coder
I
Delay
Q
Comparator
BER
Fig. 15.1. Circuit for measuring the pre-Viterbi bit error ratio
The Viterbi decoder can only correct a proportion of the bit errors.
There is thus a residual bit error ratio remaining before the Reed-Solomon
decoder. Counting the correction processes of the Reed-Solomon decoder
15.2 Measuring Bit Error Ratios
295
and relating them to the number of bits transmitted within the corresponding period of time provides the pre-Reed-Solomon bit error ratio. The limit
pre-Reed-Solomon bit error ratio is about 2•10-4. Up to there, the Reed Solomon decoder can repair all errors. At the same time, however, the
transmission is "on the brink". A little bit more interference, e.g. due to too
much attenuation due to rain, and the transmission will break down and the
picture will start to show "blocking".
But the Reed-Solomon decoder, too, cannot correct all bit errors, resulting in errored transport stream packets which are then flagged in the TS
header (transport error indicator bit = 1). If the errored transport stream
packets are counted, the post-Reed-Solomon decoder bit error ratio can be
calculated.
Fig. 15.2. Spectrum of a DVB-S signal (10 dB/Div, 10 MHz/Div, Span 100 MHz)
If very low bit error ratios (e.g. less than 10-6) are measured, long measuring times in the range of minutes or hours must be selected to detect
these with any degree of accuracy. Since there is a direct relation between
bit error ratio and the carrier/noise ratio, this can be used for determining
the latter (see diagram in Section 14.6 “Influences affecting the satellite
transmission link”, Fig. 14.24.). Virtually every DVB-S chip or DVB-S receiver contains a circuit for determining the pre-Viterbi bit error ratio because this value can be used for aligning the satellite receiving antenna and
for determining the quality of reception. The circuit itself is not very complex. In most cases, DVB-S receivers display two bar graphs in their setup
296
15 DVB-S/S2 Measuring Technology
menu, one for the signal strength and one for signal quality. The latter is
derived from the bit error ratio.
Due to the altered error protection, the following bit error ratios are defined in DVB-S2:
•
•
•
Bit error ratios before LDPC,
Bit error ratios before BCH,
Bit error ratios after BCH.
Fig. 15.3. Spectrum of a DVB-S2 signal with rolloff=0.25
15.3 Measurements on DVB-S Signals using a Spectrum
Analyzer
A spectrum analyzer is quite suitable for measuring the power in the DVBS channel, at least in the uplink. Of course, it would also be quite simple to
use a thermal power meter but a spectrum analyzer can also be used for determining the carrier/noise ratio in the uplink. Firstly, however, the power
of the DVB-S/S2 signal will be determined using the spectrum analyzer. A
DVB-S signal has the appearance of noise and has a rather large crest factor. Because of its strong similarity with white Gaussian noise, its power is
measured exactly as in the case of noise.
To determine the carrier power, the spectrum analyzer is set as follows:
At the analyzer, a resolution bandwidth of 2 MHz and a video band width
of 3 to 10 times the resolution bandwidth (10 MHz) are selected. To
15.3 Measurements on DVB-S Signals using a Spectrum Analyzer
297
achieve some averaging, a slow operating time must be set (2000 ms).
These parameters are required because of the RMS detector used in the
spectrum analyzer. The following settings are used:
•
•
•
•
•
•
•
Center frequency at the center of the DVB-S channel,
Span at 100 MHz,
Resolution bandwidth at 2 MHz,
Video bandwidth at 10 MHz (because of RMS detector and log.
representation),
Detector RMS
Slow operating time (2000 ms)
Noise marker at channel center (results in C’ in dBm/Hz)
This results in a spectrum as shown in Fig. 15.2. The RMS detector calculates the power density of the signal in a window with a bandwidth of
1 Hz, the test window being continuously pushed over the frequency window to be measured (sweep range). In principle, first the RMS (root mean
square) value of the voltage is determined from all samples in the signal
window of 1 Hz bandwidth:
U RMS =
1
u12 + u 22 + u32 + ...;
N
From this, the power in this signal window is calculated with reference
to an impedance of 50 and converted into dBm. This is then the signal
power density in a window of 1 Hz bandwidth. The slower the selected
sweep time set, the more samples can be accommodated in this window
and the smoother and better averaged will be the test result.
Because of the noise-like signal, we use the noise marker measuring
power. The noise marker is set to band center for this purpose. The prerequisite is a flat channel but this can always be assumed to be the case in the
uplink. If the channel is not flat, other suitable measuring functions must
be used for measuring the channel power but these are dependent on the
spectrum analyzer.
The analyzer provides us with the value C’ as the noise power density at
the position of the noise marker in dBm/Hz, and the filter bandwidth and
the characteristics of the logarithmic amplifier of the analyzer are automatically taken into consideration. To relate the signal power density C’ to
the Nyquist bandwidth BN of the DVB-S signal, it is necessary to calculate
the signal power C as follows:
298
15 DVB-S/S2 Measuring Technology
C = C’+ 10log BN = C’+ 10log (symbol rate/Hz) dB;
[dBm]
The Nyquist bandwidth of the signal corresponds to the symbol rate of
the DVB-S signal.
Example:
Measured value of the noise marker:
-100 dBm/Hz
Correction value at 27.5 MS/s symbol rate: + 74.4 dB
Power in the DVB-S channel:
- 25.6 dBm
15.3.1 Approximate Determination of the Noise Power N
If it were possible to switch off the DVB-S signal without changing the
noise ratios in the channel, the noise marker at the center of the band
would now provide information on the noise ratios in the channel. However, this cannot be done in such a simple way. If not an exact measurement value, then at least a “good idea”, is obtained if the noise marker is
used on the shoulder of the DVB-S signal for measuring in close proximity
of the signal. This is because it can be assumed that the noise fringe in the
wanted band continues similarly to its appearance on the shoulder.
The value N’ of the noise power density is output by the spectrum analyzer. The noise power N in the channel with the bandwidth BK of the
DVB-S transmission channel is then calculated from the noise power density N’ as follows:
N=N’+10logBK =N’+10log(channel bandwidth/ Hz)dB; [dBm]
The channel bandwidth of the signal corresponds to the symbol rate of
the DVB-S signal (DVB measurement guidelines).
Example:
Measured value of the noise marker:
-120 dBm/Hz
Correction value at 27.5 MS/s symbol rate: + 74.4 dB
Noise power in the DVB-S channel:
- 45.6 dBm
The resultant C/N is:
C/N[dB] = C[dBm] - N[dBm];
15.3 Measurements on DVB-S Signals using a Spectrum Analyzer
299
In the example: C/N[dB] = -25.6 [dBm] - (-45.6 dBm) = 20 dB;
In fact, to measure the C/N in the downlink, the noise is measured in the
gaps between the individual channels. Other possibilities of measuring
C/N would be to use a suitable constellation analyzer for DVB-S/S2 (e.g.
the Rohde&Schwarz DVM with RF option) or via the detour of measuring
the bit error rate. Naturally, such an analyzer can also be used to measure
levels.
15.3.2 C/N, S/N and Eb/N0
The carrier-to-noise ratio C/N is an important value in assessing the quality
of the satellite transmission link. From the C/N, direct conclusion can be
drawn with respect to the bit error rate to be expected. The C/N is the result of the power radiated by the satellite (< ~100W), the antenna gain at
the transmitting and receiving end (size of the receiving antenna) and the
loss in the space between. The alignment of the satellite receiving antenna
and the noise figure of the LNB also play a role. DVB-S receivers output
the C/N value as an aid for aligning the receiving antenna.
C/N[dB] = 10log(PCarrier/PNoise) ;
In addition to the carrier-to-noise ratio, there is also the signal-to-noise
ratio:
S/N[dB] = 10log(PSignal/ PNoise);
The signal power is here the power of the signal after roll-off filtering.
Pnoise is the noise power within the Nyquist bandwidth (symbol rate).
The signal-to-noise ratio S/N is thus obtained from the carrier-to-noise
ratio as:
S/N[dB] = C/N[dB] + 10log (1-r/4);
where r is the roll-off factor (= 0.35 in DVB-S); i.e., in DVB-S:
S/N[dB] = C/N[dB] -0.3977 dB;
300
15 DVB-S/S2 Measuring Technology
Fig. 15.4. Constellation diagram of an undisturbed DVB-S signal with MER and
BER measurement [DVM]
15.3.3 Finding the EB/N0 Ratio
In DVB-S, the term EB/N0 is often mentioned. This is the energy per bit
with respect to the noise power density.
EB = energy per bit;
N0 = noise power density in dBm/Hz;
The EB/N0 can be calculated from the C/N ratio:
EB/N0 [dB] = C/N[dB] + 10log(188/204) - 10log(m) - 10log(code rate);
where
m = 2 for QPSK/DVB-S;
m = 4 for 16QAM;
6 for 64QAM and
8 for 256QAM, and
15.4 Modulation Error Ratio (MER)
301
the code rate is 1/2, 2/3, 3/4, 5/6, 7/8.
With a code rate of 3/4 as in the case of the usual QPSK modulation,
EB/N0 [dB]3/4 = C/N[dB] + 10log(188/204) - 10log(2) - 10log(3/4)
= C/N[dB] + 0.3547 dB - 3.0103 dB + 1.2494 dB
= C/N[dB] - 1.4062 dB;
Fig. 15.5. Constellation diagram of an interfered DVB-S signal [DVM]
15.4 Modulation Error Ratio (MER)
The modulation error ratio (MER) is an aggregate parameter in which all
the interfering signals affecting a digitally modulated signal are mapped.
Each interfering event, or hit, can be described by an error vector which
pushes the constellation point out of the ideal center of a decision field.
The ratio of the measured RMS value of the signal amplitude to the quadratic mean of the error vectors is then the MER. This is defined in detail in
the chapters on DVB-C and DVB-T measuring technology. In the case of
302
15 DVB-S/S2 Measuring Technology
DVB-S/S2, the MER is almost identical to the S/N value since there are
virtually only noise effects.
Fig. 15.6. Constellation diagram of an undisturbed DVB-S2 signal (8PSK) [DVM]
Fig. 15.7. DVB-S spectrum with “shoulders“
15.5 Measuring the Shoulder Attenuation
The DVB-S/2 signal within the wanted DVB-S/2 channel should be as flat
as possible, i.e. it should not exhibit any ripple or tilt. Toward the edges of
the channel, the DVB-S/2 spectrum drops off filtered with a smooth rolloff. There are, however, still signal components outside the actual wanted
15.6 DVB-S/S2 Receiver Test
303
band and these are called the ‘shoulders’ of the DVB-S/S2 signal. The aim
is to achieve the best possible shoulder attenuation of at least 40 dB. [ETS
300421] specifies a tolerance mask for the DVB-S signal spectrum but, in
principle, the satellite operator can define a particular tolerance mask for
the shoulder attenuation.
The signal spectrum is analyzed using a spectrum analyzer and simple
marker functions.
Fig. 15.8. Testing DVB receivers using an MPEG-2 generator (Rohde&Schwarz
DVRG) and a test transmitter (Rohde&Schwarz SFU): The MPEG-2 generator
(top) supplies the MPEG-2 transport stream with test contents and feeds the DVB
test transmitter (center) which, in turn, generates a DVB-conformal IQ modulated
RF signal for the DVB receiver (bottom). The video output signal from the DVB
receiver is displayed on the TV monitor (left).
15.6 DVB-S/S2 Receiver Test
The testing of DVB-S/S2 receivers (set-top boxes, s. Fig. 15.4., and
IDTVs) is accorded great significance. For these tests, DVB-S/S2 test
transmitters are used which can simulate the satellite transmission link and
304
15 DVB-S/S2 Measuring Technology
the modulation process. Such a test transmitter (e.g. Rohde&Schwarz TV
Test Transmitter SFQ or SFU) includes, in addition to the DVB-S/S2
modulator and upconverter, an add-on noise source and possibly even a
channel simulator. The test transmitter is fed with an MPEG-2 transport
stream from an MPEG-2 generator. The test transmitter then supplies a
DVB-S/S2 signal within the range of the first satellite IF (900 - 2100
MHz). This signal can be fed directly to the input of the DVB-S/S2 receiver. It is then possible to create various adverse signal conditions for the
DVB-S/S2 receiver by changing numerous parameters in the test transmitter. It is also possible to measure the bit error ratio as a function of the C/N
ratio. Such test transmitters are used both in the development and in the
production and quality assurance of DVB-S/S2 receivers.
Bibliography: [ETS300421], [ETR290], [REIMERS], [GRUNWALD],
[FISCHER3], [SFQ], [SFU], [DVM]
16 Broadband Cable Transmission According to
DVB-C
In many countries, good radio and TV coverage is provided via broadband
cable, especially in densely populated areas. These cable links have either
a bandwidth of about 400 MHz (approx. 50 - 450 MHz) or about 800 MHz
(approx. 50 - 860 MHz). In addition to the VHF and UHF band known
from terrestrial television, special channels are occupied. Analog television
programs can be received easily with a conventional TV set without additional complexities which is why this type of TV coverage is of such great
interest to many. The only obstacle in comparison with analog satellite TV
reception is the additional monthly line charge with which a satellite receiving system would pay for itself within one year in many cases. If the
satellite dish is large enough, the picture quality is often better than via
broadband cable since intermodulation products sometimes result in visible interference due to the multiple channel allocation in broadband cable.
The decision between cable and satellite reception simply depends on
the following considerations:
•
•
•
•
•
Convenience
Cabel reception charges
Single- and multiple-channel reception
Picture quality
Personal requirements/preferences
In many areas of Europe, purely terrestrial reception has dropped to below 10%. Naturally, this does not apply to the rest of the world.
Since about 1995, many cable networks are also carrying digital TV
signals according to the DVB-C standard and many others in the higher
frequency bands above about 300 MHz. This section is intended to explain
the methods for transmitting digital TV signals via broadband cable in
greater detail. The chosen transmission methods and parameters were selected with reference to the typical characteristics of a broadband cable.
Cable exhibits a much better signal/noise ratio than in satellite transmission and there are not many problems with reflections, either, all of which
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_16, © Springer-Verlag Berlin Heidelberg 2010
306
16 Broadband Cable Transmission According to DVB-C
permits digital modulation methods of higher quality to be used, from
64QAM (coax) to 256QAM (optical fiber). A broadband cable network
consists of the cable head end, of the cable distribution links consisting of
coaxial cables and cable amplifiers, of the ‘last mile’ from the distributor
to the house connection of the subscriber and of the subscriber’s in-house
network itself. Special technical terms such as ‘network level’ are deliberately avoided here since these terms can be specific to cable operators or
countries. The cable distribution links from the head end to the last distribution box can also be run as optical fibers. This broadband cable system
then distributes radio programs and analog and digital TV programs. More
and more frequently there are also return channel links in the frequency
band below about 65 MHz.
Fig. 16.1. 64QAM (left) and 256QAM (right)
16.1 The DVB-C Standard
Digital video broadcasting for cable applications had been specified in
about 1994 in the standard [ETS 300429]. This service has been available
in the cable networks since then, or shortly after. We will see that in the
DVB-C modulator, the MPEG-2 transport stream passes through almost
the same stages of conditioning as in the DVB-S satellite standard. It is
only the last stage of convolutional coding which is missing here: it is simply not needed because the medium of propagation is so much more robust. This is followed by the 16, 32, 64, 128 or 256QAM quadrature amplitude modulation. In coax cable systems, 64QAM is used virtually ways
whereas optical fibre networks frequently use 256QAM.
16.1 The DVB-C Standard
307
Considering then a conventional coax system with a channel spacing of
8 MHz. It normally uses a 64QAM-modulated carrier signal with a symbol
rate of, for example, 6.9 MS/s. The symbol rate must be lower than the
system bandwidth of 8 MHz in the present case. The modulated signal is
rolled off smoothly towards the channel edges with a roll-off factor of r =
0.15. Given 6.9 MS/s and 64 QAM (6 bits/symbol), a gross data rate of
Gross_data_rateDVB-C = 6 bits/Symbol • 6.9 MSymbols/s = 41.4 Mbit/s;
is obtained. In DVB-C, only Reed-Solomon error protection is used
which is the same as in DVB-S, i.e. RS(188,204). Thus, an MPEG-2 transport stream packet of 188 bytes length is provided with 16 bytes of error
protection, resulting in a total packet length of 204 bytes during the transmission.
The resultant net data rate is:
Net_data_rateDVB-C = Gross_data_rate • 188/204 = 38.15 Mbit/s;
Thus, a 36-MHz-wide satellite channel with a symbol rate of 27.5 MS/s
and a code rate of 3/4 has the same net data rate, i.e. the same transport capacity as this DVB-C channel with a width of only 8 MHz.
The following generally applies for DVB-C:
Net_data_rateDVB-C = ld(m) • symbol_rate • 188/204;
As well, however, the DVB-C channel has a much better signal/noise
ratio (S/N) with about >30 dB compared with about 10 dB in the case of
DVB-S.
The constellations provided in the DVB-C standard are 16QAM,
32QAM, 64QAM, 128QAM and 256QAM. According to DVB-C, the
spectrum is roll-off filtered with a roll-off factor of r = 0.15. The transmission method specified in DVB-C is also known as the international standard ITU-T J83A. There is also the parallel standard ITU-T J83B used in
North America, which will be described later, and ITU-T J83C which is
used in 6 MHz-wide channels in Japan. In principle, J83C has the same
structure as DVB-C but it uses a different roll-off factor for 128QAM (r =
0.18) and for 256QAM (r = 0.13). Everything else is identical. ITU-T
J83B, the method found in the US and in Canada, has a completely different FEC and is described in a separate Section.
308
16 Broadband Cable Transmission According to DVB-C
16.2 The DVB-C Modulator
The DVB-C modulator does not need to be described in so much detail
since most of the stages are completely identical with the DVB-S modulator. The modulator locks to the MPEG-2 transport stream fed to it at the
baseband interface and consisting of 188 byte-long transport stream packets. The TS packets consist of a 4 byte header, beginning with the sync
byte (0x47), followed by 184 bytes of payload. Following this, every sync
byte is inverted to 0xB8 to carry long-term time markers in the data stream
to the receiver for energy dispersal and its cancellation. This is followed by
the energy dispersal stage (or randomizer) proper, and then the ReedSolomon coder which adds 16 bytes of error protection to each 188 bytelong TS packets. The packets, which are then 204 bytes long, are then supplied to the Forney interleaver to make the data stream more resistant to error bursts. The error bursts are broken up by the cancellation of the interleaving in the DVB-C demodulator which makes it easier for the Reed
Solomon block decoder to correct errors.
MPEG-2
TS in
Base
band
interface
Syncinv. &
energydisp.
ReedSolomon
coder
Conv.
interleaver
47...862
MHz
RS(204, 188)
Clock
to
cable
I
Byte to
m-tuple
converter
Differ.
encoder
Q
Roll-off
filter
(FIR)
QAMmod.
IF
IF/RF
upconv.
&
ampl.
Fig. 16.2. DVB-C modulator
The error-protected data stream is then fed into the mapper where the
QAM quadrant must be differentially coded, in contrast to DVB-S and
DVB-T. This is because the carrier can only be recovered in multiples of
90o in the 64QAM demodulator and the DVB-C receiver can lock to any
multiples of 90o carrier phase. The mapper is followed by the quadrature
amplitude modulation which is now done digitally. Usually, 64QAM is selected for coaxial links and 256QAM for fiber-optical links. The signal is
roll-off filtered with a roll-off factor of r = 0.15. This gradual roll-off to-
16.3 The DVB-C Receiver
309
wards the band edges optimizes the eye opening of the modulated signal.
After power amplification, the signal is then injected into the broadband
cable system.
RF from 47...862
MHz
cable
I
I
RF/IF
downconverter,
tuner
IF
SAW
QAM
demod.
Q
Matched
filter &
equalizer
Q
Different.
decoder
Carrier and clock recovery
Demapper
Convol.
deinterleaver
ReedSolomon
decoder
Energy
dispersal
and
sync inv.
removal
Baseband
interface
MPEG-2
TS
Fig. 16.3. DVB-C receiver
16.3 The DVB-C Receiver
The DVB-C receiver - set-top box or integrated - receives the DVB-C
channel in the 50 - 860 MHz band. The transmission has added effects due
to the transmission link such as noise, reflections and amplitude and group
delay distortion. These effects will be discussed later in a separate Section.
The first module of the DVB-C receiver is the cable tuner which is essentially identical with a tuner for analog television. The tuner converts the
8 MHz-wide DVB-C channel down to an IF with a band center at about
36 MHz. These 36 MHz also correspond to to the band center of an analog
TV IF channel according to ITU standard BG/Europe. Adjacent channel
components are suppressed by a downstream SAW filter which has a
bandwidth of exactly 8 MHz. Where 7 or 6 MHz channels are possible, the
filter must be replaced accordingly. This band-pass filtering to 8, 7 or 6
MHz is followed by further downconversion to a lower intermediate frequency in order to simplify the subsequent analog/digital conversion. Before the A/D conversion, however, all frequency components above half
the sampling rate must be removed by means of a low-pass filter. The sig-
310
16 Broadband Cable Transmission According to DVB-C
nal is then sampled at about 20 MHz with a resolution of 10 bits. The IF,
which is now digitized, is supplied to an IQ demodulator and then to a root
cosine squared matched filter operating digitally. In parallel with this, the
carrier and the clock are recovered. The recovered carrier with an uncertainty of multiples of 90 degrees is fed into the carrier input of the IQ demodulator. This is followed by a channel equalizer, partly combined with
the matched filter, a complex FIR filter in which it is attempted to correct
the channel distortion due to amplitude response and group delay errors.
This equalizer operates in accordance with the maximum likelihood principle, i.e. it is attempted to optimize the signal quality by “tweeking” digital “setscrews” which are the taps of the digital filter. The signal, thus optimized, passes into the demapper where the data stream is recovered.
This data stream will still have bit errors and, therefore, error protection is
added. Firstly, the interleaving is cancelled and error bursts are turned into
single errors. The following Reed Solomon decoder can eliminate up to 8
errors per 204-byte-long RS packet. The result is again transport stream
packets with a length of 188 bytes which, however, are still energydispersed. If there are more than 8 errors in a packet, they can no longer
be repaired and the transport error indicator in the TS header is then set to
‘one’. After the RS decoder, the energy dispersal and the inversion of every 8th sync byte are cancelled and the MPEG-2 transport stream is again
present at the physical baseband interface. In practice, all modules from
the A/D converter to the transport stream output are implemented in one
chip. The essential components in a DVB-C set-top box are the tuner, some discrete components, the DVB-C demodulator chip and the MPEG-2
decoder chip, all of which are controlled by a microprocessor.
16.4 Interference Effects on the DVB-C Transmission Link
Since, in practice, DVB-C modulators only use digital IQ modulators, IQ
errors such as amplitude imbalance, phase errors and carrier leakage can be
neglected today. These effects are simply no longer present in contrast to
first-generation transmission. The effects occurring during transmission
are essentially noise, intermodulation and cross-modulation interference
and echoes and amplitude and group delay effects. If a cable amplifier is
saturated and, at the same time, is occupied with a large number of channels, frequency conversion products are produced which will appear in the
useful signal range. Every amplifier, therefore, needs to be operated at the
correct operating point. It is, therefore, of importance that the levels on the
transmission link are correct. A too high level produces intermodulation in
16.4 Interference Effects on the DVB-C Transmission Link
311
the amplifiers, a too low level reduces the signal/noise ratio, both of which
result in noise. The levels in a house installation, for example, should be
adjusted in such a way that a maximum signal/noise ratio is obtained for
DVB-C. An amplifier which may be present is calibrated in such a way
that the signal/noise ratio is at the point of inversion at the most distant antenna socket. DVB-C signals are also very sensitive to amplitude and
group delay response.
Noise,
interferer,
echos,
amplitude
response,
group
delay
Intermodulation,
interferer,
noise
Cable
headend
DVB-C
receiver
IQ error of modulator
- IQ imbalance
- phase error
- carrier leakage,
noise,
phase jitter,
intermodulation,
interferer,
cross modulation
Fig. 16.4. Interference effects on the DVB-C transmission link
A slightly defective connecting line between the TV cable socket and
the DVB-C receiver is often sufficient to render correct reception impossible. Operation of a DVB-C transmission link which is still quasi error free
(QEF) requires a signal/noise ratio S/N of more than 26 dB for 64 QAM.
The channel bit error ratio, i.e. the bit error ratio before Reed Solomon, is
then 2•10-4. The Reed Solomon decoder then corrects errors up to a residual bit error ratio after Reed Solomon of 1•10-11. This corresponds to quasi
error free operation (1 error per hour) but is also close to the “brick wall”
(or “fall off the cliff”). A little more noise and the transmission will break
down abruptly. The S/N ratio required for the QEF case depends on the
degree of modulation. The higher the degree of quadrature amplitude
312
16 Broadband Cable Transmission According to DVB-C
modulation, the more sensitive the transmission system. Fig. 16.5. shows
the variation of the bit error ratio with respect to the S/N ratio for QPSK,
16QAM, 64QAM and 256QAM.
BER for 4 / 16 / 64 / 256 QAM
1 10 0
BER
1 10 1
1 10 2
ACTUAL C/N
1 10 3
END
1 10 4
NOISE MARGIN
1 10 5
1 10 6
1 10 7
1 10 8
1 10 9
256QAM
1 10 10
64QAM
1 10 11
16QAM
QPSK
1 10 12
4
6
8
10
12
14
16
18
20
22
24
26
28
30
32
34
36
38
40
C/N in dB
S/N [dB]
Fig. 16.5. Bit error ratio as a function of S/N in DVB-C [EFA]
Currently, the most widely used signals in coax networks are 64QAMmodulated signals. These require a S/N ratio of more than 26 dB which
then corresponds to operation close to the “brick wall” (or “fall-off-thecliff”).
16.4 Interference Effects on the DVB-C Transmission Link
MPEG-2
transport
stream
from
cable
S/N>26 dB
@ 64QAM
DVB-C
frontend
ReedSolomon
decoder
BER<2•10-4
MPEG-2
decoder
BER<1 • 10-11
QEF = errors/hour
Fig. 16.6. DVB-C Bit error ratios
Bibliography: [ETS300429], [ETR290], [EFA], [GRUNWALD],
[ITU-T J83]
313
17 Broadband Cable Transmission According to
ITU-T J83B (US)
In North America, a different standard is used for transmitting digital TV
signals over broadband cable which is ITU-T J83B.
"J83B" is a part of a document which describes a total of 4 system proposals for broadband cable standards:
•
•
•
•
System A corresponds fully to DVB-C (8 MHz bandwidth)
System B is the currently used US cable standard (6 MHz bandwidth), = J83B
System C is the Japanese cable standard (largely identical with
DVB-C, different roll-off, 6 MHz bandwidth)
System D is "ATSC for cable" (16VSB, 6 MHz bandwidth, not
in use)
In principle, J83B is comparable to J83A, C (Europe, Japan) but there
are great differences in detail, especially in the FEC. The channel bandwidth in J83B is 6 MHz as in J83C (Japan). The modulation methods used
are only 64QAM and 256QAM with a roll-off factor of r = 0.18 (64QAM)
and r = 0.12 (256QAM). The error protection (FEC) is much more elaborate than in J83A or C. This begins with the MPEG framing.
The sync byte in the MPEG-2 transport stream is replaced by a special
checksum which is also calculated continuously in parallel at the receiving
end as in ATM (asynchronous transfer mode) and is used as criterion for
synchronization if they agree.
J83B makes it possible to transmit both an MPEG-2 transport stream
and ATM. After replacing the sync byte with a CRC, this is followed by a
Reed Solomon block encoder RS(128,122) which, in contrast to J83A, is
not set up for the MPEG-2 block structure. In the RS encoder, 6 RS symbols are added to every 122 7-bit long symbols. This makes it possible to
repair 3 symbols within 128 symbols at the receiving end, forming a frame
of several RS(128,122) packets to which a 42-bit- or 40-bit-long sync trailer is added in which, among other things, the adjustable interleaver length
is signalled. The RS encoder is followed by an interleaver which condi-
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_17, © Springer-Verlag Berlin Heidelberg 2010
316
17 Broadband Cable Transmission According to ITU-T J83B (US)
conditions a data stream in order to prevent error bursts. A randomizer
provides for an advantageous spectral distribution and breaks up long sequences of zeroes and ones in the data stream. The last stage in the FEC is
a trellis encoder (convolutional coder) which inserts additional error protection and, naturally, overhead into the data stream. The data stream conditioned in this way is then 64QAM or 256QAM modulated and then
transmitted in the coaxial or fiber-optical broadband cable.
Frame
Checksum
CRC
instead of
0x47
187 byte
MPEG-2 TS packet
Sync
checksum
122
6
122
... Sync-trailer
122 symbols a 7 bit
6 RS symbols a 7 bit
RS(128,122); 3 symbols can be repaired
ReedSolomonencoder
Interleaver
Randomizer
I, J variable, signalling
via sync trailer
Not in use, if not MPEG-2-TS
I
Trellisencoder
6
Mapper
Q
QAM
mod.
IF
CR=14/15 bei 64QAM; r=0.18; 42 bit sync trailer
CR=19/20 bei 256QAM; r=0.12; 40 bit sync trailer
IF/RF
upconv.
&
ampl.
to
cable
6 MHz
bandwidth
Fig. 17.1. Block diagram of an ITU-T J83B modulator
In addition to J83A, B and C, there is also the J83D Standard described
in the same ITU document but this is not being used in practice. J83D corresponds to ATSC (discussed in Chapter 23), the only difference being that
16VSB modulation is proposed here instead of 8VSB.
17.1 J83B Transmission Parameters
317
17.1 J83B Transmission Parameters
The transmission parameters provided in J83B are:
•
•
64QAM, symbol rate = 5.05641 MS/s, r=0.18, net data rate =
26.970352 Mbit/s
256QAM, symbol rate = 5.360537 MS/s, r=0.12, net data rate =
38.810701 Mbit/s.
Fig. 17.2. shows the constellation diagrams used in J83B.
Fig. 17.2. Constellation diagrams in J83B (64QAM left and 256QAM right)
The gross data rates in J83B are calculated as follows:
Gross data rate = Symbolrate • bits/symbol;
64QAM: Gross data rate = 5.05641 MS/s • 6 bits/symbol
= 30.34 Mbit/s;
256QAM: Gross data rate = 5.360537 MS/s • 8 bits/symbol
= 42.88 Mbit/s;
Because the symbol rate is higher with 256QAM, a smaller roll-offfactor of r=0.12 is used there.
318
17 Broadband Cable Transmission According to ITU-T J83B (US)
17.2 J83B Baseband Input Signals
In contrast to DVB-C, J83B is not tied to the MPEG-2 transport stream as
input signal. J83B provides both MPEG-2 transport streams and, e.g.,
ATM data. There is no coupling between the J83B FEC layer and the input
data signal. Should there be an MPEG-2 transport stream, however, the
sync byte 0x47 is replaced by a checksum calculated over the entire transport stream packet (similar to the ATM case).
17.3 Forward Error Correction
The error protection in J83B consists of a Reed Solomon RS(128,122) error protection which is composed of 7-bits-long Reed Solomon symbols. It
is possible to repair 3 symbols per RS block. An FEC frame starts with a
sync trailer which has the following structure:
•
•
64QAM: 42-bit sync trailer, 0x752C0D6C (28 bits), 4-bit controlword, 10 reserved bits (set to zero)
256QAM: 40-bit sync trailer, 0x71E84DD4 (32 bits), 4-bit control
word, 4 reserved bits (set to zero).
The FEC frame in J83B consists of
•
•
60 RS-blocks of 128 RS symbols with a width of 7 bits with
64QAM
88 RS-blocks of 128 RS symbols with a width of 7 bits with
256QAM.
The Reed Solomon encoder is followed by the time interleaver (Fig.
17.3.). This has the task of breaking up burst errors into individual errors
in the deinterleaver in the receiver. Following that, the data stream is interleaved by mixing with a pseudo random sequence (Fig. 17.4.). The randomizer is reset during the sync trailer. The data stream is then fed to the
trellis coder. Trellis coding is a special type of convolutional coding. The
trellis coder has one input and N outputs, the number of outputs being
matched to the subsequent mapping or to the modulation, respectively. If
the selected modulation scheme allows the transmission of N bits per symbol, the trellis coder will have N outputs. Trellis coding can be traced back
to Gottfried Ungerböck (IBM, 1982) and was used for the first time in
telephone modems.
17.3 Forward Error Correction
Interleaver
De-interleaver
J
I
paths
319
J
1 step
per 7 bit
2J
7 bit
7 bit
7 bit
3J
2J
3J
(I-2)J
(I-2)J
(I-1)J
(I-1)J
Symbol delay
(I, J) = (128,1), (64,2), (32,4), (16,8), (8, 16)
(reduced interleaving modes)
I = 128, J = 1 to 8
(enhanced interleaving modes)
n
= Shift register for 7 bit, n stages
Fig. 17.3. Time interleaver
Randomized
data out
+
+
3
Data in
Reset of randomizer during sync trailer
Fig.17.4. Randomizer
The subsequent modulation is a combination of coherent and differential
modulation. Due to the N x 900 uncertainty in the QAM modulation, the
quadrant is mapped differentially. The trellis coder in J83B consists of a
parser which supplies 4 or 6 bits directly to the mapper, and a differential
precoder followed by convolutional coders which process the two bits contained in the quadrants. A total of 6 or 8 bits, respectively, are then
mapped. The input signal of the parser is formed by groups of 7 bits each
(a total of 28 or 38 bits, resp.). The parser is responsible for forming the
320
17 Broadband Cable Transmission According to ITU-T J83B (US)
groups and also alters their sequential order in accordance with a defined
arrangement.
17.4 Calculation of the Net Data Rate
In J83B, the net data rate is calculated with knowledge of the gross data
rate, the error protection used and the J83B frame structure. There is no
frame structure in DVB-S and DVB-C which is why the net data rate can
be calculated there in a relatively simple way. It is even more complex in
the case of standards like DVB-T and especially with the new DVB standards. The formula for calculating the net data rate with J83B is:
Net data rate = Gross data rate • f1 • f2 • f3;
where
f1 = 122/128 = 0.953125 = Reed Solomon factor;
f2 = frame_size_factor = f4/(f4+f5);
f3 = code_rate;
f4 = bits_per_frame_without_trailer;
f5 = bits_per_trailer;
With 64QAM, this results in:
f4 = 60 • 128 • 7 bits = 53760 bits;
f5 = 42 bits;
f2 = 53760/(53760 + 42) = 0.9992;
f3 = 14/15 = 0.93333;
Net data rate = 30.34 Mbit/s • 0.9992 • 0.93333 = 26.97 Mbit/s;
and with 256QAM:
f4 = 88 • 128 • 7 bits = 78848 bits;
f5 = 40 bits;
f2 = 78848/(78848 + 40) = 0.9995;
f3 = 19/20 = 0.95;
Net data rate = 42.88 Mbit/s • 0.953125 • 0.9995 = 38.81 Mbit/s;
17.6 Fall-off-the-Cliff
321
17.5 Roll-off Filtering
In J83B, two different roll-off factors are possible depending on the type of
modulation selected, which are:
•
•
with 64QAM, r=0.18
and with 256QAM, r=0.12.
Fig. 17.8. shows the spectra with 64QAM and 256QAM. The different
roll-off factors can be clearly seen.
Data
Trellis
coder
Mapper
IQ mod.
N
64/256QAM
Fig. 17.5. Principle of trellis coded modulation (TCM)
6/8 bit/symbol
I
1 bit Mapper
Conv.
coder
Conv.
coder
1 bit
Quadrant
Differential
precoder
Parser
Data
4/6 bits
IQ mod.
Q
Fig. 17.6. Trellis-coded modulation in J83B
17.6 Fall-off-the-Cliff
Due to the fact that J83B uses two error protection mechanisms, there are,
in principle, three possible bit error ratios in the receiver which are
322
17 Broadband Cable Transmission According to ITU-T J83B (US)
•
•
•
the bit error ratio before Viterbi,
the bit error ratio before Reed-Solomon
and the bit error ratio after Reed-Solomon.
RS symbols
in
(groups of 7 bits)
(28 / 38 bits)
5 x 4/6 bits
out
Parser
4 x 2 bits
out
Fig. 17.7. Parser
64QAM
r=0.18
256QAM
r=0.12
Fig. 17.8. Spectra of 64QAM and 256QAM
With trellis coding, the bit error ratio before Viterbi cannot be measured
technically because of ambiguities in the receiver in the Viterbi decoding.
With J83B, the transmission system is "at-the-cliff" (approx.)
•
•
64QAM with a C/N of 22 dB
256QAM with a C/N of 28 dB.
This corresponds to a bit error ratio before Reed-Solomon of
17.6 Fall-off-the-Cliff
1.4 • 10-4 (experimental values; not from the standard)
Bibliography: [ITUJ83], [EFA], [SFQ], [SFU], [ETL]
323
18 Measuring Digital TV Signals in the Broadband
Cable
In contrast to the measuring techniques used on digital TV signals transmitted via satellite, a wider range of measuring techniques is provided for
broadband cable testing and is also necessary. The influences acting on the
broadband cable signal, which can be modulated with up to 256QAM, are
more varied by far, and more critical than in the satellite domain. In this
section, the test instruments and measuring methods for measurements on
DVB-C and J83A, B, C signals will be discussed. A large amount of space
is reserved for the so-called constellation analysis of I/Q modulated signals
which is also encountered in DVB-T. The influences or parameters to be
considered in cable transmission are:
•
•
•
•
•
•
•
•
•
Signal level
C/N and S/N ratio
I/Q modulator errors
Interferers
Phase jitter
Echoes in the cable
Frequency response
Bit error ratio
Modulation error ratio and error vector magnitude
To be able to detect and evaluate these influences, the following test instruments are used:
•
•
•
An up-to-date spectrum analyzer
A test receiver with constellation analysis
A test transmitter with integrated noise generator and/or channel
simulator for stress testing DVB-C and J83A, B, C receivers
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_18, © Springer-Verlag Berlin Heidelberg 2010
326
18 Measuring Digital TV Signals in the Broadband Cable
18.1 DVB-C/J83A, B, C Test Receivers with Constellation
Analysis
The most important test instrument for measuring digital TV signals in
broadband cable networks is a DVB-C/J83A, B, C test receiver with an integrated constellation analyzer. Such a test receiver operates as follows:
The digital TV signal is received by a high-quality cable tuner which
converts it to IF. The TV channel to be received is then band limited to 8,
7 or 6 MHz by an SAW (surface acoustic wave) filter, thus suppressing adjacent channels. The TV channel is then usually down-converted to an
even lower 2nd IF in order to be able to use more inexpensive and better
A/D converters. The IF signal, which has been filtered with an antialiasing-type low-pass filter, is then sampled with an A/D converter and
demodulated in the DVB-C/J83A, B, C demodulator. During this process,
a signal processor accesses the demodulator at the I/Q level and detects the
constellation points as hit frequencies in the I and Q direction in the decision fields of the QAM constellation diagram. This provides frequency distributions (‘clouds’) around the individual constellation points - there are
64QAM clouds in the case of 64QAM. The individual QAM parameters
are then determined by mathematical analyses of the frequency distributions. In addition, the constellation diagram itself is displayed graphically
and can then be assessed visually. The signal is then also demodulated to
become the MPEG-2 transport stream which can be supplied to an MPEG2 test decoder for further analyses.
IF1
RF
47...
862
MHz
RF/IF
conv.,
tuner
IF2
SAW
filter
Mixer
Antialias.
LPF
X
A
DVB-C
dem.
D
I
MPEG-2
TS
Q
DSP
Noise
gen.
Display
Fig. 18.1. Block diagram of a DVB-C/J83A, B, C test receiver with constellation
analyzer
18.1 DVB-C/J83A, B, C Test Receivers with Constellation Analysis
327
If the correct DVB-C or J83A, B, C signal is present at the test receiver
and all settings at the receiver have been selected so that it can correctly
lock to the QAM signal, a constellation diagram with constellation points
of varying size (Fig. 18.2.) and the appearance of noise clouds is obtained.
The size of the constellation points depends on the magnitude of the interference effects. The smaller the constellation points, the better the signal
quality.
Fig. 18.2. Correctly locked 64QAM constellation diagram with noise
Fig. 18.3. No QAM signal in the selected channel, only noise
For the receiver to lock up in DVB-C and J83A, B, C, the following adjustment parameters of the test receiver must be selected correctly:
•
•
Channel frequency: channel band center, approx. 47 to 862 MHz
Standard: DVB-C/J83A, J83B or J83C
328
18 Measuring Digital TV Signals in the Broadband Cable
•
•
•
•
•
Channel bandwidth: 6, 7, 8 MHz
QAM level: 16QAM, 32QAM, 64QAM, 128QAM, 256QAM
Symbol rate: approx. 2 to 7 MS/s
SAW filter: ON with adjacent channels occupied
Input attenuation control: to AUTO if provided
If there is simply no signal in the selected RF channel, the constellation
analyzer of the test receiver will display a completely noisy constellation
diagram (Fig. 18.3.) which exhibits no regular features whatever. It appears like a giant constellation point in the center of the display, but without sharp contours.
If accidentally an analog channel has been selected instead of, e.g. a
DVB-C channel, constellation diagrams like Lissajou figures are produced
which change continuously depending on the current content of the analog
TV channel. If, however, there is a QAM signal in the selected channel but
some of the receiver parameters have been selected wrongly (RF not exactly right, maybe the wrong symbol rate, wrong QAM level etc), a giant
constellation point with much sharper contours appears.
Fig. 18.4. Constellation diagram with the wrong carrier frequency and wrong
symbol Rate selected (completely unsynchronized)
If all parameters have been selected correctly and only the carrier frequency is still divergent, the constellation diagram will rotate. It is then
possible to see concentric circles.
An ideal, completely undistorted constellation diagram would show
only a single constellation point per decision field in the exact center of the
18.1 DVB-C/J83A, B, C Test Receivers with Constellation Analysis
329
fields (Fig. 18.6.). However, such a constellation diagram can only be generated in a simulation.
Fig. 18.5. QAM signal with the carrier out of sync
Fig. 18.6. Completely undistorted constellation diagram of an ideal undisturbed
64QAM signal
Fig. 18.7. 256QAM modulated DVB-C signal
330
18 Measuring Digital TV Signals in the Broadband Cable
Today, transmissions of up to 256 QAM are also encountered, mainly in
HFC (hybrid fiber coax) networks. Such a constellation diagram is shown
in Fig. 18.7.
18.2 Detecting Interference Effects Using Constellation
Analysis
In this section, the most important interference effects on the broadband
cable transmission link are discussed, and how they are analysed by using
the constellation diagram. The following influences can be seen and distinguished directly by means of constellation analysis:
•
•
•
•
Additive white Gaussian noise
Phase jitter
Interference
Modulator I/Q errors
Apart from assessing the constellation diagram purely visually, the following parameters can also be calculated directly from it:
•
•
•
•
•
•
•
•
Signal level
C/N and S/N ratio
Phase jitter
I/Q amplitude imbalance
I/Q phase error
Carrier suppression
Modulation error ratio (MER)
Error vector magnitude (EVM)
18.2.1 Additive White Gaussian Noise (AWGN)
One interference effect which affects all types of transmission links in the
same way is the so-called white Gaussian noise (AWGN). This effect can
emanate more or less from virtually any point along the transmission link.
In the constellation diagram, noise-like effects are recognized from the
constellation points which are now of varying size (Fig. 18.8.). To measure
the RMS value of the noise-like interferer, the hits in the individual areas
are counted within the individual constellation fields, i.e., the frequency
with which the center and the areas around it are hit at ever increasing dis-
18.2 Detecting Interference Effects Using Constellation Analysis
331
tance is detected. If these hits or counts within a constellation field were to
be displayed multi-dimensionally, a two-dimensional bell-shaped Gaussian
curve would be obtained (Fig. 18.9.).
Fig. 18.8. Constellation diagram of a 64QAM signal with additive noise
f(X 1 ,X 2 )
X1
X2
Fig. 18.9. Two-dimensional bell-shaped Gaussian curve [EFA]
This two-dimensional distribution will then be found similarly in every
constellation field. To find the RMS value of the noise effect, the standard
deviation is then simply calculated from these hit results. The standard deviation corresponds directly to the RMS value of the noise signal. Relating
this RMS value N to the amplitude of the QAM signal S allows the logarithmic signal/noise ratio S/N in dB to be calculated by taking the logarithm.
A normal frequency distribution can be described by the Gaussian normal distribution function as:
332
18 Measuring Digital TV Signals in the Broadband Cable
y( x) =
1
σ ⋅ 2π
e
− 0.5⋅(
x−µ
σ
)2
;
where σ = standard deviation, µ = mean value. The standard deviation
can be calculated from the counter results as:
∞
σ=
³ (x − µ)
2
f ( x )dx;
−∞
It can be seen clearly that, in principle, the formula for determining the
standard deviation corresponds to the mathematical relationship for calculating the RMS value.
It must be noted, however, that it is not only noise but also impulse
interferers or intermodulation and cross-modulation products which, due to
non-linearities on the transmission link, can cause comparable noise-cloudlike distortions in the constellation diagram and thus can not be distinguished from actual noise.
Fig. 18.10. Two-dimensional hit-rates in the 16QAM constellation diagram
[HOFMEISTER]
In principle, there are two definitions for the signal-to-noise level: the
signal-to-noise ratio S/N and the carrier-to-noise ratio C/N. Each can be
converted to the other one. The C/N ratio is always referred to the actual
bandwidth of the channel which is 6, 7 or 8 MHz on broadband cable networks. S/N refers to the conditions after roll-off filtering and to the actual
Nyquist bandwidth of the signal. The symbol rate of the signal should be
used for signal bandwidth and for noise bandwidth. But basically it is recommended to use the symbol rate also as reference bandwidth for the noise
18.2 Detecting Interference Effects Using Constellation Analysis
333
bandwidth in measuring C/N, providing an unambiguous definition for the
C/N ratio as proposed, e.g., in the DVB Measurement Guidelines
[ETR290].
The signal power S is obtained from the carrier power C, as
S = C[dBm] + 10 log(1 - r/4);
where r is the roll-off factor.
The logarithmic signal-to-noise ratio S/N is, therefore:
S/N[dB] = C/N[dB] + 10 log(1-r/4);
Example:
Channel bandwidth: 8 MHz
Symbol rate:
6.9 MS/s
Roll-off factor:
0.15
S/N[dB] = C/N[dB] + 10 log(1-0.15/4)
= C/N[dB] - 0.1660 dB;
Fig. 18.11. State diagram of a 64QAM signal with phase jitter
18.2.2 Phase Jitter
Phase jitter or phase noise in the QAM signal is caused by converters in
the transmission path or by the I/Q modulator itself. In the constellation
334
18 Measuring Digital TV Signals in the Broadband Cable
diagram, phase jitter produces smear distortion of greater or lesser magnitude (Fig. 18.11.). The constellation diagram ‘totters’ in rotation around
the center point.
To find the phase jitter, the smear distortions of the outermost constellation points are measured which is where the phase jitter has the greatest effect. Then the frequency distribution within the decision field is considered
along the circular path whose center point is at the origin of the state diagram. Again, the standard deviation which is still affected by additional
noise can be calculated here. This noise effect must then still be calculated
out.
18.2.3 Sinusoidal Interferer
A sinusoidal interferer (Fig. 18.12.) produces circular distortions of the
constellation points. These circles are the result of the interference vector
rotating around the center of the constellation point. The diameter of the
circles corresponds to the amplitude of the sinusoidal interferer.
Fig. 18.12. Effect of a sinusoidal interferer
18.2.4 Effects of the I/Q Modulator
In the first generation of DVB-C modulators, analog I/Q modulators were
used. Errors in the I/Q modulator (Fig. 18.13.) then resulted in I/Q errors
in the QAM-modulated signal. If, e.g., the I branch has a different gain
than the Q branch of the I/Q modulator, I/Q amplitude imbalance is pro-
18.2 Detecting Interference Effects Using Constellation Analysis
335
duced. If the 90o phase shifter in the carrier feed to the Q modulator is not
exactly 90 degrees, an I/Q phase error is produced. Lack of carrier suppression was an even more frequent problem. This is caused by carrier
cross-talk or by some DC component in the I or Q modulation signal. Today, the I/Q modulators in broadband cable are exclusively digital and the
problems of the I/Q modulator described are no longer relevant. They will
only be mentioned briefly here for the sake of completeness.
DC
I
re(t)
im(t)
Q
DC
I gain
Q gain
+
iqmod(t)
90 Phase
Fig. 18.13. IQ modulator with IQ errors
Fig. 18.14. Constellation diagram with IQ imbalance
18.2.4.1 I/Q Imbalance
In the case of an I/Q imbalance, the constellation diagram is squashed in
the I or Q direction resulting in a rectangular diagram instead of a square
one (Fig. 18.14.). The amplitude imbalance can be determined by measuring the lengths of the sides of the rectangle. It is defined as
AI = (v2/v1 - 1) • 100%;
336
18 Measuring Digital TV Signals in the Broadband Cable
where v1 is the gain in the I direction or I side of the rectangle, and
v2 is the gain in the Q direction or Q side of the rectangle.
18.2.4.2 I/Q Phase Error
An I/Q phase error (PE) (Fig. 18.15.) leads to a diamond-shaped constellation diagram. The phase error in the 90° phase shifter of the I/Q modulator
can be determined from the angles of the diamond in the constellation diagram. The acute angle then has a value of 90° - PE and the obtuse angle
has a value of 90° + PE.
Fig. 18.15. Constellation diagram with an IQ phase error
Fig. 18.16. Constellation diagram with insufficient carrier suppression
18.2.4.3 Carrier Suppression
In the case of insufficient carrier suppression (Fig. 18.16.), the constellation diagram is pushed away from the center in some direction. The degree
of carrier suppression can be calculated from the magnitude of the displacement.
18.2 Detecting Interference Effects Using Constellation Analysis
337
It is defined as:
CS = -10 log(PRC/PSIG); [dB]
Q
Resultant vector
Error vector
Center of
decision field
Ideal vector
I
Fig. 18.17. Error vector for determining the modulation error ratio (MER)
18.2.5 Modulation Error Ratio (MER)
All the interference effects on a digital TV signal in broadband cable networks previously explained cause the constellation points to exhibit deviations from their nominal position in the center of the decision fields. If the
deviations are too great, the decision thresholds will be exceeded and bit
errors are produced. However, the deviations from the decision field center
can also be considered to be measurement parameters for the size of any
interference quantity. Which is precisely the object of an artificial measurement parameter like the modulation error ratio (MER). The MER
measurement assumes that the actual hits in the constellation fields have
been pushed out of the center of the respective field by interference quantities (Fig. 18.17.). The interference quantities are given error vectors and
the error vector points from the center of the constellation field to the point
of the actual hit in the constellation field. Then the lengths of all these error vectors are measured against time in each constellation field and the
quadratic mean is formed or the maximum peak value is acquired in a time
window. The exact definition of MER can be found in DVB Measurement
Guidelines [ETR290].
338
18 Measuring Digital TV Signals in the Broadband Cable
MERPEAK =
MERRMS =
max(| error _ vector |)
⋅ 100%;
U RMS
1
N
N −1
¦ (| error _ vector |)
n =0
2
⋅ 100%;
U RMS
The reference URMS is here the RMS value of the QAM signal. Usually,
however, a logarithmic scale is used:
§ MER[%] ·
MERdB = 20 lg¨
¸;
© 100 ¹
[dB]
The MER value is thus an aggregate quantity which includes all possible individual errors and thus completely describes the performance of the
transmission link.
In principle:
MER [dB] ≤ S/N [dB];
Table 18.1. MER and EVM
QAM
MER>EVM
[%]
EVM=MER
EVM=
MER/1.342
EVM>MER
[%]
MER=EVM
EVM=
MER•1.342
32
EVM=
MER/1.304
EVM=
MER•1.304
64
EVM=
MER/1.527
EVM=
MER•1.527
128
EVM=
MER/1.440
EVM=
MER•1.440
256
EVM=
MER/1.627
EVM=
MER•1.627
4
16
MER>EVM
[dB]
|EVM|=MER
|EVM|=
MER
+2.56dB
|EVM|=
MER
+2.31dB
|EVM|=
MER
+3.68dB
|EVM|=
MER
+3.17dB
|EVM|=
MER
+4.23dB
MER>EVM
[dB]
MER=|EVM|
MER=
|EVM|
-2.56dB
MER=
|EVM|
-2.31dB
MER=
|EVM|
-3.68dB
MER=
|EVM|
-3.17dB
MER=
|EVM|
-4.23dB
18.3 Measuring the Bit Error Ratio (BER)
339
18.2.6 Error Vector Magnitude (EVM)
The error vector magnitude (EVM) is closely related to the modulation error ratio (MER), the only difference being the different reference used.
Whereas in the MER, the reference is the RMS value of the QAM signal, it
is the peak value of the QAM signal which is used as reference for the
EVM.
EVM and MER can be converted from one to the other with the aid of
table 18.1.
18.3 Measuring the Bit Error Ratio (BER)
In DVB-C and in J83A, C, the transmission is protected by Reed Solomon
forward error correction RS(204,188). Using 16 error protection bytes per
transport stream packet, this protection allows 8 single errors per TS
packet to be corrected at the receiving end. Counting the correction events
performed by the Reed Solomon decoder at the receiving end and assuming that these are attributable to single errors, and relating them to the incoming bitstream in the comparable period (a transport stream packet has
188•8 useful bits and a total of 204•8 bits), provides the bit error rate, a
value between 1•10-4 and 1•10-11.
However, not all the errors can be corrected by the Reed Solomon decoder. Errors in TS packets which are no longer correctable lead to errored
packets which are then marked by the transport error indicator in the
MPEG-2 transport stream header. Counting the non-correctable errors and
relating them to the corresponding data volume allows the post-Reed
Solomon bit error ratio to be calculated.
Thus, there are two bit error ratios in DVB-C and J83A, C:
•
•
Bit error ratio before Reed Solomon - the channel bit error ratio
Bit error ratio after Reed Solomon
The bit error ratio (BER) is defined as:
BER = bit errors / transmitted bits;
The bit error ratio has a fixed relationship to the signal/noise ratio if
only noise is involved. This relationship is shown in the figure below. In
addition, the figure includes the equivalent noise degradation (END) and
the noise margin for an example.
340
18 Measuring Digital TV Signals in the Broadband Cable
Equivalent Noise Degradation (END):
The equivalent noise degradation is a measure of the ‘insertion loss’ of
the entire system from the modulator via the cable link up to the demodulator. It specifies the deviation of the real S/N ratio from the ideal for a
BER of 1•10-4 in dB. In practice, values of around 1 dB are achieved.
Noise Margin:
The noise margin is the margin between the C/N ratio leading to a BER
of 1•10-4, and the C/N value of the cable system. When the C/N value is
measured in the cable, the channel bandwidth of the QAM signal is used as
the noise bandwidth.
Fig. 18.18. Spectrum of a DVB-C signal
18.4 Using a Spectrum Analyzer for Measuring DVB-C
Signals
A spectrum analyzer is a good instrument for measuring the power of the
DVB-C channel, at least at the modulation end. A DVB-C signal looks like
noise and has quite a high crest factor. Due to its similarity with white
Gaussian noise, the power is measured the same way as in a noise power
measurement.
18.4 Using a Spectrum Analyzer for Measuring DVB-C Signals
341
To find the DVB-C/J83A,B,C carrier power, the spectrum analyzer is
set up as follows:
At the analyzer, a resolution bandwidth of 300 kHz and a video bandwidth of 3 to 10 times the width of the resolution bandwidth (3 MHz) is selected. To achieve some averaging, a slow sweep time (2000 ms) must be
set. These parameters are required because we are using the RMS detector
of the spectrum analyzer. The following settings are then used:
•
•
•
•
•
•
•
Center frequency:
Span:
Resolution BW:
Video BW:
play)
Detector:
Sweep:
Noise marker:
center of the cable channel
10 MHz
300 MHz
3 MHz (due to RMS detector and log. disRMS
slow (2000 ms)
channel center (C’ in dBm/Hz)
To measure power, the noise marker is used because of the noise-like
signal. The noise marker is set to band center for this. The prerequisite is a
flat channel which, however, can always be assumed at the modulator. If
the channel is not flat, other suitable but analyzer-dependent measuring
functions must be used for measuring the channel power.
The analyzer provides the C’ value as noise power density at the position of the noise marker in dBm/Hz, automatically taking into consideration the filter bandwidth and the characteristics of the logarithmic amplifier of the analyzer. To relate the signal power density C’ to the Nyquist
bandwidth BN of the cable signal. the signal power C must be calculated as
follows:
C = C’ + 10log BN
= C’ + 10log(symbol_ rate / Hz) dB;
[dBm]
The Nyquist bandwidth of the signal corresponds to the symbol rate of
the cable signal.
Example:
Measurement value of the noise marker:
Correction value at 6.9 MS/s symbol rate:
Power in the channel:
-100.0 dBm/Hz
+ 68.4 dB
- 31.6 dBm
342
18 Measuring Digital TV Signals in the Broadband Cable
Finding the Noise Power N by Approximation:
If it were possible to switch off the DVB-C/J83A,B,C signal without
changing the noise conditions in the channel, the noise marker in the center
of the channel would provide information about the noise conditions in the
channel. However, this can not be done so easily. A ‘good idea’at least, if
not an exact measurement value, about the noise power in the channel is
obtained if the noise marker is used for measuring quite near to the signal
on the ‘shoulder’ of the DVB-C/J83A, B, C signal. This is because it can
be assumed that the noise fringe within the useful band continues in a similar way to how it appears on the shoulder.
The value N’ of the noise power density is output by the spectrum analyzer. To calculate the noise power in the channel having the bandwidth BK
of the cable transmission channel from the noise power density N’, the
noise power N must be found as follows:
N = N’ + 10log (BN)
= N’ + 10log (signal_bandwidth / Hz) dB;
[dBm]
The noise bandwidth is recommended by [ETR290] which is, the symbol rate.
Example:
Measurement value of the noise marker:
Correction value at 8 MHz bandwidth:
Noise power in the cable channel:
-140.0 dBm/Hz
+ 68.4 dB
- 71.6 dBm
The resultant C/N value is then:
C/N[dB] = C[dBm] - N[dBm];
i.e. C/N[dB] = -31.6 dBm -(-71.6 dBm)
= 40 dB;
18.5 Measuring the Shoulder Attenuation
Out-of-band components close to the wanted DVB-C/J83A,B,C band are
recognized from the ‘shoulders’ of the QAM signal (s. Fig. 18.18. and
18.19.). These shoulders should be suppressed as well as possible so as to
18.7 DVB-C/J83A,B,C Receiver Test
343
cause the least possible interference to the adjacent channels. This is defined as required minimum shoulder attenuation (e.g. 43 dB). The shoulder
attenuation is measured by using simple marker functions of the spectrum
analyzer.
Fig. 18.19. Shoulders on the DVB-C signal
18.6 Measuring the Ripple or Tilt in the Channel
The ripple in the amplitude response of a digital TV channel should be as
low as possible (less than 0.4 dBPP). Moreover, the tilt of this channel
should not be greater than this value, either. The ripple and tilt of the
channel can be measured by using a spectrum analyzer. The correction
data of the channel equalizer in the test receiver can also be used for this
measurement. Some cable test receivers allow the channel frequency response to be calculated from this.
18.7 DVB-C/J83A,B,C Receiver Test
As in DVB-S and also in DVB-T, the testing of receivers (set-top boxes
and integrated) is very important. The test transmitters can simulate a cable
transmission link and the modulation process. Apart from the cable modulator and upconverter, such a test transmitter (e.g. Rohde&Schwarz TV
Test Transmitter SFQ, SFU) also contains an add-on noise source and possibly even a channel simulator. The test transmitter is fed with an MPEG-2
transport stream from an MPEG-2 generator. The output signal of the test
transmitter can be supplied directly to the input of the cable receiver. It is
then possible to generate various stress conditions for the receiver by altering numerous parameters. It is also possible to measure the bit error rate as
a function of the C/N ratio.
344
18 Measuring Digital TV Signals in the Broadband Cable
Fig. 18.20. Constellation analysis on a DVB-C signal from a test transmitter
(Rohde&Schwarz SFQ, bottom left) using a test transmitter (Rohde&Schwarz
EFA, top left): An MPEG-2 generator (Rohde&Schwarz DVRG, center left) supplies an MPEG-2 transport stream with test contents which is fed into the test
transmitter. The DVB-C receiver EFA displays the DVB-C Signal back into the
MPEG-2 transport stream which can then be decoded with an MPEG-2 test decoder (Rohde&Schwarz DVMD, center right). The picture also shows the video
analyzer VSA (bottom right), the TV monitor (top center) and a “601” analyzer
VCA (top right).
Bibliography: [ETR290], [EFA], [SFQ], [HOFMEISTER],
[ETS300429], [REIMERS], [GRUNWALD], [JAEGER], [FISCHER3],
[SFU]
19 Coded Orthogonal Frequency Division
Multiplex (COFDM)
Almost from the beginning of the electrical transmission of messages
about 100 years ago, single-carrier methods have been used for transmitting information. The message to be transmitted is impressed on a sinusoidal carrier by applying analog amplitude, frequency or phase modulation
techniques. Since the eighties, single-carrier transmission is more and
more by digital methods in the form of frequency shift keying (FSK) and
in many cases also by vector modulation (QPSK, QAM). The main applications for this are fax, modem, mobile radio, microwave links and satellite transmission and the transmission of data over broadband cables.
However, the characteristics of many transmission paths are such that single-carrier methods prove to be sensitive to interference, complex or inadequate. Since the days of Marconi and Hertz, however, it is precisely
these transmission links which are used most frequently. Today, every
child knows of transistor radios, television receivers and mobiles or the
simple walkie-talkies, all of which operate with a modulated carrier in a
terrestrial environment. And every car driver knows the effect of reception
of the radio program he is listening to suddenly ceasing when he stops at a
red light - he is in a ‘dead spot’. Due to multi-path reception, fading occurs
which is frequency- and location-selective. In terrestrial radio transmission, narrowband or wideband sinusoidal or impulse-type interferers must
also be expected which can adversely affect reception. Location, type and
orientation and mobility, i.e. movement, all play a role. This applies both
to radio and TV reception and to reception via mobile radios. Terrestrial
conditions of reception are the most difficult types of reception of all. This
similarly applies to the old two-wire line in the telecommunications field.
There can be echoes, crosstalk from other pairs, impulse interferers and
amplitude and group delay response. However, the demand for data links
with higher bit rates from PC to Internet is increasing more and more. The
usual single-carrier methods and also data transmission systems such as
ISDN are already reaching their limits. For many people, 64 kbit/s or 128
kbit/s with grouped channels in ISDN is not enough. In the terrestrial radio
link it is now the broadcasting services, which have always had a wide
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_19, © Springer-Verlag Berlin Heidelberg 2010
346
19 Coded Orthogonal Frequency Division Multiplex (COFDM)
bandwidth such as television with normally up to 8 MHz bandwidth,
which are ‘crying out’ for reliable digital transmission methods. Using a
multi-carrier method is one reliable approach to this. The information is
transmitted digitally not via one carrier but via many - in some cases thousands of subcarriers with multiple error protection and data interleaving.
These methods, which have been known since the seventies, are:
•
•
Coded Orthogonal frequency division multiplexing (COFDM),
Discrete multitone (DMT).
They are used in
•
•
•
•
•
•
•
•
Digital Audio Broadcasting (DAB),
Digital Video Broadcasting (DVB-T),
Asymmetrical Digital Subscriber Line (ADSL),
Transmission of data signals via power lines.
ISDB-T
DTMB
DVB-T2
DVB-C2
In this Section, the background, characteristics and generation of multicarrier modulation methods such as coded orthogonal frequency division
multiplexing (COFDM) or discrete multitone (DMT) are described.
The concept of multi-carrier modulation goes back to investigations in
the Bell Laboratories in the U.S. [CHANG] and to ideas in France in the
seventies. In those days, however, chips which were fast enough to implement these ideas were nowhere in sight. It wasn’t until many years later, at
the beginning of the nineties, that the concept was turned into reality and
applied for the first time in Digital Audio Broadcasting (DAB). Although
DAB cannot really be called an absolute marketing success, this is certainly not due to the technology but rather the inappropriate marketing (or
complete lack of it) and this, in principle, is attributable to industry and
politics. The technology itself is first class. Even today, it is difficult to
convince the consumer in many fields that this product or the other is better. It is certainly correct to leave many decisions to the consumer but he or
she must then have knowledge of the new possibilities and underlying
principles or even be able to purchase the new type of product. In the case
of DAB, this has only been possible since 2001 (availability of DAB receivers) which is unfortunate for this excellent method of transmitting audio virtually in CD quality via terrestrial channels. The situation is differ-
19.1 Why Multi-Carrier?
347
ent in the case of ADSL in telecommunications where ADSL is increasingly accepted and demanded for Internet access because of its speed, and
in the case of DVB-T which is spreading in countries where it is politically
promoted and prescribed as an appropriate technology. If applied correctly,
DAB and DVB-T make a good contribution to the conservation of energy
and frequencies whilst at the same time delivering better performance.
Fig. 19.1. The terrestrial radio channel
19.1 Why Multi-Carrier?
Multi-carrier methods belong to the most complicated transmission methods of all and are in no way inferior to the code division multiple access
(CDMA) methods. But why this complexity? The reason is simple: the
transmission medium is an extremely difficult medium (Fig. 19.1.) to deal
with.
The terrestrial transmission medium involves
•
•
terrestrial transmission paths,
difficult line-associated transmission conditions.
The terrestrial transmission paths, in particular, exhibit the following
characteristic features:
•
•
Multipath reception via various echo paths caused by reflections
from buildings, mountains, trees, vehicles;
Additive white Gaussian noise (AWGN);
348
•
•
19 Coded Orthogonal Frequency Division Multiplex (COFDM)
Narrow-band or wide-band interference sources caused by internal
combustion engines, streetcars or other radio sources;
Doppler effect, i.e. frequency shift in mobile reception.
Multipath reception leads to location- and frequency-selective fading
phenomena (Fig. 19.2.), an effect known as “red-light effect” in car radios.
The car stops at a red stop light and radio reception ceases. If one were to
select another station or move the car slightly forward, reception would be
restored. If information is transmitted by only one discrete carrier precisely
at one particular frequency, echoes will cause cancellations of the received
signal at particular locations at exactly this frequency. This effect is a function of the frequency, the intensity of the echo and the echo delay.
A(f)
f
Fig. 19.2. Transfer function of a radio channel with multipath reception, frequency-selective Fading
If high data rates of digital signals are transmitted by vector modulated
(I/Q modulated) carriers, they will exhibit a bandwidth which corresponds
to the symbol rate.
The available bandwidth is usually specified. The symbol rate is obtained from the type of modulation and the data rate. However, singlecarrier methods have a relatively high symbol rate, often within a range of
more than 1 MS/s up to 30 MS/s. This leads to very short symbol periods
of 1 µs and shorter (inverse of the symbol rate). However, echo delays can
easily be within a range of up to 50 µs or more in terrestrial transmission
channels. Such echoes would lead to inter-symbol interference between
adjacent symbols or even far distant symbols and render transmission more
or less impossible. An obvious trick would now be to make the symbol period as long as possible in order to minimize inter-symbol interference and,
19.1 Why Multi-Carrier?
349
in addition, pauses could be inserted between the symbols, so-called guard
intervals.
Symbol Symbol Symbol Symbol Symbol Symbol
Path 1
n
n+1
n+2
n+3
n+4
n+5
+
Symbol Symbol Symbol Symbol Symbol Symbol
Path 2
n
n+1
n+2
n+3
n+4
n+5
=
t
Path 1+2
Intersymbol interference
t = echo delay time
Fig. 19.3. Inter-symbol interference / intersymbol crosstalk with multipath reception
A(f)
f
Fig. 19.4. COFDM: multicarrier in radio channel with fading
However, there is still the problem of the location- and frequencyselective fading phenomena. If then the information is not transmitted via a
single carrier but is distributed over many, up to thousands of subcarriers
and a corresponding overall error protection is built in, the available channel bandwidth remaining constant, individual carriers or carrier bands will
be affected by the fading, but not all of them.
At the receiving end, sufficient error-free information could then be recovered from the relatively undisturbed carriers to be able to reconstruct an
error-free output data stream by means of the error protection measures
350
19 Coded Orthogonal Frequency Division Multiplex (COFDM)
taken. If, however, many thousands of subcarriers are used instead of one
carrier, the symbol rate is reduced by the factor of the number of subcarriers and the symbols are correspondingly lengthened several thousand times
up to a millisecond. The fading problem is solved and, at the same time,
the problem of inter-symbol interference is also solved due to the longer
symbols and the appropriate pauses between them.
A multi-carrier method is born and is called Coded Orthogonal Frequency Division Multiplex (COFDM). It is now only necessary to see that
the many adjacent carriers do not interfere with one another, i.e. are orthogonal to one another.
19.2 What is COFDM?
Orthogonal frequency division multiplex is a multi-carrier method with up
to thousands of subcarriers, none of which interfere with each other because they are orthogonal to one another. The information to be transmitted is distributed interleaved to the many subcarriers, first having added
the appropriate error protection, resulting in coded orthogonal frequency
division multiplex (COFDM). Each of these subcarriers is vector modulated, i.e. QPSK, 16QAM and often up to 64QAM modulated.
COFDM
Coded
Orthogonal
Frequency division multiplex
COFDM is a composite of orthogonal (at right angles to one another or,
in other words, not interfering with one another) and frequency division
multiplex (division of the information into many subcarriers in the frequency domain).
In a transmission channel, information can be transmitted continuously
or in time slots. It is then possible to transport different messages in the
various time slots, e.g. data streams from different sources. This timeslot
method has long been applied, mainly in telephony for the transmission of
different calls on one line, one satellite channel or also one mobile radio
channel. The typical impulse-type interference caused by a mobile telephone conforming to the GSM standard with irradiation into stereo systems and TV sets has its origin in this timeslot method, also called time division multiple access (TDMA) in this case. However, it is also possible to
subdivide a transmission channel of a certain bandwidth in the frequency
19.2 What is COFDM?
351
domain, resulting in subchannels into each one of which a subcarrier can
be placed. Each subcarrier is modulated independently of the others and
carries its own information independently of the other subcarriers. Each of
these subcarriers can be vector modulated, i.e. QPSK, 16QAM and often
up to 64QAM modulated.
All subcarriers are spaced apart by a constant interval Δf. A communication channel can contain up to thousands of subcarriers, each of which
could carry the information from a source which would have nothing at all
to do with any of the others. However, it is also possible first to provide a
common data stream with error protection and then to divide it into the
many subcarriers. This is then frequency division multiplex (FDM). Thus,
in FDM, a common data stream is split up and transmitted in one channel,
not via a single carrier but via many, up to thousands of subcarriers, digitally vector modulated. Since these subcarriers are very close to one another, e.g. with a spacing of a few kHz, great care must be taken to see that
these subcarriers do not interfere with one another. The carriers must be
orthogonal to each other. The term orthogonal normally stands for ‘at 90
degrees to each other’ but in communications engineering quite generally
means signals which do not interfere with one another due to certain characteristics. When will adjacent carriers of an FDM system then influence
each other to a greater or lesser extent? Surprisingly, one has to start with a
rectangular pulse and its Fourier transform (Fig. 19.5.). A single rectangular pulse of duration Δt provides a sin(x)/x-shaped spectrum in the frequency domain, with nulls spaced apart by a constant Δf = 1/Δt in the spectrum. A single rectangular pulse exhibits a continuous spectrum, i.e.
instead of discrete spectral lines there is a continuous sin(x)/x-shaped
curve.
A(f)
Fourier
transform
sin(x)/x
t
f
t
f
Fig. 19.5. Fourier Transform of a rectangular pulse
352
19 Coded Orthogonal Frequency Division Multiplex (COFDM)
Varying the period Δt of the rectangular pulse varies the spacing Δf of
the nulls in the spectrum. If Δt is allowed to tend towards zero, the nulls in
the spectrum will tend towards infinity. This results in a Dirac pulse which
has an infinitely flat spectrum which contains all frequencies. If Δt tends
towards infinity, the nulls in the spectrum will tend towards zero. This results in a spectral line at zero frequency which is DC. All cases in between
simply correspond to
Δf = 1/Δt;
A train of rectangular pulses of period Tp and pulse width Δt also corresponds to this sin(x)/x-shaped variation but there are now only discrete
spectral lines spaced apart by fP = 1/TP which, however, conform to this
sin(x)/x-shaped variation.
What then is the relationship between the rectangular pulse and orthogonality? The carrier signals are sinusoidal. A sinewave signal of frequency fS = 1/TS results in a single spectral line at frequency fS and –fS in
the frequency domain. However, these sinusoidal carriers carry information by amplitude- and frequency-shift keying.
f
Carrier spacing
f
Channel bandwidth
Fig. 19.6. Coded Orthogonal Frequency Division Multiplex (COFDM)
I.e., these sinusoidal carrier signals do not extend continuously from
minus infinity to plus infinity but change their amplitude and phase after a
particular time Δt. Thus one can imagine a modulated carrier signal to be
composed of sinusoidal sections cut out rectangularly, so-called burst
19.2 What is COFDM?
353
packets. Mathematically, a convolution occurs in the frequency domain,
i.e. the spectra of the rectangular window pulse and of the sinewave become superimposed. In the frequency domain there is then a sin(x)/xshaped spectrum at the fs and -fs position instead of a discrete spectral line.
The nulls of the sin(x)/x spectrum are described by the length of the rectangular window Δt. The space between the nulls is Δf = 1/Δt.
If then many adjacent carriers are transmitted simultaneously, the
sin(x)/x-shaped tails produced by the bursty transmission will interfere
with the adjacent carriers.
However, this interference is minimized if the carrier spacing is selected
in such a way that a carrier peak always coincides with a null of the adjacent carriers. This is achieved by selecting the subcarrier spacing Δf to correspond to the inverse of the length of the rectangular window, i.e. the
burst period or symbol period. Such a burst packet with many and often
thousands of modulated subcarriers is called a COFDM symbol.
COFDM
symbol
duration t
Fig. 19.7. COFDM symbol
The following holds true as COFDM orthogonality condition (Fig.
19.8.):
Δf = 1/Δt;
where Δf is the subcarrier spacing and Δt is the symbol period.
If, for example, the symbol period of a COFDM system is known, the
subcarrier spacing can be inferred directly, and vice versa.
354
19 Coded Orthogonal Frequency Division Multiplex (COFDM)
In DVB-T, the following conditions apply for the so-called 2k and 8k
mode (Table 19.1.):
Table 19.1. COFDM modes in DVB-T
Mode
No. of subcarriers
Subcarrier spacing Δf
Symbol duration Δt
2k
2048
~ 4 kHz
1/Δf = ~ 250 µs
8k
8192
~ 1 kHz
~ 1 ms
Orthogonality condition: f = 1/ t
f
Fig. 19.8. Orthogonality condition in COFDM
19.3 Generating the COFDM Symbols
In COFDM, the information to be transmitted is first error protected, i.e. a
considerable overhead is added before this data stream consisting of payload and error protection is impressed on the large number of subcarriers.
Each one of these often thousands of subcarriers must then transmit a portion of this data stream. As in the single-carrier method, each subcarrier
requires mapping by which the QPSK, 16QAM or 64QAM is generated.
Each subcarrier is modulated independently of the others. In principle, a
COFDM modulator could be imagined to be composed of up to thousands
of QAM modulators, each with a mapper. Each modulator receives its
own, precisely derived carrier. All the modulation processes are synchro-
19.3 Generating the COFDM Symbols
355
Mapper
nized with one another in such a manner that in each case a common symbol is produced which has the exact length of Δt = 1/Δf. However, this
procedure is pure theory: in practice, its costs would be astronomical and it
would be unstable but, nevertheless, it serves to illustrate the principle of
COFDM.
X
+
COFDM
symbol
.....
+
Mapper
Data
X
FEC
X
X
+
Fig. 19.9. Theoretical block diagram of a COFDM modulator
I
re(t)
Re(f)
+
IFFT
Im(f)
ofdm(t)
im(t)
Q
90°
Fig. 19.10. Practical implementation of a COFDM modulator by IFFT
In reality, a COFDM symbol is generated by a multiple mapping process in which two tables are produced, followed by an Inverse Fast Fourier
Transform (IFFT). I.e., COFDM is simply the result of applying numerical
mathematics in a high-speed computer (Fig. 19.10.).
356
19 Coded Orthogonal Frequency Division Multiplex (COFDM)
The COFDM modulation process is as follows: The error-protected data
stream, thus provided with an overhead, is split up and divided as randomly as possible into a large number of up to thousands of substreams, a
process called multiplexing and interleaving. Each substream passes
packet by packet into a mapper which generates the description of the respective subvector, divided into real and imaginary parts. Two tables are
generated with up to many thousands of entries, resulting in a real-part table and an imaginary-part table. This results in the description of the time
domain section in the frequency domain. Each subcarrier, which is now
modulated, is described as x-axis section and y-axis section or, expressed
mathematically, as cosinusoidal and sinusoidal component, or real and
imaginary part. These two tables - real table and imaginary table - are now
the input signals for the next signal processing block, the Inverse Fast Fourier Transform (IFFT). After the IFFT, the symbol is now available in the
time domain. The signal shape has a purely random, stochastic, appearance
due to the many thousands of independently modulated subcarriers it contains. From experience, many people find it difficult to visualize how the
many carrier are produced which is why the process of modulation with
the aid of IFFT will now be described step by step.
re(t)
Re(f)
ofdm(t)
im(t)
Im(f)
Frequency
domain
Time
domain
Fig. 19.11. IFFT of a symmetric spectrum
The COFDM modulator shown in Fig. 19.10., which consists of the
IFFT block followed by a complex mixer (I/Q modulator), is fed one by
one with various real-part and imaginary-part tables in the frequency do-
19.3 Generating the COFDM Symbols
357
main after which the Inverse Fast Fourier Transform is performed and the
result is considered at the outputs re(t) and im(t) after the IFFT, i.e. in the
time domain, and after the complex mixer.
re(t)
Re(f)
ofdm(t)
im(t)
Im(f)
Frequency
domain
Time
domain
Fig. 19.12. IFFT of a asymmetric spectrum
This begins with a spectrum which is symmetric with respect to the
band center of the COFDM channel (Fig. 19.11.). simply consisting of carrier No.1 and N. After the IFFT, an output signal is produced at output
re(t) which is purely cosinusoidal. At output im(t), u(t) = 0V is present. A
purely real time-domain signal is expected since the spectrum meets the
conditions of symmetry required for this. After the I/Q modulator, an amplitude modulated signal with suppressed carrier is produced which is only
generated by the real time-domain component (see Fig. 19.11.).
If, however, e.g. the spectral line in the upper range of the band, that is
to say the carrier at N, is suppressed and only the component at carrier
No.1 is left, a complex time-domain signal is obtained due to the asymmetric spectrum (Fig, 19.12. and 19.13.). At output re(t) after the IFFT, a cosinusoidal signal with half the amplitude as before is now present. In addition, the IFFT now supplies a sinusoidal output signal of the same
frequency and the same amplitude at output im(t). This produces a complex signal in the time domain. If this, i.e. re(t) and im(t), is fed into the
following I/Q modulator, the modulation disappears, resulting in a single
sinusoidal oscillation converted into the carrier frequency band. A single-
358
19 Coded Orthogonal Frequency Division Multiplex (COFDM)
sideband modulated signal is produced and the arrangement now represents an SSB modulator. Changing the frequency of the stimulating quantity at the frequency level only changes the frequency of the cosinusoidal
and sinusoidal output signals at re(t) and im(t). re(t) and im(t) have exactly
the same amplitude and frequency and a phase difference of 90 degrees as
before. The decisive factor in understanding this type of COFDM implementation is that, in principle, this mutual relationship applies to all subcarriers. For every subcarrier, im(t) is always at 90 degrees to re(t) and has
the same amplitude.
re(t)
Re(f)
ofdm(t)
im(t)
Im(f)
Frequency
domain
Time
domain
Fig. 19.13. IFFT with altered frequency
Including more and more carriers produces a signal with ever more random appearance for re(t) and im(t), the real and imaginary part-signals
having a 90° phase relation to one another in the time domain.
im(t) is said to be the Hilbert Transform of re(t). This transform can be
imagined to be a 90° phase shifter for all spectral components. If both time
domain signals are fed into the I/Q modulator following, the actual
COFDM symbol is produced. In each case, the corresponding upper or
lower COFDM subband is suppressed by this type of modulation, providing a thousandfold phase-shift-type single-sideband modulator. Many references, some of which date back more than 20 years, contain notes regarding single-sideband modulators of this phase-shifting type. It is only
19.3 Generating the COFDM Symbols
359
due to the fact that each subcarrier at re(t) and im(t) has the same amplitude and they are at precisely 90 degrees to one another that the upper
COFDM sideband does not produce crosstalk in the lower one, and vice
versa, with respect to the center frequency.
re(t)
Re(f)
im(t)
ofdm(t)
Im(f)
Frequency
domain
Time
domain
Fig. 19.14. COFDM with 3 carriers
re(t)
Re(f)
im(t)
ofdm(t)
Im(f)
Frequency
domain
Time
domain
Fig. 19.15. COFDM with 12 carriers
360
19 Coded Orthogonal Frequency Division Multiplex (COFDM)
Since nowadays analog, i.e. non-ideal, I/Q modulators are very often
used because of the direct modulation method, the effects arising can only
be explained in this way.
The more carriers (Fig. 19.14.), the more random the appearance of the
corresponding COFDM symbol. Even just 12 single carriers placed in relatively random order with respect to one another result in a COFDM symbol with stochastic appearance. The symbols are calculated and generated
section by section in pipeline fashion. The same number of data bits are
always combined and modulated onto a large number of up to thousands of
COFDM subcarriers. Firstly, real- and imaginary-part tables are produced
in the frequency domain and then, after the IFFT, tables for re(t) and im(t)
which are stored in downstream memories. Period by period, a COFDM
symbol of the exact constant length of Δt = 1/Δf is then generated. Between these symbols, a guard interval of defined but often adjustable
length is maintained.
Symbol n
Symbol n+1
Guard interval
Fig. 19.16. COFDM symbols with guard interval
Inside this guard interval, transient events due to echoes can decay
which prevents inter-symbol interference. The guard interval must be
longer than the longest echo delay time of the transmission system. At the
end of the guard interval, all transient events should have decayed. If this
is not the case, additional noise is produced due to the inter-symbol interference which, in turn, is a simple function of the intensity of the echo.
However, the guard intervals are not simply set to zero. Usually, the end
of the next symbol is keyed precisely into this time interval (Fig. 19.17.)
19.3 Generating the COFDM Symbols
361
and the guard intervals can thus not be seen in any oscillogram. Purely
from the point of view of signal processing, these guard intervals can be
generated quite easily. The signals produced after the IFFT are first written
into a memory in any case and are then read out alternately in accordance
with the pipeline principle. The guard interval is then simply created by
first reading out the end of the respective complex memory content in
corresponding guard interval length (Fig. 19.18.).
Symbol n
Symbol n+1
Guard interval
Fig. 19.17. Guard interval filled up with the end of the next symbol (CP = cyclic
prefix)
MEM1
IFFT
MEM2
Pointer
Fig. 19.18. Generating the guard interval
362
19 Coded Orthogonal Frequency Division Multiplex (COFDM)
But why not simply leave the guard interval empty instead of filling it
up with the end of the next symbol as is usually done? The reason is based
on the way in which a COFDM receiver locks onto the COFDM symbols.
If the guard interval (also often called CP = cyclic prefix) were not occupied with payload information, the receiver would have to hit the COFDM
symbols exactly at the right spot which, however, is no longer possible in
practice due to their rounding off due to multiple echoes during the transmission.
Echo
delay
time
Guard interval
G
1
S1
G
2
Symbol
S2
Path 1
Path 2
Sum
Intersymbol
interference
FFT
window
FFT
window
FFT
window
FFT
window
FFT
window
Autocorrelation function
Fig. 19.19. Multipath reception in COFDM
The beginning and the end of the symbols could only be detected with
difficulty in this case. If, however, e.g. the end of the next symbol is repeated in the preceding guard interval, the signal components existing several times in the signal can be easily found by means of the autocorrelation
function in the receiver. This makes it possible to find the beginning and
the end of the area within the symbols not affected by inter-symbol interference due to echoes. Fig. 19.19. shows this for the case of two receiving
paths. Using the autocorrelation function, the receiver positions its FFT
sampling window, which has the exact length of one symbol, within the
symbols in such a way that it always lines up with the undisturbed area
(Fig. 19.20. and 19.21.). Thus, the sampling window is not positioned precisely over the actual symbol but this only results in a phase error which
produces a turning of all constellation diagrams and must be eliminated in
19.3 Generating the COFDM Symbols
363
subsequent processing steps. However, this phase error produces a rotation
of all constellation diagrams.
It should not be thought, however, that the guard interval can be used
for eliminating fading. This is not so. There is nothing that can be done
against fading apart from adding error protection to the data stream by
means of upstream FEC (forward error correction) and distributing the
data stream as uniformly as possible over all COFDM subcarriers in the
transmission channel.
Fig. 19.20. Practical example: autocorrelation function and FFT window position,
receiving only one signal path [VIERACKER]
Fig. 19.21. Practical example: autocorrelation function and FFT window position,
receiving two paths with 0dB attenuation (0dB echo); sum autocorrelation function and autocorrelation function for both signal paths [VIERACKER]
364
19 Coded Orthogonal Frequency Division Multiplex (COFDM)
19.4 Supplementary Signals in the COFDM Spectrum
Up until now it has only been said that in orthogonal frequency division
multiplex, the information plus error protection is distributed over the
many subcarriers and these are then vector modulated and transmitted.
This gives the impression that every carrier is carrying payload. However,
this is not so, in fact. In all familiar COFDM transmission methods (DAB,
DVB-T, ISDB-T, WLAN, ADSL), the following categories of COFDM
carriers can be found to a greater or lesser extent, or not at all:
•
•
•
•
•
Payload carriers,
Unused carriers set to zero,
Fixed pilots,
Scattered pilots which are not fixed,
Special data carriers for supplementary information.
The term ‘supplementary signals’ has been deliberately kept general
since, although they have the same function everywhere, they have different designations.
In this section, the function of these supplementary signals in the
COFDM spectrum will be discussed in greater detail.
The payload carriers have already been described. They transmit the actual payload data plus error protection and are vector modulated in various
ways. Among others, coherent QPSK, 16QAM or 64QAM modulation is
often used as modulation and the combined 2, 4 or 6 bits per carrier are
then mapped directly onto the respective carrier. In the case of the noncoherent differential coding which is also frequently used, the information
is contained in the difference of the carrier constellation from one symbol
to the next. The main methods are DQPSK or DBPSK. Differential coding
has the advantage that it is ‘self-healing’, i.e. any phase errors which may
be present are corrected automatically, saving channel correction facilities
in the receiver which thus becomes simpler. However, this is at the cost of
twice the bit error ratio in comparison with coherent coding.
Thus, the data carriers can be coded as follows:
•
•
Coherent,
Differentially coded.
The edge carriers, that is to say the top and bottom carriers, are not used
in most cases and are set to zero and do not carry any information at all.
19.4 Supplementary Signals in the COFDM Spectrum
365
They are called zero-information carriers and there are two basic reasons
for the existence of these unused zero-information carriers.
•
•
Preventing adjacent channel crosstalk by facilitating the filtering of
the shoulders of the COFDM spectrum, and
Adapting the bit capacity per symbol to the input data structure.
Fig. 19.22. Real DVB-T COFDM spectrum with shoulders
A COFDM spectrum (Fig. 19.22.) has so-called ‘shoulders’ which are
simply the result of the sin(x)/x-shaped tails of each individual carrier.
These shoulders cause interference in the adjacent channels and it is, therefore, necessary to improve the so-called shoulder attenuation by applying
suitable filtering measures. These filtering measures, in turn, are made easier by simply not using the edge carriers because the filters do not need to
be so steep in this case.
Following an integral multiple of symbols, it is also often necessary to
join up with the input data structure which is also often structured in
blocks. A symbol can carry a certain number of bits due to the data carriers
present in the symbol. The data structure of the input data stream can also
supply a certain number of bits per block. The number of payload carriers
in the symbol is then selected to be only such that the calculation comes
out exactly after a certain number of complete data blocks and symbols.
366
19 Coded Orthogonal Frequency Division Multiplex (COFDM)
Because IFFT is used, however, it is necessary to select a power of two as
the number of carriers which, after subtracting all data and pilot carriers,
still leaves carriers, namely the zero-information carriers.
There are then also the following pilot carriers:
•
•
Pilot carriers with a fixed position in the spectrum, and
Pilot carriers with a variable position in the spectrum.
Pilot carriers with a fixed position in the spectrum are used for automatic frequency control (AFC) in the receiver, i.e. to lock it to the transmitted frequency. These pilot carriers are usually cosinusoidal signals and
are thus located on the real axis at fixed amplitude positions. There is usually a number of such fixed pilots in the spectrum. If the receive frequency
is not tied to the transmit frequency, all constellation diagrams will rotate
also within one symbol. At the receiving end, these fixed pilots within a
symbol are simply missed out and the receive frequency is corrected in
such a way that the phase difference from one fixed pilot to the next within
a symbol becomes zero.
The pilots with variable position in the spectrum are used as measuring
signal for channel estimation and channel correction at the receiving end in
the case of coherent modulation. One could say they represent a sweep
signal for the channel estimation in order to be able to measure the channel.
Special data carriers with supplementary information are very often
used as a fast information channel from transmitter to receiver in order to
inform the receiver of changes made in the type of modulation, e.g. switching from QPSK to 64QAM. In this way, frequently all current transmission
parameters are transmitted from transmitter to receiver, e.g. in DVB-T. It
is then only necessary to set the approximate receiving frequency at the receiver.
19.5 Hierarchical Modulation
Digital transmission methods often exhibit a hard ‘fall off the cliff’ or
‘brickwall' effect when the reception abruptly ceases because the signal/noise ratio limit has been exceeded. Naturally, this also applies to
COFDM. In some COFDM transmission methods (DVB-T, ISDB-T), socalled ‘hierarchical modulation’ is used to counteract this effect. When hierarchical modulation is switched on, the information is transmitted by
means of two different transmission methods within one COFDM spec-
19.6 Summary
367
trum. One of the transmission methods is more robust but cannot support
such a high data rate. The other one is less robust but is capable of handling a higher data rate, making it possible to transmit e.g. the same video
signal with poorer signal quality and with better signal quality in the same
COFDM stream. At the receiving end, one or the other method can then be
selected with an eye on the conditions of reception. Hierarchical modulation will not be discussed in greater detail at this point because there are
several approaches and these depend on the relevant standard.
19.6 Summary
Coded orthogonal frequency division multiplex (COFDM) is a transmission method which, instead of one carrier, uses a large number of subcarriers in one transmission channel. It is especially designed for the characteristics of a terrestrial transmission channel containing multiple echoes. The
information to be transmitted is provided with error protection (coded orthogonal frequency division multiplex - COFDM) and distributed over all
these subcarriers. The subcarriers are vector modulated and in each case
transmit a part of the information. COFDM produces longer symbols than
single-carrier transmission and, as a result, and with the aid of a guard interval, intersymbol interference due to echoes can be eliminated. Due to
the error protection and the fact that the information is distributed over the
many subcarriers it is possible to recover the original data stream free of
errors in spite of any fading due to echoes. A final note: Many references
mention both COFDM and OFDM. In practice, there is no difference between the two methods. OFDM is a part of COFDM. OFDM would never
work without the error protection contained in COFDM.
Bibliography: [REIMERS], [HOFMEISTER], [FISCHER2],
[DAMBACHER], [CHANG], [VIERACKER]
20 Terrestrial Transmission of Digital Television
Signals (DVB-T)
The particular characteristics of a terrestrial radio channel have already
been explained in the previous chapter on COFDM (Coded Orthogonal
Frequency Division Multiplex). They are mainly determined by multipath
reception which leads to location- and frequency-selective fading. In
DVB-T, i.e. in the terrestrial transmission of digital TV signals according
to the Digital Video Broadcasting standard, it was decided that the most
appropriate modulation method to cope with this problem would be
COFDM, the principles of which are explained in the previous chapter.
Fig. 20.1. shows a block diagram of the DVB-T modulator, consisting at
its heart of the COFDM modulator with the IFFT block followed by the
I/Q modulator which can be a digital or an analog type. The position of the
I/Q modulator in the circuit can vary depending on how the DVB-T modulator is implemented in practice. The COFDM modulation is preceded by
the channel coding, i.e. the error correction, which is exactly the same in
DVB-T as in DVB-S satellite transmission.
FIR
filter
Precorr.
Symbol
interleaver
Mapper
O(rthogonal) F(requency) D(ivision) M(ultiplex)
C(oded)
IF
RF
Power
ampl.
Bandpass
filter
IFFT
Guard
interv.
insert.
TS1
Demux
FEC LP
TS2
FEC HP
(option)
Bit
interleaver
Frame
adapt.
(2, 4, 6)
Pilots, TPS
Fig. 20.1. Block diagram of the DVB-T modulator – part 1
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_20, © Springer-Verlag Berlin Heidelberg 2010
370
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
As can also be seen from the block diagram, two MPEG-2 transport
stream inputs are possible which then provides for the so-called hierarchical modulation. However, hierarchical modulation is provided as an option
in DVB-T and has not as yet been put into practical use. Hierarchical
modulation was originally provided for transmitting the same TV programs with different data rate, different error correction and different quality in one DVB-T channel. The HP (high priority) path transmits a data
stream with a low data rate, i.e. a poorer picture quality due to higher compression, but allows better error protection or a more robust type of modulation (QPSK) to be used. The LP (low priority) path is used for transmitting the MPEG-2 transport stream with a higher data rate, a lower error
protection and a type of modulation of higher order (16QAM, 64QAM).
At the receiving end, HP or LP can be selected in dependence on the conditions of reception. Hierarchical modulation is intended to lessen the impact, as it were, of the “fall off the cliff”. But it is also quite conceivable to
transmit two totally independent transport streams. Both HP and LP
branches contain the same channel coder as in DVB-S but, as already mentioned, this is an option in the DVB-T modulator, not the receiver where
this involves very little additional expenditure.
x 204/188
Inv. sync.
TS in
Base
bandinterf.
Sync
invert.
Energy
disp.
FEC1/
outer
coder
ReedSolom.
enc.
x2
x (1.5-code rate)
FEC2/
inner
coder
Conv.
interleaver
Conv.
coder
= Date rate out
[2.17...(1.63)...1.36]
Puncturing
Data rate in
Coded
data
out
Synchronization
Same as in DVB-C
Code rate
1/2...(3/4)...7/8
Same as in DVB-S
Fig. 20.2. Block diagram of the DVB-T modulator – part 2, FEC
Not every COFDM carrier in DVB-T is a payload carrier. There is also
a large number of pilot carriers and special carriers. These special carriers
are used for frequency synchronization, channel estimation and channel
correction, and for implementing a fast information channel. They are inserted into their locations in the DVB-T spectrum before the IFFT.
20.1 The DVB-T Standard
371
Before discussing the DVB-T standard in greater detail, let us first ask:
“Why DVB-T?”
There are fully operational scenarios for supplying digital television via
satellite and cable, both of which paths are accessible to many households
throughout the world. Why then the need for yet another, terrestrial path,
e.g. via DVB-T which, in addition, is complex and expensive and may require a large amount of maintenance? The additional coverage with digital
terrestrial television is necessary for reasons of
•
•
•
•
•
Regional requirements (historical infrastructures, no satellite reception)
Regional geographic situations
Portable TV reception
Mobile TV reception
Local supplementary municipal services (regional/urban television)
Many countries in the world do not have satellite TV coverage, or only
inadequately so, for the most varied reasons of a political, geographic or
other nature. In many cases, substitute coverage by cable is not possible,
either, because of e.g. permafrost and also often cannot be financed because of sparse population density. This leaves only the terrestrial coverage. Countries which are far away from the equator such as those in Scandinavia naturally have more problems with satellite reception since, e.g.
the satellite receiving antennas are almost pointing at the ground. There are
also many countries which have not previously had analog satellite reception as a standard such as Australia where satellite reception plays only a
minor role. Population centers there are covered terrestrially and via cable
or satellite. In many countries, it is not permitted to gather in an uncontrollable variety of TV programs from the sky for political reasons. Even regions in Central Europe with good satellite and cable coverage require additional terrestrial TV coverage, mainly for local TV programs which are
not being broadcast via satellite. And portable and mobile reception is virtually only possible via the terrestrial path.
20.1 The DVB-T Standard
In 1995, the terrestrial standard for the transmission of digital TV programs was defined in ETS 300744 in connection with the DVB-T project.
A DVB-T channel can have a bandwidth of 8, 7 or 6 MHz. There are two
different operating modes: the 2K mode and the 8K mode where 2K stands
372
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
for a 2046-points IFFT and 8K stands for an 8192-points IFFT. As is already known from the chapter on COFDM, the number of COFDM subcarriers must be a power of two. In DVB-T, it was decided to use symbols
with a length of about 250 µs (2K mode) or 1 ms (8K mode). Depending
on requirements, one or the other mode can be selected. The 2K mode has
greater subcarrier spacing of about 4 kHz but the symbol period is much
shorter. Compared with the 8K mode with a subcarrier spacing of about 1
kHz, it is much less susceptible to spreading in the frequency domain
caused by doppler effects due to mobile reception and multiple echoes but
much more susceptible to greater echo delays. In single frequency networks, for example, the 8K mode will always be selected because of the
greater transmitter spacing possible. In mobile reception, the 2K mode is
better because of the greater subcarrier spacing. The DVB-T standard allows for flexible control of the transmission parameters.
Apart from the symbol length, which is a result of the use of 2K or 8K
mode, the guard interval can also be adjusted within a range of 1/4 to 1/32
of the symbol length. It is possible to select the type of modulation (QPSK,
16QAM or 64QAM)). The error protection (FEC) is designed to be the
same as in the DVB-S satellite standard. The DVB-T transmission can be
adapted to the respective requirements with regard to robustness or net
data rates by adjusting the code rate (1/2 ... 7/8).
In addition, the DVB-T standard provides for hierarchical coding as an
option. In hierarchical coding, the modulator has two transport stream inputs and two independently configurable but identical FECs. The idea is to
apply a large amount of error correction to a transport stream with a low
data rate and then to transmit it with a very robust type of modulation.
This transport stream path is then called the high priority (HP) path. The
second transport stream has a higher data rate and is transmitted with less
error correction and, e.g. 64QAM modulation and the path is called the
low priority (LP) path. It would then be possible, e.g. to subject the identical program packet to MPEG-2 coding, once at the higher data rate and
once at the lower data rate, and to combine the two packets in two multiplex packets transported in independent transport streams. Higher data rate
automatically means better (picture) quality. The data stream with the
lower data rate and correspondingly lower picture quality is fed to the high
priority path and that with the higher data rate is supplied to the low priority path. At the receiving end, the high priority signal is demodulated more
easily than the low priority one. Depending on the conditions of reception,
the HP path or the LP path will be selected at the receiving end. If the reception is poor, there will at least still be reception due to the lower data
rate and higher compression, even if the quality of the picture and sound is
inferior.
20.2 The DVB-T Carriers
373
In DVB-T, coherent COFDM modulation is used, i.e. the payload carriers are mapped absolutely and are not differentially coded. However, this
requires channel estimation and correction for which numerous pilot signals are provided in the DVB-T spectrum and are used as test signal for the
channel estimation.
TPS
carrier
Continual
or
scattered
pilot
TPS
carrier
Continual
or
scattered
pilot
Fig. 20.3. DVB-T carriers: payload carriers, Continual and Scattered Pilots, TPS
carriers
20.2 The DVB-T Carriers
In DVB-T, an IFFT with 2048 or 8192 points is used. In theory, 2048 or
8192 carriers would then be available for the data transmission. However,
not all of these carriers are used as payload carriers. In the 8K mode, there
are 6048 payload carriers and in the 2K mode there are 1512. The 8K
mode thus has exactly four times as many payload carriers as the 2K mode
but since the symbol rate is higher by a factor of exactly 4 in the 2K mode,
both modes will always have the same data rate, given the same conditions
of transmission. DVB-T contains the following types of carrier:
•
•
•
•
•
Inactive carriers with fixed position (set to zero amplitude)
Payload carriers with fixed position
Continual pilots with fixed position
Scattered pilots with changing position in the spectrum
TPS carriers with fixed position
374
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
The meaning of the words ‘payload carrier’ is clear: these are simply the
carriers used for the actual data transmission. The edge carriers at the upper and lower channel edge are set to zero, i.e. they are inactive and carry
no modulation at all, i.e. their amplitudes are zero. The continual pilots are
located on the real axis, i.e. the I (in-phase) axis, either at 0 degrees or at
180 degrees and have a defined amplitude. The continual pilots are
boosted by 3 dB compared with the average signal power and are used in
the receiver as phase reference and for automatic frequency control (AFC),
i.e. for locking the receive frequency to the transmit frequency. The scattered pilots are scattered over the entire spectrum of the DVB-T channel
from symbol to symbol and virtually constitute a sweep signal for the
channel estimation. Within each symbol, there is a scattered pilot every
12th carrier. Each scattered pilot jumps forward by three carrier positions
in the next symbol, i.e. in each case two intermediate payload carriers will
never become a scattered pilot whereas those at every 3rd position in the
spectrum are sometimes payload carriers and sometimes scattered pilots.
The scattered pilots are also on the I axis at 0 degrees and 180 degrees and
have the same amplitude as the continual pilots.
0
1
2
3
4
= scattered pilot
5
6
7
8
9
10 11 12
= payload carrier
Fig. 20.4. Change of position of the Scattered Pilots
The TPS carriers are located at fixed frequency positions. For example,
carrier No. 50 is a TPS carrier. TPS stands for Transmission Parameter
Signalling. These carriers represent virtually a fast information channel via
which the transmitter informs the receiver about the current transmission
parameters. They are DBPSK (differential bi-phase shift keying) modulated and are located on the I axis either at 0 degrees or at 180 degrees.
They are differentially coded, i.e. the information is contained in the difference between one and the next symbol. All the TPS carriers in one
symbol carry the same information, i.e. they are all either at 0 degrees or
all at 180 degrees on the I axis. At the receiving end, the correct TPS car-
20.2 The DVB-T Carriers
375
rier position of 0 degrees or 180 degrees is then determined by majority
voting for each symbol and is then used for the demodulation. DBPSK
means that a zero is transmitted when the state of the TPS carriers changes
from one symbol to the next, and a one if the TPS carrier phase does not
change from one symbol to the next. The complete TPS information is
broadcast over 68 symbols and comprises 67 bits. This segment over 68
symbols is called a frame and the scattered pilots within this frame also
jump over the DVB-T channel from the start of the channel right to the end
of the channel.
Q
I
Fig. 20.5. DBPSK modulated TPS carriers
17 of the 68 TPS bits are used for initialization and synchronization, 13
bits are error protection, 22 bits are used at present and 13 bits are reserved
for future applications. Table 20.1. explains how the TPS carriers are utilized.
Thus, the TPS carriers keep the receiver informed about:
•
•
•
•
•
Mode (2K, 8K)
Length of the guard interval (1/4, 1/8, 1/16, 1/32)
Type of modulation (QPSK, 16QAM, 64QAM)
Code rate (1/2, 2/3, 3/4, 5/6, 7/8)
Use of hierarchical coding
However, the receiver should have already determined the mode (2K,
8K) and the length of the guard interval which are thus actually meaningless as TPS information.
376
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
Table 20.1. Bit allocation of the TPS carriers
Symbol
number
s0
s1- s16
s17 - s22
s23, s24
S25, s26
Format
0011010111101110
or
1100101000010001
010 111
S27, s28, s29
S30, s31, s32
s33, s34, s35
s36, s37
s38, s39
s40 - s53
s54 - s67
all set to "0"
BCH code
Purpose/content
Initialization
Synchronization word
Length indicator
Frame number
Constellation
00=QPSK/01=16QAM/10=64QAM
Hierarchy information
000=Non hierarchical,
001=α=1, 010=α=2, 011=α=4
Code rate, HP stream
000=1/2, 001=2/3, 010=3/4,
011=5/6, 100=7/8
Code rate, LP stream
000=1/2, 001=2/3, 010=3/4,
011=5/6, 100=7/8
Guard interval
00=1/32, 01=1/16, 10=1/8, 11=1/4
Transmission mode
00=2K, 01=8K
Reserved for future use
Error protection
In Fig. 20.3., the position of the pilots and TPS carriers can be seen
clearly in a 64QAM constellation diagram. The two outer points on the I
axis correspond to the positions of the continual pilots and the scattered pilots. The two inner points on the I axis are the TPS carriers.
The position of the continual pilots and of the TPS carriers in the spectrum can be seen from the tables 20.2. and 20.3. In these tables, the carrier
numbers are listed at which the continual pilots and the TPS carriers can be
found. Counting begins at carrier number zero which is the first non-zero
carrier at the beginning of the channel.
The various types of carriers used in DVB-T are briefly summarized as
follows. Of the 2048 carriers in the 2K mode, only 1705 carriers are used
and all others are set to zero. Within these 1705 carriers there are 1512
payload carriers which can be QPSK, 16QAM or 64QAM modulated, 142
scattered pilots, 45 continual pilots and 17 TPS carriers.
20.2 The DVB-T Carriers
377
Table 20.2. Carrier positions of the Continual Pilots
2K mode
8K mode
0 48 54 87 141 156
0 48 54 87 141 156 192 201 255 279 282 333 432 450
192 201 255 279 282
483 525 531 618 636 714 759 765 780 804 873 888 918
333 432 450 483 525
939 942 969 984 1050 1101 1107 1110 1137 1140 1146
531 618 636 714 759
1206 1269 1323 1377 1491 1683 1704 1752 1758 1791
765 780 804 873 888
1845 1860 1896 1905 1959 1983 1986 2037 2136 2154
918 939 942 969 984
2187 2229 2235 2322 2340 2418 2463 2469 2484 2508
1050 1101 1107 1110
2577 2592 2622 2643 2646 2673 2688 2754 2805 2811
1137 1140 1146 1206
2814 2841 2844 2850 2910 2973 3027 3081 3195 3387
1269 1323 1377 1491
3408 3456 3462 3495 3549 3564 3600 3609 3663 3687
1683 1704
3690 3741 3840 3858 3891 3933 3939 4026 4044 4122
4167 4173 4188 4212 4281 4296 4326 4347 4350 4377
4392 4458 4509 4515 4518 4545 4548 4554 4614 4677
4731 4785 4899 5091 5112 5160 5166 5199 5253 5268
5304 5313 5367 5391 5394 5445 5544 5562 5595 5637
5643 5730 5748 5826 5871 5877 5892 5916 5985 6000
6030 6051 6054 6081 6096 6162 6213 6219 6222 6249
6252 6258 6318 6381 6435 6489 6603 6795 6816
Table 20.3. Carrier positions of the TPS Carriers
2K mode
8K mode
34 50 209 346 413 569 595
34 50 209 346 413 569 595 688 790 901
688 790 901 1073 1219 1262
1073 1219 1262 1286 1469 1594 1687 1738
1286 1469 1594 1687
1754 1913 2050 2117 2273 2299 2392 2494
2605 2777 2923 2966 2990 3173 3298 3391
3442 3458 3617 3754 3821 3977 4003 4096
4198 4309 4481 4627 4670 4694 4877 5002
5095 5146 5162 5321 5458 5525 5681 5707
5800 5902 6013 6185 6331 6374 6398 6581
6706 6799
378
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
Some of the scattered pilots occasionally coincide with positions of continual pilots which is why the number 131 should be used for calculating
the actual payload carriers in the case of the scattered pilots in 2K mode.
The conditions in the 8K mode are comparable. Here, too, not all the 8192
carriers are being used but only 6817 of which, in turn, only 6048 are actual payload carriers. The rest are scattered pilots (568), continual pilots
(177) and TPS carriers (68). As before, the number 524 must be used for
the scattered pilots in calculating the payload carriers since sometimes a
scattered pilot will coincide with a continual pilot. Every 12th carrier in a
symbol is a scattered pilot. It is thus easy to calculate the number of scattered pilots by dividing the number of carriers actually used by 12
(1705/12 = 142, 6817/12 = 568).
Table 20.4. Number of different carriers in DVB-T
2K mode
2048
1705
142/131
45
17
1512
8K mode
8192
6817
568/524
177
68
6048
Carrier
used carrier
Scattered pilots
Continual pilots
TPS carrier
payload carrier
The payload carriers are either QPSK, 16QAM or 64QAM modulated
and transmit the error-protected MPEG-2 transport stream. Fig. 20.6.
shows the constellation diagrams for QPSK, 16QAM and 64QAM with the
positions of the special carriers in the case of non-hierarchical modulation.
Fig. 20.6. DVB-T constellation diagrams for QPSK, 16QAM and 64QAM
20.3 Hierarchical Modulation
379
20.3 Hierarchical Modulation
To ensure that reliable reception is still guaranteed even in poor conditions, hierarchical modulation is provided as an option in DVB-T. Without
it, e.g. a signal/noise ratio which is too bad will lead to a hard “fall off the
cliff”, otherwise known as the ‘brick wall effect’. In the case of the frequently used DVB-T transmission with 64QAM modulation and a code
rate of 3/4 or 2/3, the limit of stable reception is at a signal/noise ratio of
just under 20 dB. In this section, the details of hierarchical modulation/coding will be explained more fully. If hierarchical modulation is
used, the DVB-T modulator has two transport stream inputs and two FEC
blocks. One transport stream with a low data rate is fed into the so-called
high priority path (HP) and provided with a large amount of error protection, e.g. by selecting the code rate 1/2. A second transport stream with a
higher data rate is supplied in parallel to the low priority path (LP) and is
provided with less error protection, e.g. with the code rate 3/4.
Fig. 20.7. Embedded QPSK in 64QAM with hierarchical modulation
In principle, both HP and LP transport streams can contain the same
programs but at different data rates, i.e. with different amounts of compression. However, the two can also carry completely different payloads.
On the high priority path, QPSK is used which is a particularly robust type
of modulation. On the low priority path, a higher level of modulation is
needed due to the higher data rate. In DVB-T, the individual payload carriers are not modulated with different types of modulation. Instead, each
payload carrier transmits portions both of LP and of HP. The high priority
path is transmitted as so-called embedded QPSK in 16QAM or 64QAM.
380
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
Fig. 20.7. shows the case of QPSK embedded in 64QAM. The LP information is carried by the discrete constellation point and the HP is described
by the quadrant. A cloud of 8 times 8 points in a quadrant as a whole thus
corresponds virtually to the total constellation point of the QPSK in this
quadrant.
64QAM, =1
64QAM, =2
64QAM, =4
16QAM, =1
16QAM, =2
16QAM, =4
Fig. 20.8. Possible Constellations with hierarchical modulation
A 64QAM modulation enables 6 bits per symbol to be transmitted.
However, since the quadrant information, as QPSK, diverts 2 bits per
symbol for the HP stream, only 4 bits per symbol remain for the transmission of the LP stream. The gross data rates for LP and HP thus have a fixed
ratio of 4:2 to one another. In addition, the net data rates are dependent on
the code rate used. QPSK embedded in 16QAM is also possible. The ratio
between the gross data rates of LP and HP is then 2:2. To make the QPSK
of the high priority path more robust, i.e. less susceptible to interference,
the constellation diagram can be spread at the I axis and the Q axis. A factor α of 2 or 4 increases the distance between the individual quadrants of
the 16QAM or 64QAM diagrams. The higher α is, the more insensitive the
high priority path becomes and the more sensitive the low priority path becomes since the discrete constellation points move closer together. Fig.
20.8. shows the 6 possible constellations with hierarchical modulation, i.e.
20.4 DVB-T System Parameters of the 8/7/6 MHz Channel
381
64QAM with α = 1, 2, 4 and 16QAM with α = 1, 2, 4. The information
about the presence or absence of hierarchical modulation and the α factor
and the code rates for LP and HP are transmitted in the TPS carriers. This
information is evaluated in the receiver which automatically adjusts its demapper accordingly. The decision to demodulate HP or LP in the receiver
can be made automatically in dependence on the current conditions of reception (channel bit error rate) or left to the user to select manually. Hierarchical modulation is provided as an option in modern DVB-T chipsets
and set-top boxes since, in practice, no additional hardware is required. In
many DVB-T receivers, however, no software is provided for this option
since it is currently not used in any country. At the beginning of 2002, hierarchical modulation was tested in field tests in Australia but it is currently not used there, either.
20.4 DVB-T System Parameters of the 8/7/6 MHz Channel
In the following paragraphs, the system parameters of DVB-T will be derived and explained in detail. These parameters are:
•
•
•
•
•
IFFT sampling frequencies
DVB-T signal bandwidths
Spectrum occupied by the 8/7 and 6 MHz DVB-T channel
Data rates
Signal levels of the individual carriers
The basic system parameter in DVB-T is the IFFT sampling frequency
of the 8-MHz channel which is defined as:
fsample IFFT 8MHz = 64/7 MHz = 9.142857143 MHz.
From this basic parameter, all other system parameters can be derived,
i.e. those of the 8/7 and 6 MHz channel. The IFFT sampling frequency is
the sampling rate of the COFDM symbol or, respectively, the bandwidth
within which all 2K (= 2048) and 8K (= 8192) subcarriers can be accommodated. However, many of these 2048 or 8192 subcarriers are set to zero
and the bandwidth of the DVB-T signal must be narrower than that of the
actual 8, 7 or 6 MHz wide channel. As will be seen, the signal bandwidth
of the 8 MHz channel is only about 7.6 MHz and there is thus a space of
approx. 200 kHz between the top and the bottom of this channel and its adjacent channels.
382
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
These 7.6 MHz contain the 6817 or 1705 carriers actually used. In the
case of the 7 or 6 MHz channel, the IFFT sampling frequency of these
channels can be calculated from the IFFT sampling frequency of the 6
MHz channel by simply multiplying it by 7/8 or 6/8, respectively.
fsample IFFT 7MHz = 64/7 MHz • 7/8 = 8 MHz;
fsample IFFT 6MHz = 64/7 MHz • 6/8 = 48/7 MHz = 6.857142857 MHz;
All 2048 or 8192 IFFT carriers in the 8/7 and 6 MHz channel can be
found within these IFFT bandwidths. From these bandwidths or sampling
frequencies, the respective subcarrier spacing can be easily derived by dividing the bandwidth fsample IFFT by the number of IFFT subcarriers:
Δf = fsample IFFT /Ntotal_carriers;
Δf2k = fsample IFFT /2048;
Δf8k = fsample IFFT /8192;
Therefore, the COFDM subcarrier spacing in an 8, 7 or 6 MHz-wide
DVB-T channel in 2K and 8K mode is:
Table 20.5. Subcarrier spacing in 2K and 8K mode
Channel
bandwidth
8 MHz
7 MHz
6 MHz
f of 2K mode
4.464285714 kHz
3.90625 kHz
3.348214275 kHz
f of 8K mode
1.116071429 kHz
0.9765625 kHz
0.8370535714 kHz
From the subcarrier spacing, the symbol length Δtsymbol can be determined directly. Due to the orthogonality condition, it is:
Δtsymbol = 1/Δf;
Therefore, the symbol lengths in the various modes and channel bandwidths in DVB-T are:
Table 20.6. Symbol durations in 2K and 8K mode
Channel
bandwidth
8 MHz
7 MHz
6 MHz
tsymbol of 2K mode
224 us
256 us
298.7 us
tsymbol of 8K mode
896 ms
1.024 ms
1.1947 ms
20.4 DVB-T System Parameters of the 8/7/6 MHz Channel
383
The DVB-T signal bandwidths are obtained from the subcarrier spacing
Δf of the respective channel (8, 7, 6 MHz) and the number of carriers actually used in 2K and 8K mode (1705 and 6817).
fsignal DVB-T = Nused_carriers • Δf;
Table20.7. Signal Bandwidths in DVB-T
Channel
bandwidth
8 MHz
7 MHz
6 MHz
fsignal DVB-T
of 2K mode
7.612 MHz
6.661 MHz
5.709 MHz
IFFT
bandwidth
Channel
bandwidth
8/7/6 MHz
fsignal DVB-T
of 8K mode
7.608 MHz
6.657 MHz
5.706 MHz
IFFT bandwidth
9.1429 MHz (64/7) @ 8 MHz
8.0000 MHz @ 7 MHz
6.8571 MHz (48/7) @ 6 MHz
Central carrier
3408 [852]
Signal bandwidth
7.61 MHz @ 8 MHz
6.66 MHz @ 7 MHz
5.71 MHz @ 6 MHz
Signal
bandwidth
Subcarrier spacing
1.11 [4.46] kHz @ 8 MHz
0.98 [3.91] kHz @ 7 MHz
0.84 [3.35] kHz @ 6 MHz
Carrier # 0
Carrier # 6816
[1704]
Fig. 20.9. Spectrum of a DVB-T signal in 8K and [2K] mode for the 8/7/6 MHz
channel
In principle, there are two ways of counting the COFDM subcarriers of
the DVB-T channel. The carriers can either be counted through from 0 to
2047 or from 0 to 8192 in accordance with the number of IFFT carriers or
counting can begin with carrier number zero at the first carrier actually
used in the respective mode. The latter counting method is the more usual
one, counting from 0 to 1704 in 2K mode and from 0 to 6816 in 8K mode.
In Fig. 20.9. then the position of the spectrum of the DVB-T channel is
384
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
shown and the most important DVB-T system parameters are summarized
again.. Fig. 20.9. also shows the center carrier numbers which are of particular importance in testing. This carrier number of 3408 in the 8K mode
and 852 in the 2K mode corresponds to the exact center of the DVB-T
channel. Some effects which can be caused by the DVB-T modulator can
only be observed at this point. The values provided in square brackets in
the Figure apply to the 2K mode (e.g.: 3408 [852]) and the others apply to
the 8K mode.
The gross data rate of the DVB-T signal is derived from, among other
things, the symbol rate of the DVB-T COFDM signal. The symbol rate is a
function of the length of the symbol and of the length of the guard interval.
as follows:
symbol _ rateCOFDM =
1
;
symbol _ duration + guard _ duration
The gross data rate is then the result of the symbol rate, the number of
actual payload carriers and the type of modulation (QPSK, 16QAM,
64QAM). In 2K mode, there are 1512 payload carriers and in 8K mode
there are 6048. In QPSK, 2 bits per symbol are transmitted, in 16QAM it is
4 bits per symbol and in 64QAM it is 6 bits per symbol. Since the symbols
are longer by a factor of 4 in the 8K mode but, on the other hand, there are
four times as many payload carriers in the channel, this factor cancels out
again which means that the data rates are independent of the mode (2K or
8K). The gross data rate of the DVB-T channel is thus:
gross_data_rate = symbol_rateCOFDM • no_of_payload_carriers
• bits_per_symbol;
The total length of the COFDM symbols is composed of the symbol
length and the length of the guard intervals:
Table 20.8. Total symbol durations in DVB-T
Chan.
bandw.
[MHz]
8
7
6
total_symbol_duration = symbol_duration + guard_duration [us]
2K
2K
2K
2K
8K
8K
8K
8K
1/4
1/8
1/16
1/32
1/4
1/8
1/16
1/32
280
320
373.3
252
288
336
238
272
317.3
231
264
308
1120
1008
1280
1152
1493.3 1344
952
924
1088
1056
1269.3 1232
total_symbol_duration = symbol_duration + guard_duration;
20.4 DVB-T System Parameters of the 8/7/6 MHz Channel
385
The symbol rate of the DVB-T channel is calculated as:
symbol_rate = 1 / total_ symbol_duration;
The DVB-T symbol rates are listed in Table 20.9. as a function of mode
and channel bandwidth.
Table 20.9. Symbol rates in DVB-T
Channel
bandwidth
8 MHz
7 MHz
6 MHz
Symbol rate [kS/s]
2K
2K
guard
guard
1/8
1/4
3.5714
3.9683
3.1250
3.4722
2.6786
2.9762
2K
guard
1/16
4.2017
3.6760
3.1513
2K
guard
1/32
4.3290
3.7888
3.2468
8K
guard
1/4
0.8929
0.7813
0.6696
8K
guard
1/8
0.9921
0.8681
0.7440
8K
guard
1/16
1.04504
0.9191
0.7878
8K
guard
1/32
1.0823
0.9470
0.8117
The gross data rate is then determined from:
gross_data_rate = symbol_rate • no_of_payload_carriers
• bits_per_symbol;
The gross DVB-T data rates are listed in Table 20.10. as a function of
channel bandwidth and the guard interval length.
Table 20.10. Gross data rates in DVB-T
Channel
bandwidth,
modulation
8 MHz
QPSK
8 MHz
16QAM
8 MHz
64QAM
7 MHz
QPSK
7 MHz
16QAM
7 MHz
64QAM
6 MHz
QPSK
6 MHz
16QAM
6 MHz
64QAM
Gross data rate [MBit/s]
2K
2K
2K
guard
guard
guard
1/16
1/8
1/4
2K
guard
1/32
8K
guard
1/4
8K
guard
1/8
8K
guard
1/16
8K
guard
1/32
10.800
12.000
12.706
13.091
10.800
12.000
12.706
13.091
21.6
24.0
25.412
26.182
21.6
24.0
25.412
26.182
32.4
36.0
38.118
39.273
32.4
36.0
38.118
39.273
9.45
10.5
11.118
11.455
9.45
10.5
11.118
11.455
18.9
21.0
22.236
22.91
18.9
21.0
22.236
22.91
28.35
31.5
33.354
34.365
28.35
31.5
33.354
34.365
8.1
9.0
9.530
9.818
8.1
9.0
9.530
9.818
16.2
18.0
19.06
19.636
16.2
18.0
19.06
19.636
24.3
27.0
28.59
29.454
24.3
27.0
28.59
29.454
386
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
The net data rate additionally depends on the code rate of the convolutional coding used and on the Reed Solomon error protection RS(188, 204)
as follows:
net_data_rate = gross_data_rate • 188/204 • code_rate;
Since the factor of 4 cancels out, the overall formula for determining the
net data rate of DVB-T signals is independent of the mode (2K or 8K) and
is:
net_data_rate = 188/204 • code_rate • log2(m) • 1/(1 + guard)
• channel • const1;
where
m
= 4 (QPSK), 16 (16-QAM), 64 (64-QAM);
log2(m) = 2 (QPSK), 4 (16-QAM), 6 (64-QAM);
code rate = 1/2, 2/3, 3/4, 5/6, 7/8;
guard
= 1/4, 1/8, 1/16, 1/32;
channel = 1 (8MHz), 7/8 (7MHz), 6/8 (6 MHz);
const1 = 6.75 • 106 bits/s;
Table 20.11. Net data rates with non-hierarchical modulation in the 8 MHz
DVB-T channel
Modulation
QPSK
16QAM
64QAM
Code rate
Guard 1/4
Guard 1/8
Guard 1/16
Guard 1/32
Mbit/s
Mbit/s
Mbit/s
Mbit/s
1/2
4.976471
5.529412
5.854671
6.032086
2/3
6.635294
7.372549
7.806228
8.042781
3/4
7.464706
8.294118
8.782007
9.048128
5/6
8.294118
9.215686
9.757785
10.05348
7/8
8.708824
9.676471
10.24567
10.55617
1/2
9.952941
11.05882
11.70934
12.06417
2/3
13.27059
14.74510
15.61246
16.08556
3/4
14.92941
16.58824
17.56401
18.09626
5/6
16.58824
18.43137
19.51557
20.10695
7/8
17.41765
19.35294
20.49135
21.11230
1/2
14.92941
16.58824
17.56401
18.0926
2/3
19.90588
22.11765
23.41869
24.12834
3/4
22.39412
24.88235
26.34602
27.14439
5/6
24.88235
27.64706
29.27336
30.16043
7/8
26.12647
29.02941
30.73702
31.66845
20.4 DVB-T System Parameters of the 8/7/6 MHz Channel
387
From this, the net data rates of the 8, 7 and 6 MHz channel in the various operating modes can be determined:
The net data rates in DVB-T vary between about 4 and 31 Mbit/s in dependence on the transmission parameters and channel bandwidths used. In
the 7 MHz and 6 MHz channels, the available net data rates are lower by a
factor of 7/8 or 6/8, respectively. in comparison with the 8 MHz channel.
Table 20.12. Net data rates with non-hierachical modulation in the 7 MHz
DVB-T Channel
Modulation
Code rate
Guard 1/4
Guard 1/8
Guard 1/16
Guard 1/32
Mbit/s
Mbit/s
Mbit/s
Mbit/s
QPSK
1/2
4.354412
4.838235
5.122837
5.278075
2/3
5.805882
6.450980
6.830450
7.037433
3/4
6.531618
7.257353
7.684256
7.917112
5/6
7.257353
8.063725
8.538062
8.796791
7/8
7.620221
8.466912
8.964965
9.236631
1/2
8.708824
9.676471
10.245675
10.556150
2/3
11.611475
12.901961
13.660900
14.074866
3/4
13.063235
14.514706
15.368512
15.834225
16QAM
64QAM
5/6
14.514706
16.127451
17.076125
17.593583
7/8
15.240441
16.933824
17.929931
18.473262
1/2
13.063235
14.514706
15.368512
15.834225
2/3
17.417647
19.352941
20.491350
21.112300
3/4
19.594853
21.772059
23.052768
23.751337
5/6
21.772059
24.191177
25.614187
26.390374
7/8
22.860662
25.400735
26.894896
27.709893
In hierarchical modulation, the gross data rates in 64QAM modulation
are distributed at a ratio of 2:4 between HP and LP and in16QAM the ratio
between HP and LP gross data rates is exactly 2:2. In addition, the net data
rates in the high priority and low priority paths depend on the code rates
used there.
The formulas for determining the net data rates of HP and LP are:
net_data_rateHP = 188/204 • code_rateHP • bits_per_symbolHP
• 1/(1 + guard_duration) • channel • const1;
net_data_rateLP = 188/204 • code_rateLP • bits_per_symbolLP
• 1/(1 + guard_duration) • channel • const1;
388
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
where
bits_per_symbolHP
bits_per_symbolLP
code_rate HP/LP
guard_duration
channel
const1
= 2:
= 2 (16QAM) or 4 (64QAM);
= 1/2, 2/3, 3/4, 5/6, 7/8;
= 1/4, 1/8, 1/16, 1/32;
= 1 (8MHz), 7/8 (7MHz), 6/8 (6 MHz);
= 6.75 • 106 bits/s;
Table 20.13. Net data rates with non-hierarchical modulation in the 6 MHz
DVB-T Channel
Modulation
Code rate
Guard 1/4
Guard 1/8
Guard 1/16
Guard 1/32
Mbit/s
Mbit/s
Mbit/s
Mbit/s
QPSK
1/2
3.732353
4.147059
4.391003
4.524064
2/3
4.976471
5.529412
5.854671
6.032086
16QAM
64QAM
3/4
5.598529
6.220588
6.586505
6.786096
5/6
6.220588
6.911765
7.318339
7.540107
7/8
6.531618
7.257353
7.684256
7.917112
1/2
7.464706
8.294118
8.782007
9.048128
2/3
9.952941
11.058824
11.709343
12.064171
3/4
11.197059
12.441177
13.173010
13.572193
5/6
12.441176
13.823529
14.636678
15.080214
7/8
13.063235
14.514706
15.368512
15.834225
1/2
11.197059
12.441177
13.173010
13.572193
2/3
14.929412
16.588235
17.564014
18.096257
3/4
16.795588
18.661765
19.759516
20.358289
5/6
18.661765
20.735294
21.955017
22.620321
7/8
19.594853
21.772059
23.052768
23.751337
This brings us to the last details of the DVB-T standard related to actual
field experience: the constellation and the levels of the individual carriers.
Depending on the type of constellation (QPSK, 16QAM or 64QAM, hierarchical with α = 1, 2 or 4), a mean signal value of the payload carriers is
obtained which can be calculated simply by means of the quadratic mean
(RMS value) of all possible vector lengths in their correct distribution.
This mean is then defined as 100% or simply as One. In the case of the 2K
mode, there are 1512 or 6048 payload carriers, the mean power of which is
100% or One. The TPS carrier levels are set in the same way exactly in relation to the individual payload carriers. The continual and scattered pilots
are differently arranged. Due to the need for easy detectability, these pilots
20.4 DVB-T System Parameters of the 8/7/6 MHz Channel
389
are boosted by 2.5 dB with respect to the mean signal level of the payload
carriers. I.e., the voltage level of the continual and scattered pilots is higher
by 4/3 compared with the mean level of the payload carriers and the power
level is higher by 16/9.
20 log(4/3) = 2.5 dB; voltage ratio of continual and scattered pilots with
respect to the payload carrier signal average;
and
10 log(16/9) = 2.5 dB; power ratio of continual and scattered pilots with
respect to the payload carrier signal average;
In summary, it can be said that the position of the TPS carriers in the
constellation diagram always corresponds to the 0 dB point of the mean
value of the payload carriers and that the position of the continual pilots
and of the scattered pilots always corresponds to the 2.5 dB point, regardless of the DVB-T constellation involved at the time.
Test instruments are often calibrated for carrier-to-noise ratio (C/N) and
not for signal-to-noise ratio (S/N). The signal-to-noise ratio, however, is
relevant to the calculation of the bit error ratio (BER) caused by pure noise
interference in the channel. The C/N must then be converted into S/N.
When converting C/N to S/N, the energy in the pilots must be taken into
consideration. The energy in the pure payload carrier without pilots can be
determined as follows, both for the 2K mode and for the 8K mode:
payload_to_signal = (payload / (payload + (scattered + continual)
• (4/3)2 + TPS • 1));
payload_to_signal2k = 10 log(1512/(1512 + (131 + 45) • 16/9
+ 17 • 1)) = -0.857 dB;
payload_to_signal8k = 10 log(6048/(6048 + (524 + 177)
• 16/9 + 68 • 1)) = -0.854 dB;
The level of the DVB-T payload carriers alone is thus about 0.86 dB below the total carrier level.
Mapping of the constellation diagrams for QPSK, 16QAM and 64QAM
is another DVB-T system parameter. The mapping tables describe the bit
allocation to the respective constellation diagrams. The mapping tables below are layed out with the LSB (bit 0) on the left and the respective MSB
390
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
on the right. Therefore, the order from left to right is bit 0, bit 1 for QPSK,
bit 0, bit 1, bit 2, bit 3 for 16QAM and bit 0, bit 1, bit 2, bit 3, bit 4, bit 5
for 64QAM.
QPSK
10
00
11
01
16QAM
1000
1010
0010
0000
1001
1011
0011
0001
1101
1111
0111
0101
1100
1110
0110
0100
64QAM
100000
100010
101010
101000
001000
001010
000010
000000
100001
100011
101011
101001
001001
001011
000011
000001
100101
100111
101111
101101
001101
001111
000111
000101
100100
100110
101110
101100
001100
001110
000110
000100
110100
110110
111110
111100
011100
011110
010110
010100
110101
110111
111111
111101
011101
011111
010111
010101
110001
110011
111011
111001
011001
011011
010011
010001
110000
110010
111010
111000
011000
011010
010010
010000
Fig. 20.10. DVB-T mapping tables
20.5 The DVB-T Modulator and Transmitter
Having dealt with the DVB-T standard and all its system parameters in detail, the DVB-T modulator and transmitter can now be discussed. A
DVB-T modulator can have one or two transport stream inputs followed
by forward error correction (FEC) and this only depends on whether this
modulator supports hierarchical modulation or not. If hierarchical modulation is used, both FEC stages are completely independent of one another
but are completely identical as far as their configuration is concerned. One
transport stream path with FEC is called the high priority path (HP) and
20.5 The DVB-T Modulator and Transmitter
391
the other one is the low priority path (LP). Since the two FEC stages are
completely identical with the FEC of the DVB-S satellite standard, discussed in the relevant chapter (Chapter 14), they do not need to be discussed in detail here.
Mem1
IQ
mod.
IFFT
D
A
IF
RF
RF
Mem2
Mem1
D
IFFT
D
Mem2
A
IQ
mod.
RF
A
Direct modulation
Fig. 20.11. Possible implementations of a DVB-T modulator
The modulator locks to the transport stream, present at the transport
stream input, in the baseband interface. It uses for this the sync byte which
has a constant value of 0x47 at intervals of 188 bytes. To carry also longterm time stamps in the transport stream, every eighth sync byte is then inverted and becomes 0xB8. This is followed by the energy dispersal stage
which is synchronized by these inverted sync bytes both at the transmitting
end and at the receiving end. Following this, initial error control is performed in the Reed Solomon encoder. The TS packets are now expanded
by 16 bytes error protection. After this block coding, the data stream is interleaved in order to be able to break up error bursts during the deinterleaving at the receiver end. In the convolutional encoder, additional
error protection is added which can be reduced again in the puncturing
stage.
392
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
Up to this point, both HP and LP paths are absolutely identical but may
have different code rates. The error-controlled data of the HP and LP
paths, or the data of the one TS path in the case of non-hierarchical modulation, then pass into the de-multiplexer where they are then divided into 2,
4 or 6 outgoing data streams depending on the type of modulation (2 paths
for QPSK, 4 for 16QAM and 6 for 64QAM). The divided data streams
then pass into a bit interleaver where 126-bit-long blocks are formed which
are then interleaved on each path. In the symbol interleaver following, the
blocks are then again mixed block by block and the error-controlled data
stream is distributed uniformly over the entire channel. Adequate error
control and good distribution over the DVB-T channel are the prerequisites
for COFDM to function correctly. Together, this is then COFDM - Coded
Orthogonal Frequency Division Multiplex. After that, all the payload carriers are then mapped depending on whether hierarchical or nonhierarchical modulation is used, and on the factor α being = 1, 2 or 4. This
results in two tables, namely that for the real part Re(f) and that for the
imaginary part Im(f). However, they also contain gaps into which the pilots and the TPS carriers are then inserted by the frame adaptation block.
The complete tables, comprising 2048 and 8192 values, respectively, are
then fed into the heart of the DVB-T modulator, the IFFT block.
After that, the COFDM signal is available separated into real and imaginary part in the time domain. The 2048 and 8192 values, respectively, for
real and imaginary part in the time domain are then temporarily stored in
buffers organized along the lines of the pipeline principle. I.e., they are alternately written into one buffer whilst the other one is being read out.
During read-out, the end of the buffer is read out first as a result of which
the guard interval is formed. To obtain a better understanding of this section, special reference is made to the chapter on COFDM. The signal is
then usually digitally filtered at the temporal I/Q level (FIR filter) to provide for better attenuation of the shoulders.
The signal is now pre-equalized in a power transmitter in order to compensate for nonlinearities in the output stage. At the same time it is clipped
in order to limit the DVB-T signal with respect to its crest factor since otherwise the output stages could be destroyed because of the very high crest
factor of the COFDM signal due to its very high and very low amplitudes.
The position of the I/Q modulator depends on how the DVB-T modulator or transmitter is implemented in practice. The signal is either digital/analog converted separately for I and Q at the I/Q level and then supplied to an analog I/Q modulator which allows direct mixing to RF in
accordance with the principle of direct modulation, a principle commonly
used at present. The other approach is to remain at the digital level up to
and including the I/Q modulator and then to perform the D/A conversion.
20.6 The DVB-T Receiver
393
Demapper
Time
sync
Analog frontend
Channel
decoder
This, however, requires a further converter stage from a lower intermediate
frequency to the final RF which is more complex and costs more and,
therefore, is usually avoided today. On the other hand, this advantage is
gained at the expense of the possibly unpleasant characteristics of an analog I/Q modulator, the presence of which can virtually always be detected
in the output signal. Given the correct implementation, however, it is possible to manage direct modulation from baseband to RF (Fig. 20.11.).
TS
RF
IF
SAW
filter
(BP)
A
Delay
D
FFT
FIR
90
LO
Clock
(fZF2 = fs/4)
NCO
fs = 4 * 32/7 MHz
Scatt.
pilots
Cont.
pilots
TPS
carrier
Chnnel
corr.
FFT window
Low
pass
filter
Chan.
estim.
Frequ.
corr.
TPS
dec.
Fig. 20.12. Block diagram of a DVB-T receiver (part 1)
20.6 The DVB-T Receiver
One may think that the DVB-T modulator is a rather complex device but
the receiving end is even more complicated. Due to the high packing density of modern ICs, however, most of the modules of the DVB-T receiver
(Fig. 20.12.) can be accommodated in a single chip today.
The first module of the DVB-T receiver is the tuner. It is used for converting the RF of the DVB-T channel down to IF. In its construction, a
DVB-T tuner only differs in being required to have a much better phase
noise characteristic. The tuner is followed by the DVB-T channel at 36
MHz band center. This also corresponds to the band center of an analog
394
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
TV channel with a bandwidth of 8 MHz. However, In analog television,
everything is referred to the vision carrier frequency which is 38.9 MHz at
intermediate frequency. In digital television, i.e. in DVB-S, DVB-C and
also in DVB-T, it is the channel center frequency which is considered to be
the channel frequency. At intermediate frequency, the signal is bandpass
filtered to a bandwidth of 8, 7 or 6 MHz, using surface acoustic wave
(SAW) filters. In this frequency range, the filters can be implemented easily with the characteristics required for DVB-T. Following this bandpass
filtering, the adjacent channels are suppressed to an acceptable degree. An
SAW filter has minimum phase shift, i.e. there is no group delay distortion, only amplitude and group delay ripple.
In the next step, the DVB-T signal is converted down to a lower, second
IF at approx. 5 MHz. This is frequently an IF of 32/7 MHz = 4.571429
MHz. After this mixing stage, all signal components above half the sampling frequency are then suppressed with the aid of a low-pass filter in order to avoid aliasing effects. This is followed by analog/digital conversion.
The A/D converter is usually clocked at exactly four times the second IF,
i.e. at 4 • 32/7 = 18.285714 MHz. This is necessary in order to be able to
use the so-called fs/4 method for I/Q demodulation in the DVB-T modulator (see chapter on I/Q modulation). Following the A/D converter, the data
stream, which is now available with a data rate of about 20 Megawords/s,
is supplied to the time synchronization stage, among others. In this stage,
autocorrelation is used to derive synchronization information. Using autocorrelation, signal components are detected which exist in the signal several times and in the same way. Since in the guard interval, the end of the
next symbol is repeated before each present symbol, the autocorrelation
function will supply an identification signal in the area of the guard intervals and in the area of the symbols. The autocorrelation function is then
used to position the FFT sampling window into the area of guard interval
plus symbol free of inter-symbol interference and this positioning control
signal is fed into the FFT processor in the DVB-T receiver.
In parallel with the time synchronization, the data stream coming from
the A/D converter is split into two data streams by a changeover switch.
E.g., the odd-numbered samples pass into the upper branch and the evennumbered ones pass into the lower branch, producing two data streams
with half the data rate in each case. However, these streams are offset from
one another by half a sampling clock cycle. To eliminate this offset, the intermediate values are interpolated by means of an FIR filter, e.g. in the
lower branch. This filter, in turn, causes a basic delay of, e.g. 30 clock periods or more which must be replicated in the upper branch by using simple shift registers. The two data streams are then fed to a complex mixer
which is supplied with carriers by a numerically controlled oscillator
20.6 The DVB-T Receiver
395
(NCO). This mixer and the NCO are then used for correcting the frequency
of the DVB-T signal but because the oscillators lack accuracy, the receiver
must also be locked to the transmitted frequency by means of automatic
frequency control (AFC). This is done by the AFC evaluating the continual
pilots after the Fast Fourier Transform (FFT). If the receiver frequency differs from the transmitted frequency, all the constellation diagrams will rotate more or less quickly clockwise or anticlockwise. The direction of rotation simply depends on whether the deviation is positive or negative and
the speed depends on the magnitude of the error. It is then only necessary
to measure the position of the continual pilots in the constellation diagram.
The only factor of interest with respect to the frequency correction is the
phase difference of the continual pilots from symbol to symbol, the aim
being to reduce this phase difference to zero. The phase difference is a direct controlled variable for the AFC, i.e. the NCO frequency is changed
until the phase difference becomes zero. The rotation of the constellation
diagrams is then stopped and the receiver is locked to the transmitted frequency.
The FFT signal processing block, the sampling window of which is controlled by the time synchronization, transforms the COFDM symbols back
into the frequency domain, providing again 2048 or 8192 real and imaginary parts. However, these do not as yet correspond directly to the carrier
constellations. Since the FFT sampling window is not placed precisely
over the actual symbol, there exists a phase shift in all COFDM subcarriers, i.e. all constellation diagrams are twisted. This means that the continual and scattered pilots are no longer located on the real axis, either, but
somewhere on a circle, the radius of which corresponds to the amplitude of
these pilots. Furthermore, channel distortions must be expected due to
echoes or amplitude response or group delay. This, in turn, means that the
constellation diagrams can also be distorted in their amplitude and can be
additionally twisted to a greater or lesser extent. However, the DVB-T signal carries a large quantity of pilot signals which can be used as measuring
signal for channel estimation and channel correction in the receiver. Over
the period of twelve symbols, scattered pilots will have come to rest at
every third carrier position, i.e. information about the distortion in the
channel is available at every third carrier position. Measuring the amplitudes and phase distortion of the continual and scattered pilots enables the
correction function for the channel to be calculated, rotating the constellation diagrams back to their nominal position. In addition, the amplitude
distortion is removed and the constellation diagrams are compressed or
expanded in such a way that the pilots come to rest at the correct position
at their nominal position on the real axis.
396
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
Knowing about the operation of channel estimation and correction is of
importance to understanding test problems in DVB-T. From the channel
estimation data, it is possible to deduce both a large amount of test information in the DVB-T test receiver (channel transfer function, impulse response etc) and problems in the DVB-T modulator (I/Q modulator, center
carrier).
Symb./
FEC bit dedata in interfrom leaver
demapper
Viterbidec.
Conv.
deinterleaver
ReedSolom.
dec.
Energy
disp.
rem.
Sync.
inv.
rem.
Baseband
interf.
TS
out
Code rate
1/2...(3/4)...7/8
From TPS decoder
Fig. 20.13. Block diagram of the DVB-T receiver (part 2), channel decoding
In parallel with the channel correction, the TPS carriers are decoded in
the uncorrected channel. The transmission parameter signalling carriers do
not require channel correction since they are differentially encoded, the
modulation of the TPS carriers being DBPSK (differential bi-phase shift
keying). Each symbol contains a large number of TPS carriers and each
carrier carries the same information. The respective bit to be decoded is
determined by differential decoding with respect to the previous symbol
and by majority voting within a symbol. In addition, the TPS information
is error-protected. Therefore, the TPS information can be evaluated correctly for the DVB-T transmission before the threshold to the “fall off the
cliff” is reached. The TPS information is needed by the de-mapper following the channel correction, and also by the channel decoder. The TPS carriers make it possible to derive the currently selected type of modulation
(QPSK, 16QAM or 64QAM) and the information about the presence of hierarchical modulation. The de-mapper is then correspondingly set to the
correct type of modulation, i.e. the correct de-mapping table is loaded. If
hierarchical modulation is provided, a decision about which path (high priority (HP) or low priority (LP)) is to be decoded must be made in dependence on the channel bit error ratio, either manually or automatically. Following the demapper, the data stream is available again and is provided for
channel decoding.
Apart from the symbol and bit de-interleaver, the channel decoder (Fig.
20.13.) is configured exactly the same as that for the DVB-S satellite TV
20.6 The DVB-T Receiver
397
standard. The de-mapped data pass from the de-mapper into the symbol
and bit de-interleaver where they are resorted and fed into the Viterbi decoder. At the locations where bits have been punctured, dummy bits are
inserted again. These are dealt with similarly to errored bits by the Viterbi
decoder which then attempts to correct the first errors in accordance with
methods known from the trellis decoder.
MPEG-2
TS
DVB-T
Tuner
TP
SAW
Video
MPEG-2
A/D
decoder
demod.
Audio
I2C bus
Microprocessor
Keyboard /
remote
control
Fig. 20.14. Block diagram of a DVB-T set-top box
The Viterbi decoder is followed by the convolutional deinterleaver
which breaks up error bursts by undoing the interleaving. This makes it
easier for the Reed Solomon decoder to correct bit errors. The Reed Solomon decoder corrects up to 8 bit errors per packet with the aid of the 16 error control bytes. If there are more than 8 errors per packet, the ‘transport
error indicator’ is set to one and then this transport stream packet cannot be
processed further in the MPEG-2 decoder and error masking must be carried out. As well, the energy dispersal must then be undone. This stage is
synchronized by the inverted sync bytes and this sync byte inversion must
also be undone, after which the MPEG-2 transport stream is available
again.
A practical DVB-T receiver (Fig. 20.14.) has only a few discrete components such as the tuner, SAW filter, the mixing oscillator for the 2nd IF
and the low-pass filter. These are followed by a DVB-T demodulator chip
which contains all modules of the DVB demodulator after the A/D converter. The transport stream coming out of the DVB-T demodulator is fed
398
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
into the downstream MPEG-2 decoder where it is decoded back into video
and audio. All these modules are controlled by a microprocessor via an
I2C bus.
20.7 Interference on the DVB-T Transmission Link and its
Effects
Terrestrial transmission paths are subject to numerous influences (Fig.
20.15.). Apart from additive white Gaussian noise, these are mainly the
many echoes, i.e. the multi-path reception which makes this type of transmission so very problematic. Terrestrial reception is easy or difficult depending on the echo situation.
Echos (multipath reception)
Interferer
Noise (AWGN)
Doppler shift
DVB-T
modulator
& transmitter
Crest factor limitation
Intermodulation
Noise
IQ errors
Interferer
Fig. 20.15. Interferences on the DVB-T transmission link
The quality of the transmission link is also determined by the DVB-T
modulator and transmitter. The high crest factor of COFDM transmissions
results in special requirements even at the transmitting end. In theory, the
crest factor, i.e. the ratio between the maximum peak amplitude and the
RMS value of DVB-T signals is of the order of magnitude of 35 to 41 dB
but it would not be possible to operate any practical power amplifier with
these crest factors. Sooner or later, they would lead to its destruction. In
practice, therefore, the crest factor is limited to about 12 to 13 dB before
the DVB-T signal is fed into the power amplifier. However, this leads to a
poor shoulder attenuation in the DVB-T signal and, in addition, in-band
noise of the same order of magnitude as the shoulder attenuation is pro-
20.7 Interference on the DVB-T Transmission Link and its Effects
399
duced due to intermodulation and cross-modulation. The shoulder attenuation is then about 38…40 dB. To bring this shoulder attenuation back to a
reasonable order of magnitude, passive band-pass filters tuned to the DVBT channel are connected downstream (Fig. 20.16.). This again provides a
shoulder attenuation of better than 50 dB (critical mask). But there is nothing that can be done against the in-band carrier/noise ratio of about 38…40
dB now present. These interference products are the result of the clipping
required for reducing the crest factor and will now determine the performance of the DVB-T transmitter. I.e., every DVB-T transmitter will exhibit
a C/N ratio of the order of about 38…40 dB.
Today, direct modulation is used in virtually every DVB-T modulator,
i.e. the signal is converted directly from the digital baseband into RF as a
result of which analog I/Q modulators are used. In consequence, this circuit section, too, which is now no longer operating with theoretical perfection, has adverse effects on the signal quality, resulting in I/Q errors such
as amplitude imbalance, I/Q phase errors and lack of carrier suppression.
It is the art of the makers of modulators to keep these influences to a
minimum. However, the presence of an analog I/Q modulator in the DVBT transmitter is always detectable by measuring instruments as will be seen
later in the chapter on test engineering. As well, the finite quality of the
signal processing in the DVB-T modulator also results in the creation of
noise-like interferers. Further noise occurs on the transmission link in dependence on the conditions of reception. Similarly, multiple echoes and sinusoidal or impulse-like interferers can be expected and echoes, in turn,
can lead to frequency- and location-selective fading.
S/N
Fig. 20.16. Shoulder attenuation after clipping and after bandpass filtering
To calculate the crest factor in COFDM signals:
The crest factor is usually defined as:
400
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
cfu = 20 log(Upeak/URMS);
Power meters and spectrum analyzers are sometimes also calibrated to
the following definition:
cfp = 10 log(PEP)/PAVG;
where PEP is the peak envelope power (Upeak/√2)2/ zo;
and PAVG = URMS2/ zo;
The two crest factor definitions thus differ by 3 dB:
cfu = cfp.+ 3 dB;
The crest factor of COFDM signals is calculated as follows:
The maximum peak voltage is obtained by adding together the peak
amplitudes of all single carriers:
Upeak = N • Upeak0;
where Upeak0 is the peak amplitude of a single COFDM carrier
and N is the number of COFDM carriers used.
The RMS value of a COFDM signal is calculated from the quadratic
mean as:
2
U RMS = N ⋅ U RMS 0 ;
where URMS is the RMS voltage of a single COFDM carrier
U RMS 0 =
U peak 0
2
2
;
The RMS value of the COFDM signal is then:
U RMS = N ⋅
U peak 0
2
2
;
20.7 Interference on the DVB-T Transmission Link and its Effects
401
Inserting into the equation the maximum peak value occurring when all
individual carriers are superimposed and the RMS value of the total signal
provides:
cf COFDM = 20 log(
U peak
U RMS
) = 20 log(
N ⋅ U peak 0
N⋅
U peak 0
2
);
2
This, in turn, can be transformed and simplified to become:
cf COFDM = 20 log(
N
= 20 log 2 N = 10 log(2 N );
N
2
The theoretical crest factors in DVB-T are then
cfDVB-T2K = 35 dB;
in 2K mode with 1705 carriers used, and
cfDVB-T8K = 41 dB;
in 8K mode with 6817 carriers used.
It must be noted that these are theoretical values which, due to the limited resolution of the signal processing and the clipping, cannot occur in
practice. Practical values are of the order of magnitude of 13 dB (DVB-T
power transmitter) to about 15 dB (with modulators without clipping).
In the following paragraphs, the DVB-T transmission path itself will be
considered in greater detail. In the ideal case, exactly one signal path arrives at the receiving antenna. The signal is then only attenuated to a
greater or lesser extent and is merely subjected to additive white Gaussian
noise (AWGN). This channel with a direct view of the transmitter is called
a Gaussian channel and provides the best conditions of reception for the
receiver (Fig. 20.17.).
If multiple echoes are added to this direct signal path, the conditions of
reception become much more difficult. This channel with a direct line of
sight and a defined number of multiple echoes, which can be simulated as
a mathematical channel model, is called a Ricean channel (Fig. 20.18.).
402
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
If then the direct line of sight to the transmitter, i.e. the direct signal
path, is also blocked, the channel is called a Rayleigh channel (Fig.
20.19.). This represents the worst conditions of stationary reception.
Direct view to transmitter,
no echos
Fig. 20.17. Gaussian channel
Direct view to transmitter
and multipath reception
Fig. 20.18. Ricean channel
No direct view to transmitter,
only multipath reception
Fig. 20.19. Rayleigh channel
20.7 Interference on the DVB-T Transmission Link and its Effects
403
If, for instance, the receiver is moving at a certain speed away from the
transmitter or towards the transmitter (Fig. 20.20.), a negative or positive
frequency shift Δf will occur due the Doppler effect. This frequency shift
by itself does not present any problems to the DVB-T receiver which will
compensate for it by means of its AFC. It can be calculated from the speed
of movement, the transmitting frequency and the velocity of light.
V
Fig. 20.20. Doppler effect
V
Fig. 20.21. Doppler effect in combination with multipath reception
The following applies:
Δf = v • (f/c) • cos(ϕ);
where
v is the speed,
f the transmitting frequency,
c the velocity of light (299792458 m/s) and
ϕ the angle of incidence of the echo in relation to the direction
of movement.
404
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
Example: At a transmitting frequency of 500 MHz and a speed of 200
km/h, the Doppler shift is 94 Hz.
If, however, multiple echoes are added (Fig. 20.21.), the COFDM spectrum becomes smeared. This smearing is due to the fact that the mobile receiver is both moving towards signal paths and moving away from other
sources. I.e. there are now spectral COFDM combs which are shifting upward and downward. Due to its subcarrier spacing, which is narrower by a
factor of 4, the 8K mode is much more sensitive to such smearing in the
frequency domain than the 2K mode. The 2K mode is thus the better
choice for mobile reception, although DVB-T was originally not intended
for mobile reception.
Considering then the behavior of the DVB-T receiver in the presence of
noise. More or less noise in the DVB-T channel leads to more or fewer bit
errors during the reception. The Viterbi decoder can correct more or fewer
of these bit errors depending on the code rate selected in the convolutional
encoder. In principle, the same rules apply to DVB-T as do for a single
carrier method (DVB-C or DVB-S), i.e. the same “waterfall” curves of bit
error ratio vs. signal/noise ratio apply. The only caution is advised with respect to the signal/noise ratio which is also often called carrier/noise ratio.
The two differ slightly in DVB-T, the reason being the power in the pilot
carriers and auxiliary carriers (continual and scattered pilots and TPS carriers). To determine the bit error rate in DVB-T, only the power in the actual payload carriers can be used as the signal power. In DVB-T, the difference between the overall carrier power and the power in the pure
payload carriers is 0.857 dB in the 2K mode and 0.854 dB in the 8K mode
but the noise bandwidth of the pure payload carriers is reduced with respect to the overall signal.
The reduced noise bandwidth of the payload carriers is:
10 log(1512/1705) = -0.522 dB; in 2K mode
and
10 log(6048/6917) = -0.520 dB; in 8K mode.
Thus, the difference between C/N and S/N in DVB-T is:
C/N - S/N = -0.522 dB –(-0.857 dB) = 0.34 dB; in 2K mode, and
C/N - S/N = -0.52 dB –(-0.854 dB) = 0.33 dB; in 8K mode.
20.7 Interference on the DVB-T Transmission Link and its Effects
405
From the S/N in Fig. 20.22., the bit error ratio before Viterbi, i.e. the
channel bit error ratio, can be determined. Fig. 20.22. only applies to nonhierarchical modulation since the constellation pattern can be expanded
with hierarchical modulation.
The theoretical minimum carrier-to-noise ratios for quasi error-free operation depend on the code rate both in DVB-T and in DVB-S. In addition,
the type of modulation (QPSK, 16QAM, 64QAM) and the type of channel
(Gaussian, Ricean, Rayleigh) have an influence. The theoretical minimum
C/Ns are listed below for the case of non-hierarchical coding.
BER
1E-1
1E-2
1E-3
QPSK
1E-4
1E-5
16QAM
1E-6
1E-7
64QAM
0
5
10
15
20
25
30 S/N[dB]
Fig. 20.22. Bit error ratio in DVB-T as a function of S/N in QPSK, 16QAM and
64QAM with non-hierarchical modulation
Table 20.14. Minimum C/N required with non-hierachical modulation
Modulation
Code rate
QPSK
1/2
2/3
3/4
5/6
7/8
1/2
2/3
3/4
5/6
7/8
1/2
2/3
3/4
5/6
7/8
16QAM
64QAM
Gaussian channel
[dB]
3.1
4.9
5.9
6.9
7.7
8.8
11.1
12.5
13.5
13.9
14.4
16.5
18.0
19.3
20.1
Rice channel
[dB]
3.6
5.7
6.8
8.0
8.7
9.6
11.6
13.0
14.4
15.0
14.7
17.1
18.6
20.0
21.0
Rayleigh channel
[dB]
5.4
8.4
10.7
13.1
16.3
11.2
14.2
16.7
19.3
22.8
16.0
19.3
21.7
25.3
27.9
406
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
Thus, the demands for a minimum C/N fluctuate within a wide range
from about 3 dB for QPSK with a code rate of 1/2 in a Gaussian channel
up to about 28 dB for 64QAM with a code rate of 7/8 in a Rayleigh channel. Practical values are about 18 to 20 dB (64QAM, code rate 2/3 or 3/4)
for stationary reception and about 11 to 17 dB (16QAM, code rate 2/3 or
3/4) for mobile reception.
Table 20.15. Theoretical minimum C/N with hierarchical modulation (QPSK,
64QAM, α = 2); low priority path (LP)
Modulation
Code rate
QPSK
1/2
2/3
3/4
1/2
2/3
3/4
5/6
7/8
64-QAM
Gaussian channel
[dB]
6.5
9.0
10.8
16.3
18.9
21.0
21.9
22.9
Rice channel
[dB]
7.1
9.9
11.5
16.7
19.5
21.6
22.7
23.8
Rayleigh channel
[dB]
8.7
11.7
14.5
18.2
21.7
24.5
27.3
29.6
20.8 DVB-T Single-Frequency Networks (SFN)
COFDM is well suited to single frequency operation. As the name indicates, in single frequency operation, all transmitter operate at the same frequency which makes for great economy with regard to frequency resources. All transmitters radiate the identical signal and have to operate in
complete synchronism with each other. Signals from adjacent signals are
seen by a transmitter as if they were simply echoes. Frequency synchronization is the easiest condition because frequency accuracy and stability had
to meet high demands even in analog television. In DVB-T, the transmitter
RF is locked to the best reference available: the signal from the GPS
(Global Positioning System) which is available throughout the world and
is now also used for synchronizing the transmitting frequencies of a
DVB-T single-frequency network. The GPS satellites radiate a 1 pps
(pulse per second) signal to which a 10 MHz oscillator in professional
GPS receivers is locked which, in turn, acts as reference signal for the
DVB-T transmitters.
However, there is also a strict requirement with respect to the maximum
distance between transmitters (Fig. 20.23. and Tables 20.16, 20.17. and
20.18.). This distance is related to the length of the guard interval and the
velocity of light, i.e. the associated signal delay. Intersymbol interference
can only be avoided if in the case of multipath reception, the delay on any
20.8 DVB-T Single-Frequency Networks (SFN)
407
path is no longer than the length of the guard interval. The question about
what would happen if the signal received from a more distant transmitter
violates the guard interval is easily answered: it results in intersymbol interference which becomes noticeable as noise in the receiver.
Tx3, RF1
Tx1, RF1
Tx5, RF1
Playout
center
Distance
Tx2, RF1
Tx4, RF1
Fig. 20.23. DVB-T single-frequency network (SFN)
Table 20.16. Guard interval lengths for 8K, 2K modes and transmitter distances
(8 MHz channel)
Mode
2K
2K
2K
2K
8K
8K
8K
8K
Symbol
duration
µs
224
224
224
224
896
896
896
896
Guard
interval
ratio
1/4
1/8
1/16
1/32
1/4
1/8
1/16
1/32
Guard
interval
µs
56
28
14
7
224
112
56
28
Transmitter
distance
km
16.8
8.4
4.2
2.1
67.1
33.6
16.8
8.4
Signals from transmitters at greater distances must simply be attennuated sufficiently. The threshold for quasi error free operation is formed by
the same conditions as for pure noise. It is, therefore, of particular importance that the levels in a single-frequency network are calibrated correctly.
It is not the maximum transmitting power at every transmitting site which
is required but the correct one. Planning of the network requires topographical information.
408
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
In many cases, however, network planning is relatively simple since
mostly only small regional single-frequency networks with only very few
transmitters are set up.
Table 20.17. Guard interval lengths for 8K, 2K modes and transmitter distances
(7 MHz channel)
Mode
2K
2K
2K
2K
8K
8K
8K
8K
Symbol
duration
µs
256
256
256
256
1024
1024
1024
1024
Guard
interval
ratio
1/4
1/8
1/16
1/32
1/4
1/8
1/16
1/32
Guard
interval
µs
64
32
16
8
256
128
64
32
Transmitter
distance
km
19.2
9.6
4.8
2.4
76.7
38.4
19.2
9.6
Table 20.18. Guard interval lengths for 8K, 2K modes and transmitter distances
(6 MHz channel)
Mode
2K
2K
2K
2K
8K
8K
8K
8K
Symbol
duration
µs
299
299
299
299
1195
1195
1195
1195
Guard
interval
1/4
1/8
1/16
1/32
1/4
1/8
1/16
1/32
Guard
interval
µs
75
37
19
9
299
149
75
37
Transmitter
distance
km
22.4
11.2
5.6
2.8
89.5
44.8
22.4
11.2
The velocity of light is c = 299792458 m/s which results in a signal delay per kilometer transmitter distance of t1km = 1000 m/c = 3.336 µs. Since
in the 8K mode, the guard interval is longer in absolute terms, it is mainly
this mode which is provided for single frequency operation.
Long guard intervals are provided for single frequency networks. Medium-length guard intervals are used in regional networks. The short guard
intervals, finally, are provided for local networks or used outside of single
frequency networks.
20.8 DVB-T Single-Frequency Networks (SFN)
409
In a single frequency network, all the individual transmitters must be
synchronized with one another. The program contribution is injected from
the playout center in which the MPEG-2 multiplexer is located, e.g. via
satellite, optical fiber or microwave link. It is clear that the MPEG-2 transport streams are subject to different feed line delays due to different path
lengths. However, it is necessary that in each DVB-T modulator in an SFN
network the same transport stream packets are processed into COFDM
symbols. Every modulator must perform every operating step completely
synchronously with all the other modulators in the network. The same
packets, the same bits and the same bytes must all be processed at the same
time. Every DVB-T transmitter site must broadcast absolutely identical
COFDM symbols at exactly the same time.
The DVB-T modulation is structured in frames, one frame being composed of 68 DVB-T COFDM symbols. Within a frame, the complete TPS
information is transmitted and the scattered pilots are scattered over the entire DVB-T channel. Four such frames, in turn, make up one superframe.
Frame structure of DVB-T:
•
•
68 COFDM symbols = 1 frame
4 frames = 1 superframe
One superframe in DVB-T accommodates an integer number of MPEG2 transport stream packets, as follows:
Table 20.19. Number of transport stream packets per superframe
Coderate
1/2
2/3
3/4
5/6
7/8
QPSK
2K
252
336
378
420
441
QPSK
8K
1008
1344
1512
1680
1764
16QAM
2K
504
672
756
840
882
16QAM
8K
2016
2688
3024
3360
3528
64QAM
2K
756
1008
1134
1260
1323
64QAM
8K
3024
4032
4536
5040
5292
In consequence, a superframe in a single frequency network must be
composed of absolutely identical transport stream packets and each modulator in the SFN must generate and broadcast the superframe at the same
time.
These modulators must, therefore, be synchronized with one another
and, in addition, the differences in the feed line delays must be equalized
statically and dynamically. To achieve this, packets with time stamps are
inserted into the MPEG-2 transport stream in the playout center. These
packets are special transport stream packets which are configured similarly
410
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
to an MPEG-2 table (PSI/SI). For this purpose, the transport stream is divided into sections, the lengths of which are selected to be approximately a
half second because they must correspond to a certain integral number of
transport stream packets fitting into a certain integral number of superframes. These sections are called megaframes.
A megaframe is composed of an integral number of superframes, as follows:
•
•
1 megaframe = 2 superframes in 8K mode
1 megaframe = 8 superframes in 2K mode
The 1 pps signals of the GPS satellites are also used for synchronizing
the timing of the DVB-T modulators. In the case of a single frequency
network, there is a professional GPS receiver outputting both a 10 MHz
reference signal and this 1 pps time signal at every transmitter site and at
the playout center (Fig. 20.24.) where the multiplexed stream is assembled.
Tx3, RF1
1pps pulse
GPS
GPS:
1pps
Global
Positioning
System
Playout
center
Tx1, RF1
pulse
MIP
Tx5, RF1
MPEG-2 TS
MIP
inserter
Tx2, RF1
Tx4, RF1
Fig. 20.24. DVB-T distribution network with MIP insertion
At the multiplexer site there is a so-called MIP inserter which inserts
this special transport stream packet into one megaframe in each case,
which is why this packet is called the megaframe initializing packet (MIP).
The MIP has a special PID of 0x15 so that it can be identified and it contains time reference and control information for the DVB-T modulators.
Among other things, it contains the time counting back to the time the last
1 pps pulse was received at the MIP inserter. This time stamp with a reso-
20.8 DVB-T Single-Frequency Networks (SFN)
411
lution of 100 ns steps is used for automatically measuring the feed distance. This time information is evaluated by the SFN adapter which automatically corrects the delay from the playout center to the transmitter site
by means of a buffer store. It also requires information about the maximum
delay in the network. Given this information, which can either be input
manually at every transmitter site or is carried in the MIP packet, each
SFN adapter adjusts itself to this time. The MIP packet also contains a
pointer to the start of the next megaframe in numbers of TS packets. Using
this pointer information, each modulator is then able to start a megaframe
at the same time.
GPS
1pps
Puls
Pointer
Synchronization time stamp
...
MFP#0
MFP#1
MFP#2
MIP
MFP#0
...
MPEG-2 TS Packet
Megaframe
Fig. 20.25. Megaframe structure at transport stream level
The length of a megaframe depends on the length of the guard interval
and on the bandwidth of the channel. The narrower the channel (8, 7 or 6
MHz), the longer the COFDM symbols since the subcarrier spacing becomes less. Every DVB-T modulator can now be synchronized by means
of the information contained in the MIP packet. The MIP packet can always be transmitted at a fixed position in the megaframe but this position
is also allowed to vary. Table 20.20. contains a list of the exact lengths of
one megaframe.
Table 20.20. Duration of a megaframe
Guard
interval
1/32
1/16
1/8
1/4
8 MHz
channel
0.502656 s
0.517888 s
0.548352 s
0.609280 s
7 MHz
channel
0.574464 s
0.598172 s
0.626688 s
0.696320 s
6 MHz
channel
0.670208 s
0.690517 s
0.731136 s
0.812373 s
412
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
An MIP can also be used for transmitting additional information such as
the DVB-T transmission parameters which makes it possible to control and
configure the entire DVB-T SFN from one center. For example, it can be
used for changing the type of modulation, the code rate, the guard interval
length etc. However, although this is possible, it may not be supported by
every DVB-T modulator.
If the transmission of the MIP packets stops for some reason or if the information in the MIP packets is corrupted, the single frequency network
will lose synchronization. If a DVB-T transmitter detects that it has
dropped lock or that it has not received a GPS signal for some time and the
1 pps reference and the 10 MHz reference have, therefore, drifted, it has to
go off air or it will only be a source of noise in the single frequency network. Reliable reception is then only possible with directional reception
close to the transmitter. For this reason, the MIPs in the transport stream
arriving at the transmitter are often monitored using an MPEG-2 test decoder (see Fig. 20.26.).
MIP content
188 bytes
4 byte
TS header
184 byte
payload
13 bit packet identifier = PID
1 byte sync byte = 0x47
0x15
Fig. 20.26. Megaframe initializing packet
Fig. 20.27. (MIP = Megaframe Initializing Packet) clearly shows that
the multiplexed MPEG-2 stream is now carrying a further table-like
packet, namely the MIP packet, containing the synchronization time
stamp, the pointer and the maximum delay. It also contains the transmission parameters. It can also be seen that every transmitter in the link-up
20.8 DVB-T Single-Frequency Networks (SFN)
413
can be addressed. Like a table, the content of the MIP packet is protected
by a CRC checksum.
In addition, each transmitter can also be “pushed”, i.e. it is possible to
change the time when the COFDM symbol is broadcast. This will not push
the single frequency network out of synchronization but only vary the delay of the signals of the transmitters with respect to each other and can thus
be used for optimizing the SFN network. These time offsets are found in
the ‘TX time offset’ functions in Fig. 20.27. Shifting the broadcasting time
makes it appear to the receiver as if the geographic position of the respective transmitter has changed. This may be of interest if two transmitter in
an SFN are very far apart and are approaching the limit of the guard interval (e.g. DVB-T network Southern Bavaria with the Olympic tower in
Munich and the Mount Wendelstein transmitter at a distance d of 63 km)
or if the guard interval has been chosen to be very short for reasons of data
rates (e.g. Sydney, Australia, with g=1/16).
Fig. 20.27. MIP packet analysis [DVMD]
414
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
20.9 Minimum Receiver Input Level Required with DVB-T
To obtain error-free reception of a DVB-T signal, the minimum receive
level required must be present at the DVB-T receiver input. Below a certain signal level, reception breaks off and at the threshold blocking and
freezing effects will occur, above which reproduction is faultless. This section discusses the principles for determining this minimum level.
The minimum level in DVB-T is dependent on:
•
•
•
•
•
•
•
Type of modulation (QPSK, 16QAM, 64QAM)
Error protection used (code rate 1/2, 2/3, 3/4,... 7/8)
Channel model (Gaussian, Ricean, Rayleigh)
Bandwidth (8, 7, 6 MHz)
Ambient temperature
Actual receiver characteristics (noise figure of the tuner etc)
Multipath reception conditions
In principle, a minimum signal/noise ratio S/N is required which is
mathematically a function of some of the factors listed above. The theoretical S/N limits are listed in Table 20.14. in Section 20.7. As an example,
the 2 following cases will be considered:
Case 1: Ricean channel with 16QAM and code rate = 2/3, and
Case 2: Ricean channel with 64QAM and code rate = 2/3.
Case 1 corresponds to conditions adapted for a DVB-T network designed for portable indoor use (e.g. Germany) and Case 2 corresponds to
conditions adapted for a DVB-T network with parameters designed for
roof antenna reception (e.g. Sweden, Australia). Table 20.14. shows that
Case 1 requires a S/N of 11.6 dB, and
Case 2 requires a S/N of 17.1 dB.
The noise level N present at the receiver input is obtained from the following physical relation:
N[dBW] = -228.6 + 10 log(B/Hz) + 10 log((T/0C +273)) + F;
where
B = bandwidth in Hz;
20.9 Minimum Receiver Input Level Required with DVB-T
415
T = temperature in 0C;
F = noise figure of the receiver in dB;
The constant -228.6 dBW/K/Hz is the so-called Boltzmann's constant.
Assuming that:
ambient temperature T = 200C;
noise figure of the tuner = 7 dB;
receiver bandwidth B = 8 MHz;
then
N[dBW] = -228.6 + 10 log(8000000/Hz) + 10 log((20/0C+273)) + 7;
0 dBm @ 50 ohms = 107 dB V;
0 dBm @ 75 ohms = 108.8 dB V;
N = -98.1 dBm = -98.1 dBm + 108.8 dB = 10.7 dB V; (at 75 ohms)
Thus, the noise level present at the receiver input under these conditions
is 10.7 dB V.
For Case 1 (16QAM), the minimum receiver input level required is,
therefore:
S = S/N [dB] + N [dB V] = (11.6 + 10.7)[dB V] = 22.3 dB V;
For Case 2 (64QAM), the minimum receiver input level required is:
S = S/N [dB] + N [dB V] = (17.1 + 10.7)[dB V] = 27.8 dB V;
In practice it is found that these values can be met quite confortably
with only one signal path but as soon as several signal paths are present at
the receiver input (multipath reception), the required level is often higher
by up to 10 to 15 dB and varies greatly with different types of receiver.
The received level actually present is the result of:
•
•
•
•
Received signal strength present at the receiving location
Antenna gain
Polarization losses
Losses in the feed line from the antenna to the receiver
416
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
The following applies to the conversion of the antenna output level from
the field strength present at the receiving site:
E[dB V/m] = U[dB V] + k[dB];
k[dB] = (-29.8 + 20 log(f[MHz]) - g[dB];
where:
E = electrical field strength,
U = antenna output level,
k = k factor of the antenna,
f = received frequency,
g = antenna gain.
The level present at the receiver input as a result is then:
S[dB V] = U[dB V] - loss[dB];
Where 'loss' designates the implementation losses (antenna feeder etc.)
Considering now Case 1 (16QAM) and Case 2 (64QAM) at 3 frequencies:
a) f = 200 MHz,
b) f = 500 MHz,
c) f = 800 MHz.
The antenna gain is assumed to be g = 0 dB in each case (nondirectional rod antenna).
k factors of the antenna:
a) k = (-29.8 + 48) dB = 16.2 dB;
b) k = (-29.8 + 54) dB = 24.2 dB;
c) k = (-29.8 + 58.1) dB = 28.3 dB;
Field strengths for Case 1 (16QAM; minimum required level
U = S - loss = 22.3 dB V - 0dB = 22.3 V:
a) E = (22.3 + 16.2) dB V/m = 38.5 dB V/m;
b) E = (22.3 + 24.2) dB V /m = 46.5 dB V/m;
c) E = (22.3 + 28.3) dB V/m = 50.6 dB V/m;
20.9 Minimum Receiver Input Level Required with DVB-T
417
If a directional antenna with gain is used, e.g. a roof antenna, the following conditions are obtained.
a) with f = 200 MHz (assuming g = 6 dB), E = 32,5 dB V/m;
b) with f = 500 MHz (assuming g = 10 dB), E = 36,5 dB V/m;
c) with f = 800 MHz (assuming g = 10 dB), E = 40.6 dB V/m;
Field strengths for Case 2 (64QAM; minimum required level
U = S - loss = 27.8 dB V - 0dB = 27.8 V):
a) E = (27.8 + 16.2) dB V/m = 44.0 dB V/m;
b) E = (27.8 + 24.2) dB V /m = 52.0 dB V/m;
c) E = (27.8 + 28.3) dB V/m = 56.1 dB V/m;
If a directional antenna with gain is used, e.g. a roof antenna, the following conditions are obtained.
a) with f = 200 MHz (assuming g = 6 dB), E = 38.0 dB V/m;
b) with f = 500 MHz (assuming g = 10 dB), E = 42.0 dB V/m;
c) with f = 800 MHz (assuming g = 10 dB), E = 46.1 dB V/m;
Under free space conditions, the field strength at the receiving site can
be calculated as:
E[dB V/m] = 106.9 + 10 log(ERP[kW]) - 201g(d[km]);
where:
E = electrical field strength
ERP = effective radiated power, i.e. the transmitter power
plus antenna gain;
d = transmitter - receiver distance;
Under real conditions, however, much lower field strengths must be assumed because this formula does not take into consideration shading, multipath reception etc. The reduction depends on the topological conditions
(hills, mountains, buildings etc) and can be up to about 20 - 30 dB, but also
much more with complete shading.
Example (without reduction; a reduction of at least 20 dB is recommended):
ERP = 50 kW;
418
20 Terrestrial Transmission of Digital Television Signals (DVB-T)
d = 1 km; E = (106.9 + 10 log(50) – 20 log(1)) dB V/m
= 123.9 dB V/m;
d = 10 km; E = (106.9 + 10 log(50) – 20 log(1)) dB V/m
= 103.9 dB V/m;
d = 30 km; E = 94.4 dB V/m;
d = 50 km; E = 89.9 dB V/m;
d = 100 km; E = 83.9 dB V/m;
Since the plane of polarization was frequently changed from horizontal
to vertical at the transmitter site as part of the DVB-T conversion. There
may also be polarization losses of about 10…20 dB at the receiving antenna if this has not also been changed from horizontal to vertical.
If the DVB-T signal is received with an indoor antenna inside the house
attenuation due to the building must also be considered which amounts to
another 10 to 20 dB.
In Germany, the following field strength values were assumed with
16QAM, CR=2/3 as limit values for the field strength outside the building
during the simulation of the conditions of reception:
•
•
•
Reception with a roof antenna:
Reception with an outdoor antenna:
Reception with an indoor antenna:
approx. 55 dB V/m
approx. 65 dB V/m
approx. 75…85 dB V/m
Bibliography: [ETS300744], [REIMERS], [HOFMEISTER], [EFA],
[SFQ], [TR101190], [ETR290].
20.9 Minimum Receiver Input Level Required with DVB-T
419
Fig. 20.28. Medium-power DVB-T transmitter (Rohde&Schwarz)
Fig. 20.29. DVB-T mask filter, critical mask (dual mode filter, manufacturer’s
photo Spinner)
21 Measuring DVB-T Signals
The DVB-T standard and its complicated COFDM modulation method
have now been discussed thoroughly. The present chapter deals with
methods for testing DVB-T signals in accordance with the DVB Measurement Guidelines ETR 290 and also beyond these. The requirement for
measurements is much greater in DVB-T than in the other two transmission path systems DVB-C and DVB-S due to the highly complex terrestrial
transmission path, the much more complicated DVB-T modulator and the
analog IQ modulator used there in most cases. DVB-T measuring techniques must cover the following interference effects:
•
•
•
•
•
•
•
•
Noise (AWGN)
Phase jitter
Interferers
Multipath reception
Doppler effect
Effects in the single-frequency network
Interference with the adjacent channels (shoulder attenuation)
I/Q errors of the modulator:
- I/Q amplitude imbalance
- I/Q phase errors
- lack of carrier suppression
The test instruments used in DVB-T measuring techniques are essentially comparable to those used in broadband cable measuring techniques.
The following are required for measuring DVB-T signals:
•
•
•
A modern spectrum analyzer
a DVB-T test receiver with constellation analyzer
a DVB-T test transmitter for measurements on DVB-T receivers
The DVB-T test receiver is by far the most important measuring means
in DVB-T. Due to the pilot signals integrated in DVB-T, it allows the most
extensive analyses to be performed on the signal without using other aids,
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_21, © Springer-Verlag Berlin Heidelberg 2010
422
21 Measuring DVB-T Signals
the most important one of these being the analysis of the DVB-T constellation pattern. Although extensive knowledge in the field of DVB-C constellation analysis has been gathered since the 90s, simply copying it across
into the DVB-T world is not sufficient. This chapter deals mainly with the
special features of DVB-T constellation analysis, points out problems and
provides assistance in interpreting the results of the measurements.
In comparison with DVB-C constellation analysis, DVB-T constellation
analysis is not simply a constellation analysis on many thousands of subcarriers and many things do not lend themselves to being simply copied
across.
Fig. 21.1. shows the constellation diagram of a 64QAM modulation in
DVB-T. The positions of the scattered pilots and of the continual pilots (on
the left and on the right outside the 64QAM constellation diagram on the I
axis) and of the TPS carriers (constellation points inside the constellation
diagram, also on the I axis) can be easily seen. The scattered pilots are
used for channel estimation and correction and thus represent a checkpoint
in the constellation diagram which is always corrected to the same position. The transmission parameter signalling carriers (TPS) serve as a fast
information channel from the transmitter to the receiver. Apart from noise,
there are no further influences acting on the constellation diagram shown
(Fig. 21.1.).
Fig. 21.1. Constellation diagram in 64QAM DVB-T
21.1 Measuring the Bit Error Ratio
423
A DVB-T test receiver (Fig. 21.2.) can be used for detecting all influences acting on the transmission link. A DVB-T test receiver basically differs from a set-top box in the analog signal processing being of a much
higher standard and the I/Q data and the channel estimation data being accessed by a signal processor (DSP). The DSP then calculates the constellation diagram and the measurement values. In addition, the DVB-T signal
can be demodulated down to the MPEG-2 transport stream level.
IF1
RF/IF
down
conv./
tuner
RF
IF2
SAW
filter
Mixer
X
Anti
alias.
lowpass
A
MPEG-2
TS
DVB-T
dem.
D
I
Q
DSP
Noise
gen.
Display
Fig. 21.2. Block diagram of a DVB-T test receiver
21.1 Measuring the Bit Error Ratio
In DVB-T, as in DVB-S, there are 3 bit error ratios due to the inner and
outer error protection:
•
•
•
Bit error ratio before Viterbi
Bit error ratio before Reed Solomon
Bit error ratio after Reed Solomon
The error ratio of greatest interest and providing the most information is
the pre-Viterbi bit error ratio. It can be determined by applying the postViterbi data stream to another convolutional encoder of the same configuration as that at the transmitter end. If the data stream before Viterbi is
compared with that after the convolutional encoder - taking into consideration the delay of the convolutional encoder - the two are identical provided
424
21 Measuring DVB-T Signals
there are no errors. The differences, and thus the bit errors, are then determined by a comparator for the I and Q branch.
I
Viterbi
decoder
Data
Q
Conv.
coder
I
Delay
Q
Comparator
Bit error ratio
Fig. 21.3. Circuit for determining the pre-Viterbi bit error ratio
The bit errors counted are then related to the number of bits transmitted
within the corresponding period, providing the bit error ratio (BER)
BER = bit errors/transmitted bits;
The pre-Viterbi bit error ratio range is between 10-9 (transmitter output)
and 10-2 (receiver input with poor receiving conditions).
The Viterbi decoder can only correct some of the bit errors, leaving a residual bit error ratio before Reed Solomon. Counting the corrections of the
Reed Solomon decoder and relating them to the number of bits transmitted
within the corresponding period provides the pre-Reed Solomon bit error
ratio.
However, the Reed Solomon decoder is not able to correct all bit errors,
either, and this then results in errored transport stream packets. These are
flagged in the TS header (transport error indicator bit = 1). Counting the
errored transport stream packets enables the post-Reed Solomon bit error
ratio to be calculated.
A DVB-T test receiver will detect all 3 bit error ratios and indicate them
in one of the main measurement menus. It must be noted that with the relatively low bit error ratios usually available after the Viterbi and Reed-
21.2 Measuring DVB-T Signals Using a Spectrum Analyzer
425
Solomon decoders, measuring times of corresponding length in the range
of minutes up to hours must be selected.
Fig. 21.4. Bit error ratio measurement [EFA]
The measurement menu example [EFA] shows that all the important information about the DVB-T transmission is combined here. Apart from the
RF selected, it also shows the received level, the frequency deviation, all 3
bit error ratios and the decoded TPS parameters.
21.2 Measuring DVB-T Signals Using a Spectrum
Analyzer
A spectrum analyzer is very useful for measuring the power of the DVB-T
channel, at least at the DVB-T transmitter output. Naturally, one could
simply use a thermal power meter for this purpose but, in principle, it is
also possible to use a spectrum analyzer which will provide a good estimate of the carrier/noise ratio. Firstly, however, the power of the DVB-T
signal will now be determined. A COFDM signal looks like noise and has
a crest factor which is rather high. Due to its similarity with white Gaussian noise, its power is measured in a comparable way.
To determine the carrier power, the spectrum analyzer is set as follows:
On the analyzer, a resolution bandwidth of 30 kHz and a video bandwidth
426
21 Measuring DVB-T Signals
of 3 to 10 times the resolution bandwidth, i.e. 300 kHz, is selected. To
achieve a certain amount of averaging, a slow sweep time of 2000 ms is
set. These parameters are needed because we are using the RMS detector
of the spectrum analyzer. The following settings are then used:
•
•
•
•
•
•
•
Center frequency: center of the DVB-T channel
Span: 20 MHz
Resolution bandwidth: 30 kHz
Video bandwidth: 300 kHz (due to RMS detector and logarithmic
scale)
Detector: RMS
Sweep: slow (2000 ms)
Noise marker: channel center (resultant C’ value in dBm/Hz)
C[dBm] − 10 lg(
DVB − T − signal _ bandwidth
)[dB];
resolution _ bandwidth
Fig. 21.5. Spectrum of a DVB-T signal
The level indicated in the useful band of the DVB-T spectrum (Fig.
21.5.) depends on the choice of resolution bandwidth (RBW) of the spectrum analyzer (e.g. 1, 4, 10, 20, 30 kHz) with respect to the bandwidth of
the DVB-T signal (7.61 MHz, 6.66 MHz, 5.71 MHz). In the literature
(DVB-T standard, systems specifications), 4 kHz is often quoted as reference bandwidth but is not always supported by spectrum analyzers. At
4 kHz reference bandwidth, the level shown in the useful band is 38.8 dB
(7.61 MHz) or 32.2 dB, respectively, below the level of the DVB-T signal.
21.2 Measuring DVB-T Signals Using a Spectrum Analyzer
427
Table 21.1. Level of the useful band shown by the spectrum analyzer vs. signal
level
Resolution bandwidth
[kHz]
1
4
5
10
20
30
50
100
500
Attenuation [dB] in useful
band vs. DVB-T signal
level in the 7 MHz channel
38.8
32.8
31.8
28.8
25.8
24.0
21.8
18.8
11.8
Attenuation [dB] in useful band vs. DVB-T signal level in the 8 MHz
channel
38.2
32.2
31.2
28.2
25.8
24.0
21.8
18.8
11.8
To measure the power, the noise marker is used because of the noiselike signal. The noise marker is set to band center for this but the prerequisite is a flat channel which can always be assumed to exist at the transmitter. If the channel is not flat, other suitable measuring functions must be
used for measuring channel power but these depend on the spectrum analyzer.
The analyzer provides the C’ value as noise power density at the position of the noise marker in dBm/Hz, automatically taking into consideration the filter bandwidth and the characteristics of the logarithmatic amplifier of the analyzer. To bring the signal power density C’ into relation with
the Nyquist bandwidth BN of the DVB-T signal it is necessary to calculate
the signal power C as follows:
C = C’ + 10log(signal bandwidth/Hz) [dBm]
The signal bandwidth of the DVB-T signal is
•
•
•
7.61 MHz in the 8 MHz channel,
6.66 MHz in the 7 MHz channel,
5.71 MHz in the 6 MHz channel.
Example (8 MHz channel):
Measurement value of the noise marker:
Correction value at 7.6 MHz bandwidth:
Power in the DVB-T channel:
-100 dBm/Hz
+ 68.8 dB
- 31.2 dBm
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21 Measuring DVB-T Signals
Approximate Determination of the Noise Power N:
If it were possible to switch off the DVB-T signal without changing the
noise ratios in the channel, the noise marker in the center of the band
would provide information about the noise ratios in the channel. However,
this cannot be done so easily. Using the noise marker for measuring very
closely to the signal on the shoulder of the DVB-T signal will provide at
least a “good idea” about the noise power in the channel, if not a precise
measurement value. This is because it can be assumed that in the useful
band, the noise fringe continues similarly to that found on the shoulder.
The spectrum analyzer outputs the value N’ of the noise power density.
The noise power N in the channel with the bandwidth BC of the DVB-T
transmission channel is calculated from the noise power density N’ as follows:
N = N’ + 10log(BC/Hz); [dBm]
The channel bandwidth to be used is the actual bandwidth of the DVB-T
channel, e.g. 8 MHz.
Example:
Measurement value of the noise marker:
Correction value at 8 MHz bandwidth:
Noise power in the DVB-T channel:
-140 dBm/Hz
+ 69.0 dB
- 71.0 dBm
From this, the C/N value is obtained as:
C/N[dB] = C[dBm] - N[dBm]
In the example: C/N[dB] = -31.2 dBm - (-71.2 dBm) = 40 dB.
In estimating C/N in this way by means of the shoulders of the DVB-T
signal it is important that this measurement is made directly at the output
interfaces after the power amplifier and before any passive bandpass filters. Otherwise, only the shoulders lowered by the bandpass filter will be
seen. The author has repeatedly verified the validity of this measuring
method in comparisons with measurement results from a DVB-T test receiver.
21.3 Constellation Analysis of DVB-T Signals
429
21.3 Constellation Analysis of DVB-T Signals
The great difference between the constellation analysis of DVB-T signals
and DVB-C is that in DVB-T, many thousands of COFDM subcarriers are
analyzed. The carrier range must be selectable. Often, displaying all the
constellation diagrams (carriers No. 0 to 6817 or 0 to 1705, resp.) as constellation diagrams written on top of each other is of interest. The carrier
ranges can be selected in 2 ways:
•
•
start/stop carrier no.,
center/span carrier no.
Apart from the pure payload carriers, the pilot carriers and the TPS carriers can also be considered but no mathematical constellation analysis will
be performed on these carriers. In the following paragraphs, the individual
influences and measurement parameters will be discussed.
The following measurement values can be detected by using constellation analysis:
•
•
•
•
•
Signal/noise ratio S/N
Phase jitter
I/Q amplitude imbalance
I/Q phase error
Modulation error ratio MER
Fig. 21.6. Influence of noise
430
21 Measuring DVB-T Signals
21.3.1 Additive White Gaussian Noise (AWGN)
White noise (AWGN, Additive White Gaussian Noise) leads to cloudshaped constellation points. The larger the constellation point, the greater
the noise effect. The signal/noise parameter S/N can be determined by analyzing the distribution function (normal Gaussian distribution) in the decision field. The RMS value of the noise component corresponds to the standard deviation. Noise effects affect every DVB-T subcarrier and can also
be found on every subcarrier. The effects and measuring methods are
completely identical with the DVB-C methods.
Fig. 21.7. Effect of phase jitter
21.3.2 Phase Jitter
Phase jitter leads to a striated distortion in the constellation diagram. It is
caused by the oscillators in the modulator and also affects every carrier
and can also be found on every carrier.
Here, too, the measuring methods and effects are completely identical.
with those in DVB-C.
21.3.3 Interference Sources
Interference sources affect individual carriers or carrier ranges. They can
be noise-like and the constellation points become noise clouds, but they
can also be sinusoidal when the constellation points appear as circles.
21.3 Constellation Analysis of DVB-T Signals
431
21.3.4 Echoes, Multipath Reception
Echoes, i.e. multipath reception, lead to frequency-selective fading. There
is interference in individual carrier ranges but the information lost as a result can be restored again due to the interleaving across the frequency and
the large amount of error protection (Reed Solomon and convolutional
coding) provided in DVB-T. Of course, COFDM (Coded Orthogonal Frequency Division Multiplex) was developed precisely for this purpose,
namely to cope with the effects of multipath reception in terrestrial transmission.
21.3.5 Doppler Effect
In mobile reception, a frequency shift occurs over the entire DVB-T spectrum due to the Doppler effect. By itself, the Doppler effect does not present a problem in DVB-T transmission because a shift of a few hundred
Hertz at motor vehicle speeds can be handled easily. It is when Doppler effect and multipath reception are combined that the spectrum becomes
smeared. Echoes moving towards the receiver will shift the spectrum into a
different direction from those moving away from the receiver and, as a result, the signal/noise ratio in the channel deteriorates.
21.3.6 I/Q Errors of the Modulator
The focus of this discussion will now shift to the I/Q errors of the DVB-T
modulator, the effects of which differ from those in DVB-C.
The COFDM symbol is produced by means of the mapper, the real parts
and imaginary parts of all subcarriers being set in the frequency domain
before the IFFT (Inverse Fast Fourier Transform). Each carrier is independently QAM modulated (QPSK, 16QAM, 64QAM) in accordance with
the information to be transmitted. The spectrum has no symmetries or
centro-symmetries and thus is not conjugate with respect to the IFFT band
center.
According to system theory, therefore, a complex time-domain signal
must be produced after the IFFT. Considering then the real time-domain
signal re(t) and the imaginary time-domain signal im(t) carrier by carrier, it
is found that for each carrier, re(t) has exactly the same amplitude as im(t)
and that im(t) is always shifted by exactly 90 degrees in phase with respect
to re(t). All re(t) superimposed in time are fed into the I branch of the
complex I/Q mixer and all im(t) superimposed in time are fed into the Q
branch. The I mixer is fed with 90 degrees carrier phase and the Q mixer is
432
21 Measuring DVB-T Signals
fed with 90 degrees carrier phase and the two modulation products added
together result in the COFDM signal cofdm(t).
I
Re(f)
re(t)
+
IFFT
Im(f)
cofdm(t)
im(t)
Q
90°
Fig. 21.8. COFDM modulator
Fig. 21.9. I/Q imbalance
The signal branches re(t) and im(t) must exhibit exactly the right ratios
of levels with respect to one another. The 900 phase shifter must also be set
correctly. And there must not be any DC component superimposed on the
re(t) and im(t) signals. Otherwise, so-called I/Q errors will occur. The re-
21.3 Constellation Analysis of DVB-T Signals
433
sultant phenomena appearing in the DVB-T signal will be shown in Fig.
21.9.
Fig. 21.9. shows the constellation diagram with an I/Q amplitude imbalance in the I/Q mixer of the modulator. The pattern is rectangularly distorted, i.e. compressed in one direction (horizontal or vertical). This effect
can be observed easily in DVB-C but can only be verified on the center
carrier (band center) in DVB-T where all the other carriers display noiselike interference.
An I/Q phase error leads to a rhomboid-like distortion of the constellation diagram (Fig. 21.10.). This effect can be observed without problems in
the DVB-C cable standard but can only be verified on the center carrier
(band center) in DVB-T where all the other carriers also display noise-like
interference due to this effect.
Fig. 21.10. I/Q phase error
A residual carrier present at the I/Q mixer (Fig. 21.11.) shifts the constellation diagram out of the center in some direction. The pattern itself
remains undistorted. This effect can only be observed on the center carrier
and only affects this carrier.
Today, virtually most of the DVB-T modulators operate in accordance
with the direct modulation method. An analog I/Q modulator used in this
mode usually exhibits problems in the suppression of the carrier, among
others. Although the manufacturers have managed to overcome problems
of I/Q amplitude imbalance and I/Q phase errors in most cases, there is a
remaining problem of carrier suppression to be found more or less with
every DVB-T modulator of this type and has been observed to a greater or
lesser extent at many DVB-T transmitter sites around the globe by the au-
434
21 Measuring DVB-T Signals
thor. The residual-carrier problem can only be verified at the center carrier
(3408 or 852, resp.) in the center of the band and only causes interference
there or in the areas around the center carrier. A lack of carrier suppression
can be detected right away as a dip in the display of the modulation error
ratio over the range of DVB-T subcarriers in the center of the band and an
expert in DVB-T measuring techniques can immediately tell that there is a
DVB-T modulator operating in direct modulation mode.
Fig. 21.11. Effect of residual carrier
DC
I
Re(f)
re(t)
IFFT
Im(f)
I Gain
Q Gain
+
im(t)
Q
DC
90°
Phase
Fig. 21.12. COFDM modulator with I/Q errors
cofdm(t)
21.3 Constellation Analysis of DVB-T Signals
435
21.3.7 Cause and Effect of I/Q Errors in DVB-T
What then is the cause of I/Q errors, why can these effects be observed
only at the center carrier and why do all other carriers exhibit noise-like interference in the presence of any I/Q amplitude imbalance and I/Q phase
error?
Fig. 21.12. shows the places in the I/Q modulator at which these errors
are produced. A DC component in re(t) or im(t) after the IFFT will lead to
a residual carrier in the I or Q branch or in both branches. Apart from a
corresponding amplitude, the residual carrier will, therefore, also exhibit a
phase angle.
Different gains in the I and Q branches will result in an I/Q amplitude
imbalance. If the phase angle at the I/Q mixer differs from 90 degrees, an
I/Q error (quadrature error) is produced.
The disturbances in DVB-T caused by the I/Q errors can be explained
quite clearly without much mathematics by using vector diagrams. Let us
begin with the vector diagram of a normal amplitude modulation. An AM
can be represented as a rotating carrier vector and by superimposed vectors
of the two sidebands, one sideband vector rotating counterclockwise and
one sideband vector rotating clockwise. The resultant vector is always located in the plane of the carrier vector, i.e. the carrier vector is varied
(modulated) in amplitude.
Fig. 21.13. Vector diagram of an amplitude modulation
Suppressing the carrier vector results in amplitude modulation with carrier suppression.
Correspondingly, the behaviour of an I/Q modulator can also be represented by superimposing 2 vector diagrams (Fig. 21.15.). Both mixers operate with suppressed carrier.
436
21 Measuring DVB-T Signals
Fig. 21.14. Vector diagram of an amplitude modulation with suppressed carrier
Q
I
+
I
Q
90°
Fig. 21.15. I/Q modulation
Q
I
+
90°
Q
90°
Fig. 21.16. Single sideband modulation
I
21.3 Constellation Analysis of DVB-T Signals
437
If the same signal is fed into the I branch and into the Q branch, but with
a phase difference of 90 degrees from one another, a vector diagram as
shown in Fig. 21.16. (‘single sideband modulation’) is obtained. It can be
seen clearly that two sideband vectors are added and two sideband vectors
cancel (are subtracted). One sideband is thus suppressed, resulting in single sideband amplitude modulation. A COFDM modulator can thus be interpreted as being a single sideband modulator for many thousands of subcarriers. In an ideal COFDM modulator, there is no crosstalk from the
upper COFDM band to the lower one and vice versa.
Since the IFFT is a purely mathematical process, it can be assumed to be
ideal. The I/Q mixer, however, can be implemented as a digital (ideal)
mixer or as an analog mixer and there are and will be in future analog I/Q
mixers in DVB-T modulators (direct modulation).
If then an I/Q amplitude imbalance exists, this means that the upper or
lower sideband no longer cancel completely, leaving an interference component. The same applies to an I/Q phase error. It is clear, therefore, that
all the subcarriers are subject to noise-like interference, with the exception
of the center carrier. It is also clear why a residual carrier will push the
constellation pattern away from the center at the center carrier and only interferes with the latter.
Fig. 21.17. Spectrum of a DVB-T signal
This can also be shown impressively in the spectrum of the DVB-T signal if the DVB-T modulator has the test function of switching off e.g. the
lower carrier band in the spectrum. This can be done, for example, with a
DVB test transmitter. In the center of the band (center carrier), an existing
residual carrier can be seen clearly. If the I/Q modulator is then adjusted to
produce an amplitude imbalance, crosstalk from the upper to the lower
sideband is clearly apparent. The same applies to an I/Q phase error.
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21 Measuring DVB-T Signals
Fig. 21.18. Lower band switched off
Fig. 21.19. 10% amplitude imbalance
Fig. 21.20. 10° phase error
21.3 Constellation Analysis of DVB-T Signals
439
The process of noise-like crosstalk can be described easily by means of
simple trigonometric operations which can be derived from the vector diagram.
a1
a2 = a1(1-AI)
N = a1-a2;
a2 = a1(1-AI)
a1
Noise N
Signal S
S = a1 + a2;
S/N = (a1+a2)/(a1-a2) = (a1+a1(1-AI)/(a1-a1(1-AI) = (2-AI)/AI;
S/N[dB] = 20lg((2-AI[%]/100)/(AI[%]/100));
Fig. 21.21. Determining the S/N with amplitude imbalance
In the case of amplitude imbalance, the opposing vectors no longer cancel completely (Fig. 21.21.), resulting in a noise vector causing crosstalk
from the upper DVB-T band to the lower band and vice versa. The actual
useful signal amplitude decreases by the same amount by which the
crosstalk increases.
a
x=90- /2
a
x
N
Noise N
N = 2 a cos(90- /2);
Signal S
a
S = 2 a sin(90- /2);
S/N = (2a)/(2a) (sin(90- /2)/cos(90- /2)) = tan(90- /2);
S/N[dB] = 20lg(tan(90- /2));
Fig. 21.22. Determining the S/N in the presence of an I/Q phase error
A phase error will result in a noise vector the length of which can be determined from the vector parallelogram. The useful signal amplitude also
440
21 Measuring DVB-T Signals
decreases by the same amount. Fig. 21.22. shows the conditions for the
signal/noise ratio S/N in the presence of amplitude imbalance and of a
phase error, respectively, which have now been derived as formulae. In the
practical implementation of a DVB-T modulator, the aim is an amplitude
imbalance of less than 0.5% and a phase error of less than 0.5 degrees.
S/N
[dB]
AI
AI[%]
[°]
Fig. 21.23. DVB-T signal/noise ratios with amplitude imbalance (AI) and phase
error (PE) of the IQ modulator
Central DVB-T carrier
8K: Nr. 3408 = continual pilot
2K: Nr. 852 = scattered pilot / payload
Fig. 21.24. Distortions in the vicinity of the center carrier due to the channel correction in the DVB-T receiver
Thus, I/Q errors of the DVB-T modulator can only be identified by observing the center carrier but may interfere with the entire DVB-T signal.
In addition, it will be found that in each case at least the two upper and
lower carriers adjacent to the center carrier are also distorted. This is
caused by the channel correction in the DVB-T receiver where channel estimation and correction is performed on the basis of the evaluation of the
21.3 Constellation Analysis of DVB-T Signals
441
scattered pilots. But these are only available in intervals of 3 carriers and
between them it is necessary to interpolate.
In 2K mode, the center carrier is No. 852 which is a payload carrier or
sometimes a scattered pilot. Verifying I/Q errors will not present any problems, therefore. The situation is different in 8K mode where the center carrier is No. 3408 and is always a continual pilot. I/Q errors can only be extrapolated in this case by observing the adjacent upper and lower carriers.
Each of the effects described has its own measurement parameters. In
the DVB-C cable standard, these parameters have been combined to form
an additional aggregate parameter called the modulation error ratio.
Q
Resultant vector
Error vector
Center of
decision field
Ideal vector
I
Fig. 21.25. Error vector for determining the modulation error ratio (MER)
The modulation error ratio (MER) is a measure of the sum of all interference effects occurring on the transmission link. Like the signal/noise ratio, it is usually specified in dB. If only one noise effect is present, MER
and S/N are equal.
The result of all the interference effects on a digital TV signal in broadband cable networks, explained above, is that the constellation points exhibit deviations with respect to their nominal position in the center of the
decision errors. If the deviations are too great, the decision boundaries are
crossed and bit errors occur. However, the deviations from the center of
the decision field can also be considered to be measurement parameters for
the magnitude of an arbitrary interferer. Which is precisely the aim of an
artificial measurement parameter like the MER. When measuring the
MER, it is assumed that the actual hits in the constellation fields have been
442
21 Measuring DVB-T Signals
pushed away from the center of the respective error by interferers. The
interferers are allocated error vectors, the error vector pointing from the
center of the constellation field to the point of the actual hit in the constellation field. Then the lengths of all these error vectors are measured with
respect to time and the quadratic mean is formed or the maximum peak
value is measured in a time window. The definition of MER can be found
in the DVB Measurement Guidelines [ETR290].
The MER is calculated from the error vector length, using the following
relation:
MERPEAK =
MERRMS =
max(| error _ vector |)
⋅ 100%;
U RMS
1
N
N −1
¦ (| error _ vector |)
n =0
2
⋅ 100%;
U RMS
The reference URMS is here the RMS value of the QAM signal.
Usually, however, the logarithmic scale is used:
§ MER[%] ·
MERdB = 20 ⋅ lg¨
¸;
© 100 ¹
[dB]
The MER value is, therefore, an aggregate quantity which includes all
the possible individual errors. The MER value thus completely describes
the performance of this transmission link.
In principle,
MER [dB] ≤ S/N [dB];
The representation of MER as a function of the subcarrier number
MER(f) is of particular significance in DVB-T because it allows the overall situation in the channel to be observed. It is easy to see areas with disturbed carriers. Often only an averaged single MER measurement value is
mentioned in connection with DVB-T measurements but this value does
not provide much practical information. A graphical representation of the
MER versus frequency is always of importance.
21.3 Constellation Analysis of DVB-T Signals
443
In summary, it can be said that noise and phase jitter affect all carriers to
the same extent, interferers affect carriers or ranges of carriers like noise or
sinusoidally. Echoes also affect only carrier ranges.
Fig. 21.26. Modulation error ratio (MER) vs. COFDM subcarriers MER(f) [EFA]
Table 21.2. DVB-T interference effects
Interference effect
Noise
Phase jitter
Interferer
Echoes
Effect
all carriers
all carriers
single carriers
carrier ranges
Doppler
all carriers
IQ amplitude
imbalance
IQ phase error
Residual carrier
carrier leakage
all carriers
all carriers
center carrier
and adjacent
carriers
Verification
all carriers
all carriers
carriers affected
carriers affected,
impulse response
frequency deviation,
smearing
center carrier
center carrier
center carrier
I/Q errors of the modulator partially affect the carriers as noise-like disturbance and as such can only be identified by observing the center carrier.
All the influences on the DVB-T transmission link described can be observed easily by constellation analysis in a DVB-T test receiver. In addition, a DVB-T test receiver also allows measurement of the received level,
measurement of the bit error rate, calculation of the amplitude and group
444
21 Measuring DVB-T Signals
delay response and of the impulse response from the channel estimation
data. The impulse response is of great importance in detecting multipath
reception in the field, particularly in single-frequency networks (SFNs).
Apart from the I/Q analysis discussed, therefore, a DVB-T test receiver
also enables for a large number of significant measurements to be made on
the DVB-T transmission link.
Fig. 21.27. Crest factor measurement [EFA]
21.4 Measuring the Crest Factor
DVB-T signals have a large crest factor which can be up to 40 dB in theory. In practice, however, the crest factor is limited to about 13 dB in
power transmitters. The crest factor can be measured by using a DVB-T
test receiver. For this purpose, the test receiver picks up the data stream
immediately after the A/D converter and calculates from it both the RMS
value and the maximum peak signal value occurring in a time window.
According to the definition, the crest factor is then
cf = 20 log(Umax peak/URMS);
21.5 Measuring the Amplitude, Phase and Group Delay
Response
Although DVB-T is quite tolerant with respect to linear distortion such as
amplitude, phase and group delay distortion, it is, on the other hand, no
great problem to measure these parameters. A DVB-T test receiver is eas-
21.6 Measuring the Impulse Response
445
ily able to analyse the pilot carrriers (scattered pilots and continual pilots)
contained in the signal and to calculate from these the linear distortions.
The linear distortion is thus determined from the channel estimation data.
Fig. 21.28. Amplitude and group delay response measurement using the scattered
pilots
Fig. 21.29. Impulse response measurement via an IFFT of the channel impulse response CIR
21.6 Measuring the Impulse Response
Transforming the channel estimation data, which are available in the frequency domain and from which the representation of the amplitude and
446
21 Measuring DVB-T Signals
phase response was derived, into the time domain by means of an inverse
fast Fourier transform provides the impulse response. The maximum
length of the calculable impulse response depends on the samples provided
by the channel estimation. Every third subcarrier supplies a contribution to
the channel estimation at some time, i.e. the distance between two interpolation points of the channel estimation is 3 • Δf, where Δf corresponds to
the subcarrier spacing of the COFDM. The calculable impulse response
length is thus 1/3 Δf, i.e. one third of the COFDM symbol period. In the
ideal case, the impulse response only consists of one main impulse at t = 0,
i.e. there is only one signal path. From the impulse response, multiple echoes can be easily classified in accordance with delay and path attenuation.
* RBW 10 kHz
* VBW 100 kHz
* Att
Ref -10 dBm
* SWT 10 s
15 dB
-10
-20
1 RM *
CLRWR
-30
1
Delta 2 [T1 ]
-39.71 dB
3.700000000 MHz
Marker 1 [T1 ]
-34.12
212.500000000
Delta 1 [T1 ]
-41.17
-3.700000000
dBm
MHz
A
dB
MHz
-40
-50
PRN
-60
-70
2
1
-80
-90
-100
-110
Center 212.5 MHz
Date:
19.MAY.2005
1.5 MHz/
Span 15 MHz
13:41:17
Fig. 21.30. Spectrum of a DVB-T signal at the transmitter output before the mask
filter
21.7 Measuring the Shoulder Attenuation
The system does not utilize the full channel bandwidth, i.e. some of the 2K
or 8K subcarriers are set to zero so that no interference to adjacent channels will be caused. Due to nonlinearities, however, there are still outband
components and the effect on the spectrum and its shape has given rise to
the term ‘shoulder attenuation’.
In the Standard, the permissible shoulder attenuation is defined as a tolerance mask. Fig. 21.30. the spectrum of a DVB-T signal at the power amplifier output, i.e. before the mask filter. To determine the shoulder attenuation, different methods are defined and especially a relatively
21.7 Measuring the Shoulder Attenuation
447
elaborate method in the Measurement Guidelines [ETR290]. In practice,
the DVB-T spectrum is in most cases simply measured by using three
markers, setting one marker to band center and the others to +/- (DVB-T
channel bandwidth/2 + 0.2 MHz). With an 8 MHz channel, this results in
test points at +/- 4.2 MHz relative to band center and +/- 3.7 MHz for the 7
MHz channel. Fig. 21.31, shows the spectrum of a DVB-T signal after the
mask filter (critical mask). The DVB-T standard [ETS 300 744] defines
various tolerance masks for various adjacent channel allocations.
* RBW 10 kHz
* VBW 100 kHz
* Att
Ref -10 dBm
* SWT 10 s
15 dB
-10
-20
1 RM *
CLRWR
1
-30
Delta 2 [T1 ]
-55.44 dB
3.700000000 MHz
Marker 1 [T1 ]
-30.46
212.500000000
Delta 1 [T1 ]
-52.45
-3.700000000
dBm
MHz
A
dB
MHz
-40
-50
PRN
-60
-70
1
-80
2
-90
-100
-110
Center 212.5 MHz
Date:
19.MAY.2005
1.5 MHz/
Span 15 MHz
13:42:15
Fig. 21.31. Spectrum of a DVB-T signal measured after the mask filter (critical
mask)
In practice, the following shoulder attenuations are achieved:
•
•
•
Power amplifier, undistorted:
Power amplifier, equalized:
After the output BPF:
approx. 28 dB
approx. 38 dB
approx. 52 dB (critical mask)
Usually the tolerance masks listed in Table 21.3. (uncritical mask) and
21.4. (critical mask) are used for evaluating a DVB-T signal (7 and 8 MHz
bandwidth). In the corresponding documents (DVB-T Standard
[ETS300744], System Specifications), the ratio with respect to channel
power at 4 kHz reference bandwidth is usually specified. If the spectrum
analyzer does not support this resolution bandwidth, it is possible to select
a different one (e.g. 10, 20 or 30 kHz) and the values can be converted.
10lg(4/7610) = -32.8 dB and 10lg(4/6770) = -32.2 dB correspond to the
attenuation with respect to the total signal power of the DVB-T signal with
4 kHz reference bandwidth in the useful DVB-T band. If another resolu-
448
21 Measuring DVB-T Signals
tion bandwidth of the analyzer is used, corresponding values must be inserted into the formula. The tables also show the relative attenuation compared with the useful channel independently of the reference bandwidth.
The important factor in choosing the resolution bandwidth of the spectrum analyzer is that it be not too small and not too large. Usually, 10, 20
or 30 kHz is selected.
Table 21.3. DVB-T tolerance mask (uncritical) in the 7 and 8 MHz channel
frel[MHz]
at 7MHz
channel bandwidth
frel[MHz]
at 8MHz
channel bandwidth
+/-3.4
+/-3.9
+/-3.7
+/-5.25
+/-10.5
+/-13.85
+/-4.2
+/-6.0
+/-12.0
Attenuation
[dB] vs.
channel power
at 4 kHz reference bandwidth
-32.2 (7 MHz)
-32.8 (8 MHz)
-73
-85
-110
-126
Attenuation
[dB] at 7 MHz
channel
bandwidth
Attenuation
[dB] at 8
MHz channel
bandwidth
0
0
-40.8
-52.8
-77.8
-93.8
-40.2
-52.2
-77.2
Table 21.4. DVB-T tolerance mask (critical) in the 7 and 8 MHz channel
frel[MHz]
at 7MHz
channel bandwidth
frel[MHz]
at 8MHz
channel bandwidth
+/-3.4
+/-3.9
+/-3.7
+/-5.25
+/-10.5
+/-13.85
+/-4.2
+/-6.0
+/-12.0
Attenuation
[dB]
vs.
channel power
at 4 kHz reference bandwidth
-32.2 (7 MHz)
-32.8 (8 MHz)
-83
-95
-120
-126
Attenuation
[dB] at 7 MHz
channel bandwidth
Attenuation
[dB] at 8 MHz
channel bandwidth
0
0
-50.8
-62.8
-87.8
-93.8
-50.2
-62.2
-87.2
21.7 Measuring the Shoulder Attenuation
449
Fig. 21.32. DVB-T mask filter (uncritical Mask, low power, manufactured by
Spinner) with directional test coupler at input and output
Fig. 21.33. DVB-T transmission link with MPEG-2 test generator DVRG (center
left), DVB-T test transmitter SFQ (bottom left), DVB-T test receiver EFA (top
left), MPEG-2 test decoder DVMD (center right) and TV monitor, video analyzer
VSA and “601” analyzer VCA (Rohde&Schwarz).
450
21 Measuring DVB-T Signals
Fig. 21.34. TV Analyzer ETL showing a DVB-T 64QAM constellation diagram
display [ETL]
Bibliography: [ETS300744], [ETR290],
[SFQ], [SFU], [FISCHER2], [ETL]
[HOFMEISTER],
[EFA],
22 DVB-H/DVB-SH - Digital Video Broadcasting
for Handhelds
22.1 Introduction
The introduction of 2G GSM (2nd generation Global System for Mobile
Communication) has triggered quite a boom for this wireless type of communication. If the possession of car telephones or similar mobile-type telephones was the prerogative of mostly special circles of people at the beginning of the nineties, at least every second person had his own personal
mobile telephone by the end of the nineties and in most cases it was used
only either for telephoning or for sending and receiving short messages SMS - until then. By then, however, people also wanted to be able to send
and receive data via a mobile telephone, e.g. from a PC. To be able to
check one's e-mail database was initially a pleasant way of keeping oneself
up-to-date, especially in the professional field; today, this is standard usage. In the GSM standard, developed mainly for mobile telephony, however, the data rates are about 9600 bit/s. This is quite adequate for simple
text e-mails without attachments but becomes rather troublesome when
long files are attached to the original message. It can also be used for surfing the Internet but is a cumbersome and expensive way of doing this.
With the introduction of 2.5G mobile telephony, the GPRS (General
Packet Radio System), the data rate was increased to 171.2 kbit/s by forming packets, i.e. combining time slots of the GSM system. It was only with
the 3rd generation, the UMTS (Universal Mobile Telecommunication System), that the data rate could be increased to 144 - 384 kbit/s and 2 Mbit/s,
respectively which, however, greatly depends on the respective conditions
of reception and coverage. Using higher-level modulation (8PSK), the
EDGE (Enhanced Data Rates for GSM Evolution) standard, too, allows
higher data rates of up to 345.6 kbit/s (ECSD) and 473.6 kbit/s (EGPRS),
repectively.
Due to their nature, all mobile radio standards are designed for bidirectional communication between the terminal and the base station. The
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_22, © Springer-Verlag Berlin Heidelberg 2010
452
22 DVB-H/DVB-SH - Digital Video Broadcasting for Handhelds
modulation methods such as, e.g. GMSK (Gaussian Minimum Shift Keying) in GSM or WCDMA (Wideband Code Division Multiple Access) in
UMTS have been designed for these "rough" receiving conditions in mobile reception.
Today, mobile telephones are no longer mere telephones but can be used
as cameras or as games consoles or organizers, evolving more and more to
become multimedia terminals. Equipment manufacturers and network operators are continuously searching for more new applications.
In parallel with the evolution of mobile radio, the transition of analog television to digital television took place. If at the end of the eighties, it still
appeared to be impossible to be able to send moving pictures digitally via
existing transmission paths such as satellite, cable TV or the old-fashioned,
terrestrial way, this is an accepted fact today. It is made possible by
modern compression methods such as MPEG (Moving Picture Experts
Group) and modern modulation methods and matching error protection
(FEC). A key event in this area can be considered to be the first use of the
so-called DCT (Discrete Cosine Transform) in the JPEG (Joint Photographic Experts Group) standard. JPEG is a method for compressing still
pictures used in digital cameras. At the beginning of the nineties, experience gained with the DCT was also applied to the compression of moving
pictures in the MPEG standard. First, the MPEG-1 standard developed for
CD data rates and applications was produced. With MPEG-2 it became
possible to compress SDTV (Standard Definition TV) moving pictures
from originally 270 Mbit/s to less than 5 Mbit/s), the data rate of the associated lip-synchronized audio channel being in most cases 200 - 400 kbit/s.
Even HDTV (High Definition TV) signals could now be reduced to tolerable data rates of about 15 Mbit/s. Knowledge of the anatomy of the human eye made it possible to perform a so-called irrelevance reduction in
combination with redundancy reduction. Signal components, information,
not perceived by the eye and ear are removed from the signal before
transmission.
The compression methods have been refined further in MPEG-4 (H.264,
MPEG-4 Part 10 AVC) and today even lower data rates are possible, with
better picture and sound quality.
During the development of DVB, three different transmission methods
were developed: DVB-S (Satellite), DVB-C (Cable) and DVB-T (Terrestrial). In DVB-T, digital television is broadcast terrestrially in 6, 7 or 8
MHz-wide radio channels in a gapped frequency band of about 47 MHz to
862 MHz at net data rates of either approx. 15 Mbit/s or 22 Mbit/s. In
some countries such as the UK, Sweden or Australia, DVB-T is designed
for pure roof aerial reception and the possible data rate is correspondingly
high at about 22 Mbit/s. Countries like Germany have selected the "Port-
22.2 Convergence between Mobile Radio and Broadcasting
453
able Indoor" option providing the possibility of receiving more than 20
programmes "free to air" via a (passive or active) indoor antenna. Due to
the higher degree of error protection (FEC) required and the more rugged
modulation method (16QAM instead of 64QAM), only lower data rates
such as, e.g. about 15 Mbit/s are possible.
If DVB-T is operated as a network which can be received by portable
receivers, the data rates are about 15 Mbit/s and correspondingly only
about 4 programmes or services can be accomodated in a DVB-T channel.
This is still four times as many as could previously be received in a comparable analog TV channel. The data rates available per programme are,
therefore, about 2.5 to 3.5 Mbit/s, present as variable data rates in a socalled statistical multiplex in most cases.
End user
terminal
Video/audio
services
Delivery
system
Down stream
DVB-(T)H
frontend
Application
DVB-(T)H
mod./Tx
IP/MPEG-2
encapsulator
Interactivity channel
MPEG-2
MUX
Gateway
UMTS/
GSM/
GPRS
Up & down stream
MPE
demux
UMTS/
GSM/
GPRS
(~15 Mbit/s, COFDM, 16QAM,
8K, 4K, 2K carrier, 8/7/6/5 MHz channels,
47…860 MHz, 1.5 GHz)
Fig. 22.1. Convergence between mobile radio and DVB
22.2 Convergence between Mobile Radio and
Broadcasting
Mobile radio networks are networks in which bi-directional (point-topoint) connections are possible at relatively low data rates. Modulation
methods, error protection and hand-over procedures are correspondingly
adapted to the mobile environment. Billing etc. are also found to be system
454
22 DVB-H/DVB-SH - Digital Video Broadcasting for Handhelds
related in the standard. The type of services to be selected, be it a telephone call, an SMS or a data link, is determined by the end user and he is
billed accordingly.
Broadcasting networks are unidirectional networks in which contents
are distributed point-to-multipoint jointly to a large number of parties at
relatively high data rates. On-demand contents are relatively rare, a predetermined content being distributed to many parties from one transmitter site or nowadays also by single-frequency networks ever a number of
transmitters. This content is usually a radio or television program. The data
rates are much higher than in mobile radio networks. Modulation methods
and error protection are often designed only for portable or roof antenna
reception. Mobile reception is provided for only as part of DAB (Digital
Audio Broadcasting) in the Standard. DVB-T has been developed only for
stationary or portable reception.
As part of DVB-H (Digital Video Broadcasting for Hand-held mobile
terminals), attempts are now being made to merge the mobile radio world
with that of broadcasting (Fig. 22.1.) and to combine the advantages of
both network systems, combining the bi-directionality of mobile radio
networks at relatively low data rates with the uni-directionality of broadcast networks at relatively high data rates. If the same services, such as
certain video/audio services on demand are demanded by many subscribers, the data service is mapped from the mobile radio network onto the
point-to-multipoint broadcast rail, depending on the demand and on the
amount of information involved.
The type of information diverted from the mobile radio network to the
broadcasting network depends only on the current requirements. It is currently still undecided what services will be offered to the mobile telephones in future over the DVB-H service. They can be purely IP-based
services or also video/audio over IP. In every case, however, DVB-H will
be a UDP/IP-based service in connection with MPEG/DVB-T/-H. Conceivable applications are current sports programs, news and other services
which could be of interest to the public using mobile telephones. It is certain that, given the right conditions of reception, a DVB-H-enabled mobile
will also be able to receive pure, non-chargeable DVB-T transmissions.
22.3 Essential Parameters of DVB-H
The essential parameters of DVB-H correspond to those of the DVB-T
Standard. The physical layer of DVB-T has been expanded only slightly.
In addition to the 8K and 2K mode, already present in DVB-T, the 4K
22.4 DSM-CC Sections
455
mode was introduced as a good compromise between the two, allowing
single-frequency networks of reasonable size to be formed whilst at the
same time being more suitable for mobile use. The 8K mode is not very
suitable for mobile use because of its small subcarrier spacing and the 2K
mode allows for only short distances of about 20 km between transmitters.
The 8K mode requires more memory space for data interleaving and deinterleaving than do the 4K and 2K modes. Space becoming available in
4K and 2K mode can now be used for deeper interleaving in DVB-H, i.e.
the interleaver can be selected between 'native' and 'in-depth' in 4K and 2K
mode. For signalling additional parameters, TPS (Transmission Parameter
Signalling) bits, reserved or already used otherwise, are used in DVB-H.
The parameters additionally introduced in DVB-H are listed as an appendix in the DVB-T Standard [ETS300744]. All other changes or extensions relate to the MPEG-2 transport stream. These, in turn, can be found
in the DVB Data Broadcast Standard [ETS301192]. The MPEG-2 transport stream, as DVB baseband signal, is the input signal for a DVB-H
modulator. In DVB-H, Multiprotocol Encapsulation (MPE), already defined in the context of DVB Data Broadcasting before DVB-H, is used as a
time-slicing method so that energy can be saved in the mobile part. Both
length and spacing of time slots must be signalled. The IP packets packaged in MPE time slots can be optionally provided with additional FEC
(forward error correction) in DVB-H. This is a Reed Solomon error protection at IP packet level. Everything else corresponds directly to DVB-T or
MPEG-2, respectively. DVB-H is a method for time slot IP packet transmission over an MPEG-2 transport stream. The physical layer used is
DVB-T with some extensions. Its aim is the convergence between a mobile radio network and a DVB-H broadcast network. Data services are
transmitted to the mobile either via the mobile radio network or via the
DVB-H network, depending on traffic volume.
22.4 DSM-CC Sections
In the MPEG-2 Standard ISO/IEC 13818 Part 6, mechanisms for transmitting data, data services and directory structures were created early on.
There are the so-called DSM-CC sections, where DSM-CC stands for
Digital Storage Media Command and Control. In principle, DSM-CC Sections have a comparable structure to the PSI/SI Tables. They begin with a
Table ID which is always within a range from 0x3A to 0x3E. DSM-CC
Sections have a length of up to 4 kbytes and are also divided into transport
stream packets and broadcast multiplexed into the transport stream. Using
456
22 DVB-H/DVB-SH - Digital Video Broadcasting for Handhelds
object carousels (cyclically repeated broadcasting of data), entire directory
trees with different file-scan be transmitted to the DVB receiver via DSMCC sections. This is done, e.g. in MHP (Multimedia Home Platform),
where HTML and Java files are transmitted which can then be executed in
the MHP-enabled DVB receiver.
6 Byte MAC address
LSB
MSB
table_id =0x3E
section_syntax_indicator
private_indicator=1
reserved =11
section_length
MAC_address_6
MAC_address_5
reserved
payload_scrambling_control
address_scrambling_control
LLC_SNAP_FLAG
current_next_indicator
section_number
8
last_section_number
MAC_address_4
MAC_address_3
MAC_address_2
MAC_address_1
IP_data()
CRC
8 Bit
1
1
2
12
8
8
2
2
2
1
1
8
8
8
8
8
32
Fig. 22.2. DSM-CC section for IP transmission (Table_ID=0x3E)
DSM-CC sections with Table ID=0x3E (Fig. 22.2.) can be used for
transmitting Internet (IP) packets in the MPEG-2 transport stream. In an
IP packet, a TCP (Transport Control Protocol) packet or a UDP (User
Datagram Protocol) packet is transmitted. TCP packets carry out a controlled transmission between transmitter and receiver via a handshake procedure. In contrast, UDP packets are sent out without any return message.
Since in most cases, there is no return channel in broadcast operation
(hence the term 'broadcasting'), TCP packets do not make any sense. For
this reason, only UDP protocols are used in DVB during the IP transmission in the so-called Multiprotocol Encapsulation (MPE). Although there
is a return channel in DVB-H via the mobile radio network, an IP packet
cannot be newly requested since the messages must go simultaneously to
many addressees in DVB-H.
22.5 Multiprotocol Encapsulation
457
22.5 Multiprotocol Encapsulation
In DVB Multiprotocol Encapsulation, contents such as, e.g. HTML files or
even MPEG-4 video and audio streams are transported in UDP (User
Datagram Protocol) packets. Windows Media 9 applications can also be
transmitted by this means and can also be reproduced in devices equipped
accordingly. The UDP packets contain the port address of the destination
(DST Port) (Fig. 22.3.), a 16-bit-wide numerical value via which the destination application is addressed. For example, the World Wide Web
(WWW) always communicates via port No. 0x80. Ports are blocked and
controlled by a firewall.
Data stream
DST port
DST IP
SRC IP
H
H
UDP packet
IP packet
CRC checksum
DST MAC
H
DSM-CC section
MPEG-2 TS
Fig. 22.3. Multiprotocol Encapsulation (MPE)
The UDP packets, in turn, are then embedded into the payload part of IP
packets. The header of the IP packets then contains the source and destination (SRC and DST) IP address via which an IP packet is looped through
the network from transmitter to receiver in controlled fashion.
If the IP packets are transmitted via a normal computer network, they
are mostly transported in Ethernet packets. The header of the Ethernet packets again contains the hardware addresses of the network components
communicating with one another, the so-called MAC (Media Access
Command) addresses.
When IP packets are transmitted via DVB networks, the Ethernet layer
is replaced by the MPEG-2 transport stream and the physical DVB layer
458
22 DVB-H/DVB-SH - Digital Video Broadcasting for Handhelds
(DVB-C, -S, -T). The IP packets are first packaged in DSM-CC sections
which are divided into many transport stream packets, in turn. This is called Multiprotocol Encapsulation: UDP divided into IP, IP into DSM-CC,
DSM-CC into TS packets. The header of the DSM-CC sections contains
the destination (DST) MAC address. It has a length of 6 bytes as in the
Ethernet layer. There is no source MAC address.
22.6 DVB-H Standard
DVB-H stands for "Digital Video Broadcasting for Handheld mobile terminals" and is an attempt at convergence between mobile radio networks
and broadcasting networks. The downstream from the mobile radio network (GSM/GPRS, UMTS) is remapped onto the broadcasting network in
dependence on the traffic volume. If, e.g., only a single subscriber requests
a service via UMTS, for example, this downstream continues to pass via
UMTS. If a large number of subscribers request the same service at approximately the same time, it makes sense to offer this service, e.g. a
video, point to multipoint via the broadcast network. The services intended
to be implemented via DVB-H are all IP based.
2K Mode
f~4kHz,
ts~250us
2048 carriers
1705 used
carriers
continual pilots
scattered pilots
TPS carrier
1512 data carrier
in-depth interleaving on/off
4K Mode
f~2kHz,
ts~500us
4096 carriers
3409 used
carriers
continual pilots
scattered pilots
TPS carrier
3024 data carrier
in-depth interleaving on/off
8K Mode
f~1kHz,
ts~1000us
8192 carriers
6817 used
carriers
continual pilots
scattered pilots
TPS carrier
6048 data carrier
Fig. 22.4. Overview of the 2K, 4K and 8K modes in DVB-H
DVB-H is intended to provide the framework for a modified DVB-T
network to broadcast IP services in time slots in an MPEG-2 transport
22.6 DVB-H Standard
459
stream. The physical modulation parameters are very similar or almost
identical to those of a DVB-T network. The MPEG-2 transport stream requires greater modifications.
Length
I
SYNC
Length
Data
res.
FEC
Cell ID
DVB-H
67 TPS bits
over 68 COFDM
symbols
Bit 27...29: hierarchical mode 000, 001, 010
Bit 27: 0 = native Interleaver, 1 = in-depth
interleaver (only in 2K and 4K mode)
Bit 38, 39: 00 = 2K, 01 = 8K, 10 = 4K mode
Bit 40...47: Cell ID
2 new TPS bits:
Bit 48: DVB-H (time slicing) on/off
Bit 49: IP FEC on/off
Fig. 22.5. TPS bits in a DVB-T frame (Transmission Parameter Signalling)
A system overview of DVB-H is provided in ETSI document
[TM2939]. The relevant details are described in the DVB Data Broadcasting Standard [ETS301192] and in the DVB-T Standard [ETS300744].
The physical layer DVB-T has been modified or influenced least. In addition to the 8K mode especially well suited to single-frequency (SFN)
networks and the 2K mode which more suitable for mobile reception, the
4K mode was introduced additionally as optional compromise. Using the
4K mode, twice the transmitter spacing can be achieved compared with the
2K mode and the mobile capability is distinctly improved compare with
the 8K mode. Memory capacities becoming available in the interleaver and
de-interleaver should provide for in-depth interleaving which, in turn,
would make DVB-H more resistant to burst errors, i.e. multibit errors and
the data stream is distributed better over time.
Some additional parameters must also be signalled via TPS carriers in
DVB-H.
460
22 DVB-H/DVB-SH - Digital Video Broadcasting for Handhelds
These are:
•
•
•
•
Time slicing on/off in the MPEG-2 transport stream (=DVB-H)
IP FEC on/off
In-depth interleaving on/off
4K mode
For this purpose, 2 additional bits from the reserved TPS (Transmission
Parameter Signalling) bits, bits 42 and 43, and bits already used are used.
The details can be found in Fig. 22.5.
Using the 4K mode and in-depth interleaving in the 4K and 2K mode allows a better RF performance to be achieved in the mobile channel. At the
same time, the achievable transmitter spacing in 4K mode (approx. 35 km)
is greater by a factor of 2 compared with the 2K mode (approx. 17 km) in
an SFN network.
Apart from the 8, 7 or 6 MHz channel known from DVB-T, a bandwidth
of 5 MHz (L band, USA) can now be selected in DVB-H.
The other modifications are found in the structure of the MPEG-2 transport stream.
Burst n
Delta t
Burst n + 1
MPEG-2 TS
MPE sections (DSM-CC),
MPE-FEC sections
Fig. 22.6. Time slicing in DVB-H
In DVB-H, IP transmission is achieved via the MPEG-2 transport
stream by means of the Multiprotocol Encapsulation (MPE) already described. Compared with conventional MPE, however, there are some special features in DVB-H: the IP packets can be protected with an additional
Reed Solomon FEC (Fig. 22.7.). The Reed Sololomon FEC of an IP datagram is transmitted in its own MPE FEC sections. These sections have the
22.6 DVB-H Standard
461
value 0x78 as Table_ID. The header of these FEC sections has the same
structure as that of the MPE sections. Due to the separate transmission of
the FEC, a receiver is capable to retrieve the IP packet even without FEC
evaluation if there are no errors. Furthermore, the IP information to be
transmitted is combined into time slots in the MPEG-2 transport stream.
In the time slots, the time t until the beginning of the next time slot is
signalled in the DSM-CC header. After receiving a time slot, the mobile
telephone can then "go to sleep" again until shortly before the next time
slot in order to save battery power. On average, the data rates in the time
slots will be up to about 400 kbit/s, depending on application. This is IP information requested simultaneously by many users. To signal the time t
until the next time slot, 4 of the total of 6 bytes provided for the destination
MAC address in the DSM-CC header are used. The end of a time slot is
signalled via the frame boundary and table boundary bit in the MPE and
FEC sections (Fig. 22.8.). The mobile receiver is notified about where an
IP service can be found by means of a new SI table, the IP MAC Notification Table (INT) in the MPEG-2 transport stream. The time slot parameters are also transmitted there (Fig. 22.8.).
n
rows
191
columns
64
columns
IP datagrams
ReedSolomon
DSM-CC sections
(Table_ID=0x3E)
MPE-FEC
DSM-CC
sections
(Table_ID=0x78)
Fig. 22.7. MPE and FEC sections in DVB-H
Instead of the least significant 4 MAC address bytes, the MPE section in
DVB-H contains the time slot parameters, the time t until the beginning
of a new time slot in 10-ms steps, and the two "table_boundary" and "frame_boundary" bits. "table_boundary" marks the last section within a time
slice and "frame_boundary" marks the real end of a time slice, especially
when MPE FEC sections are used.
22 DVB-H/DVB-SH - Digital Video Broadcasting for Handhelds
6 Byte MAC Address
table_id =0x3E
section_syntax_indicator
private_indicator=1
reserved =11
section_length
datagram_section_body()
CRC
LSB
MSB
8 Bit
1
1
2
12
32 Bit
datagram_section_body()
{
MAC_address_6
8 Bit
MAC_address_5
8
reserved
2
payload_scrambling_control 2
address_scrambling_control 2
LLC_SNAP_FLAG
1
current_next_indicator
1
section_number
8
last_section_number
8
MAC_address_4
8
replaced
MAC_address_3
8
@
MAC_address_2
8
MAC_address_1 DVB-H
8 Bit
IP_data()
}
real_time_parameters()
{
delta_t
table_boundary
frame_boundary
address
}
Real
time
parameters
462
12 Bit
1
1
18
Fig. 22.8. Structure of an MPE section with time slot parameters according to
DVB-H
22.7 Summary
DVB-H represents a convergence between GSM/UMTS and DVB. The
GSM/UMTS mobile radio network is used as the interactive channel via
which high-rate services such as, e.g. video streaming (H.264/MPEG-4
22.7 Summary
463
Part 10 AVC Advanced Video Coding or Windows Media 9) are requested
which are then transmitted either via the mobile radio network (UMTS) or
are remapped onto the DVB-H network. In DVB-H, a DVB-T network is
virtually used physically, with some modifications of the DVB-T Standard.
As part of DVB-H, additional operating modes were introduced:
•
•
•
•
•
•
The 4K mode as a good compromise between the 2K and 8K
mode with 3409 carriers now used.
In-depth interleaving is possible in the 4K and 2K modes
2 new TPS bits for additional signalling and additional signalling
via TPS bits already used,
Time slicing to save power
IP packets with FEC protection
Introduction of a 5 MHz channel (US L band)
table_id = 0x78 = MPE-FEC
table_id =0x78
section_syntax_indicator
private_indicator=1
reserved =11
section_length
MPE_FEC_section_body()
CRC
MPE_FEC_section_body()
{
padding_columns
reserved_for_future_use
reserved
reserved_for_future_use
current_next_indicator
section_number
last_section_number
real_time parameters()
RS_data()
}
8 Bit
1
1
2
12
32 Bit
8 Bit
8
2
5
1
8
8
42 Bit
Fig. 22.9. Structure of a DVB-H MPE FEC section with time slot parameters
464
22 DVB-H/DVB-SH - Digital Video Broadcasting for Handhelds
In the MPEG/2 transport stream, Multiprotocol Encapsulation is applied
in a time slicing method. The IP packets to be transmitted can be protected
by an additional Reed Solomon FEC code. The end user terminal is notified via a new DVB-SI table about where it can find the IP service.
At the end of 2003, a first prototype of a DVB-H-enabled terminal was
presented which has a DVB-H receiver integrated in a modified battery
pack.
Fig. 22.10. Representation of a DVB-H Section on an MPEG-2 Analyzer [DVM]
22.8 DVB-SH
DVB-SH stands for Mobile TV via satellite and terrestrial paths and is ultimately a virtual combination of DVB-H and DVB-S2 with altered technical parameters. Terrestrial broadcasting is intended for population centers and satellite coverage in the S band (21270 to 2200 GHz) is for rural
areas. This MSS (mobile satellite services) band is next to the UMTS
band. Terrestrial broadcasting uses the COFDM multicarrier technology,
known from DVB-T/H, in the UHF band and the satellite link uses a sin-
22.8 DVB-SH
465
gle-carrier modulation method in the near GHz range where mobile GPS is
also working very well. Satellite reception is more difficult in the population centers because of the buildings but terrestrial coverage from the TV
towers is cost-effective; the opposite holds true in rural areas. The DVBSH proposal originated with Alcatel is now an ETSI standard [EN302583]
which was published in August 2007. In the technical parameters, changes
were made in the error protection; the time interleaver was extended to
about 300 ms and the convolutional coding was replaced by turbo coding.
On the direct and indirect path from the satellite or via repeaters or TV
towers (Fig. 22.11.), the technical parameters of DVB-SH are:
•
•
•
•
•
•
•
•
Derived from DVB-T/H
COFDM mode
QPSK
16QAM
FEC = 3GPP2 turbo encoder (modified and extended)
1k, 2k, 4k, 8k mode
Hierarchical modulation via 16QAM, α=1, 2, 4
Bandwidths of 8, 7, 6, 5, 1.7 MHz.
On the direct path from the satellite to the terminal alone, the technical
parameters can also be (Fig. 22.11.):
•
•
•
•
•
•
•
Derived from DVB-S2
Single carrier TDM mode
QPSK
8PSK
16APSK
FEC = 3GPP2 turbo encoder (modified and extended)
Bandwidths of 8, 7, 6, 5, 1.7 MHz.
As in DVB-H, the baseband signal used is the MPEG-2 transport stream
with time slicing but generic streams are not excluded (are considered as
an option). In DVB-SH, two operating modes are provided:
•
•
Type 1: SH-A-SFN: The same frequency is used both for satellite
and terrestrial paths.
Type 2: SH-B or SH-A-MFN: Different frequencies are used for
satellite and terrestrial paths.
466
22 DVB-H/DVB-SH - Digital Video Broadcasting for Handhelds
With regard to the time interleaver depth, two receiver types have been
defined in DVB-SH:
•
•
Class 1 receiver: 4 Mbit interleaver memory only
Class 2 receiver: 512 Mbit interleaver memory.
COFDM COFDM/
TDM
Playout
Gap-Filler
COFDM
Fig. 22.11. DVB-SH transmission scenario
Bibliography: [ETS300744], [TM2939], [ETSA301192],
[ISO/IEC13818-6], [R&S_APPL_1MA91], [EN302583]
23 Digital Terrestrial TV to North American ATSC
Standard
Although terrestrial radio transmission poses a variety of problems due to
multipath reception and is best handled using multicarrier methods (Coded
Orthogonal Frequency Division Multiplex - COFDM), North America
opted in favour of a single carrier method under the Advanced Television
Systems Committee (ATSC). In the years 1993 to 1995, the Advanced
Television Systems Committee – with the participation of AT&T, Zenith,
General Instruments, MIT, Philips, Thomson and Sarnoff – developed a
method for the terrestrial, and also cable, transmission of digital TV signals. The cable transmission method proposed by ATSC was not put into
practice, and the J.83/B Standard was introduced instead. As in all other
digital TV transmission methods, the baseband signal is in the form of an
MPEG-2 transport stream. The video signal is MPEG-2 coded (MPEG:
Moving Picture Experts Group); the audio signal is Dolby digital AC-3
coded. In contrast to DVB, high definition television (HDTV) was favoured in ATSC. The input signal to an ATSC modulator, therefore, is a
transport stream with MPEG-2 coded video and Dolby AC-3 coded audio
information (AC-3: digital audio compression). Video signals are either
SDTV (Standard Definition Television) or HDTV signals. The modulation
mode used is eight-level trellis-coded vestigial sideband (8VSB). This is a
single-carrier method based on IQ modulation using only the I axis. Eight
equidistant constellation points are distributed along the I axis. The 8VSB
baseband signal has eight discrete amplitude modulation levels. First, however, an 8ASK signal is generated (ASK: amplitude shift keying).
The ASK signal is a staircase signal (Fig. 23.2.). The bit information to
be transmitted is contained in the step height. One step width corresponds
to one symbol or symbol duration; three bits can be transmitted per symbol. The reciprocal of the step width is the symbol rate. The ASK staircase
signal is amplitude-modulated on a sinusoidal carrier. As a result, a double-sideband spectrum is obtained.
To reduce bandwidth, one sideband is partially suppressed in 8VSB
modulation, same as with analog TV. In other words, the amplitudemodulated signal is subjected to vestigial sideband filtering, hence the des-
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_23, © Springer-Verlag Berlin Heidelberg 2010
468
23 Digital Terrestrial TV to North American ATSC Standard
ignation 8VSB. The upper sideband and a vestigial lower sideband remain.
Vestigial sideband filtering at the transmitter end calls for Nyquist filtering
at the receiver end. The 8VSB signal is subjected to soft Nyquist filtering
at the original band center at the receiver end.
Q
I
I
Fig. 23.1. Constellation diagram of an 8ASK signal
Q
8VSB time domain signal
t
Symbol rate = 1/ts
ts
Symbol duration ts
Fig. 23.2. 8VSB/8ASK baseband signal
The area below the Nyquist slope left of the previous band center (Fig.
23.5.) corresponds exactly to the area above the Nyquist edge right of the
previous band center, and so compensates for the missing part to complete
the upper sideband. As a result, a flat amplitude frequency response is ob-
23 Digital Terrestrial TV to North American ATSC Standard
469
tained. If the Nyquist slope is not properly adjusted, amplitude frequency
response at low frequencies will be the consequence.
Lower
sideband
Upper
sideband
f
Carrier
frequency
Carrier
Fig. 23.3. 8ASK modulation at RF domain
Vestigial
sideband
(VSB)
upper
sideband
VSB
filter
f
Carrier
frequency
Carrier
Receiver filter
with Nyquist
slope
VSB
Fig. 23.4. Vestigial sideband filtering
Upper
sideband
f
Carrier
frequency
Fig. 23.5. IF filtering with Nyquist slope
470
23 Digital Terrestrial TV to North American ATSC Standard
With a double-sided spectrum, the vectors representing the upper and
the lower sideband (each starting from the tip of the carrier vector) rotate
in opposite directions, thus varying the length of the carrier vector, i.e.
modulating the carrier. The carrier vector itself remains on the I axis. Even
if the carrier is suppressed, the sum vector yielded by the upper and the
lower sideband still remains on the I axis (Fig. 23.6.).
Q
Upper sideband
Carrier
vector
AM with
suppressed
carrier
Q
Upper sideband
I
I
Lower sideband
Lower sideband
Resulting vector always on I axis
Resulting vector always on I axis
Fig. 23.6. Vector diagram showing amplitude modulation with and without carrier
Q
Q
I
I
Upper sideband
I and Q compents caused
by vestigial sideband filtering
8VSB constellation diagram
after vestigial sideband filter
Fig. 23.7. Vector diagram and constellation diagram of an 8VSB signal
However, if one sideband is suppressed in part or completely, the resulting vector will swing about the I axis. Vestigial sideband filtering produces
a Q component. Such a Q component is also contained in analog vestigial
sideband filtered TV signals (Fig. 23.7.). Analog TV test receivers usually
23 Digital Terrestrial TV to North American ATSC Standard
471
have a Q output in addition to the video output (I output). The Q output is
used for measuring incidental carrier phase modulation (ICPM). Due to
vestigial sideband filtering, the constellation diagram of an 8VSB signal
also includes a Q component, and modulation is no longer shown by
points, but by vertical lines. The 8VSB constellation diagram output by an
ATSC test receiver, therefore, exhibits vertical lines (Fig. 23.7., 23.8.).
Fig. 23.8. Constellation diagram produced by an ATSC test receiver [EFA]
Q
I
u(t)
+
ssb(t)
90
Q
90
lo(t)
I
Hilbert transformer
Fig. 23.9. Vestigial sideband or single sideband modulation by means of a Hilbert
transformer
472
23 Digital Terrestrial TV to North American ATSC Standard
8VSB is no longer effected by means of a simple analog vestigial sideband filter as it used to be in analog TV. Today a Hilbert transformer and
an IQ modulator are used (Fig. 23.9.). The 8VSB baseband signal is split
into two paths. One path is directly applied to the I mixer, the other one is
taken via a Hilbert transformer to the Q mixer. A Hilbert transformer is a
90° phase shifter for all frequencies of the band to be filtered. Together
with the IQ modulator, it acts as a single sideband modulator; part of the
frequencies of the lower sideband are suppressed. Vestigial sideband filtering of modern analog TV transmitters today follows the same principle. A
vital prerequisite for the quality of vestigial sideband filtering is the correct
setting and operation of the IQ modulator. This means identical gain in the
I and Q paths; moreover, the carrier supplied to the Q path must have a
phase of exactly 90°. Otherwise the unwanted part of the lower sideband
will not be fully suppressed, so that a residual carrier is obtained at the
band center.
MPEG-2
TS
VSB
mod.
FEC
IF
RF
MUX
Sync
gen.
Segment and
field sync
Fig. 23.10. 8VSB modulator and transmitter
23.1 The 8VSB Modulator
After discussing the principle of ATSC modulation, let us take a closer
look at the 8VSB modulator (Fig. 23.10.). The ATSC-conformant
MPEG-2 transport stream, including PSIP tables, MPEG-2 video elementary streams and Dolby digital AC-3 audio elementary streams, is fed to
the forward error correction (FEC) block of the 8VSB modulator at a data
rate of exactly 19.3926585 Mbit/s. In the baseband interface, the input
23.1 The 8VSB Modulator
473
transport stream synchronizes to the MPEG-2 188-byte packet structure by
means of a sync byte.
The 188 bytes include the transport stream packet header with the sync
byte, which has a constant value of 0x47. The transport stream packet
clock and the byte clock, which are derived in the baseband interface, are
used in the FEC block and also taken to the sync generator for the 8VSB
modulator. From the transport stream packet clock and the byte clock, the
sync generator generates the data segment sync and the data field sync.
RS(208,188)
MPEG-2
TS
ReedSolomon
encoder
Data
randomizer
Baseband
interface
Data
interleaver
Trellisencoder
Clock
Synchronization
Fig. 23.11. ATSC 8VSB FEC
Initialization word:
+
+
D0
D1
+
+
+
D2
D3
D4 D5 D6 D7
+
+
X16
+
X15
+
1 1 1
X14
+
1
X13
0
X12
0
X11
X10
+
X9
+
1 1 0
X8
0
X7
X6
X5
+
X4
+
0 0 0
X3
0 0
X2
X1
0
+
Field and segment sync not scrambled;
initialization during field sync interval
Fig. 23.12. Shift register for randomization
In the FEC block (Fig. 23.11.), the data is fed to a randomizer (Fig.
23.12.) to break up any long sequences of 1s or 0s that may be contained in
the transport stream. The data randomizer XORs the incoming data bytes
474
23 Digital Terrestrial TV to North American ATSC Standard
with a pseudo random binary sequence (PRBS). The PRBS generator consists of a 16-bit feedback shift register; it is reset to a defined initialization
word at a defined time during the field sync interval. The sync information
(e.g. data field sync, data segment sync), which will be discussed in greater
detail below, is not randomized and is used, among other things, for receiver-to-modulator coupling. At the receiver end, there is a complementary PRBS generator and randomizer, i.e. of exactly the same design and
running exactly in synchronism with the generator/randomizer at the
transmitter end.
MPEG-2
RS
TS
ATSCmod.
Transmission
link
ATSC
demod.
RS
MPEG-2
TS
Reed-Solomon coder RS(208,188) = outer coder;
first forward error correction (1st FEC)
208 byte
4 byte
header
184 byte
payload
20 byte RS
error protection
188 byte
Fig. 23.13. Reed-Solomon FEC
The randomizer at the receiver end reverses the procedure that takes
place at the transmitter end, i.e. it restores the original data stream. Randomizing is necessary since long sequences of 1s or 0s may occur. During
such sequences, there is no change in the 8VSB symbols and therefore no
clock information. This would cause synchronization problems in the receiver and, during the transmission of long sequences of 1s or 0s, produce
discrete spectral lines in the transmission channel. This effect is cancelled
by randomizing, which causes energy dispersal, i.e. it creates an evenly
distributed power density spectrum. The randomizer is followed by the
Reed-Solomon block encoder. An ATSC RS encoder (Fig. 23.12.) adds 20
error protection bytes to the 188-byte transport stream (TS) packet (compared with 16 bytes in DVB), yielding a total packet size of 208 bytes. The
20 error protection bytes allow up to 10 errored bytes per TS packet (including FEC part) to be corrected at the receiver end. If more than 10 er-
23.1 The 8VSB Modulator
475
rored bytes are contained in a TS packet, Reed-Solomon error correction
fails, and the transport stream packet concerned is marked as errored.
To mark a TS packet as errored, the transport error indicator bit (Fig.
23.14.) in the TS packet header is set to 1. The packet in question will then
be discarded by the MPEG-2 decoder following the 8VSB demodulator in
the receiver, and the error will be concealed.
Sync byte 0x47
1 Bit Transport error indicator
184 byte
payload
4 byte
header
188 byte
Fig. 23.14. Transport error indicator in TS header
Precoder
Mapper
+
Z2
D
Z1
....
....
D
+
D
Data in
Z0
Trellis coder
Coderate = datarate in / datarate out = 2/3
Fig. 23.15. Trellis encoder
Z2 Z1 Z0 R
000 –7
001 –5
010 –3
011 –1
100 +1
101 +3
110 +5
111 +7
R
Data
out
476
23 Digital Terrestrial TV to North American ATSC Standard
The Reed-Solomon encoder RS(188,208) is followed by a data interleaver, which changes the time sequence of the data, i.e. it scrambles the
data. At the receiver end, the de-interleaver restores the original time sequence of the data. With interleaving, even long burst errors can be corrected as they are distributed over several frames and can thus be handled
more easily by the Reed-Solomon decoder. Interleaving is followed by a
second error correction in the form of a trellis encoder. The trellis encoder
can be compared to the convolutional encoder used in DVB-S and DVB-T.
The ATSC system employs a trellis encoder (Fig. 23.15.) with two signal paths. From the incoming bit stream, one bit is taken to a precoder with
a code rate of 1, and the second bit to a trellis encoder with a code rate of
1/2. This yields an overall code rate of 2/3. The three data streams generated by the precoder and the trellis encoder are fed to a symbol mapper,
which outputs the 8-level VSB baseband signal. The counterpart of the
trellis encoder at the receiver end is the Viterbi decoder.
Symbol rate = 10.76 MS/s
Data
segment
sync
Data + FEC
828 symbols, 207 byte
Data
segment
sync
+7
+5
+3
+1
-1
-3
-5
-7
Level
before
pilot
addition
4 symbols
(372 ns)
4 symbols
Data segment
832 symbols
208 byte (77.3 µs)
Fig. 23.16. 8VSB data segment
The Viterbi decoder corrects bit errors by retracing the path through the
trellis diagram that has with the highest probability been followed through
the encoder (see also the chapter on DVB-S). Parallel to the FEC block, a
sync generator is provided in the 8VSB modulator. This generator produces, at defined intervals, special sync patterns that are transmitted in-
23.1 The 8VSB Modulator
477
stead of data in the 8VSB signal as sync information for the receiver. The
FEC encoded data and the segment sync and field sync produced by the
sync generator are combined in the multiplexer. The 8VSB signal is divided into data segments. Each data segment starts with a data segment
sync.
The data segment sync consists of 4 symbols which are assigned defined
8VSB signal levels: The first symbol is at signal level +5, the two middle
symbols at signal level –5, and the last symbol at +5. The data segment
sync can be compared to the analog TV sync pulse. It marks the beginning
of a data segment consisting of 828 symbols and carrying a total of 207 data bytes. A complete data segment – including the sync – comprises 832
symbols and has a length of 77.3µs (s. It is followed by the next data segment, which likewise starts with the 4-symbol data segment sync. A total
of 313 data segments each combine to form a field. In 8VSB transmission,
a distinction is made between field 1 and field 2. With either field comprising 313 data segments, field 1 and field 2 contain a total of 626 data segments. Each field starts with a field sync. This is a special data segment
that likewise starts with a 4 symbol data segment sync but contains special
data. Each field of 313 data segments is 24.2 ms long, yielding an overall
length of 48.4 ms for field 1 and field 2.
Segment sync
Field sync 1
Data + FEC
313 segments
24.2 ms
Field sync 2
Data + FEC
313 segments
24.2 ms
1 segment
832 symbols
77.3 µs
Fig. 23.17. 8VSB data frame with two fields
478
23 Digital Terrestrial TV to North American ATSC Standard
The field sync, like a data segment, starts with a data segment sync. Instead of normal data, however, this data segment sync contains a number
of pseudo random sequences, the VSB mode information, and some special, reserved symbols. The VSB mode bits carry the 8VSB/16VSB mode
information. 16VSB was intended for cable transmission, but has not been
implemented in practice.
Data
segment
sync
Level
before
pilot
addition
4 symbols
(372 ns)
Precode (12 symbols)
Reserved (92 sym.)
VSB mode (24 sym.)
PN63 (63 symbols)
PN63 (63 symbols)
PN63 (63 symbols)
PN511 (511 symbols)
+7
+5
+3
+1
-1
-3
-5
-7
Data
segment
sync
4 symbols
Data segment
832 symbols
208 byte (77.3 µs)
Fig. 23.18. 8VSB field sync
Terrestrial transmission employs the 8VSB mode. The pseudo random
sequences contained in the field sync are used as training sequences by the
channel equalizer in the receiver. Moreover, it is the pseudo random sequences by which the receiver detects the field sync and is thus able to
synchronize to the frame structure. During the field sync, the randomizer
block is reset in the modulator and in the receiver. The resulting 8VSB
baseband signal, consisting of field syncs and data segments, is taken to
the 8VSB modulator. Prior to amplitude modulation, a relative DC component of +1.25 is added to the 8 level signal. Prior to this addition, the
8VSB signal has discrete amplitude stages of -7, -5, -3, -1, +1, +3, +5 and
+7. Adding the DC component shifts all 8VSB levels by a relative value of
+1.25.
Amplitude modulation of a baseband signal, no longer free of DC but
with a mixer signal actually free of carrier, however, produces a signal
with a carrier component. This carrier component is referred to as an
23.1 The 8VSB Modulator
479
8VSB pilot signal, and is found exactly at the center of the 8VSB modulation product before it is subjected to vestigial sideband filtering. As a double-sided spectrum, the modulation product would occupy bandwidth at
least as wide as the symbol rate. The symbol rate is 10.76 MS/s, so the
minimum required bandwidth is 10.76 MHz. The channel bandwidth in the
North American ATSC TV system is, however, only 6 MHz. As in analog
TV, therefore, the 8VSB signal is vestigial sideband filtered after amplitude modulation, i.e. the major part of the lower sideband is suppressed.
This could be done by means of a conventional analog vestigial sideband
filter; this method is today no longer employed however, not even by modern analog TV transmitters. Instead, the 8VSB baseband signal with its pilot DC component is split into two signals: One is taken directly to an I
mixer and the other first to a Hilbert transformer and then to a Q mixer
(Fig. 23.20.).
Pilot addition
DC +1.25
VSB
filter
+
+7
+5
+3
+1
-1
-3
-5
-7
Carrier
Fig. 23.19. 8VSB modulation with pilot
A Hilbert transformer is a 90° phase shifter for all frequencies of a band.
The Hilbert transformer in conjunction with the IQ modulator causes partial suppression of the lower sideband, which is obtained due to the configuration of the amplitudes and phases involved. The resulting 8VSB
spectrum only contains the upper sideband and a vestigial lower sideband.
Moreover, a spectral line is found at the previous band center, i.e. the band
center before vestigial sideband filtering. This spectral line results from the
added DC component and is referred to as pilot carrier. The 8VSB spectrum is Nyquist filtered with a roll-off factor of r = 0.115. After VSB
modulation, the signal is converted to RF. This conversion is today usually
effected by direct modulation simultaneously with VSB modulation. An
480
23 Digital Terrestrial TV to North American ATSC Standard
analog IQ modulator is therefore normally used in VSB modulation which
directly converts the baseband signal to RF. As an analog component, the
IQ modulator no longer operates as perfectly as does a digital device. It
must therefore be ensured that the gain in the I and Q paths is identical,
and that the phase of the carrier supplied to the Q path is exactly 90 °. Otherwise the unwanted part of the lower sideband will be inadequately suppressed. After RF conversion, the signal passes through the preequalization and power amplifier stages and is then taken to the antenna. A
passive bandpass filter in the antenna feeder line suppresses out-of-band
components.
Pilot addition
DC +1.25
+
+
Hilbert
transformer
90°
Carrier
Fig. 23.20. Typical 8VSB modulator with Hilbert transformer
23.2 8VSB Gross Data Rate and Net Data Rate
The symbol rate employed in 8VSB is calculated as follows:
symbol_rate = 4.5/286 • 684 MS/s = 10.76223776 MS/s;
This yields the following gross data rate:
gross_data_rate = 3 bit/symbol • 10.76 MS/s = 32.2867 Mbit/s;
The net data rate is then:
23.2 8VSB Gross Data Rate and Net Data Rate
net_data_rate = 188/208 • 2/3 • 312/313 • gross_data_rate
= 19.39265846 Mbit/s;
The above equations are based on the following parameter values:
•
•
•
•
8VSB = 3 bit/symbol
Reed-Solomon = 188/208
Code rate = 2/3 (trellis)
Field sync = 312/313
Fig. 23.21. 8VSB spectrum (roll-off filtered with r=0.115)
Tuner
SAW
ADC
Equalizer
8VSB
demod.
FEC
Video
MPEG-2
decod.
Audio
Fig. 23.22. ATSC receiver
481
482
23 Digital Terrestrial TV to North American ATSC Standard
23.3 The ATSC Receiver
In the ATSC receiver, a tuner converts the signal from RF to IF. Then the
adjacent channels are suppressed by a SAW filter with a Nyquist slope.
The band-limited ATSC signal is converted to a second, lower IF for simplified A/D conversion after the anti-aliasing lowpass filter. A/D conversion is followed by a digital channel equalizer that corrects transmission
errors. The channel equalizer block also includes a matched filter which
performs roll-off filtering with a roll-off factor of r = 0.115. The 8VSB
signal is then demodulated, and errored bytes are corrected in the FEC
block. This again yields the original transport stream, which is applied to
the MPEG-2 decoder to restore the original video and audio signals.
23.4 Causes of Interference on the ATSC Transmission
Path
ATSC transmission paths are subject to the same types of interference as
DVB-T transmission paths. Terrestrial transmission channels are
characterized by interference as follows:
•
•
•
•
•
Noise
Interferers
Multipath reception (echoes)
Amplitude response, group delay
Doppler effect in mobile reception
ATSC/8VSB)
(not
considered
in
Of the above types of interference, noise is the only one that can be well
predicted and relatively easily handled in ATSC transmission. All other effects, especially multipath reception, are difficult to manage. This is due to
the principle of single carrier transmission employed by ATSC. While the
equalizer in 8VSB/ATSC is capable of correcting echo, 8VSB is more susceptible to interference compared with COFDM. Mobile reception is virtually impossible.
The "brickwall effect" occurs at an S/N of about 14.9 dB in ATSC. This
corresponds to about 2.5 segment errors per second or to a segment error
rate of 1.93 • 10-4, respectively. The pre-Reed Solomon bit error rate is
then 2 • 10-3 and the post-Reed Solomon bit error rate is 2 • 10-6.
23.5 ATSC-M/H Mobile DTV
483
Assuming that the noise power at the tuner input is about 10 dB V (see
chapter on DVB-T), the minimum required receiver input voltage is about
25 dB V in ATSC.
23.5 ATSC-M/H Mobile DTV
Two proposals by the companies Samsung/Rohde&Schwarz ("A-VSB" Advanced VSB) and Harris/LG ("ATSC-MHP") resulted in the creation of
a so-called Candidate Standard ATSC M/H as an extension to the ATSC
standard for mobile TV in 2008. The ATSC M/H standard is called
[A/153] and consists of a number of documents. Part 2 describes the
transmission part (Transmission System Characteristics). The intention is
to make ATSC more portable and receivable by mobile by employing new
technologies. The extensions are backward compatible and, therefore, do
not interfere with existing ATSC receivers. The services inserted for the
use of mobiles are virtually invisible to normal ATSC receivers. In the
MPEG-2 data stream supplied to the ATSC M/H modulator, the mobile
services run on a PID which is not signalled via the PSI tables, a so-called
unreferenced PID. However, this PID is known to an ATSC M/H modulator which inserts the contents into the ATSC frame in a special way. The
"normal" ATSC services and the mobile services then share the constant
total data rate of 19.39 Mbit/s. The MPEG-2 transport stream is correspondingly edited by an ATSC M/H multiplexer which inserts the mobile
DTV contents into the data stream. The mobile DTV services are MPEG-4
AVC and MPEG-4 AAC coded video and audio at total data rates of
approx. 0.5 Mbit/s net per service at display resolutions corresponding to
mobile telephones (416 pixels x 40 lines (16:9)). For reasons of compatibility, the contents are embedded in UDP and IP protocols similar to
DVB-H. In addition, the ATSC M/H compatible modulator is supplied
with signalling and control signals. The possibility of forming SFNs (single-frequency networks) does not form a part of the proposed ATSC but is
also provided for in parallel. There is a relevant standard [A/110B] which
is surrently not used for reasons of expenditure and licensing, and also
proprietary solutions for SFN synchronization (e.g. Rohde&Schwarz).
23.5.1 Compatibility with the Existing Frame Structure
The key in ATSC M/H is the backward compatibility with normal ATSC.
Normal ATSC receivers must not sense any disturbance from the additional mobile contents. To achieve this, the format must be completely
484
23 Digital Terrestrial TV to North American ATSC Standard
matched to the existing ATSC segment, field and frame structure. An
ATSC segment contains a time interleaved and doubly error protected
(Reed Solomon and trellis) MPEG-2 transport stream packet. The original
MPEG-2 sync bytes which have not been displaced in time have been replaced by the segment sync (4 symbols). The 187 bytes - without sync byte
- of an MPEG-2 transport stream packet now become 2484 bytes of an
ATSC segment by means of RS(188, 208) and 2/3 trellis coding. Due to
the time interleaving, however, the error protected data can no longer be
found transparently in a segment but are distributed over 52 segments or
transport stream packets, respectively.
ATSC has allocated 313 segments to one field and 2 fields to one frame.
If it is intended to incorporate new mobile contents compatibly into ATSC,
this frame structure must be adhered to. The following considerations are
of importance for the adaptation of ATSC M/H:
•
•
•
•
52 transport stream packets or 52 adjacent segments contain interleaved coherent data
3 x 52 transport stream packets result in a number of 156 packets
or 156 segments, respectively
2 x 156 = 312 segments result in one ATSC field
Together with the field sync, the ATSC field has a total length of
313 segments
If single frequency networks are to be formed, the basic prerequisite is
that all modulators operate completely synchronously with respect to frequency, time and data. I.e. it is necessary to have synchronization of the
frame structure, starting with the ATSC M/H multiplexer, the multiplexer
informing the ATSC modulator of the ARSC frame start and the emission
time of the ATSC symbols.
In ATSC M/H - "Mobile DTV" - the content for the mobile service,
•
•
•
•
•
•
MPEG-4 AVC - H.264 coded (video), image resolution 426 pixels
x 240 lines (16:9)
and MPEG-4 AAC coded (audio)
embedded in an IP environment
provided with additional Reed Solomon and convolutional code
with additional TPC (transmission parameter channel) information
(about the physical layer, error protection)
extended with additional FIC (fast information channel) information (number of services)
23.5 ATSC-M/H Mobile DTV
•
•
•
485
rendered transparently operable with additional SSC (service signalling channel)
better received with additional training sequences for the mobile
receiver
optionally equipped with ESG (Electronic Service Guide) via
OMA BCAST (Open Mobile Alliance Broadcast)
thus prepared, is keyed into the MPEG-2 transport stream supplied to
the ATSC modulator, in such a way that the interleaving taking place there
and the error protection are virtually "outwitted", i.e. are precalculated.
The additional contents are running on a special PID (0x1FF9 or userdefined) known to the ATSC M/H multiplexer and modulator. This also
includes any type of signalling. The data for ATSC M/H are preinterleaved (52 transport stream packets) in advance in such a manner that
the correct order is restored by the interleaving in the modulator (52 segments). The main tasks such as additional error protection, inserting of additional training sequences etc. have already been accomplished in the
ATSC M/H multiplexer.
MPEG-4 AVC and AAC streaming with RTP
Data stream
DST port
DST IP
SRC IP
H
H
UDP packet
IP packet
Fig. 23.23. MPEG-4 AVC and AAC streaming via IP
23.5.2 MPEG-4 Video and Audio Streaming
In ATSC M/H, as in other mobile TV standards, the content transmitted
for the mobile applications is embedded via IP protocols. MPEG-4 AVC
and MPEG-4 AAC streaming material is initially inserted into the
MPEG-2 transport stream in UDP (User Datagram Protocol) packets by
way of DSM-CC sections. This variant of data transmission provides for
the compatibility with other transmission variants (s.a. the DVB-H chap-
486
23 Digital Terrestrial TV to North American ATSC Standard
ter). MPEG-4 AVC (= H.264) is currently the most modern and most effective type of video compression. The same applies to MPEG-4 AAC+
for the audio coding.
SFN
signalling
MPEG-2 TS
MPEG-4
AVC IP
AAC
MH
MUX
TS
TPC
FIC
additional
training
sequences
ATSC-M/H
modulator
RF
TPC = Transmission
Parameter Channel
FIC = Fast Information
Channel
Fig. 23.24. ATSC-M/H multiplexer and ATSC-M/H modulator
Field sync 1
Segment sync
M/H slot
M/H slot
Field sync 2
M/H slot
313 segments
24.2 ms
1 M/H slot
= 156
segments
313 segments
24.2 ms
M/H slot
1 segment
832 symbols
77.3 us
Fig. 23.25. Forming ATSC-M/H slots
23.5 ATSC-M/H Mobile DTV
487
23.5.3 ATSC M/H Multiplexer
The ATSC M/H multiplexer is actually the key element for the ATSC M/H
signal processing. It conditions the supplementary ATSC M/H data in a
"digestible" way for the ATSC M/H modulator. It provides for the IP encapsulation and for the error protection and for the pre-interleaving and inserting of the additional training sequences for the M/H receivers.
Considered in a simplified way, the ATSC M/H slots begin precisely at
a field boundary. In reality, however, they have an offset of 37 transport
stream packets or segments before the field sync. This is not shown in the
Figure and initially might only cause confusion.
968 ms
M/H
frame
1 M/H frame = 5 subframes
193.6 ms
M/H
sub-frame 0
M/H
sub-frame 1
M/H
sub-frame 2
M/H
sub-frame 3
M/H
sub-frame 4
12.1 ms
M/H
slot 0
M/H
slot 1
M/H
slot 2
M/H
slot 3
M/H
slot 4
M/H
slot 5
M/H
slot 6
M/H
M/H
M/H
M/H
slot 12 slot 13 slot 14 slot 15
main ATSC
M/H
group
1 M/H slot = 156 TS packets
1 M/H group = 118 TS packets
Fig. 23.26. ATSC-M/H framing
23.5.3.1 Compatible ATSC M/H Framing
An ATSC M/H field is here divided into two M/H slots. Each M/H slot has
a length of 156 segments (to recall: 3 x 52 interleaved TS packets = 156
packets). The actual ATSC M/H component is located in the first 118
segments whilst the rest of the 156 segments is always used for the ATSC
main stream. One to eight M/H slots make up a parade. A parade can carry
one or two ATSC M/H ensembles. A parade is thus nothing else but a series of M/H slots transmitting the same M/H contents. The data rate share
of a parade in the total of 19.38 Mbit/s of the net ATSC channel is 0.9 to
488
23 Digital Terrestrial TV to North American ATSC Standard
7.3 Mbit/s. However, the actual useful data rate of a parade is lower due to
the additional error protection.
In ATSC M/H, an M/H subframe consisting of 16 adjacent M/H slots is
first formed. 5 M/H subframes make up one M/H frame. It depends on the
number of slots (1 to 8) allocated to a parade which M/H slots are really
occupied with M/H content and this is defined in detail in the standard. If
an M/H slot is occupied with M/H content, only the first 118 segments or
transport stream packets are used for this purpose, the rest is reserved for
the main ATSC. The intention is that the MPEG buffers for main ATSC in
the receiver should not be empty.
M/H data
Main ATSC
Fig. 23.27. M/H data and main ATSC data
Data
in
ReedSolomon
Convol.
coder
Interleaver
Fig. 23.28. Additional Error Protection
23.5.3.2 Additional Error Protection
The ATSC M/H content is additionally error-protected, the error protection
already being added in the ATSC M/H multiplexer. The additional error
protection consists of a Reed Solomon error protection and of convolutional coding. The additional convolutional coding together with the trellis
coding with the ATSC modulator then results in a turbo code. After the
convolutional coding, the ATSC M/H content is additionally interleaved.
23.5 ATSC-M/H Mobile DTV
489
variable
187
lines
Data
CRC16
(depends on no. of allocated
slots and on FEC)
Reed-Solomon
(24, 36, 48 bytes)
1 CRC for
each line
Fig. 23.29. Additional Reed-Solomon FEC
SSC = Service Signalling Channel
OMA BCAST = Open Mobile Alliance Broadcast
FLUTE = File Delivery over Unidirectional Transport
Fig. 23.30. ATSC-M/H layer (source: Rohde&Schwarz)
23.5.3.3 Additional Training Sequences
The ATSC multiplexer also inserts 6 additional trainng sequences for the
equalizer in the receiver. This enables the receiver to adapt itself better to
unfavourable portable and mobile receiving conditions. It also enables a
receiver to handle single-frequency network conditions more easily.
490
23 Digital Terrestrial TV to North American ATSC Standard
23.5.3.4 Supplementary Data
Apart from the MPEG-4 streaming data, additional information of importance for the receiver is also transmitted in different layers in ATSC M/H,
which is
•
•
•
•
TPC - Transmission Parameter Channel,
FIC - Fast Information Channel,
SSC - Service Signalling Channel,
OMA BCAST - Open Mobile Alliance Broadcast.
The Transmission Parameter Channel and the Fast Information Channel
are included relatively closely at the lowest physical layer; both of them
being entered permanently in certain bytes in the M/H slots. The Transmission Parameter Channel signals the mapping of the M/H slots and the
additional error protection. Via the Fast Information Channel, the number
of services and their IDs are transmitted. The actual service names are
found in the Service Signalling Channel which is transmitted at the IP level. In addition, EPG data can be emitted in the OMA BCAST.
23.5.4 ATSC M/H Modulator
The ATSC M/H modulator receives the preconditioned data on a special
PID including the "normal" ATSC data on the usual PIDs signalled via PSI
and inserts the M/H Mobile DTV data into M/H slots. An M/H slot has a
length of half an ATSC field. An M/H time slot consists of a certain number of adjacent ATSC segments. The ATSC modulator is controlled by
signalling from the ATSC M/H multiplexer. The ATSC framing can now
no longer be freely selected in the ATSC modulator but is linked to the
ATSC M/H data. In addition, the ATSC modulator must now perform a
trellis reset at certain times. But for a "normal" ATSC receiver, the result
after the modulation only looks as if a part of the transmitted data were
contained in unknown, i.e. unreferenced PIDs. However, ATSC M/H receivers are well able to utilize these data, too.
23.5.5 Forming Single Frequency Networks
Due to the scarcity of frequencies, single frequency networks provide a
very economic way of reusing the same frequency over a relatively large
area. This possibility is used very intensively mainly in COFDM-based
transmission standards. ATSC is now attempting to use the same approach,
23.5 ATSC-M/H Mobile DTV
491
applying same rule that all transmitters in a single frequency ATSC network must be
•
•
•
•
synchronous in frequency,
synchronous in time,
synchronous in their data and
meeting emission times.
There are no guard intervals in single-carrier modulation. It is necessary
that appropriate equalizers in the receiver assist in recovering as much as is
possible from the conditions of reception. The essential factor in ATSC
with respect to single frequency networks is the frame synchronization of
all modulators involved and the emission time of the symbols. The modulators are synchronized via the multiplexer.
23.5.6 Summary
ATSC M/H "Mobile DTV" is an extension in the existing ATSC standard
to make ATSC more portable and useful in mobile applications. Numerous
backward compatible extension, simply ignored by a "normal" ATSC receiver, have been inserted into the ATSC signal. To be able to broadcast
ATSC M/H, an ATSC M/H multiplexer is required in addition to an ATSC
M/H-compatible modulator. New ATSC receivers can compensate for delay differences of approximately up to 40 s within certain limits in the
case of multipath reception and thus also provide for single-frequency networks in this order of magnitude; however this is mainly dependent on the
level differences of the receiving paths and on the number of paths, and
mainly on the receiver.
Fig. 23.31. Closed Captioning insert on a TV screen
492
23 Digital Terrestrial TV to North American ATSC Standard
23.6 Closed Captioning
It is, or was, normal practice in NTSC to transmit closed captioning (CC)
data in the vertical blanking interval in line 21 of the first and second field
of a TV frame. In principle, CC is used for sending subtitles, possibly in
various languages, in two bytes per field each to the TV receiver; i.e. texts
which are related to the current program. The relevant data rate can be calculated as:
•
•
2 bytes per field at 60 fields per second are
120 bytes per second per 8 bits = 820 bits/sec.
Fig. 23.32. Closed Captioning data in line 21 of an analog TV signal
Fig. 23.31. shows an example of the insertion of closed captioning on a
TV screen. The insert can be switched on or off by the viewer via the CC
key on their remote control. Fig. 23.32. shows line 21 with closed captioning data.
The relevant standard is EIA 608 A. Broadcasting of closed captioning
data is regulated by law in the US. The compatible transmission of such
data has also been standardized in ATSC in the EIA 708 B standard. The
CC data are not sent out in private PES streams as in DVB but via the optional user data after the picture header in the video PES steam. The data
are thus automatically synchronous with the video stream. The data rate is
ten times that of analog television. The data rate of the CC data in ATSC is
thus
23.7 Analog Switch-off
•
•
493
9600 bits,
or 20 bytes per picture user data field at 60 frames per second.
Fig. 23.3. shows the position of the picture user data in the video PES
stream. A CC decoder, which can be a component of an MPEG decoder
chip, fetches these data from the data field after the picture header and inserts the corresponding text information into the decoded image.
User data
bits insertion
possible
on
sequence,
GOP and
picture
Level,
behind the
header
Quantizer
Scale Factor
Quantizer Matrix
Extension
Sequence GOP
Pointer to
1st Picture
in PES
Packet
Sequence
Header
Picture
Slice
Macro Block
Picture Header
GOP Header
Slice Header
Packetized Elementary Stream
DTS
PTS
Macro Block
PES
Header
Length Indicator
up to 64 kByte,
PES
longer packets possible
Header (indicator = 0)
Transport Stream
Payload
Unit Start
Indicator = 1
4 Byte
184 Byte
TS Header Payload
Fig. 23.33. Position of the picture user data in the video PES stream
23.7 Analog Switch-off
The original plan was to switch off the analog high-power transmitters in
the US on 17th January 2009. However, the legislators moved this date to
12th June 2009 because of a scarcity of available digital TV receivers.
From this date on, analog terrestrial television by high-power transmitter
was past history and only low-power transmitters and analog services by
cable continued in operation.
Canada is planning its complete switch-over to ATSC for the 31st of
August 2011. In Mexico the date for the planned switch-off of analog television is in 2022 and South Korea wants to switch its analog TV services
494
23 Digital Terrestrial TV to North American ATSC Standard
from NTSC to ATSC by 31st December 2012, which is also the year for
the proposed analog switch-off in the UK and Australia.
Fig. 23.34. ATSC-M/H multiplexer Rohde&Schwarz AEM100
Bibliography: [A53], [EFA], [SFQ],
[7EB01_APP], [EIA608A], [EIA708B]
[SFU], [A153], [A110B],
24 ATSC/8VSB Measurements
In the following section, the measurements required at the air interface to
the North American terrestrial digital TV transmission system will be discussed in detail. The ATSC – Advanced Television Systems Committee –
standard employs a modulation method with a single carrier, that is 8VSB,
which stands for 8-level vestigial sideband modulation. The 8VSB constellation diagram does not exhibit points but lines. Due to the Q component
resulting from vestigial sideband filtering, eight lines are formed from the
originally eight points. As a basic rule in 8VSB, it can be said that the narrower the eight lines, the better the signal quality. While 8VSB modulation
appears relatively simple compared to the COFDM multicarrier method, it
exhibits correspondingly higher susceptibility to the various types of interference from the terrestrial environment.
The following causes of interference will, therefore, be discussed below:
•
•
•
•
•
•
•
Additive white Gaussian noise
Echoes
Amplitude and group-delay distortion
Phase jitter
IQ errors of modulator
Insufficient shoulder attenuation
Interferers
All of the above types of interference manifest themselves as bit errors
in the demodulated 8VSB signal. Bit errors can be corrected to a certain
extent by means of forward error correction (FEC). Vital in this context are
measurement of the bit error ratio and a detailed analysis of the causes of
bit errors.
24.1 Bit Error Ratio (BER) Measurement
In ATSC/8VSB, three different bit error ratios are known. These result
from two error protection methods being combined, i.e. Reed-Solomon
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_24, © Springer-Verlag Berlin Heidelberg 2010
496
24 ATSC/8VSB Measurements
block coding, and convolutional coding. The bit error ratios (BER) are as
follows:
•
•
•
Bit error ratio before Viterbi
Bit error ratio before Reed-Solomon
Bit error ratio after Reed-Solomon
MPEG-2
transport stream
RF
ATSC
frontend
Viterbi
decoder
BER before
Viterbi
RS
decoder
BER
after
Viterbi
MPEG-2
decoder
BER after RS
Fig. 24.1. Bit error ratios in ATSC
The most significant BER is the BER before Viterbi as it represents the
channel bit error ratio. But there is a problem: with trellis coding, the bit
error ratio before Viterbi cannot be measured technically because of ambiguities in the receiver in the Viterbi decoding.
The BER after Viterbi, i.e. before Reed-Solomon, is derived directly
from the Reed-Solomon decoder. The BER after Reed-Solomon, then, indicates non-correctable bit errors, i.e. more than 10 bit errors occurring in a
208-byte RS block coded transport stream packet. The BER after Reed
Solomon is likewise derived from the Reed-Solomon decoder. Noncorrectable bit errors are marked by transport error indicator bits (set to 1)
in the MPEG-2 transport stream. Bit error ratio measurement is performed
by means of an ATSC/8VSB test receiver.
24.2 8VSB Measurements Using a Spectrum Analyzer
By means of a spectrum analyzer, both in-band and – most importantly –
out-of-band measurements can be performed on the 8VSB signal. The parameters to be measured with a modern spectrum analyzer are as follows:
24.3 Constellation Analysis on 8VSB Signals
•
•
•
•
497
Shoulder attenuation
Amplitude frequency response
Pilot carrier amplitude
Harmonics
Make the following settings on a modern spectrum analyzer:
•
•
•
•
•
•
•
Center frequency at center of band
Span 20 MHz
RMS detector
Resolution bandwidth 20 kHz
Video bandwidth 200 kHz
Slow sweep time (>1 s) to allow averaging by RMS detector
No averaging function activated
Fig. 24.2. 8VSB signal spectrum with appropriate (left) and poor (right) vestigial
sideband suppression
Then the shoulder attenuation and, most importantly, the suppression of
the unwanted part of the lower sideband can be measured, as well as the
pilot amplitude and the amplitude distortion in the passband.
24.3 Constellation Analysis on 8VSB Signals
In contrast to a quadrature amplitude modulation (QAM) diagram, which
shows points, the constellation diagram of an 8VSB signal exhibits lines.
An ATSC test receiver usually comprises a constellation analyzer, which
displays the 8VSB diagram by 8 parallel vertical lines that should in the
ideal case be extremely narrow.
498
24 ATSC/8VSB Measurements
Fig. 24.3. Undistorted constellation diagram of an ATSC/8VSB signal
Fig. 24.4. 8VSB constellation diagram revealing noise impairment
The constellation diagram in Fig. 24.3. with very narrow lines reveals
only a slight impairment by noise, such as caused already in the ATSC
modulator or transmitter. As a basic rule, it can be said that the narrower
the lines, the less significant the signal distortion. In the event of pure
noise distortion, the lines are uniformly widened over their entire length.
The wider the lines, the greater the impairment due to noise. In the constellation analysis, the RMS value of the noise is determined. Based on a statistical function, i.e. the Gaussian distribution (normal distribution), the
standard deviation of the I/Q points obtained in the decision fields of the
constellation diagram is determined. From the RMS noise value, the test
24.3 Constellation Analysis on 8VSB Signals
499
receiver calculates the signal-to-noise ratio (S/N ratio) in dB referenced to
the signal power, which is likewise calculated by the test receiver.
In the event of phase jitter, the lines in the decision fields of the constellation diagram are trumpet-shaped, i.e. they become increasingly wider as
the distance from the horizontal center line increases (Fig. 24.5.).
Fig. 24.5. 8VSB constellation diagram revealing phase jitter
Q
Error
vector
Real
vector
Decision
border
Ideal
vector
I
Fig. 24.6. Determining the MER of an 8VSB signal
The Modulation Error Ratio (MER) parameter summarizes all errors
that can be measured within a constellation diagram. For each type of error
(interference), an error vector is continually calculated. The sum of the
squares (RMS value) of all error vectors is calculated. The ratio of the error-vector RMS value and the signal amplitude yields the MER, which is
500
24 ATSC/8VSB Measurements
usually specified in dB. In the event of pure noise impairment, the MER is
equal to the S/N ratio.
The following applies:
MER[dB] <= S/N[dB];
MERRMS[dB] = -10 log(1/n • (|error_vector|2/Psignal_without_pilot);
Fig. 24.7. Numerical results output by an 8VSB test receiver
Many test parameters are also output as numerical results by the 8VSB
test receiver. These include the signal amplitude, bit error rate, pilot amplitude, symbol rate, phase jitter, S/N ratio, and MER.
24.4 Measuring Amplitude Response and Group Delay
Response
Although the ATSC/8VSB signal carries no pilot signals that would provide information on channel quality, the amplitude, group-delay and phase
24.4 Measuring Amplitude Response and Group Delay Response
501
response can be roughly determined – with the aid of the test receiver
equalizer – from the PRBS sequences contained in the 8VSB signal. The
signal characteristics output by the 8VSB test receiver can be used to align
an ATSC modulator or transmitter, for example. The equalizer data also
provides information on echoes in the transmission channel and allows
calculation of the impulse response.
Fig. 24.8. Amplitude and phase response measurement using an 8VSB test receiver [EFA]
Fig. 24.9. Ghost pattern/impulse response
Bibliography: [A53], [EFA], [SFQ], [SFU], [ETL]
25 Digital Terrestrial Television according to
ISDB-T
25.1 Introduction
The Japanese standard for digital terrestrial television is called ISDB-T,
i.e. Integrated Services Digital Broadcasting – Terrestrial, which was
adopted in 1999, quite a long time after DVB-T and ATSC. This delay
made it possible to take into account also the experience gained with the
older standards. Unlike ATSC, where a single-carrier method is used, it
was decided (correctly) for ISDB-T to use a COFDM multicarrier system
as in DVB-T. ISDB-T is even more complex than DVB-T; it is also more
robust because of the greater interleaving with time. The first pilot station
was installed on the Tokyo Tower and overall, ISDB-T started with eleven
pilot stations throughout Japan.
QPSK
16QAM
64QAM
DQPSK
Mode I, II, III:
f=~4kHz, ~2kHz,
f
~1kHz
Channel bandwidth
6, 7, 8 MHz
Data rate: 3.7 ... 23.4 Mbit/s
f
Fig. 25.1. COFDM in ISDB-T
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_25, © Springer-Verlag Berlin Heidelberg 2010
504
25 Digital Terrestrial Television according to ISDB-T
25.2 ISDB-T Concept
In ISDB-T, COFDM (coded orthogonal frequency division multiplex) is
used in 2K, 4K and 8K mode (Fig. 25.1.). The 6 MHz-wide channel can be
subdivided into 13 subbands (Fig. 25.2.) in which different modulation parameters can be selected and contents transmitted. Time interleaving can
be optionally switched on in various stages. With an actual channel bandwidth of 6 MHz, the useful band only has a width of 5.57 MHz, i.e. there is
a guard band of about 200 kHz each for the upper and lower adjacent channels. One subband of the ISDB-T channel has a width of 430 kHz.
It is possible to select different types of modulation in ISDB-T:
•
•
•
•
QPSK with channel correction
16QAM with channel correction
64QAM with channel correction
DQPSK without channel correction (not required with DQPSK).
There are 3 possible modes (6 MHz channel as example):
•
Mode I, with
108 carriers per subband
3.968 kHz subcarrier spacing
1404 carriers within the channel
2048-points IFFT
• Mode II, with
216 carriers per subband
1.9841 kHz subcarrier spacing
2808 carriers within the channel
4196-points IFFT
• Mode III, with
432 carriers per subband
0.99206 kHz subcarrier spacing
5616 carriers within the channel
8192-points IFFT
As already mentioned, the full 6 MHz channel can be subdivided into 13
subbands of precisely 3000/7 kHz = 428.7 kHz (Fig. 25.2.) each.
Not all of the 2048, 4192 or 8192 COFDM carriers in mode I, II or III
are actually used as payload carriers. In ISDB-T, there are
•
Zero carriers, i.e. those which are not used,
25.2 ISDB-T Concept
•
•
•
•
•
505
Data carriers, i.e. real payload,
Scattered pilots (but not with DQPSK),
Continual pilots
TMCC (Transmission and Multiplexing Configuration Control)
carriers,
AC (Auxiliary Channels).
428.7 kHz @ 6 MHz
13
subchannels
6, 7, 8 MHz
channel bandwidth
Convolutional
coder
Time interleaver
RS(188, 204)
coder
MPEG-2
TS
in
Scrambler
Fig. 25.2. Subchannels in ISDB-T
Data
out
Fig. 25.3. ISDB-T FEC
The net data rates are between 280.85 kbit/s per segment or 3.7 Mbit/s
per channel and 1787.28 kbit/s per segment or 23.2 Mbit/s per channel.
Due to the subband or segment concept, it is possible to build both narrow-band receivers which receive only one or a number of subbands and
broadband receivers which receive the entire 6-MHz-wide channel.
In principle, the ISDB-T modulator configuration is similar to that of a
DVB-T modulator. It has outer error protection, implemented as Reed
506
25 Digital Terrestrial Television according to ISDB-T
Solomon RS(204,188) coder, an energy dispersal unit, an interleaver, an
inner coder implemented as convolutional coder, a configurable time interleaver which can be switched on or off, a frequency interleaver, the
COFDM frame adapter, the IFFT etc.. The basic configuration of the error
protection corresponds directly to that of DVB-T. The selectable code rates
are like those of DVB-T:
•
•
•
•
•
1/2
2/3
3/4
5/6
7/8
but the time interleaving is much deeper and can also be configured in
stages:
•
•
•
•
0
1
2
4
The following guard interval lengths can be set:
•
•
•
•
1/4
1/8
1/16
1/32
25.1 Forming Layers
The individual segments in ISDB-T can be combined to form a total of 3
layers in which different transmission parameters (type of modulation and
error protection) can be selected. In the 3 hierarchical layers, different contents can then be error-protected to different degrees and transmitted with
modulation of different robustness. The number of segments to be combined in one layer is selectable but the same transmission parameters are
used in each segment of a layer. In the case of 3 layers, in principle, 3 associated, mutually independent data streams must then be supplied. However, by using a “trick”, this can be done by supplying one common transport stream (see also Chapter 25.3).
25.3 Changes in the Transport Stream Structure
A
507
Layer C
B
13
subchannels
(segments)
Fig. 25.4. Forming layers
25.2 Baseband Encoding
In the baseband encoding/source encoding area, of course, ISDB-T is just
as open as any other standard, too. MPEG-2 video and audio are just as
possible here as are the new, more optimal MPEG-4 video and audio codecs. Currently, MPEG-2 video is used for baseband coding (SDTV and
HDTV), and MPEG-2 AAC for audio in Japan.
204 byte
188 byte
TS paket
Header
Payload
8 byte
8 byte
Multiplex
position
Parity
(option)
Fig. 25.5. Layer signalling in the 16 overhead bytes in the transport stream packet
in the transport stream supplied
25.3 Changes in the Transport Stream Structure
In addition to activating the 3 layers via the transport stream structure,
there are also extensions and new tables which, although they largely correspond to those of DVB-SI, differ in their detail or are completely different in some cases. These tables are called ARIB tables (Association of Radio Industries and Business) in ISDB-T. Control information for assigning
508
25 Digital Terrestrial Television according to ISDB-T
data to the individual layers of ISDB=T is found in the transport stream in
a 16-bit extension in the transport steam packets which are usually 188
bytes long (Fig. 25.5.). The “multiplex position” informs the ISDB-T
modulator about the layer for which the transport stream currently contains
information. This extension bears no relationship to the error protection in
DVB or ISDB-T. However, it has already been common practice to provide the 188-byte-long packets with 16 bytes of dummy information at the
baseband level in DVB. The possibility of using this dummy information
for layer signalling has been taken up in ISDB-T.
Fig. 25.6. Possible adjustments on the ISDB-T encoder of the Rohde&Schwarz
test transmitter SFU to illustrate the diversity of ISDB-T; It is possible to select
different transmission parameters in layers A, B and C, called “portion” here.
The order of segments is counted not from left to right, i.e. from channel
start to channel end, but starts in the center with segment S0. Segment S1
is then on the left of S0 and S2 is on the right of S0. S0 is the segment used
in the 1-segment mode. ISDB-Tsb (Integrated Services Digital Broadcast
Terrestrial – sound broadcast) is a narrowband version where only 1 … 3
segments are used. And this is the order before the frequency interleaver.
After the frequency interleaver the carriers from the different segments are
distributed over the complete channel to avoid frequency selective problems. Only in case of the narrow band ISDB-T versions the subbands are
placed in the center.
25.4 Channel Tables
509
S10 S8 S7 S5 S3 S1 S0 S2 S4 S6 S7 S9 S11
S1 S0 S2
Fig. 25.7. Order of segments in ISDB-T (top) and ISDB-Tsb (bottom)
Fig. 25.8. ISDB-T spectrum
25.4 Channel Tables
In ISDB-T, the TV channels have been shifted upward by 1/7 MHz. As an
example, Channel 7 is originally located at 177 MHz. The new center frequency is here now 177 + 1/7 MHz = 177.143 MHz.
510
25 Digital Terrestrial Television according to ISDB-T
25.5 Performance of ISDB-T
Table 25.1 shows the minimum signal/noise ratios as a function of the
transmission parameters, specified in the ISDB-T standard. They correspond to a bit error rate of 2 • 10-4 before Reed-Solomon and to a quasierror-free data stream after Reed-Solomon.
Table 25.1. Transmission parameters of ISDB-T and theoretical minimum signal/noise ratio according to the standard
Modulation
DQPSK
QPSK
16QAM
64QAM
CR=1/2
[dB]
6.2
4.9
11.5
16.5
CR=2/3
[dB]
7.7
6.6
13.5
18.7
CR=3/4
[dB]
8.7
7.5
14.6
20.1
CR=5/6
[dB]
9.6
8.5
15.6
21.3
CR=7/8
[dB]
10.4
9.1
16.2
22.0
Table 25.2 shows the data rates reproduced in the ISDB-T standard as a
function of the transmission parameters.
Table 25.2. Data rates of ISDB-T as a function of the ISDB-T transmission parameters according to the standard (at 6 MHz bandwidth)
Modulation Code rate
DQPSK
QPSK
DQPSK
QPSK
DQPSK
QPSK
DQPSK
QPSK
DQPSK
QPSK
16QAM
16QAM
16QAM
16QAM
16QAM
64QAM
64QAM
64QAM
64QAM
64QAM
½
g=1/4
[Mbit/s]
3.651
g=1/8
[Mbit/s]
4.056
g=1/16
[Mbit/s]
4.295
g=1/32
[Mbit/s]
4.425
2/3
4.868
5.409
5.727
5.900
¾
5.476
6.085
6.443
6.638
5/6
6.085
6.761
7.159
7.376
7/8
6.389
7.099
7.517
7.744
½
2/3
¾
5/6
7/8
½
2/3
¾
5/6
7/8
7.302
9.736
10.953
12.170
12.779
10.953
14.604
16.430
18.255
19.168
8.133
10.818
12.170
13.522
14.198
12.170
16.227
18.255
20.284
21.298
8.590
11.454
12.886
14.318
15.034
12.886
17.181
19.329
21.477
22.551
8.851
11.801
13.276
14.752
15.489
13.276
17.702
19.915
22.128
23.234
25.6 Other ISDB Standards
511
Table 25.3. Data rates of ISDB-T per segment in dependence of the ISDB-T
transmission parameters according to the standard (at 6 MHz bandwidth)
Modulation Code rate
DQPSK
QPSK
DQPSK
QPSK
DQPSK
QPSK
DQPSK
QPSK
DQPSK
QPSK
16QAM
16QAM
16QAM
16QAM
16QAM
64QAM
64QAM
64QAM
64QAM
64QAM
½
g=1/4
[kbit/s]
280.25
g=1/8
[kbit/s]
312.06
g=1/16
[kbit/s]
330.42
g=1/32
[kbit/s]
340.43
2/3
374.47
416.08
440.56
453.91
¾
421.28
468.09
495.63
510.65
5/6
468.09
520.10
550.70
567.39
7/8
491.50
546.11
578.23
595.76
½
2/3
¾
5/6
7/8
½
2/3
¾
5/6
7/8
561.71
748.95
842.57
936.19
983.00
842.57
1123.43
1263.86
1404.29
1474.50
624.13
832.17
936.19
1040.21
1092.22
936.19
1248.26
1404.29
1560.32
1638.34
660.84
881.12
991.26
1101.40
1156.47
991.26
1321.68
1486.90
1652.11
1734.71
680.87
907.82
1021.30
1134.78
1191.52
1021.30
1361.74
1531.95
1702.17
1787.28
25.6 Other ISDB Standards
Apart from ISDB-T, there are some other ARIB standards which are:
•
•
•
•
ISDB-S (satellite)
ISDB-Tsb (terrestrial sound broadcast)
ISDB-C (cable)
ISDB-Tmm (terrestrial mobile multi-media).
ISDB-S is more effective than DVB-S by a factor of 1.5 and also uses,
among other things, 8PSK in single-carrier modulation.
ISDB-C corresponds to a 6 MHz variant of DVB-C and is ITU-T J83C,
to put it precisely. Apart from its bandwidth and the roll-off factor, ITU-T
J83C is virtually identical with DVB-C and reference is made here to the
corresponding chapter.
512
25 Digital Terrestrial Television according to ISDB-T
ISDB-Tsb and ISDB-Tmm are the narrowband versions of ISDB-T, occupying only 1 to 3 segments. There, too, the more effective MPEG-4
baseband coding is provided.
25.7 ISDB-T measurements
ISDB-T measurements are very similar to DVB-T measurements. But
there are different constellation diagrams possible in the different layers or
portions. Fig. 25.9. shows the different constellation diagrams in the different layers. In case of differential modulation the MER is about 3 dB
lower in comparison to coherent modulation (Fig 25.10. center subchannels).
ISDB-T measurements is
•
•
•
•
•
•
Constellation analysis (Fig. 25.9.)
MER measurement (Fig. 25.10. and Fig. 25.11.)
BER measurement (in all 3 layers)
IQ impairment measurement
Impulse response measurement (especially in SFN’s)
MPEG transport stream analysis
For more details please see chapter 21 – DVB-T measurements.
Fig. 25.9. ISDB-T constellation diagrams in the 3 different layers [ETL]
25.8 Summary
513
Fig. 25.10. MER(f) measurement in ISDB-T
Fig. 25.11. ISDB-T measurement list [ETL]
25.8 Summary
In addition to the 6 MHz channel normally used in Japan, ISDB-T is also
defined for 7 and 8 MHz channels. However, it is doubtful that it will be
514
25 Digital Terrestrial Television according to ISDB-T
widely used in 7 MHz and 8 MHz countries because DVB-T has found
wide-spread acceptance, in the meantime.
ISDB-T is certainly the more flexible standard and, because of the possibility of long time interleaving, also the more robust standard in some
applications.
By its very nature, ISDB-T has SFN capability, of course, and single
frequency networks are also being formed.
Brazil is the first country outside Japan which has decided to adopt
ISDB-T. In Brazil, MPEG-4 AVC is used as baseband coding, plus
MPEG-4 LC AAC or MPEG-4 HE AAC. The Brazilian terrestrial digital
TV standard is called SBTVD, i.e. Sistema Brasileiro de Televisao Digital.
Bibliography: [ISDB-T], [SFU], [ETL]
26 Digital Audio Broadcasting - DAB
Although DAB (Digital Audio Broadcasting) was introduced back in the
early days of the nineties, well before DVB, it is still relatively unknown to
the public in many countries and it is only in a few countries such as, e.g.
the UK that some measure of success of DAB in the market can be registered. This chapter deals with the principles of the digital sound radio standard DAB.
At first, however, let us consider the history of sound radio. The age of
the transmission of audio signals for broadcasting purposes began in the
year 1923 with medium-wave broadcasting (AM). In 1948, the first FM
transmitter was taken into operation, developed and manufactured by
Rohde&Schwarz. The first FM home receivers were also developed and
produced by Rohde&Schwarz. 1983 was the year when everyone took the
step from analog audio to digital audio with the introduction of the compact disk, the audio CD. In 1991, digital audio signals intended for the
public at large were broadcast for the first time via satellite in Europe,
DSR (Digital Satellite Radio). This method, operating without compression, did not last long, however, and was little known in public. 1993 then,
ADR (Astra Digital Radio) started operation which is broadcast on subcarriers of the ASTRA satellite system on which analog TV programs are also
transmitted. The MUSICAM method, used up to the present for audio
compression in MPEG-1 and MPEG-2 layer II and is also used in DAB or,
to put it more precisely, was developed for DAB as part of the DAB project, was laid down in 1989. Digital Audio Broadcasting, DAB, was developed at the beginning of the nineties and used the then revolutionary new
techniques of MPEG-1 and MPEG-2 audio and the COFDM (Coded Orthogonal Frequency Division Multiplex) modulation method. In the midnineties then the DVB-S, DVB-C and DVB-T standards for digital television were finalized and thus the age of digital television had also begun.
Since 2001, there is a further digital sound radio standard DRM (Digital
Radio Mondiale), intended for digital short- and medium wave use, which
is also based on COFDM but uses MPEG-4 AAC audio coding.
The first DAB pilot test was carried out in 1991 in Munich. Germany
currently has a DAB coverage of about 80%, mainly in Band III. There are
also L-Band transmitters for local programs. As ever, DAB is almost unW. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_26, © Springer-Verlag Berlin Heidelberg 2010
516
26 Digital Audio Broadcasting - DAB
known to the public in Germany, one of the reasons being that there have
been no receivers available for a long time, and also that the contents
broadcast do not really cover the variety obtainable in FM radio. None of
this has anything to do with technical reasons, however. In the UK, DAB
was greatly expanded in 2003/2004, Singapore has 100% coverage, Belgium 90%. DAB is broadcast in France, Spain, Portugal and Canada.
DAB activities are taking place in 27 countries and DAB frequencies are
available in 44 countries.
Synchronous transfer mode (PDH, SDH, DAB)
...
Ch. 1 Ch. 2 Ch. 3
...
Ch. n Ch. 1 Ch. 2 Ch. 3
...
Ch. n ...
Asynchronous transfer mode (ATM, MPEG-TS / DVB)
...
Ch. 3 Ch. 2 Ch. n
...
Ch. 2 unused Ch. 8 Ch. n
...
Ch. 7 ...
Fig. 26.1. Synchronous and asynchronous transfer mode
(E1,
physical data rate
= 2.048 Mbit/s,
G.703/G.704)
Playout
ETI
DAB
mod.
RF
COFDM
Net data rate =
1.2 … 1.73 Mbit/s
Gross data rate =
2.4 Mbit/s
ETI = Ensemble Transport Interface
Fig. 26.2. DAB transmission link
RF
26.1 Comparing DAB and DVB
517
26.1 Comparing DAB and DVB
In a comparison of DAB and DVB, the basic characteristics of both methods will first be compared, pointing out properties and differences. In principle, it is possible to transmit data synchronously or asynchronously (Fig.
16.1.). In synchronous transmission, the data rate is constant for each data
channel and the time slots of the individual data channels are fixed. In
asynchronous transmission, the data rate of the individual data channels
can be constant or it can vary. The time slots have no fixed allocation.
They are allocated as required and their order in the individual channels
can thus be completely random. Examples of synchronous data transmission are PDH (Plesiochronous Digital Hierarchy), SDH (Synchronous
Digital Hierarchy) and DAB (Digital Audio Broadcasting). Examples of
asynchronous data transmission are ATM (Asynchronous Transfer Mode)
and the MPEG-2 transport stream/Digital Video Broadcasting (DVB).
DAB is a completely synchronous system, a completely synchronous
data stream being produced right back in the playout center, i.e. at the
point where the DAB multiplex signal is generated. The data rates of the
individual contents are constant and are always a multiple of 8 kbit/s. The
time slots in which the contents from the individual sources are transmitted
are permanently allocated and vary only when there is a complete change
in the multiplex, i.e. in the composition of the data stream. The data signal
coming from the multiplexer which is supplied to the DAB modulator and
transmitter is called ETI (Ensemble Transport Interface) (Fig. 26.2.). The
multiplexed data stream, or multiplex, itself is called ensemble. The ETI
signal uses E1 transmission paths known from telecommunication which
have a physical data rate of 2.048 Mbit/s. E1 would correspond to 30
ISDN channels and 2 signalling channels of 64 kbit/s each, also called
G.703 and G.704 interface. Physically, these are PDH interfaces but DAB
uses a different protocol. Although the physical data rate is 2048 kbit/s, the
actual net data rate of the DAB signal transported across it is between (0.8)
1.2 ... 1.73 Mbit/s. The ETI signal is transmitted either without error protection, or with a Reed Solomon error protection code which, however, is
removed again at the input of the DAB modulator. The error protection of
the DAB system itself is added only in the DAB modulator although this is
often wrongly shown to be different in various references. The modulation
method used in DAB is COFDM (Coded Orthogonal Frequency Division
Multiplex) and the subcarriers are /4-shift DQPSK modulated. After the
error protection has been added the gross data rate of the DAB signal is 2.4
Mbit/s. A special feature of DAB consists in that the different contents can
be error protected to a different degree (unequal FEC).
518
26 Digital Audio Broadcasting - DAB
MPEG-2, and thus DVB, is a completely asynchronous system.
The MPEG-2 transport stream is a baseband signal forming the input
signal to a DVB modulator. The MPEG-2 transport stream is generated in
the playout center by encoding and multiplexing the individual programs
(services) and is then supplied to the modulator via various transmission
paths (Fig. 26.3.). In the DVB modulator, it must be decided how, i.e. by
which transmission path, the MPEG-2 transport stream is to be emitted:
terrestrial (DVB-T), by cable (DVB-C) or by satellite (DVB-S). Naturally,
the transmission rates and modulation methods differ for the individual
transmission methods. In DVB-T, COFDM is used in conjunction with
QPSK, 16QAM or 64QAM. In DVB-C, it is either 64QAM or 256QAM
depending on the type of cable link (coaxial cable or optical fibre). In
DVB-S, the modulation method of choice has been QPSK because of the
poor signal/noise ratio in the channel.
Playout
DVB
mod.
MPEG-2
transport
stream
approx. 13…40 Mbit/s
RF
RF
COFDM @ DVB-T (terrestrical)
64QAM/256QAM @ DVB-C (cable)
QPSK @ DVB-S (satellite)
Fig. 26.3. DVB transmission link
In DVB, all contents transmitted carry the same degree of error protection (equal FEC).
As a rule, the data rate in DVB-S is about 38 Mbit/s. It only depends on
the symbol rate selected and on the code rate, i.e. the error protection. Using QPSK 2 bits/symbol can be transmitted. The symbol rate is mostly
27.5 Msymbols/s. If 3/4 is selected as code rate, the resultant data rate is
38.01 Mbit/s.
If, e.g., 64QAM (coax networks) is selected in DVB-C, and a symbol
rate of 6.9 Msymbols/s, the resultant net data rate is 38.15 Mbit/s.
In DVB-T, the possible data rate is between 4 and about 32 Mbit/s depending on operating mode (type of modulation - QPSK, 16QAM,
26.1 Comparing DAB and DVB
519
64QAM, error protection, guard interval, bandwidth). The usual data rate
is, however, approx. 15 Mbit/s in applications allowing portable reception
and approx. 22 Mbit/s in stationary applications with a roof antenna. A
DVB-T broadcast network is designed either for portable reception or for
roof antenna reception, i.e. if a roof antenna is used in a DVB-T network
designed for portable reception, this will not produce an increase in data
rate.
The MPEG-2 transport stream is the data signal supplied to the DVB
modulators. It consists of packets with a constant length of 188 bytes. The
MPEG-2 transport stream represents asynchronous transmission, i.e. the
individual contents to be transmitted are keyed into the payload area of the
transport stream packets purely randomly as required. The contents contained in the transport stream can have completely different data rates
which do not need to be absolutely constant, either. The only rule relating
to data rates is that the aggregate data rate provided by the channel must
not be exceeded. And, naturally, the data rate of the MPEG-2 transport
stream must correspond absolutely to the input data rate of the DVB
modulators resulting from the modulation parameters.
Table 26.1. DAB and DVB comparison
Transfer mode
Forward error
correction (FEC)
Modulation
Transmission link
Digital Audio
Broadcasting DAB
synchronous
unequal
COFDM mit /4shift DQPSK
terrestrical
Digital Video Broadcasting DVB
asynchronous
equal
single carrier QPSK, 64QAM,
256QAM or COFDM with
QPSK, 16QAM, 64QAM
satellite, cable, terrestrical
In summary: DAB is a completely synchronous transmission system and
DVB is a completely asynchronous one. Remembering this will make it
easier to gain a better understanding of the characteristics of both systems
The error protection in DAB is unequal, meaning it can be selected to be
different for different contents, whereas it is equal for all contents to be
transmitted in DVB and, because of the asynchronous mode, could not
even be selected to be different since it is not known what content is being
transmitted when.
The DAB modulator demultiplexes the current content in the ETI signal
and takes it into consideration. The DVB modulator is not interested in the
520
26 Digital Audio Broadcasting - DAB
current content transmitted. In DAB, the modulation method is COFDM
with /4-shift DQPSK. DVB uses single-carrier transmission or COFDM
depending on transmission path. DAB is intended for terrestrial applications whereas DVB provides terrestrial, cable and satellite transmission
standards. Satellite transmission is provided for in DAB but currently not
used.
“Digital Radio 1“
“P1“
Audio
1
SC1
“BR1“
Audio
2
SC2 SC3 SC4
Ensemble
“BR3“
Data
1
SC5
Audio
3
SC6
“P2“
Data
2
SC32
Audio
4
…
Services
Service
components
Common
Interleaved
Frame
SC = subchannel (capacity = n * 8 kbit/s per subchannel)
Fig. 26.4. DAB ensemble
26.2 An Overview of DAB
The following sections will provide a brief overview of DAB - Digital Audio Broadcasting. The DAB Standard is the ETSI Standard ETS300401.
In the Standard, the data structure, the FEC and the COFDM modulation
of the DAB Standard is described. In addition, the ETI (Ensemble Transport Interface) supply signal is described in ETS300799, and in
ETS300797 the supply signals for the ensemble multiplexer STI (Service
Transport Interface) are described. A further important document is
TR101496 which contains guidelines and rules for the implementation and
operation of DAB. Furthermore, ETS301234 describes how multimedia
objects (data broadcasting) can be transmitted in DAB.
Fig. 26.4. shows an example of the composition of a multiplexed DAB
data stream. The term "Ensemble" covers several programs which are
26.2 An Overview of DAB
521
combined to form one data stream. In the present case, the ensemble given
the exemplary name "Digital Radio 1" is composed of 4 programs, the socalled services, here having the designations "P1", "BR1", "BR3" and
"P2". These services, in turn, can be composed of a number of service
components. A service component can be, e.g. an audio stream or a data
stream. In the example, the service "P1" contains an audio stream, "Audio1". This audio stream is physically transmitted in subchannel SC1.
"BR" is composed of an audio stream "Audio2" and a data stream "Data1"
which are broadcast in subchannels SC2 and SC3. Each subchannel has a
capacity of n • 8 kbit/s. The transmission in the subchannels is completely
synchronous, i.e. the order of the subchannels is always the same and the
data rates in the subchannels are always constant. All subchannels together
- up to a possible maximum of 64 - result in the so-called Common Interleaved Frame. Service components can be associated with a number of
services, e.g. as in the example "Data2".
During their transmission in the DAB system, the different subchannels
can be provided with different degrees of error protection (unequal FEC).
SC1
ETI
SC2
FEC 2
…
OFDM
FEC 1
RF
SCn
FEC n
DAB modulator
SC =
subchannel
(up to 64)
Fig. 26.5. DAB modulator
The data stream generated in the DAB multiplexer is called ETI (Ensemble Transport Interface). It contains all programs and contents to be
broadcast later via the DAB transmitter. The ETI signal can be supplied to
the modulator from the playout center, e.g. via optical fiber links via existing telecommunication networks or by satellite. A suitable link for this
purpose is an E1 link having a data rate of 2.048 Mbit/s.
522
26 Digital Audio Broadcasting - DAB
SC1
SC2
Scrambling
Convolutional
coding
Time
interleaving
Scrambling
Convolutional
coding
Time
interleaving
…
SCn
Scrambling
Convolutional
coding
OFDM
In the DAB modulator, the COFDM is carried out (Fig. 26.5.). The data
stream is first provided with error protection and then COFDM-modulated.
After the modulator, the RF signal power is amplified and then radiated via
the antenna.
In DAB, all subchannels are error-protected individually and to different
degrees. Up to 64 subchannels are possible. The FEC is provided in the
DAB modulator. In many block diagrams, FEC is often described in conjunction with the DAB multiplexer which, although it is not wrong in principle, does not correspond to reality. The DAB multiplexer forms the ETI
data signal in which the subchannels are transmitted synchronously and
unprotected.
The ETI, however, carries the information about how much protection is
to be provided for the individual channels. The ETI data stream is then
split up in the DAB modulator and each subchannel is then error-protected
to a different degree in accordance with the signalling in the ETI. The subchannels provided with FEC are then supplied to the COFDM modulator.
Time
interleaving
SC = subchannel (up to 64)
Fig. 26.6. Forward error correction (FEC) in DAB
The error protection in DAB (Fig. 16.6.) is composed of scrambling followed by convolutional coding. In addition, the DAB signal is then subjected to long time interleaving, i.e. the data are interleaved over time so
that they are more resistant to block errors during the transmission. Each
subchannel can be error-protected to a different degree (unequal forward
error correction). The data from all subchannels are then supplied to the
COFDM modulator which first carries out frequency interleaving and then
modulates them onto a large number of COFDM subcarriers.
26.2 An Overview of DAB
523
There are 4 different selectable modes in DAB. These modes are provided for different applications and frequency bands. Mode I is used in the
VHF band and Mode II to IV are used in the L band, depending on frequency and application. The number of carriers is between 192 and 1536
and the bandwidth of the DAB signal is always 1.536 MHz. The difference
between the modes is simply the symbol length and the number of subcarriers used.
Mode I has the longest symbol and the most subcarriers and thus the
smallest subcarrier spacing. This is followed by Mode IV, Mode II and finally Mode III with the shortest symbol period and the least carriers and
thus the largest subcarrier spacing. In principle, however, it holds true that
the longer the COFDM symbol, the better the echo tolerance and the
smaller the subcarrier spacing, the poorer the suitability for mobile applications.
The modes actually used in practice are Mode I for the VHF band and
Mode II for the L band
Table 26.2. DAB modes
Subcarrier
spacing
[kHz]
1
No. of
COFDM
carriers
Used for
Symbol
duration
[µs]
1536
1000
L band
(<1.5
GHz)
4
384
250
62
24 ms
76
symbols
L band
(<3 GHz)
L band
(<1.5
GHz)
8
192
singlefrequency
network
(SFN)
multifrequency
network
(MFN)
satellite
Guard
interval
duration
[µs]
246
125
31
2
768
500
123
24 ms
152 symbols
48 ms
76 symbols
Mode
Frequency
range
I
Band III
VHF
II
III
IV
small
singlefrequency
network
(SFN)
Frame
length
96 ms
76 symbols
The audio signals in DAB are coded to MPEG-1 or MPEG-2 (Layer II),
i.e. compressed from about 1.5 Mbit/s to 64 ... 384 kbit/s. During this
process, the audio signal is divided into 24 or 48 ms long sections which
are then individually compressed, using a type of perceptual coding in
which audio signal components inaudible to the human ear are omitted.
These methods are based on the MUSICAM (Masking pattern adapted
Universal Subband Integrated Coding And Multiplexing) principle described in ISO/IEC Standards 11172-3 (MPEG-1) and 13818-3 (MPEG-2)
524
26 Digital Audio Broadcasting - DAB
Bit
allocation
SCFSI
Header
CRC
Scale
factors
Stuffing
MPEG-1, 2
compatible
part
DAB
extension
and actually developed for DAB as part of the DAB project. In MPEG-1
and -2 it is possible to transmit audio in mono, stereo, dual sound and joint
stereo modes. The frame length is 24 ms in MPEG-1 and 48 ms in MPEG2. These frame lengths are also found in the DAB Standard and also affect
the length of the COFDM frames. The same applies as before: DAB is a
completely synchronous transmission system where all processes are synchronized with one another.
Sub-band samples
X-PAD
SCF-CRC
F-PAD
Fig. 26.7. DAB audio frame
Fig. 26.7. shows the structure of a DAB audio frame. An MPEG-1compatible frame has a length of 24 ms. The frame begins with a header
containing 32 bits of system information. The header is protected by a 16
bit long CRC checksum. This is followed by the block with the bit allocation in the individual sub-bands, followed by the scale factors and subband samples. In addition, ancillary data can be optionally transmitted.
The sampling rate of the audio signal is 48 kHz in MPEG-1 and thus does
not correspond to the 44.1 kHz of the audio CD. The data rates are between 32 and 192 kbit/s for a single channel or between 64 and 384 kbit/s
for stereo, joint stereo or dual sound. The data rates are multiples of 8
kbit/s. In MPEG-2, the MPEG-1 frame is supplemented by an MPEG-2 extension. In MPEG-2 Layer II, the frame length is 48 ms and the sampling
rate of the audio signal is 24 kHz.
This audio frame structure of the MPEG-1 and -2 Standards is repeated
in DAB. The MPEG-1- and MPEG-2-compatible part is supplemented by
a DAB extension in which program-associated data (PAD) are transmitted. Between these, stuffing bytes (padding) are used, if necessary. In the
PAD, a distinction is made between the extended PAD "X-PAD" and the
fixed PAD "F-PAD". Among other things, the PAD include an identifier
for music/voice, program-related text and additional error protection.
DAB audio data rates normally used in practice are:
26.3 The Physical Layer of DAB
•
•
525
Germany:
mostly 192 kbit/s, PL3
60 kbit/s or 192 kbit/s in some cases, PL4 (one additional per
gram)
UK:
256 kbit/s, classical music
128 kbit/s, popular music,
64 kbit/s, voice
Q
/4-shift-DQPSK
I
Signal bandwidth =
1.536 MHz
f
f=1, 2, 4, 8kHz
Channel bandwidth
7/4 MHz = 1.75 MHz
Fig. 26.8. DAB COFDM channel
26.3 The Physical Layer of DAB
In the section following, the implementation of COFDM in Digital Audio
Broadcasting DAB will be discussed in detail. The main item of concern
are the DAB details at the modulation end. COFDM is a multicarrier
transmission method in which, in the case of DAB, between 192 and 1536
carriers are combined to form one symbol. Due to DQPSK, each carrier
can carry 2 bits in DAB. A symbol is the superposition of all these individual carriers. A guard interval with a length of about 1/4 of the symbol
length is added to the symbol which has a length of between 125 s and
1 ms. In the guard interval, the end of the following symbol is repeated
526
26 Digital Audio Broadcasting - DAB
where echoes due to multipath reception can "wear themselves down".
This prevents intersymbol interference as long as a maximum echo interval
is not exceeded.
IFFT
bandwidth
Channel
bandwidth
Central carrier
= 0 (set ot zero)
Signal
bandwidth
1.536 MHz
Subcarrier
spacing
1 / 4 / 8 / 2 kHz
Carrier # -k/2
Carrier # +k/2
-768/-192/-96/768/192/96/384
384
No. of carrier: 1535 /
383 / 191 / 767
Fig. 26.9. DAB spectrum
Instead of one carrier, COFDM involves hundreds to thousands of subcarriers in one channel (Fig. 26.8.). The carriers are equidistant from one
another. All carriers in DAB are /4-shift DQPSK (Differential Quadrature
Phase Shift Keying) modulated. The bandwidth of a DAB signal is 1.536
MHz, the channel bandwidth available, e.g. in VHF Band 12 (223 ... 230
MHz) is 1.75 MHz which corresponds to exactly 1/4 of a 7 MHz channel.
Firstly, however, let us turn to the principle of differential QPSK: The
vector can take up four positions, which are 45, 135, 225 and 315 degrees.
However, the vector is not mapped in absolute values but differentially.
I.e., the information is contained in the difference between one symbol and
the next. The advantage of this type of modulation lies in the fact that no
channel correction is necessary. It is also irrelevant how the receiver is locked in phase, the decoding will always operate correctly. There is also a
disadvantage, however: the arrangement requires a signal to noise ratio
which is better by about 3 dB than in the case of absolute mapping (coherent modulation) since in the case of an errored symbol, the difference with
respect to the preceding symbol and the following symbol is false and will
lead to bit errors. Any interference event will then cause 2 bit errors.
26.3 The Physical Layer of DAB
527
In reality, however, DAB does not use DQPSK but /4-shift DQPSK,
which will be discussed in detail later. Many references wrongly mention
only DQPSK in DAB. If the DAB standard is analyzed in detail, however,
and especially the COFDM frame structure, this special type of DQPSK is
encountered automatically via the phase reference symbol (TFPR).
Fig. 26.10. Real DAB spectrum after the mask filter
COFDM signals are generated with the aid of an Inverse Fast Fourier
Transform (IFFT) (s. COFDM chapter) which requires a number of carriers corresponding to a power of two. In the case of DAB, either a 2048point IFFT, a 512-point IFFT, a 256-point IFFT or a 1024-point IFFT is
performed. The cumulative IFFT bandwidth of all these carriers is greater
than the channel bandwidth but the edge carriers are not used and are set to
zero (guard band), making the actual bandwidth of DAB 1.536 MHz. The
channel bandwidth is 1.75 MHz. The subcarrier spacing is 1, 4, 8 or 2 kHz
depending on DAB mode (Mode I, II, II or IV) (see Fig. 26.8. and 26.9.).
Fig. 26.10. shows a real DAB spectrum as it would be measured with a
spectrum analyzer at the transmitter output after the mask filter. The width
of the spectrum is 1.536 MHz. There are also signal components which extend into the adjacent channels, the relevant terms being shoulders and
shoulder attenuation. The shoulders are lowered by using mask filters.
In DAB, a COFDM frame (Fig. 26.11.) consists of 77 COFDM symbols. The length of a COFDM symbol depends on the DAB mode and is
between 125 s and 1 ms, to which the guard interval is added which is
528
26 Digital Audio Broadcasting - DAB
1
Data symbols (DQPSK) (2.4 Mbit/s)
2
3
73 74 75 76
TFPR
76
Phase ref.
Null symbol
about 1/4 of the symbol length. The total length of a symbol is thus between about 156 s and 1.246 ms. Symbol No. 0 is the so-called null symbol. During this time, the RF carrier is completely gated off. The null symbol starts the DAB frame and is followed by the time frequency phase
reference (TFPR) used for frequency and phase synchronisation in the receiver. It does not contain any data.
Guard
FIC (96 kbit/s)
Symbol
(152)
MSC (2.304 Mbit/s)
FIC = Fast Information Channel
MSC = Main Service Channel
Frame
(24, 48, 96 ms)
Fig. 26.11. DAB frame
All COFDM carriers are set to defined amplitude and phase values in
the phase reference symbol. The actual data transmission starts with the
second symbol. In contrast to DVB, the data stream in DAB is completely
synchronous with the COFDM frame. In the first symbols of the DAB
frame, the Fast Information Channel (FIC) is transmitted, the length of
which is dependent on the DAB mode. The data rate of the FIC is
96 kbit/s. In the FIC, important information for the DAB receiver is transmitted. Following the FIC, the transmission of the Main Service Channel
(MSC) starts in which the actual payload data are found. The data rate of
the MSC is a constant 2.304 Mbit/s and is mode-independent. Both FIC
and MSC additionally contain FEC gated in by the DAB COFDM modulator. The FEC in DAB is very flexible and can be configured differently for
the various subchannels, resulting in net data rates of (0.8) 1.2 to 1.73
Mbit/s for the actual payload (audio and data). The type of modulation
used in DAB is differential QPSK. The aggregate gross data rate of FIC
and MSC is 2.4 Mbit/s. The length of a DAB frame is between 24 and
96 ms (mode-dependently).
26.3 The Physical Layer of DAB
529
In the further description, the details of the COFDM implementation in
DAB will be discussed in greater detail. In DAB, a COFDM frame starts
with a null symbol. All carriers are simply set to zero in this symbol.
However, Fig. 26.12. only shows a single carrier over a number of symbols. The first symbol shown at the left-hand edge of the picture is the null
symbol where the vector has the amplitude zero. This is followed by the
phase reference symbol to which the phase of the first data symbol (symbol no. 2) i referred. The difference between the phase reference symbol
and symbol no. 2 and, continuing from there, the difference between two
adjacent symbols provides the coded bits. I.e., the information is contained
in the phase change.
-90o
Null
symbol
-90o
+180o
+90o
+180o
0o
Phase Symbol Symbol Symbol Symbol Symbol Symbol
ref.
2
3
4
5
6
7
symbol
Fig. 26.12. DQPSK sequence with null symbol and phase reference symbol
The principle shown in Fig. 26.12. does still not correspond to the precise reality in DAB which, however, we are approaching step by step.
Fig. 26.12. shows the mapping and the state transitions in the case of
simple QPSK or simple DQPSK. It can be seen clearly that phase shifts of
+/-90 degrees and +/-180 degrees are possible. In the case of +/-180 degree
phase shifts, however, the voltage curve passes through zero which leads
to the envelope curve being pinched in. In single carrier methods it is
usual, therefore, to carry out so-called /4-shift DQPSK instead of
DQPSK, thus avoiding this problem. In this type of modulation, the carrier
phase is shifted by 45 degrees from phase to phase, i.e. by /4. The receiver is informed about this and cancels out this process. An example of
/4-shift DQPSK is the TETRA mobile radio standard. In DAB, too, this
530
26 Digital Audio Broadcasting - DAB
modulation method was adopted, but in this case in conjunction with the
COFDM multicarrier method.
Q
I
Fig. 26.13. Mapping of an “normal” QPSK or a “normal” DQPSK, with state transitions which also pass trough the zero point
I
I
I
Symbol
i, i+2,i+4,…
Q
Q
Q
/4-shift
DQPSK
Symbol
i+1 i+3,i+5,…
Fig. 26.14. Transition from DQPSK to π/4-shift DQPSK
Considering now the transition from DQPSK to /4-shift DQPSK (Fig.
26.14.). On the left, the constellation pattern of simple QPSK is shown.
On the right, QPSK rotated by 45 degrees, i.e. by /4, is shown. /4-shift
DQPSK is composed of both. The carrier phase is shifted on by 45 degrees
from symbol to symbol. If only 2 bits per vector transition are to be represented, the 180-degree phase shifts can be avoided. It can be shown that
phase shifts of +/-45 degrees (+/- /4) and +/-135 degrees (+/-3/4 ) are suf-
26.3 The Physical Layer of DAB
531
ficient for transmitting 2 bits per symbol difference by differential mapping. The constellation pattern of /4-shift DQPSK (Fig. 26.14., center)
shows the state transitions used. It can be seen that there is no 180-degree
shift.
In DAB, /4-shift DQPSK is used in conjunction with COFDM. The
COFDM frame starts with the null symbol in DAB. During this time, all
carriers are set to zero, i.e. u(t) = 0 for the period of a COFDM symbol.
This is followed by the phase reference symbol, or more precisely by the
time frequency phase reference (TFPR) symbol where all carriers are
mapped onto n•90 degrees corresponding to the so-called CAZAC (Constant Amplitude Zero Autocorrelation) sequence. This means that the carriers are mapped onto the I or Q axis differently for each carrier according
to a particular pattern, i.e. assume the phase space of 0, 90, 180, 270 degrees. The phase reference symbol is the reference for the /4-shift
DQPSK of the first data symbol, i.e. symbol no. 2. The carriers in symbol
No. 2 thus accupy the phase space of n•45 degrees. Symbol no. 3 gets its
phase reference from symbol no. 2 and occupies the phase space of n•90
degrees etc. The same applies to all other carriers.
Q
Q
00
01
01
00
I
I
10
11
DQPSK
I
10
11
/4-shift-DQPSK
Fig. 26.15. Constellation pattern of DQPSK compared with π/4-shift DQPSK
Fig. 26.15. shows the comparison of a DQPSK with a /4-shift DQPSK.
The selected mapping rule has been selected arbitrarily here and could easily be selected differently.
If it is intended to transmit the bit combination 00 by using the DQPSK
in the example, the phase angle will not change. Bit combination 01 is signalled by a +45 degrees phase shift, bit combination 11 corresponds to a
–45 degrees phase shift. A 10, in turn, corresponds to 180 degrees phase
shift.
532
26 Digital Audio Broadcasting - DAB
In the right-hand drawing of Fig. 26.15., the state transitions of a /4shift DQPSK are shown with phase shifts of +/-45 degrees and +/-135 degrees. The carrier never dwells on a constant phase, neither are there any
180-degree phase shifts.
Null
symbol
without
TII
Null
symbol
with
TII
Null
symbol
TII
Time
frequ.
phase
ref.
Time
frequ.
phase
ref.
Fig. 26.16. Null symbol with and without TII
without TII
TII
without TII
Fig. 26.17. Oscillogram of a DAB frame with null symbols (only each second
null symbol includes TII = Transmitter Indentification Information)
26.3 The Physical Layer of DAB
533
The null symbol is the very first symbol of a DAB frame, called symbol
no. 0 in numerical order. During this time, the amplitude of the COFDM
signal is zero. The length of a null symbol corresponds approximately to
the length of a normal symbol plus guard interval. In reality, however, it is
slightly longer because it is used for adjusting the DAB frame length to
exactly 14, 48 or 96 ms to match the frame length of the MPEG-1 or –2
audio layer II. The null symbol marks the beginning of a DAB COFDM
frame. It is the first symbol of this frame and can be easily recognized
since all carriers are zeroed during this time (Fig. 26.16., Fig. 26.17., Fig.
26.19. and Fig. 26.20.). It is thus used for roughly synchronizing the receiver timing. During the null symbol, a transmitter ID, a so-called TII
(transmitter identification information) (Fig. 26.16. and Fig. 26.17.), can
also be transmitted. In the case of a TII, certain carrier pairs in the null
symbol are set and can be used for signalling the transmitter ID (Fig.
26.18.).
Fig. 26.18. FFT of a null symbol with TII; carrier pairs are set for signalling the
TII Main ID and TII Sub-ID
The frame lengths, the symbol lengths and thus also the zero symbol
lengths depend on the DAB mode and are listed in Table 26.2.
The phase reference symbol or TFPR (Time Frequency Phase Reference) symbol is the symbol following directly after the null symbol.
Within this symbol, all carriers are set to certain fixed phase positions ac-
534
26 Digital Audio Broadcasting - DAB
cording to the CAZAC (Constant Amplitude Zero Autocorrelation) sequence. This symbol is used for receiver AFC (automatic frequency control), on the one hand, and, on the other hand as starting phase reference
for the /4-shift DQPSK.
Fig. 26.19. Spectrum of DAB signal with the null symbol running through
30 ms / Div
Fig. 26.20. Spectrum of a DAB signal with zero span; the DAB frame with the
null symbol can easily be seen (Mode I)
Phase ref.
26.3 The Physical Layer of DAB
535
ETI from ensemble MUX
FIC
SC1 SC2
SC3
SC4
…
SCn
Header
MST (Main Stream Data)
TFPR
1
+FEC
+FEC
SC1
SC2
SC3
SC4
…
SCn
FIC (96 kbit/s)
MSC (2.304 Mbit/s)
SC = subchannel
net data rate
= n * 8 kbit/s
FIC = Fast Information Channel
MSC = Main Service Channel
Frame
(24, 48, 96 ms)
Fig. 26.21. DAB frame
The receiver can also use this symbol for calculating the impulse response of the channel in order to carry out accurate time synchronisation,
among other things for positioning the FFT sampling window in the receiver. The impulse response allows the individual echo paths to be identified. During the TFPR symbol, the carriers are set to 0, 90, 180 or 270 degrees, differently for each carrier. The relevant rule is defined in tables in
the standard (CAZAC Sequence).
Returning now to the DAB data signal, the gross data rate of a DAB
channel is 2.4 Mbit/s. Subtracting the FIC (Fast Information Channel)
which is used for receiver configuration, and the error protection (convolutional coding), a net data rate of (0.8) 1.2 … 1.73 Mbit/s is obtained. In
contrast to DVB, DAB operates completely synchronously. Whereas in
DVB-T, no COFDM frame structure can be recognized in the data signal,
the MPEG-2 transport stream, the DAB data signal also consists of frames.
A DAB COFDM frame (Fig. 26.21.) begins with a null symbol.
During this time, the RF signal is zeroed. This is followed by the reference symbol. There is no data transmission during the time of the null
symbol and the reference symbol. Data transmission starts with COFDM
symbol no. 2 with the transmission of the FIC (Fast Information Channel),
536
26 Digital Audio Broadcasting - DAB
followed by the MSC, the Main Service Channel. The FIC and MSC already contain error protection (FEC) inserted by the modulator. The error
protection used in the FIC is equal and that used in the MSC is unequal.
Equal error protection means that all data are provided with equal error
protection, unequal error protection means that more important data are
protected better than unimportant ones. The data rate of the FIC is
96 kbit/s, that of the MSC is 2.304 Mbit/s. Together, a gross data rate of
2.4 Mbit/s is obtained. A DAB frame is 77 COFDM symbols long in mode
I, II, IV and 153 COFDM symbols long in mode III. The frame consists of
1536•2•76 bits = 233472 bits in DAB mode I, of 384•2•76 bits = 58638
bits in mode II, 152•2•151 bits = 57984 bits in mode III and 768•76 bits =
116736 bits in mode IV.
STI
Audio
Enc.
STI
Audio
Info
Enc.
STI
Audio
Info
Enc.
Audio
Info
Audio
Enc.
Ensemble
multiplexer
Info
ETI
DAB
mod.
RF
COFDM
STI
Enc.
Info
ETI = Ensemble
Transport Interface
STI = Service
Transport Interface
Fig. 26.22. DAB feed via ETI
The DAB data are fed from the ensemble multiplexer to the DAB modulator and transmitter via a data signal called ETI (Ensemble Transport Interface) (Fig. 27.22.). The data rate of the ETI signal is lower than that of
the DAB frame since it does not yet contain error protection. Error protection is only added in the modulator (convolutional coding and interleaving). However, the ETI signal already contains the frame structure of DAB
(Fig. 26.21.). An ETI frame starts with a header. This is followed by the
data of the Fast Information Channel (FIC). After that comes the mainstream (MST). The mainstream is sub divided into subchannels. Up to 64
subchannels are possible. The information about the structure of the main-
26.4 DAB – Forward Error Correction
537
stream and the error protection to be added in the modulator is found in the
Fast Information Channel (FIC). The FIC is intended for automatic configuration of the receiver.
The modulator obtains its information for the composition and configuration of the multiplexed data stream from the ETI header however.
Reset after 24 ms frame
P(x)=x9+x5+1;
1
1
1
1
1
1
1
1
1
1 2
3
4
5
6
7
8
9
+
EXOR
+
Data in
Scrambled data out
EXOR
Fig. 26.23. Data scrambling
Out
Puncturing
In
+
+
+
+
+
+
+
+
Coderate
+
+
+
Coderate = in / out;
8/9, 8/10, 8/11, …8/32
Fig. 26.24. DAB convolutional coding with puncturing
26.4 DAB – Forward Error Correction
In this section, error protection, the Forward Error Correction (FEC) used
in DAB will be discussed in greater detail.
538
26 Digital Audio Broadcasting - DAB
In DAB, all subchannels are error protected individually and to different
degrees (Fig. 26.5. and 26.6.). Up to 64 subchannels are possible. Error
protection (FEC) is carried out in the DAB modulator.
Before the data stream is provided with error protection, it is scrambled
(Fig. 26.23.). This is done by mixing with a pseudo random binary sequence (PRBS). The PRBS is generated with the aid of a shift register with
feedback. The data stream is then mixed with this PRBS by using an exclusive-OR gate. This breaks up long sequences of ones and zeroes which
maybe present in the data stream. This is called energy dispersal. In singlecarrier methods, energy dispersal is required for preventing the carrier vector from staying at constant positions. This would lead to discrete spectral
lines. But error protection, too, only operates correctly if there is movement in the data signal. This is the reason why this scrambling is carried
out at the beginning of the FEC also in the COFDM method. Every 24 ms,
the shift register arrangement is loaded with all ones and thus reset.
Such an arrangement is also found in the receiver and must be synchronised with the transmitter. Mixing again with the same PRBS in the receiver restores the original data stream.
This is followed by the convolutional coding. The convolutional coder
used in DAB (followed by puncturing) is shown in Fig. 26.24. The data
signal passes through a 6-stage shift register. In parallel to this, it is exclusive-OR-ed with the information stored in the shift registers at different delay times in three branches. The shift register content delayed by six clock
cycles and the three data signals manipulated by EXOR operations are serially combined to form a new data stream now having four times the data
rate of the input data rate.
This is called ¼ code rate. The code rate is the ratio of input data rate to
output data rate.
After the convolutional coding, the data stream had been expanded by a
factor of four. However, the output data stream now carries 300% overhead, i.e. error protection. This lowers the available net data rate. This
overhead, and thus the error protection, can be controlled in the puncturing
unit. The data rate can be lowered again by selectively omitting bits.
Omitting, i.e. puncturing, is done in accordance with a scheme known to
the transmitter and the receiver: a puncturing scheme. The code rate describes the puncturing and thus provides a measure for the error protection.
The code rate is simply calculated from the ratio of input data rate to output data rate. In DAB, it can be varied between 8/9, 8/10, 8/11…8/32.
8/32 provides the best error protection at the lowest net data rate, 8/9 provides the lowest error protection at the highest net data rate. Various data
contents are protected to a different degree in DAB. Frequently, however,
burst errors occur during a transmission. If the burst errors last longer, the
26.4 DAB – Forward Error Correction
539
error protection will fail. For this reason, the data are interleaved in a further operating step, i.e. distributed over a certain period of time. Long interleaving over 384 ms makes the system very robust and suitable for mobile use. During the de-interleaving at the receiving end, any burst errors
which maybe present are then broken up and distributed more widely in
the data stream. It is now easier to repair these burst errors, which have become single errors, and this without any additional data overhead. In DAB
there are two types of error protection being used, namely equal error protection and unequal error protection.
Equal error protection means that all components are provided with the
same FEC overhead. This applies to the Fast Information Channel (FIC)
and to the case of pure data transmission.
Audio contents, i.e. the components of an MPEG-1 or -2 audio frame
carry unequal protection. Some components in the audio frame are more
important because bit errors would cause greater disruption there and these
parts are protected more therefore. These different components in the audio frame are provided with different code rates.
In many transmission methods, constant equal error protection is used.
An example of this is DVB. In DAB, only parts of the information to be
transmitted are provided with equal error protection. This includes the following data: the FIC is protected equally with a mean code rate of 1/3. The
data of the packet mode can be provided with a code rate of 2/8, 3/8, 4/8 or
6/8.
PI = Puncturing Index
24
PAD
Puncturing index PI1, PI2, PI3, PI4 depending
on protection level (1,2,3,4,5) and audio bitrate
(ETS 300401, page 158)
protection level 1 = highest protection,
5 = lowest protection
PI1
1
Scale factors
Header
PI2
PI4
PI3
Subband samples
DAB audio frame
Coderate = 8/(8+PI); PI = 1…24;
Fig. 26.25. Unequal forward error correction of a DAB audio frame
time
540
26 Digital Audio Broadcasting - DAB
The MPEG audio packets are protected with unequal error protection
which is also controllable in DAB. Some components of the MPEG audio
packet are more sensitive to bit errors than other ones.
The components in the DAB audio frame which are provided with different error protection are:
•
•
•
•
Header
Scale factors
Subband samples
Program-associated data (PAD)
The header must be protected particularly well. If errors occur in the
header, this will lead to serious synchronization problems. The scale factors must also be well protected since bit errors in this area would make for
very unpleasant listening. The subband samples are less sensitive and their
error protection is correspondingly lower.
Fig. 26.25. shows an example of the unequal error protection within a
DAB audio frame. The puncturing index describes the quality of the error
protection. From the puncturing index, the code rate in the relevant section
can be easily calculated using the following formula:
code_rate = 8/(8+PI);
where PI = 1 … 24, the puncturing index.
The puncturing index, in turn, is obtained from the protection level,
which is in the range 1, 2, 3, 4 or 5, and the audio bit rate. Table 26.3. lists
the mean code rates as a function of protection level and the audio bit
rates. PL1 offers the highest error protection and PL5 offers the lowest error protection.
Table 26.3. DAB protection levels and mean code rates
Audio
bitrate
[kbit/s]
32
48
56
64
80
96
Mean code
rate
protection
level 1
0.34
0.35
X
0.34
0.36
0.35
Mean code
rate
protection
level 2
0.41
0.43
0.40
0.41
0.43
0.43
Mean code
rate
protection
level 3
0.50
0.51
0.50
0.50
0.52
0.51
Mean code
rate
protection
level 4
0.57
0.62
0.60
0.57
0.58
0.62
Mean
code rate
protection
level 5
0.75
0.75
0.72
0.75
0.75
0.75
26.4 DAB – Forward Error Correction
112
128
160
192
224
256
320
384
X
0.34
0.36
0.35
0.36
0.34
X
0.35
0.40
0.41
0.43
0.43
0.40
0.41
0.43
X
0.50
0.50
0.52
0.51
0.50
0.50
X
0.51
0.60
0.57
0.58
0.62
0.60
0.57
0.58
X
541
0.72
0.75
0.75
0.75
0.72
0.75
0.75
0.75
Table 26.4. shows the minimum signal/noise ratio S/N needed and the
number of programs which can be accommodated in a multiplexed DAB
data stream on the basis of a data rate of 192 kbit/s per program, in dependence on the protection level. If, e.g. PL3 is used, 6 programs of 196
kbit/s each can be accommodated in a multiplexed DAB data stream and
the minimum signal/noise ratio then needed is 11 dB. The gross data rate
of the DAB signal (including error protection) is 2.4 Mbit/s and the net
data rate is between (0.8) 1.2 and 1.7 Mbit/s depending on the error protection selected.
Table 26.4. DAB channel capacity and minimum S/N
Protection level (FEC)
PL1 (highest)
PL2
PL3
PL4
PL5 (lowest)
No. of programs at 196 S/N [dB]
kbit/s
4
7.4
5
9.0
6
11.0
7
12.7
8
16.5
Table 26.5. DAB parameters and quality
Program
Format
type
music/voice mono
music/voice 2-channel
stereo
music/voice multichannel
voice
mono
news
mono
data
--
Quality
broadcast
broadcast
broadcast
acceptable
intelligible
--
Sampling
rate [kHz]
48
48
Protection
level
PL2 oder 3
PL2 oder 3
Bitrate
[kbit/s]
112...160
128...224
48
24 oder 48
24 oder 48
PL2 oder 3
PL3
PL4
PL4
384...640
64...112
32 or 64
32 or 64
The unequal error protection in DAB has the effect that the DAB receivability does not abruptly break off when the signal drops below a certain minimum S/N ratio. At first, audible disturbances arise and receivabil-
542
26 Digital Audio Broadcasting - DAB
ity ceases only about 2 dB later. Table 26.5. shows frequently selected protection levels and audio rates in DAB [HOEG_LAUTERBACH].
GPS
ETI
Input
interface
Frame
gen.
Delay
comp.
FEC
Freq.
interleaver
Different.
mapper
IFFT
Guard
interval
insertion
FIR
Precorrection
IQ mod.
IF/RF up.
Fig. 26.26. Block diagram of a DAB modulator and transmitter
26.5 DAB Modulator and Transmitter
Let us now consider the overall block diagram of a DAB modulator (Fig.
26.26.) and transmitter. The ETI (Ensemble Transport Interface) is present
at the input interface where the modulator synchronizes itself to the ETI
signal. In the case of a single-frequency network, delay compensation is
carried out in the modulator controlled via the TIST (Time Stamp) in the
ETI signal. This is followed by the error protection (FEC) which is different for each signal content.
The error-protected data stream is then frequency interleaved, i.e. distributed. Each COFDM carrier is assigned a part of the data stream which
is always 2 bits per carrier in DAB. In the differential mapper the real-and
imaginary-part table is then formed, i.e. the current vector position is determined for each carrier. Following this the DAB frame with null symbol,
TFPR symbol and data symbols is formed and the completed real-and
imaginary-part tables are then supplied to the IFFT, the Inverse Fast Fourier Transform. After that, we are back in the time domain where the guard
interval is added to the symbol by repeating the end of the symbol following.
After FIR filtering, pre-correction is carried out in the power transmitter
for compensating for the non-linearities of amount and phase of the ampli-
26.5 DAB Modulator and Transmitter
543
fier characteristic. The IQ modulator following is then usually the IF/RF
up converter at the same time. Today, direct modulation is normally used,
i.e. direct conversion from baseband up into RF. This is followed by power
amplification in transistor output stages. The remaining non-linearities and
the necessary clipping of voltage peaks to about 13 dB result in the socalled shoulders of the DAB signal. These are out-off-band components
which would interfere with the adjacent channels.
216.928 218.640 220.352 222.064 223.936 225.648 227.360 229.072
MHz
MHz
MHz
MHz
MHz
MHz
MHz
MHz
A
B
C
D
A
C
Channel 12
Channel 11
216
MHz
B
223
MHz
D
f
[MHz]
230
MHz
Band III: 174 – 240 MHz
L Band: 1452 – 1492 MHz
Fig. 26.27. DAB channel allocation with channel 11 and 12 as example
Table 26.6. DAB channel table band III VHF
Channel
Center frequency [MHz]
5A
5B
5C
5D
6A
6B
6C
6D
7A
7B
7C
7D
8A
8B
8C
8D
9A
9B
9C
9D
174.928
176.640
178.352
180.064
181.936
183.648
185.360
187.072
188.928
190.640
192.352
194.064
195.936
197.648
199.360
201.072
202.928
204.640
206.352
208.064
544
26 Digital Audio Broadcasting - DAB
10A
10N
10B
10C
10D
11A
11N
11B
11C
11D
12A
12N
12B
12C
12D
13A
13B
13C
13D
13E
13F
209.936
210.096
211.648
213.360
215.072
216.928
217.088
218.640
220.352
222.064
223.936
224.096
225.648
227.360
229.072
230.784
232.496
234.208
235.776
237.488
239.200
Table 26.7. DAB channel table L band
Channel
LA
LB
LC
LD
LF
LG
LH
LI
LJ
LK
LL
LM
LN
LO
LP
LQ
LR
LS
LT
LU
LV
LW
Center frequency [MHz]
1452.960
1454.672
1456.384
1458.096
1461.520
1463.232
1464.944
1466.656
1468.368
1470.080
1471.792
1473.504
1475.216
1476.928
1478.640
1480.352
1482.064
1483.776
1485.488
1487.200
1488.912
1490.624
Table 26.8. DAB channel table L band, Canada
Channel
1
2
3
4
5
Center frequency [MHz]
1452.816
1454.560
1456.304
1458.048
1459.792
26.5 DAB Modulator and Transmitter
6
7
8
9
10
11
12
13
14
15
16
17
18
19
20
21
22
23
545
1461.536
1463.280
1465.024
1466.768
1468.512
1470.256
1472.000
1473.744
1475.488
1477.232
1478.976
1480.720
1482.464
1484.464
1485.952
1487.696
1489.440
1491.184
32…384 kbit/s
STI
32…384 kbit/s
n * 8 kbit/s
Data 1
PAD 1
Audio 1
H
SC
Config 1
SC FIG
Service c
Service b
Service a
Audio 2
...
H
32…384 kbit/s
Config 2
SC
SC FIG FIG
Up to 64
subchannels
each n * 8 kbit/s
FIC
ETI
SC1 SC2
Audio 3
SC3
SC4
Config 3
H = Header
SC = Subchannel
FIG = Fast
Information Group
PAD = Program
Associated Data
FIC = Fast
Information
Channel
…
SCn
Header
MST (Main Stream Data)
Fig. 26.28. Composition of the ETI data stream
For this reason there is another passive bandpass filter (mask filter).
Without pre-correction a DAB signal would have a shoulder attenuation of
about 30 dB. If the pre-correction has been properly set the shoulder attenuation will be about 40 dB. This would still interfere with the adjacent
channels and would not be authorised by the Authorities. Following the
mask filter the shoulders are then lowered by another 10 dB.
Fig. 26.27. shows frequently used DAB blocks. A VHF channel (7 MHz
bandwidth) is divided into 4 DAB blocks. The blocks are then called e.g.
block 12A, 12B, 12C or 12D.
546
26 Digital Audio Broadcasting - DAB
Tables 26.6., 26.7. and 26.8. list the channel tables used in DAB. Each
DAB channel has a width of 7/4 MHz = 1.75 MHz. However, the COFDM
signal bandwidth is only 1.536 MHz and there is thus a guard band for the
adjacent channels.
26.6 DAB Data Structure
In the section following the essential features of the data structure of DAB
will be explained. In DAB, a number of MPEG-1 or –2 Audio Layer II
coded audio signals (MUSICAM) combined to form an ensemble (Fig.
26.28.) are transmitted in a 1.75 MHz-wide DAB channel. The maximum
net data rate of the DAB channel is about 1.7 Mbit/s and the gross data rate
is 2.4 Mbit/s. The data rate of an audio channel is between 32 and 384
kbit/s.
The details described in the following section can be found in the
[ETS300401] (DAB), [ETS300799] (ETI) and [ETS300797] (STI) standards.
DAB data signal
Fast Information
Channel FIC
Multiplex
Configuration
Information
MCI
Fast Information
Data Channel
Main Service
Channel MSC
Up to 64
Subchannels
Audio
StreamMode
Data
StreamMode
Packet
Mode
Fig. 26.29. DAB data structure
A DAB data signal (ETI) is composed of the Fast Information Channel
(FIC) and the Main Service channel (MSC) (Fig. 26.29.). In the Fast Information Channel, the modulator and the receiver are informed about the
composition of the multiplexed data stream by means of the Multiplex
26.6 DAB Data Structure
547
Configuration Information (MCI). The Main Service Channel contains up
to 64 subchannels with a data rate of n•8 kbit/s each (Fig. 26.30.). In the
subchannels audio signals and data are transmitted. The modulator and receiver obtain the information about the composition of the Main Service
Channel from the Multiplex Configuration Information (MCI).
FIC
ETI
SC1 SC2
SC3
SC4
…
Header
MST (Main Stream Data)
...
Packet
header
with
packet ID
24/48/72/92 byte long packets
n * 8 kbit/s
Subchannel packet
Fig. 26.30. DAB data structure in packet mode
Non-time
interleaved
Time interleaved
SC1 SC2
SC3
SC4
…
SCn
FIC
MSC (Main Service Channel)
FIB
…
FIB
FIB=Fast Information Block
FIC=Fast Information Channel
MCI
Multiplex
Configuration
Information
SI
Service
Information
FIDC
Fast
Information
Data
Channel
Fig. 26.31. Structure of the DAB Fast Information Channel (FIC)
SCn
548
26 Digital Audio Broadcasting - DAB
The transmission in the subchannels can be carried out in Stream Mode
and in packet mode. In Stream Mode, data are transmitted continuously. In
Packet Mode, the subchannel is additionally sub divided into sub-packets
with a constant length. Audio is always transmitted in Stream Mode. The
data structure is here pre-determined by the audio coding (24/48 ms pattern). Data can be transmitted in Packet Mode (e.g. MOT - Multimedia
Object Transfer) or in Stream Mode (e.g. T-DMB). In Packet Mode, the
most varied data streams can be transmitted within a subchannel.
FIC
FIB
…
FIC=Fast Information Channel
FIB
FIB=Fast Information Block
FIB data field
FIG n FIG m
…
CRC
FIB =
256 bit with
16 bit CRC
Padding
FIG=Fast Information Group
FIG type Length
FIG data field
Fig. 26.32. Structure of the DAB FIC
In Stream Mode, a subchannel is used completely for a continuous data
stream. This is the case e.g. during audio transmission. Data can also be
transmitted in stream mode. This is the case e.g. in the T-DMB method
(South Korea). In Packet Mode, a subchannel is additionally sub divided
into packets of a constant length of 24, 48, 72 or 92 bytes. A packet begins
with a 5-byte-long packet header which contains the packet ID, among
other things. The packet ID can be used for identifying the contents. A
packet ends with a CRC checksum. This provides for flexible use of the
subchannel. It is possible to embed different data services and to provide
variable data rates.
In the following section, the structure and content of the Fast Information Channel (FIC) (Fig. 26.31. and 26.32.) and of the Main Service Channel (MSC) will be considered in greater detail. The information transmitted
in the fast information channel and main service channel come from the
Main Stream Data (MST) from the Ensemble Transport Interface (ETI).
26.6 DAB Data Structure
549
FIC and MSC are provided with error protection (FEC) in the modulator,
the FIC being given the strongest protection. The error protection in the
MSC is configurable. The strength of the error protection in the MSC is
signalled to the receiver in the FIC.
Non-time
interleaved
Time interleaved
SC1 SC2
FIC
CIF
= 24 ms
SC3
SC4
…
SCn
MSC (Main Service Channel)
CIF
=24 ms
…
CIF=Common
Interleaved Frame
CU=Capacity Unit
CU … CU
…
CU … CU
SC=
Subchannel
SC1
1 CIF = 864 CUs
SCn
1 CU = 64 Byte Æ 1 CIF = 55296 Byte
Fig. 26.33. DAB main service channel
In the MSC, the individual subchannels are transmitted, a total of 64
subchannels being possible. Each subchannel can be error-protected to a
different degree which is also signalled in the FIC. The subchannels are
combined or more precisely allocated to services.
The Fast Information Channel is not time interleaved but transmitted error protected in so-called Fast Information Blocks (FIB).
In the FIC the Multiplex Configuration Information (MCI) is transmitted which is information about the composition of the multiplex data
stream, as well as the Service Information (SI) and the Fast Information
Data Channel (FIDC).
The SI transmits information about the programs transmitted, the services. In the FIDC, fast supplementary multi-program data are transmitted.
The Fast Information Channel (FIC) is composed of Fast Information
Blocks with a length of 256 bits. An FIB consists of an FIB data field and
a 16-bit-wide CRC checksum. In the data area of the FIB the messages are
transmitted in so-called Fast Information Groups (FIG). Each FIG is iden-
550
26 Digital Audio Broadcasting - DAB
tified by its FIG type. An FIG is composed of the FIG type, of the length
and the FIG data field in which the actual messages are transmitted.
ETI Frame
ETI
Header
FC
Frame
characterization
-Frame count
-FIC flag
-Number of
streams
-frame phase
-DAB mode
-frame length
STC
(24, 48, 96 ms)
Main Stream
Data
EOH
Stream
End
characteri- of
zation
header
-subCh-Id
-Start address
in MSC
-type and
protection
level
-stream length
-MNSC
-CRC
MST
End
of
Frame
Time
Stamp
EOF TIST
-FIC data (if FIC
flag set) per
stream
-MSC sub-channel
stream data
-CRC
- time
stamp
Fig. 26.34. DAB ETI frame structure
GPS
1pps
pulse
1s
Time stamp
...
ETI
header
Main Stream Data
TIST
...
Frame (24, 48, 96 ms)
Fig. 26.35. Synchronization of DAB modulators via the TIST in the ETI frame
In the main service channel (Fig. 26.33.), the individual subchannels are
broadcast. A total of 64 subchannels are possible. Each subchannel has a
data rate of n•8 kbit/s. The subchannels are associated with services (pro-
26.7 DAB Single-Frequency Networks
551
grams). The MSC is composed of so-called Common Interleaved Frames
(Fig. 26.33.) which have a length of 24 ms and consist of Capacity Units
(CU) with a length of 64 bytes. Overall, 864 CUs result in one CIF which
then has a length of 55296 bytes. A number of CUs make up one subchannel in which the audio frames or data are transmitted.
An ETI frame (Fig. 26.34.) is composed of the header, the Main Stream
Data (MST), and End of Frame (EOF) and the Time Stamp (TIST). An
ETI frame has a length of 24, 48 or 96 ms.
26.7 DAB Single-Frequency Networks
In the further text, DAB single-frequency networks (SFN) and their synchronisation will be discussed.
COFDM is optimally suited to single-frequency operation. In singlefrequency operation, all transmitters are operating at the same frequency
which is why single-frequency operation is very economical with regard to
frequencies. All transmitters are broadcasting an absolutely identical signal
and must operate completely synchronously for this reason. Signals from
adjacent transmitters look to a DAB receiver as if they were simply echoes.
The condition which can be met most simply is the frequency synchronisation because frequency accuracy and stability already had to meet high
requirements in analog terrestrial radio. In DAB, the RF of the transmitter
is tied to the best possible reference. Since the signal of the GPS (Global
Positioning System) satellites is available throughout the world, it is used
as reference for synchronizing the transmitting frequency of a DAB singlefrequency network.
The GPS satellites radiate a 1pps signal to which a 10 MHz oscillator is
tied in professional GPS receivers which is used as reference signal for the
DAB transmitters.
However, there is also a strict requirement with regard to the maximum
transmitter spacing. The maximum possible transmitter spacing is a result
of the length of the guard interval and the velocity of light and the associated propagation time. Inter-symbol interference can only be avoided if in
multi-path reception no path has a longer propagation time than the guard
interval length. The question about what would happen if a signal of a
more remote transmitter violating the guard interval is received can be easily answered. Inter-symbol interference is produced which becomes noticeable as disturbing noise in the receiver. Signals from more remote
transmitters must simply be attenuated sufficiently well. The threshold for
552
26 Digital Audio Broadcasting - DAB
virtually error-free operation is set by the same conditions as in the case of
pure noise. It is of particular importance therefore that a single-frequency
network has the correct levels. It is not the maximum transmitting power
which is required at every site but the correct one. Network planning requires topographical information.
With the velocity of light of C=299792458 m/s, a signal delay of
3.336 µs per kilometer transmitter distance is obtained.
The maximum distances between adjacent transmitters possible with
DAB in a single-frequency network are shown in Table 26.9.
Table 26.9. SFN parameters in DAB
Symbol duration
Guard interval
Symbol+guard
Max. transmitter distance
Mode I
1 ms
Mode IV
500 µs
Mode II
250 µs
Mode III
125 µs
246 ms
1246 µs
73.7 km
123 µs
623 µs
36.8 km
62 µs
312 µs
18.4 km
31 µs
15 6µs
9.2 km
In a single-frequency network, all individual transmitters must operate
synchronised with one another. The contributions are supplied by the playout center in which the DAB multiplexer is located, e.g. via satellite, optical fibre or microwave link. It is obvious that due to different path lengths
the ETI signals fed in will carry different delays.
However, in each DAB modulator in a single-frequency network the
same data packets must be processed to form COFDM symbols. Each
modulator must perform all operating steps in complete synchronism with
all other modulators in the network. The same packets, the same bits and
the same bytes must be processed at the same time. At each DAB transmitter site, absolutely identical COFDM symbols must be radiated at the same
time.
The DAB modulation is organized in frames.
To carry out delay compensation in the DAB SFN, Time Stamps (TIST)
(Fig. 26.34.) derived from the GPS signal are added to the ETI signal in
the multiplexer.
At the end of an ETI frame the TIST is transmitted which is derived by
the DAB ensemble multiplexer by GPS reception and is keyed into the ETI
signal. It specifies the time back to the reception of the last GPS 1pps signal (Fig. 26.35.). The time information in the TIST is then compared in the
modulator with the GPS signal also received at the transmitter site and
used for performing a controlled ETI signal delay.
26.8 DAB Data Broadcasting
553
26.8 DAB Data Broadcasting
In the following section, the possibility of data broadcasting in DAB will
be briefly discussed. In DAB data broadcasting (Fig. 26.36.), a distinction
is made between the MOT (Multimedia Object Transfer ) standard as defined in the [ETS301234] Standard, and the IP transmission via DAB. In
both cases, a DAB subchannel is operated in packet mode, i.e. the data
packets to be transmitted are divided into short constant-length packets.
Each of these packets has a packet ID in the header section by means of
which the transmitted content can be identified.
In the Multimedia Object Transfer (MOT) according to [ETS301234], a
distinction is made between file transmission, a slide show and the
“Broadcast Web Page” operation. In file transmission, only files are fed
out cyclically. A slide show can be configured with respect to its display
speed. It is possible to transmit JPEG or GIF files.
In the “Broadcast Web Page”, a directory of HTML pages is cyclically
transmitted and a starting page can be defined. The resolution corresponds
to ¼ VGA.
Fig. 26.37. shows the MOT data structure. The files to be transmitted,
the slide show or the HTML data are transmitted in the payload segment of
an MOT packet. The MOT packet plus header is inserted into the payload
segment of an MSC data group, the MOT header coming first followed by
a CRC checksum. The entire MOT packet is divided into short constantlength packets of the packet mode. These packets are then transmitted in
subchannels.
Data Broadcasting over DAB
MOT (ETS 301 234)
Multimedia Object
Transfer
Files
Slide show
JPG/GIF
Broadcast
Webpage
Fig. 26.36. Data broadcasting over DAB
IP over DAB
Directory with
HTML files with
start page
554
26 Digital Audio Broadcasting - DAB
Files, slide show or broadcast webpage
MOT
header
Header
MOT payload
MSC data group - payload
CRC
Packets
with header
Fig. 26.37. MOT data structure
The category DAB Data Broadcasting should also include T-DMB (Terrestrial Digital Multimedia Broadcasting). In this South Korean method,
DAB is operated in the data stream mode.
26.9 DAB+
One of the more recent developments in audio broadcasting is DAB+. In
DAB+, MPEG-4 AAC Plus is used instead of MPEG-1 or -2 Layer II Audio. The unequal error protection originally provided in DAB is then no
longer possible since it is coupled directly to the MPEG-1 or MPEG-2
Layer II frame. However, it is now possible to accommodate three times as
many services, i.e. programs, per DAB multiplex. Similar to a T-DMB
transmission, there are no changes in the physical layer of DAB. Australia
has decided to adopt DAB+ and is now also setting up these networks.
26.10 DAB Measuring Technology
The DAB measuring technology can be copied directly from the world of
DVB-T. It is necessary both to test DAB receivers and to measure DAB
transmitters. For these purposes, test transmitters are now available which
26.10 DAB Measuring Technology
555
deliver a DAB signal [SFU] and test receivers which are capable of analyzing DAB signals [ETL].
26.10.1 Testing DAB Receivers
In the DAB receiver test, the reality of DAB reception must be simulated
for the DAB receiver. This requires multi-path reception, noise, minimum
receiver input level, interferers etc. as necessary test scenarios. Similar to
DVB-T, the source of these inputs is provided by a corresponding test
transmitter with fading simulator [SFU]. This can also be used for T-DMB
and DAB+ since the physical layer is the same.
Fig. 26.38. Relatively undisturbed differentially demodulated DAB constellation
diagram [ETL]
26.10.2 Measuring the DAB Signal
In DAB, as in DVB-T, the following measurements can be performed on
the DAB signal:
•
•
•
Detecting the bit error ratios,
Measurements on the DAB spectrum,
Constellation analysis.
Due to the unequal error protection, it is more difficult to measure the
bit error ratios. Measuring the bit errors is relatively simple only at the Fast
556
26 Digital Audio Broadcasting - DAB
Information Channel (FIC) since a constant error protection with a code
rate of 1/3 is present there.
In the DAB constellation analysis [ETL], the constellation diagram is
first differentially demodulated and produces 4 points again. The smaller
the appearance of these points in the constellation diagram, the more undisturbed was their transmission (Fig. 26.38.).
Fig. 26.39. DAB constellation diagram with noise
Fig. 26.40. DAB constellation diagram with superimposed sinusoidal interferer
If noise effects are affecting a DAB signal, a DAB constellation diagram
will appear as shown in Fig. 26.39. Similar to DVB-T, phase jitter will result in striation-like distortions of the constellation diagram. Sinusoidal
interferers will generate circular constellation points (Fig. 26.40.). A
wrongly calibrated IQ modulator will produce carrier cross-talk from the
lower DAB sub-band into the upper one and conversely and will lead to a
poorer S/N ratio just as in DVB-T. In DAB, a modulation error ratio
26.10 DAB Measuring Technology
557
(MER) can also be defined (see also the chapter on DVB-T Measuring
Technology). In DAB, too, an MER can also be defined and measured as a
function of the subcarriers (Figs. 26.41. and 26.42.).
A further measurement necessary in DAB is measuring the channel impulse response (Fig. 26.43.). The channel impulse response, which can be
calculated by analyzing the TFPR symbol, can be used for verifying if a
DAB single-frequency network is running synchronously and that there are
no guard interval violations.
Fig. 26.41. MER(f) in undisturbed DAB (Mode I)
Fig. 26.42. MER(f) in DAB with fading
Naturally, the evaluation of the data contents in the DAB signal is also
of interest. The analysis of an ETI signal at the output of the playout center
or at the transmitter input, respectively, corresponds to the MPEG-2 analysis in DVB. By now there are analysis tools available also for this purpose. This delay in the availability of DAB test technology is one of the
558
26 Digital Audio Broadcasting - DAB
consequences of the DAB market delays and does not have any basis in
technical reasons.
Fig. 26.43. DAB channel impulse response (3 paths) [ETL]
Bibliography: [FISCHER7], [HOEG_LAUTERBACH], [ETS300401],
[ETS300799], [ETS300797], [TR101496], [ETS301234], [ETL], [SFU]
27 DVB Data Services: MHP and SSU
Apart from DVB-H, there are also other DVB data services. These are the
Multimedia Home Platform, or MHP in short, and the System Software
Update (SSU) for DVB receivers. In parallel with these, there is also
MHEG (the Multimedia and Hypermedia Information Coding Experts
Group) running over DVB-T in the UK. All these data services have in
common that they are broadcast via so-called object carousels in DSM-CC
sections. Applications are transmitted to the receiver via MHP and MHEG
and can be stored and run by a receiver especially equipped for this purpose. In the case of MHP, these are HTML files and Java applications
transmitted to the terminal in complete directory structures. MHEG allows
HTML and XML files to be transmitted and started.
Video/audio/
data
Data
PSI/SI
tables
asynchronous
or synchronous
data streaming
direct copy of
PTS
data into
payload of
H PES packets
an MPEG-2
transportstream
~64 max.
packet
kByte
=
data piping
Data
DSM-CC
Sections
table_ID
Object
carousel's,
IP over
MPEG
(MPE)
Sections
max.
4 kByte
MPEG-2 transport stream
Fig. 27.1. Data transmission via an MPEG-2 transport stream: data piping, data
streaming and DSM-CC sections
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_27, © Springer-Verlag Berlin Heidelberg 2010
560
27 DVB Data Services: MHP and SSU
27.1 Data Broadcasting in DVB
In MPEG-2/DVB, data transmission can take place as (Fig. 27.1.):
•
•
•
•
•
Data piping
Asynchronous or synchronous data streaming
via object carousels in DSM-CC sections
as datagram transmission in DSM-CC sections
as IP transmission in DSM-CC sections
In data piping, the data to be transmitted are copied directly into the
payload part of MPEG-2 transport stream packets asynchronously to all
other contents and without any other defined intermediate protocol. In data
streaming, in contrast, the familiar PES (packetized elementary stream)
packet structures are used which allow the contents to be synchronized
with one another through the presentation time stamps (PTS). Another
mechanism for asynchronous data transmission, defined in MPEG-2, are
DSM-CC (Digital Storage Media Command and Control) sections (Fig.
27.2.).
table_id (=0x3A …0x3E)
8 Bit
section_syntax_indicator
1
private_indicator=1
1
reserved =11
2
section_length
12
{
table_id_extension
16
reserved
2
version_number
5
current_next_indicator
1
section_number
8
last_section_number
8
switch(table_id)
{
case 0x3A: LLCSNAP(); break;
case 0x3B: userNetworkMessage(); break;
case 0x3C: downloadDataMessage(); break;
case 0x3D: DSMCC_descriptor_list(); break;
case 0x3E: for (i=0; i<dsmcc_section_length-9;i++)
private_data_byte;
}
}
CRC
Fig. 27.2. Structure of a DSM-CC section
8 Bit
32 Bit
27.2 Object Carousels
561
27.2 Object Carousels
DSM-CC sections have already been discussed in detail in the DVB-H section. DSM-CC sections are table-like structures and are considered to be
private sections according to MPEG-2 Systems. The basic structure of a
DSM-CC (Fig. 27.2.) section corresponds to the structure of a so-called
long section with a checksum at the end. A DSM-CC section has a length
of up to 4 kbytes and begins with a table_ID in the range of 0x3A ... 0x3E.
This is followed by the section header with version administration, already
discussed in detail in other chapters. Data services such as object carousels
or general datagrams or IP packets as in DVB-H (MPE, multiprotocol encapsulation) are then transmitted in the actual trunk of the section. The table_ID shows the type of data services involved.
Table_ID’s:
•
0x3A and 0x3C provide for the broadcasting of object/data carousels
0x3D provides for the signalling of stream events
0x3E provides for the transmission of datagrams or IP packets
•
•
DDB
DDB
DII
DDB
DDB
DDB
DDB
DII
Logical
entry
point
DSI
DDB
DDB
DSI = Download Server Initializing
DII = Download Info Identification
DDB = Data Download Block
Transmission
sequency
Fig. 27.3. Principle of an object carousel
562
27 DVB Data Services: MHP and SSU
Object carousels (Fig. 27.3.) allow complete file and directory structures
to be transmitted from a server to the terminal via the MPEG-2 transport
stream. A restriction imposed by the data carousels is that they only allow
a relatively flat directory structure and flat logical structure. Object and
data carousels are described both in the standard [ISO/IEC 13818-6] (a
part of MPEG-2) and in the DVB data broadcasting document
[EN301192].
Firstly, data/object carousels have a logical structure which owes nothing to the content actually to be transmitted (directory tree plus files). The
entry point into the carousel is via the DSI (Download Server Initializing)
message, or via a DII (Download Information Identification) message in
the case of the data carousel. It is retransmitted cyclically with a table_ID=0x3B in a DSM-CC section. Cyclically because this is broadcasting and it must be possible to reach a large number of terminals time and
again and the terminals are unable to request messages from the server.
The DSI packet then uses IDs to refer to one or more DII messages (Fig.
27.4.) which are also retransmitted cyclically in DSM-CC sections with a
table_ID = 0x3B. The DII messages, in turn, refer to modules in which the
actual data are then repeatedly broadcast cyclically via many data
download blocks (DBB) with a table_ID=0x3C in DSM-CC sections.
DSI: Download Server Initializing
gi: Group Info Bytes
DII: Download Info Identification
mi: Module Info Bytes
DDB: Data Download Blocks
PSI/SI
data_broadcast_desc
Super Group
transaction_ID
DSI
gi
gi
transaction_ID
DII
transaction_ID
mi
mi
DDB
DDB
DII
mi
mi
DDB
DDB
DDB
DDB
DDB
DDB
DDB
DDB
DDB
DDB
DDB
Block
Module
Group
Fig. 27.4. Logical structure of an object carousel
27.3 The Multimedia Home Platform MHP
563
The transmission of a directory tree can take up to several minutes depending on the volume of data and the available data rate.
The presence of an object/data carousel must be announced via PSI/SI
tables. Such a data service is allocated to a program service and entered in
the respective program map table (PMT) where the PIDs of the object/data
carousels are to be found. In the case of a data carousel, entry takes place
directly by DII.
Additional items such as a more detailed description of the contents in
the carousels are broadcast in separate, new SI tables like the AIT (Application Information Table) and the UNT (Update Notification Table). The
AIT belongs to the Multimedia Home Platform and the UNT belongs to
the System Software Update and both - AIT and UNT - must also be announced via PSI/SI. The AIT is entered in the PMT of the associated program and the UNT is entered in the NIT.
Java binary code
Java binary code
DSM-CC
PID
DSM-CC
PID
DSM-CC
PID
Obj.
car.
Java binary code
Start file
PMT
HTML
PID
AIT
Start file
Obj.
car.
HTML
Table_ID
=0x74
Start file
PMT = Program Map Table
AIT = Application Identifcation Table
Obj.
car.
Fig. 27.5. MHP structure
27.3 The Multimedia Home Platform MHP
The Multimedia Home Platform has been provided in DVB as supplementary service for MHP-enabled receivers. The standard, with about 1000
pages, is [ETS101812] and has been released in the year 2000. There are
two versions which are MHP 1.1. and MHP 1.2. MHP is used for trans-
564
27 DVB Data Services: MHP and SSU
mitting HTML (Hypertext Multimedia Language) files familiar from the
Internet, and Java applications. Starting the HTML and Java applications
requires special software (or middleware) in the receiver. MHP-capable receivers are more expensive and not available in great numbers on the market. MHP applications are broadcast in many countries but are currently
really successful only in Italy.
Fig. 27.6. MHP file structure of an object carousel as analyzed on an MPEG analyzer [DVM]
The contents broadcast by MHP are:
•
•
•
•
•
Games
Electronic programme guides
News
Interactive program-associated services
"Modern" teletext
The entry point into the MHP directory structure (Fig. 27.5., 27.6.,
27.7.), the starting file and the name and type of the MHP application are
27.4 System Software Update SSU
565
signalled via the AIT (Application Information Table, Fig. 27.5.). The AIT
is entered in a PMT as PID with the value of 0x74 as table_ID.
27.4 System Software Update SSU
Since the software of DVB receivers is also subject to continuous updates,
it makes sense to provide these to the customer in a relatively simple manner. This can be done "by air" in the case of DVB-S and DVB-T and, of
course, via cable in the case of DVB-C. If the software is transmitted in
object carousels embedded in the MPEG-2 transport stream according to
DVB it is called SSU (System Software Update) and is defined in the
[TS102006] standard. Currently, however, mainly proprietary software
updates are used.
In SSU, the available software updates are announced via another table,
the Update Notification Table (UNT). The PID of the UNT is entered in
the NIT and the table_ID of the UNT is 0x4B.
Fig. 27.7. Entry of an MHP object carousel in a Program Map Table (PMT) as
analyzed on an MPEG analyzer [DVM]
566
27 DVB Data Services: MHP and SSU
Bibliography: [ISO/IEC13818/6], [EN301192], [ETS101812],
[TS102006]
28 T-DMB
The idea for T-DMB - Terrestrial Digital Multimedia Broadcasting comes from Germany, it was developed in South Korea, and its physical
parameters are identical to the European DAB (Digital Audio Broadcasting) standard. T-DMB is intended for the mobile reception of broadcasting
services similar to DVB-H. T-DMB corresponds wholly to DAB which itself supports the data stream mode also used in T-DMB (Fig. 28.1.). However, the "unequal forward error correction" possible in DAB is no longer
possible in this case because the entire subchannel used for the T-DMB
channel must be equally error protected.
MPEG-2
PSI
(PAT, PMT)
Section
generator
MPEG-4
SL
MPEG-2
PES
Video
MPEG-4
part 10
H.264
AVC
Audio
MPEG-4
part 3
BSAC
AAC
MPEG-4
SL
MPEG-4 ISO/IEC 14496
part 1, 3, 10
MPEG-2
multiplexer
MPEG-4 part 1
object descriptor
stream
MPEG-2
PES
to
DAB
data
stream
mode
RS
(204,
188)
Conv.
interleaver
Similar to DVB
outer coder
MPEG-2 ISO/IEC 13818-1
Fig. 28.1. T-DMB modulator block diagram
In T-DMB, the video and audio contents are MPEG-4-AVC- and AACcoded. The video coding uses the new H.264 method. Video and audio are
then packaged in PES packets and are then assembled to form an MPEG-2
transport stream (Fig. 28.1.) which also contains the familiar PSI/SI tables.
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_28, © Springer-Verlag Berlin Heidelberg 2010
568
28 T-DMB
The transport stream is then error protected similarly to DVB-C, i.e. with
Reed Solomon RS(204, 188) error protection plus Forney interleaving, after which the data stream is docked onto DAB in data stream mode (Fig.
28.2.).
Up to 64
subchannels
Audio
Stream
mode
unequal
FEC
MPEG-1/2
Layer II
Data
Stream
mode
Packet
mode
equal
FEC
T-DMB
Fig. 28.2. DAB data structure
Bibliography: [ETS300401], [T-DMB]
29 IPTV – Television over the Internet
Thanks to new technologies, the traditional transmission paths for television of terrestrial, broadband cable and satellite transmission have been
joined by an additional propagation path, the two-wire line, conventionally
known as telephone cable. VDSL (Very-high-bit-rate Digital Subscriber
Line, [ITU-T G.993]) now provides for data rates on these lines which also
allow television, IPTV – Internet Protocol Television, i.e. television over
the Internet. IPTV is now provided, e.g. by the German T-COM/Deutsche
Telekom or the Telekom Austria under the new slogan "Triple Play“.
"Triple Play“ is telephone, Internet and television out of one socket. The
term has also been applied for some time to broadband cable where all 3
media are also available from one socket.
TTXT, VPS
ITU
601
Matrix
Camera
R
G
B
Y
Cb
Cr
MPEG-2
TS
MPEG-2
encoder
SDI
Distribution
network
(microwave,
coax,
fibre)
MUX
MPEG-2
TS
Studio
L
R
RF
signal
Receiver
Modulator&
transmitter Transmission
link
(terrestrial,
satellite,
cable,
VDSL/IPTV)
Fig. 29.1. Distribution paths for digital television
The contents are here MPEG-4-coded to compress the input material optimally to the lowest possible data rates, using MPEG-4 AVC (or possibly
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_29, © Springer-Verlag Berlin Heidelberg 2010
570
29 IPTV – Television over the Internet
VC-1 (Windows Media 9)) and AAC. However, MPEG-2-coded video
streams and MPEG-1-coded audio streams are also still transmitted via IP.
Currently, four possibilities exist for transmitting DTV over IP. The first
one of these is proprietary where MPEG-4 Video or possibly. Windows
Media 9 (VC-1) is simply embedded, together with MPEG-4 Audio
(AAC), in UDP-packets (i.e. without handshake). The UDP-packets, in
turn, are then placed in IP-packets and are then transmitted via Ethernet,
WLAN, WiMAX or xDSL.
TV services over xDSL based IP networks
DVBIP
ISMA
proprietary
proprietary
ETS102034
„Streaming“
Video (MPEG-4 AVC, VC1)
Audio (MPEG-4 AAC)
MPEG-2 TS
RTP
Real Time Transport Protocol
UDP
User Datagram Protocol
IP
Ethernet / xDSL / …
Fig. 29.2. IPTV-Protocols
A further approach, also not standardized at the moment, is to insert
video and audio-streams into an MPEG-2-transport stream as specified in
the MPEG-2 and MPEG-4-standards, and then to transport this transport
stream in UDP- and IP-packets, e.g. also via xDSL. In the method specified as part of DVB-IP in the ETS 102034 standard, the RTP (Real Time
Transport Protocol) is additionally inserted between the transport stream
and the UDP layer. In ISMA (Internet Streaming Media Alliance) streaming the transport stream layer is missing but the RTP is used here, also.
All methods have in common that in each case only one program is trans-
29.1 DVB-IP
571
mitted on-demand. In the MPEG-2-transport stream, PAT and PMT-tables
are inserted for signalling purposes.
Fig. 29.3. Examples of DVB-IP-compliant transport streams, recorded in the network of Telekom Austria; on the left – MPEG-2 contents, on the right – MPEG-4
AVC and Dolby Digital
29.1 DVB-IP
In DVB-IP [ETS 102034], the MPEG-2 transport stream with either
MPEG-4- or MPEG-2-coded video signals and MPEG-4-, MPEG-2- or
MPEG-1-coded audio signals (Fig. 29.3.) is embedded via RTP (Real
Time Protocol) in UDP packets and then transmitted in an IP network via
DXL with instantaneous data rates of 8 or 16 Mbit/s. The main purpose of
the RTP is to assist in the restoration of the original order of the packets in
an IP network. The RTP also contains mechanisms for managing the timing (s.a. PCR jitter). The DVB IP has provisions for sending the MPEG-2
transport stream either with all PSI/SI tables or only sending the PSI tables
along. On registration, the delivery system sends to the DVB-IP receiver a
list of available services with the associated socket. A socket consists of an
IP address consisting of 4 bytes, and the associated UDP port. This address
has the following syntax:
a.b.c.d:port
572
29 IPTV – Television over the Internet
where a,, b, c and d have a value of between 0 ... 255 and port comprises
a range of from 0 ... 65535. "Normal" IP TV runs on multicast addresses
within a range of
224.0.0.0 ... 239.255.255.255.
It is only in the case of video on demand that unicast addresses make
sense which, with exceptions, can comprise almost the entire address range
of from
0.0.0.0 ... 255.255.255.255
According to the Internet Protocol, exceptions are:
127.0.0.1 (= local host),
x.x.x.0 (= current network),
x.x.x.255 (= broadcast),
244.0.0.0 ... 239.255.255.255 (= multicast).
On registering, an IP address, by means of which it can then receive
both multicast and unicast services, is assigned to the IPTV receiver. When
the service or program is selected by the user, the receiver then signals the
corresponding socket to the nearest network node (DSLAM) and is then
fed the MPEG-2 transport stream via precisely this address plus UDP port
via UDP protocol.
29.2 IP Interface Replaces TS-ASI
It has been noted that the TS-ASI interface is being replaced more and
more by a Gigabit Ethernet interface, especially in the head end and in the
playout center. This is being mentioned because it fits into the present
chapter. If head-end components are connected to one another via gigabit
IP, the transport stream, which is otherwise distributed via TS-ASI, must
be embedded fully compatibly, i.e. completely, in IP and be provided with
all associated PSI/SI tables. In the IP network , the required transport
stream is also addressed via a socket, i.e. via the 4-byte-long IP address
and the UDP port.
29.3 Summary
573
29.3 Summary
Replacing the TS-ASI interface with a Gigabit Ethernet interface appears
to be more and more prevalent. It remains to be seen how successful the
new IPTV offer via two-wire line will be compared with the other three
previous TV propagation paths. However, Triple Play - telephone, Internet
and television from one "connector", is a very interesting alternative to
what has hitherto been on offer with respect to communications, either by
broadband cable or by the two-wire line being discussed here. Like Mobile
TV, IPTV had been an important subject in the broadcasting exhibitions in
2007/2008.
Bibliography: [ITU-T G.993], [ETS102034]
30 DRM – Digital Radio Mondiale
In 2000, a further digital broadcasting standard called DRM - Digital Radio Mondiale [ETS 101980] was created. DRM is intended for the frequency band from 30 kHz ... 30 MHz, in which the AM service was normally transmitted. The broadcasting frequency bands were basically
divided in accordance with their propagation characteristics, as follows:
•
•
•
•
•
LW (Long Wave)
MW (Medium Wave)
SW (Short Wave)
VHF:
UHF:
~30 kHz ... 300 kHz
~300 kHz ... 3 MHz
~3 MHz ... 30 MHz
~30 MHz ... 300 MHz
~300 MHz ... 3 GHz
VHF is split into three bands:
•
•
•
VHF I:
VHF II:
VHF III:
47 ... 85 MHz
87.5 ... 108 MHz
174 … 230 MHz
UHF has two frequency bands, which are:
•
•
UHF IV:
UHF V:
470 ... 606 MHz
606 ... 826 MHz
In the frequency band below 30 MHz, very long-range reception is
sometimes possible which, however, is greatly dependent on diurnal
(day/night) variations and on solar activity. The channel bandwidths specified here are 9 kHz (ITU-Region 1 (Europe, Africa) and Region 3
(Asia/Pacific)) and 10 kHz (ITU-Region 2 (North and South America)).
DRM is the attempt to replace more and more unused frequency bands
in which amplitude modulation has hitherto been used, with modern digital
transmission methods. The modulation method applied is COFDM, using
MPEG-4 AAC for compressing the audio signals. The net data rates are
usually approx. 10 to 20 kbit/s.
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_30, © Springer-Verlag Berlin Heidelberg 2010
576
30 DRM – Digital Radio Mondiale
The channel bandwidths specified for DRM are derived from the bandwidths normally used in the frequency bands provided. The DRM bandwidths are between 4.5 kHz und 20 kHz (Fig. 30.2.) and are defined via
the parameter of "spectrum occupancy“. Table 30.1 shows the possible
bandwidths. As in other standards, too, which define COFDM as the
modulation method, modes are defined here. The DRM modes are designated as Robustness Mode A, B, C and D. The mode determines the carrier
spacing and the symbol duration. The physical parameters of the DRMmodes can be seen in Table 30.2. The number of carriers in an COFDMsymbol depends on the mode and on the DRM bandwidth. The number of
carriers which can be accommodated in a symbol is listed in Table 30.3.
Normal
protection
Source
encoder(s)
Data
FAC
info
SDC
info
Normal
protection
Energy
dispersal
MUX
Channel
coder
Cell
interleaver
Main service channel (MSC)
Precoder
Pilot
generator
[high
protection]
Fast access channel (FAC)
Precoder
Energy
dispersal
Channel
coder
Precoder
Energy
dispersal
Channel
coder
Service description channel (SDC)
Fig. 30.1. Block diagram of a DRM modulator
Table 30.1. DRM bandwidths
Spectrum 0
occupancy
Channel
4.5
bandwidth
[kHz]
1
2
3
4
5
5
9
10
18
20
OFDM
Mod.
1…4
services
[high
protection]
MSC: 16QAM/64QAM
FAC: QPSK
SDC: QPSK/16QAM
OFDM cell mapper
Audio
Normal/
[high]
protection
RF
30 DRM – Digital Radio Mondiale
4.5 kHz
5 kHz
9 kHz
10 kHz
18 kHz
20 kHz
577
Fig. 30.2. DRM spectra at 4.5, 5, 9, 10, 18 and 20 kHz bandwidth with the same
channel frequency in each case; it must be noted that the channel frequency does
not always correspond to the band center of the DRM spectrum; compare also Table 30.3. (Kmin/Kmax).
Table 30.2. DRM modes and their physical parameters
DRM
robustness
mode
A
B
C
D
Symbol
duration
[ms]
24
21.33
14.66
9.33
Carrier
spacing
[Hz]
41 2/3
46 7/8
68 2/11
107 1/7
tguard
[ms]
tguard/tsymbol
2.66
5.33
5.33
7.33
1/9
1/4
4/11
11/14
No.
of symbols
per frame
15
15
20
24
578
30 DRM – Digital Radio Mondiale
Table 30.3. Number of DRM carrier per COFDM symbol (Kmin = lowest carrier
no., Kmax = highest carrier no., Kunused = unused carrier numbers, SO = spectrum
occupancy)
Robustness
mode
A
A
A
B
B
B
C
C
C
D
D
D
Carrier
SO 0
4.5 kHz
SO 1
5 kHz
SO 2
9 kHz
SO 3
10 kHz
SO 4
18 kHz
Kmin
Kmax
Kunused
Kmin
Kmax
Kunsed
Kmin
Kmax
Kunused
Kmin
Kmax
Kunused
2
102
-1,0,1
1
91
0
-
2
114
-1,0,1
1
103
0
-
-102
102
-1,0,1
-91
91
0
-
-114
114
-1,0,1
-103
103
0
-69
69
0
-44
44
0
-98
314
-1,0,1
-87
279
0
-
SO 5
20
kHz
-110
350
-1,0,1
-99
311
0
-67
213
0
-43
135
0
Fig. 30.1. shows the block diagram of a DRM modulator. Up to 4 services (audio or data) can be combined to form one DRM multiplex and to
be transmitted in the so-called MSC (Main Service Channel). A DRM signal contains the following subchannels:
•
•
•
the MSC = Main Service Channel (16QAM/64QAM modulation)
the FAC = Fast Information Channel (QPSK)
the SDC = Service Description Channel (QPSK/16QAM)
The FAC is used for signalling the following information to the receiver:
•
•
•
•
•
•
Robustness mode
Spectrum occupancy
Interleaving depth
MSC mode (16QAM/64QAM)
SDC mode (QPSK/16QAM)
Number of services
The SDC is used for transmitting information such as
•
•
Protection level of the MSC
Stream description
30.2 Forward Error Correction
•
•
•
•
579
Service label
Conditional access information
Audio coding information
Time and date
30.1 Audio source encoding
DRM transmits MPEG-4-coded audio signals which can be compressed
with the following algorithms:
•
•
•
MPEG-4 AAC (= Advanced Audio Coding),
MPEG-4 CELP speech coding (= Code Excited Linear Prediction),
MPEG-4 HVXC speech coding (= Harmonic Vector Excitation
Coding)
30.2 Forward Error Correction
The forward error correction in DRM is composed of the following:
•
•
•
an energy dispersal block
a convolutional coder
a puncturing block
In DRM, it is possible to chose between
•
•
Equal FEC and
Unequal FEC
Parts of the audio frame can be error-protected to different degrees by
this means. The degree of error protection is determined via the protection
level and can be chosen as
•
•
•
•
PL = 0 (maximum error protection)
PL = 1
PL = 2
PL = 3 (lowest error protection)
The PL then results in a particular code rate.
580
30 DRM – Digital Radio Mondiale
30.3 Modulation Method
The Fast Access Channel (FAC) is permanently QPSK-modulated (Fig.
30.3.) since it is virtually the first "entry point“ for the DRM receiver and
must, therefore, be modulated firmly and very robustly.
FAC
MSC
SDC
MSC/SDC/FAC
Fig. 30.3. Modulation methods in DRM
In the case of the Service Description Channel (SDC) it is possible to
chose between QPSK and 16QAM as modulation method which is again
signalled to the receiver via the FAC. The types of modulation possible in
the MSC are either 16QAM or 64QAM (Fig. 30.3.) which is also signalled
to the receiver via the FAC. Apart from the modulated data carriers which
transmit the information of the MSC, FAC and SDC, there are also pilots
which are not responsible for any information transport. They have special
tasks and are mapped onto fixed constellation schemes known to the
modulator and receiver. These pilots are used for:
•
•
•
Frame, frequency and time synchronization
Channel estimation and correction
Robustness-mode-signalling
30.4 Frame structure
581
In DRM it is possible to chose, apart from "simple modulation“ (SM),
also "hierarchical modulation“ (HM), similar to DVB-T. Different levels
of error protection can then be used on the two paths of the hierarchical
modulation.
Transmission
super frame
Symbols containing MSC cells only
Transmission
frame
SDC block
Symbols containing MSC and FAC cells
Fig. 30.4. Frame structure in DRM
30.4 Frame structure
Like other transmission standards such as DVB-T or DAB, DRM, too, has
a frame structure (Fig. 30.4.) for arranging the COFDM symbols which is
organized as follows:
•
•
a certain number of COFDM symbols Ns results in an COFDMtransmission frame
3 transmission frames produce one transmission superframe
An COFDM frame, in turn, is composed of:
•
•
•
Pilot cells
Control cells (FAC, SDC)
Data cells (MSC)
582
30 DRM – Digital Radio Mondiale
In this context, cells are understood to be carriers allocated to various
uses. Control cells are used for transmitting the FAC and the SDC. Data
cells are used for transporting the MSC.
The pilot cells are simply the pilots already mentioned. Table 30.4.
shows how many CODFM symbols make up a transmission frame.
Table 30.4. Number of symbols Ns per frame
Robustness mode
A
B
C
D
Number of symbols Ns per transmission frame
15
15
20
24
At the beginning of a transmission super frame, the so-called SDCblock is transmitted in symbol no. 0 and 1 in Mode A and B and in symbol
no. 0, 1 and 2 in Mode C and D. After that, only MSC and FAC cells are
transported until the beginning of the next super frame (Fig. 30.4.).
Pilot carriers or pilot cells are distributed over the entire range of
COFDM carriers. Depending on the mode, they are spaced apart by 20, 6,
4 or 3 carriers from one another and skip forward by 4, 2 or 1 carrier from
symbol to symbol.
Table 30.5. Pilot Carriers
Mode
A
B
C
D
Pilot carrier spacing
in the symbol
20
6
4
3
Carrier skip distance
from symbol to symbol
4
2
2
1
30.5 Interference on the transmission link
DRM is operated in a frequency band in which atmospheric disturbances
and diurnal fluctuations of the transmission characteristics (ground and sky
wave) are particularly pronounced. In the frequency band below 30 MHz
there is mainly also the presence of man-made noise to be considered.
According to the standard, DRM has a bit error ratio of 1 • 106 in the
MSC after the channel decoder with a signal/noise ratio (S/N) of 14.9 dB
with 64QAM and a code rate of 0.6. In practice, the "fall off-the cliff“
30.6 DRM data rates
583
phenomenon (also known as "brickwall effect") was actually observed
with an approximate S/N of 16 dB at a CR=0.5. With 16QAM modulation,
this effect occurred with an SNR of about 5 dB (receiver: mixer DRT1 by
Sat Schneider and DREAM Software).
Table 30.6. “Fall-off-the-Cliff“ (Receiver: Mixer DRT1 by Sat Schneider, Germany and DREAM Software from the Technical University of Darmstadt, Germany
Transmission parameters
MSC=64QAM, CR=0.5
MSC=16QAM, CR=0.5
S/N at “fall-off-the-cliff”
16 dB
5 dB
30.6 DRM data rates
The DRM data rates depend on the DRM bandwidth (spectrum
occupancy), on the mode, on the selected type of modulations and on the
forward error correction. They are between about 5 and 72 kbit/s.
Table 30.7. MSC net data rates at a code rate of CR=0.6 (equal FEC, simple
modulation) with 64QAM
Robustness
mode
A
B
C
D
SO 0
4.5 kHz
[kbit/s]
11.3
8.7
-
SO 1
5 kHz
[kbit/s]
12.8
10.0
-
SO 2
9 kHz
[kbit/s]
23.6
18.4
-
SO 3
10 kHz
[kbit/s]
26.6
21.0
16.6
11.0
SO 4
18 kHz
[kbit/s]
49.1
38.2
-
SO 5
20 kHz
[kbit/s]
55.0
43.0
34.8
23.4
Table 30.8. MSC net data rates at a code rate of CR=0.62 (equal FEC, simple
modulation) with 16QAM
Robustness
mode
A
B
C
D
SO 0
4.5 kHz
[kbit/s]
7.8
6.0
-
SO 1
5 kHz
[kbit/s]
8.9
6.9
-
SO 2
9 kHz
[kbit/s]
16.4
12.8
-
SO 3
10 kHz
[kbit/s]
18.5
14.6
11.5
7.6
SO 4
18 kHz
[kbit/s]
34.1
26.5
-
SO 5
20 kHz
[kbit/s]
38.2
29.8
24.1
16.3
584
30 DRM – Digital Radio Mondiale
The lowest possible data rate (CR=0.5, 16QAM, Mode B, 4.5 kHz) 4.8
kbit/s. The highest possible data rate (CR=0.78, 64QAM, Mode A, 20
kHz) is 72 kbit/s.
30.7 DRM transmitting stations and DRM receivers
Numerous transmitting stations throughout the world have already been
converted from AM to DRM. Relevant information is available from the
Internet. Apart from software-based DRM receivers, compact receivers are
also available now. Software-based solutions are in most cases based on a
DRM signal down-converted at 12 kHz which is fed into the line-in socket
of a PC. A suitable example which can be mentioned is the DREAM software from the Technical University of Darmstadt (see also Fig. 30.5.).
Fig. 30.5. Constellation diagram of a DRM signal (MSC, FAC and SDC superimposed), recorded using the DREAM software
30.8 DRM+
585
30.8 DRM+
DRM+, an extension of DRM, is being developed and is intended for frequencies above 30 MHz. DRM+ could be a possible alternative to DAB.
Like DRM, DRM+ would work with the latest AAC+ codec and could be
used both in VHF band II, where VHF FM technology is currently employed, and in VHF band I which, as now, is empty.
Bibliography: [ETS101980], [DREAM]
31 Digital Terrestrial TV Networks in Practice
This chapter is intended to provide the practical engineer with an overview
of the configuration of TV transmitting stations and of the structure of single-frequency DVB-T networks, using as examples the DVB-T SFN networks of Southern and Eastern Bavaria with some TV transmitting stations
of the Bayerischer Rundfunk (BR) and T-Systems/Deutsche Telekom
(now Media Broadcast GmbH). The author has been able to experience the
commissioning of both networks at close quarters, both during the initial
training of some of the operating personnel, during visits in the installation
phase, and when the networks were being switched on. In addition, both
networks are located in the region in which the author has grown up and is
still living. And all the TV transmitting stations are completely equipped
throughout with "Bavarian Technology" by the companies
Rohde&Schwarz, Spinner and Kathrein. Beginning with the playout center, the entire feed link of the single-frequency networks (SFN) and especially the transmitting stations themselves, from the mask filter and the
combiner to the transmitting antenna will be described in this chapter. The
author was even granted the privilege of experiencing the "flight" of the
antenna by helicopter at the Olympic Tower, Munich, and at the Mt.
Wendelstein transmitter during the installation. This chapter also describes
measurements in a single-frequency network and explains how coverage is
measured in an SFN. In addition, the practical requirements for an SFNenabled receiver are discussed and how they can be verified. All this information is derived from practical experience and cannot be found in
some standards or documents: information from practical engineer to practical engineer.
31.1 The DVB-T SFNs Southern and Eastern Bavaria
The DVB-T single-frequency networks (SFNs) used as the example are
networks in the South of Germany, in Germany's largest federal state
which has a geography of high (Alps) and low mountain ranges and in-
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_31, © Springer-Verlag Berlin Heidelberg 2010
588
31 Digital Terrestrial TV Networks in Practice
cludes the gentle foothills. This topography was of the greatest significance in the planning of the networks.
The Southern Bavaria DVB-T network consists of the two transmitters
Olympic Tower Munich and Mt. Wendelstein. The Olympic Tower is a
typical telecommunications tower to the North-West of Munich with a
height of 292 m at about 450 m above sea level. It was originally used as
microwave tower for telephoning and was built in 1968. Microwave has
been largely replaced by optical fiber today and is no longer of the same
significance as before. Thus there are hardly any more microwave dishes
in operation on the Olympic Tower. At the upper end of the Olympic
Tower there are the transmitting antennas for FM radio, DAB and now
also for DVB-T.
Ch 34, 35, 48, 54, 56, 66 V
Olympic Tower
63 km
Mt. Wendelstein
Ch 34, 35, 48, 54, 56, 66 V
Fig. 31.1. DVB-T single-frequency network Southern Bavaria (DTK500; © Landesamt für Vermessung und Geoinformation Bayern, Nr. 4385/07)
The Mt. Wendelstein transmitter is located at about 1750 m above sea
level on the mountain of the same name which has a total height of 1850
31.1 The DVB-T SFNs Southern and Eastern Bavaria
589
m. Although it is not the highest mountain in Bavaria, or Germany, for that
matter, it has certainly one of the most beautiful panoramic views in Bavaria. The TV transmitter there is the oldest one in Bavaria, and possibly also
the one with the most beautiful location. Anyone who has been able to experience a sunrise or sunset there - which is something not many people
are able to do because of the lack of an hotel at the top - will be able to
confirm this.
Hoher Bogen
7, 53, 28 V
Hohe Linie
7, 33, 28 V
65 km
54 km
84 km
Brotjacklriegel
7, 33, 27 V
88 km
97 km
48 km
Pfarrkirchen
40, 33, 27 H
Fig. 31.2. DVB-T single-frequency network Eastern Bavaria (DTK500; © Landesamt für Vermessung und Geoinformation Bayern, Nr. 4385/07)
The Wendelstein transmitter belongs to the Bayerischer Rundfunk. The
two Munich Olympic Tower and Wendelstein transmitters form the SFN
Southern Bavaria which was taken into operation in the night of the 30th
May 2005. At the same time, this was the end of analog terrestrial television in Southern Bavaria. The Munich Olympic Tower and the Wendelstein transmitters broadcast 6 DVB-T channels completely synchronously
on the same frequencies as DVB-T single frequency network. The data
rates are about 13 Mbit/s each and each carries about 4 TV programs per
590
31 Digital Terrestrial TV Networks in Practice
data stream. Altogether, the viewer is thus provided with about 22 programs over terrestrial digital antenna TV. These TV programs, which are
both public service programs and private programs, form a viable alternative to the satellite and cable media. The transmitting frequencies are now
only located in the UHF band.
The DVB-T single-frequency network Eastern Bavaria consists of the 4
TV transmitting stations Pfarrkirchen (T-Systems/Deutsche Telekom, now
Media Broadcast GmbH), Brotjacklriegel (BR), Hoher Bogen (BR) and
Hohe Linie (BR). Two of these transmitters (Brotjacklriegel and Hoher
Bogen) are located in the low mountain ranges of the Bavarian Forrest at
approx. 1000 m above sea level. The 4 transmitting stations broadcast 3
DVB-T transport streams, some of them at the same frequencies. Only
public service programs are distributed. The data rate per data stream is
also approx. 13 Mbit/s. Alltogether, 12 programs are distributed. Fig. 31.1.
shows the sites of the transmitters in the DVB-T single-frequency network
Southern Bavaria and Fig. 31.2. shows the sites of the transmitters in the
DVB-T network Eastern Bavaria. With 63 km, the distances between
transmitters in the DVB-T single-frequency network Southern Bavaria,
consisting of the Olympic tower and Mt. Wendelstein, are right on the
limit of what is still allowed. In the DVB-T single-frequency network
Eastern Bavaria, the permissible distances between transmitters have been
greatly exceeded in some cases and, without inbuilt delay times, would
lead to guard interval violations at some locations. Table 31.1. and 31.2.
list the technical parameters of both networks.
31.2 Playout Center and Feed Networks
The Playout Center of the Bayerischer Rundfunk is located on the site of
the München-Freimann television studio. Here, 2 multiplexed data streams
are formed consisting of the ARD stream ("Das Erste, ..."), and the BR
stream ("Bayrisches Fernsehen“, "BR-Alpha“, ...). Throughout the country,
the ZDF stream comes directly from Mainz and contains (“ZDF, ...). The
other transport streams in the Playout Center directly at the Olympic
Tower in Munich are formed by Media Broadcast GmbH, partially by reception back via DVB-S. These are 3 further transport streams containing
private TV programs. These 3 transport streams are only broadcast in the
DVB-T network Southern Bavaria and not in Eastern Bavaria at the moment. The Olympic Tower is linked to the BR Playout Center by fibre optic, the transport streams of the Playout Center from Media Broadcast
GmbH being supplied to the broadcast studio via a few meters of cable.
31.3 Technical Configuration of the Transmitter Sites
591
The ZDF transport stream arrives from Mainz by optical fiber. All the
transport streams are then beamed up by microwave from the Olympic
tower to the Mt. Wendelstein by way of Schnaitsee-Hochries (Media
Broadcast GmbH) or via Freimann-Wendelstein (BR). The MIP inserters
for synchronizing the transmitters in the single-frequency network are located at the output of the respective Playout Center. The feed links and the
components in the Playout Center are all provided redundantly. The transport streams are fed to the DVB-T SFN Eastern Bavaria by microwave
(ARD-MUX, BR-MUX) or optical fibre (ZDF-MUX) via an ATM network.
Transmitter
MIP
inserter
MUX
1+2
Distribution
link 1
MPEG-2
TS
Distribution
network
(ATM/SDH)
MUX
MPEG-2 TS
(Spare)
MIP
inserter
(Spare)
Distribution
link 2
(Spare)
1+2
Transmitter
Fig. 31.3. Feed of a transport stream from the playout center to the DVB-T transmitters in the single-frequency network
31.3 Technical Configuration of the Transmitter Sites
In this section, the configuration of a DVB-T site will be explained by
means of three examples. These are the Olympic tower Munich (Media
Broadcast GmbH), Wendelstein (BR), and Brotjacklriegel (BR) transmitters. As described in the previous chapter, the MPEG-2-transport stream is
592
31 Digital Terrestrial TV Networks in Practice
supplied to the TS-ASI input interface over the most varied networks, but
finally via 75 Ohm coaxial cable. There is always a spare link provided.
The important factor is that, naturally, the same MIP inserter always feeds
the spare link. Each MIP inserter per se operates completely asynchronously and only creates synchronous information in the MPEG-2 transport
stream by itself inserting MIP packets. Each MIP inserter inserts these
special transport stream packets virtually completely freely into gaps in the
MPEG-2 transport stream instead of null packets (PID=0x1FFF).
The DVB-T transmitters at all three sites are liquid cooled solidstate
transmitters of the NX7000 and NX8000 series by Rohde&Schwarz, constructed in various power classes and in various spare capacities.
Table 31.1. Channel allocation and power Mt. Wendelstein transmitter
Channel
Frequency
34
578 MHz
Mean transmitter
output power
5 kW
35
586 MHz
5 kW
48
690 MHz
5 kW
54
738 MHz
5 kW
56
754 MHz
5 kW
66
834 MHz
5 kW
ERP
100 kW
directional
100 kW
directional
100 kW
directional
100 kW
directional
100 kW
directional
100 kW
directional
31.3.1 Mount Wendelstein Transmitter
The Mt. Wendelstein transmitter is the oldest television transmitter site in
Bavaria and may well be one of the oldest in all of Germany. It was taken
into operation as television transmitter in 1953, is sited at 1740 m above
sea level and its antenna is at approx. 1840m. The Wendelstein mountain
itself is very popular as destination for outings by wanderers and skiers.
From there, six transport streams are being broadcast by DVB-T since
May 30th, 2005. Like from the Olympic Tower, these are channels 34, 35,
48, 54, 56 and 66. In addition to FM transmitters and one DAB transmitter,
an analog TV transmitter had been operating there on VHF channel 10 for
many decades. At the place where the analog TV transmitter was located,
all 6 UHF-transmitters are now installed. From all 6 UHF transmitters, outof-band components, which are not permissible and could interfere both
31.3 Technical Configuration of the Transmitter Sites
593
with the adjacent channels and other channels, are first removed by in each
case one mask filter (critical mask). The required shoulder attenuation is
generally about 52 dB.
Table 31.2. Program offer and technical parameters, DVB-T Network Southern
Bavaria (channel 10 has been moved to channel 54 in July 2009)
Channel
Modulation
Code
rate
Guard
Data rate
[Mbit/s]
54
16QAM
2/3
1/4
13.27
Number of
programs
4
34
16QAM
2/3
1/4
13.27
4
35
16QAM
2/3
1/4
13.27
3
+ MHP
48
16QAM
2/3
1/4
13.27
4
56
16QAM
2/3
1/4
13.27
3
+ MHP
66
16QAM
2/3
1/4
13.27
4
Programs
Das Erste,
Phoenix,
arte,
1plus
RTL,
RTL2,
VOX,
SuperRTL
ZDF,
3sat,
Doku/Kika
Sat1,
Pro7,
Kabel 1,
N24
Bayr.
Fernsehen,
BR alpha,
SWR
Tele5,
Eurosport,
HSE24,
München
TV
The mask filters are made by the company Spinner and are designed as
dual mode and cavities filters. The pass-band attenuation of such a filters is
about 0.6 dB. The group delay of this filter can be easily pre-equalized,
and thus compensated for, by the TV transmitter. Like the transmitters
themselves, these mask filters are here located in the same room and are
coupled to the transmitters "from the top" via 50 Ohm coaxial copper tubing. I.e., the transmitter output is at the top, which is not always the case.
From the mask filters it continues in this case to the combiners which are
594
31 Digital Terrestrial TV Networks in Practice
located underneath and via which in each case one transmitter is connected, decoupled from the other ones, to the antenna line.
The combiner is also made by Spinner. The configuration of the combiners for one channel is shown in Fig. 31.10. and consists of two 3 dB
couplers and two band-pass filters tuned to the supplied channel in each
case. The channel to be coupled in is thus supplied separately from the
others.
Thus, 6 UHF channels are coupled together and conducted up to the
transmitting antenna via a coax cable. The length of the line to the antenna
is about 280m. Depending on frequency, the cable attenuation can be assumed to be about 0.5 dB per 100 m length of transmitting cable.
RSource
50 Ohm
-3dB 0o 50 Ohm
RLoad
/4
RTerm
50 Ohm
-3dB -90o 50 Ohm
RLoad
Fig. 31.4. 3-dB splitter
31.3.1.1 Transmitter Technology Used
All of the 6 DVB-T transmitters on the Mt. Wendelstein are liquid-cooled
high-power solid-state transmitters of the NX7000 and now also NX8000
series by Rohde&Schwarz. Such a transmitter can be imagined simply as a
high-power amplifier driven by an exciter and produced by connecting together a large number of power amplifier modules. Each transistor in these
single power amplifier stages generates a average power of about 25 W.
The output signals of the amplifiers are added together via 3 dB couplers.
Using 3 dB couplers, more and more amplifiers are coupled together one
by one so that a total power of about 450 W per amplifier module is ob-
31.3 Technical Configuration of the Transmitter Sites
595
tained in the case of the NX7000 series by Rohde&Schwarz. The output
powers of the various amplifier modules are then combined again via couplers to give the total output power of the transmitter of about 5 kW (mean
power) in the case of the Mt. Wendelstein transmitter. These couplers can
be implemented as Wilkinson coupler or again as 3dB coupler. A Wilkinson coupler is a 0 degree coupler, whereas the 3dB coupler is a 90 degree
coupler.
50 Ohm 0 dB 0o
50 Ohm
P=0
RSource
RTerm
/4
50 Ohm
50 Ohm
-90o
0 dB
RSource
+3dB -90o
RLoad
Fig. 31.5. 3-db combiner
31.3.1.1.1 3dB Coupler as Splitter and Combiner
In principle, a directional coupler consists of two closely adjacent parallel
lines with a length of λ/4. The spacing between the lines determines the
overcoupling attenuation; if this is 3 dB, it is called a 3 dB coupler. If a
signal is fed into an input of a 3 dB coupler, 3 dB of it are coupled out with
0 degree phase shift at the output opposite the input, and another 3 dB of it
are coupled out with 90 degree phase shift at the output of the λ/4 line section connected electrically to the input.
A 3 dB coupler can be used for adding powers by feeding a signal into
one input with 0 degree phase shift and feeding it with a 90 degree phase
shift into the input of the parallel λ/4 line section (3 dB combiner). The
signals then cancel at one output and the aggregate power is present at the
other output with 90 degrees phase rotation. The unused output is terminated with 50 Ohms (load matching resistor). In the case of phase errors or
596
31 Digital Terrestrial TV Networks in Practice
power differences in the signals supplied, power is also absorbed in the
load matching resistor.
Fig. 31.6. shows a simplified circuit symbol of a 3 dB coupler which
will be used in the following sections.
Fig. 31.6. Simplified circuit symbol of a 3-dB coupler
Vdc=approx. 30 V DC
Vdc
Vdc
50 Ohm
Push pull
MOS FET
See above
50 Ohm
Fig. 31.7. Principle of a single power amplifier stage (50 W), used to build up a
power amplifier module
31.3.1.1.2 Single Power Amplifier Stage (appr. 50 W)
Firstly, the basic principle of a single power amplifier stage for
DVB-T/ATSC/Analog TV of the 50 W averaged power will be explained.
Such power amplifiers also form the high-power transmitter for
DVB/ATSC/Analog TV. The input signal is split into two signals of –3dB
31.3 Technical Configuration of the Transmitter Sites
597
power each and 90 degrees phase difference by means of a 3 dB coupler.
Each amplifier path consists of a class AB amplifier which, in turn, consists of a dual transistor operated in push-pull mode. The operating point is
set in such a way that similar conditions are achieved for all transistors in
class AB mode and the transfer distortion is minimal. In contrast to FM
transmitters, the amplifiers used here are very linear transmitters but must
still be pre-corrected. In FM transmitters, class C amplifiers are used
which are quite nonlinear but have much greater efficiency.
Fig. 31.8. Principle of a power amplifier module for VHF or UHF
In the FM transmitter, the required power can even be chosen by controlling the supply voltage of the class C amplifiers. In television transmitters, where very good linearity is required, power is controlled via the amplifier feed power. This applies especially to analog television, but also to
digital television (DVB-T, ATSC, ISDB-T). These amplifiers operate with
VMOS transistors (VHF band), or LDMOS transistors (UHF band) and are
already "pre-equalized" via the exciter, i.e. the characteristic is simulated
in the equalizer and used for comparison.
In principle, a class AB amplifier consists of a push-pull stage which is
set by the quiescent current of the transistor in such a way that the transfer
distortion is already minimized. The transistor supply voltage is about
30 V.
To build up a power amplifier module, the input signal is first amplified
from approx. 0 dBm to a reasonable order of magnitude and then split with
the correct power and phase over a number of 3 dB couplers and supplied
598
31 Digital Terrestrial TV Networks in Practice
to the respective individual amplifiers. The output powers of the individual
amplifiers are then combined again and again via 3 dB couplers to form
the total output signal of an amplifier module. The average total power of
an power amplifier module is then approx. 450 W. Then the power amplifier modules are coupled together building a power amplifier unit.
31.3.1.2 Mask Filter
The mask can be implemented as an uncritical mask or as a critical mask,
depending on the requirements of the relevant regulating authorities. In the
case of the Mt. Wendelstein and Olympic Tower transmitters, they are filters with a critical mask, i.e. the "shoulder" of the DVB-T signal must be
lower by more than 51 dB in the adjacent channel. The manufacturer of
these filters is the company Spinner in Munich. The filters are so-called
dual mode filters. The filters are passive mechanical cavity resonators and
are relatively large because of their power rating and weigh more than 100
kg. The mask filters are simply used for lowering or suppressing adjacent
channel emissions. The mask filters are tuned to the respective channel and
have an attenuation of about 0.3 to 0.6 dB in the passband which is also
easily noticed as heat.
31.3.1.2.1 Filter Technology
In the band-pass filters used here, a technological distinction is made between coaxial filters and waveguide filters. In both technologies, band-pass
filters with 3 to 10 sections are then constructed as required. They both
have their advantages and disadvantages. A coaxial filter has the following
characteristics:
•
•
•
continuous tuning in Band III or IV/V
greater attenuation (e.g. 0.31 dB in mid-band)
good temperature stability
The waveguide filter characteristics are:
•
•
•
•
less attenuation (e.g. 0.17 dB in mid-band)
poorer temperature stability
tunable only over some channels (5 - 6 channels)
larger dimensions
31.3 Technical Configuration of the Transmitter Sites
599
In combiners, 3- or 4-section filters are used. In mask filters, 6-section filters are needed for the uncritical mask and 8-section filters for the critical
mask.
31.3.1.2.1.1 Coaxial Filter
In principle, coaxial filters are nothing else than λ/4-long coaxial lines
which are short-circuited at one end and open at the other end. These can
be used for constructing resonators. The resonant frequency of the resonators can be varied by varying the length of the line. The impedance of a
coaxial line with round inner conductor and round outer conductor is:
z L = 377Ω ⋅ ln( D / d )(2π ε r ) = 60Ω ⋅ ln( D / d ) / ε r ;
where
d = diameter of the inner conductor;
D = diameter of the outer conductor;
εr = relative dielectric constant.
The impedance for the combination of round inner conductors and rectangular outer conductors is:
z L ≈ 60Ω ⋅ ln(1.07 ⋅ D / d ) / ε r ;
The following applies to coaxial systems:
Minimum attenuation at ZL = 77 / εr
maximum dielectric strength at ZL = 60 / εr
maximum transferable power at ZL = 30 / εr
The mask filter and the antenna combiner require mainly a low attenuation
to keep the losses as low as possible. The dielectric is air, resulting in εr =
1. The selected impedance is about 77 since the attenuation is lowest
there.
The wavelength is calculated as:
λ = c0 /( f ε r );
where
600
31 Digital Terrestrial TV Networks in Practice
c0 = velocity of light = 3 x 108 m/s
f = frequency;
εr = relative dielectric constant.
This results in a λ/4 resonator line length of about 36 cm to about 8 cm in
the frequency band of between about 200 and 900 MHz with an air dielectric. The cavity diameter depends on the power class, choosing round or
square cavities. The line can also be shortened by loading it capacitively at
its open end. Coupling into and out of the cavity can be capacitive or inductive. Fig. 31.9. shows the basic configuration of a two-section coaxial
filter. The signal is coupled in capacitively by a plunger the engaged depth
of which is adjustable. The resonator lengths l can be adjusted by inserting
plungers which are short-circuited at one end, to a greater or lesser depth.
The resonator length then determines the resonant frequency which can be
influenced additionally by trimming screws.
Cavity 1 + 2
D
Capacitive
coupling
(input),
adjustable
Inductive
coupling
of cavities
using a conductor loop
(length adjustable)
Screw for
fine trimming
d
Resonator length l
adjustable
Capacitive
coupling
(output),
adjustable
Screw for fine trimming
Fig. 31.9. Structure of a coaxial filter
Coupling from cavity 1 to cavity 2 is carried out here by a conductor loop
which can be inserted to a greater or lesser depth, making the coupling between the two resonant cavities controllable. The degree of coupling determines the bandwidth of the filter. The output coupling from cavity 2 is
also capacitive and is also adjustable. To make the sides of multi-section
31.3 Technical Configuration of the Transmitter Sites
601
filters even steeper, further non-adjacent cavities are also adjustably connected.
The linear expansion of the resonators must be compensated for as far as
possible by a suitable choice of materials since otherwise the resonant frequency will become detuned.
Conductor loops
Resonators
Screws for fine adjustment
Fig. 31.10. Trimmers on a 6-section coaxial filter (Spinner)
31.3.1.2.1.2 Waveguide Filters
Waveguide filters are hollow chambers, or cavities, which are really hollow without inner conductor and in which real waveguide modes are generated. Cavities can also be used for creating resonators the resonant frequency of which ultimately depends on the volume of the waveguide. The
shape of the filters used in mask filters is cylindrical; the two parameters D
= diameter and h = height of the cylinder then determine the resonant frequency and also the Q factor. In the cylindrical cavity, two modes which
do not interfere with one another can be excited orthogonally to one another (by 90 degrees). This makes it possible to use this cavity resonator
virtually twice, thus saving space. It is then called a dual-mode filter, implementing two sections with one cavity. A 6-section filter thus requires 3
602
31 Digital Terrestrial TV Networks in Practice
cavities and an 8-section filter requires 4. Fig. 31.11. shows the basic configuration of a dual-mode waveguide filter.
D
Coupling
screws
Trimming screws for
resonance frequencies
Trimming
screws
for
resonance
frequencies
Capazitive
coupling
(output)
Cavity 2
Coupling slot
between
cavity 1 and 2
Cavity 1
Capacitive
coupling
(input)
Fig. 31.11. Structure of a dual-mode waveguide filter
Coupling
slot
adjustement
Coupling
screws
at 45 degree
Trimming screws for
resonance frequencies
Fig. 31.12. Trimmers of a dual-mode filter (Spinner)
h
31.3 Technical Configuration of the Transmitter Sites
603
The signal is coupled capacitively into cavity 1 by a plunger which can be
inserted to a greater or lesser depth. Opposite to it there are trimming bolts
which can be screwed in more or less deeply and can be used for tuning
the resonant frequency. A coupling screw which rotates the field and generates an orthogonal waveguide mode at an angle of 90 degrees from the
first one is arranged at an angle of 45 degrees to the bolts. Coupling between cavities 1 and 2 is via a variable-length coupling slot. In the second
cavity there are also trimming bolts, and a couling screw. The degree of
coupling between the orthogonal waveguide modes is determined by the
depth of insertion of the coupling screws. When the waveguide filter becomes hot, the cavities expand and the resonant frequency thus becomes
detuned towards lower frequencies. This can be prevented by using metal
alloys (Invar) which have a particularly low coefficient of expansion.
Waveguide filters are larger than coaxial filters; both the cavity height and
its diameter must accommodate a half wavelength. The waveguide mode
excited is called an H111 wave.
0º,
-3dB
0º, 0dB
50
-90º, 0dB
50
-90º,
-3dB
Fig. 31.13. Cascading two 3-dB couplers
31.3.1.3 Combiner
The combiner has the task of combining the various TV channels to form
one signal which is then supplied to the respective TV transmitting antenna
via one cable in the VHF band and one cable in the UHF band. The transmitters themselves must be well decoupled from one another which is precisely what the filters in the combiner are doing. The combiner has an approximate passband attenuation of 0.3 dB.
Each channel separating filter of the combiner consists of two channel
filters which are tuned to the respective TV channel to be supplied. This is
preceded and followed by a 3-dB coupler. To understand the operation,
two cascaded 3 dB couplers can first be considered (Fig. 31.13.).
604
31 Digital Terrestrial TV Networks in Practice
If a signal is fed into the first coupler, it divides into 2 signals of 0 degrees
and 90 degrees in phase. The second coupler adds these signals again to
form a signal which is now shifted by 90 degrees compared with the input
signal. In the channel separating filter, band-pass filters are connected
between the two couplers.
f2
Narrow-band input
f1
f1
Broad-band input
f1+f2
Output
50 Ohm
f1
Fig. 31.14. Principle of an antenna combiner
Transmitter 1
f1
Transmitter 2
f2
f1
Air-cooled/liquid-cooled
dummy load
f2
f1
f1
f2
To/from further
combiners
f2
Jumpers
Jumpers
f1
f2
Input
for
further
extensions
To
antenna
and
further
combiners
Fig. 31.15. Transmitter, mask filter and antenna combiner
The channel separating filter has a narrow-band input and a wide-band
input and is basically constructed in exactly the same way as a vision/sound diplexer in analog television. The channel to be supplied passes
through the filter with 0 degrees and 90 degrees phase shift and the signal
31.3 Technical Configuration of the Transmitter Sites
605
of the other transmitters is immediately totally reflected at the filters in the
wideband input after the coupler and comes out again in aggregate with the
supplied channel at the wideband output. Both filters of the channel separating filter must be tuned at least relatively identically.
Fig. 31.15. shows a combiner with two transmitters, two mask filters, two
combiners, and patch panels at which the combiner can be bridged and the
transmitter output can be switched to a dummy load, if required, in order to
be able to carry out tuning and measuring work without applying the signal
to the antenna.
31.3.1.4 Antenna and Feed System
From the transmitter building on the Wendelstein mountain to the
transmitting antenna, a total of three lines are run which are “flexible” 50
Ohm coaxial cables. The very first transmitter cable was pulled up unwound on the rail by the rack railway in a "special action" at the beginning
of the fifties. The new cables were flown up the mountain by helicopter,
among them the last UHF cable which, with a diameter of almost 20 cm, is
the thickest cable ever used on the Wendelstein mountain. One cable is
used for the VHF band and one is used for the UHF band, and the third
cable is a spare for emergency cases.
Such coax cables have approximately the following attenuations over
100 m length in each case:
Table 31.3. Technical parameters of HELIFLEX (¤RFS) coax cables
Diameter of
the coax
cable
4-1/8’’
5’’
6-1/8’’
8’’
Max. avg.
power
at 500 MHz
35 kW
55 kW
75 kW
120 kW
Att. [dB] at
200 MHz
Att. [dB] at
500 MHz
Att. [dB] at
800 MHz
0.4 dB
0.3 dB
0.3 dB
0.2 dB
0.7 dB
0.5 dB
0.4 dB
0.4 dB
0.9 dB
0.7 dB
0.6 dB
0.5 dB
At the top, at the solar observatory and the meteorological observatory
of the German Meteorological Service (Deutscher Wetterdienst), there is
also the so-called antenna house from which the cables are then run to the
actual transmitting antenna. It contains another patch panel via which the
upper and lower half antenna of the transmitting antenna can be selectively
fed or disconnected both in the VHF band and in the UHF band. If necessary, this is done via 20 cm-large coax-type U-links.
The antenna itself consists of the following components by Kathrein,
Rosenheim, which are housed in a FRP (fiber-reinforced plastic) cylinder.
606
31 Digital Terrestrial TV Networks in Practice
The FRP cylinder has a total height of approx. 24 m and the antenna as a
whole is appr. 65 m high; the tip of the antenna being about 1900 m above
sea level.
The VHF antenna is composed of 6 levels with 6 vertically polarized
Band-III dipole antenna arrays. The bottom 3 levels form the lower half
VHF antenna, the upper 3 levels form the upper half VHF antenna. Both
half antennas are supplied via their own feed cable each. The UHF antenna
consists of 12 levels with 8 Band-V/V antenna arrays each which, just like
the VHF antenna, is built up from the lower and upper half antenna (with 6
levels each in this case). Here, too, each half antenna is driven by its own
coaxial cable. Above the UHF antenna, there is the mechanical vibration
absorber which is intended to prevent oscillations in the case of wind loading.
~ 1.2 m
~2m
FRP
cylinder,
wall strength ~ 24 m
~ 1cm,
diameter
~ 1.6 m
~ 12m
Lightning arrester grid
Mechanical vibration absorber
(system Prof. Dr. Nonnhoff)
Upper half antenna
12 levels with
UHF
8 band IV/V antenna
antenna
arrays in
each level
Lower half antenna
Upper half antenna
~ 9m
VHF
antenna
6 levels with
6 band III antenna
arrays in each level
Lower half antenna
Adapter
Tower
Fig. 31.16. Structure of a VHF/UHF transmitting antenna constructed in FRP
31.3.2 Olympic Tower Transmitter, Munich
The Munich Olympic Tower has originally been built in 1968 as telecommunications tower. Since April 2005, it has at its top the DVB-T transmit-
31.3 Technical Configuration of the Transmitter Sites
607
ting antennas which, together with the Mt. Wendelstein transmitter, form
the DVB-T single-frequency network Southern Bavaria since May 30th,
2005. From here, too, a total of 6 channels are broadcast in single frequency completely synchronized with the Mt. Wendelstein transmitter.
The output power of the DVB-T-transmitters is approximately twice as
high as that of the transmitters on the Wendelstein mountain. It is about 10
kW per channel in the UHF band. In contrast to the Mt. Wendelstein
transmitter, where the main pattern of the antenna points to the North, the
Olympic Tower has an omni-directional pattern. The ERP is about 100 kW
in the UHF band from the Olympic Tower.
Fig. 31.17. Olympic Tower Munich; antenna installation for DVB-T using a helicopter (left) and Wendelstein antenna (right)
The high-power transmitters are also liquid-cooled NX7000/8000 series
solid state transmitters by Rohde&Schwarz, but with much higher power
capacity. As far as the installation is concerned, the system is set up for
heating the Olympic pool close-by with the waste heat of the 6 high power
transmitters – which, in spite of their very effective efficiency, is not inconsiderable – instead of dissipating it to the environment via heat exchangers.
608
31 Digital Terrestrial TV Networks in Practice
Table 31.4. Technical parameters Olympic Tower transmitter
Channel
Frequency
34
578 MHz
Average
transmitter
output
power
9.3 kW
35
586 MHz
9.3 kW
48
690 MHz
9.3 kW
54
738 MHz
9.3 kW
56
754 MHz
9.3 kW
66
834 MHz
9.3 kW
ERP
100 kW
omnidir.
100 kW
omnidir.
100 kW
omnidir.
100 kW
omnidir.
100 kW
omnidir.
100 kW
omnidir.
Fig. 31.18. Wendelstein transmitter hall; antenna combiner (left) and UHF transmitters (right)
31.3 Technical Configuration of the Transmitter Sites
609
Mask filter and combiner are made by Spinner, but are also of much larger dimensions. The transmitting antenna is of similar design as that on the
Wendelstein mountain and has also been manufactured by Kathrein.
Table 31.5. Technical parameters Brotjacklriegel transmitter
Channel
Frequency
7
33
28
191.5 MHz
570 MHz
522 MHz
Average
transmitter
output
power
4.6 kW
5 kW
3.4 kW
ERP
25 kW
50 kW
100 kW
Table 31.6. Range of programs distributed by the Brotjacklriegel transmitter
Channel
Modulation
Code rate
Guard
Data rate
[Mbit/s]
Number
of programs
4
7
16QAM
3/4
1/4
13.06
33
16QAM
2/3
1/4
13.27
3
+ MHP
28
16QAM
2/3
1/4
13.27
3
+ MHP
Programs
Das Erste,
Phoenix,
arte,
1plus
Bayr.
Fernsehen,
BRAlpha,
SWR
ZDF,
Info/3sat,
Doku
/Kika
31.3.3 Brotjacklriegel Transmitter
The Brojacklriegel transmitter belongs to the DVB-T network Eastern Bavaria and, like the other transmitter sites in Eastern Bavaria, has 3 DVB-T
transmitters. The DVB transmitter is thus currently much smaller than the
Wendelstein and Olympic Tower installations but its performance category
corresponds to that of the Wendelstein site. Brotjacklriegel currently only
broadcasts public service programs which is why 3 data streams or trans-
610
31 Digital Terrestrial TV Networks in Practice
mitting frequencies are sufficient here. The Brotjacklriegel antenna is also
of similar construction to the Wendelstein antenna.
The Hoher Bogen (Bayerischer Wald, Furth im Wald) and Hohe Linie
(Regensburg) transmitting stations are comparable sites of the Bayerischer
Rundfunk in the Eastern Bavarian DVB-T network.
Channel 7 of the "Brotjacklriegel" transmitter runs synchronously in a
single-frequency network with the "Hohe Linie" transmitter at 84 km distance near Regensburg and the "Hoher Bogen" transmitter at 54 km distance. However, the distance to the "Hohe Linie“ would violate the guard
interval constraint (77 km in the VHF band) which is why the "Brotjacklriegel" signal in channel 7 is radiated 20 s earlier which effectively
moves the transmitter towards the "Hohe Linie" and also towards the
"Hoher Bogen" by 84 km. Because of the mountain chain of the intervening Bavarian Forrest, the "Brotjacklriegel“ and "Hoher Bogen“ transmitters are already well decoupled from one another.
Fig. 31.19. Liquid cooled mask filter (8 kW) for DTV (DVB/ATSC); manufacturer's photo, Spinner.
31.3 Technical Configuration of the Transmitter Sites
611
Fig. 31.20. Antenna combiner Olympic Tower Munich, rear view; manufacturer's
photo, Spinner.
Fig. 31.21. DVB-T transmitters Olympic Tower Munich; manufacturer's photo,
Rohde&Schwarz.
612
31 Digital Terrestrial TV Networks in Practice
31.4 Measurements in DVB-T Single-Frequency Networks
Single-frequency networks (SFNs) are "special" networks. They must be
•
•
•
synchronous in frequency,
synchronous in time, and
meet the requirements for the guard interval.
To ensure that these conditions are met in the region of Bavaria, as in the
previous examples, they must be measured and monitored during the
commissioning and also in later operation. This chapter is the result of
numerous inputs from readers and participants in seminars.
An SFN must firstly be planned correctly with knowledge of the topographical and geographical structure. The transmitter distances must not
violate the guard interval condition, i.e. they must not exceed a particular
maximum distance from one another. Should this be the case, nevertheless,
it may help to "shift" one or the other transmitter by advancing or delaying
the radiation of the COFDM signal. It is then possible to guarantee reliable
reception in areas where otherwise signal paths would have exceeded the
guard intervals. However, this gain may be at the cost of problems in other
regions. Naturally, the antenna pattern also plays an important role. An
SFN can be modelled by "drawing in" the pattern, i.e. reducing the radiated power in a certain direction.
Most of the DVB-T matters discussed in this chapter can be easily applied
also to other standards using COFDM as a modulation method, such as,
e.g., DAB or ISDB-T.
31.4.1 Test Parameters
Now to answer the question "What is actually measured how in DVB-T
single-frequency networks?". Naturally, the matters applicable to measuring SFN coverage are also applicable to the special case of an SFN, the
MFN - Multi-Frequency Network, where each transmitter has its own frequency. In comparison with an SFN, the receiver can only expect one signal path from the transmitter, and possibly "correct" echo paths, in this
case. But the delay differences are only within a range of about 1- 10 s
instead of up to about 200 s, by comparison. The parameters to be measured in the field are:
•
•
Level or field strength
Modulation error ratio
31.4 Measurements in DVB-T Single-Frequency Networks
•
•
•
613
Bit error ratios
Channel impulse response
Constellation diagram (visual assessment).
Naturally, the most important test parameter is firstly the signal level or
field strength present on site. The signal level is measured as the output
signal of a known test antenna. Its K factor or antenna gain can then be
used for calculating the field strength. The formula for this is:
E[dBµV/m] = U[dBµV] + k[dB/m];
k[dB] = (-29.8 + 20•log(f[MHz]) – g[dB]);
where
E = electrical field strength
U = antenna output level
k = antenna k factor
F = received frequency
g = antenna gain
The required minimum receiver input level depends on the selected modulation parameters and on the quality of the receiver. As shown in Chapter
20, a noise level of about 10 dB V can be expected at the receiver input,
which leads to a minimum level of about 22 dB V with 16QAM, code rate
2/3 and of about 28 dB V with 64 QAM, code rate 2/3. This corresponds
quite well to reality in an AWGN (Additive White Gaussian Noise) channel. It won't hurt to add a margin of about 3 dB to this, however. There are
implementation losses (antenna, cable) and there are differences in the
quality of receivers. However, these minimum receiver input levels only
apply to the case of one-way reception, i.e. the pure AWGN channel. In
practice, multi-path reception often requires an input level which is higher
by 5 to 10 dB. This is due to the actual characteristics of the DVB-T demodulator chips built into the DVB-T receivers. It can be demonstrated,
however, that there are distinct differences here and that the latest generations of DVB-T receivers and chips come much closer to expectation. Calculating then firstly the minimum field strengths with an antenna gain of 0
dB, using the theoretical minimum levels without deductions, and then
also adding QPSK, with code rate 2/3 (-6 dB compared with 16QAM = 16
dB V):
614
31 Digital Terrestrial TV Networks in Practice
Table 31.7. Theoretically required minimum receiver input level (CR=2/3) in an
AWGN channel with DVB-T with 0 dB antenna gaain and no implementation
losses, receiver noise figure = 7 dB, ambient temperature = 20 ºC.
k factor at
0dB gain
Min. level
with QPSK
Min. received
field strength
with
QPSK
Min. level
with 16QAM
Min. received
field strength
with
16QAM
Min. level
with 64QAM
Min. received
field strength
with
64QAM
200
MHz
16.4 dB
500
MHz
24.2 dB
600
MHz
25.8 dB
700
MHz
27.1 dB
800
MHz
28.3 dB
16 dB V
16 dB V
16 dB V
16 dB V
32.4
dB V/m
40.2
dB V/m
41.8
dB V/m
43.1
dB V/m
16
dB V
44.3
dB V/m
22 dB V
22 dB V
22 dB V
22 dB V
38.4
dB V/m
46.2
dB V/m
47.8
dB V/m
49.1
dB V/m
28 dB V
28 dB V
28 dB V
28 dB V
44.4
dB V/m
52.2
dB V/m
53.8
dBuV/m
55.1
dB V/m
22
dB V
50.3
dB V/m
28
dB V
56.3
dB V/m
In reality, the implementation losses must be added to this and these, in
turn, depend on the chosen receiving situation. There are ultimately four
receiving situations which are:
•
•
•
•
reception by fixed outdoor antenna,
reception by portable outdoor antenna,
reception by portable indoor antenna,
mobile reception.
With reception by fixed outdoor antenna, the antenna gain of about 6 to
12 dB is added to this and ensures that correspondingly less field strength
is required. Field strength is here defined at a corresponding height above
ground, in most cases 10 m, and the measurements are therefore also taken
under these conditions (mast, 10 m, directional antenna). However, it is
advisable also to take into consideration the line losses etc. to the receiver
(e.g. 6 dB) and to include these in the calculations. With reception by portable outdoor antenna there is no antenna gain. The reception situation to be
31.4 Measurements in DVB-T Single-Frequency Networks
615
considered is then 0 dB antenna gain and e.g. 2 m above ground. With indoor reception, attenuation losses of the walls and windows of up to about
20 dB must be added. Concrete buildings with metallized windows produce especially high attenuation. Polarization losses of 5 to 15 dB are another factor to be considered. Reception with a vertical rod antenna in a
horizontally polarized DVB-T network, e.g., leads to a loss of about 15 dB.
Portable indoor reception covers a very wide range with respect to minimum field strength. The antenna gain may also exhibit negative values.
The most difficult case is mobile reception, DVB-T being a system which
was originally not designed for this purpose. The time interleaver value is
very short. Although the mobile field strength values measured do no differ from the stationary ones, the Doppler effect plays a very large role and
the changing receiving situations play havoc with the receiver. The signalto noise ratio (SNR) and the modulation error ratio (MER) differ greatly
under mobile and stationary conditions and also depend on location. The
MER is the aggregate parameter in which all interference effects on the
DVB-T reception can be mapped. As explained in Chapter 21, it is the
logarithmic ratio of the RMS value of the signal to the RMS value of the
error vector in the constellation diagram. If only a noise effect is present,
the MER corresponds to the SNR. If the SNR or the MER are measured in
mobile operation, the Doppler effect additionally affects a deterioration of
the MERs or SNRs in dependence on the speed of travel due to the different types of local reception-related effects and signal paths. This will also
be illustrated later in this chapter by providing practical examples from the
exemplary DVB-T networks. The MER is thus measured under stationary
conditions in accordance with the required nominal conditions of reception, e.g. with a directional antenna at 5 or 10 m height, or with a nondirectional antenna. The minimum MER required for reception also depends on the modulation parameters.
Table 31.8. Minimum required MER at code rate = 2/3.
MER
QPSK
6 dB
16QAM
12 dB
64QAM
18 dB
It is important to know that, as a simple fact of physics, the MER measured under mobile conditions can never correspond even approximately to
that measured under stationary conditions. The MER is always, and also in
every standard, a function of the speed of travel and of the multi-path reception conditions. The same also applies to the bit error ratios (BER).
These are also not only dependent on the received level but can be derived
directly from the MER. The minimum required BER before Reed Solomon
616
31 Digital Terrestrial TV Networks in Practice
or after Viterbi with quasi error free DVB-T reception is 2 • 10-4. There are
three bit error ratios in DVB-T:
•
•
•
BER before Viterbi, or the channel bit error rate ratio,
BER after Viterbi or before Reed Solomon, and the
BER after Reed Solomon.
It makes sense to measure all three BERs during the field test. The engineer obtains appropriate information especially from the BER after
Viterbi, i.e. before Reed Solomon. Naturally, the measurement of the
BERs in mobile operation will also result in quite different values in comparison with stationary measurements. The test results also depend on
where in the SFN one is moving, the reason again being simply the effect
of Doppler on the multipath reception in the various regions of reception.
Fig. 31.22. Channel impulse response with one signal path; measured with the TV
test receiver ETL
The channel impulse response (Fig. 31.22.) provides reliable information
about the multi-path receiving conditions in the various regions in the
SFN. It also tells one whether an SFN is running synchronously or not. In
addition, the channel impuls response can be used for estimating whether
the receiving situation could represent critical states with respect to synchronisation to symbol and guard interval. Critical states are:
•
•
•
•
•
violation of the guard interval,
the pre-echo,
the 0-dB echo,
the quasi mobile receiving situation
radiation of different TPS bits.
31.4 Measurements in DVB-T Single-Frequency Networks
617
Fig. 31.23. Channel impulse response with multi-path reception with post-echoes,
measured here in an SFN with 3 transmitters
Fig. 31.24. Typical constellation diagram for a guard interval violation (outer
points are larger than the inner ones)
31.4.1.1 Guard Interval Violation
The guard interval violation, among the critical receiving states, is the
simplest to explain and is considered to constitute an absolute infringement
of the SFN conditions. It simply involves the reception of signal paths
which are outside the guard interval and still have sufficient energy. Such a
problem can arise when transmitters are spaced too far apart and delays
have been selected wrongly or unfourably at the transmitter sites, or simply with propagation overshoots. The energy of such a signal path becomes
critical if it passes from the attenuation with respect to the main path into
618
31 Digital Terrestrial TV Networks in Practice
the order of magnitude of minimum S/N or minimum MER (fall-off-thecliff), depending on the selected transmission parameters.
This problem can be solved by a suitable choice of the delay times, i.e.
the transmission times, of the COFDM symbols to the transmitter sites, by
adapting the antenna patterns and the ERPs of the transmitters. In the case
of a guard interval violation, a typical constellation diagram has larger
constellation points outside than inside (Fig. 31.41.).
Fig. 31.25. Channel impulse response with pre-echoes
Fig. 31.26. Channel impulse response with pre-echo and 0-dB echo
31.4.1.2 Pre-echoes
Pre-echoes are signal paths in a single-frequency network which appear
with a lower level and earlier than the main path (Fig. 31.25.). In theory, it
should be possible for all receivers to handle this situation without any
problem. In practice, however, it is found that many, mainly older DVB-T
receivers cannot manage this. This also depends on the delay time between
the 0-dB path, i.e. the main path, and the reduced pre-echo. The problem
of the pre-echo can be explained simply by the fact that the receiver simply
31.4 Measurements in DVB-T Single-Frequency Networks
619
places the FFT sampling window symmetrically over the main path and
thus pushes the pre-echo beyond the guard interval.
31.4.1.3 The 0-dB Echo
It is called a 0-dB echo if 2 or more signal paths having the same level but
different delays appear at the receiver (Fig. 31.43.). This can also lead to
receiver synchronization problems, mainly in the case of longer delay differences from half the guard interval onward. In theory, a receiver should
be able to cope also with this receiving situation without any problem. This
problem, too, is explained by the receiver placing the FFT window so
badly that one signal path is located outside the guard interval.
31.4.1.4 Quasi Mobile Receiving Situation
It is called a quasi mobile receiving situation if the channel continuously
changes due to continuously changing conditions of reflection. The behavior of a receiver in this situation depends on the characteristics of the
channel correction of the receiver and, naturally, on the receiving situation.
Quasi mobile receiving situations are encountered when e.g. there is no direct line of sight to the transmitters and the reception "lives" mainly from
reflections, but these reflections are influenced by cars, trains, trams etc.
Fig. 31.27. Constellation diagrams in an SFN with differently transmitted TPS bits
31.4.1.5 Transmission of Different TPS Bits
In DVB-T, a total of 67 bits are transmitted as so-called TPS (transmission
parameter signalling) bits via 68 symbols. These bits represent a fast information channel from transmitter to receiver for conveying the transmission parameters. These transmission parameters are, among other
things, modulation method, error protection etc. Apart from the TPS bits
620
31 Digital Terrestrial TV Networks in Practice
already defined originally in the DVB-T standard, there are the reserved
bits, more and more of which are used, e.g., for cell ID and DVB-H.
A length indicator transmitted before the actual TPS payload bits tells
how many of the reserved TPS bits are actually currently being used. It is
important that all transmitters in an SFN transmit all TPS bits identically
and completely synchronously. It has happened a number of times that
transmiters in an SFN were configured differently and had transmitted
length indicator and reserved bits differently. Depending on their location,
the receivers which had then actually evaluated the TPS bits were unable
to cope with the receiving situation and could not lock up. The TPS carriers work with DBPSK modulation, i.e. with differential BPSK modulation.
The information is transferred in the difference from one symbol to the
next. However, this means that from the point in time at which a TPS bit is
transmitted differently than at other transmitter sites, the carrier vectors are
pointing in the opposite direction and the DBPSK modulation no longer
works, causing the circular distortions in the constellation diagram at the
TPS points (see Fig. 31.27.) It is strongly recommended always to have
one or more test vehicles in the field for all changes being carried out in an
SFN and to determine the situation also at the TPS carriers (e.g. carrier No.
50).
31.4.1.6 Frequency Accuracy of the Transmitters
It is important that all transmitters in an SFN transmit at the same frequency, as accurately as possible. The accuracy to be aimed for is 1 • 10-9
or better. The frequency accuracy can be easily verified with a suitable test
receiver by measuring the impulse response. This provides a frequency accuracy of somewhat better than 0.5 Hz, a condition which normally can be
easily met.
31.4.2 Practical Examples
31.4.2.1 Pre-echoes
The pre-echoes described occur mainly in regions in which the closer path
in terms of distance is attenuated more compared with the 0-dB path due to
geographical obstacles (hills, mountains). In the region of the SFN Southern Bavaria described this occurs, e.g, to the North of the Munich airport
in the vicinity of the course of the river Isar, where the Olympia Tower is
shaded by hills and more distant Wendelstein transmitter thus dominates.
31.4 Measurements in DVB-T Single-Frequency Networks
621
33.4.2.2 The 0-dB Echoes
0-dB echoes occur whenever two or more paths appear with the same
power level at the receiver due to the propagation conditions. In the SFN
Southern Bavaria, such a situation occurs mainly in the region of the
"Erdinger Moos" around Munich airport. It is very flat there and the
Wendelstein and Olympia Tower transmitters are received partly with the
same level, but with an extreme delay difference of about 140 s.
31.4.2.3 Quasi Mobile Channel
A quasi mobile channel exists in regions where there is no direct line of
sight to the transmitters. This is the case where the transmitters are
shielded by obstacles and the reception survives with reflections partly
from "variable" obstacles such as cars, trains or trams.
31.4.2.4 TPS Bits
When SFNs are commissioned or re-organized, it may happen that not all
transmitters (of one or of different transmitter manufacturers) are identically configured and that the transmitters thus transmit different TPS information. This occurred several times during the commissioning or conversion of the SFN.
Fig. 31.28. Spectrum of a DVB-T signal in the AWGN channel
622
31 Digital Terrestrial TV Networks in Practice
31.4.2.5 Mobile DVB-T Reception
A question often asked is "Up to what speed does DVB-T work?", a question which is not easily answered. In principle, it must be said at this point
that DVB-T was never intended for mobile reception and, therefore, does
not have any characteristics especially provided for this purpose succh as,
e.g., a long time interleaver. Mobile reception depends mainly on the
multi-path receiving situation. If only one signal path is received, mobile
reception does not present a problem. The Doppler effect then only shifts
the DVB-T spectrum in the direction of higher or lower frequencies, depending on whether one is moving towards the transmitter or away from it.
At the usual travelling speeds, the frequency shift is of the order of 50 to
100 Hz. This frequency shift does not present a problem for DVB-T receivers receiving one signal path. In the case of multi-path reception and
Doppler shift, the problem is one of spectrum smearing with all possible
intermediate stages which will be presented in examples in the following
paragraphs.
Fig. 31.29. Single COFDM carrier in an AWGN Channel
AWGN channel
In the AWGN channel, the carriers are only affected by noise, as shown in
Fig. 31.28. The noise pedestal at about 20 dB below the payload signal can
be seen clearly at the shoulder.
31.4 Measurements in DVB-T Single-Frequency Networks
623
The single COFDM carriers are at exactly the right frequency positions in
the AWGN channel (Fig. 31.29.). Each carrier is only affected by a greater
or lesser "noise fringe".
Fig. 31.30. Single frequency-shifted DVB-T carrier in a mobile channel
Fig. 31.31. Stationary multi-path reception of two signal paths (0 dB / 0 s, -5 dB
/1 s)
624
31 Digital Terrestrial TV Networks in Practice
Doppler Shift
During movement in the mobile channel, the complete DVB-T signal is
frequency shifted. All single COFDM carriers are shifted towards higher
or lower frequencies depending on whether one is moving towards the
transmitter or away from it. Fig. 31.30. shows a single carrier, shifted by
70 Hz, of a DVB-T signal at a speed of 150 km/h moving towards the
transmitter.
0 dB
Tx1
v=150 km/h
Tx2
-10 dB
Fig. 31.32. Mobile multi-path reception of two signal paths (0 dB and –10 dB at
150 km/h)
Stationary Multi-path Reception
In stationary multi-path reception the only problem is fading. Depending
on the difference in echo delay and echo attenuation, more or less deep
dips occur in the signal spectrum as shown in Fig. 31.31. The spacing of
the dips corresponds to the inverse of the echo delay difference.
Mobile Multi-path Reception
In mobile multi-path reception, the DVB-T subcarriers are shifted simultaneously upwards and downwards in frequency (Fig. 31.32.) or may not be
shifted at all. Depending on the receiving conditions, this frequency smearing results in unwanted amplitude modulation of the DVB-T signal.
31.4 Measurements in DVB-T Single-Frequency Networks
625
Mobile Rice Channel
The model of the Rice channel simulates the case of multiple multi-path
reception and dominant main path. Fig. 31.34. shows the spectrum of a
single DVB-T carrier in the mobile Rice channel. The dominant main
channel can be clearly seen at -10 dB.
Fig. 31.33. DVB-T constellation diagram with unwanted amplitude modulation
caused by mobile multi-path reception (500 MHz, 3 paths, -20 dB/-150 km/h,
0 dB/0 km/h, 150 km/h/-20 dB)
Fig. 31.34. Mobile Rice channel, v=150 km/h, power ratio = 10 dB
626
31 Digital Terrestrial TV Networks in Practice
Rayleigh Channel
In the Rayleigh channel there is no longer a main path. It corresponds to
the Rice channel with a power ratio = 0 dB. Fig. 31.35. shows an example
of a single DVB-T carrier in the mobile Rayleigh channel at a speed of 150
km/h.
Fig. 31.35. Mobile Rayleigh channel, v=150 km/h, power ratio = 10 dB
Fig. 31.36. Unwanted amplitude modulation caused by a dried-out electrolytic capacitor in an antenna amplifier
31.4 Measurements in DVB-T Single-Frequency Networks
627
Comparable "mobile situations" can be created in a DVB-T receiver
even by dried-out electrolytic capacitors in an antenna amplifier. Superimposed AC hum produced by these can create an unwanted amplitude
modulation at 50 or 100 Hz. Fig.31.36. shows a corresponding constellation diagram.
31.4.3 Response of DVB-T Receivers
The response of DVB-T receivers in one or the other receiving situation is
greatly dependent on the characteristics of the respective receiver, i.e.
mainly on the characteristics of the installed tuner, the DVB-T chip, the
MPEG decoder and the firmware of the receiver. In the next section, testing of the receiver will be discussed. The characteristics of the tuner can be
differentiated as follows:
•
•
•
•
Noise figure
Phase noise
RF and IF selectivity characteristics
Linearity/intermodulation
The characteristics of the tuner essentially determine the minimum received level required and the adjacent-channel compatibility, especially
with a high adjacent-channel level.
The DVB-T chip is mainly responsible for how well a receiver can handle different receiving situations such as
•
•
•
•
•
•
•
Pre-echo,
0-dB echo,
Multi-path reception in general,
Mobile reception and quasi mobile reception,
Adjacent-channel occupancy,
TPS bits set differently,
Hierarchical modulation.
The MPEG decoder and the firmware determine how the receiver responds
to different transport stream contents. This relates to:
•
•
the channel search (speed and characteristics under critical conditions,
the PSI/SI tables (e.g. response to dynamic PMT),
628
31 Digital Terrestrial TV Networks in Practice
•
•
•
•
•
•
•
response to network overlap (identical service in different TS)
decoding of the elementary stream,
signalling of the source characteristics (4:3/16:9, mono/stereo),
error concealment,
switching rate,
stability,
receiver configuration such as teletext, VPS, MHP.
31.4.4 Receiver Test and Simulation of Receiving Conditions in
Single-Frequency Networks
The characteristics of TV receivers must be tested comprehensively especially in terrestrial broadcasting in order to find out how well they are capable of handling the problem situations described in the previous sections.
Receivers are tested in
•
•
•
•
the development of receivers,
production handover (EMI, EMC,...),
receiver production for final testing,
comparing receivers in test houses and at network operators.
Experience has shown that DVB-T receivers especially have not been adequately "stress tested". The maximum amount of tests should be performed
at least during the receiver development, the production handover and the
receiver comparison. These maximum tests are:
•
•
•
•
•
•
•
•
•
•
•
•
•
detecting the minimum receiver input level at some frequencies,
detecting the minimum SNR at some frequencies,
the response at high adjacent-channel levels (close or more distant),
the response with co-channel reception of analog TV,
the response of the receiver during channel search,
the response of the receiver with network overlap,
measuring the booting speed,
measuring the switching speed,
testing the teletext function,
testing the VPS function,
testing the response of dynamic PSI/SI tables,
testing the firmware configuration and quality,
EMC tests,
31.4 Measurements in DVB-T Single-Frequency Networks
•
629
mechanical construction.
The minimum tests in production must be suitably selected for the respective product by the manufacturer.
33.4.4.1 Minimum Receiver Input Level in the AWGN Channel
The minimum receiver input level should first be determined in the
AWGN channel with different transmission parameters (64QAM, 16QAM,
QPSK, different code rates) at least 3 frequencies (one VHF and two UHF
frequencies). The receiver is supplied with one signal path in this case.
Starting with a level of abut 50 dB V, this level is reduced until the visual
and aural assessment of decoded video and audio shows that the receiver is
no longer operating correctly. It is important that, when the precise point
when the "fall-off-the-cliff" occurs is determined, one always waits for a
sufficiently long time (at least 1 minute) to see whether the receiver is
really still operating in a stable mode.
31.4.4.2 Minimum SNR
Apart from determining the minimum receiver input level, it is of interest
to determine the minimum signal/noise ratio in the AWGN channel. The
results should then be compared with the minimum receiver input level
measurement and discussed. These tests should also be performed at at
least 3 frequencies (the same ones as in paragraph 3.4.4.1, of course), selecting, e.g., a "sensible" DVB-T receiver input level of 50 to 60 dB V so
that the receiver is neither supplied too poorly nor caused to go into attenuating mode. More and more noise is then added progressively until the
"fall off the cliff" condition is reached again. This, too, is then determined
carefully as in 31.4.4.1.
31.4.4.3 Adjacent-Channel Occupancy
In the adjacent-channel test, the response of a DVB-T receiver with a high
adjacent-channel level is determined placing an adjacent DVB-T channel
below or above a payload channel. The level of the adjacent channel or
channels is then increased more and more until no further reliable reception is possible. This test, too, is performed with different transmission parameters as in 31.4.4.1 and 31.4.4.2. The aim should be to be able to handle an adjacent-channel level which is at least 20 dB above the useful
level. Such conditions could easily arise especially with a mixed DVB-
630
31 Digital Terrestrial TV Networks in Practice
T/DVB-H/DAB scenario. This test should also be performed at 3 frequencies, at the least.
31.4.4.4 Co-channel Reception
Checking the co-channel reception of DVB-T with DVB-T is essentially
already done by 31.4.4.2 since a non-synchronous DVB-T interference
signal virtually looks like noise. Testing with analog TV in the co-channel,
however, is definitely a noteworthy measurement but depends greatly on
the ATV image content chosen.
31.4.4.5 Multi-path Reception
In the multi-path reception test, the receiver is presented with a situation
which can occur in real life in an MFN or SFN, using a test transmitter
with a channel simulator (fading simulator).
31.4.4.6 Channel Search
Testing the channel search function of a receiver mainly tests the search
rate and also the search action under different conditions (incl. adjacentchannel occupancy). The test should also involve checking of the performance of a receiver with wrong NIT entries. This also includes its performance in the case of network overlaps, i.e. when receiving the same services
from different transport streams. This occurs in regions where the receiver
is seeing two or more networks, i.e. at the edges of SFNs.
31.4.4.7 Booting Speed and Action
In the world of computers, "booting" is known to be the initialization of a
computer. Since a DVB-T receiver is also nothing else than a computer, it
takes a certain time until it is ready for operation and a user will be interested to know how long this will take and how it takes place.
31.4.4.8 Program Change
Users find it particularly bothersome if a program change takes a long time
and is "untidy". This test checks the receiver reaction to "zapping".
31.4.4.9 Teletext
DVB provides for the "tunneling" of teletext via private PES packets and
this is done mainly by the program providers operating under public law.
31.4 Measurements in DVB-T Single-Frequency Networks
631
In this arrangement, teletext is gated back into the vertical blanking interval of the video signal by the receiver at the analog output interface
(SCART or cinch connector). A TV receiver connected there can then decode this teletext. It is also possible for the DVB-T receiver itself to decode the teletext and to output it as a frame signal, storing a number of
pages in its buffer. The teletext modes supported by a receiver, either gated
into the vertical blanking interval or self-decoded, are a criterion for testing and comparing receivers.
31.4.4.10 VPS Functions
In data line 16 in the vertical blanking interval of the analog TV signal, the
VPS information for controlling video recorders has hitherto been transmitted, among other items. This signal, too, can be "tunneled" in DVB in
private PES packets in the MPEG-2 data stream and used directly in the
receiver (hard disk receiver) and/or gated back into line 16 at the CCVS interface. A video recorder connected there can then respond to this signal
and control the recording. These functions, too, must be covered in a receiver comparison test.
31.4.4.11 Dynamic PSI/SI Tables
Dynamic PSI/SI tables means the change of these tables with time. EIT
and TOT/TDT are clearly always dynamic but there are also so-called
"window programs" which are transmitted only at particular times of the
day and are signalled by changing PMTs, so-called dynamic PMTs. A
change in the PMT is not recognized by all receivers which is why the response to changes in PAT, PMTs and the SDT should be tested.
31.4.4.12 Firmware Configuration
The way in which a DVB receiver can be operated and how especially the
electronic program guide is handled depends greatly on the firmware installed in the receiver. This is another matter to which attention should be
paid in a receiver comparison test.
31.4.4.13 Miscellaneous
Naturally, the receiver test also includes adherence to the EMC regulations
but this will not be discussed in greater detail at this point. As well, an assessment of the mechanical construction of the receiver case is of importance in a comparison of receivers in test establishments but this again will
not be discussed any further here.
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31 Digital Terrestrial TV Networks in Practice
31.5 Network Planning
Naturally, a network expansion is preceded by network planning which today is done with the support of software tools. This involves simulation
and determination of the network data such as antenna patterns, transmitter
powers, error protection. guard interval, delays etc and calculation of the
coverage of the regions on the basis of geographical, topographical and
morphological data and with knowledge of the possible transmitter sites.
Firstly, the frequencies and powers, direction of radiation etc are assigned
by the regulatory authorities. In border regions this requires international
coordination. Examples of planning tools in the German-speaking area are:
•
•
•
tools by the Deutsche Telekom,
tools by the Institut für Rundfunktechnik (IRT),
tools by the company LStelcom.
The software CHIRplus_BC© by the company LStelcom (Lichtenau near
Baden-Baden, Germany), in particular, is encountered throughout the
world. By now, the author also has gained the experience that with the appropriate use of planning tools, the problem areas described above for receivers (0-dB echo, pre-echoes) can be unambiguously identified, e.g. by
clicking on corresponding buttons in the planning software.
31.6 Filling the Gaps in the Coverage
Even in the days of analog television, gaps in the coverage were normally
filled by so-called gap fillers, called translators. In analog television, alternative channels with guard band without adjacent-channels being occupied
were selected which supply a signal received from a master transmitter for
covering a limited region. In this set-up, TV signals of a master transmitter, received by rebroadcasting reception, were translated into another TV
channel and then retransmitted via a transmitter, thus covering a region
which was otherwise shaded. In digital terrestrial television it is firstly assumed that many areas are automatically covered by the characteristics of
digital television. Nevertheless, depending on the required coverage which,
in turn, is dependent on the country concerned, additional gap fillers cannot be avoided. This is because, in analog television, reflections have not
only led to reception not being possible at all, but in moderate cases have
simply caused unsightly "ghost images". There are no longer any "ghost
images" in digital television and, because of COFDM, neither do echoes
31.6 Filling the Gaps in the Coverage
633
present problems to the same extent as in analog television. In theory, a
COFDM system should be able to handle such a situation quite easily, of
course. But if the received field strength is too low because of shading,
such regions must still be covered by gap fillers even with digital terrestrial television. In digital television, these gap fillers can be operated both
at the same frequency and at other frequencies. In analog television, these
transmitters had to be operated at other frequencies. There are:
•
•
•
transmitters transmitting at the same frequency (gap fillers, SFN),
and
frequency-converting transmitters (transposers, MFN), and
frequency-converting transmitters with remodulation (retransmitters, MFN).
Reception
antenna
Transmission
antenna
BP
Signal back-coupling for
for measurement and control purposes,
and at a gap-filler for
echo-cancellation
Exciter
BP
Exciter: rebroadcasting at the same frequency,
with or without echo-cancellation, or
frequency-translating or a remodulating
Fig. 31.37. Principle of a gap-filler or translator
To fill gaps in the coverage of SFNs, only gap fillers are used. In these
transmiters transmitting at the same frequency, remodulation is impossible
since going back to data stream level (demodulating) and remodulating
would involve too much delay. This is why in this case the approach of
downconverting to a low intermediate frequency and upconverting again to
RF and amplifying was adopted. It is important here that receiving and
transmitting antenna must be sufficiently well decoupled. Up to a certain
extent, an equalizer can be of assistance here in providing echo cancella-
634
31 Digital Terrestrial TV Networks in Practice
tion. The minimum isolation necessary between receiving and transmitting
antenna is (current state of the art):
•
•
+ 10 dB gain without echo cancellation,
- 10 dB gain with echo cancellation.
Fig. 31.38. Practical example of a translator or gap-filler site; the 2 (log periodic)
receiving antennas are located in the lower part of the extended pinnacle of the
tower, the 3 transmitting antennas (8-element bays) are located in its upper part
Apart from the transmitting antenna and the receiving antenna being sufficiently well decoupled, the correct orientation of the receiving antenna is
31.7 Fall-off-the-Cliff
635
also of great importance. If possible, only one signal path, and not several
as in a multi-path situation, should be forwarded from the SFN. Under no
circumstances should pre-echo situations or a 0-dB echo path be radiated
since this will lead to problems in many receivers as is well known. Otherwise, receiver problems are created not only over a small area at some
locations but over a large area in the gap-filling region and possibly beyond. In the case of a frequency-converting transmitter, the approach via
the IF can be selected as in the case of the gap filler, or one can choose a
remodulation process. Remodulation is more expensive and means delay,
of course. This is also the reason why remodulation is not possible in the
gap filler because otherwise the SFN timing would be violated completely.
However, remodulation is more stable and, above all, results in a better
signal quality. When the transmitter powers are greater, however, the recommended approach is always that of remodulation or of using the retransmitter, respectively.
Fig. 31.39. „Fall off the Cliff” artifacts in digital television
31.7 Fall-off-the-Cliff
Blocking artifacts caused by the compression are too often mistaken for artifacts caused by the transmission link. An image at the output of a re-
636
31 Digital Terrestrial TV Networks in Practice
ceiver which, due to bit errors, has been brought to the limit of decodability, i.e. the "fall-off-the-cliff" state, looks quite different from an image
which has been rendered "unsightly" due to too much MPEG compression.
In the case of bit errors, entire slices are missing or the entire image
freezes or no image can be seen at all. Fig. 31.39. shows an image in which
entire groups of blocks, so-called slices, are missing within a line.
31.8 Summary
The empirical values described in this chapter with reference to DVB-T
can also be easily applied to other terrestrial transmission standards. The
details in error protection and the modulation methods are different in
DAB, ISDB-T or other standards, but the principle always remains the
same. Although this chapter is thus tailored for DVB-T, due to the author's
experience in this field, it is not limited to DVB-T alone. In this section,
experiences and problems from real networks were presented. It must be
hoped that these experiences will assist in recognizing, and solving, most
of the real problems with DVB-T, and in removing the apprehension felt
about the new digital television. Digital television differs from analog television but, with the appropriate experience, it is not unfathomable.
Bibliography:
[NX7000],
[KATHREIN1],
[VIERACKER], [LSTELCOM], [NX8000]
[LVGB],
[RFS],
32 DTMB
32.1 DMB-T, or now DTMB
DTMB - Digital Terrestrial Multimedia Broadcasting - is a Chinese standard which, like DVB-T - has the aim of broadcasting television economically terrestrially by digital means and with modern supplementary services. DMB-T was published in 2006 – at least in excerpts, as “GB206002006 – Framing Structure, Channel Coding and Modulation for Digital
Terrestrial Broadcasting System”. It was renamed DTMB, having combined two proposals to form one standard in 2007. In one proposal, a multicarrier method was stipulated, in the other one a single-carrier method is
suggested. The favoured proposal of the multi-carrier method comes from
Tsinghua University in Beijing and was called DMB-T for a long time.
The single-carrier method is called ADTB-T and originates from Jiaotong
University in Shanghai. DTMB has similarities with DVB-T whilst
ADTB-T is derived from the North American ATSC.
Multi carrier
TD-COFDM
Tshinghua University
Beijing
„DMB-T“
Single carrier
modulation
Jiaotong University
Shanghai
„ADTB-T“
„DTMB“
Digital Terrestrial
Multimedia Broadcasting
Fig. 32.1. Joining two proposals to form DTMB
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_32, © Springer-Verlag Berlin Heidelberg 2010
638
32 DTMB
32.2 Some more Details
PN sequency
PN sequency
It is used single carrier modulation and TD-COFDM (Time Domain Coded
Orthogonal Frequency Division Multiplex), among other things. In multicarrier mode the guard interval is here not filled with the end of the next
symbol following but with a PRBS. This symbol preamble is called frame
header and has a length of 56.6 s, 78.7 s or 125 s with a channel bandwidth of 8 MHz. In multicarrier mode DTMB runs in 4K mode with 3780
used carriers which are spaced apart at 2 kHz in the 8 MHz channel. The
symbol period is therefore 500 s. 3744 of these 3780 carriers are modulated data carriers and 36 are signalling carriers, i.e. virtually TPS carriers.
DTMB supports channel bandwidths of 8, 7 and 6 MHz. The useful spectrum is 7.56 MHz wide in the 8 MHz channel. The net data rate is between
4.813 Mbit/s and 32.486 Mbit/s. The spectrum is roll-off filtered with a
roll-off factor of r=0.05. The transmission method is intended for SDTV
and HDTV transmissions and should work both in stationary and in mobile
operation. It is possible to implement both MFN and SFN networks.
Symbol n
Symbol n+1
Guard interval
Fig. 32.2. DTMB TD-COFDM
The following can be selected as modulation methods on the 3744 data
carriers:
•
•
•
•
•
64QAM
32QAM
16QAM
4QAM
4QAM=NR (Nordstrom Robinson).
Time interleaver
BCH coder
Scrambler
TS
in
LDPC coder
32.2 Some more Details
Data
out
Fig. 32.3. DTMB Forward Error Correction
4K mode:
3780 carriers
f=2kHz
f
f
Channel bandwidth
6, 7, 8 MHz
Fig. 32.4. Characteristics of the DTMB multi-carrier mode
The error protection in DTMB (Fig. 32.3.) consists of a
•
•
•
•
Scrambler
BCH coder
LDPC coder
Time interleaver
639
640
32 DTMB
The DMB-T signal is made up out of a
•
•
•
•
Signal frame (frame header + frame body = virtually guard +
symbol)
Super frame = N1 • signal frame
Minute frame = N2 • super frame
Calendar day frame = N3 • minute frame
As in other transmission methods too, the input signal of a DTMB
transmitter is the MPEG-2 transport stream.
Unfortunately, it is difficult to provide more details of DTMB. Not
much has been published and not all of the published papers appear to
agree with one another, either. At this stage it appears to be prudent to say
nothing rather than to provide false information. It is not really clear what
advantages are to be gained by a guard interval filled with a PN sequence.
The only thing that is clear is that licensing rights have moved towards being less binding with regard to DVB and ATSC and that some details of
standards may well have something to do with this. Neither is it clear from
where and to what purpose the roll-off characteristic has been introduced.
The roll-off characteristic in multi-carrier mode may well come from the
guard interval filled with a PN sequence (single-carrier method in the
guard interval?).
Fig. 32.5. DTMB spectrum
Bibliography: [DTMB]
33 Return Channel Techniques
Return channels have been provided for some time in digital television,
having been defined both in DVB-T, in DVB-S and in DVB-C (DVBRCT, DVB-RCS and DVB-RCC, resp.), but only the cable return channel
has gained any real significance. It serves to provide rapid Internet access
by broadband cable. There are two standards for return channel arrangements in cable TV: DVB-RCC/DAVIC or DOCSIS/EURO-DOCSIS. All
the networks known to the author in Europe use EURO-DOCSIS for the
return channel which is located here within the 5 - 65 MHz band. To implement the return channel, this frequency band was completely emptied,
i.e. the channels located there were moved into the special-channel band.
Today, both the traditional telephone companies and TV cable network
operators are promoting "Triple Play", i.e. everything out of one socket telephone, television and the Internet. This combination can also bring
with it a distinct price advantage for the end user.
DOCSIS stands for "Data Over Cable Service Interface Specification"
and has its origin in the US. It has also been standardized by ETSI in the
[ETSI201488] document and is provided in several versions. Communication with the cable modem takes place both in the downstream and in the
upstream direction. Downstream is a continuous MPEG-2 transport stream
transmitted physically via ITU-TJ83B (DOCSIS) and ITU-TJ83A (EURODOCSIS). The modulation method used there is 64QAM or 256QAM.
Downstream, in the frequency band between 5 and 65 MHz, is divided into
burst packets (time slots). The modulation method used there is either
QPSK or 16QAM. Due to man-made ingress noise, the frequency band between 5 and 65 MHz is not uncritical and the house cabling must be installed with the appropriate care. Thanks to the HFC (hybrid fibre coax)
networks provided in most cases, where only the last 1000 meters are still
implemented in coax, the distribution network has become quite uncritical.
By now, the optical fibre cable can also be run right into one's house. Ingress noise can be measured by means of a spectrum analyzer by comparing the burst peaks in the return channel with the burst intervals
(Fig.33.1.).
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_33, © Springer-Verlag Berlin Heidelberg 2010
642
33 Return Channel Techniques
Fig. 33.1. Example of a return channel spectrum (internet and telephony) (picture
supplied by UPC Telekabel Klagenfurt, Austria)
Bibliography: [ETSI201488], [ETSI300800], [KELLER]
34 Display Technologies
The cathode ray tube (CRT) has long been dominant as an essential electronic component both on the recording side and on the reproduction side.
It forms the basis for many parameters and characteristics of a video baseband signal (composite video signal) such as, e.g., the horizontal and vertical blanking interval and the interlaced scanning method for reducing
flicker, none of which would have been necessary with the new technologies where, in fact, they prove to be troublesome.
Before the CRT, however, there had already been attempts, since 1883,
in fact, to transmit images electronically from one place to another. Paul
Nipkow had invented the rotating Nipkow disk which provided the stimulus for thinking about sending picture information - moving picture
information, what's more - from place to place. The Nipkow disk already
slices images into transmittable components which are basically lines.
Present-day modern display technologies are pixel-oriented, in so-called
progressive scanning, i.e. without line interlacing, and exhibit distinctly
higher resolutions. Today, we have basically
•
•
•
partly still the "old" cathode ray tube (CRT),
the flat screen, and
projection systems such as beamers and back-projectors.
The underlying technologies for these are
•
•
•
•
•
the cathode ray tube (CRT),
liquid crystal displays (LCD),
plasma displays,
digital micro-mirror chips (DLP=Digital Light Processing chips),
and
organic light-emitting diodes (OLEDs).
Video compression methods such as MPEG-1 and MPEG-2 had still
been developed for the cathode ray tube as reproduction medium at the beginning of the 90s. On a CRT, MPEG-2-coded video material still looks
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_34, © Springer-Verlag Berlin Heidelberg 2010
644
34 Display Technologies
quite tolerable even at relatively low data rates (below 3 Mbit/s), whereas
it produces clearly visible blocking effects, blurring and so-called mosquito noise (visible DCT structures) on a large modern flat screen.
MPEG-4 video, however, still looks clean on these modern displays, even
at distinctly lower data rates. Interlaced material must be de-interlaced before being displayed on flat screens if it is not to result in disturbing "line
feathering" (Fig. 34.10.). This can be seen most clearly in the case of continuous caption (text) inserts. New technologies result in new problems
and artefacts. The new effects in image reproduction, caused by the reproduction system, are:
•
•
•
•
•
•
•
burn-in,
resolution conversion problems,
motion blur,
the appearance of compression artefacts,
phosphor lag,
dynamic false contours,
and the rainbow effect.
In this chapter, the operation of the various display technologies and
their characteristics and the artefacts which occur, and the respective attempts at compensating for them, will be described.
Video 60 Hz
Film 24 Hz
A
A odd
A even
Video 50 Hz
Film 25 Hz
A odd
A
A odd
B
B even
B odd
B
B odd
C
C even
C
C even
B even
C odd
C even
C odd
D
A even
D
D odd
D even
D odd
D even
3:2 pulldown
2:2 pulldown
Fig. 34.1. Conversion from cinema-film to video (3:2 and 2:2 pulldown)
34.1 Previous Converter Systems - the Nipkow Disk
645
The source material for many films transmitted on television was, and
still is, original cine-film. This is produced with 24 frames per second, independently of where it is produced. When it is reproduced in television, it
must be adapted to the 25 or 30 frames per second used there and the frames must be converted into fields in the line interlace method (Fig. 34.1.).
In the 25 frames/second technology, the films are simply run at 25 frames
per second instead of at 24 frames per second and each frame is scanned
twice by the film scanner, with half a video line offset in each case. The
film is thus running imperceptibly slightly faster in television than in the
cinema. This type of conversion does not present any problems and does
not lead to any visible conversion artefacts. In the case of the 30 (59.94)
frames/second standard, a so-called 3:2 pull-down conversion takes place
(Fig. 34.1.). Instead of 24 frames per second, the film is here run at 23.976
frames per second, i.e. by 0.1 % more slowly, and 4 frames are then
stretched to 5 frames using the line interlace method. For each frame, 2 or
3 fields are then generated alternately. Such converted image material can
be juddery and is not easily converted again. A conversion from 30 to 25
frames/s is called an inverse 3:2 pull-down conversion. All this also affects
the image processing in the respective TV displays.
Fig. 34.2. Nipkow disk
34.1 Previous Converter Systems - the Nipkow Disk
Paul Nipkow had the idea of splitting images into "lines" back in 1883, using a rapidly rotating disk both as pickup and for replaying purposes. The
646
34 Display Technologies
Nipkow disk had holes drilled spirally into the disk so that an original picture was scanned by the holes "line by line". In front of the Nipkow disk
(Fig. 34.2.) there was an optical system which projected the original picture onto a selenium cell located behind the disk. This selenium cell was
then supplied virtually line by line with the luminance variation of the
original picture. At the receiving end, there was a reverse arrangement of a
controllable light source and a replay disk, rotating in synchronism with
the pickup disk, plus replay optics. Nipkow disks were used until well into
the 1940s. The greatest problem was the synchronization of pickup and replay disks. The first trials were conducted with disks mounted on a common axis.
Televisors based on John Logie Baird's principles also used Nipkow
disks and were used in so-called television rooms until the 30s. The narrow-band television signal was transmitted by wire or wirelessly by radio.
Demonstration models still available today (see, e.g., NBTV - Narrow
Band Television Association) show that the signals used then can be handled by narrow-band audio transmission channels. The devices used line
numbers of about 30 lines per frame. To simplify synchronization, synchronous motors were used which virtually tied the system to the alternating mains frequency of 25 or 30 Hz. It is interesting to note that the link of
the frame rate with the AC frequency of the respective national grid has
been maintained to the present day. To synchronize transmitter and receiver, synchronization signals were additionally keyed into the narrowband video signal which is similar to a current composite color video signal. Modern Nipkow disk demonstration models use black/white line patterns on the disks in conjunction with phase locked loops. The signals derived from these are compared with the synchronization pulses and the
error signal is used for correcting the speed of rotation of the disk. Between the sync pulses there is a linearly modulated narrow-band video signal which controls the brightness of the replay element (an LED today, a
neon tube in those days) while the respective associated hole in the Nipkow disk scans the visible screen area. The image quality of these simple
systems does not require much comment but the principle employed is still
fascinating even for the experienced video specialist. It simply illustrates
the basics and the history of television technology. Nipkow disk equipment
operates in accordance with the principle of stroboscopy, i.e. of a shorttime display without image retention, just like its successor, the cathode
ray tube.
34.2 The Cathode Ray Tube (CRT)
647
34.2 The Cathode Ray Tube (CRT)
Cathode ray tubes (CRT) are based on the principle of the Braun tube in
which an electron beam which is deflected by two orthogonal magnetic
fields and the intensity of which can be controlled by a grid writes an image line by line onto a luminous phosphor coating on the back of the display screen (Fig. 34.3. and 34.4.). The first display tubes, but also the first
camera tubes, were monochrome devices. The camera tubes operated inversely, i.e. the electron beam in these read out an optical storage layer line
by line. On the pickup side, the cathode ray tubes have been replaced by
charge-coupled devices, so-called CCD chips, back in the 1980s. On the
display side, they have been dominant until about 2005. Today, there are
virtually only the "new screen technologies" in existence. All the characteristics of an analog video baseband signal, the so-called composite colour
video signal, are based on the cathode ray tube.
Magnetic deflection system
Focus
G3
Cathode
Elektron
beam
Heater
G2
G1
Wehnelt Brightness
cylinder
Anode
20kV
Electron beam gun
Fig. 34.3. Cathode ray tube with magnetic electron beam deflection
Horizontal and vertical blanking intervals provided for beam retrace
blanking and this could only be done in finite time. This is because magnetic fields can only be re-polarized within a finite time with manageable
energy consumption. To reduce the flickering effect, the frame was additionally split into fields, i.e. into odd- and even-numbered lines which were
then reproduced with an offset. It was thus possible to create virtually a
648
34 Display Technologies
sequence of fields with twice the number of images (50 or 60 fields) from
25 frames or 30 frames per second. Persistence and filtering characteristics
of the controllable deflected electron beam were significant components of
the display characteristics. The first video compression methods (MPEG-1
and MPEG-2) used in most cases to the present day were developed with
these reproduction technologies and for these reproduction technologies.
However, the monochrome screens differed from the color CRTs in significant details. In the color CRTs, there is a so-called shadow mask located immediately behind the luminous phosphor coating; at the point
where they pass through the slotted mask, the three Red, Green and Blue
beams intersect if the convergence is set correctly.
Anode 20 kV
Electron beam gun
Magnetic deflection unit
Degaussing coil
Fig. 34.4. Cathode ray tube (CRT)
I.e., in the case of the color CRT screens, there was already virtually a
pixel structure in existence. This pixel structure can also be seen if one approaches a CRT-type monitor very closely (Fig. 34.5.). If it was still very
complicated to establish convergence, i.e. the targeting precision of the
three electron beams with the delta-type shadow mask, it was much simpler with the slotted-mask or in-line tube. The three electron guns were arranged here in one row (Fig. 34.6. right) so that the beam systems only had
to be brought into convergence in the horizontal plane. From more than 30
controls in delta-type shadow mask systems, the convergence adjustments
were reduced initially to just a few and later to none in slotted-mask or inline tubes.
34.2 The Cathode Ray Tube (CRT)
649
Fig. 34.5. Pixel structure of a slotted, in-line-type shadow mask tube
Fig. 34.6. Electron beam gun of a delta-type (left) and in-line-type (right) shadow
mask tube
Picture tube monitors have completely different characteristics from
modern flat displays. The CRT monitors operate
•
•
•
with line interlace (fields and frames),
with low-pass filtering due to their physics,
stroboscopically, i.e. with short-time reproduction.
For these reasons, they do not show certain artefacts which become visible in connection with compressed image material on modern flat
650
34 Display Technologies
screens. And these artefacts on modern flat displays are noticed not only
by the experts; it is mainly the consumer, the viewer, who notices these
unsightly image patterns which are now visible whereas they were not on
picture tubes.
Fig. 34.7. Delta-type (left) and in-line-type (slotted) shadow mask (right)
Fig. 34.8. Deflection systems with delta-type (left) and in-line-type (slotted)
shadow mask (right)
This will increase the pressure on the program providers to progress
more quickly towards HDTV, the high-resolution television which will be
broadcast in any case in the new MPEG-4 AVC technology. In summary,
it can be said that the picture tube was very tolerant - it did not show things
which we can now see. It even filtered out most of the noise. But the picture tube had been developed for television with simple resolution, SDTV
- standard definition television. And now it is to be hoped that we will
soon really enter the age of high-definition television - HDTV. In the
good old "picture tube TV", the power consumption depends on the active
image content. The more light there is to be displayed, the more current is
consumed.
34.3 The Plasma Screen
651
34.3 The Plasma Screen
A plasma screen operates in accordance with the principles of a gas discharge in a more or less completely evacuated environment. A gas discharge of a so-called plasma - an ionised, rarefied air mixture such as, e.g.
in a fluorescent tube, is ignited by a high voltage, producing ultraviolet
light. Coloured light can be generated by converting this ultraviolet light of
the gas discharge by means of the appropriate phosphors. Each pixel of a
plasma screen consists of three chambers of the colours Red, Green and
Blue. The three colours can be switched on or off by applying a high voltage and igniting the corresponding cell or can also be switched on or off
simultaneously. The deciding factor is only that they can only be switched
on or off and not incrementally more or less. To be able to achieve gradations in the respective brightness levels of the colours, a trick has to be
used: the respective Red, Green or Blue cell is fired only for a certain
length of time, i.e. applying pulse duration modulation. Short firing means
darker, long firing means brighter. This principle can also lead to display
artefacts (phosphor lag, wrong colours due to different response times of
the phosphors). The plasma screen differs from the picture tube in this respect. They are very similar, however, in respect to their energy consumption. More light means more power and the energy consumption is thus
dependent on the active picture content both in the picture tube TV and in
the plasma screen TV. For a long time, large displays could only be implemented in plasma technology. Current indications are that this technology is fading into the background again in favour of LCD screens. The essential characteristics of plasma screens are:
•
•
•
•
•
•
•
•
•
•
•
power consumption depends on image content,
less weight than a CRT,
less overall depth ("flat screen")
high contrast,
wide viewing angle,
tendency to burn-in effects with static images (aging of phosphors),
progressive scanning, i.e. tendency to line feathering with interlaced material,
service life used to be a problem, is 100,000 hours today,
retentive and non-stroboscopic system due to the drive system,
insensitive to magnetic fields,
possible short wave radiation with poor shielding,
652
34 Display Technologies
•
•
possible phosphor lag, wrong colour display due to phosphor characteristics,
dynamic false contours.
Glass,
horizontally
polarized
Electrode
Liquid
crystal,
rotating
the polarization
of light,
influenced
by an electrical
field
Glass,
vertically
polarized
Light
Electrode
Fig. 34.9. Liquid crystal technology
34.4 The Liquid Crystal Display Screen
LCD displays (Fig. 34.9.) have been on the market for consumers since the
1970s. Today there are scarcely any applications in which LCDs are not
found. Simple liquid crystal displays are cheap to produce and very energy-saving. If they are statically controlled they use virtually no energy,
which is part of the reason why especially wrist watches have long been
equipped with LCDs. Following the monochrome variant, the colour variant has also been in use since about the middle of the 1990s. Since this
time, especially TFT - thin film transistor - displays have been increasingly
used. Modern computer monitors are employing this technology almost
without exception. The CRT monitor has virtually faded away since the
beginning of the millenium and it no longer exists in the computer field.
And the CRT monitor would not have any advantages there, either, neither
in image quality nor in price, weight or size. In the computer field, the
main reason for this is the technology used from the beginning - the framesequential scanning technique. Fields and line interlace are not used in
34.4 The Liquid Crystal Display Screen
653
computers and were never necessary since the images were relatively static
and flickering was never a problem there. Computers and computer monitors have always operated with progressive scanning.
In LCD displays, the quantity of light through the display is controlled
by more or less rotation of polarized light between two polarizing filters
(Fig. 34.9.). The speed of control was a major item for discussion for a
long time. As a result, these displays were initially very inert. In LCD displays, the quantity of light through the display can be controlled relatively
linearly by the applied cell voltage by more or less rotation of the polarization of the light. If the polarizing filters in front of and behind the liquid
crystal are identically polarized, the light will pass through unimpeded
when no drive is applied and is attenuated only by applying a control voltage; if the polarizing filters are crossed, light will only pass when a control
voltage is applied and the crystal thus rearranges itself and thus rotates the
polarisation of the light. The power consumption of LCD displays essentially depends on the background illumination and is constant and thus independent of the active picture content. LCD displays belong to the category of retentive, i.e. non-stroboscopic display systems and may, therefore,
exhibit so-called motion blur. The essential characteristics of an LCD
screen are
•
•
•
•
•
•
•
•
•
•
•
•
•
small overall depth ("flat screen"),
less weight compared with previous screen technologies,
less contrast compared with CRT and plasma displays,
slower response times,
constant, image-independent power consumption,
possible motion blur,
service life of approx. 100,000 hours,
smaller, but now not inadequate viewing angle compared with
CRT and plasma displays,
retentive and non-stroboscopic system due to the drive system,
insensitive to magnetic fields,
progressive scanning, i.e. tendency to line feathering with interlaced material,
faulty pixels,
pixel response time depends on the step height (signal change).
654
34 Display Technologies
34.5 Digital Light Processing Systems
At the beginning of the current millenium, Texas Instruments marketed a
new display technology for projection systems called Digital Light Processing (DLP), involving so-called digital micro mirror chips - DMM chips.
The concept itself goes back to the year 1987 (Dr. Larry Hornbeck, Texas
Instruments). In this technology, the light intensity per pixel is controlled
by infinitesimally small movable mirrors. Each pixel is a mirror which can
be tilted by about 10 degrees. There is either one chip per colour or only
one chip for all colours which is then divided for Red, Blue and Green in
time-division multiplex via a rotating color disk. There are systems in use
which have three mirror systems for Red, Green and Blue. A mirror can
only be switched into or out of the beam path which is why pulse duration
modulation is used here, too, for controlling the light intensity. When rotating color disks are used, a so-called rainbow effect occurs, i.e. a color
spectrum appears, possibly due to the physiological perception characteristics of the human eye. DLP systems are used in projection systems such as
beamers or back-projectors. They cannot be used for constructing real flat
screens. They are applied in home cinema systems and may also be used in
professional movie theatres because of their good light yield and high resolution and color rendition properties.
34.6 Organic Light-Emitting Diodes
Organic Light-Emitting Diodes (OLEDs) could become the most modern
display technology. They are currently being offered only for relatively
small displays. Their stability or service life still presents a problem. However, they can be applied in very thin layers even to flexible material, thus
making it possible to implement displays which can be rolled up. There are
already first small TV monitors with this technology on the market. For
each pixel, there are three OLEDs the intensity of which can be controlled
linearly.
34.7 Effects on Image Reproduction
For many years, cathode ray tube television was adapted relatively optimally to the characteristics of the human optical system of perception and
had been developed further. But now there are new revolutionary technologies which are replacing the picture tube but have quite different char-
34.7 Effects on Image Reproduction
655
acteristics. The decisive factor for image reproduction is the type of display technology in conjunction with the drive technology, i.e. the signal
processing. And it must be said that new is not always better. A distinction
can be made here between
•
•
stroboscopic reproduction systems with short frame retention time,
and
hold-type (retention) systems with frame retention times within the
range of one frame.
However, the perception characteristics of a moving picture depend not
only on the physics of the display but also on the anatomy of the human
optical perception system (eye, eye tracking and brain).
A distinction must also be made between the type of color reproduction
where there are systems
•
•
which share one reproduction element for all colors (e.g. via rotating color filters) or
which have discrete color reproduction elements for RGB.
And, naturally, the response time of the screen plays an important role.
This is greatly dependent on the display technology. LED systems and
plasma screens have an inherently very short switch-over time between
states of brightness. They also exhibit the largest ratio between light which
is switched on and that which is switched off, and thus have the greatest
contrast ratio.
In addition, there is a difference between display technologies which
operate in
•
•
line interlace scanning mode, and in
progressive scanning mode.
All modern displays can be categorized as "progressive". When using a
display operating with progressive scan it is absolutely necessary to carry
out de-interlacing before the scan if unsightly line feathering known from
the early 100 Hz TV technology is to be avoided. The reason can be found
in the mismatch of fields due to movement between them. Pure cinematographic material is non-critical in this respect since there is no movement
between the two fields.
Displays can also be classified as
656
34 Display Technologies
•
•
linearly driven display elements, and
switchable display elements.
The display elements which can only be switched on and off are then intensity-controlled by pulse duration modulation.
The display technology thus gives rise to the following problems:
•
•
•
•
•
•
•
•
•
•
•
•
blurring due to display delay times,
blurring due to interpolation of unmatched resolution of broadcast
material and display system,
motion blur with non-stroboscopic displays,
blurring due to the display frame rate,
line feathering (line tearing) due to poor de-interlacing (Fig. 34.10.
left),
color shift (plasma, phosphor lag),
different contrast with different steps in brightness (LCD),
rainbow effect (chromatic separation),
obvious appearance of compression artefacts due to the high resolution,
losses in contrast,
power consumption depending on picture content,
burn-in effects (plasma).
Fig. 34.10. Different picture quality after de-interlacing (e.g. feathering left)
34.9 Test Methods
657
34.8 Compensation Methods
In modern electronics, there are countermeasures to act as improvement
for every shortcoming. Their effectiveness depends on the state of the art.
All modern display technologies operate with progressive scanning technology. The simplest remedy for interlace artefacts is not to use interlaced
material. This is the aim especially in HDTV - s. a. 720p. But in the SDTV
field, interlaced material must be offered because there are still countless
television sets with cathode ray tubes. And the archives still contain any
amount of interlaced material. De-interlacing is, therefore, an absolute necessity with modern displays if unsightly line feathering effects are to be
avoided. In de-interlacing, it is necessary to interpolate between the fields.
This is not necessary with material which is inherently progressive such as,
e.g., the good old cine-film which only has frames, the fields being produced by repeated scanning in the intermediate lines. In the cine-projector,
the film is interrupted twice per frame by a rotating aperture wheel to reduce flicker. However, in material recorded with electronic cameras there
is already movement between the two fields. From these fields, virtually
motion-compensated interpolated frames must now be generated otherwise
the lines will become "frayed". To counter motion blur due to nonstroboscopic displays there are so-called 100 Hz or 200 Hz systems which,
in contrast to earlier 100 Hz CRT displays, do into need this for reducing
flicker but virtually simulate for the human eye a movement of an object
on the display by interpolating fewer or a greater number of intermediate
images. Otherwise, the eye and the entire "human optical pickup system"
will dwell on an intermediate state of the image display which, time and
again, is virtually static which then leads to smearing or, in other words,
blurring.
34.9 Test Methods
Video signal generators [DVSG] contain test sequences for detecting the
problems caused by the display technology. These sequences deliberately
generate stress signals for the displays in order to reveal , e.g., resolution,
interpolation characteristics, motion blur, rainbow effect and line feathering. It is also of major interest to see how a display system behaves when it
adjusts itself to a source resolution not corresponding to the system (e.g. a
laptop with unmatched screen resolution), requiring interpolation between
pixels (scaling up or down). It is also of interest to see how the display devices handle compression artefacts such as
658
34 Display Technologies
•
•
•
blurring,
blocking, or
mosquito noise.
The good old picture tube (CRT) was relatively tolerant in this respect
as can be seen impressively in demonstrations. But we are not concerned
here with hanging on to a proven old system but are only comparing new
systems with it and are correspondingly astonished or disappointed. Certain reproduction systems simply do not show some of the things because
these are concealed by them. The normal Gaussian noise evident in analog
TV channels, in particular, produces different effects in different display
systems which causes viewers of analog TV with flat screens in broadband
cable networks increasingly to protest even though the comfort of not requiring a special DVB-C receiver is appreciated so much in these quarters.
But there are also screens which allow noise to be suppressed but this can
only be done at the cost of image sharpness. Noise can only be eliminated
by averaging between frames which again leads to blurring. New interfaces such as HDMI also exchange data between display and receiver, e.g.
for determining the correct resolution from the possible features provided
by the display. These things must also be tested. Another interesting test is
how the various displays behave in a comparison of
•
•
•
25/50 Hz interlaced material,
30/60 Hz material (3:2 pull-down) (Fig. 34.1.), and
inverse 3:2 pull-down (material reconverted from 30 to 25 frames/s) (Fig. 34.1.).
34.10 Current State of the Technology
Conclusion: The quality of reproduction on displays is a matter not only
of the installed display technology but mainly also of the type of compensation measures applied. Some consume a great amount of memory space
and computer power. It is expected that LCD and LED technologies will
displace the plasma systems. Cathode ray tube systems have provided
good service for many decades but their end has come. It cannot be predicted what significance will be accorded to e.g. DLP systems or other,
mainly new technologies. They may well be the active element in cineprojectors of the future. And it can also be said that for most of the picture
material available at present (status of 2009 in Europe), the cathode ray tu-
34.10 Current State of the Technology
659
tube still offers the best solution in most cases. However, this will change
as SDTV becomes HDTV.
Bibliography: [ERVER], [NBTV], [DVSG]
35 The New Generation of DVB Standards
In 2003, DVB-S2 appeared as the first new transmission standard for Digital Video Broadcasting - DVB. Because the transmission of High Definition Television (HDTV) now requires a much higher data rate, the transmission capacity, i.e. the net data rate per satellite transponder, had to be
increased by at least 30% compared with DVB-S. On the other hand, the
hardware now available is much more powerful in comparison with that of
the early 90s. Both the memory capacity and the computing speed of the
chips have increased significantly which made it possible to use Forward
Error Correction (FEC) which, although it swallowed up an enormous
amount of resources on the receiver side, also yielded a significantly
higher net data rate (approx. 30%). This error protection has been known
since 1963 and is based on Robert Gallager's work. Today, all the new
transmission standards, either for mobile or for broadcasting applications,
are using either turbo codes (1993) or LDPC (low density parity check)
codes. DVB-S2 now enables 48 to 49 Mbit/s to be transmitted over a satellite transponder which previously provided a data rate of 38 Mbit/s via
DVB-S. Using MPEG-4 Part 10 AVC (H.264) video coding, 4 HDTV
programs will then fit into one satellite transponder. But with DVB-S2, the
direct tie with the MPEG-2 transport stream as baseband input signal was
also relinquished for the first time. The catchphrase is now GS - Generic
Stream. I.e., apart from the MPEG-2 transport stream, quite general data
streams such as, e.g. IP, are now also provided as the input signal for a
DVB-S2 modulator, making it possible to feed either one or even several
streams into the modulator. At present, this capability is not being exploited for DVB-S2, however, where only pure, standard MPEG-2 transport streams are currently being transmitted. As well, only parts of the capabilities provided in the Standard are currently utilized. The DVB-S2
Standard has already been discussed in Chapter 14 "Transmitting Digital
TV Signals by Satellite - DVB-S/S2". In 2008, the new, very powerful
standard DVB-T2 - "Second Generation Digital Video Broadcasting Terrestrial" was then published which also contains this new error protection,
with Generic Streams - GS - and Multiple Inputs also being a subject. The
possibility of having multiple input streams, either transport streams or generic streams, is here called Physical Layer Pipes (PLP) and will probably
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_35, © Springer-Verlag Berlin Heidelberg 2010
662
35 The New Generation of DVB Standards
also be made use of. The ability to chose between different modulation parameters (error protection and type of modulation) for different contents
will probably be applied mainly with DVB-T2, the keywords being Variable Coding and Modulation (VCM). It will thus be possible to broadcast
HDTV programs, e.g. less robustly but equipped with a higher net data rate
than, e.g., SDTV programs. DVB-T2 was being promoted by the BBC; the
first field trials in the so-called BBC mode have been running since 2008
as Single PLP, i.e. with one (transport stream) input. In 2009, the new, also
very powerful DVB-C2 cable standard then appeared as draft standard.
DVB-C2 is based on DVB-T2 and also has the capability of GSE and Multiple PLP with VCM. The LDPC coding is also used as FEC. The novel
feature in DVB-C2 is the possibility of combining several 8- or 6-MHzwide channels to form channel groups. This is intended to increase the effectiveness by avoiding gaps between the channels. Like DVB-T2, DVBC2 uses COFDM, i.e. a multicarrier method, whereas DVB-S2 only uses a
single-carrier method.
35.1 Overview of the DVB Standards
The following DVB standards exist at present:
•
•
•
•
•
•
•
•
•
•
•
DVB-S - first generation satellite transmission standard
DVB-C - first generation cable transmission standard
DVB-T - first generation terrestrial transmission standard
DVB-SI - Service Information
DVB-SNG - satellite transmission standard for professional
applications
DVB-H - extension for handheld mobiles in DVB-T
DVB-IP - transport stream transmission via IP networks, e.g.
VDSL,
DVB-SH - hybrid method for handheld mobiles via terrestrial and
satellite
DVB-S2 - second generation satellite transmission standard
DVB-C2 - second generation cable transmission standard
DVB-T2 - second generation terrestrial transmission standard.
Naturally, there are numerous other DVB documents but those listed
above are the ones which are most meaningful in the present context.
35.2 Characteristics of the Old and the New Standards
663
35.2 Characteristics of the Old and the New Standards
In this section, the essential characteristics of the old and new DVB
standards are compared with one another. The essential factor is that the
old DVB standards were firmly tied to the MPEG-2 transport stream as an
input signal and a combination of Reed Solomon coding and convolutional
coding was used as error protection. In the new DVB standards, the tie
with the transport stream is relinquished and the error protection has been
modernized.
Table 35.1. DVB standards
Standard
Applicatio
n
Input signal
DVB-S
Satellite
MPEG-2 TS
DVB-C
Cable
MPEG-2 TS
DVB-T
Terrestrial
MPEG-2 TS
DVBSNG
Satellite
MPEG-2 TS
DVB-H
Terrestrial
MPEG-2 TS
with MPE
DVB-IP
Twisted
pair lines
or
Ethernet
Terrestrial
and
MPEG-2 TS
DVB-SH
MPEG-2 TS
with MPE
Forward Error
Correction
(FEC)
Reed-Solomon
and
convolutional
coding
Reed-Solomon
Reed-Solomon
and
convolutional
coding
Reed-Solomon
and Trelliscoding
Reed-Solomon
and
convolutional
coding and
additional
Reed-Solomon
over IP
--
Turbo coding
Modulation
Single Carrier,
QPSK
Single Carrier,
64QAM,
256QAM,
additional QAM
orders below 256
are possible
CODFM,
QPSK, 16QAM,
64QAM
Single Carrier,
QPSK, 8PSK,
16QAM
like DVB-T
--
largely like
DVB-T and
664
35 The New Generation of DVB Standards
DVB-S2
satellite
Satellite
DVB-T2
Terrestrial
DVB-C2
Cable
Single or
multiple
MPEG-2-TS
or GS
Single or
Multiple
MPEG-2-TS
or GS
Single oder
Multiple
MPEG-2-TS
oder GS
BCH and
LDPC
BCH and
LDPC
BCH and
LDPC
DVB-S2
Single Carrier,
QPSK, 8PSK,
16APSK,
32APSK
COFDM,
QPSK, 16QAM,
64QAM,
256QAM
COFDM,
QPSK, 16QAM,
64QAM,
256QAM,
1024QAM,
4096QAM
35.3 Capabilities and Aims of the New DVB Standards
It is especially the new error protection which provides about 30% increase
in data rate from DVB-S2 onward. The Shannon limit is thus coming
closer. DVB-S2 also enables several different input data streams to be
radiated via one satellite. And it is intended to break the tie with the
MPEG-2 transport stream. DVB-T2 additionally exploits other
possibilities for gaining even more data rate compared with DVB-T. The
symbols used are longer, thus reducing, e.g., the overhead in the guard
interval; widening of the useful spectrum more in the direction of the
adjacent channels is being offered etc.. In addition, variable coding and
modulation can be activated in the multiple PLP mode, i.e. contents can be
transmitted with different degrees of robustness similar to DAB and ISDBT. DVB-T2 provides for much more such as, e.g., the application of multiantenna systems at the transmitting end, the active reduction of the crest
factor and much else besides. The details are described in the "DVB-T2"
chapter. DVB-C2 is based on DVB-T2 and uses similar "tricks" for raising
the data rate. It provides multiple PLP as well as VCM. DVB-T2 and
DVB-C2 provide for up to about 50% more data rate compared with the
comparable old DVB standards. DVB/T2 and DVB/C2 also provide higher
modulation types such as, e.g. 256QAM with DVB-T2 and up to
4096QAM with DVB-C2.
The essential features and capabilities of the new DVB standards are
thus:
35.3 Capabilities and Aims of the New DVB Standards
•
•
•
665
30 - 50% increase in net data rate,
multiple inputs (TS or GS),
with T2 and C2, the possibility of variable coding and modulation.
With the change from SDTV to HDTV looming ahead, increased
amounts of net data rate will become necessary. DVB-S2 will eventually
assert itself over DVB-S, but not as yet. There are only a few transponders
with DVB-S2 on air and the future will tell how quickly DVB-T2 and
DVB-C2 will be employed.
In the chapters following, the baseband signals of DVB-x2 and then the
DVB-T2 and DVB-C2 standards will be described. DVB-S2 has already
been discussed in the chapter on DVB-S/S2 and the associated DVB-S2
test methods are already found in the joint chapter "DVB-S/S2
Measurements".
Bibliography: [EN302307], [TR102376].
[TS102773], [TS102606], [TS102771], [A133]
[EN302755],
[A138],
36 Baseband Signals for DVB-x2
In the first generation DVB standards (DVB-S, DVB-C and DVB-T), the
format of the input data was confined only precisely to MPEG-2 transport
streams where all the modulation and demodulation steps are firmly synchronously linked to the 188-bytes-long transport stream packet structure.
An MPEG-2 transport stream packet begins with a 4-bytes-long header
which, in turn, begins with a sync byte having the value 0x47. This limitation to the transport stream structure no longer exists in the new DVB-x2
standards. In the first generation DVB standards, it was also only possible
to feed precisely one transport stream into the modulator, the only exception being DVB-T in its "hierarchical modulation" mode of operation
where the modulator could be supplied with up to two transport streams. In
the new DVB-x2 standards, up to 255 transport streams or generic streams,
or both, can be fed into the modulator and transmitted. The present chapter
deals with the input signals for the new DVB-x2 standards and how they
are processed and conditioned in the input interfaces of the DVB-x2 modulators.
36.1 Input Signal Formats
The new DVB-x2 modulators can accept four different input signal formats, which are:
•
•
•
•
MPEG-2 transport streams (TS),
Generic Fixed Packetized Streams (GFPS),
Generic Continuous Streams (GCS),
and Generic Encapsulated Streams (GSE).
At the time of publication of the DVB-S2 standards, the DVB-GSE
standard did not yet exist. DVB-S2 only provided for GFPS streams and
GCE streams as generic input streams, supporting both a single input
stream (TS or generic) and multiple input streams. Multiple input streams
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_36, © Springer-Verlag Berlin Heidelberg 2010
668
36 Baseband Signals for DVB-x2
can also have different formats. The multiple input streams are called PLPs
- Physical Layer Pipes - in DVB-T2 and DVB-C2.
188 bytes
4 byte
TS header
184 byte
payload
13 bit packet identifier = PID
1 bit transport error indicator
1 byte sync byte = 47 hex
Fig. 36.1. MPEG-2 Transport Stream (TS)
36.1.1 MPEG-2 Transport Streams - TS
An MPEG-2 transport stream consists of packets having a constant length
of 188 bytes (Fig. 36.1.). The packet itself is divided into a header component of 4 bytes length and a payload component of 184 bytes. The first
byte of the header is the sync byte which has a constant value of 0x47. The
next three bytes in the header are used for signalling important transport
parameters. The rest is described in detail in Chapter 3.
Payload
UPL = constant and UPL <= 64 kbyte
Fig. 36.2. Generic Fixed Packetized Stream (GFPS)
Sync
Payload
Sync
Sync
UPL
36.1 Input Signal Formats
669
36.1.2 Generic Fixed Packetized Streams - GFPS
Generic Fixed Packetized Streams (GFPS) are data streams which have a
packet structure and the packet length of which is known and is constant
(Fig. 36.2.). The beginning of a packet is marked with a special sync byte.
An example of a relevant application would be ATM data signals with a
constant length of 53 bytes. The length of a GFPS must not exceed 64
kbytes for else it would be a Generic Continuous Stream - a GCE.
Payload
Sync
Payload
Sync
Sync
UPL
Payload
UPL = variable or UPL > 64 kbyte or no sync
Fig. 36.3. Generic Continuous Stream (GCS)
CRC
GSE
header
GSE
header
GSE
header
PDU = Protocol Data Unit
Fig. 36.4. Generic Stream Encapsulation (GSE)
36.1.3 Generic Continuous Streams - GCS
Generic Continuous Streams (GCS) do not have any packet structure (Fig.
36.3.). Thus, the modulator interface does not recognize any boundaries in
the data stream. Generic continuous streams are the most generalized form
of data streams. Generic continuous streams are also data streams which
have a packet structure which don’t differs from that of GFPS but the
packet length of which varies or is longer than 64 kbytes.
670
36 Baseband Signals for DVB-x2
36.1.4 Generic Encapsulated Streams - GSE
Generic Encapsulated Streams (GSE) have a packet structure the packet
length of which varies. The beginning of the packet is provided with a special GSE header as defined in the DVB GSE standard [TS102606]. The
user data packet is prefixed by a GSE header in the GSE encapsulator and
a CRC is formed over the entire data packet, which CRC is then appended.
The user data packets can also be divided into a number of packets where
each packet starts with a GSE header (Fig. 36.4.). This type of data did not
yet exist at the time when DVB-S2 became a standard.
Single
input
stream
Input
interface
BB header
insertion
CRC-8
encoder
Mode adaptation
Padding
insertion
BB
scrambler
To BICM
module
Stream adaptation
Fig. 36.5. Input Processing with a single input stream
Only if input data = TS or GFPS and UPL<=64kbyte
Calculate
CRC-8
Payload
Sync
Payload
UPL = User packet length
Sync
Sync
UPL
Replace
next Sync-byte
Fig. 36.6. CRC-8 encoding
36.2 Signal Processing and Conditioning in the Modulator
Input Section
The following paragraphs describe how one or more DVB-x2 input
streams are conditioned in the input section of the modulator. The difference lies in the conditioning of a single input stream in comparison with
multiple input streams. Conditioning a single data stream, either a trans-
36.2 Signal Processing and Conditioning in the Modulator Input Section
671
Payload
CRC-8
CRC-8
CRC-8
port stream or a generic stream is clearly easier than conditioning a
number of input streams.
Payload
SYNCD
80 bits
Baseband
header
MATYPE
(2 bytes)
UPL
(2 bytes)
DFL
Data field
DFL
(2 bytes)
SYNC
(1 byte)
SYNCD
(2 bytes)
CRC-8
(1 byte)
Fig. 36.7. Baseband header insertion with TS or GFPS
Payload
80 bits
Baseband
header
MATYPE
(2 bytes)
UPL
(2 bytes)
DFL
Data field
DFL
(2 bytes)
SYNC
(1 byte)
SYNCD
(2 bytes)
CRC-8
(1 byte)
Fig. 36.8. Baseband header insertion with GCS
36.2.1 Single Input Stream
If only one data stream is fed into the DVB-x2 modulator (called Mode A
in DVB-T2), the modulator first synchronizes itself in the input interface
to the data stream supplied (Fig. 36.5.). This is followed by the CRC-8 encoder (Fig. 36.6.) which inserts a checksum into the data stream at a particular point unless this is a continuous data stream. In the case of a transport stream, a CRC-8 is formed over the 187 bytes preceding the next sync
byte and the subsequent sync byte is then replaced by this checksum. If it
is a GFPS, the CRC-8 is formed over all data except the sync byte and the
sync byte following the packet is also replaced by a CRC-8.
672
36 Baseband Signals for DVB-x2
Following this, a piece of corresponding length is cut out of the data
stream and a ten-byte-long, i.e. 80-bit-long, baseband header is placed in
front (Fig. 36.7.). This is done continuously piece by piece and occurs
completely asynchronously with the data steam supplied, whether this is a
TS, GFPS, GCS or GSE. In the case of a TS or GFPS, the distance in bytes
from the beginning of the packet cut out to the next sync byte is entered in
the SYNCD part of the baseband header (Fig. 36.7.).
Components of the baseband header:
MATYPE (2 bytes) – Mode Adaptation Type
MATYPE-1
•
•
•
•
•
•
TS/GS field (2 bits), input stream format: Generic Fixed Packetized Stream, Transport Stream, Generic Continuous Stream, Generic Encapsulated Stream
SIS/MIS field (1 bit): Single or multiple input streams
CCM/ACM field (1 bit): Constant coding and modulation or variable coding and modulation (or adaptive coding and modulation in
DVB-S2)
ISSY (1 bit): Input stream synchronization indicator
NPD (1 bit): Null-packet deletion active/not active
EXT (2 bits): media specific, reserved for future use
MATYPE-2
•
If multiple input streams, MATYPE-2 = ISI = Input stream identifier (up to 255 input streams).
Other components of the baseband header:
•
•
•
•
•
•
UPL (2 bytes): User packet length in bits (0…65535)
DFL (2 bytes): Data field length in bits (0 …53760)
SYNC (1 byte): A copy of the user packet sync-byte
SYNCD (2 bytes): Distance between beginning of data field to the
beginning of the first user packet which starts in the data field
CRC-8 mode (1 byte): The XOR of the CRC-8 field with the
MODE field
CRC-8 is the CRC over the first 9 bytes of the baseband header
36.2 Signal Processing and Conditioning in the Modulator Input Section
•
•
673
MODE: 0 = Normal Mode (NM), 1 = High Effiency Mode
(HEM), other
values reserved for future use
The first 3 blocks of input processing, namely input interface, CRC-8encoding und baseband header insertion are called Mode Adaptation (Fig.
36.5.). This is followed by the stream adaptation (Fig. 36.5.) consisting of
padding und baseband scrambler.
80 bits
DFL
Baseband
header
Data field
Padding
DVB-x2 outer FEC input data size
Fig. 36.9. Padding
Padding (Fig. 36.9.) means filling up or stuffing, i.e. if there are not sufficient user data available, a DVB-x2 FEC frame is filled up with stuffing
or padding bytes until it is completely filled with data. This corresponds to
an adaptation to the FEC frame structure of the outer DVB-x2 (BCH) error
protection and is dependent on the code rate set for the error protection.
Reset at the beginning of each baseband frame
1
0
0
1
0
1
0
1
0
0
0
0
0
0
0
1 2
3
4
5
6
7
8
9 10 11 12 13 14 15
=1
=1
Randomized
data out
Data in
Fig. 36.10. Baseband scrambler
This is followed by the baseband scrambler (Fig. 36.10.) which has the
task of randomizing the data as much as possible, i.e. adjacent long se-
674
36 Baseband Signals for DVB-x2
quences of zeroes and ones are broken up and a pseudo-random data
stream is generated. For this purpose, the data are Exclusive-ORed with a
pseudo-random sequence. The PRBS generator has exactly the same structure as the energy dispersal stage of the first generation DVB standard and
is always reset at the beginning of a baseband frame.
The DVB-x2 input data are now ready for the next signal processing
steps which depend on the respective DVB-x2 standard.
36.2.2 Multiple Input Streams
In the sections following, the much more complex signal processing in the
input stage of a DVB-x2 modulator in the case of multiple input streams is
described. In DVB-T2, the operating mode with multiple input streams,
whether they are transport streams or generic streams, is called Mode B or
Multiple PLP.
Nullpacket
delet.
CRC-8
encod.
BB
header
Insert.
Input
interface
Input
stream
sychronizer
Nullpacket
delet.
CRC-8
encod.
BB
header
Insert.
Input
interface
Input
stream
sychronizer
Nullpacket
delet.
CRC-8
encod.
BB
header
Insert.
…
Multiple
input
PLP1
streams
Input
stream
sychronizer
…
PLP0
Input
interface
PLPn
To
stream
adaptation
Fig. 36.11. Mode adaptation with multiple input streams
The multiple input streams (TS, GFPS, GCS or GSE) are present at the
respective inputs of the input interface where synchronization to the input
streams takes place. The input streams are then supplied to the further signal processing blocks of the mode adaptation section (Fig. 36.11.). The
first one is the optional input stream synchronizer (ISSY) (Fig. 36.12.).
The ISSY can be used for removing jitter from the input streams in the receiver; which is done via an internal clock similar to the STC/PCR in the
case of the MPEG-2 transport stream. However, this clock is controlled via
the modulator clock. The clock consists of a 22-bit counter the current
value of which is continuously repeatedly written into optional ISSY fields
36.2 Signal Processing and Conditioning in the Modulator Input Section
675
which are appended to the end of the user packet (UP) and have a length of
2 or 3 bytes.
Modulator
clock
22 bit
counter
ISSY
Sync
UPL
Payload
2 byte or
3 byte
Fig. 36.12. Input stream synchronization (ISSY)
DNP
counter
Data out
Data in
DNP (1 byte)
DNP
Payload
ISSY
Sync
DNP
Payload
ISSY
Sync
DNP
Payload
ISSY
Sync
Insertion after
next useful
packet
Fig. 36.13. Null Packet Deletion
The use of ISSY is signalled via the baseband header, followed by the
"Null Packet Deletion" block (Fig. 36.13.). This block is optional and is
only used if an MPEG-2 transport stream is present as input stream. An
MPEG-2 transport stream contains null packets which run on
PID=0x1FFF, i.e. on the highest PID. These packets do not carry any payload data but only pad the transport stream to a constant total data rate. The
"Null Packet Deletion" processing block has the aim of removing these
676
36 Baseband Signals for DVB-x2
null packets from the transport stream. Null packets are unnecessary ballast which does not have to be transmitted. However, the null packets must
be removed in such a way that the receiver is able to add them again to the
transport stream in the correct number and at the correct position. For this
purpose, a DNP byte (Fig. 36.14.) is inserted after each transport stream
packet (after the ISSY field or, if there is none, directly after the transport
stream packet). In the DNP (Delete Null Packet) field, the number of null
packets removed from in front of this packet is entered. If the value in the
DNP field is zero, no null packets have been removed from in front of this
packet. To this end, the removal of the null packets is tracked in parallel by
a null packet counter which is incremented with each removal. After the
next transport stream packet which is not a null packet, the count is then
entered in the DNP field of this transport stream packet carrying "real
data" and the counter is reset. If the counter reaches its maximum of 255,
this value is entered in the DNP field of the next transport stream packet
and this is also transmitted even if it is a null packet. Following this, the
DNP counter is also reset.
Payload
Payload
ISSY
Sync
Payload
ISSY
n+2
ISSY
Payload
ISSY
Sync
n
Sync
PID=0x1FFF
Null packet
n+1
Delete
DNP=0
DNP
Sync
DNP
Payload
n+2
ISSY
Sync
n
DNP=1
Fig. 36.14. Deletion of Null Packets
As in the case of the single input stream, too, the CRC-8 checksum and
the baseband header are then inserted. The operation of these two blocks
has already been described in the "Single Input Streams" section (Fig.
36.6.).
In the next block "Stream Adaptation" (Fig. 36.15.), all the streams prepared in the Mode Adaptation block are then combined. The block which
combines the streams is called either merger or scheduler, depending on
the standard, DVB-S2, DVB-T2 or DVB-C2. The "Stream Adaptation"
block also differs slightly in detail with these standards. In the present sec-
36.3 Standard-related Special Features
677
……..
PLP1
PLPn
Padding
BB
scrambler
Padding
BB
scrambler
Padding
BB
scrambler
…..
PLP0
Scheduler / Merger
tion, only the common features of all standards will be discussed. The details and differences will be described later.
As in the case of the single input stream, this is followed by the padding
and the baseband scrambler. These two blocks have also been described already in the "Single Input Stream" chapter,
The data streams are now ready for the "Bit Interleaved Coding and
Modulation" signal processing block following, which, however, differs
greatly in the different standards and will be explained in the respective
chapter on DVB-S2, DVB-T2 and DVB-C2.
To
BICM
module
Fig. 36.15. Stream Adaptation in multiple PLP
36.3 Standard-related Special Features
We will now briefly discuss standard-related special features in the input
signal processing of DVB-S2, DVB-T2 and DVB-C2.
36.3.1 DVB-S2
The signal processing in DVB-S2 corresponds very much to that already
described in this chapter. At the time when the DVB-S2 standard was being fixed, the "Generic Encapsulated Stream" (GSE) data format had not
yet been defined. Neither is there any mention of the term PLP (Physical
Layer Pipe) in the DVB-S2 standard where simply multiple input streams
are referred to. The signal processing section in the "Stream Adaptation"
678
36 Baseband Signals for DVB-x2
block is called "merger/slicer" in DVB-S2 and "Multiple Input Streams"
has not as yet appeared as an operating mode in any DVB-S2 application.
36.3.2 DVB-T2
The term PLP (Physical Layer Pipe) was used for the first time in
DVB-T2. And in DVB-T2, the Multiple Input Mode is called Mode B and
will also be used as such. It is especially the possibility of being able to use
this standard for transmitting contents with different robustness and with
different data rates which will be made use of and is here called VCM Variable Coding and Modulation.
The Stream Adaptation block (Fig. 36.16.) contains further processing
steps such as
•
•
•
dynamic scheduling information,
frame delay and
in-band signalling.
Frame m
PLP0
Frame m-1
Frame
delay
In-band
signalling/
padding
BB
scrambler
In-band
signalling/
padding
BB
scrambler
PLPn
Frame
delay
L1 dynPLP1(m)
Frame
delay
L1 dynPLPn(m)
…..
……..
PLP1
Scheduler
L1 dynPLP0(m)
In-band
signalling/
padding
To
BICM
module
BB
scrambler
Dynamic
scheduling
information
L1 dynPLP0-n(m)
Fig. 36.16. Stream Adaptation Block in DVB-T2
Instead of padding data, the padding field can also contain in-band signalling data. This can be used for dynamic Layer-1 (L1) signalling for subsequent frames. I.e. it can be used for dynamically signalling and altering
36.3 Standard-related Special Features
679
e.g. modulation parameters and error protection. Since the signalling information relates to subsequent frames, each PLP path will require a frame
delay block.
Multiple
TS or GS
streams
Input
preprocessor(s)
Input
processing
Bit interleaved
coding &
modulation
Frame
builder
COFDM
generation
„T2-MI“
Fig. 36.17. Interface between DVB-T2 multiplexer and modulator
Frame m
PLP0
Frame m-1
Frame
delay
In-band
signalling/
padding
BB
scrambler
In-band
signalling/
padding
BB
scrambler
PLPn
Frame
delay
L1 dynPLP1(m)
Frame
delay
L1 dynPLPn(m)
…..
……..
PLP1
Scheduler
L1 dynPLP0(m)
In-band
signalling/
padding
To
BICM
module
BB
scrambler
Dynamic
scheduling
information
L1 dynPLP0-n(m)
T2-MI
Fig. 36.18. Precise interface between DVB-T2 gateway and modulator (T2-MI)
Since DVB-T2 is also intended for forming single-frequency networks,
these multiple input streams must be supplied completely synchronously to
all modulators. This would never be possible over n feed lines which is
why the PLPs are combined in the DVB-T2 multiplexer/DVB-T2 gateway
680
36 Baseband Signals for DVB-x2
outside the modulator, a separate interface, the DVB-T2 modulator interface, T2-MI in brief, being defined for this purpose (Fig 36.17.).
In principle, the interface between DVB-T2 multiplexer or DVB-T2
gateway and modulator is located between the input processing block and
Bit Interleaved Coding and Modulation (Fig. 36.17.). The T2-MI signal
contains all PLPs. The DVB-T2 modulator is only supplied with a single
special input signal. In precise terms, the T2-MI interface is located after
the scheduler (Fig. 36.18.). However, the padding is still carried out in the
DVB-T2 gateway.
Payload length (bits)
48 bits
T2-MI
header
Pad
Payload
0…7 bits
Packet
type
Packet
count
Superframe
Reserved
index
CRC32
32 bits
Payload
length
Fig. 36.19. T2-MI packet structure
Table 36.1. Packet Type in the T2-MI header
T2-MI Packet Type
0x00
0x01
0x10
0x11
0x20
0x21
0x30
0x31
All other values
Description
Baseband frame
Auxiliary stream I/Q data
L1 current
L1 future
DVB-T2 timestamp
Individual addressing
FEF part: Null
FEF part: I/Q data
Reserved for future use
For the T2-MI DVB-T2 modulator interface, a separate packet structure
was defined, namely T2-MI packets (Fig. 36.19.) with header and payload.
After the payload field, the frame is padded with bits to provide an integral
number of bytes overall. This is followed at the end with a CRC-32 checksum. The T2-MI header contains the following components:
•
•
•
•
Packet Type (8 bits)
Packet Count (8 bits)
Superframe Index (4 bits)
Reserved for future use (12 bits)
36.3 Standard-related Special Features
•
681
Payload Length (16 bits)
The Packet Type is used for signalling which data are currently being
transmitted in the paylaod-field.
Packet count is a counter which is incremented by one continuously and
independently of payload always from T2-MI packet to T2-MI packet. The
counter runs from 00 to FF and then starts again at 00. Packet Count can
be used to determine at the modulator input, e.g., if packets have been lost.
The superframe index is constant for all T2-MI packets belonging to the
same superframe. The "reserved for future use" bits are currently meaningless. "Payload length" signals the length of the payload component in bits.
L1, SFN info,
Aux Data
(BB frames) Streams
MPEG-2 TS/GSE
ETSI TS 102 773
TS data
T2-MI packets
DVB Data Piping
„T2-MI“
DVB/MPEG-2 TS
Modulator Interface
RTP
UDP
DVB-IP Phase 1
(MPEG TS over IP)
IP
ASI
Ethernet
Fig. 36.20. Physical T2-MI interface
The T2-MI packets are physically packaged into MPEG-2 transport
stream packets via DVB Data Piping (Fig. 36.21.). i.e. the T2-MI packets
are cut into 184-byte-long pieces and then packaged into the payload component of the MPEG-2 transport stream packets. To utilize the transport
stream interface particularly effectively, it is intended to work with a
pointer field immediately after the transport stream header similar to
MPEG-2 sections. If the transport error indicator in the TS header is set to
One, it marks the beginning of a new T2-MI packet in the payload component of the transport stream packet when the T2-MI packet is embedded.
However, it is also possible to transmit the rest of the preceding T2-MI
packet at the beginning in the TS packet. The pointer then points to the beginning of the next T2-MI packet in the payload proportion of the TS
packet. This saves stuffing in the transport stream packet containing the
682
36 Baseband Signals for DVB-x2
end part of the last T2-MI packet and thus gains transmission capacity on
the feed link.
The interface type provided is then either TS-ASI or DVB-IP (Fig.
36.20.). The Gigabit Ethernet interface is becoming more and more popular as a TS interface and it makes sense, therefore, to provide both currently very popular TS interfaces TS-ASI and DVB-IP as the physical interface for T2-MI, making it possible to use the existing TS infrastructure
also for the distribution of T2-MI signals.
T2-MI packet
Payload
H
T2-MI packet
H
Payload
H
Payload
Pointer
Payload unit start indicator = 1
Fig. 36.21. Data piping – T2-MI packets are transmitted in MPEG-2 transport
stream packets
In 0
In 1
In n
DVB-T2
gateway
DVB-T2
mod.&Tx
RF
DVB-T2
mod.&Tx
RF
DVB-T2
mod.&Tx
RF
T2-MI
Fig. 36.22. DVB-T2 network with DVB-T2 gateway and DVB-T2 modulators and
transmitters
36.3 Standard-related Special Features
PLP0
Input
stream
sychronizer
Compensat.
delay
Nullpacket
delet.
CRC-8
encod.
BB
header
Insert.
Input
interface
Input
stream
sychronizer
Compensat.
delay
Nullpacket
delet.
CRC-8
encod.
BB
header
Insert.
Input
interface
Input
stream
sychronizer
Nullpacket
delet.
CRC-8
encod.
BB
header
Insert.
PLPn
…
…
…
Multiple
input
PLP1
streams
Input
interface
Compensat.
delay
683
To
stream
adaptation
Payload
DNP
DNP
Payload
No CRC
ISSY in
BBHeader
Fig. 36.23. Extended Mode Adaptation Block in DVB-T2
SYNCD
DFL
80 bits
UPL
(2 bytes)
ISSY
MATYPE
(2 bytes)
Data field
DFL
(2 bytes)
SYNC
(1 byte)
ISSY
Baseband
header
SYNCD
(2 bytes)
CRC-8
(1 byte)
optional
Fig. 36.24. High Efficiency Mode in DVB-T2 for TS
T2-MIP packets for synchronizing DVB-T2 single-frequency networks
(SFNs) are also transmitted over the MPEG-2 transport stream interface
serving as a T2-MI interface, where MIP stands for Modulator Information
Packet. This is a normal MPEG-2 transport stream packet and not a T2-MI
packet. The DVB-T2 modulators are also configured (Layer-1 signalling)
via special T2-MI packets over the T2-MI interface. Further details are described in the DVB-T2 chapter since this requires more prior DVB-T2
knowledge.
DVB-T2 contains more special features. Firstly, there is also the concept
of a Common PLP. This is a physical layer pipe which carries information
for several PLPs. The mode adaptation block also contains the compensat-
684
36 Baseband Signals for DVB-x2
ing delay circuit section. In this section, delay differences between the
PLPs are compensated for. Together with the ISSY block, a Common PLP
can be synchronized here with the other PLPs. Although this also provides
for synchronisation between the PLPs, this does not appear to be relevant
from the current point of view. It can be expected that a receiver will demodulate exactly one PLP plus Common PLP at the same time.
Furthermore, a Normal Mode (NM) and a High Efficiency Mode
(HEM) have also been defined in DVB-T2. These two modes only relate to
the input signal processing. The Normal Mode actually does not require
any further comment since it corresponds precisely to the subject matter
discussed before. The NM is also the mode which is compatible with
DVB-S2. It applies to all four input signal formats. The High Efficiency
Mode is restricted to only TS and GSE. In this mode, no CRC-8 is formed
and transmitted. In addition, the ISSY field is transported in the baseband
header in the UPL and SYNC fields, which are now free. UPL and SYNC
are both known in the signal formats TS (UPL = 188 bytes and SYNC =
0x47) and GSE (UPL signalled in the GSE header). It is thus possible to
save a few more bytes at this point.
UPL
UPL in GSE
header
GSE UP
GSE user packet
SYNCD
DFL
80 bits
UPL
(2 bytes)
ISSY
MATYPE
(2 bytes)
Data field
DFL
(2 bytes)
SYNC
(1 byte)
ISSY
Baseband
header
SYNCD
(2 bytes)
CRC-8
(1 byte)
optional
Fig. 36.25. High Efficiency Mode in DVB-T2 for GSE
36.3.3 DVB-C2
As far as DVB-C2 is concerned, no additional features can be currently reported. The signal processing is almost exactly the same as that discussed
in the present chapter. DVB-C2 also mentions PLPs. At present, no modu-
36.3 Standard-related Special Features
685
lator interface is defined. DVB-C2 also contains the Normal Mode (NM)
and the High Efficiency Mode (HEM).
Bibliography: [EN303307], [TR102376],
[TS102773], [TS102606], [TS102771], [A133]
[EN302755],
[A138],
37 DVB-T2
DVB-T2 – „Second Generation Digital Terrestrial Video Broadcasting“
[DVB A122r1], [ETSI EN 302755] is a completely new DVB-T-Standard
which no longer has anything in common with the conventional DVB-Tstandard. Just like DVB-T, DVB-T2 was mainly promoted by the BBC. In
the UK, it was intended to use it to push the terrestrial HDTV-coverage in
conjunction with MPEG-4-source coding. DVB-T2 is intended to achieve
an approximately at least 30% to 50% higher net data rate compared with
DVB-T, as well as better suitability for mobile use. There are applications
which, on the one hand, require either a higher data rate, but also applications which must be very rugged in the mobile environment and must be
able to manage with very narrow channels in some cases. It was not possible to cover both with DVB-T itself. It wasn't without reason that the
bandwidth defined for DVB-T2 was also 1.7 MHz which is why it will be
interesting to see what this will mean for DAB.
37.1 Introduction
Just as in DVB-T, the modulation method used is also COFDM, but with
altered and extended constellation diagrams. The error protection used is
the FEC defined in DVB-S2, i.e. BCH coding in the outer error protection
and LDPC coding in the inner error protection, followed by bit interleaving.
The FEC frame structure as a whole corresponds to the frame structure
of DVB-S2. The LDPC-coding (Low Density Parity Check Codes) has already been known since the 1960s but requires very much more computing
power in the receiver and could only be implemented recently due to the
chip technologies now available. Since the Spring of 2006 until March
2008, there have been seven DVB-T2-meetings lasting several days each.
In the March 2008 meeting, a preliminary paper was adopted which was
published by ETSI as a Draft in May 2008. In the autumn of 2008, the Implementation Guidelines and the T2-MI (T2-Modulator Interface) then appeared. DVB-T2 is probably coming too early or too late for many coun-
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_37, © Springer-Verlag Berlin Heidelberg 2010
688
37 DVB-T2
tries – depending on your point of view -, since DVB-T has already been
introduced in these countries and a change-over to DVB-T2 would not be
appropriate at the present time or would not find any acceptance. The
DVB-T networks in the UK are the very first ones and have been running
since 1998. A jump in technology from SDTV to HDTV would probably
justify a parallel introduction of DVB-T2 in this country, especially since
terrestrial propagation is dominant here. In countries like Germany where
terrestrial TV coverage has no longer been the main path for TV propagation for years, HDTV by terrestrial means, and thus DVB-T2, does not
seem to make any sense at the moment. In principle, the scenarios for the
introduction of DVB-T2 can be subdivided as follows:
•
•
•
into countries which are already operating DVB-T,
countries which are still operating analog TV country-wide, and
countries wishing to introduce additional new applications which
cannot be implemented using DVB-T.
New applications are, e.g. applications in frequency bands which are now
free and which were or are intended for DAB.
37.2 Theoretical Maximum Channel Capacity
However, before discussing the DVB-T2-standard in detail, the theoretical
limits of the terrestrial transmission channel will first be considered on the
basis of an 8-MHz-wide channel, looking at various receiving conditions
from portable indoor antenna to fixed outdoor antenna, with characteristics
known from DVB-T. The maximum possible data rate in theory is expressed in approximation by the Shannon-limit via the following formula
if the signal-to-noise ratio is about or more than 10 dB:
C = 1/3 • B • SNR;
where C = channel capacity (in bits/s);
B = bandwidth (in Hz);
SNR = signal/noise ratio (in dB);
An 8-MHz-wide terrestrial TV channel will then provide the following
theoretical maximum channel capacity:
37. 2 Theoretical Maximum Channel Capcity
689
Table 37.1. Theoretical maximum channel capacity of an 8-MHz-wide TV channel
SNR[dB]
10
12
15
18
20
25
30
Theor. max.
channel capacity
[Mbit/s]
26.7
32
40
48
53.3
66.7
80
Comment
Poor portable indoor reception
Portable indoor reception
Good portable indoor reception
Poor reception with outdoor antenna
Good roof antenna reception
Very good roof antenna reception
In DVB-T, the data rates in an 8-MHz channel in DVB-T networks designed for portable indoor-reception are frequently about
13.27 Mbit/s (16QAM, CR = 2/3, g= 1/4, SFN, limit SNR = 12 dB)
and in DVB-T networks designed for roof antenna reception they are in
most cases about
22.39 Mbit/s (64QAM, CR = 3/4, g=1/4, SFN, limit SNR = 18 dB).
The aim in DVB-T2 is to achieve data rates which are higher by at least 30
to 50%. Without familiarity with the DVB-T2 Standard, it can thus be expected that, given comparable conditions, the following data rates can be
achieved:
Portable indoor-reception (SFN, long guard interval):
17.3 to 19.9 Mbit/s
Roof antenna reception (SFN, long guard interval):
29.1 to 33.6 Mbit/s.
The error protection alone will bring 30 %. Additional features such as
•
•
•
•
the 16K- and 32K-mode,
the extended carrier mode,
256QAM modulation,
the rotated Q-delayed constellation diagrams
will bring further improvements in the data rate.
690
37 DVB-T2
37.3 DVB-T2 - Overview
The essential core parameters of DVB-T2 are:
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
Several MPEG-2 transport stream inputs or possibly genericstreams as baseband signals (up to 255)
approx. at least 30% higher net data rate mainly due to the improved BCH+LDPC error protection already used in DVB-S2
compatibility with the Geneva 2007 Frequency Plan (8, 7, 6 MHz
bandwidths)
additional bandwidths 1.7 MHz and 10 MHz
stationary, but also mobile applications
COFDM
1K, 2K, 4K, 8K, 16K and 32K-Mode
guard interval 1/4, 1/8, 1/16, 1/32, 19/256 and 1/128
modulation scheme QPSK, 16QAM, 64QAM and 256QAM
Q-delayed „rotated“ constellation diagrams
RF frame-structure with P1 and P2-symbol at the beginning of the
frame
flexible pilot structures with fixed and distributed pilots
PAPR reduction (Peak to Average Power Ratio) reduction, i.e. reduction of the crest-factor (2 different methods)
variable coding and modulation (the transmission parameters can
be changed in operation)
time interleaving
time slicing
optional MISO-principle (Multiple Input, Single Output)
inbuilt FEFs (Future Extension Frames) for later extensions
auxiliary data streams as an option
time frequency slicing (TFS) mentioned in the Appendix of the
Standard.
The details of DVB-T2 will now be discussed in the following sections.
37.4 Baseband Interface
The DVB-T2 baseband interface provides one or more data inputs. DVBT2 is no longer orientated only towards MPEG-2 transport streams but also
provides for generic-streams as possible input streams. Up to 255 input
37. 5 Forward Error Correction
691
streams are possible. Initially it was left open where these streams will be
multiplexed. The answer came with the "T2-MI“ standard, the modulatorinterface for DVB-T2. The streams are combined in the playout center and
the DVB-T2 modulator is supplied with only one data stream via DVB-T2MI. Similar to the ETI stream in DAB, this data stream is provided with all
necessary information for the modulator. It also contains the time stamp
for synchronizing single-frequency networks. The baseband interface has
already been described in Chapter 36. There are two modes in DVB-T2,
namely Mode A = Single PLP (Physical Layer Pipe) and Mode B = Multiple PLPs. It is only in the Mode A case that all processing steps take place
in the modulator itself, whereas in the case of Mode B, the T2-MI interface
follows immediately after the scheduler.
There is one special feature of DVB-T2 which also influences the input
signal processing. In Mode B, it is possible to work with variable coding
and modulation and this can also be done dynamically. I.e., in the next
DVB-T2 frame, the transmission parameters could change and this may
have to be signalled dynamically. This is done in the padding field of the
baseband header. And there is an optional Common PLP which contains
information for all or some PLPs. In Mode B, an additional distinction is
made between the
•
•
HEM = High Efficiency Mode (for MPEG-2 transport streams and
GSE) and the
NM = Normal Mode (compatible with DVB-S2).
Further details of the signal processing in the baseband interface can be
found in Chapter 36 “Baseband Signals für DVB-x2”.
37.5 Forward Error Correction
Just as in DVB-S2, the modified error protection (Fig. 37.1.) leads to a gain in S/N, thus coming closer to the Shannon-Limit overall. The net data
rate is increased by 30% by this measure alone. As in DVB-S2, the error
protection in DVB-T2 consists of a baseband scrambler, a BCH-block encoder, and an LDPC-block encoder followed by the bit-interleaver. In the
DVB-T2 modulator, the baseband-frame including baseband header and
padding-block are first scrambled (baseband scrambler) and then supplied
to the FEC-block where the BCH code is first added. After that, a further
error protection, the length of which depends on a selectable code rate, is
appended in the LDPC encoder, the possible code rates being:
692
37 DVB-T2
•
•
•
•
•
•
1/2,
3/5,
2/3,
3/4,
4/5 or
5/6.
Code rate 1/2 means maximum error protection and minimum net data rate
and code rate 5/6 means minimum error protection and maximum net data
rate.
Baseband
scrambler
BCH
encoder
LDPC
encoder
Bit
interleaver
Fig. 37.1. DVB-T2-Error protection (BCH = Bose-Chaudhuri-Hoquenghem,
LDPC = Low Density Parity Check Code)
80 bits
Baseband
header
Data from MPEG-2 TS or generic data
Data field
tBCH=8,10,12
DFL
Padding
kBCH
Outer FEC: BCH coding
16*tBCH bits
BCH
kLDPC = code rate * FEC frame
Inner FEC: LDPC coding
Code
rate
LDPC
64800 or 16200 bits FEC frame
Fig. 37.2. DVB-T2 FEC frame
Just as in DVB-S2, it is possible to use a short (16K) or a long FEC
frame (64K) in DVB-T2 (Fig. 37.2.). The differences in performance with
37. 5 Forward Error Correction
693
respect to the required signal/noise ratio are minimum and are within a
range of a few tenths dB. The short FEC frame is possibly more advantageous for low-rate data streams and the long frame is better for higher-rate
data streams. The data rates now possible in DVB-T2 lie between 7.49
Mbit/s (QPSK, CR = 1/2) and 50.32 Mbit/s (256QAM, CR = 5/6). The
lowest required signal/noise ratios are between 0.4 and 25.9 dB (Table
37.2. and 37.3.). In comparison with DVB-T, the fall-off-the-cliff effect is
much steeper in DVB-T2. The transition from Go- to No-Go takes place
within only a few steps of a hundredth dB each. The reason for this lies in
the concatenation of two block-codes. Examples of data rates are listed in
Table 37.4. for the case of an 8-MHz-channel, 32K mode, g = 1/128, PP7.
Other transmission parameters will lead to numerous other data rates.
There are many possible combinations which would fill many pages.
Table 37.2. C/N limits for a BER of 1•10-4 after LDPC, long 64K FEC frame
Modulation
Code rate
C/N
Gaussian
channel
[dB]
QPSK
1/2
3/5
2/3
3/4
4/5
5/6
1/2
3/5
2/3
3/4
4/5
5/6
1/2
3/5
2/3
3/4
4/5
5/6
1/2
3/5
2/3
3/4
4/5
5/6
0.8
2.1
2.9
3.9
4.5
5.0
5.7
7.4
8.6
9.8
10.6
11.2
9.6
11.7
13.2
14.9
15.9
16.6
12.8
15.6
17.5
19.7
21.1
21.8
16QAM
64QAM
256QAM
C/N
Rice
channel
[dB]
1.0
2.4
3.3
4.3
5.0
5.6
6.1
7.7
8.9
10.3
11.1
11.8
10.0
12.1
13.6
15.3
16.4
17.2
13.3
16.0
17.8
20.2
21.5
22.3
C/N
Rayleigh
channel
[dB]
1.8
3.4
4.6
5.9
6.8
7.2
7.3
9.1
10.5
12.2
13.4
14.4
11.7
13.8
15.4
17.5
19.0
19.9
15.4
18.1
20.0
22.5
24.2
25.3
C/N
0dB echo
channel
@ 90%
GI [dB]
1.5
3.0
4.2
5.5
6.4
7.2
7.0
8.8
10.2
11.9
13.2
14.2
11.5
13.6
15.1
17.3
18.9
20.1
15.3
18.2
20.0
22.5
24.4
25.7
694
37 DVB-T2
Table 37.3. C/N limits for a BER of 1•10-4 after LDPC, short 16K FEC frame
Modulation
Code rate
C/N
Gaussian
channel
[dB]
QPSK
1/2
3/5
2/3
3/4
4/5
5/6
1/2
3/5
2/3
3/4
4/5
5/6
1/2
3/5
2/3
3/4
4/5
5/6
1/2
3/5
2/3
3/4
4/5
5/6
0.4
2.2
3.1
4.0
4.6
5.1
5.2
7.5
8.8
10.0
10.8
11.4
8.7
12.0
13.4
15.2
16.1
16.8
12.1
16.5
17.7
19.9
21.2
22.0
16QAM
64QAM
256QAM
C/N
Rice
channel
[dB]
0.7
2.4
3.4
4.5
5.1
5.7
5.5
7.9
9.1
10.5
11.3
12.0
9.1
12.4
13.8
15.6
16.6
17.4
12.4
16.9
18.1
20.4
21.7
22.5
C/N
Rayleigh
channel
[dB]
1.5
3.5
4.7
6.0
6.9
7.8
6.6
9.3
10.7
12.4
13.6
14.6
10.7
14.2
15.7
17.8
19.1
20.3
14.4
18.8
20.3
22.6
24.2
25.6
C/N
0dB echo
channel
@ 90%
GI [dB]
1.2
3.2
4.4
5.7
6.5
7.4
6.3
9.0
10.4
12.1
13.3
14.4
10.5
14.0
15.5
17.6
18.9
20.3
14.3
18.8
20.3
22.7
24.3
25.9
Table 37.4. DVB-T2 channel capacity in the 8-MHz channel, 32K mode,
g=1/128, PP7 (Source: DVB-T2 Implementation Guidelines, February 2009)
Modulation
Code rate
QPSK
1/2
3/5
2/3
3/4
4/5
5/6
1/2
3/5
2/3
16QAM
Bit
[Mbit/s]
7.44
8.94
9.95
11.20
11.95
12.46
15.04
18.07
20.11
rate Frame length
[symbols]
60
60
60
60
60
60
60
60
60
FEC blocks
per frame
50
50
50
50
50
50
101
101
101
37. 6 COFDM Parameters
64QAM
256QAM
3/4
4/5
5/6
1/2
3/5
2/3
3/4
4/5
5/6
1/2
3/5
2/3
3/4
4/5
5/6
22.62
24.14
25.16
22.48
27.02
30.06
33.82
36.09
37.62
30.08
36.14
40.21
45.24
48.27
50.32
60
60
60
60
60
60
60
60
60
60
60
60
60
60
60
695
101
101
101
151
151
151
151
151
151
202
202
202
202
202
202
f
f
Channel bandwidth
10, 8, 7, 6, 5, 1.7 MHz
Fig. 37.3. DVB-T2 channel
37.6 COFDM Parameters
DVB-T2 supports channel bandwidths of 1.7, 5, 6, 7, 8 and 10 MHz (Fig.
37.3.). The actual signal bandwidth is slightly narrower because of the
guard band at the upper and lower end of the DVB-T2 channel (Table
37.5.). Table 37.3. shows the CODFM parameters in the 8-MHz channel
possible in DVB-T2. In the case of the 7- and 6-MHz and 1.7- and 10-
696
37 DVB-T2
MHz channel, respectively, (Fig. 37.3.), the parameters must be adapted
correspondingly by a factor of 7/8 and 6/8 etc.. As can be seen in Table
37.6., not every guard interval is possible in every COFDM mode. Apart
from one exception (P1 symbol), the guard interval (Fig. 37.4.) is also a
cyclic prefix (CP) in DVB-T2, i.e. a repetition of the symbol end in the
corresponding length.
Symbol n
Symbol n+1
Guard interval
Fig. 37.4. Symbol and guard interval as a cyclic prefix (CP)
Table 37.5. DVB-T2 channel- and signal bandwidths
Bandwidth
[MHz]
Elementary
period
[ s]
Signal
bandwidth
[MHz]
1.7
5
6
7
8
10
71/131
7/40
7/48
1/8
7/64
7/80
1.54
4.76
5.71
6.66
7.61
9.51
Table 37.6. DVB-T2 COFDM parameters in the 8-MHz channel
FFT
32K
16K
8K
4K
2K
1K
Symbol Carrier
period spacing
[ms]
[kHz]
f
3.584 0.279
1.792 0.558
0.896 1.116
0.448 2.232
0.224 4.464
0.112 8.929
g=
g=
1/128 1/32
g=
1/16
g=
g= g=
g=
19/256 1/8 19/128 1/4
X
X
X
----
X
X
X
X
X
X
X
X
X
----
X
X
X
X
X
--
X
X
X
X
X
X
X
X
X
----
-X
X
X
X
X
37. 6 COFDM Parameters
697
A 16K and a 32K mode was provided in order to achieve less time
overhead and thus a higher net data rate (6% overhead in the 32K mode,
25% in the 8K mode) with the same absolute guard interval length. The
longest guard interval is now more than twice as long as the longest guard
interval in DVB-T (0.224 ms); 0.532 ms in DVB-T2, 32K, g=19/128, correspond to a maximum transmitter distance of almost 160 km (Table
37.17.). Narrower signal bandwidths (7, 6, 5, 1.7 MHz) lead to even longer
symbols and thus also lead to longer guard intervals. With these guard interval-lengths nationwide single-frequency networks can be implemented.
The 32K mode is the mode providing the longest symbols and thus has the
least overhead in the guard interval; at the same time it serves to implement the largest single frequency networks. However, the 32K mode, due
to its much narrower subcarrier spacing, is the mode least suitable for mobile use. The mode most suitable for mobiles is the 1K mode which has the
largest subcarrier spacing. But it thus also has the shortest symbols and is,
therefore, the one least suitable for forming large SFNs.
f
Fig. 37.5. sin(x)/x-shaped spectra of the OFDM carriers
37.6.1 Normal Carrier Mode
In the DVB-T2 Normal Carrier Mode, the bandwidths of the useful signal
approximately correspond to the bandwidths of DVB-T. Between the useful signal spectrum and the beginning of the adjacent channel there is at
the lower and the upper end of the DVB-T2 channel the so-called guard
band which has a width of up to about 200 kHz. The guard-band has several tasks, the most important of which is to protect the adjacent channels
(Fig. 37.7.). The shoulders of the OFDM signal must decay within the
guard band. One cause of the shoulders is the superimposition of the tails
698
37 DVB-T2
of the sin(x)/x-functions of each modulated single carrier (Fig. 37.5.). It
can be demonstrated, however, that the more carriers are used, the more
the resultant shoulders are suppressed; i.e. the 1K mode inherently has
higher shoulders than the 32K mode. These shoulders are lowered as much
as possible by digital filtering measures in the modulator. Nevertheless, it
can also be demonstrated with a very good test transmitter that in the short
range, the shoulders are much lower in the 32K mode than in the 1K-Mode
(Fig. 37.6.). There are simple mathematical reasons for this. The more carriers are used, the better the sin(x)/x tails will cancel.
Fig. 37.6. Shoulders of the DVB-T2 signal in 32K and 1K mode in comparison
IFFT
bandwidth
Channel
bandwidth
Center carrier
Signal
bandwidth
Carrier # 0
Carrier # n
Fig. 37.7. DVB-T2 spectrum with top and bottom guard bands
37. 6 COFDM Parameters
699
37.6.2 Extended Carrier Mode
Since the sin(x)/x tails, and thus the shoulders, drop more towards the adjacent channels in the modes having more carriers, it was provided in
DVB-T2 to support a wider spectrum above the 8K mode in the Extended
Carrier Mode. The advantage is that this increases the data rate of the
DVB-T2 signal.
Fig. 37.8. and 37.9. clearly show the wider DVB-T2 spectrum of the
32K mode in the Extended Carrier Mode compared with the Normal Carrier Mode. In the selected example, the difference in the net data rate is
about 1 Mbit/s.
256QAM
CR=3/5
PP7
32K:
35.246Mbit/s
32K extended:
36.140 Mbit/s
Fig. 37.8. DVB-T2 spectrum in Normal and Extended Carrier Mode (32K)
Fig. 37.9. Overall DVB-T2 spectrum in Normal and Extended Carrier Mode
(32K)
700
37 DVB-T2
Table 37.7. DVB-T2 OFDM parameters in the 8 MHz channel
Parameter
Number of
carriers in
Normal Mode
Number of carriers in Extended
Carrier Mode
Additional carriers in Extended
Carrier Mode
IFFT
Symbol period
[ s]
Carrier spacing
f [kHz]
Signal bandwidth
in Normal
Mode [MHz]
Signal
bandwidth
in Extended
Carrier Mode
[MHz]
1K
Mode
853
2K
Mode
1705
4K
Mode
3409
8K
Mode
6817
16K
Mode
13633
32K
Mode
27265
--
--
--
6913
13921
27265
0
0
0
96
288
596
1024
112
2048
224
4096
448
8192
896
16384
1792
32768
3584
8.929
4.464
2.232
1.116
0.558
0.279
7.61
7.61
7.61
7.61
7.61
7.61
--
--
--
7.71
7.77
7.77
37.7 Modulation Patterns
The modulation patterns (Fig. 37.10.) used in DVB-T2 are coherent Graycoded QAM orders. The constellations possible in DVB-T2 are:
•
•
•
•
QPSK,
16QAM,
64QAM and
256QAM.
Differential modulation is not supported in DVB-T2. The first three QAM
orders fully correspond to the mapping used in DVB-T. It is a special feature that the constellation diagrams can be either reversed or rotated
("flipped") to the left by a certain angle; so-called rotated, Q-delayed constellation diagrams (Fig. 37.12.).
37. 7 Modulation Patterns
701
Fig. 37.10. Non-rotated "normal“ constellation diagrams in DVB-T2 (QPSK,
16QAM, 64QAM and 256QAM)
37.7.1 Normal Constellation Diagrams
In the case of QPSK, 16QAM and 64QAM, the normal non-rotated constellation diagrams exactly correspond to the constellation diagrams of
DVB-T. In addition, 256QAM was also defined as a possible constellation
in DVB-T2. 256QAM makes sense because of the improved error protection. Fig. 37.10. shows the non-rotated variants of the constellation diagrams possible in DVB-T2.
A cell is a mapped IQ value
n=2, 4, 6, 8
n bits
Mapping
I
2n
Q
= N constellation points
Cell
N=4, 16, 64, 256
Fig. 37.11. Definition of 'cell'
37.7.2 Definition of 'Cell'
The term "cell“ (Fig. 37.11.) will now have to be defined. This term is
mentioned time and again in the DVB-T2 Standard. Thus, there is also,
e.g. a so-called cell interleaver. A cell is simply the result of mapping a
later carrier. In contrast to DVB-T, mapping is not carried out after all interleaving processes but relatively early, after the error protection and after
the bit interleaver. However, this is still followed by the cell-interleaver,
the time interleaver and the frequency interleaver which is why the map-
702
37 DVB-T2
ping result cannot yet be allocated to a carrier and why the term "cell“ was
introduced. A cell is, therefore, a complex number consisting of an Icomponent and a Q-component, i.e. a real part and an imaginary part (Fig.
37.11.).
Q
I
Fig. 37.12. Rotated Q-delayed constellation diagram
Q
I
Fig. 37.13. Discrete mapping of constellation points on the I and Q axis with a rotated constellation diagram
37.7.3 Rotated Q-delayed Constellation Diagrams
If rotated constellation diagrams (Fig. 37.18.) are used, the information
about the position of a constellation point is contained both in the I component and in the Q component of the signal (Fig. 37.13.). In a case of dis-
37. 7 Modulation Patterns
703
turbance, this can be used for providing more reliable information about
the position of the constellation point, in contrast to a non-rotated diagram
(Fig. 37.14.), contributing to better decodability. In contrast to a nonrotated constellation diagram, the IQ information, which is now discrete,
can be used for soft-decisions if necessary. Practice will show how much
actual benefit can be derived from this. Table 37.5. lists the angles of rotation of the various DVB-T2-constellations as a function of the QAM
mode.
Table 37.8. Angles of rotation of the constellation diagrams
Mod.
[degrees]
QPSK
29.0
16QAM
16.8
64QAM
8.6
256QAM
atan(1/16)
Q
I
Fig. 37.14. The constellation points on the I and Q axis in a non-rotated constellation diagram
In reality, however, the whole process is slightly more complex. With a rotated diagram, the Q component is not transmitted on the same carrier, or
more precisely in the same "cell“, but with delay on another carrier (Fig.
37.15. and 37.16.) or better in another cell. From one QAM, virtually two
ASKs (Amplitude Shift Keying modulations) in the I and Q-direction are
then produced which are then transmitted on independent carriers – "cells“
which are disturbed differently in practice and are thus intended to contribute to the reliability of demodulation.
704
37 DVB-T2
n=2, 4, 6, 8
n bits
Mapping
I
Q
2n = N constellation points
Cell
from
k-1
N=4, 16, 64, 256
Rotation
I
I
Q
Q-delay
Cellk
Q
Cellk+1
Fig. 37.15. Mapping, rotation and Q delay
Cell 4
I4
Q3
Cell 5
I5
Q4
Cell 6
I4
Q5
Cell 7
I4
Q6
Fig. 37.16. Cyclic Q delay between adjacent cells
37.8 Frame Structure
A Physical Layer Frame (Fig. 37.17.) in DVB-T2 begins with a P1 symbol
used for synchronization and frame-finding, followed by one or more P2symbols containing Layer-1 (L1)-signalling data for the receiver. This is
followed by symbols which carry the actual payload data. The theoretically
up to 255 input data streams are transmitted in so-called Physical Layer
Pipes (PLPs) in which the different contents can be transmitted with higher
or lower data rate and more or less robustly (error protection and modulation). This is called Variable Coding and Modulation (VCM). In addition,
the transmission parameters of the PLPs can also be changed dynamically
from T2 frame to T2 frame. The current transmission parameters of all
PLPs are signalled in the P2 symbols; dynamic L1 signalling for the receiver takes place in the padding-field of the baseband frame.
37. 8 Frame Structure
705
T2 frame
P1
P2
PLP0 PLP1
Payload symbols PLPx
P2 symbols containing signalling data
P1 symbol for synchronization and frame detection
Fig. 37.17. Structure of a DVB-T2 frame
Table 37.9. Maximum length of a DVB-T2 frame in numbers of OFDM symbols
FFT
32K
16K
8K
4K
2K
1K
Symbol
duration
in a
8 MHz
channel
[ s]
3584
1792
896
448
224
112
g=
1/128
g=
1/32
g=
1/16
g=
19/256
g=
1/8
g=
g=
19/128 1/4
68
138
276
----
66
135
270
540
1081
--
64
131
262
524
1049
2098
64
129
259
----
60
123
247
495
991
1982
60
121
242
----
-111
223
446
892
1784
Table 37.10. Number of P2 symbols per DVB-T2 frame as a function of the
DVB-T2 FFT mode
FFT mode
1K
2K
4K
8K
16K
32K
Number of P2symbols per DVBT2 frame
16
8
4
2
1
1
Apart from the FFT mode, almost all transmission parameters can be
changed from Physical Layer Pipe (PLP) to Physical Layer Pipe. As already mentioned, their signalling and the addressing of the PLP (Start,
Length etc.) is handled via the P2 symbols; called L1-signalling. The num-
706
37 DVB-T2
ber of P2 symbols depends on the FFT mode (Table 37.10.); the reason being simply the different data capacity of the symbols, depending on the
FFT mode. The 1K mode has the shortest symbols and thus the lowest data
capacity per symbol. In the 32K mode, it is possible to transmit more data
per symbol due to the much longer symbols. This is possible to the extent
that the data transmission can even begin in the P2 symbols due to unused
capacity. Table 37.10. lists the number of P2-Symbols per DVB-T2 frame.
A DVB-T2 frame thus consists of
•
•
•
a P1 symbol
1 … 16 P2 symbols (depending on FFT mode)
N data symbols (PLP data, FEF, auxiliary data, dummy cells)
A frame can have a maximum length of 250 ms, resulting in a maximum
number of data symbols which, in turn, is dependent on the FFT mode and
the guard interval. The net data rate per PLP can fluctuate due to different
transmission parameters. A data stream carrying HDTV services, e.g., requires a higher data rate than a data stream transporting SDTV services or
a data stream transporting pure audio broadcasting services (Fig. 37.18.).
Radio1
HDTV2
SDTV1
HDTV1
Data
rate
Time
Fig. 37.18. Variable Coding and Modulation (VCM)
37.8.1 P1 Symbol
A P1 symbol (preamble symbol 1) marks the beginning of a frame, similar
to the null symbol in DAB. Overall, the P1 symbol is used for
•
•
marking the beginning of the DVB-T2 frame,
time and frequency synchronization,
37. 8 Frame Structure
•
707
signalling the basic transmission parameters (FFT mode,
SISO/MISO),
The P1 symbol has the following characteristics:
•
•
•
•
FFT mode = 1K,
1/2 guard interval with frequency offset before and after the P1
symbol,
carrier DBPSK modulated,
7-bit signalling data (SISO/MISO/Future Use (3 bits), use of FEF
(1 bit), FFT (3 bits)).
So that the P1 symbol (Fig. 37.19.) could be identified easily and reliably,
two guard intervals were appended – one in front and one behind. This results virtually in a double correlation during the autocorrelation. In addition, the carriers are all shifted upward in frequency in the cyclical preand post-fix. Pre- and post-fix are not exactly of the same length.
Guard
+ f
542 samples
P1
1024 samples
Guard
+ f
482 samples
Fig. 37.19. P1 symbol
37.8.2 P2 Symbols
The Layer 1 signalling (L1 signalling) is transmitted from the modulator to
the receiver in 1 ... 16 P2 symbols (preamble symbols 2) per DVB-T2
frame. Physically, a preamble symbol 2 has almost the same structure as
later data symbols. The FFT mode corresponds to that of the data symbols
and is already signalled in the P1 symbol. However, the pilot density is
greater.
A P2 symbol (Fig. 37.20.) consists of a pre- and post-signalling component. Both components are differently modulated and error protected. The
708
37 DVB-T2
pre-signalling-component is permanently BPSK-modulated and protected
with a constant error protection known to the receiver. The transmission
parameters of the P2-pre-signalling component are:
•
•
•
BSPK modulation,
FEC = BCH+16K LDPC,
LDPC code rate =1/2.
The transmission parameters of the P2 post-signalling component are:
•
•
•
•
BPSK, QPSK, 16QAM or 64QAM,
FEC = BCH+16K LDPC,
LDPC code rate =1/2 or 1/4 with BPSK,
LDPC code rate =1/2 with QPSK, 16QAM or 64QAM.
P2 data in part 1 (constant length, L1 pre-signalling):
•
•
•
•
•
•
•
Guard interval
Pilot pattern
Cell ID
Network ID
PAPR use
Number of data symbols
L1 post-signalling parameters (FEC and mod. of L1 post)
P2 data in part 2 (variable length, L1 post-signalling):
•
•
•
•
Number of PLPs
RF frequency
PLP IDs
PLP signalling parameters (FEC and mod. of PLP)
In the higher FFT modes, not all the carriers are needed for the L1 signalling. This free capacity can then be used already for the actual data
transmission. I.e., the transmission of the PLPs can start already in the P2
symbols. Although this sounds somewhat adventurous to someone with
years of experience and probably doesn't simplify its implementation, either, it does bring additional capacity.
37. 9 Block Diagram
709
37.8.3 Symbol, Frame, Superframe
A DVB-T2 frame is composed of a P1 symbol, 1 to 16 P2 symbols and N
data symbols which can contain PLP data, Future Extension Frames and
Auxiliary Data , as well as dummy cells. Several frames, in turn, become
one superframe.
P2
P2
L1 postsignalling
L1 presignalling
Fig. 37.20. P2-Symbols
TSF
Super frame
Super frame
…
Super frame
TF
T2-frame 0
P1
P2
0
TP1
TS
T2-frame 1
…
T2-frame 2
Data Data
P2
NP2-1 symb. symb.
1
0
…
FEF
Data
symb.
Ldata-1
… T2-frame N
T2-1
FEF
FEF = Future Extension Frame
TS
Fig. 37.21. Symbol, frame and superframe
37.9 Block Diagram
The time has come to turn to the complete block diagram of the DVB-T2
modulator. A comparison with the DVB-T modulator will demonstrate the
colossal scale of the DVB-T2 Standard (Fig. 37.22.).
T2-MI
PLP n
Cell
builder
Mapper
Cell
interleaver
Time
interleaver
Cell
builder
Mapper
Cell
interleaver
Time
interleaver
......
FEC
......
Stream adaptation part 2
Mode/stream adaptation
PLP 0
FEC
T2gateway
Modulator
Pilot
insert.
IFFT
PAPR
red.
Guard
interv.
insert.
P1
symbol
Insert.
D
Pilot
insert.
IFFT
PAPR
red.
Guard
interv.
insert.
P1
symbol
Insert.
D
MISO
proc.
Fig. 37.22. Block diagram of the DVB-T2 modulator
PLP0
Mapping
Time
interleaver
Cell
interleaver
Time
interleaver
Cell
interleaver
Time
interleaver
Q
I
PLP1
Bit
interleaver
FEC
Cell
interleaver
Mapping
Q
PLPn
I
Bit
interleaver
Mapping
Frame builder
I
Bit
interleaver
FEC
FEC
Q
Frequency
interleaver
Fig. 37.23. Interleavers in DVB-T2
37.10 Interleavers
In DVB-T2, interleaving (Fig. 37.23.) is separated into
•
•
Frequency
interleaver
37 DVB-T2
Frame
builder
710
bit interleaving,
cell interleaving,
IFFT
A
Tx1
To
transmitters
A
Tx2
(optional)
37.10 Interleavers
•
•
711
time interleaving, and
frequency interleaving.
The bit interleaver is a very short interleaver. It corresponds to the
DVB-S2 bit interleaver and like that operates at the FEC frame level. The
bit interleaver has the task of optimizing the characteristics of the error
protection for immunity against burst errors, independently of the other interleavers. The cell interleaver also operates at FEC frame level and interleaves the cells already mapped, i.e. the IQ values. It improves the performance mainly in conjunction with the rotated and Q-delayed
constellations. The time interleaver then distributes the information over a
time which can be adjusted within wide ranges. The time interleaver is of
assistance mainly with mobile reception, with long burst errors and with
impulsive noise when the frequency interleaver distributes the information
as randomly as possible to the various DVB-T2 OFDM carriers. Notches
due to multipath reception will then lead to reduced losses.
37.10.1 Types of Interleaver
Back in the 1960s, David Forney had the idea of improving error protection with regard to susceptibility to burst errors by applying time interleaving. Interleavers are intended to distribute a data stream in time or frequency during transmission as randomly as possible, but so as to be
recoverable by receivers. There are
•
•
block interleavers and
PRBS interleavers.
The interleavers used in DVB-T2 are either block interleavers or PRBS interleavers.
Table 37.11. Interleaver types in DVB-T2
Interleaver
Bit interleaver
Cell interleaver
Time interleaver
Frequency interleaver
Interleaver type
Block interleaver
PRBS interleaver
Block interleaver
PRBS interleaver
A block interleaver will first read the data, e.g. line by line, into a block
and then read them out again, e.g. column by column, or in zig-zag form.
712
37 DVB-T2
A PRBS interleaver is controlled by a pseudo random sequence and distributes the data even more randomly.
FEC blocks
of one PLP
TI block
Time
interleaver
Interleaving frame
Fig. 37.24. DVB-T2 time interleaving
TIME_IL_TYPE = 0
k FEC blocks
TI block
TI block
TI block
Interleaving
frame
Interleaving
frame
Interleaving
frame
T2 frame
T2 frame
T2 frame
0 < k < PLP_NUM_BLOCKS_MAX
k is a varying number
Fig. 37.25. Time interleaver, type 1
37.10.2 DVB-T2 Time Interleaver Configuration
The time interleaver (Fig. 37.24.) has the task of distributing the data of a
PLP over a very long period if possible (several hundred milliseconds).
37.10 Interleavers
713
This increases robustness against burst errors. Burst errors can occur
mainly in mobile reception and with impulsive noise. In the DVB-T2 time
interleaver, several FEC blocks are combined to form one or more time interleaving blocks and these are then interleaved and result in an interleaving frame. The time interleaver in DVB-T2 can be set by the following
configuration parameters:
•
•
•
•
TIME_IL_TYPE (1 Bit) 0 or 1
TIME_IL_LENGTH (8 Bit) in blocks per interleaving frame
FRAME_INTERVAL (8 Bit) in frames
PLP_NUM_BLOCKS_MAX (10 Bit) in blocks 0 … 1023
k FEC blocks
TIME_IL_TYPE = 1
TI block
Interleaving
frame
T2 frame
T2 frame
T2 frame
FRAME_INTERVAL = 2
0 < k < PLP_NUM_BLOCKS_MAX
k is a varying number
Fig. 37.26. Time interleaver, type 2
These enable the time interleaver to be set within wide ranges. The TI
configuration parameters of each PLP are signalled via the L1 postsignalling part in the P2 symbols.
The TIME_IL_TYPE can be used for deciding whether the time interleaving is to take place within one T2 frame (TIME_IL_TYPE = 0) or distributed over several T2 frames (TIME_IL_TYPE = 1).
TIME_IL_LENGTH specifies the number of interleaving blocks per
time interleaving frame.
FRAME_INTERVAL defines the interval between two frames carrying
the time interleaving data of a PLP. I.e., gaps or T2 frames can now be inserted between interleaving data of PLPs and thus greater interleaving intervals can be achieved.
714
37 DVB-T2
PLP_NUM_BLOCKS_MAX specifies how many FEC frames may be
maximally combined to form one time interleaving block.
TIME_IL_TYPE = 0
TI
block 0
TI
2
TI
block 3
TIME_IL_LENGTH = 3
Interleaving
frame
T2 frame
T2 frame
T2 frame
Fig. 37.27. Time Interleaver, Typ 3
Having selected these TI configuration parameters, 3 time interleaver
types can now be implemented, namely
•
•
•
Typ 1: a time interleaver block in one time interleaving frame,
mapped in exactly one T2 frame (TIME_IL_TYPE=0,
TIME_IL_LENGTH=1) (Fig. 37.25.),
Typ 2: a time interleaver block in one time interleaving frame,
mapped in several (n) T2 frames with a definable frame interval
between them (TIME_IL_TYPE=1, FRAME_INTERVAL=n)
(Fig. 37.26.) and
Typ 3: a definable number (m) in one time interleaving frame
mapped
in
one
T2
frame
(TIME_IL_TYPE=0,
TIME_IL_LENGTH=m) (Fig. 37.27.).
Each of these 3 time interleaver-types has different TI characteristics
which are listed in Table 37.12.
Table 37.12. Characteristics of the TI types in DVB-T2
TI type
1
2
3
Characteristics/Application
Better interleaving at medium data rates than with type 3
Better interleaving at low data rates
Application at higher data rates, but less interleaving than with Type 1
37.11 Pilots
715
Frequency (carrier no.)
d2
0
i
Time
….
….
d3
d1
….
….
d1 = distance between scattered pilot carrier positions
d2 = distance between scattered pilots in one symbol
d3 = symbols forming one scattered pilot sequence
Fig. 37.28. Parameters for pilot pattern in DVB-T2
37.11 Pilots
In COFDM systems, the following tasks must basically always be implemented by special-carriers:
•
•
•
Frequency lock (AFC = Automatic Frequency Control),
Channel estimation – and channel correction, and the
Signalling of the transmission parameters.
For this purpose, DVB-T has the following pilot signals, namely
•
•
•
the Continual Pilots for the AFC,
the Scattered Pilots for the channel estimation and
the TPS carriers for the signalling.
DVB-T2 has the following pilots, namely
•
•
Edge pilots at the beginning and the end of the channel,
Continual pilots,
716
37 DVB-T2
•
•
•
Scattered pilots,
P2 pilots at every 3rd carrier position,
Frame-closing pilots for cleanly closing a frame.
Table 37.13. Pilot pattern in DVB-T2
Pilot pattern
PP1
PP2
PP3
PP4
PP5
PP6
PP7
PP8
Distance d1 between pilot
carrier positions (and distance d2 of the pilots within
a symbol)
3 (12)
6 (12)
6 (24)
12 (24)
12 (48)
24 (48)
24 (96)
6 (96)
Number of symbols d3 forming a
pilot sequence
4
2
4
2
4
2
4
16
Table 37.14. Scattered pilot pattern in SISO mode
FFT
Mode
32K
g=1/128 g=1/32
g=1/16
g=19/256 g=1/8
g=19/128 g=1/4
PP7
PP4,
PP6
PP2,
PP8
--
PP7
PP7,
PP4,
PP6
PP2,
PP3,
PP8
PP2,
PP3,
PP8
PP1,
PP8
8K
PP7
PP7,
PP4
PP2,
PP8,
PP4
PP2,
PP8,
PP4,
PP5
PP8,
PP4,
PP5
--
PP2,
PP8
16K
PP2,
PP8,
PP4
PP2,
PP8,
PP4,
PP5
PP8,
PP4,
PP5
PP4,
PP5
PP4,
PP5
PP2,
PP3,
PP8
PP2,
PP3
PP2,
PP3
PP2,
PP3,
PP8
--
PP1,
PP8
--
PP1
4K, 2K -1K
--
PP7,
PP4
--
--
PP1
In DVB-T2, the information of the previous TPS carriers is contained in
the P2 symbols. There are no longer any TPS carriers, but there are continual pilots and scattered pilots. In addition, the term Edge Pilot was introduced; it is not actually a special feature and was already provided in
DVB-T where it fell under the category of Continual Pilot. The edge pilots
37.12 Sub-Slicing
717
are simply the first and last pilot in the spectrum. The scattered pilots have
several selectable, more or less dense, pilot patterns. Less dense pilot patterns means that there are more payload carriers, resulting in a higher net
data rate. Denser pilot patterns (Fig. 37.28. and Tab. 37.13.), however, allow for a better channel estimation especially in the presence of difficult
reception conditions such as multipath reception and mobile reception.
When planning the network, the corresponding pilot pattern can then be
selected in dependence on the planned coverage. Not all pilot patterns
(called PP1 to PP8) can be used in all mode and guard interval configurations.
Table 37.15. Scattered pilot pattern in MISO mode
FFT
Mode
32K
g=1/128 g=1/32
PP8,
PP4,
PP6
16K
PP8,
PP4,
PP5
8K
PP8,
PP4,
PP5
4K, 2K --
1K
--
g=1/16
g=19/256 g=1/8
g=19/128 g=1/4
PP8,
PP4
PP2,
PP8
PP2,
PP8
--
--
--
PP8,
PP4,
PP5
PP8,
PP4,
PP5
PP4,
PP5
--
PP3,
PP8
PP3,
PP8
PP1,
PP8
PP1,
PP8
--
PP3,
PP8
PP3,
PP8
PP1,
PP8
PP1,
PP8
--
PP3
--
PP1
--
--
PP3
--
PP1
--
--
Table 37.16. Amplitudes of the scattered pilots
Scattered pilot pattern
PP1, PP2
PP3, PP4
PP5, PP6, PP7, PP8
Amplitude
4/3
7/4
7/3
Equivalent boost [dB]
2.5
4.9
7.4
37.12 Sub-Slicing
Without sub-slicing (Fig. 37.29.), a PLP will arrive in the receiver in one
piece in a timeslot in DVB-T2. I.e. the peak data rate may be relatively
high for a PLP and then there may be no further reception of these data for
a relatively long period. Sub-slicing divides the PLPs into smaller "morsels“ which are then transmitted synchronously from PLP to PLP in the
DVB-T2 frame. A PLP can be divided into 2 to 6480 subslices in sub-
718
37 DVB-T2
slicing. The subslices of the various PLPs then follow one another synchronously within a T2 frame. More subslices means more time diversity
and less buffer memory demand, and fewer subslices means less time diversity, but offers the possibility of saving more energy in the receiver.
Sub-slice
1
2
3
Sub-slice
interval
…
M
1
2
…
3
M
1
2
T2-frame
Fig. 37.29. Sub-slicing
37.13 Time-Frequency-Slicing (TFS)
Time-Frequency-Slicing (TFS) is mentioned as an option in the Appendix
to the DVB-T2 Standard. This would make it possible to radiate the PLPs
or their subslices in up to 8 different RF channels. The complexity would
be very high at the transmitting end since up to 8 transmitting trains would
have to be implemented and the receiver would then require at least two
tuners. It is questionable whether TFS will become reality, although this is
already contained as a requirement in the first version of the NorDig-Spec
for DVB-T2 receivers.
Peak to Average Power Reduction
ACE
Active Constellation
Extension
TR
Tone Reservation
Modification
of constellation
Using addional
reserved carriers
Fig. 37.30. PAPR (Peak to Average Power Ratio Reduction)
37.14 PAPR Reduction
719
37.14 PAPR Reduction
PAPR reduction stands for Peak to Average Power Ratio Reduction (Fig.
37.30.) and means nothing else than the reduction of the crest factors. The
crest factor is the ratio of the maximum peak voltage to the RMS value. In
theory, the crest factor can assume very high values in COFDM systems.
In practice it is maximally about 12 to 15 dB, clipped at about 12 to 13 dB
in the power transmitter. There have been relevant discussions and
contributions since the beginning of the applications of COFDM. In order
to be able to limit this crest factor in DVB-T2, two methods of PAPR are
provided, namely Active Constellation Extension (ACE) (Fig. 37.31.) and
Tone Reservation (TR).
Fig. 37.31. PAPR – ACE – Active Constellation Extension
In the case of the Active Constellation Extension, the fact is used that
the outermost constellation points could be shifted further out within certain limits, without restriction in the demodulation, in order to reduce the
current crest factor by the summation of all carriers and by suitably adapting certain carrier amplitudes. However, the ACE is not possible with rotated constellation diagrams which is why this method will probably not be
used in DVB-T2. In the case of the Tone Reservation, certain carrier bands
are not intended for payload data transmission and also not for pilot tones.
If necessary, these carriers, which are normally not switched on, can be
activated so that they reduce the crest-factor. It is then up to the respective
DVB-T2 modulator in its characteristics determined by the respective
manufacturer to set these carriers in amplitude and phase in such a way
that they really decisively reduce the crest-factor at the transmitter site.
720
37 DVB-T2
This is of minor importance in the interplay with neighboring sites in an
SFN.
PAPR can be used mainly for increasing the efficiency of the transmitter output stages which would be a real cost saving for the operation. It
remains to be seen whether the gain is so great with regard to the dielectric
strength of the downstream passive transmitter components. However, due
to the crest factor, there has always been a great need for discussion with
respect to the correct dimensioning especially in the case of the mask filters, the antenna combiner, the antenna cable and the antenna itself.
Data
Tx
Rx
Data
Fig. 37.32. SISO = Single Input – Single Output
Data
Tx1
Rx
Data
Fig. 37.33. SIMO = Single Input – Multiple Output
37.15 SISO/MISO Multi-Antenna Systems
DVB-T2 contains MISO (Multiple Input/Single Output) as an option. This
means that possibly two transmitting antennas may be used which, however, do not radiate the same transmitted signal as in an SFN. Instead, adjacent symbols are transmitted repeatedly once by one and once by the
other transmitting antenna in accordance with the modified Alamouti principle. This is an attempt to come closer to the Shannon limit, especially in
the mobile channel. SISO (Single Input/Single Output) is the traditional
case of a terrestrial transmission link (Fig. 32.32.).
This arrangement uses exactly one transmitting and receiving antenna
(Fig. 32.32.). SIMO (Single Input/Multiple Output) corresponds to diversity-reception with one transmitting antenna and several receiving antennas (Fig. 32.33.) In motor vehicles, in some cases 2 – 4 receiving antennas
37.15 SISO/MISO Multi-Antenna Systems
721
bonded to the vehicle's windows are employed for mobile DVB-T reception.
Tx1
Rx
Data
Data
Tx2
Fig. 37.34. MISO = Multiple Input – Single Output
37.15.1 MISO according to Alamouti
In the case of MISO according to Alamouti, however, two transmitting antennas and one receiving antenna are used. The aim is to save on antenna
expenditure in the receiver by changing to transmit diversity (Fig. 37.34.).
This is also called space/time diversity according to [ALAMOUTI]. A further possibility would also be MIMO (Multiple Input/ ultiple Output) with
several transmitting and receiving antennas (Fig. 37.35.). The idea of the
MISO principle goes back to [ALAMOUTI], 1998. This principle is already being used in mobile radio (WiMAX) where adjacent symbols
(COFDM symbols) are repeated at the two transmitting antennas. At the
receiving antenna, a superimposed grouping of adjacent symbols always
arrives which, without modification, would result in mutual interference
and could thus no longer be separated in the receiver. In the case of the
Alamouti principle, on the other hand. the adjacent symbols are not radiated unmodified at the various transmitting antennas, but in accordance
with the Alamouti code (Fig. 37.36.). According to [ALAMOUTI], the two
adjacent symbols sn are first present at antenna 1 and sn+1 at antenna 2.
Then symbol sn+1 is applied in negative conjugate complex form to antenna
1 and at the same time symbol sn is radiated in conjugate complex form at
transmitting antenna 2.
This will enable the receiver (Fig.32.37.) to separate two adjacent symbols again by means of suitable complex mathematical operations on these
symbols (Fig. 37.38.).
722
37 DVB-T2
In addition, the channel transfer function from transmitting antenna 1 to
the receiver and transmitting antenna 2 to the receiver must be known. I.e.,
it is necessary to perform a channel estimation over all transmitting and receiving paths.
Tx1
Data
Rx
Data
Tx2
Fig. 37.35. MIMO - Multiple Input – Multiple Output
S1
-S2*
S2
S1*
S1
-S2*
1
-()*
S3
-S4*
Tx1
Path 1
S1
S2
Alamouti Matrix
-S2*
S1*
Symbol
Data
FEC
Mapper
Sn
Sn+1
Data
Rx
Path 2
()*=conjugate complex
1
Tx2
()*
Symbol
S2
S1*
S4
S3*
Fig. 37.36. MISO principle according to Alamouti
37.15.2 Modified Alamouti in DVB-T2
It is intended to employ the MISO principle both at only one site and distributed over SFN sites. Used at one site, this can be done by horizontal
and vertical polarization. However, DVB-T2 uses a modified Alamouti
37.15 SISO/MISO Multi-Antenna Systems
723
coding (Fig. 32.39.). At antenna 1, the cells c1, c2, c3, c4, ... are present unchanged. It is only at antenna 2 that correspondingly changed cells are radiated. This has the advantage that the DVB-T2 system can be easily
reduced to SISO by simply omitting the second transmitting path.
r1 r2
S1
S2
-S2*
S1 *
channel
estimation
r1
Rx
r2
combiner
max. likelihood
detector
data
Fig. 37.37. MISO signal reception in the receiver
time
t1
t2
path1
S1
-S2*
path2
S2
S1*
received symbols:
r1=s1+s2
r2=-s2*+s1*
Alamouti Matrix
Combining rule in the receiver:
s1~=r1+r2*=(s1+s2)+(-s2*+s1*)*=s1+s1=2s1;
s2~=r1-r2*=(s1+s2)-(-s2*+s1*)*=s2+s2=2s2;
()*=conjugate complex
Fig. 37.38. MISO signal processing in the receiver
724
37 DVB-T2
C1
C1
C2
C2*
-C1*
C2
C3
C4
Tx1
1
C1 C2
C2* -C1*
1
Modified Alamouti Matrix
Cell
Data
FEC
Mapper
Cn
Cn+1
Data
Rx
()*
-()*
Tx2
Cell
C2* -C1* C4* -C3*
Fig. 37.39. Modified Alamouti in DVB-T2
Transmitter 1
+
Transmitter 2
Result: deep notches
Fig. 37.40. Reception of two signal paths in an SFN without MISO, with fading
notches
This alone is not all, however. In DVB-T2, MISO is not applied via
space/time diversity but via space/frequency diversity (based on adjacent
cells in the spectrum). I.e. at transmitting antenna 2, adjacent pairs of car-
37.15 SISO/MISO Multi-Antenna Systems
725
riers are radiated interchanged compared with those radiated at transmitting antenna 1. One great advantage of this modified Alamouti principle in
DVB-T2 is that the signals from transmitting antenna 1 and 2 are no longer
correlated with one another. This makes it possible to avoid the notches
prevalent in DVB-T and DAB, especially when using "distributed MISO"
in an SFN (Fig. 37.40. and 37.41.). The principle of MISO by H- and Vpolarization is called co-located MISO in DVB-T2.
No correlation
between
signal M1
and M2
+
Transmitter 1
MISO 1
Transmitter 2
MISO 2
Distributed
MISO
Result: no notches
Fig. 37.41. Reception of two signal paths in a distributed DVB-T2-MISO, without
fading notches
T2 frame
P1
P2
Common
PLP‘s
data shared by
multiple PLP‘s
Data PLP‘s
type 1
one slice per
T2-frame
Data PLP‘s
type 2
Auxillary
data
two or more
sub-slices
per T2-frame
Fig. 37.42. DVB-T2 frame with auxiliary stream data
Dummy
cells
726
37 DVB-T2
37.16 Future Extension Frames
DVB-T2 inherently already provides for possible expansion in so-called
Future Extension Frames. These are special frames with as yet undefined
transmission parameters which can be tied into the DVB-T2-frame structure. They are signalled via the P2 symbols.
37.17 Auxiliary Data Streams
At the end of a DVB-T2 frame (Fig. 37.42.), auxiliary stream data can still
be appended. These are customer-designed error-protected and mapped IQ
values. A normal DVB-T2 receiver does not need to be able to evaluate
these data.
37.18 DVB-T2-MI
In order to be able to conduct several (up to 255) data streams synchronously to the DVB-T2 modulators and transmitters in Mode B (Multiple
Physical Layer Pipes), the T2-MI modulator interface was defined. Apart
from supplying the data streams, it also handles the control and signalling.
The DVB-T2-MI interfaces are described in Chapter 36.
37.19 SFNs in DVB-T2
Naturally, it should be possible to implement single frequency networks
also in DVB-T2, the main reason being the economic use of frequencies.
Frequencies are expensive and are becoming ever more scarce. Being able
to reuse the same frequency is, therefore, important. Single frequency networks allow this at several adjacent transmitter sites in isolated, singlefrequency networks. DVB-T2 additionally allows larger interference-free
single frequency networks to be formed.
Single frequency networks must meet the following conditions:
•
•
•
Frequency synchronism,
Time synchronism,
Data synchronism,
37.19 SFN S in DVB-T2
•
727
Guard interval condition, i.e. maximum transmitter spacing must
not be exceeded (see table 37.17 for a 8 MHz wide DVB-T2 channel).
Table 37.17. Guard interval sizes in DVB-T2 (8 MHz-channel)
Mode Symbol
duration
[ms]
32K
3.584
16K
1.792
8K
0.896
4K
0.448
g=
1/128
t [ms]
d [km]
0.028
8.4
0.014
4.2
0.007
2.1
--
2K
0.224
--
1K
0.112
--
g=
1/32
t [ms]
d [km]
0.112
33.6
0.056
16.8
0.028
8.4
0.014
4.2
0.07
2.1
--
g=
1/16
t [ms]
d [km]
0.224
67.2
0.112
33.6
0.061
16.8
0.031
8.4
0.016
g=
19/256
t [ms]
d [km]
0.266
79.7
0.133
39.9
0.067
19.8
--
4.2
--
--
g=
1/8
t [ms]
d [km]
0.448
134.3
0.224
67.2
0.112
33.6
0.056
16.8
0.028
8.4
0.014
4.2
g=
19/128
t [ms]
d [km]
0.532
159.5
0.266
79.75
0.133
39.89
----
g=
1/4
t [ms]
d [km]
-0.448
134.3
0.224
67.2
0.112
33.6
0.056
16.8
0.028
8.4
With d = t • 299792458 m/s;
correction factor for
10 MHz = 8/10, 8 MHz = 1, 7 MHz = 8/7, 6 MHz = 8/6,
5 MHz = 8/5 and 1.7 MHz = 8/1.7.
Frequency synchronism is achieved by frequency standards at the transmitter site, generally a professional GPS receiver providing a 10-MHz reference. Time and data synchronism is achieved by time stamps in the baseband feed signal. This is the T2-MI signal in DVB-T2. The DVB-T2
modulator synchronizes its frame structure to these time stamps. The guard
interval condition is met by suitable network planning with planning software [LStelcom]. The new factor in DVB-T2 is the possibility of distributed MISO. There are transmitter sites which radiate either MISO Mode 1
or 2. The advantage of this method is that destructive fading no longer occurs between two adjacent transmitter sites. However, the appropriate
choice and simulation of the MISO modes is also a new challenge for the
network planning.
728
37 DVB-T2
37.20 Transmitter Identification Information in DVB-T2
A the time the present edition went to press, an ETSI Draft was being prepared which is intended to enable transmitters in an SFN to be identified
by using measurement techniques as is done in DAB. DAB has the TII
signal in the Null symbol for this purpose. Transmitter identification was
not possible in DVB-T since this would have violated synchronism and
would have led to severe disturbances in an SFN. The cell ID in DVB-T
only allowed an SFN cell to be identified. The cell ID is here built into reserved TPS bits.
DVB-T2 has the capability of inserting Auxiliary Streams and FEFs
(Future Extension Frames) into a T2-frame. Normal DVB-T2 receivers
will ignore the Auxiliary Streams and FEFs. DVB-T2 provides for such
transmitter identifiers or transmitter signatures to be inserted into the T2
frames either via the Auxiliary Streams or via the FEFs. Similar to DAB,
these would be certain active carriers which are activated only at individual carriers within the Auxiliary Streams and would be switched off at all
other transmitters in an SFN. By this means, the signals of the channel impulse response in an SFN can then be correlated with the individual transmitters. The signalling of the transmitter identification via the FEFs provides for broadcasting certain different signatures or sequences over the
transmitters which can be identified by correlation and thus also enable the
individual transmitters in an SFN to be recognized. The additional signalling of the transmitter identification is an overhead which, however, should
occupy only very little payload data rate. In DVB-T, transmitter identification has always been wished for by the maintenance technicians who had
to perform the coverage tests, but it had been a wish which could not be
fulfilled. In DVB-T2, as with DAB, however, this provides both for more
comfortable calibration and balancing of delays in a single-frequency network, and for monitoring these.
37.21 Capacity
The aim was to achieve 30% better capacity compared with DVB-T. In
some cases, the data rate will be up to 50% higher under comparable conditions. And that is already enormous. To achieve this, this Standard is also
more complex than DVB-T by about 150%. In combination with the new
video and audio compression standards such as MPEG-4 H.264/AVC and
MPEG-4 AAC this results in a huge increase in effectiveness in respect to
program variety and quality. It may well be that this Standard is so far
37.22 Outlook
729
ahead of its time that there will be peace and quiet in this corner for some
years to come.
37.22 Outlook
In areas where DVB-T has just been successfully introduced and analog
television belongs to the past it will be initially difficult to apply new models of coverage without harming the existing one. And that applies to many
countries in Central Europe, but also worldwide. In a lecture at the IRT in
2008, it was quoted, e.g. "For Germany, DVB-T2 is either 5 years too
early or 5 years too late" [IRT2008_KUNERT]. But there may also be applications which do not have the aim of initially replacing traditional
DVB-T but to find a replacement for applications which were planned for
DAB. There are standards which have been on air longer than DVB-T and
have not really been able to gain the attention of the consumer. DVB-T2
can also be used for broadcasting pure audio.
Bibliography:
[DVB_A122r1],
[DVB-T2],
[FKTG2008_GUNKEL], [IRT2008_KUNERT]
[ALAMOUTI],
38 DVB-C2 – the New DVB Broadband Cable
Standard
It has been apparent for some time that after DVB-T2, there would also be
a new DVB broadband cable standard, the "Call for Papers" having been
issued at the end of 2007. At the IBC 2008, rough outlines were then published on a DIN A4 sheet. But it was not quite clear then whether this
would only amount to a multi-carrier method or whether single-carrier
modulation would also be a component of DVB-C2. DVB-C2 working
documents based on DVB-T2 were then available at the end of 2008 and in
the spring of 2009 the time had come: DVB-C2 was published as a Draft.
38.1 Introduction
There are many formulations in DVB-C2 which are derived straight from
DVB-T2. And it is also true that it is relatively easy to find one's way
around DVB-C2 if one knows DVB-T2. Like DVB-T2, DVB-C2 uses
COFDM as a modulation method, the only difference being that the guard
intervals are very short and there is only a 4k mode. And the constellation
diagrams extend from QPSK up to 4096QAM. At first glance, it appears to
be surprising that up to 4096QAM is provided but rough estimates of parameters known from DVB-T2 show that 4096QAM is possible with signal-to-noise ratios which are easily achieved in modern broadband cables.
Such high-level types of modulation are possible mainly because of the error protection used in DVB-C2, which corresponds to the error protection
used in DVB-S2 and DVB-T2. DVB-C2 thus also uses BCH and LDPC
coding. And modern broadband cable networks are continuously improving with respect to the signal-to-noise ratio. Fiber optics are coming closer
and closer to the end user terminal, i.e. only the last few meters up to perhaps 1000 meters are still run in coaxial cable technology. Even in purely
coaxial cable networks, signal-to-noise ratios within a range of more than
30 dB are easily possible in most cases and modern broadband cable networks provide signal-to-noise ratios within a range of up to 40 dB. The
"old" DVB-C standard operates with a single-carrier modulation method
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_38, © Springer-Verlag Berlin Heidelberg 2010
732
38 DVB-C2 – the New DVB Broadband Cable Standard
and normally uses either 64QAM or 256QAM. However, modulation
schemes down to QPSK would also be possible but are not normally used.
In DVB-C, the "fall-off-the-cliff" occurs at pre-Reed Solomon bit error ratios of more than 2•10-4, corresponding to a signal-to-noise-ratio of about
•
•
26dB with 64QAM or
32dB with 256QAM.
Before considering a new standard which replaces another one, it is always of interest to know the limitations of the previous standard. Broadband cable networks operate within a frequency range of about 30 to 860
MHz. However, the range below 65 MHz is in most cases used for the return channel (Internet, telephony) today. The upper end of the broadbandchannel frequency band is followed closely by the GSM900 band. To protect other radio services, certain frequency bands in the cable must be kept
free due to possible leaks in the cable. There are currently attempts in the
terrestrial domain to utilize 9 previous TV channels for mobile radio applications (WiMAX, LTE) at the upper end of the TB band ("Digital Dividend"). This will correspond to a bandwidth of about 80 MHz. However,
investigations have shown that it may then no longer be possible to use this
frequency band in the cable since even fractions of the transmitting powers
provided for these mobile radio applications will have such severe effects,
especially on the receivers, thus making reception impossible. But there
have previously been frequency bands in the cable which could be disturbed by other services. The reason could be especially radiation-induced
interference on the final meters at the end user but the broadband cable
also produces its own interference. Multi-channel allocation and non-linear
amplifiers generate intermodulation products. And the end user's wiring
can also be quite eccentric at times. Wrong terminations and simple
T-junctions are no rarities. Amplifiers which are overdriven or set at the
wrong level can be frequently found at the end user's. The cable network
operator does not always have a compulsory influence on the correct state
of the network at level 4, i.e. the end user, and this especially is the main
cause of the limitations of DVB-C. The reason for this is the relatively
simple error protection (only Reed-Solomon block coding) and the singlecarrier modulation method used. Fitted with a channel equalizer, DVB-C
receivers can only compensate for frequency response errors to a limited
extent. It is easier to detect and compensate for frequency response errors
with the aid of a multi-carrier modulation method and the assistance of pilot carriers which is why COFDM is also the correct approach in cable applications. At the moment it appears in any case that the majority of modern transmission standards back COFDM and will continue to do so. This
38.2 Theoretical Maximum Channel Capacity
733
also applies more and more to the mobile radio field. And it is clear, therefore, that COFDM will also be used in DVB-C2 where a single-carrier
modulation method is not provided for. It can be said that the DVB-T2
standard was taken as the basis for DVB-C2, modes and features irrelevant
for DVB-C2 were initially deleted from it and then new applications were
implemented for broadband cable applications and the modulation
schemes were also adapted to the world of cable.
38.2 Theoretical Maximum Channel Capacity
Before considering the DVB-C2 standard in greater detail, the theoretical
limits of broadband cable will first be discussed. The Shannon Limit specifies that with signal-to-noise ratios from 10 dB, approximately the following formula for the theoretical maximum channel capacity applies:
C = 1/3 • B • SNR;
where
C = channel capacity (in bit/s);
B = bandwidth (in Hz);
SNR = signal/noise ratio (in dB);
A 8-MHz-wide broadband channel then provides the following theoretical maximum channel capacity:
Table 38.1. Shannon-Limits in a broadband cable
SNR[dB]
26
30
32
35
40
45
Theoretical
max. channel
capacity
[Mbit/s]
69.3
80
85.3
93.3
106.7
120
Remarks
Fall off the cliff with DVB-C at 64QAM
Poor coax cable system
Fall-off-the-cliff with DVB-C at 256QAM
Good HFC
In the headend
Assuming that a symbol rate of 6.9 MS/s is used, the data rates in an 8MHz channel in DVB-C were in most cases:
734
38 DVB-C2 – the New DVB Broadband Cable Standard
•
•
38.15 Mbit/s with 64QAM, and
50.87 Mbit/s with 256QAM.
In DVB-C2, the minimum aim was to achieve a 30% higher data rate.
Without having any knowledge of the DVB-C2 standard, it can be expected, therefore that, instead of the data rates shown above, either at least
about 50 Mbit/s should be achieved with signal/noise ratios of around 26
dB, or 66 Mbit/s with a signal/noise ratio of around 32 dB, in an 8-MHzwide channel. The aims were even exceeded. The reasons for the increase
in channel capacity in DVB-C2 are:
•
•
•
Better forward error correction,
higher-order types of modulation,
channel bundling and, as a result, no use of guard bands.
38.3 DVB-C2 – An Overview
In the following section, an overview of the most important DVB-C2 details will be given briefly.
•
•
•
•
•
•
•
•
•
•
•
•
Based on DVB-T2
COFDM (4K Mode, 2 short guard intervals)
FEC based on LDPC (… DVB-S2, DVB-T2, …)
multiple TS (transport streams) and GS (generic streams)
single and multiple PLPs (physical layer pipes)
data slices (fixed mapping of certain PLPs onto certain fixed carrier groups)
QPSK … 4096QAM (16K QAM under discussion)
Variable Coding and Modulation
channel-raster bandwidth of 6 or 8 MHz
channel bundling up to approx. 450 MHz bandwidth
supports notches (notching out disturbed or interfering frequency
bands)
reserved carriers for PAPR reduction (Peak to Average Power Ratio reduction)
38.6 COFDM Parameters
735
38.4 Baseband Interface
The DVB-C2 input interface allows use of a simple MPEG-2 transport
stream, to feed in several transport streams, a generic stream and several
generic streams. The input signal formats fully correspond to those of
DVB-T2 and have already been described in Chapter 36 and 37. Like
DVB-T2, DVB-C2 supports the Normal Mode and High Efficiency Mode
(HEM). The term Physical Layer Pipe (PLP) is also used in DVB-C2 and
corresponds to the physical transport of one of these input signals. There is
provision for common PLPs for a group of PLPs in order to save on signalling. A group of up to 255 PLPs can be combined to form data slices.
38.5 Forward Error Correction
The forward error correction in DVB-C2 corresponds to that of DVB-S2
and DVB-T2. It consists of a baseband scrambler, a BCH-block coder, an
LDPC block coder followed by a bit interleaver. The error protection in
DVB-C2 can be adjusted by selecting one of 5 code rates, listed in Table
38.2.
Table 38.2. Code rates in DVB-C2
Long frame (64800 bits)
Code rate
2/3
3/4
4/5
5/6
8/9
Short frame (16200 bits)
Code rate
2/3
3/4
4/5
5/6
8/10
38.6 COFDM Parameters
In DVB-C, only the 4k mode is possible, corresponding to 3408 carriers
used (Fig. 38.1.). However, the IFFT operates with carriers to a power of
two, i.e. 4096 carriers overall. The channel raster bandwidths are 6 or 8
MHz. The subcarrier spacings and the actual signal bandwidths are then:
736
38 DVB-C2 – the New DVB Broadband Cable Standard
Table 38.3. COFDM parameters in DVB-C2
Channel raster bandwidth [MHz]
6
8
Subcarrier spacing [kHz]
Signal bandwidth [MHz]
1.674
2.232
5.71
7.61
f
f
Channel raster
bandwidth
8 and 6 MHz
Fig. 38.1. COFDM parameters
Symbol n
Symbol n+1
Guard interval
Fig. 38.2. COFDM symbol and guard interval
Only very short guard intervals (Fig. 38.2.) will be used since only very
short reflections can be expected in a broadband cable. The selectable
38.7 Modulation Pattern
737
guard intervals are 1/64 or 1/128 of the symbol period. The guard interval
is a cyclic prefix, i.e. a prefixed copy of the symbol end.
Fig. 38.3. QAM orders QPSK, 16QAM, 64QAM and 256QAM in DVB-C2
Fig. 38.4. QAM order 1024 in DVB-C2 (simulated)
38.7 Modulation Pattern
DVB-C2 operates with coherent Gray-coded modulation. The following
QAM orders (Fig. 38.3., 38.4. and 38.5.) are supported:
•
•
QPSK
16QAM
738
38 DVB-C2 – the New DVB Broadband Cable Standard
•
•
•
•
64QAM
256QAM
1024QAM
4096QAM.
Fig. 38.5. 4096QAM (simulated)
Although 4096QAM (Fig. 38.5.) initially appears to be an interesting or
eccentric choice, it is quite practicable when considered more closely and
compared with the minimum signal/noise ratios necessary known from
DVB-T2. Although no tables about the minimum signal/noise ratios necessary with DVB-C2 have been publishes as yet, a minimum SNR of about
30 dB can be expected with 4096QAM on the basis of estimates from parameters of the DVB-T2 standard. Such QAM orders are made possible by
the signal/noise ratios of 30 dB up to almost 40 dB in cables which have
become quite good in the meantime. DVB-C2 mentions even modulation
methods of up to 16k QAM.
38.9 Interleavers
739
38.8 Definition of a Cell
Similar to DVB-T2, the term "cell" (Fig. 38.6.) was introduced in DVBC2. The reason is that bit groups are combined relatively early to form
later carriers which are mapped, i.e. modulated, but some interleaving
processes still have to be carried out. In DVB-T2, a cell is an IQ value
whereas in DVB-C2 a cell corresponds to a 2-, 4-, 8-, 10- or 12-bit-wide
bit group depending on the selected QAM order transported later on a carrier.
n=2, 4, 6, 8, 10, 12
n bits = cell
Mapping
I
Q
2n = N constellation points
Mapped cell
N=4, 16, 64, 256,
1024, 4096
Fig. 38.6. Definition of a cell in DVB-C2
38.9 Interleavers
There are altogether 3 interleavers (Fig. 38.7.) in DVB-C2, which are:
•
•
•
a bit interleaver which belongs directly to the FEC,
a time interleaver, and
a frequency interleaver.
There is no cell interleaver here as there is in DVB-T2.
The bit interleaver operates at FEC level and optimizes the characteristics of the error protection. Exactly as in DVB-S2 and DVB-T2, however,
it is a short interleaver. The time interleaver makes DVB-C2 more robust
against relatively long burst errors. The time interleaver is adjustable in its
interleaving depth deep time interleaving as in DVB-T2 is thus not necessary here. And the frequency interleaver distributes the data as randomly
as possible over many COFDM carriers. Time and frequency interleaving
740
38 DVB-C2 – the New DVB Broadband Cable Standard
takes place in DVB-C2 within a data slice to which a certain number of
PLPs are allocated.
......
PLP 0
Bit
interl.
Data
slice
builder
0
Time
interleaver
Frequ.
interleaver
......
PLP n
Bit
interl.
......
PLP 0
PLP n
L1
header
L1
data
Bit
interl.
Bit
interl.
Data
slice
builder
m
Time interl.
Frame
builder
Time
interleaver
L1
Block
builder
RF
OFDM
Frequ.
interleaver
Frequ.
interl.
Fig. 38.7. Interleavers in DVB-C2
38.10 Variable Coding and Modulation (VCM)
In DVB-C2, each input stream is transmitted in its own Physical Layer
Pipe (PLP). All interleaving-processes are restricted not to a PLP but to a
group of PLPs which are combined to form a data slice. Within each PLP,
however, various transmission parameters such as error protection and
modulation method can be selected. This is called Variable Coding and
Modulation (VCM), a term which is also known from DVB-T2. However,
due to the uniformity of the broadband cable, it must be assumed that
VCM, i.e. the selection of different transmission parameters in different
frequency bands or PLPs in the cable will not be used so intensively.
38.11 Frame Structure
Just like other standards, DVB-C2, too, has the concept of a frame. A
DVB-C2 frame begins with preamble symbols which are repeated every
7.61 MHz and have a width of 3408 carriers each. These are followed by
the data symbols, a total of 448 data symbols. The preamble symbols are
used both for time and frequency synchronization and for signalling of the
38.12 Channel Bundling and Slice Building
741
Layer-1 (L1) parameters. The preamble symbols are arranged with respect
to frequency in such a way that a receiver with a receiver bandwidth of
7.61 MHz will get all the data necessary for finding the Layer-1 parameters.
Preamble
7.61 MHz
L1 block
L1 block
…
L1 block
Approx. 200 ms
Frequency
…
Data slice n-1
Time
7.61 MHz
tuner window
7.61 MHz
Data slice n-2
Data slice 3
Data slice 2
Data slice 1
Data slice 0
448
data
symbols
7.61 MHz
Max.
7.61
MHz
Max. 450 MHz
Fig. 38.8. Framing and channel bundling
38.12 Channel Bundling and Slice Building
There are actually no longer any channels in DVB-C2 but only two channel rasters of either 6 or 8 MHz. Channels can be bundled together to form
a channel with a total width of approx. 450 MHz (Fig. 38.8.). There are
then no longer any gaps between the original channels. The lack of any
more gaps (guard bands) then enables the frequency spectrum to be used
more effectively and allows a higher data rate overall. However, there are
very frequently also disturbed frequency bands in the cable or frequency
bands which could interfere with other radio services (e.g. aircraft radio).
These can be notched out in DVB-C2 by simply switching off certain
OFDM carriers (Fig. 38.10.). This is called nothing which produces gaps
in the frequency spectrum. The channel bundling leads to increased demands on the modulator but not on the receiver. The receiver bandwidth is
limited to 7.6 MHz in DVB-C2, i.e. the receiver only needs to be capable
of demodulating channel rasters with a maximum width of 7.6 MHz. In
DVB-C2, frequency slices with a maximum width of up to 7.6 MHz are
742
38 DVB-C2 – the New DVB Broadband Cable Standard
formed for this purpose in which a whole number of Physical Layer Pipes
(PLPs) are mapped. During the demodulating, the receiver selects the frequency slice containing the data stream to be demodulated, i.e. the relevant
PLP. Every 7.6 MHz there is a Layer-1 signalling so that the receiver will
find the slices and know the transmission parameters in the slices. I.e.,
every 7.6 MHz, special signalling symbols are inserted at every beginning
of a DVB-C2 frame. Since the symbols are repeated every 7.6 MHz, the
receiver can arbitrarily locate itself over the bundled DVB-C2 channel
within the 7.6-MHz receiver-bandwidth.
The first L1 block begins mathematically at 0 MHz; and this L1 block is
repeated every 7.6 MHz at the beginning of a DVB-C2 frame. Regardless
of where the receiver, having a bandwidth of at least 7.6 MHz, places itself
it will capture the complete signalling of the Layer-1 parameters.
L1 block
7.61 MHz
Preamble
H
H
H
L1 part 2 data
Time
L1 header
Fig. 38.9. Preamble symbols
38.13 Preamble Symbols
The preamble symbols (Fig. 38.9.) are used for
•
•
•
time synchronization,
frequency synchronization and for
signalling the Layer-1 parameters (guard -interval, modulation, error protection etc.).
They are located at the beginning of a DVB-C2-frame in the L1 block
and also mark the beginning of the frame. The preamble symbols consist
of a header and an L1 time interleaving block. The header contains over 32
bits, consisting of 16 data bits and error protection, the basic signalling of
the most important basic L1 parameters.
38.13 Preamble Symbols
743
The preamble header is 32 OFDM cells wide. The following are transmitted in the header:
•
•
L1_INFO_SIZE (14 bits), and
L1_TI_MODE (2 bits).
Preamble
7.61 MHz
L1 block
L1 block
7.61 MHz
…
Approx. 200 ms
Data slice n-1
…
L1 block
Data slice n-2
Data slice 3
Data slice 2
Notch
Data slice 1
Data slice 0
448
data
symbols
7.61 MHz
Time
Max.
7.61
MHz
Frequency
Max. 450 MHz
Fig. 38.10. Forming of notches (carrier bands switched off) in DVB-C2
Altogether, the preamble header consists of 32 bits which are composed
of the 16 payload data bits and additional 32 FEC bits (Reed-Muellercoded). The 32 OFDM carriers of the preamble header are QPSKmodulated.
L1_INFO_SIZE signals in 14 bits half the width of L1-Part 2 (L1 time
interleaving block, consisting of data and stuffing bits).
L1_TI_MODE provides information about the Time Interleaving Mode
of L1 Part 2 (Layer 1 Time Interleaving Block) used. These are 2 bits
which convey the following:
•
•
•
•
00 = no time interleaving
01 = best fit
10 = 4 OFDM symbols
11 = 8 OFDM symbols.
744
38 DVB-C2 – the New DVB Broadband Cable Standard
The OFDM data carriers in the Layer-1 Part 2 block are 16QAMmodulated; the error protection consists of BCH and 16K-LDPC coding
with a code rate of ½.
In Part 2, the following L1 parameters are then signalled:
•
•
•
•
•
•
•
•
•
•
•
Network ID,
C2 system ID,
Start frequency,
Guard interval,
C2 frame length,
No. of bundled channels,
No. of data slices,
No. of notches,
Data slice parameters in the data slice loop,
PLP parameters in PLP parameter loops,
Notch parameters.
All data slices are described by the following parameters in the data
slice loop:
•
•
•
•
•
•
•
•
Data slice ID,
Data slice tune position,
Data slice offset left,
Data slice offset right,
Data slice time interleaver depth,
Data slice type,
No. of PLPs per data slice,
PLP loop with all PLP descriptions per data slice.
All PLPs of a data slice are described in PLP loops. These loops contain
the following parameters:
•
•
•
•
•
•
•
PLP ID,
PLP bundled,
PLP type,
PLP payload type,
PLP start,
PLP FEC type,
PLP modulation.
38.14 Pilots in DVB-C2
745
L1 signalling header and L1 time interleaving block can be repeated
several times within 7.6 MHz in the direction of frequency depending on
the length of the L1 time interleaving blocks.
38.14 Pilots in DVB-C2
Like DVB-T2, DVB-C2 has
•
•
•
edge pilots,
continual pilots and
scattered pilots.
Edge-pilots and continual pilots are used for frequency synchronization
(AFC). Additionally, denser pilot structures are formed by inserting jumping scattered pilots (Fig. 38.11.) for channel estimation and channel correction. In DVB-C2, the pilot structures do not need to be as dense as in
DVB-T2 due to the cable channel being physically simpler for the receiver. DVB-C2 supports a pilot structure for each of the two guard interval lengths. All DVB-C2 pilots are boosted by 7/3 compared with the data
carriers.
Frequency (carrier no.)
d2
0
i
Time
….
….
d3
d1
….
….
d1 = distance between scattered pilot carrier positions
d2 = distance between scattered pilots in one symbol
d3 = symbols forming one scattered pilot sequence
Fig. 38.11. Pilot structures in DVB-C2
746
38 DVB-C2 – the New DVB Broadband Cable Standard
Table 38.4. Scattered pilot structures in DVB-C2
Guard interval
1/64
1/128
Distance between carriers
carrying pilot carriers (and
distance between the pilot
carriers within a COFDM
symbol
12 (48)
24 (96)
Number of symbols
forming one pilot sequence
4
4
Table 38.5. Continual pilot positions in DVB-C2
Continual Pilot No.
96 216 306 390 450 486 780 804
924 1026 1224 1422 1554 1620 1680 1902
1956 2016 2142 2220 2310 2424 2466 2736
3048 3126 3156 3228 3294 3366
38.15 PAPR
Like DVB-T2, DVB-C2 also provides for the reservation of carriers in order to reduce the crest factor. If needed, these carriers can then be switched
on by the modulator and set in such a way that they reduce the current
crest-factor of the signal. This method corresponds to the tone reservation
in DVB-T2.
38.16 Block Diagram
The DVB-C2-block diagram is much more powerful than that of DVB-C
but still less complex than that in DVB-T2 although some parts correspond
to those of DVB-T2 (Fig. 38.12.).
38.17 Levels in Broadband Cables
The levels in DVB-C channels (Fig. 38.13.) are usually adjusted in such a
way that 64QAM-modulated channels are 12 dB below the analog TV
reference level (vision carrier - sync peak power) and 256QAMmodulated channels are 6 dB below that (RMS), in order to obtain
sufficient distance from interfering noise in the broadband cable. DVB-C2
channels with 1024QAM-modulation will probably be 4 ... 6 dB below the
38.17 Levels in Broadband Cables
747
Mapper
FEC
frame
header
insert.
FEC
Cell
builder
Mapper
FEC
frame
header
insert.
FEC
Cell
builder
Mapper
FEC
frame
header
insert.
Cell
builder
Mapper
FEC
frame
header
insert.
FEC
L1 config.
L1
sign.
FEC
Mapper
Time
interl.
Time & frequency
interleaver
block
PLP n
......
Mode/
stream
Slice m adaptation
......
PLP 0
Data slice
builder
....
PLP n
......
Mode/
stream
adaptation
block
......
Slice 0
L1
block
builder
Frequ.
interl.
4K OFDM, pilots, guard
interv. , PAPR red., DAC
Frame builder
Cell
builder
FEC
Time & frequency
interleaver
PLP 0
Data slice
builder
1024QAM-modulation will probably be 4 ... 6 dB below the ATV reference level and with 4096QAM, the level is then approx. 0 ... +2 dB above
the analog-TV-reference level.
RF
Level
Fig. 38.12. DVB-C2 block diagram
-12 dB
-6 dB
-4…-6 dB
0…+2 dB
ATV system level
DVB-C2
4096QAM
DVB-C
64QAM
DVB-C
256QAM
DVB-C2
1024QAM
ATV
Frequency
Fig. 38.13. Levels in broadband cables
748
38 DVB-C2 – the New DVB Broadband Cable Standard
38.18 Capacity
DVB-C2 can now handle up to more than 70 Mbit/s in an former 8-MHz
channel which could previously handle only 51 Mbit/s. The reason for this
is mainly the modern LDPC forward error correction which can now be
applied, in conjunction with channel bundling, i.e. the avoidance of gaps
between the channels. Using COFDM modulation, the broadband cable
can now be used even more effectively, since frequency-selective problems can be eliminated more easily. It is intended to bundle up to approx.
ca. 450-MHz-wide channels. This will result in about 380000 individually
usable carriers in 862 MHz bandwidth.
38.19 Outlook
With DVB-C2, as with DVB-T2 and DVB-S2, a very modern, highcapacity new DVB transmission standard has been created. Its applications
will lie in the field of HDTV and fast downstreams for broadband Internet
via broadband cable. It is not yet known which countries or cable network
operators will be the first to employ this standard. DVB-C is still fighting
for customer acceptance in many broadband cable networks. Analog television (ATV) by cable is still very comfortable since no special receivers
are needed. Every normal television set or flat panel screen still has a conventional analog TV tuner. On the other hand, these will not provide
HDTV and many customers now have a flat panel screen which, in any
case, reproduces ATV only with modest quality and displays artefacts even
with SDTV material.
Bibliography: [DVB_A138]
39 DVB-x2 Measuring Techniques
In the DVB-x2-Standards, some revolutionary new approaches were implemented. It was clear from the beginning that this will also require new
and enhanced DVB-x2 measuring techniques. The changes in the new
Standards, which will have to be considered as having an influence on the
new measuring techniques, are:
•
•
•
•
•
•
•
•
•
•
The high degree of complexity of DVB-x2,
a new optimized but computationally highly complex error protection,
multiple input streams,
generic streams,
higher-level modulation methods,
rotated, Q-delayed QAM,
variable coding and modulation,
MISO,
TFS (Time Frequency Slicing),
grouping of channels.
The points of high complexity and new error protection apply similarly to
all new DVB-x2 Standards. This requires simply more computing power in
the test instruments, i.e. hardware. of even higher power. But this can be
achieved simply by "work" and has been applied in practice in DVB-S2
for some time. A huge innovation compared with the old DVB Standards
is the provision for multiple input streams and also the departure from the
MPEG-2-transport stream format. The details will now be discussed in the
standard-specific chapters.
39.1 DVB-S2
DVB-S2 has been established firmly since 2004 and there are now also
numerous satellite transponders which radiate DVB-S2-modulated HDTV
programs. A chapter on the subject of DVB-S2 measuring techniques is al-
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_39, © Springer-Verlag Berlin Heidelberg 2010
750
39 DVB-x2 Measuring Techniques
ready contained in this book under the heading of DVB-S/S2 Measuring
Technology and is referred to here. Multiple input streams and data formats deviating from MPEG-2-TS are not being used up to the present day.
Measuring the bit error ratios differs from DVB-S and there are now
•
•
•
the pre-LDPC bit error ratio,
the pre-BCH bit error ratio
and the post-BCH bit error ratio or better the post-BCH packet error rate.
In comparison with DVB-S, the MER must now be measured on higherlevel modulation methods but it does not require any further rethinking.
There are test transmitters (e.g. Rohde&Schwarz SFU, SFE) and also instruments with integrated DVB-S2 test receivers (e.g. Rohde&Schwarz
DVM400).
Fig. 39.1. DVB-S2 Constellation Diagram [DVM] and DVB-S2 Measurement Parameters
39.2 DVB-T2
DVB-T2 represents a new and greater challenge for measuring techniques.
This is where almost all the new approaches of the Standard are transformed into reality. In the signal generation, the new input stream formats
39.2 DVB-T2
751
must now be supported. However, using only precompiled stream libraries
makes testing of the output units too inflexible, i.e. the requirement is for
tools by means of which meaningful scenarios for input stream combinations can be edited. In the case of multiple PLPs, the transmission parameters of the modulators are also controlled via the T2-MI modulator input
stream. I.e., the modulator can then no longer be controlled in such a simple way directly in its modulation parameters such as mode, error protection and modulation method as in DAB. If TFS (Time Frequency Slicing)
is used at some time, a test transmitter must also be capable of simulating
several channels. MISO, too, requires at least one second RF path with its
own fading simulator. And MISO will be used because, above all, distributed MISO costs nothing extra in the SFN but avoids the destructive fading notches or at least renders them minimisable in a large SFN. Incidentally, the planning of such MISO-SFNs also represents a new challenge for
the planning tools. The transmitter sites can then operate in two different
modes, i.e. these two types of coverage must be simulated. All new test receivers must now be capable of handling the new error protection which
requires very much additional computing power, and thus hardware.. The
BER definitions are the same as in DVB-S2. QAM orders of up to
256QAM are nothing special and have been known already since DVB-C
times at the latest, i.e. since 1995.
Fig. 39.2. Rotated DVB-T2 Constellation Diagram [ETL]
However, rotated and especially Q-delayed constellation diagrams will
change the way in which the MER is calculated. Since the I and Q value of
a cell are then no longer transported by the same cell but are transmitted on
752
39 DVB-x2 Measuring Techniques
quite different pairs of carriers by multiple interleaving, new calculation
methods, definitions and forms of representation must be considered, e.g.
MER of I and Q separately as a function of frequency or number of carriers. If TFS is used (which is currently improbable), two receiving sections
will be needed. In MISO operation, each signal path must be channelestimated, i.e. the signal path from MISO Mode 1 and MISO Mode 2 must
be evaluated. The results must also be presented correspondingly to the
test technician. This may also have an influence on the MER calculation
and its definition. All this will only become evident through practical experience. In the case of multiple PLPs, the test receiver is confronted by a
different modulation method and a different FEC code rate in every PLP.
And the transmission parameters can also change dynamically. There are
already test transmitters available which support especially the Single PLP
mode and will soon support other features (Rohde&Schwarz SFU).
39.3 DVB-C2
In DVB-C2, too, multiple PLPs will be used. The requirements for the test
transmitter and test receiver will here be comparable to DVB-T2. The special feature of DVB-C2 is the provision for channel grouping in a total
bandwidth of 450 MHz. However, the receiver bandwidth is limited to a
maximum of 8 MHz but the channel tuning is no longer necessarily fixed
on channels; there are narrow data slices and also notches which are
masked out. The extended requirements arise here on the modulator side or
on the test transmitter side, respectively. Both the modulator and a test
transmitter must be capable of generating not only one 8-MHz-wide channel but two and more continuous channels with a maximum width of 450
MHz. The new requirement on the receiver side is the new error protection
and especially the modulation method with up to 4096QAM. It is the bit
resolutions in the receiver and also in the test receiver which must be designed for this. Another challenge is increasingly the representation of
constellation diagrams of greater than 256QAM. Without having a zoomfunction, artifacts in the constellation diagram will no longer be recognizable at all. The BER definitions correspond to those of DVB-S2 and
DVB-T2; The MER definition will not need to be changed compared with
DVB-C but is only performed on higher-level modulation methods. At the
time when the third edition of this book goes to press, DVB-C2 is still too
"young" for measuring techniques to be available for this technology. It is
a safe bet, however, that here, too, test transmitters will be the first to provide signals.
39.4 Summary
753
39.4 Summary
DVB-S2 has been operating for years; the associated measuring techniques
are in place. The additional demands made on the measuring techniques
have previously been approximately comparable to those of DVB-S. DVBT2 and DVB-C2, on the other hand, necessitate increased demands also on
the measuring techniques. At the time of this book going to press, the
range of measuring techniques required is available only in parts or not at
all.
Bibliography: [SFU], [DVM]
40 CMMB – Chinese Multimedia Mobile
Broadcasting
CMMB – Chinese Multimedia Mobile Broadcasting is also a mobile TV
standard. CMMB is comparable to DVB-SH; it is a hybrid system and
supports the terrestrial service route and the service route via satellite. Just
as in DVB-SH, gap-fillers are provided which are fed via satellite. CMMB
has nothing to do with DTMB, and the baseband signal is not the MPEG-2
transport stream, either, in this case. The radiated data signal can consist
of up to 39 service channels and a logical control channel. These are
broadcast in up to 40 time slots. The modulation method used is OFDM,
the types of modulation are BPSK, QPSK and 16QAM. The error protection used is a combination of Reed-Solomon coding and LDPC coding and
bit interleaving. The Standard is published only in excerpts and only under
the seal of „Confidential“. The channel bandwidths supported are 8 MHz
and 2 MHz, the signal bandwidths are 7.512 MHz and 1.536 MHz. In the
8-MHz channel, the 4k mode is used with 3077 carriers actually being
used. The 2-MHz channel uses the 1k mode with 629 carriers actually being set. The subcarrier spacing is 2.441 kHz, similar to the other OFDMbased systems, there are Continual Pilots, Scattered Pilots and data carriers. The contents broadcast are certainly MPEG-4 AVC and AAC signals.
Unfortunately it is not possible to report further details at present.
Bibliography:
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_40, © Springer-Verlag Berlin Heidelberg 2010
41 Other Transmission Standards
Apart from the standards for Digital Video and Audio Broadcasting and
Mobile TV, there are by now other methods which are only touched upon
to some extent in this chapter but are partly also discussed in greater detail
as far as possible. The situation is frequently that it is impossible to discuss
details since they have not been published, or are only known by their
status. Naturally, it is not intended to violate any rights.
The standards discussed in this chapter are:
•
•
•
MediaFLO
IBOC/HD Radio
FMextra
41.1 MediaFLO
MediaFLO stands for Media Forward Link Only and is a proprietary US
standard for mobile TV developed by Qualcomm.
The technical parameters of MediaFLO are:
•
•
•
•
•
•
•
•
COFDM
4k mode, QPSK or 16QAM
Guard interval = 1/8
Channel bandwidth = 5, 6, 7 or 8 MHz
Concatenated Reed-Solomon(16, k = 12, 14 or 16) and turbo code
forward error correction (code rate = 1/3, 2/3 1/2)
Net data rate (6 MHz, CRInner=2/3, CROuter=12/16, 16QAM) = 8.4
Mbit/s
Data rates of up to 11.2 Mbit/s
Source encoding: MPEG-4 Part 10 AVC Viddeo, MPEG-4 AAC+
Audio or IP
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_41, © Springer-Verlag Berlin Heidelberg 2010
758
41 Other Transmission Standards
41.2 IBOC - HD Radio
IBOC (In Band on Channel) or HD Radio (Hybrid Radio) (iBiquity Digital
Coporation, US) are synonymous terms for digital radio in combination
with analog FM radio COFDM-modulated bands (Fig. 41.1 and 41.2.), in
which digital audio signals are transmitted, are here added above and below an FM carrier. IBOC has its origin in the US and is currently also being tested in some regions in Europe. It represents a current possibility of
operating FM and digital radio adjacently to one another.
Lower
COFDM
sideband
FM modulated
carrier
Upper
COFDM
sideband
f
Center frequency
Fig. 41.1. IBOC spectrum
In VHF sound broadcasting, the channel spacing is 300 kHz, or 200 kHz
(US), and the frequency deviation is typically 75 kHz. The VHF multiplex
signal consists of the L+R signal with a width of 15 kHz, the pilot with 19
kHz and the L-R component in two sidebands around a suppressed AM
carrier at 38 MHz, and a modulated auxiliary carrier at 57 kHz (RDS, previously ARI); the baseband bandwidth is thus 57 kHz. According to Carson's formula, the required RF bandwidth is a little over 250 kHz with a
deviation of 75 kHz; the gaps towards the adjacent channels can be filled
up with COFDM-modulated signals. It is also possible here to play with
the deviation of the FM carrier, i.e. it may be possibly reduced in order to
be able to widen the COFDM spectra. In this context, channel spacing
does not necessarily imply available RF bandwidth. The baseband band-
41.4 Effects of the Digital Dividend on Cable and Terrestrial TV Networks
759
width actually available in VHF sound broadcasting, too, is up to 100 kHz
and is being utilized in the FMextra standard described later.
Fig. 41.2. Real HD-Radio spectrum
41.3 FMextra
The baseband bandwidth of a VHF FM channel is 100 kHz, of which
about 60 kHz are currently being utilized. In the US, the range between 60
and 100 kHz is partly being used for additional information. VHF
transmitters have SCA inputs which allow for a spectrum occupancy
between 60 and 100 kHz. This is where FMextra comes into play and
enables this range to be used in the baseband. The channel spacing in the
RF band remains unchanged.
41.4 Effects of the Digital Dividend on Cable and
Terrestrial TV Networks
Due to the switchover from analog terrestrial television to digital terrestrial
television (DVB-T) in Europe, much fewer frequencies are needed for
propagating the same number of programs since DVB-T can accommodate
760
41 Other Transmission Standards
at least 4 SDTV programs per channel. At the last World Radiocommunication Conference (WRC07, Geneva, October/November 2007), the upper
TV channels 61 – 69 (790 – 862 MHz) were therefore reserved for mobile
radio applications (UMTS, LTE, WiMAX) under the heading of "Digital
Dividend“. Naturally, this has effects on applications operating in the same
frequency band like DVB-T-broadcasts in these channels, but also broadband cable networks using the full frequency band from 5 to 862 MHz.
The effects differ and have now also been verified in practice by trials. The
author was actively involved in some of the first investigations. In the following sections, the influences of the use by mobile radio of channels 61 69 on DVB-T networks and on broadband cable networks will be discussed.
41.4.1 Anatomy of the Mobile Radio Signals
One might think at first that the anatomy of the mobile radio signals, i.e.
the modulation methods used, which with the digital dividend are now creating interference signals for DVB-T and DVB-C networks, would play a
role, but this is not the case. All mobile radio standards use digital modulation methods, just like digital television. And these are not much different
from one another. Regardless of whether it is a single-carrier modulation
method, WCDMA (Wide-Band Code Division Multiple Access) or OFDM
which is used, the signals all look like band-limited noise signals with a
more or less large crest factor. GSM uses a single-carrier modulation
method, UMTS uses WCDMA, and LTE and WiMAX operate with
OFDM methods. And the general trend in all areas is in any case towards
multi-carrier methods. Anatomically, these signals can be distinguished by
the following parameters:
•
•
•
bandwidth,
envelope shape (rectangular, or roll-off or Gaussian filtering),
crest factor.
The main influence on applications in the same frequency band is produced by the bandwidth of these mobile radio signals. And only the uplink,
i.e. the return channel from the mobile radio to the base station, can have
an interfering effect. As far as the power budget is concerned, the
downlink is too weak to have any interfering effect at the location where
terrestrial and cable receivers are used. Neither does a dense cable network
exert any interfering effect on uplink or downlink. DVB-T transmitters and
mobile radio applications cannot coexist in the same channel; a DVB-T
41.4 Effects of the Digital Dividend on Cable and Terrestrial TV Networks
761
transmitter would displace a mobile radio downlink and a mobile radio uplink would radiate into a DVB-T antenna and make it impossible to receive
DVB-T channels in the same frequency band. There are, therefore, effects
which do not even need to be investigated since their interactions with applications in the same frequency band can be explained or estimated simply by physics, mathematics and experience.
The permissible transmitting power in the uplink of mobile radios is up
to 24 dBm (250 mW) in this frequency band. This needs to be taken into
consideration.
41.4.2 Terrestrial TV Networks and Mobile Radio
Co-channel operation of mobile radio and DVB-T is not possible. DVB-T
would interfere with the mobile radio downlink since the DVB-T field
strength is much higher than the field strength of the mobile radio channel
at the receiving location. This applies especially to DVB-T networks
which are designed for portable indoor reception. The mobile radio uplink
would interfere with the DVB-T reception since the uplink channel would
radiate either directly into the DVB-T receiving antenna; and the DVB-T
receivers are also certainly leaky and would admit stray radiation. Neither
is there any need to confirm this by investigations since the characteristics
of DVB-T receivers are well known. The extent to which a mobile radio
uplink is noticed in the adjacent channel greatly depends on the characteristics of the DVB-T receiver. Virtually all DVB-T receivers can handle adjacent channels which are up to 20 dB higher, but many can also handle
adjacent channels which are up to 40 dB and more higher. The greater the
separation from the frequency of an adjacent channel the easier it becomes
for the DVB-T receiver. An adjacent mobile radio channel is virtually like
an adjacent DVB-T channel for a DVB-T receiver. Analyses and investigations of the interference effects between mobile radio network and analog
terrestrial television network are no longer required since there will soon
be no more analog terrestrial television (ATV) networks in Europe.
41.4.3 Broadband Cable TV Networks and Mobile Radio
The situation is quite different in the case of broadband cable networks.
Broadband cable networks currently handle the following four applications:
•
analog television,
762
41 Other Transmission Standards
•
•
digital television (now DVB-C, later DVB-C2),
fast Internet access (Euro-DOCSIS with DVB-C signals in the
downlink), and telephony.
It is not clear for how long analog television will still be supplied. Many
broadband customers love especially this simple type of reception without
additional equipment apart from the actual screen with integrated analog
TV tuner. A broadband provider will have to think long and hard before he
switches off the last analog TV channels even if it would be more economical for him to do so.
It was found in trials that up to network level 4 (= subscriber network
level), a broadband cable network using modern multi-shielded cables,
modern amplifiers and termination boxes is leakproof to such an extent
that only slight or no influences were detectable with an external irradiation of up to approx. 24 dBm (250 mW) power. It was also found that most
of the terminals were not leakproof enough with co-channel irradiation.
And this applies to all types of broadband reception. In the adjacent channel, up to 20 dB more interfering radiation can be accepted in some cases
which was shown in tests of modern DVB-C receivers. I.e., the problem is
mainly the terminals themselves and naturally also the cables from the
socket to the terminal.
41.4.3.1 Influence of Co-channel Mobile Radio on DVB-C
Reception
Tests have shown that with a modern network level 4, the terminals are the
only problem. Virtually all the DVB-C receivers showed symptoms of interference from about 8 to 15 dBm irradiated power from 1 to 2 meters distance and displayed slicing, freezing and picture loss. The plane of polarization (horizontal or vertical) played a great part in this and amounts to
between 3 and 10 dB. Naturally, the quality of the transmitting antenna in
the mobile unit also plays a great role and easily accounts for another 5 dB
difference. From about 5 to 10 m distance and with properly attenuating
walls, no further interference effects can be expected for the co-channel
mobile radio downlink. However, the variance of immunity from interference of the terminals is very great and lies within a range of around 10 dB.
The quality of the connecting cables used is always decisive and does not
need to be investigated. The immunity of the plugs is also a problem. The
starting point of the disturbances depends on the type of QAM (64QAM or
256QAM) and on the RF level present. The worst-case constellation is
256QAM with a level of 54 dB V (minimum level) and the best is
41.4 Effects of the Digital Dividend on Cable and Terrestrial TV Networks
763
64QAM and a level of 74 dB V (maximum level). The standard should be
assumed to be 256QAM with a level of 64 dB V (usual average level).
41.4.3.2 Influence of Co-channel Mobile Radio on Analog TV
Reception
Analog TV reception in the broadband cable network also showed that the
only leakage is to be expected at the terminal or in the terminal wiring.
Terminals show symptoms of interference more or less from about 1 to 2
m distance and depending on polarization The interference is visible as
noise or moire in the picture. At more than 5 m distance or with sufficiently leakproof walls, there will be no further detectable interference but
this, too, depends on the terminal.
41.4.3.3 Influence of Co-channel Mobile Radio on Other Digital
Broadband Services
The influence of co-channel interferences of mobile radio leakages in the
broadband cable network on the other digital broadband cable services can
be seen as being particularly dramatic. Depending on polarization, transmitting powers of approx. 10 to 15 dBm here lead to shorter dropouts or
lower data rates, respectively, and, in addition, to resynchronization pauses
of the cable modem lasting minutes. The end user will not especially enjoy
the resynchronization pauses, in particular. The major leakage in cochannel reception corresponds to that experienced in pure DVB-C reception. It is the downlink which is affected and this is also pure DVB-C with
Euro-DOCSIS. A similar effect will be experienced in telephony via
broadband cable.
41.4.4 Electromagnetic Field Immunity Standard for Sound and
Television Broadcast Receivers
The electromagnetic field immunity of sound and television broadcast receivers and related terminals is regulated in standards EN 55020 and
CISPR 20, resp. [CISPR20]. Such receiving units must not display any
symptoms of interference at noise field intensities of up to 3 V/m, apart
from the operating frequency itself ("tuned frequency excluded“). Naturally, this is appropriate for terrestrial receivers but this standard is also
applied to broadband cable receivers.
764
41 Other Transmission Standards
41.4.5 Summary
The broadband cable networks are leakproof – both with respect to the
power radiated and to the incident power - as long as they are of the correct and modern design. Naturally, this will not apply to all networks, especially the older ones of network level 4. Only new networks have been
tested. New DVB-C receivers and cable modems have also been identified
as particular leakage points. And this also applies to analog TV receivers.
DVB terminals already show symptoms of interference at a radiated interference of 10 dB less than what a mobile receiver is allowed to radiate in
the uplink. This should not be understood to be a wholesale opposition
against the "Digital Dividend“ but simply represents the current problems
and facts. The electromagnetic field immunity standard for broadcast receivers explicitly allows susceptibility to interference in the operating
channel set. The co-existence of broadband cable networks or DVB-T
networks with mobile radio networks in the same frequency band was
never intended or planned. If the broadband cable terminals are made leakproof, this will not be a problem for broadband cable networks of the more
modern type. To achieve this, however, a corresponding electromagnetic
field immunity in the operating channel should be backed up with a corresponding electromagnetic field immunity standard. If only adjacent channels are occupied in DVB-T and mobile radio networks, this will not be a
problem, either.
Bibliography: [SFU], [LESNIK], [CISPR20]
42 Digital Television throughout the World - an
Overview
The numerous technical details of the various digital television standards
have now been discussed. The only thing that is still missing is a report
about the current development and spread of these technologies, and a look
at the future. Digital satellite television (DVB-S) is available in Europe
over numerous transponders of the ASTRA and Eutelsat satellites. Many
streams can be received unencrypted. Complete receiving systems for
DVB-S are available at low cost in many department stores. DVB-C, too,
has become well established in the meantime. Digital terrestrial television
has also become widely used in numerous countries and above all in Great
Britain, where DVB-T started in 1998. DVB-T first spread in Scandinavia
where Sweden is covered completely by DVB-T. Australia, too, was one
of the first countries to have introduced DVB-T. In Australia, DVB-T is
available mainly in the population centers along the Eastern and Southern
coast. DVB-T is also being built up in South Africa and in India. In
Europe, the current status is as follows: Autumn 2002 saw the start of
DVB-T in Berlin and in August 2003, 7 data streams with more than 20
programs were on the air and analog television was being operated in parallel for only a brief period in simulcast mode and then switched off completely in August 2003, which certainly represented a minor revolution!
DVB-T was designed to implement portable indoor reception. Reception is
possible using simple indoor antennas from the heart of Berlin out to the
outer suburbs in some cases. Naturally, there are restrictions in indoor reception due to the attenuation of buildings and other shadowing. In the
years of 2003, 2004 and 2005, this type of reception known as "Anywhere
Television" then also spread to the North-Rhine-Westphalia, Hamburg,
Bremen, Hanover and Frankfurt regions and since May 30th 2005 also to
the Munich and Nuremberg conurbation areas in Germany. The data rates
per DVB-T channel are 13.27 Mbit/s, providing space for about 4 programs per channel. In most cases there are about 4 - 6 frequencies in the
air at any time. Mecklenburg followed in autumn 2005, and Stuttgart in
2006. The networks implemented in Germany are all designed as small
isolated SFN regions with few transmitters. In the meantime, the upgrad-
W. Fischer, Digital Video and Audio Broadcasting Technology, Signals and Communication
Technology, 3rd ed., DOI 10.1007/978-3-642-11612-4_42, © Springer-Verlag Berlin Heidelberg 2010
766
42 Digital Television throughout the World - an Overview
ing to DVB-T has been completed in Germany since November 2008,
apart from a few additional low-power transmitters. Analog terrestrial
television no longer exists in Germany.
In Italy, DVB-T was expanded greatly in 2004. MHP is not only "in the
air" but is also accepted there. Switzerland followed in 2004/2005 and in
Austria, DVB-T started in 2006. In Greenland, for example, DVB-T is a
very inexpensive alternative for offering TV to the population in the small
towns, each of which is an isolated self-contained community. Satellite reception is very difficult in Greenland because of its location and requires
very large antennas, making it a logical choice to rebroadcast the received
channels inexpensively terrestrially by means of DVB-T, using transmitting powers of around 100 W. DVB-T is also transmitted in Belgium and
Holland and is being expanded greatly, especially in Holland.
In the US, Canada, Mexico and South Korea, ATSC is used and analog
terrestrial television was switched off in June 2009 in the US. ATSC will
probably remain restricted to these countries.
Japan has its own ISDB-T standard which has also been adopted by
Brazil.
On the other hand, it appears that the initial attempts to introduce mobile
TV by means of DVB-H and T-DMB in Europe seem to have failed in
most countries.
Europe is currently also in the initial stages of introducing HDTV (High
Definition Television). This is based on new technologies such as MPEG4 AVC / H.264, and the new satellite standard DVB-S2. The German public-law broadcasters will commence regular DVB operations in February
2010.
It is only digital audio broadcasting which is still having problems, the
main reason being the very good quality and acceptance of FM audio
broadcasting. Australia is currently engaged in starting up DAB+ and some
transmitter sites have already been set up. In the UK, DAB has been operating successfully for some years.
There are other TV or mobile TV standards and extensions to standards
which were or are being developed since the first (English and German)
editions and information about these has been included in this book as far
as possible. But somewhere there must be a cut-off point. It already includes reports about the new terrestrial digital DVB-T2 TV standard and
about the new DVB-C2 cable TV standard. At some stage, DVB-T2 and
DVB-C2 will replace the DVB-T and DVB-C standards developed in the
mid-nineties. The same applies to DVB-S2. The new DVB-x2 standards
provide a distinctly higher net data ate.
Following suggestions by many participants of seminars on digital television all over the world, this book was created to provide the man in the
42 Digital Television throughout the World – an Overview
767
field, be he a transmitter network planner, a service technician at a transmitter site, a technician responsible for MPEG-2 encoders and multiplexers in a studio or in a playout center, an engineer working in a development laboratory or even a student, with an insight into the technology and
measuring techniques of digital television. It concentrates deliberately on
the practical things of importance and attempts to include as little mathematical “ballast” as possible.
The author of this book was able to participate personally in the introduction of digital television both in his work in the TV test instrumentation
development department of Rohde & Schwarz and especially also later in
his many seminar trips including eleven journeys to Australia alone. Direct
participation in the installation of the DVB-T network of Southern Bavaria
with the Olympic Tower transmitter in Munich and the Mt. Wendelstein
transmitter right from the start up to switch-on at 1.00 am on May 30th,
2005 at the Mt. Wendelstein transmitter also left a deep impression. Many
subsequent field measurements and experiences from trips ranging from
Australia to Greenland have thus found their way into this book.
The author's assistance with smaller or greater technical problems or
problems involving test technology has always been readily welcomed and
the resultant insights can also be found in this book. The continuing very
close contact with the development departments of TV test instrumentation
and transmitter technology of Rohde & Schwarz, is still of especially great
value to me. Just copying a TV standard would never result in a useful
book.
Greetings and many heartfelt thanks to the many participants in these
courses throughout the world, for the discussions and suggestions in the
seminars and for the lively interest shown in this work. Lots of feedback
has shown that this book is really being used in practice. Is is now considered to be one of the standard textbooks for television and radio engineering which is by far the most rewarding compensation for the many hours
of work spent on producing it.
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Definition of Terms
AAL0
AAL1
AAL5
ASI
ATM
ATSC
BCH
BAT
CA
CAT
CI
COFDM
CRC
CVCT
DAB
DDB
DII
DMB-T
DRM
DSI
DSM-CC
DTS
DVB
ECM
EIT
EMM
ES
ETT
FEC
IRD
ISDB-T
J83
LDPC
LVDS
ATM Adaptation Layer 0
ATM Adaptation Layer 1
ATM Adaptation Layer 5
Asynchronous Serial Interface
Asynchronous Transfer Mode
Advanced Television Systems Committee
Bose-Chaudhuri-Hocquenghem Code
Bouquet Association Table
Conditional Access
Conditional Access Table
Common Interface
Coded Orthogonal Frequency Division Multiplex
Cyclic Redundancy Check
Cable Virtual Channel Table
Digital Audio Broadcasting
Data Download Block
Download Information Identification
Digital Multimedia Broadcasting Terrestrial
Digital Radio Mondiale
Download Server Initializing
Digital Storage Media Command and Control
Decoding Time Stamp
Digital Video Broadcasting
Entitlement Control Messages
Event Information Table
Entitlement Management Messages
Elementary Stream
Extended Text Table
Forward Error Correction
Integrated Receiver Decoder
Integrated Services Digital Broadcasting Terrestrial
ITU-T J83
Low Density Parity Check Code
Low Voltage Differential Signalling
778
Definition of Terms
MGT
MHEG
MHP
MIP
MP@ML
MPEG
NIT
OFDM
PAT
PCR
PCMCIA
PDH
PES
PID
PMT
Profile
PS
PSI
PSIP
PTS
QAM
QPSK
RRT
SDH
SDT
SI
SONET
SSU
ST
STD
STT
T-DMB
TDT
TOT
TS
TVCT
VSB
Master Guide Table
Multimedia and Hypermedia Information Coding Experts
Group
Multimedia Home Platform
Megaframe Initialization Packet
Main Profile at Main Level
Moving Picture Experts Group
Network Information Table
Orthogonal Frequency Division Multiplex
Program Association Table
Program Clock Reference
PCMCIA
Plesiochronous Digital Hierarchy
Packetized Elementary Stream
Packet Identity
Program Map Table
MP@ML
Program Stream
Program-Specific Information
Program and System Information Protocol
Presentation Time Stamp
Quadrature Amplitude Modulation
Quadrature Phase Shift Keying
Rating Region Table
Synchronous Digital Hierarchy
Service Description Table
Service Information
Synchronous Optical Network
System Software Update
Stuffing Table
System Target Decoder
System Time Table
Terrestrial Digital Multimedia Broadcasting
Time and Date Table
Time Offset Table
Transport Stream
Terrestrial Virtual Channel Table
Vestigial Sideband Modulation
Definition of Terms
779
Adaptation Field
The adaptation field is an extension of the TS header and contains ancillary data for a program. The program clock reference (PCR) is of special
importance. The adaptation field must never be scrambled when it is to be
transmitted (see Conditional Access).
Advanced Television Systems Committee (ATSC)
The North American Standards Committee which determined the standard of the same name for the digital transmission of TV signals. Like
DVB, ATSC is also based on MPEG-2 systems as far as transport stream
multiplexing is concerned and on MPEG-2 video for video compression.
However, instead of MPEG-2, standard AC-3 is used for audio compression. ATSC specifies terrestrial transmission and transmission via cable
while transmission via satellite is not taken into account.
Asynchronous Serial Interface (ASI)
The ASI is an interface for the transport stream. Each byte of the transport stream is expanded to 10 bits (energy dispersal) and is transmitted
with a fixed bit clock of 270 MHz (asynchronous) irrespective of the data
rate of the transport stream. The fixed data rate is obtained by adding
dummy data without information content. Useful data is integrated into the
serial data stream either as individual bytes or as whole TS packets. This is
necessary to avoid PCR jitter. A variable buffer memory at the transmitter
end is therefore not permissible.
Asynchronous Transfer Mode (ATM)
Connection-oriented wideband transmission method with fixed-length
53-byte cells. Both payload and signalling information is transmitted.
ATM Adaptation Layer 0 (AAL0)
The ATM AAL0 Layer is a transparent ATM interface. The ATM cells
are forwarded here directly without having been processed by the ATM
Adaptation Layer.
ATM Adaptation Layer 1 (AAL1)
The ATM Adaption Layer AAL1 is used for MPEG-2 with and without
FEC. The payload is 47 bytes, the remaining 8 bytes are used for the
header with the forward error correction and the sequence number. This
makes it possible to check the order of incoming data units and the transmission. The FEC allows transmission errors to be corrected.
ATM Adaptation Layer 5 (AAL5)
The ATM Adaption Layer AAL5 is basically used for MPEG-2 without
FEC. The payload is 48 bytes, the remaining 7 bytes are used for the
header. Data with transmission errors cannot be corrected on reception.
Bose-Chaudhuri-Hocquenghem Code (BCH)
Cyclic block code used in the FEC of the DVB-S2 satellite transmission
standard.
780
Definition of Terms
Bouquet Association Table (BAT)
The BAT is an SI table (DVB). It contains information about the different programs (bouquet) of a broadcaster. It is transmitted in TS packets
with PID 0x11 and indicated by table_ID 0x4A.
Cable Virtual Channel Table (CVCT)
CVCT is a PSIP table (ATSC) which comprises the characteristic data
(eg channel number, frequency, modulation type) of a program (= virtual
channel) in the cable (terrestrial transmission → TVCT). TVCT is transmitted with the PID 0x1FFB in TS packets and indicated by the table_id
0xC9.
Channel Coding
The channel coding is performed prior to the modulation and transmission of a transport stream. The channel coding is mainly used for forward
error correction (FEC), allowing bit errors occurring during transmission
to be corrected in the receiver.
Coded Orthogonal Frequency Division Multiplex (CODFM)
COFDM is basically OFDM with error protection (coding - C), which
always precedes OFDM.
Common Interface (CI)
The CI is an interface at the receiver end for a broadcaster-specific, exchangeable CA plug-in card. This interface allows scrambled programs
from different broadcasters to be de-scrambled with the same hardware despite differences in CA systems.
Conditional Access (CA)
The CA is a system allowing programs to be scrambled and for providing access to these programs at the receiver end only to authorized users.
Broadcasters can thus charge fees for programs or individual broadcasts.
Scrambling can be performed at one of the two levels provided by an
MPEG-2 multiplex stream, e.g. the transport stream or the packetized elementary stream level. The relevant headers remain unscrambled. The PSI
and SI tables also remain unscrambled except for the EIT.
Conditional Access Table (CAT)
The CAT is a PSI table (MPEG-2) and comprises information required
for descrambling programs. It is transmitted in TS packets with PID
0x0002 and indicated by table_ID 0x01.
Continuity Counter
A continuity counter for each elementary stream (ES) is provided as a
four-bit counter in the fourth and last byte of each TS header. It counts the
TS packets of a PES, determines the correct order and checks whether the
packets of a PES are complete. The counter (fifteen is followed by zero) is
incremented with each new packet of the PES. Exceptions are permissible
under certain circumstances.
Definition of Terms
781
Cyclic Redundancy Check (CRC)
The CRC serves to verify whether data transmission was error-free. To
this effect, a bit pattern is calculated in the transmitter based on the data to
be monitored. This bit pattern is added to the data in such a way that an
equivalent computation in the receiver yields a fixed bit pattern in case of
error-free transmission after processing of the data. Every transport stream
contains a CRC for the PSI tables (PAT, PMT, CAT, NIT) as well as for
some SI tables (EIT, BAT, SDT, TOT).
Decoding Time Stamp (DTS)
The DTS is a 33-bit value in the PES header and represents the decoding time of the associated PES packet. The value refers to the 33 most significant bits of the associated program clock reference. A DTS is only
available if it differs from the presentation time stamp (PTS). For video
streams this is the case if delta frames are transmitted and if the order of
decoding does not correspond to that of output.
Digital Audio Broadcasting
A standard for digital radio in VHF band III, and the L band, defined as
part of the EUREKA Project 147. The audio is coded to MPEG-1 or
MPEG-2 Layer II. The modulation method used is COFDM with DSPSK
modulation.
Digital Multimedia Broadcasting Terrestrial (DMB-T)
Chinese standard for digital terrestrial television.
Digital Radio Mondiale
Digital standard for audio broadcasting in the medium- and short-wave
bands. The audio signals are MPEG-4 AAC coded. The modulation
method used is COFDM.
Digital Storage Media Command and Control (DSM-CC)
Private sections according to MPEG-2 which are used for the transmission of data services in object carousels or for datagrams such as IP packets in the MPEG-2 transport stream.
Digital Video Broadcasting (DVB)
The European DVB project stipulates methods and regulations for the
digital transmission of TV signals. Abbreviations such as DVB-C (for
transmission via cable), DVB-S (for transmission via satellite) and DVB-T
(for terrestrial transmission) are frequently used as well.
Data Download Block (DDB)
Data transmission blocks of an object carousel, logically organized into
modules.
Download Information Identification (DII)
Logical entry point into modules of an object carousel.
Download Server Initializing (DSI)
Logical entry point into an object carousel.
782
Definition of Terms
Elementary Stream (ES)
The elementary stream is a ‘continuous’ data stream for video, audio or
user-specific data. The data originating from video and audio digitization
are compressed by means of methods defined in MPEG-2 Video and
MPEG-2 Audio.
Entitlement Control Messages (ECM)
ECM comprise information for the descrambler in the receiver of a CA
system providing further details about the descrambling method.
Entitlement Management Messages (EMM)
EMM comprise information for the descrambler in the receiver of a CA
system providing further details about the access rights of the customer to
specific scrambled programs or broadcasts.
Event Information Table (EIT)
EIT is defined both as an SI table (DVB) and a PSIP table (ATSC). It
provides information about program contents like a TV guide.
In DVB the EIT is transmitted in TS packets with PID 0x0012 and indicated by a table_ID from 0x4E to 0x6F. Depending on the table_ID, it contains different information:
Table_ID 0x4E
actual TS / present+following
Table_ID 0x4E
actual TS / present+following
Table_ID 0x4F
other TS / present+following
Table_ID 0x50...0x5F
actual TS / schedule
Table_ID 0x60...0x6F
other TS / schedule
EIT-0 to EIT-127 are defined in ATSC. Each of the EIT-k comprises information on program contents of a three-hour section where EIT-0 is the
current time window. EIT-4 to EIT-127 are optional. Each EIT can be
transmitted in a PID defined by the MGT with table_id 0xCB.
Extended Text Table (ETT)
ETT is a PSIP table (ATSC) and comprises information on a program
(channel ETT) or on individual transmissions (ETT-0 to ETT-127) in the
form of text. ETT-0 to ETT-127 are assigned to ATSC tables EIT-0 to
EIT-127 and provide information on the program contents of a three-hour
section. ETT-0 is with reference to the current time window, the other
ETTs refer to later time sections. All ETTs are optional. Each ETT can be
transmitted in a PID defined by MGT with table_id 0xCC.
Forward Error Correction (FEC)
Error protection in data transmission, channel coding.
Integrated Receiver Decoder (IRD)
The IRD is a receiver with integrated MPEG-2 decoder. A more colloquial expression would be set-top box.
Definition of Terms
783
Integrated Services Digital Broadcasting Terrestrial (ISDB-T)
Japanese standard for digital terrestrial television. The modulation
method used is OFDM. The baseband signal is an MPEG-2 transport
stream.
ITU-T J83
Collection of various standards for digital television over broadband cable.
J83A = DVB-C
J83B = North American standard for digital television over broadband
cable (64QAM, 256QAM).
J83C = Japanese standard for digital television over broadband cable (6MHz variant of DVB-C)
J83D = ATSC variant for digital television over broadband cable
(16VSB); not used.
Low Density Parity Check Code
Block code used in the FEC of the DVB-S2 satellite transmission standard.
Low Voltage Differential Signalling (LVDS)
LVDS is used for the parallel interface of the transport stream. It is a
positive differential logic. The difference voltage is 330 mV into 100 Ω.
Master Guide Table (MGT)
MGT is a reference table for all other PSIP tables (ATSC). It lists the
version number, the table length and the PID for each PSIP table with the
exception of the STT. MGT is always transmitted with a
Section in the PID 0x1FFB and indicated by the table_ID=0xC7.
Main Profile at Main Level (MP@ML)
MP@ML is a type of source coding for video signals. The profile determines the source coding methods that may be used while the level defines the picture resolution.
Megaframe Initialization Packet (MIP)
The MIP is transmitted with the PID of 0x15 in transport streams of terrestrial single frequency networks (SFNs) and is defined by DVB. The
MIP contains timing information for GPS (Global Positioning System) and
modulation parameters. Each megaframe contains exactly one MIP. A
megaframe consists of n TS packets, n being dependent on the modulation
parameters. The transmission period of a megaframe is about 0.5 seconds.
Moving Picture Experts Group (MPEG)
MPEG is an international standardization committee working on the
coding, transmission and recording of (moving) pictures and sound.
MPEG-2
MPEG-2 is a standard consisting of three main parts and written by the
Moving Picture Experts Group (ISO/IEC 13818). It describes the coding
784
Definition of Terms
and compression of video (Part 2) and audio (Part 3) to obtain an elementary stream, as well as the multiplexing of elementary streams to form a
transport stream (Part 1).
Multimedia Home Platform (MHP)
Program-associated DVB data service. HTML files and JAVA applications are broadcast via object carousels for MHP-enabled receivers and can
then be started in the receiver.
Multimedia and Hypermedia Information Group (MHEG)
Program-associated data service in MPEG-2 transport streams, based on
object carousels and HTML applications Broadcast in the UK as part of
DVB-T.
Network Information Table (NIT)
The NIT is a PSI table (MPEG-2/DVB). It comprises technical data
about the transmission network (eg orbit positions of satellites and transponder numbers). The NIT is transmitted in TS packets with PID 0x0010
and indicated by table_ID 0x40 or 0x41.
Null Packet
Null packets are TS packets with which the transport stream is filled to
obtain a specific data rate. Null packets do not contain any payload and
have the packet identity 0x1FFF. The continuity counter is undefined.
Orthogonal Frequency Division Multiplex (OFDM)
The modulation method is used in DVB systems for broadcasting transport streams with terrestrial transmitters. It is a multicarrier method and is
suitable for the operation of single-frequency networks.
Packet Identity (PID)
The PID is a 13 bit value in the TS header. It shows that a TS packet belongs to a substream of the transport stream. A substream may contain a
packetized elementary stream (PES), user-specific data, program specific
information (PSI) or service information (SI). For some PSI and SI tables
the associated PID values are predefined (see 1.3.6). All other PID values
are defined in the PSI tables of the transport stream.
Packetized Elementary Stream (PES)
For transmission, the "continuous" elementary stream is subdivided into
packets. In the case of video streams one frame constitutes the PES,
whereas with audio streams, the PES is an audio frame which may represent an audio signal between 16 ms and 72 ms. Each PES packet is preceded by a PES header.
Payload
Payload signifies useful data in general. With reference to the transport
stream all data except for the TS header and the adaptation field are payload. With reference to an elementary stream (ES) only the useful data of
the ES without the PES header are payload.
Definition of Terms
785
Payload Unit Start Indicator
The payload unit start indicator is a 1 bit flag in the second byte of a TS
header. It indicates the beginning of a PES packet or of a section of PSI or
SI tables in the corresponding TS packet.
PCMCIA (PC Card)
PCMCIA is a physical interface standardized by the Personal Computer
Memory Card International Association for the data exchange between
computers and peripherals. A model of this interface is used for the common interface.
PCR Jitter
The value of a PCR refers to the exact beginning of a TS packet in
which it is located. The reference to the 27 MHz system clock yields an
accuracy of approx. ±20 ns. If the difference of the transferred values deviates from the actual difference of the beginning of the packets concerned,
this is called PCR jitter. It can be caused, for example, by an inaccurate
PCR calculation during transport stream multiplexing or by the subsequent
integration of null packets on the transmission path without PCR correction.
PES Header
Each PES packet in the transport stream starts with a PES header. The
PES header contains information for decoding the elementary stream. The
presentation time stamp (PTS) and decoding time stamp (DTS) are of vital
importance. The beginning of a PES header and thus also the beginning of
a PES packet is indicated in the associated TS packet by means of the set
payload unit start indicator. If the PES header is to be scrambled, it is
scrambled at the transport stream level. It is not affected by scrambling at
the elementary stream level.
PES Packet
The PES packet (not to be confused with TS packet) contains a transmission unit of a packetized elementary stream (PES). In a video stream,
for example, this is a source-coded image. The length of a PES packet is
normally limited to 64 kbytes. It may exceed this length only if a video
image requires more capacity. Each PES packet is preceded by a PES
header.
Plesiochronous Digital Hierarchy (PDH)
The Plesiochronous Digital Hierarchy was originally developed for the
transmission of digitized voice calls. In this method, high-bit-rate transmission systems are generated by time-interleaving the digital signals of
low-bit-rate subsystems. In PDH, the clock rates of the individual subsystems are allowed to fluctuate and these fluctuations are compensated for by
appropriate stuffing methods. The PDH includes E3 and DS3, among others.
786
Definition of Terms
Presentation Time Stamp (PTS)
The PTS is a 33 bit value in the PES header and represents the output
time of the contents of a PES packet. The value refers to the 33 most significant bits of the associated program clock reference. If the order of output does not correspond to the order of decoding, a decoding time stamp
(DTS) is additionally transmitted. This is the case for video streams containing delta frames.
Program and System Information Protocol (PSIP)
PSIP is the summary of tables defined by ATSC for sending transmission parameters, program descriptions etc. They contain the structure defined by MPEG-2 systems for 'private' sections. The following tables exist:
Master Guide Table (MGT),
Terrestrial Virtual Channel Table (TVCT),
Cable Virtual Channel Table (CVCT),
Rating Region Table (RRT),
Event Information Table (EIT),
Extended Text Table (ETT),
System Time Table (STT).
Program Association Table (PAT)
The PAT is a PSI Table (MPEG-2). It lists all the programs contained in
a transport stream and refers to the associated PMTs containing further information about the programs. The PAT is transmitted in TS packets with
PID 0x0000 and indicated by table_ID 0x00.
Program Clock Reference (PCR)
The PCR is a 42-bit value contained in an adaptation field and helps the
decoder to synchronize its system clock (27 MHz) to the clock of the encoder or TS multiplexer by means of PLL. In this case, the 33 most significant bits refer to a 90 kHz clock while the 9 least significant bits count
from 0 to 299 and thus represent a clock of 300 x 90 kHz (= 27 MHz).
Each program of a transport stream relates to a PCR which is transmitted
in the adaptation field by TS packets with a specific PID. The presentation
time stamps (PTS) and decoding time stamps (DTS) of all the elementary
streams of a program refer to the 33 most significant bits of the PCR.
PCRs have to be transmitted at intervals of max. 100 ms according to
MPEG-2 and at intervals of max. 40 ms according to the DVB regulations.
Program Map Table (PMT)
The PMT is a PSI table (MPEG-2). The elementary streams (video, audio, data) belonging to the individual programs are described in a PMT. A
PMT consists of one or several sections each containing information about
a program. The PMT is transmitted in TS packets with a PID from 0x0020
to 0x1FFE (referenced in the PAT) and indicated in table_ID 0x02.
Definition of Terms
787
Program Stream (PS)
Like the transport stream, the program stream is a multiplex stream but
only contains elementary streams for a program and is only suitable for the
transmission in ‘undisturbed’ channels (e.g. recording in storage media).
Program Specific Information (PSI)
The four tables below defined by MPEG-2 are summed up as program
specific information:
Program Association Table (PAT),
Program Map Table (PMT),
Conditional Access Table (CAT),
Network Information Table (NIT).
Quadrature Amplitude Modulation (QAM)
QAM is the modulation method used for transmitting a transport stream
via cable. The channel coding is performed prior to QAM.
Quadrature Phase Shift Keying (QPSK)
QPSK is the modulation method used for transmitting a transport stream
via satellite. The channel coding is performed prior to QPSK.
Rating Region Table (RRT)
The RRT is a PSIP table (ATSC). It comprises reference values for different geographical regions for the classification of transmissions (e.g.
'suitable for children older than X years'). RRT is transmitted with a section in the PID 0x1FFB and indicated by the table_ID=0xCA.
Running Status Table (RST)
The RST is an SI table (DVB) and contains status information about the
individual broadcasts. It is transmitted in TS packets with PID 0x0013 and
indicated by table_ID=0x71.
Section
Each table (PSI and SI) may comprise one or a number of sections. A
section may have a length of up to 1 kbyte (for EIT and ST up to
4 Kbytes). Most of the tables have 4 bytes at the end of each section for the
CRC.
Service Description Table (SDT)
The SDT is an SI table (DVB) and contains the names of programs and
broadcasters. It is transmitted in TS packets with PID 0x0011 and indicated by table_ID 0x42 or 0x46.
Service Information (SI)
The following tables defined by DVB are called service information.
They have the structure for 'private' sections defined by MPEG-2 systems:
Bouquet Association Table (BAT),
Service Description Table (SDT),
Event Information Table (EIT),
Running Status Table (RST),
788
Definition of Terms
Time and Date Table (TDT),
Time Offset Table (TOT).
Sometimes, the Program Specific Information (PSI) is also included.
Source Coding
The aim of source coding is data reduction by eliminating redundancy to
the greatest possible extent whilst affecting the relevance in a video or audio signal as little as possible. The methods to be applied are defined in
MPEG-2. They are the precondition for the bandwidth required for the
transmission of digital signals being narrower than that for the transmission of analog signals.
Stuffing Table (ST)
The ST is an SI table (DVB). It has no relevant content and is obtained
by overwriting tables that are no longer valid on the transmission path (eg
at cable headends). It is transmitted in TS packets with a PID of 0x0010 to
0x0014 and indicated by table_ID 0x72.
Sync byte
The sync byte is the first byte in the TS header and thus also the first
byte of each TS packet. Its value is 0x47.
Synchronous Digital Hierarchy (SDH)
The Synchronous Digital Hierarchy (SDH) is an international standard
for the digital transmission of data in a uniform frame structure (containers). All bit rates of the PDH can be transmitted, like ATM, by means of
SDH. Although SDH differs due to the pointer management, it is compatible with the American PDH and SONET standards.
Synchronous Optical NETwork (SONET)
The Synchronous Optical NETwork (SONET) is an American standard
for the digital transmission of data in a uniform frame structure (containers). All bit rates of the PDH can be transmitted, like ATM, by means off
SONET. SONET differs due to the pointer management and is thus not
compatibel with the European SDH standard.
System Software Update (SSU)
Standardized system software update for DVB receivers according to
ETSI TS102006.
System Target Decoder (STD)
The system target decoder describes the (theoretical) model for a decoder of MPEG-2 transport streams. A 'real' decoder has to fulfil all the
conditions based on STD if it is to be guaranteed that the contents of all
transport streams created to MPEG-2 are decoded error-free.
System Time Table (STT)
STT is a PSIP table (ATSC). It comprises date and time (UTC) as well
as the local time difference. STT is transmitted in TS packets with the PID
0x1FFB and indicated by the table_ID 0xCD.
Definition of Terms
789
Table_ID
The table_identity defines the type of table (eg PAT, NIT, SDT, etc) and
is always located at the beginning of a section of the table. The table_ID is
necessary especially because different tables can be transmitted with one
PID in one substream (eg BAT and SDT with PID 0x11, see Table 1-3).
Terrestrial Digital Multimedia Broadcasting (T-DMB)
South Korean standard for digital TV reception for mobile receivers,
based on DAB and MPEG-4 AVC and AAC
Terrestrial Virtual Channel Table (TVCT)
TVCT is a PSIP table (ATSC) comprising the characteristic data of a
program (eg channel number, frequency, modulation method) for terrestrial emission (transmission in cable → CVCT). TVCT is transmitted in TS
packets with the PID of 0x1FFB and indicated by the table_id 0xC8.
Time and Date Table (TDT)
The TDT is an SI table (DVB) and contains date and time (UTC). It is
transmitted in TS packets with PID 0x0014 and indicated by table_ID
0x70.
Time Offset Table (TOT)
The TOT is an SI table (DVB) and contains information about the local
time offset in addition to date and time (UTC). It is transmitted in TS packets with PID 0x0014 and indicated by table_ID 0x73.
Transport Error Indicator
The transport error indicator is contained in the TS header and is the
first bit after the sync byte (MSB of the second byte). It is set during channel decoding if channel decoding could not correct all the bit errors generated in the corresponding TS packet on the transmission path. As it is basically not possible to find the incorrect bits (e.g. the PID could also be
affected), the errored packet must not be processed any further. The frequency of occurrence of a set transport error indicator is not a measure of
the bit error rate on the transmission path. A set transport error indicator
shows that the quality of the transmission path is not sufficient for an error-free transmission despite error control coding. A slight drop in transmission quality will quickly increase the frequency of occurrence of a set
transport error indicator and finally transmission will cease.
Transport Stream (TS)
The transport stream is a multiplex data stream defined by MPEG-2
which may contain several programs that may consist of a number of elementary streams. A program clock reference (PCR) is carried along for
each program. Multiplexing is done by forming TS packets for each elementary stream and by stringing together these TS packets originating
from different elementary streams.
790
Definition of Terms
TS Header
The TS header is provided at the beginning of each TS packet and has a
length of four bytes. The TS header always begins with the sync byte
0x47. Further important elements are the PID and the continuity counter.
The TS header must never be scrambled when it is to be transmitted (see
Conditional Access).
TS Packet
The transport stream is transmitted in packets of 188 bytes (204 bytes
after channel coding). The first four bytes form the TS header which is followed by the 184 payload bytes.
Vestigial Sideband Modulation (VSB)
The vestigial sideband amplitude modulation method is used in ATSC
systems. For terrestrial transmission, 8VSB with 8 amplitude levels is used
while 16VSB is mainly for cable transmission.
TV Channel Tables
The channels listed in the following tables are possible examples for analog television and for DVB-C, DVB-T, ATSC and J83B.
Analog TV:
Vision carrier at 7 MHz bandwidth 2.25 MHz below center frequency,
vision carrier at 8 MHz bandwidth 2.75 MHz below center frequency,
vision carrier at 6 MHz bandwidth 1.75 MHz below center frequency.
ATSC:
Pilot carrier at ATSC (6 MHz bandwidth) 2.69 MHz below center frequency.
Europe, Terrestrial and Cable
Table 45.1. TV channel occupancy, Europe
Channel
Band
2
3
4
VHF I
VHF I
VHF I
VHF II
5
6
7
8
9
10
11
12
S1
VHF III
VHF III
VHF III
VHF III
VHF III
VHF III
VHF III
VHF III
special channel
Center
frequency
[MHz]
50.5
57.5
64.5
Bandwidth
[MHz]
Remarks
7
7
7
FM 87.5...108.0
MHz
177.5
184.5
191.5
198.5
205.5
212.5
219.5
226.5
107.5
7
7
7
7
7
7
7
7
7
not in use (FM)
792
S2
S3
S4
S5
S6
S7
S8
S9
S 10
S 11
S 12
S 13
S 14
S 15
S 16
S 17
S 18
S 19
S 20
S 21
S 22
S 23
S 24
S 25
S 26
S 27
S 28
S 29
S 30
S 31
S 32
S 33
S 34
S 35
S 36
S 37
S 38
S 39
S40
S41
21
22
23
24
25
26
TV Channel Tables
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
special channel
UHF IV
UHF IV
UHF IV
UHF IV
UHF IV
UHF IV
114.5
121.5
128.5
135.5
142.5
149.5
156.5
163.5
170.5
233.5
240.5
247.5
254.5
261.5
268.5
275.5
282.5
289.5
296.5
306
314
322
330
338
346
354
362
370
378
386
394
402
410
418
426
434
442
450
458
466
474
482
490
498
506
514
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
cable, midband
cable, midband
cable, midband
cable, midband
cable, midband
cable, midband
cable, midband
cable, midband
cable, midband
cable, superband
cable, superband
cable, superband
cable, superband
cable, superband
cable, superband
cable, superband
cable, superband
cable, superband
cable, superband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
cable, hyperband
Europe, Terrestrial and Cable
27
28
29
30
31
32
33
34
35
36
37
38
39
40
41
42
43
44
45
46
47
48
49
50
51
52
53
54
55
56
57
58
59
60
61
62
63
64
65
66
67
68
69
UHF IV
UHF IV
UHF IV
UHF IV
UHF IV
UHF IV
UHF IV
UHF IV
UHF IV
UHF IV
UHF IV
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
522
530
538
546
554
562
570
578
586
594
602
610
618
626
634
642
650
658
666
674
682
690
698
706
714
722
730
738
746
754
762
770
778
786
794
802
810
818
826
834
842
850
858
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
8
793
794
TV Channel Tables
Australia, Terrestrial
Table 45.2. TV terrestrial channel occupancy, Australia (terrestrial)
Channel
Band
VHF I
VHF I
VHF I
Center
frequency
[MHz]
48.5
59.5
66.5
0
1
2
7
7
7
3
4
5
5A
6
VHF II
VHF II
VHF II
VHF II
VHF III
88.5
97.5
104.5
140.5
177.5
7
7
7
7
7
7
VHF III
184.5
7
8
9
9A
10
11
12
28
VHF III
VHF III
VHF III
VHF III
VHF III
VHF III
UHF IV
191.5
198.5
205.5
211.5
219.5
226.5
529.5
7
7
7
7
7
7
7
29
30
31
32
33
34
UHF IV
UHF IV
UHF IV
UHF IV
UHF IV
UHF IV
536.5
543.5
550.5
557.5
564.5
571.5
7
7
7
7
7
7
35
36
37
38
39
40
41
42
43
44
45
UHF IV
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
578.5
585.5
592.5
599.5
606.5
613.5
620.5
627.5
634.5
641.5
648.5
7
7
7
7
7
7
7
7
7
7
7
Bandwidth
[MHz]
Remarks
”ABC Analog“
Sydney
s.t. “Seven Digital“
s.t. ”Seven Analog“
s.t. ”Nine Digital“
s.t. ”Nine Analog“
s.t. “Ten Analog“
s.t. “Ten Digital“
s.t. “ABC Digital“
“SBS Analog“
Sydney
“SBS Digital“
Sydney
North America, Terrestrial
46
47
48
49
50
51
52
53
54
55
56
57
58
59
60
61
62
63
64
65
66
67
68
69
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
UHF V
655.5
662.5
669.5
676.5
683.5
690.5
697.5
704.5
711.5
718.4
725.5
732.5
739.5
746.5
753.5
760.5
767.5
774.5
781.5
788.5
795.5
802.5
809.5
816.5
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
7
North America, Terrestrial
Table 45.3. TV terrestrial channel occupancy, North America
Channel
Band
2
3
4
5
6
7
8
9
10
11
12
VHF
VHF
VHF
VHF
VHF
VHF
VHF
VHF
VHF
VHF
VHF
Center
frequency
[MHz]
57
63
69
79
85
177
183
189
195
201
207
Bandwidth
[MHz]
6
6
6
6
6
6
6
6
6
6
6
Remarks
795
796
13
14
15
16
17
18
19
20
21
22
23
24
25
26
27
28
29
30
31
32
33
34
35
36
37
38
39
40
41
42
43
44
45
46
47
48
49
50
51
52
53
54
55
56
57
58
TV Channel Tables
VHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
213
473
479
485
491
497
503
509
515
521
527
533
539
545
551
557
563
569
575
581
587
593
599
605
611
617
623
629
635
641
647
653
659
665
671
677
683
689
695
701
707
713
719
725
731
737
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
North America, Cable
59
60
61
62
63
64
65
66
67
68
69
70
71
72
73
74
75
76
77
78
79
80
81
82
83
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
UHF
743
749
755
761
767
773
779
785
791
797
803
809
815
821
827
833
839
845
851
857
863
869
875
881
887
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
North America, Cable
Especially in the cable the current occupancy can not be guaranteed.
Table 45.4. Channel occupancy North America, Cable
Channel
2
3
4
5
6
7
8
9
Band
Center
frequency
[MHz]
57
63
69
79
85
177
183
189
Bandwidth
6
6
6
6
6
6
6
6
Remarks
797
798
10
11
12
13
14
15
16
17
18
19
20
21
22
23
24
25
26
27
28
29
30
31
32
33
34
35
36
37
38
39
40
41
42
43
44
45
46
47
48
49
50
51
52
53
54
55
TV Channel Tables
195
201
207
213
123
129
135
141
147
153
159
165
171
219
225
231
237
243
249
255
261
267
273
279
285
291
297
303
309
315
321
327
333
339
345
351
357
363
369
375
381
387
393
399
405
411
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
North America, Cable
56
57
58
59
60
61
62
63
64
65
66
67
68
69
70
71
72
73
74
75
76
77
78
79
80
81
82
83
84
85
86
87
88
89
90
91
92
93
94
95
96
97
98
99
100
101
417
423
429
435
441
447
453
459
465
471
477
483
489
495
501
507
513
519
525
531
537
543
549
555
561
567
573
579
585
591
597
603
609
615
621
627
633
639
645
93
99
105
111
117
651
657
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
799
800
102
103
104
105
106
107
108
109
110
111
112
113
114
115
116
117
118
119
120
121
122
123
124
125
126
127
128
129
130
131
132
133
134
135
136
137
138
139
140
141
142
143
144
145
146
147
TV Channel Tables
663
669
675
681
687
693
699
705
711
717
723
729
735
741
747
753
759
765
771
777
783
789
795
801
807
813
819
825
831
837
843
849
855
861
867
873
879
885
891
897
903
909
915
921
927
933
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
6
Europe, Satellite
148
149
150
151
152
153
154
155
156
157
158
939
945
951
957
963
969
975
981
987
993
999
801
6
6
6
6
6
6
6
6
6
6
6
Europe, Satellite
Fig. 45.1. shows the occupancy of the Ku band for direct broadcasting TV
satellite–reception.
f
f = 29.5 MHz / 39 MHz;
B = 26 MHz / 33 MHz;
V
H
B
FSS band
BSS band
11.7 GHz
10.95 GHz
Low band
Ku band
SMS band
12.5 GHz
High band
Fig. 35.1. Ku band for direct broadcasting TV satellite-reception
Bibliography:
12.75 GHz
Index
16APSK 286
16K 697
16QAM 232
16VSB 478
20T pulse 26
256QAM 306, 701
2K mode 372
2T K factor 24
3 dB coupler 595
32APSK 286
32K 697
3dB coupler 595
3-dB coupler 603
4096QAM 731
5.1 multichannel audio 161
64QAM 232, 306
8K mode 372
8PSK 264, 286
8VSB 467, 477, 495, 497
AAL0 777
AAL1 777
AAL5 777
AC-3 149
ACE 719
adaptation field 40, 47, 50, 79, 80
additive white gaussian noise 281
ADR 515
ADSL 347
Advanced Television Systems
Committee 467, 495
Advanced Video Coding 179
AIT 563
Alamouti 721
AM 219, 223
amplitude modulation 219, 223
analog video signals 24
ARIB 76
ASI 777
ASK 220, 467
Association of Radio Industries and
Business 76
ASTRA 262
ATM 40, 517, 777
ATSC 4, 52, 73, 204, 467, 471,
482, 495, 777
audibility 152
audio 147
Australia (terrestrial) 794
auxiliary stream data 726
AVC 180
AWGN 281, 330, 421, 430
baseband frame 674
baseband header 672
baseband scrambler 673
BAT 66, 777
Bayerischer Rundfunk 589
BCH 288, 777
BER 424, 496
bit error ratio 293, 339, 405, 423,
424, 496
Blackman 99
blocking 208, 658
Blocking Effects 209
blur 657, 658
804
Index
Boltzmann 415
Bose-Chaudhuri-Hocquenghem 288
Bouquet Association Table 65
BPSK 228, 290
broadband cable 305
Broadcast Web Page 553
Brotjacklriegel 591
burst 26
C/N 281, 293, 299, 325, 332
CA 777
Capacity Unit 551
carrier suppression 330, 336, 433
carrier/noise ratio 295
CAT 47, 777
CAT_error 200
cathode ray tube 643
CAZAC 531, 534
CB 81
CCD 647
CCIR 17 20
CCIR 601 81
CCVS 7, 10
CDMA 347
cell 701, 739
cell interleaver 701
center carrier 441
channel bundling 734
channel coding 242
chrominance 81
CI 777
CMMB 755
Coded Orthogonal Frequency
Division Multiplex 350, 369
COFDM 346, 350, 355, 367, 369,
395, 421, 517, 525, 576, 581, 777
COFDM modulator 354
COFDM receiver 362
COFDM spectrum 365
COFDM subcarriers 363
COFDM symbol 355, 360, 409
color cubcarrier 21
combiner 603, 605
Common Interleaved Frame 551
conditional access table 47
Constant Amplitude Zero
Autocorrelation 531
constellation analysis 422
constellation analyzer 326, 421
constellation diagram 328, 330, 429
Continual pilots 373
Continual Pilots 377
continuity counter 80
Continuity_count_error 195
convolutional code 243
convolutional coder 270, 272
convolutional coding 266, 538
CR 81
CRC 777
CRC_error 197
CRC-8 676
CRC-8 encoder 671
crest factor 444, 746
critical mask 447
Critical Mask 419
CRT 643, 647, 648
CU 551
CVBS 7, 10, 15
CVCT 777
D2MAC 2
DAB 6, 346, 515, 522, 523, 524,
528, 777
data broadcasting 520, 553
data line 170
Data piping 560
data slice 744
data stream mode 568
data streaming 560
DBB 562
DBPSK 374
DCT 2, 101, 175
DDB 777
decoding time stamps 35
de-interleaver 270
deinterleaving 280
Delete Null Packet 676
demapper 234
DFT 97
differential gain 19, 24
differential phase 19, 24
Index
Digital Audio Broadcasting 346,
515
Digital Dividend 732, 760
digital micro-mirror chips 643
digital modulation 219
Digital Radio Mondiale 575
Digital Storage Media Command
and Control 53, 455
DII 562, 777
Dirac 100, 352
direct modulation 437
Discrete Cosine Transform 101
Discrete Fourier Transform 97, 99,
101
discrete multitone 346
DLP 643, 654
DMB-T 78, 777
DMT 346
DNP 676
Dolby digital 149
Dolby Digital AC-3 159
Doppler 403
Doppler effect 421
Double Stimulus Continual Quality
Scale method 211
DQPSK 520, 525, 527, 529, 531
DRM 575, 777
DSCQS 209
DSI 562, 777
DSLAM 572
DSM-CC 53, 55, 455, 461, 560,
561, 777
DSNG 285
DSR 515
DTMB 4
DTS 35, 51, 777
DVB 3, 39, 61, 189, 777
DVB Measurement Guidelines 421,
442
DVB-C 305, 306, 325
DVB-C modulator 308
DVB-C receiver 309
DVB-C2 6, 662, 677, 684, 731
DVB-GSE 667
DVB-H 454, 458, 463, 559, 561
DVB-IP 571
805
DVB-S 263, 264
DVB-S measuring technology 293
DVB-S receiver 295
DVB-S2 285, 286, 661
DVB-SH 6, 464
DVB-T 347, 369, 371, 421
DVB-T constellation 389
DVB-T demodulator 397
DVB-T receiver 393
DVB-T standard 390
DVB-T test receiver 396
DVB-T Transmitter 419
DVB-T2 6, 662, 677, 687
DVB-T2 frame 706
DVB-x2 667
DVCPro 177
DVD 4, 176
DVI 186
DVQ 211
dynamic PMT 59
EAV 83
EB/N0 300
EBU TTXT 167
ECM 47, 777
EDGE 451
EIRP 283
EIT 53, 67, 777
EIT_other_error 204
electronic program guide 1
Elementary Stream 35
EMM 47, 777
END 339
energy dispersal 267
Ensemble 520
Ensemble Transport Interface 517,
520, 521
EOF 551
EPG 51
equivalent isotropic radiated power
283
equivalent noise degradation 339
error vector magnitude 330, 339
ES 777
ETI 517, 519, 520, 521, 536, 551
ETT 74, 777
806
Index
Eutelsat 264
EVM 339
Extended Carrier Mode 699
Extended Text Table 75
FAC 578, 580
fading 349
fall off the cliff 366
Fast Access Channel 580
Fast Fourier Transform 99, 101
Fast Information Block 549
Fast Information Channel 528
Fast Information Data Channel 549
FEC 242, 245, 521, 537, 579, 777
FEC frame 289
FFT 99, 395
FFT sampling window 394
FIB 549
FIC 528, 535, 539, 549
FIDC 549
field strength 416
Field strength 416
flat screen 643
FM 219
FMextra 757, 759
Forney interleaver 269
Forward Error Correction 242
Fourier 94
Fourier analysis 95
Fourier transform 95, 238
frame_boundary 461
free space attenuation 283
free space condition 417
free-space loss 277
frequency modulation 219
FSK 220
Full HD 187
Future Extension Frame 709
Future Extension Frames 726
G.703 517
G.704 517
Gallager 661
Gaussian channel 401
Gaussian Minimum Shift Keying
452
Gaussian noise 398
GCS 667
Generic Continuous Stream 667,
669
Generic Encapsulated Stream 667,
670
Generic Fixed Packetized Stream
667, 669
geostationary satellite 261, 262
GFPS 667
Ghost Pattern 501
Gigabit Ethernet 188, 572, 573
Global Positioning System 406
Global System for Mobile
Communication 451
GMSK 452
GPRS 451
GPS 406, 551, 552
GSE 667
GSM 451
guard interval 360, 361, 408
H.264 179, 452
Hamming 99
Hanning 99
HD Radio 757, 758
HD Ready 187
HDMI 187
HD-SDI 89, 188
HDTV 2, 87, 176, 662
HEM 673, 691
hierarchical modulation 286, 366,
370, 379, 380, 581
High Definition Television 87
High Definition TV 1
Hilbert transform 238, 239, 241,
358
Hilbert transformer 472, 479
HM 581
horizontal sync pulse 12
HP 370, 379
HTML 564
I/Q amplitude imbalance 330, 433
I/Q errors 431, 435
I/Q imbalance 335
Index
I/Q modulator 334
I/Q phase error 330, 336, 433
IBOC 757, 758
ICPM 25
IDFT 97
IFFT 99, 355, 431, 527
IFFT sampling frequency 381
impulse response 446
impulsive noise 713
INT 461
interferer 325
interlace 655
interleaver 270, 711, 739
Internet 6
intersymbol interference 406
inter-symbol interference 348, 362
Inverse Discrete Fourier Transform
97
Inverse Fast Fourier Transform 99,
355
IP 572
IPTV 6
IQ demodulation 233
IQ error 310
IQ modulation 219, 225
IQ modulator 225
IRD 777
ISDB-T 4, 76, 777
ISMA 570
ISSY 672, 674
ITU 17 20
ITU 601 183
ITU-BT.R601 85
ITU-T J83A 307
ITU-T J83B 307
ITU-T J83C 307
J83 777
J83B 307
Java 564
JPEG 2, 175
L band 523
LCD 643, 652
LDPC 288, 289, 687, 731, 777
liquid crystal display 643
807
LNB 278
low density parity check 285, 288
Low Density Parity Check Codes
687
LP 370, 379
luminance 8
LVDS 777
MAC 457
macroblock 208
Main Service Channel 528, 578
mapper 225, 230, 231
Mask Filter 419
masking threshold 153
Master Guide Table 74
MATYPE 672
MCI 547
Measurement Group 189
Measurement Guidelines 189, 333,
337
MediaFLO 757
megaframe 410
MER 337, 441, 442, 499
MER(f) 443
MGT 74, 778
MHEG 559, 778
MHP 559, 563, 778
Microsoft Windows Media 9 181
MiniDV 177
Minimum Receiver Input Level 414
MIP 410, 412, 778
MISO 720, 722
mixer 221
modulation error ratio 330, 337,
441
mosquito noise 658
MOT 548, 553
motion blur 644
Moving Picture Experts Group 178
Moving Pictures Experts Group 31
MP@ML 778
MP3 148
MPE 455, 456, 460, 561
MPEG 3, 31, 778
MPEG-1 3, 33, 175
MPEG-2 3, 33, 175
808
Index
MPEG-2 transport stream 38, 668
MPEG-21 178
MPEG-4 88, 178
MPEG-4 AAC 575, 579
MPEG-4 Part 10 452
MPEG-7 178
MSC 528, 536, 546, 548, 578
MST 551
Multimedia Home Platform 563
Multimedia Object Transfer 548
multipath 526
multipath reception 348, 431
multiple input streams 674
Multiple Input/Single Output 720
Multiple PLP 674
Multiplex Configuration
Information 547
Multiprotocol Encapsulation 455,
457, 464
MUSICAM 148, 515, 523, 546
Nipkow disc 2
Nipkow disk 643, 646
NIT 56, 60, 62, 778
NIT_other_error 204
NM 673, 691
noise marker 341
Noise marker 426
noise power density 428
Normal Carrier Mode 697
North America 795
North America, Cable 797
notch 744
NRZ 219, 227
NTSC 7, 10
Null Packet Deletion 675
null symbol 529
Nyquist 219, 468, 482
object carousel 561
Objective Picture Quality Analysis
211
OFDM 93, 367, 778
OLED 643, 654
Olympic Tower Munich 588
Orthogonal Frequency Division
Multiplex 93
orthogonality 352
orthogonality condition 353
P1 symbol 706, 707
P2 symbol 707
Packet Mode 548
packetized elementary stream 33
PAD 524
PAL 2, 7, 10
PALplus 2
PAPR 719, 734
PAT 44, 56, 778
PAT_error 192
Payload Unit Start Indicator 78
PCMCIA 778
PCR 49, 198, 778
PCR jitter 49, 199
PCR_accuracy_error 199
PCR_error 198
PCR_PID 59
PDH 517, 778
PES 33, 35, 778
phase jitter 330, 334, 430
phosphor lag 644
physical layer frame 289
Physical Layer Frame 704
Physical Layer Pipe 668, 691
picture freeze 216
picture loss 216
picture quality 207
PID 38, 42, 44, 46, 778
PID_Error 194
pilot carriers 366
pilot cell 582
pilot pattern 717
pilot structure 745
pilots 715
plasma display 643
plasma screen 651
playout 411
Playout Center 590
PLP 668, 678, 691, 735, 742
PMT 44, 56, 58, 778
PMT_error 193
Index
polarization 418
Portable indoor 689
PRBS 244, 268, 538
Presentation Time Stamp 35, 199
Profile 778
Program Association Table 44, 57
Program Clock Reference 49, 198
Program Map Table 44
progressive 655
PS 778
pseudo random binary sequence
244
PSI 53, 778
PSIP 74, 472, 778
PSK 220
psychoacoustic model 148
PTS 35, 51, 199, 778
PTS_error 199
pull-down 645
puncturing 275
QAM 220, 778
QPSK 229, 232, 264, 286, 778
quadrature error 435
quadrature phase shift keying 264
Quantization 157
rainbow effect 644
Rayleigh channel 402
Reed Solomon 266, 280, 423
Reed Solomon decoder 397
Reed-Solomon 265, 269, 294, 307,
475
residual carrier 433
Residual picture carrier 25
RGB 10
Ricean channel 401
RMS detector 297
roll-off factor 277, 280, 287, 333
roll-off filtering 277, 299
Roof antenna reception 689
rotated constellation 702
RRT 76, 778
RST 60, 67
RTP 570
809
S/N 325, 332
SA 214
satellite 261
satellite receiver 279
SAV 83
SAW 326, 394
SAW filter 309
Scattered pilots 373
scrambling 47, 244
SDC 578, 580
SDH 517, 778
SDI 85, 182
SDT 60, 64, 778
SDT_other_error 204
SDTV 32, 87, 176
SECAM 7, 10
Second Generation Digital Video
Broadcasting 661
section 52
Serial Digital Interface 85
Service Description Channel 580
Service Information 549
Service Transport Interface 520
SFN 406, 412, 551, 587
Shannon 242, 244, 688, 733
shoulder attenuation 293, 303, 343,
365, 399, 446
SI 51, 53, 54, 60, 61, 70, 72, 73, 74,
76, 78, 549, 778
SI Other 204
SI Tables 202
SI_repetition_error 201
SIMO 720
simple modulation 581
Single frequency network 726
Single Input Stream 677
Single Stimulus Continual Quality
Evaluation 209
Single Stimulus Continual Quality
Evaluation method 211
single-frequency network 408, 551
single-frequency networks 406, 587
sinusoidal interferer 334
SM 581
SMPTE 310 186
810
Index
SONET 778
sound loss 216
spatial activity 214
spectrum 293
spectrum analyzer 296, 340, 400,
425
SSCQE 209
SSU 559, 565, 778
ST 61, 778
Standard Definition Television 32,
87
Standard Definition TV 1
STC 48, 50
STD 778
STI 520
Stream Adaptation 676
Stream Mode 548
STT 75, 778
subband coding 157
subcarrier 351, 354
subcarrier spacing 353, 382
subjective picture quality analysis
210
sub-slicing 717
subtitles 165
superframe 409, 581
Symbol rate 333
sync byte 44
sync byte inversion 267
sync_byte_error 192
synchronization time stamp 412
System Software Update 559, 565
System Time Clock 198
T2-MI 680, 683, 687
T2-MIP 683
T2-Modulator Interface 687
TA 214
table_boundary 461
table_ID 59
TCP 456
T-DMB 548, 554, 778
TDT 60, 68, 778
teletext 1, 20, 22, 165, 167
teletext PES 169
Terrestrial Digital Multimedia
Broadcasting 554
test receiver 326, 423
test transmitter 293, 343, 421
TFPR 527, 531, 533
TFS 718
the temporal activity 214
TII 533, 728
time frequency phase reference 531
time interleaver 712
Time Slicing 460
Time-Frequency-Slicing 718
TIST 542, 551, 552
TOT 68, 778
TPS 375, 376, 422, 459
TPS carriers 373
TPS Carriers 377
TR 719
transform coding 161
Transmitter identification 728
transmitter identification
information 533
Transport Error Indicator 196, 280
Transport Scrambling Control 200
Transport Scrambling Control Bits
79
transport stream 37, 38, 41
Transport_Error 196
traveling wave tube amplifier 264
trellis 476
trellis coder 270
trellis diagram 274
Triple Play 569, 573
TS 667, 778
TS_sync_loss 191
TS-ASI 182, 185
TVCT 75, 778
TV-Kanalbelegung 791
TWA 264, 277
UDP 454, 456, 572
UHF 8
UMTS 451
uncritical mask 447
unequal error protection 540
unicast 572
Index
Universal Mobile
Telecommunication System 451
unreferenced PID 203
unsharp picture areas 207
UNT 563, 565
UPL 672
Variable Coding and Modulation
678, 704
VBI 170
VC-1 181
VCD 176
VCM 662, 678, 704, 740
VCT 75
VDSL 569
vertical insertion test signals 20
vertical sync pulse 13
vertical synchronization pulses 12
Vestigial sideband 470
VHF 8, 545
Video CD 176
video program system 23
Video Program System 165
video quality analyzer 208
Video Quality Experts Group 209
811
Virtual Channel Table 75
Viterbi 243, 294, 397, 404, 423,
424, 476, 496
Viterbi decoder 275, 280
VITS 20
VPS 23, 165, 169, 171
VQEG 209
VSB 476, 778
VSB-AM 18
WCDMA 452
Wendelstein 588, 592, 594, 598,
605, 607
White bar amplitude 24
Wideband Code Division Multiple
Access 452
WiMAX 721
window function 110
Windows Media 9 463
xDSL 570
Y/C 10
Typical Test Instruments and Broadcasting Systems for
TV Signals
R&S®SFU, R&S®SFE and R&S®SFE100 Broadcast Test
Transmitters
Test transmitters provide reference RF signals for testing digital receivers.
The intentional degradation of the ideal signal by superimposed noise as
well as the simulation of mobile reception scenarios help to make the receivers operational under any conditions and to ensure the interferencefree reception of TV programs.
R&S®DVSG Digital Video Signal Generator
Comprehensive tests are required to assess the functioning of the picture
processing unit of a TV set or the quality of built in display panels: All
resolution, timing and color depth modes must be verified. To make sure
that the quality of the display unit under test is assessed, the test signal
source must satisfy extremely high quality requirements. Digital video signal generators are reference signal sources for development and quality assurance applications of latest-generation TV sets and projectors.
R&S®ETL TV Analyzer and R&S®ETH Handheld TV
Analyzers
TV analyzers measure high-precision RF signals of both analog and digital
TV systems. Measurements performed directly on an antenna, a TV transmitter, or cable headend make it possible to clearly assess the quality of
digital transmission. Error sources can be identified and specifically eliminated.
Monitoring, Analysis, Recording, and Generation of
MPEG Transport Streams with the R&S®DVM Family
An MPEG transport stream is modified at many places in the transmission
chain. For example, after the satellite signal has arrived at a cable headend,
various programs are taken from the transport stream multiplex and replaced by local programs. This is a deep intervention in the transport
stream structure. MPEG analyzers check the entire MPEG protocol syntax
and indicate any errors and discrepancies, thus ensuring the secure transmission of signals.
Transmitters, Transposers and Gap Fillers for ATV, DTV
and Mobile TV from 1 W to 60 kW
TV transmitters are one of the most important components of terrestrial
transmitter networks. They convert the MPEG transport stream signal into
high-quality RF signals of different power classes from low power to high
power. These signals are then distributed via antennas. Reliability, small
footprint, and high efficiency are key parameters for securely providing
viewers with digital TV signals via antenna.
R&S AEM100 Multiplexer for ATSC Mobile DTV
As a key component for ATSC Mobile DTV, the R&S AEM100 inserts the
mobile data into the existing ATSC multiplex. It generates the required robust transport stream, which is output via ASI or Ethernet. The
R&S AEM100 includes all of the required functions for ATSC Mobile
DTV. The complete functionality can be configured, controlled and monitored using a web browser locally or remotely. All commands for automatic monitoring and for the instrument settings are available over an
SNMP interface.