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WO2022048287A1 - 一种dc/dc变换器及通信电源 - Google Patents

一种dc/dc变换器及通信电源 Download PDF

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Publication number
WO2022048287A1
WO2022048287A1 PCT/CN2021/103422 CN2021103422W WO2022048287A1 WO 2022048287 A1 WO2022048287 A1 WO 2022048287A1 CN 2021103422 W CN2021103422 W CN 2021103422W WO 2022048287 A1 WO2022048287 A1 WO 2022048287A1
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WIPO (PCT)
Prior art keywords
converter
resonant
dead point
winding
point
Prior art date
Application number
PCT/CN2021/103422
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English (en)
French (fr)
Inventor
叶琼
程杰斌
王庆海
郭甫龙
Original Assignee
华为技术有限公司
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Application filed by 华为技术有限公司 filed Critical 华为技术有限公司
Priority to EP21863355.0A priority Critical patent/EP4080751A4/en
Publication of WO2022048287A1 publication Critical patent/WO2022048287A1/zh
Priority to US17/879,579 priority patent/US12119740B2/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
    • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
    • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0064—Magnetic structures combining different functions, e.g. storage, filtering or transformation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from AC input or output
    • H02M1/123—Suppression of common mode voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from AC input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/01—Resonant DC/DC converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/01—Resonant DC/DC converters
    • H02M3/015—Resonant DC/DC converters with means for adaptation of resonance frequency, e.g. by modification of capacitance or inductance of resonance circuit
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
    • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
    • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
    • H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33571—Half-bridge at primary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
    • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
    • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
    • H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33573—Full-bridge at primary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
    • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
    • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
    • H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
    • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
    • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
    • H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only

Definitions

  • the present application relates to the technical field of power electronics, and in particular, to a DC/DC converter and a communication power supply.
  • switching power converters are widely used in communication power supplies due to their high efficiency and small size.
  • the power density of the power module is required to be higher and higher, but the volume of magnetic devices in the current switching power converter is relatively large.
  • the operating frequency of the switching power supply can be increased from 200kHz to 1M-10MHz.
  • increasing the switching frequency increases the switching noise (the naked noise increases by 20dB+), which puts a lot of pressure on the pre-filter circuit. Therefore, how to reduce the noise of the switching power supply and reduce the cost of the pre-stage filter circuit has become an important direction of current research.
  • FIG. 1 this figure is a schematic diagram of suppressing common mode interference provided by the prior art.
  • the common mode current output by the AC power supply is suppressed by the common mode inductor 100 .
  • the common mode inductor 100 presents a high resistance characteristic to the common mode current, which can attenuate the common mode noise, so that a better common mode filtering effect can be achieved.
  • the common mode inductor should be enlarged to prevent saturation of the magnetic core, and the volume of the magnetic core will increase accordingly, which will eventually lead to a very large volume of the common mode inductance, which is not conducive to the miniaturization of the filter circuit and limits the miniaturization of the switching power supply. change.
  • the present application provides a DC/DC converter and a communication power supply, which can suppress common mode noise and facilitate the miniaturization of the switching power supply.
  • An embodiment of the present application provides a DC/DC converter, which is an LLC resonant DC/DC converter, which can be a full-bridge LLC resonant DC/DC converter or a half-bridge LLC resonant DC converter
  • the working principle of the /DC converter is to convert the input DC power into AC power, and pass it to the rectifier circuit at the output end through the transformer. Interference signals from the primary winding of the transformer are not transmitted to the secondary winding of the transformer.
  • the DC/DC converter includes: a primary winding, a secondary winding, a resonant inductor, a resonant capacitor and a noise suppression network; the primary winding, the resonant inductor and the resonant capacitor are connected in series at the first node and the second node node; the secondary winding is connected to the third node and the fourth node; the primary winding and the secondary winding form a transformer; the noise suppression network is connected to the primary static point and the secondary static point of the transformer between points; the primary side dead point is the DC stable potential point at the input end of the DC/DC converter, and the secondary side dead point is the DC stabilized potential point at the output end of the DC/DC converter; the primary side The first parasitic capacitance of the first end of the winding relative to the dead point of the secondary side is equal to the second parasitic capacitance of the second end of the primary winding relative to the dead point of the secondary side; the noise suppression network is used for A suppression current is generated.
  • the direction of the total noise current generated by the resonant inductor and the resonant capacitor is opposite to the direction of the suppression current.
  • the suppression current is used to suppress the total noise current.
  • the voltage of the noise suppression network varies with the size of the load. and change.
  • the first parasitic capacitance C AD ⁇ the second parasitic capacitance C BD , so a common mode noise channel is formed between the primary winding and the secondary winding of the transformer.
  • C AD C BD
  • the balance capacitor can be added between point A and point D , or between point B and point D. Specifically, it is necessary to analyze the relationship between CAD and CBD .
  • the dead point means that the DC potential of the dead point is stable relative to GND.
  • the dead point exists on the primary side of the transformer and also exists on the secondary side of the transformer. As the name implies, the potential of the moving point fluctuates relative to the static point.
  • the technical solution adds a noise suppression network on the basis of including a balance capacitor in the DC/DC converter.
  • the noise suppression network can generate a suppression current, and the direction of the suppression current is opposite to the direction of the total noise current generated by the resonant inductor and the resonant capacitor. Therefore, The total noise current generated by the resonant inductor and the resonant capacitor can be suppressed.
  • the current source of the noise suppression network can be additionally increased or derived from the energy inside the DC/DC converter.
  • the noise suppression network is connected to the primary side of the transformer. and the secondary side dead point, so that the common mode noise formed by the resonant circuit on the parasitic capacitance between the primary winding and the secondary winding of the transformer can be effectively reduced.
  • the voltage of the noise suppression network varies with the size of the load, so it can be applied to common mode noise suppression in the entire load range. As the load changes, the current will change. Since the voltage amplitude on the inductance (resonant inductance Lr and leakage inductance Lk) is proportional to the current, as the load changes, the voltage amplitude on Lk and Lr and the resonant capacitor Cr will change.
  • the voltage of the noise suppression network provided in this embodiment varies with the size of the load, so common-mode noise suppression within the full load range can be achieved, so that regardless of the size of the load, the noise suppression network can automatically adjust its own output according to the size of the load , thus canceling out common-mode noise over the full load range.
  • the noise suppression network includes: a signal source and an impedance network; the impedance network is used to adjust the magnitude of the voltage amplitude injected by the signal source between the primary side dead point and the secondary side dead point.
  • the signal source is connected in series with the impedance network between the primary side dead point and the secondary side dead point.
  • the signal source can be implemented in various ways, an additional signal source can be added, or a signal source can be formed by directly coupling energy from the converter, for example, coupling energy from a resonant inductor, or coupling energy from a resonant capacitor.
  • the signal source couples energy from the resonant inductor; the voltage magnitude of the signal source is proportional to the voltage magnitude of the resonant inductor. That is, when the voltage on the resonant inductor increases, the voltage of the signal source also increases.
  • the inductance can be used to couple energy from the resonant inductance through a magnetic field, that is, the signal source includes a cancellation inductance; the cancellation inductance is used for coupling energy from the resonant inductance, the A cancellation inductance is connected in series with the impedance network between the primary side dead point and the secondary side dead point.
  • the offset inductance and the resonant inductance are wound with a common magnetic core.
  • the energy can also be coupled from the resonant inductance through a capacitor, that is, the signal source includes: a first capacitive coupling circuit; the first capacitive coupling circuit is used for From the resonant inductive coupling energy, the first capacitive coupling circuit is connected in series with the impedance network between the primary dead point and the secondary dead point.
  • the signal source couples energy from the resonant capacitor; the voltage amplitude of the signal source is proportional to the voltage amplitude of the resonant capacitor. That is, when the voltage on the resonance capacitor increases, the voltage of the signal source also increases.
  • the signal source couples energy from the resonant inductor, and the following describes the signal source coupling energy from the resonant capacitor.
  • Coupling energy from the resonant capacitor can be achieved in various ways.
  • the energy can be coupled from the resonant capacitor through a capacitive coupling circuit, or the energy can be coupled from the resonant capacitor through a coupling transformer, which will be introduced separately below.
  • the signal source includes: a second capacitive coupling circuit; the second capacitive coupling capacitor is used for coupling energy from the resonance capacitor, and the second capacitive coupling circuit and the impedance network are connected in series to the original between the side dead point and the secondary side dead point.
  • the signal source includes: a signal coupling transformer; the primary winding of the signal coupling transformer is used for coupling energy from the resonant capacitance, and the secondary winding of the signal coupling transformer is connected in series with the impedance network and then connected to the between the static point of the primary side and the static point of the secondary side.
  • the sum of the total noise current generated by the resonant inductance and the resonant capacitor on the parasitic capacitance and the current of the cancellation impedance is zero; the parasitic capacitance is the primary winding and parasitic capacitance between the secondary windings.
  • V C /Z1+V L /Z2+V/Z 0;
  • VC represents the voltage across the resonant capacitor
  • VL represents the inductance across the resonant inductor
  • Z1 represents the impedance presented by CAD
  • Z2 represents the impedance presented by CBD.
  • the DC/DC converter includes at least one of the following: a full-bridge LLC resonant DC/DC converter and a half-bridge LLC resonant DC/DC converter.
  • the secondary winding includes: a first secondary winding and a second secondary winding; the first secondary winding is connected between the third node and the fifth node, the second secondary winding is connected between the fifth node and the fourth node; the third node and the fourth node are connected to the full-wave rectifier circuit; the output end of the full-wave rectifier circuit is the output end of the DC/DC converter.
  • the signal source couples energy from the resonant inductance, no additional signal source is required, and it is simple and easy to implement.
  • By winding an offset inductor on the magnetic core of the resonant inductor the energy can be coupled from the resonant inductor.
  • the network Z can adjust the amplitude of the injected signal. When the load changes, the currents passing through Cr and Lr will change. Therefore, as the current changes, the amplitude of the voltage that cancels the inductive coupling also changes. Therefore, the The scheme is suitable for noise suppression over the full load range.
  • it further includes: a full-bridge rectifier circuit; the number of secondary windings is one; the third node and the fourth node are the positive and negative electrodes of the input end of the full-bridge rectifier circuit, respectively; the full-bridge rectifier The output end of the circuit is the output end of the DC/DC converter.
  • the impedance network includes at least one of the following: resistance, capacitance and inductance, that is, at least one or more of the above three items, for example, the impedance network may include one, two, or three of the above items.
  • Embodiments of the present application further provide a communication power supply, including the DC/DC converter described above, and further comprising: a rectifier circuit; a first end of the rectifier circuit is used to connect an AC power supply; the rectifier circuit is used to convert the The alternating current of the alternating current power supply is converted into direct current; the second end of the rectifier circuit is used to connect the first end of the DC/DC converter; the second end of the DC/DC converter is used to supply power to the load; The DC/DC converter is used for converting the direct current and then supplying it to the load.
  • a communication power supply including the DC/DC converter described above, and further comprising: a rectifier circuit; a first end of the rectifier circuit is used to connect an AC power supply; the rectifier circuit is used to convert the The alternating current of the alternating current power supply is converted into direct current; the second end of the rectifier circuit is used to connect the first end of the DC/DC converter; the second end of the DC/DC converter is used to supply power to the load; The
  • the load can be any electrical equipment, and the voltage output by the DC/DC converter 200 can be different according to the load, for example, it can output voltages such as 48V, 12V, 5V, and 3.3V.
  • the noise suppression network can generate a suppression current, and the direction of the suppression current is opposite to the current direction of the resonant inductor and the resonant capacitor, so it can The total noise current generated by the resonant inductor and resonant capacitor is suppressed.
  • the current source of the noise suppression network can be additionally increased, or it can be derived from the energy inside the DC/DC converter.
  • the noise suppression network is connected to the primary side of the transformer. Between the secondary side static points, the common mode noise formed by the resonant circuit on the parasitic capacitance between the primary winding and the secondary winding of the transformer can be effectively reduced.
  • the voltage of the noise suppression network varies with the size of the load, so it can be applied to common mode noise suppression in the entire load range. Therefore, the communication power supply having the DC/DC converter can achieve a better common mode noise suppression effect, thereby providing a higher quality power supply to the load.
  • the DC/DC converter provided by this technical solution is an LLC resonant DC/DC converter, including a transformer and a resonant circuit, the resonant circuit is connected in series with the primary winding of the transformer, and the resonant circuit includes a series-connected resonant inductor and resonant capacitor. Therefore, when the load size is different, the current flowing through the leakage inductance is also different, and then the impact of the current size on the leakage inductance is different. When the inductance flowing through the leakage inductance is larger, the leakage inductance has a greater influence as a noise source , so it cannot be ignored.
  • the first parasitic capacitance at the first end of the primary winding of the transformer relative to the secondary static point is equal to the second end of the primary winding relative to the secondary static point.
  • a noise suppression network is added to the DC/DC converter.
  • the noise suppression network can generate a suppression current.
  • the direction of the suppression current is opposite to the direction of the total noise current generated by the resonant inductor and the resonant capacitor. Therefore, the total noise current generated by the resonant inductor and the resonant capacitor can be suppressed.
  • the current source of the noise suppression network can be additionally increased or derived from the energy inside the DC/DC converter.
  • the noise suppression network is connected to the primary side of the transformer. It can effectively reduce the common mode noise formed by the resonant circuit on the parasitic capacitance between the primary winding and the secondary winding of the transformer.
  • the voltage of the noise suppression network varies with the size of the load, so it can be applied to common mode noise suppression in the entire load range.
  • FIG. 2 is a schematic diagram of a full-bridge LLC resonant DC/DC converter provided by an embodiment of the present application
  • FIG. 3 is a schematic diagram of a half-bridge LLC resonant DC/DC converter provided by an embodiment of the present application
  • FIG. 4 is a schematic diagram of another half-bridge LLC resonant DC/DC converter provided by an embodiment of the present application.
  • FIG. 5 is a schematic diagram of another full-bridge LLC resonant DC/DC converter provided by an embodiment of the present application.
  • FIG. 6 is a schematic diagram of a DC/DC converter provided by an embodiment of the present application.
  • FIG. 7 is a schematic diagram of a DC/DC converter in which a signal source couples energy from a resonant inductance according to an embodiment of the present application;
  • FIG. 8 is a schematic diagram of a DC/DC converter for coupling energy from a resonant inductance from another signal source according to an embodiment of the present application
  • FIG. 9A is a schematic diagram of a small signal model corresponding to FIG. 7 and FIG. 8 provided by an embodiment of the present application;
  • FIG. 9B is a schematic diagram of the capacitive coupling circuit coupling energy from the resonance inductance according to an embodiment of the present application.
  • FIG. 10 is a schematic diagram of a DC/DC converter in which a signal source couples energy from a resonant capacitor according to an embodiment of the present application;
  • 11A is a schematic diagram of another DC/DC converter in which a signal source couples energy from a resonant capacitor according to an embodiment of the present application;
  • 11B is a schematic diagram of a signal coupling transformer coupling energy from a resonant capacitor according to an embodiment of the present application
  • FIG. 13 is an equivalent schematic diagram of a half-bridge LLC resonant DC/DC converter provided by an embodiment of the application;
  • FIG. 14 is a schematic diagram of a communication power supply provided by an embodiment of the present application.
  • the DC/DC converter provided in the embodiment of the present application is an isolated switching power supply.
  • the LLC resonant DC/DC converter is taken as an example for introduction. According to the circuit topology of the primary side connection of the transformer, it can be a full-bridge LLC resonant DC/DC converter.
  • the DC converter can also be a half-bridge LLC resonant DC/DC converter.
  • the form of the rectifier circuit connected to the secondary winding of the transformer is not limited.
  • FIG. 2 a schematic diagram of a full-bridge LLC resonant DC/DC converter provided in an embodiment of the present application.
  • the working principle of the LLC resonant DC/DC converter is to convert the input DC power into AC power, and pass it to the rectifier circuit at the output end through the transformer. As a result, the interference signal of the primary winding of the transformer will not be transmitted to the secondary winding of the transformer.
  • the primary winding of the transformer in Fig. 2 includes the excitation winding Lm, and the leakage inductance Lk of the primary winding and the secondary winding of the transformer is also drawn at the position of the primary winding of the transformer.
  • the resonant inductor Lr and the resonant capacitor Cr are connected in series with the primary winding of the transformer to form LLC resonance.
  • the secondary winding of the transformer is connected to the full-wave rectifier circuit as an example for illustration.
  • the primary winding, Lr and Cr of the transformer are connected in series to connect the first node and the second node.
  • the first node is the midpoint V1N of the first bridge arm of the full bridge
  • the second node is the second bridge arm of the full bridge. The midpoint of V2N.
  • the secondary winding of the transformer is connected between the third node C and the fourth node E.
  • the secondary winding of the transformer also includes a center tap, that is, point D is The terminal of the center tap of the secondary winding.
  • Point D and point G in Fig. 2 are the positive pole and negative pole of the output end of the DC/DC converter, respectively, and the negative pole is grounded.
  • the static point of the primary side of the transformer is the positive pole (point M) or negative pole (N) of the input end of the DC/DC converter
  • the static point of the secondary side of the transformer is the positive pole (point D) or ground of the output end of the DC/DC converter
  • the static point in the embodiments of the present application refers to the static point for small AC signals, that is, for the ground potential GND, the potential of the static point is the DC stable potential point, that is, the DC potential of the static point is stable relative to GND.
  • the potential of the moving point fluctuates relative to the static point.
  • the potentials of points C and E are fluctuating and belong to AC potential, so points C and E are the secondary moving points of the transformer, while points A and B are the primary moving points of the transformer.
  • FIG. 3 it is a schematic diagram of a half-bridge LLC resonant DC/DC converter according to an embodiment of the present application.
  • the conversion circuit connected to the primary side of the transformer has only one bridge arm, one end of the primary winding of the transformer is connected to the midpoint V1N of the bridge arm, and the other side of the primary winding of the transformer is connected One end is connected to the common terminal V2N of the capacitors C1 and C2.
  • FIG. 4 it is a schematic diagram of another half-bridge LLC resonant DC/DC converter provided in an embodiment of the present application.
  • Figures 3 and 4 respectively show two topological forms of the half-bridge LLC resonant DC/DC converter connected to the primary winding. The difference between Figure 4 and Figure 3 is only the location of the V2N connection.
  • FIG. 5 it is a schematic diagram of another full-bridge LLC resonant DC/DC converter provided in an embodiment of the present application.
  • the rectifier circuits connected to the secondary winding shown in FIG. 2 to FIG. 4 are all full-wave rectifier circuits, and the rectifier circuit shown in FIG. 5 is a full-bridge rectifier circuit.
  • the first end C of the secondary winding is connected to the first input end of the full bridge rectifier circuit
  • the second end E of the secondary winding is connected to the second input end of the full bridge rectifier circuit.
  • the positive pole of the output terminal of the full-bridge rectifier circuit is point D
  • the negative pole of the output terminal of the full-bridge rectifier circuit is point G, that is, the ground.
  • Points D and G can be connected to loads, that is, electrical equipment.
  • the DC/DC converters of the above topologies due to the existence of switching tubes, resonant inductances, resonant capacitors and transformer leakage inductances, are the sources of common mode noise.
  • the transformer transfers the common mode noise from the original in the process of transferring energy.
  • the side winding is passed to the secondary side winding as the transmission path of the disturbance.
  • the following is a detailed introduction with reference to the accompanying drawings. It should be noted that the technical solutions provided in all the embodiments of this application are applicable to the 2- Figure 5 DC/DC converter in any topology.
  • Embodiment 1 of the DC/DC converter is a DC/DC converter
  • FIG. 6 this figure is a schematic diagram of a DC/DC converter according to an embodiment of the present application.
  • the DC/DC converter is a full-bridge LLC resonant DC/DC converter
  • the rectifier circuit connected to the secondary winding is a full-wave rectifier circuit as an example for description.
  • the DC/DC converter provided in this embodiment includes: a primary winding, a secondary winding, a resonant inductor Lr, a resonant capacitor Cr, and a noise suppression network;
  • the noise suppression network can include a signal source V and an impedance network Z as an example.
  • the signal source V is used to generate voltage and current signals
  • the impedance network Z is used to match the suppression current output by the noise suppression network. .
  • the primary winding, the resonant inductor Lr and the resonant capacitor Cr are connected in series between the first node V1N and the second node V2N;
  • the secondary winding is connected between the third node D and the fourth node E;
  • the primary winding and the secondary winding form a transformer
  • the signal source V is connected between the primary side dead point and the secondary side dead point of the transformer; the primary side dead point is the DC stable potential point of the input end of the DC/DC converter, and the secondary side dead point is the DC
  • the DC stable potential point at the output end of the DC/DC converter for example, the dead point of the primary side is the positive pole M or the negative pole N of the input end of the DC/DC converter, and the dead point of the secondary side is the positive pole D or the ground G of the output end of the DC/DC converter;
  • the first parasitic capacitance of the first end of the primary winding relative to the secondary dead point is equal to the second parasitic capacitance of the second end of the primary winding relative to the secondary dead point;
  • the noise suppression network is used to generate a suppression current, the direction of the suppression current is opposite to the direction of the total noise current generated by the resonant inductor and the resonant capacitor, and used to suppress the resonant inductor and the resonant capacitor from generating The total noise current, the voltage of the noise suppression network varies with the size of the load.
  • the signal source V in FIG. 6 is introduced by being connected to the dead point N of the primary side and the dead point D of the secondary side as an example.
  • the signal source V can also be connected between the dead point M of the primary side and the dead point G of the secondary side.
  • the parasitic capacitance is equivalent to the point A of the first end of the primary winding shown in FIG. 6 relative to the secondary winding.
  • the first parasitic capacitance C AD of the side dead point D and the second end B point of the primary winding are relative to the second parasitic capacitance C BD of the secondary side dead point D, that is, the parasitic capacitance is equivalent to the point A of the primary winding of the transformer , between point B and point D of the secondary winding, in practice, the parasitic capacitance may be evenly distributed between the primary winding and the secondary winding.
  • the first parasitic capacitance C AD ⁇ the second parasitic capacitance C BD , so a common mode noise channel is formed between the primary winding and the secondary winding of the transformer.
  • C AD C BD
  • the technical solution adds a noise suppression network on the basis of including a balance capacitor in the DC/DC converter.
  • the noise suppression network can generate a suppression current, and the direction of the suppression current is opposite to the direction of the total noise current generated by the resonant inductor and the resonant capacitor. Therefore, It can suppress the total noise current generated by the resonant inductor and the resonant capacitor.
  • the current source of the noise suppression network can be additionally increased, or it can be derived from the energy inside the DC/DC converter.
  • the noise suppression network is connected to the primary side of the transformer. and the secondary side dead point, so that the common mode noise formed by the resonant circuit on the parasitic capacitance between the primary winding and the secondary winding of the transformer can be effectively reduced.
  • the voltage of the noise suppression network varies with the size of the load, so it can be applied to common mode noise suppression in the entire load range.
  • the voltage of the noise suppression network provided in this embodiment varies with the size of the load, so common-mode noise suppression within the full load range can be achieved, so that regardless of the size of the load, the noise suppression network can automatically adjust its own output according to the size of the load , thus canceling out common-mode noise over the full load range.
  • the following describes how the signal source couples energy from the inside of the DC/DC converter.
  • the energy can be coupled from the resonant inductor or from the resonant capacitor.
  • the following describes in detail with reference to the accompanying drawings.
  • the signal source coupling energy from the resonant inductor is taken as an example for introduction.
  • FIG. 7 is a schematic diagram of a DC/DC converter in which a signal source couples energy from a resonant inductance according to an embodiment of the present application.
  • the signal source provided in this embodiment includes an impedance network Z and a cancellation inductance; as shown in the figure, the inductance coupled to the cancellation inductance is V.
  • the cancellation inductance may be a winding, and the cancellation inductance couples energy from the resonant inductance Lr.
  • it can be wound on the same magnetic core as Lr, that is, the same magnetic core with Lr.
  • the cancellation inductance is connected in series with the impedance network Z between the primary side dead point N and the secondary side dead point D.
  • the signal source is connected between N and D.
  • the signal source can also be connected between the dead point M on the primary side and the dead point G on the secondary side.
  • the voltage amplitude V of the cancellation inductance is proportional to the voltage amplitude of the resonant inductance Lr.
  • the impedance network Z is to adjust the magnitude of the voltage amplitude injected by the signal source between the dead point of the primary side and the dead point of the secondary side.
  • the impedance network Z is a two-port network, which can be connected in series with the cancelling inductance.
  • the impedance network Z may include at least any one of the following: resistance, inductance and capacitance. That is, it may be at least one, two, or three of the above three.
  • Z may include inductance, inductance and resistance, capacitance and resistance, inductance and capacitance, or both inductance, resistance and capacitance.
  • the voltage V coupled from Lr to the offset inductance is opposite to the voltage across Lr.
  • the same name terminal of the offset inductor is connected to the primary static point M, and the opposite terminal of the offset inductor is connected to the secondary side static point through the impedance network Z.
  • Point G that is, the same-named end of the canceling inductance is close to the dead point of the primary side, and the synonymous end of the canceling inductance is close to the dead point of the secondary side.
  • the LISN in Figure 7 is grounded, indicating that the Line Impedance Stabilization Network (LISN, Line Impedance Stabilization Network) is grounded in the Electromagnetic Compatibility (EMC, Electromagnetic Compatibility) test.
  • EMC Electromagnetic Compatibility
  • this figure is a schematic diagram of a small signal model corresponding to FIG. 7 and FIG. 8 .
  • the following analysis is performed in conjunction with the small signal model shown in FIG. 9A .
  • the small signal model shown in FIG. 9A short-circuits V1N and V2N together, and does not affect the calculation of V.
  • the parasitic capacitance refers to the sum of the primary winding and the secondary winding
  • V C /Z1+V L /Z2+V/Z 0;
  • VC represents the voltage across the resonant capacitor
  • VL represents the inductance across the resonant inductor
  • Z1 represents the impedance presented by CAD
  • Z2 represents the impedance presented by CBD.
  • n1 represents the turns ratio of the cancellation inductance and the resonant inductance Lr.
  • the DC/DC converter does not necessarily work completely at the resonant frequency point, but only approximately at the resonant frequency point, for example, VL +V Lk -VC is approximately equal to 0.
  • n VLk / VL .
  • the magnitude of the impedance presented by the impedance network can be obtained.
  • the signal source couples energy from the resonant inductance, and no additional signal source is required, which is simple and easy to implement.
  • the energy can be coupled from the resonant inductance.
  • the amplitude of the injected signal can be adjusted through the impedance network Z in series with the cancellation inductance. Since the current through Cr and Lr will change when the load changes, as the current changes, the amplitude of the voltage coupled by the cancellation inductance will change. The value also follows, so this scheme is suitable for noise suppression over the full load range.
  • energy is coupled from the resonant inductance Lr through an offset inductance.
  • the energy can also be coupled from the resonant inductance through a capacitive coupling circuit.
  • FIG. 9B is a schematic diagram of coupling energy from a resonant inductance by a capacitive coupling capacitor provided by an embodiment of the present application.
  • the signal source may include: a first capacitive coupling circuit Z2;
  • the first capacitive coupling circuit Z2 couples energy from the resonant inductor Lr, and the first capacitive coupling circuit Z2 is connected in series with the impedance network Z at the primary side dead point N and the secondary side dead point D. between.
  • Z2 and Z can also be coupled between other dead points on the primary side and other dead points on the secondary side after being connected in series, and the connection method is not limited to that shown in FIG. 9B .
  • FIG. 10 is a schematic diagram of a DC/DC converter in which a signal source couples energy from a resonant capacitor according to an embodiment of the present application.
  • This embodiment introduces that a signal source couples energy from the resonant capacitor Cr; the voltage amplitude V of the signal source is proportional to the voltage amplitude of the resonant capacitor Cr.
  • the signal source may include: a second capacitive coupling circuit Z3;
  • Z3 of the second capacitive coupling circuit couples energy from the resonant capacitor Cr.
  • the two ends of the second capacitive coupling circuit Z3 are connected to the two ends of Cr respectively, and Z3 and the impedance network Z are connected in series to the primary static point and the secondary side In between.
  • FIG. 11A is a schematic diagram of another DC/DC converter in which a signal source couples energy from a resonant capacitor according to an embodiment of the present application.
  • the parasitic capacitance refers to the sum of the primary winding and the secondary winding
  • V C /Z1+V L /Z2+V/Z 0;
  • VC represents the voltage across the resonant capacitor
  • VL represents the inductance across the resonant inductor
  • Z1 represents the impedance presented by CAD
  • Z2 represents the impedance presented by CBD.
  • n2 represents the ratio of the coupling voltage of the cancellation impedance to the voltage across the Cr.
  • V L +V Lk -V C 0.
  • n VLk / VL .
  • the magnitude of the presented impedance of the impedance network can be obtained from the above formula.
  • the implementation method described above is to couple the energy from the resonant capacitor through the capacitive coupling capacitor.
  • the energy can also be coupled from the resonant capacitor through the transformer. Please refer to FIG. 11B for details. Schematic diagram of coupled energy.
  • the signal source includes: a signal coupling transformer T;
  • the primary winding of the signal coupling transformer T couples energy from the resonant capacitor Cr, and the secondary winding of the signal coupling transformer T is connected in series with the impedance network Z and then connected to the primary dead point and the secondary side. In between.
  • the secondary winding of T is connected in series with the impedance network Z and then connected to the primary dead point M and the secondary dead point D as an example. It can also be connected to other dead points, which will not be repeated here.
  • the signal source couples energy from the resonant capacitor, and no additional signal source is required, which is simple and easy to implement.
  • the energy can be coupled from the resonant capacitor.
  • the impedance network Z in series with the canceling impedance can adjust the amplitude of the injected signal. Since the current through Cr and Lr will change when the load changes, the amplitude of the voltage coupled by the canceling impedance will also change with the change of the current. Following the change, therefore, the scheme is suitable for noise suppression in the full load range.
  • the parasitic capacitances in Fig. 12 and Fig. 13 include two, respectively C AC and C BC , that is, the parasitic capacitance C AC is between the first end of the primary winding and the first end of the secondary winding, and the parasitic capacitance C BC is between the first end of the primary winding and the first end of the secondary winding. between the second end of the side winding and the first end of the secondary side winding.
  • the transformer is symmetrical if the first parasitic capacitance of the first end of the primary winding relative to the dead point of the secondary winding is equal to the second end of the primary winding
  • the embodiment of the present application adds a noise suppression network on the basis of adding a balance capacitor, the noise suppression network can generate a suppression current, and the direction of the suppression current is related to the total amount generated by the resonant inductor and the resonant capacitor.
  • the direction of the noise current is opposite, so the total noise current generated by the resonant inductor and the resonant capacitor can be suppressed.
  • the current source of the noise suppression network can be additionally increased, or it can be derived from the energy inside the DC/DC converter.
  • the noise suppression network is connected in Between the primary side dead point and the secondary side dead point of the transformer, the common mode noise formed by the resonant circuit on the parasitic capacitance between the primary side winding and the secondary side winding of the transformer can be effectively reduced.
  • the voltage of the noise suppression network changes with the change of the load size, so it can be applied to the common mode noise suppression in the whole load range.
  • the embodiment of the present application further provides a communication power supply, which will be described in detail below with reference to the accompanying drawings.
  • FIG. 14 is a schematic diagram of a communication power supply provided by an embodiment of the present application.
  • the communication power supply provided in this embodiment includes the DC/DC converter 200 described in any one of the above embodiments, and further includes: a rectifier circuit 300;
  • the first end of the rectifier circuit 300 is used for connecting the alternating current power source AC, and the rectifying circuit 300 is used for converting the alternating current of the alternating current power source AC into direct current;
  • the second end of the rectifier circuit 300 is used to connect to the first end of the DC/DC converter 200; the second end of the DC/DC converter 200 is used to supply power to the load;
  • the DC/DC converter 200 is configured to convert the direct current and then provide it to the load.
  • the load can be any electrical equipment, and the voltage output by the DC/DC converter 200 can be different according to the load, for example, it can output voltages such as 48V, 12V, 5V, and 3.3V.
  • the noise suppression network can generate a suppression current, and the direction of the suppression current is opposite to the current direction of the resonant inductor and the resonant capacitor, so it can The total noise current generated by the resonant inductor and the resonant capacitor is suppressed.
  • the current source of the noise suppression network can be additionally increased, or it can be derived from the energy inside the DC/DC converter.
  • the noise suppression network is connected to the primary side of the transformer. Between the secondary side static points, the common mode noise formed by the resonant circuit on the parasitic capacitance between the primary winding and the secondary winding of the transformer can be effectively reduced.
  • the voltage of the noise suppression network varies with the size of the load, so it can be applied to common mode noise suppression in the entire load range. Therefore, the communication power supply having the DC/DC converter can achieve a better common mode noise suppression effect, thereby providing a higher quality power supply to the load.
  • At least one (item) refers to one or more, and "a plurality” refers to two or more.
  • At least one item(s) below” or similar expressions thereof refer to any combination of these items, including any combination of single item(s) or plural items(s).
  • at least one (a) of a, b or c can mean: a, b, c, "a and b", “a and c", “b and c", or "a and b and c” ", where a, b, c can be single or multiple.

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Abstract

本申请公开了一种DC/DC变换器及通信电源，变换器包括：原边绕组、副边绕组、谐振电感、谐振电容和噪声抑制网络；原边绕组和副边绕组形成变压器；噪声抑制网络连接在原边静点和副边静点之间；原边静点为DC/DC变换器输入端的直流稳定电位点，副边静点为DC/DC变换器输出端的直流稳定电位点；原边绕组的第一端相对于副边静点的第一寄生电容等于原边绕组的第二端相对副边静点的第二寄生电容；噪声抑制网络产生抑制电流，方向与谐振电感和谐振电容产生的总噪声电流的方向相反，抑制总噪声电流，噪声抑制网络的电压随负载大小的变化而变化，实现全负载范围的共模噪声抑制，有效降低在原边绕组和副边绕组之间的形成的共模噪声。

Description

一种DC/DC变换器及通信电源
本申请要求于2020年09月07日提交中国国家知识产权局的申请号为202010930447.4、申请名称为“一种DC/DC变换器及通信电源”的中国专利申请的优先权，其全部内容通过引用结合在本申请中。
技术领域
本申请涉及电力电子技术领域，尤其涉及一种DC/DC变换器及通信电源。
背景技术
目前，开关电源变换器，由于其效率高、体积小等优点，广泛应用于通信电源。随着用电负载容量的不断扩大，对电源模块的功率密度要求越来越高，但是目前开关电源变换器中磁性器件的体积占比比较大。
为了降低磁性器件所占的体积，可以提高其工作频率，例如将开关电源的工作频率从200kHz提升到1M～10MHz。但是提高开关频率增加了开关噪声(裸噪增大20dB+)，给前级滤波电路带来很大压力。因此如何降低开关电源的噪声，降低前级滤波电路的成本，成为目前研究的重要方向。
目前，开关电源面临的主要问题是共模干扰，可以使用共模电感来滤波。参见图1，该图为现有技术提供的抑制共模干扰的示意图。
图1中利用共模电感100对交流电源输出的共模电流进行抑制。
共模电感100对共模电流呈现高阻特性，可以对共模噪声进行衰减，因此可以达到较好的共模滤波效果。
但是，随着开关电源的功率增大，开关电源输入侧的电流也越来越大，这就要求共模电感具有足够大的通流能力。另外，还要增大共模电感的磁芯来防止磁芯饱和，磁芯体积随之增大，最终导致共模电感的体积非常大，不利于滤波电路的小型化，限制了开关电源的小型化。
发明内容
为了解决以上的技术问题，本申请提供一种DC/DC变换器及通信电源，能够抑制共模噪声，而且利于开关电源的小型化。
本申请实施例提供一种DC/DC变换器，该DC/DC变换器为一种LLC谐振DC/DC变换器，可以为全桥LLC谐振DC/DC变换器，也可以为半桥LLC谐振DC/DC变换器，工作原理是将输入的直流电转换为交流电，经过变压器传递给输出端的整流电路，整流电路整流为直流电再输出，由于经过了变压器隔离，因此，可以起到干扰信号隔离的作用，变压器的原边绕组的干扰信号不会传递到变压器的副边绕组。
DC/DC变换器包括：原边绕组、副边绕组、谐振电感、谐振电容和噪声抑制网络；所述原边绕组、所述谐振电感和所述谐振电容串联后连接在第一节点和第二节点；所述副边绕组连接在第三节点和第四节点；所述原边绕组和所述副边绕组形成变压器；所述噪声抑制网络连接在所述变压器的原边静点和副边静点之间；所述原边静点为所述DC/DC变换器输入端的直流稳定电位点，所述副边静点为所述DC/DC变换器输出端的直流稳定电位点；所述原边绕组的第一端相对于所述副边静点的第一寄生电容等 于所述原边绕组的第二端相对于所述副边静点的第二寄生电容；所述噪声抑制网络，用于产生抑制电流，所述谐振电感和所述谐振电容产生的总噪声电流的方向与所述抑制电流的方向相反，抑制电流用于抑制总噪声电流，所述噪声抑制网络的电压随负载大小的变化而变化。
由于实际设计时，难以做到完全对称，因此第一寄生电容C AD≠第二寄生电容C BD，因此变压器的原边绕组和副边绕组之间，形成共模噪声的通道。但是，可以通过增加平衡电容，使C AD＝C BD，在变压器对称的基础上，做到共模噪声源信号的平衡，即可实现共模噪声抵消，同时变压器漏感产生的共模噪声在对称的变压器上也可以实现抵消。平衡电容可以在A点和D点之间添加，也可以在B点和D点之间添加，具体需要分析C AD和C BD的大小关系，添加平衡电容后，使C AD＝C BD即可。静点是指即相对于GND，静点的直流电位是稳定的，静点在变压器的原边存在，也在变压器的副边存在。顾名思义，相对于静点而言，动点的电位是波动的。
该技术方案在DC/DC变换器中包括平衡电容的基础上，增加噪声抑制网络，该噪声抑制网络可以产生抑制电流，抑制电流的方向与谐振电感和谐振电容产生的总噪声电流方向相反，因此可以抑制谐振电感和谐振电容产生的总噪声电流，该噪声抑制网络的电流来源可以是额外增加的，也可以来源于DC/DC变换器内部的能量，噪声抑制网络连接在变压器的原边静点和副边静点之间，从而可以有效降低谐振电路在变压器原边绕组和副边绕组之间的寄生电容上形成的共模噪声。并且，该噪声抑制网络的电压随着负载大小的变化而变化，因此，可以适用于全负载范围的共模噪声抑制。由于负载变化时，电流会发生变化，由于电感(谐振电感Lr和漏感Lk)上的电压幅值与电流成比例关系，因此随着负载变化，Lk和Lr以及谐振电容Cr上的电压幅值均会发生变化。本实施例提供的噪声抑制网络的电压随负载大小的变化而变化，因此可以实现全负载范围内的共模噪声抑制，从而不论负载的大小，噪声抑制网络可以根据负载的大小自动调节自身的输出，因此可以抵消全负载范围的共模噪声。
优选地，所述噪声抑制网络包括：信号源和阻抗网络；阻抗网络是为了调节信号源注入原边静点和副边静点之间的电压幅值的大小。所述信号源与所述阻抗网络串联后连接在所述原边静点和所述副边静点之间。
其中，信号源可以有多种实现方式，可以额外增加信号源，也可以直接从变换器中耦合能量形成信号源，例如从谐振电感上耦合能量，也可以从谐振电容上耦合能量。
例如，所述信号源从所述谐振电感上耦合能量；所述信号源的电压幅值与所述谐振电感的电压幅值成正比。即，谐振电感上的电压增大时，信号源的电压也增大。
具体地，从谐振电感上耦合能量，可以通过电感通过磁场来从谐振电感上耦合能量，即所述信号源包括抵消电感；所述抵消电感，用于从所述谐振电感上耦合能量，所述抵消电感与所述阻抗网络串联后连接在所述原边静点和所述副边静点之间。
具体实现时，为了绕制方便，节省磁芯，所述抵消电感与所述谐振电感共磁芯绕制。
从谐振电感上耦合能量，除了利用以上的抵消电感以外，还可以通过电容从谐振 电感上耦合能量，即，所述信号源包括：第一电容耦合电路；所述第一电容耦合电路，用于从所述谐振电感耦合能量，所述第一电容耦合电路与所述阻抗网络串联后连接在所述原边静点和所述副边静点之间。
所述信号源从所述谐振电容上耦合能量；所述信号源的电压幅值与所述谐振电容的电压幅值成正比。即，谐振电容上的电压增大时，信号源的电压也增大。
以上介绍的是信号源通过从谐振电感上耦合能量，下面介绍信号源从谐振电容上耦合能量。从谐振电容上耦合能量，可以通过多种方式来实现，例如，可以通过电容耦合电路从谐振电容上耦合能量，也可以通过耦合变压器从谐振电容上耦合能量，下面分别介绍。
即，所述信号源包括：第二电容耦合电路；所述第二电容耦合电容，用于从所述谐振电容耦合能量，所述第二电容耦合电路和所述阻抗网络串联后连接所述原边静点和所述副边静点之间。
即，所述信号源包括：信号耦合变压器；所述信号耦合变压器的原边绕组，用于从所述谐振电容耦合能量，所述信号耦合变压器的副边绕组与所述阻抗网络串联后连接在所述原边静点和所述副边静点之间。
为了实现更好的降噪效果，所述谐振电感和所述谐振电容在寄生电容上产生的总噪声电流与所述抵消阻抗的电流之和为零；所述寄生电容为所述原边绕组和所述副边绕组之间的寄生电容。
即表达式如下：
V C/Z1+V L/Z2+V/Z＝0；
其中，V C表示谐振电容两端的电压，V L表示谐振电感两端的电感，Z1表示C AD呈现的阻抗，Z2表示C BD呈现的阻抗。
优选地，所述DC/DC变换器包括以下至少一种：全桥LLC谐振DC/DC变换器和半桥LLC谐振DC/DC变换器。
优选地，还包括：全波整流电路；所述副边绕组包括：第一副边绕组和第二副边绕组；所述第一副边绕组连接所述第三节点和第五节点之间，所述第二副边绕组连接所述第五节点和所述第四节点之间；所述第三节点和所述第四节点连接所述全波整流电路；所述全波整流电路的输出端为所述DC/DC变换器的输出端。
信号源从谐振电感上耦合能量，不需要额外增加信号源，简单易于实现，通过在谐振电感的磁芯上绕制一个抵消电感，从谐振电感上耦合能量即可，通过与抵消电感串联的阻抗网络Z可以调节注入的信号的幅值大小，由于负载变化时，通过Cr和Lr的电流均会变化，因此，随着电流的变化，抵消电感耦合的电压的幅值也跟随变化，因此，该方案适用于全负载范围内的噪声抑制。
优选地，还包括：全桥整流电路；所述副边绕组为一个；所述第三节点和所述第四节点分别为所述全桥整流电路的输入端的正极和负极；所述全桥整流电路的输出端为所述DC/DC变换器的输出端。
另外，所述阻抗网络包括以下至少一项：电阻、电容和电感，即包括以上三项中 至少一个或多个，例如阻抗网络包括以上的一项、两项、三项均可以。
本申请实施例还提供一种通信电源，包括以上介绍的DC/DC变换器，还包括：整流电路；所述整流电路的第一端用于连接交流电源；所述整流电路，用于将所述交流电源的交流电转换为直流电；所述整流电路的第二端用于连接所述DC/DC变换器的第一端；所述DC/DC变换器的第二端用于为负载供电；所述DC/DC变换器，用于将所述直流电进行转换后提供给所述负载。
负载可以为任何用电设备，具体可以根据负载的不同，DC/DC变换器200输出的电压不同，例如可以输出48V、12V、5V、3.3V等电压。
本申请实施例提供的通信电源，由于其中的DC/DC变换器中增加了噪声抑制网络，该噪声抑制网络可以产生抑制电流，抑制电流的方向与谐振电感和谐振电容的电流方向相反，因此可以抑制谐振电感和谐振电容产生的总噪声电流，该噪声抑制网络的电流来源可以是额外增加的，也可以来源于DC/DC变换器内部的能量，噪声抑制网络连接在变压器的原边静点和副边静点之间，从而可以有效降低谐振电路在变压器原边绕组和副边绕组之间的寄生电容上形成的共模噪声。并且，该噪声抑制网络的电压随着负载大小的变化而变化，因此，可以适用于全负载范围的共模噪声抑制。因此，拥有该DC/DC变换器的通信电源可以实现较好的共模噪声抑制的效果，从而给负载提供更高质量的电源。
与现有技术相比，本申请实施例提供的技术方案具有以下优点：
该技术方案提供的DC/DC变换器为LLC谐振DC/DC变换器，包括变压器和谐振电路，谐振电路与变压器的原边绕组串联，谐振电路包括串联的谐振电感和谐振电容，由于变压器存在漏感，因此，负载大小不同时，流过漏感的电流大小也不同，进而电流大小不同对于漏感的影响不同，当流过漏感的电感越大时，漏感作为噪声源的影响越大，因此不可忽略。因此，为了在全负载范围内均抑制共模噪声的影响，在变压器的原边绕组的第一端相对于副边静点的第一寄生电容等于原边绕组的第二端相对于副边静点的第二寄生电容的前提下，在DC/DC变换器中增加噪声抑制网络，该噪声抑制网络可以产生抑制电流，抑制电流的方向与谐振电感和谐振电容产生的总噪声电流的方向相反，因此可以抑制谐振电感和谐振电容产生的总噪声电流，该噪声抑制网络的电流来源可以是额外增加的，也可以来源于DC/DC变换器内部的能量，噪声抑制网络连接在变压器的原边静点和副边静点之间，从而可以有效降低谐振电路在变压器原边绕组和副边绕组之间的寄生电容上形成的共模噪声。并且，该噪声抑制网络的电压随着负载大小的变化而变化，因此，可以适用于全负载范围的共模噪声抑制。
附图说明
图1为现有技术提供的抑制共模干扰的示意图；
图2为本申请实施例提供的一种全桥LLC谐振DC/DC变换器的示意图；
图3为本申请实施例提供的一种半桥LLC谐振DC/DC变换器的示意图；
图4为本申请实施例提供的另一种半桥LLC谐振DC/DC变换器的示意图；
图5为本申请实施例提供的另一种全桥LLC谐振DC/DC变换器的示意图；
图6为本申请实施例提供的一种DC/DC变换器的示意图；
图7为本申请实施例提供的一种信号源从谐振电感耦合能量的DC/DC变换器的示意图；
图8为本申请实施例提供的另一种信号源从谐振电感耦合能量的DC/DC变换器的示意图；
图9A为本申请实施例提供的与图7和图8对应的小信号模型示意图；
图9B为本申请实施例提供的电容耦合电路从谐振电感耦合能量的示意图；
图10为本申请实施例提供的一种信号源从谐振电容耦合能量的DC/DC变换器的示意图；
图11A为本申请实施例提供的另一种信号源从谐振电容耦合能量的DC/DC变换器的示意图；
图11B为本申请实施例提供的信号耦合变压器从谐振电容耦合能量的示意图；
图12为本申请实施例提供的全桥LLC谐振DC/DC变换器的等效示意图；
图13为本申请实施例提供的半桥LLC谐振DC/DC变换器的等效示意图；
图14为本申请实施例提供的一种通信电源的示意图。
具体实施方式
为了使本领域技术人员更好地理解本申请实施例提供的技术方案，下面先介绍DC/DC变换器的工作原理。
本申请实施例中提供的DC/DC变换器为隔离式开关电源，以LLC谐振DC/DC变换器为例进行介绍，按照变压器原边连接的电路拓扑来区分，可以为全桥LLC谐振DC/DC变换器，也可以为半桥LLC谐振DC/DC变换器。变压器的副边绕组连接的整流电路形式不限定。
如图2所示，为本申请实施例提供的一种全桥LLC谐振DC/DC变换器的示意图。
LLC谐振DC/DC变换器的工作原理是将输入的直流电转换为交流电，经过变压器传递给输出端的整流电路，整流电路整流为直流电再输出，由于经过了变压器隔离，因此，可以起到干扰信号隔离的作用，变压器的原边绕组的干扰信号不会传递到变压器的副边绕组。
图2中的变压器的原边绕组包括励磁绕组Lm，另外将变压器的原边绕组和副边绕组的漏感Lk也画在变压器的原边绕组的位置。
谐振电感Lr和谐振电容Cr与变压器的原边绕组相串联，形成LLC谐振。
图2中以变压器的副边绕组连接全波整流电路为例进行说明。
变压器的原边绕组、Lr和Cr串联后连接第一节点和第二节点，如图所示，第一节点为全桥第一桥臂的中点V1N，第二节点为全桥第二桥臂的中点V2N。
变压器的副边绕组连接在第三节点C和第四节点E之间，对于DC/DC变换器输出端的整流电路为全波整流电路时，变压器的副边绕组还包括中心抽头，即D点为副边绕组的中心抽头的引出端。图2中D点和G点分别为DC/DC变换器的输出端的正极和负极，负极接地。
变压器的原边静点为DC/DC变换器的输入端的正极(M点)或负极(N)，变压 器的副边静点为DC/DC变换器的输出端的正极(即D点)或地；
本申请实施例中的静点是指对于交流小信号为静点，即对于地电位GND，静点的电位是直流稳定电位点，即相对于GND，静点的直流电位是稳定的。顾名思义，相对于静点而言，动点的电位是波动的。例如对于变压器来说，C点和E点的电位是波动的，属于交流电位，因此C点和E点是变压器的副边动点，而A点和B点是变压器的原边动点。
以上介绍的是全桥LLC谐振DC/DC变换器的结构，下面介绍两种半桥LLC谐振DC/DC变换器的结构。
如图3所示，为本申请实施例提供的一种半桥LLC谐振DC/DC变换器的示意图。
图3所示的半桥LLC谐振DC/DC变换器，变压器的原边连接的变换电路只有一个桥臂，变压器的原边绕组的一端连接桥臂的中点V1N，变压器的原边绕组的另一端连接电容C1和C2的公共端V2N。
其他的连接关系与图2中的相同，在此不再赘述。
如图4所示，为本申请实施例提供的另一种半桥LLC谐振DC/DC变换器的示意图。
图4所示的半桥LLC谐振DC/DC变换器与图3的区别是，原边绕组的两端连接的均是同一个桥臂，原边绕组的一端连接桥臂中点V1N，原边绕组的另一端连接桥臂的另一个输出端V2N。
图3和图4分别为原边绕组连接的均是半桥LLC谐振DC/DC变换器的两种拓扑形式，图4和图3的区别仅是V2N连接的位置有所区别。
如图5所示，为本申请实施例提供的另一种全桥LLC谐振DC/DC变换器的示意图。
图2-图4中所示的副边绕组连接的整流电路均为全波整流电路，图5所示的整流电路为全桥整流电路。
副边绕组的第一端C点连接全桥整流电路的第一输入端，副边绕组的第二端E点连接全桥整流电路的第二输入端。全桥整流电路的输出端的正极为D点，全桥整流电路的输出端的负极为G点，即地。
D点和G点可以连接负载，即用电设备。
以上几种拓扑的DC/DC变换器，由于存在开关管、谐振电感、谐振电容和变压器漏感，这些均是产生共模噪声的来源，变压器在传递能量的过程中将该共模噪声从原边绕组传递到副边绕组，作为干扰的传递路径。
因此，本申请实施例提供的技术方案是在变压器的原边绕组的第一端相对于副边静点之间的第一寄生电容C AD与原边绕组的第二端相对于副边绕组的静点之间的第二寄生电容C BD相等的前提下，即C AD＝C BD时，变压器的原边绕组的两个动点A和B相分别对于副边绕组的静点D点对称，再通过增加信号源，来实现全负载范围内抵消DC/DC变换器中的共模干扰，下面结合附图进行详细介绍，需要说明的是，本申请所有实施例提供的技术方案均适用于图2-图5任意一个拓扑形式的DC/DC变换器。
DC/DC变换器实施例一：
参见图6，该图为本申请实施例提供的一种DC/DC变换器的示意图。
本实施例中，以DC/DC变换器为全桥LLC谐振DC/DC变换器，且副边绕组连接的整流电路为全波整流电路为例进行介绍。
本实施例提供的DC/DC变换器，包括：原边绕组、副边绕组、谐振电感Lr、谐振电容Cr、噪声抑制网络；
为了便于分析其工作原理，以噪声抑制网络可以包括信号源V和阻抗网络Z为例进行介绍；其中信号源V用来产生电压和电流信号，阻抗网络Z用来匹配噪声抑制网络输出的抑制电流。
原边绕组、谐振电感Lr和谐振电容Cr串联后连接在第一节点V1N和第二节点V2N之间；
副边绕组连接在第三节点D和第四节点E之间；
原边绕组和副边绕组形成变压器；
信号源V连接在变压器的原边静点和副边静点之间；所述原边静点为所述DC/DC变换器输入端的直流稳定电位点，所述副边静点为所述DC/DC变换器输出端的直流稳定电位点，例如原边静点为DC/DC变换器的输入端正极M或负极N，副边静点为DC/DC变换器的输出端的正极D或地G；
所述原边绕组的第一端相对于所述副边静点的第一寄生电容等于所述原边绕组的第二端相对于所述副边静点的第二寄生电容；
所述噪声抑制网络，用于产生抑制电流，所述抑制电流的方向与所述谐振电感和所述谐振电容产生的总噪声电流的方向相反，用于抑制所述谐振电感和所述谐振电容产生的总噪声电流，所述噪声抑制网络的电压随负载大小的变化而变化。
图6中的信号源V是以连接在原边静点N和副边静点D为例进行的介绍。另外，信号源V也可以连接在原边静点M和副边静点G之间。
实际应用中，变压器的原边绕组和副边绕组之间存在寄生电容，为了方便介绍将寄生电容等效为图6中所示的所述原边绕组的第一端A点相对于所述副边静点D的第一寄生电容C AD和原边绕组的第二端B点相对于副边静点D的第二寄生电容C BD，即将寄生电容等效在了变压器的原边绕组A点、B点与副边绕组的D点之间，实际中，可能寄生电容平均分布在原边绕组和副边绕组之间。
由于实际设计时，难以做到完全对称，因此第一寄生电容C AD≠第二寄生电容C BD，因此变压器的原边绕组和副边绕组之间，形成共模噪声的通道。但是，可以通过增加平衡电容，使C AD＝C BD，在变压器对称的基础上，做到共模噪声源信号的平衡，即可实现共模噪声抵消，同时变压器漏感产生的共模噪声在对称的变压器上也可以实现抵消。平衡电容可以在A点和D点之间添加，也可以在B点和D点之间添加，具体需要分析C AD和C BD的大小关系，添加平衡电容后，使C AD＝C BD即可。
LLC谐振DC/DC变换器工作在谐振状态时，Lk+Lr的总噪声与Cr的总噪声幅值相等，相位相反，VLr+VCr+VLk＝0；实际上，Lk均匀分布在变压器中，且Lk不可忽 略，因此，Lr与Cr的噪声幅值不等，不能实现抵消，VLr+VCr≠0，并且，Lk上的电流分量幅值随着负载的大小不同而变化，不是一个固定值，即负载大小会影响Lk上的电流分量，从而影响VLr+VCr的幅值。当流过Lk的电流越大时，Lk在(Lk+Lr)中占比越大，Lk作为噪声源的影响越不可以被忽略。
该技术方案在DC/DC变换器中包括平衡电容的基础上，增加噪声抑制网络，该噪声抑制网络可以产生抑制电流，抑制电流的方向与谐振电感和谐振电容产生的总噪声电流方向相反，因此可以抑制谐振电感和谐振电容产生的总噪声电流，该噪声抑制网络的电流来源可以是额外增加的，也可以来源于DC/DC变换器内部的能量，噪声抑制网络连接在变压器的原边静点和副边静点之间，从而可以有效降低谐振电路在变压器原边绕组和副边绕组之间的寄生电容上形成的共模噪声。并且，该噪声抑制网络的电压随着负载大小的变化而变化，因此，可以适用于全负载范围的共模噪声抑制。
由于负载变化时，电流会发生变化，由于电感(Lr和Lk)上的电压幅值与电流成比例关系，因此随着负载变化，Lk和Lr以及Cr上的电压幅值均会发生变化。本实施例提供的噪声抑制网络的电压随负载大小的变化而变化，因此可以实现全负载范围内的共模噪声抑制，从而不论负载的大小，噪声抑制网络可以根据负载的大小自动调节自身的输出，因此可以抵消全负载范围的共模噪声。
下面介绍信号源从DC/DC变换器内部耦合能量的实现方式，可以从谐振电感上耦合能量，也可以从谐振电容上耦合能量，下面分别结合附图进行详细介绍。
DC/DC变换器实施例二：
本实施例中以信号源从谐振电感上耦合能量为例进行介绍。
参见图7，该图为本申请实施例提供的信号源从谐振电感耦合能量的DC/DC变换器的示意图。
本实施例提供的信号源包括阻抗网络Z和抵消电感；如图所示，抵消电感上耦合的电感为V。
其中抵消电感可以是一个绕组，抵消电感从谐振电感Lr上耦合能量，具体实现时，可以与Lr绕制在同一个磁芯上，即与Lr共磁芯。
抵消电感与阻抗网络Z串联后连接在所述原边静点N和所述副边静点D之间。图7中是以信号源连接在N和D之间，另外，还可以如图8所示，信号源也可以连接在原边静点M和副边静点G之间。
其中，所述抵消电感的电压幅值V与所述谐振电感Lr的电压幅值成正比。
阻抗网络Z是为了调节信号源注入原边静点和副边静点之间的电压幅值的大小。
本实施例中不具体限定阻抗网络Z的具体实现方式，阻抗网络Z是一个二端口网络，与抵消电感串联即可。例如阻抗网络Z可以至少包括以下任意一项：电阻、电感和电容。即可以为以上三者中的至少一个、两个，也可以为三个。例如Z可以包括电感，也可以包括电感和电阻，也可以包括电容和电阻，也可包括电感和电容，也可以同时包括电感、电阻和电容。
图7中可以看出，抵消电感从Lr上耦合的电压V与Lr两端的电压反向，抵消电 感的同名端连接原边静点M，抵消电感的异名端通过阻抗网络Z连接副边静点G，即抵消电感的同名端靠近原边静点，抵消电感的异名端靠近副边静点。
图7中的LISN接地，表示电磁兼容(EMC，Electromagnetic Compatibility)测试中的线路阻抗稳定网络(LISN，Line Impedance Stabilization Network)接地。
参见图9A，该图为与图7和图8对应的小信号模型示意图。
下面结合图9A所示的小信号模型进行分析，图9A所示的小信号模型将V1N和V2N短接在一起，不影响V的计算。
第一、由于V1N和V2N两点的信号幅值相等，相位相反，因此当C AD＝C BD时，V1N和V2N产生噪声相互抵消。
第二、由于Lk在变压器的内部均匀分布，且变压器两端的寄生电容平衡，即C AD＝C BD时，则Lk产生的噪声抵消。
第三、为了实现更好的降噪效果，谐振电感和谐振电容在寄生电容上产生的总噪声电流与抵消阻抗的电流之和为零；其中，寄生电容是指原边绕组和副边绕组之间的寄生电容，即表达式如下：
V C/Z1+V L/Z2+V/Z＝0；
其中，V C表示谐振电容两端的电压，V L表示谐振电感两端的电感，Z1表示C AD呈现的阻抗，Z2表示C BD呈现的阻抗。
由于抵消电感与谐振电感Lr反向耦合，因此，V＝-n1V L。
其中，n1表示抵消电感与谐振电感Lr的匝数比。
实际工作中，当DC/DC变换器工作在谐振频率点时，所述谐振电感的电压V L、所述谐振电容的电压V C与所述变压器漏感的电压V Lk满足以下关系，V L+V Lk-V C＝0。
需要说明的是，实际工作时，DC/DC变换器不一定完全工作在谐振频率点，近似工作在谐振频率点即可，例如V L+V Lk-V C约等于0。
由于变压器的漏感Lk与谐振电感Lr串联，因此满足V C＝V L+V Lk＝(1+n)V L。
其中，n＝V Lk/V L。
又由于C AD＝C BD，因此，Z＝[n1/(n+2)]Z1。
因此，利用以上公式，可以获得阻抗网络呈现的阻抗大小。
本实施例提供的技术方案，信号源从谐振电感上耦合能量，不需要额外增加信号源，简单易于实现，通过在谐振电感的磁芯上绕制一个抵消电感，从谐振电感上耦合能量即可，通过与抵消电感串联的阻抗网络Z可以调节注入的信号的幅值大小，由于负载变化时，通过Cr和Lr的电流均会变化，因此，随着电流的变化，抵消电感耦合的电压的幅值也跟随变化，因此，该方案适用于全负载范围内的噪声抑制。
本实施例提供的技术方案相对于没有设置信号源进行抵消降噪时的对照表如下。
表1
Figure PCTCN2021103422-appb-000001
Figure PCTCN2021103422-appb-000002
分析表1的数据可知，利用本申请实施例提供的信号源进行抵消降噪时，滤波电路所占的电路板的面积降低至之前的近三分之一，即降低了2/3的面积，即本方案在抑制共模噪声的同时，降低了电路板的面积，而且效果显著，利于DC/DC变换器的小型化，而且有利于整个通信电源的小型化。
以上实施例是以通过抵消电感从谐振电感Lr耦合能量，另外，还可以通过电容耦合电路来从谐振电感来耦合能量。
参见图9B，该图为本申请实施例提供的电容耦合电容从谐振电感耦合能量的示意图。
本实施例提供的DC/DC变换器，信号源可以包括：第一电容耦合电路Z2；
所述第一电容耦合电路Z2从所述谐振电感Lr耦合能量，所述第一电容耦合电路Z2与所述阻抗网络Z串联后连接在所述原边静点N和所述副边静点D之间。另外，Z2和Z串联后也可以耦合在原边其他的静点和副边其他的静点之间，不局限与图9B所示的连接方式。
以上介绍的是从谐振点电感耦合能量，下面介绍从谐振电容耦合能量。
DC/DC变换器实施例三：
参见图10，该图为本申请实施例提供的一种信号源从谐振电容耦合能量的DC/DC变换器的示意图。
本实施例介绍信号源从所述谐振电容Cr上耦合能量；所述信号源的电压幅值V与所述谐振电容Cr的电压幅值成正比。
当信号源从谐振电容Cr上耦合能量时，信号源可以包括：第二电容耦合电路Z3；
第二电容耦合电路的Z3从谐振电容Cr耦合能量，具体地，第二电容耦合电路Z3的两端分别连接Cr的两端，Z3和阻抗网络Z串联后连接原边静点和所述副边静点之间。
图10中以Z3和Z串联后连接原边静点M和副边静点G为例，另外，Z3和Z串联后还可以如图11A所示连接原边静点N和副边静点D。参见图11A，该图为本申请实施例提供的另一种信号源从谐振电容耦合能量的DC/DC变换器的示意图。
下面分析具体的工作原理。
第一、由于V1N和V2N两点的信号幅值相等，相位相反，因此当C AD＝C BD时，V1N和V2N产生噪声相互抵消。
第二、由于Lk在变压器的内部均匀分布，且变压器两端的寄生电容平衡，即C AD＝C BD时，则Lk产生的噪声抵消。
第三、为了实现更好的降噪效果，谐振电感和谐振电容在寄生电容上产生的总噪 声电流与抵消阻抗的电流之和为零；其中，寄生电容是指原边绕组和副边绕组之间的寄生电容，即表达式如下：
V C/Z1+V L/Z2+V/Z＝0；
其中，V C表示谐振电容两端的电压，V L表示谐振电感两端的电感，Z1表示C AD呈现的阻抗，Z2表示C BD呈现的阻抗。
由于抵消阻抗与谐振电容Cr反向耦合，因此，V＝-n2V C。
其中，n2表示抵消阻抗的耦合电压与Cr两端电压的比值。
实际工作中，当DC/DC变换器工作在谐振频率点时，V L+V Lk-V C＝0。
由于变压器的漏感Lk与谐振电感Lr串联，因此满足V C＝V L+V Lk＝(1+n)V L。
其中，n＝V Lk/V L。
又由于C AD＝C BD，因此，Z＝[n2(n+1)/(n+2)]Z1。
从以上公式可以获得阻抗网络的呈现的阻抗大小。
以上介绍的实现方式，是通过电容耦合电容从谐振电容上耦合能量，另外也可以通过变压器从谐振电容上耦合能量，具体参见图11B，该图为本申请实施例提供的信号耦合变压器从谐振电容耦合能量的示意图。
所述信号源包括：信号耦合变压器T；
所述信号耦合变压器T的原边绕组从所述谐振电容Cr耦合能量，所述信号耦合变压器T的副边绕组与所述阻抗网络Z串联后连接在所述原边静点和所述副边静点之间。
图11B中以T的副边绕组与阻抗网络Z串联后连接在原边静点M和副边静点D为例，还可以连接在其他静点，在此不再赘述。
本实施例提供的技术方案，信号源从谐振电容上耦合能量，不需要额外增加信号源，简单易于实现，通过在谐振电容的两端并联一个抵消阻抗，从谐振电容上耦合能量即可，通过与抵消阻抗串联的阻抗网络Z可以调节注入的信号的幅值大小，由于负载变化时，通过Cr和Lr的电流均会变化，因此，随着电流的变化，抵消阻抗耦合的电压的幅值也跟随变化，因此，该方案适用于全负载范围内的噪声抑制。
以上是以变压器的副边绕组连接全波整流电路为例介绍的工作原理，当变压器的副边绕组连接全桥整流电路时，变压器的原边绕组与副边绕组之间的寄生电容的等效形式如图12和图13所示，其中图12为变压器的原边绕组连接的是全桥LLC谐振电路，图13为变压器的原边绕组连接的是半桥LLC谐振电路。
图12和图13中的寄生电容均包括两个，分别为C AC和C BC，即寄生电容C AC在原边绕组的第一端和副边绕组的第一端之间，寄生电容C BC在原边绕组的第二端和副边绕组的第一端之间。
与图6类似，对于图12和图13所示的拓扑，变压器对称的前提是，原边绕组的第一端相对于副边绕组的静点的第一寄生电容等于原边绕组的第二端对于副边绕组的静点的第二寄生电容，即C AC＝C BC。可以通过增加平衡电容来实现C AC＝C BC，即A与C之间，或者B与C之间增加平衡电容。在C AC＝C BC的基础上，本申请实施例通过增加包括平衡电容的基础上，增加噪声抑制网络，该噪声抑制网络可以产生抑制电流，抑制 电流的方向与谐振电感和谐振电容产生的总噪声电流方向相反，因此可以抑制谐振电感和谐振电容产生的总噪声电流，该噪声抑制网络的电流来源可以是额外增加的，也可以来源于DC/DC变换器内部的能量，噪声抑制网络连接在变压器的原边静点和副边静点之间，从而可以有效降低谐振电路在变压器原边绕组和副边绕组之间的寄生电容上形成的共模噪声。并且，该噪声抑制网络的电压随着负载大小的变化而变化，因此，可以适用于全负载范围的共模噪声抑制。
以上各个实施例的技术方案，如果不是并列的实现方式，可以交互使用，不做具体限定。
通信电源实施例
基于以上实施例提供的一种DC/DC变换器，本申请实施例还提供一种通信电源，下面结合附图进行详细介绍。
参见图14，该图为本申请实施例提供的一种通信电源示意图。
本实施例提供的通信电源包括以上任意一个实施例介绍的DC/DC变换器200，还包括：整流电路300；
所述整流电路300的第一端用于连接交流电源AC，所述整流电路300，用于将所述交流电源AC的交流电转换为直流电；
所述整流电路300的第二端用于连接所述DC/DC变换器200的第一端；所述DC/DC变换器200的第二端用于为负载供电；
所述DC/DC变换器200，用于将所述直流电进行转换后提供给所述负载。
负载可以为任何用电设备，具体可以根据负载的不同，DC/DC变换器200输出的电压不同，例如可以输出48V、12V、5V、3.3V等电压。
本申请实施例提供的通信电源，由于其中的DC/DC变换器中增加了噪声抑制网络，该噪声抑制网络可以产生抑制电流，抑制电流的方向与谐振电感和谐振电容的电流方向相反，因此可以抑制谐振电感和谐振电容产生的总噪声电流，该噪声抑制网络的电流来源可以是额外增加的，也可以来源于DC/DC变换器内部的能量，噪声抑制网络连接在变压器的原边静点和副边静点之间，从而可以有效降低谐振电路在变压器原边绕组和副边绕组之间的寄生电容上形成的共模噪声。并且，该噪声抑制网络的电压随着负载大小的变化而变化，因此，可以适用于全负载范围的共模噪声抑制。因此，拥有该DC/DC变换器的通信电源可以实现较好的共模噪声抑制的效果，从而给负载提供更高质量的电源。
应当理解，在本申请中，“至少一个(项)”是指一个或者多个，“多个”是指两个或两个以上。“以下至少一项(个)”或其类似表达，是指这些项中的任意组合，包括单项(个)或复数项(个)的任意组合。例如，a，b或c中的至少一项(个)，可以表示：a，b，c，“a和b”，“a和c”，“b和c”，或“a和b和c”，其中a，b，c可以是单个，也可以是多个。
以上所述，仅是本申请的较佳实施例而已，并非对本申请作任何形式上的限制。虽然本申请已以较佳实施例揭露如上，然而并非用以限定本申请。凡是未脱离本申请技术方案的内容，依据本申请的技术实质对以上实施例所做的任何简单修改、等同变 化及修饰，均仍属于本申请技术方案保护的范围内。

Claims (15)

  1. 一种DC/DC变换器，其特征在于，包括：原边绕组、副边绕组、谐振电感、谐振电容和噪声抑制网络；
    所述原边绕组、所述谐振电感和所述谐振电容串联后连接在第一节点和第二节点；
    所述副边绕组连接在第三节点和第四节点；
    所述原边绕组和所述副边绕组形成变压器；
    所述噪声抑制网络连接在所述变压器的原边静点和副边静点之间；所述原边静点为所述DC/DC变换器输入端的直流稳定电位点，所述副边静点为所述DC/DC变换器输出端的直流稳定电位点；
    所述原边绕组的第一端相对于所述副边静点的第一寄生电容等于所述原边绕组的第二端相对于所述副边静点的第二寄生电容；
    所述噪声抑制网络，用于产生抑制电流，所述抑制电流的方向与所述谐振电感和所述谐振电容产生的总噪声电流的方向相反，用于抑制所述谐振电感和所述谐振电容产生的总噪声电流，所述噪声抑制网络的电压随负载大小的变化而变化。
  2. 根据权利要求1所述的DC/DC变换器，其特征在于，所述噪声抑制网络包括：信号源和阻抗网络；
    所述信号源与所述阻抗网络串联后连接在所述原边静点和所述副边静点之间。
  3. 根据权利要求2所述的DC/DC变换器，其特征在于，所述信号源从所述谐振电感上耦合能量；
    所述信号源的电压幅值与所述谐振电感的电压幅值成正比。
  4. 根据权利要求3所述的DC/DC变换器，其特征在于，所述信号源包括抵消电感；
    所述抵消电感，用于从所述谐振电感上耦合能量，所述抵消电感与所述阻抗网络串联后连接在所述原边静点和所述副边静点之间。
  5. 根据权利要求4所述的DC/DC变换器，其特征在于，所述抵消电感与所述谐振电感共磁芯绕制。
  6. 根据权利要求3所述的DC/DC变换器，其特征在于，所述信号源包括：第一电容耦合电路；
    所述第一电容耦合电路，用于从所述谐振电感耦合能量，所述第一电容耦合电路与所述阻抗网络串联后连接在所述原边静点和所述副边静点之间。
  7. 根据权利要求2所述的DC/DC变换器，其特征在于，所述信号源从所述谐振电容上耦合能量；
    所述信号源的电压幅值与所述谐振电容的电压幅值成正比。
  8. 根据权利要求7所述的DC/DC变换器，其特征在于，所述信号源包括：第二电容耦合电路；
    所述第二电容耦合电容，用于从所述谐振电容耦合能量，所述第二电容耦合电路和所述阻抗网络串联后连接所述原边静点和所述副边静点之间。
  9. 根据权利要求7所述的DC/DC变换器，其特征在于，所述信号源包括：信号耦 合变压器；
    所述信号耦合变压器的原边绕组，用于从所述谐振电容耦合能量，所述信号耦合变压器的副边绕组与所述阻抗网络串联后连接在所述原边静点和所述副边静点之间。
  10. 根据权利要求3-7任一项所述的DC/DC变换器，其特征在于，所述谐振电感和所述谐振电容在寄生电容上产生的总噪声电流与所述抵消阻抗的电流之和为零；
    所述寄生电容为所述原边绕组和所述副边绕组之间的寄生电容。
  11. 根据权利要求1-10任一项所述的DC/DC变换器，其特征在于，所述DC/DC变换器包括以下至少一种：全桥LLC谐振DC/DC变换器和半桥LLC谐振DC/DC变换器。
  12. 根据权利要求11所述的DC/DC变换器，其特征在于，还包括：全波整流电路；
    所述副边绕组包括：第一副边绕组和第二副边绕组；
    所述第一副边绕组连接所述第三节点和第五节点之间，所述第二副边绕组连接所述第五节点和所述第四节点之间；
    所述第三节点和所述第四节点连接所述全波整流电路；所述全波整流电路的输出端为所述DC/DC变换器的输出端。
  13. 根据权利要求11所述的DC/DC变换器，其特征在于，还包括：全桥整流电路；
    所述副边绕组为一个；
    所述第三节点和所述第四节点分别为所述全桥整流电路的输入端的正极和负极；
    所述全桥整流电路的输出端为所述DC/DC变换器的输出端。
  14. 根据权利要求2所述的DC/DC变换器，其特征在于，所述阻抗网络包括以下至少一项：电阻、电容和电感。
  15. 一种通信电源，其特征在于，包括权利要求1-14任一项所述的DC/DC变换器，还包括：整流电路；
    所述整流电路的第一端用于连接交流电源；
    所述整流电路，用于将所述交流电源的交流电转换为直流电；
    所述整流电路的第二端用于连接所述DC/DC变换器的第一端；
    所述DC/DC变换器的第二端用于为负载供电；
    所述DC/DC变换器，用于将所述直流电进行转换后提供给所述负载。
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