[go: up one dir, main page]

WO2021048935A1 - Circuit de commutation de puissance - Google Patents

Circuit de commutation de puissance Download PDF

Info

Publication number
WO2021048935A1
WO2021048935A1 PCT/JP2019/035612 JP2019035612W WO2021048935A1 WO 2021048935 A1 WO2021048935 A1 WO 2021048935A1 JP 2019035612 W JP2019035612 W JP 2019035612W WO 2021048935 A1 WO2021048935 A1 WO 2021048935A1
Authority
WO
WIPO (PCT)
Prior art keywords
phase
voltage
battery
mode
current
Prior art date
Application number
PCT/JP2019/035612
Other languages
English (en)
Japanese (ja)
Inventor
田中　正一
Original Assignee
田中　正一
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by 田中　正一 filed Critical 田中　正一
Priority to JP2020573554A priority Critical patent/JP7113095B2/ja
Priority to PCT/JP2019/035612 priority patent/WO2021048935A1/fr
Publication of WO2021048935A1 publication Critical patent/WO2021048935A1/fr

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using DC to AC converters or inverters

Definitions

  • the present invention relates to a power switching circuit, particularly to a power switching circuit of an electric propulsion vehicle.
  • an electric propulsion vehicle such as a battery electric vehicle (BEV) has a power switching circuit including a three-phase inverter.
  • a common DC bus-based dual inverter has been proposed as one type of this three-phase inverter.
  • a switched battery that switches between series and parallel connections of two batteries has been proposed as another power switching circuit for electric propulsion vehicles.
  • the low energy density of batteries (kWh / kg) is well known as an essential drawback of BEVs.
  • the weight and manufacturing cost of the BEV will increase as the battery weight increases to extend the mileage.
  • the weight of the traction motor and the power consumption of the inverter are increased by this weight increase of the BEV.
  • a common quick charging method is an offboard current controlled DC charging method in which the quick charging spot adjusts the DC charging current according to the demands of the BEV battery.
  • the construction of this high density fast charging network consisting of a large number of current controlled DC charging spots is economically difficult due to the high spot cost.
  • Patent Document 1 proposes a grid-connected onboard integrated charger that uses a common DC bus-based dual inverter.
  • the dual inverter allows the neutral point open contactor to be omitted, which is required by a normal onboard integrated charger.
  • the first problem is that the grid-connected onboard integrated charger needs to have a built-in PFC (power factor correction circuit) to improve the power factor.
  • PFC power factor correction circuit
  • a three-phase PFC circuit formed by a three-phase boost chopper consisting of an open-ended three-phase coil and a common DC bus-based dual inverter has not yet been proposed.
  • the second problem is that many off-board current-controlled DC charging spots have already been built. Therefore, the grid-connected onboard integrated charger needs to have both an AC power receiving function from the three-phase grid and a DC power receiving function from the offboard current control DC charging spot. It is possible for a grid-connected onboard integrated charger to accept DC power.
  • the three-phase coil of the traction motor as part of the boost chopper causes high resistance loss. For example, when the equivalent resistance value of the three-phase coil is 0.15 ohms and the DC charging current is 100 A, a resistance loss of 1500 W occurs.
  • the third problem is that in the near future it is expected that BEV batteries will be charged by external DC power sources such as wind power plants, solar power plants, fuel cell power plants, and their composite plants. Power loss occurs when the power of these external DC power sources is converted to grid power. It is possible to charge the BEV battery directly with an external DC power source. However, the external DC power supply requires a DCDC converter for controlling the charging current for each charging spot. As a result, the cost of constructing a fast charging network will rise. After all, traditional grid-connected onboard integrated chargers have not yet envisioned the acceptance of current-controlled DC power or current-uncontrolled DC power.
  • the fourth problem is that when the battery is hot, the fast charging time of the BEV is extended. For example, in quick charging after driving on a highway in summer, the quick charging time is extended to avoid high temperature deterioration of the battery. This problem is remedied by reducing the motor loss and inverter loss of the BEV. However, conventional traction motor systems do not fully anticipate a reduction in quick charging time by reducing these power losses.
  • boost choppers and known switched batteries reduce the loss of the traction motor system.
  • these boost choppers and switched batteries greatly fluctuate the DC link voltage applied to the inverter.
  • the step-down DCDC converter for charging the low-voltage battery that steps down the DC-link voltage is adversely affected by the fluctuation of the DC-link voltage.
  • Conventional step-down DCDC converters for charging low-voltage batteries have not yet assumed this fluctuation in DC link voltage.
  • the fifth problem is that the fast charging time of the BEV is extended by the low battery temperature.
  • the quick charge time in the midwinter must be extended to avoid permanent deterioration of the lithium-ion battery.
  • An object of the present invention is to provide a power switching circuit for an electric propulsion vehicle having excellent convenience.
  • a three-phase grid-connected on-board integrated charger using a common DC bus-based dual inverter has a three-phase grid charging mode and a current-controlled DC charging mode.
  • the dual inverter is connected to a three-phase grid or external DC power supply through a common plug.
  • the battery-side three-phase inverter of the dual inverter executes a step-up rectification operation.
  • the three-phase grid has a phase-to-phase voltage lower than the DC link voltage, which is the voltage of the DC bus to which the dual inverters are connected.
  • the battery-side three-phase inverter that operates as a step-up rectifier in the three-phase grid charging mode can control the battery charging current according to the charging state of the battery.
  • the battery can be charged by both the existing current-controlled DC quick charging spot and the three-phase grid outlet, and the convenience of the electric propulsion vehicle is improved while avoiding an increase in manufacturing cost.
  • the three-phase coil of the traction motor does not consume power.
  • the plug can also be connected to a current uncontrolled external DC power supply.
  • the battery-side three-phase inverter as a boost chopper can adjust the battery charging current.
  • the battery-side three-phase inverter is driven as a three-phase phase correction (PFC) circuit in a three-phase grid charging mode.
  • PFC three-phase phase correction
  • the external DC power supply does not need to have a built-in three-phase phase correction (PFC) circuit.
  • PFC three-phase grid-connected on-board quick charging network with a good power factor can be economically constructed.
  • the battery-side three-phase inverter driven as a three-phase PFC circuit in the three-phase grid charging mode has a step-down phase period driven as a step-down chopper in addition to a step-up phase period driven as a step-up chopper.
  • a high power factor can be realized.
  • This three-phase PFC circuit which alternates between step-up phase periods and step-down phase periods, can be employed in other applications that require rectifying the three-phase grid voltage.
  • the battery is connected to the DC bus through a front end boost converter that boosts the battery voltage.
  • This front end boost converter adjusts the battery charging current in the three-phase grid charging mode and the current uncontrolled DC charging method. As a result, the battery can be charged even if the grid voltage or DC voltage applied from the outside is higher than the battery voltage.
  • a deeply discharged battery has a voltage lower than its rated voltage.
  • the three-phase grid voltage in China and Europe has high interphase voltage values.
  • wind farms, solar power plants, and fuel cell plants can have a DC voltage higher than the battery voltage of an electric vehicle to reduce loss.
  • the step-down charging operation of the front-end boost converter enables quick charging under these conditions.
  • the step-down charging operation of the front end boost converter is stopped. As a result, the loss of the front end boost converter can be reduced.
  • the front end boost converter comprises a single reactor type boost switched battery.
  • This single-reactor boost-switched battery can be adopted by other electric propulsion vehicles.
  • the front end boost converter has a voltage balance mode that reduces the voltage difference between the two batteries before connecting the two batteries in parallel when the voltage difference between the two batteries exceeds a predetermined value. .. As a result, the loss of the front end boost converter can be reduced.
  • a smoothing capacitor that applies a DC link voltage to the inverter is connected to the low voltage battery through an isolated bidirectional DCDC converter that can act as a chopper.
  • This converter has two H-bridges that are separately connected to the two coils of the transformer.
  • the H-bridge on the primary side connected to the smoothing capacitor has a precharge mode for charging the smoothing capacitor in addition to a step-down charging mode for charging the low voltage battery. This makes it possible to prevent the so-called inrush current of the smoothing capacitor.
  • the leakage inductance of the primary side H-bridge and primary coil is driven as a step-down chopper in step-down charging mode. This makes it possible to absorb fluctuations in the DC link voltage with a simple circuit topology.
  • the leakage inductance of the primary side H-bridge and primary coil is driven as a boost chopper in precharge mode.
  • inrush current can be prevented by a simple circuit topology.
  • the inverter when the battery temperature is low, supplies a single-phase alternating current from the battery to the three-phase coil.
  • the three-phase coil of a stationary three-phase motor can be regarded as a mere inductance load.
  • the battery is efficiently heated by alternately repeating the charging operation and the discharging operation.
  • the electrolyte in the battery rises and the battery resistance drops rapidly.
  • This single-phase alternating current forms two rotating magnetic fields that rotate in opposite directions. Therefore, the three-phase motor does not generate starting torque.
  • the three H-bridges of the common DC bus-based dual inverter consist of a fixed potential leg and a PWM leg, respectively.
  • the fixed potential leg and PWM leg of each H-bridge are alternated every 180 degrees of electrical angle.
  • the switching loss of the dual inverter is almost halved.
  • the upper arm transistor on the fixed potential leg is always turned on.
  • This PWM control of the dual inverter is called the upper arm conduction single PWM method. As a result, the inverter loss is reduced because the ringing surge voltage is reduced.
  • the H-bridge that outputs the smallest amplitude phase voltage is stopped and is referred to as a dormant H-bridge.
  • the two legs of the dormant H-bridge each have an upper arm switch and a lower arm switch that are turned off.
  • This PWM control method for dual inverters is called double H-bridge mode. Thereby, the battery loss can be reduced.
  • This double H-bridge mode can be adopted by other dual inverters that drive a three-phase motor.
  • the H-bridge that outputs the phase voltage of maximum amplitude is called a fixed potential H-bridge.
  • This potential-fixed H-bridge consists of two fixed-potential legs that output opposite voltages.
  • the front end boost converter connecting the battery and the dual inverter outputs a DC link voltage equal to the phase voltage of the maximum amplitude.
  • This DC link voltage is applied to one phase coil through a fixed potential H-bridge.
  • the output voltages of the two fixed potential legs of the fixed potential H-bridge are periodically switched.
  • This inverter control method is called a step-up double H-bridge mode. As a result, the power loss of the battery and the inverter can be reduced.
  • This step-up double H-bridge mode can be adopted by other dual inverters that drive a three-phase motor.
  • the double H-bridge mode and the boosted double H-bridge mode are performed together.
  • This mode is called single H-bridge mode.
  • this single H-bridge mode only one H-bridge of the dual inverter is PWM controlled. As a result, the power loss of the battery and the inverter is further reduced.
  • This single H-bridge mode can be adopted by other dual inverters that drive a three-phase motor.
  • the three H-bridges of the dual inverter are each controlled by the space vector PWM method (SVPWM), and the current supply periods of the three H-bridges are arranged so as not to overlap as much as possible. Thereby, the battery loss can be reduced.
  • This current distribution method can be adopted by other dual inverters that drive three-phase motors.
  • FIG. 1 is a circuit diagram showing a power switching circuit of an embodiment.
  • FIG. 2 is a waveform diagram of a three-phase grid current.
  • FIG. 3 is a timing chart showing a power factor correction operation of a dual inverter operating as a three-phase PFC circuit.
  • FIG. 4 is a circuit diagram showing the current flow in the dual inverters in the phase periods P1 and P2.
  • FIG. 5 is a circuit diagram showing a current flow in the dual inverter during the phase period P3.
  • FIG. 6 is a circuit diagram showing a current flow in the dual inverter during the phase period P4.
  • FIG. 7 is a circuit diagram showing the current flow in the dual inverters in the phase periods P5 and P6.
  • FIG. 8 is a circuit diagram showing a current flow in the dual inverter during the phase period P7.
  • FIG. 9 is a circuit diagram showing a current flow in the dual inverter during the phase period P8.
  • FIG. 10 is a circuit diagram showing the current flow in the dual inverters in the phase periods P9 and P10.
  • FIG. 11 is a circuit diagram showing a current flow in the dual inverter during the phase period P11.
  • FIG. 12 is a circuit diagram showing a current flow in the dual inverter during the phase period P12.
  • FIG. 13 is a circuit diagram showing a power switching circuit in a modified mode.
  • FIG. 14 is a block diagram showing an electromagnetic brake device.
  • FIG. 15 is a flowchart showing the control of the electromagnetic brake device.
  • FIG. 15 is a flowchart showing the control of the electromagnetic brake device.
  • FIG. 16 is a circuit diagram showing an isolated bidirectional DCDC converter.
  • FIG. 17 is a block circuit diagram for explaining a single-phase battery heating method.
  • FIG. 18 is a timing chart showing waveforms of fundamental frequency components of a single-phase AC current and a single-phase AC voltage adopted in the single-phase battery heating method.
  • FIG. 19 is a diagram showing a state of dual inverters in the upper arm conduction type single PWM method.
  • FIG. 20 is a diagram showing an output voltage of an H bridge driven by the upper arm conduction type single PWM method.
  • FIG. 21 is a schematic circuit diagram showing a current supply period of the H bridge driven by the double PWM method.
  • FIG. 22 is a schematic circuit diagram showing the current regeneration period of the H bridge driven by the double PWM method.
  • FIG. 23 is a schematic circuit diagram showing a current supply period of the H bridge driven by the single PWM method.
  • FIG. 24 is a schematic circuit diagram showing a freewheeling period of an H-bridge driven by the single PWM method.
  • FIG. 25 is a timing chart showing the fundamental frequency components of the U-phase voltage applied to the U-phase coil and the U-phase current supplied to the U-phase coil.
  • FIG. 26 is a schematic circuit diagram showing a U-phase current flowing through a U-phase coil during the phase period P1.
  • FIG. 27 is a schematic circuit diagram showing a U-phase current flowing through a U-phase coil during the phase period P2.
  • FIG. 28 is a schematic circuit diagram showing a U-phase current flowing through a U-phase coil during the phase period P3.
  • FIG. 29 is a schematic circuit diagram showing a U-phase current flowing through a U-phase coil during the phase period P4.
  • FIG. 30 is a vector diagram showing a phase region in which the combined rotation voltage vector formed by the double H bridge mode rotates.
  • FIG. 31 is a diagram showing a state of the dual inverter in the double H bridge mode.
  • FIG. 32 is a diagram showing a waveform of a fundamental frequency component of a three-phase grid voltage in a step-up double H bridge mode.
  • FIG. 33 is a diagram showing a state of the dual inverter in the step-up double H bridge mode.
  • FIG. 34 is a diagram showing a state of each leg in the single H bridge mode.
  • FIG. 35 is a timing chart showing the arrangement of the current supply period of each phase given to the dual inverter driven by the current distribution method.
  • FIG. 36 is a timing chart showing a modification of the current dispersion method.
  • FIG. 1 is a circuit diagram showing a power switching circuit that also serves as an onboard integrated charger.
  • This power switching circuit called a three-phase motor drive, has a single reactor type step-up switched battery 100, a common DC bus-based dual inverter 200, and an isolated bidirectional DCDC converter 300, a controller 9, a smoothing capacitor 13, and a smoothing capacitor 13. It further includes a three-phase plug 400.
  • the boost-switched battery 100 is connected to the high-level DC bus 81 through the high-side main relay 61, and is connected to the low-level DC bus 82 through the low-side main relay 62.
  • the smoothing capacitor 13 connected to the DC buses 81 and 82 has a DC link voltage VC.
  • the main relays 61 and 62 are safety switches for disconnecting the step-up switched battery 100 from other circuit sections of the power switching circuit.
  • Each switch of the step-up switched battery 100 and the dual inverter 200 consists of an IGBT having an antiparallel diode.
  • the controller 9 controls the step-up switched battery 100, the dual inverter 200, and the DCDC converter 300.
  • the boost-switched battery 100 which is a DC power supply circuit, comprises a low-side battery 1, a high-side battery 2, and a front-end boost converter 10.
  • the front end boost converter 10 called a connection switching circuit has both a connection switching function for batteries 1 and 2 and a bidirectional DCDC converter function.
  • the converter 10 has a series switch 3, a low-side parallel switch 4, a high-side parallel switch 5, an output switch 6, a reactor 7, a surge absorbing diode 8, and three safety fuses (501-503).
  • the high-level DC bus 81 is connected to the low-level DC bus 82 through the output switch 6, the battery 2, the reactor 7, the series switch 3, and the battery 1.
  • the connection point between the negative electrode of the battery 2 and the reactor 7 is connected to the negative electrode of the battery 1 through the parallel switch 4.
  • the connection point between the positive electrode of the battery 1 and the series switch 3 is connected to the positive electrode of the battery 2 through the parallel switch 5.
  • the connection point between the reactor 7 and the series switch 3 is connected to the cathode electrode of the diode 8, and the anode electrode of the diode 8 is connected to the negative electrode of the battery 1.
  • Batteries 1 and 2 have the same rated voltage.
  • the operation mode of the front end boost converter 10 includes a parallel mode, a series mode, a transient mode, a voltage balance mode, a step-up discharge mode, and a step-down charge mode.
  • Parallel mode and series mode can be adopted in both the discharging operation and the charging operation of the batteries 1 and 2.
  • parallel mode the series switch 3 is turned off and the parallel switches 4 and 5 are turned on.
  • the DC link voltage VC is equal to the voltage of one battery.
  • series mode the series switch 3 is turned on, the parallel switches 4 and 5 are turned off, and the batteries 1 and 2 are connected in series. Therefore, the DC link voltage VC is the sum of the voltages of the two batteries.
  • the transient mode executed in the short transient period between the parallel mode and the series mode consists of a transient discharge mode and a transient charge mode.
  • the parallel switches 4 and 5 are turned off and the series switch 3 is PWM controlled.
  • the PWM duty ratio of the series switch 3 is slowly changed from 0 to 1.
  • the PWM duty ratio of the series switch 3 is slowly changed from 1 to 0.
  • the DC link voltage VC is a value obtained by subtracting the counter electromotive force of the reactor 7 from the sum of the voltages of the batteries 1 and 2.
  • the series switch 3 is turned off, the magnetic energy of the reactor 7 is consumed by the current flowing through the diode 8, the reactor 7, the battery 2, and the output switch 6. After all, the transient discharge mode slowly changes the DC link voltage VC.
  • the parallel switches 4 and 5 are synchronously PWM controlled.
  • the PWM duty ratios of the parallel switches 4 and 5 are slowly changed from 1 to 0.
  • the PWM duty ratios of the parallel switches 4 and 5 are slowly changed from 0 to 1.
  • the magnetic energy of the reactor 7 is increased by the charging current that charges the batteries 1 and 2 in series. In other words, the reactor 7 suppresses an increase in the charging current flowing through the batteries 1 and 2.
  • the DC link voltage VC is applied to the batteries 1 and 2, respectively. This causes a sharp increase in the charging current of batteries 1 and 2.
  • the magnetic energy of the reactor 7 causes a freewheeling current to flow through the antiparallel diode of the series switch 3, the parallel switch 5, and the battery 2.
  • the magnetic energy of the reactor 7 causes a freewheeling current to flow through the antiparallel diode of the series switch 3, the battery 1, and the parallel switch 4.
  • the voltage balance mode is executed when the batteries 1 and 2 have different voltages.
  • a short circuit current circulates through batteries 1 and 2 through the two parallel switches 4 and 5.
  • the high voltage battery charges the low voltage battery.
  • this short-circuit current exceeds a predetermined value, the power loss of the batteries 1 and 2 increases.
  • the voltage difference between the two batteries 1 and 2 is detected before the parallel mode is started.
  • this voltage difference exceeds a predetermined value, turning on the transistor of at least one of the parallel switches is prohibited at least in the initial period of the parallel mode. For example, a case where the battery 1 is higher than the battery 2 by a predetermined voltage value or more before the start of the parallel mode will be described below.
  • the transistor of the parallel switch 4 In the battery discharge mode, the transistor of the parallel switch 4 is prohibited from being turned on. As a result, only the battery 1 supplies the current to the dual inverter 200. After the voltages of the batteries 1 and 2 become substantially equal due to the voltage drop of the battery 1, the transistor of the parallel switch 4 is turned on. However, the transistor of the parallel switch consisting of the IGBT does not need to be turned on because it is a parasitic diode.
  • the transistor of the parallel switch 5 In the battery charge mode, the transistor of the parallel switch 5 is prohibited from being turned on. As a result, the battery 2 is charged through the parallel switch 4. After the voltages of the batteries 1 and 2 become substantially equal due to the voltage rise of the battery 2, the transistor of the parallel switch 5 is turned on. As a result, the voltages of the batteries 1 and 2 are equalized.
  • a control opposite to the above control is executed. According to this voltage balance mode, it is possible to prevent a useless circulating current (short-circuit current) flowing immediately after the start of the parallel mode.
  • Boost-discharge mode In the boost-discharge mode in which the boost-switched battery 100 applies a boost voltage to the dual inverter 200, the series switch 4 is turned on, and the parallel switches 4 and 5 are synchronously PWM-controlled.
  • the parallel switches 4 and 5 When the parallel switches 4 and 5 are turned on, the batteries 1 and 2 each supply a circulating current to the reactor 7. As a result, the magnetic energy of the reactor 7 increases.
  • the parallel switches 4 and 5 are turned off, the DC link voltage VC is the sum of the sum of the voltages of the batteries 1 and 2 and the voltage of the reactor 7.
  • the boost ratio (VC / (V1 + V2)) is controlled by adjusting the PWM duty ratios of the parallel switches 4 and 5.
  • the short-circuit current circulates through the battery 1, parallel switch 5, battery 2, and parallel switch 4 immediately after the parallel switches 4 and 5 are turned on. To do. As a result, the higher voltage battery charges the lower voltage battery.
  • the voltage balance mode is executed in the step-up / discharge mode. For example, when the voltage of the battery 2 is higher than the voltage of the battery 1 by a predetermined value or more, the PWM switching of the parallel switch 5 is suspended. As a result, only the battery 1 is discharged. After the voltages of the batteries 1 and 2 become substantially equal, the parallel switches 4 and 5 are synchronously PWM controlled.
  • Step-down charging mode In the step-down charging mode for charging the batteries 1 and 2, the series switch 4 is turned on, the parallel switches 4 and 5 are turned off, and the output switch 6 is PWM-controlled.
  • the output switch 6 When the output switch 6 is turned on, charging current flows through the battery 2, reactor 7, series switch 3, and battery 1, and batteries 1 and 2 are charged in series.
  • the reactor 7 stores magnetic energy.
  • the magnetic energy of the reactor 7 circulates the charging current through the reactor 7, the series switch 3, the parallel switch 5, and the battery 2. Further, this magnetic energy circulates the charging current through the reactor 7, the series switch 3, the battery 1, and the parallel switch 4. After all, the magnetic energy of the reactor 7 charges the batteries 1 and 2 in parallel.
  • the step-down ratio ((V1 + V2) / VC) is adjusted by the PWM duty ratio of the output switch 6.
  • Boost Switched Battery 100 In battery electric vehicle (BEV) applications, parallel mode is adopted in the low speed region, series mode is adopted in the medium speed region, and boost discharge mode and step down charge mode are adopted in the high speed region.
  • the step-up discharge mode realizes an increase in the number of turns of the three-phase coil 50 and a decrease in the inverter current.
  • inverter loss is significantly reduced compared to non-boost switched batteries. For example, when the maximum boost ratio is 2, the inverter current is halved as compared with the non-boost type switched battery, and the inverter loss is 1/4.
  • this inverter current reduction reduces battery loss in parallel and series modes.
  • the maximum boost ratio is 2
  • the amplitude of the power supply current supplied from the batteries 1 and 2 to the inverter is halved.
  • the energization time of the power supply current is almost doubled due to the decrease of the DC link voltage VC.
  • the step-up switched battery 100 can have about half the battery loss in the non-boost region including the parallel mode and the series mode as compared with the non-boost switched battery.
  • the parallel mode of the switched battery has approximately 1/4 of the battery resistance compared to its series mode.
  • the energization time of the power supply current supplied from the batteries 1 and 2 to the inverter is almost doubled due to the decrease in the DC link voltage VC.
  • the parallel mode can have about half the battery loss as the series mode.
  • the reactor 7 of the step-up switched battery 100 shown in FIG. 1 does not generate resistance loss in parallel mode.
  • the single reactor type boost switched battery 100 shown in FIG. 1 may have about 70% weight and loss as compared to the reactor as compared with the conventional double reactor type boost switched battery having two reactors. it can.
  • a traction motor composed of a three-phase synchronous motor has an open-end three-phase coil 50 as a stator coil.
  • the three-phase coil 50 includes a U-phase coil 5U, a V-phase coil 5V, and a W-phase coil 5W.
  • the common DC bus-based dual inverter 200 includes a grid-side three-phase inverter 30 and a battery-side three-phase inverter 40.
  • the inverter 30 includes a U-phase leg 3U, a V-phase leg 3V, and a W-phase leg 3W.
  • the inverter 40 includes a U-phase leg 4U, a V-phase leg 4V, and a W-phase leg 4W. In normal operation, the upper arm switch and the lower arm switch of each leg are switched complementarily.
  • the three AC terminals of the inverter 30 are separately connected to the three terminals (X, Y, Z) of the plug 400 for grid connection.
  • the inverter 40 is individually connected to each end of each of the three phase coils (5U, 5V, and 5W), and the inverter 30 is individually connected to each other end of each of the three phase coils (5U, 5V, and 5W). ..
  • the inverter 30 includes three upper arm switches (31, 33, 35) and three lower arm switches (32, 34, 36).
  • the inverter 40 includes three upper arm switches (41, 43, 45) and three lower arm switches (42, 44, 46).
  • the U-phase H-bridge consisting of legs 3U and 4U controls the U-phase current IU supplied to the phase coil 5U.
  • the V-phase H-bridge consisting of legs 3V and 4V controls the V-phase current IV supplied to the phase coil 5V.
  • the W-phase H-bridge consisting of legs 3W and 4W controls the W-phase current IW supplied to the W-phase coil 5W.
  • the torque control operation of the dual inverter 200 for generating the motor torque will be described later.
  • the step-up switched battery 100, dual inverter 200, and three-phase coil 50 form an in-vehicle integrated charger.
  • the operation mode of this in-vehicle integrated charger will be described below. This operating mode includes a current controlled DC charging mode, a current uncontrolled DC charging mode, a grid charging mode, and a reverse power transmission mode.
  • the conventional current control DC quick charging spot has a built-in DCDC converter that can control the DC charging current according to the BEV's request.
  • This current-controlled DC charging mode is executed when the two terminals (X, Y) of the charging plug 400 are connected to the current-controlled DC quick charging spot.
  • the series switch 3 and the output switch 6 of the step-up switched battery 100 are turned on, and the batteries 1 and 2 are charged in series.
  • the switching operation of the two three-phase inverters 30 and 40 of the dual inverter 200 is essentially stopped. However, synchronous rectification operation is possible.
  • the DC charging current flows from the terminal X of the charging plug 400 to the high-level DC bus 81 through the upper arm switch 31, and then charges the two batteries 1 and 2.
  • the DC charging current returned from the battery 1 to the low-level DC bus 82 returns to the terminal Y of the charging plug 400 through the lower arm switch 34.
  • Current control The DCDC converter of the DC quick charge spot controls the DC charge current in response to the request of the controller 9.
  • the advantage of this current controlled DC charging mode is that the DC charging current does not flow through the three-phase coil 50. As a result, it is possible to improve the DC charging efficiency and suppress the temperature rise of the three-phase coil 50. For example, when the average resistance value of the three-phase coil 50 is 0.15 ohms and the average charging current is 100 A, the power loss of the three-phase coil 50 is 1500 W.
  • This low-voltage current-uncontrolled DC charging mode is operated when the DC voltage of the current-uncontrolled power plant is lower than the sum of the voltages of the batteries 1 and 2.
  • This DC voltage is boosted by a boost chopper composed of a U-phase coil 5U and a leg 4U, and then applied to the boost-switched battery 100.
  • the boost chopper composed of a U-phase coil 5U and a leg 4U, and then applied to the boost-switched battery 100.
  • the U-phase current IU returns to the power plant through the upper arm switch 41, the step-up switched battery 100, and the lower arm switch 34. As a result, batteries 1 and 2 are charged.
  • the charging current is adjusted by controlling the PWM duty ratio of the lower arm switch 42.
  • This low voltage current uncontrolled DC charging mode is suitable for charging a BEV from a low voltage DC power plant, such as a home solar panel.
  • the high voltage current uncontrolled DC charging mode is suitable for charging BEVs from high voltage DC power plants such as large solar cell plants.
  • the three-phase grid charging mode When a three-phase grid voltage is applied to the plug 400, the three-phase grid charging mode is executed.
  • the three-phase sinusoidal grid voltage applied to the plug 400 includes a U-phase voltage VU1, a V-phase voltage VV1, and a W-phase voltage VW1. Batteries 1 and 2 are connected in series.
  • the three interphase voltages (VU1-VV1, VV1-VW1, and VW1-VU1) are lower than the DC link voltage VC.
  • the dual inverter 200 and the open-ended three-phase coil 50 form a three-phase power factor correction circuit (PFC circuit).
  • FIG. 2 shows waveforms of U-phase current IU, V-phase current IV, and W-phase current IW flowing from the plug 400 to the three-phase coil 50 in one cycle period of the three-phase grid voltage.
  • the U-phase current IU, V-phase current IV, and W-phase current IW are called charge-phase currents.
  • Each charging phase current (IU, IV, and IW) generally has a different amplitude value than each phase grid voltage (VU1, VV1, and VW1) of the three-phase grid voltage.
  • each charge phase current is shown to have an amplitude equal to each phase grid voltage.
  • a U-phase current IU having a sinusoidal waveform in phase with the U-phase voltage VU1 flows through the U-phase coil 5U.
  • the V-phase current IV which has a sinusoidal waveform in phase with the V-phase voltage VV1, flows through the V-phase coil 5V.
  • a W-phase current IW having a sinusoidal waveform in phase with the W-phase voltage VW1 flows through a W-phase coil 5W.
  • This one cycle period is divided into 12 phase periods P1-P12.
  • the inverter 40 forms a chopper circuit together with the three-phase coil 50.
  • the legs 4U, 4V, and 4W of the inverter 40 have a step-up mode B, a pause mode OFF, and a step-down mode D in the three-phase PFC mode.
  • FIG. 3 is a diagram showing mode selection of legs (4U, 4V, and 4W) in the phase period P1-P12.
  • 4 to 12 are diagrams showing the flow of phase current during the phase periods P1-P12. When the hibernation mode is OFF, the PWM switching of the leg is stopped.
  • FIG. 4 shows the current flow during the phase period (P1, P2).
  • the U-phase coil 5U and the U-phase leg 4U form a U-phase boost chopper.
  • the switch 42 is PWM controlled.
  • the U-phase current IU is supplied to the smoothing capacitor 13 during the off period of the switch 42.
  • the U-phase current IU returning from the smoothing capacitor 13 returns to the grid through the V-phase lower arm switch 34.
  • the W-phase coil 5W and the W-phase leg 4W form a W-phase boost chopper.
  • the switch 46 is PWM controlled.
  • the W-phase current IW is supplied to the smoothing capacitor 13 during the off period of the switch 46.
  • the W-phase current IW returning from the smoothing capacitor 13 returns to the grid through the V-phase lower arm switch 34.
  • FIG. 5 shows the current flow during the phase period P3.
  • the switch 42 continues PWM control.
  • the W-phase coil 5W and the W-phase leg 4W form a W-phase step-down chopper.
  • the switch 45 is PWM controlled.
  • the W-phase current IW is supplied to the grid from the smoothing capacitor 13 through the W-phase coil 5W during the on period of the switch 45.
  • the magnetic energy of the W-phase coil 5W supplies a freewheeling current through the switch 46.
  • FIG. 6 shows the current flow during the phase period P4.
  • the switch 42 continues PWM control.
  • the V-phase coil 5V and the V-phase leg 4V form a V-phase step-down chopper.
  • the switch 43 is PWM controlled.
  • the V-phase current IV is supplied to the grid from the smoothing capacitor 13 through the V-phase coil 5V while the switch 43 is turned on.
  • the magnetic energy of the V-phase coil 5V supplies a freewheeling current through the switch 44.
  • FIG. 7 shows the current flow during the phase period (P5, P6).
  • the switch 42 continues PWM control.
  • the V-phase coil 5V and the V-phase leg 4V form a V-phase boost chopper.
  • the switch 44 is PWM controlled.
  • the V-phase current IV is supplied to the smoothing capacitor 13 during the off period of the switch 44.
  • the V-phase current IV returning from the smoothing capacitor 13 returns to the grid through the W-phase lower arm switch 36.
  • FIG. 8 shows the current flow during the phase period P7.
  • the switch 44 continues PWM control.
  • the U-phase coil 5U and the U-phase leg 4U form a U-phase step-down chopper.
  • the switch 41 is PWM controlled.
  • the U-phase current IU is supplied to the grid from the smoothing capacitor 13 through the U-phase coil 5U while the switch 41 is on.
  • the magnetic energy of the U-phase coil 5U supplies a freewheeling current through the switch 42.
  • FIG. 9 shows the current flow during the phase period P8.
  • the switch 44 continues PWM control.
  • the W-phase coil 5W and the W-phase leg 4W form a W-phase step-down chopper.
  • the switch 45 is PWM controlled.
  • the W-phase current IW is supplied to the grid from the smoothing capacitor 13 through the W-phase coil 5W while the switch 45 is turned on.
  • the magnetic energy of the W-phase coil 5W supplies a freewheeling current through the switch 46.
  • FIG. 10 shows the current flow during the phase period (P9, P10).
  • the switch 44 continues PWM control.
  • the W-phase coil 5W and the W-phase leg 4W form a W-phase boost chopper.
  • the switch 46 is PWM controlled.
  • the W-phase current IW is supplied to the smoothing capacitor 13 during the off period of the switch 46.
  • the W-phase current IW returning from the smoothing capacitor 13 returns to the grid through the U-phase lower arm switch 32.
  • FIG. 11 shows the current flow during the phase period P11.
  • the switch 46 continues PWM control.
  • the V-phase coil 5V and the V-phase leg 4V form a V-phase step-down chopper.
  • the switch 43 is PWM controlled.
  • the V-phase current IV is supplied to the grid from the smoothing capacitor 13 through the V-phase coil 5V while the switch 43 is turned on.
  • the magnetic energy of the V-phase coil 5V supplies a freewheeling current through the switch 44.
  • FIG. 12 shows the current flow during the phase period P12.
  • the switch 46 continues PWM control.
  • the U-phase coil 5U and the U-phase leg 4U form a U-phase step-down chopper.
  • the switch 41 is PWM controlled.
  • the U-phase current IU is supplied to the grid from the smoothing capacitor 13 through the U-phase coil 5U while the switch 41 is on.
  • the magnetic energy of the U-phase coil 5U supplies a freewheeling current through the switch 42.
  • the charging current for charging the batteries 1 and 2 is adjusted by the step-up chopper operation of the three-phase inverter 40 or the step-down charging mode of the step-up switched battery 100.
  • this charging current can be adjusted by the step-down charging mode of the step-up switched battery 100. Suitable.
  • the charging current adjusted by the PWM control of the output switch 6 matches the step-up current output by the three-phase inverter 40. As a result, the ripple of the DC link voltage VC can be reduced.
  • This charging current adjustment by the step-up switched battery 100 is suitable in a three-phase grid charging mode using a 400V class three-phase grid.
  • a single-phase grid voltage is applied between the two terminals (X and Y) of the plug 400. This single-phase grid voltage is lower than the DC link voltage VC.
  • the dual inverter 200 and the three-phase coil 50 form a single-phase PFC circuit.
  • the front end boost converter 10 that executes the step-down charging mode controls the charging current. The step-down operation of the front end boost converter 10 is the same in the three-phase grid charging mode and the single-phase grid charging mode.
  • a single-phase PFC mode operated by a single-phase PFC circuit is described.
  • the phase voltage VU1 is higher than the phase voltage VV1.
  • the phase voltage VU1 is lower than the phase voltage VV1.
  • the U-phase boost chopper composed of the U-phase coil 5U and the U-phase leg 4U is operated, and the lower arm switch 42 is PWM-controlled.
  • the switch 42 is turned on, a current IU flows through the U-phase coil 5U, the switch 42, and the switch 34, and the U-phase coil 5U stores magnetic energy.
  • the V-phase boost chopper composed of the V-phase coil 5V and the V-phase leg 4V is operated, and the lower arm switch 44 is PWM-controlled.
  • the switch 44 When the switch 44 is turned on, a current IV flows through the V-phase coil 5V, the switch 44, and the switch 32, and the V-phase coil 5V stores magnetic energy.
  • the switch 44 When the switch 44 is turned off, a current IV flows through the V-phase coil 5U, the switch 43, the smoothing capacitor 13, and the switch 32, and the smoothing capacitor 13 is charged. Due to the PWM control of the switch 44, the waveform of the charging current becomes a sinusoidal waveform that matches the negative half-cycle sinusoidal waveform of the single-phase grid voltage.
  • Three-Phase Reverse Power Transmission Mode A three-phase reverse power transmission mode for reverse power transmission from a step-up switched battery 100 to a three-phase grid will be described.
  • the DC link voltage VC is higher than the peak value of the interphase voltage of the three-phase grid.
  • the front end boost converter 10 executes the boost discharge mode when the sum of the voltages of the batteries 1 and 2 is lower than this peak value, and executes the series mode when the sum of the voltages of the batteries 1 and 2 is higher than this peak value. ..
  • the three-phase step-down chopper including the three-phase coil 50 and the three-phase inverter 40 supplies a three-phase alternating current to the three-phase grid through the plug 400.
  • the three-phase inverter 30 is stopped. This three-phase step-down chopper doubles as a PFC circuit.
  • the leg 4U and the U-phase coil 5U form a U-phase step-down chopper
  • the leg 4V and the V-phase coil 5V form a V-phase step-down chopper
  • the leg 4W and the W-phase coil 5W form a W-phase step-down chopper.
  • the operation of the three-phase buck chopper will be described with reference to FIG.
  • the three-phase step-down chopper supplies the plug 400 with a three-phase sinusoidal current (IU, IV, IW) that is in phase opposite to the three-phase grid voltage.
  • the three-phase currents (IU, IV, IW) have phases that differ from the three-phase grid voltages (VU1, VV1, VW1) by an electrical angle of 180 degrees.
  • phase period (P12, P1-P3) shown in FIG. 2 the lower arm switch 44 of the V-phase leg 4V is turned on, and the upper arm switch 41 of the U-phase leg 4U and the upper arm switch 45 of the W-phase leg 4W are PWM. Be controlled.
  • the phase period (P4-P7) the lower arm switch 46 of the W phase leg 4W is turned on, and the upper arm switch 41 of the U phase leg 4U and the upper arm switch 43 of the V phase leg 4V are PWM controlled.
  • phase period (P8-P11) the lower arm switch 42 of the U-phase leg 4U is turned on, and the upper arm switch 43 of the V-phase leg 4V and the upper arm switch 45 of the W-phase leg 4W are PWM-controlled.
  • the DC link voltage VC selects a suitable discharge mode according to the single-phase grid voltage.
  • the terminals (X, Y) of the plug 400 are connected to a single-phase grid.
  • the U-phase coil 5U and the U-phase leg 4U form a U-phase step-down chopper.
  • the V-phase coil 5V and the V-phase leg 4V form a V-phase step-down chopper.
  • the three-phase inverter 30 and the W-phase leg 4W are stopped.
  • the U-phase step-down chopper and the V-phase step-down chopper also serve as a single-phase PFC circuit.
  • a U-phase boost chopper consisting of a U-phase coil 5U and a U-phase leg 4U is operated, and the upper arm switch 41 is PWM-controlled. Will be done.
  • the lower arm switch 44 of the V-phase leg 4V is turned on.
  • the V-phase boost chopper consisting of V-phase coil 5V and V-phase leg 4V is operated, and the upper arm switch 43 is PWM-controlled. Will be done.
  • the lower arm switch 42 of the U-phase leg 4U is turned on.
  • the vehicle-mounted integrated charger shown in FIG. 13 uses one battery 1 instead of the step-up switched battery 100 of the vehicle-mounted integrated charger shown in FIG.
  • the vehicle-mounted integrated charger of FIG. 13 performs essentially the same operation as the vehicle-mounted integrated charger of FIG.
  • the interphase voltage (VU1-VV1, VV1-VW1, and VW1-VU1) of the three-phase grid is lower than the voltage Vb of the battery 1.
  • the PWM duty ratio of the battery-side three-phase inverter 40 is controlled in order to adjust the charging current flowing into the battery 1.
  • the DC voltage applied to the plug 400 is lower than the voltage Vb of the battery 1 in the current uncontrolled DC charging mode.
  • This in-vehicle integrated charger can operate the dual inverter 200 as a PFC circuit in the three-phase and single-phase grid charging modes.
  • the in-vehicle integrated charger can be connected to both current controlled DC fast charging spots and current uncontrolled DC quick charging spots. Further, this in-vehicle integrated charger can reduce the resistance loss of the three-phase coil 50 to zero when connected to the DC quick charging spot. For example, when a current uncontrolled DC charging method is adopted, it is possible to charge the battery from the solar panel with high efficiency.
  • the grid-side three-phase inverter is turned off. This prevents the DC power supply battery and semiconductor switch from being destroyed by the short-circuit current.
  • FIG. 14 of the electric brake device 90 is a block diagram showing the electromagnetic brake device 90 used in the grid charging mode.
  • the controller 9 controls the electric braking device 90 that restrains the wheels 91.
  • the brake control routine executed by the controller 9 will be described with reference to FIG. First, it is determined whether or not the battery charge mode is commanded (S100), and when it is determined that the battery charge mode is commanded, the wheel 91 is restrained by the electric brake device 90 (S102). When the battery charge mode is not commanded, it is determined whether or not a three-phase grid voltage is applied from the plug 400 to the dual inverter 200 (S104), and when it is determined that a three-phase grid voltage is applied, the electric brake device. The wheel 91 is restrained by 90 (S102). When it is determined that the three-phase grid voltage is not applied, the restraint of the wheel 91 by the electric braking device 90 is released (S106).
  • the three-phase coil 50 In the three-phase grid charging mode, the three-phase coil 50 generates a rotating magnetic field synchronized with the grid frequency.
  • the rotor of the stationary three-phase synchronous motor generates acceleration torque in the first half of the cycle period of the grid frequency, and generates deceleration torque in the latter half of the cycle period.
  • the three-phase synchronous motor After all, the three-phase synchronous motor generates vibrations synchronized with the grid frequency. This vibration is reduced when the electric braking device 90 constrains the wheels 91.
  • the isolated bidirectional DCDC converter 300 includes a primary H-bridge 301, a transformer 302, and a secondary H-bridge 303. Details of the converter 300 will be described with reference to FIGS. 1 and 16.
  • the H-bridge 301 is composed of a first leg and a second leg.
  • the first leg consists of an upper arm switch 311 and a lower arm switch 312.
  • the second leg consists of an upper arm switch 313 and a lower arm switch 314.
  • the H-bridge 303 is composed of a third leg and a fourth leg.
  • the third leg consists of an upper arm switch 315 and a lower arm switch 316.
  • the fourth leg consists of an upper arm switch 317 and a lower arm switch 318.
  • the transformer 302 having the primary coil C1 and the secondary coil C2 has a leakage inductance L1 on the primary side and a leakage inductance L2 on the secondary side. Since the primary coil C1 has a much larger number of turns than the secondary coil C2, the leakage inductance L1 is much higher than the leakage inductance L2.
  • the H bridge 301 is connected to the primary coil C1 through the leakage inductance L1.
  • the H bridge 303 is connected to the secondary coil C2 through the leakage inductance L2.
  • the H-bridge 301 is connected in parallel with the smoothing capacitor 13 through the DC buses 81 and 82.
  • the H-bridge 303 is connected in parallel with the low voltage battery 304 connected to the low voltage electrical load.
  • the switches 311-318 consist of MOS transistors.
  • the converter 300 has a step-down charge mode and a precharge mode.
  • Step-down charging mode In the step-down charging mode for charging the low-voltage battery 304, the H-bridge 301 is PWM-controlled and an AC voltage is applied to the primary coil C1. A positive half cycle period of this AC voltage is explained.
  • the lower arm switch 314 is turned on. Switches 311 and 312 are complementarily PWM switched. During the period when the switch 311 is turned on, the primary current flowing into the primary coil C1 through the leakage inductance L1 increases. During the period when the switch 311 is turned off, the magnetic energy of the leakage inductance L1 causes a freewheeling current circulating through the switch 312, the leakage inductance L1, the primary coil C1, and the lower arm switch 314 to flow. This freewheeling current decays rapidly. The secondary voltage induced in the secondary coil C2 by this change in the primary current is rectified by the H bridge 303 to charge the battery 304. The charging current is controlled by controlling the PWM duty ratio of the switch 311.
  • the negative half cycle period of this AC voltage is explained.
  • the lower arm switch 312 is turned on.
  • Switches 313 and 314 are complementarily PWM switched.
  • the primary current flowing into the leakage inductance L1 through the primary coil C1 increases.
  • the magnetic energy of the leakage inductance L1 causes a freewheeling current circulating through the switch 314, the primary coil C1, the leakage inductance L1 and the lower arm switch 312.
  • This freewheeling current decays rapidly.
  • the secondary voltage induced in the secondary coil C2 by this change in the primary current is rectified by the H bridge 303 to charge the battery 304.
  • the charging current is controlled by controlling the PWM duty ratio of the switch 313.
  • the H bridge 301 and the leakage inductance L1 operate as an insulating step-down chopper. As a result, this isolated step-down chopper allows for changes in the DC link voltage VC.
  • the two legs of the H-bridge 303 are complementarily PWM controlled at a predetermined frequency.
  • an alternating current flows through the secondary coil C2, and the secondary voltage is induced in the primary coil C1.
  • the lower arm switch 312 is PWM controlled.
  • the lower arm switch 312 When the lower arm switch 312 is turned on, the secondary current circulates through the primary coil C1, the leakage inductance L1, the lower arm switch 312, and the lower arm switch 314, and the magnetic energy of the leakage inductance L1 increases.
  • the lower arm switch 312 is turned off, this magnetic energy supplies the precharge current through the upper arm switch 311, the smoothing capacitor 13, the lower arm switch 314, and the primary coil C1.
  • the lower arm switch 314 is PWM controlled.
  • the secondary current circulates through the primary coil C1, the lower arm switch 314, the lower arm switch 312, and the leakage inductance L1, and the magnetic energy of the leakage inductance L1 increases.
  • the lower arm switch 314 is turned off, this magnetic energy supplies a precharge current through the leakage inductance L1, the primary coil C1, the upper arm switch 313, the smoothing capacitor 13, and the lower arm switch 312.
  • the smoothing capacitor 13 is precharged.
  • the H bridge 301 and the leakage inductance L1 operate as an insulating boost chopper.
  • This converter 300 can absorb changes in the DC link voltage VC between parallel mode, series mode, and step-up / discharge mode in charging the low voltage battery 304. Further, the converter 300 can prevent an inrush current flowing into the smoothing capacitor 13.
  • FIG. 17 is a block circuit diagram showing the flow of current flowing through the dual inverter 200.
  • the dual inverter 200 connected to the three-phase coil 50 is supplied with power from the battery 1 through the DC buses 81 and 82.
  • the dual inverter 200 includes a U-phase H-bridge 710, a V-phase H-bridge 720, and a W-phase H-bridge 730.
  • the controller 9 instructs the dual inverter 200 to operate the single-phase battery under cold conditions where the battery temperature is less than a predetermined value. In this single-phase energization heating, the H-bridge 710 applies a U-phase voltage VU to the U-phase coil 5U, and the H-bridge 720 applies a U-phase voltage VU to the V-phase coil 5V.
  • the H-bridge 730 applies a -U phase voltage (-VU) to the W phase coil 5W.
  • the fundamental frequency component of the U-phase voltage VU and the -U-phase voltage (-VU) is the sinusoidal voltage.
  • the -U phase voltage (-VU) has the opposite phase to the U phase voltage VU.
  • a single-phase AC voltage is applied to the three-phase coil 50.
  • the U-phase current IU flows through the U-phase coil 5U and the V-phase coil 5V, respectively, and the -U-phase current (-IU) flows through the W-phase coil 5W.
  • the -U-phase current (-IU) has the opposite phase to the U-phase current IU. Therefore, the three-phase coil 50 forms a single-phase alternating magnetic field in the three-phase motor.
  • the single-phase alternating magnetic field is a U-phase alternating magnetic field.
  • the three-phase motor does not generate motor torque due to this single-phase alternating magnetic field.
  • the three-phase coil 50 to which the single-phase alternating current is supplied can be regarded as an inductor in essence.
  • the battery 1 supplies the dual inverter 200 with a U-phase current IU having three times the amplitude.
  • FIG. 18 shows the waveforms of the fundamental wave components of the U-phase voltage VU and the U-phase current IU applied to the U-phase coil 5U.
  • the phase difference between the U-phase current IU and the U-phase voltage VU is an electrical angle of 90 degrees.
  • the fact that the U-phase alternating current IU, which is a single-phase alternating current, flows through the battery 1 means that the battery 1 periodically repeats charging and discharging. As a result, the battery 1 generates heat due to its internal resistance.
  • the electrolyte resistance becomes the main resistance component of the battery 1. Therefore, the electrolyte is heated rapidly and the battery resistance is rapidly reduced.
  • the dual inverter 200 can supply a non-sinusoidal single-phase alternating current to the three-phase coil 50.
  • the sinusoidal current is effective in reducing iron loss.
  • the two phase coils (5U, 5V) form a magnetic field in the opposite direction to the phase coil 5W. This makes it possible to prevent the three phase magnetic fields formed by the three phase coils (5U, 5V, and 5W') from canceling each other in the magnetic circuit of the three-phase motor.
  • the dual inverter 200 can supply another alternating current component to the three-phase coil 50 in addition to the single-phase alternating current.
  • the single-phase alternating current is suitable for heating the battery 1, the single-phase alternating current should be the main component of the current supplied by the dual inverter 200 to the three-phase coil 50.
  • This single-phase battery heating method can also be performed immediately after the start of running on a cold winter morning.
  • the dual inverter 200 supplies both the torque generation current and the single-phase alternating current to the three-phase coil 50 after the start of traveling. As a result, the battery 1 can be rapidly heated without generating excessive motor torque.
  • the single-phase battery heating method is performed by a single three-phase inverter connected to a conventional star-connected three-phase coil.
  • This single three-phase inverter consisting of a U-phase leg, a V-phase leg, and a W-phase leg is connected to a star-shaped three-phase coil consisting of a U-phase coil, a V-phase coil, and a W-phase coil.
  • the U-phase leg applies a U-phase voltage VU to the U-phase coil
  • the V-phase leg applies a -U-phase voltage (-VU) to the V-phase coil
  • the W-phase leg applies a -U-phase voltage (-VU) to the W-phase coil.
  • VU) is applied.
  • twice the U-phase voltage is applied to the star-shaped three-phase coil
  • twice the U-phase current is supplied to the star-shaped connected three-phase coil.
  • U-phase voltage VU and -U-phase voltage (-VU) are sinusoidal voltages, both of which have opposite phases.
  • the U-phase current IU which is a single-phase alternating current, flows through the battery 1, and the electrolytic solution of the battery 1 is heated.
  • This single three-phase inverter is essentially equal to the U-phase H-bridge 710, and this star-shaped three-phase coil is essentially equal to the U-phase coil 5U.
  • a single-phase alternating current is supplied to the battery 1 by a conventional single-phase three-phase inverter connected to a star-shaped three-phase coil.
  • the conventional quick charger cannot perform quick charging when the battery 1 is at a low temperature.
  • the electrolyte of the low temperature battery can be rapidly warmed before the start of rapid charging. As a result, the total charging time can be shortened.
  • the conventional external electric heater indirectly heats the electrolytic solution of the battery through a plurality of heat conductive members. Therefore, the temperature rise of the electrolytic solution is delayed.
  • the electrolytic solution is directly heated without heat loss, so that the electrolytic solution can be heated quickly.
  • a plurality of torque control methods for the dual inverter 200 Next, a torque control method for the dual inverter 200 for generating motor torque will be described.
  • the drawback of the grid-connected onboard integrated charger that uses the dual inverter 200 as described above as a three-phase PFC circuit is that the dual inverter 200 requires a complicated circuit topology as compared with a normal single three-phase inverter. Therefore, the dual inverter 200 is required to realize a new merit that overcomes this drawback, for example, reduction of inverter loss.
  • the upper arm conduction single PWM method as the first torque control method will be described with reference to FIG.
  • the leg 3U applies a U-phase voltage VU1 to the phase coil 5U, and the leg 4U applies a U-phase voltage VU2 to the phase coil 5U.
  • the leg 3V applies the V-phase voltage VV1 to the phase coil 5V, and the leg 4V applies the U-phase voltage VV2 to the phase coil 5V.
  • the leg 3W applies a W-phase voltage VW1 to the phase coil 5W
  • the leg 4W applies a W-phase voltage VW2 to the phase coil 5W.
  • the phase difference between the three phase voltages (VU, VV, and VW) is an electrical angle of 120 degrees.
  • the legs 3U and 4U form a U-phase H-bridge that applies a phase voltage VU to the phase coil 5U.
  • the legs 3V and 4V form a V-phase H-bridge that applies a phase voltage VV to the phase coil 5V.
  • the legs 3W and 4W form a W-phase H-bridge that applies a phase voltage VW to the phase coil 5W.
  • each H bridge is composed of a PWM leg and a fixed potential leg, respectively.
  • a PWM leg is a leg that is PWM controlled.
  • the fixed potential leg is a leg that outputs either of the two DC link potentials.
  • FIG. 19 is a timing chart showing the states of the six legs of the dual inverter 200 in the triple H-bridge mode in which the three H-bridges are controlled by the upper arm conduction type single PWM method.
  • the three phase voltage vectors applied to the three-phase coil 50 form a combined rotational voltage vector for forming a rotating magnetic field in the three-phase motor.
  • FIG. 20 shows the U-phase voltages VU1 and VU2.
  • the U-phase voltage VU1 is the fundamental frequency component of the output voltage of the leg 3U
  • the U-phase voltage VU2 is the fundamental frequency component of the output voltage of the leg 4U.
  • One cycle of the U-phase voltage VU is divided into a positive half-wave period PA and a negative half-wave period PB, which are equal to an electric angle of 180 degrees, respectively.
  • the leg 4U in the half-wave period PA, the leg 4U becomes a PWM leg, and the leg 3U becomes a fixed potential leg that outputs a high level DC link voltage (1).
  • the leg 3U In the negative half-wave period PB, the leg 3U becomes a PWM leg, and the leg 4U becomes a fixed potential leg that outputs a high level DC link voltage (1).
  • the upper arm switch of the fixed potential leg is constantly turned on.
  • the fixed potential leg and the PWM leg are alternated every 180 degrees of electrical angle.
  • the PWM leg outputs a low level DC link voltage (0)
  • the PWM leg outputs a high level DC link voltage (1). Is output.
  • the fixed potential leg does not generate switching loss. Therefore, the upper arm conduction single PWM method has half the switching loss as compared with the conventional double PWM method in which the two legs of the H bridge are PWM legs.
  • the upper arm switch of the fixed potential leg is always turned on. Therefore, the ringing surge voltage caused by turning off the upper arm transistor of the inverter can be reduced. As a result, the switching loss can be reduced by shortening the switching transition period.
  • FIGS. 21-24 show the phase period in which the U-phase current IU flows through the U-phase coil 5U.
  • 21 and 22 show a conventional double PWM method in which the legs 3U and 4U are complementarily PWM controlled.
  • 23 and 24 show an upper arm conduction single PWM method in which only the leg 4U is PWM controlled. The upper arm switch and lower arm switch of the PWM leg are switched complementarily.
  • the U-phase current IU consists of a power supply current IP flowing from the battery 1 to the phase coil 5U.
  • switches 32 and 41 are turned on.
  • the U-phase current IU is the regenerative current IR flowing from the phase coil 5U to the battery 1.
  • the U-phase voltage VU applied to the U-phase coil 5U is VU1-VU2
  • the U-phase current IU is IP-IR.
  • the regenerative current IR returning to batteries 1 and 2 for each PWM cycle period increases the high frequency resistance loss of battery 1.
  • the U-phase current IU consists of a power supply current IP flowing from the battery 1 to the phase coil 5U.
  • switches 31 and 41 are turned on.
  • the U-phase current IU is a freewheeling current If that circulates through the phase coil 5U, the switch 41, and the switch 31.
  • the U-phase voltage VU applied to the U-phase coil 5U is VU1-VU2
  • the U-phase current IU is IP + If.
  • the regenerative current IR is prevented from returning to the battery 1. As a result, the loss of the battery 1 is reduced.
  • FIG. 25 shows the fundamental frequency components of the U-phase voltage VU and the U-phase current IU.
  • the U-phase voltage VU is the phase voltage applied to the U-phase coil 5U
  • the U-phase current IU is the phase current flowing through the U-phase coil 5U.
  • the one-cycle period corresponding to the electric angle of 360 degrees is divided into a positive half-wave period PA and a negative half-wave period PB.
  • the amplitude value of the U-phase voltage VU is positive in the period PA and negative in the period PB.
  • the leg 3U becomes a fixed potential leg in the period PA and a PWM leg in the period PB.
  • the leg 4U becomes a PWM leg in the period PA and a fixed potential leg in the period PB.
  • the upper arm switch of the fixed potential leg is turned on, and the lower arm switch of the fixed potential leg is turned off.
  • the U-phase current IU lags the U-phase voltage VU by a predetermined phase period due to the inductance of the U-phase coil 5U.
  • the period PA is divided into phase periods P1 and P2, and the period PB is divided into phase periods P3 and P4.
  • phase periods P1 and P3 the U-phase voltage VU has the opposite direction to the U-phase current IU.
  • the U-phase voltage VU has the same direction as the U-phase current IU.
  • FIG. 26 shows the phase period P1.
  • the freewheeling current If flows, and when the upper arm switch 41 is turned off, the power supply current IP, which is a regenerative current, charges the battery 1.
  • the upper arm switch 41 is turned off, a ringing voltage is generated by the inductance of the high level internal DC bus 810 connecting the two upper arm switches 31 and 41.
  • this high-level internal DC bus 810 has a lower inductance value than the high-level DC bus 81. As a result, this ringing voltage is reduced.
  • FIG. 27 shows the phase period P2.
  • the power supply current IP is supplied to the U-phase coil 5U, and when the lower arm switch 42 is turned off, the freewheeling current If circulates through the U-phase coil 5U. ..
  • FIG. 28 shows the phase period P3.
  • the freewheeling current IP which is a regenerative current
  • the power supply current IP which is a regenerative current
  • FIG. 29 shows the phase period P4.
  • the power supply current IP is supplied to the U-phase coil 5U, and when the lower arm switch 32 is turned off, the freewheeling current If circulates through the U-phase coil 5U. ..
  • Double H-bridge mode A double-H-bridge mode as a second torque control method will be described with reference to FIGS. 30 and 31.
  • this double H-bridge mode one of the three H-bridges of the dual inverter 200 is suspended in turn every 60 degrees of electrical angle.
  • a dormant H-bridge is called a dormant H-bridge.
  • the upper and lower arm switches on each of the two legs of the dormant H-bridge are turned off.
  • This double H-bridge mode which suspends one of the three H-bridges, reduces the loss of the dual inverter 200.
  • the dual inverter 200 applies a combined rotational voltage vector to the three-phase coil 50 in order to generate motor torque.
  • This combined rotational voltage vector forms a combined rotational current vector, and this combined rotational current vector forms a rotating magnetic field vector.
  • the combined rotational voltage vector is a vector of the U-phase voltage vector VU applied to the U-phase coil 5U, the V-phase voltage vector VV applied to the V-phase coil 5V, and the W-phase voltage vector VW applied to the W-phase coil 5W. It is a sum.
  • the combined rotational voltage vector is formed by the vector sum of the phase voltage vectors of two adjacent phases.
  • the double H-bridge mode is adopted in the low voltage operating region where the maximum amplitude of the combined rotational voltage vector is less than or equal to the maximum amplitude of each phase voltage vector. Therefore, the double H bridge mode is suitable in the medium and low speed region of the three-phase motor in which the back electromotive force Vemmf of the three-phase coil 50 is relatively low.
  • one of the three H-bridges is paused in turn every 60 degrees of electrical angle, and the remaining two H-bridges are PWM controlled.
  • FIG. 30 is a vector diagram for showing the existing region of the combined rotational voltage vector in the double H bridge mode.
  • the existing region of the combined rotational voltage vector is divided into seven phase regions Z0-Z6.
  • the combined rotational voltage vector is the vector sum of the U-phase voltage vector (VU) and the -W-phase voltage vector (-VW).
  • the V-phase H-bridge is suspended.
  • the combined rotation voltage vector is the vector sum of the V-phase voltage vector (VV) and the -W-phase voltage vector (-VW).
  • the U-phase H-bridge is suspended.
  • the combined rotational voltage vector is the vector sum of the V-phase voltage vector (VV) and the -U-phase voltage vector (-VU).
  • the W-phase H-bridge is suspended.
  • the combined rotational voltage vector is the vector sum of the W phase voltage vector (VW) and the -U phase voltage vector (-VU).
  • the V-phase H-bridge is suspended.
  • the combined rotation voltage vector is the vector sum of the W phase voltage vector (VW) and the -V phase voltage vector (-VV).
  • the U-phase H-bridge is suspended.
  • the combined rotational voltage vector is the vector sum of the U-phase voltage vector (VU) and the -V-phase voltage vector (-VV).
  • the W-phase H-bridge is suspended.
  • the U-phase H-bridge consisting of legs 3U and 4U outputs the phase voltage vectors VU and -VU.
  • the V-phase H-bridge consisting of legs 3V and 4V outputs the phase voltage vectors VV and -VV.
  • the W-phase H-bridge consisting of legs 3W and 4W outputs the phase voltage vectors VW and -VW.
  • each of the six phase voltage vectors (VU, -VW, VV, -VU, VW, -VV) has a maximum amplitude value.
  • the maximum amplitude of the combined rotational voltage vector equal to the radius of the inner circle shown by the dashed line in FIG. 30 is equal to the maximum amplitude of one phase voltage vector in this double H-bridge mode. Therefore, the double H-bridge mode is performed inside this inner circle.
  • the triple H-bridge mode, in which each of the three H-bridges is PWM controlled, is executed in the phase region Z0 between the inner circle and the outer circle shown by the solid line.
  • each of the three H-bridges forms a phase voltage vector, and the vector sum of the three phase voltage vectors becomes the combined rotational voltage vector.
  • Hibernation H-bridge PWM control outage reduces battery loss. Switching between each phase region can be performed slowly to suppress abrupt changes in phase current. In other words, the triple H-bridge mode may be executed for a short time during the transient period in which switching between each phase region is performed.
  • FIG. 31 is a timing chart showing this double H bridge mode.
  • Each of the three H-bridges is controlled by an upper arm conduction single PWM.
  • the U-phase H-bridge is rested during a rest period with an electrical angle of 330-30 degrees and an electrical angle of 150 ° -210 degrees.
  • the V-phase H-bridge is rested during a rest period with an electrical angle of 90-150 degrees and an electrical angle of 270-330 degrees.
  • the W-phase H-bridge is rested during a rest period with an electrical angle of 30-90 degrees and an electrical angle of 210-270 degrees.
  • the H-bridge that outputs the phase voltage of the minimum amplitude becomes the dormant H-bridge.
  • the upper arm switch and lower arm switch of the dormant H-bridge are turned off, and the boost-switched battery 100 does not supply power current to the dormant H-bridge.
  • Boosted Double H-Bridge Mode A boosted double H-bridge mode as a third torque control method will be described with reference to FIGS. 32 and 33.
  • PWM switching of the fixed potential H-bridge is stopped.
  • the fixed potential H-bridge outputs the phase voltage with the maximum amplitude.
  • This double H-bridge mode which suspends one of the three H-bridges, reduces the loss of the dual inverter 200.
  • the step-up switched battery 100 shown in FIG. 1 applies a step-up DC link voltage VC to the dual inverter 200.
  • the U-phase H-bridge of the dual inverter 200 applies a U-phase voltage VU to the U-phase coil 5U.
  • the V-phase H-bridge of the dual inverter 200 applies a V-phase voltage VV to the V-phase coil 5U.
  • the W-phase H-bridge of the dual inverter 200 applies a W-phase voltage VW to the W-phase coil VW.
  • the two legs of the fixed-potential H-bridge that output the phase voltage of maximum amplitude are both fixed-potential legs. These two fixed potential legs output opposite voltages.
  • One of the two fixed potential legs outputs a high level DC link voltage and the other outputs a low level DC link voltage.
  • the output potentials of the two fixed potential legs are alternated every 180 degrees of electrical angle.
  • FIG. 32 is a timing chart showing the fundamental frequency components of the three phase voltages VU, VV, and VW.
  • the U-phase fixed phase period (electrical angles 60 ° -120 ° and 240 ° -300 °) where the amplitude of the U-phase voltage VU is maximized, the U-phase H-bridge becomes the potential-fixed H-bridge.
  • the V-phase fixed phase period (electrical angle 180 ° -240 ° and 0 ° -60 °) where the amplitude of the V-phase voltage VV is maximized, the V-phase H-bridge becomes the potential-fixed H-bridge.
  • the W phase H bridge becomes the potential fixed H bridge.
  • the length of the fixed phase period of each phase arranged in order is an electrical angle of 60 degrees.
  • the upper arm switch 31 and the lower arm switch 42 are turned on during the U-phase fixed phase period in which the U-phase voltage VU has a positive value.
  • the upper arm switch 41 and the lower arm switch 32 are turned on during the U-phase fixed potential period in which the U-phase voltage VU has a negative value.
  • the upper arm switch 33 and the lower arm switch 44 are turned on during the V-phase fixed phase period in which the V-phase voltage VV has a positive value.
  • the upper arm switch 43 and the lower arm switch 34 are turned on during the V-phase fixed potential period in which the V-phase voltage VV has a negative value.
  • the upper arm switch 35 and the lower arm switch 46 are turned on in the W phase fixed phase period in which the W phase voltage VW has a positive value.
  • the upper arm switch 45 and the lower arm switch 36 are turned on during the W phase fixed potential period in which the W phase voltage VW has a negative value.
  • the front end boost converter 10 boosts the battery voltage.
  • the boosted DC link voltage VC is equal to the U-phase voltage VU in the U-phase fixed phase period, equal to the V-phase voltage VV in the V-phase fixed phase period, and equal to the W-phase voltage VW in the W-phase fixed phase period. ..
  • the front end boost converter 10 applies a DC link voltage VC having a waveform equal to the rectified three-phase sinusoidal voltage to the dual inverter 200.
  • the front end boost converter 10 is PWM controlled instead of the fixed potential H bridge.
  • the DC link voltage VC has ripples. Therefore, the PWM duty ratios of the other two H-bridges except the fixed potential H-bridge are corrected to eliminate the influence of the ripple of this DC link voltage VC.
  • FIG. 33 is a timing chart showing the states of the three H bridges.
  • the U-phase H-bridge becomes the potential-fixed H-bridge.
  • the V-phase H-bridge becomes the potential-fixed H-bridge.
  • the W-phase H-bridge becomes the potential-fixed H-bridge.
  • the other two H-bridges except the fixed potential H-bridge are operated by the upper arm conduction single PWM method described above.
  • This step-up double H-bridge mode utilizing the step-up operation of the converter 10 is executed in a high-speed operation region where the output voltage of the fixed potential H-bridge is higher than the battery voltage.
  • FIG. 34 is a timing chart showing the state of each leg of the dual inverter 200 driven in the single H bridge mode.
  • the high level period'H' is the phase period in which each leg outputs the high level potential
  • the low level period'L' is the phase period in which each leg outputs the low level potential.
  • the phase period'PWM' is the period during which each leg is PWM controlled.
  • Each H-bridge is controlled by the upper arm conduction single PWM method in each phase period'PWM'.
  • the boost-switched battery 100 applies a DC link voltage VC equal to the phase voltage of maximum amplitude to the fixed potential H-bridge.
  • the double H-bridge mode is executed, and the paused H-bridge that should output the phase voltage of the minimum amplitude is paused. After all, only one of the three H-bridges and the converter 10 are PWM controlled. This reduces the switching loss of the three-phase motor drive.
  • the dual inverter 200 is controlled by the space vector PWM method (SVPWM method). This current distribution method reduces battery loss.
  • the three H-bridges of the dual inverter 200 are PWM controlled based on a PWM carrier signal having a predetermined frequency value. For example, when the frequency of the PWM carrier signal is 20kHz, the common PWM cycle period TC has a length of 50 ⁇ sec.
  • each H-bridge has a current supply period and a freewheeling period arranged within a common PWM cycle period TC.
  • This current supply period is a period in which the step-up switched battery 100 supplies the power supply current to the H bridge.
  • the freewheeling period is the period in which the freewheeling current circulates between the H-bridge and the phase coil.
  • the U-phase voltage vector VU is approximately proportional to the length of the current supply period of the U-phase H-bridge
  • the V-phase voltage vector VV is approximately proportional to the length of the current supply period of the V-phase H-bridge
  • the W-phase voltage vector VW is. It is almost proportional to the length of the current supply period of the W-phase H-bridge.
  • each H-bridge can be PWM-controlled independently.
  • the current supply periods of each phase can be freely arranged with each other within a common PWM cycle period TC.
  • the current supply periods of each phase are arranged in a common PWM cycle period TC so as not to overlap each other as much as possible. Thereby, the loss of the step-up switched battery 100 can be reduced.
  • FIG. 35 is a timing chart showing one PWM cycle period TC in the triple H-bridge mode in which three H-bridges are PWM-controlled.
  • the current supply period TXV of the V-phase H-bridge, the current supply period TXW of the W-phase H-bridge, and the current supply period TXU of the U-phase H-bridge are sequentially arranged in the common PWM cycle period TC. This essentially avoids the overlap of the three current supply periods TXV, TXW, TXU.
  • the boost-switched battery 100 supplies the U-phase power supply current IUP to the U-phase H-bridge during the period TXU, supplies the V-phase power supply current IVP to the V-phase H-bridge during the period TXV, and W to the W-phase H-bridge during the period TXW.
  • Supply phase power current IWP Supply phase power current
  • the current supply period of each phase is arranged continuously. As a result, the high frequency current component included in the power supply current supplied from the DC power supply 100 to the dual inverter 200 is reduced.
  • the longest current supply period TXW is sandwiched between the other two current supply periods TXV and TXU. As a result, the high frequency current component flowing through the step-up switched battery 100 is reduced.
  • the two adjacent current supply periods overlap each other by a short transient period Tt. As a result, the high frequency current component flowing through the step-up switched battery 100 is reduced.
  • FIG. 36 is a timing chart showing one PWM cycle period TC in the double H bridge mode or the step-up double H bridge mode.
  • the resistance loss of the step-up switched battery 100 is reduced.
  • the shorter current supply period is placed immediately after the longer current supply period. As a result, the sudden decrease in the power supply current is suppressed, and the ringing surge voltage is reduced. After all, according to the current distribution method, the resistance loss of the step-up switched battery 100 is reduced.
  • Each torque control method described above is preferably switched according to the motor rotation speed. It is also possible to select the optimum torque control method according to other parameters such as torque command value and efficiency.
  • the current dispersion method is very effective in the low speed region where the back electromotive force (back EMF) of the three-phase coil 50 is low.
  • the PWM duty ratio of each phase is reduced by this low back EMF. Therefore, the current supply periods of the respective phases can be arranged within each PWM cycle period without overlapping each other.

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Inverter Devices (AREA)
  • Control Of Ac Motors In General (AREA)
  • Electric Propulsion And Braking For Vehicles (AREA)
  • Charge And Discharge Circuits For Batteries Or The Like (AREA)

Abstract

Dans la présente invention, un onduleur double est connecté à la bobine triphasée à extrémité ouverte d'un moteur triphasé. La bobine triphasée est connectée à un réseau triphasé et à une alimentation en courant continu. L'onduleur triphasé côté batterie de l'onduleur double fonctionne comme un redresseur élévateur pour charger une batterie par un courant de réseau triphasé. L'onduleur triphasé côté réseau de l'onduleur double fonctionne comme un redresseur pour charger la batterie par courant continu. Un convertisseur élévateur frontal disposé entre la batterie et l'onduleur double peut être utilisé comme hacheur abaisseur pour commander le courant de charge de la batterie. Le convertisseur élévateur frontal précharge un condensateur de lissage par le courant continu d'une batterie basse tension. L'onduleur commande un courant alternatif monophasé qui circule à partir de la batterie à une température très basse vers la bobine triphasée.
PCT/JP2019/035612 2019-09-11 2019-09-11 Circuit de commutation de puissance WO2021048935A1 (fr)

Priority Applications (2)

Application Number Priority Date Filing Date Title
JP2020573554A JP7113095B2 (ja) 2019-09-11 2019-09-11 パワースイッチング回路
PCT/JP2019/035612 WO2021048935A1 (fr) 2019-09-11 2019-09-11 Circuit de commutation de puissance

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
PCT/JP2019/035612 WO2021048935A1 (fr) 2019-09-11 2019-09-11 Circuit de commutation de puissance

Publications (1)

Publication Number Publication Date
WO2021048935A1 true WO2021048935A1 (fr) 2021-03-18

Family

ID=74866217

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/JP2019/035612 WO2021048935A1 (fr) 2019-09-11 2019-09-11 Circuit de commutation de puissance

Country Status (2)

Country Link
JP (1) JP7113095B2 (fr)
WO (1) WO2021048935A1 (fr)

Cited By (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
KR20230173901A (ko) * 2022-06-20 2023-12-27 비테스코 테크놀로지스 게엠베하 차량용 인버터 시스템 및 그 제어 방법
JP7407247B1 (ja) * 2022-07-28 2023-12-28 正一 田中 バッテリ保護回路
WO2024087398A1 (fr) * 2022-10-26 2024-05-02 广汽埃安新能源汽车股份有限公司 Circuit de chauffage de batterie à base de transformateur, et véhicule électrique

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2004248335A (ja) * 2002-12-19 2004-09-02 Honda Motor Co Ltd モータ制御装置
JP2012509656A (ja) * 2008-11-18 2012-04-19 ヴァレオ システム ドゥ コントロール モトゥール 給電および充電に共用される電気装置
JP2013077452A (ja) * 2011-09-30 2013-04-25 Toyota Central R&D Labs Inc 電源システム
JP2017208954A (ja) * 2016-05-19 2017-11-24 田中　正一 巻数切換型電気機械

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2004248335A (ja) * 2002-12-19 2004-09-02 Honda Motor Co Ltd モータ制御装置
JP2012509656A (ja) * 2008-11-18 2012-04-19 ヴァレオ システム ドゥ コントロール モトゥール 給電および充電に共用される電気装置
JP2013077452A (ja) * 2011-09-30 2013-04-25 Toyota Central R&D Labs Inc 電源システム
JP2017208954A (ja) * 2016-05-19 2017-11-24 田中　正一 巻数切換型電気機械

Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
KR20230173901A (ko) * 2022-06-20 2023-12-27 비테스코 테크놀로지스 게엠베하 차량용 인버터 시스템 및 그 제어 방법
KR102748614B1 (ko) * 2022-06-20 2024-12-30 비테스코 테크놀로지스 게엠베하 차량용 인버터 시스템 및 그 제어 방법
JP7407247B1 (ja) * 2022-07-28 2023-12-28 正一 田中 バッテリ保護回路
WO2024087398A1 (fr) * 2022-10-26 2024-05-02 广汽埃安新能源汽车股份有限公司 Circuit de chauffage de batterie à base de transformateur, et véhicule électrique

Also Published As

Publication number Publication date
JPWO2021048935A1 (ja) 2021-09-27
JP7113095B2 (ja) 2022-08-04

Similar Documents

Publication Publication Date Title
JP7191951B2 (ja) 電気自動車のパワーシステム
US11799292B2 (en) On-board bidirectional AC fast charger for electric vehicles
US7859201B2 (en) Charge control apparatus, electrically powered vehicle and electric storage charge control method
US11970067B2 (en) Constant current fast charging of electric vehicles via DC grid using dual inverter drive
US10562404B1 (en) Integrated onboard chargers for plug-in electric vehicles
JP7110099B2 (ja) １又は複数の誘導巻線を有する誘導負荷用のコントローラ
EP3628109B1 (fr) Systèmes et procédés pour chargeur rapide embarqué
US7891451B2 (en) Power controller and vehicle equipped with power controller
CN110971173A (zh) 动力电池的充电方法、电机控制电路及车辆
US9914362B2 (en) Energy storage for power factor correction in battery charger for electric-powered vehicles
WO2018227307A1 (fr) Charge rapide à courant constant de véhicules électriques par le biais d'un réseau de distribution de courant continu en utilisant un circuit d'attaque à double onduleur
WO2021019608A1 (fr) Entraînement de moteur triphasé
US20240186914A1 (en) Device for creating a dc voltage bus for a polyphase electrical system, motor vehicle and renewable energy generator comprising such a device
WO2021048935A1 (fr) Circuit de commutation de puissance
JP2023114972A (ja) モーター駆動システムを用いた車両用バッテリー充電システム
WO2024207830A1 (fr) Système de chauffage de batterie et camion électrique
Shah et al. Integrated power converter with G2V and V2G capabilities for 4-phase SRM drive based EV application
KR20090097030A (ko) 전기 자동차용 충전 및 구동 장치와 그의 충전방법
Niakinezhad et al. A new modular asymmetrical half-bridge switched reluctance motor integrated drive for electric vehicle application
JP3306326B2 (ja) コンデンサ電力貯蔵装置
WO2024172050A1 (fr) Dispositif de charge embarqué et procédé de charge embarqué
Nakatomi et al. Integration of Motor Drive and Charger System with Two-phase Motor and H-Bridge Inverter for Electric Vehicle
JP2024115455A (ja) 車載充電装置
JP2023160230A (ja) 低電圧バッテリ充電システム
CN117730477A (zh) 利用电机绕组的动态可重构电力转换器

Legal Events

Date Code Title Description
ENP Entry into the national phase

Ref document number: 2020573554

Country of ref document: JP

Kind code of ref document: A

121 Ep: the epo has been informed by wipo that ep was designated in this application

Ref document number: 19944800

Country of ref document: EP

Kind code of ref document: A1

NENP Non-entry into the national phase

Ref country code: DE

122 Ep: pct application non-entry in european phase

Ref document number: 19944800

Country of ref document: EP

Kind code of ref document: A1