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WO2019021983A1 - High frequency filter, multiplexer, high frequency front-end circuit, and communication device - Google Patents

High frequency filter, multiplexer, high frequency front-end circuit, and communication device Download PDF

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Publication number
WO2019021983A1
WO2019021983A1 PCT/JP2018/027433 JP2018027433W WO2019021983A1 WO 2019021983 A1 WO2019021983 A1 WO 2019021983A1 JP 2018027433 W JP2018027433 W JP 2018027433W WO 2019021983 A1 WO2019021983 A1 WO 2019021983A1
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WO
WIPO (PCT)
Prior art keywords
circuit
band
filter
high frequency
node
Prior art date
Application number
PCT/JP2018/027433
Other languages
French (fr)
Japanese (ja)
Inventor
浩司 野阪
Original Assignee
株式会社村田製作所
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by 株式会社村田製作所 filed Critical 株式会社村田製作所
Priority to CN201880049640.8A priority Critical patent/CN110999081B/en
Priority to CN202311040641.5A priority patent/CN117060880A/en
Publication of WO2019021983A1 publication Critical patent/WO2019021983A1/en
Priority to US16/746,996 priority patent/US11283428B2/en

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Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/64—Filters using surface acoustic waves
    • H03H9/6423—Means for obtaining a particular transfer characteristic
    • H03H9/6433—Coupled resonator filters
    • H03H9/6483—Ladder SAW filters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/64—Filters using surface acoustic waves
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H7/00—Multiple-port networks comprising only passive electrical elements as network components
    • H03H7/01—Frequency selective two-port networks
    • H03H7/0153—Electrical filters; Controlling thereof
    • H03H7/0161—Bandpass filters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
    • H03H9/02—Details
    • H03H9/125—Driving means, e.g. electrodes, coils
    • H03H9/145—Driving means, e.g. electrodes, coils for networks using surface acoustic waves
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/54—Filters comprising resonators of piezoelectric or electrostrictive material
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/64—Filters using surface acoustic waves
    • H03H9/6403—Programmable filters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
    • H03H9/46—Filters
    • H03H9/64—Filters using surface acoustic waves
    • H03H9/6423—Means for obtaining a particular transfer characteristic
    • H03H9/6433—Coupled resonator filters
    • H03H9/6436—Coupled resonator filters having one acoustic track only
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
    • H03H9/66—Phase shifters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
    • H03H9/70—Multiple-port networks for connecting several sources or loads, working on different frequencies or frequency bands, to a common load or source
    • H03H9/703—Networks using bulk acoustic wave devices
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
    • H03H9/70—Multiple-port networks for connecting several sources or loads, working on different frequencies or frequency bands, to a common load or source
    • H03H9/72—Networks using surface acoustic waves
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H7/00—Multiple-port networks comprising only passive electrical elements as network components
    • H03H7/18—Networks for phase shifting
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H7/00—Multiple-port networks comprising only passive electrical elements as network components
    • H03H7/18—Networks for phase shifting
    • H03H7/19—Two-port phase shifters providing a predetermined phase shift, e.g. "all-pass" filters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
    • H03H9/66—Phase shifters
    • H03H9/68—Phase shifters using surface acoustic waves

Definitions

  • the present invention relates to a high frequency filter, a multiplexer, a high frequency front end circuit and a communication device.
  • a wireless receiving circuit including a plurality of band pass filters having different pass bands connected in parallel to one another and a switch element connected to a plurality of band pass filters connected in parallel.
  • a switch element connected to a plurality of band pass filters connected in parallel.
  • An object of the present invention is to provide a low loss wide band high frequency filter, multiplexer, high frequency front end circuit and communication device in which ripple (insertion loss deviation) in the passband is reduced.
  • a high frequency filter is a high frequency filter having a first band as a pass band, and is provided on a path connecting a first input / output terminal and a second input / output terminal.
  • a first node provided, and a first circuit connected to a second node provided between the first node on the path and the second input / output terminal, the first node and the second
  • a second circuit connected to a node, wherein the first circuit is a band including a part of the frequency of the first band and passes through a second band having a narrower bandwidth than the first band
  • a second filter is a band including a part of the frequency of the first band and has a narrower bandwidth than the first band, and the second band.
  • a high frequency filter having a first wide band as a pass band by arranging a first filter and a second filter in parallel between a first node and a second node and combining the first filter and the second filter. It is assumed that ripples occur in the first band due to the difference in amplitude, phase and impedance of the two filters.
  • the phase difference and the impedance can be made to correspond to the amplitude difference between the two filters. It becomes possible to adjust. Therefore, it is possible to realize a low loss, wide band high frequency filter in which the first band is a pass band and ripples in the pass band are reduced.
  • the frequency at which the impedance of the second circuit is maximized among the singular points at which the impedance of the second circuit is maximized is the second frequency.
  • the phases of the first phase shifter and the second phase shifter are adjusted so as to be included in the range from the frequency at the lower end of the two bands to the cutoff frequency on the lower side of the pass band of the second filter. May be
  • a frequency at which the first circuit and the second circuit have the same phase in the second band there may be at least one location, and there may be at least one frequency where the first circuit and the second circuit are in phase in the third band.
  • the sum of the phase difference between the first filter and the second filter and the sum of phase shifts of the first phase shifter and the second phase shifter is The phase shift of the first phase shifter and the second phase shifter may be adjusted so as to be 312 ° or more and 362 ° or less.
  • the path is a first path passing through the first node, the first circuit, and the second node, and a second path passing through the first node, the second circuit, and the second node.
  • the first filter includes: a first series arm circuit provided on the first path; and a first parallel arm circuit connected to a node provided on the first path and a ground.
  • a second series arm circuit provided on the second path connecting the first phase shifter and the second phase shifter, and the second filter provided on the second path A second parallel arm circuit connected to the selected node and the ground, the first series arm circuit, the first parallel arm circuit, the second series arm circuit, and the second parallel arm circuit At least one of the plurality of elastic wave resonators, and at least one of the elastic wave resonators And an IDT (InterDigital Transducer) electrode formed on a substrate having piezoelectricity, and a reflector, and the wavelength of the elastic wave determined by the electrode period of the IDT electrode in at least one of the elastic wave resonators In the case of ⁇ , the pitch between the IDT electrode and the reflector may be 0.42 ⁇ or more and less than 0.50 ⁇ .
  • An elastic wave resonator having an IDT electrode and a reflector is constituted by a periodic structure consisting of periodically arranged electrode fingers, and has a frequency band that reflects a surface acoustic wave in a specific frequency range with a high reflection coefficient.
  • This frequency band is generally referred to as a stop band, and is defined by the repetition period of the periodic structure or the like.
  • ripples in which the reflection coefficient locally increases are generated.
  • the pitch (IR pitch) between the IDT electrode and the reflector is 0.5 ⁇ or more, the ripple at the high end of the stop band becomes large, the ripple in the pass band of the filter becomes large, and the insertion loss Becomes larger.
  • the IR pitch is 0.42 ⁇ or more and less than 0.50 ⁇ , it is possible to reduce the ripple at the high end of the stop band. From the above point of view, by setting the IR pitch to be 0.42 ⁇ or more and less than 0.50 ⁇ , the insertion loss in the pass band of the high frequency filter can be reduced.
  • the first circuit further includes a first switch element connected between the first node and the first filter, and a second switch element connected between the second node and the first filter.
  • a switch element may be provided, and switching between conduction and non-conduction of the first switch element and the second switch element may switch between the first band and the third band.
  • variable-frequency high frequency filter capable of switching between the first band and the third band by switching between conduction and non-conduction of the first switch element and the second switch element.
  • the first circuit is further connected between a connection node between the first switch element and the first filter, and the ground, and conduction and non-conduction are exclusively performed with the first switch element.
  • the third switch element to be switched, the connection node between the second switch element and the first filter, and the ground are connected, and conduction and non-conduction are switched exclusively to the second switch element.
  • a fourth switch element is provided.
  • the second circuit is further connected between a fifth switch element connected between the first node and the first phase shifter, and between the second node and the second phase shifter.
  • the first band and the second band may be switched by switching between conduction and non-conduction of the fifth switch element and the sixth switch element.
  • variable-frequency high frequency filter capable of switching between the first band and the second band by switching between conduction and non-conduction of the fifth switch element and the sixth switch element.
  • the second circuit is further connected between a connection node between the fifth switch element and the first phase shifter and the ground, and is conductive and non-conductive exclusively with the fifth switch element. It is connected between a seventh switch element whose conduction is switched, a connection node between the sixth switch element and the second phase shifter, and the ground, and is conductive and non-conductive exclusively with the sixth switch element. And an eighth switch element whose conduction is switched.
  • a third circuit connected to the first node and the second node is provided, and the third circuit is a band including a part of the frequency of a fourth band different from the first band,
  • a third filter that has a narrower bandwidth than the fourth band, is located on a higher frequency side than the center frequency of the second band, and uses a fifth band different from the third band as a pass band;
  • a third phase shifter connected to one terminal of the third filter and a fourth phase shifter connected to the other terminal of the third filter may be provided.
  • the first phase shifter includes one or more inductors arranged in a series arm path connecting the first node and one terminal of the second filter, a node on the series arm path, and a ground
  • a low pass filter circuit having a capacitor connected between the one or more capacitors arranged in the series arm path, and an inductor connected to a node on the series arm path and a ground It may be a high pass filter circuit.
  • the circuit configuration of the first phase shifter can be appropriately selected as either the low pass filter circuit or the high pass filter circuit according to the required attenuation characteristic.
  • the second phase shifter includes at least one inductor disposed in a series arm path connecting the second node and the other terminal of the second filter, and a node and a ground on the series arm path.
  • a low pass filter circuit composed of connected capacitors, or one or more capacitors arranged in the series arm path, and an inductor connected to a node on the series arm path and the ground It may be a high pass filter circuit.
  • the circuit configuration of the second phase shifter can be appropriately selected as either the low pass filter circuit or the high pass filter circuit according to the required attenuation characteristic.
  • the first phase shifter and the second phase shifter may be impedance elements formed of a capacitor or an inductor.
  • the circuit configuration of the high frequency filter can be simplified and miniaturization can be achieved.
  • At least one of the first filter and the second filter is connected to a series arm circuit disposed on the path connecting the first node and the second node, and to a node on the path and a ground.
  • a parallel arm circuit, at least one of the series arm circuit and the parallel arm circuit includes a resonator and a ninth switch element, and switching between conduction and non-conduction of the ninth switch element The resonant frequency and / or the antiresonant frequency of at least one of the series arm circuit and the parallel arm circuit may be switched.
  • At least one of the first filter and the second filter is any of a surface acoustic wave filter, a boundary acoustic wave filter, and an elastic wave filter using a bulk acoustic wave (BAW). It is also good.
  • At least one of the first filter and the second filter may have a longitudinally coupled resonator.
  • the sharpness can be improved and the attenuation can be improved.
  • phase of the first phase shifter may be different from the phase of the second phase shifter.
  • a multiplexer includes a plurality of filters including the high frequency filter according to any one of the above, and the input terminals or the output terminals of the plurality of filters are connected directly or indirectly to a common terminal. It is done.
  • a high frequency front end circuit includes the high frequency filter according to any one of the above or the multiplexer according to the above, and an amplifier circuit directly or indirectly connected to the high frequency filter or the multiplexer. Equipped with
  • a communication apparatus includes: an RF signal processing circuit processing high frequency signals transmitted and received by an antenna element; and transmitting the high frequency signal between the antenna element and the RF signal processing circuit. And the described high frequency front end circuit.
  • FIG. 1A is a circuit block diagram of a high frequency filter according to a first embodiment.
  • FIG. 1B is a diagram showing the concept of increasing the bandwidth of the high frequency filter according to the first embodiment.
  • FIG. 2 is a circuit diagram of the high frequency filter according to the first embodiment.
  • FIG. 3A is a circuit block diagram of a high frequency filter according to Comparative Example 1.
  • FIG. 3B is a circuit block diagram of a high frequency filter according to Comparative Example 2.
  • FIG. 4A is a graph showing the pass characteristic, the amplitude characteristic, the phase characteristic, and the impedance characteristic of the high frequency filter according to Comparative Example 1.
  • FIG. 4B is a graph showing the pass characteristic, the amplitude characteristic, the phase characteristic, and the impedance characteristic of the high frequency filter according to Comparative Example 2.
  • FIG. 5 is a graph showing pass characteristics, amplitude characteristics, phase characteristics, and impedance characteristics of the high frequency filter according to the first embodiment.
  • FIG. 6 is a graph for explaining the factor of broadening the band of the high frequency filter according to the first embodiment.
  • FIG. 7 is a graph showing pass characteristics, phase characteristics, and impedance characteristics when the phase of the phase shifter of the high frequency filter according to the first embodiment is changed.
  • FIG. 8 is a graph showing the relationship between the phase, the insertion loss, and the maximum impedance of the phase shifter of the high frequency filter according to the first embodiment.
  • FIG. 9 is a circuit diagram of a high frequency filter according to Examples 1a, 1b, 1c, 1d and 1e.
  • FIG. 10 is an example of a diagram schematically showing the structure of the parallel arm resonator in the first embodiment.
  • FIG. 11A is a graph comparing the pass characteristic and the reflection characteristic of the high frequency filters according to Example 1a and Example 1e.
  • FIG. 11B is a graph comparing impedance characteristics and reflection characteristics of series arm resonators according to Example 1a and Example 1e.
  • FIG. 12A is a graph comparing the pass characteristics of the high frequency filters according to Example 1 b and Example 1 e.
  • FIG. 12B is a graph comparing impedance characteristics and reflection characteristics of parallel arm resonators according to Example 1 b and Example 1 e.
  • FIG. 13A is a graph comparing the pass characteristics of the high frequency filters according to Example 1c and Example 1e.
  • FIG. 11A is a graph comparing the pass characteristic and the reflection characteristic of the high frequency filters according to Example 1a and Example 1e.
  • FIG. 11B is a graph comparing impedance characteristics and reflection characteristics of series arm
  • FIG. 13B is a graph comparing impedance characteristics and reflection characteristics of parallel arm resonators according to Example 1 c and Example 1 e.
  • FIG. 14 is a graph comparing the pass characteristic and the reflection characteristic of the high frequency filters according to Example 1 d and Example 1 e.
  • FIG. 15A is a graph showing a change in characteristics when the IR pitch is changed at 0.40 ⁇ to 0.50 ⁇ in the resonator of the typical example.
  • FIG. 15B is a graph showing a change in characteristics when the IR pitch is changed at 0.50 ⁇ to 0.60 ⁇ in the resonator of the typical example.
  • FIG. 16 is a circuit block diagram of the high frequency filter according to the second embodiment.
  • FIG. 17A is a circuit configuration diagram of a high frequency filter according to a second embodiment.
  • FIG. 17B is a graph showing a change in pass characteristics due to the switching of the switch of the high frequency filter according to the second embodiment.
  • FIG. 18 is a graph for explaining the factor of increasing the bandwidth of the high frequency filter according to the second embodiment.
  • FIG. 19 is a graph showing the pass characteristic when the switch-off capacity of the high frequency filter according to the second embodiment is changed.
  • FIG. 20 is a circuit block diagram of the high frequency filter according to the third embodiment.
  • FIG. 21 is a graph showing the pass characteristic of the high frequency filter according to the third embodiment.
  • FIG. 22A is a circuit configuration diagram of a high frequency filter according to Modification 1 of Embodiment 2.
  • FIG. 22A is a circuit configuration diagram of a high frequency filter according to Modification 1 of Embodiment 2.
  • FIG. 22B is a circuit configuration diagram of a high frequency filter according to a second modification of the second embodiment.
  • FIG. 23 is a graph comparing the pass characteristics of the high frequency filters according to the first and second modifications of the second embodiment.
  • FIG. 24A is a circuit configuration diagram of a high frequency filter according to a fourth embodiment.
  • FIG. 24B is a graph showing the change of the passage characteristic due to the switch of the high frequency filter according to the fourth embodiment.
  • FIG. 25 is a graph for explaining the factor of increasing the band in the first band of the high frequency filter according to the fourth embodiment.
  • FIG. 26 is a graph for explaining the factor of increasing the band in the 1a band of the high frequency filter according to the fourth embodiment.
  • FIG. 27A is a circuit configuration diagram of a high frequency filter according to a fifth embodiment.
  • FIG. 27B is a graph showing a change in pass characteristics due to switching of a switch of the high frequency filter according to the fifth embodiment.
  • FIG. 28 is a graph for explaining the factor of increasing the bandwidth of the high frequency filter according to the fifth embodiment.
  • FIG. 29A is a circuit configuration diagram of a high frequency filter according to a sixth embodiment.
  • FIG. 29B is a graph showing a change in passage characteristics due to the switch of the high frequency filter according to the sixth embodiment.
  • FIG. 30 is a graph for explaining the factor of increasing the band in the first band of the high frequency filter according to the sixth embodiment.
  • FIG. 31 is a graph for explaining the factor of increasing the band in the 1b band of the high frequency filter according to the sixth embodiment.
  • 32A is a circuit configuration diagram of Modification Example 1 of each filter configuring the high-frequency filter according to Example 6.
  • 32B is a circuit configuration diagram of Modification Example 2 of each of the filters forming the high-frequency filter according to Example 6.
  • FIG. 32C is a circuit configuration diagram of Modification Example 3 of each of the filters forming the high-frequency filter according to Example 6.
  • FIG. 32D is a circuit configuration diagram of Modification Example 4 of each filter configuring the high-frequency filter according to Example 6.
  • FIG. 33A is a circuit configuration diagram of Modified Example 5 of each filter configuring the high frequency filter according to Example 6.
  • FIG. 33B is a circuit configuration diagram of Modified Example 6 of each filter configuring the high frequency filter according to Example 6.
  • FIG. 33C is a circuit configuration diagram of Modified Example 7 of each filter configuring the high frequency filter according to Example 6.
  • FIG. 33D is a circuit configuration diagram of Modified Example 8 of the filters configuring the high frequency filter according to Example 6.
  • FIG. 33E is a circuit configuration diagram of Modification 9 of the filters configuring the high-frequency filter according to the sixth embodiment.
  • FIG. 33F is a circuit configuration diagram of Modified Example 10 of the filters configuring the high-frequency filter according to Example 6.
  • FIG. 33G is a circuit configuration diagram of Modification 11 of the filters configuring the high-frequency filter according to Embodiment 6.
  • FIG. 34 is a block diagram of the communication device and its peripheral circuits according to the sixth embodiment.
  • the resonant frequency in the resonator or circuit is a resonant frequency for forming an attenuation pole in the pass band or near the pass band of the resonator or the filter including the circuit, and the resonator Alternatively, it is the frequency of the “resonance point” which is a singular point at which the impedance of the circuit becomes minimum (ideally, the point at which the impedance is 0).
  • the antiresonance frequency in the resonator or the circuit is an antiresonance frequency for forming an attenuation pole in the pass band or near the pass band of the resonator or the filter including the circuit, unless otherwise specified. It is the frequency of the "anti-resonance point" which is a singular point at which the impedance of the resonator or the circuit concerned is maximal (ideally, the point at which the impedance is infinite).
  • a series arm (resonance) circuit and a parallel arm (resonance) circuit are defined as follows.
  • the parallel arm (resonance) circuit is a circuit disposed between one node on a path connecting the first input / output terminal and the second input / output terminal and the ground.
  • the series arm (resonance) circuit is a circuit disposed between the first input / output terminal or the second input / output terminal and a node on the above path to which the parallel arm (resonance) circuit is connected, or The circuit is disposed between one node on the path to which the parallel arm (resonance) circuit is connected and the other node on the path to which the other parallel arm (resonance) circuit is connected.
  • passband lower end means “the lowest frequency in the passband”.
  • pass band high end means “the highest frequency in the pass band”.
  • low pass band side means “outside of pass band and lower frequency side than pass band”.
  • high pass side of pass band means “outside of pass band and higher frequency side than pass band”.
  • the “low frequency side” may be referred to as the “low frequency side”
  • the “high frequency side” may be referred to as the “high frequency side”.
  • FIG. 1A is a circuit block diagram of the high frequency filter 10 according to the first embodiment.
  • the high frequency filter 10 shown in the figure includes filters 11 and 12, phase shifters 21 and 22, and input / output terminals 110 and 120.
  • the filter 11 is a first filter that constitutes a first circuit.
  • the first circuit includes a node X1 (first node) provided on a path connecting the input / output terminal 110 (first input / output terminal) and the input / output terminal 120 (second input / output terminal), and a node X1 and It is connected to a node X2 (second node) provided between the input and output terminals 120.
  • the filter 12 is a second filter that constitutes a second circuit together with the phase shifters 21 and 22.
  • the second circuit is connected to nodes X1 and X2.
  • the phase shifter 21 is a first phase shifter connected to one terminal of the filter 12.
  • the phase shifter 22 is a second phase shifter connected to the other terminal of the filter 12. That is, the phase shifter 21, the filter 12, and the phase shifter 22 are connected in series between the node X1 and the node X2 in this order.
  • FIG. 1B is a diagram showing the concept of broadening the band of the high-frequency filter 10 according to the first embodiment.
  • the high frequency filter 10 has a first wide band as a pass band.
  • the filter 11 is a band including a part of the frequency of the first band, and a second band having a narrower bandwidth than the first band is a pass band.
  • the filter 12 is a band including a portion of the frequency of the first band, has a narrower bandwidth than the first band, and passes through a third band located higher than the center frequency of the second band. It is a band.
  • the first band is not defined as a band constituted by a plurality of discrete bands, but is defined as one band continuous in frequency.
  • FIG. 2 is a circuit configuration diagram of the high frequency filter 10A according to the first embodiment.
  • the high frequency filter 10A shown in the figure includes filters 11A and 12A, phase shifters 21 and 22, and input / output terminals 110 and 120.
  • High frequency filter 10A is a specific circuit configuration example of high frequency filter 10 according to the first embodiment
  • filter 11A is a specific circuit configuration example of filter 11
  • filter 12A is a specific circuit configuration example of filter 12 .
  • the filter 11A includes series arm resonators s11, s12 and s13 disposed on a first path connecting the node X1 and the node X2, and a parallel arm disposed between each node on the first path and the ground.
  • a resonator p11, p12, p13 and p14, and a parallel arm resonator p13 and an inductor Lp13 connected to the ground are provided.
  • the filter 11A constitutes a ladder type band pass filter.
  • the filter 12A includes series arm resonators s21 and s22 arranged on a second path connecting the node X1 and the node X2, and parallel arm resonators arranged between each node on the second path and the ground. and p21, p22 and p23.
  • the filter 12A constitutes a ladder type band pass filter.
  • Table 1 shows circuit parameters of the high frequency filter 10A according to the first embodiment.
  • the phase shifter 21 is disposed at one end of the second filter, and the phase shifter 22 is disposed at the other end of the second filter. It features an arrangement.
  • the features and problems of the high frequency filter according to the comparative example will be described.
  • FIG. 3A is a circuit block diagram of the high frequency filter 500 according to the first comparative example.
  • FIG. 3B is a circuit block diagram of the high frequency filter 600 according to the second comparative example.
  • the high frequency filter 500 according to Comparative Example 1 shown in FIG. 3A includes the filters 11 and 12 and the input / output terminals 110 and 120.
  • the high frequency filter 500 shown in FIG. 3A differs from the high frequency filter 10 shown in the embodiment only in that the phase shifters 21 and 22 are not arranged.
  • the filters 11 and 12 constituting the high frequency filter 500 have the same configuration as the filters 11 and 12 constituting the high frequency filter 10, and the filter 11 of the high frequency filter 500 has a second band as a pass band.
  • Reference numeral 12 designates the third band as the pass band.
  • the high frequency filter 600 according to the comparative example 2 shown in FIG. 3B includes the filters 11 and 12, the phase shifter 21, and the input and output terminals 110 and 120.
  • the high frequency filter 600 according to Comparative Example 2 shown in FIG. 3B is different from the high frequency filter 10 shown in the embodiment only in that the phase shifter 22 is not disposed.
  • the filters 11 and 12 constituting the high frequency filter 600 have the same configuration as the filters 11 and 12 constituting the high frequency filter 10, and the filter 11 of the high frequency filter 500 has a second band as a pass band.
  • Reference numeral 12 designates the third band as the pass band.
  • the second band is a Band 3 reception pass band (1805-1880 MHz) of LTE (Long Term Evolution), and the third band is a Band 39 pass band of LTE
  • the first band is (Band 3 reception pass band + Band 39 pass band) (1805-1920 MHz).
  • FIG. 4A is a graph showing the pass characteristic, the amplitude characteristic, the phase characteristic, and the impedance characteristic of the high frequency filter 500 according to the first comparative example.
  • the pass characteristics of the high frequency filter 500 are shown in (a) and (f) of FIG. 4A.
  • In (b) and (c) of FIG. 4A an amplitude difference and a phase difference between the first circuit alone between the nodes X1 and X2 and the second circuit alone between the nodes X1 and X2 are shown.
  • impedance characteristics are shown when the first circuit alone and the second circuit alone are viewed from the node X1.
  • FIG. 4A shows impedance characteristics when the first circuit single unit and the second circuit single unit are viewed from the node X2.
  • G shows the pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2.
  • phase characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2 are shown.
  • the “impedance” of the circuit refers to the impedance of reflection
  • the “phase” of the circuit represents the phase of passage
  • FIG. 4B is a graph showing the pass characteristic, the amplitude characteristic, the phase characteristic, and the impedance characteristic of the high frequency filter 600 according to the second comparative example.
  • the pass characteristics of the high frequency filter 600 are shown in (a) and (f) of FIG. 4B.
  • In (b) and (c) of FIG. 4B an amplitude difference and a phase difference between the first circuit alone between the nodes X1 and X2 and the second circuit alone between the nodes X1 and X2 are shown.
  • the impedance characteristics are shown when the first circuit alone and the second circuit alone are viewed from the node X1.
  • FIG. 4B (e) shows impedance characteristics when the first circuit alone and the second circuit alone are viewed from the node X2.
  • FIG. 4B (g) shows the pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2.
  • (h) of FIG. 4B phase characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2 are shown.
  • FIG. 5 is a graph showing the pass characteristic, the amplitude characteristic, the phase characteristic, and the impedance characteristic of the high frequency filter 10A according to the first embodiment.
  • the pass characteristics of the high frequency filter 10A are shown in (a) and (f) of FIG.
  • FIGS. 5B and 5C respectively show an amplitude difference and a phase difference between the first circuit alone between the nodes X1 and X2 and the second circuit alone between the nodes X1 and X2.
  • FIG. 5D shows impedance characteristics when the first circuit unit and the second circuit unit are viewed from the node X1.
  • FIG. 5E shows impedance characteristics when the first circuit unit and the second circuit unit are viewed from the node X2.
  • FIG. 5 (g) shows the pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2.
  • FIG. 5H shows phase characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2.
  • the phase shifters 21 and 22 are connected to both input / output terminals (previous stage and subsequent stage) of the filter 12A, as shown in (a) and (f) of FIG. As shown, in the first band which is the pass band of the high frequency filter 10A, the ripple is reduced to reduce the insertion loss.
  • the high frequency filter 10A according to the first embodiment reduces the in-passband ripple and the in-passband insertion loss as compared to the high frequency filter 500 according to the first comparative example and the high frequency filter 600 according to the second comparative example are described below. explain.
  • FIG. 6 is a graph for explaining the factor of increasing the bandwidth of the high frequency filter 10A according to the first embodiment.
  • (c), (d), (e), (g) and (h) are shown in order to explain the above factors.
  • frequency f 11 which is the maximum of the impedance of the singular point as a maximum is the low frequency end of the second band
  • the phases of the phase shifters 21 and 22 are adjusted so as to be included in the frequency band from the frequency f A to the cut-off frequency f C on the low pass band side of the filter 12A.
  • frequency f 22 which is the maximum of the impedance of the singular point as a maximum is the frequency f A of the low-frequency end of the second band, the filter 12A
  • the phases of the phase shifters 21 and 22 are adjusted so as to be included in the frequency band up to the cutoff frequency f C on the lower side of the pass band.
  • the cut-off frequency on the low pass side of the filter is the lowest pass band among the frequencies on the low side of the low pass band which is increased by a predetermined insertion loss from the insertion loss at the low end of the pass band of the filter. It is a frequency close to the end.
  • the cut-off frequency on the lower side of the passband is defined as the frequency on the lower side of the passband which is increased by 10 dB from the insertion loss at the lower end of the passband.
  • the phase change is large, the phase difference with the filter 11A is large, and the amplitude difference is also large. Therefore, by the phase adjustment of the phase shifters 21 and 22, the impedance of the second circuit is in the frequency range from the frequency f A at the low band end of the second band to the cutoff frequency f C at the low band side of the filter 12A.
  • the impedance of the second circuit is in the frequency range from the frequency f A at the low band end of the second band to the cutoff frequency f C at the low band side of the filter 12A.
  • the amplitude difference and the phase difference between the filter 11A and the second circuit become large also at the frequency of the attenuation pole on the high band side of the pass band (second band) of the filter 11A, but the attenuation pole concerned is the filter 11A. It is comprised by the antiresonance frequency of the series arm circuit to comprise. For this reason, since the impedance of the filter 11A is maximum, even if there is no phase shifter, it does not become a ripple in the pass band (first band) of the high frequency filter 10A.
  • of the series arm resonator, the parallel arm resonator, the series arm circuit and the parallel arm circuit is minimized is the resonance frequency
  • is the maximum is the antiresonance frequency.
  • the phase shifters 21 and 22 are adjusted such that the phase difference between the first circuit and the second circuit is in the same phase in a small frequency band.
  • the high frequency signals of the frequencies included in the first band do not cancel each other after passing through the first circuit and the second circuit.
  • the ripple in the 1st zone can be reduced in the frequency contained in the 1st zone of high frequency filter 10A.
  • the first circuit and the second circuit are the same in the second band.
  • FIG. 7 is a graph showing pass characteristics, phase characteristics, and impedance characteristics when the phases of the phase shifters 21 and 22 of the high frequency filter 10A according to the first embodiment are changed.
  • FIG. 8 is a graph showing the relationship between the phase of the phase shifters 21 and 22 of the high frequency filter 10A according to the first embodiment and the frequency of the insertion loss and the maximum impedance. Note that the frequency of the maximum impedance is a frequency at which the impedance is maximized among the singular points at which the impedance is maximized. More specifically, FIG. 7 shows changes in pass characteristics, phase characteristics and impedance characteristics when the phase of phase shifters 21 and 22 is changed from 0 ° to 150 ° (synonymous with 180 ° to 330 °). Is shown.
  • FIG. 8 shows the change in the insertion loss and the maximum impedance when the phase of the phase shifters 21 and 22 is changed in steps of 5 ° at 0 ° to 175 ° (equivalent to 180 ° to 355 °). There is.
  • phase difference between the first circuit and the second circuit is opposite (a phase of 180
  • the ripples increase, and when the phase difference (0 ° or 360 ° phase difference) is reached, the ripples decrease. Therefore, by eliminating the phase difference between the first circuit and the second circuit in the frequency band where the amplitude difference in the first band, which is the pass band, is small, the ripple in the first band of the high frequency filter 10A is reduced, and the pass band Internal insertion loss can be reduced.
  • the phase difference (A) between the filters 11A and 12A is 152.2 as shown in (h) of FIG. °
  • the phase difference (B) between the first circuit and the second circuit is 349.2 ° according to (h) in FIG.
  • the sum (C) of the phases of the phase shifters 21 and 22 is 197 °. .
  • the ripple of the high frequency filter 10A is 3 dB or less when the phase of each of the phase shifters 21 and 22 is 80 to 105 °.
  • the range of the phase difference (B) between the first circuit and the second circuit is defined. That is, the phase difference (B) of the first circuit and the second circuit is 312.2 ° (152.2 ° + 80 ° ⁇ 2) to 362.2 ° (152.2 ° + 105 ° ⁇ 2 °).
  • the phase shifters 21 and 22 are adjusted.
  • changing the phase of phase shifters 21 and 22 changes the frequency at which impedance
  • the impedance of the second circuit is increased, and the signal does not go to the second circuit in the lower attenuation band of the second circuit.
  • the ripple in the passband is suppressed to reduce the insertion loss in the passband, or the ripple in the passband is suppressed to reduce the insertion loss.
  • the frequency at which the impedance at the maximum of the singular point at which the impedance of the second circuit is maximal is the second band is low. It is desirable that the phases of the phase shifters 21 and 22 be adjusted so as to be included in the frequency band from the frequency of the band edge to the cut-off frequency on the low band side of the pass band of the filter 12A. In particular, in the present embodiment, by adjusting the phase of the phase shifters 21 and 22 on the low pass band side of the filter 12A, the ripple of the high frequency filter 10A can be made within 3 dB.
  • FIG. 9 is a circuit diagram of the high frequency filter 10B according to the embodiments 1a, 1b, 1c, 1d and 1e.
  • the high frequency filter 10B shown in the figure includes filters 11D and 12D, phase shifters 21 and 22, and input / output terminals 110 and 120.
  • the high frequency filter 10B is a specific circuit configuration example of the high frequency filter 10 according to the first embodiment
  • the filter 11D is a specific circuit configuration example of the filter 11
  • the filter 12D is a specific circuit configuration example of the filter 12 .
  • the filter 11D includes series arm resonators s11 and s12 arranged on a first path connecting the node X1 and the node X2, and a node and a ground on a path connecting the series arm resonator s11 and the series arm resonator s12. And a parallel arm resonator p11 disposed between the two.
  • the filter 11D constitutes a ladder type band pass filter.
  • the filter 12D includes a series arm resonator s21 disposed on a second path connecting the node X1 and the node X2, a parallel arm resonator p21 disposed between each node on the second path and the ground, and and p22.
  • the filter 12D constitutes a ladder type band pass filter.
  • each resonator constituting the high frequency filter 10B will be described in more detail focusing on the parallel arm resonator p11.
  • the other resonators have almost the same structure as the parallel arm resonator p11 except that the IR pitch is about 0.5 times the wavelength ⁇ of the elastic wave, and so on. I omit explanation.
  • FIG. 10 is an example of a diagram schematically illustrating the structure of the parallel arm resonator p11 according to the first embodiment, in which (a) is a plan view and (b) is a cross-sectional view of (a).
  • the parallel arm resonator p11 shown in FIG. 10 is for describing a typical structure of each of the resonators constituting the high frequency filter 10B. For this reason, the number, length, etc. of the electrode fingers constituting the IDT (InterDigital Transducer) electrode of each resonator of the high frequency filter 10B are not limited to the number or the length of the electrode fingers of the IDT electrode shown in FIG.
  • IDT InterDigital Transducer
  • the parallel arm resonator p11 includes an electrode film 301 constituting an IDT electrode 321 and a reflector 322, and a piezoelectric substrate 302 on which the electrode film 301 is formed. And a protective layer 303 covering the electrode film 301.
  • these components will be described in detail.
  • the comb-tooth electrode 301a is composed of a plurality of parallel electrode fingers 310a and a bus bar electrode 311a connecting the plurality of electrode fingers 310a.
  • the comb-tooth electrode 301 b is configured of a plurality of parallel electrode fingers 310 b and a bus bar electrode 311 b connecting the plurality of electrode fingers 310 b.
  • the plurality of electrode fingers 310a and 310b are formed along a direction orthogonal to the propagation direction of the elastic wave, and are periodically formed along the propagation direction.
  • the thus configured IDT electrode 321 excites a surface acoustic wave in a specific frequency range defined by the electrode pitch and the like of the plurality of electrode fingers 310 a and 310 b constituting the IDT electrode 321.
  • Each of the comb-tooth electrodes 301a and 301b may be referred to as an IDT electrode alone. However, in the following, for convenience, the description will be made assuming that one IDT electrode 321 is configured by the pair of comb electrodes 301a and 301b.
  • the reflector 322 is disposed in the propagation direction of the elastic wave with respect to the IDT electrode 321. Specifically, the pair of reflectors 322 is disposed so as to sandwich the IDT electrode 321 from both sides in the propagation direction of the elastic wave.
  • the reflector 322 includes a plurality of parallel electrode fingers 410, a bus bar electrode 411 connecting one end of the plurality of electrode fingers 410, and a bus bar electrode 411 connecting the other ends of the plurality of electrode fingers 410. A pair of bus bar electrodes 411 is formed.
  • the plurality of electrode fingers 410 are formed along the direction orthogonal to the propagation direction of the elastic wave, and are periodically formed along the propagation direction. ing.
  • the reflector 322 configured in this way reflects the surface acoustic wave with a high reflection coefficient in a frequency band (stop band) defined by the electrode pitch and the like of the plurality of electrode fingers 410 constituting the reflector 322. That is, when the electrode pitch of the IDT electrode 321 and the electrode pitch of the reflector 322 are equal, the reflector 322 reflects the surface acoustic wave excited by the IDT electrode 321 with a high reflection coefficient.
  • the parallel arm resonator p11 can confine the excited surface acoustic wave inside and make it difficult to leak it to the outside. Therefore, the parallel arm resonator p11 can improve Q of the resonance point and the antiresonance point defined by the electrode pitch, the logarithm, the crossing width, and the like of the IDT electrode 321.
  • the reflector 322 only needs to have the electrode finger 410, and may not have the bus bar electrode 411.
  • the number of the electrode fingers 410 may be one or more, and is not particularly limited. However, if the number of the electrode fingers 410 is too small, the elastic wave leaks more, and the filter characteristics may be degraded. On the other hand, when the number of the electrode fingers 410 is too large, the reflector 322 is increased in size, so the entire high frequency filter 10B may be increased in size. For this reason, the number of electrode fingers 410 can be appropriately determined in consideration of the filter characteristics, the size, and the like required of the high frequency filter 10B.
  • the IDT electrode 321 and the reflector 322 are constituted by an electrode film 301 shown in (b) of FIG.
  • the electrode film 301 has a laminated structure of the adhesion layer 301 g and the main electrode layer 301 h.
  • the IDT electrode 321 and the reflector 322 are formed of the same electrode film 301, but they may be formed of electrode films having different structures, compositions, and the like.
  • the adhesion layer 301g is a layer for improving the adhesion between the piezoelectric substrate 302 and the main electrode layer 301h, and, for example, Ti is used as a material.
  • the film thickness of the adhesion layer 301 g is, for example, 12 nm.
  • the material of the main electrode layer 301h for example, Al containing 1% of Cu is used.
  • the film thickness of the main electrode layer 301 h is, for example, 162 nm.
  • the piezoelectric substrate 302, the electrode film 301 (i.e., IDT electrodes 321 and reflectors 322) substrate which is formed, for example, LiTaO 3 piezoelectric single crystal, LiNbO 3 piezoelectric single crystal, KNbO 3 piezoelectric single crystal, quartz or, It consists of piezoelectric ceramics.
  • the protective layer 303 is formed to cover the comb-tooth electrodes 301a and 301b.
  • the protective layer 303 is a layer for protecting the main electrode layer 301 h from the external environment, adjusting frequency temperature characteristics, enhancing moisture resistance, and the like, and is, for example, a film containing silicon dioxide as a main component. .
  • each resonator which the high frequency filter 10B has is not limited to the structure described in FIG.
  • the electrode film 301 may be a single layer of a metal film instead of the laminated structure of the metal film.
  • the materials forming the adhesion layer 301 g, the main electrode layer 301 h, and the protective layer 303 are not limited to the above-described materials.
  • the electrode film 301 may be made of, for example, a metal or alloy such as Ti, Al, Cu, Pt, Au, Ag, Pd, etc., and is made of a plurality of laminates made of the above metals or alloys. May be
  • the protective layer 303 may not be formed.
  • the wavelength of the elastic wave to be excited is defined by the design parameters of the IDT electrode 321 and the like.
  • design parameters of the IDT electrode 321 that is, design parameters of the comb-tooth electrode 301a and the comb-tooth electrode 301b will be described.
  • the wavelength of the elastic wave is defined by the repetition period ⁇ of the plurality of electrode fingers 310a or 310b constituting the comb-tooth electrodes 301a and 301b shown in FIG.
  • the electrode pitch is a half of the repetition period ⁇
  • the line width of the electrode fingers 310a and 310b constituting the comb electrodes 301a and 301b is W
  • the adjacent electrode fingers 310a and the electrodes are When the space width between the finger 310 b is S, it is defined by (W + S). Further, as shown in (a) of FIG.
  • the cross width L of the IDT electrode 321 is the electrode finger 310a of the comb electrode 301a and the electrode finger 310b of the comb electrode 301b viewed from the propagation direction of the elastic wave.
  • the electrode duty (duty ratio) is a line width occupancy rate of the plurality of electrode fingers 310a and 310b, and is a ratio of the line width to an added value of the line width and the space width of the plurality of electrode fingers 310a and 310b. , W / (W + S).
  • the electrode pitch (electrode period) of the reflector 322 is defined by (W REF + S REF ), where the line width of the electrode finger 410 is W REF and the space width between adjacent electrode fingers 410 is S REF. .
  • the electrode duty (duty ratio) of the reflector 422 is a line width occupancy rate of the plurality of electrode fingers 410 and is a ratio of the line width to the addition value of the line width of the electrode finger 410 and the space width. It is defined by REF / (W REF + S REF ).
  • the film thickness of the reflector 322 is the thickness of the plurality of electrode fingers 410.
  • the electrode pitch and electrode duty of the reflector 322 are equal to the electrode pitch and electrode duty of the IDT electrode 321. Further, the reflector 322 is arranged such that the pair of bus bar electrodes 411 overlap the bus bar electrodes 311 a and 311 b of the IDT electrode 321 when viewed in the propagation direction of the elastic wave.
  • the above configuration is preferable from the viewpoint of suppressing the leak of the elastic wave, but the configuration may be different from the above configuration.
  • the pitch (IR pitch) between the IDT electrode 321 and the reflector 322 is (i) the electrode finger closest to the reflector 322 among the plurality of electrode fingers 310 a or 310 b constituting the IDT electrode 321 and (ii) It is defined as a center-to-center distance with the electrode finger 410 closest to the IDT electrode 321 among the plurality of electrode fingers 410 constituting the reflector 322.
  • This IR pitch can be expressed using a repetition period ⁇ of a plurality of electrode fingers 310a or 310b constituting the comb-tooth electrodes 301a and 301b (ie, a wavelength ⁇ of an elastic wave determined by the electrode pitch of the IDT electrode 321). For example, in the case where the repetition period ⁇ is 0.50 times, it is expressed as 0.50 ⁇ .
  • the high frequency filter 10B includes the filters 11D and 12D.
  • the filter 11D is connected to a first series arm circuit (series arm resonator s11 or s12) provided on a first path connecting the node X1 and the node X2, a node provided on the first path, and the ground.
  • the first parallel arm circuit parallel arm resonator p11.
  • the filter 12D includes a second series arm circuit (series arm resonator s21) provided on a second path connecting the phase shifters 21 and 22, a node provided on the second path, and a ground.
  • a second parallel arm circuit (parallel arm resonator p21 or p22) connected to At least one of the first series arm circuit, the first parallel arm circuit, the second series arm circuit, and the second parallel arm circuit has an elastic wave resonator, and at least one of the elastic wave resonators.
  • the wavelength of the elastic wave determined by the electrode period of the IDT electrode in at least one of the elastic wave resonators is ⁇
  • the pitch between the IDT electrode and the reflector is 0.42 ⁇ or more and less than 0.50 ⁇ . It is desirable to have.
  • the elastic wave resonator having the IDT electrode 321 and the reflector 322 is constituted by a periodic structure consisting of periodically arranged electrode fingers, and reflects the surface acoustic wave of a specific frequency region with a high reflection coefficient.
  • Frequency band. This frequency band is generally referred to as a stop band, and is defined by the repetition period of the periodic structure or the like.
  • the stop band At this time, at the high end of the stop band, ripples in which the reflection coefficient locally increases are generated.
  • the pitch (IR pitch) between the IDT electrode 321 and the reflector 322 is 0.5 ⁇ or more, the ripple at the high end of the stop band becomes large, and the ripple in the pass band of the filter becomes large. Insertion loss increases.
  • the IR pitch is 0.42 ⁇ or more and less than 0.50 ⁇ , it is possible to reduce the ripple at the high end of the stop band. From the above viewpoint, in the filters 11D and 12D constituting the high frequency filter 10B, by setting the IR pitch to be 0.42 ⁇ or more and less than 0.50 ⁇ , the insertion loss in the pass band of the high frequency filter 10B is further reduced. be able to.
  • the high frequency filter 10B according to the embodiments 1a to 1d and the high frequency filter according to the embodiment 1e will be compared.
  • the configuration of the IR pitch is as follows. It is as it is.
  • Embodiment 1e The IR pitch of all the elastic wave resonators (series arm resonators s11, s12, s21 and parallel arm resonators p11, p21, p22) is 0.50 ⁇ .
  • Example 1a The IR pitch of the series arm resonators s11 and s12 is 0.44 ⁇ , and the IR pitch of the other elastic wave resonators is 0.50 ⁇ .
  • Example 1b The IR pitch of the parallel arm resonator p11 is 0.44 ⁇ , and the IR pitch of the other elastic wave resonators is 0.50 ⁇ .
  • Example 1c The IR pitch of the parallel arm resonators p21 and p22 is 0.44 ⁇ , and the IR pitch of the other elastic wave resonators is 0.50 ⁇ .
  • Embodiment 1d The IR pitch of all the elastic wave resonators (series arm resonators s11, s12, s21 and parallel arm resonators p11, p21, p22) is 0.44 ⁇ .
  • FIG. 11A is a graph comparing the pass characteristic and the reflection characteristic of the high frequency filters according to Example 1a and Example 1e.
  • FIG. 11B is a graph comparing impedance characteristics and reflection characteristics of the series arm resonators s11 and s12 alone according to Example 1a and Example 1e.
  • the pass characteristics of the high frequency filters according to Example 1a and Example 1e are shown.
  • the pass characteristic and the reflection characteristic of the single filter 11D according to Example 1a and Example 1e are shown.
  • ripples in which the reflection loss is locally increased occur at the high frequency end of the stop band.
  • the ripple of the reflection loss is reduced at the high end of the stop band (areas A4 and A5 in the figure).
  • the high frequency filter 10B includes the filters 11D and 12D.
  • the filter 11D is connected to a first series arm circuit (series arm resonator s11 or s12) provided on a first path connecting the node X1 and the node X2, a node provided on the first path, and the ground.
  • the first parallel arm circuit (parallel arm resonator p11).
  • the filter 12D is connected to a second series arm circuit (series arm resonator s21) provided on the second path of the phase shifters 21 and 22, a node provided on the second path, and a ground.
  • a parallel arm circuit parallel arm resonator p21 or p22).
  • the first series arm circuit includes an elastic wave resonator, and the elastic wave resonator includes an IDT electrode formed on a substrate having piezoelectricity, and a reflector.
  • the pitch between the IDT electrode of the elastic wave resonator and the reflector is 0.42 ⁇ or more and less than 0.50 ⁇ .
  • the series arm resonators s11 and s12 in the series arm circuit of the filter 11D have a resonant frequency in the pass band of the filter 11D and an antiresonance frequency in the pass band of the filter 12D on the high band side of the pass band of the filter 11D.
  • the stop bands of the series arm resonators s11 and s12 become the pass band of the filter 12D.
  • the IR pitch to 0.42 ⁇ or more and less than 0.50 ⁇ , it is possible to reduce the ripple at the high end of the stop band. Therefore, by making the IR pitch 0.42 ⁇ or more and less than 0.50 ⁇ , the insertion loss in the pass band of the high frequency filter 10B can be further reduced.
  • FIG. 12A is a graph comparing the pass characteristics of the high frequency filters according to Example 1 b and Example 1 e.
  • FIG. 12B is a graph comparing impedance characteristics and reflection characteristics of the parallel arm resonator p11 and the single body according to Example 1b and Example 1e.
  • the pass characteristics of the high frequency filter according to Example 1 b and Example 1 e are shown.
  • the pass characteristics of the filter 11D alone according to Example 1b and Example 1e are shown.
  • the high frequency filter 10B includes the filters 11D and 12D.
  • the filter 11D is connected to a first series arm circuit (series arm resonator s11 or s12) provided on a first path connecting the node X1 and the node X2, a node provided on the first path, and the ground.
  • the first parallel arm circuit (parallel arm resonator p11).
  • the filter 12D is connected to a second series arm circuit (series arm resonator s21) provided on the second path of the phase shifters 21 and 22, a node provided on the second path, and a ground.
  • a parallel arm circuit parallel arm circuit (parallel arm resonator p21 or p22).
  • the first parallel arm circuit includes an elastic wave resonator, and the elastic wave resonator includes an IDT electrode formed on a substrate having piezoelectricity, and a reflector.
  • the pitch between the IDT electrode of the elastic wave resonator and the reflector is 0.42 ⁇ or more and less than 0.50 ⁇ .
  • the parallel arm resonator p11 in the parallel arm circuit of the filter 11D has a resonant frequency on the low frequency side of the passband of the filter 11D and an antiresonant frequency in the passband of the filter 11D.
  • the stop band of the parallel arm resonator p11 is located within the pass band of the filter 11D or on the high frequency side of the pass band of the filter 11D.
  • FIG. 13A is a graph comparing the pass characteristic and the reflection characteristic of the high frequency filters according to Example 1 c and Example 1 e.
  • FIG. 13B is a graph comparing impedance characteristics and reflection characteristics of the parallel arm resonators p21 and p22 according to the example 1c and the example 1e.
  • the pass characteristics of the high frequency filters according to Example 1c and Example 1e are shown.
  • the pass characteristics of the filter 12D alone according to Example 1c and Example 1e are shown.
  • the high frequency filter 10B includes the filters 11D and 12D.
  • the filter 11D is connected to a first series arm circuit (series arm resonator s11 or s12) provided on a first path connecting the node X1 and the node X2, a node provided on the first path, and the ground.
  • the first parallel arm circuit (parallel arm resonator p11).
  • the filter 12D is connected to a second series arm circuit (series arm resonator s21) provided on the second path of the phase shifters 21 and 22, a node provided on the second path, and a ground.
  • a parallel arm circuit parallel arm circuit (parallel arm resonator p21 or p22).
  • the second parallel arm circuit includes an elastic wave resonator, and the elastic wave resonator includes an IDT electrode formed on a substrate having piezoelectricity, and a reflector.
  • the pitch between the IDT electrode of the elastic wave resonator and the reflector is 0.42 ⁇ or more and less than 0.50 ⁇ .
  • the parallel arm resonators p21 and p22 in the parallel arm circuit of the filter 12D have a resonant frequency on the low frequency side of the pass band of the filter 12D and an antiresonant frequency in the pass band of the filter 12D.
  • the stop bands of the parallel arm resonators p21 and p22 are located in the pass band of the filter 12D or on the high frequency side of the pass band of the filter 12D.
  • the IR pitch to 0.42 ⁇ or more and less than 0.50 ⁇ , it is possible to reduce the ripple at the high end of the stop band. Therefore, by making the IR pitch 0.42 ⁇ or more and less than 0.50 ⁇ , the insertion loss in the pass band of the high frequency filter 10B can be further reduced.
  • FIG. 14 is a graph comparing the pass characteristic and the reflection characteristic of the high frequency filters according to Example 1 d and Example 1 e.
  • the pass characteristics of the high frequency filter according to Example 1 d and Example 1 e are shown.
  • the pass characteristic and the reflection characteristic of the single filter 11D according to Example 1 d and Example 1 e are shown.
  • the pass characteristic and the reflection characteristic of the single filter 12D according to Example 1 d and Example 1 e are shown.
  • the high frequency filter 10B includes the filters 11D and 12D.
  • the filter 11D is connected to a first series arm circuit (series arm resonator s11 or s12) provided on a first path connecting the node X1 and the node X2, a node provided on the first path, and the ground.
  • the first parallel arm circuit (parallel arm resonator p11).
  • the filter 12D is connected to a second series arm circuit (series arm resonator s21) provided on the second path of the phase shifters 21 and 22, a node provided on the second path, and a ground.
  • a parallel arm circuit parallel arm circuit (parallel arm resonator p21 or p22).
  • Each of the first series arm circuit, the second series arm circuit, the first parallel arm circuit, and the second parallel arm circuit has an elastic wave resonator, and the elastic wave resonator is formed on a substrate having piezoelectricity. It has the formed IDT electrode and a reflector.
  • the pitch between the IDT electrode of the elastic wave resonator and the reflector is 0.42 ⁇ or more and less than 0.50 ⁇ .
  • the IR pitch of the series arm resonator s21 of the filter 12D does not directly affect the insertion loss of the high frequency filter 10B, but the high frequency filter 10B and a filter having a passband on the higher frequency side than the high frequency filter 10B
  • the multiplexer is configured by a filter in which the generation frequency of the stop band ripple of the high frequency filter 10B and the pass band overlap with each other, the insertion loss of the filter can be reduced.
  • FIG. 15A is a graph showing a change in characteristics when the IR pitch is changed at 0.40 ⁇ to 0.50 ⁇ in the resonator of the typical example.
  • FIG. 15B is a graph showing a change in characteristics when the IR pitch is changed at 0.50 ⁇ to 0.60 ⁇ in the resonator of the typical example.
  • FIG. 15A is a graph showing the absolute value of the impedance
  • (b) is a graph showing the phase characteristic
  • (c-1) is a graph in which the impedance is represented by a Smith chart.
  • (C-2) is a graph showing reflection loss (return loss).
  • FIG. 15A shows the characteristics of the resonator when the IR pitch is changed from 0.40 ⁇ to 0.50 ⁇ in steps of 0.02 ⁇ , as shown in the legend in the figure.
  • FIG. 15B shows the characteristics of the resonator when the IR pitch is changed from 0.50 ⁇ to 0.60 ⁇ in steps of 0.02 ⁇ , as indicated by the legend in the figure.
  • the IR pitch of the resonator by setting the IR pitch of the resonator to 0.42 ⁇ or more and 0.50 ⁇ or less, it is possible to suppress the ripple that may occur on the high frequency side of the resonance frequency of the resonator, so the passage by the ripple It is possible to suppress the deterioration of the insertion loss of the band.
  • FIG. 16 is a circuit block diagram of the high frequency filter 20 according to the second embodiment.
  • the high frequency filter 20 shown in the figure includes filters 11 and 12, phase shifters 21 and 22, switches SW1, SW2, SW5 and SW6, and input / output terminals 110 and 120.
  • the high frequency filter 20 according to the present embodiment is different from the high frequency filter 10 according to the first embodiment in that the first circuit and the second circuit include switch elements.
  • the high frequency filter 20 according to the present embodiment will not be described the same as the high frequency filter 10 according to the first embodiment, and different points will be mainly described.
  • the first circuit further includes a switch SW1 and a switch SW2.
  • the second circuit further includes a switch SW5 and a switch SW6.
  • the switch SW1 is a first switch element connected between the node X1 and the filter 11.
  • the switch SW2 is a second switch element connected between the node X2 and the filter 11.
  • the switch SW5 is a fifth switch element connected between the node X1 and the phase shifter 21.
  • the switch SW6 is a sixth switch element connected between the node X2 and the phase shifter 22.
  • FIG. 17A is a circuit configuration diagram of a high frequency filter 20A according to a second embodiment.
  • the high frequency filter 20A shown in the figure includes filters 11A and 12A, phase shifters 21 and 22, switches SW1, SW2, SW5 and SW6, and input / output terminals 110 and 120.
  • the high frequency filter 20A is a specific circuit configuration example of the high frequency filter 20 according to the second embodiment
  • the filter 11A is a specific circuit configuration example of the filter 11
  • the filter 12A is a specific circuit configuration example of the filter 12 .
  • the high frequency filter 20A according to the present embodiment is located between the first circuit and the nodes X1 and X2, and between the second circuit and the nodes X1 and X2.
  • the configuration is different in that switch elements are disposed.
  • the high frequency filter 20A according to the present embodiment will not be described the same as the high frequency filter 10A according to the first embodiment, and differences will be mainly described.
  • the first circuit further includes a switch SW1 and a switch SW2.
  • the second circuit further includes a switch SW5 and a switch SW6.
  • the switch SW1 is a first switch element connected between the node X1 and the filter 11.
  • the switch SW2 is a second switch element connected between the node X2 and the filter 11.
  • the switch SW5 is a fifth switch element connected between the node X1 and the phase shifter 21.
  • the switch SW6 is a sixth switch element connected between the node X2 and the phase shifter 22.
  • Table 2 shows circuit parameters of the high frequency filter 20A according to the second embodiment.
  • the second band is set to Band 3 reception pass band (1805-1880 MHz) of LTE
  • the third band is set to Band 39 pass band (1880 to 1920 MHz) of LTE.
  • the first band is (Band 3 reception pass band + Band 39 pass band) (1805-1920 MHz).
  • the high frequency filter 20A has two filters of the filter 11A whose band 2 is the Band 3 reception passband and the filter 12A whose band 3 is the band 39 passband, and it is between the filter 12A and the node X1.
  • the phase shifter 21 is disposed, and the phase shifter 22 is disposed between the filter 12A and the node X2.
  • FIG. 17B is a graph showing a change in passage characteristics due to the switching of the switch of the high frequency filter 20A according to the second embodiment.
  • the high frequency filter 20A is a filter that uses the first band (Band 3 reception pass band + Band 39 pass band) as the pass band when all of the switches SW1, SW2, SW5 and SW6 are conductive. It becomes.
  • the filter becomes a pass band of the second band (Band 3 reception pass band).
  • the third band (Band 39 pass band) serves as a pass band.
  • the switches SW1 and SW2 are disposed on the first circuit side, and the switches SW5 and SW6 are disposed on the second circuit side.
  • the switches are connected only to the first circuit side. It may be a configuration or a configuration in which a switch is connected only to the second circuit side.
  • the high frequency filter serves as a filter that switches between the first band and the third band. Further, in the case where the switch is connected only to the second circuit side, the high frequency filter serves as a filter that switches between the first band and the second band.
  • the phase shifters 21 and 22 are connected to both input and output terminals of the filter 12A, thereby In one band, the reduction of the ripple reduces the insertion loss.
  • FIG. 18 is a graph for explaining the factor of increasing the bandwidth of the high frequency filter 20A according to the second embodiment.
  • the pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2 are shown in FIG.
  • the impedance characteristics of the first circuit viewed from the node X1 when the switch SW1 is in the conductive state and the second circuit alone from the node X1 when the switch SW5 is in the conductive state are illustrated in FIG. Impedance characteristics are shown.
  • the impedance characteristics of the first circuit viewed from the node X2 when the switch SW2 is in the conductive state and the second circuit alone from the node X2 when the switch SW6 is in the conductive state are illustrated in FIG. Impedance characteristics are shown.
  • unit between node X1-X2 is shown by (d) of the figure.
  • the phase difference between the first circuit and the second circuit is shown in (e)
  • frequency f 11 which is the maximum of the impedance of the singular point as a maximum is the low frequency end of the second band
  • the phases of the phase shifters 21 and 22 are adjusted so as to be included in the frequency band from the frequency f A to the pass band low frequency of the filter 12 A (the cut-off frequency f C on the side).
  • at the maximum viewed from the node X 2 is the cut-off on the lower band side of the filter 12 A from the frequency f A at the lower band end of the second band
  • the phases of the phase shifters 21 and 22 are adjusted so as to be included in the frequency band up to the frequency f C.
  • the phase change is large, the phase difference with the filter 11A is large, and the amplitude difference is also large. Therefore, by the phase adjustment of the phase shifters 21 and 22, in the frequency range from the frequency f A to the cut-off frequency f C , the frequencies f 11 and f 22 that become the largest impedance among the singular points of the impedance maximum of the second circuit.
  • the high frequency signal in the above frequency range can be prevented from flowing into the filter 12A of the second circuit. That is, it is possible to suppress the wraparound of the signal to the second circuit in the frequency band in which the second circuit is attenuated. As a result, the ripple in the first band of the high frequency filter 20A is suppressed, and the insertion loss can be reduced.
  • the high frequency signal of the frequency included in the first band is adjusted by adjusting the phase difference of the first circuit and the second circuit to be in phase in a small frequency band by adjusting the phase of the phase shifters 21 and 22. Are not canceled each other after passing through the first circuit and the second circuit. Thereby, the ripple in the first band can be reduced and the insertion loss can be reduced.
  • the high frequency filter 20A sets the first band as the pass band due to the conduction state of the switches SW1, SW2, SW5, and SW6, it is possible to reduce the insertion loss of the first band.
  • FIG. 19 is a graph showing the pass characteristic when the off capacitance C off of the switch of the high frequency filter 20A according to the second embodiment is changed.
  • the switch element When the switch element is nonconductive, the impedance of the switch element is ideally infinite, but actually, the switch element is nonconductive due to the parasitic capacitance of the semiconductor element (FET or the like) constituting the switch element. It has a capacity component. As the off capacitance C off increases, the impedance of the switch element decreases.
  • the characteristics of the attenuation band in the vicinity of the high pass side of the pass band are degraded. More specifically, when the filter 11 is selected in which the switches SW1 and SW2 are turned on and the switches SW5 and SW6 are turned off and the second band is set to the pass band, the larger the off capacitance C off, the more signal is sent to the second circuit. Get around. For this reason, the attenuation characteristics on the second band high band side deteriorate, and a ripple is generated in the second band to deteriorate the insertion loss.
  • the pass band (third band) and the pass of the second circuit are affected by the influence of the first circuit disposed in the direction of the switches SW1 and SW2 to be turned off as shown in FIG.
  • the characteristics of the attenuation band near the lower band side deteriorate. More specifically, when the filter 12 is selected in which the switches SW1 and SW2 are nonconductive and the switches SW5 and SW6 are conductive and the third band is a pass band, the larger the off capacitance C off, the more signal is transmitted to the first circuit. Get around. As a result, the attenuation characteristics on the lower side of the third band deteriorate, and ripples occur in the third band to deteriorate the insertion loss.
  • the configuration shown in FIG. 20 can be mentioned as a configuration for suppressing the deterioration of the filter characteristics due to the off capacitance C off of the switch element as described above.
  • FIG. 20 is a circuit block diagram of the high frequency filter 30 according to the third embodiment.
  • the high frequency filter 30 shown in the figure includes the filters 11 and 12, the phase shifters 21 and 22, the switches SW 1, SW 2, SW 3, SW 4, SW 5, SW 6, SW 7 and SW 8, and the input and output terminals 110 and 120. And.
  • the high frequency filter 30 differs from the high frequency filter 20 according to the second embodiment in that switches SW3, SW4, SW7 and SW8 are arranged.
  • the high frequency filter 30 according to the present embodiment will not be described the same as the high frequency filter 20 according to the second embodiment, and different points will be mainly described.
  • the first circuit further includes a switch SW3 and a switch SW4.
  • the second circuit further includes a switch SW7 and a switch SW8.
  • the switch SW3 is a third switch element connected to a connection node between the filter 11 and the switch SW1 and to the ground, and switched between conduction and non-conduction exclusively with the switch SW1.
  • the switch SW4 is a fourth switch element connected to the connection node between the filter 11 and the switch SW2 and to the ground, and switched between conduction and non-conduction exclusively with the switch SW2.
  • the switch SW7 is a connection node between the phase shifter 21 and the switch SW5, and is a seventh switch element connected to the ground and switched between conduction and non-conduction exclusively with the switch SW5.
  • the switch SW8 is an eighth switch element connected to the connection node between the phase shifter 22 and the switch SW6 and to the ground, and switched between conduction and non-conduction exclusively with the switch SW6.
  • the high frequency filter 30 has two filters, the filter 11 whose second band is the Band 3 reception passband and the filter 12 whose third band is the Band 39 passband, and the phase shift is made between the filter 12 and the node X1.
  • the phase shifter 22 is disposed between the filter 12 and the node X2.
  • FIG. 21 is a graph showing the pass characteristic of the high frequency filter 30 according to the third embodiment.
  • the switches SW7 and SW8 are turned on, whereby the signal can be prevented from looping around in the second circuit. Therefore, the high frequency filter 30 (located SW7 & SW8) switches SW7 and SW8 are arranged, the same pass characteristic and the ideal SW (no C off). That is, the influence of the second circuit in which the switches SW5 and SW6 are nonconductive is reduced, the ripple in the second band is suppressed to reduce the insertion loss, and the attenuation characteristic in the vicinity of the second band can be improved.
  • the high frequency filter 30 (with SW3 & SW4) in which the switches SW3 and SW4 are arranged has a passing characteristic equivalent to that of the ideal SW (without Coff ). That is, the influence of the first circuit in which the switches SW1 and SW2 are nonconductive is reduced, the ripple in the third band is suppressed to reduce the insertion loss, and the attenuation characteristics in the vicinity of the third band can be improved.
  • phase shifters 21 and 22 are ideal elements, but here, specific configurations of the phase shifters 21 and 22 will be illustrated.
  • FIG. 22A is a circuit configuration diagram of a high frequency filter 20B according to a first modification of the second embodiment.
  • the high frequency filter 20B shown in the figure includes filters 11A and 12A, phase shifters 21B and 22B, switches SW1, SW2, SW5 and SW6, and input / output terminals 110 and 120.
  • the high frequency filter 20B according to the present modification differs from the high frequency filter 20A according to the second embodiment in that a specific circuit configuration of the phase shifter is shown.
  • the high frequency filter 20B according to the present modification will not be described the same as the high frequency filter 20A according to the second embodiment, and different points will be mainly described.
  • the phase shifter 21B includes two inductors Ls21 arranged in a series arm path connecting the node X1 and one terminal of the filter 12A, and a capacitor Cp21 connected to the node on the series arm path and the ground. It is a T-type first phase shifter.
  • the phase shifter 22B includes two inductors Ls22 arranged in a series arm path connecting the node X2 and the other terminal of the filter 12A, and a capacitor Cp22 connected to the node on the series arm path and the ground. It is a T-type second phase shifter.
  • the phase shifter 21 B and the phase shifter 22 B constitute a low pass filter circuit (LPF phase shifter).
  • the inductance values of the two inductors Ls 21 may be different. Further, two inductances Ls 21 may not be disposed, and may be one.
  • FIG. 22B is a circuit configuration diagram of a high frequency filter 20C according to Modification 2 of Embodiment 2.
  • the high frequency filter 20C shown in the figure includes filters 11A and 12A, phase shifters 21C and 22C, switches SW1, SW2, SW5 and SW6, and input / output terminals 110 and 120.
  • the high frequency filter 20C according to the present modification differs from the high frequency filter 20A according to the second embodiment in that a specific circuit configuration of the phase shifter is shown.
  • the high frequency filter 20C according to the present modification will not be described the same as the high frequency filter 20A according to the second embodiment, and different points will be mainly described.
  • the phase shifter 21C includes two capacitors Cs21 arranged in a series arm path connecting the node X1 and one terminal of the filter 12A, and an inductor Lp21 connected to the node on the series arm path and the ground. It is a T-type first phase shifter.
  • the phase shifter 22C includes two capacitors Cs22 arranged in a series arm path connecting the node X2 and the other terminal of the filter 12A, and an inductor Lp22 connected to the node on the series arm path and the ground. It is a T-type second phase shifter.
  • the phase shifters 21C and 22C constitute a high-pass filter circuit (HPF phase shifter).
  • the capacitance values of the two capacitors Cs21 may be different. Further, two capacitors Cs21 may not be disposed, and may be one.
  • the same in-passband characteristics as the high frequency filter 20A according to the second embodiment having the phase shifters 21 and 22 as ideal elements are obtained. That is, the phase shifters 21B and 22B of the second circuit, and the phase shifters 21C and 22C are the attenuation bands on the third band lower side of the second circuit, which is a frequency band where the amplitude difference between the first bands is large.
  • the phase is adjusted so that the frequency at which the impedance is maximized in the second band is located within the range of the lowest attenuation pole frequency in the second circuit to the 10 dB cutoff frequency on the second band lower side. There is.
  • FIG. 23 is a graph comparing the pass characteristics of the high frequency filters according to the first and second modifications of the second embodiment.
  • the circuit configuration of the phase shifter can be appropriately selected according to the required characteristics of the high frequency filter.
  • one of the first phase shifter and the second phase shifter is a low pass filter circuit, and the other is a high pass. It may be a type filter circuit.
  • a ⁇ -type phase shift having one of a capacitor or an inductor in a series arm circuit and the other of a capacitor or an inductor in a parallel arm circuit It may be a container. That is, the first phase shifter and the second phase shifter may be configured by two or more elements (secondary). Also, the first phase shifter and the second phase shifter may be delay lines composed of strip lines or micro strip lines.
  • a high frequency filter in addition to the first circuit and the second circuit, a high frequency filter will be described in which a circuit configured of a third filter and a switch element is connected to the node X1 and the node X2.
  • FIG. 24A is a circuit configuration diagram of a high frequency filter 40A according to a fourth embodiment.
  • the high frequency filter 40A shown in the figure includes filters 11A, 12A and 12Aa, phase shifters 21, 22, 21a and 22a, switches SW1, SW2, SW5, SW6, SW35 and SW36, input / output terminals 110 and 110. And 120.
  • the high frequency filter 40A according to the present embodiment has a second circuit (third circuit) including the filter 12Aa, the switches SW35 and SW36 between the node X1 and the node X2. The point of being added in parallel is different as a configuration.
  • the high frequency filter 40A according to the present embodiment will not be described the same as the high frequency filter 20A according to the second embodiment, and differences will be mainly described.
  • the filter 12Aa is a third filter that constitutes a circuit 2a together with the phase shifters 21a and 22a.
  • the switch SW35 is a switch element connected between the node X1 and the phase shifter 21a.
  • the switch SW36 is a switch element connected between the node X2 and the phase shifter 22a.
  • the 2a circuit composed of the filter 12Aa, the phase shifters 21a and 22a, and the switches SW35 and SW36 is connected to the nodes X1 and X2.
  • the phase shifter 21a is a third phase shifter connected to one terminal of the filter 12Aa.
  • the phase shifter 22a is a fourth phase shifter connected to the other terminal of the filter 12Aa. That is, the switch SW35, the phase shifter 21a, the filter 12Aa, the phase shifter 22a, and the switch SW36 are connected in series between the node X1 and the node X2 in this order.
  • the filter 12Aa includes series arm resonators s31 and s32 disposed on a path connecting the node X1 and the node X2, and parallel arm resonators p31 and p32 disposed between each node on the path and the ground. and p33.
  • the filter 12Aa constitutes a ladder type band pass filter.
  • the filter 12Aa is a band including a part of the frequency of the first band a (the fourth band) different in frequency from the first band, and has a narrower bandwidth than the first band a and the center frequency of the second band
  • a third band (fifth band) which is located on the higher side and has a frequency different from that of the third band, is used as a pass band.
  • Table 3 shows circuit parameters of the high frequency filter 40A according to the fourth embodiment.
  • the second band is the Band 3 reception passband (1805-1880 MHz) of LTE
  • the third band is the Band 39 passband of LTE (1880-1920 MHz)
  • the third a band is 1880-
  • the first band is (Band 3 reception pass band + Band 39 pass band) (1805-1920 MHz)
  • the first a band is 1805-1950 MHz.
  • FIG. 24B is a graph showing a change in pass characteristics due to the switch of the high frequency filter 40A according to the fourth embodiment.
  • the high frequency filter 40A is a variable frequency high frequency filter that can be changed to a filter of a plurality of bandwidths by appropriately switching on and off of the switches SW1, SW2, SW5, SW6, SW35 and SW36. In the present embodiment, five bandwidths can be switched.
  • the high frequency filter 40A becomes a filter that sets the first band as a pass band.
  • the switches SW1, SW2, SW35 and SW36 are made conductive and the switches SW5 and SW6 are made nonconductive, the filter becomes a pass band of the 1a band.
  • the switches SW1 and SW2 are turned on and the switches SW5, SW6, SW35 and SW36 are turned off as shown in (c) of FIG. 24B, the second band is a pass band. Further, as shown in (d) of FIG.
  • the phase shifters 21 and 22 are connected to both input and output terminals of the filter 12A, and the phase shifters 21a and 22a are connected to both input and output terminals of the filter 12Aa.
  • the insertion loss is reduced by reducing the ripple.
  • the high frequency filter 40A according to the fourth embodiment reduces the ripple and the insertion loss in the first band and the 1a band.
  • FIG. 25 is a graph for explaining the cause of wide band in the first band of the high-frequency filter 40A according to the fourth embodiment.
  • switches SW1, SW2, SW5 and SW6 are in a conducting state
  • switches SW35 and SW36 are in a non-conducting state.
  • the pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2 are shown in FIG.
  • the impedance characteristics of the first circuit viewed from the node X1 when the switch SW1 is in the conductive state and the second circuit alone from the node X1 when the switch SW5 is in the conductive state are illustrated in FIG. Impedance characteristics are shown.
  • the impedance characteristics of the first circuit viewed from the node X2 when the switch SW5 is in the conductive state and the second circuit alone from the node X2 when the switch SW6 is in the conductive state are illustrated in FIG. Impedance characteristics are shown.
  • unit between node X1-X2 is shown by (d) of the figure.
  • the phase difference between the first circuit and the second circuit is shown in (e) of FIG.
  • frequency f 11 which is the maximum of the impedance of the singular point as a maximum is The phases of the phase shifters 21 and 22 are adjusted so as to be included in the frequency band from the frequency f A at the low band end of the second band to the cutoff frequency f C on the low band side of the pass band of the filter 12A. There is.
  • frequency f 22 which is the maximum of the impedance of the singular point as a maximum is the frequency f A of the low-frequency end of the second band, the filter 12A
  • the phases of the phase shifters 21 and 22 are adjusted so as to be included in the frequency band up to the cutoff frequency f C on the lower side of the pass band.
  • the phase change is large, the phase difference with the filter 11A is large, and the amplitude difference is also large. Therefore, by the phase adjustment of the phase shifters 21 and 22, the impedance maximum of the second circuit is in the frequency range from the frequency f A of the attenuation pole on the low pass side of the filter 12A to the low 10 dB cut off frequency f C on the low frequency side.
  • the high frequency signal of the frequency included in the first band is adjusted by adjusting the phase difference of the first circuit and the second circuit to be in phase in a small frequency band by adjusting the phase of the phase shifters 21 and 22. Are not canceled each other after passing through the first circuit and the second circuit. Thereby, in the frequency included in the first band of the high frequency filter 40A, it is possible to suppress the ripple in the first band and reduce the insertion loss.
  • the first circuit is generated in the second band.
  • the second circuit have the same phase (0 ° or -360 °) in at least one point (broken line circle mark: three points in the same figure), and in the third band, the first circuit and the second circuit There may be at least one frequency (dotted line circle in the same figure: one place) in which the phase is the same phase (0 ° or -360 °).
  • variable-frequency high frequency filter 40A capable of switching the first band, the second band, and the third band.
  • FIG. 26 is a graph for explaining the factor of broadening the band in the 1a band of the high frequency filter 40A according to the fourth embodiment.
  • band 1a is selected as the pass band of high-frequency filter 40A
  • switches SW1, SW2, SW35 and SW36 are in a conducting state
  • switches SW5 and SW6 are in a non-conducting state.
  • unit between node X1-X2 is shown by (a) of the figure.
  • the impedance characteristics are shown in (b) of the figure when the first circuit single-piece when the switch SW1 is in the conductive state from the node X1 and the 2a-th single circuit when the switch SW35 is in the conductive state.
  • the impedance characteristics are shown in (c) of the figure when the first circuit single-piece when the switch SW2 is in a conductive state from the node X2 and the 2a-th single circuit when the switch SW36 is in a conductive state.
  • unit between node X1-X2 is shown by (d) of the figure.
  • the phase difference between the first circuit and the 2a circuit is shown in (e) of FIG.
  • frequency f 11 which is the maximum of the impedance of the singular point as a maximum is
  • the phases of the phase shifters 21a and 22a are adjusted so as to be included in the frequency band from the frequency f A at the low band end of the second band to the cutoff frequency f C at the low band side of the pass band of the filter 12A. There is.
  • frequency f 22 which is the maximum of the impedance of the singular point as a maximum is the frequency f A of the low-frequency end of the second band, the filter 12A
  • the phases of the phase shifters 21a and 22a are adjusted so as to be included in the frequency band up to the cutoff frequency f C on the lower side of the pass band.
  • the phase change is large, the phase difference with the filter 11A becomes large, and the amplitude difference also becomes large. Therefore, the phase adjustment of the phase shifter 21a and 22a, the frequency range from the frequency f A of the attenuation pole of the passband low frequency side of the filter 12Aa to low frequency side 10dB cut-off frequency f C, the impedance maximum of the 2a circuit
  • the phase adjustment of the phase shifter 21a and 22a the frequency range from the frequency f A of the attenuation pole of the passband low frequency side of the filter 12Aa to low frequency side 10dB cut-off frequency f C, the impedance maximum of the 2a circuit
  • a high frequency signal of the frequency included in the 1a band are not canceled after passing through the first circuit and the 2a circuit.
  • the ripple in the 1a band can be suppressed and the insertion loss can be reduced.
  • variable-frequency high frequency filter 40A capable of switching the 1a band, the 2nd band, and the 3rd band.
  • the high frequency filter 40A it is possible to reduce the insertion loss by reducing the ripples in the first band and the 1a band while increasing the variation of switching of the pass band.
  • Embodiment 4 In the present embodiment, a high frequency filter configured of a filter having a longitudinally coupled resonator will be described.
  • FIG. 27A is a circuit configuration diagram of a high frequency filter 50A according to a fifth embodiment.
  • the high frequency filter 50A shown in the figure includes filters 11B and 12A, phase shifters 21D and 22D, switches SW1, SW2, SW5 and SW6, and input / output terminals 110 and 120.
  • the high frequency filter 50A according to the present embodiment differs from the high frequency filter 20A according to the second embodiment in the circuit configurations of the first circuit and the phase shifter.
  • the high frequency filter 50A according to the present embodiment will not be described the same as the high frequency filter 20A according to the second embodiment, and differences will be mainly described.
  • the filter 11B includes series arm resonators s11, s12 and s13 disposed on a first path connecting the node X1 and the node X2, and parallel arms disposed between the respective nodes on the first path and the ground.
  • Resonators p11 and p12 and a longitudinally coupled resonator 15A connected to series arm resonators s12 and s13 are provided.
  • the filter 11B constitutes a band pass filter.
  • the longitudinally coupled resonator 15A is configured of five IDT electrodes and reflectors disposed at both ends in the arrangement direction of the five IDT electrodes.
  • the number of IDT electrodes of the longitudinally coupled resonator may be two or more.
  • the phase shifter 21D is a first phase shifter connected to one terminal of the filter 12A.
  • the phase shifter 22D is a second phase shifter connected to the other terminal of the filter 12A. That is, the phase shifter 21D, the filter 12A, and the phase shifter 22D are connected in series between the node X1 and the node X2 in this order.
  • the phase shifter 21D includes one capacitor Cs21 arranged in a series arm path connecting the node X1 and one terminal of the filter 12A, and one inductor Lp21 connected to the node on the series arm path and the ground. It is configured.
  • the phase shifter 22D includes one capacitor Cs22 arranged in a series arm path connecting the node X2 and the other terminal of the filter 12A, and one inductor Lp22 connected to the node on the series arm path and the ground. It is configured.
  • Table 4 shows circuit parameters of the high frequency filter 50A according to the fifth embodiment.
  • the filter 11B since the filter 11B includes the longitudinally coupled resonator 15A, the filter 11B has filter characteristics that are advantageous for attenuation characteristics. Therefore, high attenuation characteristics are not required for the phase shifters 21D and 22D. From this point of view, each of the phase shifters 21D and 22D can be configured with one capacitor and one inductor.
  • the circuit configuration of the phase shifters 21D and 22D is not limited to the above configuration.
  • a plurality of capacitors and inductors may be disposed, and connection configurations such as T-type and ⁇ -type are also arbitrary.
  • FIG. 27B is a graph showing the change of the passage characteristic due to the switch of the high frequency filter 50A according to the fifth embodiment.
  • the high frequency filter 50A uses the first band (Band 3 reception pass band + Band 39 pass band) as a pass band. It becomes.
  • the filters SW1 and SW2 are made conductive and the switches SW5 and SW6 are not made conductive, the filter becomes a pass band that is the second band (Band 3 reception pass band).
  • the switches SW1 and SW2 are turned off and the switches SW5 and SW6 are turned on, the filter becomes a pass band that is the third band (Band 39 pass band).
  • the switches SW1 and SW2 are disposed on the first circuit side, and the switches SW5 and SW6 are disposed on the second circuit side.
  • the switches are connected only to the first circuit side. It may be a configuration or a configuration in which a switch is connected only to the second circuit side.
  • the high frequency filter serves as a filter that switches between the first band and the third band. Further, in the case where the switch is connected only to the second circuit side, the high frequency filter serves as a filter that switches between the first band and the second band.
  • the phase shifters 21D and 22D are connected to both input and output terminals of the filter 12A, thereby the second high frequency filter 50A In one band, the reduction of the ripple reduces the insertion loss.
  • FIG. 28 is a graph for explaining the factor of increasing the bandwidth of the high frequency filter 50A according to the fifth embodiment.
  • the pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2 are shown in FIG.
  • the impedance characteristics of the first circuit viewed from the node X1 when the switch SW1 is in the conductive state and the second circuit alone from the node X1 when the switch SW5 is in the conductive state are illustrated in FIG. Impedance characteristics are shown.
  • the impedance characteristics of the first circuit viewed from the node X2 when the switch SW2 is in the conductive state and the second circuit alone from the node X2 when the switch SW6 is in the conductive state are illustrated in FIG. Impedance characteristics are shown.
  • unit between node X1-X2 is shown by (d) of the figure.
  • the phase difference between the first circuit and the second circuit is shown in (e)
  • frequency f 11 which is the maximum of the impedance of the singular point as a maximum is the low frequency end of the second band
  • the phases of the phase shifters 21D and 22D are adjusted so as to be included in the frequency band from the frequency f A to the cut-off frequency f C on the low pass side of the filter 12A.
  • frequency f 22 which is the maximum of the impedance of the singular point as a maximum is the frequency f A of the low-frequency end of the second band, the filter 12A
  • the phases of the phase shifters 21D and 22D are adjusted so as to be included in the frequency band up to the cutoff frequency f C on the lower side of the pass band.
  • the phase change is large, the phase difference with the filter 11B is large, and the amplitude difference is also large. Therefore, the phase adjustment of the phase shifter 21D and 22D, in the frequency range from the frequency f A of the attenuation pole of the passband low frequency side of the filter 12A to the low-frequency side 10dB cut-off frequency f C, the impedance maximum of the second circuit by matching the frequency f 11 and f 22 with the maximum of the impedance of the singular point, the high-frequency signal of the frequency range can be prevented from flowing into the filter 12A of the second circuit. That is, it is possible to suppress the wraparound of the signal to the second circuit in the frequency band in which the second circuit is attenuated. This makes it possible to suppress the ripple in the first band of the high frequency filter 50A and reduce the insertion loss.
  • the high frequency signal of the frequency included in the first band is adjusted by adjusting the phase difference of the first circuit and the second circuit to be in phase in a small frequency band by adjusting the phase of the phase shifters 21D and 22D. Are not canceled each other after passing through the first circuit and the second circuit. Thereby, at the frequency included in the first band of the high frequency filter 50A, the ripple in the first band can be suppressed and the insertion loss can be reduced.
  • the high frequency filter 50A sets the first band as the pass band due to the conduction state of the switches SW1, SW2, SW5, and SW6, it is possible to reduce the insertion loss of the first band.
  • the first circuit is generated in the second band.
  • the second circuit have the same phase (0 ° or -360 °) in at least one point (two dotted circle marks in the figure), and in the third band, the first circuit and the second circuit There may be at least one frequency (dotted line circle in the same figure: one place) in which the phase is the same phase (0 ° or -360 °).
  • the filter 11B is configured by the longitudinally coupled resonator 15A, the sharpness can be improved and the amount of attenuation can be improved on the low frequency band side of the high frequency filter 50A.
  • a high frequency filter configured of a first filter and a second filter having a parallel arm circuit to which an impedance circuit having a switch element is added will be described.
  • FIG. 29A is a circuit configuration diagram of a high frequency filter 60A according to a sixth embodiment.
  • the high frequency filter 60A shown in the figure includes filters 11C and 12B, phase shifters 21C and 22C, switches SW1, SW2, SW5 and SW6, and input / output terminals 110 and 120.
  • the high frequency filter 60A according to the present embodiment is different from the high frequency filter 20C according to the second modification of the second embodiment in the circuit configuration of the first filter and the second filter.
  • the high frequency filter 60A according to the present embodiment will not be described the same as the high frequency filter 20C according to the second modification of the second embodiment, and differences will be mainly described.
  • the filter 11C includes three series arm circuits (series arm resonators s11, s12 and s13) and parallel arm circuits 101, 102, 103 and 104.
  • the series arm resonators s11 to s13 are provided on a first path connecting the nodes X1 and X2.
  • Each of parallel arm circuits 101 to 104 is a parallel arm circuit connected to each node on the first path connecting nodes X1 and X2 and to the ground.
  • the parallel arm circuit 101 includes a parallel arm resonator p11 and an impedance circuit 111 connected in series to the parallel arm resonator p11.
  • the impedance circuit 111 has a capacitor Cp11 and a switch SW11.
  • the switch SW11 is a ninth switch element connected in parallel to the capacitor Cp11. Since the impedance circuit 111 is configured by a parallel circuit of the capacitor Cp11 and the switch SW11, the resonance frequency of the parallel arm circuit 101 is switched to the high frequency side by switching the switch SW11 from conduction to non-conduction.
  • the parallel arm circuit 102 includes a parallel arm resonator p12 and an impedance circuit 112 connected in series to the parallel arm resonator p12.
  • the impedance circuit 112 includes a capacitor Cp12 and a switch SW12.
  • the switch SW12 is a ninth switch element connected in parallel to the capacitor Cp12. Since the impedance circuit 112 is constituted by a parallel circuit of the capacitor Cp12 and the switch SW12, the resonance frequency of the parallel arm circuit 102 is switched to the high frequency side by switching the switch SW12 from conduction to non-conduction.
  • the parallel arm circuit 104 includes a parallel arm resonator p14 and an impedance circuit 114 connected in series to the parallel arm resonator p14.
  • the impedance circuit 114 has a capacitor Cp14 and a switch SW14.
  • the switch SW14 is a ninth switch element connected in parallel to the capacitor Cp14. Since the impedance circuit 114 is formed of a parallel circuit of the capacitor Cp14 and the switch SW14, the resonance frequency of the parallel arm circuit 104 is switched to the high frequency side by switching the switch SW14 from conduction to non-conduction.
  • the parallel arm circuit 103 has a parallel arm resonator p13 and an impedance circuit 113 connected in series to the parallel arm resonator p13.
  • the impedance circuit 113 includes an inductor Lp13 and a switch SW13.
  • the switch SW13 is a ninth switch element connected in parallel to the inductor Lp13. Since the impedance circuit 113 is constituted by a parallel circuit of the inductor Lp13 and the switch SW13, the resonance frequency of the parallel arm circuit 103 is switched to the low frequency side by switching the switch SW13 from conduction to non-conduction.
  • the filter 11C switches the conduction and non-conduction of the switches SW11, SW12, SW13, and SW14 to lower the passband.
  • the steepness on the band side and the frequency at the low band edge of the pass band are variable.
  • the filter 12B includes two series arm circuits (series arm resonators s21 and s22) and parallel arm circuits 201, 202 and 203.
  • the series arm resonators s21 and s22 are provided on a second path connecting the nodes X1 and X2.
  • Each of the parallel arm circuits 201 to 203 is a parallel arm circuit connected to each node on the second path connecting the nodes X1 and X2 and the ground.
  • the parallel arm circuit 201 includes a parallel arm resonator p21 and an impedance circuit 211 connected in series to the parallel arm resonator p21.
  • the impedance circuit 211 includes a capacitor Cp21 and a switch SW21.
  • the switch SW21 is a ninth switch element connected in parallel to the capacitor Cp21. Since the impedance circuit 211 is configured by a parallel circuit of the capacitor Cp21 and the switch SW21, the resonance frequency of the parallel arm circuit 201 is switched to the high frequency side by switching the switch SW21 from conduction to non-conduction.
  • the parallel arm circuit 202 includes a parallel arm resonator p22 and an impedance circuit 212 connected in series to the parallel arm resonator p22.
  • the impedance circuit 212 includes a capacitor Cp22 and a switch SW22.
  • the switch SW22 is a ninth switch element connected in parallel to the capacitor Cp22. Since the impedance circuit 212 is configured by a parallel circuit of the capacitor Cp22 and the switch SW22, the resonance frequency of the parallel arm circuit 202 is switched to the high frequency side by switching the switch SW22 from conduction to non-conduction.
  • the parallel arm circuit 203 has a parallel arm resonator p23 and an impedance circuit 213 connected in series to the parallel arm resonator p23.
  • the impedance circuit 213 includes a capacitor Cp23 and a switch SW23.
  • the switch SW23 is a ninth switch element connected in parallel to the capacitor Cp23. Since the impedance circuit 213 is formed of a parallel circuit of the capacitor Cp23 and the switch SW23, the resonance frequency of the parallel arm circuit 203 is switched to the high frequency side by switching the switch SW23 from conduction to non-conduction.
  • the sharpness of the low pass band and the frequency of the low end of the pass band are varied by switching between conduction and non-conduction of the switches SW21, SW22, and SW23.
  • Table 5 shows circuit parameters of the high frequency filter 60A according to the sixth embodiment.
  • FIG. 29B is a graph showing the change of the passage characteristic due to the switching of the switch of the high frequency filter 60A according to the sixth embodiment.
  • the high frequency filter 60A is a variable-frequency high frequency filter that can be changed to a filter of a plurality of bandwidths by appropriately switching between conduction and non-conduction of the switches SW1, SW2, SW5, SW6, SW11 to SW14, and SW21 to SW23. It becomes. In the present embodiment, six pass bands can be switched.
  • the third band It becomes a filter which makes Band 39 pass band) a pass band.
  • the switches SW1, SW2, SW5, SW6, SW13, and SW21 to SW23 are made conductive and the switches SW11, SW12, and SW14 are made nonconductive, for the first band.
  • the attenuation pole at the low band side of the pass band and the low band end of the pass band become a filter having the 1 b band shifted to the high frequency side as the pass band.
  • the passband lower band end is a filter that uses the 3b band shifted to the high frequency side as the passband.
  • phase shifters 21C and 22C are connected to both input and output terminals of the filter 12B, whereby ripples are reduced in the first band and the 1b band of the high frequency filter 60A. Insertion loss is reduced.
  • the high frequency filter 60A according to the sixth embodiment reduces the ripple and the insertion loss in the first band and the 1b band.
  • FIG. 30 is a graph for explaining the factor of broadening the band in the first band of the high-frequency filter 60A according to the sixth embodiment.
  • the pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2 are shown in FIG.
  • the impedance characteristics of the first circuit viewed from the node X1 when the switch SW1 is in the conductive state and the second circuit alone from the node X1 when the switch SW5 is in the conductive state are illustrated in FIG. Impedance characteristics are shown.
  • the impedance characteristics of the first circuit viewed from the node X2 when the switch SW2 is in the conductive state and the second circuit alone from the node X2 when the switch SW6 is in the conductive state are illustrated in FIG. Impedance characteristics are shown.
  • unit between node X1-X2 is shown by (d) of the figure.
  • frequency f 11 which is the maximum of the impedance of the singular point as a maximum is
  • the phases of the phase shifters 21C and 22C are adjusted so as to be included in the frequency band from the frequency f A at the low band end of the second band to the cutoff frequency f C at the low band side of the pass band of the filter 12B. There is.
  • frequency f 22 which is the maximum of the impedance of the singular point as a maximum is the frequency f A of the low-frequency end of the second band, the filter 12B
  • the phases of the phase shifters 21C and 22C are adjusted so as to be included in the frequency band up to the cutoff frequency f C on the lower side of the pass band.
  • the phase change is large, the phase difference with the filter 11C is large, and the amplitude difference is also large. Therefore, by the phase adjustment of the phase shifters 21C and 22C, in the frequency range from the frequency f A to the cut-off frequency f C , the frequencies f 11 and f 22 that become the largest impedance among the singular points of the impedance maximum of the second circuit.
  • the high frequency filter 60A sets the first band as the pass band due to the conduction state of the switches SW1, SW2, SW5, and SW6, it is possible to reduce the insertion loss of the first band.
  • the high frequency signal of the frequency included in the first band is adjusted by adjusting the phase difference of the first circuit and the second circuit to be in phase in a small frequency band by phase adjustment of the phase shifters 21C and 22C.
  • the ripple in the first band can be suppressed and the insertion loss can be reduced.
  • the first circuit and the second circuit are the same in the second band.
  • variable-frequency high frequency filter 60A capable of switching the first band, the second band, and the third band.
  • FIG. 31 is a graph for explaining the factor of broadening the band in the 1b band of the high frequency filter 60A according to the sixth embodiment.
  • the pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2 are shown in FIG.
  • the impedance characteristics of the first circuit viewed from the node X1 when the switch SW1 is in the conductive state and the second circuit alone from the node X1 when the switch SW5 is in the conductive state are illustrated in FIG. Impedance characteristics are shown.
  • the impedance characteristics of the first circuit viewed from the node X2 when the switch SW2 is in the conductive state and the second circuit alone from the node X2 when the switch SW6 is in the conductive state are illustrated in FIG. Impedance characteristics are shown.
  • unit between node X1-X2 is shown by (d) of the figure.
  • frequency f 11 which is the maximum of the impedance of the singular point as a maximum is
  • the phases of the phase shifters 21C and 22C are adjusted so as to be included in the frequency band from the frequency f A at the low band end of the second band to the cutoff frequency f C at the low band side of the pass band of the filter 12B. There is.
  • frequency f 22 which is the maximum of the impedance of the singular point as a maximum is the frequency f A of the low-frequency end of the second band, the filter 12B
  • the phases of the phase shifters 21C and 22C are adjusted so as to be included in the frequency band up to the cutoff frequency f C on the lower side of the pass band.
  • the phase change is large, the phase difference with the filter 11C is large, and the amplitude difference is also large. Therefore, by the phase adjustment of the phase shifters 21C and 22C, in the frequency range from the frequency f A to the cut-off frequency f C , the frequencies f 11 and f 22 that become the largest impedance among the singular points of the impedance maximum of the second circuit.
  • the high frequency filter 60A sets the band 1b as the pass band due to the conduction state of the switches SW1, SW2, SW5, and SW6, it is possible to reduce the insertion loss of the band 1b.
  • the phase difference of the first circuit and the second circuit so that the amplitude difference between the first circuit and the second circuit becomes the same phase in a small frequency band, the high frequency signal of the frequency included in the 1b band Are not canceled each other after passing through the first circuit and the second circuit. Thereby, it is possible to suppress the ripple in the 1b band and reduce the insertion loss.
  • the first circuit and the second circuit are the same in the 2b band.
  • variable-frequency high frequency filter 60A capable of switching the 1b band, the 2b band, and the 3b band.
  • the high frequency filter 60A it is possible to reduce the insertion loss by reducing the ripples in the first band and the 1b band while increasing the variation of switching of the pass band.
  • series arm resonators and the number of parallel arm circuits are appropriately determined according to the required specification of the filter characteristics. Also, instead of the series arm resonator, circuit elements such as an inductor and a capacitor may be arranged.
  • the high frequency filter 60A as a unit circuit constituted by one series arm resonator and one parallel arm circuit connected to the series arm, which constitute each of the filters 11C and 12B, There are modifications as shown in FIGS. 32A-32D and FIGS. 33A-33G.
  • FIG. 32A is a circuit configuration diagram of Modification Example 1 of each filter constituting the high frequency filter 60A according to the sixth embodiment.
  • the filter 13A shown in the figure is composed of a series arm circuit 151 and a parallel arm circuit.
  • the parallel arm circuit has a parallel arm resonator p1 and an impedance circuit connected in series with each other.
  • the impedance circuit has a switch (ninth switch element) and a capacitor connected in parallel to one another. Switching between conduction and non-conduction of the switch switches the resonant frequency of the parallel arm circuit.
  • FIG. 32B is a circuit configuration diagram of Modification Example 2 of each filter constituting the high frequency filter 60A according to the sixth embodiment.
  • the filter 13B shown in the figure is composed of a series arm circuit 151 and a parallel arm circuit.
  • the parallel arm circuit has a parallel arm resonator p1 and an impedance circuit connected in series with each other.
  • the impedance circuit has a switch (ninth switch element) and an inductor connected in parallel to each other. Switching between conduction and non-conduction of the switch switches the resonant frequency of the parallel arm circuit.
  • FIG. 32C is a circuit configuration diagram of Modification Example 3 of each filter constituting the high-frequency filter 60A according to Example 6.
  • FIG. The filter 13C shown in the same drawing is configured of a series arm circuit 151 and a parallel arm circuit.
  • the parallel arm circuit has a parallel arm resonator p1 and an impedance circuit connected in series with each other.
  • the impedance circuit has a series circuit of a switch (ninth switch element) and an inductor connected in parallel to each other and a capacitor. Switching between conduction and non-conduction of the switch switches the resonant frequency of the parallel arm circuit.
  • FIG. 32D is a circuit configuration diagram of Modification Example 4 of each filter configuring the high-frequency filter 60A according to the sixth embodiment.
  • the filter 13D shown in the figure is constituted by a series arm circuit 151 and a parallel arm circuit.
  • the parallel arm circuit has a parallel arm resonator p1 and an impedance circuit connected in series with each other.
  • the impedance circuit has a series circuit of a switch (ninth switch element) and a capacitor connected in parallel to each other and an inductor. Switching between conduction and non-conduction of the switch switches the resonant frequency of the parallel arm circuit.
  • FIG. 33A is a circuit configuration diagram of Modification Example 5 of each filter configuring the high-frequency filter 60A according to the sixth embodiment.
  • the filter 14A shown in the figure is configured of a series arm circuit 151 and a parallel arm circuit.
  • the parallel arm circuit includes a parallel arm resonator p1 and a parallel arm resonator p2 and a switch (ninth switch element) connected in series with each other.
  • the resonance frequency of the parallel arm resonator p1 is lower than the resonance frequency of the parallel arm resonator p2, and the antiresonance frequency of the parallel arm resonator p1 is set lower than the antiresonance frequency of the parallel arm resonator p2. Switching between conduction and non-conduction of the switch switches at least one of the resonant frequency and the antiresonant frequency of the parallel arm circuit.
  • FIG. 33B is a circuit configuration diagram of Modification Example 6 of the filters configuring the high-frequency filter 60A according to the sixth embodiment.
  • the filter 14B shown in the figure is composed of a series arm circuit 151 and a parallel arm circuit.
  • the parallel arm circuit includes a parallel arm resonator p1, a parallel arm resonator p2 connected in series with one another, and an impedance circuit.
  • the resonance frequency of the parallel arm resonator p1 is different from the resonance frequency of the parallel arm resonator p2, and the antiresonance frequency of the parallel arm resonator p1 is different from the antiresonance frequency of the parallel arm resonator p2.
  • the impedance circuit has a switch (ninth switch element) and a capacitor connected in parallel to one another.
  • the parallel arm circuit has two resonance frequencies and two antiresonance frequencies, and switching between conduction and non-conduction of the switch allows at least one of the resonance frequencies of the parallel arm circuit and at least one of the antiresonance frequencies to be both. Switch.
  • FIG. 33C is a circuit configuration diagram of Modification Example 7 of each of the filters forming the high-frequency filter 60A according to the sixth embodiment.
  • the filter 14C shown in the same drawing is configured of a series arm circuit 151 and a parallel arm circuit.
  • the parallel arm circuit includes a parallel arm resonator p1 and a first impedance circuit connected in series with each other, and a parallel arm resonator p2 and a second impedance circuit connected with each other in series.
  • the resonance frequency of the parallel arm resonator p1 is different from the resonance frequency of the parallel arm resonator p2, and the antiresonance frequency of the parallel arm resonator p1 is different from the antiresonance frequency of the parallel arm resonator p2.
  • the first impedance circuit includes a switch SW1 (ninth switch element) and a capacitor connected in parallel to each other.
  • the second impedance circuit includes a switch SW2 (ninth switch element) and a capacitor connected in parallel to each other.
  • the parallel arm circuit has two resonant frequencies and two antiresonant frequencies, and switching between conduction and non-conduction of each switch causes at least one of the resonant frequencies of the parallel arm circuit and at least one of the antiresonant frequencies to It switches together.
  • FIG. 33D is a circuit configuration diagram of Modified Example 8 of the filters constituting the high frequency filter 60A according to the sixth embodiment.
  • the filter 14D shown in the figure is configured of a series arm circuit 151 and a parallel arm circuit.
  • the parallel arm circuit has a parallel arm resonator p1 and an impedance circuit connected in series with each other.
  • the impedance circuit includes a switch SW1 (ninth switch element) and a parallel arm resonator p2 connected in parallel to each other.
  • the resonance frequency of the parallel arm resonator p1 is different from the resonance frequency of the parallel arm resonator p2, and the antiresonance frequency of the parallel arm resonator p1 is different from the antiresonance frequency of the parallel arm resonator p2. Switching between conduction and non-conduction of the switch switches the frequency and number of resonant frequencies and antiresonant frequencies of the parallel arm circuit.
  • FIG. 33E is a circuit configuration diagram of Modification 9 of the filters constituting the high-frequency filter 60A according to the sixth embodiment.
  • the filter 14E shown in the figure is configured of a series arm circuit 151 and a parallel arm circuit.
  • the parallel arm circuit has a configuration in which a circuit in which parallel arm resonators p1 and p2 are connected in parallel and an impedance circuit are connected in series.
  • the resonance frequency of the parallel arm resonator p1 is lower than the resonance frequency of the parallel arm resonator p2, and the antiresonance frequency of the parallel arm resonator p1 is set lower than the antiresonance frequency of the parallel arm resonator p2.
  • the impedance circuit has a switch SW1 (ninth switch element) and a capacitor connected in parallel to each other.
  • the parallel arm circuit has two resonant frequencies and two antiresonant frequencies, and switching between conduction and non-conduction of the switch switches the two resonant frequencies of the parallel arm circuit.
  • FIG. 33F is a circuit configuration diagram of Modification Example 10 of the filters configuring the high-frequency filter 60A according to the sixth embodiment.
  • the filter 14F shown in the figure is configured of a series arm circuit 151 and a parallel arm circuit.
  • the parallel arm circuit has a configuration in which a circuit in which parallel arm resonators p1 and p2 are connected in parallel and an impedance circuit are connected in series.
  • the resonance frequency of the parallel arm resonator p1 is lower than the resonance frequency of the parallel arm resonator p2, and the antiresonance frequency of the parallel arm resonator p1 is set lower than the antiresonance frequency of the parallel arm resonator p2.
  • the impedance circuit has a switch SW1 (ninth switch element) and an inductor connected in parallel to each other.
  • the parallel arm circuit has two resonant frequencies and two antiresonant frequencies, and switching between conduction and non-conduction of the switch switches the two resonant frequencies of the parallel arm circuit.
  • FIG. 33G is a circuit configuration diagram of Modification 11 of the filters constituting the high frequency filter 60A according to Embodiment 6.
  • the filter 14 ⁇ / b> G shown in the figure is configured of a series arm circuit and a parallel arm circuit 161.
  • the series arm circuit has a configuration in which a series arm resonator s1 and a series circuit of a capacitor and a switch (ninth switch element) are connected in parallel. Switching between conduction and non-conduction of the switch switches the antiresonant frequency of the series arm circuit.
  • the high frequency filters described in the first to sixth embodiments can also be applied to multiplexers and high frequency front end circuits corresponding to a system having a large number of used bands. Therefore, in the present embodiment, such a multiplexer, a high frequency front end circuit, and a communication apparatus will be described.
  • FIG. 34 is a block diagram of the communication device 9 and its peripheral circuits according to the sixth embodiment.
  • the communication apparatus 9 includes a switch group 210 formed of a plurality of switches, a filter group 220 formed of a plurality of filters, a transmission side switch 231, and reception side switches 251, 252 and 253. , Switches 271 and 272, transmission amplification circuits 241, 242 and 243, reception amplification circuits 261, 262, 263 and 264, RF signal processing circuit (RFIC) 7, and baseband signal processing circuit (BBIC) 8 , And an antenna element 5.
  • RFIC RF signal processing circuit
  • BBIC baseband signal processing circuit
  • Switch group 210 connects antenna element 5 and a signal path corresponding to a predetermined band in accordance with a control signal from a control unit (not shown), and is formed of, for example, a plurality of SPST type switches.
  • the number of signal paths connected to the antenna element 5 is not limited to one, and may be plural. That is, the communication device 9 may support carrier aggregation.
  • the filter group 220 is composed of, for example, a plurality of filters (including duplexers) having the next band in the pass band.
  • the band is a transmission band of (i-Tx) Band 3, 4 and 66, a reception band of (i-Rx) Band 3 or Band 3, 39, a transmission band of (ii-Tx) Band 25, (ii- Rx) Band 25 reception band, (iii-Tx) Band 1 or 65 transmission band, (iii-Rx) Band 1, 4, 65, 66 reception band, (iv-Tx) Band 30 transmission band, (iv-Rx) (B) Band 7 transmission band, (v) Band 7 or Band 7 or 38 reception band, (vii) Band 39 transmission / reception band, (viii) Band 34 transmission / reception band, (ix) Band 40 transmission / reception band, x) Band 41 or 38 transmission / reception band.
  • the transmission side switch 231 is a switch circuit having a plurality of selection terminals connected to a plurality of transmission side signal paths on the low band side where the center frequency of the filter group 220 is low and a common terminal connected to the transmission amplification circuit 241.
  • the reception side switch 251 is a switch circuit having a plurality of selection terminals connected to a plurality of reception side signal paths and a common terminal connected to the reception amplification circuit 261.
  • the reception side switch 252 is a switch circuit having a plurality of selection terminals connected to a plurality of reception side signal paths and a common terminal connected to the reception amplification circuit 262.
  • the reception side switch 253 is a switch circuit having a reception side signal path, a plurality of selection terminals connected to the selection terminal of the switch 271, and a common terminal connected to the reception amplification circuit 263.
  • the switch 271 is a switch circuit having a common terminal connected to the transmission / reception path of a predetermined band (here, Band 39), and two selection terminals connected to the selection terminal of the reception side switch 253 and the transmission amplification circuit 242.
  • the switch 272 is a switch circuit having a plurality of selection terminals connected to a plurality of transmission and reception paths, and two selection terminals connected to the transmission amplification circuit 243 and the reception amplification circuit 264.
  • These switches are provided in the subsequent stage (in this case, the subsequent stage in the reception side signal path) of the filter group 220, and the connection state is switched according to the control signal from the control unit (not shown).
  • the transmission amplification circuits 241, 242 and 243 are power amplifiers for power-amplifying high frequency transmission signals in a predetermined frequency band.
  • the reception amplifier circuits 261, 262, 263 and 264 are each a low noise amplifier for power-amplifying a high frequency reception signal of a predetermined frequency band.
  • the RF signal processing circuit (RFIC) 7 is a circuit that processes a high frequency signal transmitted and received by the antenna element 5. Specifically, the RF signal processing circuit (RFIC) 7 performs signal processing of a high frequency signal (here, a high frequency received signal) input from the antenna element 5 via the receiving signal path by down conversion or the like, and the signal The received signal generated by processing is output to the baseband signal processing circuit (BBIC) 8. Further, the RF signal processing circuit (RFIC) 7 performs signal processing on the transmission signal input from the baseband signal processing circuit (BBIC) 8 by up conversion and the like, and generates a high frequency signal (high frequency in this case) generated by the signal processing. The transmission signal is output to the transmission side signal path.
  • a high frequency signal here, a high frequency received signal
  • BBIC baseband signal processing circuit
  • the communication apparatus 9 configured in this manner includes the filter according to any one of the first to fifth embodiments as at least one of the filters having any of the above (i-Tx) to (x) in the pass band. .
  • the reception amplifier circuits 261, 262, 263 and 264 and the control unit constitute a high frequency front end circuit 6.
  • the switch group 210 and the filter group 220 constitute a multiplexer.
  • the filter group 220 may be connected to the common terminal through the switch group 210, and one of the first to fifth embodiments. A plurality of filters may be directly connected to the common terminal.
  • control unit may be included in an RF signal processing circuit (RFIC), or may constitute a switch IC together with each switch controlled by the control unit. Good.
  • RFIC RF signal processing circuit
  • ripple (insertion loss deviation) in the passband is reduced. It is possible to realize a low loss and wide band high frequency front end circuit and communication device.
  • the transmission side switch 231, the reception side switches 251, 252 and 253, and the switch provided in the front stage or the rear stage of the filter group 220 (a plurality of high frequency filters). 271 and 272 (switch circuits).
  • the transmission amplification circuits 241, 242 and 243 corresponding to a plurality of high frequency filters, and the reception amplification circuits 261, 262, 263 and 264 (amplification circuits) can be shared. Therefore, miniaturization and cost reduction of the high frequency front end circuit can be realized.
  • the transmission side switch 231, the reception side switches 251, 252 and 253, and the switches 271 and 272 may be provided.
  • the number of transmission side switches and the number of reception side switches are not limited to the number described above.
  • one transmission side switch and one reception side switch may be provided.
  • the number of selection terminals and the like of the transmission side switch and the reception side switch is not limited to the present embodiment, and may be two.
  • the second band is applied to the Band 3 reception passband (1805-1880 MHz) of LTE
  • the third band is applied to the Band 39 passband (1880 to 1920 MHz) of LTE.
  • the band is applied to the band (1805-1920 MHz) including the Band 3 reception pass band and the Band 39 pass band
  • the present invention is not limited thereto.
  • the second band is applied to the Band 28A reception passband (758 to 788 MHz) of LTE
  • the third band is applied to the Band 20 reception passband (791 to 821 MHz) of LTE
  • the first band is the Band 28A reception passband and Band 20 reception passband.
  • the present invention may be applied to a band (758-821 MHz) including the band.
  • the second band is applied to Band 38 pass band (2570-2620 MHz) of LTE
  • the third band is applied to Band 7 reception pass band (2620-2690 MHz) of LTE
  • the first band is Band 38 pass band and Band 7 reception pass
  • the present invention may be applied to a band (2570-2690 MHz) including the band.
  • each of the series arm resonator and the parallel arm resonator constituting each filter is not limited to one resonator, and may be constituted by a plurality of divided resonators in which one resonator is divided.
  • phase shifters described in the first to sixth embodiments have a function of impedance conversion as well as phase conversion, and may be replaced with “impedance converter”.
  • control unit may be provided outside the RF signal processing circuit (RFIC) 7, and may be provided, for example, in the high frequency front end circuit 6. That is, the high frequency front end circuit 6 is not limited to the above-described configuration, and the high frequency filter according to any one of the first to fifth embodiments, and a control unit for controlling on and off of the switch element of the high frequency filter. You may have
  • an inductor or a capacitor may be connected between each component.
  • the inductor may include a wiring inductor formed by wiring connecting the components.
  • each switch element included in the high frequency filter according to any one of the first to fifth embodiments may be, for example, a field effect transistor (FET) switch or a diode switch formed of GaAs or complementary metal oxide semiconductor (CMOS). . Since such a switch is small, the high frequency filter according to any one of the first to fifth embodiments can be miniaturized.
  • FET field effect transistor
  • CMOS complementary metal oxide semiconductor
  • the series arm resonator and the parallel arm resonator included in the high frequency filter according to any one of the first to fifth embodiments are elastic wave resonators using elastic waves, and use, for example, SAW (Surface Acoustic Wave) Or a resonator using BAW (Bulk Acoustic Wave), or a film bulk acoustic resonator (FBAR).
  • SAW Surface Acoustic Wave
  • BAW Bulk Acoustic Wave
  • FBAR film bulk acoustic resonator
  • the parallel arm resonator is a resonator or a circuit represented by an equivalent circuit model (for example, a BVD model etc.) composed of an inductance component and a capacitance component, and a resonator having a resonant frequency and an antiresonant frequency or It may be a circuit.
  • an equivalent circuit model for example, a BVD model etc.
  • the present invention is widely applicable to communication devices such as mobile phones as wide-band low-loss filters, multiplexers, front end circuits and communication devices.
  • RFIC radio frequency front end circuit
  • BBIC Baseband Signal Processing Circuit
  • Switch group 220 Filter group 231 Transmission side switch 241, 242 243 transmission amplification circuit 251, 252, 253

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  • Filters And Equalizers (AREA)
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Abstract

A high frequency filter (10) having a first band as a passband is provided with a first circuit connected to a node (X1) and a node (X2), and a second circuit connected to the node (X1) and the node (X2). The first circuit includes a filter (11) which has, as a passband, a second band which is a band including some of the frequencies of the first band and having a bandwidth narrower than that of the first band. The second circuit includes: a filter (12) which is a band including some of the frequencies of the first band and having a bandwidth narrower than that of the first band, and which has, as a passband, a third band positioned on the higher frequency side of a central frequency of the second band; a phase shifter (21) connected to one terminal of the filter (12); and a phase shifter (22) connected to another terminal of the filter (12).

Description

高周波フィルタ、マルチプレクサ、高周波フロントエンド回路および通信装置High frequency filter, multiplexer, high frequency front end circuit and communication device
　本発明は、高周波フィルタ、マルチプレクサ、高周波フロントエンド回路および通信装置に関する。 The present invention relates to a high frequency filter, a multiplexer, a high frequency front end circuit and a communication device.
　広帯域の高周波フィルタとして、互いに並列接続された異なる通過帯域を有する複数の帯域通過フィルタと、並列接続された複数の帯域通過フィルタに接続されたスイッチ素子とを備えた無線受信回路が知られている（例えば、特許文献１参照）。この無線受信回路の構成によれば、スイッチ素子が導通（オン）の場合、通過帯域の周波数の異なる複数の帯域通過フィルタが互いに並列接続された回路となり、通過帯域が広帯域化されたフィルタを実現できるとしている。 As a wide band high frequency filter, there is known a wireless receiving circuit including a plurality of band pass filters having different pass bands connected in parallel to one another and a switch element connected to a plurality of band pass filters connected in parallel. (See, for example, Patent Document 1). According to the configuration of the wireless reception circuit, when the switch element is conductive (on), a plurality of band pass filters having different pass band frequencies are connected in parallel to realize a filter with a wider pass band. It is supposed to be possible.
特開２００８－１６０６２９号公報JP 2008-160629 A
　しかしながら、特許文献１に開示された構成において、スイッチ素子を導通にして複数の帯域通過フィルタからなる広帯域のフィルタが選択された場合、通過帯域内にリップル(挿入損失偏差)が入るため、通過帯域内の挿入損失が大きくなるという問題がある。 However, in the configuration disclosed in Patent Document 1, when the switch element is made conductive and a wide band filter including a plurality of band pass filters is selected, a ripple (insertion loss deviation) is included in the pass band, so the pass band There is a problem that the insertion loss inside is increased.
　そこで、本発明は、通過帯域内のリップル(挿入損失偏差)が低減された低損失かつ広帯域の高周波フィルタ、マルチプレクサ、高周波フロントエンド回路および通信装置を提供することを目的とする。 SUMMARY OF THE INVENTION An object of the present invention is to provide a low loss wide band high frequency filter, multiplexer, high frequency front end circuit and communication device in which ripple (insertion loss deviation) in the passband is reduced.
　上記目的を達成するために、本発明の一態様に係る高周波フィルタは、第１帯域を通過帯域とする高周波フィルタであって、第１入出力端子と第２入出力端子とを結ぶ経路上に設けられた第１ノード、ならびに、前記経路上の前記第１ノードおよび前記第２入出力端子の間に設けられた第２ノードに接続された第１回路と、前記第１ノードおよび前記第２ノードに接続された第２回路と、を備え、前記第１回路は、前記第１帯域の一部の周波数を含む帯域であって、前記第１帯域より狭い帯域幅を有する第２帯域を通過帯域とする第１フィルタを有し、前記第２回路は、前記第１帯域の一部の周波数を含む帯域であって、前記第１帯域より狭い帯域幅を有し、かつ、前記第２帯域の中心周波数より高周波数側に位置する第３帯域を通過帯域とする第２フィルタと、前記第２フィルタの一方の端子に接続された第１移相器と、前記第２フィルタの他方の端子に接続された第２移相器と、を有する。 In order to achieve the above object, a high frequency filter according to one aspect of the present invention is a high frequency filter having a first band as a pass band, and is provided on a path connecting a first input / output terminal and a second input / output terminal. A first node provided, and a first circuit connected to a second node provided between the first node on the path and the second input / output terminal, the first node and the second A second circuit connected to a node, wherein the first circuit is a band including a part of the frequency of the first band and passes through a second band having a narrower bandwidth than the first band And a second filter is a band including a part of the frequency of the first band and has a narrower bandwidth than the first band, and the second band. Pass band in the third band located higher than the center frequency of To a second filter, a first phase shifter which is connected to one terminal of the second filter, and a second phase shifter connected to the other terminal of the second filter, the.
　第１フィルタと第２フィルタとを第１ノードと第２ノードとの間に並列配置して、第１フィルタと第２フィルタとの合成回路により、広帯域の第１帯域を通過帯域とする高周波フィルタを実現する場合、２つのフィルタの振幅差、位相差、およびインピーダンスの差により、第１帯域内にリップルが発生することが想定される。 A high frequency filter having a first wide band as a pass band by arranging a first filter and a second filter in parallel between a first node and a second node and combining the first filter and the second filter. It is assumed that ripples occur in the first band due to the difference in amplitude, phase and impedance of the two filters.
　これに対して、上記構成によれば、第２フィルタの両端に第１移相器および第２移相器が配置されるので、２つのフィルタの振幅差に対応させて、位相差およびインピーダンスを調整することが可能となる。よって、第１帯域を通過帯域とし、当該通過帯域内のリップルが低減された低損失かつ広帯域の高周波フィルタを実現できる。 On the other hand, according to the above configuration, since the first phase shifter and the second phase shifter are disposed at both ends of the second filter, the phase difference and the impedance can be made to correspond to the amplitude difference between the two filters. It becomes possible to adjust. Therefore, it is possible to realize a low loss, wide band high frequency filter in which the first band is a pass band and ripples in the pass band are reduced.
　また、前記第１ノードおよび前記第２ノードの少なくとも一方から前記第２回路を単体で見た場合、前記第２回路のインピーダンスが極大となる特異点のうち最大のインピーダンスとなる周波数が、前記第２帯域の低域端の周波数から前記第２フィルタの通過帯域低域側のカットオフ周波数までの範囲に含まれるように、前記第１移相器および第２移相器の位相が調整されていてもよい。 Further, when the second circuit is viewed alone from at least one of the first node and the second node, the frequency at which the impedance of the second circuit is maximized among the singular points at which the impedance of the second circuit is maximized is the second frequency. The phases of the first phase shifter and the second phase shifter are adjusted so as to be included in the range from the frequency at the lower end of the two bands to the cutoff frequency on the lower side of the pass band of the second filter. May be
　これにより、第１帯域におけるリップルが低減された低損失かつ広帯域の高周波フィルタを実現できる。 As a result, it is possible to realize a low loss, wide band high frequency filter in which the ripple in the first band is reduced.
　また、前記第１ノードおよび前記第２ノードの少なくとも一方から前記第２回路を単体で見た場合、前記第２帯域において、前記第１回路と前記第２回路とが同位相になっている周波数が少なくとも１箇所あり、前記第３帯域において、前記第１回路と前記第２回路とが同位相になっている周波数が少なくとも１箇所あってもよい。 Further, when the second circuit is viewed alone from at least one of the first node and the second node, a frequency at which the first circuit and the second circuit have the same phase in the second band. There may be at least one location, and there may be at least one frequency where the first circuit and the second circuit are in phase in the third band.
　これにより、第１帯域におけるリップルが低減された低損失かつ広帯域の高周波フィルタを実現できる。 As a result, it is possible to realize a low loss, wide band high frequency filter in which the ripple in the first band is reduced.
　また、前記第３帯域の低域端の周波数において、前記第１フィルタおよび前記第２フィルタの位相差と、前記第１移相器および前記第２移相器の移相和との和が、３１２°以上かつ３６２°以下となるように、前記第１移相器および前記第２移相器の移相が調整されていてもよい。 Further, at the frequency of the low band end of the third band, the sum of the phase difference between the first filter and the second filter and the sum of phase shifts of the first phase shifter and the second phase shifter is The phase shift of the first phase shifter and the second phase shifter may be adjusted so as to be 312 ° or more and 362 ° or less.
　これにより、第１帯域におけるリップルが低減された低損失かつ広帯域の高周波フィルタを実現できる。 As a result, it is possible to realize a low loss, wide band high frequency filter in which the ripple in the first band is reduced.
　また、前記経路は、前記第１ノード、前記第１回路、および前記第２ノードを経由する第１経路と、前記第１ノード、前記第２回路、および前記第２ノードを経由する第２経路と、を含み、前記第１フィルタは、前記第１経路上に設けられた第１直列腕回路と、前記第１経路上に設けられたノードとグランドとに接続された第１並列腕回路と、を有し、前記第２フィルタは、前記第１移相器と前記第２移相器とを結ぶ前記第２経路上に設けられた第２直列腕回路と、前記第２経路上に設けられたノードとグランドとに接続された第２並列腕回路と、を有し、前記第１直列腕回路、前記第１並列腕回路、前記第２直列腕回路、および、前記第２並列腕回路のうちの少なくとも１つは、弾性波共振子を有し、前記弾性波共振子の少なくとも１つは、圧電性を有する基板上に形成されたＩＤＴ（ＩｎｔｅｒＤｉｇｉｔａｌ　Ｔｒａｎｓｄｕｃｅｒ）電極と、反射器と、を有し、前記弾性波共振子の少なくとも１つにおける前記ＩＤＴ電極の電極周期で定まる弾性波の波長をλとした場合、前記ＩＤＴ電極と前記反射器とのピッチは、０．４２λ以上かつ０．５０λ未満であってもよい。 Further, the path is a first path passing through the first node, the first circuit, and the second node, and a second path passing through the first node, the second circuit, and the second node. And the first filter includes: a first series arm circuit provided on the first path; and a first parallel arm circuit connected to a node provided on the first path and a ground. A second series arm circuit provided on the second path connecting the first phase shifter and the second phase shifter, and the second filter provided on the second path A second parallel arm circuit connected to the selected node and the ground, the first series arm circuit, the first parallel arm circuit, the second series arm circuit, and the second parallel arm circuit At least one of the plurality of elastic wave resonators, and at least one of the elastic wave resonators And an IDT (InterDigital Transducer) electrode formed on a substrate having piezoelectricity, and a reflector, and the wavelength of the elastic wave determined by the electrode period of the IDT electrode in at least one of the elastic wave resonators In the case of λ, the pitch between the IDT electrode and the reflector may be 0.42 λ or more and less than 0.50 λ.
　ＩＤＴ電極および反射器を有する弾性波共振子は、周期的に並べられた電極指からなる周期構造によって構成され、特定の周波数領域の弾性表面波を高い反射係数で反射する周波数帯域を有する。この周波数帯域は、一般的にストップバンドと称され、上記周期構造の繰り返し周期等によって規定される。このとき、ストップバンドの高域端では、反射係数が局所的に大きくなるリップルが生じる。さらに、ＩＤＴ電極と反射器とのピッチ（Ｉ－Ｒピッチ）を０．５λ以上にすると、ストップバンドの高域端でのリップルが大きくなり、フィルタの通過帯域内のリップルが大きくなり、挿入損失が大きくなる。一方、Ｉ－Ｒピッチを０．４２λ以上かつ０．５０λ未満とすることにより、ストップバンドの高域端でのリップルを低減できる。上記観点から、Ｉ－Ｒピッチを０．４２λ以上かつ０．５０λ未満とすることにより、高周波フィルタの通過帯域内の挿入損失を小さくすることができる。 An elastic wave resonator having an IDT electrode and a reflector is constituted by a periodic structure consisting of periodically arranged electrode fingers, and has a frequency band that reflects a surface acoustic wave in a specific frequency range with a high reflection coefficient. This frequency band is generally referred to as a stop band, and is defined by the repetition period of the periodic structure or the like. At this time, at the high end of the stop band, ripples in which the reflection coefficient locally increases are generated. Furthermore, when the pitch (IR pitch) between the IDT electrode and the reflector is 0.5 λ or more, the ripple at the high end of the stop band becomes large, the ripple in the pass band of the filter becomes large, and the insertion loss Becomes larger. On the other hand, by setting the IR pitch to be 0.42 λ or more and less than 0.50 λ, it is possible to reduce the ripple at the high end of the stop band. From the above point of view, by setting the IR pitch to be 0.42 λ or more and less than 0.50 λ, the insertion loss in the pass band of the high frequency filter can be reduced.
　また、前記第１回路は、さらに、前記第１ノードと前記第１フィルタとの間に接続された第１スイッチ素子と、前記第２ノードと前記第１フィルタとの間に接続された第２スイッチ素子と、を有し、前記第１スイッチ素子および前記第２スイッチ素子の導通および非導通の切り替えにより、前記第１帯域と前記第３帯域とが切り替わってもよい。 The first circuit further includes a first switch element connected between the first node and the first filter, and a second switch element connected between the second node and the first filter. A switch element may be provided, and switching between conduction and non-conduction of the first switch element and the second switch element may switch between the first band and the third band.
　これにより、第１スイッチ素子および第２スイッチ素子の導通および非導通の切り替えにより、第１帯域と第３帯域を切り替えることができる周波数可変型の高周波フィルタ（チューナブルフィルタ）を実現できる。 Thus, it is possible to realize a variable-frequency high frequency filter (tunable filter) capable of switching between the first band and the third band by switching between conduction and non-conduction of the first switch element and the second switch element.
　また、前記第１回路は、さらに、前記第１スイッチ素子と前記第１フィルタとの間の接続ノードと、グランドとの間に接続され、前記第１スイッチ素子と排他的に導通および非導通が切り替えられる第３スイッチ素子と、前記第２スイッチ素子と前記第１フィルタとの間の接続ノードと、グランドとの間に接続され、前記第２スイッチ素子と排他的に導通および非導通が切り替えられる第４スイッチ素子と、を有してもよい。 Further, the first circuit is further connected between a connection node between the first switch element and the first filter, and the ground, and conduction and non-conduction are exclusively performed with the first switch element. The third switch element to be switched, the connection node between the second switch element and the first filter, and the ground are connected, and conduction and non-conduction are switched exclusively to the second switch element. And a fourth switch element.
　これにより、第３帯域に切り替えられた場合の通過帯域（第３帯域）内のリップルを低減できるとともに、通過帯域低域側の減衰特性を向上できる。 As a result, it is possible to reduce ripples in the pass band (third band) when switching to the third band, and to improve attenuation characteristics on the low pass band side.
　また、前記第２回路は、さらに、前記第１ノードと前記第１移相器との間に接続された第５スイッチ素子と、前記第２ノードと前記第２移相器との間に接続された第６スイッチ素子と、を有し、前記第５スイッチ素子および前記第６スイッチ素子の導通および非導通の切り替えにより、前記第１帯域と前記第２帯域とが切り替わってもよい。 Further, the second circuit is further connected between a fifth switch element connected between the first node and the first phase shifter, and between the second node and the second phase shifter. The first band and the second band may be switched by switching between conduction and non-conduction of the fifth switch element and the sixth switch element.
　これにより、第５スイッチ素子および第６スイッチ素子の導通および非導通の切り替えにより、第１帯域と第２帯域を切り替えることができる周波数可変型の高周波フィルタ（チューナブルフィルタ）を実現できる。 Thus, it is possible to realize a variable-frequency high frequency filter (tunable filter) capable of switching between the first band and the second band by switching between conduction and non-conduction of the fifth switch element and the sixth switch element.
　また、前記第２回路は、さらに、前記第５スイッチ素子と前記第１移相器との間の接続ノードと、グランドとの間に接続され、前記第５スイッチ素子と排他的に導通および非導通が切り替えられる第７スイッチ素子と、前記第６スイッチ素子と前記第２移相器との間の接続ノードと、グランドとの間に接続され、前記第６スイッチ素子と排他的に導通および非導通が切り替えられる第８スイッチ素子と、を有してもよい。 The second circuit is further connected between a connection node between the fifth switch element and the first phase shifter and the ground, and is conductive and non-conductive exclusively with the fifth switch element. It is connected between a seventh switch element whose conduction is switched, a connection node between the sixth switch element and the second phase shifter, and the ground, and is conductive and non-conductive exclusively with the sixth switch element. And an eighth switch element whose conduction is switched.
　これにより、第２帯域に切り替えられた場合の通過帯域（第２帯域）内のリップルを低減できるとともに、通過帯域高域側の減衰特性を向上できる。 As a result, it is possible to reduce the ripple in the pass band (second band) when switching to the second band, and to improve the attenuation characteristics on the high band side of the pass band.
　また、さらに、前記第１ノードおよび前記第２ノードに接続された第３回路を備え、前記第３回路は、前記第１帯域と異なる第４帯域の一部の周波数を含む帯域であって、前記第４帯域より狭い帯域幅を有し、かつ、前記第２帯域の中心周波数より高周波数側に位置し、かつ、前記第３帯域と異なる第５帯域を通過帯域とする第３フィルタと、前記第３フィルタの一方の端子に接続された第３移相器と、前記第３フィルタの他方の端子に接続された第４移相器と、を有してもよい。 Furthermore, a third circuit connected to the first node and the second node is provided, and the third circuit is a band including a part of the frequency of a fourth band different from the first band, A third filter that has a narrower bandwidth than the fourth band, is located on a higher frequency side than the center frequency of the second band, and uses a fifth band different from the third band as a pass band; A third phase shifter connected to one terminal of the third filter and a fourth phase shifter connected to the other terminal of the third filter may be provided.
　これにより、通過帯域の切り替えのバリエーションが増える。 This increases the variation of switching of the passband.
　また、前記第１移相器は、前記第１ノードと前記第２フィルタの一方の端子とを結ぶ直列腕経路に配置された１以上のインダクタ、および、前記直列腕経路上のノードとグランドとの間に接続されたキャパシタを有する低域通過型フィルタ回路、または、前記直列腕経路に配置された１以上のキャパシタ、および、前記直列腕経路上のノードおよびグランドに接続されたインダクタで構成された高域通過型フィルタ回路であってもよい。 Further, the first phase shifter includes one or more inductors arranged in a series arm path connecting the first node and one terminal of the second filter, a node on the series arm path, and a ground A low pass filter circuit having a capacitor connected between the one or more capacitors arranged in the series arm path, and an inductor connected to a node on the series arm path and a ground It may be a high pass filter circuit.
　これにより、要求される減衰特性に応じて、第１移相器の回路構成を低域通過型フィルタ回路および高域通過型フィルタ回路のいずれかに適宜選択することが可能となる。 Thus, the circuit configuration of the first phase shifter can be appropriately selected as either the low pass filter circuit or the high pass filter circuit according to the required attenuation characteristic.
　また、前記第２移相器は、前記第２ノードと前記第２フィルタの他方の端子とを結ぶ直列腕経路に配置された１以上のインダクタ、および、前記直列腕経路上のノードおよびグランドに接続されたキャパシタで構成された低域通過型フィルタ回路、または、前記直列腕経路に配置された１以上のキャパシタ、および、前記直列腕経路上のノードおよびグランドに接続されたインダクタで構成された高域通過型フィルタ回路であってもよい。 In addition, the second phase shifter includes at least one inductor disposed in a series arm path connecting the second node and the other terminal of the second filter, and a node and a ground on the series arm path. A low pass filter circuit composed of connected capacitors, or one or more capacitors arranged in the series arm path, and an inductor connected to a node on the series arm path and the ground It may be a high pass filter circuit.
　これにより、要求される減衰特性に応じて、第２移相器の回路構成を低域通過型フィルタ回路および高域通過型フィルタ回路のいずれかに適宜選択することが可能となる。 Thus, the circuit configuration of the second phase shifter can be appropriately selected as either the low pass filter circuit or the high pass filter circuit according to the required attenuation characteristic.
　また、前記第１移相器および前記第２移相器は、キャパシタまたはインダクタからなるインピーダンス素子であってもよい。 The first phase shifter and the second phase shifter may be impedance elements formed of a capacitor or an inductor.
　これにより、高周波フィルタの回路構成を簡素化でき、小型化が可能となる。 As a result, the circuit configuration of the high frequency filter can be simplified and miniaturization can be achieved.
　また、前記第１フィルタおよび第２フィルタの少なくとも一方は、前記第１ノードと前記第２ノードとを結ぶ前記経路上に配置された直列腕回路と、前記経路上のノードおよびグランドに接続された並列腕回路と、を有し、前記直列腕回路および前記並列腕回路の少なくとも一方は、共振子、および、第９スイッチ素子、を有し、前記第９スイッチ素子の導通および非導通の切り替えにより、前記直列腕回路および前記並列腕回路の少なくとも一方の共振周波数および反共振周波数の少なくとも一方の周波数が切り替わってもよい。 Further, at least one of the first filter and the second filter is connected to a series arm circuit disposed on the path connecting the first node and the second node, and to a node on the path and a ground. A parallel arm circuit, at least one of the series arm circuit and the parallel arm circuit includes a resonator and a ninth switch element, and switching between conduction and non-conduction of the ninth switch element The resonant frequency and / or the antiresonant frequency of at least one of the series arm circuit and the parallel arm circuit may be switched.
　これにより、通過帯域の切り替えのバリエーションが増える。 This increases the variation of switching of the passband.
　また、前記第１フィルタおよび前記第２フィルタの少なくとも一方は、弾性表面波フィルタ、弾性境界波フィルタ、および、バルク弾性波（ＢＡＷ：Ｂｕｌｋ　Ａｃｏｕｓｔｉｃ　Ｗａｖｅ）を用いた弾性波フィルタのいずれかであってもよい。 Further, at least one of the first filter and the second filter is any of a surface acoustic wave filter, a boundary acoustic wave filter, and an elastic wave filter using a bulk acoustic wave (BAW). It is also good.
　これにより、選択度の高い高周波フィルタを実現できる。 Thereby, a high frequency filter with high selectivity can be realized.
　また、前記第１フィルタおよび前記第２フィルタの少なくとも一方は、縦結合型共振器を有してもよい。 Also, at least one of the first filter and the second filter may have a longitudinally coupled resonator.
　これにより、通過帯域低域側において、急峻性が向上するとともに、減衰量を向上できる。 As a result, on the low pass band side, the sharpness can be improved and the attenuation can be improved.
　また、第１移相器の位相と第２移相器の位相とは異なってもよい。 Also, the phase of the first phase shifter may be different from the phase of the second phase shifter.
　また、本発明の一態様に係るマルチプレクサは、上記いずれかに記載の高周波フィルタを含む複数のフィルタを備え、前記複数のフィルタの入力端子または出力端子は、共通端子に直接的または間接的に接続されている。 A multiplexer according to one aspect of the present invention includes a plurality of filters including the high frequency filter according to any one of the above, and the input terminals or the output terminals of the plurality of filters are connected directly or indirectly to a common terminal. It is done.
　これにより、通過帯域内のリップルが低減された低損失かつ広帯域の高周波フィルタ有するマルチプレクサを実現できる。 This makes it possible to realize a multiplexer having a low loss and wide band high frequency filter with reduced ripple in the pass band.
　また、本発明の一態様に係る高周波フロントエンド回路は、上記いずれかに記載の高周波フィルタまたは上記記載のマルチプレクサと、前記高周波フィルタまたは前記マルチプレクサに直接的または間接的に接続された増幅回路と、を備える。 A high frequency front end circuit according to one aspect of the present invention includes the high frequency filter according to any one of the above or the multiplexer according to the above, and an amplifier circuit directly or indirectly connected to the high frequency filter or the multiplexer. Equipped with
　これにより、通過帯域内のリップルが低減された低損失かつ広帯域の高周波フィルタ有する高周波フロントエンド回路を実現できる。 As a result, it is possible to realize a high frequency front end circuit having a low loss, wide band high frequency filter with reduced ripple in the pass band.
　また、本発明の一態様に係る通信装置は、アンテナ素子で送受信される高周波信号を処理するＲＦ信号処理回路と、前記アンテナ素子と前記ＲＦ信号処理回路との間で前記高周波信号を伝達する上記記載の高周波フロントエンド回路と、を備える。 Further, a communication apparatus according to an aspect of the present invention includes: an RF signal processing circuit processing high frequency signals transmitted and received by an antenna element; and transmitting the high frequency signal between the antenna element and the RF signal processing circuit. And the described high frequency front end circuit.
　これにより、通過帯域内のリップルが低減された低損失かつ広帯域の高周波フィルタ有する通信装置を実現できる。 As a result, it is possible to realize a communication apparatus having a low loss, wide band high frequency filter with reduced ripple in the pass band.
　本発明によれば、通過帯域内のリップル(挿入損失偏差)が低減された低損失かつ広帯域の高周波フィルタ、マルチプレクサ、高周波フロントエンド回路および通信装置を提供することが可能となる。 According to the present invention, it is possible to provide a low loss and wide band high frequency filter, multiplexer, high frequency front end circuit and communication device in which ripple (insertion loss deviation) in the pass band is reduced.
図１Ａは、実施の形態１に係る高周波フィルタの回路ブロック図である。FIG. 1A is a circuit block diagram of a high frequency filter according to a first embodiment. 図１Ｂは、実施の形態１に係る高周波フィルタの広帯域化の概念を表す図である。FIG. 1B is a diagram showing the concept of increasing the bandwidth of the high frequency filter according to the first embodiment. 図２は、実施例１に係る高周波フィルタの回路構成図である。FIG. 2 is a circuit diagram of the high frequency filter according to the first embodiment. 図３Ａは、比較例１に係る高周波フィルタの回路ブロック図である。FIG. 3A is a circuit block diagram of a high frequency filter according to Comparative Example 1. 図３Ｂは、比較例２に係る高周波フィルタの回路ブロック図である。FIG. 3B is a circuit block diagram of a high frequency filter according to Comparative Example 2. 図４Ａは、比較例１に係る高周波フィルタの通過特性、振幅特性、位相特性およびインピーダンス特性を表すグラフである。FIG. 4A is a graph showing the pass characteristic, the amplitude characteristic, the phase characteristic, and the impedance characteristic of the high frequency filter according to Comparative Example 1. 図４Ｂは、比較例２に係る高周波フィルタの通過特性、振幅特性、位相特性およびインピーダンス特性を表すグラフである。FIG. 4B is a graph showing the pass characteristic, the amplitude characteristic, the phase characteristic, and the impedance characteristic of the high frequency filter according to Comparative Example 2. 図５は、実施例１に係る高周波フィルタの通過特性、振幅特性、位相特性およびインピーダンス特性を表すグラフである。FIG. 5 is a graph showing pass characteristics, amplitude characteristics, phase characteristics, and impedance characteristics of the high frequency filter according to the first embodiment. 図６は、実施例１に係る高周波フィルタの広帯域化の要因を説明するグラフである。FIG. 6 is a graph for explaining the factor of broadening the band of the high frequency filter according to the first embodiment. 図７は、実施例１に係る高周波フィルタの移相器の位相を変化させた場合の通過特性、位相特性およびインピーダンス特性を表すグラフである。FIG. 7 is a graph showing pass characteristics, phase characteristics, and impedance characteristics when the phase of the phase shifter of the high frequency filter according to the first embodiment is changed. 図８は、実施例１に係る高周波フィルタの移相器の位相と挿入損失および最大インピーダンスとの関係を表すグラフである。FIG. 8 is a graph showing the relationship between the phase, the insertion loss, and the maximum impedance of the phase shifter of the high frequency filter according to the first embodiment. 図９は、実施例１ａ、１ｂ、１ｃ、１ｄ、１ｅに係る高周波フィルタの回路構成図である。FIG. 9 is a circuit diagram of a high frequency filter according to Examples 1a, 1b, 1c, 1d and 1e. 図１０は、実施の形態１における並列腕共振子の構造を模式的に表す図の一例である。FIG. 10 is an example of a diagram schematically showing the structure of the parallel arm resonator in the first embodiment. 図１１Ａは、実施例１ａおよび実施例１ｅに係る高周波フィルタの通過特性および反射特性を比較したグラフである。FIG. 11A is a graph comparing the pass characteristic and the reflection characteristic of the high frequency filters according to Example 1a and Example 1e. 図１１Ｂは、実施例１ａおよび実施例１ｅに係る直列腕共振子のインピーダンス特性および反射特性を比較したグラフである。FIG. 11B is a graph comparing impedance characteristics and reflection characteristics of series arm resonators according to Example 1a and Example 1e. 図１２Ａは、実施例１ｂおよび実施例１ｅに係る高周波フィルタの通過特性を比較したグラフである。FIG. 12A is a graph comparing the pass characteristics of the high frequency filters according to Example 1 b and Example 1 e. 図１２Ｂは、実施例１ｂおよび実施例１ｅに係る並列腕共振子のインピーダンス特性および反射特性を比較したグラフである。FIG. 12B is a graph comparing impedance characteristics and reflection characteristics of parallel arm resonators according to Example 1 b and Example 1 e. 図１３Ａは、実施例１ｃおよび実施例１ｅに係る高周波フィルタの通過特性を比較したグラフである。FIG. 13A is a graph comparing the pass characteristics of the high frequency filters according to Example 1c and Example 1e. 図１３Ｂは、実施例１ｃおよび実施例１ｅに係る並列腕共振子のインピーダンス特性および反射特性を比較したグラフである。FIG. 13B is a graph comparing impedance characteristics and reflection characteristics of parallel arm resonators according to Example 1 c and Example 1 e. 図１４は、実施例１ｄおよび実施例１ｅに係る高周波フィルタの通過特性および反射特性を比較したグラフである。FIG. 14 is a graph comparing the pass characteristic and the reflection characteristic of the high frequency filters according to Example 1 d and Example 1 e. 図１５Ａは、典型例の共振子において、Ｉ－Ｒピッチを０．４０λ～０．５０λで変化させた場合の特性の変化を表すグラフである。FIG. 15A is a graph showing a change in characteristics when the IR pitch is changed at 0.40 λ to 0.50 λ in the resonator of the typical example. 図１５Ｂは、典型例の共振子において、Ｉ－Ｒピッチを０．５０λ～０．６０λで変化させた場合の特性の変化を表すグラフである。FIG. 15B is a graph showing a change in characteristics when the IR pitch is changed at 0.50 λ to 0.60 λ in the resonator of the typical example. 図１６は、実施の形態２に係る高周波フィルタの回路ブロック図である。FIG. 16 is a circuit block diagram of the high frequency filter according to the second embodiment. 図１７Ａは、実施例２に係る高周波フィルタの回路構成図である。FIG. 17A is a circuit configuration diagram of a high frequency filter according to a second embodiment. 図１７Ｂは、実施例２に係る高周波フィルタのスイッチの切り替えによる通過特性の変化を表すグラフである。FIG. 17B is a graph showing a change in pass characteristics due to the switching of the switch of the high frequency filter according to the second embodiment. 図１８は、実施例２に係る高周波フィルタの広帯域化の要因を説明するグラフである。FIG. 18 is a graph for explaining the factor of increasing the bandwidth of the high frequency filter according to the second embodiment. 図１９は、実施例２に係る高周波フィルタのスイッチオフ容量を変化させた場合の通過特性を表すグラフである。FIG. 19 is a graph showing the pass characteristic when the switch-off capacity of the high frequency filter according to the second embodiment is changed. 図２０は、実施例３に係る高周波フィルタの回路ブロック図である。FIG. 20 is a circuit block diagram of the high frequency filter according to the third embodiment. 図２１は、実施例３に係る高周波フィルタの通過特性を表すグラフである。FIG. 21 is a graph showing the pass characteristic of the high frequency filter according to the third embodiment. 図２２Ａは、実施の形態２の変形例１に係る高周波フィルタの回路構成図である。FIG. 22A is a circuit configuration diagram of a high frequency filter according to Modification 1 of Embodiment 2. 図２２Ｂは、実施の形態２の変形例２に係る高周波フィルタの回路構成図である。FIG. 22B is a circuit configuration diagram of a high frequency filter according to a second modification of the second embodiment. 図２３は、実施の形態２の変形例１および２に係る高周波フィルタの通過特性を比較したグラフである。FIG. 23 is a graph comparing the pass characteristics of the high frequency filters according to the first and second modifications of the second embodiment. 図２４Ａは、実施例４に係る高周波フィルタの回路構成図である。FIG. 24A is a circuit configuration diagram of a high frequency filter according to a fourth embodiment. 図２４Ｂは、実施例４に係る高周波フィルタのスイッチの切り替えによる通過特性の変化を表すグラフである。FIG. 24B is a graph showing the change of the passage characteristic due to the switch of the high frequency filter according to the fourth embodiment. 図２５は、実施例４に係る高周波フィルタの第１帯域における広帯域化の要因を説明するグラフである。FIG. 25 is a graph for explaining the factor of increasing the band in the first band of the high frequency filter according to the fourth embodiment. 図２６は、実施例４に係る高周波フィルタの第１ａ帯域における広帯域化の要因を説明するグラフである。FIG. 26 is a graph for explaining the factor of increasing the band in the 1a band of the high frequency filter according to the fourth embodiment. 図２７Ａは、実施例５に係る高周波フィルタの回路構成図である。FIG. 27A is a circuit configuration diagram of a high frequency filter according to a fifth embodiment. 図２７Ｂは、実施例５に係る高周波フィルタのスイッチの切り替えによる通過特性の変化を表すグラフである。FIG. 27B is a graph showing a change in pass characteristics due to switching of a switch of the high frequency filter according to the fifth embodiment. 図２８は、実施例５に係る高周波フィルタの広帯域化の要因を説明するグラフである。FIG. 28 is a graph for explaining the factor of increasing the bandwidth of the high frequency filter according to the fifth embodiment. 図２９Ａは、実施例６に係る高周波フィルタの回路構成図である。FIG. 29A is a circuit configuration diagram of a high frequency filter according to a sixth embodiment. 図２９Ｂは、実施例６に係る高周波フィルタのスイッチの切り替えによる通過特性の変化を表すグラフである。FIG. 29B is a graph showing a change in passage characteristics due to the switch of the high frequency filter according to the sixth embodiment. 図３０は、実施例６に係る高周波フィルタの第１帯域における広帯域化の要因を説明するグラフである。FIG. 30 is a graph for explaining the factor of increasing the band in the first band of the high frequency filter according to the sixth embodiment. 図３１は、実施例６に係る高周波フィルタの第１ｂ帯域における広帯域化の要因を説明するグラフである。FIG. 31 is a graph for explaining the factor of increasing the band in the 1b band of the high frequency filter according to the sixth embodiment. 図３２Ａは、実施例６に係る高周波フィルタを構成する各フィルタの変形例１の回路構成図である。FIG. 32A is a circuit configuration diagram of Modification Example 1 of each filter configuring the high-frequency filter according to Example 6. 図３２Ｂは、実施例６に係る高周波フィルタを構成する各フィルタの変形例２の回路構成図である。32B is a circuit configuration diagram of Modification Example 2 of each of the filters forming the high-frequency filter according to Example 6. FIG. 図３２Ｃは、実施例６に係る高周波フィルタを構成する各フィルタの変形例３の回路構成図である。32C is a circuit configuration diagram of Modification Example 3 of each of the filters forming the high-frequency filter according to Example 6. FIG. 図３２Ｄは、実施例６に係る高周波フィルタを構成する各フィルタの変形例４の回路構成図である。FIG. 32D is a circuit configuration diagram of Modification Example 4 of each filter configuring the high-frequency filter according to Example 6. 図３３Ａは、実施例６に係る高周波フィルタを構成する各フィルタの変形例５の回路構成図である。FIG. 33A is a circuit configuration diagram of Modified Example 5 of each filter configuring the high frequency filter according to Example 6. 図３３Ｂは、実施例６に係る高周波フィルタを構成する各フィルタの変形例６の回路構成図である。FIG. 33B is a circuit configuration diagram of Modified Example 6 of each filter configuring the high frequency filter according to Example 6. 図３３Ｃは、実施例６に係る高周波フィルタを構成する各フィルタの変形例７の回路構成図である。FIG. 33C is a circuit configuration diagram of Modified Example 7 of each filter configuring the high frequency filter according to Example 6. 図３３Ｄは、実施例６に係る高周波フィルタを構成する各フィルタの変形例８の回路構成図である。FIG. 33D is a circuit configuration diagram of Modified Example 8 of the filters configuring the high frequency filter according to Example 6. 図３３Ｅは、実施例６に係る高周波フィルタを構成する各フィルタの変形例９の回路構成図である。FIG. 33E is a circuit configuration diagram of Modification 9 of the filters configuring the high-frequency filter according to the sixth embodiment. 図３３Ｆは、実施例６に係る高周波フィルタを構成する各フィルタの変形例１０の回路構成図である。FIG. 33F is a circuit configuration diagram of Modified Example 10 of the filters configuring the high-frequency filter according to Example 6. 図３３Ｇは、実施例６に係る高周波フィルタを構成する各フィルタの変形例１１の回路構成図である。FIG. 33G is a circuit configuration diagram of Modification 11 of the filters configuring the high-frequency filter according to Embodiment 6. 図３４は、実施の形態６に係る通信装置およびその周辺回路の構成図である。FIG. 34 is a block diagram of the communication device and its peripheral circuits according to the sixth embodiment.
　以下、本発明の実施の形態について、実施例および図面を用いて詳細に説明する。なお、以下で説明する実施の形態は、いずれも包括的または具体的な例を示すものである。以下の実施の形態で示される数値、形状、材料、構成要素、構成要素の配置および接続形態などは、一例であり、本発明を限定する主旨ではない。以下の実施の形態における構成要素のうち、独立請求項に記載されていない構成要素については、任意の構成要素として説明される。また、図面に示される構成要素の大きさ、または大きさの比は、必ずしも厳密ではない。また、各図において、実質的に同一の構成に対しては同一の符号を付しており、重複する説明は省略または簡略化する場合がある。また、共振子等の回路素子については要求仕様等に応じて定数が適宜調整され得る。このため、回路素子については、同一の符号であっても定数が異なる場合もある。 Hereinafter, embodiments of the present invention will be described in detail using examples and drawings. The embodiments described below are all inclusive or specific examples. Numerical values, shapes, materials, components, arrangements of components, connection configurations and the like shown in the following embodiments are merely examples, and are not intended to limit the present invention. Among the components in the following embodiments, components not described in the independent claims are described as optional components. Also, the sizes or ratio of sizes of components shown in the drawings are not necessarily exact. Further, in the drawings, substantially the same configuration is given the same reference numeral, and overlapping description may be omitted or simplified. In addition, constants of circuit elements such as resonators can be appropriately adjusted according to required specifications and the like. Therefore, the circuit elements may have different constants even if they have the same code.
　また、共振子または回路における共振周波数とは、特に断りの無い限り、当該共振子または当該回路を含むフィルタの通過帯域または通過帯域近傍の減衰極を形成するための共振周波数であり、当該共振子または当該回路のインピーダンスが極小となる特異点（理想的にはインピーダンスが０となる点）である「共振点」の周波数である。 Further, unless otherwise specified, the resonant frequency in the resonator or circuit is a resonant frequency for forming an attenuation pole in the pass band or near the pass band of the resonator or the filter including the circuit, and the resonator Alternatively, it is the frequency of the “resonance point” which is a singular point at which the impedance of the circuit becomes minimum (ideally, the point at which the impedance is 0).
　また、共振子または回路における反共振周波数とは、特に断りの無い限り、当該共振子または当該回路を含むフィルタの通過帯域または通過帯域近傍の減衰極を形成するための反共振周波数であり、当該共振子または当該回路のインピーダンスが極大となる特異点（理想的にはインピーダンスが無限大となる点）である「反共振点」の周波数である。 Further, the antiresonance frequency in the resonator or the circuit is an antiresonance frequency for forming an attenuation pole in the pass band or near the pass band of the resonator or the filter including the circuit, unless otherwise specified. It is the frequency of the "anti-resonance point" which is a singular point at which the impedance of the resonator or the circuit concerned is maximal (ideally, the point at which the impedance is infinite).
　なお、以下の実施の形態において、直列腕（共振）回路および並列腕（共振）回路は、以下のように定義される。 In the following embodiments, a series arm (resonance) circuit and a parallel arm (resonance) circuit are defined as follows.
　並列腕（共振）回路は、第１入出力端子および第２入出力端子を結ぶ経路上の一のノードと、グランドと、の間に配置された回路である。 The parallel arm (resonance) circuit is a circuit disposed between one node on a path connecting the first input / output terminal and the second input / output terminal and the ground.
　直列腕（共振）回路は、第１入出力端子または第２入出力端子と、並列腕（共振）回路が接続される上記経路上のノードと、の間に配置された回路、または、一の並列腕（共振）回路が接続される上記経路上の一のノードと、他の並列腕（共振）回路が接続される上記経路上の他のノードと、の間に配置された回路である。 The series arm (resonance) circuit is a circuit disposed between the first input / output terminal or the second input / output terminal and a node on the above path to which the parallel arm (resonance) circuit is connected, or The circuit is disposed between one node on the path to which the parallel arm (resonance) circuit is connected and the other node on the path to which the other parallel arm (resonance) circuit is connected.
　また、以下において、「通過帯域低域端」は、「通過帯域内の最も低い周波数」を意味する。また、「通過帯域高域端」は、「通過帯域内の最も高い周波数」を意味する。また、以下において、「通過帯域低域側」は、「通過帯域外かつ通過帯域より低周波数側」を意味する。また「通過帯域高域側」は、「通過帯域外かつ通過帯域より高周波数側」を意味する。また、以下では、「低周波数側」を「低域側」と称し、「高周波数側」を「高域側」と称する場合がある。 Also, in the following, “passband lower end” means “the lowest frequency in the passband”. Also, “pass band high end” means “the highest frequency in the pass band”. Also, in the following, “low pass band side” means “outside of pass band and lower frequency side than pass band”. Also, "high pass side of pass band" means "outside of pass band and higher frequency side than pass band". Also, in the following, the “low frequency side” may be referred to as the “low frequency side”, and the “high frequency side” may be referred to as the “high frequency side”.
　（実施の形態１）
　［１．１　実施の形態１に係る高周波フィルタ１０の基本構成］
　図１Ａは、実施の形態１に係る高周波フィルタ１０の回路ブロック図である。同図に示された高周波フィルタ１０は、フィルタ１１および１２と、移相器２１および２２と、入出力端子１１０および１２０と、を備える。 Embodiment 1
[1.1 Basic Configuration of High-Frequency Filter 10 According to Embodiment 1]
FIG. 1A is a circuit block diagram of the high frequency filter 10 according to the first embodiment. The high frequency filter 10 shown in the figure includes filters 11 and 12, phase shifters 21 and 22, and input / output terminals 110 and 120.
　フィルタ１１は、第１回路を構成する第１フィルタである。第１回路は、入出力端子１１０（第１入出力端子）と入出力端子１２０（第２入出力端子）とを結ぶ経路上に設けられたノードＸ１（第１ノード）、ならびに、ノードＸ１および入出力端子１２０の間に設けられたノードＸ２（第２ノード）に接続されている。 The filter 11 is a first filter that constitutes a first circuit. The first circuit includes a node X1 (first node) provided on a path connecting the input / output terminal 110 (first input / output terminal) and the input / output terminal 120 (second input / output terminal), and a node X1 and It is connected to a node X2 (second node) provided between the input and output terminals 120.
　フィルタ１２は、移相器２１および２２とともに第２回路を構成する第２フィルタである。第２回路は、ノードＸ１およびＸ２に接続されている。 The filter 12 is a second filter that constitutes a second circuit together with the phase shifters 21 and 22. The second circuit is connected to nodes X1 and X2.
　移相器２１は、フィルタ１２の一方の端子に接続された第１移相器である。また、移相器２２は、フィルタ１２の他方の端子に接続された第２移相器である。つまり、移相器２１、フィルタ１２、および移相器２２は、この順で、ノードＸ１とノードＸ２との間に直列接続されている。 The phase shifter 21 is a first phase shifter connected to one terminal of the filter 12. The phase shifter 22 is a second phase shifter connected to the other terminal of the filter 12. That is, the phase shifter 21, the filter 12, and the phase shifter 22 are connected in series between the node X1 and the node X2 in this order.
　図１Ｂは、実施の形態１に係る高周波フィルタ１０の広帯域化の概念を表す図である。同図には、フィルタ１１および１２、ならびに、高周波フィルタ１０の通過帯域が概念的に示されている。高周波フィルタ１０は、広帯域の第１帯域を通過帯域としている。フィルタ１１は、第１帯域の一部の周波数を含む帯域であって、第１帯域より狭い帯域幅を有する第２帯域を通過帯域としている。フィルタ１２は、第１帯域の一部の周波数を含む帯域であって、第１帯域より狭い帯域幅を有し、かつ、第２帯域の中心周波数より高域側に位置する第３帯域を通過帯域としている。 FIG. 1B is a diagram showing the concept of broadening the band of the high-frequency filter 10 according to the first embodiment. In the figure, the pass bands of the filters 11 and 12 and the high frequency filter 10 are conceptually shown. The high frequency filter 10 has a first wide band as a pass band. The filter 11 is a band including a part of the frequency of the first band, and a second band having a narrower bandwidth than the first band is a pass band. The filter 12 is a band including a portion of the frequency of the first band, has a narrower bandwidth than the first band, and passes through a third band located higher than the center frequency of the second band. It is a band.
　なお、第１帯域とは、複数の離散的な帯域で構成された帯域ではなく、周波数的に連続した１つの帯域であると定義される。 The first band is not defined as a band constituted by a plurality of discrete bands, but is defined as one band continuous in frequency.
　［１．２　実施例１に係る高周波フィルタ１０Ａの回路構成］
　図２は、実施例１に係る高周波フィルタ１０Ａの回路構成図である。同図に示された高周波フィルタ１０Ａは、フィルタ１１Ａおよび１２Ａと、移相器２１および２２と、入出力端子１１０および１２０と、を備える。高周波フィルタ１０Ａは、実施の形態１に係る高周波フィルタ１０の具体的回路構成例であり、フィルタ１１Ａはフィルタ１１の具体的回路構成例であり、フィルタ１２Ａはフィルタ１２の具体的回路構成例である。 [1.2 Circuit configuration of the high frequency filter 10A according to the first embodiment]
FIG. 2 is a circuit configuration diagram of the high frequency filter 10A according to the first embodiment. The high frequency filter 10A shown in the figure includes filters 11A and 12A, phase shifters 21 and 22, and input / output terminals 110 and 120. High frequency filter 10A is a specific circuit configuration example of high frequency filter 10 according to the first embodiment, filter 11A is a specific circuit configuration example of filter 11, and filter 12A is a specific circuit configuration example of filter 12 .
　フィルタ１１Ａは、ノードＸ１とノードＸ２とを結ぶ第１経路上に配置された直列腕共振子ｓ１１、ｓ１２およびｓ１３と、当該第１経路上の各ノードとグランドとの間に配置された並列腕共振子ｐ１１、ｐ１２、ｐ１３およびｐ１４と、並列腕共振子ｐ１３およびグランドに接続されたインダクタＬｐ１３と、を備える。これにより、フィルタ１１Ａは、ラダー型の帯域通過型フィルタを構成している。 The filter 11A includes series arm resonators s11, s12 and s13 disposed on a first path connecting the node X1 and the node X2, and a parallel arm disposed between each node on the first path and the ground. A resonator p11, p12, p13 and p14, and a parallel arm resonator p13 and an inductor Lp13 connected to the ground are provided. Thus, the filter 11A constitutes a ladder type band pass filter.
　フィルタ１２Ａは、ノードＸ１とノードＸ２とを結ぶ第２経路上に配置された直列腕共振子ｓ２１およびｓ２２と、当該第２経路上の各ノードとグランドとの間に配置された並列腕共振子ｐ２１、ｐ２２およびｐ２３と、を備える。これにより、フィルタ１２Ａは、ラダー型の帯域通過型フィルタを構成している。 The filter 12A includes series arm resonators s21 and s22 arranged on a second path connecting the node X1 and the node X2, and parallel arm resonators arranged between each node on the second path and the ground. and p21, p22 and p23. Thus, the filter 12A constitutes a ladder type band pass filter.
　表１に、実施例１に係る高周波フィルタ１０Ａの回路パラメータを示す。 Table 1 shows circuit parameters of the high frequency filter 10A according to the first embodiment.
Figure JPOXMLDOC01-appb-T000001
Figure JPOXMLDOC01-appb-T000001
　実施の形態１に係る高周波フィルタ１０および実施例１に係る高周波フィルタ１０Ａは、第２フィルタの一方端に移相器２１が配置され、第２フィルタの他方端に移相器２２が配置される構成を特徴としている。以下、本構成の特徴および効果を説明するために、まず、比較例に係る高周波フィルタの特徴および問題点を説明する。 In the high frequency filter 10 according to the first embodiment and the high frequency filter 10A according to the first embodiment, the phase shifter 21 is disposed at one end of the second filter, and the phase shifter 22 is disposed at the other end of the second filter. It features an arrangement. Hereinafter, in order to explain the features and effects of the present configuration, first, the features and problems of the high frequency filter according to the comparative example will be described.
　［１．３　比較例に係る高周波フィルタの回路構成およびフィルタ特性］
　図３Ａは、比較例１に係る高周波フィルタ５００の回路ブロック図である。また、図３Ｂは、比較例２に係る高周波フィルタ６００の回路ブロック図である。 [1.3 Circuit configuration and filter characteristics of high frequency filter according to comparative example]
FIG. 3A is a circuit block diagram of the high frequency filter 500 according to the first comparative example. FIG. 3B is a circuit block diagram of the high frequency filter 600 according to the second comparative example.
　図３Ａに示された比較例１に係る高周波フィルタ５００は、フィルタ１１および１２と、入出力端子１１０および１２０と、を備える。図３Ａに示された高周波フィルタ５００は、実施の形態に示された高周波フィルタ１０と比較して、移相器２１および２２が配置されていない点のみが構成として異なる。なお、高周波フィルタ５００を構成するフィルタ１１および１２は、高周波フィルタ１０を構成するフィルタ１１および１２と同じ構成であり、高周波フィルタ５００のフィルタ１１は第２帯域を通過帯域とし、高周波フィルタ５００のフィルタ１２は第３帯域を通過帯域としている。 The high frequency filter 500 according to Comparative Example 1 shown in FIG. 3A includes the filters 11 and 12 and the input / output terminals 110 and 120. The high frequency filter 500 shown in FIG. 3A differs from the high frequency filter 10 shown in the embodiment only in that the phase shifters 21 and 22 are not arranged. The filters 11 and 12 constituting the high frequency filter 500 have the same configuration as the filters 11 and 12 constituting the high frequency filter 10, and the filter 11 of the high frequency filter 500 has a second band as a pass band. Reference numeral 12 designates the third band as the pass band.
　図３Ｂに示された比較例２に係る高周波フィルタ６００は、フィルタ１１および１２と、移相器２１と、入出力端子１１０および１２０と、を備える。図３Ｂに示された比較例２に係る高周波フィルタ６００は、実施の形態に示された高周波フィルタ１０と比較して、移相器２２が配置されていない点のみが構成として異なる。なお、高周波フィルタ６００を構成するフィルタ１１および１２は、高周波フィルタ１０を構成するフィルタ１１および１２と同じ構成であり、高周波フィルタ５００のフィルタ１１は第２帯域を通過帯域とし、高周波フィルタ５００のフィルタ１２は第３帯域を通過帯域としている。 The high frequency filter 600 according to the comparative example 2 shown in FIG. 3B includes the filters 11 and 12, the phase shifter 21, and the input and output terminals 110 and 120. The high frequency filter 600 according to Comparative Example 2 shown in FIG. 3B is different from the high frequency filter 10 shown in the embodiment only in that the phase shifter 22 is not disposed. The filters 11 and 12 constituting the high frequency filter 600 have the same configuration as the filters 11 and 12 constituting the high frequency filter 10, and the filter 11 of the high frequency filter 500 has a second band as a pass band. Reference numeral 12 designates the third band as the pass band.
　以下、実施例１、比較例１および比較例２に係る高周波フィルタでは、第２帯域をＬＴＥ（Ｌｏｎｇ　Ｔｅｒｍ　Ｅｖｏｌｕｔｉｏｎ）のＢａｎｄ３受信通過帯域（１８０５－１８８０ＭＨｚ）とし、第３帯域をＬＴＥのＢａｎｄ３９通過帯域（１８８０－１９２０ＭＨｚ）とし、第１帯域を（Ｂａｎｄ３受信通過帯域＋Ｂａｎｄ３９通過帯域）（１８０５－１９２０ＭＨｚ）としている。 Hereinafter, in the high frequency filters according to Example 1 and Comparative Examples 1 and 2, the second band is a Band 3 reception pass band (1805-1880 MHz) of LTE (Long Term Evolution), and the third band is a Band 39 pass band of LTE The first band is (Band 3 reception pass band + Band 39 pass band) (1805-1920 MHz).
　図４Ａは、比較例１に係る高周波フィルタ５００の通過特性、振幅特性、位相特性およびインピーダンス特性を表すグラフである。図４Ａの（ａ）および（ｆ）には、高周波フィルタ５００の通過特性が示されている。図４Ａの（ｂ）および（ｃ）には、それぞれ、ノードＸ１－Ｘ２間の第１回路単体とノードＸ１－Ｘ２間の第２回路単体との振幅差および位相差が示されている。図４Ａの（ｄ）には、ノードＸ１から第１回路単体および第２回路単体を見たインピーダンス特性が示されている。図４Ａの（ｅ）には、ノードＸ２から第１回路単体および第２回路単体を見たインピーダンス特性が示されている。図４Ａの（ｇ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の通過特性が示されている。図４Ａの（ｈ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の位相特性が示されている。 FIG. 4A is a graph showing the pass characteristic, the amplitude characteristic, the phase characteristic, and the impedance characteristic of the high frequency filter 500 according to the first comparative example. The pass characteristics of the high frequency filter 500 are shown in (a) and (f) of FIG. 4A. In (b) and (c) of FIG. 4A, an amplitude difference and a phase difference between the first circuit alone between the nodes X1 and X2 and the second circuit alone between the nodes X1 and X2 are shown. In (d) of FIG. 4A, impedance characteristics are shown when the first circuit alone and the second circuit alone are viewed from the node X1. FIG. 4A (e) shows impedance characteristics when the first circuit single unit and the second circuit single unit are viewed from the node X2. (G) of FIG. 4A shows the pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2. In (h) of FIG. 4A, phase characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2 are shown.
　なお、本実施の形態および以降の実施の形態において、回路の「インピーダンス」とは、反射のインピーダンスを指し、回路の「位相」とは、通過の位相を表すものとしている。 In the present embodiment and the following embodiments, the “impedance” of the circuit refers to the impedance of reflection, and the “phase” of the circuit represents the phase of passage.
　まず、図４Ａの（ａ）に示すように、比較例１に係る高周波フィルタ５００は、通過帯域である第１帯域内で大きなリップル（挿入損失偏差）が見られ、第１帯域内挿入損失を悪化させている。上記リップルの主原因としては、以下の２つが挙げられる。 First, as shown in (a) of FIG. 4A, in the high frequency filter 500 according to Comparative Example 1, a large ripple (insertion loss deviation) is observed in the first band which is the pass band, and the first in-band insertion loss It is getting worse. The following two are mentioned as the main cause of the said ripple.
　（１）フィルタ１２Ａの通過帯域低域側の減衰極が現れる周波数（図４Ａの（１））において、図４Ａの（ｄ）および（ｅ）に示すように、フィルタ１１Ａのインピーダンスに対してフィルタ１２Ａ（第２回路）のインピーダンスが低い。また、上記減衰極が現れる周波数において、図４Ａの（ｂ）に示すように、第１回路と第２回路との振幅差が大きい。これらのため、上記減衰極が現れる周波数の高周波信号は、第２回路のフィルタ１２Ａから（並列腕共振子を経由して）グランドへ流れこむこととなる。これにより、高周波フィルタ５００の第１帯域に含まれる、上記減衰極が現れる周波数では、第１帯域内においてリップルを発生させることとなる。 (1) At the frequency ((1) in FIG. 4A) at which the attenuation pole on the lower side of the pass band of the filter 12A appears, as shown in (d) and (e) of FIG. The impedance of 12A (second circuit) is low. Further, at the frequency at which the attenuation pole appears, as shown in (b) of FIG. 4A, the difference in amplitude between the first circuit and the second circuit is large. For these reasons, a high frequency signal of a frequency at which the attenuation pole appears is caused to flow from the filter 12A of the second circuit (via the parallel arm resonator) to the ground. As a result, at the frequency at which the attenuation pole appears and is included in the first band of the high frequency filter 500, ripples are generated in the first band.
　（２）第１帯域に含まれる２つの周波数（図４Ａの（２））において、図４Ａの（ｃ）に示すように、フィルタ１１Ａ（第１回路）とフィルタ１２Ａ（第２回路）との位相差が－１８０°となる。また、図４Ａの（ｂ）に示すように、上記２つの周波数において、フィルタ１１Ａ（第１回路）とフィルタ１２Ａ（第２回路）との振幅差が略０ｄＢとなる。つまり、第１帯域に含まれる上記２つの周波数において、フィルタ１１Ａ（第１回路）とフィルタ１２Ａ（第２回路）とが逆位相かつ同振幅となる。このため、第１帯域に含まれる上記２つの周波数の高周波信号は、第１回路および第２回路を経由した後、打ち消し合うこととなる。これにより、高周波フィルタ５００の第１帯域に含まれる、上記２つの周波数では、第１帯域内においてリップルを発生させることとなる。 (2) At two frequencies ((2) in FIG. 4A) included in the first band, as shown in (c) in FIG. 4A, between the filter 11A (first circuit) and the filter 12A (second circuit) The phase difference is -180 °. Further, as shown in (b) of FIG. 4A, the amplitude difference between the filter 11A (first circuit) and the filter 12A (second circuit) is approximately 0 dB at the two frequencies. That is, at the two frequencies included in the first band, the filter 11A (first circuit) and the filter 12A (second circuit) have opposite phases and the same amplitude. Therefore, the high frequency signals of the two frequencies included in the first band cancel each other after passing through the first circuit and the second circuit. Thus, at the two frequencies included in the first band of the high frequency filter 500, ripples are generated in the first band.
　図４Ｂは、比較例２に係る高周波フィルタ６００の通過特性、振幅特性、位相特性およびインピーダンス特性を表すグラフである。図４Ｂの（ａ）および（ｆ）には、高周波フィルタ６００の通過特性が示されている。図４Ｂの（ｂ）および（ｃ）には、それぞれ、ノードＸ１－Ｘ２間の第１回路単体とノードＸ１－Ｘ２間の第２回路単体との振幅差および位相差が示されている。図４Ｂの（ｄ）には、ノードＸ１から第１回路単体および第２回路単体を見たインピーダンス特性が示されている。図４Ｂの（ｅ）には、ノードＸ２から第１回路単体および第２回路単体を見たインピーダンス特性が示されている。図４Ｂの（ｇ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の通過特性が示されている。図４Ｂの（ｈ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の位相特性が示されている。 FIG. 4B is a graph showing the pass characteristic, the amplitude characteristic, the phase characteristic, and the impedance characteristic of the high frequency filter 600 according to the second comparative example. The pass characteristics of the high frequency filter 600 are shown in (a) and (f) of FIG. 4B. In (b) and (c) of FIG. 4B, an amplitude difference and a phase difference between the first circuit alone between the nodes X1 and X2 and the second circuit alone between the nodes X1 and X2 are shown. In (d) of FIG. 4B, the impedance characteristics are shown when the first circuit alone and the second circuit alone are viewed from the node X1. FIG. 4B (e) shows impedance characteristics when the first circuit alone and the second circuit alone are viewed from the node X2. FIG. 4B (g) shows the pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2. In (h) of FIG. 4B, phase characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2 are shown.
　図４Ｂの（ａ）に示すように、比較例２に係る高周波フィルタ６００は、通過帯域である第１帯域内で大きなリップル（挿入損失偏差）が見られ、第１帯域内挿入損失を悪化させている。比較例２に係る高周波フィルタ６００では、移相器２１を配置したため、比較例１に係る高周波フィルタ５００に見られたような上記（２）に起因したリップルは低減されているが、下記（１）に起因したリップルが発生している。 As shown in (a) of FIG. 4B, in the high frequency filter 600 according to the comparative example 2, a large ripple (insertion loss deviation) is observed in the first band which is the pass band, and the first in-band insertion loss is deteriorated. ing. In the high frequency filter 600 according to the comparative example 2, since the phase shifter 21 is disposed, the ripple due to the above (2) as seen in the high frequency filter 500 according to the comparative example 1 is reduced. There is a ripple due to).
　（１）フィルタ１２Ａの通過帯域低域側の減衰極が現れる周波数（図４Ｂの（１））において、図４Ｂの（ｅ）に示すように、フィルタ１２Ａ（第２回路）のノードＸ２から見たインピーダンス｜Ｚ２２｜が低い。また、上記減衰極が現れる周波数において、図４Ｂの（ｂ）に示すように、第１回路と第２回路との振幅差が大きい。これらのため、上記減衰極が現れる周波数の高周波信号は、第２回路のフィルタ１２Ａから（並列腕共振子を経由して）グランドへ流れこむこととなる。これにより、高周波フィルタ６００の第１帯域に含まれる、上記減衰極が現れる周波数では、第１帯域内においてリップルを発生させることとなる。 (1) At a frequency ((1) in FIG. 4B) at which the attenuation pole on the lower side of the pass band of the filter 12A appears, as shown in (e) of FIG. 4B, viewed from the node X2 of the filter 12A (second circuit). Impedance | Z22 | is low. Further, at the frequency at which the attenuation pole appears, as shown in (b) of FIG. 4B, the difference in amplitude between the first circuit and the second circuit is large. For these reasons, a high frequency signal of a frequency at which the attenuation pole appears is caused to flow from the filter 12A of the second circuit (via the parallel arm resonator) to the ground. As a result, at the frequency at which the attenuation pole appears and included in the first band of the high frequency filter 600, ripples are generated in the first band.
　［１．４　実施例１に係る高周波フィルタ１０Ａのフィルタ特性］
　図５は、実施例１に係る高周波フィルタ１０Ａの通過特性、振幅特性、位相特性およびインピーダンス特性を表すグラフである。図５の（ａ）および（ｆ）には、高周波フィルタ１０Ａの通過特性が示されている。図５の（ｂ）および（ｃ）には、それぞれ、ノードＸ１－Ｘ２間の第１回路単体とノードＸ１－Ｘ２間の第２回路単体との振幅差および位相差が示されている。図５の（ｄ）には、ノードＸ１から第１回路単体および第２回路単体を見たインピーダンス特性が示されている。図５の（ｅ）には、ノードＸ２から第１回路単体および第２回路単体を見たインピーダンス特性が示されている。図５の（ｇ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の通過特性が示されている。図５の（ｈ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の位相特性が示されている。 [1.4 Filter Characteristics of High-Frequency Filter 10A According to First Embodiment]
FIG. 5 is a graph showing the pass characteristic, the amplitude characteristic, the phase characteristic, and the impedance characteristic of the high frequency filter 10A according to the first embodiment. The pass characteristics of the high frequency filter 10A are shown in (a) and (f) of FIG. FIGS. 5B and 5C respectively show an amplitude difference and a phase difference between the first circuit alone between the nodes X1 and X2 and the second circuit alone between the nodes X1 and X2. FIG. 5D shows impedance characteristics when the first circuit unit and the second circuit unit are viewed from the node X1. FIG. 5E shows impedance characteristics when the first circuit unit and the second circuit unit are viewed from the node X2. FIG. 5 (g) shows the pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2. FIG. 5H shows phase characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2.
　実施例１に係る高周波フィルタ１０Ａのように、フィルタ１２Ａの両方の入出力端子（前段および後段）に移相器２１および２２が接続されることで、図５の（ａ）および（ｆ）に示すように、高周波フィルタ１０Ａの通過帯域である第１帯域において、リップルが低減されることで挿入損失が低減する。 As in the high frequency filter 10A according to the first embodiment, the phase shifters 21 and 22 are connected to both input / output terminals (previous stage and subsequent stage) of the filter 12A, as shown in (a) and (f) of FIG. As shown, in the first band which is the pass band of the high frequency filter 10A, the ripple is reduced to reduce the insertion loss.
　実施例１に係る高周波フィルタ１０Ａが、比較例１に係る高周波フィルタ５００および比較例２に係る高周波フィルタ６００と比較して、通過帯域内リップルおよび通過帯域内挿入損失を低減する要因について、以下に説明する。 The reasons why the high frequency filter 10A according to the first embodiment reduces the in-passband ripple and the in-passband insertion loss as compared to the high frequency filter 500 according to the first comparative example and the high frequency filter 600 according to the second comparative example are described below. explain.
　図６は、実施例１に係る高周波フィルタ１０Ａの広帯域化の要因を説明するグラフである。同図には、上記要因を説明するため、図５の（ｃ）、（ｄ）、（ｅ）、（ｇ）、（ｈ）が示されている。 FIG. 6 is a graph for explaining the factor of increasing the bandwidth of the high frequency filter 10A according to the first embodiment. In FIG. 5, (c), (d), (e), (g) and (h) are shown in order to explain the above factors.
　実施例１に係る高周波フィルタ１０Ａでは、第２回路単体をノードＸ１から見たインピーダンス｜Ｚ１１｜が極大となる特異点のうち最大のインピーダンスとなる周波数ｆ１１が、第２帯域の低域端の周波数ｆＡから、フィルタ１２Ａの通過帯域低域側のカットオフ周波数ｆＣまでの周波数帯に含まれるように、移相器２１および２２の位相が調整されている。また、第２回路単体をノードＸ２から見たインピーダンス｜Ｚ２２｜が極大となる特異点のうち最大のインピーダンスとなる周波数ｆ２２が、第２帯域の低域端の周波数ｆＡから、フィルタ１２Ａの通過帯域低域側のカットオフ周波数ｆＣまでの周波数帯に含まれるように、移相器２１および２２の位相が調整されている。 In the high frequency filter 10A according to the first embodiment, the impedance viewed second circuit itself from the node X1 | Z11 | frequency f 11 which is the maximum of the impedance of the singular point as a maximum is the low frequency end of the second band The phases of the phase shifters 21 and 22 are adjusted so as to be included in the frequency band from the frequency f A to the cut-off frequency f C on the low pass band side of the filter 12A. The impedance seen second circuit itself from the node X2 | Z22 | frequency f 22 which is the maximum of the impedance of the singular point as a maximum is the frequency f A of the low-frequency end of the second band, the filter 12A The phases of the phase shifters 21 and 22 are adjusted so as to be included in the frequency band up to the cutoff frequency f C on the lower side of the pass band.
　なお、フィルタの通過帯域低域側のカットオフ周波数とは、当該フィルタの通過帯域低域端の挿入損失から所定の挿入損失だけ増加した通過帯域低域側の周波数のうち、最も通過帯域低域端に近い周波数である。本実施の形態および以降の実施の形態では、通過帯域低域側のカットオフ周波数を、通過帯域低域端の挿入損失から１０ｄＢ増加した通過帯域低域側の周波数と定義している。 The cut-off frequency on the low pass side of the filter is the lowest pass band among the frequencies on the low side of the low pass band which is increased by a predetermined insertion loss from the insertion loss at the low end of the pass band of the filter. It is a frequency close to the end. In the present embodiment and the following embodiments, the cut-off frequency on the lower side of the passband is defined as the frequency on the lower side of the passband which is increased by 10 dB from the insertion loss at the lower end of the passband.
　フィルタ１２Ａの減衰極の周波数ｆＡでは位相変化が大きく、フィルタ１１Ａとの位相差が大きくなるとともに、振幅差も大きくなる。そのため、移相器２１および２２の位相調整によって、第２帯域の低域端の周波数ｆＡからフィルタ１２Ａの通過帯域低域側のカットオフ周波数ｆＣまでの周波数範囲に、第２回路のインピーダンス極大の特異点の周波数を合わせることにより、上記周波数範囲の高周波信号が、第２回路のフィルタ１２Ａに流れこむことを抑制できる。つまり、第２回路が減衰している周波数帯域で第２回路への信号の回り込みを抑制できる。これにより、比較例１および２で見られた、上記（１）の要因を解消でき、高周波フィルタ１０Ａの第１帯域内のリップルを抑制することが可能となる。 At the frequency f A of the attenuation pole of the filter 12A, the phase change is large, the phase difference with the filter 11A is large, and the amplitude difference is also large. Therefore, by the phase adjustment of the phase shifters 21 and 22, the impedance of the second circuit is in the frequency range from the frequency f A at the low band end of the second band to the cutoff frequency f C at the low band side of the filter 12A. By matching the frequency of the maximum singular point, it is possible to suppress the high frequency signal in the above frequency range from flowing into the filter 12A of the second circuit. That is, it is possible to suppress the wraparound of the signal to the second circuit in the frequency band in which the second circuit is attenuated. Thereby, the factor of the above (1) seen in Comparative Examples 1 and 2 can be eliminated, and it is possible to suppress the ripple in the first band of the high frequency filter 10A.
　なお、フィルタ１１Ａの通過帯域（第２帯域）高域側の減衰極の周波数でも、同様に、フィルタ１１Ａと第２回路との振幅差および位相差が大きくなるが、当該減衰極はフィルタ１１Ａを構成する直列腕回路の反共振周波数で構成される。このため、フィルタ１１Ａのインピーダンスは極大になっているため、移相器が無くても高周波フィルタ１０Ａの通過帯域（第１帯域）内のリップルとはならない。 Similarly, the amplitude difference and the phase difference between the filter 11A and the second circuit become large also at the frequency of the attenuation pole on the high band side of the pass band (second band) of the filter 11A, but the attenuation pole concerned is the filter 11A. It is comprised by the antiresonance frequency of the series arm circuit to comprise. For this reason, since the impedance of the filter 11A is maximum, even if there is no phase shifter, it does not become a ripple in the pass band (first band) of the high frequency filter 10A.
　なお、本実施の形態および以降の実施の形態において、直列腕共振子、並列腕共振子、直列腕回路および並列腕回路のインピーダンス｜Ｚ｜が極小となる周波数が共振周波数であり、インピーダンス｜Ｚ｜が極大となる周波数が反共振周波数である。 In the present embodiment and the following embodiments, the frequency at which the impedance | Z | of the series arm resonator, the parallel arm resonator, the series arm circuit and the parallel arm circuit is minimized is the resonance frequency, and the impedance | Z The frequency at which | is the maximum is the antiresonance frequency.
　さらに、比較例１で見られた、上記（２）の要因については、第１回路と第２回路との振幅差が小さい周波数帯で同位相となるように移相器２１および２２を位相調整することにより、第１帯域に含まれる周波数の高周波信号は、第１回路および第２回路を経由した後で、互いに打ち消し合うことがない。これにより、高周波フィルタ１０Ａの第１帯域に含まれる周波数において、第１帯域内のリップルを低減できる。 Furthermore, with regard to the factor (2) described in Comparative Example 1, the phase shifters 21 and 22 are adjusted such that the phase difference between the first circuit and the second circuit is in the same phase in a small frequency band. By doing this, the high frequency signals of the frequencies included in the first band do not cancel each other after passing through the first circuit and the second circuit. Thereby, the ripple in the 1st zone can be reduced in the frequency contained in the 1st zone of high frequency filter 10A.
　本実施例では、図６の（ｃ）に示すように、第２回路のノードＸ１およびＸ２から第２回路を単体で見た場合、第２帯域において、第１回路と第２回路とが同位相（０°または－３６０°）になっている周波数が少なくとも１箇所（同図の破線丸印：３箇所）あり、第３帯域において、第１回路と第２回路とが同位相（０°または－３６０°）になっている周波数が少なくとも１箇所（同図の破線丸印：２箇所）あってもよい。 In the present embodiment, as shown in FIG. 6C, when the second circuit is viewed alone from the nodes X1 and X2 of the second circuit, the first circuit and the second circuit are the same in the second band. There are at least one frequency (0 circle or -360 °) at which the phase (0 ° or -360 °) is present (3 dotted circles), and in the third band, the first and second circuits are in phase (0 °) Alternatively, there may be at least one frequency (two dashed circle marks in the same figure) which is set to -360 °.
　図７は、実施例１に係る高周波フィルタ１０Ａの移相器２１および２２の位相を変化させた場合の通過特性、位相特性およびインピーダンス特性を表すグラフである。また、図８は、実施例１に係る高周波フィルタ１０Ａの移相器２１および２２の位相と挿入損失および最大インピーダンスの周波数との関係を表すグラフである。なお、上記最大インピーダンスの周波数とは、インピーダンスが極大となる特異点のうち、最大のインピーダンスとなる周波数のことである。より具体的には、図７は、移相器２１および２２の位相を、０°～１５０°(１８０°～３３０°と同義)まで変化させた場合の通過特性、位相特性およびインピーダンス特性の変化を示している。また、図８は、移相器２１および２２の位相を、０°～１７５°(１８０°～３５５°と同義)において５°ステップで変化させた場合の挿入損失および最大インピーダンスの変化を示している。 FIG. 7 is a graph showing pass characteristics, phase characteristics, and impedance characteristics when the phases of the phase shifters 21 and 22 of the high frequency filter 10A according to the first embodiment are changed. FIG. 8 is a graph showing the relationship between the phase of the phase shifters 21 and 22 of the high frequency filter 10A according to the first embodiment and the frequency of the insertion loss and the maximum impedance. Note that the frequency of the maximum impedance is a frequency at which the impedance is maximized among the singular points at which the impedance is maximized. More specifically, FIG. 7 shows changes in pass characteristics, phase characteristics and impedance characteristics when the phase of phase shifters 21 and 22 is changed from 0 ° to 150 ° (synonymous with 180 ° to 330 °). Is shown. Also, FIG. 8 shows the change in the insertion loss and the maximum impedance when the phase of the phase shifters 21 and 22 is changed in steps of 5 ° at 0 ° to 175 ° (equivalent to 180 ° to 355 °). There is.
　まず、当然の結果であるが、移相器２１および２２の位相を変化させることで、全周波数で第２回路の位相が変化する。そのため、図７の（ａ）および（ｂ）に示すように、通過帯域である第１帯域の振幅差が小さい周波数帯で、第１回路と第２回路の位相が逆相（１８０°の位相差）になるとリップルが大きくなり、同相（０°または３６０°の位相差）になるとリップルが小さくなる。よって、通過帯域である第１帯域の振幅差が小さい周波数帯で、第１回路と第２回路との位相差を無くすことで、高周波フィルタ１０Ａの第１帯域内のリップルが低減し、通過帯域内挿入損失を低減できる。 First, as a matter of course, changing the phase of the phase shifters 21 and 22 changes the phase of the second circuit at all frequencies. Therefore, as shown in (a) and (b) of FIG. 7, the phase difference between the first circuit and the second circuit is opposite (a phase of 180 When the phase difference is reached, the ripples increase, and when the phase difference (0 ° or 360 ° phase difference) is reached, the ripples decrease. Therefore, by eliminating the phase difference between the first circuit and the second circuit in the frequency band where the amplitude difference in the first band, which is the pass band, is small, the ripple in the first band of the high frequency filter 10A is reduced, and the pass band Internal insertion loss can be reduced.
　実施例１に係る高周波フィルタ１０Ａにおいて、フィルタ１２Ａの通過帯域（第３帯域）低域端の周波数において、フィルタ１１Ａおよび１２Ａの位相差（Ａ）は、図４Ａの（ｈ）より、１５２．２°である。一方、第１回路および第２回路の位相差（Ｂ）は、図５の（ｈ）より、３４９．２°である。また、表１より、移相器２１および２２の位相は、それぞれ、９８．９°および９８．１°であるので、移相器２１および２２の位相の和（Ｃ）は、１９７°である。 In the high-frequency filter 10A according to the first embodiment, the phase difference (A) between the filters 11A and 12A is 152.2 as shown in (h) of FIG. ° On the other hand, the phase difference (B) between the first circuit and the second circuit is 349.2 ° according to (h) in FIG. Further, according to Table 1, since the phases of the phase shifters 21 and 22 are 98.9 ° and 98.1 °, respectively, the sum (C) of the phases of the phase shifters 21 and 22 is 197 °. .
　ここで、（Ａ）＋（Ｃ）＝（Ｂ）の関係が成立している。 Here, the relationship of (A) + (C) = (B) is established.
　これに対して、図８の（ａ）より、高周波フィルタ１０Ａのリップルが３ｄＢ以下となるのは、移相器２１および２２のそれぞれの位相が、８０～１０５°である場合となっている。この移相器２１および２２の位相範囲を、上記（Ｃ）に代入することで、第１回路および第２回路の位相差（Ｂ）の範囲が規定される。すなわち、第１回路および第２回路の位相差（Ｂ）が、３１２．２°（１５２．２°＋８０°×２）～３６２．２°（１５２．２°＋１０５°×２°）となるように、移相器２１および２２が調整されている。 On the other hand, according to FIG. 8A, the ripple of the high frequency filter 10A is 3 dB or less when the phase of each of the phase shifters 21 and 22 is 80 to 105 °. By substituting the phase range of the phase shifters 21 and 22 into (C), the range of the phase difference (B) between the first circuit and the second circuit is defined. That is, the phase difference (B) of the first circuit and the second circuit is 312.2 ° (152.2 ° + 80 ° × 2) to 362.2 ° (152.2 ° + 105 ° × 2 °). The phase shifters 21 and 22 are adjusted.
　一方、移相器２１および２２の位相を変化させることで、第２回路のノードＸ１のインピーダンス|Ｚ１１|、および、第２回路のノードＸ２側のインピーダンス|Ｚ２２|の最大になる周波数が変化する。そのため、第２回路のインピーダンスが極大となる特異点のうち最大のインピーダンスとなる周波数が、第２帯域の低域端の周波数からフィルタ１２Ａの通過帯域低域側のカットオフ周波数までの周波数帯から外れると、第２回路のインピーダンスが低くなり、第２回路の低域側減衰帯域で第２回路側に信号が回り込み、高周波フィルタの第１帯域におけるリップルとなり、通過帯域内挿入損失を悪化させる。一方、上記低域側減衰帯域にインピーダンス最大になる周波数があると、第２回路のインピーダンスが高くなり、第２回路の低域側減衰帯域で第２回路側に信号が回り込まないため、高周波フィルタの通過帯域内リップルは抑制されて通過帯域内挿入損失が低減する、もしくは、通過帯域におけるリップルは抑制されて挿入損失が低減する。 On the other hand, changing the phase of phase shifters 21 and 22 changes the frequency at which impedance | Z11 | of node X1 of the second circuit and impedance | Z22 | of node X2 of the second circuit become maximum. . Therefore, from the frequency band from the frequency at the low band end of the second band to the cutoff frequency at the low band side of the pass band on the low band side of the second band from the frequency band at the maximum impedance among the singular points where the impedance of the second circuit is maximum When out of the range, the impedance of the second circuit becomes low, and the signal wraps around to the second circuit side in the lower attenuation band of the second circuit, resulting in ripples in the first band of the high frequency filter, thereby deteriorating the insertion loss in the passband. On the other hand, if there is a frequency at which the impedance is maximized in the lower attenuation band, the impedance of the second circuit is increased, and the signal does not go to the second circuit in the lower attenuation band of the second circuit. The ripple in the passband is suppressed to reduce the insertion loss in the passband, or the ripple in the passband is suppressed to reduce the insertion loss.
　以上により、第２回路の少なくとも一方の入出力端子から第２回路を単体で見た場合、第２回路のインピーダンスが極大となる特異点のうち最大のインピーダンスとなる周波数が、第２帯域の低域端の周波数からフィルタ１２Ａの通過帯域低域側のカットオフ周波数までの周波数帯に含まれるように、移相器２１および２２の位相が調整されていることが望ましい。特に、本実施例では、フィルタ１２Ａの通過帯域低域側の移相器２１および２２の位相を調整することで、高周波フィルタ１０Ａのリップルを３ｄＢ以内にすることが可能となる。 From the above, when the second circuit is viewed alone from at least one input / output terminal of the second circuit, the frequency at which the impedance at the maximum of the singular point at which the impedance of the second circuit is maximal is the second band is low. It is desirable that the phases of the phase shifters 21 and 22 be adjusted so as to be included in the frequency band from the frequency of the band edge to the cut-off frequency on the low band side of the pass band of the filter 12A. In particular, in the present embodiment, by adjusting the phase of the phase shifters 21 and 22 on the low pass band side of the filter 12A, the ripple of the high frequency filter 10A can be made within 3 dB.
　［１．５　実施例１ａ－１ｅに係る高周波フィルタ１０Ｂの構成］
　図９は、実施例１ａ、１ｂ、１ｃ、１ｄ、１ｅに係る高周波フィルタ１０Ｂの回路構成図である。同図に示された高周波フィルタ１０Ｂは、フィルタ１１Ｄおよび１２Ｄと、移相器２１および２２と、入出力端子１１０および１２０と、を備える。高周波フィルタ１０Ｂは、実施の形態１に係る高周波フィルタ１０の具体的回路構成例であり、フィルタ１１Ｄはフィルタ１１の具体的回路構成例であり、フィルタ１２Ｄはフィルタ１２の具体的回路構成例である。 [1.5 Configuration of High-Frequency Filter 10B According to Embodiments 1a-1e]
FIG. 9 is a circuit diagram of the high frequency filter 10B according to the embodiments 1a, 1b, 1c, 1d and 1e. The high frequency filter 10B shown in the figure includes filters 11D and 12D, phase shifters 21 and 22, and input / output terminals 110 and 120. The high frequency filter 10B is a specific circuit configuration example of the high frequency filter 10 according to the first embodiment, the filter 11D is a specific circuit configuration example of the filter 11, and the filter 12D is a specific circuit configuration example of the filter 12 .
　フィルタ１１Ｄは、ノードＸ１とノードＸ２とを結ぶ第１経路上に配置された直列腕共振子ｓ１１およびｓ１２と、直列腕共振子ｓ１１と直列腕共振子ｓ１２とを結ぶ経路上のノードとグランドとの間に配置された並列腕共振子ｐ１１と、を備える。これにより、フィルタ１１Ｄは、ラダー型の帯域通過型フィルタを構成している。 The filter 11D includes series arm resonators s11 and s12 arranged on a first path connecting the node X1 and the node X2, and a node and a ground on a path connecting the series arm resonator s11 and the series arm resonator s12. And a parallel arm resonator p11 disposed between the two. Thus, the filter 11D constitutes a ladder type band pass filter.
　フィルタ１２Ｄは、ノードＸ１とノードＸ２とを結ぶ第２経路上に配置された直列腕共振子ｓ２１と、当該第２経路上の各ノードとグランドとの間に配置された並列腕共振子ｐ２１およびｐ２２と、を備える。これにより、フィルタ１２Ｄは、ラダー型の帯域通過型フィルタを構成している。 The filter 12D includes a series arm resonator s21 disposed on a second path connecting the node X1 and the node X2, a parallel arm resonator p21 disposed between each node on the second path and the ground, and and p22. Thus, the filter 12D constitutes a ladder type band pass filter.
　以下、高周波フィルタ１０Ｂを構成する各共振子の構造について、並列腕共振子ｐ１１に着目してより詳細に説明する。なお、他の共振子については、Ｉ－Ｒピッチが弾性波の波長λの０．５倍程度で構成されている点等を除き、並列腕共振子ｐ１１と概ね同じ構造を有するため、詳細な説明を省略する。 Hereinafter, the structure of each resonator constituting the high frequency filter 10B will be described in more detail focusing on the parallel arm resonator p11. The other resonators have almost the same structure as the parallel arm resonator p11 except that the IR pitch is about 0.5 times the wavelength λ of the elastic wave, and so on. I omit explanation.
　図１０は、実施の形態１における並列腕共振子ｐ１１の構造を模式的に表す図の一例であり、（ａ）は平面図、（ｂ）は（ａ）の断面図である。なお、図１０に示された並列腕共振子ｐ１１は、高周波フィルタ１０Ｂを構成する各共振子の典型的な構造を説明するためのものである。このため、高周波フィルタ１０Ｂの各共振子のＩＤＴ（ＩｎｔｅｒＤｉｇｉｔａｌ　Ｔｒａｎｓｄｕｃｅｒ）電極を構成する電極指の本数や長さなどは、同図に示すＩＤＴ電極の電極指の本数や長さに限定されない。 FIG. 10 is an example of a diagram schematically illustrating the structure of the parallel arm resonator p11 according to the first embodiment, in which (a) is a plan view and (b) is a cross-sectional view of (a). The parallel arm resonator p11 shown in FIG. 10 is for describing a typical structure of each of the resonators constituting the high frequency filter 10B. For this reason, the number, length, etc. of the electrode fingers constituting the IDT (InterDigital Transducer) electrode of each resonator of the high frequency filter 10B are not limited to the number or the length of the electrode fingers of the IDT electrode shown in FIG.
　同図の（ａ）及び（ｂ）に示すように、並列腕共振子ｐ１１は、ＩＤＴ電極３２１及び反射器３２２を構成する電極膜３０１と、当該電極膜３０１が形成された圧電基板３０２と、当該電極膜３０１を覆う保護層３０３と、を備える。以下、これらの構成要素について、詳細に説明する。 As shown in (a) and (b) of the figure, the parallel arm resonator p11 includes an electrode film 301 constituting an IDT electrode 321 and a reflector 322, and a piezoelectric substrate 302 on which the electrode film 301 is formed. And a protective layer 303 covering the electrode film 301. Hereinafter, these components will be described in detail.
　図１０の（ａ）に示すように、圧電基板３０２の上には、ＩＤＴ電極３２１を構成する互いに対向する一対の櫛歯電極３０１ａ及び３０１ｂが形成されている。櫛歯電極３０１ａは、互いに平行な複数の電極指３１０ａと、複数の電極指３１０ａを接続するバスバー電極３１１ａとで構成されている。また、櫛歯電極３０１ｂは、互いに平行な複数の電極指３１０ｂと、複数の電極指３１０ｂを接続するバスバー電極３１１ｂとで構成されている。複数の電極指３１０ａ及び３１０ｂは、弾性波の伝搬方向と直交する方向に沿って形成され、当該伝搬方向に沿って周期的に形成されている。 As shown in (a) of FIG. 10, on the piezoelectric substrate 302, a pair of mutually opposing comb-shaped electrodes 301a and 301b which constitute the IDT electrode 321 are formed. The comb-tooth electrode 301a is composed of a plurality of parallel electrode fingers 310a and a bus bar electrode 311a connecting the plurality of electrode fingers 310a. Further, the comb-tooth electrode 301 b is configured of a plurality of parallel electrode fingers 310 b and a bus bar electrode 311 b connecting the plurality of electrode fingers 310 b. The plurality of electrode fingers 310a and 310b are formed along a direction orthogonal to the propagation direction of the elastic wave, and are periodically formed along the propagation direction.
　このように構成されたＩＤＴ電極３２１は、当該ＩＤＴ電極３２１を構成する複数の電極指３１０ａ及び３１０ｂの電極ピッチ等によって規定される特定の周波数領域の弾性表面波を励振する。 The thus configured IDT electrode 321 excites a surface acoustic wave in a specific frequency range defined by the electrode pitch and the like of the plurality of electrode fingers 310 a and 310 b constituting the IDT electrode 321.
　なお、櫛歯電極３０１ａ及び３０１ｂは、それぞれが単体でＩＤＴ電極と称される場合もある。ただし、以下では、便宜上、一対の櫛歯電極３０１ａ及び３０１ｂによって１つのＩＤＴ電極３２１が構成されているものとして説明する。 Each of the comb- tooth electrodes 301a and 301b may be referred to as an IDT electrode alone. However, in the following, for convenience, the description will be made assuming that one IDT electrode 321 is configured by the pair of comb electrodes 301a and 301b.
　反射器３２２は、ＩＤＴ電極３２１に対して弾性波の伝搬方向に配置されている。具体的には、一組の反射器３２２は、ＩＤＴ電極３２１を弾性波の伝搬方向両側から挟みこむように配置されている。反射器３２２は、互いに平行な複数の電極指４１０と、複数の電極指４１０の一方の端部を接続するバスバー電極４１１及び複数の電極指４１０の他方の端部を接続するバスバー電極４１１からなる一組のバスバー電極４１１とで構成されている。複数の電極指４１０は、ＩＤＴ電極３２１を構成する複数の電極指３１０ａ及び３１０ｂと同様に、弾性波の伝搬方向と直交する方向に沿って形成され、当該伝搬方向に沿って周期的に形成されている。 The reflector 322 is disposed in the propagation direction of the elastic wave with respect to the IDT electrode 321. Specifically, the pair of reflectors 322 is disposed so as to sandwich the IDT electrode 321 from both sides in the propagation direction of the elastic wave. The reflector 322 includes a plurality of parallel electrode fingers 410, a bus bar electrode 411 connecting one end of the plurality of electrode fingers 410, and a bus bar electrode 411 connecting the other ends of the plurality of electrode fingers 410. A pair of bus bar electrodes 411 is formed. Similar to the plurality of electrode fingers 310 a and 310 b constituting the IDT electrode 321, the plurality of electrode fingers 410 are formed along the direction orthogonal to the propagation direction of the elastic wave, and are periodically formed along the propagation direction. ing.
　このように構成された反射器３２２は、当該反射器３２２を構成する複数の電極指４１０の電極ピッチ等によって規定される周波数帯域（ストップバンド）において、弾性表面波を高い反射係数で反射する。つまり、ＩＤＴ電極３２１の電極ピッチと反射器３２２の電極ピッチとが等しい場合、反射器３２２は、ＩＤＴ電極３２１によって励振された弾性表面波を高い反射係数で反射する。 The reflector 322 configured in this way reflects the surface acoustic wave with a high reflection coefficient in a frequency band (stop band) defined by the electrode pitch and the like of the plurality of electrode fingers 410 constituting the reflector 322. That is, when the electrode pitch of the IDT electrode 321 and the electrode pitch of the reflector 322 are equal, the reflector 322 reflects the surface acoustic wave excited by the IDT electrode 321 with a high reflection coefficient.
　このような反射器３２２を有することにより、並列腕共振子ｐ１１は、励振した弾性表面波を内部に閉じ込めて外部に漏らしにくくすることができる。よって、並列腕共振子ｐ１１は、ＩＤＴ電極３２１の電極ピッチ、対数及び交叉幅等で規定される共振点および反共振点のＱを向上させることができる。 By having such a reflector 322, the parallel arm resonator p11 can confine the excited surface acoustic wave inside and make it difficult to leak it to the outside. Therefore, the parallel arm resonator p11 can improve Q of the resonance point and the antiresonance point defined by the electrode pitch, the logarithm, the crossing width, and the like of the IDT electrode 321.
　なお、反射器３２２は電極指４１０を有していればよく、バスバー電極４１１を有していなくてもかまわない。また、電極指４１０の本数は１以上であればよく、特に限定されない。ただし、電極指４１０の本数が少なすぎると弾性波の漏れが多くなるため、フィルタ特性が劣化し得る。一方、電極指４１０の本数が多すぎると反射器３２２が大型化になるため、高周波フィルタ１０Ｂ全体が大型化し得る。このため、電極指４１０の本数は、高周波フィルタ１０Ｂに要求されるフィルタ特性及びサイズ等を考慮して、適宜決定され得る。 The reflector 322 only needs to have the electrode finger 410, and may not have the bus bar electrode 411. The number of the electrode fingers 410 may be one or more, and is not particularly limited. However, if the number of the electrode fingers 410 is too small, the elastic wave leaks more, and the filter characteristics may be degraded. On the other hand, when the number of the electrode fingers 410 is too large, the reflector 322 is increased in size, so the entire high frequency filter 10B may be increased in size. For this reason, the number of electrode fingers 410 can be appropriately determined in consideration of the filter characteristics, the size, and the like required of the high frequency filter 10B.
　これらＩＤＴ電極３２１及び反射器３２２は、図１０の（ｂ）に示す電極膜３０１によって構成されている。本実施例では、電極膜３０１は、図１０の（ｂ）に示すように、密着層３０１ｇと主電極層３０１ｈとの積層構造となっている。なお、本実施例では、ＩＤＴ電極３２１と反射器３２２とが同じ電極膜３０１で構成されているが、これらは構造または組成等が互いに異なる電極膜で構成されていてもかまわない。 The IDT electrode 321 and the reflector 322 are constituted by an electrode film 301 shown in (b) of FIG. In the present embodiment, as shown in FIG. 10B, the electrode film 301 has a laminated structure of the adhesion layer 301 g and the main electrode layer 301 h. In the present embodiment, the IDT electrode 321 and the reflector 322 are formed of the same electrode film 301, but they may be formed of electrode films having different structures, compositions, and the like.
　密着層３０１ｇは、圧電基板３０２と主電極層３０１ｈとの密着性を向上させるための層であり、材料として、例えば、Ｔｉが用いられる。密着層３０１ｇの膜厚は、例えば、１２ｎｍである。 The adhesion layer 301g is a layer for improving the adhesion between the piezoelectric substrate 302 and the main electrode layer 301h, and, for example, Ti is used as a material. The film thickness of the adhesion layer 301 g is, for example, 12 nm.
　主電極層３０１ｈは、材料として、例えば、Ｃｕを１％含有したＡｌが用いられる。主電極層３０１ｈの膜厚は、例えば１６２ｎｍである。 As the material of the main electrode layer 301h, for example, Al containing 1% of Cu is used. The film thickness of the main electrode layer 301 h is, for example, 162 nm.
　圧電基板３０２は、電極膜３０１（すなわち、ＩＤＴ電極３２１及び反射器３２２）が形成された基板であり、例えば、ＬｉＴａＯ３圧電単結晶、ＬｉＮｂＯ３圧電単結晶、ＫＮｂＯ３圧電単結晶、水晶、または圧電セラミックスからなる。 The piezoelectric substrate 302, the electrode film 301 (i.e., IDT electrodes 321 and reflectors 322) substrate which is formed, for example, LiTaO 3 piezoelectric single crystal, LiNbO 3 piezoelectric single crystal, KNbO 3 piezoelectric single crystal, quartz or, It consists of piezoelectric ceramics.
　保護層３０３は、櫛歯電極３０１ａ及び３０１ｂを覆うように形成されている。保護層３０３は、主電極層３０１ｈを外部環境から保護する、周波数温度特性を調整する、及び、耐湿性を高めるなどを目的とする層であり、例えば、二酸化ケイ素を主成分とする膜である。 The protective layer 303 is formed to cover the comb- tooth electrodes 301a and 301b. The protective layer 303 is a layer for protecting the main electrode layer 301 h from the external environment, adjusting frequency temperature characteristics, enhancing moisture resistance, and the like, and is, for example, a film containing silicon dioxide as a main component. .
　なお、高周波フィルタ１０Ｂが有する各共振子の構造は、図１０に記載された構造に限定されない。例えば、電極膜３０１は、金属膜の積層構造でなく、金属膜の単層であってもよい。また、密着層３０１ｇ、主電極層３０１ｈ及び保護層３０３を構成する材料は、上述した材料に限定されない。また、電極膜３０１は、例えば、Ｔｉ、Ａｌ、Ｃｕ、Ｐｔ、Ａｕ、Ａｇ、Ｐｄなどの金属又は合金から構成されてもよく、上記の金属又は合金から構成される複数の積層体から構成されてもよい。また、保護層３０３は、形成されていなくてもよい。 In addition, the structure of each resonator which the high frequency filter 10B has is not limited to the structure described in FIG. For example, the electrode film 301 may be a single layer of a metal film instead of the laminated structure of the metal film. Further, the materials forming the adhesion layer 301 g, the main electrode layer 301 h, and the protective layer 303 are not limited to the above-described materials. Also, the electrode film 301 may be made of, for example, a metal or alloy such as Ti, Al, Cu, Pt, Au, Ag, Pd, etc., and is made of a plurality of laminates made of the above metals or alloys. May be In addition, the protective layer 303 may not be formed.
　以上のように構成された並列腕共振子ｐ１１では、ＩＤＴ電極３２１の設計パラメータ等によって、励振される弾性波の波長が規定される。以下、ＩＤＴ電極３２１の設計パラメータ、すなわち櫛歯電極３０１ａ及び櫛歯電極３０１ｂの設計パラメータについて説明する。 In the parallel arm resonator p11 configured as described above, the wavelength of the elastic wave to be excited is defined by the design parameters of the IDT electrode 321 and the like. Hereinafter, design parameters of the IDT electrode 321, that is, design parameters of the comb-tooth electrode 301a and the comb-tooth electrode 301b will be described.
　上記弾性波の波長は、図１０に示す櫛歯電極３０１ａ及び３０１ｂを構成する複数の電極指３１０ａまたは３１０ｂの繰り返し周期λで規定される。また、電極ピッチ（電極周期）とは、当該繰り返し周期λの１／２であり、櫛歯電極３０１ａ及び３０１ｂを構成する電極指３１０ａ及び３１０ｂのライン幅をＷとし、隣り合う電極指３１０ａと電極指３１０ｂとの間のスペース幅をＳとした場合、（Ｗ＋Ｓ）で定義される。また、ＩＤＴ電極３２１の交叉幅Ｌとは、図１０の（ａ）に示すように、櫛歯電極３０１ａの電極指３１０ａと櫛歯電極３０１ｂの電極指３１０ｂとを弾性波の伝搬方向から見た場合の重複する電極指長さである。また、電極デューティ（デューティ比）は、複数の電極指３１０ａ及び３１０ｂのライン幅占有率であり、複数の電極指３１０ａ及び３１０ｂのライン幅とスペース幅との加算値に対する当該ライン幅の割合であり、Ｗ／（Ｗ＋Ｓ）で定義される。また、対数とは、櫛歯電極３０１ａ及び３０１ｂのうち、対をなす電極指３１０ａ及び電極指３１０ｂの数であり、電極指３１０ａ及び電極指３１０ｂの総数の概ね半数である。例えば、対数をＮとし、電極指３１０ａ及び電極指３１０ｂの総数をＭとすると、Ｍ＝２Ｎ＋１を満たす。また、ＩＤＴ電極３２１の膜厚とは、複数の電極指３１０ａ及び３１０ｂの厚みｈである。 The wavelength of the elastic wave is defined by the repetition period λ of the plurality of electrode fingers 310a or 310b constituting the comb- tooth electrodes 301a and 301b shown in FIG. In addition, the electrode pitch (electrode period) is a half of the repetition period λ, and the line width of the electrode fingers 310a and 310b constituting the comb electrodes 301a and 301b is W, and the adjacent electrode fingers 310a and the electrodes are When the space width between the finger 310 b is S, it is defined by (W + S). Further, as shown in (a) of FIG. 10, the cross width L of the IDT electrode 321 is the electrode finger 310a of the comb electrode 301a and the electrode finger 310b of the comb electrode 301b viewed from the propagation direction of the elastic wave. In the case of overlapping electrode finger length. The electrode duty (duty ratio) is a line width occupancy rate of the plurality of electrode fingers 310a and 310b, and is a ratio of the line width to an added value of the line width and the space width of the plurality of electrode fingers 310a and 310b. , W / (W + S). The term "logarithm" means the number of the electrode fingers 310a and the electrode fingers 310b forming a pair out of the comb- tooth electrodes 301a and 301b, which is approximately half the total number of the electrode fingers 310a and the electrode fingers 310b. For example, assuming that the logarithm is N and the total number of the electrode fingers 310a and the electrode fingers 310b is M, M = 2N + 1 is satisfied. Further, the film thickness of the IDT electrode 321 is the thickness h of the plurality of electrode fingers 310 a and 310 b.
　次いで、反射器３２２の設計パラメータについて説明する。 Next, design parameters of the reflector 322 will be described.
　反射器３２２の電極ピッチ（電極周期）とは、電極指４１０のライン幅をＷＲＥＦとし、隣り合う電極指４１０間のスペース幅をＳＲＥＦとした場合、（ＷＲＥＦ＋ＳＲＥＦ）で定義される。また、反射器４２２の電極デューティ（デューティ比）は、複数の電極指４１０のライン幅占有率であり、電極指４１０のライン幅とスペース幅との加算値に対する当該ライン幅の割合であり、ＷＲＥＦ／（ＷＲＥＦ＋ＳＲＥＦ）で定義される。また、反射器３２２の膜厚とは、複数の電極指４１０の厚みである。 The electrode pitch (electrode period) of the reflector 322 is defined by (W REF + S REF ), where the line width of the electrode finger 410 is W REF and the space width between adjacent electrode fingers 410 is S REF. . Further, the electrode duty (duty ratio) of the reflector 422 is a line width occupancy rate of the plurality of electrode fingers 410 and is a ratio of the line width to the addition value of the line width of the electrode finger 410 and the space width. It is defined by REF / (W REF + S REF ). Further, the film thickness of the reflector 322 is the thickness of the plurality of electrode fingers 410.
　本実施例では、反射器３２２の電極ピッチ及び電極デューティは、ＩＤＴ電極３２１の電極ピッチ及び電極デューティと同等である。また、反射器３２２は、弾性波の伝搬方向から見た場合に、一組のバスバー電極４１１がＩＤＴ電極３２１のバスバー電極３１１ａ及び３１１ｂに重なるように配置されている。 In the present embodiment, the electrode pitch and electrode duty of the reflector 322 are equal to the electrode pitch and electrode duty of the IDT electrode 321. Further, the reflector 322 is arranged such that the pair of bus bar electrodes 411 overlap the bus bar electrodes 311 a and 311 b of the IDT electrode 321 when viewed in the propagation direction of the elastic wave.
　なお、反射器３２２は、弾性波の漏れを抑制する観点から上記構成が好ましいが、上記構成と異なる構成であってもかまわない。 The above configuration is preferable from the viewpoint of suppressing the leak of the elastic wave, but the configuration may be different from the above configuration.
　次に、ＩＤＴ電極３２１と反射器３２２との相対的な配置に関する設計パラメータについて説明する。 Next, design parameters relating to the relative arrangement of the IDT electrode 321 and the reflector 322 will be described.
　ＩＤＴ電極３２１と反射器３２２との間のピッチ（Ｉ－Ｒピッチ）は、（ｉ）ＩＤＴ電極３２１を構成する複数の電極指３１０ａまたは３１０ｂのうち最も反射器３２２側の電極指と（ｉｉ）反射器３２２を構成する複数の電極指４１０のうち最もＩＤＴ電極３２１側の電極指４１０との、中心間距離で定義される。このＩ－Ｒピッチは、櫛歯電極３０１ａ及び３０１ｂを構成する複数の電極指３１０ａまたは３１０ｂの繰り返し周期λ（すなわちＩＤＴ電極３２１の電極ピッチで定まる弾性波の波長λ）を用いて表すことができ、例えば、当該繰り返し周期λの０．５０倍の場合には０．５０λと表される。 The pitch (IR pitch) between the IDT electrode 321 and the reflector 322 is (i) the electrode finger closest to the reflector 322 among the plurality of electrode fingers 310 a or 310 b constituting the IDT electrode 321 and (ii) It is defined as a center-to-center distance with the electrode finger 410 closest to the IDT electrode 321 among the plurality of electrode fingers 410 constituting the reflector 322. This IR pitch can be expressed using a repetition period λ of a plurality of electrode fingers 310a or 310b constituting the comb- tooth electrodes 301a and 301b (ie, a wavelength λ of an elastic wave determined by the electrode pitch of the IDT electrode 321). For example, in the case where the repetition period λ is 0.50 times, it is expressed as 0.50 λ.
　本実施例に係る高周波フィルタ１０Ｂは、フィルタ１１Ｄおよび１２Ｄを備える。フィルタ１１Ｄは、ノードＸ１とノードＸ２とを結ぶ第１経路上に設けられた第１直列腕回路（直列腕共振子ｓ１１またはｓ１２）と、第１経路上に設けられたノードとグランドとに接続された第１並列腕回路（並列腕共振子ｐ１１）とを有する。フィルタ１２Ｄは、移相器２１と移相器２２とを結ぶ第２経路上に設けられた第２直列腕回路（直列腕共振子ｓ２１）と、第２経路上に設けられたノードとグランドとに接続された第２並列腕回路（並列腕共振子ｐ２１またはｐ２２）とを有する。第１直列腕回路、第１並列腕回路、第２直列腕回路、および、第２並列腕回路のうちの少なくとも１つは、弾性波共振子を有し、当該弾性波共振子の少なくとも１つは、圧電性を有する基板上に形成されたＩＤＴ電極と、反射器とを有する。ここで、上記弾性波共振子の少なくとも１つにおけるＩＤＴ電極の電極周期で定まる弾性波の波長をλとした場合、ＩＤＴ電極と反射器とのピッチは、０．４２λ以上かつ０．５０λ未満であることが望ましい。 The high frequency filter 10B according to the present embodiment includes the filters 11D and 12D. The filter 11D is connected to a first series arm circuit (series arm resonator s11 or s12) provided on a first path connecting the node X1 and the node X2, a node provided on the first path, and the ground. And the first parallel arm circuit (parallel arm resonator p11). The filter 12D includes a second series arm circuit (series arm resonator s21) provided on a second path connecting the phase shifters 21 and 22, a node provided on the second path, and a ground. And a second parallel arm circuit (parallel arm resonator p21 or p22) connected to At least one of the first series arm circuit, the first parallel arm circuit, the second series arm circuit, and the second parallel arm circuit has an elastic wave resonator, and at least one of the elastic wave resonators. Has an IDT electrode formed on a substrate having piezoelectricity, and a reflector. Here, assuming that the wavelength of the elastic wave determined by the electrode period of the IDT electrode in at least one of the elastic wave resonators is λ, the pitch between the IDT electrode and the reflector is 0.42 λ or more and less than 0.50 λ. It is desirable to have.
　上記のように、ＩＤＴ電極３２１および反射器３２２を有する弾性波共振子は、周期的に並べられた電極指からなる周期構造によって構成され、特定の周波数領域の弾性表面波を高い反射係数で反射する周波数帯域を有する。この周波数帯域は、一般的にストップバンドと称され、上記周期構造の繰り返し周期等によって規定される。このとき、ストップバンドの高域端では、反射係数が局所的に大きくなるリップルが生じる。さらに、ＩＤＴ電極３２１と反射器３２２とのピッチ（Ｉ－Ｒピッチ）を０．５λ以上にすると、ストップバンドの高域端でのリップルが大きくなり、フィルタの通過帯域内のリップルが大きくなり、挿入損失が大きくなる。一方、Ｉ－Ｒピッチを０．４２λ以上かつ０．５０λ未満とすることにより、ストップバンドの高域端でのリップルを低減できる。上記観点から、高周波フィルタ１０Ｂを構成するフィルタ１１Ｄおよび１２Ｄにおいて、Ｉ－Ｒピッチを０．４２λ以上かつ０．５０λ未満とすることにより、高周波フィルタ１０Ｂの通過帯域内の挿入損失を、より小さくすることができる。 As described above, the elastic wave resonator having the IDT electrode 321 and the reflector 322 is constituted by a periodic structure consisting of periodically arranged electrode fingers, and reflects the surface acoustic wave of a specific frequency region with a high reflection coefficient. Frequency band. This frequency band is generally referred to as a stop band, and is defined by the repetition period of the periodic structure or the like. At this time, at the high end of the stop band, ripples in which the reflection coefficient locally increases are generated. Furthermore, when the pitch (IR pitch) between the IDT electrode 321 and the reflector 322 is 0.5 λ or more, the ripple at the high end of the stop band becomes large, and the ripple in the pass band of the filter becomes large. Insertion loss increases. On the other hand, by setting the IR pitch to be 0.42 λ or more and less than 0.50 λ, it is possible to reduce the ripple at the high end of the stop band. From the above viewpoint, in the filters 11D and 12D constituting the high frequency filter 10B, by setting the IR pitch to be 0.42 λ or more and less than 0.50 λ, the insertion loss in the pass band of the high frequency filter 10B is further reduced. be able to.
　以下では、実施例１ａ～１ｄに係る高周波フィルタ１０Ｂと実施例１ｅに係る高周波フィルタとを比較する。 Hereinafter, the high frequency filter 10B according to the embodiments 1a to 1d and the high frequency filter according to the embodiment 1e will be compared.
　なお、実施例１ａ～１ｄに係る高周波フィルタ１０Ｂ、および、実施例１ｅに係る高周波フィルタ１０Ｂは、全て、図９に示された回路構成を有しているが、Ｉ－Ｒピッチの構成が以下のとおりとなっている。 Although all the high frequency filters 10B according to the embodiments 1a to 1d and the high frequency filter 10B according to the embodiment 1e have the circuit configuration shown in FIG. 9, the configuration of the IR pitch is as follows. It is as it is.
　（１）実施例１ｅ：全ての弾性波共振子（直列腕共振子ｓ１１、ｓ１２、ｓ２１および並列腕共振子ｐ１１、ｐ２１、ｐ２２）のＩ－Ｒピッチは０．５０λである。 (1) Embodiment 1e: The IR pitch of all the elastic wave resonators (series arm resonators s11, s12, s21 and parallel arm resonators p11, p21, p22) is 0.50 λ.
　（２）実施例１ａ：直列腕共振子ｓ１１およびｓ１２のＩ－Ｒピッチは０．４４λであり、それ以外の弾性波共振子のＩ－Ｒピッチは０．５０λである。 (2) Example 1a: The IR pitch of the series arm resonators s11 and s12 is 0.44 λ, and the IR pitch of the other elastic wave resonators is 0.50 λ.
　（３）実施例１ｂ：並列腕共振子ｐ１１のＩ－Ｒピッチは０．４４λであり、それ以外の弾性波共振子のＩ－Ｒピッチは０．５０λである。 (3) Example 1b: The IR pitch of the parallel arm resonator p11 is 0.44 λ, and the IR pitch of the other elastic wave resonators is 0.50 λ.
　（４）実施例１ｃ：並列腕共振子ｐ２１およびｐ２２のＩ－Ｒピッチは０．４４λであり、それ以外の弾性波共振子のＩ－Ｒピッチは０．５０λである。 (4) Example 1c: The IR pitch of the parallel arm resonators p21 and p22 is 0.44 λ, and the IR pitch of the other elastic wave resonators is 0.50 λ.
　（５）実施例１ｄ：全ての弾性波共振子（直列腕共振子ｓ１１、ｓ１２、ｓ２１および並列腕共振子ｐ１１、ｐ２１、ｐ２２）のＩ－Ｒピッチは０．４４λである。 (5) Embodiment 1d: The IR pitch of all the elastic wave resonators (series arm resonators s11, s12, s21 and parallel arm resonators p11, p21, p22) is 0.44λ.
　［１．６　実施例１ａ－１ｅに係る高周波フィルタ１０Ｂのフィルタ特性］
　図１１Ａは、実施例１ａおよび実施例１ｅに係る高周波フィルタの通過特性および反射特性を比較したグラフである。また、図１１Ｂは、実施例１ａおよび実施例１ｅに係る直列腕共振子ｓ１１およびｓ１２単体のインピーダンス特性および反射特性を比較したグラフである。なお、図１１Ａの左側には、実施例１ａおよび実施例１ｅに係る高周波フィルタの通過特性が示されている。また、図１１Ａの中央および右側には、実施例１ａおよび実施例１ｅに係るフィルタ１１Ｄ単体の通過特性および反射特性が示されている。 [1.6 Filter Characteristics of High-Frequency Filter 10B According to Embodiments 1a-1e]
FIG. 11A is a graph comparing the pass characteristic and the reflection characteristic of the high frequency filters according to Example 1a and Example 1e. FIG. 11B is a graph comparing impedance characteristics and reflection characteristics of the series arm resonators s11 and s12 alone according to Example 1a and Example 1e. On the left side of FIG. 11A, the pass characteristics of the high frequency filters according to Example 1a and Example 1e are shown. Further, in the center and the right side of FIG. 11A, the pass characteristic and the reflection characteristic of the single filter 11D according to Example 1a and Example 1e are shown.
　まず、図１１Ｂに示すように、実施例１ｅに係る高周波フィルタの直列腕共振子ｓ１１およびｓ１２では、ストップバンドの高域端において、反射損失が局所的に大きくなるリップルが生じる。これに対して、実施例１ａに係る高周波フィルタ１０Ｂの直列腕共振子ｓ１１およびｓ１２では、ストップバンドの高域端において、反射損失の上記リップルが低減されている（図中の領域Ａ４およびＡ５）。 First, as shown in FIG. 11B, in the series arm resonators s11 and s12 of the high frequency filter according to the example 1e, ripples in which the reflection loss is locally increased occur at the high frequency end of the stop band. On the other hand, in the series arm resonators s11 and s12 of the high frequency filter 10B according to the embodiment 1a, the ripple of the reflection loss is reduced at the high end of the stop band (areas A4 and A5 in the figure). .
　これに伴い、図１１Ａの右側の拡大グラフに示すように、実施例１ｅに係るフィルタ１１Ｄでは、ストップバンドの高域端において、反射損失が局所的に大きくなるリップルが生じる。これに対して、実施例１ａに係るフィルタ１１Ｄでは、ストップバンドの高域端において、反射損失の上記リップルが低減されている（図中の領域Ａ２およびＡ３）。 Along with this, as shown in the enlarged graph on the right side of FIG. 11A, in the filter 11D according to the example 1e, a ripple in which the reflection loss locally increases occurs at the high frequency end of the stop band. On the other hand, in the filter 11D according to the example 1a, the ripple of the reflection loss is reduced at the high frequency end of the stop band (areas A2 and A3 in the drawing).
　この結果、図１１Ａの左側のグラフに示すように、実施例１ｅに係る高周波フィルタ１０Ｂでは、通過帯域高域端において、挿入損失が局所的に大きくなるリップルが生じる。これに対して、実施例１ａに係る高周波フィルタ１０Ｂでは、通過帯域高域端において、挿入損失が低減している（図中の領域Ａ１）。 As a result, as shown in the graph on the left side of FIG. 11A, in the high frequency filter 10B according to the example 1e, ripples in which the insertion loss locally increases occur at the high end of the pass band. On the other hand, in the high frequency filter 10B according to the example 1a, the insertion loss is reduced at the high end of the pass band (area A1 in the drawing).
　すなわち、実施例１ａに係る高周波フィルタ１０Ｂは、フィルタ１１Ｄおよび１２Ｄを備える。フィルタ１１Ｄは、ノードＸ１とノードＸ２とを結ぶ第１経路上に設けられた第１直列腕回路（直列腕共振子ｓ１１またはｓ１２）と、第１経路上に設けられたノードとグランドとに接続された第１並列腕回路（並列腕共振子ｐ１１）とを有する。フィルタ１２Ｄは、移相器２１および２２第２経路上に設けられた第２直列腕回路（直列腕共振子ｓ２１）と、第２経路上に設けられたノードとグランドとに接続された第２並列腕回路（並列腕共振子ｐ２１またはｐ２２）とを有する。第１直列腕回路は、弾性波共振子を有し、当該弾性波共振子は、圧電性を有する基板上に形成されたＩＤＴ電極と、反射器とを有する。ここで、上記弾性波共振子のＩＤＴ電極と反射器とのピッチは、０．４２λ以上かつ０．５０λ未満である。 That is, the high frequency filter 10B according to the embodiment 1a includes the filters 11D and 12D. The filter 11D is connected to a first series arm circuit (series arm resonator s11 or s12) provided on a first path connecting the node X1 and the node X2, a node provided on the first path, and the ground. And the first parallel arm circuit (parallel arm resonator p11). The filter 12D is connected to a second series arm circuit (series arm resonator s21) provided on the second path of the phase shifters 21 and 22, a node provided on the second path, and a ground. And a parallel arm circuit (parallel arm resonator p21 or p22). The first series arm circuit includes an elastic wave resonator, and the elastic wave resonator includes an IDT electrode formed on a substrate having piezoelectricity, and a reflector. Here, the pitch between the IDT electrode of the elastic wave resonator and the reflector is 0.42 λ or more and less than 0.50 λ.
　フィルタ１１Ｄの直列腕回路における直列腕共振子ｓ１１およびｓ１２は、フィルタ１１Ｄの通過帯域内に共振周波数を有し、フィルタ１１Ｄの通過帯域高域側のフィルタ１２Ｄの通過帯域に反共振周波数を有する。そして、直列腕共振子ｓ１１およびｓ１２のストップバンドは、フィルタ１２Ｄの通過帯域となる。これに対して、Ｉ－Ｒピッチを０．４２λ以上かつ０．５０λ未満とすることにより、ストップバンドの高域端でのリップルを低減できる。このため、Ｉ－Ｒピッチを０．４２λ以上かつ０．５０λ未満とすることにより、高周波フィルタ１０Ｂの通過帯域内の挿入損失を、より小さくすることができる。 The series arm resonators s11 and s12 in the series arm circuit of the filter 11D have a resonant frequency in the pass band of the filter 11D and an antiresonance frequency in the pass band of the filter 12D on the high band side of the pass band of the filter 11D. The stop bands of the series arm resonators s11 and s12 become the pass band of the filter 12D. On the other hand, by setting the IR pitch to 0.42 λ or more and less than 0.50 λ, it is possible to reduce the ripple at the high end of the stop band. Therefore, by making the IR pitch 0.42 λ or more and less than 0.50 λ, the insertion loss in the pass band of the high frequency filter 10B can be further reduced.
　図１２Ａは、実施例１ｂおよび実施例１ｅに係る高周波フィルタの通過特性を比較したグラフである。また、図１２Ｂは、実施例１ｂおよび実施例１ｅに係る並列腕共振子ｐ１１および単体のインピーダンス特性および反射特性を比較したグラフである。なお、図１２Ａの左側には、実施例１ｂおよび実施例１ｅに係る高周波フィルタの通過特性が示されている。また、図１２Ａの右側には、実施例１ｂおよび実施例１ｅに係るフィルタ１１Ｄ単体の通過特性が示されている。 FIG. 12A is a graph comparing the pass characteristics of the high frequency filters according to Example 1 b and Example 1 e. FIG. 12B is a graph comparing impedance characteristics and reflection characteristics of the parallel arm resonator p11 and the single body according to Example 1b and Example 1e. On the left side of FIG. 12A, the pass characteristics of the high frequency filter according to Example 1 b and Example 1 e are shown. Further, on the right side of FIG. 12A, the pass characteristics of the filter 11D alone according to Example 1b and Example 1e are shown.
　まず、図１２Ｂに示すように、実施例１ｅに係る高周波フィルタ１０Ｂの並列腕共振子ｐ１１では、ストップバンドの高域端において、反射損失が局所的に大きくなるリップルが生じる。これに対して、実施例１ｂに係る高周波フィルタ１０Ｂの並列腕共振子ｐ１１では、ストップバンドの高域端において、反射損失の上記リップルが低減されている（図中の領域Ｂ３）。 First, as shown in FIG. 12B, in the parallel arm resonator p11 of the high frequency filter 10B according to the example 1e, ripples in which the reflection loss locally increases occur at the high frequency end of the stop band. On the other hand, in the parallel arm resonator p11 of the high frequency filter 10B according to the example 1b, the ripple of the reflection loss is reduced at the high end of the stop band (region B3 in the drawing).
　これに伴い、図１２Ａの右側のグラフに示すように、実施例１ｅに係るフィルタ１１Ｄでは、ストップバンドの高域端に相当する通過帯域内において、挿入損失が局所的に大きくなるリップルが生じる。これに対して、実施例１ｂに係るフィルタ１１Ｄでは、ストップバンドの高域端に相当する通過帯域内おいて、挿入損失の上記リップルが低減されている（図中の領域Ｂ２）。 Along with this, as shown in the graph on the right side of FIG. 12A, in the filter 11D according to the example 1e, a ripple in which the insertion loss locally increases occurs in the passband corresponding to the high end of the stop band. On the other hand, in the filter 11D according to the example 1b, the ripple of the insertion loss is reduced in the pass band corresponding to the high end of the stop band (area B2 in the drawing).
　この結果、図１２Ａの左側のグラフに示すように、実施例１ｅに係る高周波フィルタ１０Ｂでは、通過帯域内において、挿入損失が局所的に大きくなるリップルが生じる。これに対して、実施例１ｂに係る高周波フィルタ１０Ｂでは、通過帯域内において、挿入損失が低減している（図中の領域Ｂ１）。 As a result, as shown in the graph on the left side of FIG. 12A, in the high frequency filter 10B according to the example 1e, ripples in which the insertion loss is locally increased occur in the passband. On the other hand, in the high frequency filter 10B according to the example 1b, the insertion loss is reduced in the pass band (region B1 in the drawing).
　すなわち、実施例１ｂに係る高周波フィルタ１０Ｂは、フィルタ１１Ｄおよび１２Ｄを備える。フィルタ１１Ｄは、ノードＸ１とノードＸ２とを結ぶ第１経路上に設けられた第１直列腕回路（直列腕共振子ｓ１１またはｓ１２）と、第１経路上に設けられたノードとグランドとに接続された第１並列腕回路（並列腕共振子ｐ１１）とを有する。フィルタ１２Ｄは、移相器２１および２２第２経路上に設けられた第２直列腕回路（直列腕共振子ｓ２１）と、第２経路上に設けられたノードとグランドとに接続された第２並列腕回路（並列腕共振子ｐ２１またはｐ２２）とを有する。第１並列腕回路は、弾性波共振子を有し、当該弾性波共振子は、圧電性を有する基板上に形成されたＩＤＴ電極と、反射器とを有する。ここで、上記弾性波共振子のＩＤＴ電極と反射器とのピッチは、０．４２λ以上かつ０．５０λ未満である。 That is, the high frequency filter 10B according to the embodiment 1b includes the filters 11D and 12D. The filter 11D is connected to a first series arm circuit (series arm resonator s11 or s12) provided on a first path connecting the node X1 and the node X2, a node provided on the first path, and the ground. And the first parallel arm circuit (parallel arm resonator p11). The filter 12D is connected to a second series arm circuit (series arm resonator s21) provided on the second path of the phase shifters 21 and 22, a node provided on the second path, and a ground. And a parallel arm circuit (parallel arm resonator p21 or p22). The first parallel arm circuit includes an elastic wave resonator, and the elastic wave resonator includes an IDT electrode formed on a substrate having piezoelectricity, and a reflector. Here, the pitch between the IDT electrode of the elastic wave resonator and the reflector is 0.42 λ or more and less than 0.50 λ.
　フィルタ１１Ｄの並列腕回路における並列腕共振子ｐ１１は、フィルタ１１Ｄの通過帯域低周波側に共振周波数を有し、フィルタ１１Ｄの通過帯域内に反共振周波数を有する。そして、並列腕共振子ｐ１１のストップバンドは、フィルタ１１Ｄの通過帯域内、または、フィルタ１１Ｄの通過帯域高周波数側に位置する。これに対して、Ｉ－Ｒピッチを０．４２λ以上かつ０．５０λ未満とすることにより、ストップバンドの高域端でのリップルを低減できる。このため、Ｉ－Ｒピッチを０．４２λ以上かつ０．５０λ未満とすることにより、高周波フィルタ１０Ｂの通過帯域内の挿入損失を、より小さくすることができる。 The parallel arm resonator p11 in the parallel arm circuit of the filter 11D has a resonant frequency on the low frequency side of the passband of the filter 11D and an antiresonant frequency in the passband of the filter 11D. The stop band of the parallel arm resonator p11 is located within the pass band of the filter 11D or on the high frequency side of the pass band of the filter 11D. On the other hand, by setting the IR pitch to 0.42 λ or more and less than 0.50 λ, it is possible to reduce the ripple at the high end of the stop band. Therefore, by making the IR pitch 0.42 λ or more and less than 0.50 λ, the insertion loss in the pass band of the high frequency filter 10B can be further reduced.
　図１３Ａは、実施例１ｃおよび実施例１ｅに係る高周波フィルタの通過特性および反射特性を比較したグラフである。また、図１３Ｂは、実施例１ｃおよび実施例１ｅに係る並列腕共振子ｐ２１およびｐ２２単体のインピーダンス特性および反射特性を比較したグラフである。なお、図１３Ａの左側には、実施例１ｃおよび実施例１ｅに係る高周波フィルタの通過特性が示されている。また、図１３Ａの右側には、実施例１ｃおよび実施例１ｅに係るフィルタ１２Ｄ単体の通過特性が示されている。 FIG. 13A is a graph comparing the pass characteristic and the reflection characteristic of the high frequency filters according to Example 1 c and Example 1 e. FIG. 13B is a graph comparing impedance characteristics and reflection characteristics of the parallel arm resonators p21 and p22 according to the example 1c and the example 1e. On the left side of FIG. 13A, the pass characteristics of the high frequency filters according to Example 1c and Example 1e are shown. Further, on the right side of FIG. 13A, the pass characteristics of the filter 12D alone according to Example 1c and Example 1e are shown.
　まず、図１３Ｂに示すように、実施例１ｅに係る高周波フィルタの並列腕共振子ｐ２１およびｐ２２では、ストップバンドの高域端において、反射損失が局所的に大きくなるリップルが生じる。これに対して、実施例１ｃに係る高周波フィルタ１０Ｂの並列腕共振子ｐ２１およびｐ２２では、ストップバンドの高域端において、反射損失の上記リップルが低減されている（図中の領域Ｃ３およびＣ４）。 First, as shown in FIG. 13B, in the parallel arm resonators p21 and p22 of the high frequency filter according to the example 1e, ripples in which the reflection loss becomes locally large occur at the high frequency end of the stop band. On the other hand, in the parallel arm resonators p21 and p22 of the high frequency filter 10B according to the example 1c, the ripple of the reflection loss is reduced at the high end of the stop band (areas C3 and C4 in the figure). .
　これに伴い、図１３Ａの右側のグラフに示すように、実施例１ｅに係るフィルタ１２Ｄでは、ストップバンドの高域端に相当する通過帯域内において、挿入損失が局所的に大きくなるリップルが生じる。これに対して、実施例１ｃに係るフィルタ１２Ｄでは、ストップバンドの高域端に相当する通過帯域内において、挿入損失の上記リップルが低減されている（図中の領域Ｃ２）。 Along with this, as shown in the graph on the right side of FIG. 13A, in the filter 12D according to the example 1e, a ripple in which the insertion loss locally increases occurs in the passband corresponding to the high end of the stop band. On the other hand, in the filter 12D according to the example 1c, the ripple of the insertion loss is reduced in the pass band corresponding to the high end of the stop band (region C2 in the drawing).
　この結果、図１３Ａの左側のグラフに示すように、実施例１ｅに係る高周波フィルタ１０Ｂでは、通過帯域内において、挿入損失が局所的に大きくなるリップルが生じる。これに対して、実施例１ｃに係る高周波フィルタ１０Ｂでは、通過帯域内において、挿入損失が低減している（図中の領域Ｃ１）。 As a result, as shown in the graph on the left side of FIG. 13A, in the high frequency filter 10B according to the example 1e, ripples in which the insertion loss becomes locally large occur in the pass band. On the other hand, in the high frequency filter 10B according to the example 1c, the insertion loss is reduced in the pass band (area C1 in the drawing).
　すなわち、実施例１ｃに係る高周波フィルタ１０Ｂは、フィルタ１１Ｄおよび１２Ｄを備える。フィルタ１１Ｄは、ノードＸ１とノードＸ２とを結ぶ第１経路上に設けられた第１直列腕回路（直列腕共振子ｓ１１またはｓ１２）と、第１経路上に設けられたノードとグランドとに接続された第１並列腕回路（並列腕共振子ｐ１１）とを有する。フィルタ１２Ｄは、移相器２１および２２第２経路上に設けられた第２直列腕回路（直列腕共振子ｓ２１）と、第２経路上に設けられたノードとグランドとに接続された第２並列腕回路（並列腕共振子ｐ２１またはｐ２２）とを有する。第２並列腕回路は、弾性波共振子を有し、当該弾性波共振子は、圧電性を有する基板上に形成されたＩＤＴ電極と、反射器とを有する。ここで、上記弾性波共振子のＩＤＴ電極と反射器とのピッチは、０．４２λ以上かつ０．５０λ未満である。 That is, the high frequency filter 10B according to the example 1c includes the filters 11D and 12D. The filter 11D is connected to a first series arm circuit (series arm resonator s11 or s12) provided on a first path connecting the node X1 and the node X2, a node provided on the first path, and the ground. And the first parallel arm circuit (parallel arm resonator p11). The filter 12D is connected to a second series arm circuit (series arm resonator s21) provided on the second path of the phase shifters 21 and 22, a node provided on the second path, and a ground. And a parallel arm circuit (parallel arm resonator p21 or p22). The second parallel arm circuit includes an elastic wave resonator, and the elastic wave resonator includes an IDT electrode formed on a substrate having piezoelectricity, and a reflector. Here, the pitch between the IDT electrode of the elastic wave resonator and the reflector is 0.42 λ or more and less than 0.50 λ.
　フィルタ１２Ｄの並列腕回路における並列腕共振子ｐ２１およびｐ２２は、フィルタ１２Ｄの通過帯域低周波側に共振周波数を有し、フィルタ１２Ｄの通過帯域内に反共振周波数を有する。そして、並列腕共振子ｐ２１およびｐ２２のストップバンドは、フィルタ１２Ｄの通過帯域内、または、フィルタ１２Ｄの通過帯域高周波側に位置する。これに対して、Ｉ－Ｒピッチを０．４２λ以上かつ０．５０λ未満とすることにより、ストップバンドの高域端でのリップルを低減できる。このため、Ｉ－Ｒピッチを０．４２λ以上かつ０．５０λ未満とすることにより、高周波フィルタ１０Ｂの通過帯域内の挿入損失を、より小さくすることができる。 The parallel arm resonators p21 and p22 in the parallel arm circuit of the filter 12D have a resonant frequency on the low frequency side of the pass band of the filter 12D and an antiresonant frequency in the pass band of the filter 12D. The stop bands of the parallel arm resonators p21 and p22 are located in the pass band of the filter 12D or on the high frequency side of the pass band of the filter 12D. On the other hand, by setting the IR pitch to 0.42 λ or more and less than 0.50 λ, it is possible to reduce the ripple at the high end of the stop band. Therefore, by making the IR pitch 0.42 λ or more and less than 0.50 λ, the insertion loss in the pass band of the high frequency filter 10B can be further reduced.
　図１４は、実施例１ｄおよび実施例１ｅに係る高周波フィルタの通過特性および反射特性を比較したグラフである。なお、図１４の左側には、実施例１ｄおよび実施例１ｅに係る高周波フィルタの通過特性が示されている。また、図１４の中央には、実施例１ｄおよび実施例１ｅに係るフィルタ１１Ｄ単体の通過特性および反射特性が示されている。また、図１４の右側には、実施例１ｄおよび実施例１ｅに係るフィルタ１２Ｄ単体の通過特性および反射特性が示されている。 FIG. 14 is a graph comparing the pass characteristic and the reflection characteristic of the high frequency filters according to Example 1 d and Example 1 e. On the left side of FIG. 14, the pass characteristics of the high frequency filter according to Example 1 d and Example 1 e are shown. Further, in the center of FIG. 14, the pass characteristic and the reflection characteristic of the single filter 11D according to Example 1 d and Example 1 e are shown. Further, on the right side of FIG. 14, the pass characteristic and the reflection characteristic of the single filter 12D according to Example 1 d and Example 1 e are shown.
　まず、図１４の中央のグラフに示すように、実施例１ｅに係るフィルタ１１Ｄの直列腕共振子および並列腕共振子では、ストップバンドの高域端において、反射損失が局所的に大きくなるリップルが生じる。これに対して、実施例１ｄに係る高周波フィルタ１０Ｂの直列腕共振子および並列腕共振子では、ストップバンドの高域端において、反射損失の上記リップルが低減されている（図中の領域Ｄ５およびＤ６）。 First, as shown in the center graph of FIG. 14, in the series arm resonator and the parallel arm resonator of the filter 11D according to the example 1e, ripples in which the reflection loss locally increases at the high end of the stop band It occurs. On the other hand, in the series arm resonator and the parallel arm resonator of the high frequency filter 10B according to the embodiment 1d, the ripple of the reflection loss is reduced at the high frequency end of the stop band (region D5 in FIG. D6).
　また、図１４の右側のグラフに示すように、実施例１ｅに係るフィルタ１２Ｄの直列腕共振子および並列腕共振子では、ストップバンドの高域端において、反射損失が局所的に大きくなるリップルが生じる。これに対して、実施例１ｄに係る高周波フィルタ１０Ｂの直列腕共振子および並列腕共振子では、ストップバンドの高域端において、反射損失の上記リップルが低減されている（図中の領域Ｄ９およびＤ１０）。 Further, as shown in the graph on the right side of FIG. 14, in the series arm resonator and the parallel arm resonator of the filter 12D according to the example 1e, ripples in which the reflection loss locally increases at the high frequency end of the stop band It occurs. On the other hand, in the series arm resonator and the parallel arm resonator of the high frequency filter 10B according to the embodiment 1d, the ripple of the reflection loss is reduced at the high frequency end of the stop band (region D9 in FIG. D10).
　この結果、図１４の左側のグラフに示すように、実施例１ｅに係る高周波フィルタでは、通過帯域内および高域端において、挿入損失が局所的に大きくなるリップルが生じる。これに対して、実施例１ｄに係る高周波フィルタ１０Ｂでは、通過帯域内および高域端において、挿入損失が低減している（図中の領域Ｄ１およびＤ２）。 As a result, as shown in the graph on the left side of FIG. 14, in the high frequency filter according to the example 1e, ripples in which the insertion loss is locally increased occur in the pass band and the high band end. On the other hand, in the high frequency filter 10B according to the embodiment 1d, the insertion loss is reduced in the pass band and at the high frequency end (areas D1 and D2 in the drawing).
　すなわち、実施例１ｄに係る高周波フィルタ１０Ｂは、フィルタ１１Ｄおよび１２Ｄを備える。フィルタ１１Ｄは、ノードＸ１とノードＸ２とを結ぶ第１経路上に設けられた第１直列腕回路（直列腕共振子ｓ１１またはｓ１２）と、第１経路上に設けられたノードとグランドとに接続された第１並列腕回路（並列腕共振子ｐ１１）とを有する。フィルタ１２Ｄは、移相器２１および２２第２経路上に設けられた第２直列腕回路（直列腕共振子ｓ２１）と、第２経路上に設けられたノードとグランドとに接続された第２並列腕回路（並列腕共振子ｐ２１またはｐ２２）とを有する。第１直列腕回路、第２直列腕回路、第１並列腕回路、および第２並列腕回路のそれぞれは、弾性波共振子を有し、当該弾性波共振子は、圧電性を有する基板上に形成されたＩＤＴ電極と、反射器とを有する。ここで、上記弾性波共振子のＩＤＴ電極と反射器とのピッチは、０．４２λ以上かつ０．５０λ未満である。高周波フィルタ１０Ｂを構成する全ての弾性波共振子のＩ－Ｒピッチを０．４２λ以上かつ０．５０λ未満とすることにより、ストップバンドの高域端でのリップルを低減できる。このため、Ｉ－Ｒピッチを０．４２λ以上かつ０．５０λ未満とすることにより、高周波フィルタ１０Ｂの通過帯域内の挿入損失を、より小さくすることができる。 That is, the high frequency filter 10B according to the embodiment 1d includes the filters 11D and 12D. The filter 11D is connected to a first series arm circuit (series arm resonator s11 or s12) provided on a first path connecting the node X1 and the node X2, a node provided on the first path, and the ground. And the first parallel arm circuit (parallel arm resonator p11). The filter 12D is connected to a second series arm circuit (series arm resonator s21) provided on the second path of the phase shifters 21 and 22, a node provided on the second path, and a ground. And a parallel arm circuit (parallel arm resonator p21 or p22). Each of the first series arm circuit, the second series arm circuit, the first parallel arm circuit, and the second parallel arm circuit has an elastic wave resonator, and the elastic wave resonator is formed on a substrate having piezoelectricity. It has the formed IDT electrode and a reflector. Here, the pitch between the IDT electrode of the elastic wave resonator and the reflector is 0.42 λ or more and less than 0.50 λ. By setting the IR pitches of all the elastic wave resonators constituting the high frequency filter 10B to 0.42 λ or more and less than 0.50 λ, it is possible to reduce the ripple at the high end of the stop band. Therefore, by making the IR pitch 0.42 λ or more and less than 0.50 λ, the insertion loss in the pass band of the high frequency filter 10B can be further reduced.
　なお、フィルタ１２Ｄの直列腕共振子ｓ２１のＩ－Ｒピッチに関しては、高周波フィルタ１０Ｂの挿入損失には直接影響しないが、高周波フィルタ１０Ｂと、当該高周波フィルタ１０Ｂより高周波側の通過帯域を有するフィルタ（高周波フィルタ１０Ｂのストップバンドリップルの発生周波数と通過帯域が重複するフィルタ）とでマルチプレクサを構成する場合、当該フィルタの挿入損失を低減することが可能となる。 The IR pitch of the series arm resonator s21 of the filter 12D does not directly affect the insertion loss of the high frequency filter 10B, but the high frequency filter 10B and a filter having a passband on the higher frequency side than the high frequency filter 10B In the case where the multiplexer is configured by a filter in which the generation frequency of the stop band ripple of the high frequency filter 10B and the pass band overlap with each other, the insertion loss of the filter can be reduced.
　ここで、Ｉ－Ｒピッチと弾性波共振子特性との関係を詳細に説明する。 Here, the relationship between the IR pitch and the elastic wave resonator characteristics will be described in detail.
　図１５Ａは、典型例の共振子において、Ｉ－Ｒピッチを０．４０λ～０．５０λで変化させた場合の特性の変化を表すグラフである。また、図１５Ｂは、典型例の共振子において、Ｉ－Ｒピッチを０．５０λ～０．６０λで変化させた場合の特性の変化を表すグラフである。 FIG. 15A is a graph showing a change in characteristics when the IR pitch is changed at 0.40 λ to 0.50 λ in the resonator of the typical example. FIG. 15B is a graph showing a change in characteristics when the IR pitch is changed at 0.50 λ to 0.60 λ in the resonator of the typical example.
　図１５Ａおよび図１５Ｂの双方において、（ａ）はインピーダンスの絶対値を表すグラフであり、（ｂ）は位相特性を表すグラフであり、（ｃ－１）はインピーダンスをスミスチャート表記したグラフであり、（ｃ－２）は反射損失（リターンロス）を表すグラフである。具体的には、図１５Ａには、図中の凡例に示すように、Ｉ－Ｒピッチを０．０２λ刻みで０．４０λから０．５０λまで変化させた場合の共振子の特性が示されている。また、図１５Ｂには、図中の凡例に示すように、Ｉ－Ｒピッチを０．０２λ刻みで０．５０λから０．６０λまで変化させた場合の共振子の特性が示されている。 In both FIG. 15A and FIG. 15B, (a) is a graph showing the absolute value of the impedance, (b) is a graph showing the phase characteristic, and (c-1) is a graph in which the impedance is represented by a Smith chart. , (C-2) is a graph showing reflection loss (return loss). Specifically, FIG. 15A shows the characteristics of the resonator when the IR pitch is changed from 0.40 λ to 0.50 λ in steps of 0.02 λ, as shown in the legend in the figure. There is. Further, FIG. 15B shows the characteristics of the resonator when the IR pitch is changed from 0.50 λ to 0.60 λ in steps of 0.02 λ, as indicated by the legend in the figure.
　図１５Ｂに示すように、Ｉ－Ｒピッチが０．５λから大きくなるほど、ストップバンド高域端（具体的には反共振周波数の高域側）のリップル（同図の（ｂ）、（ｃ－１）及び（ｃ－２）の破線囲み内）が大きくなることが分かる。 As shown in FIG. 15B, as the IR pitch increases from 0.5λ, the ripples at the high end of the stop band (specifically, the high side of the antiresonance frequency) ((b), (c− in the same figure). It can be seen that 1) and (c-2) within the dashed box become larger.
　これに対して、図１５Ａに示すように、Ｉ－Ｒピッチが０．５λから小さくなるほど、ストップバンド高域端（具体的には反共振周波数の高域側）のリップル（同図の（ｂ）、（ｃ－１）及び（ｃ－２）の実線囲み内）が抑制されることが分かる。しかし、一方で、Ｉ－Ｒピッチが小さくなると共振周波数高域側（具体的には共振周波数と反共振周波数との間）に新たなリップル（同図の（ｂ）、（ｃ－１）及び（ｃ－２）の破線囲み内）が発生し、このリップルはＩ－Ｒピッチが狭くなるほど大きくなることが分かる。 On the other hand, as shown in FIG. 15A, as the IR pitch becomes smaller from 0.5λ, the ripple at the high end of the stop band (specifically, the high side of the antiresonance frequency) ((b of FIG. 15B). ), (C-1) and (c-2) are suppressed. However, on the other hand, when the IR pitch becomes smaller, a new ripple ((b), (c-1) and (c-1) in the same figure) on the higher side of the resonant frequency (specifically, between the resonant frequency and the antiresonant frequency). (C-2) (in the broken line box) occurs, and it can be seen that this ripple increases as the IR pitch narrows.
　つまり、共振子のＩ－Ｒピッチを０．４２λ以上かつ０．５０λ以下とすることにより、共振子の共振周波数の高域側に発生し得るリップルを抑制することができるので、当該リップルによる通過帯域の挿入損失の悪化を抑制することができる。 That is, by setting the IR pitch of the resonator to 0.42λ or more and 0.50λ or less, it is possible to suppress the ripple that may occur on the high frequency side of the resonance frequency of the resonator, so the passage by the ripple It is possible to suppress the deterioration of the insertion loss of the band.
　また、共振子のＩ－Ｒピッチを０．４４λ以上かつ０．４６λ以下とすることにより、（ｉ）ストップバンド高域端のリップル、及び、（ｉｉ）共振周波数の高域側に発生し得るリップルの双方を抑制することができるので、これら双方のリップルによる通過帯域内の挿入損失を抑制することができる。 In addition, by setting the IR pitch of the resonator to be 0.44 λ or more and 0.46 λ or less, (i) ripple at the high end of the stop band, and (ii) may be generated on the high frequency side of the resonance frequency. Since both of the ripples can be suppressed, it is possible to suppress the insertion loss in the passband due to these two ripples.
　（実施の形態２）
　本実施の形態では、実施の形態１に係る高周波フィルタ１０に対して、スイッチ素子の導通（オン）および非導通（オフ）の切り替えにより、通過帯域を可変する機能が付加された高周波フィルタについて説明する。 Second Embodiment
In the present embodiment, a high frequency filter having a function of changing the passband by switching between conduction (on) and non-conduction (off) of the switch element to the high frequency filter 10 according to the first embodiment will be described. Do.
　［２．１　実施の形態２に係る高周波フィルタ２０の基本構成］
　図１６は、実施の形態２に係る高周波フィルタ２０の回路ブロック図である。同図に示された高周波フィルタ２０は、フィルタ１１および１２と、移相器２１および２２と、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６と、入出力端子１１０および１２０と、を備える。本実施の形態に係る高周波フィルタ２０は、実施の形態１に係る高周波フィルタ１０と比較して、第１回路および第２回路が、スイッチ素子を有している点が構成として異なる。以下、本実施の形態に係る高周波フィルタ２０について、実施の形態１に係る高周波フィルタ１０と同じ点は説明を省略し、異なる点を中心に説明する。 [2.1 Basic Configuration of High-Frequency Filter 20 According to Second Embodiment]
FIG. 16 is a circuit block diagram of the high frequency filter 20 according to the second embodiment. The high frequency filter 20 shown in the figure includes filters 11 and 12, phase shifters 21 and 22, switches SW1, SW2, SW5 and SW6, and input / output terminals 110 and 120. The high frequency filter 20 according to the present embodiment is different from the high frequency filter 10 according to the first embodiment in that the first circuit and the second circuit include switch elements. Hereinafter, the high frequency filter 20 according to the present embodiment will not be described the same as the high frequency filter 10 according to the first embodiment, and different points will be mainly described.
　第１回路は、さらに、スイッチＳＷ１と、スイッチＳＷ２とを有する。 The first circuit further includes a switch SW1 and a switch SW2.
　第２回路は、さらに、スイッチＳＷ５と、スイッチＳＷ６とを有する。 The second circuit further includes a switch SW5 and a switch SW6.
　スイッチＳＷ１は、ノードＸ１とフィルタ１１との間に接続された第１スイッチ素子である。スイッチＳＷ２は、ノードＸ２とフィルタ１１との間に接続された第２スイッチ素子である。スイッチＳＷ５は、ノードＸ１と移相器２１との間に接続された第５スイッチ素子である。スイッチＳＷ６は、ノードＸ２と移相器２２との間に接続された第６スイッチ素子である。 The switch SW1 is a first switch element connected between the node X1 and the filter 11. The switch SW2 is a second switch element connected between the node X2 and the filter 11. The switch SW5 is a fifth switch element connected between the node X1 and the phase shifter 21. The switch SW6 is a sixth switch element connected between the node X2 and the phase shifter 22.
　［２．２　実施例２に係る高周波フィルタ２０Ａの回路構成］
　図１７Ａは、実施例２に係る高周波フィルタ２０Ａの回路構成図である。同図に示された高周波フィルタ２０Ａは、フィルタ１１Ａおよび１２Ａと、移相器２１および２２と、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６と、入出力端子１１０および１２０と、を備える。高周波フィルタ２０Ａは、実施の形態２に係る高周波フィルタ２０の具体的回路構成例であり、フィルタ１１Ａはフィルタ１１の具体的回路構成例であり、フィルタ１２Ａはフィルタ１２の具体的回路構成例である。本実施例に係る高周波フィルタ２０Ａは、実施例１に係る高周波フィルタ１０Ａと比較して、第１回路とノードＸ１およびＸ２との間、ならびに、第２回路とノードＸ１およびＸ２との間に、スイッチ素子が配置されている点が構成として異なる。以下、本実施例に係る高周波フィルタ２０Ａについて、実施例１に係る高周波フィルタ１０Ａと同じ点は説明を省略し、異なる点を中心に説明する。 [2.2 Circuit configuration of the high frequency filter 20A according to the second embodiment]
FIG. 17A is a circuit configuration diagram of a high frequency filter 20A according to a second embodiment. The high frequency filter 20A shown in the figure includes filters 11A and 12A, phase shifters 21 and 22, switches SW1, SW2, SW5 and SW6, and input / output terminals 110 and 120. The high frequency filter 20A is a specific circuit configuration example of the high frequency filter 20 according to the second embodiment, the filter 11A is a specific circuit configuration example of the filter 11, and the filter 12A is a specific circuit configuration example of the filter 12 . Compared to the high frequency filter 10A according to the first embodiment, the high frequency filter 20A according to the present embodiment is located between the first circuit and the nodes X1 and X2, and between the second circuit and the nodes X1 and X2. The configuration is different in that switch elements are disposed. Hereinafter, the high frequency filter 20A according to the present embodiment will not be described the same as the high frequency filter 10A according to the first embodiment, and differences will be mainly described.
　第１回路は、さらに、スイッチＳＷ１と、スイッチＳＷ２とを有する。 The first circuit further includes a switch SW1 and a switch SW2.
　第２回路は、さらに、スイッチＳＷ５と、スイッチＳＷ６とを有する。 The second circuit further includes a switch SW5 and a switch SW6.
　スイッチＳＷ１は、ノードＸ１とフィルタ１１との間に接続された第１スイッチ素子である。スイッチＳＷ２は、ノードＸ２とフィルタ１１との間に接続された第２スイッチ素子である。スイッチＳＷ５は、ノードＸ１と移相器２１との間に接続された第５スイッチ素子である。スイッチＳＷ６は、ノードＸ２と移相器２２との間に接続された第６スイッチ素子である。 The switch SW1 is a first switch element connected between the node X1 and the filter 11. The switch SW2 is a second switch element connected between the node X2 and the filter 11. The switch SW5 is a fifth switch element connected between the node X1 and the phase shifter 21. The switch SW6 is a sixth switch element connected between the node X2 and the phase shifter 22.
　表２に、実施例２に係る高周波フィルタ２０Ａの回路パラメータを示す。 Table 2 shows circuit parameters of the high frequency filter 20A according to the second embodiment.
Figure JPOXMLDOC01-appb-T000002
Figure JPOXMLDOC01-appb-T000002
　以下、実施例２および後述する実施例３に係る高周波フィルタでは、第２帯域をＬＴＥのＢａｎｄ３受信通過帯域（１８０５－１８８０ＭＨｚ）とし、第３帯域をＬＴＥのＢａｎｄ３９通過帯域（１８８０－１９２０ＭＨｚ）とし、第１帯域を（Ｂａｎｄ３受信通過帯域＋Ｂａｎｄ３９通過帯域）（１８０５－１９２０ＭＨｚ）としている。 Hereinafter, in the high frequency filter according to Example 2 and Example 3 to be described later, the second band is set to Band 3 reception pass band (1805-1880 MHz) of LTE, and the third band is set to Band 39 pass band (1880 to 1920 MHz) of LTE. The first band is (Band 3 reception pass band + Band 39 pass band) (1805-1920 MHz).
　つまり、高周波フィルタ２０Ａは、第２帯域をＢａｎｄ３受信通過帯域としたフィルタ１１Ａ、および、第３帯域をＢａｎｄ３９通過帯域としたフィルタ１２Ａの２つのフィルタを有し、フィルタ１２ＡとノードＸ１との間に移相器２１が配置され、フィルタ１２ＡとノードＸ２との間に移相器２２が配置されている。 That is, the high frequency filter 20A has two filters of the filter 11A whose band 2 is the Band 3 reception passband and the filter 12A whose band 3 is the band 39 passband, and it is between the filter 12A and the node X1. The phase shifter 21 is disposed, and the phase shifter 22 is disposed between the filter 12A and the node X2.
　［２．３　実施例２に係る高周波フィルタ２０Ａのフィルタ特性］
　図１７Ｂは、実施例２に係る高周波フィルタ２０Ａのスイッチの切り替えによる通過特性の変化を表すグラフである。図１７Ｂの（ａ）に示すように、高周波フィルタ２０Ａは、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６の全てを導通とした場合、第１帯域（Ｂａｎｄ３受信通過帯域＋Ｂａｎｄ３９通過帯域）を通過帯域とするフィルタとなる。また、図１７Ｂの（ｂ）に示すように、スイッチＳＷ１およびＳＷ２を導通とし、ＳＷ５およびＳＷ６を非導通とした場合、第２帯域（Ｂａｎｄ３受信通過帯域）を通過帯域とするフィルタとなる。また、図１７Ｂの（ｃ）に示すように、スイッチＳＷ１およびＳＷ２を非導通とし、ＳＷ５およびＳＷ６を導通とした場合、第３帯域（Ｂａｎｄ３９通過帯域）を通過帯域とするフィルタとなる。 [2.3 Filter Characteristics of High-Frequency Filter 20A According to Second Embodiment]
FIG. 17B is a graph showing a change in passage characteristics due to the switching of the switch of the high frequency filter 20A according to the second embodiment. As shown in (a) of FIG. 17B, the high frequency filter 20A is a filter that uses the first band (Band 3 reception pass band + Band 39 pass band) as the pass band when all of the switches SW1, SW2, SW5 and SW6 are conductive. It becomes. Further, as shown in (b) of FIG. 17B, when the switches SW1 and SW2 are made conductive and the switches SW5 and SW6 are not made conductive, the filter becomes a pass band of the second band (Band 3 reception pass band). When the switches SW1 and SW2 are turned off and the switches SW5 and SW6 are turned on as shown in (c) of FIG. 17B, the third band (Band 39 pass band) serves as a pass band.
　なお、本実施例では、第１回路側にスイッチＳＷ１およびＳＷ２が配置され、第２回路側にスイッチＳＷ５およびＳＷ６が配置された回路構成としたが、第１回路側のみにスイッチが接続された構成でもよいし、第２回路側のみにスイッチが接続された構成でもよい。第１回路側のみにスイッチが接続された構成の場合、高周波フィルタは第１帯域と第３帯域とを切り替えるフィルタとなる。また、第２回路側のみにスイッチが接続された構成の場合、高周波フィルタは第１帯域と第２帯域とを切り替えるフィルタとなる。 In this embodiment, the switches SW1 and SW2 are disposed on the first circuit side, and the switches SW5 and SW6 are disposed on the second circuit side. However, the switches are connected only to the first circuit side. It may be a configuration or a configuration in which a switch is connected only to the second circuit side. When the switch is connected only to the first circuit side, the high frequency filter serves as a filter that switches between the first band and the third band. Further, in the case where the switch is connected only to the second circuit side, the high frequency filter serves as a filter that switches between the first band and the second band.
　実施例２に係る高周波フィルタ２０Ａにおいても、実施例１に係る高周波フィルタ１０Ａと同様に、フィルタ１２Ａの両方の入出力端子に移相器２１および２２が接続されることで、高周波フィルタ２０Ａの第１帯域において、リップルが低減されることで挿入損失が低減する。 In the high frequency filter 20A according to the second embodiment, as in the high frequency filter 10A according to the first embodiment, the phase shifters 21 and 22 are connected to both input and output terminals of the filter 12A, thereby In one band, the reduction of the ripple reduces the insertion loss.
　実施例２に係る高周波フィルタ２０Ａが通過帯域内リップルおよび通過帯域内挿入損失を低減する要因について、以下に説明する。 The factors that the high frequency filter 20A according to the second embodiment reduces in-passband ripple and in-passband insertion loss will be described below.
　図１８は、実施例２に係る高周波フィルタ２０Ａの広帯域化の要因を説明するグラフである。同図の（ａ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の通過特性が示されている。同図の（ｂ）には、スイッチＳＷ１が導通状態である場合のノードＸ１から第１回路単体を見たインピーダンス特性およびスイッチＳＷ５が導通状態である場合のノードＸ１から第２回路単体を見たインピーダンス特性が示されている。同図の（ｃ）には、スイッチＳＷ２が導通状態である場合のノードＸ２から第１回路単体を見たインピーダンス特性およびスイッチＳＷ６が導通状態である場合のノードＸ２から第２回路単体を見たインピーダンス特性が示されている。同図の（ｄ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の位相特性が示されている。同図の（ｅ）には、第１回路と第２回路との位相差が示されている。 FIG. 18 is a graph for explaining the factor of increasing the bandwidth of the high frequency filter 20A according to the second embodiment. The pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2 are shown in FIG. The impedance characteristics of the first circuit viewed from the node X1 when the switch SW1 is in the conductive state and the second circuit alone from the node X1 when the switch SW5 is in the conductive state are illustrated in FIG. Impedance characteristics are shown. The impedance characteristics of the first circuit viewed from the node X2 when the switch SW2 is in the conductive state and the second circuit alone from the node X2 when the switch SW6 is in the conductive state are illustrated in FIG. Impedance characteristics are shown. The phase characteristic of the 1st circuit single-piece | unit and 2nd circuit single-piece | unit between node X1-X2 is shown by (d) of the figure. The phase difference between the first circuit and the second circuit is shown in (e) of FIG.
　実施例２に係る高周波フィルタ２０Ａでは、第２回路単体をノードＸ１から見たインピーダンス｜Ｚ１１｜が極大となる特異点のうち最大のインピーダンスとなる周波数ｆ１１が、第２帯域の低域端の周波数ｆＡからフィルタ１２Ａの通過帯域低周波（側のカットオフ周波数ｆＣまでの周波数帯に含まれるように、移相器２１および２２の位相が調整されている。また、第２回路単体をノードＸ２から見たインピーダンス｜Ｚ２２｜が極大となる特異点のうち最大のインピーダンスとなる周波数ｆ２２が、第２帯域の低域端の周波数ｆＡからフィルタ１２Ａの通過帯域低域側のカットオフ周波数ｆＣまでの周波数帯に含まれるように、移相器２１および２２の位相が調整されている。 In the high frequency filter 20A according to the second embodiment, the impedance viewed second circuit itself from the node X1 | Z11 | frequency f 11 which is the maximum of the impedance of the singular point as a maximum is the low frequency end of the second band The phases of the phase shifters 21 and 22 are adjusted so as to be included in the frequency band from the frequency f A to the pass band low frequency of the filter 12 A (the cut-off frequency f C on the side). The frequency f 22 at which the maximum impedance is obtained at the singular point at which the impedance | Z 22 | at the maximum viewed from the node X 2 is the cut-off on the lower band side of the filter 12 A from the frequency f A at the lower band end of the second band The phases of the phase shifters 21 and 22 are adjusted so as to be included in the frequency band up to the frequency f C.
　フィルタ１２Ａの減衰極の周波数ｆＡでは位相変化が大きく、フィルタ１１Ａとの位相差が大きくなるとともに、振幅差も大きくなる。そのため、移相器２１および２２の位相調整によって、周波数ｆＡからカットオフ周波数ｆＣまでの周波数範囲に、第２回路のインピーダンス極大の特異点のうち最大のインピーダンスとなる周波数ｆ１１およびｆ２２を合わせることにより、上記周波数範囲の高周波信号が、第２回路のフィルタ１２Ａに流れこむことを抑制できる。つまり、第２回路が減衰している周波数帯域で第２回路への信号の回り込みを抑制できる。これにより、高周波フィルタ２０Ａの第１帯域内のリップルが抑制されて挿入損失を低減することが可能となる。 At the frequency f A of the attenuation pole of the filter 12A, the phase change is large, the phase difference with the filter 11A is large, and the amplitude difference is also large. Therefore, by the phase adjustment of the phase shifters 21 and 22, in the frequency range from the frequency f A to the cut-off frequency f C , the frequencies f 11 and f 22 that become the largest impedance among the singular points of the impedance maximum of the second circuit. As a result, the high frequency signal in the above frequency range can be prevented from flowing into the filter 12A of the second circuit. That is, it is possible to suppress the wraparound of the signal to the second circuit in the frequency band in which the second circuit is attenuated. As a result, the ripple in the first band of the high frequency filter 20A is suppressed, and the insertion loss can be reduced.
　さらに、移相器２１および２２の位相調整により、第１回路と第２回路との振幅差が小さい周波数帯で同位相となるように調整することにより、第１帯域に含まれる周波数の高周波信号は、第１回路および第２回路を経由した後で、互いに打ち消し合うことがない。これにより、第１帯域内のリップルを低減し、挿入損失を低減できる。 Further, the high frequency signal of the frequency included in the first band is adjusted by adjusting the phase difference of the first circuit and the second circuit to be in phase in a small frequency band by adjusting the phase of the phase shifters 21 and 22. Are not canceled each other after passing through the first circuit and the second circuit. Thereby, the ripple in the first band can be reduced and the insertion loss can be reduced.
　以上のように、高周波フィルタ２０Ａが、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６の導通状態により第１帯域を通過帯域とする場合、第１帯域の挿入損失を低減することが可能となる。 As described above, when the high frequency filter 20A sets the first band as the pass band due to the conduction state of the switches SW1, SW2, SW5, and SW6, it is possible to reduce the insertion loss of the first band.
　本実施例では、図１８の（ｅ）に示すように、スイッチＳＷ５およびＳＷ６が導通状態である場合にノードＸ１およびＸ２から第２回路を単体で見た場合、第２帯域において、第１回路と第２回路とが同位相（０°または－３６０°）になっている周波数が少なくとも１箇所（同図の破線丸印：３箇所）あり、第３帯域において、第１回路と第２回路とが同位相（０°または－３６０°）になっている周波数が少なくとも１箇所（同図の破線丸印：１箇所）あってもよい。 In the present embodiment, as shown in (e) of FIG. 18, when the switches SW5 and SW6 are in the conductive state, when the second circuit is viewed alone from the nodes X1 and X2, the first circuit is generated in the second band. And the second circuit have the same phase (0 ° or -360 °) in at least one point (broken line circle mark: three points in the same figure), and in the third band, the first circuit and the second circuit There may be at least one frequency (dotted line circle in the same figure: one place) in which the phase is the same phase (0 ° or -360 °).
　上記構成によれば、第１帯域、第２帯域、および第３帯域を切り替えることができる周波数可変型の高周波フィルタ２０Ａにおいて、第１帯域内のリップルを抑制し、挿入損失を低減することが可能となる。 According to the above configuration, it is possible to suppress the ripple in the first band and reduce the insertion loss in the variable-frequency high frequency filter 20A capable of switching the first band, the second band, and the third band. It becomes.
　［２．４　実施例３に係る高周波フィルタ３０のフィルタ特性］
　図１９は、実施例２に係る高周波フィルタ２０Ａのスイッチのオフ容量Ｃｏｆｆを変化させた場合の通過特性を表すグラフである。スイッチ素子が非導通の場合、当該スイッチ素子のインピーダンスは理想的には無限大であるが、実際には、スイッチ素子を構成する半導体素子（ＦＥＴ等）の寄生容量により、非導通のスイッチ素子は容量成分を有する。オフ容量Ｃｏｆｆが大きくなるにつれ、スイッチ素子のインピーダンスが低くなる。 [2.4 Filter Characteristics of High-Frequency Filter 30 According to Third Embodiment]
FIG. 19 is a graph showing the pass characteristic when the off capacitance C off of the switch of the high frequency filter 20A according to the second embodiment is changed. When the switch element is nonconductive, the impedance of the switch element is ideally infinite, but actually, the switch element is nonconductive due to the parasitic capacitance of the semiconductor element (FET or the like) constituting the switch element. It has a capacity component. As the off capacitance C off increases, the impedance of the switch element decreases.
　このため、図１９の（ａ）に示すように、オフ状態となるスイッチＳＷ５およびＳＷ６の方に配置されている第２回路の影響を受けて、第１回路の通過帯域（第２帯域）および通過帯域高域側近傍の減衰帯域の特性が劣化する。より具体的には、スイッチＳＷ１およびＳＷ２を導通とし、スイッチＳＷ５およびＳＷ６を非導通として第２帯域を通過帯域とするフィルタ１１を選択した場合、オフ容量Ｃｏｆｆが大きくなるほど第２回路に信号が回り込む。このため、第２帯域高域側の減衰特性が悪化するとともに、第２帯域内にリップルが発生して挿入損失を悪化させる。 Therefore, as shown in (a) of FIG. 19, the passband (second band) of the first circuit and the second circuit under the influence of the second circuit disposed in the direction of the switches SW5 and SW6 to be turned off as shown in FIG. The characteristics of the attenuation band in the vicinity of the high pass side of the pass band are degraded. More specifically, when the filter 11 is selected in which the switches SW1 and SW2 are turned on and the switches SW5 and SW6 are turned off and the second band is set to the pass band, the larger the off capacitance C off, the more signal is sent to the second circuit. Get around. For this reason, the attenuation characteristics on the second band high band side deteriorate, and a ripple is generated in the second band to deteriorate the insertion loss.
　また、図１９の（ｂ）に示すように、オフ状態となるスイッチＳＷ１およびＳＷ２の方に配置されている第１回路の影響を受けて、第２回路の通過帯域（第３帯域）および通過帯域低域側近傍の減衰帯域の特性が劣化する。より具体的には、スイッチＳＷ１およびＳＷ２を非導通とし、スイッチＳＷ５およびＳＷ６を導通として第３帯域を通過帯域とするフィルタ１２を選択した場合、オフ容量Ｃｏｆｆが大きくなるほど第１回路に信号が回り込む。このため、第３帯域低域側の減衰特性が悪化するとともに、第３帯域内にリップルが発生して挿入損失を悪化させる。 Further, as shown in (b) of FIG. 19, the pass band (third band) and the pass of the second circuit are affected by the influence of the first circuit disposed in the direction of the switches SW1 and SW2 to be turned off as shown in FIG. The characteristics of the attenuation band near the lower band side deteriorate. More specifically, when the filter 12 is selected in which the switches SW1 and SW2 are nonconductive and the switches SW5 and SW6 are conductive and the third band is a pass band, the larger the off capacitance C off, the more signal is transmitted to the first circuit. Get around. As a result, the attenuation characteristics on the lower side of the third band deteriorate, and ripples occur in the third band to deteriorate the insertion loss.
　なお、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６の全てが導通の場合については、オフ容量Ｃｏｆｆの影響は無いため、記載していない。 The case where all of the switches SW1, SW2, SW5 and SW6 are conductive is not described because there is no influence of the off capacitance Coff .
　上記のようなスイッチ素子のオフ容量Ｃｏｆｆによるフィルタ特性の劣化を抑制する構成として図２０に示される構成が挙げられる。 The configuration shown in FIG. 20 can be mentioned as a configuration for suppressing the deterioration of the filter characteristics due to the off capacitance C off of the switch element as described above.
　図２０は、実施例３に係る高周波フィルタ３０の回路ブロック図である。同図に示された高周波フィルタ３０は、フィルタ１１および１２と、移相器２１および２２と、スイッチＳＷ１、ＳＷ２、ＳＷ３、ＳＷ４、ＳＷ５、ＳＷ６、ＳＷ７およびＳＷ８と、入出力端子１１０および１２０と、を備える。高周波フィルタ３０は、実施の形態２に係る高周波フィルタ２０と比較して、スイッチＳＷ３、ＳＷ４、ＳＷ７およびＳＷ８が配置されている点が構成として異なる。以下、本実施例に係る高周波フィルタ３０について、実施の形態２に係る高周波フィルタ２０と同じ点は説明を省略し、異なる点を中心に説明する。 FIG. 20 is a circuit block diagram of the high frequency filter 30 according to the third embodiment. The high frequency filter 30 shown in the figure includes the filters 11 and 12, the phase shifters 21 and 22, the switches SW 1, SW 2, SW 3, SW 4, SW 5, SW 6, SW 7 and SW 8, and the input and output terminals 110 and 120. And. The high frequency filter 30 differs from the high frequency filter 20 according to the second embodiment in that switches SW3, SW4, SW7 and SW8 are arranged. Hereinafter, the high frequency filter 30 according to the present embodiment will not be described the same as the high frequency filter 20 according to the second embodiment, and different points will be mainly described.
　第１回路は、さらに、スイッチＳＷ３と、スイッチＳＷ４とを有する。 The first circuit further includes a switch SW3 and a switch SW4.
　第２回路は、さらに、スイッチＳＷ７と、スイッチＳＷ８とを有する。 The second circuit further includes a switch SW7 and a switch SW8.
　スイッチＳＷ３は、フィルタ１１とスイッチＳＷ１との接続ノード、および、グランドに接続され、スイッチＳＷ１と排他的に導通および非導通が切り替えられる第３スイッチ素子である。スイッチＳＷ４は、フィルタ１１とスイッチＳＷ２との接続ノード、および、グランドに接続され、スイッチＳＷ２と排他的に導通および非導通が切り替えられる第４スイッチ素子である。スイッチＳＷ７は、移相器２１とスイッチＳＷ５との接続ノード、および、グランドに接続され、スイッチＳＷ５と排他的に導通および非導通が切り替えられる第７スイッチ素子である。スイッチＳＷ８は、移相器２２とスイッチＳＷ６との接続ノード、および、グランドに接続され、スイッチＳＷ６と排他的に導通および非導通が切り替えられる第８スイッチ素子である。 The switch SW3 is a third switch element connected to a connection node between the filter 11 and the switch SW1 and to the ground, and switched between conduction and non-conduction exclusively with the switch SW1. The switch SW4 is a fourth switch element connected to the connection node between the filter 11 and the switch SW2 and to the ground, and switched between conduction and non-conduction exclusively with the switch SW2. The switch SW7 is a connection node between the phase shifter 21 and the switch SW5, and is a seventh switch element connected to the ground and switched between conduction and non-conduction exclusively with the switch SW5. The switch SW8 is an eighth switch element connected to the connection node between the phase shifter 22 and the switch SW6 and to the ground, and switched between conduction and non-conduction exclusively with the switch SW6.
　高周波フィルタ３０は、第２帯域をＢａｎｄ３受信通過帯域としたフィルタ１１、および、第３帯域をＢａｎｄ３９通過帯域としたフィルタ１２の２つのフィルタを有し、フィルタ１２とノードＸ１との間に移相器２１が配置され、フィルタ１２とノードＸ２との間に移相器２２が配置されている。 The high frequency filter 30 has two filters, the filter 11 whose second band is the Band 3 reception passband and the filter 12 whose third band is the Band 39 passband, and the phase shift is made between the filter 12 and the node X1. The phase shifter 22 is disposed between the filter 12 and the node X2.
　図２１は、実施例３に係る高周波フィルタ３０の通過特性を表すグラフである。 FIG. 21 is a graph showing the pass characteristic of the high frequency filter 30 according to the third embodiment.
　同図の（ａ）に示すように、スイッチＳＷ５およびＳＷ６がオフ状態となっても、スイッチＳＷ７およびＳＷ８がオン状態となることで、第２回路に信号が回り込むことを抑制できる。このため、スイッチＳＷ７およびＳＷ８が配置されている高周波フィルタ３０（ＳＷ７＆ＳＷ８あり）は、理想ＳＷ（Ｃｏｆｆなし）と同等の通過特性となる。つまり、スイッチＳＷ５およびＳＷ６が非導通である第２回路の影響が低減され、第２帯域内のリップルを抑制して挿入損失を低減するとともに、第２帯域近傍の減衰特性を向上できる。 As shown in (a) of the figure, even when the switches SW5 and SW6 are turned off, the switches SW7 and SW8 are turned on, whereby the signal can be prevented from looping around in the second circuit. Therefore, the high frequency filter 30 (located SW7 & SW8) switches SW7 and SW8 are arranged, the same pass characteristic and the ideal SW (no C off). That is, the influence of the second circuit in which the switches SW5 and SW6 are nonconductive is reduced, the ripple in the second band is suppressed to reduce the insertion loss, and the attenuation characteristic in the vicinity of the second band can be improved.
　また、同図の（ｂ）に示すように、スイッチＳＷ１およびＳＷ２がオフ状態となっても、スイッチＳＷ３およびＳＷ４がオン状態となることで、第１回路に信号が回り込むことを抑制できる。このため、スイッチＳＷ３およびＳＷ４が配置されている高周波フィルタ３０（ＳＷ３＆ＳＷ４あり）は、理想ＳＷ（Ｃｏｆｆなし）と同等の通過特性となる。つまり、スイッチＳＷ１およびＳＷ２が非導通である第１回路の影響が低減され、第３帯域内のリップルを抑制して挿入損失を低減するとともに、第３帯域近傍の減衰特性を向上できる。 Further, as shown in (b) of the figure, even when the switches SW1 and SW2 are turned off, the switches SW3 and SW4 are turned on, so that the signal can be prevented from looping into the first circuit. For this reason, the high frequency filter 30 (with SW3 & SW4) in which the switches SW3 and SW4 are arranged has a passing characteristic equivalent to that of the ideal SW (without Coff ). That is, the influence of the first circuit in which the switches SW1 and SW2 are nonconductive is reduced, the ripple in the third band is suppressed to reduce the insertion loss, and the attenuation characteristics in the vicinity of the third band can be improved.
　［２．５　移相器２１および２２の回路構成例］
　実施例１および実施例２では、移相器２１および２２は理想素子であるものとしたが、ここでは、移相器２１および２２の具体的構成を例示する。 [2.5 Example of Circuit Configuration of Phase Shifters 21 and 22]
In the first and second embodiments, the phase shifters 21 and 22 are ideal elements, but here, specific configurations of the phase shifters 21 and 22 will be illustrated.
　図２２Ａは、実施の形態２の変形例１に係る高周波フィルタ２０Ｂの回路構成図である。同図に示された高周波フィルタ２０Ｂは、フィルタ１１Ａおよび１２Ａと、移相器２１Ｂおよび２２Ｂと、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６と、入出力端子１１０および１２０と、を備える。本変形例に係る高周波フィルタ２０Ｂは、実施例２に係る高周波フィルタ２０Ａと比較して、移相器の具体的回路構成が示されている点が異なる。以下、本変形例に係る高周波フィルタ２０Ｂについて、実施例２に係る高周波フィルタ２０Ａと同じ点は説明を省略し、異なる点を中心に説明する。 FIG. 22A is a circuit configuration diagram of a high frequency filter 20B according to a first modification of the second embodiment. The high frequency filter 20B shown in the figure includes filters 11A and 12A, phase shifters 21B and 22B, switches SW1, SW2, SW5 and SW6, and input / output terminals 110 and 120. The high frequency filter 20B according to the present modification differs from the high frequency filter 20A according to the second embodiment in that a specific circuit configuration of the phase shifter is shown. Hereinafter, the high frequency filter 20B according to the present modification will not be described the same as the high frequency filter 20A according to the second embodiment, and different points will be mainly described.
　移相器２１Ｂは、ノードＸ１とフィルタ１２Ａの一方の端子とを結ぶ直列腕経路に配置された２つのインダクタＬｓ２１、および、当該直列腕経路上のノードおよびグランドに接続されたキャパシタＣｐ２１で構成されたＴ型の第１移相器である。移相器２２Ｂは、ノードＸ２とフィルタ１２Ａの他方の端子とを結ぶ直列腕経路に配置された２つのインダクタＬｓ２２、および、当該直列腕経路上のノードおよびグランドに接続されたキャパシタＣｐ２２で構成されたＴ型の第２移相器である。移相器２１Ｂおよび移相器２２Ｂは、低域通過型フィルタ回路（ＬＰＦ型移相器）を構成している。なお、２つのインダクタＬｓ２１のインダクタンス値は異なっていてもよい。また、インダクタンスＬｓ２１は２つ配置されていなくてもよく、１つであってもよい。 The phase shifter 21B includes two inductors Ls21 arranged in a series arm path connecting the node X1 and one terminal of the filter 12A, and a capacitor Cp21 connected to the node on the series arm path and the ground. It is a T-type first phase shifter. The phase shifter 22B includes two inductors Ls22 arranged in a series arm path connecting the node X2 and the other terminal of the filter 12A, and a capacitor Cp22 connected to the node on the series arm path and the ground. It is a T-type second phase shifter. The phase shifter 21 B and the phase shifter 22 B constitute a low pass filter circuit (LPF phase shifter). The inductance values of the two inductors Ls 21 may be different. Further, two inductances Ls 21 may not be disposed, and may be one.
　図２２Ｂは、実施の形態２の変形例２に係る高周波フィルタ２０Ｃの回路構成図である。同図に示された高周波フィルタ２０Ｃは、フィルタ１１Ａおよび１２Ａと、移相器２１Ｃおよび２２Ｃと、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６と、入出力端子１１０および１２０と、を備える。本変形例に係る高周波フィルタ２０Ｃは、実施例２に係る高周波フィルタ２０Ａと比較して、移相器の具体的回路構成が示されている点が異なる。以下、本変形例に係る高周波フィルタ２０Ｃについて、実施例２に係る高周波フィルタ２０Ａと同じ点は説明を省略し、異なる点を中心に説明する。 FIG. 22B is a circuit configuration diagram of a high frequency filter 20C according to Modification 2 of Embodiment 2. The high frequency filter 20C shown in the figure includes filters 11A and 12A, phase shifters 21C and 22C, switches SW1, SW2, SW5 and SW6, and input / output terminals 110 and 120. The high frequency filter 20C according to the present modification differs from the high frequency filter 20A according to the second embodiment in that a specific circuit configuration of the phase shifter is shown. Hereinafter, the high frequency filter 20C according to the present modification will not be described the same as the high frequency filter 20A according to the second embodiment, and different points will be mainly described.
　移相器２１Ｃは、ノードＸ１とフィルタ１２Ａの一方の端子とを結ぶ直列腕経路に配置された２つのキャパシタＣｓ２１、および、当該直列腕経路上のノードおよびグランドに接続されたインダクタＬｐ２１で構成されたＴ型の第１移相器である。移相器２２Ｃは、ノードＸ２とフィルタ１２Ａの他方の端子とを結ぶ直列腕経路に配置された２つのキャパシタＣｓ２２、および、当該直列腕経路上のノードおよびグランドに接続されたインダクタＬｐ２２で構成されたＴ型の第２移相器である。移相器２１Ｃおよび移相器２２Ｃは、高域通過型フィルタ回路（ＨＰＦ型移相器）を構成している。なお、２つのキャパシタＣｓ２１の容量値は異なっていてもよい。また、キャパシタＣｓ２１は２つ配置されていなくてもよく、１つであってもよい。 The phase shifter 21C includes two capacitors Cs21 arranged in a series arm path connecting the node X1 and one terminal of the filter 12A, and an inductor Lp21 connected to the node on the series arm path and the ground. It is a T-type first phase shifter. The phase shifter 22C includes two capacitors Cs22 arranged in a series arm path connecting the node X2 and the other terminal of the filter 12A, and an inductor Lp22 connected to the node on the series arm path and the ground. It is a T-type second phase shifter. The phase shifters 21C and 22C constitute a high-pass filter circuit (HPF phase shifter). The capacitance values of the two capacitors Cs21 may be different. Further, two capacitors Cs21 may not be disposed, and may be one.
　変形例１に係る高周波フィルタ２０Ｂおよび変形例２に係る高周波フィルタ２０Ｃにおいても、移相器２１および２２を理想素子とした実施例２に係る高周波フィルタ２０Ａと同様の通過帯域内特性が得られる。つまり、第２回路の移相器２１Ｂおよび２２Ｂ、ならびに、移相器２１Ｃおよび２２Ｃは、第１帯域の振幅差が大きい周波数帯である第２回路の第３帯域低域側の減衰帯域、すなわち、第２回路において最も低い減衰極の周波数から、第２帯域低域側の１０ｄＢカットオフ周波数の範囲内に、第２帯域においてインピーダンスが最大となる周波数が位置するように、位相が調整されている。 Also in the high frequency filter 20B according to the first modification and the high frequency filter 20C according to the second modification, the same in-passband characteristics as the high frequency filter 20A according to the second embodiment having the phase shifters 21 and 22 as ideal elements are obtained. That is, the phase shifters 21B and 22B of the second circuit, and the phase shifters 21C and 22C are the attenuation bands on the third band lower side of the second circuit, which is a frequency band where the amplitude difference between the first bands is large. The phase is adjusted so that the frequency at which the impedance is maximized in the second band is located within the range of the lowest attenuation pole frequency in the second circuit to the 10 dB cutoff frequency on the second band lower side. There is.
　図２３は、実施の形態２の変形例１および２に係る高周波フィルタの通過特性を比較したグラフである。 FIG. 23 is a graph comparing the pass characteristics of the high frequency filters according to the first and second modifications of the second embodiment.
　移相器として高域通過型フィルタ回路が適用された場合には、図２３の（ａ）および（ｃ）に示すように、第１帯域および第３帯域選択時に、通過帯域低域側の減衰特性が向上する。 When a high-pass filter circuit is applied as a phase shifter, as shown in (a) and (c) of FIG. 23, attenuation in the low pass band side when the first and third bands are selected. The characteristics are improved.
　一方、移相器として低域通過型フィルタ回路が適用された場合には、図２３の（ａ）および（ｃ）に示すように、第１帯域および第３帯域選択時に、通過帯域高域側（例えば、中心周波数の２倍および３倍の周波数）の減衰特性が向上する。 On the other hand, when a low pass filter circuit is applied as a phase shifter, as shown in (a) and (c) of FIG. The attenuation characteristics (for example, frequencies twice and three times the center frequency) are improved.
　変形例１および２に係る高周波フィルタによれば、要求される高周波フィルタの特性に応じて、移相器の回路構成を適宜選択することが可能となる。 According to the high frequency filter according to the first and second modifications, the circuit configuration of the phase shifter can be appropriately selected according to the required characteristics of the high frequency filter.
　なお、変形例１に係る高周波フィルタ２０Ｂおよび変形例２に係る高周波フィルタ２０Ｃにおいて、第１移相器および第２移相器の一方が低域通過型フィルタ回路であって、他方が高域通過型フィルタ回路であってもよい。 In the high frequency filter 20B according to the first modification and the high frequency filter 20C according to the second modification, one of the first phase shifter and the second phase shifter is a low pass filter circuit, and the other is a high pass. It may be a type filter circuit.
　また、実施例１および２では、Ｔ型の移相器としたが、直列腕回路にキャパシタまたはインダクタの一方を有し、並列腕回路にキャパシタまたはインダクタの他方を有した、π型の移相器であってもよい。つまり、第１移相器および第２移相器は、２素子（２次）以上の素子で構成されていればよい。また、第１移相器および第２移相器は、ストリップラインまたはマイクロストリップラインなどで構成される遅延線であってもよい。 In Examples 1 and 2, although the T-type phase shifter is used, a π-type phase shift having one of a capacitor or an inductor in a series arm circuit and the other of a capacitor or an inductor in a parallel arm circuit It may be a container. That is, the first phase shifter and the second phase shifter may be configured by two or more elements (secondary). Also, the first phase shifter and the second phase shifter may be delay lines composed of strip lines or micro strip lines.
　（実施の形態３）
　本実施の形態では、第１回路および第２回路に加えて、第３フィルタおよびスイッチ素子で構成された回路が、ノードＸ１およびノードＸ２に接続された高周波フィルタについて説明する。 Third Embodiment
In this embodiment, in addition to the first circuit and the second circuit, a high frequency filter will be described in which a circuit configured of a third filter and a switch element is connected to the node X1 and the node X2.
　［３．１　実施例４に係る高周波フィルタ４０Ａの回路構成］
　図２４Ａは、実施例４に係る高周波フィルタ４０Ａの回路構成図である。同図に示された高周波フィルタ４０Ａは、フィルタ１１Ａ、１２Ａおよび１２Ａａと、移相器２１、２２、２１ａおよび２２ａと、スイッチＳＷ１、ＳＷ２、ＳＷ５、ＳＷ６、ＳＷ３５およびＳＷ３６と、入出力端子１１０および１２０と、を備える。本実施例に係る高周波フィルタ４０Ａは、実施例２に係る高周波フィルタ２０Ａと比較して、フィルタ１２Ａａ、スイッチＳＷ３５およびＳＷ３６を有する第２ａ回路（第３回路）が、ノードＸ１とノードＸ２との間に並列付加されている点が構成として異なる。以下、本実施例に係る高周波フィルタ４０Ａについて、実施例２に係る高周波フィルタ２０Ａと同じ点は説明を省略し、異なる点を中心に説明する。 [3.1 Circuit configuration of the high frequency filter 40A according to the fourth embodiment]
FIG. 24A is a circuit configuration diagram of a high frequency filter 40A according to a fourth embodiment. The high frequency filter 40A shown in the figure includes filters 11A, 12A and 12Aa, phase shifters 21, 22, 21a and 22a, switches SW1, SW2, SW5, SW6, SW35 and SW36, input / output terminals 110 and 110. And 120. As compared with the high frequency filter 20A according to the second embodiment, the high frequency filter 40A according to the present embodiment has a second circuit (third circuit) including the filter 12Aa, the switches SW35 and SW36 between the node X1 and the node X2. The point of being added in parallel is different as a configuration. Hereinafter, the high frequency filter 40A according to the present embodiment will not be described the same as the high frequency filter 20A according to the second embodiment, and differences will be mainly described.
　フィルタ１２Ａａは、移相器２１ａおよび２２ａとともに第２ａ回路を構成する第３フィルタである。 The filter 12Aa is a third filter that constitutes a circuit 2a together with the phase shifters 21a and 22a.
　スイッチＳＷ３５は、ノードＸ１と移相器２１ａとの間に接続されたスイッチ素子である。スイッチＳＷ３６は、ノードＸ２と移相器２２ａとの間に接続されたスイッチ素子である。 The switch SW35 is a switch element connected between the node X1 and the phase shifter 21a. The switch SW36 is a switch element connected between the node X2 and the phase shifter 22a.
　フィルタ１２Ａａ、移相器２１ａおよび２２ａ、ならびにスイッチＳＷ３５およびＳＷ３６で構成された第２ａ回路は、ノードＸ１およびＸ２に接続されている。 The 2a circuit composed of the filter 12Aa, the phase shifters 21a and 22a, and the switches SW35 and SW36 is connected to the nodes X1 and X2.
　移相器２１ａは、フィルタ１２Ａａの一方の端子に接続された第３移相器である。また、移相器２２ａは、フィルタ１２Ａａの他方の端子に接続された第４移相器である。つまり、スイッチＳＷ３５、移相器２１ａ、フィルタ１２Ａａ、移相器２２ａ、およびスイッチＳＷ３６は、この順で、ノードＸ１とノードＸ２との間に直列接続されている。 The phase shifter 21a is a third phase shifter connected to one terminal of the filter 12Aa. The phase shifter 22a is a fourth phase shifter connected to the other terminal of the filter 12Aa. That is, the switch SW35, the phase shifter 21a, the filter 12Aa, the phase shifter 22a, and the switch SW36 are connected in series between the node X1 and the node X2 in this order.
　フィルタ１２Ａａは、ノードＸ１とノードＸ２とを結ぶ経路上に配置された直列腕共振子ｓ３１およびｓ３２と、当該経路上の各ノードとグランドとの間に配置された並列腕共振子ｐ３１、ｐ３２およびｐ３３と、を備える。これにより、フィルタ１２Ａａは、ラダー型の帯域通過型フィルタを構成している。フィルタ１２Ａａは、第１帯域と周波数が異なる第１ａ帯域（第４帯域）の一部の周波数を含む帯域であって、第１ａ帯域より狭い帯域幅を有し、かつ、第２帯域の中心周波数より高域側に位置し、かつ、第３帯域と周波数が異なる第３ａ帯域（第５帯域）を通過帯域としている。 The filter 12Aa includes series arm resonators s31 and s32 disposed on a path connecting the node X1 and the node X2, and parallel arm resonators p31 and p32 disposed between each node on the path and the ground. and p33. Thus, the filter 12Aa constitutes a ladder type band pass filter. The filter 12Aa is a band including a part of the frequency of the first band a (the fourth band) different in frequency from the first band, and has a narrower bandwidth than the first band a and the center frequency of the second band A third band (fifth band), which is located on the higher side and has a frequency different from that of the third band, is used as a pass band.
　表３に、実施例４に係る高周波フィルタ４０Ａの回路パラメータを示す。 Table 3 shows circuit parameters of the high frequency filter 40A according to the fourth embodiment.
Figure JPOXMLDOC01-appb-T000003
Figure JPOXMLDOC01-appb-T000003
　以下、実施例４に係る高周波フィルタでは、第２帯域をＬＴＥのＢａｎｄ３受信通過帯域（１８０５－１８８０ＭＨｚ）とし、第３帯域をＬＴＥのＢａｎｄ３９通過帯域（１８８０－１９２０ＭＨｚ）とし、第３ａ帯域を１８８０－１９５０ＭＨｚ）とし、第１帯域を（Ｂａｎｄ３受信通過帯域＋Ｂａｎｄ３９通過帯域）（１８０５－１９２０ＭＨｚ）とし、第１ａ帯域を１８０５－１９５０ＭＨｚ）としている。 Hereinafter, in the high frequency filter according to the fourth embodiment, the second band is the Band 3 reception passband (1805-1880 MHz) of LTE, the third band is the Band 39 passband of LTE (1880-1920 MHz), and the third a band is 1880- The first band is (Band 3 reception pass band + Band 39 pass band) (1805-1920 MHz), and the first a band is 1805-1950 MHz.
　［３．２　実施例４に係る高周波フィルタ４０Ａのフィルタ特性］
　図２４Ｂは、実施例４に係る高周波フィルタ４０Ａのスイッチの切り替えによる通過特性の変化を表すグラフである。高周波フィルタ４０Ａは、スイッチＳＷ１、ＳＷ２、ＳＷ５、ＳＷ６、ＳＷ３５およびＳＷ３６の導通および非導通を適宜切り替えることにより、複数の帯域幅のフィルタへと可変できる周波数可変型の高周波フィルタとなる。なお、本実施例では、５つの帯域幅を切り替えることが可能となる。 [3.2 Filter Characteristics of High-Frequency Filter 40A According to Fourth Embodiment]
FIG. 24B is a graph showing a change in pass characteristics due to the switch of the high frequency filter 40A according to the fourth embodiment. The high frequency filter 40A is a variable frequency high frequency filter that can be changed to a filter of a plurality of bandwidths by appropriately switching on and off of the switches SW1, SW2, SW5, SW6, SW35 and SW36. In the present embodiment, five bandwidths can be switched.
　図２４Ｂの（ａ）に示すように、高周波フィルタ４０Ａは、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６を導通とし、スイッチＳＷ３５およびＳＷ３６を非導通とした場合、第１帯域を通過帯域とするフィルタとなる。また、図２４Ｂの（ｂ）に示すように、スイッチＳＷ１、ＳＷ２、ＳＷ３５およびＳＷ３６を導通とし、スイッチＳＷ５およびＳＷ６を非導通とした場合、第１ａ帯域を通過帯域とするフィルタとなる。また、図２４Ｂの（ｃ）に示すように、スイッチＳＷ１およびＳＷ２を導通とし、スイッチＳＷ５、ＳＷ６、ＳＷ３５およびＳＷ３６を非導通とした場合、第２帯域を通過帯域とするフィルタとなる。また、図２４Ｂの（ｄ）に示すように、スイッチＳＷ５およびＳＷ６を導通とし、スイッチＳＷ１、ＳＷ２、ＳＷ３５およびＳＷ３６を非導通とした場合、第３帯域を通過帯域とするフィルタとなる。また、図２４Ｂの（ｅ）に示すように、スイッチＳＷ３５およびＳＷ３６を導通とし、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６を非導通とした場合、第３ａ帯域を通過帯域とするフィルタとなる。 As shown in (a) of FIG. 24B, when the switches SW1, SW2, SW5, and SW6 are turned on and the switches SW35 and SW36 are turned off, the high frequency filter 40A becomes a filter that sets the first band as a pass band. . Also, as shown in (b) of FIG. 24B, when the switches SW1, SW2, SW35 and SW36 are made conductive and the switches SW5 and SW6 are made nonconductive, the filter becomes a pass band of the 1a band. When the switches SW1 and SW2 are turned on and the switches SW5, SW6, SW35 and SW36 are turned off as shown in (c) of FIG. 24B, the second band is a pass band. Further, as shown in (d) of FIG. 24B, when the switches SW5 and SW6 are made conductive and the switches SW1, SW2, SW35 and SW36 are made non-conductive, the filter becomes a pass band in the third band. Further, as shown in (e) of FIG. 24B, when the switches SW35 and SW36 are made conductive and the switches SW1, SW2, SW5 and SW6 are made non-conductive, the filter becomes a pass band of the 3a band.
　実施例４に係る高周波フィルタ４０Ａにおいて、フィルタ１２Ａの両方の入出力端子に移相器２１および２２が接続され、フィルタ１２Ａａの両方の入出力端子に移相器２１ａおよび２２ａが接続されることで、高周波フィルタ４０Ａの第１帯域および第１ａ帯域において、リップルが低減されることで挿入損失が低減する。 In the high frequency filter 40A according to the fourth embodiment, the phase shifters 21 and 22 are connected to both input and output terminals of the filter 12A, and the phase shifters 21a and 22a are connected to both input and output terminals of the filter 12Aa. In the first band and the 1a band of the high frequency filter 40A, the insertion loss is reduced by reducing the ripple.
　実施例４に係る高周波フィルタ４０Ａが、第１帯域および第１ａ帯域においてリップルおよび挿入損失を低減する要因について、以下に説明する。 The factor by which the high frequency filter 40A according to the fourth embodiment reduces the ripple and the insertion loss in the first band and the 1a band will be described below.
　図２５は、実施例４に係る高周波フィルタ４０Ａの第１帯域における広帯域化の要因を説明するグラフである。なお、高周波フィルタ４０Ａの通過帯域として第１帯域が選択されている場合には、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６は導通状態、かつ、スイッチＳＷ３５およびＳＷ３６は非導通状態である。同図の（ａ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の通過特性が示されている。同図の（ｂ）には、スイッチＳＷ１が導通状態である場合のノードＸ１から第１回路単体を見たインピーダンス特性およびスイッチＳＷ５が導通状態である場合のノードＸ１から第２回路単体を見たインピーダンス特性が示されている。同図の（ｃ）には、スイッチＳＷ５が導通状態である場合のノードＸ２から第１回路単体を見たインピーダンス特性およびスイッチＳＷ６が導通状態である場合のノードＸ２から第２回路単体を見たインピーダンス特性が示されている。同図の（ｄ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の位相特性が示されている。同図の（ｅ）には、第１回路と第２回路との位相差が示されている。 FIG. 25 is a graph for explaining the cause of wide band in the first band of the high-frequency filter 40A according to the fourth embodiment. When the first band is selected as the pass band of high-frequency filter 40A, switches SW1, SW2, SW5 and SW6 are in a conducting state, and switches SW35 and SW36 are in a non-conducting state. The pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2 are shown in FIG. The impedance characteristics of the first circuit viewed from the node X1 when the switch SW1 is in the conductive state and the second circuit alone from the node X1 when the switch SW5 is in the conductive state are illustrated in FIG. Impedance characteristics are shown. The impedance characteristics of the first circuit viewed from the node X2 when the switch SW5 is in the conductive state and the second circuit alone from the node X2 when the switch SW6 is in the conductive state are illustrated in FIG. Impedance characteristics are shown. The phase characteristic of the 1st circuit single-piece | unit and 2nd circuit single-piece | unit between node X1-X2 is shown by (d) of the figure. The phase difference between the first circuit and the second circuit is shown in (e) of FIG.
　実施例４に係る高周波フィルタ４０Ａでは、第１帯域が選択された場合、第２回路単体をノードＸ１から見たインピーダンス｜Ｚ１１｜が極大となる特異点のうち最大のインピーダンスとなる周波数ｆ１１が、第２帯域の低域端の周波数ｆＡから、フィルタ１２Ａの通過帯域低域側のカットオフ周波数ｆＣまでの周波数帯に含まれるように、移相器２１および２２の位相が調整されている。また、第２回路単体をノードＸ２から見たインピーダンス｜Ｚ２２｜が極大となる特異点のうち最大のインピーダンスとなる周波数ｆ２２が、第２帯域の低域端の周波数ｆＡから、フィルタ１２Ａの通過帯域低域側のカットオフ周波数ｆＣまでの周波数帯に含まれるように、移相器２１および２２の位相が調整されている。 In the high frequency filter 40A according to the fourth embodiment, when the first band is selected, the impedance viewed second circuit itself from the node X1 | Z11 | frequency f 11 which is the maximum of the impedance of the singular point as a maximum is The phases of the phase shifters 21 and 22 are adjusted so as to be included in the frequency band from the frequency f A at the low band end of the second band to the cutoff frequency f C on the low band side of the pass band of the filter 12A. There is. The impedance seen second circuit itself from the node X2 | Z22 | frequency f 22 which is the maximum of the impedance of the singular point as a maximum is the frequency f A of the low-frequency end of the second band, the filter 12A The phases of the phase shifters 21 and 22 are adjusted so as to be included in the frequency band up to the cutoff frequency f C on the lower side of the pass band.
　フィルタ１２Ａの減衰極の周波数ｆＡでは位相変化が大きく、フィルタ１１Ａとの位相差が大きくなるとともに、振幅差も大きくなる。そのため、移相器２１および２２の位相調整によって、フィルタ１２Ａの通過帯域低域側の減衰極の周波数ｆＡから低周波数側１０ｄＢカットオフ周波数ｆＣまでの周波数範囲に、第２回路のインピーダンス極大の特異点を合わせることにより、上記周波数範囲の高周波信号が、第２回路のフィルタ１２Ａに流れこむことを抑制できる。つまり、第２回路が減衰している周波数帯域で第２回路への信号の回り込みを抑制できる。これにより、高周波フィルタ４０Ａの第１帯域内のリップルを抑制し挿入損失を低減することが可能となる。 At the frequency f A of the attenuation pole of the filter 12A, the phase change is large, the phase difference with the filter 11A is large, and the amplitude difference is also large. Therefore, by the phase adjustment of the phase shifters 21 and 22, the impedance maximum of the second circuit is in the frequency range from the frequency f A of the attenuation pole on the low pass side of the filter 12A to the low 10 dB cut off frequency f C on the low frequency side. By matching the singular points of the above, it is possible to suppress the high frequency signal in the above frequency range from flowing into the filter 12A of the second circuit. That is, it is possible to suppress the wraparound of the signal to the second circuit in the frequency band in which the second circuit is attenuated. This makes it possible to suppress the ripple in the first band of the high frequency filter 40A and reduce the insertion loss.
　さらに、移相器２１および２２の位相調整により、第１回路と第２回路との振幅差が小さい周波数帯で同位相となるように調整することにより、第１帯域に含まれる周波数の高周波信号は、第１回路および第２回路を経由した後で、互いに打ち消し合うことがない。これにより、高周波フィルタ４０Ａの第１帯域に含まれる周波数において、第１帯域内のリップルを抑制し挿入損失を低減できる。 Further, the high frequency signal of the frequency included in the first band is adjusted by adjusting the phase difference of the first circuit and the second circuit to be in phase in a small frequency band by adjusting the phase of the phase shifters 21 and 22. Are not canceled each other after passing through the first circuit and the second circuit. Thereby, in the frequency included in the first band of the high frequency filter 40A, it is possible to suppress the ripple in the first band and reduce the insertion loss.
　本実施例では、図２５の（ｅ）に示すように、スイッチＳＷ５およびＳＷ６が導通状態である場合にノードＸ１およびＸ２から第２回路を単体で見た場合、第２帯域において、第１回路と第２回路とが同位相（０°または－３６０°）になっている周波数が少なくとも１箇所（同図の破線丸印：３箇所）あり、第３帯域において、第１回路と第２回路とが同位相（０°または－３６０°）になっている周波数が少なくとも１箇所（同図の破線丸印：１箇所）あってもよい。 In the present embodiment, as shown in (e) of FIG. 25, when the switches SW5 and SW6 are in the conductive state and the second circuit is viewed alone from the nodes X1 and X2, the first circuit is generated in the second band. And the second circuit have the same phase (0 ° or -360 °) in at least one point (broken line circle mark: three points in the same figure), and in the third band, the first circuit and the second circuit There may be at least one frequency (dotted line circle in the same figure: one place) in which the phase is the same phase (0 ° or -360 °).
　上記構成によれば、第１帯域、第２帯域、および第３帯域を切り替えることができる周波数可変型の高周波フィルタ４０Ａにおいて、第１帯域内のリップルを抑制することが可能となる。 According to the above configuration, it is possible to suppress the ripple in the first band in the variable-frequency high frequency filter 40A capable of switching the first band, the second band, and the third band.
　図２６は、実施例４に係る高周波フィルタ４０Ａの第１ａ帯域における広帯域化の要因を説明するグラフである。なお、高周波フィルタ４０Ａの通過帯域として第１ａ帯域が選択されている場合には、スイッチＳＷ１、ＳＷ２、ＳＷ３５およびＳＷ３６は導通状態、かつ、スイッチＳＷ５およびＳＷ６は非導通状態である。同図の（ａ）には、ノードＸ１－Ｘ２間の第１回路単体および第２ａ回路単体の通過特性が示されている。同図の（ｂ）には、ノードＸ１からスイッチＳＷ１が導通状態である場合の第１回路単体およびスイッチＳＷ３５が導通状態である場合の第２ａ回路単体を見たインピーダンス特性が示されている。同図の（ｃ）には、ノードＸ２からスイッチＳＷ２が導通状態である場合の第１回路単体およびスイッチＳＷ３６が導通状態である場合の第２ａ回路単体を見たインピーダンス特性が示されている。同図の（ｄ）には、ノードＸ１－Ｘ２間の第１回路単体および第２ａ回路単体の位相特性が示されている。同図の（ｅ）には、第１回路と第２ａ回路との位相差が示されている。 FIG. 26 is a graph for explaining the factor of broadening the band in the 1a band of the high frequency filter 40A according to the fourth embodiment. When band 1a is selected as the pass band of high-frequency filter 40A, switches SW1, SW2, SW35 and SW36 are in a conducting state, and switches SW5 and SW6 are in a non-conducting state. The pass characteristic of the 1st circuit single-piece | unit and 2a circuit single-piece | unit between node X1-X2 is shown by (a) of the figure. The impedance characteristics are shown in (b) of the figure when the first circuit single-piece when the switch SW1 is in the conductive state from the node X1 and the 2a-th single circuit when the switch SW35 is in the conductive state. The impedance characteristics are shown in (c) of the figure when the first circuit single-piece when the switch SW2 is in a conductive state from the node X2 and the 2a-th single circuit when the switch SW36 is in a conductive state. The phase characteristic of the 1st circuit single-piece | unit and 2a circuit single-piece | unit between node X1-X2 is shown by (d) of the figure. The phase difference between the first circuit and the 2a circuit is shown in (e) of FIG.
　実施例４に係る高周波フィルタ４０Ａでは、第１ａ帯域が選択された場合、第２ａ回路単体をノードＸ１から見たインピーダンス｜Ｚ１１｜が極大となる特異点のうち最大のインピーダンスとなる周波数ｆ１１が、第２帯域の低域端の周波数ｆＡから、フィルタ１２Ａの通過帯域低域側のカットオフ周波数ｆＣまでの周波数帯に含まれるように、移相器２１ａおよび２２ａの位相が調整されている。また、第２ａ回路単体をノードＸ２から見たインピーダンス｜Ｚ２２｜が極大となる特異点のうち最大のインピーダンスとなる周波数ｆ２２が、第２帯域の低域端の周波数ｆＡから、フィルタ１２Ａの通過帯域低域側のカットオフ周波数ｆＣまでの周波数帯に含まれるように、移相器２１ａおよび２２ａの位相が調整されている。 In the high frequency filter 40A according to the fourth embodiment, when the first 1a band is selected, the impedance viewed first 2a circuit itself from the node X1 | Z11 | frequency f 11 which is the maximum of the impedance of the singular point as a maximum is The phases of the phase shifters 21a and 22a are adjusted so as to be included in the frequency band from the frequency f A at the low band end of the second band to the cutoff frequency f C at the low band side of the pass band of the filter 12A. There is. The impedance viewed first 2a circuit itself from the node X2 | Z22 | frequency f 22 which is the maximum of the impedance of the singular point as a maximum is the frequency f A of the low-frequency end of the second band, the filter 12A The phases of the phase shifters 21a and 22a are adjusted so as to be included in the frequency band up to the cutoff frequency f C on the lower side of the pass band.
　フィルタ１２Ａａの減衰極の周波数ｆＡでは位相変化が大きく、フィルタ１１Ａとの位相差が大きくなるとともに、振幅差も大きくなる。そのため、移相器２１ａおよび２２ａの位相調整によって、フィルタ１２Ａａの通過帯域低域側の減衰極の周波数ｆＡから低周波数側１０ｄＢカットオフ周波数ｆＣまでの周波数範囲に、第２ａ回路のインピーダンス極大の特異点を合わせることにより、上記周波数範囲の高周波信号が、第２ａ回路のフィルタ１２Ａａに流れこむことを抑制できる。つまり、第２ａ回路が減衰している周波数帯域で第２ａ回路への信号の回り込みを抑制できる。これにより、高周波フィルタ４０Ａの第１ａ帯域内のリップルを抑制し挿入損失を低減することが可能となる。 At the frequency f A of the attenuation pole of the filter 12Aa, the phase change is large, the phase difference with the filter 11A becomes large, and the amplitude difference also becomes large. Therefore, the phase adjustment of the phase shifter 21a and 22a, the frequency range from the frequency f A of the attenuation pole of the passband low frequency side of the filter 12Aa to low frequency side 10dB cut-off frequency f C, the impedance maximum of the 2a circuit By matching the singular points of the above, it is possible to suppress the high frequency signal of the above frequency range from flowing into the filter 12Aa of the circuit 2a. That is, it is possible to suppress the wraparound of the signal to the circuit 2a in the frequency band in which the circuit 2a is attenuated. As a result, it is possible to suppress the ripple in the 1a band of the high frequency filter 40A and reduce the insertion loss.
　さらに、移相器２１ａおよび２２ａの位相調整により、第１回路と第２ａ回路との振幅差が小さい周波数帯で同位相となるように調整することにより、第１ａ帯域に含まれる周波数の高周波信号は、第１回路および第２ａ回路を経由した後で、互いに打ち消し合うことがない。これにより、高周波フィルタ４０Ａの第１ａ帯域に含まれる周波数において、第１ａ帯域内のリップルを抑制し挿入損失を低減できる。 Further, by adjusting the phase difference of the first circuit and the 2a circuit to be in phase in a small frequency band by adjusting the phase of the phase shifters 21a and 22a, a high frequency signal of the frequency included in the 1a band Are not canceled after passing through the first circuit and the 2a circuit. Thereby, at the frequency included in the 1a band of the high frequency filter 40A, the ripple in the 1a band can be suppressed and the insertion loss can be reduced.
　本実施例では、図２６の（ｅ）に示すように、スイッチＳＷ３５およびＳＷ３６が導通状態である場合にノードＸ１およびＸ２から第２ａ回路を単体で見た場合、第２帯域において、第１回路と第２ａ回路とが同位相（０°または－３６０°）になっている周波数が少なくとも１箇所（同図の破線丸印：３箇所）あり、第３帯域において、第１回路と第２ａ回路とが同位相（０°または－３６０°）になっている周波数が少なくとも１箇所（同図の破線丸印：１箇所）あってもよい。 In this embodiment, as shown in (e) of FIG. 26, when switches SW 35 and SW 36 are in a conductive state, and the second a circuit is viewed alone from nodes X 1 and X 2, the first circuit is generated in the second band. And the circuit 2a are in phase (0 ° or -360 °) in at least one place (circled with broken lines in the same figure: 3 places), and in the third band, the first circuit and the 2a circuit There may be at least one frequency (dotted line circle in the same figure: one place) in which the phase is the same phase (0 ° or -360 °).
　上記構成によれば、第１ａ帯域、第２帯域、および第３帯域を切り替えることができる周波数可変型の高周波フィルタ４０Ａにおいて、第１ａ帯域内のリップルを抑制することが可能となる。 According to the above configuration, it is possible to suppress the ripple in the 1a band in the variable-frequency high frequency filter 40A capable of switching the 1a band, the 2nd band, and the 3rd band.
　本実施例に係る高周波フィルタ４０Ａによれば、通過帯域の切り替えのバリエーションを増やしつつ、第１帯域内および第１ａ帯域内のリップルを低減させて挿入損失を低減させることが可能となる。 According to the high frequency filter 40A according to the present embodiment, it is possible to reduce the insertion loss by reducing the ripples in the first band and the 1a band while increasing the variation of switching of the pass band.
　（実施の形態４）
　本実施の形態では、縦結合型共振器を有するフィルタで構成された高周波フィルタについて説明する。 Embodiment 4
In the present embodiment, a high frequency filter configured of a filter having a longitudinally coupled resonator will be described.
　［４．１　実施例５に係る高周波フィルタ５０Ａの回路構成］
　図２７Ａは、実施例５に係る高周波フィルタ５０Ａの回路構成図である。同図に示された高周波フィルタ５０Ａは、フィルタ１１Ｂおよび１２Ａと、移相器２１Ｄおよび２２Ｄと、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６と、入出力端子１１０および１２０と、を備える。本実施例に係る高周波フィルタ５０Ａは、実施例２に係る高周波フィルタ２０Ａと比較して、第１回路ならびに移相器の回路構成が異なる。以下、本実施例に係る高周波フィルタ５０Ａについて、実施例２に係る高周波フィルタ２０Ａと同じ点は説明を省略し、異なる点を中心に説明する。 [4.1 Circuit configuration of the high frequency filter 50A according to the fifth embodiment]
FIG. 27A is a circuit configuration diagram of a high frequency filter 50A according to a fifth embodiment. The high frequency filter 50A shown in the figure includes filters 11B and 12A, phase shifters 21D and 22D, switches SW1, SW2, SW5 and SW6, and input / output terminals 110 and 120. The high frequency filter 50A according to the present embodiment differs from the high frequency filter 20A according to the second embodiment in the circuit configurations of the first circuit and the phase shifter. Hereinafter, the high frequency filter 50A according to the present embodiment will not be described the same as the high frequency filter 20A according to the second embodiment, and differences will be mainly described.
　フィルタ１１Ｂは、ノードＸ１とノードＸ２とを結ぶ第１経路上に配置された直列腕共振子ｓ１１、ｓ１２およびｓ１３と、当該第１経路上の各ノードとグランドとの間に配置された並列腕共振子ｐ１１およびｐ１２と、直列腕共振子ｓ１２およびｓ１３に接続された縦結合型共振器１５Ａと、を備える。これにより、フィルタ１１Ｂは、帯域通過型フィルタを構成している。縦結合型共振器１５Ａは、５つのＩＤＴ電極と当該５つのＩＤＴ電極の並び方向の両端に配置された反射器とで構成されている。なお、縦結合型共振器のＩＤＴ電極の数は２つ以上であればよい。 The filter 11B includes series arm resonators s11, s12 and s13 disposed on a first path connecting the node X1 and the node X2, and parallel arms disposed between the respective nodes on the first path and the ground. Resonators p11 and p12 and a longitudinally coupled resonator 15A connected to series arm resonators s12 and s13 are provided. Thus, the filter 11B constitutes a band pass filter. The longitudinally coupled resonator 15A is configured of five IDT electrodes and reflectors disposed at both ends in the arrangement direction of the five IDT electrodes. The number of IDT electrodes of the longitudinally coupled resonator may be two or more.
　移相器２１Ｄは、フィルタ１２Ａの一方の端子に接続された第１移相器である。また、移相器２２Ｄは、フィルタ１２Ａの他方の端子に接続された第２移相器である。つまり、移相器２１Ｄ、フィルタ１２Ａ、および移相器２２Ｄは、この順で、ノードＸ１とノードＸ２との間に直列接続されている。 The phase shifter 21D is a first phase shifter connected to one terminal of the filter 12A. The phase shifter 22D is a second phase shifter connected to the other terminal of the filter 12A. That is, the phase shifter 21D, the filter 12A, and the phase shifter 22D are connected in series between the node X1 and the node X2 in this order.
　移相器２１Ｄは、ノードＸ１とフィルタ１２Ａの一方の端子とを結ぶ直列腕経路に配置された１つのキャパシタＣｓ２１、および、当該直列腕経路上のノードおよびグランドに接続された１つのインダクタＬｐ２１で構成されている。移相器２２Ｄは、ノードＸ２とフィルタ１２Ａの他方の端子とを結ぶ直列腕経路に配置された１つのキャパシタＣｓ２２、および、当該直列腕経路上のノードおよびグランドに接続された１つのインダクタＬｐ２２で構成されている。 The phase shifter 21D includes one capacitor Cs21 arranged in a series arm path connecting the node X1 and one terminal of the filter 12A, and one inductor Lp21 connected to the node on the series arm path and the ground. It is configured. The phase shifter 22D includes one capacitor Cs22 arranged in a series arm path connecting the node X2 and the other terminal of the filter 12A, and one inductor Lp22 connected to the node on the series arm path and the ground. It is configured.
　表４に、実施例５に係る高周波フィルタ５０Ａの回路パラメータを示す。 Table 4 shows circuit parameters of the high frequency filter 50A according to the fifth embodiment.
Figure JPOXMLDOC01-appb-T000004
Figure JPOXMLDOC01-appb-T000004
　本実施例では、フィルタ１１Ｂは、縦結合型共振器１５Ａを有しているため、フィルタ１１Ｂは、減衰特性に有利なフィルタ特性となっている。このため、移相器２１Ｄおよび２２Ｄに、高い減衰特性は要求されない。この観点から、移相器２１Ｄおよび２２Ｄは、それぞれ、１つのキャパシタおよび１つのインダクタで構成することが可能となる。 In the present embodiment, since the filter 11B includes the longitudinally coupled resonator 15A, the filter 11B has filter characteristics that are advantageous for attenuation characteristics. Therefore, high attenuation characteristics are not required for the phase shifters 21D and 22D. From this point of view, each of the phase shifters 21D and 22D can be configured with one capacitor and one inductor.
　なお、移相器２１Ｄおよび２２Ｄの回路構成は、上記構成に限定されない。キャパシタおよびインダクタは、それぞれ複数配置されていてもよく、Ｔ型およびπ型などの接続構成も任意である。 The circuit configuration of the phase shifters 21D and 22D is not limited to the above configuration. A plurality of capacitors and inductors may be disposed, and connection configurations such as T-type and π-type are also arbitrary.
　［４．２　実施例５に係る高周波フィルタ５０Ａのフィルタ特性］
　図２７Ｂは、実施例５に係る高周波フィルタ５０Ａのスイッチの切り替えによる通過特性の変化を表すグラフである。図２７Ｂの（ａ）に示すように、高周波フィルタ５０Ａは、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６の全てを導通とした場合、第１帯域（Ｂａｎｄ３受信通過帯域＋Ｂａｎｄ３９通過帯域）を通過帯域とするフィルタとなる。また、図２７Ｂの（ｂ）に示すように、スイッチＳＷ１およびＳＷ２を導通とし、ＳＷ５およびＳＷ６を非導通とした場合、第２帯域（Ｂａｎｄ３受信通過帯域）を通過帯域とするフィルタとなる。また、図２７Ｂの（ｃ）に示すように、スイッチＳＷ１およびＳＷ２を非導通とし、ＳＷ５およびＳＷ６を導通とした場合、第３帯域（Ｂａｎｄ３９通過帯域）を通過帯域とするフィルタとなる。 [4.2 Filter Characteristics of High-Frequency Filter 50A According to Fifth Embodiment]
FIG. 27B is a graph showing the change of the passage characteristic due to the switch of the high frequency filter 50A according to the fifth embodiment. As shown in (a) of FIG. 27B, when all the switches SW1, SW2, SW5, and SW6 are turned on, the high frequency filter 50A uses the first band (Band 3 reception pass band + Band 39 pass band) as a pass band. It becomes. Further, as shown in (b) of FIG. 27B, when the switches SW1 and SW2 are made conductive and the switches SW5 and SW6 are not made conductive, the filter becomes a pass band that is the second band (Band 3 reception pass band). Further, as shown in (c) of FIG. 27B, when the switches SW1 and SW2 are turned off and the switches SW5 and SW6 are turned on, the filter becomes a pass band that is the third band (Band 39 pass band).
　なお、本実施例では、第１回路側にスイッチＳＷ１およびＳＷ２が配置され、第２回路側にスイッチＳＷ５およびＳＷ６が配置された回路構成としたが、第１回路側のみにスイッチが接続された構成でもよいし、第２回路側のみにスイッチが接続された構成でもよい。第１回路側のみにスイッチが接続された構成の場合、高周波フィルタは第１帯域と第３帯域とを切り替えるフィルタとなる。また、第２回路側のみにスイッチが接続された構成の場合、高周波フィルタは第１帯域と第２帯域とを切り替えるフィルタとなる。 In this embodiment, the switches SW1 and SW2 are disposed on the first circuit side, and the switches SW5 and SW6 are disposed on the second circuit side. However, the switches are connected only to the first circuit side. It may be a configuration or a configuration in which a switch is connected only to the second circuit side. When the switch is connected only to the first circuit side, the high frequency filter serves as a filter that switches between the first band and the third band. Further, in the case where the switch is connected only to the second circuit side, the high frequency filter serves as a filter that switches between the first band and the second band.
　実施例５に係る高周波フィルタ５０Ａにおいても、実施例２に係る高周波フィルタ２０Ａと同様に、フィルタ１２Ａの両方の入出力端子に移相器２１Ｄおよび２２Ｄが接続されることで、高周波フィルタ５０Ａの第１帯域において、リップルが低減されることで挿入損失が低減する。 Also in the high frequency filter 50A according to the fifth embodiment, as in the high frequency filter 20A according to the second embodiment, the phase shifters 21D and 22D are connected to both input and output terminals of the filter 12A, thereby the second high frequency filter 50A In one band, the reduction of the ripple reduces the insertion loss.
　実施例５に係る高周波フィルタ５０Ａの第１帯域におけるリップルおよび挿入損失を低減する要因について、以下に説明する。 The factors for reducing the ripple and insertion loss in the first band of the high frequency filter 50A according to the fifth embodiment will be described below.
　図２８は、実施例５に係る高周波フィルタ５０Ａの広帯域化の要因を説明するグラフである。同図の（ａ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の通過特性が示されている。同図の（ｂ）には、スイッチＳＷ１が導通状態である場合のノードＸ１から第１回路単体を見たインピーダンス特性およびスイッチＳＷ５が導通状態である場合のノードＸ１から第２回路単体を見たインピーダンス特性が示されている。同図の（ｃ）には、スイッチＳＷ２が導通状態である場合のノードＸ２から第１回路単体を見たインピーダンス特性およびスイッチＳＷ６が導通状態である場合のノードＸ２から第２回路単体を見たインピーダンス特性が示されている。同図の（ｄ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の位相特性が示されている。同図の（ｅ）には、第１回路と第２回路との位相差が示されている。 FIG. 28 is a graph for explaining the factor of increasing the bandwidth of the high frequency filter 50A according to the fifth embodiment. The pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2 are shown in FIG. The impedance characteristics of the first circuit viewed from the node X1 when the switch SW1 is in the conductive state and the second circuit alone from the node X1 when the switch SW5 is in the conductive state are illustrated in FIG. Impedance characteristics are shown. The impedance characteristics of the first circuit viewed from the node X2 when the switch SW2 is in the conductive state and the second circuit alone from the node X2 when the switch SW6 is in the conductive state are illustrated in FIG. Impedance characteristics are shown. The phase characteristic of the 1st circuit single-piece | unit and 2nd circuit single-piece | unit between node X1-X2 is shown by (d) of the figure. The phase difference between the first circuit and the second circuit is shown in (e) of FIG.
　実施例５に係る高周波フィルタ５０Ａでは、第２回路単体をノードＸ１から見たインピーダンス｜Ｚ１１｜が極大となる特異点のうち最大のインピーダンスとなる周波数ｆ１１が、第２帯域の低域端の周波数ｆＡから、フィルタ１２Ａの通過帯域低域側のカットオフ周波数ｆＣまでの周波数帯に含まれるように、移相器２１Ｄおよび２２Ｄの位相が調整されている。また、第２回路単体をノードＸ２から見たインピーダンス｜Ｚ２２｜が極大となる特異点のうち最大のインピーダンスとなる周波数ｆ２２が、第２帯域の低域端の周波数ｆＡから、フィルタ１２Ａの通過帯域低域側のカットオフ周波数ｆＣまでの周波数帯に含まれるように、移相器２１Ｄおよび２２Ｄの位相が調整されている。 In the high frequency filter 50A according to Embodiment 5, the impedance viewed second circuit itself from the node X1 | Z11 | frequency f 11 which is the maximum of the impedance of the singular point as a maximum is the low frequency end of the second band The phases of the phase shifters 21D and 22D are adjusted so as to be included in the frequency band from the frequency f A to the cut-off frequency f C on the low pass side of the filter 12A. The impedance seen second circuit itself from the node X2 | Z22 | frequency f 22 which is the maximum of the impedance of the singular point as a maximum is the frequency f A of the low-frequency end of the second band, the filter 12A The phases of the phase shifters 21D and 22D are adjusted so as to be included in the frequency band up to the cutoff frequency f C on the lower side of the pass band.
　フィルタ１２Ａの減衰極の周波数ｆＡでは位相変化が大きく、フィルタ１１Ｂとの位相差が大きくなるとともに、振幅差も大きくなる。そのため、移相器２１Ｄおよび２２Ｄの位相調整によって、フィルタ１２Ａの通過帯域低域側の減衰極の周波数ｆＡから低周波数側１０ｄＢカットオフ周波数ｆＣまでの周波数範囲に、第２回路のインピーダンス極大の特異点のうち最大のインピーダンスとなる周波数ｆ１１およびｆ２２を合わせることにより、上記周波数範囲の高周波信号が、第２回路のフィルタ１２Ａに流れこむことを抑制できる。つまり、第２回路が減衰している周波数帯域で第２回路への信号の回り込みを抑制できる。これにより、高周波フィルタ５０Ａの第１帯域内のリップルを抑制し挿入損失を低減することが可能となる。 At the frequency f A of the attenuation pole of the filter 12A, the phase change is large, the phase difference with the filter 11B is large, and the amplitude difference is also large. Therefore, the phase adjustment of the phase shifter 21D and 22D, in the frequency range from the frequency f A of the attenuation pole of the passband low frequency side of the filter 12A to the low-frequency side 10dB cut-off frequency f C, the impedance maximum of the second circuit by matching the frequency f 11 and f 22 with the maximum of the impedance of the singular point, the high-frequency signal of the frequency range can be prevented from flowing into the filter 12A of the second circuit. That is, it is possible to suppress the wraparound of the signal to the second circuit in the frequency band in which the second circuit is attenuated. This makes it possible to suppress the ripple in the first band of the high frequency filter 50A and reduce the insertion loss.
　さらに、移相器２１Ｄおよび２２Ｄの位相調整により、第１回路と第２回路との振幅差が小さい周波数帯で同位相となるように調整することにより、第１帯域に含まれる周波数の高周波信号は、第１回路および第２回路を経由した後で、互いに打ち消し合うことがない。これにより、高周波フィルタ５０Ａの第１帯域に含まれる周波数において、第１帯域内のリップルを抑制し挿入損失を低減できる。 Furthermore, the high frequency signal of the frequency included in the first band is adjusted by adjusting the phase difference of the first circuit and the second circuit to be in phase in a small frequency band by adjusting the phase of the phase shifters 21D and 22D. Are not canceled each other after passing through the first circuit and the second circuit. Thereby, at the frequency included in the first band of the high frequency filter 50A, the ripple in the first band can be suppressed and the insertion loss can be reduced.
　以上のように、高周波フィルタ５０Ａが、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６の導通状態により第１帯域を通過帯域とする場合、第１帯域の挿入損失を低減することが可能となる。 As described above, when the high frequency filter 50A sets the first band as the pass band due to the conduction state of the switches SW1, SW2, SW5, and SW6, it is possible to reduce the insertion loss of the first band.
　本実施例では、図２８の（ｅ）に示すように、スイッチＳＷ５およびＳＷ６が導通状態である場合にノードＸ１およびＸ２から第２回路を単体で見た場合、第２帯域において、第１回路と第２回路とが同位相（０°または－３６０°）になっている周波数が少なくとも１箇所（同図の破線丸印：２箇所）あり、第３帯域において、第１回路と第２回路とが同位相（０°または－３６０°）になっている周波数が少なくとも１箇所（同図の破線丸印：１箇所）あってもよい。 In the present embodiment, as shown in (e) of FIG. 28, when the switches SW5 and SW6 are in the conductive state, and the second circuit is viewed alone from the nodes X1 and X2, the first circuit is generated in the second band. And the second circuit have the same phase (0 ° or -360 °) in at least one point (two dotted circle marks in the figure), and in the third band, the first circuit and the second circuit There may be at least one frequency (dotted line circle in the same figure: one place) in which the phase is the same phase (0 ° or -360 °).
　上記構成によれば、第１帯域、第２帯域、および第３帯域を切り替えることができる周波数可変型の高周波フィルタ５０Ａにおいて、第１帯域内のリップルを抑制することが可能となる。また、フィルタ１１Ｂが縦結合型共振器１５Ａで構成されているので、高周波フィルタ５０Ａの通過帯域低域側において、急峻性が向上するとともに減衰量を向上できる。 According to the above configuration, it is possible to suppress the ripple in the first band in the variable-frequency high frequency filter 50A capable of switching the first band, the second band, and the third band. Further, since the filter 11B is configured by the longitudinally coupled resonator 15A, the sharpness can be improved and the amount of attenuation can be improved on the low frequency band side of the high frequency filter 50A.
　（実施の形態５）
　本実施の形態では、スイッチ素子を有するインピーダンス回路が付加された並列腕回路を有する第１フィルタおよび第２フィルタで構成された高周波フィルタについて説明する。 Fifth Embodiment
In the present embodiment, a high frequency filter configured of a first filter and a second filter having a parallel arm circuit to which an impedance circuit having a switch element is added will be described.
　［５．１　実施例６に係る高周波フィルタ６０Ａの回路構成］
　図２９Ａは、実施例６に係る高周波フィルタ６０Ａの回路構成図である。同図に示された高周波フィルタ６０Ａは、フィルタ１１Ｃおよび１２Ｂと、移相器２１Ｃおよび２２Ｃと、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６と、入出力端子１１０および１２０と、を備える。本実施例に係る高周波フィルタ６０Ａは、実施の形態２の変形例２に係る高周波フィルタ２０Ｃと比較して、第１フィルタおよび第２フィルタの回路構成が異なる。以下、本実施例に係る高周波フィルタ６０Ａについて、実施の形態２の変形例２に係る高周波フィルタ２０Ｃと同じ点は説明を省略し、異なる点を中心に説明する。 [5.1 Circuit Configuration of High-Frequency Filter 60A According to Sixth Embodiment]
FIG. 29A is a circuit configuration diagram of a high frequency filter 60A according to a sixth embodiment. The high frequency filter 60A shown in the figure includes filters 11C and 12B, phase shifters 21C and 22C, switches SW1, SW2, SW5 and SW6, and input / output terminals 110 and 120. The high frequency filter 60A according to the present embodiment is different from the high frequency filter 20C according to the second modification of the second embodiment in the circuit configuration of the first filter and the second filter. Hereinafter, the high frequency filter 60A according to the present embodiment will not be described the same as the high frequency filter 20C according to the second modification of the second embodiment, and differences will be mainly described.
　フィルタ１１Ｃは、３つの直列腕回路（直列腕共振子ｓ１１、ｓ１２およびｓ１３）と、並列腕回路１０１、１０２、１０３および１０４と、を備える。 The filter 11C includes three series arm circuits (series arm resonators s11, s12 and s13) and parallel arm circuits 101, 102, 103 and 104.
　直列腕共振子ｓ１１～ｓ１３は、ノードＸ１およびＸ２を結ぶ第１経路上に設けられている。 The series arm resonators s11 to s13 are provided on a first path connecting the nodes X1 and X2.
　並列腕回路１０１～１０４のそれぞれは、ノードＸ１およびＸ２を結ぶ第１経路上の各ノードおよびグランドに接続された並列腕回路である。 Each of parallel arm circuits 101 to 104 is a parallel arm circuit connected to each node on the first path connecting nodes X1 and X2 and to the ground.
　並列腕回路１０１は、並列腕共振子ｐ１１と、並列腕共振子ｐ１１に直列接続されたインピーダンス回路１１１とを有している。インピーダンス回路１１１は、キャパシタＣｐ１１と、スイッチＳＷ１１とを有する。スイッチＳＷ１１は、キャパシタＣｐ１１と並列接続された第９スイッチ素子である。インピーダンス回路１１１がキャパシタＣｐ１１とスイッチＳＷ１１との並列回路で構成されているので、スイッチＳＷ１１が導通から非導通へと切り替わることにより、並列腕回路１０１の共振周波数が高域側へと切り替えられる。 The parallel arm circuit 101 includes a parallel arm resonator p11 and an impedance circuit 111 connected in series to the parallel arm resonator p11. The impedance circuit 111 has a capacitor Cp11 and a switch SW11. The switch SW11 is a ninth switch element connected in parallel to the capacitor Cp11. Since the impedance circuit 111 is configured by a parallel circuit of the capacitor Cp11 and the switch SW11, the resonance frequency of the parallel arm circuit 101 is switched to the high frequency side by switching the switch SW11 from conduction to non-conduction.
　並列腕回路１０２は、並列腕共振子ｐ１２と、並列腕共振子ｐ１２に直列接続されたインピーダンス回路１１２とを有している。インピーダンス回路１１２は、キャパシタＣｐ１２と、スイッチＳＷ１２とを有する。スイッチＳＷ１２は、キャパシタＣｐ１２と並列接続された第９スイッチ素子である。インピーダンス回路１１２がキャパシタＣｐ１２とスイッチＳＷ１２との並列回路で構成されているので、スイッチＳＷ１２が導通から非導通へと切り替わることにより、並列腕回路１０２の共振周波数が高域側へと切り替えられる。 The parallel arm circuit 102 includes a parallel arm resonator p12 and an impedance circuit 112 connected in series to the parallel arm resonator p12. The impedance circuit 112 includes a capacitor Cp12 and a switch SW12. The switch SW12 is a ninth switch element connected in parallel to the capacitor Cp12. Since the impedance circuit 112 is constituted by a parallel circuit of the capacitor Cp12 and the switch SW12, the resonance frequency of the parallel arm circuit 102 is switched to the high frequency side by switching the switch SW12 from conduction to non-conduction.
　並列腕回路１０４は、並列腕共振子ｐ１４と、並列腕共振子ｐ１４に直列接続されたインピーダンス回路１１４とを有している。インピーダンス回路１１４は、キャパシタＣｐ１４と、スイッチＳＷ１４とを有する。スイッチＳＷ１４は、キャパシタＣｐ１４と並列接続された第９スイッチ素子である。インピーダンス回路１１４がキャパシタＣｐ１４とスイッチＳＷ１４との並列回路で構成されているので、スイッチＳＷ１４が導通から非導通へと切り替わることにより、並列腕回路１０４の共振周波数が高域側へと切り替えられる。 The parallel arm circuit 104 includes a parallel arm resonator p14 and an impedance circuit 114 connected in series to the parallel arm resonator p14. The impedance circuit 114 has a capacitor Cp14 and a switch SW14. The switch SW14 is a ninth switch element connected in parallel to the capacitor Cp14. Since the impedance circuit 114 is formed of a parallel circuit of the capacitor Cp14 and the switch SW14, the resonance frequency of the parallel arm circuit 104 is switched to the high frequency side by switching the switch SW14 from conduction to non-conduction.
　並列腕回路１０３は、並列腕共振子ｐ１３と、並列腕共振子ｐ１３に直列接続されたインピーダンス回路１１３とを有している。インピーダンス回路１１３は、インダクタＬｐ１３と、スイッチＳＷ１３とを有する。スイッチＳＷ１３は、インダクタＬｐ１３と並列接続された第９スイッチ素子である。インピーダンス回路１１３がインダクタＬｐ１３とスイッチＳＷ１３との並列回路で構成されているので、スイッチＳＷ１３が導通から非導通へと切り替わることにより、並列腕回路１０３の共振周波数が低域側へと切り替えられる。 The parallel arm circuit 103 has a parallel arm resonator p13 and an impedance circuit 113 connected in series to the parallel arm resonator p13. The impedance circuit 113 includes an inductor Lp13 and a switch SW13. The switch SW13 is a ninth switch element connected in parallel to the inductor Lp13. Since the impedance circuit 113 is constituted by a parallel circuit of the inductor Lp13 and the switch SW13, the resonance frequency of the parallel arm circuit 103 is switched to the low frequency side by switching the switch SW13 from conduction to non-conduction.
　並列腕回路の共振周波数は、ラダー型フィルタにおける通過帯域低域側の減衰極を規定することから、フィルタ１１Ｃは、スイッチＳＷ１１、ＳＷ１２、ＳＷ１３、ＳＷ１４の導通および非導通の切り替えにより、通過帯域低域側の急峻度および通過帯域低域端の周波数が可変する。 Since the resonance frequency of the parallel arm circuit defines the attenuation pole on the low pass band side of the ladder type filter, the filter 11C switches the conduction and non-conduction of the switches SW11, SW12, SW13, and SW14 to lower the passband. The steepness on the band side and the frequency at the low band edge of the pass band are variable.
　フィルタ１２Ｂは、２つの直列腕回路（直列腕共振子ｓ２１およびｓ２２）と、並列腕回路２０１、２０２、および２０３と、を備える。 The filter 12B includes two series arm circuits (series arm resonators s21 and s22) and parallel arm circuits 201, 202 and 203.
　直列腕共振子ｓ２１およびｓ２２は、ノードＸ１およびＸ２を結ぶ第２経路上に設けられている。 The series arm resonators s21 and s22 are provided on a second path connecting the nodes X1 and X2.
　並列腕回路２０１～２０３のそれぞれは、ノードＸ１およびＸ２を結ぶ第２経路上の各ノードおよびグランドに接続された並列腕回路である。 Each of the parallel arm circuits 201 to 203 is a parallel arm circuit connected to each node on the second path connecting the nodes X1 and X2 and the ground.
　並列腕回路２０１は、並列腕共振子ｐ２１と、並列腕共振子ｐ２１に直列接続されたインピーダンス回路２１１とを有している。インピーダンス回路２１１は、キャパシタＣｐ２１と、スイッチＳＷ２１とを有する。スイッチＳＷ２１は、キャパシタＣｐ２１と並列接続された第９スイッチ素子である。インピーダンス回路２１１がキャパシタＣｐ２１とスイッチＳＷ２１との並列回路で構成されているので、スイッチＳＷ２１が導通から非導通へと切り替わることにより、並列腕回路２０１の共振周波数が高域側へと切り替えられる。 The parallel arm circuit 201 includes a parallel arm resonator p21 and an impedance circuit 211 connected in series to the parallel arm resonator p21. The impedance circuit 211 includes a capacitor Cp21 and a switch SW21. The switch SW21 is a ninth switch element connected in parallel to the capacitor Cp21. Since the impedance circuit 211 is configured by a parallel circuit of the capacitor Cp21 and the switch SW21, the resonance frequency of the parallel arm circuit 201 is switched to the high frequency side by switching the switch SW21 from conduction to non-conduction.
　並列腕回路２０２は、並列腕共振子ｐ２２と、並列腕共振子ｐ２２に直列接続されたインピーダンス回路２１２とを有している。インピーダンス回路２１２は、キャパシタＣｐ２２と、スイッチＳＷ２２とを有する。スイッチＳＷ２２は、キャパシタＣｐ２２と並列接続された第９スイッチ素子である。インピーダンス回路２１２がキャパシタＣｐ２２とスイッチＳＷ２２との並列回路で構成されているので、スイッチＳＷ２２が導通から非導通へと切り替わることにより、並列腕回路２０２の共振周波数が高域側へと切り替えられる。 The parallel arm circuit 202 includes a parallel arm resonator p22 and an impedance circuit 212 connected in series to the parallel arm resonator p22. The impedance circuit 212 includes a capacitor Cp22 and a switch SW22. The switch SW22 is a ninth switch element connected in parallel to the capacitor Cp22. Since the impedance circuit 212 is configured by a parallel circuit of the capacitor Cp22 and the switch SW22, the resonance frequency of the parallel arm circuit 202 is switched to the high frequency side by switching the switch SW22 from conduction to non-conduction.
　並列腕回路２０３は、並列腕共振子ｐ２３と、並列腕共振子ｐ２３に直列接続されたインピーダンス回路２１３とを有している。インピーダンス回路２１３は、キャパシタＣｐ２３と、スイッチＳＷ２３とを有する。スイッチＳＷ２３は、キャパシタＣｐ２３と並列接続された第９スイッチ素子である。インピーダンス回路２１３がキャパシタＣｐ２３とスイッチＳＷ２３との並列回路で構成されているので、スイッチＳＷ２３が導通から非導通へと切り替わることにより、並列腕回路２０３の共振周波数が高域側へと切り替えられる。 The parallel arm circuit 203 has a parallel arm resonator p23 and an impedance circuit 213 connected in series to the parallel arm resonator p23. The impedance circuit 213 includes a capacitor Cp23 and a switch SW23. The switch SW23 is a ninth switch element connected in parallel to the capacitor Cp23. Since the impedance circuit 213 is formed of a parallel circuit of the capacitor Cp23 and the switch SW23, the resonance frequency of the parallel arm circuit 203 is switched to the high frequency side by switching the switch SW23 from conduction to non-conduction.
　フィルタ１２Ｂは、スイッチＳＷ２１、ＳＷ２２、ＳＷ２３の導通および非導通の切り替えにより、通過帯域低域側の急峻度および通過帯域低域端の周波数が可変する。 In the filter 12B, the sharpness of the low pass band and the frequency of the low end of the pass band are varied by switching between conduction and non-conduction of the switches SW21, SW22, and SW23.
　表５に、実施例６に係る高周波フィルタ６０Ａの回路パラメータを示す。 Table 5 shows circuit parameters of the high frequency filter 60A according to the sixth embodiment.
Figure JPOXMLDOC01-appb-T000005
Figure JPOXMLDOC01-appb-T000005
　［５．２　実施例６に係る高周波フィルタ６０Ａのフィルタ特性］
　図２９Ｂは、実施例６に係る高周波フィルタ６０Ａのスイッチの切り替えによる通過特性の変化を表すグラフである。高周波フィルタ６０Ａは、スイッチＳＷ１、ＳＷ２、ＳＷ５、ＳＷ６、ＳＷ１１～ＳＷ１４、および、ＳＷ２１～ＳＷ２３の導通および非導通を適宜切り替えることにより、複数の帯域幅のフィルタへと可変できる周波数可変型の高周波フィルタとなる。なお、本実施例では、６つの通過帯域を切り替えることが可能となる。 5.2 Filter Characteristics of High-Frequency Filter 60A According to Sixth Embodiment
FIG. 29B is a graph showing the change of the passage characteristic due to the switching of the switch of the high frequency filter 60A according to the sixth embodiment. The high frequency filter 60A is a variable-frequency high frequency filter that can be changed to a filter of a plurality of bandwidths by appropriately switching between conduction and non-conduction of the switches SW1, SW2, SW5, SW6, SW11 to SW14, and SW21 to SW23. It becomes. In the present embodiment, six pass bands can be switched.
　図２９Ｂの（ａ）に示すように、高周波フィルタ６０Ａは、スイッチＳＷ１、ＳＷ２、ＳＷ１１、ＳＷ１２、ＳＷ１４、ＳＷ２１～ＳＷ２３を導通とし、スイッチＳＷ１３を非導通とした場合、第１帯域（Ｂａｎｄ３受信通過帯域およびＢａｎｄ３９通過帯域）を通過帯域とするフィルタとなる。また、図２９Ｂの（ｂ）に示すように、スイッチＳＷ１、ＳＷ２、ＳＷ１１、ＳＷ１２、ＳＷ１４、および、ＳＷ２１～ＳＷ２３を導通とし、スイッチＳＷ５、ＳＷ６およびＳＷ１３を非導通とした場合、第２帯域（Ｂａｎｄ３受信通過帯域）を通過帯域とするフィルタとなる。また、図２９Ｂの（ｃ）に示すように、スイッチＳＷ５、ＳＷ６、ＳＷ１１、ＳＷ１２、ＳＷ１４、および、ＳＷ２１～ＳＷ２３を導通とし、スイッチＳＷ１、ＳＷ２およびＳＷ１３を非導通とした場合、第３帯域（Ｂａｎｄ３９通過帯域）を通過帯域とするフィルタとなる。また、図２９Ｂの（ｄ）に示すように、スイッチＳＷ１、ＳＷ２、ＳＷ５、ＳＷ６、ＳＷ１３、ＳＷ２１～ＳＷ２３を導通とし、スイッチＳＷ１１、ＳＷ１２、ＳＷ１４を非導通とした場合、第１帯域に対して通過帯域低域側の減衰極および通過帯域低域端が高周波側へシフトした第１ｂ帯域を通過帯域とするフィルタとなる。なお、この場合、第２回路は選択されていないため、ＳＷ２１－ＳＷ２３の導通および非導通は、どちらであってもフィルタ特性に影響しない。また、図２９Ｂの（ｅ）に示すように、スイッチＳＷ１、ＳＷ２、ＳＷ１３を導通とし、スイッチＳＷ５、ＳＷ６、ＳＷ１１、ＳＷ１２、ＳＷ１４を非導通とした場合、第２帯域に対して通過帯域低域側の減衰極および通過帯域低域端が高周波側へシフトした第２ｂ帯域を通過帯域とするフィルタとなる。なお、この場合、第１回路は選択されていないため、ＳＷ１１－ＳＷ１４の導通および非導通は、どちらであってもフィルタ特性に影響しない。また、図２９Ｂの（ｆ）に示すように、スイッチＳＷ５、ＳＷ６を導通とし、スイッチＳＷ１、ＳＷ２、ＳＷ２１～ＳＷ２３を非導通とした場合、第３帯域に対して通過帯域低域側の減衰極および通過帯域低域端が高周波側へシフトした第３ｂ帯域を通過帯域とするフィルタとなる。 As shown in (a) of FIG. 29B, when the high frequency filter 60A turns on the switches SW1, SW2, SW11, SW12, SW14, and SW21 to SW23 and turns off the switch SW13, the first band (Band 3 reception pass) It becomes a filter which makes a band and a Band 39 pass band) a pass band. Further, as shown in (b) of FIG. 29B, when the switches SW1, SW2, SW11, SW12, SW14, and SW21 to SW23 are made conductive, and the switches SW5, SW6, and SW13 are made nonconductive, the second band It becomes a filter which makes Band 3 reception passband) a passband. Further, as shown in (c) of FIG. 29B, in the case where the switches SW5, SW6, SW11, SW12, SW14 and SW21 to SW23 are made conductive and the switches SW1, SW2 and SW13 are made nonconductive, the third band It becomes a filter which makes Band 39 pass band) a pass band. Further, as shown in (d) of FIG. 29B, when the switches SW1, SW2, SW5, SW6, SW13, and SW21 to SW23 are made conductive and the switches SW11, SW12, and SW14 are made nonconductive, for the first band. The attenuation pole at the low band side of the pass band and the low band end of the pass band become a filter having the 1 b band shifted to the high frequency side as the pass band. In this case, since the second circuit is not selected, the conduction or non-conduction of SW21-SW23 does not affect the filter characteristics regardless of which is selected. Further, as shown in (e) of FIG. 29B, when the switches SW1, SW2 and SW13 are turned on and the switches SW5, SW6, SW11, SW12 and SW14 are turned off, the pass band lower than the second band The attenuation pole on the side and the low band end of the pass band become a filter having the second band b shifted to the high frequency side as the pass band. In this case, since the first circuit is not selected, the conduction or non-conduction of SW11 to SW14 does not affect the filter characteristics regardless of which is selected. Further, as shown in (f) of FIG. 29B, when the switches SW5 and SW6 are turned on and the switches SW1, SW2 and SW21 to SW23 are turned off, the attenuation pole on the lower side of the pass band with respect to the third band. And, the passband lower band end is a filter that uses the 3b band shifted to the high frequency side as the passband.
　実施例６に係る高周波フィルタ６０Ａにおいて、フィルタ１２Ｂの両方の入出力端子に移相器２１Ｃおよび２２Ｃが接続されることで、高周波フィルタ６０Ａの第１帯域および第１ｂ帯域において、リップルが低減されることで挿入損失が低減する。 In the high frequency filter 60A according to the sixth embodiment, the phase shifters 21C and 22C are connected to both input and output terminals of the filter 12B, whereby ripples are reduced in the first band and the 1b band of the high frequency filter 60A. Insertion loss is reduced.
　実施例６に係る高周波フィルタ６０Ａが、第１帯域および第１ｂ帯域においてリップルおよび挿入損失を低減する要因について、以下に説明する。 The factor by which the high frequency filter 60A according to the sixth embodiment reduces the ripple and the insertion loss in the first band and the 1b band will be described below.
　図３０は、実施例６に係る高周波フィルタ６０Ａの第１帯域における広帯域化の要因を説明するグラフである。同図の（ａ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の通過特性が示されている。同図の（ｂ）には、スイッチＳＷ１が導通状態である場合のノードＸ１から第１回路単体を見たインピーダンス特性およびスイッチＳＷ５が導通状態である場合のノードＸ１から第２回路単体を見たインピーダンス特性が示されている。同図の（ｃ）には、スイッチＳＷ２が導通状態である場合のノードＸ２から第１回路単体を見たインピーダンス特性およびスイッチＳＷ６が導通状態である場合のノードＸ２から第２回路単体を見たインピーダンス特性が示されている。同図の（ｄ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の位相特性が示されている。同図の（ｅ）には、第１回路と第２回路との位相差が示されている。 FIG. 30 is a graph for explaining the factor of broadening the band in the first band of the high-frequency filter 60A according to the sixth embodiment. The pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2 are shown in FIG. The impedance characteristics of the first circuit viewed from the node X1 when the switch SW1 is in the conductive state and the second circuit alone from the node X1 when the switch SW5 is in the conductive state are illustrated in FIG. Impedance characteristics are shown. The impedance characteristics of the first circuit viewed from the node X2 when the switch SW2 is in the conductive state and the second circuit alone from the node X2 when the switch SW6 is in the conductive state are illustrated in FIG. Impedance characteristics are shown. The phase characteristic of the 1st circuit single-piece | unit and 2nd circuit single-piece | unit between node X1-X2 is shown by (d) of the figure. The phase difference between the first circuit and the second circuit is shown in (e) of FIG.
　実施例６に係る高周波フィルタ６０Ａでは、第１帯域が選択された場合、第２回路単体をノードＸ１から見たインピーダンス｜Ｚ１１｜が極大となる特異点のうち最大のインピーダンスとなる周波数ｆ１１が、第２帯域の低域端の周波数ｆＡから、フィルタ１２Ｂの通過帯域低域側のカットオフ周波数ｆＣまでの周波数帯に含まれるように、移相器２１Ｃおよび２２Ｃの位相が調整されている。また、第２回路単体をノードＸ２から見たインピーダンス｜Ｚ２２｜が極大となる特異点のうち最大のインピーダンスとなる周波数ｆ２２が、第２帯域の低域端の周波数ｆＡから、フィルタ１２Ｂの通過帯域低域側のカットオフ周波数ｆＣまでの周波数帯に含まれるように、移相器２１Ｃおよび２２Ｃの位相が調整されている。 In the high frequency filter 60A according to the sixth embodiment, when the first band is selected, the impedance viewed second circuit itself from the node X1 | Z11 | frequency f 11 which is the maximum of the impedance of the singular point as a maximum is The phases of the phase shifters 21C and 22C are adjusted so as to be included in the frequency band from the frequency f A at the low band end of the second band to the cutoff frequency f C at the low band side of the pass band of the filter 12B. There is. The impedance seen second circuit itself from the node X2 | Z22 | frequency f 22 which is the maximum of the impedance of the singular point as a maximum is the frequency f A of the low-frequency end of the second band, the filter 12B The phases of the phase shifters 21C and 22C are adjusted so as to be included in the frequency band up to the cutoff frequency f C on the lower side of the pass band.
　フィルタ１２Ｂの減衰極の周波数ｆＡでは位相変化が大きく、フィルタ１１Ｃとの位相差が大きくなるとともに、振幅差も大きくなる。そのため、移相器２１Ｃおよび２２Ｃの位相調整によって、周波数ｆＡからカットオフ周波数ｆＣまでの周波数範囲に、第２回路のインピーダンス極大の特異点のうち最大のインピーダンスとなる周波数ｆ１１およびｆ２２を合わせることにより、上記周波数範囲の高周波信号が、第２回路のフィルタ１２Ｂに流れこむことを抑制できる。つまり、第２回路が減衰している周波数帯域で第２回路への信号の回り込みを抑制できる。これにより、高周波フィルタ６０Ａの第１帯域内のリップルを抑制し挿入損失を低減することが可能となる。 At the frequency f A of the attenuation pole of the filter 12B, the phase change is large, the phase difference with the filter 11C is large, and the amplitude difference is also large. Therefore, by the phase adjustment of the phase shifters 21C and 22C, in the frequency range from the frequency f A to the cut-off frequency f C , the frequencies f 11 and f 22 that become the largest impedance among the singular points of the impedance maximum of the second circuit. By combining the above, it is possible to suppress the high frequency signal of the above frequency range from flowing into the filter 12B of the second circuit. That is, it is possible to suppress the wraparound of the signal to the second circuit in the frequency band in which the second circuit is attenuated. As a result, it is possible to suppress the ripple in the first band of the high frequency filter 60A and reduce the insertion loss.
　以上のように、高周波フィルタ６０Ａが、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６の導通状態により第１帯域を通過帯域とする場合、第１帯域の挿入損失を低減することが可能となる。 As described above, when the high frequency filter 60A sets the first band as the pass band due to the conduction state of the switches SW1, SW2, SW5, and SW6, it is possible to reduce the insertion loss of the first band.
　さらに、移相器２１Ｃおよび２２Ｃの位相調整により、第１回路と第２回路との振幅差が小さい周波数帯で同位相となるように調整することにより、第１帯域に含まれる周波数の高周波信号は、第１回路および第２回路を経由した後で、互いに打ち消し合うことがない。これにより、第１帯域内のリップルを抑制し挿入損失を低減できる。 Further, the high frequency signal of the frequency included in the first band is adjusted by adjusting the phase difference of the first circuit and the second circuit to be in phase in a small frequency band by phase adjustment of the phase shifters 21C and 22C. Are not canceled each other after passing through the first circuit and the second circuit. Thereby, the ripple in the first band can be suppressed and the insertion loss can be reduced.
　本実施例では、図３０の（ｅ）に示すように、第２回路のノードＸ１およびＸ２から第２回路を単体で見た場合、第２帯域において、第１回路と第２回路とが同位相（０°または－３６０°）になっている周波数が少なくとも１箇所（同図の破線丸印：２箇所）あり、第３帯域において、第１回路と第２回路とが同位相（０°または－３６０°）になっている周波数が少なくとも１箇所（同図の破線丸印：１箇所）あってもよい。 In this embodiment, as shown in (e) of FIG. 30, when the second circuit is viewed alone from the nodes X1 and X2 of the second circuit, the first circuit and the second circuit are the same in the second band. There are at least one frequency (0 dotted line circles: 2 locations) at which the phase (0 ° or -360 °) is in phase, and in the third band, the first circuit and the second circuit have the same phase (0 ° Alternatively, there may be at least one frequency (dotted line circle in the same figure: one place) which is at -360 °.
　上記構成によれば、第１帯域、第２帯域、および第３帯域を切り替えることができる周波数可変型の高周波フィルタ６０Ａにおいて、第１帯域内のリップルを抑制することが可能となる。 According to the above configuration, it is possible to suppress the ripple in the first band in the variable-frequency high frequency filter 60A capable of switching the first band, the second band, and the third band.
　図３１は、実施例６に係る高周波フィルタ６０Ａの第１ｂ帯域における広帯域化の要因を説明するグラフである。同図の（ａ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の通過特性が示されている。同図の（ｂ）には、スイッチＳＷ１が導通状態である場合のノードＸ１から第１回路単体を見たインピーダンス特性およびスイッチＳＷ５が導通状態である場合のノードＸ１から第２回路単体を見たインピーダンス特性が示されている。同図の（ｃ）には、スイッチＳＷ２が導通状態である場合のノードＸ２から第１回路単体を見たインピーダンス特性およびスイッチＳＷ６が導通状態である場合のノードＸ２から第２回路単体を見たインピーダンス特性が示されている。同図の（ｄ）には、ノードＸ１－Ｘ２間の第１回路単体および第２回路単体の位相特性が示されている。同図の（ｅ）には、第１回路と第２回路との位相差が示されている。 FIG. 31 is a graph for explaining the factor of broadening the band in the 1b band of the high frequency filter 60A according to the sixth embodiment. The pass characteristics of the first circuit unit and the second circuit unit between the nodes X1 and X2 are shown in FIG. The impedance characteristics of the first circuit viewed from the node X1 when the switch SW1 is in the conductive state and the second circuit alone from the node X1 when the switch SW5 is in the conductive state are illustrated in FIG. Impedance characteristics are shown. The impedance characteristics of the first circuit viewed from the node X2 when the switch SW2 is in the conductive state and the second circuit alone from the node X2 when the switch SW6 is in the conductive state are illustrated in FIG. Impedance characteristics are shown. The phase characteristic of the 1st circuit single-piece | unit and 2nd circuit single-piece | unit between node X1-X2 is shown by (d) of the figure. The phase difference between the first circuit and the second circuit is shown in (e) of FIG.
　実施例６に係る高周波フィルタ６０Ａでは、第１ｂ帯域が選択された場合、第２回路単体をノードＸ１から見たインピーダンス｜Ｚ１１｜が極大となる特異点のうち最大のインピーダンスとなる周波数ｆ１１が、第２帯域の低域端の周波数ｆＡから、フィルタ１２Ｂの通過帯域低域側のカットオフ周波数ｆＣまでの周波数帯に含まれるように、移相器２１Ｃおよび２２Ｃの位相が調整されている。また、第２回路単体をノードＸ２から見たインピーダンス｜Ｚ２２｜が極大となる特異点のうち最大のインピーダンスとなる周波数ｆ２２が、第２帯域の低域端の周波数ｆＡから、フィルタ１２Ｂの通過帯域低域側のカットオフ周波数ｆＣまでの周波数帯に含まれるように、移相器２１Ｃおよび２２Ｃの位相が調整されている。 In the high frequency filter 60A according to the sixth embodiment, when the first 1b band is selected, the impedance viewed second circuit itself from the node X1 | Z11 | frequency f 11 which is the maximum of the impedance of the singular point as a maximum is The phases of the phase shifters 21C and 22C are adjusted so as to be included in the frequency band from the frequency f A at the low band end of the second band to the cutoff frequency f C at the low band side of the pass band of the filter 12B. There is. The impedance seen second circuit itself from the node X2 | Z22 | frequency f 22 which is the maximum of the impedance of the singular point as a maximum is the frequency f A of the low-frequency end of the second band, the filter 12B The phases of the phase shifters 21C and 22C are adjusted so as to be included in the frequency band up to the cutoff frequency f C on the lower side of the pass band.
　フィルタ１２Ｂの減衰極の周波数ｆＡでは位相変化が大きく、フィルタ１１Ｃとの位相差が大きくなるとともに、振幅差も大きくなる。そのため、移相器２１Ｃおよび２２Ｃの位相調整によって、周波数ｆＡからカットオフ周波数ｆＣまでの周波数範囲に、第２回路のインピーダンス極大の特異点のうち最大のインピーダンスとなる周波数ｆ１１およびｆ２２を合わせることにより、上記周波数範囲の高周波信号が、第２回路のフィルタ１２Ｂに流れこむことを抑制できる。つまり、第２回路が減衰している周波数帯域で第２回路への信号の回り込みを抑制できる。これにより、高周波フィルタ６０Ａの第１ｂ帯域内のリップルを抑制し挿入損失を低減することが可能となる。 At the frequency f A of the attenuation pole of the filter 12B, the phase change is large, the phase difference with the filter 11C is large, and the amplitude difference is also large. Therefore, by the phase adjustment of the phase shifters 21C and 22C, in the frequency range from the frequency f A to the cut-off frequency f C , the frequencies f 11 and f 22 that become the largest impedance among the singular points of the impedance maximum of the second circuit. By combining the above, it is possible to suppress the high frequency signal of the above frequency range from flowing into the filter 12B of the second circuit. That is, it is possible to suppress the wraparound of the signal to the second circuit in the frequency band in which the second circuit is attenuated. As a result, it is possible to suppress the ripple in the 1b band of the high frequency filter 60A and reduce the insertion loss.
　以上のように、高周波フィルタ６０Ａが、スイッチＳＷ１、ＳＷ２、ＳＷ５およびＳＷ６の導通状態により第１ｂ帯域を通過帯域とする場合、第１ｂ帯域の挿入損失を低減することが可能となる。 As described above, when the high frequency filter 60A sets the band 1b as the pass band due to the conduction state of the switches SW1, SW2, SW5, and SW6, it is possible to reduce the insertion loss of the band 1b.
　さらに、移相器２１Ｃおよび２２Ｃの位相調整により、第１回路と第２回路との振幅差が小さい周波数帯で同位相となるように調整することにより、第１ｂ帯域に含まれる周波数の高周波信号は、第１回路および第２回路を経由した後で、互いに打ち消し合うことがない。これにより、第１ｂ帯域内のリップルを抑制し挿入損失を低減できる。 Furthermore, by adjusting the phase difference of the first circuit and the second circuit so that the amplitude difference between the first circuit and the second circuit becomes the same phase in a small frequency band, the high frequency signal of the frequency included in the 1b band Are not canceled each other after passing through the first circuit and the second circuit. Thereby, it is possible to suppress the ripple in the 1b band and reduce the insertion loss.
　本実施例では、図３１の（ｅ）に示すように、第２回路のノードＸ１およびＸ２から第２回路を単体で見た場合、第２ｂ帯域において、第１回路と第２回路とが同位相（０°または－３６０°）になっている周波数が少なくとも１箇所（同図の破線丸印：２箇所）あり、第３ｂ帯域において、第１回路と第２回路とが同位相（０°または－３６０°）になっている周波数が少なくとも１箇所（同図の破線丸印：１箇所）あってもよい。 In this embodiment, as shown in (e) of FIG. 31, when the second circuit is viewed alone from the nodes X1 and X2 of the second circuit, the first circuit and the second circuit are the same in the 2b band. There are at least one frequency (two dotted circles in the same figure: two positions) at which the phase (0 ° or -360 °) is in phase, and the first circuit and the second circuit have the same phase (0 ° in the 3b band Alternatively, there may be at least one frequency (dotted line circle in the same figure: one place) which is at -360 °.
　上記構成によれば、第１ｂ帯域、第２ｂ帯域、および第３ｂ帯域を切り替えることができる周波数可変型の高周波フィルタ６０Ａにおいて、第１ｂ帯域内のリップルを抑制することが可能となる。 According to the above configuration, it is possible to suppress the ripple in the 1b band in the variable-frequency high frequency filter 60A capable of switching the 1b band, the 2b band, and the 3b band.
　本実施例に係る高周波フィルタ６０Ａによれば、通過帯域の切り替えのバリエーションを増やしつつ、第１帯域内および第１ｂ帯域内のリップルを低減させて挿入損失を低減させることが可能となる。 According to the high frequency filter 60A according to this embodiment, it is possible to reduce the insertion loss by reducing the ripples in the first band and the 1b band while increasing the variation of switching of the pass band.
　なお、直列腕共振子の数および並列腕回路の数は、フィルタ特性の要求仕様に応じて適宜決定される。また、直列腕共振子の替わりに、インダクタやキャパシタなどの回路素子が配置されていてもよい。 The number of series arm resonators and the number of parallel arm circuits are appropriately determined according to the required specification of the filter characteristics. Also, instead of the series arm resonator, circuit elements such as an inductor and a capacitor may be arranged.
　また、本実施例に係る高周波フィルタ６０Ａにおいて、フィルタ１１Ｃおよび１２Ｂのそれぞれを構成する、１つの直列腕共振子およびその直列腕に接続された１つの並列腕回路で構成される単位回路としては、図３２Ａ～図３２Ｄ、および、図３３Ａ～図３３Ｇのような変形例が挙げられる。 Further, in the high frequency filter 60A according to the present embodiment, as a unit circuit constituted by one series arm resonator and one parallel arm circuit connected to the series arm, which constitute each of the filters 11C and 12B, There are modifications as shown in FIGS. 32A-32D and FIGS. 33A-33G.
　図３２Ａは、実施例６に係る高周波フィルタ６０Ａを構成する各フィルタの変形例１の回路構成図である。同図に示されたフィルタ１３Ａは、直列腕回路１５１と並列腕回路とで構成されている。並列腕回路は、互いに直列接続された並列腕共振子ｐ１およびインピーダンス回路を有する。インピーダンス回路は、互いに並列接続されたスイッチ（第９スイッチ素子）およびキャパシタを有する。スイッチの導通および非導通の切り替えにより、並列腕回路の共振周波数が切り替わる。 FIG. 32A is a circuit configuration diagram of Modification Example 1 of each filter constituting the high frequency filter 60A according to the sixth embodiment. The filter 13A shown in the figure is composed of a series arm circuit 151 and a parallel arm circuit. The parallel arm circuit has a parallel arm resonator p1 and an impedance circuit connected in series with each other. The impedance circuit has a switch (ninth switch element) and a capacitor connected in parallel to one another. Switching between conduction and non-conduction of the switch switches the resonant frequency of the parallel arm circuit.
　図３２Ｂは、実施例６に係る高周波フィルタ６０Ａを構成する各フィルタの変形例２の回路構成図である。同図に示されたフィルタ１３Ｂは、直列腕回路１５１と並列腕回路とで構成されている。並列腕回路は、互いに直列接続された並列腕共振子ｐ１およびインピーダンス回路を有する。インピーダンス回路は、互いに並列接続されたスイッチ（第９スイッチ素子）およびインダクタを有する。スイッチの導通および非導通の切り替えにより、並列腕回路の共振周波数が切り替わる。 FIG. 32B is a circuit configuration diagram of Modification Example 2 of each filter constituting the high frequency filter 60A according to the sixth embodiment. The filter 13B shown in the figure is composed of a series arm circuit 151 and a parallel arm circuit. The parallel arm circuit has a parallel arm resonator p1 and an impedance circuit connected in series with each other. The impedance circuit has a switch (ninth switch element) and an inductor connected in parallel to each other. Switching between conduction and non-conduction of the switch switches the resonant frequency of the parallel arm circuit.
　図３２Ｃは、実施例６に係る高周波フィルタ６０Ａを構成する各フィルタの変形例３の回路構成図である。同図に示されたフィルタ１３Ｃは、直列腕回路１５１と並列腕回路とで構成されている。並列腕回路は、互いに直列接続された並列腕共振子ｐ１およびインピーダンス回路を有する。インピーダンス回路は、互いに並列接続されたスイッチ（第９スイッチ素子）とインダクタとの直列回路およびキャパシタを有する。スイッチの導通および非導通の切り替えにより、並列腕回路の共振周波数が切り替わる。 32C is a circuit configuration diagram of Modification Example 3 of each filter constituting the high-frequency filter 60A according to Example 6. FIG. The filter 13C shown in the same drawing is configured of a series arm circuit 151 and a parallel arm circuit. The parallel arm circuit has a parallel arm resonator p1 and an impedance circuit connected in series with each other. The impedance circuit has a series circuit of a switch (ninth switch element) and an inductor connected in parallel to each other and a capacitor. Switching between conduction and non-conduction of the switch switches the resonant frequency of the parallel arm circuit.
　図３２Ｄは、実施例６に係る高周波フィルタ６０Ａを構成する各フィルタの変形例４の回路構成図である。同図に示されたフィルタ１３Ｄは、直列腕回路１５１と並列腕回路とで構成されている。並列腕回路は、互いに直列接続された並列腕共振子ｐ１およびインピーダンス回路を有する。インピーダンス回路は、互いに並列接続されたスイッチ（第９スイッチ素子）とキャパシタとの直列回路およびインダクタを有する。スイッチの導通および非導通の切り替えにより、並列腕回路の共振周波数が切り替わる。 FIG. 32D is a circuit configuration diagram of Modification Example 4 of each filter configuring the high-frequency filter 60A according to the sixth embodiment. The filter 13D shown in the figure is constituted by a series arm circuit 151 and a parallel arm circuit. The parallel arm circuit has a parallel arm resonator p1 and an impedance circuit connected in series with each other. The impedance circuit has a series circuit of a switch (ninth switch element) and a capacitor connected in parallel to each other and an inductor. Switching between conduction and non-conduction of the switch switches the resonant frequency of the parallel arm circuit.
　図３３Ａは、実施例６に係る高周波フィルタ６０Ａを構成する各フィルタの変形例５の回路構成図である。同図に示されたフィルタ１４Ａは、直列腕回路１５１と、並列腕回路とで構成されている。並列腕回路は、並列腕共振子ｐ１と、互いに直列接続された並列腕共振子ｐ２およびスイッチ（第９スイッチ素子）と、を有する。並列腕共振子ｐ１の共振周波数は並列腕共振子ｐ２の共振周波数より低く、並列腕共振子ｐ１の反共振周波数は並列腕共振子ｐ２の反共振周波数より低く設定されている。スイッチの導通および非導通の切り替えにより、並列腕回路の共振周波数および反共振周波数の少なくとも一方が切り替わる。 FIG. 33A is a circuit configuration diagram of Modification Example 5 of each filter configuring the high-frequency filter 60A according to the sixth embodiment. The filter 14A shown in the figure is configured of a series arm circuit 151 and a parallel arm circuit. The parallel arm circuit includes a parallel arm resonator p1 and a parallel arm resonator p2 and a switch (ninth switch element) connected in series with each other. The resonance frequency of the parallel arm resonator p1 is lower than the resonance frequency of the parallel arm resonator p2, and the antiresonance frequency of the parallel arm resonator p1 is set lower than the antiresonance frequency of the parallel arm resonator p2. Switching between conduction and non-conduction of the switch switches at least one of the resonant frequency and the antiresonant frequency of the parallel arm circuit.
　図３３Ｂは、実施例６に係る高周波フィルタ６０Ａを構成する各フィルタの変形例６の回路構成図である。同図に示されたフィルタ１４Ｂは、直列腕回路１５１と並列腕回路とで構成されている。並列腕回路は、並列腕共振子ｐ１と、互いに直列接続された並列腕共振子ｐ２およびインピーダンス回路を有する。並列腕共振子ｐ１の共振周波数は並列腕共振子ｐ２の共振周波数と異なり、並列腕共振子ｐ１の反共振周波数は並列腕共振子ｐ２の反共振周波数と異なる。インピーダンス回路は、互いに並列接続されたスイッチ（第９スイッチ素子）およびキャパシタを有する。並列腕回路は、２つの共振周波数と２つの反共振周波数を有し、スイッチの導通および非導通の切り替えにより、並列腕回路の共振周波数の少なくとも１つと、反共振周波数の少なくとも１つとが、ともに切り替わる。 FIG. 33B is a circuit configuration diagram of Modification Example 6 of the filters configuring the high-frequency filter 60A according to the sixth embodiment. The filter 14B shown in the figure is composed of a series arm circuit 151 and a parallel arm circuit. The parallel arm circuit includes a parallel arm resonator p1, a parallel arm resonator p2 connected in series with one another, and an impedance circuit. The resonance frequency of the parallel arm resonator p1 is different from the resonance frequency of the parallel arm resonator p2, and the antiresonance frequency of the parallel arm resonator p1 is different from the antiresonance frequency of the parallel arm resonator p2. The impedance circuit has a switch (ninth switch element) and a capacitor connected in parallel to one another. The parallel arm circuit has two resonance frequencies and two antiresonance frequencies, and switching between conduction and non-conduction of the switch allows at least one of the resonance frequencies of the parallel arm circuit and at least one of the antiresonance frequencies to be both. Switch.
　図３３Ｃは、実施例６に係る高周波フィルタ６０Ａを構成する各フィルタの変形例７の回路構成図である。同図に示されたフィルタ１４Ｃは、直列腕回路１５１と、並列腕回路とで構成されている。並列腕回路は、互いに直列接続された並列腕共振子ｐ１および第１インピーダンス回路と、互いに直列接続された並列腕共振子ｐ２および第２インピーダンス回路とを有する。並列腕共振子ｐ１の共振周波数は並列腕共振子ｐ２の共振周波数と異なり、並列腕共振子ｐ１の反共振周波数は並列腕共振子ｐ２の反共振周波数と異なる。第１インピーダンス回路は、互いに並列接続されたスイッチＳＷ１（第９スイッチ素子）およびキャパシタを有する。第２インピーダンス回路は、互いに並列接続されたスイッチＳＷ２（第９スイッチ素子）およびキャパシタを有する。並列腕回路は、２つの共振周波数と２つの反共振周波数を有し、それぞれのスイッチの導通および非導通の切り替えにより、並列腕回路の共振周波数の少なくとも１つと、反共振周波少なくとも１つとが、ともに切り替わる。 FIG. 33C is a circuit configuration diagram of Modification Example 7 of each of the filters forming the high-frequency filter 60A according to the sixth embodiment. The filter 14C shown in the same drawing is configured of a series arm circuit 151 and a parallel arm circuit. The parallel arm circuit includes a parallel arm resonator p1 and a first impedance circuit connected in series with each other, and a parallel arm resonator p2 and a second impedance circuit connected with each other in series. The resonance frequency of the parallel arm resonator p1 is different from the resonance frequency of the parallel arm resonator p2, and the antiresonance frequency of the parallel arm resonator p1 is different from the antiresonance frequency of the parallel arm resonator p2. The first impedance circuit includes a switch SW1 (ninth switch element) and a capacitor connected in parallel to each other. The second impedance circuit includes a switch SW2 (ninth switch element) and a capacitor connected in parallel to each other. The parallel arm circuit has two resonant frequencies and two antiresonant frequencies, and switching between conduction and non-conduction of each switch causes at least one of the resonant frequencies of the parallel arm circuit and at least one of the antiresonant frequencies to It switches together.
　図３３Ｄは、実施例６に係る高周波フィルタ６０Ａを構成する各フィルタの変形例８の回路構成図である。同図に示されたフィルタ１４Ｄは、直列腕回路１５１と、並列腕回路とで構成されている。並列腕回路は、互いに直列接続された並列腕共振子ｐ１およびインピーダンス回路を有する。インピーダンス回路は、互いに並列接続されたスイッチＳＷ１（第９スイッチ素子）および並列腕共振子ｐ２を有する。並列腕共振子ｐ１の共振周波数は並列腕共振子ｐ２の共振周波数と異なり、並列腕共振子ｐ１の反共振周波数は並列腕共振子ｐ２の反共振周波数と異なる。スイッチの導通および非導通の切り替えにより、並列腕回路の共振周波数および反共振周波数の周波数と個数が切り替わる。 FIG. 33D is a circuit configuration diagram of Modified Example 8 of the filters constituting the high frequency filter 60A according to the sixth embodiment. The filter 14D shown in the figure is configured of a series arm circuit 151 and a parallel arm circuit. The parallel arm circuit has a parallel arm resonator p1 and an impedance circuit connected in series with each other. The impedance circuit includes a switch SW1 (ninth switch element) and a parallel arm resonator p2 connected in parallel to each other. The resonance frequency of the parallel arm resonator p1 is different from the resonance frequency of the parallel arm resonator p2, and the antiresonance frequency of the parallel arm resonator p1 is different from the antiresonance frequency of the parallel arm resonator p2. Switching between conduction and non-conduction of the switch switches the frequency and number of resonant frequencies and antiresonant frequencies of the parallel arm circuit.
　図３３Ｅは、実施例６に係る高周波フィルタ６０Ａを構成する各フィルタの変形例９の回路構成図である。同図に示されたフィルタ１４Ｅは、直列腕回路１５１と、並列腕回路とで構成されている。並列腕回路は、並列腕共振子ｐ１およびｐ２が並列接続された回路と、インピーダンス回路とが直列接続された構成を有する。並列腕共振子ｐ１の共振周波数は並列腕共振子ｐ２の共振周波数より低く、並列腕共振子ｐ１の反共振周波数は並列腕共振子ｐ２の反共振周波数より低く設定されている。インピーダンス回路は、互いに並列接続されたスイッチＳＷ１（第９スイッチ素子）およびキャパシタを有する。並列腕回路は、２つの共振周波数と２つの反共振周波数を有し、スイッチの導通および非導通の切り替えにより、並列腕回路の２つの共振周波数が切り替わる。 FIG. 33E is a circuit configuration diagram of Modification 9 of the filters constituting the high-frequency filter 60A according to the sixth embodiment. The filter 14E shown in the figure is configured of a series arm circuit 151 and a parallel arm circuit. The parallel arm circuit has a configuration in which a circuit in which parallel arm resonators p1 and p2 are connected in parallel and an impedance circuit are connected in series. The resonance frequency of the parallel arm resonator p1 is lower than the resonance frequency of the parallel arm resonator p2, and the antiresonance frequency of the parallel arm resonator p1 is set lower than the antiresonance frequency of the parallel arm resonator p2. The impedance circuit has a switch SW1 (ninth switch element) and a capacitor connected in parallel to each other. The parallel arm circuit has two resonant frequencies and two antiresonant frequencies, and switching between conduction and non-conduction of the switch switches the two resonant frequencies of the parallel arm circuit.
　図３３Ｆは、実施例６に係る高周波フィルタ６０Ａを構成する各フィルタの変形例１０の回路構成図である。同図に示されたフィルタ１４Ｆは、直列腕回路１５１と、並列腕回路とで構成されている。並列腕回路は、並列腕共振子ｐ１およびｐ２が並列接続された回路と、インピーダンス回路とが直列接続された構成を有する。並列腕共振子ｐ１の共振周波数は並列腕共振子ｐ２の共振周波数より低く、並列腕共振子ｐ１の反共振周波数は並列腕共振子ｐ２の反共振周波数より低く設定されている。インピーダンス回路は、互いに並列接続されたスイッチＳＷ１（第９スイッチ素子）およびインダクタを有する。並列腕回路は、２つの共振周波数と２つの反共振周波数を有し、スイッチの導通および非導通の切り替えにより、並列腕回路の２つの共振周波数が切り替わる。 FIG. 33F is a circuit configuration diagram of Modification Example 10 of the filters configuring the high-frequency filter 60A according to the sixth embodiment. The filter 14F shown in the figure is configured of a series arm circuit 151 and a parallel arm circuit. The parallel arm circuit has a configuration in which a circuit in which parallel arm resonators p1 and p2 are connected in parallel and an impedance circuit are connected in series. The resonance frequency of the parallel arm resonator p1 is lower than the resonance frequency of the parallel arm resonator p2, and the antiresonance frequency of the parallel arm resonator p1 is set lower than the antiresonance frequency of the parallel arm resonator p2. The impedance circuit has a switch SW1 (ninth switch element) and an inductor connected in parallel to each other. The parallel arm circuit has two resonant frequencies and two antiresonant frequencies, and switching between conduction and non-conduction of the switch switches the two resonant frequencies of the parallel arm circuit.
　図３３Ｇは、実施例６に係る高周波フィルタ６０Ａを構成する各フィルタの変形例１１の回路構成図である。同図に示されたフィルタ１４Ｇは、直列腕回路と、並列腕回路１６１とで構成されている。直列腕回路は、直列腕共振子ｓ１とキャパシタおよびスイッチ（第９スイッチ素子）の直列回路とが並列接続された構成を有する。スイッチの導通および非導通の切り替えにより、直列腕回路の反共振周波数が切り替わる。 FIG. 33G is a circuit configuration diagram of Modification 11 of the filters constituting the high frequency filter 60A according to Embodiment 6. The filter 14 </ b> G shown in the figure is configured of a series arm circuit and a parallel arm circuit 161. The series arm circuit has a configuration in which a series arm resonator s1 and a series circuit of a capacitor and a switch (ninth switch element) are connected in parallel. Switching between conduction and non-conduction of the switch switches the antiresonant frequency of the series arm circuit.
　（実施の形態６）
　以上の実施の形態１～６で説明した高周波フィルタは、使用バンド数が多いシステムに対応するマルチプレクサおよび高周波フロントエンド回路に適用することもできる。そこで、本実施の形態では、このようなマルチプレクサおよび高周波フロントエンド回路および通信装置について説明する。 Sixth Embodiment
The high frequency filters described in the first to sixth embodiments can also be applied to multiplexers and high frequency front end circuits corresponding to a system having a large number of used bands. Therefore, in the present embodiment, such a multiplexer, a high frequency front end circuit, and a communication apparatus will be described.
　図３４は、実施の形態６に係る通信装置９およびその周辺回路の構成図である。 FIG. 34 is a block diagram of the communication device 9 and its peripheral circuits according to the sixth embodiment.
　同図に示すように、通信装置９は、複数のスイッチにより構成されるスイッチ群２１０と、複数のフィルタにより構成されるフィルタ群２２０と、送信側スイッチ２３１と、受信側スイッチ２５１、２５２および２５３と、スイッチ２７１および２７２と、送信増幅回路２４１、２４２および２４３と、受信増幅回路２６１、２６２、２６３および２６４と、ＲＦ信号処理回路（ＲＦＩＣ）７と、ベースバンド信号処理回路（ＢＢＩＣ）８と、アンテナ素子５と、を備える。 As shown in the figure, the communication apparatus 9 includes a switch group 210 formed of a plurality of switches, a filter group 220 formed of a plurality of filters, a transmission side switch 231, and reception side switches 251, 252 and 253. , Switches 271 and 272, transmission amplification circuits 241, 242 and 243, reception amplification circuits 261, 262, 263 and 264, RF signal processing circuit (RFIC) 7, and baseband signal processing circuit (BBIC) 8 , And an antenna element 5.
　スイッチ群２１０は、制御部（図示せず）からの制御信号にしたがって、アンテナ素子５と所定のバンドに対応する信号経路とを接続し、例えば、複数のＳＰＳＴ型のスイッチによって構成される。なお、アンテナ素子５と接続される信号経路は１つに限らず、複数であってもかまわない。つまり、通信装置９は、キャリアアグリゲーションに対応してもかまわない。 Switch group 210 connects antenna element 5 and a signal path corresponding to a predetermined band in accordance with a control signal from a control unit (not shown), and is formed of, for example, a plurality of SPST type switches. The number of signal paths connected to the antenna element 5 is not limited to one, and may be plural. That is, the communication device 9 may support carrier aggregation.
　フィルタ群２２０は、例えば次の帯域を通過帯域に有する複数のフィルタ（デュプレクサを含む）によって構成される。具体的には、当該帯域は、（ｉ－Ｔｘ）Ｂａｎｄ３、４および６６の送信帯域、（ｉ－Ｒｘ）Ｂａｎｄ３またはＢａｎｄ３、３９の受信帯域、（ｉｉ－Ｔｘ）Ｂａｎｄ２５の送信帯域、（ｉｉ－Ｒｘ）Ｂａｎｄ２５の受信帯域、（ｉｉｉ－Ｔｘ）Ｂａｎｄ１または６５の送信帯域、（ｉｉｉ－Ｒｘ）Ｂａｎｄ１、４、６５、６６の受信帯域、（ｉｖ－Ｔｘ）Ｂａｎｄ３０の送信帯域、（ｉｖ－Ｒｘ）Ｂａｎｄ３０の受信帯域、（ｖ）Ｂａｎｄ７の送信帯域、（ｖｉ）Ｂａｎｄ７またはＢａｎｄ７、３８の受信帯域、（ｖｉｉ）Ｂａｎｄ３９の送受信帯域、（ｖｉｉｉ）Ｂａｎｄ３４の送受信帯域、（ｉｘ）Ｂａｎｄ４０の送受信帯域、（ｘ）Ｂａｎｄ４１または３８の送受信帯域、である。 The filter group 220 is composed of, for example, a plurality of filters (including duplexers) having the next band in the pass band. Specifically, the band is a transmission band of (i-Tx) Band 3, 4 and 66, a reception band of (i-Rx) Band 3 or Band 3, 39, a transmission band of (ii-Tx) Band 25, (ii- Rx) Band 25 reception band, (iii-Tx) Band 1 or 65 transmission band, (iii-Rx) Band 1, 4, 65, 66 reception band, (iv-Tx) Band 30 transmission band, (iv-Rx) (B) Band 7 transmission band, (v) Band 7 or Band 7 or 38 reception band, (vii) Band 39 transmission / reception band, (viii) Band 34 transmission / reception band, (ix) Band 40 transmission / reception band, x) Band 41 or 38 transmission / reception band.
　送信側スイッチ２３１は、フィルタ群２２０の中心周波数が低いローバンド側の複数の送信側信号経路に接続された複数の選択端子と送信増幅回路２４１に接続された共通端子とを有するスイッチ回路である。 The transmission side switch 231 is a switch circuit having a plurality of selection terminals connected to a plurality of transmission side signal paths on the low band side where the center frequency of the filter group 220 is low and a common terminal connected to the transmission amplification circuit 241.
　受信側スイッチ２５１は、複数の受信側信号経路に接続された複数の選択端子と受信増幅回路２６１に接続された共通端子とを有するスイッチ回路である。受信側スイッチ２５２は、複数の受信側信号経路に接続された複数の選択端子と受信増幅回路２６２に接続された共通端子とを有するスイッチ回路である。受信側スイッチ２５３は、受信側信号経路およびスイッチ２７１の選択端子に接続された複数の選択端子と受信増幅回路２６３に接続された共通端子とを有するスイッチ回路である。 The reception side switch 251 is a switch circuit having a plurality of selection terminals connected to a plurality of reception side signal paths and a common terminal connected to the reception amplification circuit 261. The reception side switch 252 is a switch circuit having a plurality of selection terminals connected to a plurality of reception side signal paths and a common terminal connected to the reception amplification circuit 262. The reception side switch 253 is a switch circuit having a reception side signal path, a plurality of selection terminals connected to the selection terminal of the switch 271, and a common terminal connected to the reception amplification circuit 263.
　スイッチ２７１は、所定のバンド（ここではＢａｎｄ３９）の送受信経路に接続された共通端子と、受信側スイッチ２５３の選択端子および送信増幅回路２４２に接続された２つの選択端子とを有するスイッチ回路である。スイッチ２７２は、複数の送受信経路に接続された複数の選択端子と、送信増幅回路２４３、受信増幅回路２６４に接続された２つの選択端子とを有するスイッチ回路である。 The switch 271 is a switch circuit having a common terminal connected to the transmission / reception path of a predetermined band (here, Band 39), and two selection terminals connected to the selection terminal of the reception side switch 253 and the transmission amplification circuit 242. . The switch 272 is a switch circuit having a plurality of selection terminals connected to a plurality of transmission and reception paths, and two selection terminals connected to the transmission amplification circuit 243 and the reception amplification circuit 264.
　これらのスイッチは、フィルタ群２２０の後段（ここでは受信側信号経路における後段）に設けられ、制御部（図示せず）からの制御信号にしたがって接続状態が切り替えられる。 These switches are provided in the subsequent stage (in this case, the subsequent stage in the reception side signal path) of the filter group 220, and the connection state is switched according to the control signal from the control unit (not shown).
　送信増幅回路２４１、２４２および２４３は、それぞれ、所定の周波数帯域の高周波送信信号を電力増幅するパワーアンプである。 The transmission amplification circuits 241, 242 and 243 are power amplifiers for power-amplifying high frequency transmission signals in a predetermined frequency band.
　受信増幅回路２６１、２６２、２６３および２６４は、それぞれ、所定の周波数帯域の高周波受信信号を電力増幅するローノイズアンプである。 The reception amplifier circuits 261, 262, 263 and 264 are each a low noise amplifier for power-amplifying a high frequency reception signal of a predetermined frequency band.
　ＲＦ信号処理回路（ＲＦＩＣ）７は、アンテナ素子５で送受信される高周波信号を処理する回路である。具体的には、ＲＦ信号処理回路（ＲＦＩＣ）７は、アンテナ素子５から受信側信号経路を介して入力された高周波信号（ここでは高周波受信信号）を、ダウンコンバートなどにより信号処理し、当該信号処理して生成された受信信号をベースバンド信号処理回路（ＢＢＩＣ）８へ出力する。また、ＲＦ信号処理回路（ＲＦＩＣ）７は、ベースバンド信号処理回路（ＢＢＩＣ）８から入力された送信信号をアップコンバートなどにより信号処理し、当該信号処理して生成された高周波信号（ここでは高周波送信信号）を送信側信号経路に出力する。 The RF signal processing circuit (RFIC) 7 is a circuit that processes a high frequency signal transmitted and received by the antenna element 5. Specifically, the RF signal processing circuit (RFIC) 7 performs signal processing of a high frequency signal (here, a high frequency received signal) input from the antenna element 5 via the receiving signal path by down conversion or the like, and the signal The received signal generated by processing is output to the baseband signal processing circuit (BBIC) 8. Further, the RF signal processing circuit (RFIC) 7 performs signal processing on the transmission signal input from the baseband signal processing circuit (BBIC) 8 by up conversion and the like, and generates a high frequency signal (high frequency in this case) generated by the signal processing. The transmission signal is output to the transmission side signal path.
　このように構成された通信装置９は、上記（ｉ－Ｔｘ）～（ｘ）のいずれかを通過帯域に有するフィルタの少なくとも１つとして、実施の形態１～５のいずれかに係るフィルタを備える。 The communication apparatus 9 configured in this manner includes the filter according to any one of the first to fifth embodiments as at least one of the filters having any of the above (i-Tx) to (x) in the pass band. .
　なお、通信装置９のうち、スイッチ群２１０と、フィルタ群２２０と、送信側スイッチ２３１と、受信側スイッチ２５１、２５２および２５３と、スイッチ２７１および２７２と、送信増幅回路２４１、２４２および２４３と、受信増幅回路２６１、２６２、２６３および２６４と、上記制御部とは、高周波フロントエンド回路６を構成する。また、スイッチ群２１０とフィルタ群２２０とは、マルチプレクサを構成する。なお、本発明に係るマルチプレクサは、本実施の形態のように、フィルタ群２２０が、スイッチ群２１０を介して共通端子に接続されていてもよいし、実施の形態１～５のいずれかに係る複数のフィルタが、共通端子に直接接続されている構成であってもよい。 Of the communication device 9, the switch group 210, the filter group 220, the transmission side switch 231, the reception side switches 251, 252 and 253, the switches 271 and 272, and the transmission amplification circuits 241, 242 and 243, The reception amplifier circuits 261, 262, 263 and 264 and the control unit constitute a high frequency front end circuit 6. The switch group 210 and the filter group 220 constitute a multiplexer. In the multiplexer according to the present invention, as in the present embodiment, the filter group 220 may be connected to the common terminal through the switch group 210, and one of the first to fifth embodiments. A plurality of filters may be directly connected to the common terminal.
　ここで、上記制御部は、図３４には図示していないが、ＲＦ信号処理回路（ＲＦＩＣ）が有していてもよいし、制御部が制御する各スイッチとともにスイッチＩＣを構成していてもよい。 Here, although the control unit is not shown in FIG. 34, it may be included in an RF signal processing circuit (RFIC), or may constitute a switch IC together with each switch controlled by the control unit. Good.
　以上のように構成された高周波フロントエンド回路６および通信装置９によれば、上記実施の形態１～５のいずれかに係るフィルタを備えることにより、通過帯域内のリップル(挿入損失偏差)が低減された低損失かつ広帯域の高周波フロントエンド回路および通信装置を実現できる。 According to the high frequency front end circuit 6 and the communication device 9 configured as described above, by providing the filter according to any one of the first to fifth embodiments, ripple (insertion loss deviation) in the passband is reduced. It is possible to realize a low loss and wide band high frequency front end circuit and communication device.
　また、本実施の形態に係る高周波フロントエンド回路６によれば、フィルタ群２２０（複数の高周波フィルタ）の前段または後段に設けられた送信側スイッチ２３１、受信側スイッチ２５１、２５２および２５３、ならびにスイッチ２７１および２７２（スイッチ回路）を備える。これにより、高周波信号が伝達される信号経路の一部を共通化することができる。よって、例えば、複数の高周波フィルタに対応する送信増幅回路２４１、２４２および２４３、ならびに、受信増幅回路２６１、２６２、２６３および２６４（増幅回路）を共通化することができる。したがって、高周波フロントエンド回路の小型化および低コスト化が可能となる。 Further, according to the high frequency front end circuit 6 according to the present embodiment, the transmission side switch 231, the reception side switches 251, 252 and 253, and the switch provided in the front stage or the rear stage of the filter group 220 (a plurality of high frequency filters). 271 and 272 (switch circuits). Thus, part of the signal path through which the high frequency signal is transmitted can be shared. Therefore, for example, the transmission amplification circuits 241, 242 and 243 corresponding to a plurality of high frequency filters, and the reception amplification circuits 261, 262, 263 and 264 (amplification circuits) can be shared. Therefore, miniaturization and cost reduction of the high frequency front end circuit can be realized.
　なお、送信側スイッチ２３１、受信側スイッチ２５１、２５２および２５３、ならびにスイッチ２７１および２７２は、少なくとも１つが設けられていればよい。また、送信側スイッチの個数、ならびに、受信側スイッチの個数は、上記説明した個数に限らず、例えば、１つの送信側スイッチと１つの受信側スイッチとが設けられていてもかまわない。また、送信側スイッチおよび受信側スイッチの選択端子等の個数も、本実施の形態に限らず、それぞれ２つであってもかまわない。 Note that at least one of the transmission side switch 231, the reception side switches 251, 252 and 253, and the switches 271 and 272 may be provided. Further, the number of transmission side switches and the number of reception side switches are not limited to the number described above. For example, one transmission side switch and one reception side switch may be provided. Further, the number of selection terminals and the like of the transmission side switch and the reception side switch is not limited to the present embodiment, and may be two.
　（その他の実施の形態）
　以上、本発明に係る高周波フィルタ、マルチプレクサ、高周波フロントエンド回路および通信装置について、実施の形態１～６を挙げて説明したが、本発明は、上記実施の形態に限定されるものではない。上記実施の形態における任意の構成要素を組み合わせて実現される別の実施の形態や、上記実施の形態に対して本発明の主旨を逸脱しない範囲で当業者が思いつく各種変形を施して得られる変形例や、本発明に係る高周波フィルタ、マルチプレクサ、高周波フロントエンド回路および通信装置を内蔵した各種機器も本発明に含まれる。 (Other embodiments)
The high frequency filter, the multiplexer, the high frequency front end circuit, and the communication apparatus according to the present invention have been described above with reference to the first to sixth embodiments, but the present invention is not limited to the above embodiments. Other embodiments realized by combining arbitrary components in the above embodiments, and modifications obtained by applying various modifications to the above embodiments without departing from the scope of the present invention will occur to those skilled in the art Examples of the present invention include various devices incorporating the high frequency filter, multiplexer, high frequency front end circuit, and communication device according to the present invention.
　例えば、上記実施の形態１～６では、第２帯域をＬＴＥのＢａｎｄ３受信通過帯域（１８０５－１８８０ＭＨｚ）に適用し、第３帯域をＬＴＥのＢａｎｄ３９通過帯域（１８８０－１９２０ＭＨｚ）に適用し、第１帯域をＢａｎｄ３受信通過帯域とＢａｎｄ３９通過帯域とを含む帯域（１８０５－１９２０ＭＨｚ）に適用した例を示したが、これに限られない。 For example, in the first to sixth embodiments, the second band is applied to the Band 3 reception passband (1805-1880 MHz) of LTE, and the third band is applied to the Band 39 passband (1880 to 1920 MHz) of LTE. Although the example in which the band is applied to the band (1805-1920 MHz) including the Band 3 reception pass band and the Band 39 pass band has been shown, the present invention is not limited thereto.
　第２帯域をＬＴＥのＢａｎｄ２８Ａ受信通過帯域（７５８－７８８ＭＨｚ）に適用し、第３帯域をＬＴＥのＢａｎｄ２０受信通過帯域（７９１－８２１ＭＨｚ）に適用し、第１帯域をＢａｎｄ２８Ａ受信通過帯域とＢａｎｄ２０受信通過帯域とを含む帯域（７５８－８２１ＭＨｚ）に適用してもよい。 The second band is applied to the Band 28A reception passband (758 to 788 MHz) of LTE, the third band is applied to the Band 20 reception passband (791 to 821 MHz) of LTE, and the first band is the Band 28A reception passband and Band 20 reception passband. The present invention may be applied to a band (758-821 MHz) including the band.
　また、第２帯域をＬＴＥのＢａｎｄ３８通過帯域（２５７０－２６２０ＭＨｚ）に適用し、第３帯域をＬＴＥのＢａｎｄ７受信通過帯域（２６２０－２６９０ＭＨｚ）に適用し、第１帯域をＢａｎｄ３８通過帯域とＢａｎｄ７受信通過帯域とを含む帯域（２５７０－２６９０ＭＨｚ）に適用してもよい。 Also, the second band is applied to Band 38 pass band (2570-2620 MHz) of LTE, the third band is applied to Band 7 reception pass band (2620-2690 MHz) of LTE, and the first band is Band 38 pass band and Band 7 reception pass The present invention may be applied to a band (2570-2690 MHz) including the band.
　また、各フィルタを構成する直列腕共振子および並列腕共振子の各々は、１つの共振子に限らず、１つの共振子が分割された複数の分割共振子によって構成されていてもかまわない。 Further, each of the series arm resonator and the parallel arm resonator constituting each filter is not limited to one resonator, and may be constituted by a plurality of divided resonators in which one resonator is divided.
　また、実施の形態１～６で示された移相器は、位相変換とともにインピーダンス変換の機能も有しており、「インピーダンス変換器」と読み替えてもよい。 In addition, the phase shifters described in the first to sixth embodiments have a function of impedance conversion as well as phase conversion, and may be replaced with “impedance converter”.
　また、例えば、制御部は、ＲＦ信号処理回路（ＲＦＩＣ）７の外部に設けられていてもよく、例えば、高周波フロントエンド回路６に設けられていてもかまわない。つまり、高周波フロントエンド回路６は、上記説明した構成に限らず、実施の形態１～５のいずれかに係る高周波フィルタと、当該高周波フィルタが備えるスイッチ素子のオンおよびオフを制御する制御部と、を備えてもかまわない。 Also, for example, the control unit may be provided outside the RF signal processing circuit (RFIC) 7, and may be provided, for example, in the high frequency front end circuit 6. That is, the high frequency front end circuit 6 is not limited to the above-described configuration, and the high frequency filter according to any one of the first to fifth embodiments, and a control unit for controlling on and off of the switch element of the high frequency filter. You may have
　また、例えば、高周波フロントエンド回路６または通信装置９において、各構成要素の間に、インダクタやキャパシタが接続されていてもかまわない。なお、インダクタには、各構成要素間を繋ぐ配線による配線インダクタが含まれてもよい。 Further, for example, in the high frequency front end circuit 6 or the communication device 9, an inductor or a capacitor may be connected between each component. Note that the inductor may include a wiring inductor formed by wiring connecting the components.
　また、実施の形態１～５のいずれかに係る高周波フィルタが備える各スイッチ素子は、例えば、ＧａＡｓもしくはＣＭＯＳ（Ｃｏｍｐｌｅｍｅｎｔａｒｙ　Ｍｅｔａｌ　Ｏｘｉｄｅ　Ｓｅｍｉｃｏｎｄｕｃｔｏｒ）からなるＦＥＴ（Ｆｉｅｌｄ　Ｅｆｆｅｃｔ　Ｔｒａｎｓｉｓｔｏｒ）スイッチ、または、ダイオードスイッチが挙げられる。このようなスイッチは小型であるため、実施の形態１～５のいずれかに係る高周波フィルタを小型化することができる。 In addition, each switch element included in the high frequency filter according to any one of the first to fifth embodiments may be, for example, a field effect transistor (FET) switch or a diode switch formed of GaAs or complementary metal oxide semiconductor (CMOS). . Since such a switch is small, the high frequency filter according to any one of the first to fifth embodiments can be miniaturized.
　また、実施の形態１～５のいずれかに係る高周波フィルタが備える直列腕共振子および並列腕共振子は、弾性波を用いた弾性波共振子であり、例えば、ＳＡＷ（Ｓｕｒｆａｃｅ　Ａｃｏｕｓｔｉｃ　Ｗａｖｅ）を利用した共振子、ＢＡＷ（Ｂｕｌｋ　Ａｃｏｕｓｔｉｃ　Ｗａｖｅ）を利用した共振子、もしくは、ＦＢＡＲ（Ｆｉｌｍ　Ｂｕｌｋ　Ａｃｏｕｓｔｉｃ　Ｒｅｓｏｎａｔｏｒ）等である。なお、ＳＡＷには、表面波だけでなく境界波も含まれる。さらに、並列腕共振子は、インダクタンス成分およびキャパシタンス成分で構成された等価回路モデル（例えば、ＢＶＤモデルなど）で表現される共振子または回路であって、共振周波数および反共振周波数を有する共振子または回路であればよい。 In addition, the series arm resonator and the parallel arm resonator included in the high frequency filter according to any one of the first to fifth embodiments are elastic wave resonators using elastic waves, and use, for example, SAW (Surface Acoustic Wave) Or a resonator using BAW (Bulk Acoustic Wave), or a film bulk acoustic resonator (FBAR). The SAW includes not only surface waves but also boundary waves. Furthermore, the parallel arm resonator is a resonator or a circuit represented by an equivalent circuit model (for example, a BVD model etc.) composed of an inductance component and a capacitance component, and a resonator having a resonant frequency and an antiresonant frequency or It may be a circuit.
　本発明は、広帯域かつ低損失なフィルタ、マルチプレクサ、フロントエンド回路および通信装置として、携帯電話などの通信機器に広く利用できる。 INDUSTRIAL APPLICABILITY The present invention is widely applicable to communication devices such as mobile phones as wide-band low-loss filters, multiplexers, front end circuits and communication devices.
　５　　アンテナ素子
　６　　高周波フロントエンド回路
　７　　ＲＦ信号処理回路（ＲＦＩＣ）
　８　　ベースバンド信号処理回路（ＢＢＩＣ）４
　９　　通信装置
　１０、１０Ａ、１０Ｂ、２０、２０Ａ、２０Ｂ、２０Ｃ、３０、４０Ａ、５０Ａ、６０Ａ、５００、６００　　高周波フィルタ
　１１、１１Ａ、１１Ｂ、１１Ｃ、１１Ｄ、１２、１２Ａ、１２Ａａ、１２Ｂ、１２Ｄ、１３Ａ、１３Ｂ、１３Ｃ、１３Ｄ、１４Ａ、１４Ｂ、１４Ｃ、１４Ｄ、１４Ｅ、１４Ｆ、１４Ｇ　　フィルタ
　１５Ａ　　縦結合型共振器
　２１、２１ａ、２１Ｂ、２１Ｃ、２１Ｄ、２２、２２ａ、２２Ｂ、２２Ｃ、２２Ｄ　　移相器
　１０１、１０２、１０３、１０４、１６１、２０１、２０２、２０３　　並列腕回路
　１１０、１２０　　入出力端子
　１１１、１１２、１１３、１１４、２１１、２１２、２１３　　インピーダンス回路
　１５１　　直列腕回路
　２１０　　スイッチ群
　２２０　　フィルタ群
　２３１　　送信側スイッチ
　２４１、２４２、２４３　　送信増幅回路
　２５１、２５２、２５３　　受信側スイッチ
　２６１、２６２、２６３、２６４　受信増幅回路
　２７１、２７２　　スイッチ
　３２１　　ＩＤＴ電極
　３０１ａ、３０１ｂ　　櫛歯電極
　３２２　　反射器
　３０１　　電極膜
　３０２　　圧電基板
　３０１ｇ　　密着層
　３０１ｈ　　主電極層
　３０３　　保護層
　３１０ａ、３１０ｂ、４１０　　電極指
　３１１ａ、３１１ｂ、４１１　　バスバー電極
　Ｃｐ１１、Ｃｐ１２、Ｃｐ１４、Ｃｐ２１、Ｃｐ２２、Ｃｐ２３、Ｃｓ２１、Ｃｓ２２　　キャパシタ
　Ｌｐ１３、Ｌｐ２１、Ｌｐ２２、Ｌｓ２１、Ｌｓ２２　　インダクタ
　ｐ１１、ｐ１２、ｐ１３、ｐ１４、ｐ２１、ｐ２２、ｐ２３、ｐ３１、ｐ３２、ｐ３３　　並列腕共振子
　ｓ１１、ｓ１２、ｓ１３、ｓ２１、ｓ２２、ｓ３１、ｓ３２　　直列腕共振子
　ＳＷ１、ＳＷ１１、ＳＷ１２、ＳＷ１３、ＳＷ１４、ＳＷ２、ＳＷ２１、ＳＷ２２、ＳＷ２３、ＳＷ３、ＳＷ３５、ＳＷ３６、ＳＷ４、ＳＷ５、ＳＷ６、ＳＷ７、ＳＷ８　　スイッチ 5 antenna element 6 high frequency front end circuit 7 RF signal processing circuit (RFIC)
8 Baseband Signal Processing Circuit (BBIC) 4
9 communication apparatus 10, 10A, 10B, 20, 20A, 20B, 20C, 30, 40A, 50A, 60A, 500, 600 high-frequency filters 11, 11A, 11B, 11C, 11D, 12, 12A, 12Aa, 12B, 12D, 13A, 13B, 13C, 13D, 14A, 14B, 14C, 14D, 14F, 14G filters 15A, longitudinally coupled resonators 21, 21a, 21B, 21C, 21D, 22, 22a, 22B, 22C, 22D phase shifters 101, 102, 103, 104, 161, 201, 202, 203 Parallel arm circuit 110, 120 Input / output terminals 111, 112, 113, 114, 211, 212, 213 Impedance circuit 151 Series arm circuit 210 Switch group 220 Filter group 231 Transmission side switch 241, 242 243 transmission amplification circuit 251, 252, 253 reception side switch 261, 262, 263, 264 reception amplification circuit 271, 272 switch 321 IDT electrode 301a, 301b comb electrode 322 reflector 301 electrode film 302 piezoelectric substrate 301g contact layer 301h main electrode Layer 303 Protective layer 310a, 310b, 410 Electrode finger 311a, 311b, 411 Busbar electrode Cp11, Cp12, Cp14, Cp21, Cp22, Cp23, Cs21, Cs22 Capacitor Lp13, Lp21, Lp22, Ls21, Ls22 Inductor p11, p12, p13 p14, p21, p22, p23, p32, p33 parallel arm resonators s11, s12, s13, s21, s22, s31, s32 series arm resonators SW1, SW11, SW12, SW13, SW14, SW2, SW21, SW22, SW23, SW3, SW35, SW36, SW4, SW5, SW6, SW7, SW8 Switches

Claims (20)

  1. 　第１帯域を通過帯域とする高周波フィルタであって、
    　第１入出力端子と第２入出力端子とを結ぶ経路上に設けられた第１ノード、ならびに、前記経路上の前記第１ノードおよび前記第２入出力端子の間に設けられた第２ノードに接続された第１回路と、
    　前記第１ノードおよび前記第２ノードに接続された第２回路と、を備え、
    　　前記第１回路は、
    　　　前記第１帯域の一部の周波数を含む帯域であって、前記第１帯域より狭い帯域幅を有する第２帯域を通過帯域とする第１フィルタを有し、
    　　前記第２回路は、
    　　　前記第１帯域の一部の周波数を含む帯域であって、前記第１帯域より狭い帯域幅を有し、かつ、前記第２帯域の中心周波数より高周波数側に位置する第３帯域を通過帯域とする第２フィルタと、
    　　　前記第２フィルタの一方の端子に接続された第１移相器と、
    　　　前記第２フィルタの他方の端子に接続された第２移相器と、を有する、
    　高周波フィルタ。     A high frequency filter having a first band as a pass band,
    A first node provided on a path connecting a first input / output terminal and a second input / output terminal, and a second node provided between the first node on the path and the second input / output terminal A first circuit connected to
    And a second circuit connected to the first node and the second node,
    The first circuit is
    It has a first filter including a second band having a bandwidth narrower than the first band, which is a band including a partial frequency of the first band, as a pass band,
    The second circuit is
    A band including a partial frequency of the first band, having a narrower bandwidth than the first band, and a third band located higher than the center frequency of the second band as a pass band And the second filter
    A first phase shifter connected to one terminal of the second filter;
    And a second phase shifter connected to the other terminal of the second filter.
    High frequency filter.
  2. 　前記第１ノードおよび前記第２ノードの少なくとも一方から前記第２回路を単体で見た場合、前記第２回路のインピーダンスが極大となる特異点のうち最大のインピーダンスとなる周波数が、前記第２帯域の低域端の周波数から前記第２フィルタの通過帯域低域側のカットオフ周波数までの範囲に含まれるように、前記第１移相器および第２移相器の位相が調整されている、
    　請求項１に記載の高周波フィルタ。     When the second circuit is viewed alone from at least one of the first node and the second node, the frequency at which the maximum impedance among the singular points at which the impedance of the second circuit is maximized is the second band The phases of the first phase shifter and the second phase shifter are adjusted so as to fall within the range from the frequency of the low band end of the second filter to the cut-off frequency of the low band side of the second filter,
    The high frequency filter according to claim 1.
  3. 　前記第１ノードおよび前記第２ノードの少なくとも一方から前記第２回路を単体で見た場合、
    　前記第２帯域において、前記第１回路と前記第２回路とが同位相になっている周波数が少なくとも１箇所あり、
    　前記第３帯域において、前記第１回路と前記第２回路とが同位相になっている周波数が少なくとも１箇所ある、
    　請求項１または２に記載の高周波フィルタ。     When the second circuit is viewed alone from at least one of the first node and the second node,
    In the second band, there is at least one frequency at which the first circuit and the second circuit are in phase with each other,
    In the third band, there is at least one frequency at which the first circuit and the second circuit are in phase.
    The high frequency filter according to claim 1 or 2.
  4. 　前記第３帯域の低域端の周波数において、
    　前記第１フィルタおよび前記第２フィルタの位相差と、前記第１移相器および前記第２移相器の移相和との和が、３１２°以上かつ３６２°以下となるように、前記第１移相器および前記第２移相器の移相が調整されている、
    　請求項１～３のいずれか１項に記載の高周波フィルタ。     At the lower end frequency of the third band,
    The first of the first filter and the second filter such that the sum of the phase difference of the first filter and the second filter and the sum of phase shifts of the first phase shifter and the second phase shifter is 312 ° or more and 362 ° or less The phase shift of one phase shifter and the second phase shifter is adjusted,
    The high frequency filter according to any one of claims 1 to 3.
  5. 　前記経路は、
    　前記第１ノード、前記第１回路、および前記第２ノードを経由する第１経路と、
    　前記第１ノード、前記第２回路、および前記第２ノードを経由する第２経路と、を含み、
    　前記第１フィルタは、
    　前記第１経路上に設けられた第１直列腕回路と、
    　前記第１経路上に設けられたノードとグランドとに接続された第１並列腕回路と、を有し、
    　前記第２フィルタは、
    　前記第１移相器と前記第２移相器とを結ぶ前記第２経路上に設けられた第２直列腕回路と、
    　前記第２経路上に設けられたノードとグランドとに接続された第２並列腕回路と、を有し、
    　前記第１直列腕回路、前記第１並列腕回路、前記第２直列腕回路、および、前記第２並列腕回路のうちの少なくとも１つは、弾性波共振子を有し、
    　前記弾性波共振子の少なくとも１つは、
    　圧電性を有する基板上に形成されたＩＤＴ（ＩｎｔｅｒＤｉｇｉｔａｌ　Ｔｒａｎｓｄｕｃｅｒ）電極と、
    　反射器と、を有し、
    　前記弾性波共振子の少なくとも１つにおける前記ＩＤＴ電極の電極周期で定まる弾性波の波長をλとした場合、前記ＩＤＴ電極と前記反射器とのピッチは、０．４２λ以上かつ０．５０λ未満である、
    　請求項１～４のいずれか１項に記載の高周波フィルタ。     The route is
    A first path passing through the first node, the first circuit, and the second node;
    Including the first node, the second circuit, and a second path passing through the second node,
    The first filter is
    A first series arm circuit provided on the first path;
    A first parallel arm circuit connected to a node provided on the first path and a ground;
    The second filter is
    A second series arm circuit provided on the second path connecting the first phase shifter and the second phase shifter;
    A second parallel arm circuit connected to a node provided on the second path and the ground;
    At least one of the first series arm circuit, the first parallel arm circuit, the second series arm circuit, and the second parallel arm circuit has an elastic wave resonator,
    At least one of the elastic wave resonators is
    IDT (Inter Digital Transducer) electrodes formed on a substrate having piezoelectricity;
    And a reflector,
    When the wavelength of the elastic wave determined by the electrode period of the IDT electrode in at least one of the elastic wave resonators is λ, the pitch between the IDT electrode and the reflector is 0.42 λ or more and less than 0.50 λ. is there,
    The high frequency filter according to any one of claims 1 to 4.
  6. 　前記第１回路は、さらに、
    　前記第１ノードと前記第１フィルタとの間に接続された第１スイッチ素子と、
    　前記第２ノードと前記第１フィルタとの間に接続された第２スイッチ素子と、を有し、
    　前記第１スイッチ素子および前記第２スイッチ素子の導通および非導通の切り替えにより、前記第１帯域と前記第３帯域とが切り替わる、
    　請求項１～５のいずれか１項に記載の高周波フィルタ。     The first circuit further comprises
    A first switch element connected between the first node and the first filter;
    A second switch element connected between the second node and the first filter;
    The first band and the third band are switched by switching between conduction and non-conduction of the first switch element and the second switch element,
    The high frequency filter according to any one of claims 1 to 5.
  7. 　前記第１回路は、さらに、
    　前記第１スイッチ素子と前記第１フィルタとの間の接続ノードと、グランドとの間に接続され、前記第１スイッチ素子と排他的に導通および非導通が切り替えられる第３スイッチ素子と、
    　前記第２スイッチ素子と前記第１フィルタとの間の接続ノードと、グランドとの間に接続され、前記第２スイッチ素子と排他的に導通および非導通が切り替えられる第４スイッチ素子と、を有する、
    　請求項６に記載の高周波フィルタ。     The first circuit further comprises
    A third switch element connected between a connection node between the first switch element and the first filter, and the ground, and selectively switched between conduction and non-conduction with the first switch element;
    It has a connection node between the second switch element and the first filter, and a fourth switch element connected between the ground and the fourth switch element to switch between conduction and non-conduction exclusively with the second switch element. ,
    The high frequency filter according to claim 6.
  8. 　前記第２回路は、さらに、
    　前記第１ノードと前記第１移相器との間に接続された第５スイッチ素子と、
    　前記第２ノードと前記第２移相器との間に接続された第６スイッチ素子と、を有し、
    　前記第５スイッチ素子および前記第６スイッチ素子の導通および非導通の切り替えにより、前記第１帯域と前記第２帯域とが切り替わる、
    　請求項１～７のいずれか１項に記載の高周波フィルタ。     The second circuit further includes
    A fifth switch element connected between the first node and the first phase shifter;
    A sixth switch element connected between the second node and the second phase shifter;
    The first band and the second band are switched by switching between conduction and non-conduction of the fifth switch element and the sixth switch element.
    The high frequency filter according to any one of claims 1 to 7.
  9. 　前記第２回路は、さらに、
    　前記第５スイッチ素子と前記第１移相器との間の接続ノードと、グランドとの間に接続され、前記第５スイッチ素子と排他的に導通および非導通が切り替えられる第７スイッチ素子と、
    　前記第６スイッチ素子と前記第２移相器との間の接続ノードと、グランドとの間に接続され、前記第６スイッチ素子と排他的に導通および非導通が切り替えられる第８スイッチ素子と、を有する、
    　請求項８に記載の高周波フィルタ。     The second circuit further includes
    A seventh switch element connected between a connection node between the fifth switch element and the first phase shifter, and the ground, and selectively switched between conduction and non-conduction with the fifth switch element;
    An eighth switch element connected between a connection node between the sixth switch element and the second phase shifter, and the ground, and switched between conduction and non-conduction exclusively with the sixth switch element; Have
    The high frequency filter according to claim 8.
  10. 　さらに、
    　前記第１ノードおよび前記第２ノードに接続された第３回路を備え、
    　　前記第３回路は、
    　　　前記第１帯域と異なる第４帯域の一部の周波数を含む帯域であって、前記第４帯域より狭い帯域幅を有し、かつ、前記第２帯域の中心周波数より高周波数側に位置し、かつ、前記第３帯域と異なる第５帯域を通過帯域とする第３フィルタと、
    　　　前記第３フィルタの一方の端子に接続された第３移相器と、
    　　　前記第３フィルタの他方の端子に接続された第４移相器と、を有する、
    　請求項８または９に記載の高周波フィルタ。     further,
    A third circuit connected to the first node and the second node;
    The third circuit is
    A band including a partial frequency of a fourth band different from the first band, having a narrower bandwidth than the fourth band, and located higher than the center frequency of the second band, And a third filter having a fifth band different from the third band as a pass band,
    A third phase shifter connected to one terminal of the third filter;
    And a fourth phase shifter connected to the other terminal of the third filter.
    The high frequency filter according to claim 8 or 9.
  11. 　前記第１移相器は、
    　　前記第１ノードと前記第２フィルタの一方の端子とを結ぶ直列腕経路に配置された１以上のインダクタ、および、前記直列腕経路上のノードとグランドとの間に接続されたキャパシタを有する低域通過型フィルタ回路、または、
    　　前記直列腕経路に配置された１以上のキャパシタ、および、前記直列腕経路上のノードおよびグランドに接続されたインダクタで構成された高域通過型フィルタ回路である、
    　請求項１～１０のいずれか１項に記載の高周波フィルタ。     The first phase shifter is
    A low having one or more inductors arranged in a series arm path connecting the first node and one terminal of the second filter, and a capacitor connected between the node on the series arm path and the ground Low pass filter circuit, or
    A high pass filter circuit configured of one or more capacitors disposed in the series arm path, and an inductor connected to a node on the series arm path and the ground;
    The high frequency filter according to any one of claims 1 to 10.
  12. 　前記第２移相器は、
    　　前記第２ノードと前記第２フィルタの他方の端子とを結ぶ直列腕経路に配置された１以上のインダクタ、および、前記直列腕経路上のノードおよびグランドに接続されたキャパシタで構成された低域通過型フィルタ回路、または、
    　前記直列腕経路に配置された１以上のキャパシタ、および、前記直列腕経路上のノードおよびグランドに接続されたインダクタで構成された高域通過型フィルタ回路である、
    　請求項１～１１のいずれか１項に記載の高周波フィルタ。     The second phase shifter is
    A low pass comprised of one or more inductors arranged in a series arm path connecting the second node and the other terminal of the second filter, and a capacitor connected to the node on the series arm path and the ground Pass-through filter circuit, or
    A high pass filter circuit configured of one or more capacitors disposed in the series arm path, and an inductor connected to a node on the series arm path and the ground;
    The high frequency filter according to any one of claims 1 to 11.
  13. 　前記第１移相器および前記第２移相器は、キャパシタまたはインダクタからなるインピーダンス素子である、
    　請求項１～１０のいずれか１項に記載の高周波フィルタ。     The first phase shifter and the second phase shifter are impedance elements comprising a capacitor or an inductor.
    The high frequency filter according to any one of claims 1 to 10.
  14. 　前記第１フィルタおよび第２フィルタの少なくとも一方は、
    　　前記第１ノードと前記第２ノードとを結ぶ前記経路上に配置された直列腕回路と、
    　　前記経路上のノードおよびグランドに接続された並列腕回路と、を有し、
    　前記直列腕回路および前記並列腕回路の少なくとも一方は、
    　　共振子、および、第９スイッチ素子、を有し、
    　前記第９スイッチ素子の導通および非導通の切り替えにより、前記直列腕回路および前記並列腕回路の少なくとも一方の共振周波数および反共振周波数の少なくとも一方の周波数が切り替わる、
    　請求項１～１３のいずれか１項に記載の高周波フィルタ。     At least one of the first filter and the second filter is
    A series arm circuit disposed on the path connecting the first node and the second node;
    A parallel arm circuit connected to a node on the path and to ground,
    At least one of the series arm circuit and the parallel arm circuit is:
    A resonator and a ninth switch element,
    Switching of conduction and non-conduction of the ninth switch element switches at least one of the resonant frequency and the antiresonant frequency of at least one of the series arm circuit and the parallel arm circuit.
    The high frequency filter according to any one of claims 1 to 13.
  15. 　前記第１フィルタおよび前記第２フィルタの少なくとも一方は、弾性表面波フィルタ、弾性境界波フィルタ、および、バルク弾性波（ＢＡＷ：Ｂｕｌｋ　Ａｃｏｕｓｔｉｃ　Ｗａｖｅ）を用いた弾性波フィルタのいずれかである、
    　請求項１～１４のいずれか１項に記載の高周波フィルタ。     At least one of the first filter and the second filter is any of a surface acoustic wave filter, a boundary acoustic wave filter, and an elastic wave filter using a bulk acoustic wave (BAW).
    The high frequency filter according to any one of claims 1 to 14.
  16. 　前記第１フィルタおよび前記第２フィルタの少なくとも一方は、縦結合型共振器を有する、
    　請求項１５に記載の高周波フィルタ。     At least one of the first filter and the second filter has a longitudinally coupled resonator,
    The high frequency filter according to claim 15.
  17. 　第１移相器の位相と第２移相器の位相とは異なる、
    　請求項１６に記載の高周波フィルタ。     The phase of the first phase shifter is different from the phase of the second phase shifter,
    The high frequency filter according to claim 16.
  18. 　請求項１～１７のいずれか１項に記載の高周波フィルタを含む複数のフィルタを備え、
    　前記複数のフィルタの入力端子または出力端子は、共通端子に直接的または間接的に接続されている、
    　マルチプレクサ。     A plurality of filters including the high frequency filter according to any one of claims 1 to 17,
    Input terminals or output terminals of the plurality of filters are connected directly or indirectly to a common terminal,
    Multiplexer.
  19. 　請求項１～１７のいずれか１項に記載の高周波フィルタまたは請求項１８に記載のマルチプレクサと、
    　前記高周波フィルタまたは前記マルチプレクサに直接的または間接的に接続された増幅回路と、を備える、
    　高周波フロントエンド回路。     A high frequency filter according to any one of claims 1 to 17 or a multiplexer according to claim 18;
    An amplification circuit directly or indirectly connected to the high frequency filter or the multiplexer;
    High frequency front end circuit.
  20. 　アンテナ素子で送受信される高周波信号を処理するＲＦ信号処理回路と、
    　前記アンテナ素子と前記ＲＦ信号処理回路との間で前記高周波信号を伝達する請求項１９に記載の高周波フロントエンド回路と、を備える、
    　通信装置。     An RF signal processing circuit that processes a high frequency signal transmitted and received by the antenna element;
    20. The high frequency front end circuit according to claim 19, wherein the high frequency signal is transmitted between the antenna element and the RF signal processing circuit.
    Communication device.
PCT/JP2018/027433 2017-07-25 2018-07-23 High frequency filter, multiplexer, high frequency front-end circuit, and communication device WO2019021983A1 (en)

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CN117060880A (en) 2023-11-14

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