WO2010004586A2

WO2010004586A2 - Method and system for signal transmission and reception

Info

Publication number: WO2010004586A2
Application number: PCT/IN2009/000391
Authority: WO
Inventors: Kiran Kumar Kuchi; Deviraj Klutto Milleth Jeniston; Vinod Ramaswamy; Baskaran Dhivagar; Krishnamurthi Giridhar; Bhaskar Ramamurthi; Padmanabhan Madampu Suryasarman
Original assignee: Centre Of Excellence In Wireless Technology
Priority date: 2008-07-10
Filing date: 2009-07-09
Publication date: 2010-01-14
Also published as: WO2010004586A3

Abstract

Embodiments herein disclose a digitally precoded SC-FDMA scheme with low PAPR, where the scheme applies a constellation rotation to the M-ary input data sequence, before circularly convolving the rotated data sequence with a polynomial decoder. The data is then transformed to the frequency domain and then mapped. An N-point IIFT is then applied to the data to produce time domain samples.

Description

Method and System for Signal Transmission and Reception

TECHNICAL FIELD

[001] The embodiments herein relate to communication and, more particularly, to design and use of power efficient modulation techniques with low peak-to-average power ratio in wireless communication, and interference suppression in wireless systems

BACKGROUND

[002] Broadband wireless standards such as DEEE 802.16e and 3GPP LTE use Orthogonal Frequency-Division Multiple Access (OFDMA) technology in the downlink. OFDMA systems use multiple carriers to modulate and transmit data. An IDFT is applied on the modulated data tone to generate the transmitted time domain signal. The transmitted signal is a sum of several sinusoids modulated by random modulation symbols. Due to the summation of different sinusoids, the transmitted signal exhibits high peaks and nulls resulting in a high Peak- to-Average Power Ratio (PAPR). For power efficiency, it is generally desirable to drive the signals in the non-linear part of the Power Amplifier (PA) near the saturation point. High power back-off has to be applied to operate the PA in the linear range if OFDM signals having high PAPR are fed as input to non-linear PA' s. With insufficient power back-off, the PA output signals exhibits significant distortion and spectral regrowth occurs and out-of band emissions become significant. [003] Portable mobile terminals have limited power to transmit and receive signals. In cells with large cell radius, the uplink generally limits the coverage and data rate due to limited transmitted signal power available in portable mobile terminals. SUMMARY

[004] Embodiments herein disclose a method and system for precoding in a communication network, the method comprising steps of applying a constellation rotation to an input data sequence to create a constellation rotated data sequence; performing convolution on the constellation rotated data sequence using a polynomial precoder to create a precoded data sequence; transforming the precoded data sequence into frequency domain using an M-point DFT (Discrete Fourier Transform) to create a DFT output data sequence; performing mapping on the DFT output data sequence to create a mapped data sequence; and performing N-point IDFT (Inverse Discrete Fourier Transform) on the mapped data sequence. [005] Embodiments herein disclose a method and system for precoding in a communication network, the method comprising steps of applying a constellation rotation to an input data sequence to create a constellation rotated data sequence; transforming the constellation rotated data sequence into frequency domain using an M-point DFT (Discrete Fourier Transform) to create a DFT output data sequence; multiplying the DFT output data sequence with DFT of a polynomial precoder to create a precoded data sequence; performing mapping on the precoded data sequence to create a mapped data sequence; and performing N-point IDFT (Inverse Discrete Fourier Transform) on the mapped data sequence.

[006] Embodiments herein disclose a method and system for precoding in a communication network, the method comprising steps of transforming an input data sequence into frequency domain using an M-point DFT (Discrete Fourier Transform) to create a DFT output data sequence; shifting the DFT output data sequence by no samples to create a shifted data sequence; multiplying the shifted data sequence with DFT of a polynomial precoder to create a precoded data sequence; performing mapping on the precoded data sequence to create a mapped data sequence; and performing N-point IDFT (Inverse Discrete Fourier Transform) on the mapped data sequence.

[007] Embodiments herein disclose a method and system for transmitting a plurality of ASK (Amplitude Shift Keying) signals in a communication scheme, wherein the plurality of ASK signals are transmitted simultaneously in a fixed block, wherein the block is fixed in frequency and time.

[008] Embodiments herein disclose a method and a receiver for processing a received communication signal, the method comprising steps of applying an N-point DFT to the received signal to create a frequency domain signal; de-mapping the frequency domain signal to create a de-mapped frequency domain signal; de-shifting the de-mapped frequency domain signal to create a de-shifted frequency domain signal; taking complex conjugate and frequency reversal of the de-shifted frequency domain signal to create a modified frequency domain signal; filtering the de-shifted frequency domain signal and the modified frequency domain signal with a plurality of filter weights to obtain a filtered signal; and applying L-point IDFT to the filtered signal.

[009] Embodiments herein disclose a method and a receiver for processing a received communication signal, the method comprising steps of de-rotating the received signal to create a de-rotated signal; applying an N-point DFT to the de-rotated signal to create a frequency domain signal; de-mapping the frequency domain signal to create a de-mapped frequency domain signal; taking complex conjugate and frequency reversal of the de-mapped frequency domain signal to create a modified frequency domain signal; filtering the de-mapped frequency domain signal and the modified frequency domain signal with a plurality of filter weights to obtain a filtered signal; and applying L-point IDFT to the filtered signal. [0010] Embodiments herein disclose a method and receiver for processing pilot data in a communication signal, the method comprising steps of performing circular shifting on a first pilot data to create a circular shifted first pilot data; taking conjugate of the first pilot data to create a conjugate pilot data; frequency reversing the conjugate pilot data to create a frequency reversed pilot data; performing circular shifting of the frequency reversed pilot data to create a circular shifted second pilot data; and transmitting the circular shifted first pilot data and the circular shifted second pilot data.

[0011] Embodiments herein disclose a method and receiver for processing received pilot data in a received communication signal, the method comprising steps of performing circular de- shifting on a first received pilot data and a second received pilot data to create a first de-shifted received pilot data and a de-shifted second pilot data; performing channel estimation using the first received pilot data and the second received pilot data to obtain an estimated channel; taking conjugate of the de-shifted second pilot data to create a conjugate received pilot data; frequency reversing the conjugate received pilot data to create a frequency reversed received pilot data; estimating a first Noise and Interference Co- variance Matrix (NICM) of background noise and interference from the received pilot data using the de-shifted first pilot data and the frequency reversed received pilot data; and multiplying elements of the NICM using a frequency dependent weight.

[0012] These and other aspects of the embodiments herein will be better appreciated and understood when considered in conjunction with the following description and the accompanying drawings. BRIEF DESCRIPTION OF THE FIGURES

[0013] The embodiments herein will be better understood from the following detailed description with reference to the drawings, in which:

[0014] FIG. 1 illustrates a block diagram of an SC-FDMA transmitter with discrete data precoding, according to an embodiment herein;

[0015] FIG. 2 is a flowchart depicting a method to precode data in an SC-FDMA transmitter, according to an embodiment herein;

[0016] FIG. 3 illustrates a block diagram of an SC-FDMA transmitter with discrete data precoding done in frequency domain, according to an embodiment herein; [0017] FIG. 4 is a flowchart depicting a method to precode data in the frequency domain, in an SC-FDMA transmitter, according to an embodiment herein;

[0018] FIG. 5 illustrates a block diagram of a receiver for precoded SC-FDMA signals, according to an embodiment herein;

[0019] FIG. 6 is a flowchart depicting a method to de-modulate precoded SC-FDMA data, according to an embodiment herein;

[0020] FIG. 7 illustrates a block diagram of a receiver with signal de-rotation done in time domain for precoded SC-FDMA signals, according to an embodiment herein;

[0021] FIG. 8 illustrates an SC-FDMA pilot slot structure, according to an embodiment herein; [0022] FIG. 9 illustrates a block diagram of a receiver used for channel estimation, according to an embodiment herein; and

[0023] FIG. 10 is a flowchart depicting a method to estimate the channel from pilot sequences, according to an embodiment herein. DETAILED DESCRIPTION OF EMBODIMENTS

[0024] The embodiments herein and the various features and advantageous details thereof are explained more fully with reference to the non-limiting embodiments that are illustrated in the accompanying drawings and detailed in the following description. Descriptions of well- known components and processing techniques are omitted so as to not unnecessarily obscure the embodiments herein. The examples used herein are intended merely to facilitate an understanding of ways in which the embodiments herein may be practiced and to further enable those of skill in the art to practice the embodiments herein. Accordingly, the examples should not be construed as limiting the scope of the embodiments herein.

[0025] The embodiments herein disclose a digitally precoded Single Carrier-Frequency Division Multiple Access (SC-FDMA) scheme with low Peak-to-Average Power Ratio (PAPR) by a constellation rotation of the input data sequence and circularly convolving the constellation rotated data. Referring now to the drawings, and more particularly to FIGS. 1 through 10, where similar reference characters denote corresponding features consistently throughout the figures, there are shown embodiments.

[0026] FIG. 1 illustrates a block diagram of an SC-FDMA transmitter with discrete data precoding. The input data sequence a_k 101 is applied to a constellation rotation block 102. The input data sequence may be Binary Phase Shift Keying (BPSK) data, Q-ary Pulse Amplitude Modulation (PAM)/Amplitude Shift Keying (ASK) data or complex-valued Quadrature Amplitude Modulation (QAM) data. The constellation rotation block 102 rotates the signal constellation by a specific degree. For example, a_k may take values from the set b_k e [- (2Q-l),..-,3,-lX3,..,(2Q-l)] where Q is the constellation size and the constellation rotation block 102 may rotate the signal constellation by multiplying a_k 101 by j^k , where k is

the data index and j = V-TT The constellation rotated data may then be represented asc_k = j^kα_k . The constellation rotated data sequence is then circularly convolved using a polynomial precoder 103. The coefficients of the precoder 103 may take real or complex values. For example, the precoder 103 may have two taps and the tap weights may be given by:

_ 1, for k = 0,1

0, elsewhere.

Where k is the data index. If the precoder 103 introduces a delay to the data signal, then the tap weights of the precoder 103 may be given by:

1, foτ k = n,n + l

Pk ⁼ 0, elsewhere. Where n is delay introduced by the precoder 103 to the data signal. The precoded data can be represented as x_k = c_k ® p_k , where x_k is the precoded data, p_k are the precoder 103 tap weights

and c_k is the constellation rotated data. The precoded data is then transformed to frequency domain by taking an M-point Discrete Fourier Transform (DFT) 104 of the precoded data, where M is the data length. The M-point DFT 104 of the precoded data can be represented as

Where d,is the M-point DFT 104 output. The M-point DFT 104 output data is then mapped to specific subcarriers in the subcarrier mapping block 105. The DFT output may be mapped to a contiguous set of subcarriers and if there are any subcarriers remaining, then zero padding would be used to fill the remaining subcarriers. The DFT output may also be mapped to subcarriers distributed in the entire frequency domain with zeros filled in the unused subcarriers. After 0391

mapping the M-point DFT 104 output data to subcarriers, an N-Point Inverse DFT (IDFT) of the subcarriers is taken using an N-Point IDFT block 106. The N-Point IDFT 106 output can be represented as

s(t) = ∑ d_ιe ^J2πlAft , t G [o,τ], / G [N_{1 5} N₂]

Where / denotes the subcarrier index, Δ/ is the subcarrier width, and the signal spans over the time interval "T". Ν may also be chosen to be greater then M. 5(0 is the transmitted signal that is sent to the receiver. A cyclic prefix may also be added to -s(t) before being transmitted.

[0027] FIG. 2 is a flowchart depicting a method to precode data in an SC-FDMA transmitter. The input data sequence a_k 101 is applied to the constellation rotation block 102. The input data sequence may be BPSK data, Q-ary-PAM/ASK data or complex- valued QAM data. The constellation rotation block 102 rotates (201) the signal constellation by a specific degree. For example, a_k may take values from the set b_k e [- (2β -l),.. -,3,-1,1,3,..,(2Q- 1)] where Q is the constellation size and the constellation may be rotated (201) by multiplying a_k 101 by j^k , where k is the data index and j = V-I. The constellation rotated data may then be represented as c_k = j^ka_k . The constellation rotated data sequence is then circularly convolved (202) using the polynomial precoder 103. The precoded data can be represented as x_k = c_k ® p_k , where x_k is the precoded data, p_k are the precoder 103 tap weights and c_k is the constellation rotated data. The precoded data is then transformed to frequency domain by taking (203) an M- point DFT 104 of the precoded data, where M is the data length. The M-point Discrete Fourier Transform (DFT) 104 of the precoded data can be represented as

-j2πkl d, = ∑x_ke ^M , k = 0,1,2,..,M -I Where Ci₁ Is the M-point DFT 104 output. The M-point DFT 104 output data is then mapped

(204) to specific subcarriers in the subcarrier mapping block 105. The DFT output may be mapped to a contiguous set of subcarriers and if there are any subcarriers remaining, then zero padding would be used to fill the remaining subcarriers. The DFT output may also be mapped to subcarriers distributed in the entire frequency domain with zeros filled in the unused subcarriers. After mapping the M-point DFT 104 output data to subcarriers, an N-Point Inverse DFT (IDFT) of the subcarriers is taken (205). The N-Point IDFT 106 output can be represented as

s(0 = ∑ <V^;2;riΔ\ r e [O_f .T], Z e [N_{1 5} N₂ ]

Where / denotes the subcarrier index, Af is the subcarrier width, and the signal spans over the time interval "T". Ν may also be chosen to be greater then M. s(t) is the transmitted signal that is sent to the receiver. A cyclic prefix may also be added to s(t) before being transmitted. The various actions in the method may be performed in the order presented, in a different order, or simultaneously. Further, in some embodiments, some actions listed in FIG. 2 may be omitted. [0028] FIG. 3 illustrates a block diagram of an SC-FDMA transmitter with discrete data precoding done in frequency domain. The input data sequence a_k 101 is applied to a constellation rotation block 102. The input data sequence may be BPSK data, Q-ary- PAM/ASK data or complex- valued QAM data. The constellation rotation block 102 rotates the signal constellation by a specific degree. For example, a_k may take values from the set b_k e [- (2β-l),..-,3,-l,l,3,..,(2β-l)] where Q is the constellation size and the constellation rotation block 102 may rotate the signal constellation by multiplying a_k 101 by/ , where k is the data index and j = V-I. The constellation rotated data may then be represented as c_k = j^ka_k . The constellation rotated data sequence is then transformed to frequency domain by taking an M-point DFT 104 of the constellation rotated data, where M is the data length. The M-point DFT 104 of the constellation data can be represented as

Where d, is the output of M-point DFT 104. The M-point DFT 104 output data is then precoded using a frequency domain pulse shaping block 301. The frequency domain pulse shaping block 301 multiplies the elements of the M-point DFT 104 output using the coefficients of the DFT of the precoder. The coefficients of the frequency domain pulse shaping block 301 may take real or complex values. For example, the frequency domain pulse shaping block 301 coefficients may be represented as:

-jlTtd

P_ι =∑P_ke ^M / =0,l,2,.M-l k

Mathematically, the frequency domain pulse shaping is represented as:

d, =d_lPl, l=0XZ..M-l

The frequency domain pulse shaped data d_t is then mapped to specific subcarriers in the subcarrier mapping block 105. The pulse shaped data may be mapped to a contiguous set of subcarriers and if there are any subcarriers remaining, then zero padding would be used to fill the remaining subcarriers. The pulse shaped data may also be mapped to subcarriers distributed in the entire frequency domain with zeros filled in the unused subcarriers. After mapping the pulse shaped data to subcarriers, an N-Point IDFT of the subcarriers is taken using an N-Point IDFT block 106. The N-Point IDFT 106 signal is the transmitted signal that is sent to the receiver. A cyclic prefix may also be added to N-Point IDFT 106 signal before being transmitted.

[0029] FTG. 4 is a flowchart depicting a method to precode data in the frequency domain, in an SC-FDMA transmitter. The input data sequence a_k 101 is applied to the constellation rotation block 102. The input data sequence may be BPSK data, Q-ary-PAM/ASK data or complex- valued QAM data. The constellation rotation block 102 rotates (401) the signal constellation by a specific degree. For example, a_k may take values from the set b_k € [- (2β — 1),..— ,3,-l,l,3,..,(2β — 1)] where Q is the constellation size and the constellation rotation block 102 may rotate (401) the signal constellation by multiplying a_k 101 by j^k , where k is the data index and j = V-IT The constellation rotated data may then be represented as c_k = j^ka_k . The constellation rotated data sequence is then transformed to frequency domain by taking (402) an M-point DFT 104 of the constellation rotated data, where M is the data length. The M-point DFT 104 of the precoded data can be represented as

-jlTkl d, =∑c_ke ^M , fc =0,U-.,M-l k

Where d, is the M-point DFT 104 output. The M-point DFT 104 output data is then precoded using a frequency domain pulse shaping block 301. The frequency domain pulse shaping block 301 multiplies the elements of the M-point DFT 104 output using the coefficients of the DFT of the precoder. The coefficients of the frequency domain pulse shaping block 301 may take real or complex values. For example, the frequency domain pulse shaping block 301 coefficients may be represented as: Pι =∑P_ke ^M l = 0X2,.M-l k

Mathematically, the frequency domain pulse shaping is represented as:

O₁ =(I₁P₁₉ l=0XZ..M-l

The frequency domain pulse shaped data d_t is then mapped (404) to specific subcarriers in the subcarrier mapping block 105. The pulse shaped data may be mapped to a contiguous set of subcarriers and if there are any subcarriers remaining, then zero padding would be used to fill the remaining subcarriers. The pulse shaped data may also be mapped to subcarriers distributed in the entire frequency domain with zeros filled in the unused subcarriers. After mapping the pulse shaped data to subcarriers, an N-Point IDFT of the subcarriers is taken (405) using the N- Point IDFT block 106. The N-Point IDFT 106 signal is the transmitted signal that is sent to the receiver. A cyclic prefix may also be added to N-Point IDFT 106 signal before being transmitted. The various actions in the method may be performed in the order presented, in a different order, or simultaneously. Further, in some embodiments, some actions listed in FIG. 4 may be omitted.

[0030] FIG. 5 illustrates a block diagram of a receiver for precoded DFT-S-OFDMA signals. The transmitted DFT-S-OFDMA signals would be received at the receiver and the transmitted data can be obtained from the received signal. The desired signal component of the sampled time domain signal received at the receiver can be represented as:

s(k) = ∑j^ka_k ® p_k ® h_k (1) Where "k" is the discrete time index. An N-point DFT of the received signal is taken using an N-point DFT block 501 at the receiver. N may be chosen to be greater than the number of sub- carriers used at the transmitter. The N-point DFT signal is then passed to a sub-carrier de- mapping block 502 to retrieve the data that was mapped onto the subcarriers at the transmitter. The de-mapped low pass signal can be represented in frequency domain as: s(n) = x{n)p(ή)h{ή)

Where n is the discrete frequency index ranging from (0, M-I), p(n) is the precoding done at the transmitter, h(n) is the frequency domain propagation channel coefficients andx(ra) is the DFT of the constellation rotated data. The constellation rotation of the data was done at the transmitter. x(n) can be represented as

M-I -j2*n x(n) = ∑fa_ke ^M , « = 0,2,,,N-l. ₍₂)

Ic=O

x(n) can also be represented as

Σ M -I -~2Z.π/IkK ["-"ol M IY£ a_ke ^M , n_o= — (3)

<fc=0 ⁴ x(ή) can then be represented as : x(ή) = a(n - n₀) , or x(n + n₀) = a(n) (4)

Where addition is modulo-M addition and subtraction is modulo-M subtraction. a_k is the input data sequence at the transmitter and the DFT of the input data sequence can be represented as

M _i - jlπkn a (n) = ∑ a_ke ^M (5) fc = 0 The input data sequence at the transmitter has a real- valued constellation and a(ή) has conjugate symmetry. Since a(ή) has conjugate symmetry, a(ή) can also be represented as a (M -ή) = a(ή) (6)

If two a precoder with equal taps is used at the transmitter i.e.,

1, for k = 0,1

Pk - 0, elsewhere.

The frequency shifted precoder p(n) can be defined as

-]2m p(n) = p(n + n_o) = l - je ^M (7)

-j2m

Where P_n =I +e ^M , n = 0X2,.M-l In practice, the de-mapped low pass signal contains the desired signal transmitted from the transmitter, the Co-channel interference components which are generated by other signals and the noise components. The de-mapped low pass signal can be represented in terms of the desired signal, the Co-channel interference components and the noise components as:

Where X₁(H) denotes the DFT of /-th pi/2 rotated interference component, h,(n) denotes the propagation channel vectors of the CCI components, h(n) denotes the channel vector of the desired signal and x(n) is the DFT of the desired signal. If the receiver has 'Nr' antennae then the vectors y(n) , h(n) , jc(n) , p(n) ,h,(n) , jc,(n) and n(n) would be of size equal to 'Nr'. [0031] The de-mapped signal is then applied to a constellation de-rotation block 503. The constellation de-rotation block 503 de-rotates the signal constellation by a specific degree. For example, the constellation de-rotation block 503 may de-rotate the signal constellation by a factor j^~k , where k is the data index and j = V-I. In frequency domain, de-rotation can be done by applying a circular frequency de-shifting operation. If the frequency is de-shifted by H₀ tones, the frequency de-shifted signal can be represented as

y(n) = y(n + n_o) = h(n + n_o)x(n + n_o)p(n + n_o) + ∑h_ι(n + n_o)x,(n + n_o)p(n + n_o) + n(n + n_D)^'lϊ the

;=2 frequency de-shifting operation is defined as h(n) = h(n + n₀)

then the frequency de-shifted signal can be represented as

M y(n) = h(n)a(n)p(n) + ∑h, (n)α, (n)p(n) + ή(n) (8)

1=2

The frequency de-shifted signal is then applied to a conjugation and frequency reversal block 504. A complex-conjugation operation is performed on the frequency de-shifted signal and a frequency reversal is then performed on the signal. Frequency reversal is performed by changing the sign of the frequency value in the signal. Positive values of the frequency are changed to negative values and negative values are changed to positive values under moduIo-M operation. The frequency reversed and complex conjugated signal is sent to a filter 505. Also, a copy of the constellation de-rotated signal, without frequency reversal and complex conjugation, is applied to the filter 505. The constellation de-rotated signal and the frequency reversed and complex conjugated signal can be represented in vector form as:

-jlm

Where p^* (M - n) = 1 + je ^lr (10)

The signal can be written in compact form as:

^Mk ~ y(n) = h(n)a(n) + ∑h . (n)α, (ή) + n(ή) (H)

Desired Signal 1=2 l (n)

Where y(«) ,h(n) ,h_J(n)and n(n) denote corresponding vectors in equation (9). y(n)is the signal sent to the filter 505. The elements of y(n) would then be weighed by the filter 505 weights and then all the elements of the signal would be added. The filter weights are obtained by estimating the channel and noise-plus-interference covariance (NICM) by the use of pilots. Pilot sequences may be transmitted for supervisory, control, equalization, continuity, synchronization or reference purposes. A channel and Noise-plus Interference-Covariance- Matrix (NICM) estimation block 506 estimates the channel from the pilots received at the receiver from the transmitter. After the channel and NICM estimation block 506 estimates the channel, the filter weights would be determined. For example, the filter 505 used may be a Minimum Mean Square Error (MMSE) filter and the filter weights may be represented as:

w(n) = Q + h^* (n)R-¹ (n)h (n)]^"1 h^* (n)R:¹ (n) (12)

Where R₁., (n) is the estimated NICM. The NICM is a matrix whose elements are a measure of how much the noise and interference variables in the communication system change with respect to each other. By using the filter weights given by equation (12) the minimum Mean- Square-Error (MSE) can be represented as

^MSB - ⁼ ]ir∑ i ₊ h^*(nrR_Jk-ⁱ(»)h(») ⁽¹³⁾

The filtered signal is then applied to an M-Point IDFT block 507 and an M-Point IDFT of the filtered signal is taken. The M-Point IDFT gives the decision variable which is used to demodulate the original transmitted data. The bias introduced by the MMSE filter may also be corrected before demodulation. By including the full NICM, the receiver is able to suppress interference. The number of receiver antennas mentioned here is exemplary, and do not restrict the embodiments herein. It is obvious to a person of ordinary skill in the art that the embodiments disclosed above can be extended to a receiver with any number of antennas.

[0032] FIG. 6 is a flowchart depicting a method to de-modulate precoded SC-FDMA data. The transmitted SC-FDMA signals would be received at the receiver and the transmitted data can be obtained from the received signal. The desired signal component of the sampled time domain signal received at the receiver can be represented as:

s(k) = ∑j^ka_k ® p_k ® h_k k

Where "k" is the discrete time index. An N-point DFT of the received signal is taken (601) using an N-point DFT block 501 at the receiver. N may be chosen to be greater than the number of sub-carriers used at the transmitter. The N-point DFT signal is then passed (602) to a sub- carrier de-mapping block 502 to retrieve the data that was mapped onto the subcarriers at the transmitter. The de-mapped low pass signal can be represented in frequency domain as:

Where n is the discrete frequency index ranging from (O, M-I), p(n) is the precoding done at the transmitter, /i(n) is the frequency domain propagation channel coefficients andjc(n) is the DFT of the constellation rotated data. The constellation rotation of the data was done at the transmitter. x{n) can be represented as:

x(ή) = , B = (UJV-I.

x(n) can also be represented as:

M-I __2π*fa] _M x{n) = 2_^ a_ke ^M , n_o= —

Jt=O 4 x(n) can then be represented as : x(ri) = a(n - H₀) , or x(n + n_o) = a(n) Where addition is modulo-M addition and subtraction is modulo-M subtraction. a_k is the input data sequence at the transmitter and the DFT of the input data sequence taken along with the received signal at the receiver can be represented as:

_M _i - j2πkn

Jt = 0

The input data sequence at the transmitter has a real-valued constellation and a{ή) is conjugate symmetric. Since a{ή) is conjugate symmetric, a(n) can also be represented as: a\M - n) = a{ή)

If two tap precoder with equal taps is used at the transmitter i.e., 1, for ^ = 0,1

Pk ⁼ 0, elsewhere. The frequency shifted precoder p(ή) can be defined as: -j2m p(n) = p(n + n_o) = l - je ^M

The de-mapped low pass signal contains the desired signal transmitted from the transmitter, the Co-channel interference components which are generated by other signals and the noise components. The de-mapped low pass signal can be represented in terms of the desired signal, the Co-Channel Interference components and the noise components as:

M₁ y (n) = h(n)x(ri)p(ή)+ ∑h_t (ή)x, in)p{n) + n(n)

Desired Signal ^ ^I=2 _v , Noise

CCI

Where X₁(Ji) denotes the DFT of /-th pi/2 rotated interference component, h,(«) denotes the propagation channel vectors of the CCI components, h(/z) denotes the channel vector of the desired signal and x{n) is the DFT of the desired signal. If the receiver has 'Nr' antennae then the vectors y (n) , h(/z) , x(n) , p(ή) , h, (n) , x, (n) and n(n) would be of size equal to 'Nr' .

[0033] The de-mapped signal is then de-rotated using a constellation de-rotation block 503. The constellation de-rotation block 503 de-rotates (603) the signal constellation by a specific degree. For example, the constellation de-rotation block 503 may de-rotate (603) the signal constellation by a factor j^'k , where k is the data index and j = V-I. In frequency domain, the de-rotation (603) can be done by applying a circular frequency de-shifting operation. If the frequency is de-shifted by H₀ tones, the frequency de-shifted signal can be represented as:

M₁ y(n) = y(n + n₀) = h(n + n_o)x(n + n_o)p(n + n_o) + ∑h,(n + n_Q)x,(n + n_o)p(n + n_o) + n(n + n₀) If the

1=2 frequency de-shifting operation is defined as: h(n) = h(n + n₀) then the frequency de-shifted signal can be represented as:

y (n) = h(n)a(n)p(ή) + ∑h, (n)a, (n)p(n) + ή(n)

1=2

The frequency de-shifted signal is then conjugated and frequency reversed (604) using a conjugation and frequency reversal block 504. A complex-conjugation operation is performed (604) on the frequency de-shifted signal and a frequency reversal is then performed (604) on the signal. Frequency reversal is performed by changing the sign of the frequency value in the signal. Positive values of the frequency are changed to negative values and negative values are changed to positive values. The frequency reversed and complex conjugated signal is then filtered (605) using a filter 505. Also, a copy of the constellation de-rotated signal, without frequency reversal and complex conjugation, is applied to the filter 505. The constellation de- rotated signal and the frequency reversed and complex conjugated signal can be represented in vector form as:

Where p (M -ή) = \ + je M

The signal can be written in compact form as:

M₁ y (n) = h(ή)a(ή) + ∑h, (K)O₁ (n) + n(n)

Desired Signal 1=2

Hn)

Where y(n) ,h(n) ,h_j(n) and n(n) denote corresponding vectors in equation (14). y(n)is the signal sent to the filter 505. The elements of y(n) would then be weighed by the filter 505 weights and then all the elements of the signal would be added. The filter weights are obtained by estimating the channel by the use of pilots. A channel estimation block 506 estimates the channel from the pilots received at the receiver from the transmitter. After the channel and NICM estimation block 506 estimates the channel, the filter 505 weights would be determined. For example, the filter 505 used may be a Minimum Mean Square Error (MMSE) filter and the filter weights may be represented as:

w(n) = [l + h^*(n)R-¹(/i)h (n)]^'1^ (n)Rz\n) (15)

Where R₁₇(W)Is the estimated NICM. By using the filter weights given by equation (15) the minimum Mean-Square-Error (MSE) can be represented as:

The filtered signal is then applied to an M-Point IDFT block 507 and an M-Point IDFT of the filtered signal is taken (606). The M-Point IDFT gives the decision variable which is used to demodulate the original transmitted data. In certain implementations, the bias introduced by the MMSE filter may be corrected before demodulation. The various actions in the method may be performed in the order presented, in a different order, or simultaneously. Further, in some embodiments, some actions listed in FIG. 6 may be omitted.

[0034] FIG. 7 illustrates a block diagram of a receiver with signal de-rotation done in time domain for precoded SC-FDMA signals. The transmitted SC-FDMA signals would be received at the receiver and the transmitted data can be obtained from the received signal. The received data signal is applied to a constellation de-rotation block 503. The constellation de- rotation block 503 de-rotates the signal constellation by a specific degree. For example, the constellation de-rotation block 503 may de-rotate the signal constellation by multiplying the received signal by;^"* , where k is the data index and j = 4-T. The de-rotated signal is collected and applied to an N-point DFT block 701. The output of the N-point DFT block 701 is passed to a sub-carrier de-mapping block 502 to retrieve the data that was mapped onto the subcarriers at the transmitter. The de-mapped low pass signal contains the desired signal transmitted from the transmitter, the Co-Channel Interference components and the noise components. The de-mapped signal is then applied to a conjugation and frequency reversal block 504. A complex-conjugation operation is performed on the de-mapped signal and a frequency reversal is then performed on the signal. Frequency reversal is performed by changing the sign of the frequency value in the signal. Positive values of the frequency are changed to negative values using a Module-M operation. The negative values of the frequency are changed to positive values using a Module-M operation. The frequency reversed and complex conjugated signal is sent to a filter 505. Also, a copy of the constellation de-rotated signal, without frequency reversal and complex conjugation, is applied to the filter 505. The signal components would be weighed by the filter weights and then all the elements of the signal would be added. The filter weights are obtained by estimating the channel by the use of pilots. A channel estimation block 506 estimates the channel from the pilots received at the receiver from the transmitter. After the channel estimation block 506 estimates the channel, the filter 505 weights would be determined. The filtered signal is then applied to an M-Point IDFT block 507 and an M-Point IDFT of the filtered signal is taken. The M-Point IDFT gives the decision variable which is used to demodulate the original transmitted data. In certain implementations, the bias introduced by the MMSE filter may be corrected before demodulation. [0035] FIG. 8 illustrates an SC-FDMA pilot slot structure. Slot one and slot two show two slot structures used for transmitting the pilot sequence. Pilot sequences are transmitted for estimating the channel. Pilot sequences may also be transmitted for supervisory, control, equalization, continuity, synchronization or reference purposes. In slot one, Pl is the pilot sequence transmitted in the specific slot and Dl, D2, D3, D4, D5 and D6 are the data signals that are transmitted in the specific slots. Pl is a frequency domain pilot sequence and Pl can be represented as b, (ή) where / is the signal index. For example, the frequency domain pilot sequence used may also be a CAZAC type pilot sequence. The pilot sequence is circularly shifted to produce another sequence that can be represented as: £>,(!!) = £, (H-ZI₀). b, (n) is transmitted as the first pilot symbol. In slot two, P2 is the pilot sequence transmitted in the specific slot and D7, D8, D9, DlO, DIl and D12 are the data signals that are transmitted in the specific slots. A conjugated and frequency reversed version of b_t (n)is taken and the conjugated and frequency reversed signal can be represented as: b, = b] (M - ή) . The signal is then circularly shifted by n₀ tones to produce another sequence

represented as: b,(n — n₀) . The circularly shifted signal is then transmitted in the second pilot. The pilot sequence may also be transmitted without circularly shifting the conjugated and frequency reversed signal.

[0036] In another embodiment, a slot structure can be used for DFT-S-OFDMA signals wherein the same modulator is used for the pilots and the data. Time domain pilot sequences would be used as training sequences and the pilot sequences would have good auto correlation, cross correlation and low PAPR. The pilots and data would be transmitted using DFT-S- OFDMA systems and low PAPR would be maintained across all OFDM symbols including the pilots and the data. If the receiver jointly filters the complex and complex-conjugate parts of the baseband received signal, then efficient interference suppression could be achieved using the slot structure. [0037] FTG. 9 illustrates a block diagram of a receiver used for channel estimation. Pilot sequences are transmitted for estimating the channel. Pilot sequences may also be transmitted for supervisory, control, equalization, continuity, synchronization or reference purposes. At the receiver the received pilot sequence is applied to an N-point DFT block 501 and an N-point DFT of the received pilot sequence is obtained. The signal is then applied to a sub-carrier de- mapping block 901 to retrieve the pilot codes that were mapped onto the subcarriers at the transmitter. The pilot codes are then applied to a channel estimation block 902. The channel estimation block 902 estimates the propagation channel vectors from the pilots. If there is no CCI present in the channel, the propagation channel vectors in matrix form can be represented as:

Where h(«) is the propagation channel vector.

If CCI is present in the channel, then, to estimate the NICM the two pilots are sent to the NICM estimation block 903. The two pilots Pl and P2 applied to the NICM estimation block 903 can be represented as:

M₁ y_bl(n) = h(n)b(n-n₀) + γh_J(ή)b_l(n-n₀)+n_bl(ή), an_d

/=2

y _b2 (n) = h(n)b^* (M - n - n₀ ) + £ h _} (n)b^* (M - n - n₀ ) + n_b2 (n)

1=2 y_M(«) and y_fr2(«) are then circularly de-shifted by n_o, and the circularly de-shifted signals can be represented as:

M, y_b2(n) =y_b2{n+nJ =Hn)b\M-n)+∑%(n)b;(M-ή)+ή_b2(ή)

1=2

The second pilot sequence is then conjugated and frequency reversed and the conjugated and frequency reversed pilot sequence can be represented as:

The pilot sequences can be represented in vector form as:

NICM can then be estimated by

collecting the interference components from the pilot sequence. Interference components can be obtained by subtracting the desired signal components from the pilot signals. The interference components can be represented as:

Where ε is the error caused by the imperfect interference sample estimation. The NICM is can be estimated from the interference samples. If the channel has the same frequency response for all the sub-carriers spanning a resource block, then an estimate of R₁₁ («) can be represented as:

(17) The averaging operation in equation 17 is done by using samples collected from the entire resource block.

An estimate of R₁, (n) can be obtained by multiplying the two matrices R₍.,(n)andP(n) .

R_ώ (n) can then be represented as:

R,,.(n) = R,(n) o P(n) (18) Where P(n) = p(n)p^* (n) and

Where pis a column vector of size 2Nr and p(ή), p^*(M -ri) are repeated 'Nr' times. 'Nr' is the number of antennae at the receiver. The values of h(n)from equation 16 and R_n (ri) from equation 18 can be used in equation 12 to obtain the filter weights.

[0038] FIG. 10 is a flowchart depicting a method to estimate the channel from pilot sequences. Pilot sequences are transmitted for estimating the channel. Pilot sequences may also be transmitted for supervisory, control, equalization, continuity, synchronization or reference purposes. At the receiver an N-point DFT of the received pilot sequence is taken (1001) using the N-point DFT block 501. The signal is then applied to a sub-carrier de-mapping block 901 to retrieve (1002) the pilot codes that were mapped onto the subcarriers at the transmitter. The pilot codes are then applied to a channel estimation block 902. The channel estimation block 902 estimates (1003) the propagation channel vectors from the pilots. If there is no CCI present in the channel, the propagation channel vectors in matrix form can be represented as:

Where h(n) is the propagation channel vector. If CCI is present in the channel, then, to estimate the NICM the two pilots are sent to the NICM estimation block 903. The two pilots Pl and P2 applied to the NICM estimation block 903 can be represented as:

y _M (n) = h(ή)b(n - H₀ ) + J]h_;. («)£_z (n - n₀ ) + n_M (ή) , and

1=1

y_b2(n) = (ή)b](M -n -n_ϋ) + n_b2(ή)

y_fcl(n) and y_fc2(n) are then circularly de-shifted (1004) by n_o, and the circularly de-shifted signals can be represented as:

Af, yw(^rt) =y_M("+"_o) =h(n)Kn)+∑h_/(Φ_/(n)+n_M(n) , _an_d

1=2

The second pilot sequence is then conjugated and frequency reversed (1005) and the conjugated and frequency reversed pilot sequence can be represented as:

The pilot sequences can be represented in vector form as:

NICM can then be estimated

(1006) by collecting the interference components from the pilot sequence. Interference components can be obtained by subtracting the desired signal components from the pilot signals. The interference components can be represented as:

Where ε is the error caused by the imperfect interference sample estimation. The NICM can be estimated (1007) from the interference samples. If the channel has the same frequency response for all the sub-carriers spanning a resource block, then an estimate of R_n. («) can be represented as:

K - ⁽O = TrZ [^{i ( « )i * ( n )}] (20)

The averaging operation in equation 20 is done by using samples collected from the entire resource block. An estimate of R₁, (n) can be obtained by multiplying the two matrices

R₁₇ (ή) and P(n) . R₁₁ (n) can then be represented as:

R,,(n) = R,(n) o P(n) (21)

Where P(n) = p(n)p^*(n) and

Where pis a column vector of size 2Nr and pin) , p' (M -n) are repeated 'Nr' times. 'Nr' is the number of antennae at the receiver. The values of h(n) from equation 19 and R_n. (n) from equation 21 can be used in equation 12 to obtain the filter weights. The various actions in the method may be performed in the order presented, in a different order, or simultaneously. Further, in some embodiments, some actions listed in FIG. 10 may be omitted.

[0039] Low PAPR signaling and interference cancellation (IC) features can be implemented in SC-FDMA, DFT-S-OFDMA or OFDMA networks in an IC region allocated to serve cell edge users and/or control channel transmission. The communication network assigns a pre-defined IC region used exclusively for SC-FDMA, Q-ary PAM/ASK, DFT-S-OFDMA or OFDMA transmission either in Down Link (DL) or in the Up Link (UL). IC region may be composed of a predefined set of resource units. For example, the IC region may be a predefined set of PRUs or slots. The basic IC resource unit may be composed of a pair of slots which may be contiguous or distributed in time-frequency plane. The communication network assigns a pre specified time-frequency resource in the IC region. In the IC region, the co-channel base stations and mobile stations transmit signals in a synchronous manner. Different base stations and mobile stations may use different modulation sizes but the signal constellation used would have rectilinear signals. In an embodiment, the transmitter may use pi/2 rotated Q-ary PAM/ASK data and digital precoding. Information about the IC region may be communicated to each mobile station in a broadcast control channel. In the IC region, the receiver uses IC receivers. The receiver uses interference suppression MMSE algorithms to increase the data rate and reliability of reception. The slot structure which is depicted in FIG. 8 can be used for Q-ary PAM/ASK DFT-S-OFDMA or OFDMA transmission. [0040] The embodiments herein reduce the PAPR and allow the PA to be used near the saturation region. PA' s can be operated in the linear range without the need to have a high power back-off to operate the PA. Transmitted signal power can thus be increased in communication terminals having limited reserve power. Signals with greater power can be used in the Up-link and Down Link. The data rate of signals transmitted can also be increased. Embodiments herein can be implemented without using DFT and DDFT blocks. Also, the disclosed modulation and de-modulation schemes can be used for DFT-S-OFDMA systems.

[0041] The embodiments disclosed herein can be implemented through at least one software program running on at least one hardware device and performing network management functions to control the network elements. The network elements shown in Fig. 1 include blocks which can be at least one of a hardware device, or a combination of hardware device and software module.

[0042] The embodiment disclosed herein specifies a system for a digitally precoded SC- FDMA scheme with low PAPR. The mechanism allows transmission and reception of signals having low PAPR providing a system thereof. Therefore, it is understood that the scope of the protection is extended to such a program and in addition to a computer readable means having a message therein, such computer readable storage means contain program code means for implementation of one or more steps of the method, when the program runs on a server or mobile device or any suitable programmable device. The method is implemented in a preferred embodiment through or together with a software program written in e.g. Very high speed integrated circuit Hardware Description Language (VHDL) another programming language, or implemented by one or more VHDL or several software modules being executed on at least one hardware device. The hardware device can be any kind of device which can be programmed including e.g. any kind of computer like a server or a personal computer, or the like, or any combination thereof, e.g. one processor and two FPGAs. The device may also include means which could be e.g. hardware means like e.g. an ASIC, or a combination of hardware and software means, e.g. an ASIC and an FPGA, or at least one microprocessor and at least one memory with software modules located therein. Thus, the means are at least one hardware means and/or at least one software means. The method embodiments described herein could be implemented in pure hardware or partly in hardware and partly in software. The device may also include only software means. Alternatively, the invention may be implemented on different hardware devices, e.g. using a plurality of CPUs.

[0043] The foregoing description of the specific embodiments will so fully reveal the general nature of the embodiments herein that others can, by applying current knowledge, readily modify and/or adapt for various applications such specific embodiments without departing from the generic concept, and, therefore, such adaptations and modifications should and are intended to be comprehended within the meaning and range of equivalents of the disclosed embodiments. It is to be understood that the phraseology or terminology employed herein is for the purpose of description and not of limitation. Therefore, while the embodiments herein have been described in terms of preferred embodiments, those skilled in the art will recognize that the embodiments herein can be practiced with modification within the spirit and scope of the claims as described herein.

Claims

CLAIMS What is claimed is:

1. A method for precoding in a communication network, said method comprising steps of applying a constellation rotation to an input data sequence to create a constellation rotated data sequence; performing convolution on said constellation rotated data sequence using a polynomial precoder to create a precoded data sequence; transforming said precoded data sequence into frequency domain using an M-point DFT (Discrete Fourier Transform) to create a DFT output data sequence; performing mapping on said DFT output data sequence to create a mapped data sequence; and performing N-point IDFT (Inverse Discrete Fourier Transform) on said mapped data sequence.

2. The method, as claimed in claim 1, wherein said communication scheme is a DFT-S-FDMA (Discrete Fourier Transform - Spread - Frequency Division Multiple Access) scheme.

3. The method, as claimed in claim 1, wherein said input data sequence is a BPSK (Binary Phase Shift Keying) sequence.

4. The method, as claimed in claim 1, wherein said input data sequence is a real valued sequence.

5. The method, as claimed in claim 1, wherein said input data sequence is a Q-ary ASK (Amplitude Shift Keying) sequence, where Q is constellation size of said input data sequence.

6. The method, as claimed in claim 5, wherein said ASK sequence takes values from a set a_ki said set a_k comprising of values [-(2Q-I), -3,-1,1,3, ,(2Q-I)].

7. The method, as claimed in claim 1, wherein said input data sequence is a complex valued QAM (Quadrature Amplitude Modulation) constellation.

8. The method, as claimed in claim 1, wherein said constellation rotation is j\ where k is index of said data sequence and J = V-I.

9. The method, as claimed in claim 1, wherein said convolution is a circular convolution.

10. The method, as claimed in claim 1, wherein coefficients of said polynomial encoder take complex values.

11. The method, as claimed in claim 1, wherein coefficients of said polynomial encoder take real values.

12. The method, as claimed in claim 1, wherein coefficients of said polynomial encoder take real and complex values.

13. The method, as claimed in claim 1, wherein said polynomial encoder is a two tap polynomial encoder.

14. The method, as claimed in claim 13, wherein said two tap polynomial encoder takes values pk has the same value, for k=n, n+1 ; p_k = 0, elsewhere, where n is the filter delay.

15. The method, as claimed in claim 13, wherein said two tap polynomial encoder takes values Pk has the same value, for k=0, 1 ; p^ = 0 elsewhere.

16. The method, as claimed in claim 1, wherein said mapping is done to a contiguous set of M subcarriers.

17. The. method, as claimed in claim 1, wherein said mapping is done by distributing M subcarriers with equal spacing over entire frequency domain.

18. The method, as claimed in claim 1, wherein said mapping is done by distributing M subcarriers with unequal spacing over entire frequency domain.

19. The method, as claimed in claim 1, wherein said M is length of said input data sequence.

20. A method for precoding in a communication network, said method comprising steps of applying a constellation rotation to an input data sequence to create a constellation rotated data sequence; transforming said constellation rotated data sequence into frequency domain using an M- point DFT (Discrete Fourier Transform) to create a DFT output data sequence; multiplying said DFT output data sequence with DFT of a polynomial precoder to create a precoded data sequence; performing mapping on said precoded data sequence to create a mapped data sequence; and performing N-point IDFT (Inverse Discrete Fourier Transform) on said mapped data sequence.

21. The method, as claimed in claim 18, wherein said communication scheme is a DFT-S-FDMA (Discrete Fourier Transform - Spread - Frequency Division Multiple Access) scheme.

22. The method, as claimed in claim 18, wherein said input data sequence is a BPSK (Binary Phase Shift Keying) sequence.

23. The method, as claimed in claim 18, wherein said input data sequence is a Q-ary ASK (Amplitude Shift Keying) sequence, where Q is constellation size of said input data sequence.

24. The method, as claimed in claim 23, wherein said ASK sequence takes values from a set a^ said set a_k comprising of values [-(2Q-I), -3,-1,1,3, ,(2Q-I)].

25. The method, as claimed in claim 18, wherein said input data sequence is a real valued sequence.

26. The method, as claimed in claim 18, wherein said input data sequence is a complex valued QAM (Quadrature Amplitude Modulation) constellation.

27. The method, as claimed in claim 18, wherein said constellation rotation is j^k, where k is index of said data sequence and J = V-I.

28. The method, as claimed in claim 18, wherein coefficients of said polynomial encoder take complex values.

29. The method, as claimed in claim 18, wherein coefficients of said polynomial encoder take real values.

30. The method, as claimed in claim 18, wherein coefficients of said polynomial encoder take real and complex values.

31. The method, as claimed in claim 18, wherein said polynomial encoder is a two tap polynomial encoder.

32. The method, as claimed in claim 31, wherein said two tap polynomial encoder takes values p_k has the same value, for k=n, n+1 ; p_k = 0, elsewhere, where n is the filter delay.

33. The method, as claimed in claim 31, wherein said two tap polynomial encoder takes values P_k has the same value, for k=0, 1; pk= 0 elsewhere.

34. The method, as claimed in claim 18, wherein said mapping is done by distributing M subcarriers with equal spacing over entire frequency domain.

35. The method, as claimed in claim 18, wherein said mapping is done by distributing M subcarriers with unequal spacing over entire frequency domain.

36. The method, as claimed in claim 18, wherein said M is length of said input data sequence.

37. A method for precoding in a communication network, said method comprising steps of transforming an input data sequence into frequency domain using an M-point DFT (Discrete

Fourier Transform) to create a DFT output data sequence; shifting said DFT output data sequence by no samples to create a shifted data sequence; multiplying said shifted data sequence with DFT of a polynomial precoder to create a precoded data sequence; performing mapping on said precoded data sequence to create a mapped data sequence; and performing N-point IDFT (Inverse Discrete Fourier Transform) on said mapped data sequence.

38. The method, as claimed in claim 37, wherein said communication scheme is a DFT-S-FDMA (Discrete Fourier Transform - Spread - Frequency Division Multiple Access) scheme.

39. The method, as claimed in claim 37, wherein said input data sequence is a BPSK (Binary Phase Shift Keying) sequence.

40. The method, as claimed in claim 37, wherein said input data sequence is a Q-ary ASK (Amplitude Shift Keying) sequence, where Q is constellation size of said input data sequence.

41. The method, as claimed in claim 40, wherein said ASK sequence takes values from a set a^ said set a_k comprising of values [-(2Q-I), -3,-1,1,3, ,(2Q-I)].

42. The method, as claimed in claim 37, wherein said input data sequence is a complex valued QAM (Quadrature Amplitude Modulation) constellation.

43. The method, as claimed in claim 37, wherein said input data sequence is a real valued sequence.

44. The method, as claimed in claim 37, wherein said no is equal to M/4.

45. The method, as claimed in claim 37, wherein said shifting is a circular shifting.

46. The method, as claimed in claim 37, wherein coefficients of said polynomial encoder take complex values.

47. The method, as claimed in claim 37, wherein coefficients of said polynomial encoder take real values.

48. The method, as claimed in claim 37, wherein coefficients of said polynomial encoder take real and complex values.

49. The method, as claimed in claim 37, wherein said polynomial encoder is a two tap polynomial encoder.

50. The method, as claimed in claim 49, wherein said two tap polynomial encoder takes values pk has the same value, for k=n, n+1; p_k= 0, elsewhere, where n is the filter delay.

51. The method, as claimed in claim 49, wherein said two tap polynomial encoder takes values Pk has the same value, for k=0, 1 ; pk = 0 elsewhere.

52. The method, as claimed in claim 49, wherein said mapping is done by distributing M subcarriers with equal spacing over entire frequency domain.

53. The method, as claimed in claim 49, wherein said mapping is done by distributing M subcarriers with unequal spacing over entire frequency domain.

54. The method, as claimed in claim 49, wherein said M is length of said input data sequence.

55. A method for transmitting a plurality of ASK (Amplitude Shift Keying) signals in a communication scheme, wherein said plurality of ASK signals are transmitted simultaneously in a fixed block, wherein said block is fixed in frequency and time.

56. The method, as claimed in claim 55, wherein said ASK signal is a precoded and constellation rotated ASK signal.

57. The method, as claimed in claim 55, wherein said ASK signal is a DFT-S-OFDMA signal.

58. The method, as claimed in claim 57, wherein said DFT-S-OFDMA signal is a precoded and constellation rotated DFT-S-OFDMA signal.

59. The method, as claimed in claim 55, wherein said ASK signal is an OFDMA signal.

60. The method, as claimed in claim 55, wherein said plurality of ASK signals are transmitted from a plurality of base stations.

61. The method, as claimed in claim 55, wherein said plurality of ASK signals are transmitted from a plurality of mobile stations, wherein each of said plurality of mobile stations transmit only one ASK signal selected from said plurality of ASK signals.

62. A method for processing a received communication signal, said method comprising steps of applying an N-point DFT to said received signal to create a frequency domain signal; de-mapping said frequency domain signal to create a de-mapped frequency domain signal; de-shifting said de-mapped frequency domain signal to create a de-shifted frequency domain signal; taking complex conjugate and frequency reversal of said de-shifted frequency domain signal to create a modified frequency domain signal; filtering said de-shifted frequency domain signal and said modified frequency domain signal with a plurality of filter weights to obtain a filtered signal; and applying L-point IDFT to said filtered signal.

63. The method, as claimed in claim 62, wherein said communication signal is a DFT-S-FDMA (Discrete Fourier Transform - Spread - Frequency Division Multiple Access) signal.

64. The method, as claimed in claim 62, wherein said de-mapped frequency domain signal is de- shifted by applying a circular frequency de-shift of n_o tones.

65. The method, as claimed in claim 62, wherein said n₀ tones is M/4, where M is length of said received signal.

66. The method, as claimed in claim 62, wherein said de-shifted frequency domain signal contains a conjugate symmetric signal.

67. The method, as claimed in claim 62, wherein filtering of said de-shifted frequency domain signal and said modified frequency domain signal is done by a filter, wherein coefficients of said filter are obtained by minimizing mean square error between filtered sequence and input data.

68. The method, as claimed in claim 62, wherein filtering of said de-shifted frequency domain signal and said modified frequency domain signal is done by a filter, wherein coefficients of said filter are obtained by maximizing Signal to Interference Noise Ratio (SINR) at output of said filter.

69. The method, as claimed in claim 62, wherein said plurality of filter weights are obtained by estimating channel response from pilots present in said received signal.

70. The method, as claimed in claim 62, wherein said plurality of filter weights are obtained by estimating equivalent channel response and a modified channel response, wherein estimating said equivalent channel response and said modified channel response comprises steps of obtaining precoded estimated channel by applying said precoding to estimated channel; and applying constellation de-rotation to said precoded estimated channel to obtain said equivalent channel response; applying conjugation and frequency reversal to said equivalent channel response to obtain said modified channel response.

71. The method, as claimed in claim 62, wherein said plurality of filter weights are obtained by estimating covariance of background interference and thermal noise.

72. A method for processing a received communication signal, said method comprising steps of de-rotating said received signal to create a de-rotated signal; applying an N-point DFT to said de-rotated signal to create a frequency domain signal; de-mapping said frequency domain signal to create a de-mapped frequency domain signal; taking complex conjugate and frequency reversal of said de-mapped frequency domain signal to create a modified frequency domain signal; filtering said de-mapped frequency domain signal and said modified frequency domain signal with a plurality of filter weights to obtain a filtered signal; and applying L-point IDFT to said filtered signal.

73. The method, as claimed in claim 72, wherein said communication signal is a DFT-S-FDMA (Discrete Fourier Transform - Spread - Frequency Division Multiple Access) signal.

74. The method, as claimed in claim 72, wherein said received signal is de-rotated by π/2.

75. The method, as claimed in claim 72, wherein said received signal is de-rotated by j^"k.

76. The method, as claimed in claim 72, wherein said de-mapped frequency domain signal contains a conjugate symmetric signal.

77. The method, as claimed in claim 72, wherein filtering of said de-shifted frequency domain signal and said modified frequency domain signal is done by a filter, wherein coefficients of said filter are obtained by minimizing mean square error between filtered sequence and input data.

78. The method, as claimed in claim 72, wherein filtering of said de-shifted frequency domain signal and said modified frequency domain signal is done by a filter, wherein coefficients of said filter are obtained by maximizing Signal to Interference Noise Ratio (SINR) at output of said filter.

79. The method, as claimed in claim 72, wherein said plurality of filter weights are obtained by estimating channel response from pilots present in said received signal.

80. The method, as claimed in claim 72, wherein said plurality of filter weights are obtained by estimating equivalent channel response and a modified channel response, wherein estimating said equivalent channel response and said modified channel response comprises steps of obtaining precoded estimated channel by applying said precoding to estimated channel; and applying constellation de-rotation to said precoded estimated channel to obtain said equivalent channel response; applying conjugation and frequency reversal to said equivalent channel response to obtain said modified channel response.

81. The method, as claimed in claim 72, wherein said plurality of filter weights are obtained by estimating covariance of background interference and thermal noise.

82. A method for processing pilot data in a communication signal, said method comprising steps of performing circular shifting on a first pilot data to create a circular shifted first pilot data; taking conjugate of said first pilot data to create a conjugate pilot data; frequency reversing said conjugate pilot data to create a frequency reversed pilot data; performing circular shifting of said frequency reversed pilot data to create a circular shifted second pilot data; and transmitting said circular shifted first pilot data and said circular shifted second pilot data.

83. A method for processing received pilot data in a received communication signal, said method comprising steps of performing circular de-shifting on a first received pilot data and a second received pilot data to create a first de-shifted received pilot data and a de-shifted second pilot data; performing channel estimation using said first received pilot data and said second received pilot data to obtain an estimated channel; taking conjugate of said de-shifted second pilot data to create a conjugate received pilot data; frequency reversing said conjugate received pilot data to create a frequency reversed received pilot data; estimating a first Noise and Interference Co- variance Matrix (NICM) of background noise and interference from said received pilot data using said de-shifted first pilot data and said frequency reversed received pilot data; and multiplying elements of said NICM using a frequency dependent weight.

84. The method, as claimed in claim 83, wherein said second received pilot data is multiplied with a polynomial precoder, wherein said polynomial precoder is used in a transmitter used to transmit said received signal.

85. The method, as claimed in claim 83, wherein said frequency weights are obtained using DFT of said polynomial precoder.

86. The method, as claimed in claim 83, wherein estimating NICM comprises steps of collecting interference samples by subtracting desired signal contribution from said first de- shifted pilot data and said frequency reversed received pilot data; and estimating NICM from said interference samples.

87. A system for precoding in a communication network, said system comprising atleast one means adapted for applying a constellation rotation to an input data sequence to create a constellation rotated data sequence; performing convolution on said constellation rotated data sequence using a polynomial precoder to create a precoded data sequence; transforming said precoded data sequence into frequency domain using an M-point DFT

(Discrete Fourier Transform) to create a DFT output data sequence; performing mapping on said DFT output data sequence to create a mapped data sequence; and performing N-point IDFT (Inverse Discrete Fourier Transform) on said mapped data sequence.

88. The system, as claimed in claim 87, wherein said system is adapted to use DFT-S-FDMA (Discrete Fourier Transform - Spread - Frequency Division Multiple Access) scheme.

89. The system, as claimed in claim 87, wherein said system is adapted to use a BPSK (Binary Phase Shift Keying) sequence as said input data sequence.

90. The system, as claimed in claim 87, wherein said system is adapted to use a real valued sequence as said input data sequence.

91. The system, as claimed in claim 87, wherein said system is adapted to use a Q-ary ASK (Amplitude Shift Keying) sequence, where Q is constellation size of said input data sequence.

92. The system, as claimed in claim 91, wherein said system is adapted to take values from a set a_kfor said ASK sequence, said set a_t comprising of values [-(2Q-I), -3,-1,1,3, ,(2Q-I)].

93. The system, as claimed in claim 87, wherein said system is adapted to use a complex valued QAM (Quadrature Amplitude Modulation) constellation as said input data sequence input data sequence.

94. The system, as claimed in claim 87, wherein said system is adapted to use circular convolution.

95. The system, as claimed in claim 87, wherein said system is adapted to perform mapping to a contiguous set of M subcarriers.

96. The system, as claimed in claim 87, wherein said system is adapted to perform mapping by distributing M subcarriers with equal spacing over entire frequency domain.

97. The system, as claimed in claim 87, wherein said system is adapted to perform mapping by distributing M subcarriers with unequal spacing over entire frequency domain.

98. A system for precoding in a communication network, said system comprising atleast one means adapted for applying a constellation rotation to an input data sequence to create a constellation rotated data sequence; transforming said constellation rotated data sequence into frequency domain using an M- point DFT (Discrete Fourier Transform) to create a DFT output data sequence; multiplying said DFT output data sequence with DFT of a polynomial precoder to create a precoded data sequence; performing mapping on said precoded data sequence to create a mapped data sequence; and performing N-point IDFT (Inverse Discrete Fourier Transform) on said mapped data sequence.

99. The system, as claimed in claim 98, wherein said system is adapted to use DFT-S-FDMA (Discrete Fourier Transform - Spread - Frequency Division Multiple Access) scheme.

100. The system, as claimed in claim 98, wherein said system is adapted to use a BPSK (Binary Phase Shift Keying) sequence as said input data sequence.

101. The system, as claimed in claim 98, wherein said system is adapted to use a real valued sequence as said input data sequence.

102. The system, as claimed in claim 98, wherein said system is adapted to use a Q-ary ASK (Amplitude Shift Keying) sequence, where Q is constellation size of said input data sequence.

103. The system, as claimed in claim 102, wherein said system is adapted to take values from a set a_k for said ASK sequence,, said set a_k comprising of values [-(2Q-I), -3,-

1,1,3, ,(2Q-I)].

104. The system, as claimed in claim 98, wherein said system is adapted to use a complex valued QAM (Quadrature Amplitude Modulation) constellation as said input data sequence input data sequence.

105. The system, as claimed in claim 98, wherein said system is adapted to use circular convolution.

106. The system, as claimed in claim 98, wherein said system is adapted to perform mapping to a contiguous set of M subcarriers.

107. The system, as claimed in claim 98, wherein said system is adapted to perform mapping by distributing M subcarriers with equal spacing over entire frequency domain.

108. The system, as claimed in claim 98, wherein said system is adapted to perform mapping by distributing M subcarriers with unequal spacing over entire frequency domain.

109. A system for precoding in a communication network, said system comprising atleast one means adapted for transforming an input data sequence into frequency domain using an M-point DFT (Discrete Fourier Transform) to create a DFT output data sequence; shifting said DFT output data sequence by n₀ samples to create a shifted data sequence; multiplying said shifted data sequence with DFT of a polynomial precoder to create a precoded data sequence; performing mapping on said precoded data sequence to create a mapped data sequence; and performing N-point IDFT (Inverse Discrete Fourier Transform) on said mapped data sequence.

110. The system, as claimed in claim 109, wherein said system is adapted to use DFT-S- FDMA (Discrete Fourier Transform - Spread - Frequency Division Multiple Access) scheme.

111. The system, as claimed in claim 109, wherein said system is adapted to use a BPSK (Binary Phase Shift Keying) sequence as said input data sequence.

112. The system, as claimed in claim 109, wherein said system is adapted to use a real valued sequence as said input data sequence.

113. The system, as claimed in claim 109, wherein said system is adapted to use a Q-ary ASK (Amplitude Shift Keying) sequence, where Q is constellation size of said input data sequence.

114. The system, as claimed in claim 113, wherein said system is adapted to take values from a set ak for said ASK sequence, said set ak comprising of values [-(2Q-I), -3,-

1,1,3, ,(2Q-I)].

115. The system, as claimed in claim 109, wherein said system is adapted to use a complex valued QAM (Quadrature Amplitude Modulation) constellation as said input data sequence input data sequence.

116. The system, as claimed in claim 109, wherein said system is adapted to use circular convolution.

117. The system, as claimed in claim 109, wherein said system is adapted to perform mapping to a contiguous set of M subcarriers.

118. The system, as claimed in claim 109, wherein said system is adapted to perform mapping by distributing M subcarriers with equal spacing over entire frequency domain.

119. The system, as claimed in claim 109, wherein said system is adapted to perform mapping by distributing M subcarriers with unequal spacing over entire frequency domain.

120. The system, as claimed in claim 109, wherein said system is adapted to perform circular shifting.

121. A transmitting station for transmitting a plurality of ASK (Amplitude Shift Keying) signals in a communication scheme, wherein said transmitting station is adapted to transmit said plurality of ASK signals simultaneously in a fixed block, wherein said block is fixed in frequency and time.

122. The transmitting station, as claimed in claim 121, wherein said transmitting station is a base station.

123. The transmitting station, as claimed in claim 121, wherein said transmitting station is a mobile station.

124. A receiver comprising atleast one means adapted for applying an N-point DFT to a received signal to create a frequency domain signal; de-mapping said frequency domain signal to create a de-mapped frequency domain signal; de-shifting said de-mapped frequency domain signal to create a de-shifted frequency domain signal; taking complex conjugate and frequency reversal of said de-shifted frequency domain signal to create a modified frequency domain signal; filtering said de-shifted frequency domain signal and said modified frequency domain signal with a plurality of filter weights to obtain a filtered signal; and applying L-point EDFT to said filtered signal.

125. The receiver, as claimed in claim 124, wherein said receiver is adapted to receive a DFT- S-FDMA (Discrete Fourier Transform - Spread - Frequency Division Multiple Access) signal.

126. The receiver, as claimed in claim 124, wherein said receiver is adapted to de-shift said de-mapped frequency domain signal by applying a circular frequency de-shift of n_<, tones.

127. The receiver, as claimed in claim 124, wherein said receiver is adapted to obtain said plurality of filter weights by estimating channel response from pilots present in said received signal.

128. The receiver, as claimed in claim 124, wherein said receiver is adapted to obtain said plurality of filter weights by estimating equivalent channel response and a modified channel response, wherein for estimating said equivalent channel response and said modified channel response, said receiver is further configured to perform steps of obtaining precoded estimated channel by applying said precoding to estimated channel; and applying constellation de-rotation to said precoded estimated channel to obtain said equivalent channel response; applying conjugation and frequency reversal to said equivalent channel response to obtain said modified channel response.

129. The receiver, as claimed in claim 124, wherein said receiver is adapted to obtain said plurality of filter weights by estimating covariance of background interference and thermal noise.

130. A receiver comprising atleast one means adapted for de-rotating said received signal to create a de-rotated signal; applying an N-point DFT to said de-rotated signal to create a frequency domain signal; de-mapping said frequency domain signal to create a de-mapped frequency domain signal; taking complex conjugate and frequency reversal of said de-mapped frequency domain signal to create a modified frequency domain signal; filtering said de-mapped frequency domain signal and said modified frequency domain signal with a plurality of filter weights to obtain a filtered signal; and applying L-point IDFT to said filtered signal.

131. The receiver, as claimed in claim 130, wherein said receiver is adapted to receive a DFT- S-FDMA (Discrete Fourier Transform - Spread - Frequency Division Multiple Access) signal.

132. The receiver, as claimed in claim 130, wherein said receiver is adapted to obtain said plurality of filter weights by estimating channel response from pilots present in said received signal.

133. The receiver, as claimed in claim 130, wherein said receiver is adapted to obtain said plurality of filter weights by estimating equivalent channel response and a modified channel response, wherein for estimating said equivalent channel response and said modified channel response, said receiver is further configured to perform steps of obtaining precoded estimated channel by applying said precoding to estimated channel; and applying constellation de-rotation to said precoded estimated channel to obtain said equivalent channel response; applying conjugation and frequency reversal to said equivalent channel response to obtain said modified channel response.

134. The receiver, as claimed in claim 130, wherein said receiver is adapted to obtain said plurality of filter weights by estimating covariance of background interference and thermal noise.

135. A receiver comprising atleast one means adapted for performing circular shifting on a first pilot data to create a circular shifted first pilot data; taking conjugate of said first pilot data to create a conjugate pilot data; frequency reversing said conjugate pilot data to create a frequency reversed pilot data; performing circular shifting of said frequency reversed pilot data to create a circular shifted second pilot data; and transmitting said circular shifted first pilot data and said circular shifted second pilot data.

136. A receiver comprising atleast one means adapted for performing circular de-shifting on a first received pilot data and a second received pilot data to create a first de-shifted received pilot data and a de-shifted second pilot data; performing channel estimation using said first received pilot data and said second received pilot data to obtain an estimated channel; taking conjugate of said de-shifted second pilot data to create a conjugate received pilot data; frequency reversing said conjugate received pilot data to create a frequency reversed received pilot data; estimating a first Noise and Interference Co- variance Matrix (NICM) of background noise and interference from said received pilot data using said de-shifted first pilot data and said frequency reversed received pilot data; and multiplying elements of said NICM using a frequency dependent weight.

137. The receiver, as claimed in claim 136, wherein said receiver is adapted to multiply said first received pilot data with a polynomial precoder, wherein said polynomial precoder is used in a transmitter used to transmit said received signal.

138. The receiver, as claimed in claim 136, wherein said receiver is adapted to obtain said frequency weights using DFT of said polynomial precoder.