WO2002021712A1

WO2002021712A1 - A method of channel estimation and a system for implementing the method

Info

Publication number: WO2002021712A1
Application number: PCT/CN2000/000262
Authority: WO
Inventors: Yonghui Li; Daoben Li
Original assignee: Linkair Communications, Inc.
Priority date: 2000-09-05
Filing date: 2000-09-05
Publication date: 2002-03-14
Also published as: CN1208910C; AU2000269780A1; CN1460331A

Abstract

The present invention has disclosed a method of channel estimation used in high rate data transmission and high speed mobile environment as well as a system for implementing the method. The method comprise: data source inputted from sending terminal is spreaded spectrum by a pair of spread spectrum code groups which are orthogonal to each other so as to form two branch of signals which are transmitted by two channels, being sent after combination; at the receiving terminal, the received signals are separated into two channels by using the same pair of orthogonal spread spectrum code groups, and make channel estimation of the multiple phase matching in the way of multiple phase diversity through the two channels this can improve the accuracy of the system while not increasing the complexity of the system.

Description

Method for channel estimation and system for implementing the method

The present invention relates to the technical field of wireless spread-spectrum communication and digital mobile communication, and in particular, to a method for channel estimation in a high data rate transmission and high-speed mobile environment, and a signal transmission and reception system for implementing the method. Background of the invention

In modern high-speed mobile communications, due to the requirements for high data rates, research on high-dimensional phase shift keying (PSK) modulation systems has gradually become one of the core contents of wireless communications. However, multipath and large Doppler frequency shifts have been the main bottlenecks that limit the application of multilevel and multiphase modulation methods in high-speed mobile environments. In a high-speed mobile environment, the main factor affecting the performance of a PSK system is the effect of fast fading on the amplitude and phase of the PSK system. Especially in the deep fading environment, the amplitude deep fading will seriously degrade the performance of the system. To overcome these effects of fading channels, effective channel estimation must be performed. Especially in the third-generation mobile communication systems, mobile stations are required to adapt to the requirements of high data rate transmission and to withstand a deep fading environment with a Doppler frequency shift of about 500 Hz. For this reason, there are already several solutions in the prior art. For example, the main solution proposed by Qualcomm in the third-generation mobile communication standard IS-2000 is to use continuous pilot channel estimation, and the European Nokia, Errisson proposed in WCDMA is to use continuous pilot channels. And dedicated pilot channel joint estimation (for the specific scheme of Qualcomm continuous pilot, please refer to Physical Layer Standard for ANSI / TIA / EIA-95-B; for the joint pilot channel estimation scheme in WCDMA, please refer to 3GPP TS25.211).

The continuous pilot is only used in the downlink communication of the CDMA communication system from the base station to the mobile station. At this time, the transmitting end uses a dedicated channel to conduct the frequency signal, and sends it with the signals of other channels. Shoot. The signals of other channels at the receiving end are related to this pilot signal to eliminate the phase offset caused by the channel transmission process, so that each channel demodulates the original information. Continuous pilots can to some extent eliminate the effects of deep fading caused by high-speed movement. However, the pilot channel is required to be transmitted together with the signals of all other channels, and a dedicated channel is occupied. Therefore, the transmit power of the transmitter must be increased. Therefore, continuous pilot is only used in downlink communication (from base station to mobile station) of synchronous CDMA system. The dedicated pilot is to transmit a pilot symbol at a certain interval in each channel of the transmitting end, and use the channel parameters estimated by the pilot symbol to perform channel compensation on the data symbols following the pilot symbol to eliminate channel pairing. Effects of transmitted signals. It is easy to see from the principle of dedicated pilots that in a high-speed mobile environment, because the correlation between adjacent symbols is reduced, the compensation of data symbols following the pilot symbol channel estimation value is obviously Inaccurate, so this solution cannot effectively overcome the effects of deep fading on signal amplitude and phase, and cannot guarantee the application of higher-dimensional modulation methods in high-speed mobile environments. Summary of the invention

An object of the present invention is to provide a channel estimation method applied in a high-data-rate transmission and high-speed mobile environment, so as to overcome the technical problems and defects existing in the foregoing prior art solutions.

Compared with the continuous pilot channel estimation method and the dedicated pilot channel estimation method in the prior art, the method of the present invention can save the transmission power of the signal, can effectively overcome the influence of deep fading on the signal amplitude and phase, and can It guarantees the advantages of applying higher-dimensional modulation methods in high-speed mobile environments.

A channel estimation method applied to a high data rate transmission and high speed mobile environment includes the following steps: The data source input by a transmitting end is spread by a pair of mutually orthogonal spreading code groups to form a dual-channel transmission The two signals are sent after being combined; The received signal is separated into two channels by using the same orthogonal spreading code pair generated locally, and multi-phase matched channel estimation is performed by the dual-channel signal in a multi-phase diversity manner.

According to the technical solution of the present invention, the performing multi-phase matching channel estimation through dual-channel signals in a multi-phase diversity manner includes the following steps:

(1) despread the intermediate frequency received signal received by an antenna and output after radio frequency demodulation, using the orthogonal spreading code pair generated locally to separate the signal into two channels;

(2) Demodulate the signals of the two separated channels by using two locally generated 2M + 1 orthogonal carrier signals with a uniform phase interval, where M ranges from less than or equal to the modulation dimension;

(3) Separate the amplitude and phase of each demodulated signal separately;

(4) average the amplitude to determine the amplitude, and operate the phase to select the phase that minimizes the absolute value of the estimated difference;

(5) Combine the maximum ratio using the amplitude and phase determined in (4), judge and output the data.

This receiving method is actually a maximum likelihood receiver. Considering the complexity of the implementation, the multi-phase matching channel estimation in a multi-phase diversity manner through a two-channel signal may further include the following steps:

(1) despreading an intermediate frequency received signal received by an antenna and output after radio frequency demodulation, using the orthogonal spreading code group generated locally to separate the signal into two channels;

(2) using a pair of locally generated zero-phase orthogonal carrier signals to separately demodulate the signals of the two separated channels;

(3) Separate the amplitude and phase of the demodulated two-channel signal respectively;

(4) Average the amplitude to determine the amplitude; multiply the phases by 2M + 1 constant cos (/? _M ), sin ( _m ), me [-M, M] with uniform phase spacing, The value of Μ The range is less than or equal to the modulation dimension, and the phase that minimizes the value of | sin (^-^ ₂ + 2 ^) | is selected as the compensation phase, where ^ is the phase obtained by separating the Chinese channel signals; combining the compensation phases Compensate to the phase of any channel output, so as to obtain the phase required for decision;

(5) Use the amplitude and phase determined in (4) to perform maximum ratio combining, judge and output the data.

A signal transmitting and receiving system implementing the above method of the present invention includes a signal transmitting end and a receiving end, where the transmitting end includes a signal input part, a spread spectrum modulation part, a radio frequency modulation part, and an antenna, and the receiving end includes an antenna and a radio frequency demodulation part. A channel estimation part and a decision and output data part; the spread-spectrum modulation part of the transmitting end is a two-channel spread-spectrum modem, and the data source output by the signal input part is spread by a pair of mutually orthogonal spreading code groups To form two transmission signals; the two signals are sent to a signal combining device, and are combined and sent to an antenna for transmission; the channel estimation part of the receiving end is a polyphase matching channel estimator, which will be received by the antenna, The intermediate frequency received signal output after radio frequency demodulation is separated into two signal channels by using the same orthogonal "frequency code group", and the multi-phase matched channel estimation is performed through the two channel signals.

The above-mentioned multi-phase matched channel estimator includes a dual-channel despread demodulation device, a dual-channel amplitude and phase separation device, an arithmetic device, and a merging device; the signals received by the antenna, demodulated by radio frequency, and demodulated by the local intermediate frequency are output by The channel de-spreading and demodulation device performs channel separation and de-spreading and demodulation; the dual-channel amplitude and phase separation device separately separates the amplitude and phase of the dual-channel signal; the amplitude is determined by the arithmetic device, and the phase is determined by multiple phase diversity; The merging device 4 performs maximum ratio merging according to the determined amplitude and phase.

The novel channel estimation method for high data rate transmission and high speed mobile environment proposed by the present invention is implemented by orthogonal demodulation and Appropriate mathematical combination operations are used to obtain the parameters of some channels such as phase difference. The more types of diversity, the more obvious the performance improvement of the system. System performance is scored The effect of the number of times of the set can finally reach the performance of BPSK, so it can effectively overcome the effects of deep fading caused by high speed on the signal amplitude and phase. At the same time, applying this method can ensure the application of 16PSK, 32PSK in high-speed mobile environments 64PSK These higher-dimensional modulations.

The channel estimation method of the present invention implements channel estimation by using two independent channels at a base station and a mobile station, and adopting multiple phases for matching reception at the receiving end, so this channel estimation method can be called a multi-phase matching channel estimation The method is called CEMPM (Channel Estimation by using Multiple Phase Matching) method. The following describes the implementation of the CEMPM method through the expression of the formula and the accompanying drawings. From the derivation process, a person of ordinary skill in the prior art can easily understand the principle of channel estimation by the CEMPM method, and from the final estimation result, the superior performance of the CEMPM channel estimation method can be seen. Brief description of the drawings

FIG. 1 is a basic block diagram of a wireless communication system according to the method of the present invention.

Fig. 2 is a block diagram of a preferred embodiment of the method according to the present invention.

Fig. 3 is a block diagram of another preferred embodiment of the method according to the present invention. Mode of Carrying Out the Invention

Referring to FIG. 1, after the data source 10 input by the transmitting end passes through the two-channel signal modulators 11 and 12, the two signals formed are:

Λ (0 Q (/ ― kT) (J _k cos (m (t-kT) + p ₀ )-Q _k sin (fl) (t― kT) + φ ₀ 》 f ₂ (0 ^c 2 if― kT) (I _k cos ((t-kT) + (p ₀ ) + Q _k s {m {t-kT) + φ ₀ ;))

Where is the modulation information, E. Is the transmitted signal energy and is the initial phase. C, (t), C ₂ (t), and te [0, r] are mutually orthogonal spreading code groups with a code length of M. The two-channel signal modulators 11, 12 may be a baseband modulator or an intermediate frequency modulator.

To facilitate the following derivation, let us now:

Where ^ is the amplitude and phase of the signal points in the signal constellation, respectively. At this point, you have:

+ φ ₀ ) ― sin () sin (cy (t-kT) + φ ₀ ))

(0 = ∑ a _k C ₂ (t-kH _k ) cos (iy (t-^) + φ ₀ ) + ήχν (φ,) sin («(t-kT) + φ ₀ )) k

= ¾ "∑ ^a kC ₂ (t-kT) cos (» (t-kT) + φ ₀ -φ,)

k

The dual-channel signal formed at the transmitting end is transmitted by the transmitting antenna 15 after passing through the adder 13 and the radio frequency modulator 14.

Referring also to FIG. 1, after the transmitted signal passes through the fading channel, the signal is received by the receiving antenna 20, passes through the radio frequency demodulator 21, and enters the channel estimator 22. The signal is the sum of the following two formulas:

^ (= ^ ₀ ∑∑ <^ (卜-: r) ¾cos tz _c -kT) + ₀ +

i = \ k

r ₂ (= V ^ ∑ ∑ ^a kC ₂ (t-iT _c -kT) h _m cos {(t-iT _c- ^) + φ _0- , + φ ^ + η ^ ί)

'' ^{= 1 k}

Where is the effect of the fading channel on the signal amplitude and phase, £ 7 is the maximum multipath spread, I is the width of the spreading chip, T = MT _C , LT _C <T. Referring to FIG. 2, in the channel estimator 22 at the receiving end, the received signals are separated into two channels by two locally generated spreading codes, and then the two locally generated orthogonal carrier signals are used to compare the two signals. The signals of the channels are demodulated. Two channels pass through the same fading channel and are interfered by the same additive white Gaussian noise.

At the receiving end, the 2M + 1 sets of two orthogonal signals generated locally are used to demodulate the received signal. The value of M is less than or equal to the modulation dimension.

The locally generated 2M + 1 groups of two orthogonal signals are: ∑ C (t -kT- iT _c ) cos («(t-kT) + β, _η ), JC (t -kT- iT _c ) sin (6> (t-kT) + β _η ) kk

Among them, i represents the ith diversity signal, ζ · = 1,2, ..., 4 ^ =-M, ...,-1,0,1,2, ..., M for the NPSK signal, 0 p _m = ml ^ 3 = m27tlM, _M is the phase of the local carrier, so 2M + 1 carriers with a phase interval of 2πIM are used for tracking locally.

The expression of the signal after passing through two orthogonal despreading demodulator 31, 33 is:

(0 = 2E∑ a _k K∞s (fk + + β _ηι ) + ¾ (0

i = l, 2, ..., L

r _2i (0 =-2E∑ A. sin (^ + Αγ _Μ + β _ηι ) + (t)

k = \ r _u (0 = ¾ "∑" Α · cos (-Αγ _Μ -β „,) + ¾ · (

k = l i = l, 2, ..., L

r _2i (0 =-^ ∑ ^a kK sin (-Ay _ki -β _ηι ) + · (0

k = l

After despreading and demodulating the signals, after passing through the amplitude and phase separators 32 and 34, the estimated amplitude and phase of each signal are solved, and the expression is:

ξ _χι (t) = ∑ cos (^ + Δ¾ + β _ιη ) + n _u (t)

/ = 1,2, ..., (1) k = l (2)

Ccu (0 = ∑ cos (^-Δ¾ -β _ηι ) + n _u '(ί) (3)

k = l

ζ · = 1,2,…,

^{^{s2i (= Σ Sin (- Δ}} ¾ - m) + i (0 (4)

among them:

Then, the arithmetic unit 36 is used to find the average amplitude By (1) * (3) + (2) * (4), we get:

cos (2 (A ^ _;. + J3 _m )) = cos (2 (^ _fo . + _m )) + n _u (t) cos ( _-Δ. -β _ηι )

+ Κ (cos + Αφ _Μ + β, _η ) + n _2i (t) sin (-Δ¾-β _η ) (5) + n ₂ ' _i (t) sin (^ + Αφ _Μ + β „,) + n _u (t) * n _u '(t) + n _2i (t) * n ₂ ' _i (t)

ζ · = 1,2,…,

Put (2 "(3)-(1) * (4), at this time:

sin (2 (A _¾ + β _ηι )) = sin (2 (^. + β _Μ ))-n _u (t) sin (^-A _ki -β „,)

+ K (t) sin (^ + Αφ _Μ + β _η ) + n _2i (t) cos (^ 一 Α ^-β _η ) (6)-n ₂ ' _i (t) cos (^ + Αφ _Μ + β _Μ ) + n _2i (t) * n _u '(t)-n _u (t) * n ₂ ' _i (t)

= 1,2, ...,

Here, if we remember

A0 _mik = Δ <¾ + β _η (i = 1, 2 ,, .., L, m =-Μ, ...,-1,0, 1, ... M) (7) represents each local carrier The residual phase difference between the signal and the received signal estimates. In order to make a correct decision, we need to select a local carrier that can minimize the absolute value of the estimated difference to demodulate. Since the sine function is a monotonically increasing function between them, we only need to choose to make (7)

twenty two

The absolute value of the sine value of the equation can be demodulated by the local carrier with the smallest absolute value. This task is performed by the operator 35.

Finally, the arithmetic unit 37 combines the estimation results of each path with a maximum ratio,

Where Anin is the ^ value that minimizes the absolute value of the sine value of (7).

Assuming that the received signal has undergone the initial phase correction of the pilot signal, it is at this time that each data symbol after the pilot symbol has an estimated difference from the previous symbol. Because at this moment Ground has a total of 2M + 1 phases for tracking, so as long as the phase jump between two adjacent symbols does not exceed MΔ ^, no data error will occur at this time. Therefore, after the received signal is transformed by this algorithm, the phase jump of the tolerated signal is expanded to MΔ, that is, the performance of the NPSK system at this time.

N

Performance has been improved to i¾: For example, for 32PSK, as long as M = 16, use 33

M

In heavy phase diversity, the performance of the system is about the same as that of the BPSK system.

For the i-th received signal, a total of 2M + 1 signals are demodulated locally. We take the local carrier with the smallest absolute difference in the estimated difference between the received signals to demodulate. At this time, as long as the minimum Δ does not exceed the NPSK system The tolerance error is 2WN. At this time, no bit error occurs in the system, which is actually a maximum likelihood receiver. It can be seen that within the range of possible values, the larger the value of M, the higher the accuracy of the system. Referring to FIG. 3, we will explain from another preferred embodiment of the present invention. Although the phase diversity used at this time is large, it does not increase the complexity of the receiver, but only adds a limited number of decision variables. . That is, the technical solution of the present invention does not increase the complexity of the system while improving the performance of the system.

First, expand equation (2),

, ₂ ,) ⁼ ∑ ^sin + ^A ¾ + ^) = ∑ ^sin ½ + A¾) cos (^ _ff! ) + Cos ^ + A¾) sin (^ _ffi ) (8) k = lk = l

From the above formula, we will see that the estimated phase difference is actually only a combination of cos (+ Αφ,), sin (+ Αφ,) and cos (), sin (^), and _CO s (/? _M ) , sin (/? _m ) is just a constant, so as long as we know (^ (^ + ^ ₁ ), 5¾1 (^ + ^)), we can find the value of the above formula.

So in fact we only need to demodulate with a pair of zero-phase local orthogonal carriers, and the two-channel despreading demodulator 41, 43 and phase separator 42, 44 are used to complete this function. In the decision, the arithmetic unit 45 multiplies the output results of the arithmetic units 42 and 44 by some regular expressions (8). Number cos ( _m ), sin (5 _m ), and select the compensation phase that minimizes the absolute value of the formula (6). The arithmetic unit 47 is used to combine the compensation phase calculated by the arithmetic unit 45 with the phase output by the arithmetic unit 42 To form the phase required for decision. In this embodiment, the compensation phase is obtained by calculation, and the value of | _S i _n (A-% + ² ) | is selected to be the smallest, and is denoted as _min . The merging is performed by operations _C0S ^ + /? _Min :), ^! ^ + ^ Achieved.

Obviously, although we have adopted 2M + 1 phase diversity, it does not increase the complexity of the receiver, but only increases the limited decision variables, so it does not increase the complexity of the system.

After the correct phase compensation is performed, the judgment can be performed. When the feedback phase correction is performed, the phase error of the zero-phase carrier estimation still needs to be corrected.

Finally, the arithmetic unit 48 combines the estimation results of each path with a maximum ratio.

Claims

Claim

1. A channel estimation method applied in spread spectrum communication technology, characterized in that the method includes at least the following steps: spreading a data source input by a transmitting end by a pair of mutually orthogonal spreading code groups to form a silent The two signals transmitted in the channel are transmitted after being combined; at the receiving end, the same orthogonal spreading code pair generated locally is used to separate the received signal into two channels, and the multi-phase diversity is used to pass through the two signals. Channel signals are polyphase-matched for channel estimation.

2. The method according to claim 1, characterized in that said performing multi-phase matching channel estimation through dual-channel signals in a multi-phase diversity manner comprises the following steps:

(1) despreading an intermediate frequency received signal received by an antenna and output after radio frequency demodulation using the orthogonal spreading code pair generated locally to separate the signal into two channels;

(2) using two locally generated 2M + 1 groups of orthogonal carrier signals with uniform phase spacing to demodulate the signals of the two separated channels, where the value of M is less than or equal to the modulation dimension;

(3) Separate the amplitude and phase of each demodulated signal separately;

(4) averaging the amplitude to determine the amplitude, and operating the phase to select a phase that minimizes the absolute value of the estimated difference;

3. The method according to claim 2, characterized in that: in step (2), for an NPSK modulation signal, the phase interval is 2 M.

4. The method according to claim 2, characterized in that: the phase operation in step (4) is to determine the phase value by calculating | sii½ _lm -and selecting _cos ^, sin φ _1η which minimizes its value, Among them, φ _{2η is} each of the two-channel signals after demodulation. Phase of signal separation, m [-M, M].

5. The method according to claim 1, wherein the performing multi-phase matching channel estimation through dual-channel signals in a multi-phase diversity manner comprises the following steps:

(4) average the amplitude to determine the amplitude; multiply the phases by 2M + 1 constants ^C0S (A «), sin (A«), mG [-M, M) The value range of _M is less than or equal to the modulation dimension, and

The phase with the smallest value is the compensation phase, where are the phases obtained from the separation of the two channel signals, respectively; combining the compensation phases to the phase of any channel output to obtain the phase required for the decision;

6. The method according to claim 5, wherein: in step (4), for an NPSK modulation signal, the phase interval is 2WM.

7. The method according to claim 5, characterized in that: in step (5), when the simple compensation phase is merged and compensated to the phase of the first channel output, it is calculated by operation cos (^ + ^ _min ), ^ + ^ Is achieved, where 5 _min is the compensation phase.

8. The method according to claim 1, wherein: spreading the input data source may be baseband modulation, that is, spreading first and then modulating.

9. The method according to claim 1, wherein: spreading the input data source may be intermediate frequency modulation, that is, first modulating and then spreading.

10. The method according to claim 1, wherein: the combining comprises sending two signals transmitted by a silent channel into an adder, and the output of the adder is sent by an antenna through radio frequency modulation.

11. A signal transmitting and receiving system for implementing the method according to claim 1, comprising a signal transmitting end and a receiving end, wherein the transmitting end includes a signal input part, a spread spectrum modulation part, a radio frequency modulation part and an antenna, and the receiving end includes an antenna, The radio frequency demodulation part, the channel estimation part, and the decision and output data part are characterized by:

The spread-spectrum modulation part at the transmitting end is a two-channel spread-spectrum modem. The data source output by the signal input part is spread by a pair of mutually orthogonal spreading code groups to form two transmission signals. A signal combining device is input, and the combined signal is sent to the antenna for transmission. The channel estimation part of the receiving end is a polyphase matching channel estimator, which will receive the intermediate frequency received signal received by the antenna and output after RF demodulation, and then use the same orthogonality. The spreading code group is separated into two signal channels, and multi-phase matched channel estimation is performed through the two-channel signals.

12. The system according to claim 11, wherein:

The multi-phase matched channel estimator includes a dual-channel despread demodulation device, a dual-channel amplitude and phase separation device, an arithmetic device, and a merging device. The signals received by the antenna, demodulated by radio frequency, and demodulated by local intermediate frequency are processed by Chinese channel despreading and demodulation device performs channel separation and despreading and demodulation; dual-channel amplitude and phase separation device separates the amplitude and phase of the dual-channel signal; determines the amplitude by the operation device, and determines the phase by means of multiple phase diversity ; At the combining device, a maximum ratio combining is performed according to the determined amplitude and phase.

13. The system according to claim 11, wherein: the signal combining device is an adder.