TWI858446B - Direct current converter - Google Patents

Abstract

A direct current (DC) converter includes a boost converting circuit and a resonant converting circuit. The boost converting circuit is coupled to a DC power and boosts an input DC voltage of the DC power to a first output DC voltage. The resonant converting circuit is coupled to the boost converting circuit and includes an inverter circuit, a resonant circuit and a rectifier circuit. The inverter circuit converts the first output DC voltage into an alternating current (AC) voltage. The resonant circuit and the rectifier circuit convert the AC voltage into a second output DC voltage to supply the second output DC voltage to a load. The inverter circuit has a switching frequency, and the resonant circuit has a first resonant frequency and a second resonant frequency. When the switching frequency is between the first resonant frequency and the second resonant frequency, the inverter circuit operates in a zero voltage switching (ZVS), and the rectifier circuit operates in a zero current switching (ZCS). Therefore, the DC converter can be applied to the wide input voltage range and has the effect of the soft switching.

Description

DC Converter

The present disclosure relates to a DC conversion device, and more particularly to a DC conversion device with a high transformation ratio.

With the development of economy and the advancement of science and technology, the requirements for environmental protection and energy conservation are getting higher and higher in today's society. As the storage content of traditional fossil energy, such as oil, natural gas and coal, continues to decrease and it also causes great pollution to the environment, the development of renewable energy has gradually received attention. Renewable energy includes solar energy, wind power, geothermal energy and hydrogen energy. Solar energy and wind energy are easily restricted by the natural environment, and geothermal energy lacks good heat exchange technology, resulting in low thermal efficiency. Therefore, hydrogen energy is a more promising option among renewable energy.

In recent years, fuel-powered vehicles have gradually turned to electrification, but because pure electric systems are not fully popularized, many manufacturers have turned to fuel cells as an alternative. Fuel cells have the advantages of low environmental pollution, high conversion efficiency, low noise, fast startup, and large current output capacity, so fuel cells have become the first choice for power supply systems in recent years. Even though fuel cells have the above advantages, unlike general storage batteries, fuel cells themselves do not have storage capacity. The output voltage range will be affected by the output load size, resulting in a wide range of output voltage characteristics. In addition, fuel cells are mostly low-voltage outputs, which are not suitable for directly providing general loads. It can be seen from this that there is currently a lack of a conversion device on the market that can enable fuel cells to provide stable voltage to DC loads, so relevant industries are looking for solutions.

Therefore, the purpose of the present disclosure is to provide a DC converter, wherein the front stage uses a boost converter circuit to reduce input current ripples, and the rear stage uses a resonant converter circuit to have a flexible switching effect, and the resonant converter circuit is coupled in series with the boost converter circuit to achieve a high conversion ratio and be applicable to a wide input voltage range, while reducing the switching loss of the switch. This can solve the problem that the resonant converter in the prior art has a voltage regulation function but still cannot be directly used in the wide range of input voltages of current fuel cells.

According to an implementation method of the present disclosure, a DC conversion device is provided, which includes a boost conversion circuit and a resonant conversion circuit. The boost conversion circuit is coupled to a DC power source, and boosts an input DC voltage of the DC power source to a first output DC voltage. The resonant conversion circuit is coupled to the boost conversion circuit, and includes an inverter circuit, a resonant circuit, and a rectifier circuit. The inverter circuit converts the first output DC voltage to an AC voltage. The resonant circuit is coupled to the inverter circuit. The rectifier circuit is coupled to the resonant circuit. The resonant circuit and the rectifier circuit convert the AC voltage to a second output DC voltage to supply power to a load. The inverter circuit has a switch switching frequency, and the resonant circuit has a first resonant frequency and a second resonant frequency. When the switch switching frequency is between the first resonant frequency and the second resonant frequency, the inverter circuit operates in a zero voltage switching, and the rectifier circuit operates in a zero current switching.

Thus, when the switch switching frequency of the DC converter disclosed in the present invention is between the first resonant frequency and the second resonant frequency, the inverter circuit and the rectifier circuit can have a flexible switching function, thereby improving the loss caused by the rigid switching of the existing circuit and improving the overall conversion efficiency.

Other embodiments of the aforementioned embodiment are as follows: The aforementioned DC power source may have a first voltage terminal and a second voltage terminal, and the boost conversion circuit may include an input capacitor, a first boost inductor, a second boost inductor, a first power switch, a second power switch, a first diode, a second diode and an output capacitor. The two ends of the input capacitor are coupled to the first voltage terminal and the second voltage terminal respectively. One end of the first boost inductor is coupled to the first voltage terminal. One end of the second boost inductor is coupled to the first voltage terminal. A drain end of the first power switch is coupled to the other end of the first boost inductor. A drain end of the second power switch is coupled to the other end of the second boost inductor. An anode end of the first diode is coupled to the other end of the first boost inductor. An anode terminal of the second diode is coupled to the other terminal of the second boost inductor. One terminal of the output capacitor is coupled to a cathode terminal of the first diode and a cathode terminal of the second diode, and the other terminal of the output capacitor is coupled to a source terminal of the first power switch, a source terminal of the second power switch and the second voltage terminal.

Other embodiments of the aforementioned embodiment are as follows: a switch duty ratio of the aforementioned first power switch and the second power switch can be controlled by two first switch drive signals, and a phase difference between the two first switch drive signals is 180°. When the switch duty ratio is 0.5, a first current ripple flowing through the first boost inductor and a second current ripple flowing through the second boost inductor cancel each other out.

Other embodiments of the above-mentioned embodiment are as follows: The above-mentioned inverter circuit may include two first switches and two second switches. The two first switches are connected in series to form a first bridge arm, and the first bridge arm is coupled to the resonant circuit. The two second switches are connected in series to form a second bridge arm, and the second bridge arm is coupled to the resonant circuit. There is an AC voltage between the first bridge arm and the second bridge arm.

Other embodiments of the aforementioned embodiment are as follows: the aforementioned first switch and the second switch can be controlled by two second switch driving signals respectively and operate at a switch duty ratio of 50% at a switch switching frequency, and a phase difference between the first switch and the second switch is 180°.

Other embodiments of the aforementioned embodiment are as follows: The aforementioned resonant circuit may include a resonant capacitor, a resonant inductor, a magnetizing inductor and a transformer. One end of the resonant capacitor is coupled to the first bridge arm. One end of the resonant inductor is coupled to the other end of the resonant capacitor. One end of the magnetizing inductor is coupled to the other end of the resonant inductor, and the other end of the magnetizing inductor is coupled to the second bridge arm. The transformer has a primary side and a secondary side, the magnetizing inductor is connected in parallel to the primary side, and the secondary side is coupled to the rectifier circuit.

Other embodiments of the aforementioned embodiment are as follows: when the aforementioned switch switching frequency is between the first resonant frequency and the second resonant frequency, a resonant current flowing through the resonant inductor is equal to an excitation current flowing through the excitation inductor, and the transformer enters a decoupling region so that the rectifier circuit is turned off and operates at zero current switching.

Other embodiments of the aforementioned embodiment are as follows: The aforementioned rectifier circuit may include two first rectifier diodes, two second rectifier diodes and an output capacitor. The two first rectifier diodes are connected in series to form a third bridge arm, and the third bridge arm is coupled to a starting end of the secondary side of the transformer. The two second rectifier diodes are connected in series to form a fourth bridge arm, and the fourth bridge arm is coupled to an ending end of the secondary side of the transformer. The two ends of the output capacitor are respectively coupled to the starting end and the ending end of the secondary side.

Other embodiments of the aforementioned implementation method are as follows: The aforementioned DC conversion device may further include a voltage sampling module, a digital control module and a switch drive circuit. The voltage sampling module is coupled to the boost conversion circuit and the rectifier circuit, and captures the first output DC voltage and the second output DC voltage to generate a first feedback analog signal and a second feedback analog signal respectively. The digital control module is electrically connected to the voltage sampling module. The digital control module generates a plurality of first control signals according to the first feedback analog signal, and generates a plurality of second control signals according to the second feedback analog signal. The switch drive circuit is electrically connected to the digital control module, the boost conversion circuit and the inverter circuit. The switch driving circuit converts the first control signals into a plurality of first switch driving signals, and converts the second control signals into a plurality of second switch driving signals to control the boost conversion circuit and the inverter circuit.

Other embodiments of the aforementioned implementation method are as follows: The aforementioned digital control module may include a signal converter, a subtractor, and a proportional integral controller. The signal converter converts the first feedback analog signal into a first feedback digital signal, and converts the second feedback analog signal into a second feedback digital signal. The subtractor is electrically connected to the signal converter. The subtractor subtracts the first feedback digital signal from a plurality of first reference voltage signals to generate a plurality of first difference signals, and subtracts the second feedback digital signal from a plurality of second reference voltage signals to generate a plurality of second difference signals. The proportional integral controller is electrically connected to the subtractor. The proportional-integral controller receives the first difference signals and the second difference signals to generate the first control signals and the second control signals respectively.

The following will describe multiple embodiments of the present disclosure with reference to the drawings. For the sake of clarity, many practical details will be described together in the following description. However, it should be understood that these practical details should not be used to limit the present disclosure. In other words, in some embodiments of the present disclosure, these practical details are not necessary. In addition, in order to simplify the drawings, some commonly known structures and components will be depicted in a simple schematic manner in the drawings; and repeated components may be represented by the same number.

In addition, in this article, when a certain component (or unit or module, etc.) is "connected/linked" to another component, it may refer to that the component is directly connected/linked to the other component, or it may refer to that the component is indirectly connected/linked to the other component, that is, there are other components between the component and the other component. When it is explicitly stated that a certain component is "directly connected/linked" to another component, it means that there are no other components between the component and the other component. The terms first, second, third, etc. are only used to describe different components, and there is no restriction on the components themselves. Therefore, the first component can also be renamed as the second component. Moreover, the combination of components/units/circuits in this article is not a generally known, conventional or known combination in this field. Whether the components/units/circuits themselves are known cannot be used to determine whether their combination relationship is easy to be completed by ordinary knowledgeable people in the technical field.

Please refer to FIG. 1 and FIG. 2 together, wherein FIG. 1 is a block diagram of a DC converter 100 according to a first embodiment of the present disclosure, and FIG. 2 is a circuit diagram of the DC converter 100 of FIG. 1. As shown in FIG. 1 and FIG. 2, the DC converter 100 is coupled between a DC power source 10 and a load 20, wherein the DC power source 10 may be a fuel cell or an energy storage device for supplying power to an electric vehicle, and the load 20 may be a storage battery or a power supply system of the electric vehicle. The DC converter 100 includes a boost converter circuit 110 used as a front stage and a resonant converter circuit 120 used as a rear stage. The boost converter circuit 110 is coupled to the DC power source 10 and boosts an input DC voltage V _IN1 of the DC power source 10 to a first output DC voltage V _OUT1 . The resonant converter circuit 120 is coupled in series to the boost converter circuit 110 and includes an inverter circuit 121, a resonant circuit 122 and a rectifier circuit 123. The inverter circuit 121 converts the first output DC voltage V _OUT1 to an AC voltage v _AB . The resonant circuit 122 and the rectifier circuit 123 convert the AC voltage _vAB into a second output DC voltage _VOUT2 to power the load 20, and the load 20 includes a load resistor _RL2 , whose voltage is the second output DC voltage _VOUT2 . The inverter circuit 121 has a switching frequency, and the resonant circuit 122 has a first resonant frequency and a second resonant frequency. When the DC converter 100 is controlled by a digital control module and the switching frequency of the inverter circuit 121 is between the first resonant frequency and the second resonant frequency of the resonant circuit 122, the inverter circuit 121 operates in a zero voltage switching (Zero Voltage Switching; ZVS), and the rectifier circuit 123 operates in a zero current switching (Zero Current Switching; ZCS).

Specifically, the boost converter circuit 110 may be an interleaved boost converter. The resonant converter circuit 120 may be a full-bridge LLC resonant converter, a half-bridge LLC resonant converter, or a phase-shifted full-bridge LLC resonant converter. Therefore, the DC converter 100 of the present disclosure is a staggered boost converter cascade LLC resonant converter, which reduces the total input current ripple through the boost converter circuit 110 and controls the resonant converter circuit 120 to operate at an appropriate frequency (i.e., the switch switching frequency is between the first resonant frequency and the second resonant frequency), so that the inverter circuit 121 and the rectifier circuit 123 can have a flexible switching function, thereby improving the loss caused by the hard switching of the existing circuit and improving the overall conversion efficiency. The following will describe in detail the various circuit structures of the DC converter 100 of the present disclosure.

Specifically, the input DC voltage V _IN1 of the DC power source 10 may have a first voltage terminal (i.e., a positive voltage terminal) and a second voltage terminal (i.e., a negative voltage terminal). The boost conversion circuit 110 may include an input capacitor C _IN1 , a first boost inductor L ₁ , a second boost inductor L ₂ , a first power switch S ₁ , a second power switch S ₂ , a first diode D ₁ , a second diode D ₂ and an output capacitor C _OUT1 . Two ends of the input capacitor C _IN1 are respectively coupled to the first voltage terminal and the second voltage terminal of the input DC voltage V _IN1 . One end of the first boost inductor _L1 and one end of the second boost inductor _L2 are both coupled to the first voltage terminal of the input DC voltage V _IN1 . A drain end of the first power switch _S1 is coupled to the other end of the first boost inductor _L1 . A drain end of the second power switch _S2 is coupled to the other end of the second boost inductor _L2 . An anode end of the first diode _D1 is coupled to the other end of the first boost inductor _L1 . An anode end of the second diode _D2 is coupled to the other end of the second boost inductor _L2 . One end of the output capacitor C _OUT1 is coupled to a cathode end of the first diode D ₁ and a cathode end of the second diode D ₂ , and the other end of the output capacitor C _OUT1 is coupled to a source end of the first power switch S ₁ , a source end of the second power switch S ₂ and a second voltage end of the input DC voltage V _IN1 .

In the boost conversion circuit 110, the DC conversion device 100 adopts pulse width modulation (PWM) technology. A switch duty ratio (Duty Ratio) of the first power switch _S1 and the second power switch _S2 can be controlled by two first switch drive signals v _GS1 and v _GS2 (shown in FIG. 3 ) from a switch drive circuit. After PWM modulation, a phase difference between the two first switch drive signals v _GS1 and v _GS2 can be 180°, thereby controlling the energy stored in the first boost inductor _L1 and the second boost inductor _L2 to achieve the function of regulating the first output DC voltage V _OUT1 . When the switch duty ratio is 0.5, a first current ripple flowing through the first boost inductor _L1 and a second current ripple flowing through the second boost inductor _L2 cancel each other, i.e., ripple cancellation occurs, thereby reducing the ripple of a total input current i _IN1 (shown in FIG. 3 ) of the DC power source 10 .

In addition, the inverter circuit 121 may include an input capacitor C _IN2 , two first switches S ₃ , S ₄ and two second switches S ₅ , S ₆ . The input capacitor C _IN2 is connected in parallel with the output capacitor C _OUT1 . The input capacitor C _IN2 is coupled to a drain terminal of the first switch S ₃ and a drain terminal of the second switch S ₅ , and is coupled to a source terminal of the first switch S ₄ and a source terminal of the second switch S ₆ . The two first switches S ₃ , S ₄ are connected in series to form a first bridge arm B ₁ , and the first bridge arm B ₁ is coupled to the resonant circuit 122 . The two second switches S ₅ , S ₆ are connected in series to form a second bridge arm B ₂ , and the second bridge arm B ₂ is coupled to the resonant circuit 122 . There is an AC voltage v _AB between the first bridge arm _B1 and the second bridge arm _B2 ; in other words, the DC conversion device 100 converts the first output DC voltage V _OUT1 into an AC voltage v AB by controlling a switch duty ratio of the two first switches S ₃ and S ₄ and a switch duty ratio of the two second switches S ₅ and S ₆ , and outputs the AC voltage v _AB from the first bridge arm _B1 and the second bridge arm _B2 , wherein the AC voltage v _AB can be a square wave input voltage, which can be used as an input voltage of the rear-end resonant tank (i.e., the resonant circuit 122).

The resonant circuit 122 may include a resonant capacitor _Cr , a resonant inductor _Lr , a magnetizing inductor _Lm and a transformer TR. One end of the resonant capacitor _Cr is coupled to the first bridge arm _B1 . One end of the resonant inductor _Lr is coupled to the other end of the resonant capacitor _Cr . One end of the magnetizing inductor _Lm is coupled to the other end of the resonant inductor _Lr , and the other end of the magnetizing inductor _Lm is coupled to the second bridge arm _B2 . The transformer TR has a primary side and a secondary side, the primary side is connected in parallel with the magnetizing inductor L _m , and the secondary side is coupled to the rectifier circuit 123 , wherein the number of turns _NP of the primary side can be 16 turns, and the number of turns _NS of the secondary side can be 40 turns, so the turns ratio of the transformer TR can be 0.4, but the present disclosure is not limited thereto.

The rectifier circuit 123 may include two first rectifier diodes D ₃ , D ₄ , two second rectifier diodes D ₅ , D ₆ and an output capacitor C _OUT2 . The two first rectifier diodes D ₃ , D ₄ are connected in series to form a third bridge arm B ₃ , and the third bridge arm B ₃ is coupled to a starting end (i.e., a dot end) of the secondary side of the transformer TR. The two second rectifier diodes D ₅ , D ₆ are connected in series to form a fourth bridge arm B ₄ , and the fourth bridge arm B ₄ is coupled to an ending end of the secondary side of the transformer TR. The two ends of the output capacitor C _OUT2 are respectively coupled to the starting end and the ending end of the secondary side, and are respectively coupled to the two ends of the load resistor RL ₂ .

In the part of the resonant conversion circuit 120, the DC conversion device 100 adopts a pulse frequency modulation (PFM) technique. The first switch _S3 and the second switch _S6 can be respectively controlled by two second switch driving signals from the switch driving circuit to operate at a switching duty ratio of 50% at the switch switching frequency, and a phase difference between the first switch _S3 and the second switch _S6 can be 180°. Similarly, the first switch _S4 and the second switch _S5 can be controlled by the other two second switch driving signals from the switch driving circuit respectively and work at a switch duty ratio of 50% at the switch switching frequency, and a phase difference between the first switch _S4 and the second switch _S5 can be 180°, wherein the aforementioned switch duty ratio can include a dead time to prevent all switches from being turned on at the same time and causing circuit short circuit damage. Therefore, the DC conversion device 100 cuts the first output DC voltage V _OUT1 into an AC square wave (i.e., AC voltage v _AB ) to feed the resonant circuit ₁₂₂ through a full-bridge inverter (i.e., inverter circuit 121 ₎ composed of two first switches _S3 , _S4 and two second switches S5, S6. The resonant converter circuit 120 transmits the energy of the AC voltage v _AB from the primary side to the secondary side through the transformer TR, and then rectifies the other AC voltage on the secondary side into a second output DC voltage V _OUT2 for use by the load 20 through the rectifier circuit 123. In the resonant converter circuit 120, the inverter circuit 121 and the rectifier circuit 123 each achieve soft switching, thereby reducing switch switching loss and electromagnetic interference (EMI).

On the other hand, a total voltage gain of the DC converter 100 can be divided into a first voltage gain and a second voltage gain, wherein the first voltage gain is the conversion of the input DC voltage V _IN1 to the first output DC voltage V _OUT1 , and the second voltage gain is the conversion of the first output DC voltage V _OUT1 to the second output DC voltage V _OUT2 .

In the boost conversion circuit 110, an inductor current of the first boost inductor _L1 and an inductor current of the second boost inductor _L2 are both in continuous conduction mode (CCM), and the phase difference between the two inductor currents can be 180°. Therefore, when analyzing the first voltage gain, only one of the inductors needs to be analyzed. When the first power switch _S1 is turned on, the first boost inductor _L1 can be in an energy storage state, and the inductor current of the first boost inductor _L1 increases linearly; when the first power switch _S1 is turned off, the energy stored in the first boost inductor _L1 is released to the output capacitor C _OUT1 through the first diode _D1 , and the inductor current of the first boost inductor _L1 decreases linearly. When the boost converter circuit 110 is in a steady state, within a switching cycle, the following equations (1) and (2) can be derived according to the volt-second balance principle: (1); (2).

Wherein, V _IN1 is the input DC voltage, D is the switching duty ratio of the first power switch S ₁ and the second power switch S ₂ (0＜D＜1), _TS is the switching period, and G ₁ is the voltage gain of the boost converter circuit 110 (i.e., the first voltage gain). It can be seen from equation (2) that the voltage gain of the boost converter circuit 110 is related to the switching duty ratio. The voltage gain increases as the switching duty ratio increases, and is always greater than or equal to 1.

In the resonant converter circuit 120, a fundamental harmonic approximation (FHA) is most commonly used to analyze the equivalent model of the resonant circuit 122 to obtain the relationship between the voltage gain and the switching frequency of the resonant circuit 122. When the resonant circuit 122 operates at the resonant frequency point, a resonant current of the resonant inductor _Lr is similar to a sine wave, so only the fundamental component of the AC voltage _vAB and the excitation inductor _Lm and the primary voltage of the transformer TR need to be considered. According to FHA, the voltage gain of the resonant circuit 122 meets the following equation (3), and the voltage gain of the resonant converter circuit 120 meets the following equation (4): (3); (4).

Wherein, M _vr is the voltage gain of the resonant circuit 122, f _n is the normalized frequency, which is the ratio of the switch switching frequency to the resonant frequency, k is the ratio of the magnetizing inductance L _m to the resonant inductance L _r (the present disclosure sets k=6), Q is the circuit quality factor (the present disclosure sets Q=0.45), N is the turns ratio of the transformer TR, and G ₂ is the voltage gain of the resonant conversion circuit 120 (i.e., the second voltage gain). It can be seen that the second voltage gain is affected by various variables (f _n , k, Q), but the second output DC voltage V _OUT2 can still be stabilized by changing the switching frequency of the two first switches S ₃ , S ₄ and the two second switches S ₅ , S ₆ to achieve a voltage stabilization effect. Thus, the DC converter 100 of the present disclosure can boost the wide-range input (i.e., input DC voltage V _IN1 ) of the DC power source 10 to the first output DC voltage V _OUT1 through the front-stage boost converter circuit 110, and then boost it through the transformer TR of the rear-stage resonant converter circuit 120, and finally stabilize the second output DC voltage V _OUT2 through frequency modulation control, thereby achieving the effect of a DC-DC converter with a high conversion ratio. The following will use FIG. 2 in conjunction with subsequent figures to illustrate the detailed operation of each circuit of the DC converter 100 of the present disclosure.

Please refer to FIG. 2, FIG. 3, FIG. 4 and FIG. 5 together, wherein FIG. 3 is a circuit diagram showing the first power switch _S1 and the second power switch _S2 of the DC converter 100 of FIG. 2 being turned on, FIG. 4 is a circuit diagram showing the first power switch _S1 of the DC converter 100 of FIG. 2 being turned on and the second power switch _S2 being turned off, and FIG. 5 is a circuit diagram showing the first power switch S1 of the DC converter 100 of FIG. ₂ being turned off and the second power switch _S2 being turned on. In the circuit action analysis, the DC converter 100 of FIG. 2 operates in a steady state. The input capacitor C _IN1 and the output capacitor C _OUT1 are large capacitors, and the DC power source 10 is an ideal DC power source. Each switch has an intrinsic diode and a parasitic capacitance, but the on-resistance and the parasitic capacitance are both negligible. The transformer TR, each inductor, each capacitor and each diode are all ideal components, and parasitic components and series equivalent resistance are all negligible. An inductance value of the first boost inductor _L1 is the same as an inductance value of the second boost inductor _L2 .

The boost converter circuit 110 is a combination of two boost circuits. The switch duty ratio (hereinafter referred to as D) of the first power switch _S1 and the second power switch _S2 is the same. The two first switch drive signals v _GS1 and v _GS2 have a 180° phase difference. When the input DC voltage V _IN1 is twice greater than the first output DC voltage V _OUT1 , D is less than 0.5; when the input DC voltage V _IN1 is twice less than the first output DC voltage V _OUT1 , D is greater than 0.5. The operation state of the boost converter circuit 110 can be divided into three parts. Since the circuit operation corresponding to D less than 0.5 is the same as the circuit operation corresponding to D greater than 0.5, the following only describes D greater than 0.5.

As shown in FIG. 3 , the first power switch S ₁ and the second power switch S ₂ are both in the on state, and the total input current i _IN1 can be split into a first inductor current i _L1 and a second inductor current i _L2 , and they flow from the input DC voltage V _IN1 through the first boost inductor L ₁ and the second boost inductor L ₂ to store energy. The first diode D ₁ and the second diode D ₂ are both reverse biased and cut off, and the output capacitor C _OUT1 releases energy to an output resistor R _OUT1 . As shown in FIG. 4 , the first power switch S ₁ is in the on state, and the second power switch S ₂ is in the off state. The second diode D ₂ is forward biased and turned on, and the second inductor current i _L2 flowing through the second boost inductor L ₂ releases energy to the output capacitor C _OUT1 and the output resistor R _OUT1 . The first boost inductor L ₁ stores energy. As shown in FIG. 5 , the first power switch S ₁ is in the off state, and the second power switch S ₂ is in the on state. The first diode D ₁ is forward biased and turned on, and the first inductor current i _L1 flowing through the first boost inductor L ₁ releases energy to the output capacitor C _OUT1 and the output resistor R _OUT1 . The second boost inductor L ₂ stores energy.

Furthermore, since the two inductors have the same value, the first current ripple flowing through the first boost inductor _L1 and the second current ripple flowing through the second boost inductor _L2 are also the same and meet the following equation (5): (5).

in, is the current ripple of each phase, is the first current wave, is the second current ripple, V _IN1 is the input DC voltage, D is the switching duty ratio of the first power switch S ₁ to the second power switch S ₂ , and f _s is the switching frequency of the first power switch S ₁ to the second power switch S _2. The total input current i _IN1 is the sum of the first inductor current i _L1 and the second inductor current i _L2 (i.e., i _IN1 =i _L1 +i _L2 ). When D ≤ 0.5, the total input current ripple meets the following formula (6); when D > 0.5, the total input current ripple meets the following formula (7): (6); (7).

in, is the total input current ripple, _Ts is the switching period of the first power switch _S1 and the second power switch _S2 . Then, according to equations (5) to (7), the following equation (8) can be derived: (8).

Wherein, λ is a ripple coefficient, which indicates the ability to suppress the total input current ripple. When λ is smaller, it means the suppression ability is better, and the total input current ripple is smaller. It can be seen that the ripple coefficient first decreases and then increases as the switch duty ratio gradually increases. When the switch duty ratio is 0.5, the first current ripple and the second current ripple cancel each other. In this way, the total input current ripple of the DC power supply 10 can be reduced, thereby increasing the output power.

Please refer to FIG. 2 and FIG. 6 together, wherein FIG. 6 is a voltage gain curve diagram of the resonant circuit 122 of the DC converter 100 of FIG. 2. As shown in FIG. 6, the present disclosure can be used to perform theoretical analysis on the resonant circuit 122 through a mathematical simulation software (Mathcad) to obtain a voltage gain curve diagram of the resonant circuit 122 at different values of the circuit quality factor Q, wherein the vertical axis represents the voltage gain of the resonant circuit 122, and the horizontal axis represents the normalized frequency (f _n ), and the normalized frequency is the switch switching frequency f _{s of the inverter circuit 121 divided by the first resonant frequency f r (} i.e., f _n =f _s /f _r ). It should be noted that the first resonant frequency f _r can be formed by the resonant capacitor _Cr and the resonant inductor L _r , and the second resonant frequency f _m can be formed by the resonant capacitor _Cr , the resonant inductor L _r and the magnetizing inductor L _m , and they meet the following equations (9) and (10): (9); (10).

The resonant converter circuit 120 of the present disclosure adopts a resonant parameter configuration and can be divided into four working regions by adjusting the switch switching frequency _fs and in combination with the voltage gain curve of FIG. 6 .

First, when the switch switching frequency _fs is greater than the first resonant frequency _fr (i.e., _fs > _fr ), the voltage gain curve operates in the first region R1. At this time, the input impedance of the resonant converter circuit 120 is inductive, and the two first switches _S3 , _S4 and the two second switches _S5 , _S6 of the inverter circuit 121 operate in ZVS. However, because the switching cycle is shorter than the resonant cycle, the resonant current of the resonant inductor _Lr is not equal to the excitation current of the excitation inductor _Lm , and the energy is continuously transmitted to the secondary side of the transformer TR, resulting in the current on the secondary side being in a continuous state, so the secondary side cannot achieve the ZCS function.

Secondly, when the switch switching frequency _fs is equal to the first resonant frequency _fr (i.e., _fs = _fr ), the voltage gain curve operates in the second region R2. At this time, the resonant inductor _Lr and the resonant capacitor _Cr resonate in series, the input impedance of the resonant converter circuit 120 is inductive, the inverter circuit 121 operates in ZVS, and the rectifier circuit 123 operates in ZCS. The efficiency of the resonant converter circuit 120 is the highest. The voltage gain can be 1 and is not affected by the load 20.

Third, when the switching frequency _fs is between the first resonant frequency _fr and the second resonant frequency _fm (i.e. _fm < _fs < _fr ), the voltage gain curve operates in the third region R3. At this time, the input impedance of the resonant converter circuit 120 is inductive, and the inverter circuit 121 operates in ZVS. However, because the switching cycle is longer than the resonant cycle, when the magnetizing inductor _Lm participates in the resonance, the resonant current of the resonant inductor _Lr is equal to the magnetizing current of the magnetizing inductor _Lm . The transformer TR enters the decoupling region, and in the decoupling region, the primary side of the transformer TR cannot transfer energy to the secondary side, making the current on the secondary side discontinuous. Therefore, the current flowing through the two first rectifying diodes _D3 , _D4 and the two second rectifying diodes _D5 , _D6 will naturally drop to zero, and the rectifying circuit 123 reaches ZCS. Thus, the resonant converter circuit 120 of the present disclosure can have a soft switching characteristic when operating in the third region R3, thereby reducing switching loss.

Fourth, when the switch switching frequency _fs is less than the first resonant frequency _fr (i.e., _fs < _fr ), the voltage gain curve operates in the fourth region R4. At this time, the input impedance of the resonant converter circuit 120 exhibits capacitive properties, and the inverter circuit 121 cannot achieve ZVS. In the fourth region R4, the voltage gain will vary greatly with the change of the switch switching frequency _fs , so the resonant converter circuit 120 should be avoided from operating in the fourth region R4. The resonant converter circuit 120 of the present disclosure has a unidirectional fixed output input voltage. In order to enable both the primary side and the secondary side to have flexible switching characteristics, the resonant converter circuit 120 should be operated in the third region R3 (i.e., when the switch switching frequency _fs is between the first resonant frequency _fr and the second resonant frequency _fm ). The following will describe in detail the multiple working states of the resonant converter circuit 120 operating in the third region R3 after being controlled by frequency modulation.

Please refer to FIG. 2, FIG. 6, FIG. 7, FIG. 8, FIG. 9, FIG. 10, FIG. 11 and FIG. 12, wherein FIG. 7 is a waveform diagram showing the components of the resonant conversion circuit 120 of the DC converter 100 of FIG. 2 operating in the third region R3, FIG. 8 is a circuit diagram showing the resonant conversion circuit 120 of the DC converter 100 of FIG. 2 operating in the first working state, and FIG. 9 is a circuit diagram showing the resonant conversion of the DC converter 100 of FIG. 2. FIG. 10 is a circuit diagram showing the resonant conversion circuit 120 of the DC converter 100 of FIG. 2 operating in the third working state, FIG. 11 is a circuit diagram showing the resonant conversion circuit 120 of the DC converter 100 of FIG. 2 operating in the fourth working state, and FIG. 12 is a circuit diagram showing the resonant conversion circuit 120 of the DC converter 100 of FIG. 2 operating in the fifth working state. It should be noted that, according to the switching states of the two first switches S ₃ , S ₄ and the two second switches S ₅ , S ₆ , one cycle of each second switch driving signal v _GS3 , v _GS4 , v _GS5 , v _GS6 for controlling each switch can be divided into 10 working states. Since the working state of the positive half cycle is similar to the working state of the negative half cycle, the present disclosure only describes the five working states of the positive half cycle, and the five working states of the negative half cycle are not described separately. The following will be used in conjunction with Figures 8 to 12 to illustrate the operating waveforms of each component of the DC converter 100 in Figure 7 in multiple time segments. In addition, the two first switches S ₃ and S ₄ have two currents i _DS3 and i _DS4 respectively, and the two second switches S ₅ and S ₆ have two currents i _DS5 and i _DS6 respectively. The two first rectifying diodes D ₃ and D ₄ have two voltages v _D3 and v _D4 respectively, and the two second rectifying diodes D ₅ and D ₆ have two voltages v _D5 and v _D6 respectively.

As shown in FIG. 8 , when time t is time t ₀ (t=t ₀ ), the first switch S ₄ and the second switch S ₅ are turned off, and the two first switches S ₃ , S ₄ and the two second switches S ₅ , S ₆ on the primary side are all turned off and enter the dead zone state. The resonant current i _Lr begins to store energy in a parasitic capacitor _Coss4 of the first switch S ₄ and a parasitic capacitor _Coss5 of the second switch S ₅ (i.e., the voltages v _DS4 , v _DS5 rise), and simultaneously releases energy in a parasitic capacitor _Coss3 of the first switch S ₃ and a parasitic capacitor _Coss6 of the second switch S ₆ (i.e., the voltages v _DS3 , v _DS6 drop). The first rectifier diode _D3 and the second rectifier diode _D6 on the secondary side are turned on, and the first rectifier diode _D4 and the second rectifier diode _D5 are turned off. The present disclosure configures a dead time in the switch duty ratio, and the energy storage time of the parasitic capacitors C _oss4 and C _oss5 and the energy release time of the parasitic capacitors C _oss3 and C _oss6 are both less than the dead time, thereby ensuring that the two first switches S ₃ and S ₄ and the two second switches S ₅ and S ₆ achieve ZVS switching. When the parasitic capacitors C _oss4 and C _oss5 are charged to another input DC voltage V _IN2 (equal to the first output DC voltage V _OUT1 ), the parasitic capacitors C _oss3 and C _oss6 are discharged to 0V, and the first working state ends.

As shown in FIG. 9 , when time t is time t ₁ (t=t ₁ ), the two first switches S ₃ , S ₄ and the two second switches S ₅ , S ₆ are all turned off, and the parasitic capacitors C _oss3 , C _oss4 , C _oss5 , and C _oss6 are charged and discharged. After the resonant current i _Lr discharges the parasitic capacitors C _oss3 and C _oss6 to 0 V, it continues to flow through the intrinsic diode (not separately labeled) of the first switch S ₃ and the intrinsic diode (not separately labeled) of the second switch S ₆ , so that the first switch S ₃ and the second switch S ₆ achieve ZVS switching. At the same time, the resonant current i _Lr transmits energy to the secondary side through the transformer TR. The first rectifying diode _D3 and the second rectifying diode _D6 are turned on, and the first rectifying diode _D4 and the second rectifying diode _D5 are turned off. When time t is time _t2 (t= _t2 ), the first switch _S3 and the second switch _S6 are turned on, and the second working state enters the third working state.

As shown in FIG. 10 , when time t is time t ₂ (t=t ₂ ), the first switch S ₃ and the second switch S ₆ are turned on, the first switch S ₄ and the second switch S ₅ are turned off, and ZVS soft switching is realized. At this time, the resonant current i _Lr and the excitation current i _Lm continue to rise, and the current difference between the resonant current i _Lr and the excitation current i _Lm transfers energy to the secondary side through the transformer TR. The first rectifier diode D ₃ and the second rectifier diode D ₆ are turned on to provide energy to an output resistor R _OUT2 . Due to the clamping effect of the second output DC voltage V _OUT2 , the excitation inductance _{L m does} not participate in the resonance, so the excitation current i _Lm continues to rise. When the resonant current i _Lr is equal to 0, the third working state ends.

As shown in FIG. 11 , when time t is time t ₃ (t=t ₃ ), the first switch S ₃ and the second switch S ₆ are still turned on, and the resonant current i _Lr changes from negative to 0. At this time, the resonant capacitor _Cr and the resonant inductor L _r resonate to increase the resonant current i _Lr , and the current difference between the resonant current i _Lr and the excitation current i _Lm transfers energy to the secondary side through the transformer TR. The first rectifier diode D ₃ and the second rectifier diode D ₆ are turned on to provide energy to the output resistor R _OUT2 . The magnetizing inductor L _m is affected by the clamping of the second output DC voltage V _OUT2 , and the magnetizing inductor L _m does not participate in the resonance, so the magnetizing current i _Lm continues to rise from negative to positive until the resonance current i _Lr is equal to the magnetizing current i _Lm , and the fourth working state ends. At the same time, a current i _D3 flowing through the first rectifier diode D ₃ and a current i _D6 flowing through the second rectifier diode D ₆ also drop to 0 amperes (A), achieving ZCS switching.

As shown in FIG. 12 , when time t is time t ₄ (t=t ₄ ), the first switch S ₃ and the second switch S ₆ are still on, the first switch S ₄ and the second switch S ₅ are off, and the resonant current i _Lr is equal to the excitation current i _Lm . At this time, the transformer TR enters the decoupling region and does not transfer energy, so the currents i _D3 , i _D4 , i _D5 , and i _D6 are naturally cut off, achieving ZCS switching. Since the transformer TR does not transfer energy to the output resistor R _OUT2 , the magnetizing inductor L _m is not clamped by the second output DC voltage V _OUT2 , so the resonant capacitor _Cr , the resonant inductor L _r and the magnetizing inductor L _m resonate until the time t is time t ₅ , the first switch S ₃ and the second switch S ₆ are turned off, and the fifth working state ends. In other time segments (i.e., time t ₅ , t ₆ , t ₇ , t ₈ , t ₉ , t ₁₀ in FIG. 7 ), the operation mode of each component is the same as that of the aforementioned working states, so it is not repeated. Thus, the DC converter 100 of the present disclosure uses a boost converter circuit 110 at the front stage to reduce input current ripple, and uses a resonant converter circuit 120 at the rear stage to have a flexible switching effect, thereby achieving a high transformation ratio and being applicable to a wide input voltage range of fuel cells, while reducing switching losses of switches.

Please refer to FIG. 13 and FIG. 14 together, wherein FIG. 13 is a circuit diagram of a DC converter 100 according to a second embodiment of the present disclosure, and FIG. 14 is a schematic diagram of a digital control module 140 of the DC converter 100 of FIG. 13. As shown in FIG. 13 and FIG. 14, the DC converter 100 may further include a voltage sampling module 130, a digital control module 140, and a switch driving circuit 150. The voltage sampling module 130 is coupled to the boost converter circuit 110 and the rectifier circuit 123. The voltage sampling module 130 captures the first output DC voltage V _OUT1 from the boost converter circuit 110 and the second output DC voltage V _OUT2 from the rectifier circuit 123 to generate a first feedback analog signal _FS1 and a second feedback analog signal _FS2 respectively. The digital control module 140 is electrically connected to the voltage sampling module 130. The digital control module 140 generates a plurality of first control signals con ₁ and con ₂ according to the first feedback analog signal _FS1 , and generates a plurality of second control signals con ₃ , con ₄ , con ₅ , and con ₆ according to the second feedback analog signal _FS2 . The switch driving circuit 150 is electrically connected to the digital control module 140, the first power switch _S1 and the second power switch _S2 of the boost converter circuit 110, and the two first switches _S3 , _S4 and the two second switches _S5 , _S6 of the inverter circuit 121. The switch driving circuit 150 converts the first control signals _con1 , _con2 into a plurality of first switch driving signals _vGS1 , _vGS2 , and converts the second control signals _con3 , _con4 , _con5 , _con6 into a plurality of second switch driving signals _vGS3 , _vGS4 , _vGS5 , _vGS6 to control the boost converter circuit 110 and the inverter circuit 121. Thus, the DC converter device 100 disclosed in the present invention only uses the voltage sampling module 130, the digital control module 140 and the switch driving circuit 150 as the device control core of the boost converter circuit 110 and the resonant converter circuit 120, so there is no need to connect/hang other control loops externally, thereby reducing the device cost and the number of components, and also reducing the size of the DC converter device 100.

Specifically, the voltage sampling module 130 can be an isolated voltage sensor. The digital control module 140 can be a general purpose processor, a special purpose processor, a digital signal processor, a plurality of microprocessors, a microprocessor combined with a digital signal processor core, a controller, a microcontroller, an application specific integrated circuit (ASIC), a field programmable gate array (FPGA), any other type of integrated circuit, a state machine, an advanced RISC machine (ARM) based processor, and the like, but the present disclosure is not limited thereto.

Specifically, the digital control module 140 may include a signal converter 141, a subtractor 142, a proportional integral controller 143, a constant voltage/current controller 144, and a pulse signal generating module 145. The signal converter 141 may be an analog-to-digital converter (ADC), which converts the first feedback analog signal _FS1 into a first feedback digital signal _VADC1 and converts the second feedback analog signal _FS2 into a second feedback digital signal _VADC2 . The subtractor 142 is electrically connected to the signal converter 141. The subtractor 142 subtracts the first feedback digital signal V _ADC1 from the plurality of first reference voltage signals V _ref1 to generate a plurality of first difference signals Dif ₁ , and subtracts the second feedback digital signal V _ADC2 from the plurality of second reference voltage signals V _ref2 to generate a plurality of second difference signals Dif ₂ . The proportional-integral controller 143 is electrically connected to the subtractor 142 . The proportional-integral controller 143 receives these first difference signals Dif ₁ and these second difference signals Dif ₂ , and generates these first control signals con ₁ , con ₂ and these second control signals con ₃ , con ₄ , con ₅ , con ₆ respectively through the constant voltage/current controller 144 and the pulse signal generating module 145, wherein these first control signals con ₁ , con ₂ are output by a PWM generator of the pulse signal generating module 145, and these second control signals con ₃ , con ₄ , con ₅ , con ₆ are output by a PFM generator of the pulse signal generating module 145.

Thus, the DC converter 100 of the present disclosure uses the digital control module 140 as the device control core, and can realize that the analog DC power source 10 can have a wide input voltage range (e.g., 40 V to 125 V) and output the second output DC voltage V _OUT2 (e.g., 400 V) to the load 20 when the voltage is low. The DC converter 100 feeds back a signal to the digital control module 140 through the voltage sampling module 130, and can perform voltage regulation and signal processing control in response to changes in the load 20, thereby achieving overall system output stability. In addition, experimental results show that when the DC converter 100 operates at a low voltage input of 40 V, the maximum conversion efficiency can reach 92.69%, and when it operates at a high voltage input of 125 V, the maximum conversion efficiency can reach 93.57%.

In summary, the present disclosure has the following advantages: First, the front stage uses a boost converter circuit, and the rear stage uses a resonant converter circuit, so the DC converter can achieve a high conversion ratio and can be applied to a wide input voltage range. Second, the boost converter circuit reduces the ripple of the total input current, thereby increasing the output power of the DC converter. Third, the digital control module is used to adjust the switch switching frequency, so that the switch switching frequency is between the first resonant frequency and the second resonant frequency, so that the inverter circuit operates in ZVS and the rectifier circuit operates in ZCS, so that the DC converter can have a flexible switching effect, thereby reducing the switching loss of the switch. Fourthly, the voltage sampling module, digital control module and switch drive circuit are used as the device control core, so there is no need to connect/hang up other control loops, thereby reducing the device cost and the number of components, and also reducing the size of the DC conversion device.

Although the contents of this disclosure have been disclosed as above by way of embodiments, they are not intended to limit the contents of this disclosure. Anyone skilled in the art can make various changes and modifications without departing from the spirit and scope of the contents of this disclosure. Therefore, the protection scope of the contents of this disclosure shall be subject to the scope defined by the attached patent application.

10: DC power source 20: Load 100: DC converter 110: Boost converter circuit 120: Resonant converter circuit 121: Inverter circuit 122: Resonant circuit 123: Rectifier circuit 130: Voltage sampling module 140: Digital control module 141: Signal converter 142: Subtractor 143: Proportional integral controller 144: Constant voltage/current controller 145: Pulse signal generation module 150: Switch drive circuit V _IN1 , V _IN2 : Input DC voltage V _OUT1 : First output DC voltage V _OUT2 : Second output DC voltage v _AB : AC voltage v _D3 , v _D4 , v _D5 , v _D6 , v _DS3 , v _DS4 ,v _DS5 ,v _DS6 : voltage L ₁ : first boost inductor L ₂ : second boost inductor L _r : resonant inductor L _m : magnetizing inductor S ₁ : first power switch S ₂ : second power switch S ₃ ,S ₄ : first switch S ₅ ,S ₆ : second switch D ₁ : first diode D ₂ : second diode D ₃ ,D ₄ : first rectifier diode D ₅ ,D ₆ : second rectifier diode B ₁ : first bridge arm B ₂ : second bridge arm B ₃ : third bridge arm B ₄ : fourth bridge arm C _IN1 ,C _IN2 : input capacitor C _OUT1 ,C _OUT2 : output capacitor C _r : resonant capacitor C _oss3 , C _oss4 , C _oss5 , C _oss6 : parasitic capacitance TR: transformer N _P , N _S : number of turns i _IN1 : total input current i _L1 : first inductor current i _L2 : second inductor current i _Lr : resonance current i _Lm : magnetizing current i _D3 , i _D4 , i _D5 , i _D6 , i _DS3 , i _DS4 , i _DS5 , i _DS6 : current _RL2 : load resistance R _OUT1, R _OUT2 : output resistance F _S1 : first feedback analog signal F _S2 : second feedback analog signal V _ADC1 : first feedback digital signal V _ADC2 : second feedback digital signal V _ref1 : first reference voltage signal V _ref2 : second reference voltage signal Dif ₁ : first difference signal Dif ₂ : second difference signal v _GS1 ,v _GS2 : first switch drive signal v _GS3 ,v _GS4 ,v _GS5 ,v _GS6 : second switch drive signal con ₁ ,con ₂ : first control signal con ₃ ,con ₄ ,con ₅ ,con ₆ : second control signal R1 : first region R2 : second region R3 : third region R4 : fourth region f _s : switch switching frequency f _r : first resonant frequency f _m : second resonant frequency t, t ₀ ,t ₁ ,t ₂ ,t ₃ ,t ₄ ,t ₅ ,t ₆ ,t ₇ ,t ₈ ,t ₉ ,t ₁₀ :Time Q: Circuit quality factor

Figure 1 is a block diagram of a DC converter according to the first embodiment of the present disclosure; Figure 2 is a circuit diagram of the DC converter of Figure 1; Figure 3 is a circuit diagram of the DC converter of Figure 2 with the first power switch and the second power switch turned on; Figure 4 is a circuit diagram of the DC converter of Figure 2 with the first power switch turned on and the second power switch turned off; Figure 5 is a circuit diagram of the DC converter of Figure 2 with the first power switch turned off and the second power switch turned on; Figure 6 is a voltage gain curve of the resonant circuit of the DC converter of Figure 2; Figure 7 is a waveform diagram of each element of the resonant conversion circuit of the DC converter of Figure 2 operating in the third region; FIG. 8 is a circuit diagram showing the resonant conversion circuit of the DC conversion device of FIG. 2 operating in the first working state; FIG. 9 is a circuit diagram showing the resonant conversion circuit of the DC conversion device of FIG. 2 operating in the second working state; FIG. 10 is a circuit diagram showing the resonant conversion circuit of the DC conversion device of FIG. 2 operating in the third working state; FIG. 11 is a circuit diagram showing the resonant conversion circuit of the DC conversion device of FIG. 2 operating in the fourth working state; FIG. 12 is a circuit diagram showing the resonant conversion circuit of the DC conversion device of FIG. 2 operating in the fifth working state; FIG. 13 is a circuit diagram showing a DC conversion device according to the second embodiment of the present disclosure; and FIG. 14 is a circuit diagram showing a digital control module of the DC conversion device of FIG. 13.

10: DC power supply

20: Load

100: DC conversion device

110: Boost converter circuit

120: Resonance conversion circuit

121: Inverter circuit

122: Resonance circuit

123: Rectifier circuit

Claims (9)

A DC conversion device includes: a boost conversion circuit coupled to a DC power source and boosting an input DC voltage of the DC power source to a first output DC voltage; a resonant conversion circuit coupled to the boost conversion circuit and including: an inverter circuit converting the first output DC voltage to an AC voltage; a resonant circuit coupled to the inverter circuit; and a rectifier circuit coupled to the resonant circuit; a voltage sampling module, coupled to the boost conversion circuit and the rectifier circuit, and capturing the first output DC voltage and the second output DC voltage to generate a first feedback analog signal and a second feedback analog signal respectively; a digital control module, electrically connected to the voltage sampling module, the digital control module generates a plurality of first control signals according to the first feedback analog signal, and generates a plurality of first control signals according to the second feedback analog signal and generating a plurality of second control signals; and a switch drive circuit electrically connected to the digital control module, the boost conversion circuit and the inverter circuit, the switch drive circuit converting the first control signals into a plurality of first switch drive signals, and converting the second control signals into a plurality of second switch drive signals to control the boost conversion circuit and the inverter circuit; wherein the resonant circuit and the rectifier circuit The inverter circuit converts the AC voltage into a second output DC voltage to supply power to a load; wherein the inverter circuit has a switch switching frequency, and the resonant circuit has a first resonant frequency and a second resonant frequency. When the switch switching frequency is between the first resonant frequency and the second resonant frequency, the inverter circuit operates in a zero voltage switching, and the rectifier circuit operates in a zero current switching. A DC conversion device as described in claim 1, wherein the DC power source has a first voltage terminal and a second voltage terminal, and the boost conversion circuit comprises: an input capacitor, two ends of the input capacitor are coupled to the first voltage terminal and the second voltage terminal respectively; a first boost inductor, one end of the first boost inductor is coupled to the first voltage terminal; a second boost inductor, one end of the second boost inductor is coupled to the first voltage terminal; a first power switch, a drain end of the first power switch is coupled to the other end of the first boost inductor; a second power switch, A drain terminal of the second power switch is coupled to the other end of the second boost inductor; a first diode, an anode terminal of the first diode is coupled to the other end of the first boost inductor; a second diode, an anode terminal of the second diode is coupled to the other end of the second boost inductor; and an output capacitor, one end of the output capacitor is coupled to a cathode terminal of the first diode and a cathode terminal of the second diode, and the other end of the output capacitor is coupled to a source terminal of the first power switch, a source terminal of the second power switch and the second voltage terminal. A DC converter as described in claim 2, wherein a switch duty ratio of the first power switch and the second power switch is controlled by two first switch drive signals, and a phase difference between the two first switch drive signals is 180°; and when the switch duty ratio is 0.5, a first current ripple flowing through the first boost inductor and a second current ripple flowing through the second boost inductor cancel each other out. A DC converter as described in claim 1, wherein the inverter circuit comprises: two first switches connected in series to form a first bridge arm, wherein the first bridge arm is coupled to the resonant circuit; and two second switches connected in series to form a second bridge arm, wherein the second bridge arm is coupled to the resonant circuit; wherein the AC voltage exists between the first bridge arm and the second bridge arm. A DC converter as described in claim 4, wherein the first switch and the second switch are respectively controlled by two second switch drive signals and operate at a switch duty ratio of 50% at the switch switching frequency, and a phase difference between the first switch and the second switch is 180°. A DC converter as described in claim 4, wherein the resonant circuit comprises: a resonant capacitor, one end of which is coupled to the first bridge arm; a resonant inductor, one end of which is coupled to the other end of the resonant capacitor; an excitation inductor, one end of which is coupled to the other end of the resonant inductor, and the other end of which is coupled to the second bridge arm; and a transformer having a primary side and a secondary side, wherein the primary side is connected in parallel to the excitation inductor, and the secondary side is coupled to the rectifier circuit. A DC converter as described in claim 6, wherein when the switch switching frequency is between the first resonant frequency and the second resonant frequency, a resonant current flowing through the resonant inductor is equal to an excitation current flowing through the excitation inductor, and the transformer enters a decoupling region to turn off the rectifier circuit and operate in the zero current switching. A DC converter as described in claim 6, wherein the rectifier circuit comprises: two first rectifier diodes connected in series to form a third bridge arm, wherein the third bridge arm is coupled to a starting end of the secondary side of the transformer; two second rectifier diodes connected in series to form a fourth bridge arm, wherein the fourth bridge arm is coupled to an ending end of the secondary side of the transformer; and an output capacitor, wherein two ends of the output capacitor are respectively coupled to the starting end and the ending end of the secondary side. The DC converter device as described in claim 1, wherein the digital control module comprises: a signal converter, which converts the first feedback analog signal into a first feedback digital signal, and converts the second feedback analog signal into a second feedback digital signal; a subtractor, which is electrically connected to the signal converter, and the subtractor subtracts the first feedback digital signal from a plurality of first reference voltage signals to generate a plurality of first difference signals, and subtracts the second feedback digital signal from a plurality of second reference voltage signals to generate a plurality of second difference signals; and a proportional-integral controller, which is electrically connected to the subtractor, and the proportional-integral controller receives the first difference signals and the second difference signals to generate the first control signals and the second control signals respectively.

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