RU2695799C1 - Method of determining location parameters of location objects in radar sensors with frequency manipulation of continuous radiation of radio waves and a device for its realizing - Google Patents

Abstract

FIELD: radar ranging and radio navigation.

SUBSTANCE: invention relates to the field of radar with frequency manipulation of continuous radiation (FMCR) of radio waves and can be used for detection of moving targets, measurement of distance to object of location, speed and direction of movement. Result is achieved due to the fact that the Doppler signal obtained at different frequencies is amplified and formed, then an integer number of periods of the Doppler signal is counted and the frequency of the probing radio signal is switched from one frequency to the other at the moment when the Doppler signal passes through zero. Device implementing the disclosed method comprises an antenna, an FM Doppler receiver-transmitter, an amplifier, an amplitude-threshold shaper, a divider-decoder with a decoder, a time interval comparison unit and a measurement unit connected to each other in a certain manner.

EFFECT: technical result is expansion of radar sensors (RS) working area with FMCR of radio waves in range and speed towards their increase, enabling determination of direction of movement of location object, as well as reduction of level of out-of-band radiation.

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Description

The invention relates to the field of radar with frequency shift keying continuous radiation (CMI) radio waves. The method and device can be used in radar sensors (RLD), designed to detect moving objects of the location, measure the distance to them, as well as determine the speed and direction of movement.

The well-known method of radar with MNRI radio waves (in foreign literature it is also known as FSK - an acronym for "Frequency Shift Keying") assumes a sequential transmission of sounding radio signals at two different frequencies ƒ ₁ and ƒ ₂ . The radio signals reflected from the location object and returned back to the RLD have additional phase incursions ϕ ₁ and ϕ ₂ relative to the emitted, depending on the distance to the location object (see pages 218-221 in the book: A. Vinitsky, “Essay on the fundamentals of radar with continuous radiation of radio waves. ”M: Sov. radio, 1961. [1]). The phase difference ϕ = ϕ ₁ -ϕ ₂ between the transmitted and received radio signals for moving location objects is transferred at the output of the receiver mixer to the phase of the Doppler signals. Therefore, the current phase difference of the Doppler signals received at different frequencies ƒ ₁ and ƒ ₂ gives information about the distance to the location object: R = сϕ / 4πΔƒ, where Δƒ = ƒ ₂ -ƒ ₁ is the frequency spacing, s is the propagation velocity of the radio emission, and Δƒ << ƒ _1,2 . Here ƒ _1,2 is any of the frequencies ƒ ₁ or ƒ ₂ . In this case, the uniqueness of measuring the range of R _one to the location object is limited by the possibility of measuring the phase difference of no more than 2π and is determined by the choice of the frequency spacing Δƒ: R _one ≤с / 2Δƒ. The relative (radial) speed V _p of the location object is found by the frequency F _{D of the} Doppler signal V _p = cF _D / 2ƒ _1,2 . The direction of movement of the location object is usually determined by the sign of the phase difference ϕ.

It should be noted that frequency manipulation, unlike other types of modulation of the frequency of radiation of radio waves (for example, sinusoidal, sawtooth, etc.) and signal processing are the easiest to implement. They allow quite simply to separate in the signal the data on the phase difference, which is the result of the Doppler effect, from the data on the phase difference, which is due to the delay in the reflected radiation and is related to the distance to the location object. In this case, the determination of the parameters of the target’s motion in RLD with the MNI of radio waves during signal processing is performed at a low (Doppler) frequency, which is also an advantage of this method of radar, since low-speed and cheap digital signal processors can be used to process these signals. Another advantage of this type of modulation for many applications is the lack of linearity requirements for the modulation characteristics of the transmitter. Therefore, due to the noted advantages, RLDs with the CMNI of radio waves are widely used for solving various problems. For example, in motor vehicles in collision avoidance systems (see Stevens JE, Nagy LL, “Diplex Doppler radar for automotive obstacle detections IEEE Transactions on Vehicular Technology, vol. Vt-23, no. 2, pp. 34-44 [2]; p. 72 of the article: S. Sysoeva “Actual technologies and the use of sensors of automotive active safety systems. Part 6. Radars. Components and technologies. 2007. No. 3. P. 67-76 [3]), in security systems (see. US patent US 6677887 B2, 01/13/2004. IPC G01S 13/62. "Intrusion detection radar system" / RK Harman [4]; UK applications: GB 1554661, 10.24.1979. IPC G01S 9/42. "Dual frequency radar intruder detector "/ NI Buckley [5]; GB 2073984, 10.21.1981. IPC G01S 13/56." Microwave detecting circuit ”/ RC Birks [6]), in the equipment for controlling the movement of the cargo platform of the parachute system (see pages 60-61 of the article: Noskov V.Ya., Varavin A.V., Vasiliev A.S., Ermak G.P. ., Zakarlyuk N.M., Ignatkov K.A., Smolsky S.M. “Modern hybrid-integrated autodyne microwave and millimeter-wave generators and their application. Part 9. Radar application of autodyne ”// Successes of modern radio electronics. 2016. No3. S. 32-86 [7]) and others.

The method of radar with MNRI radio waves lies at the heart of the implementation of the devices described in a large number of publications. For example, in one of the earliest US patents US 3155972, 11/03/1964. NKP 343-12. Continuous wave radar / W.D. Boyer [8] and pp. 262-263 of the article: Nilssen O.K. "New methods of range measuring Doppler radar." IRE Transactions on Aerospace and Navigational Electronics, 1962, vol. ANE-9, no. 4, p. 255-265 [9] radar systems with frequency manipulation are proposed, in the transceiver of which a reflective klystron is used as a transmitter. Means are provided for generating a control voltage on the klystron reflector in the form of a “square wave” (meander) to obtain transmitter radiation alternately at two frequencies. The spacing between the frequencies is determined by the meander amplitude. It is noted that the frequency of the meander should be significantly higher than the maximum expected frequency of the Doppler signal. Probing and reflected radio signals are emitted and received by the antenna. A circulator is used to separate the transmitted and received radio signals, as well as mix them in the mixer. This construction of the transceiver is usually called homodyne. The radio signal reflected from the moving object of the location causes the formation of two Doppler signals at the output of the mixer of the homodyne transceiver, in the form of samples made with the switching frequency of the transmitter. After the mixer, these signals are amplified in the pre-amplifier and then synchronized with the switching frequency of the transmitter are separated using an electronic switch into two identical channels. In each channel, signals in the form of samples after passing through low-pass filters and amplifiers get rid of the high-frequency components of the switching frequency and are formed in the form of smoothed Doppler signals. Then these two signals are fed to the phase difference meter, based on the RS-trigger. This meter gives a distance indication for the location object as a linear function of the phase difference between two Doppler signals. In this case, the phase counter is configured to determine the direction of movement of the location object.

A device with frequency shift keying is also known for determining the distance to a location object, in which a reflective klystron is also used in the transceiver, but it simultaneously functions as a transmitter of sounding radio signals and a receiver of radio signals reflected from the location object. The microwave output of the klystron is directly coupled to the antenna without any decoupling elements (see article: Tereshchenko AF, Yashin VA, “Two-frequency Doppler autodyne range finder.” Electronic equipment. Ser. Instrumentation, 1966, No. 3 , pp. 109-117. [10]). Such a transceiver is called autodyne. To isolate the converted signal in the klystron-autodyne, a resistor is used that is connected in series to the resonator circuit (see Fig. 4 of [10]). The klystron-auto din frequency is manipulated by applying a modulating voltage in the form of a meander to the klystron reflector circuit. After amplification of the signal received from the klystron-autodyne, its further processing is performed as in the device described above with a homodyne transceiver. The advantage of an autodyne transceiver in relation to a homodyne is the simplicity of design and low cost of the microwave unit of the radar sensor.

The disadvantages of frequency-manipulated radar sensors using reflective klystrons and other electro-vacuum devices as transmitters and autodyne are large dimensions, weight and cost, as well as high values of supply voltages and currents, which limits their use in airborne and cheap radar sensors.

Free from these drawbacks are frequency-manipulated radar sensors, the transceivers of which are made on the basis of generator semiconductor devices, such as Gunn diodes, avalanche-span diodes (LPD) and transistors. Moreover, the transceivers of the known devices are made according to a homodyne or autodyne scheme (see US patents US 3659293, 04.25.1972. IPC G01S 9/04. "Range-detecting Doppler radar" / RR Gupta [11]; US 3697985, 10.10.1972. NPK 343-5PD. "Rear end warning system for automobiles" / WR Faris [12]; US 3750171, 07/31/1973., G01S 9/24, G01S 9/46. "Diplexed multi-frequency CW Doppler radar" / WR Faris [13]; US 3750172, 07/31/1973., G01S 9/38, G01S 9/44. "Multifrequency CW radar with range cutoff" / CP Tresselt [14]; US 3766554, 10.16.1973. IPC G01S 9/24. “Range cutoff system for dual frequency CW radar” / CP Tresselt [15]; US 3832709, 08.28.1974., IPC G01S 9/42. “Motion detection apparatus having the ability to determine the direction of motion and range of a moving object "/ CF Klein [16]; US 3898655, 05.08 / 1975., IPC G01 / S9 / 44." Variable range cut-off system for dual frequency CW radar ”/ C.P. Tresselt [17]).

Devices are also known in which two antennas are used to separate probing and reflected radio signals, one of which is connected to the transmitter and the other to the receiver of the transceiver (see, for example, US patents: US 4893125, 01/09/1990. IPC G01S 13/38 . "Vehicle diplex Doppler near-obstacle detection system" / PA May [18]; USA US 7079030, 07/18/2006. IPC G08B 13/18. "Microwave sensor" / M. Tsuji [19]; US 7167008, 23.06.2007 IPC G01S 13/00. "Microwave sensor for object detection basid on reflected microwaves" / M. Tsuji [20]; Japanese application: JPH 02181686 A, 07/16/1990. IPC G01S 13/56. "Diplex Doppler type obstacle detector for vehicle "/ FA Mei [21]; EP application: EP 0367404, 09/29/1989. IPC G01S 13/56." Vehicle diplex Doppler near-obstacle detection apparatus)) / P.A. May [22]). In this case, part of the transmitter power is supplied to the receiver mixer through a directional coupler. However, the use of two antennas increases the dimensions of the devices, which creates difficulties for their use in airborne radar sensors.

To manipulate the frequency of the probing radio signals in RLDs with the FMNI of radio waves, various microwave generators are usually used that provide electrical control of the generation frequency. Among them, microwave generators are known in which the frequency is tuned using a varicap or YIG sphere, in addition, microwave generators controlled by a power circuit are also used in simple and cheap RLDs with FMNI radio waves (see Figs. 1-3 of US Pat. No. 3,659,293, 04/25/1972. IPC G01S 9/04. "Range-detecting Doppler radar" / RR Gupta [11]). In recent years, the industry has mastered a wide range of Doppler transceivers in the form of ready-to-use modules integrated with an antenna (see, for example, “24.125 GHz, K-band FMCW radar sensor” at www.sagemillimeter.com [23]). Among them there are modules in which the frequency of the transmitter is stabilized using a digital frequency synthesizer based on phase-locked loop (PLL). Microcontrollers are used to control the frequency of these synthesizers, which set the desired frequency values according to the commands received on them (see China patent: CN 2938141, 08/22/2007. IPC G01S 13/93. "Vehicle-mounted anti-collision radar system" / XG Yang [24]).

Devices are also known in which, along with frequency manipulation, pulsed modulation of radio wave radiation is simultaneously used (see US patents US 3766554, 10.16.1973. IPC G01S 09/24. "Range cutoff system for dual frequency CW radar" / CP Tresselt [15] ; US 3889261 A, 06/10/1975. IPC G01S 09/24. Range measurement pulse radar system / J. Sirven [25]; US 4697184 A, 09/29/1987 IPC G01S 13/56. "Intrusion detection radar system with amplitude and frequency carrier modulation to eliminate targets at short and long ranges "/ J. Cheal [26]; US 5539410 A, July 23, 1996. IPC F42C 13/04." Pulse Doppler proximity sensor "/ JE Zveglich [27]; p. 60 -61 articles: Noskov V.Ya., Varavin A.V., Vasiliev A.S., Ermak G.P., Zakarlyuk N.M., Ignatkov K.A., Smolsky S.M. mennye hybrid-integrated generators autodyne microwave and millimeter wave bands and their application. Part 9. The radar application autodynes "// Successes of modern electronics. 2016. №3. pp 32-86 [7]). The use of simultaneous pulse modulation and frequency manipulation with a slight complication of devices makes it possible to increase their noise immunity to passive interference due to the possibility of forming “dead zones” in range (see p. 31-43 of the article by V. Ya. Noskov, S. M. Smolsky, “Modern hybrid-integrated autodyne microwave and millimeter-wave generators and their application. Part 6. Research of radio-pulse autodyne "// Advances in modern radio electronics. 2009. No. 6. P. 3-51 [28]).

Also known are RLDs with MNRI radio waves in which the processing of Doppler signals is partially or completely performed by digital signal processors (see US patents US 6380882, 04.30.2002. IPC G01S 13/56. "Motion detector based on the Doppler principle" / S. Hegnauer [29]; US 6677887, 06/13/2004. IPC G01S 13/62. "Intrusion detection radar system" / RK Harman [30]; US 6703967, 03/09/2004., G01S 13/38. "Distance measuring device" / H . Kuroda [31]; US 7791528, 09/07/2010., G01S 13/00, G01S 13/58. "Method and apparatus for radar signal processing" / D. Klotzbuecher [32]; EP application: EP 1067397, 01.25.2006 IPC G01S 13/56. "Bewegungsmelder nach dem Doppler-prinzip" / S. Hegnauer [33]; Chinese patent: CN 104898114, 09.09.2015. IPC G01S 13/58. "FSK-CW radar design and implementation method" / ZM Yan [34]). Modern digital signal processors (DSP) as part of the RLD perform the functions of controlling the operating modes of the units included in the device, generating and processing signals with a significant simplification of the device design and provide the flexibility to change the signal processing algorithm due to the possibility of their reprogramming (see pages 71-77, Fig. 55 and 60 articles: Noskov V.Ya., Varavin A.V., Vasiliev A.S., Ermak G.P., Zakarlyuk N.M., Ignatkov K.A., Smolsky S.M. “Modern hybrid microwave and millimeter integrated autodyne generators of ranges and their application. Part 9. The radar application autodynes "// Successes of modern electronics. 2016. №3. pp 32-86 [7]). The DSP includes blocks that perform the functions of: analog-to-digital (ADC) and digital-to-analog (DAC) converters designed to digitize signals and control the operation of external analog nodes; exchange of information with the display and control unit or with a personal computer; spectral analysis, digital filtering of signals, data generation for displaying results, and many others (see the book: “User Guide: MSP430X1XX Microcontroller Family”. Translated from English. M .: Kompel CJSC, 2004 [35]).

его движения период модуляции Т_м должен удовлетворять следующим неравенствам:An analysis of the prior art has shown that the known methods for determining the motion parameters of location objects and the technical solutions of RLD with MNII radio waves have a common drawback. It consists in the fact that for the normal reception and formation of Doppler signals it is necessary to fulfill the following conflicting conditions. On the one hand, to minimize amplitude-phase distortions of Doppler signals, the modulation period T _{m of} radiation should be significantly less than the minimum period of the Doppler signal T _D = 1 / F _D = s / 2V _p ƒ _1.2 at least an order of magnitude. On the other hand, for the normal reception of the reflected radio signal, its conversion and processing, the time τ = 2R / s of the propagation of the radio signal to the location object and back should be less than half the modulation period T _m (see page 45 of the article Votoropin SD, Noskov V .Ya., Smolsky SM "Modern hybrid-integrated autodyne microwave and millimeter-wave generators and their application. Part 5. Research of autodyne with frequency modulation" // Advances in modern radio electronics. 2009. No. 3. P. 3-50 [ 36]). It follows that for a given limit range R = R _max to the location object and maximum speed

its movement, the modulation period T _m must satisfy the following inequalities:

, а также при переходе на более высокочастотный диапазон радиолокации, данные неравенства могут не выполняться, и нормальная работа РЛД с ЧМНИ радиоволн становится невозможной. Например, при R_max=225 м,

и ƒ_1,2=24,125 ГГц левая и правая части системы неравенств становятся равными Т_м=3×10^-6 секунд и с дальнейшим увеличением перечисленных параметров система неравенств не имеет решения. Это является существенным ограничением известных технических решений, особенно в свете общих тенденций освоения радиолокацией миллиметрового и субмиллиметрового диапазонов волн. Кроме того, в известных устройствах необходимо для выделения и формирования доплеровских сигналов иметь два идентичных канала, демодулятор частотной манипуляции (аналоговый коммутатор), тактовый генератор и синхронизатор, которые излишне усложняют реализацию РЛД с ЧМНИ радиоволн. Дополнительными недостатками известных устройств является также широкая полоса спектра зондирующего радиосигнала из-за высокой частоты манипуляции и его излучение в «нерабочих» условиях, когда объекта локации в поле излучения антенны нет, создавая помехи иным радиосредствам.From this system of inequalities it is seen that with an increase in the limiting range R _max and speed

, and also when moving to a higher frequency range of radar, these inequalities may not be fulfilled, and the normal operation of the RLD with the MNI of radio waves becomes impossible. For example, when R _max = 225 m,

and ƒ _1.2 = 24.125 GHz, the left and right sides of the system of inequalities become equal to T _m = 3 × 10 ^-6 seconds and with a further increase in the above parameters, the system of inequalities has no solution. This is a significant limitation of the known technical solutions, especially in light of the general trends in the development of millimeter and submillimeter wavelength ranges by radiolocation. In addition, in known devices it is necessary for separation and formation of Doppler signals to have two identical channels, a frequency-shift keying demodulator (analog switch), a clock generator and a synchronizer, which unnecessarily complicate the implementation of RLDs with MNI radio waves. An additional disadvantage of the known devices is also a wide band of the spectrum of the probing radio signal due to the high frequency of manipulation and its emission in “non-working” conditions, when there is no object of location in the radiation field of the antenna, interfering with other radio facilities.

The closest analogue (prototype) in terms of technical nature, principle of operation and the achieved positive effect is RLD with CMNI radio waves, declared according to US patent US 3913106, 10/14/1975. IPC G01S 9/24. “Radar detection apparatus for preventing vehicular collisions)) / K. Sato [37].

The RLD with the CMNI of the prototype radio waves contains a Doppler transceiver with the possibility of electric control of its frequency (see Fig. 1 of the prototype description), made according to a homodyne circuit with separate antennas for transmitting and receiving radio signals. The output of the transmitter through a directional coupler is connected to the input of the mixer, the output of which is connected to the input of a low-noise amplifier. The output of the latter is connected to the input of the amplitude-threshold driver, connected by its output to the input of the binary counter-divider made from the cascade connection of the first and second triggers, connected by the inverting outputs of the triggers with a decoder made on the logical element And, and the non-inverting output of the first trigger is connected to the control input the frequency of the Doppler transceiver, while the non-inverting output of the second trigger and the output of the decoder are respectively connected to the first and second moves speed and distance measuring unit. The amplitude-threshold driver includes a serial connection of a comparator, a front driver and a pulse cutoff and a single-shot.

The method for determining the motion parameters of the objects of the RLD location with the CMNI radio waves of the prototype in accordance with the description of the principle of its action is as follows. The moving location object is alternately irradiated with a sounding radio signal at two frequencies, the radio signal reflected at the same frequencies is received from the moving location object, mixed with a part of the sounding radio signal and converted to the low frequency region, receiving two alternating Doppler signals in the form of fragments of instantaneous sinusoid values. Then the Doppler signal is amplified and formed in the form of pulses, and the fronts of these pulses coincide with each moment the Doppler signal passes through zero. Next, divide by two numbers of transitions of the Doppler signal through zero, decode the result of division and switch the frequency of the probing radio signal from one frequency to another at the moments of each transition of the Doppler signal through zero. In this case, the distance to the location object is determined by the duration of the time intervals corresponding to the phase jumps of the signal when switching the frequency of the probing radio signal from one frequency to another, and the relative speed of the object is determined based on the duration of the periods of the Doppler signal.

, здесь смысл обозначений Т₁, t₀ и t₂ раскрыт на эпюре фиг. 2А и в описания прототипа. Их временной интервал равен периоду доплеровского сигнала

, а скорость объекта локации в блоке измерений вычисляется по формуле:The analysis of the prototype showed that the known method for determining the motion parameters of location objects and its technical implementation of RLD with MNII radio waves have several disadvantages. The main disadvantage is that the method of determining the motion parameters of the objects of the location of the prototype ensures the normal functioning of the device only in conditions where the propagation time τ of the probing radio signal to the location object and back in the form of a reflected radio signal is negligible compared to the period T _{D of the} Doppler signal: τ <<T _D This (called first) case, pulses at the noninverting output of the second flip-flop (see. Epure FIG. 2H description of a prototype), whose period is denoted as T _spd, are in the measurement unit for determining the velocity of the location of the object. According to the description of the prototype (see the plot in Fig. 2A of the description of the prototype), the pulse duration for the first case is

, here the meaning of the designations T ₁ , t ₀ and t _{2 is} disclosed in the diagram of FIG. 2A and in the description of the prototype. Their time interval is equal to the period of the Doppler signal

, and the speed of the location object in the measurement block is calculated by the formula:

(см. обозначения t₀ и t₁ на эпюре фиг. 2 описания прототипа). По длительности

этих импульсов с учетом периода Т_Д доплеровского сигнала в блоке измерения для этого (первого) случая находится расстояние до объекта локации:In this case, the pulse duration T _dist (to determine the distance R) at the output of the decoder (element I) (see the diagram in FIG. 2J of the prototype description) corresponds to time intervals t _ϕ = t ₁ -t ₀ due to phase jumps of the Doppler signal when switching the frequency sounding radio signal from one frequency to another:

(see the designations t ₀ and t ₁ on the plot of FIG. 2 of the description of the prototype). By duration

of these pulses, taking into account the period T _{D of the} Doppler signal in the measurement unit for this (first) case, is the distance to the location object:

. Аналогично, при определении длительности импульсов Т_скор также необходимо учитывать время распространения τ. Как видно из эпюр на фиг. 2А описания прототипа, в этом случае скачков частоты за период доплеровского сигнала насчитывается четыре. Поэтому к длительности каждого импульса Т_скор на неинвертирующем выходе второго триггера прибавляется четыре интервала времени распространения т радиосигналов до объекта локации и обратно:

However, with increasing distance R to the location object and the relative speed V _{p of} its movement, the propagation time τ of the radio signals and the period T _D = s / 2ƒ _1,2 V _{p of the} Doppler signal become comparable in duration. This situation is especially characteristic of the CMNI RLD in the ranges of millimeter and submillimeter waves. For example, for values of R = 225 m, V _p = 300 m / s and ƒ _1.2 = 78 GHz, we have: τ = 2R / s = 1.5 × 10 ^-6 and T _D = 6.4 × 10 ^-6 seconds. In this case (let's call it second), when generating and processing Doppler signals, it is necessary to take into account the propagation time factor τ of radio signals to the location object and vice versa. During time τ, after a frequency jump at the mixer output, a beat signal is generated with a frequency equal to the frequency spacing Δƒ = ƒ ₂ -ƒ ₁ (see Fig. 8.17 and the description thereto on pages 218-221 of the book: A. Vinitsky, “ Essay on the basics of radar in the continuous emission of radio waves. "M .: Sov. Radio, 1961. [1]). The beat signal in frequency is much higher than the Doppler signal and it is suppressed at the mixer output and in the Doppler signal amplifier, i.e. in fact, he is not. The formation of the Doppler signal begins after the arrival of the reflected radio signal, i.e. after the propagation time of r (see Fig. 8.17 and the description thereto on pages 218-221 of the book: AS Vinitsky, “An Essay on the Basics of Radar with Continuous Radiation of Radio Waves.” M .: Sov. Radio, 1961. [1]) . Therefore, in the second case, when the pulse duration T is _{found, the distance} at the non-inverting output of the second trigger to each time interval t _ϕ corresponding to the phase jump of the Doppler signal when switching the frequency of the probe radio signal from one frequency to another, in this case the propagation time τ is added:

. Similarly, when determining the pulse duration T _{speed it is} also necessary to take into account the propagation time τ. As can be seen from the diagrams in FIG. 2A of the description of the prototype, in this case, there are four frequency jumps during the Doppler signal period. Therefore, four intervals of the propagation time t of radio signals to the location object and vice versa are added to the duration of each pulse T _speed at the non-inverting output of the second trigger:

The measurement unit of the prototype device in accordance with the calculation algorithm laid down in it in the second case determines the relative speed of the object:

этих импульсов с учетом полученных значений

в блоке измерения для второго случая находится расстояние до объекта локации:By duration

of these pulses, taking into account the obtained values

in the measurement unit for the second case is the distance to the location object:

и расстоянию ΔR=R⁽²⁾ - R⁽¹⁾:If the first case is considered true, then in the second case the prototype device during measurements gives the following methodological errors in speed

and the distance ΔR = R ⁽²⁾ - R ⁽¹⁾ :

,

,

,

,

Where

- относительная методическая ошибка определения скорости;

- relative methodological error in determining the speed;

- относительная методическая ошибка определения расстояния;

- relative methodological error in determining the distance;

τ _n = τ / T _D - normalized relative to the period T _{D of the} Doppler signal, the time τ of the propagation of radio signals to the location object and back;

t _ϕn = t _ϕn / T _D is the time jump of the Doppler signal normalized with respect to the period T _{D of the} Doppler signal when switching the frequency of the probing radio signal.

From the analysis of the formula for the relative error γ _scor it follows that with an increase in the normalized time τ _{n, the} relative methodological error in determining the speed increases rapidly, and in the direction of underestimating its true value. For example, a value of τ _n = 0.1 obtain: γ = -0.28 _spd.

An analysis of the formula for the relative error γ _dist showed that this error also depends on the value of t _ϕн , i.e. from the current distance to the location object. For example, with a value of t _ϕn = 0.25, the methodological error in determining the distance is zero. However, for other values of t _{ϕn, the} error in determining the distance increases rapidly, and increases toward the overestimation of the reference under the condition t _ϕn <0.25 and increases towards the underestimation of the reference - under the condition t _ϕn > 0.25. For example, with values of t _ϕn = 0.1 and τ _n = 0.1, we obtain: γ _dist = 0.43.

Thus, the essence of the problem of the prototype device is that it can only work at small distances and low speeds of the location object, such that the condition for the negligible propagation time of radio signals to the location object and vice versa compared with the period of the Doppler signal is fulfilled. In the case when the specified neglect is not performed, the known device does not provide satisfactory accuracy in determining the distance and speed of movement of location objects. This is a significant limitation of the known technical solution, especially in the millimeter and submillimeter ranges of radio waves.

The frequency manipulation frequency of the probing radio signal in the prototype device, as noted above, is four times higher than the frequency of the Doppler signal. In this case, the expansion of the spectrum of the radio signal in the operating mode of the device is inevitable when a location object is in the radiation field of the antenna. This is an additional disadvantage of the prototype device, since the probing radio signal can interfere with other radio means. In addition, the prototype device does not provide for determining the direction of movement of the location object.

Thus, the solution to this problem is to remove the indicated restrictions, namely, to expand the working area of RLD with MNI radio waves in range and speed in the direction of their increase, as well as to reduce the level of minor spectral components of the probing radio signals and to determine the direction of movement of the location object . This is achieved by the fact that in the method for determining the parameters of the movement of location objects in the RLD with the CMNG radio waves, in which the moving location object is irradiated with a sounding radio signal at two frequencies, the radio signal reflected from the moving location object at the same frequencies is received, it is mixed with a part of the sounding radio signal, transform it into a low-frequency region, receiving two alternating Doppler signals, which are then amplified and formed in the form of pulses, and the cut of these pulses coincides with the transition of additional of the Erov signal through zero at the same sign of the derivative of the instantaneous value of the Doppler signal, an integer number of periods of the Doppler signal are calculated and the results of the calculation are decrypted, upon completion of the calculation, the frequency of the probing radio signal is switched from one frequency to another at the moments when the Doppler signal passes through zero the same sign of the derivative of the instantaneous value of the Doppler signal, while the distance to the location object is determined by the difference in the time intervals of radiation Ndira radio signal on one and another frequency, the direction of motion of the object location is determined by the sign of the difference of time slots, and the relative speed of the object is determined based on the duration of periods of the Doppler signal obtained after decoding.

To implement the claimed method, a device has been developed for determining the motion parameters of a location object. The device includes an antenna connected to a frequency-controlled Doppler transceiver, an amplifier connected to the signal output of a frequency-controlled Doppler transceiver, and an amplitude-threshold driver connected to the output of the amplifier, as well as a counter-divider combined with a decoder, a unit for comparing time intervals and a measurement unit, in this case, a counting input of the counter-divider is connected to the first output of the amplitude-threshold former, and to the second output of the amplitude-threshold former I connected the reset input of the counter-divider, the first input of the time interval comparison unit and the “Detection” input terminal of the measurement unit, and the non-inverting and inverting outputs of the counter-divider are respectively connected to the second and third inputs of the time interval comparison unit, the output of which is connected to the input terminal “ Range "of the measurement unit, while the non-inverting output of the counter-divider is also connected to the frequency control input of the Doppler transceiver, and the output of one of the higher bits Rathore - to input terminal "Speed" measurement unit.

The technical result of the proposed method is to expand the working area of the device in range and speed in the direction of their increase due to the choice of conditions for switching the probing radio signal from one frequency to another and counting an integer number of periods of the Doppler signal. If in the prototype device the switching of the frequency of the probing radio signal is carried out at each moment of the transition of the Doppler signal through zero, then in the proposed device the switching is performed only at the moments of transition of the instantaneous values of the Doppler signal through zero at the same sign of the derivative of the instantaneous value of the Doppler signal. These conditions, as shown below in the description of the operation of the device, made it possible to determine the phase jump of the Doppler signal when switching the frequency of the probing radio signal taking into account its sign (positive or negative). Taking into account the sign of the phase jump during the formation of time intervals causes a corresponding decrease in one time interval and an increase in another time interval for sounding the location object. The operation of comparing (subtracting) the time intervals of sounding the location object at one and the other frequency of the radio signal made it possible to exclude the dependence of the processing results of Doppler signals on the propagation time of radio signals to the location object and vice versa, and thereby remove the limitation of the prototype.

The invention is illustrated by drawings, where in FIG. 1 shows a structural diagram of a device that implements the proposed method; in FIG. 2 shows a structural diagram of an amplitude-threshold driver; in FIG. 3 and 4 are diagrams explaining the principle of operation of the device. The essence of the proposed method will be discussed below when describing the operation of the device.

The device contains (see Fig. 1) an antenna 1 connected to a frequency-controlled Doppler transceiver 2, an amplifier 3, an amplitude-threshold driver 4, a counter-divider 5, combined with a decoder 6, a time interval comparison unit 7, and a measurement unit 8. In this case, the signal output 9 of the transceiver 2 is connected in series with an amplifier 3 and an amplitude-threshold driver 4, the first output 10 of which is connected to the counting input 11, and to the second output 12 is the input 13 of the reset counter-divider 5, the outputs of the counting triggers which are connected to the inputs of the decoder 6. The second output 12 of the amplitude-threshold driver 4 is connected to terminal 14 "Detection" of the measurement unit 8 and to the first input 15 of the unit 7 comparing time intervals. Non-inverting 16 and inverting 17 outputs of the counter-divider 5 are respectively connected to the second 18 and third 19 inputs of the device 7 comparing time intervals, while the output 16 of the counter-divider 5 is also connected to the input 20 of the frequency control of the transceiver 2. Terminal 21 "Speed" of the measurement unit 8 is connected to the output 22 of the decoder 6, and the terminal 23 "Range" of the measuring unit 8 is connected to the output 24 of the device 7 comparing time intervals.

Antenna 1 can have various versions depending on the requirements for the radiation pattern and the operating frequency range, for example, in the form of a slotted vibrator, a horn, dielectric rod, spiral antenna or the type of “wave channel” (see, respectively, pages 115, 149, 218 , 239, 260, books: "Antennas and microwave devices. Calculation and design of antenna arrays and their radiating elements" / Under the editorship of DI Voskresensky. - M .: Radio and communications, 1972. [38]).

Doppler transceiver 2 with the possibility of frequency manipulation of radiation has alternative technical solutions. It can be performed according to a homodyne or autodyne scheme (see figures 4, 5 and 6 in the description of the prototype; Fig. 1 and 2 in the description of US patent US 3750171, 07/31/1973., G01S 9/24, G01S 9/46. " Diplexed multi-frequency CW Doppler radar ”/ WR Faris [13]; description of the Doppler module on the website:“ 24.125 GHz, K-band FMCW radar sensor ”on the website: www.sagemillimeter.com [23]). When homodyne execution, the transceiver 2 contains separate nodes of the transmitter and receiver connected to the antenna 1 through an isolation device, for example, a circulator (see figures 4, 5 and 6 in the description of the Chinese patent: CN 104898114, 09.09.2015. IPC G01S 13/58. “FSK-CW radar design and implementation method” / ZM Yan [34]). In autodyne execution, the functions of the transmitter and receiver are performed by a single element - the oscillator (see Fig. 2 in the description of US patent US 3750171, 07/31/1973., G01S 9/24, G01S 9/46. "Diplexed multi-frequency CW Doppler radar" / WR Faris [13]). In this case, the oscillator is connected to the antenna 1 directly, without any decoupling elements. To isolate the Doppler signal, the detector section is usually connected to the generator chamber (see Fig. 1 of the article by Kotani M., Mitsui S., Shirahata K. “Load-variation detector characteristics of a detector-diode loaded Gunn oscillator” // Electronics and Communications in Japan. 1975. Vol. 58-B. No. 5. P. 60-66 [39]) or in the power supply circuit of the oscillator one of the signal recording device circuits described in the article is used: Noskov V.Ya., Smolsky S.M. . “Registration of the autodyne signal in the power circuit of generators on microwave semiconductor diodes. (Review) ”// Microwave engineering and instruments. 2009. No1. S. 14-26 [40]. With the simultaneous modulation of the frequency of the probing radio signal and the separation of the Doppler signal in the generator power circuit, a signal modulation-isolation block based on an operational amplifier and transistor is usually used (see Fig. 23, b on page 34 of the article: Votoropin SD, Noskov V.Ya., Smolsky SM "Modern hybrid-integrated autodyne microwave and millimeter-wave generators and their application. Part 5. Research of autodyne with frequency modulation" // Advances in modern radio electronics. 2009. No. 3. P. 3-50 [38]).

The amplifier 3 (see Fig. 1, a) simultaneously combines the amplification and filtering functions of the signal in the desired Doppler frequency band. Amplifier 3 can be made in the form of a conventional variable signal amplifier and contain elements of correction and formation of the frequency response (see, for example, pages 38-40, Fig. 2.9 and 2.10. Books: Scherbakov V.I., Grezdov G.I. “Electronic Circuits on Operational Amplifiers: A Handbook.” K .: Technika, 1983 [41]).

The amplitude-threshold driver 4 (see Fig. 1) can have various technical solutions aimed at isolating the Doppler signal against the background of noise and generating a signal in the form of impulses normalized by amplitude with reference to their cutoff to the transition of the Doppler signal through zero at the same sign of the derivative of the instantaneous values of the Doppler signal. One of the possible technical solutions of the amplitude-threshold device 4 is presented in the structural diagram of FIG. 2. The amplitude-threshold device 4 contains the first 25 and second 26 comparators with anti-noise hysteresis (see pages 253-262, Fig. 6.51 of Herpy M.'s book “Analog Integrated Circuits”. M: Radio and Communication, 1983 [42] ) and a linear amplitude detector 27 of the Doppler signal, made, for example, on the basis of an operational amplifier (see Fig. 3.6-7 on page 173 of the book: AA Rovdo “Semiconductor diodes and circuits with diodes. M.: Light LTD , 2000 [43]). The inputs of the first comparator 25 and the linear amplitude detector 27 are connected together and form the input of the amplitude-threshold driver 4, and the output of the first comparator 25 is the first output 10 of the amplitude-threshold driver 4. The output of the linear amplitude detector 27 is connected to the input of the second comparator 26, the output of which is the second 12 output of the amplitude-threshold driver 4.

As a counter-divider 5, it is preferable to use a microcircuit that combines the counter-divider 5 and the decoder 6 in one case (see Fig. 2.37-2.41 on pages 237-241 of the book: Shilo VL “Popular Digital Circuits. Reference” M.: Metallurgy, 1988 [44]). The outputs of the counting triggers of the counter-divider 5 are connected to the inputs of the decoder 6 inside the housing of the chip.

Block 7 comparison of time intervals can be performed on the basis of analog or digital methods for measuring time intervals. In the “analog” version, this device can be implemented using the principle of converting time intervals to voltages (see, for example, the description of the USSR auto certificate No. 683006 of 08.30.1979. (Bull. No. 32). IPC H03K 5/20. “A device for comparing time intervals” / S.Ya. Krasyukova [45]), followed by a comparison of the obtained voltages by means of a difference amplifier (see pages 219-225, Fig. 6.10 of the book: Herpy M. “Analog Integrated Circuits.” M .: Radio and Communications, 1983 [42]). More preferable is the “digital” version, in which the difference between time intervals is simply found using a digital time interval meter (see Fig. 3.1 and 3.2 on page 63 of the book: Aleshechkin AM, Musonov V.M., Romanov A.P. . "Metrology and radio measurements." Textbook. Krasnoyarsk: SibFU, 2008 [46]), based on a reversible counter. At the same time, counting pulses are first sent to the counting input “More” of the reverse counter when measuring the first time interval, and to the counting input “Less” - when measuring the second time interval (see the description on pages 242-246 of the book: Shilo V. L. "Popular digital microcircuits. Handbook." M.: Metallurgy, 1988 [44]). In this case, the counter reset input is the first input of the time interval comparison unit 7.

Block 8 measurements can be performed on the basis of known analog and digital methods for measuring time intervals, the latter is more preferable. Block 8 measurements can be performed using "hard logic", programmable logic integrated circuits (FPGA), other programmable digital circuits or microcontrollers. Methods for measuring the temporal parameters of signals and the principles for constructing such meters are described in the technical literature (see, for example, pp. 80-119 of the book: Komarov IV, Smolsky SM “Fundamentals of the theory of radar systems with continuous emission of frequency-modulated oscillations ". M.: Hotline-Telecom, 2010 [47]; pp. 280-291 of the book: Metrology and Radio Measurement, edited by V. I. Nefedov. M: Higher School, 2003 [48]; in the book: Rathor TS "Digital Measurements. Methods and Circuitry". M: Technosphere, 2004 [49]; in the book "User Guide: Microcon family Roller MSP430X1XX. »Trans. from English. M .: ZAO" Compel ", 2004 [35]).

The radar sensor with MNI radio waves works as follows.

When voltage is applied from the power source (not shown in Fig. 1), microwave oscillations occur in the transmitting device of the Doppler transceiver 2 when it is homodyne, which in the form of a sounding radio signal enters the antenna 1 and is radiated into the controlled space in accordance with its radiation pattern. Part of the microwave radiation of the transmitting device simultaneously affects the mixing element of the receiving device of the Doppler transceiver 2. When the transceiver 2 is autodyne, the mixing and frequency conversion function is performed on the intrinsic nonlinearity of the active element of the self-oscillator-autodyne.

Let us first consider the operation of the device in the absence in the radiation field of the antenna 1 of the location object. In this case, the reflected radio signal, of course, is absent, and only the internal noise of the transceiver 2 is affected by the mixing element 2. These noise are converted by the nonlinearity of the mixing element to the low (Doppler) frequency range and fed through amplifier 3 to the input of the amplitude-threshold driver 4. Threshold levels included in the amplitude-threshold driver 4 comparators 25 and 26 (see Fig. 2) are selected higher than the rms noise level. Therefore, at the first 10 and second 12 outputs of the amplitude-threshold driver 4 voltage levels are zero. At the same time, the output voltage of the low level from the second output 12 of the amplitude-threshold driver 4, supplied to the input 13 "Reset" of the counter-divider 5 and decoder 6, as well as to the first input 15 of the unit 7 comparing time intervals, "holds" the non-inverting state in the zero state the output 16 of the counter-divider 5 and the output 24 of the block 7 comparing time intervals. In addition, at the input 20 of the frequency control of the transceiver 2, the control voltage is logical zero. Moreover, the radiation frequency of the transceiver 2 does not change and is equal to ƒ ₁ .

Thus, in the described case, when the radio signal reflected from the location object is absent, the microwave frequency of the transceiver 2 is unchanged and there is no data on the location object at all information inputs “Detection” 14, “Speed” 21 and “Range” 23 of the measurement unit 8.

Now consider the operation of the device in the presence in the radiation field of the antenna 1 of the location object. In this case, the reflected radio signal in the mixer of the transceiver 2 is converted into a quasiharmonic Doppler frequency signal, which is output 9. After amplification and filtering in the amplifier 3, this signal is input to the amplitude-threshold driver 4. Here it is fed to the inputs of the first comparator 25 with anti-noise hysteresis and linear amplitude detector 27 (see Fig. 2). The detected voltage by the detector 27 is then fed to the input of the second comparator 26 with anti-noise hysteresis. If the amplitude of the Doppler signal exceeds the threshold levels of the first 25 and second 26 comparators with anti-noise hysteresis at the first output 10 of the amplitude-threshold driver 4, pulse-normalized Doppler frequency signals are generated in amplitude. Moreover, due to the selection of the nominal values of the elements included in the first comparator 25 with anti-noise hysteresis in accordance with the recommendations and formulas for calculation set forth on page 260 of the book: Herpy M. Analog Integrated Circuits. M .: Radio and communication, 1983 [42], the slice of the generated pulses coincides with the moment the Doppler signal passes through zero with the same derivative of the instantaneous value of the Doppler signal. At the second output 12 of the amplitude-threshold driver 4, the level of the logical unit is established, indicating the detection of a location object. At the same time, at input 13 “Reset”, the counter-divider 5 receives permission to count the pulses received at the counter input 11 from the first output 10 of the amplitude-threshold driver 4, and at the first input 15, the time interval comparison unit 7 is switched to the operating mode for measuring time intervals.

After completing the calculation of N periods of the Doppler signal (here N is the division ratio of the counter-divider 5, which can be selected from unity to tens of times) at the non-inverting output 16 of the counter-divider 5 and the input 20 of the frequency control of the transceiver 2, the state changes to “unit »And, accordingly, switching the radiation frequency to the second frequency ƒ ₂ . Changing the frequency of the radiation of the transceiver 2 from the first ƒ ₁ frequency to the second ƒ ₂ causes (after the arrival of the reflected radiation after a time τ) a phase jump of the Doppler signal at an angle ϕ _1-2 : ϕ _1-2 = 4π (ƒ ₂ -ƒ ₁ ) R / sec. (see formula 8.76 in the book: A. Vinitsky, Essay on the fundamentals of radar in the continuous emission of radio waves. M: Sov. radio, 1961 [1]). It can be seen from the formula for ϕ _1-2 that when the condition ƒ ₂ <ƒ ₁ is fulfilled, the sign of the phase jump is negative: ϕ _1-2 <0.

After completing the above cycle of counting N transitions of the Doppler signal through zero at a frequency of ƒ ₂ , the radiation frequency of the transceiver 2 is reversed: from the second ƒ ₂ frequency to the first ƒ ₁ . In addition, after the arrival of radiation reflected from the object of the location after a time τ, a phase jump occurs at an angle ϕ _2-1 : ϕ _2-1 = 4π (ƒ ₁ -ƒ ₂ ) R / s, which is equal in magnitude to the angle ϕ _1-2 , but in sign is the opposite of the first, i.e. ϕ _2-1 > 0.

In FIG. 3a shows time plots of Doppler signals 1 and 2 u ₁ (t) and u ₂ (t) received from a moving location object at frequencies соответственно ₁ and ƒ _{2 of the} radiation of transceiver 2, respectively, for the hypothetical case of the absence of “frequency feedback”, those. frequency manipulation of the probing radio signal is disabled. In the same figure, for the case of the presence of "frequency feedback", the arrows near the diagrams show the displacement along the time axis t of the image point of the instantaneous value of the Doppler signal. The value of the division coefficient N of the counter-divider 5 hereinafter is assumed to be equal to two. The figure 3, b presents the plot of the output voltage u _cf (t) of the counter-divider 5, shows the period T _{D of the} Doppler signal, as well as the time intervals T ₁ and ₂ T corresponding to the operating times of the device at the first ƒ ₁ and second ƒ ₂ frequencies of the transceiver 2. For clarity, we have considered the case where the time τ of radiation propagation to the location object and vice versa is negligible compared to the period T _{D of the} Doppler signal: τ << T _D.

In a real situation of comparatively large ranges and high values of the frequency of the Doppler signal, when the time τ of propagation of radio signals to the location object and vice versa is comparable with the period T _{D of the} Doppler signal, as shown above, it is necessary to take into account the time τ of propagation of radio signals to the location object and vice versa. For this case, Fig. 4a shows the plots of the resulting Doppler signal u _p (t) at the input of the amplitude-threshold driver 4 and the output voltage u _cf (t) of the counter-divider 5 (see Fig. 4, b). Callouts show the period T _{D of the} Doppler signal, the time intervals T ₁ and T ₂ of the device at the first ƒ ₁ and second ƒ ₂ frequencies, respectively. The time intervals τ caused by the propagation of radio signals to the location object and vice versa, and the time intervals τ _1-2 and τ _2-1 caused by phase jumps ϕ _1-2 and ϕ _2-1 when switching the radiation frequency from the first frequency ƒ ₁ to second ƒ ₂ and vice versa. The pulses of the time intervals T ₁ and T ₂ in this case, as already noted, are formed on the inverting 17 and non-inverting 16 outputs of the counter-divider 5, respectively. Based on the principle of operation of the device described above and the diagrams shown in FIG. 4, the formulas for calculating the intervals T ₁ and T ₂ have the form: T ₁ = NT _D + τ-t _ϕ , T ₂ = NT _D + τ + t _ϕ . Here, in these formulas, the signs at τ _1-2 and τ _2-1 and the equality of their absolute values are taken into account: t _ϕ = | τ _1-2 | = | τ _2-1 |.

The pulses of the time intervals T ₁ and T ₂ from the outputs 16 and 17 of the counter-divider 5 then go to the second 18 and third 19 inputs of block 7 comparing time intervals. In this block is the difference ΔТ of the time intervals T ₁ and T ₂ : ΔT = T ₂ -T ₁ = NT _D + τ + t _ϕ -NT _D -τ + t _ϕ = 2t _ϕ .

The value ΔТ = 2t _ϕ obtained at the output 24 of the block 7 for comparing time intervals allows, for a known period T _{D of the} Doppler signal (or its frequency), to determine the phase difference ϕ and, accordingly, the distance R to the location object:

R = K _R ΔT,

Where

K _R is the scale factor of the distance, which can be calculated by the following formulas: K _R = c / 4T _D Δƒ = cF _D / 4Δƒ.

From the formula for calculating the distance R to the location object shows that the output signal of the unit 7 comparing time intervals received at the output 23 "Range" contains information about the distance to the location object. In this case, the influence of the propagation time τ of the radio signals to the location object and vice versa on the comparison result is excluded.

When the direction of movement of the location object changes, the signs of phase jumps ϕ _1-2 and ϕ _2-1 when switching the radiation frequency from the first frequency ƒ ₁ to the second ƒ ₂ and vice versa, as well as, respectively, the polarity of the change in time intervals τ _1-2 and τ _{2 -1} , caused by phase jumps, are reversed. In this regard, the duration of the time intervals T ₁ and T ₂ formed on the inverting 17 and non-inverting 16 outputs of the counter-divider 5, respectively, change. Moreover, the polarity of the result of their comparison ΔT = T ₂ -T ₁ also changes sign. Therefore, in the block of measurements 8 by the sign of the difference ΔT time intervals, you can determine the direction of the relative movement of the location object.

The outputs of the 22 high-order bits (with the exception of the first one) of the decoder 6 are formed by pulses whose duration is equal to the period T _{D of the} Doppler signal (see the time diagrams in Fig. 2.38 and 2.41 of the book by Shilo VL “Popular digital microcircuits. Reference book.” M. : Metallurgy, 1988 [44]). Using these pulses during further processing, we obtain data on the speed of movement of the location object: V _p = s / 2T _D ƒ _1.2 .

Thus, the proposed method for determining the parameters of the movement of location objects in the RLD with the CMNI of radio waves and the device for its implementation while maintaining the functionality of the prototype ensure the operability of the device in conditions of large distances and speeds of the movement of the location object.

The switching period of the frequency of the probing radio signal in the proposed device is determined by the double value of the number of calculated periods of the Doppler signal by the counter-divider. Whereas in the prototype device, the frequency of the probing radio signal is switched four times during the Doppler signal. Reducing the switching frequency with respect to the prototype helps to reduce the radiation level of minority spectral components of the probing radio signals (see pages 245-249 of the book: A. Denisenko, “Signals. Theoretical Radio Engineering. Reference Guide.” M.: Hot Line - Telecom, 2005 [50]) and thereby reduce interference to other radio equipment. In addition, the proposed device provides the ability to determine the direction of movement of the location object, which is not provided for in the prototype device. This is an additional advantage of the proposed device.

It should be noted that the implementation of the claimed method described above is not the only possible one. Modern signal processors incorporating ADCs and DACs, clock generators, a large amount of RAM and a wide range of functionality of digital signal processing can significantly simplify the design of the device using the proposed method of generating and processing signals.

LITERATURE

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Claims (6)

1. The method of determining the motion parameters of location objects in radar sensors with frequency manipulation of continuous radiation of radio waves, in which the moving location object is irradiated with a probing radio signal at two frequencies, the radio signal reflected from the moving location object at the same frequencies is received, mixed with a part of the probing radio signal transform it into a low-frequency region, receive a Doppler signal and amplify it, characterized in that the Doppler signal obtained after amplification they are formed in the form of pulses, and the slice of these pulses coincides with the transition of the Doppler signal through zero at the same sign of the derivative of its instantaneous value, after which an integer number of periods of the generated pulses is counted and the counting result is decrypted, upon completion of the calculation, the frequency of the probing radio signal is switched from one frequency to another at the moments when the Doppler signal passes through zero at the same sign of the derivative of its instantaneous value, while the distance to the location object determined from the difference of time slots of radio signal probing radiation at one and the other frequencies, the direction of movement of the object location is determined by the sign of the difference of time slots, and the relative speed of the object is determined based on the duration of periods of the Doppler signal obtained after decoding. 2. A device that implements the method according to claim 1 and comprising an antenna connected to a frequency-controlled Doppler transceiver and an amplifier connected to a signal output of a frequency-controlled Doppler transceiver, characterized in that an amplitude-threshold driver connected to the output of the amplifier is introduced, a counter-divider combined with a decoder, a unit for comparing time intervals and a block of measurements, while the counting input of the counter-divider is connected to the first output of the amplitude-threshold former the second output of the amplitude-threshold driver is connected to the reset input of the counter-divider, the first input of the unit for comparing time intervals and the input terminal "Detection" of the measurement unit, the non-inverting and inverting outputs of the counter-divider are respectively connected to the second and third inputs of the unit for comparing time intervals, the output which is connected to the input terminal "Range" of the measurement unit, while the non-inverting output of the counter-divider is also connected to the frequency control input of the Doppler reception transmitter, and the output of one of the highest bits of the decoder to the input terminal "Speed" of the measurement unit. 3. The device according to claim 2, characterized in that the frequency-controlled Doppler transceiver is made according to a homodyne circuit, while it contains separate transmitter and receiver nodes connected to the antenna through an isolation element, for example, a circulator. 4. The device according to p. 2, characterized in that the frequency-controlled Doppler transceiver is made according to an autodyne circuit, while it contains as a transceiver an oscillator connected to the antenna coupled to the Doppler signal extraction means. 5. The device according to claim 2, characterized in that the amplitude-threshold driver includes first and second comparators with anti-noise hysteresis and a linear amplitude detector of the Doppler signal, while the inputs of the first comparator with anti-noise hysteresis and a linear amplitude detector are connected together, forming an input of amplitude threshold shaper, and the output of the first comparator with anti-noise hysteresis is the first output of the amplitude-threshold shaper, and the output of the linear amplitude detector is connected chen to the input of the second comparator, the output of which is the second output of the amplitude-threshold driver. 6. The device according to claim 2, characterized in that the time interval comparison unit comprises a digital time interval meter made on the basis of a reverse counter, in which the counting input “More” of the reverse counter is the second input of the time interval comparison unit, and the counting input “ Less ”- the third input of the time interval comparison unit, the reset counter input is the first input of the time interval comparison unit.

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