RU2608567C2 - Method of decametric radio communication with high-speed data transmission - Google Patents

Abstract

FIELD: radio engineering and communications.

SUBSTANCE: invention relates to radio communication and can be used in wide application decameter range radio networks. For this purpose in the method of decametric radio communication with high-speed data transmission at the transmitting side the data flow rate from the source of messages is increased twice, and the binary sequence coming after encoding to the input of a serial-parallel converter with the data flow rate is additionally distributed for its additional outputs in such a way, that formed at the additional outputs binary elements of corresponding additional channel sequences flow in parallel and synchronously with binary elements of channel sequences formed on its outputs with the channel flow rate in an additional unit of channel manipulators.

EFFECT: technical result is improvement of noise immunity of receiving data at the disturbing effect of concentrated in the spectrum sinusoidal and fluctuation interference.

1 cl, 3 dwg

Description

The invention relates to radio communications and can be used in decameter radio networks of widespread use, designed to transmit high-speed data (discrete messages) using signals with angular manipulation.

A known method of decameter radio communication with high-speed data transfer using a single-channel (sequential) method of transmitting discrete messages [1], p. 107.

The method consists in the fact that on the transmitting side, the transmitted data stream from the source is encoded and with a speed V _u = 1 / T _u (bit / s), determined by the duration of the binary element of the signal T _u , is fed to a modulator that modulates (manipulates) one carrier frequency.

The data transfer rate is also determined by the multiplicity of compression k of the transmitted signal [2], p. 573. For example, with k = 1, with the help of a modulator one can generate signals of a single relative phase telegraphy (OFT) or frequency telegraphy (CT) signals with a manipulation speed V _m (bit / s) equal to the transmission speed of binary information, i.e. V _m = V _u = 1 / T _e (T _e is the duration of the element (sending) generated by the modulator signal, in this case T _e = T _u ). For k = 2, signals of a double OFP (DFT) or double frequency telegraphy (DBT) with a manipulation speed 2 times lower than the information transfer rate, i.e. V _m = 1 / T _e = 1 / 2T _u , etc.

Using a radio transmitting device and a transmitting antenna, a single-channel signal with sequential transmission of parcels is broadcast.

On the receiving side, the demodulation of such signals is carried out by traditional methods described in [2] and determining the structure of constructing a demodulator that implements the operation of restoring the transmitted binary sequence, the inverse of the modulation operation.

However, such a single-channel (sequential) method of data transmission with traditional methods of processing received signals [2] has the following disadvantages:

1. Serial data transmission over a decameter radio channel with a high speed determined by the corresponding manipulation speed V _m = V _{u of a} single-channel signal is associated with serious difficulties due to the occurrence of intersymbol interference on the receiving side due to the presence of delayed rays. Moreover, the duration T _{e of the} element of the transmitted signal (sending) cannot be selected less than 2-3 ms, which limits the maximum manipulation speed of the transmitted signal V _{m to the} value of the order of 300-500 bit / s [3].

2. Increasing the data transfer rate by increasing the multiplexing factor k of the signal compression while maintaining the necessary duration T _{e of the} element of the transmitted signal leads to a significant decrease in the noise immunity of receiving discrete information [2], p. 615.

There is a method of decameter radio communication with high-speed data transmission using a single-channel (sequential) method of transmitting discrete messages with a test pulse and prediction (SIIP) [3].

The method consists in the fact that on the transmitting side the stream of transmitted information symbols is divided into packets of equal duration, between which special test pulses are regularly transmitted. Each test pulse is separated on both sides by passive guard intervals, which are necessary for the temporary separation of the channel response to the test pulse and the work package.

On the receiving side, this makes it possible to study all parameters of the received signal during the test pulse and the subsequent protective interval and, on this basis, perform both optimal coherent reception of the packages of the work package affected by multipath and synchronization [3].

Accordingly, with this method of decameter radio communication, a more complex algorithm for processing the received signal is applied with respect to the decameter radio communication method described above.

The disadvantage of this method is that increasing the data transfer rate by reducing the duration T _{e of the} element of the transmitted signal when implementing a more complex algorithm for processing the received signal to overcome the effects of intersymbol interference leads to a decrease in noise immunity due to a decrease in the energy of the transmitted signal element [2] and deterioration of the electromagnetic compatibility characteristics while working with other radio communication systems due to the expansion of the spectrum before waveform.

In addition, the expansion of the spectrum of the transmitted signal requires a corresponding increase in the bandwidth when receiving this signal, which further reduces the noise immunity of the reception due to the increased likelihood that the spectral components of the station or concentrated on the spectrum of radio interference get into a wider reception band.

A known method of decameter radio communication with multi-channel (parallel) transmission of discrete messages, according to which on the transmitting side the transmitted information data stream from the message source is encoded, after which a binary sequence with a speed V _u = 1 / T _u determined by the duration of the binary symbol T _u , fed to the input of a series-parallel converter, with which the input sequence is distributed on kN of its outputs in such a way that binary e ementy respective channel sequence followed in parallel and in synchronism with the channel speed V _ch = 1 / T _ch, the duration T _ch = kNT _u channel signal element is selected much greater than the magnitude of the average lag time rays on the radio path by selecting the required number N of orthogonal channel signals at different channel frequencies in the transmitted group signal and multiplexing factor k of each channel signal, in the block of N channel manipulators, consisting of N of the same type of channel man pulyatorov with serial numbers from 1 to N, each arm with a sequence number i, at the initial time of each clock period of duration T _CAD supplied in parallel and synchronously k symbols corresponding to the channel sequences to generate a corresponding element of the channel signal transmitted on the channel frequency f _i so the same serial number i, after which the channel signals generated by the channel manipulators are summed at the output of the block N channel manipulators and the N-channel group signal with a frequency interval between adjacent channel frequencies of Δf = 1 / T _ch emit broadcast via a radio transmission apparatus and the transmitting antenna.

On the receiving side, the group signal received by the receiving antenna from the output of the linear path of the radio receiving device is fed to a block of N channel demodulators, consisting of N of the same channel demodulators with serial numbers from 1 to N, while demodulating each of the N channel signals transmitted at the channel frequency f _i with the serial number i in the group signal is carried out using the corresponding channel demodulator with the same serial number i of the block N channel demodulators, the results of the demodulation of N channel lnyh signals in the form of binary sequences kN channel formed on the respective outputs of the block N-channel demodulators are fed to respective inputs of a parallel-serial converter, the output binary stream which was subjected to decoding, and then transmit the message to the recipient.

The duration T _kan element channel signal transmitted becomes kN times the initial duration T _u binary signal element that allows to overcome the negative effects of multipath reception signal at a relatively high data rate group.

However, the noise immunity of this method of decameter radio communication is insufficient for radio communication with a higher information transfer rate:

1. Increasing the data rate by increasing the multiplicity of sealing to each channel while maintaining the necessary duration T _kan element channel signal transmitted reduces the noise immunity of the reception of digital information in each channel [2].

2. Increasing the data rate by reducing the duration T _kan element channel signal transmitted in each frequency channel or by increasing the number N of frequency channels results in a corresponding expansion of bandwidth occupied by the transmitted baseband spectrum and a corresponding expansion of the bandwidth at its reception, resulting , as noted above, to reduce the noise immunity of the reception of discrete information and the deterioration of the characteristics of electromagnetic compatibility during joint work ote with other communication systems.

Of the known methods for decameter radiocommunication, the closest in essence to the tasks to be solved and for the majority of the matching essential features is the decameter radiocommunication method with high-speed data transmission described in [5].

This method of decameter radio communication with multichannel (parallel) transmission of discrete messages basically corresponds to the above method of decameter radio communication [4] except that on the receiving side they can carry out not only single reception of signals to one antenna, but also more noise-resistant dual reception on two antennas spaced apart in space or polarization.

Let us consider in more detail the process of receiving and processing signals in accordance with this method.

The voltages of two samples of the group signal received from the outputs of the linear paths of the respective two radio receivers, taken to the space or polarization of the antenna, are simultaneously fed to each of N = 20 channel blocks (KB), in which each of the samples of the group signal is filtered out to its channel signals and calculating values proportional to the cosines and sines of the phase difference between the received sample channel signal and the corresponding reference oscillation.

Each of the two active filters in the composition of each channel block consists of two identical correlators, which are fed with reference vibrations, 90 ° shifted in phase. The correlator contains a multiplier and an integrator, built on the basis of a DC amplifier with a large gain and an RC feedback circuit.

The output signals of the integrators of each channel block are the results of converting the corresponding two samples of the channel signal to zero frequency with the decomposition of each channel signal into two quadrature components, the voltage values of which are recorded and stored in the corresponding cells of the storage device (memory). Information on the quadrature components of each package of each channel signal in the form of analog voltage levels is stored in memory cells (on capacitors) for the duration of two adjacent packages. In this case, the amplitude and phase of the carrier wave of the corresponding channel signal are calculated from the level values of each pair of quadrature components.

In the subsequent phase difference calculation unit (BVRF), linear incoherent addition of each of the N pairs of samples of channel signals received at the respective two antennas is performed by summing the corresponding analog levels of the quadrature components recorded in the memory.

Since the decameter radio communication method under consideration for the transmission of information uses phase difference manipulation signals (relative phase telegraphy) with a multiplicity of multiplexing k = 1, or k = 2, or k = 3, to determine the true values of binary symbols when demodulating each of the N resulting channel signals (after linear addition), it is required to determine the phase difference between every two adjacent time-related bursts of the resulting channel signal. In BVRF this operation is performed by calculating the values of trigonometric functions according to the data recorded in the memory.

Depending on the multiplexing factor k, output binary information is issued from the output (s) of each channel demodulating device to the input (s) of the subsequent output device (WU), one, two, or three buses (outputs), i.e. by the number of binary subchannels (corresponding to multiplexing factor k) in one channel of the system.

In the WU, the incoming to it via kN outputs of the corresponding channel sequences is folded into one binary sequence similar to the sequence on the transmitting side.

Thus, in the BVRF linear incoherent addition of each of N pairs of identical samples of channel signals is performed and the phase difference of the signal sendings across all channels is calculated for all manipulation factors, i.e. in fact, in the BVRF, they perform the function of a block of N channel demodulators, in which, unlike the block of channel demodulators of the above system [4], they demodulate each result of the linear addition of two samples of the channel signal.

Through WU here actually perform the function of a parallel-serial Converter, as in the above method of decameter radio communication [4].

The dual diversity reception is an effective means of increasing the noise immunity in radio channels with fading signals [2] and in this case to some extent compensates for the reduction in noise immunity of the reception due to an increase in the data transfer rate.

However, the data transfer rate with this method of decameter radio communication is insufficient.

In addition, with this method of radio communication, linear incoherent addition of two samples of each channel signal is performed before its demodulation, which does not provide the required signal-to-noise ratio characterizing noise immunity during channel signal demodulation.

From [6], p. 183 it is known that, with diversity reception, linear signal addition provides a smaller gain in noise immunity (in terms of signal-to-noise power) in relation to the optimal coherent addition of these same signals, since in phase-sum addition of oscillations, the signals are added algebraically, while the noise is added geometrically .

The disadvantage of linear addition is that its implementation puts forward stringent requirements for ensuring equal gain in the diversity branches. A significant difference in the gain in the branches in the limit turns the dual reception into a single one. The allowable spread of gain should provide a spread of no more than 1-2 dB in the entire dynamic range of the linear receiving path, taking into account the influence on the characteristics of the devices destabilizing factors [6], p. 180.

Moreover, a significant drawback of linear addition in this case is the insecurity of each result of the addition of two samples of any channel signal from exposure to the input of at least one of the radio receivers of an additive spectrum-concentrated (sinusoidal) noise [6], p. 7, within the frequency band occupied by the spectrum of any channel signal.

In this case, the interference voltage will also add linearly (without reducing (suppressing) its level) with the voltages of the channel signal samples, distorting the result of the summation. Upon reaching the level of interference voltage, commensurate with the resulting voltage level of the summed samples of the channel signal, the demodulation of the result of the summation by the corresponding channel demodulator can be blocked, i.e. when regenerated binary sequence at the output of the channel demodulator (for k = 1) or in each of the k regenerated binary subsequences corresponding to k outputs of the channel demodulator (for k> 1) avalanche appear erroneously received symbols with probability P _err ≈0,5.

As a result, if one of the N channel signals is concentrated by the interference concentrated in the spectrum, the output binary sequence at the output of the parallel-serial converter (to the decoder) may be distorted, for example, for k = 1, each N-th binary symbol. When N = 20, error probability (before the decoder) could reach value P _err ≤0,05, which may be critical for correcting capability of the selected correction code data communication system and not acceptable to the recipient information (after the decoder).

When exposed to two or more concentrated at a certain level interference spectrum, radio communication obviously becomes unsuitable.

Under the influence of a comparatively broadband spectrum of additive interference, for example, station noise, the occupied frequency band of which covers the frequency band occupied by more than one channel signal, then when a certain level of interference is reached, two or more parallel channels may be affected, which is also obviously not suitable for communication.

The disadvantage is that when using the known method of decameter radio communication [5], only signals with phase difference modulation can be transmitted and received at k = 1, 2, 3, which limits its functionality.

The tasks to which the invention is directed, the method of decameter radio communication with high-speed data transmission, is to increase the data transfer rate by 2 times and increase the noise immunity of receiving binary information when exposed to additive interference and fluctuation interference concentrated in the spectrum.

In addition, an additional object of the invention is to expand the functionality of the proposed method of decameter radio communication by providing transmission and reception of signals with any angular manipulation of carrier channel signals, both in phase (OFT, DOPT, etc.) and frequency (CT, DHT).

с порядковым номером i в составе дополнительного группового сигнала, после чего выходное напряжение этого БКС подают на вход соответствующего канального демодулятора с порядковым номером i дополнительного блока N канальных демодуляторов, состоящем из N однотипных канальных демодуляторов с порядковыми номерами от 1 до N, каждый из которых с порядковым номером i осуществляет демодуляцию канального сигнала, переданного на канальной частоте

с порядковым номером i в составе дополнительного группового сигнала, результаты демодуляции N канальных сигналов второго группового сигнала в виде kN канальных двоичных последовательностей, формируемых на соответствующих выходах дополнительного блока N канальных демодуляторов с канальной скоростью V_кан=1/T_кан подают на соответствующие дополнительные входы параллельно-последовательного преобразователя, в каждом БКС напряжение входного сигнала с выхода соответствующего радиоприемного устройства подают на вход соответствующего узла фазированияя, в каждом из которых напряжение входного сигнала фильтруют канальным фильтром, нормируют по уровню нормирующим усилителем и подают на первый вход второго перемножителя и первый вход первого перемножителя, выходное напряжение которого фильтруют измерительным фильтром и подают на второй вход второго перемножителя, выходное напряжение второго перемножителя каждого узла фазирования подают на соответствующий вход сумматора, выходное напряжение которого фильтруют фильтром результирующего колебания и подают на выход БКС и через нормирующий усилитель результирующего колебания подают на второй вход первого перемножителя каждого узла фазирования.The solution of the tasks is achieved by the fact that in the known method of decameter radio communication with high-speed data transmission, in which the transmitted information data stream from the message source is encoded, after which the binary sequence with speed V _u = 1 / T _u , determined by the duration of the binary symbol T _u, is input to the serial-parallel converter, with which the input sequence is partitioned kN at its outputs so that generated at these outputs FEB Meth elements corresponding kN channel sequence followed in parallel and in synchronism with the channel speed V _ch = 1 / T _ch, the duration T _ch = kNT _u binary element of each channel sequence is selected much greater than the magnitude of the average lag time rays on the radio path by selecting the required number N of parallel orthogonal channel signals at different channel frequencies as part of the transmitted group signal and multiplexing factor k of each channel signal, in block N channel mani pulyatorov consisting of N single-type channel manipulators with serial numbers from 1 to N, each arm with a sequence number i, at the initial time of each clock period of duration T _CAD supplied in parallel and synchronously k symbols corresponding k channel sequences to generate a corresponding element of the transmitted channel _kan signal duration T on a frequency channel f _i with same sequence number i, after which the formed channel manipulators channel signals are summed n output of block N channel manipulators and N-channel baseband signal with the frequency interval between neighboring channel frequencies Δf = 1 / T _ch emit broadcast via a radio transmission apparatus and the transmission antenna emitting an electromagnetic field with a horizontal polarization, while reception is carried out of two samples at the receiving end radio signals reflected from the ionosphere using two antennas receiving electromagnetic waves polarized in mutually perpendicular directions, and two radio receiving devices, with a demod yatsiyu each of the N channel signals, transmitted on the channel frequency f _i with the sequence i number in the composition group signal is performed via the corresponding channel demodulator with same sequence number i block N channel demodulators consisting of N single-type channel demodulators with serial numbers from 1 to N, N-channel demodulation of the signals in the form of binary sequences kN channel formed on the respective outputs of the block N-channel demodulators channel speed V _ch = 1 / T _ch, the hearth to respective inputs of a parallel-serial converter, the output binary sequence which is at a speed of V _u = 1 / T _u is subjected to decoding, after which the bitstream is transmitted to the recipient of messages used over a high-speed and fail-mode decameters radio communication, wherein the transmitting side speed information flow data from the message source is doubled, and the binary sequence received after encoding to the input of a serial-parallel conversion zovatelya with data rate V _u = 2 / T _u, additionally partitioned in kN additional its outputs in a way that formed on the other outputs of the binary elements of respective additional kN channel sequence followed in parallel and in synchronism with the binary elements of the channel sequence generated in kN its outputs with duct velocity V _ch = 1 / T _ch, the additional block N channel manipulators, consisting of the N-channel manipulators with similar serial numbers from 1 to N, each ma Cranes are with sequence number i, at the initial time of each clock period of duration T _CAD supplied in parallel and synchronous to the symbol corresponding to the additional channel sequences to generate a corresponding element of the transmitted channel signal to a channel frequency f _i = f _i + Δf / 2 with same sequence number i, after which the channel signals generated by the channel manipulators are summed up at the output of an additional block of N channel manipulators and an additional group N-channel group The new signal is broadcasted using an additional radio transmitting device and an additional transmitting antenna emitting an electromagnetic field with vertical polarization, and on the receiving side, the output voltages of the radio receiving devices are supplied to the respective inputs of each of 2N coherent addition units (BCS), each BCS with serial number i from the composition of the first group of N BCS with serial numbers from 1 to N performs optimal coherent summation of the voltages of two samples of the channel signal transmitted by and the channel frequency f _i with serial number i in the group signal, after which the output voltage of this BCS is fed to the input of the corresponding channel demodulator with serial number i of block N of channel demodulators, and each BCS with serial number i from the second group of N BCS with serial numbers from 1 to N, the optimal coherent summation of the voltages of two samples of the channel signal transmitted at the channel frequency

with serial number i as part of an additional group signal, after which the output voltage of this BCS is fed to the input of the corresponding channel demodulator with serial number i of an additional block of N channel demodulators, consisting of N identical channel demodulators with serial numbers from 1 to N, each of which serial number i demodulates the channel signal transmitted at the channel frequency

with sequence number i in additional baseband, results demodulation N channel signals in a second group signal in the form kN channel binary sequences generated for respective outputs additional block N channel demodulators channel speed V _ch = 1 / T _ch is fed to the respective further inputs of parallel - a serial converter, in each BCS, the voltage of the input signal from the output of the corresponding radio receiver is fed to the input of the corresponding node phasing, in each of which the input signal voltage is filtered by a channel filter, normalized by the level of a normalizing amplifier and fed to the first input of the second multiplier and the first input of the first multiplier, the output voltage of which is filtered by a measuring filter and applied to the second input of the second multiplier, the output voltage of the second multiplier of each the phasing unit is fed to the corresponding input of the adder, the output voltage of which is filtered by the filter of the resulting oscillation and fed to the output of the BCS and h Res normalizing amplifier resulting oscillation is fed to the second input of the first multiplier of each node phasing.

Comparative analysis with the prototype shows that the introduction of significant distinguishing features is new and allows, as will be shown below, to solve the tasks.

Consider the effectiveness of the invention by the example of the operation of a decameter radio communication system with high-speed data transmission, the electrical structural diagram of which is shown in FIG. one; in FIG. 2 is a schematic representation of the system signals.

The decameter radio communication system comprises a transmitting complex 1, comprising a message source 2 connected in series, an encoder 3 and a serial-parallel converter 4, the outputs of which are connected to the corresponding inputs of the first block N of channel manipulators 5 ₁ , the output of which is connected to the first radio-transmitting device 6 ₁ and connected in series the first transmitting antenna 7 ₁ other outputs of the serial-parallel converter 4 are connected to the corresponding inputs of the second block N channel manipulators headers 5 ₂ , the output of which is connected in series with the second radio transmitting device 6 ₂ and the second transmitting antenna 7 ₂ , as well as the receiving complex 8, containing two receiving antennas 9 ₁ and 9 ₂ , the output of each of which is connected to the input of the corresponding radio receiving device 10 ₁ (10 ₂ ), and the first block of N channel demodulators 11 ₁ whose outputs are connected to the corresponding inputs of the parallel-serial converter 12, the output of which is connected to the serial connected decoder 13 and the message receiver 14.

One input of each block of coherent addition (BCS) from the composition of the first group from NBKC 15 _1-1 , ..., 15 _1-N with serial numbers from 1 to N and one input of each BCS from the composition of the second group from NBKC 15 _2-1 , ... , 15 _2-N with similar serial numbers from 1 to N is combined with the output of one radio receiver 10 ₁ , and the other input of each BCS 15 _1-1 , ..., 15 _1-N and 15 _2-1 , ..., 15 _2-N combined with the output of another radio receiving device 10 _2, the output of an i-th BCS January _15-i and from the first group of N BCS with sequence number i (i = 1, 2, ..., N) is connected to the input of the respective dem undulator with same sequence i of the first block number N channel demodulators Jan. _11, and the output of i-th BCS February _15-i from the second group of N BCS to the same sequence number i (i = 1, 2, ..., N) is coupled with the input of the corresponding demodulator with the same serial number i of the second block N of channel demodulators 11 ₂ , the outputs of which are connected to the corresponding inputs of the parallel-serial converter 12.

Each BCS 15 _1-1 , ..., 15 _1-N and 15 _2-1 , ..., 15 _2-N contains two phasing nodes 16 ₁ and 16 ₂ , each of which contains a channel filter 17 connected in series, the input of which is a corresponding input BCS, the normalizing amplifier 18, the first multiplier 19, the measuring filter 20 and the second multiplier 21, the other input of which is connected to the input of the first multiplier 19.

The output of the second multiplier 21 of each phasing unit 16 ₁ (16 ₂ ) is connected to the corresponding input of the adder 22, the output of which is connected to the input of the filter of the resulting oscillation 23, the output of which, which is the output of the BCS, is connected through the normalizing amplifier of the resulting oscillation 24 to the other input of the first multiplier 19 each phasing node is 16 ₁ (16 ₂ ).

A decameter radio communication system implementing the invention operates as follows.

In the transmitting complex 1, the transmitted information data stream from the message source 2 enters the encoder 3, the task of which is to increase the noise immunity of data transmission. Coding, as a rule, is accompanied by two effective procedures — scrambling and interleaving. Scrambling converts a digital signal into a quasi-random one in order to obtain a more uniform energy spectrum of the emitted radio signal. Simple interleaving (temporal permutation) of message symbols allows to decorrelate errors in the channel, i.e. convert long-duration error packets into a series of single ones. The last operation significantly increases the coding efficiency [4].

From the output of encoder 3, a binary sequence with a speed of V _u = 1 / T _u (bit / s), where T _u is the duration of the binary element of the transmitted sequence, is fed to the input of a serial-parallel converter 4, which provides its conversion into 2kN parallel channel sequences with channel repetition rate of binary symbols each equal to V _ch = 1 / 2kNT _u (bit / s). Here N is the number of parallel orthogonal channel signals in each of the two transmitted group signals, k is the multiplicity of multiplexing of each channel signal.

The first kN parallel channel sequences with kN outputs of the serial-parallel converter 4 are supplied to the corresponding inputs of the first block N of channel manipulators 5 ₁ for generating channel signals of the first group signal, the second kN parallel subsequences from the other kN outputs of the serial-parallel converter 4 are supplied to the corresponding inputs of the second block N channel manipulators 5 ₂ for the formation of channel signals of the second group signal.

отличающейся от частоты f_i на величину Δf/2, где Δf=1/Т_кан - частотный интервал между соседними канальными частотами [5].Each block of N channel manipulators January ₅ (5 _2), which consists of N similar channel manipulators, e.g., phase or frequency, on to the inputs of each link arm at the initial time of each clock period of duration equal to the duration of the channel signal element T _ch = 2kNT _u , in parallel and synchronously k symbols of the corresponding k channel sequences are supplied. Moreover, in each clock interval, each i-th (i = 1, ..., N) manipulator of the first block of N channel manipulators 5 ₁ generates an elementary signal (package) at the corresponding channel frequency f _i , and each i-th (i = 1 , ..., N) the manipulator of the second block of N channel manipulators 5 ₂ generates an elementary signal (package) at the corresponding frequency

differs from the frequency f _i by the value Δf / 2, where Δf = 1 / T _kan - the frequency interval between neighboring channel frequencies [5].

The output signals of the channel manipulators in each block of N channel manipulators 5 ₁ (5 ₂ ) are summed, forming at the output of each block a group N - channel (frequency) group signal. The first group signal generated by the first block N of channel manipulators 5 ₁ is radiated using the first radio transmitting device 6 ₁ and the first transmitting antenna 7 ₁ . The second group signal generated by the second block N of channel manipulators 5 ₂ is radiated into the air using a second radio transmitting device 6 ₂ and a second transmitting antenna 7 ₂ .

The first antenna 7 ₁ emitting a first group signal is a transmitting antenna emitting an electromagnetic field of horizontal polarization, and the second antenna 7 ₂ emitting a second group signal is a transmitting antenna emitting an electromagnetic field of vertical polarization [7].

The first baseband signal emitted in the air in an upper sideband (e.g., radiation class J3E or R3E), represents in each transmission interval of duration T = 2kTN amount _kan N harmonic oscillations with frequencies located schematically depicted in FIG. 2 a. These frequencies, called channel, are marked on the frequency axis by points f ₁ ..., f _N. The frequencies that are far from the carrier wave f _{n1 of the} emitted single-band group signal (in the upper sideband with a width of F = 3100 Hz) by 300 and 3400 Hz, which limit the bandwidth of the telephone channel, are also indicated here. These boundary frequencies are separated from the extreme channel frequencies by ΔF. The frequency interval between adjacent channel frequencies, as noted above, is equal to Δf = 1 / T _ch.

In addition, the dotted lines conventionally indicate the frequency spectra of individual channel signals, and within the frequency band occupied by the spectrum of each channel signal, the amplitudes of the main frequency components of the spectrum of the signal, spaced from the channel frequency of each channel signal by ± f _m , are conventionally indicated by vertical lines f _m = Ω _m / 2π is the frequency of phase manipulation with an abrupt change in the phase of the circular frequency of manipulation Ω _m from 0 to π / 2 [8], p. 129.

однополосного сигнала смещена по частоте относительно f_н1 на величину Δf/2. На эту же величину смещены и канальные частоты

относительно канальных частот f₁, …, f_N.In FIG. 2b schematically shows the second single-band group signal emitted by the second transmitting antenna 7 ₂ , similar in structure to that emitted by the antenna 7 ₁ , except that the carrier frequency

a single- _band signal is shifted in frequency relative to f _n1 by Δf / 2. Channel frequencies are shifted by the same amount

relative to the channel frequencies f ₁ , ..., f _N.

It is assumed here that the power and directivity of the radiation of antennas 7 ₁ and 7 _{2 are the} same. Since the two group signals emitted by these antennas actually occupy the frequency band of the same telephone channel with a width of F = 3100 Hz (due to the relatively small value of Δf / 2, which can be on the order of several tens of hertz [5]), electromagnetic waves of transmitted group The signals are reflected from the same areas of the ionosphere and arrive at the receiving sites along the same trajectories [9].

It is known that with diversity reception, each “diversity” branch receives its own “sample” of a signal reflected from the ionosphere with a certain implementation of interference [2], in our case, to each diversity branch, consisting of a series-connected receiving antenna 9 ₁ (9 ₂ ) and a radio receiver device (RPU) 10 ₁ (10 ₂ ), receives its own sample of the total signal, consisting of samples of the first and second group signals, as well as its own sample of interference (fluctuation, concentrated in the spectrum, etc.).

Further, for convenience of presentation, the sample of the total signal at the output of the first RPU 10 ₁ (the first diversity branch) will be called the “first sample of the total signal”, and the components of this signal will be called:

- the first sample of the first group signal;

- the first sample of the second group signal;

- the first sample of the i-th channel signal of the first group signal (at the channel frequency f _i with serial number i);

с порядковым номером i);- the first sample of the i-th channel signal of the second group signal (at the channel frequency

with serial number i);

Another sample of the total signal at the output of the second RPU 102 (second diversity branch) will be called the "second sample of the total signal", and the components of this signal will be called:

- the second sample of the first group signal;

- the second sample of the second group signal;

- the second sample of the i-th channel signal of the first group signal (at the channel frequency f _i with serial number i);

с порядковым номером i).- the second sample of the i-th channel signal of the second group signal (at the channel frequency

with serial number i).

In the receiving complex 8, the antenna 9 ₁ is a receiving antenna receiving an electromagnetic field of horizontal polarization, and antenna 9 ₂ is a receiving antenna receiving an electromagnetic field of vertical polarization [7].

The samples of the total signal and additive interference received by antennas 9 ₁ and 9 ₂ from the outputs of the linear reception paths of the corresponding RPUs 10 ₁ and 10 ₂ simultaneously arrive at the corresponding two inputs of each of 2N BCS 15 _1-1 , ..., 15 _1-N and 15 _{2- 1} , ..., 15 _2-N .

второго группового сигнала после его приема РПУ 10₁ и 10₂.In each BCS 15 _1-1 , ..., 15 _1-N and 15 _2-1 , ..., 15 _2-N channel filters 17 phasing nodes 16 ₁ and 16 _{2 are} identical, and in the first group of N BCS 15 _1-1 , ..., 15 _{1-N, the} central frequencies of the passbands of the channel filters 17 of the phasing nodes 16 ₁ and 16 ₂ correspond to the channel frequencies f ₁ , ..., f _{N of the} first group signal after its reception by RPU 10 ₁ and 10 ₂ , and in the second group of N others BCS 15 _2-1 , ..., 15 _{2-N the} center frequencies of the passbands of the channel filters 17 phasing units 16 ₁ and 16 ₂ correspond to channel frequencies

the second group signal after its reception RPU 10 ₁ and 10 ₂ .

The frequency response of each of the identical channel filters 17 phasing units 16 ₁ and 16 _{2 of} each BCS 15 _1-1 , ..., 15 _1-N and 15 _2-1 , ..., 15 _{2-N is} consistent with the frequency spectrum of the corresponding channel signal of the first or second received group signals.

Schematically, the frequency response of each of the identical channel filters 17 phasing nodes 16 ₁ and 16 _{2 of} each BCS from the first group of N BCS 15 _1-1 , ..., 15 _1-N can be represented similarly to the frequency spectrum of the corresponding channel signal of the first group signal shown on FIG. 2a, and the frequency response of each of the identical channel filters 17 phasing nodes 16 ₁ and 16 _{2 of} each BCS from the second group of N BCS 15 _2-1 , ..., 15 _2-N is similar to the frequency spectrum of the corresponding channel signal of the second group signal shown in FIG. 2, b.

и

первого образца второго группового сигнала.The channel filter 17 of the first phasing unit 16 _{1 of} any i-th BCS 15 _1-i with serial number i (i = 1, ..., N) from the first group of N BCS 15 _1-1 , ..., 15 _1-N filters ( passes the input of the subsequent normalizing amplifier 18) the voltage of the first sample of the i-th channel signal at the channel frequency f _{i of the} first group signal and the voltage of the first sample of the i-th additive inter-channel interference created by the oscillations of the individual spectral components of two adjacent channel signals at the channel frequencies

and

the first sample of the second group signal.

и

второго образца второго группового сигнала.Channel filter 17 of the second phasing unit 16 _{2 of} any i-th BCS 15 _1-i with serial number i (i = 1, ..., N) from the first group of N BCS 15 _1-1 , ..., 15 _1-N filters the voltage the second sample of the i-th channel signal at the channel frequency f _{i of the} first group signal and the voltage of the second sample of the i-th additive inter-channel interference created by oscillations of individual spectral components of two adjacent channel signals at the channel frequencies

and

the second sample of the second group signal.

с порядковым номером i второго группового сигнала и напряжение первого образца i-ой аддитивной межканальной помехи, создаваемой колебаниями отдельных спектральных составляющих двух соседних канальных сигналов на частотах f_i и f_i+1 первого образца первого группового сигнала.Channel filter 17 of the first phasing unit 16 _{1 of} any i-th BCS 15 _2-i with serial number i (i = 1, ..., N) from the second group of N BCS 15 _2-1 , ..., 15 _2-N filters the voltage the first sample of the i-th channel signal at the channel frequency

with serial number i of the second group signal and the voltage of the first sample of the i-th additive inter-channel interference created by the oscillations of the individual spectral components of two adjacent channel signals at frequencies f _i and f _{i + 1 of the} first sample of the first group signal.

с порядковым номером i второго группового сигнала и напряжение второго образца i-ой аддитивной межканальной помехи, создаваемой колебаниями отдельных спектральных составляющих двух соседних канальных сигналов на канальных частотах f_i и f_i+1 второго образца первого группового сигнала.Channel filter 17 of the second phasing unit 16 _{2 of} any i-th BCS 15 _2-i with serial number i (i = 1, ..., N) from the second group of N BCS 15 _2-1 , ..., 15 _2-N filters the voltage second sample of the i-th channel signal at the channel frequency

with serial number i of the second group signal and the voltage of the second sample of the i-th additive inter-channel interference created by the oscillations of the individual spectral components of two adjacent channel signals at channel frequencies f _i and f _{i + 1 of the} second sample of the first group signal.

In addition, the voltage of the corresponding sample concentrated along the channel interference spectrum, as well as the voltage of the fluctuation interference, can fall into the passband of the channel filter 17 of the first and second phasing nodes 16 ₁ and 16 _{2 of} any BCS.

For a more detailed analysis of the operation of the receiving complex 8 of the communication system that implements the invention, we consider the process of changing the direction of polarization of the radio waves emitted by the antennas 7 ₁ and 7 _{2 of the} transmitting complex 1, after reflection from the ionosphere.

From [9], p. 276-279, it is known that a plane-polarized wave incident on the ionosphere (in our case, an electromagnetic field (EMF) with horizontal polarization, radiated by an antenna 7 ₁ or electromagnetic field with vertical polarization, radiated by an antenna 7 ₂ ) is split in general by the Earth’s magnetic field case with two elliptically polarized beams, and the resulting field at the receiving site acquires the character of an elliptically polarized field with a very elongated polarization ellipse, which is clearly illustrated in [9], p. 277, (Fig. 5.24).

Fluctuation changes in the electron concentration along the path of propagation of radio waves manifest themselves in a continuous change in the direction of the large axis of the polarization ellipse.

Moreover, it was experimentally established [9] that if one simultaneously receives such a field on two antennas with different polarizations (in our case, antennas 9 ₁ and 9 ₂ ), then fluctuations in the direction of the plane of polarization will lead to independent fading of the signal at the outputs of the corresponding two RPUs connected to these antennas (in our case, RPU 10 ₁ and 10 ₂ ).

It is noted that an increase in the signal level when receiving a vertical antenna is accompanied by a decrease in the signal level on a horizontal antenna and vice versa, which clearly indicates the existence of continuous fluctuations in the orientation of the major axis of the polarization ellipse [9], p. 227. Such fading is called polarization. A sample recording of signal levels from the outputs of two RPUs that receive on the vertical and horizontal dipoles is given in [9], p. 278 (Fig. 5.25).

In view of the foregoing, let us consider the processes of changes in the voltage levels of samples of each channel signals in the composition of the samples of the total signals at the outputs of the RPU 10 ₁ and 10 ₂ .

The polarization planes of the electromagnetic fields emitted by the antennas 7 ₁ and 7 _{2 of the} transmitting complex 1 are mutually perpendicular. At the receiving site, as mentioned above, each of the two electromagnetic fields becomes elliptically polarized, and the small and large axes of the two polarization ellipses will also be mutually orthogonal, since the channel frequencies of the first and second emitted group signals differ insignificantly (as indicated above, by a few tens of hertz) )

второго группового сигнала, примет направление, близкое к вертикали), то электродвижущая сила (эдс), наводимая в этой антенне полем i-го канального сигнала на частоте f_i первого группового сигнала будет максимальной, а эдс, наводимое полем i-го канального сигнала на частоте

второго группового сигнала, будет минимальной (около нуля). Соответственно на выходе РПУ 10₁ уровень напряжения первого образца i-го канального сигнала на частоте f_i первого группового сигнала будет максимальным, а уровень первого образца i-го канального сигнала на частоте

второго группового сигнала будет минимальной (около нуля).When these signals are received on the antenna 9 ₁ (receiving horizontal polarization EMF), when in the process of continuous changes the large axis of the polarization ellipse ([9], p. 227), for example, the field of the ith channel signal at a frequency f _{i of the} first group signal, will take a direction close to the horizontal (in this case, the large axis of the ellipse of polarization of the field of the i-th channel signal at a frequency

of the second group signal, will take a direction close to the vertical), then the electromotive force (emf) induced in this antenna by the field of the i-th channel signal at a frequency f _{i of the} first group signal will be maximum, and the emf induced by the field of the i-th channel signal by frequency

the second group signal will be minimal (near zero). Accordingly, at the output of the RPU 10 ₁ the voltage level of the first sample of the i-th channel signal at a frequency f _{i of the} first group signal will be maximum, and the level of the first sample of the i-th channel signal at a frequency

the second group signal will be minimal (near zero).

второго группового сигнала будет максимальным.In this case, during the reception of signals to the antenna 9 ₂ (receiving the electromagnetic field of vertical polarization), the opposite picture will be observed at the output of the RPU 10 ₂ : the voltage level of the second sample of the i-th channel signal at a frequency f _{i of the} first group signal will be minimal (about zero ), and the voltage level of the second sample of the i-th channel signal at a frequency

the second group signal will be maximum.

Thus, due to polarization fading, in the communication system that implements the invention, the levels of received first samples of channel signals of the first and second group signals at the output of the RPU 10 ₁ and the levels of received second samples of the same channel signals of the first and second group signals at the output of the RPU 10 _{2 are} continuously changed as follows:

второго группового сигнала, одновременно уменьшается уровень напряжения (до минимального значения - около нуля) второго образца i-го канального сигнала на частоте f_i первого группового сигнала и возрастает уровень напряжения (до максимального значения) второго образца i-го канального сигнала на частоте

второго группового сигнала.- with increasing voltage level of the first sample of the i-th channel signal at a frequency f _{i of the} first group signal, the voltage level decreases (to a minimum value of about zero) of the first sample of the i-th channel signal at a frequency

of the second group signal, at the same time, the voltage level (to a minimum value of about zero) of the second sample of the i-th channel signal at a frequency f _{i of the} first group signal decreases and the voltage level (to the maximum value) of the second sample of the i-th channel signal at a frequency increases

second group signal.

These changes in the levels of channel signals can be represented in another form:

второго группового сигнала уменьшается уровень напряжения (до минимального значения - около нуля) первого образца i-го канального сигнала на частоте f_i первого группового сигнала, одновременно уменьшается уровень напряжения (до минимального значения - около нуля) второго образца 16₁ и 16₂ канального сигнала на частоте

второго группового сигнала и увеличивается уровень напряжения (до максимального значения) второго образца i-го канального сигнала на частоте f_i первого группового сигнала.- as the voltage level increases (up to the maximum value) of the first sample of the i-th channel signal at a frequency

of the second group signal, the voltage level decreases (to a minimum value of about zero) of the first sample of the i-th channel signal at a frequency f _{i of the} first group signal, the voltage level (to the minimum value of about zero) of the second sample of the 16 ₁ and 16 ₂ channel signals decreases at frequency

the second group signal and the voltage level increases (up to the maximum value) of the second sample of the i-th channel signal at a frequency f _{i of the} first group signal.

и

напряжений образцов i-го канального сигнала на частоте

второго группового сигнала (также по упрощенным линейным законам).To illustrate the nature of the polarization fading of the voltage samples of the i-th channel signal (as part of the samples of the total signal) received RPU 10 ₁ and 10 ₂ and recorded at the outputs of the channel filters 17 phasing nodes 16 ₁ and 16 _{2 of the} i-th BCS 15 _1-i from the composition of the first group of N BCS and by analogy with the experimental data given in [9], p. 278 (Fig. 5.25), in FIG. _3a , a model of the character of fading of the amplitudes U _1Ci and U _2Ci (according to simplified linear laws) of the voltage of samples of the i-th channel signal at a frequency f _{i of the} first group signal is presented, and FIG. 3, b - model of the character of fading amplitudes

and

voltage samples of the i-th channel signal at a frequency

second group signal (also according to simplified linear laws).

Changes in the amplitudes of the voltages are presented within a certain period of time, which can be called the "half-period" of fading T _PZ (0≤t≤t _Z ) with a duration equal to 4 conditional gradations of time.

, и

первого образца второго группового сигнала при максимальном значении их амплитуд, здесь принято равным 10% от максимального значения амплитуды U_1Cimax (фиг. 3, а), что практически может несколько превышать истинное максимальное значение величины U_1МПimax,определяемое экспертным путем или экспериментально.The voltage amplitudes of the samples of channel signals are given in the form of normalized (relative) quantities - U (t) / U _1max . It also shows the nature of the fading of the average values of the stress amplitudes of the inter-channel interference samples in relative units. The maximum average value of the voltage amplitude of the first sample of the i-th inter-channel interference U _1MPimax at the output of the channel filter 17 of the first phasing unit 16 _{1 of the} i-th BCS 15 _1-i created by the voltages of two adjacent channel signals at frequencies

, and

the first sample of the second group signal at the maximum value of their amplitudes, here it is assumed equal to 10% of the maximum value of the amplitude U _1Cimax (Fig. _3a ), which can practically slightly exceed the true maximum value of the value U _1MPimax , determined by expert or experimentally.

и

второго образца второго группового сигнала при максимальном значении их амплитуд, принято также равным 10% от максимального значения амплитуды U_2Cimax (фиг. 3, а).The maximum average voltage amplitude of the second sample of the i-th inter-channel interference U _2MPimax at the output of the channel filter 17 of the second phasing unit 16 _{2 of the} i-th BCS 15 _1-i created by the voltages of two adjacent channel signals at frequencies

and

the second sample of the second group signal at the maximum value of their amplitudes, also taken equal to 10% of the maximum amplitude U _{2 Cimax} (Fig. 3, a).

и

напряжений первого и второго образцов i-ой межканальной помехи на выходах канальных фильтров 17 узлов фазирования 16₁ и 16₂ i-го БКС 15_2-i из состава второй группы из N БКС и создаваемых напряжениями образцов двух соседних канальных сигналов на частотах f_i и f_i+1 первого группового сигнала.Similarly in FIG. 3b, the nature of the change in the average amplitudes

and

voltages of the first and second samples of the i-th inter-channel interference at the outputs of the channel filters 17 phasing nodes 16 ₁ and 16 _{2 of the} i-th BCS 15 _2-i from the second group of N BCS and generated by the voltage of the samples of two adjacent channel signals at frequencies f _i and f _{i + 1 of the} first group signal.

Consider the process of isolating the voltages of channel signals, for example, the first group signal from the voltages of the samples of the total signal at the outputs of the RPU 10 ₁ and 10 ₂ using BCS from the first group of N BCS 15 _1-1 , ..., 15 _1-N .

Since the channel signals of each group signal are orthogonal, i.e. independent of each other, it is enough to consider the process of extracting one of the N channel signals of the first group signal using the example of isolating, for example, the i-th channel signal with the channel frequency f _i using the corresponding i-th BCS 15 _1-i from the first group of N BCS 15 _1-1 , ..., 15 _1-N .

Let us analyze the operation of any i-th BCS 15 _1-i from the composition of the first group of N BCS 15 _1-1 , ..., 15 _1-N , which provides selection and optimal coherent addition of voltages of two samples of the i-th channel signal of the first group signal. The operation of the BCS will be considered when receiving at any stationarity interval of duration Δt, selected in accordance with figure 3, a, b within the duration of the conditional average "half-period" fading samples of the channel signal T _PZ (0≤t≤t _Z ).

We assume that the voltage amplitudes of the samples of channel signals within each stationarity interval of duration Δt do not change. Moreover, the duration of each selected interval Δt should be greater than the time constant of the measuring filter 20 and more than the time constant of the AGC circuit of the normalizing amplifiers 18 BCS 15 _1-i , but much less than the value of T _PZ .

Under such conditions, it is natural to assume that at the end of any arbitrarily chosen stationary interval of duration Δt, all transients in each i-th BCS 15 _{1-i are} completed with the voltage levels of the channel signal samples and additive noise at the outputs of the channel filters 17 corresponding to this interval Δt.

Let from the outputs of the linear paths of the RPU 10 ₁ and 10 ₂ to the first and second inputs of the BCS 15 _1-i come within a stationary interval of duration Δt, ending, for example, at a conditional time t = 1 (Fig. 3, a, b), samples of the first group signal in the form of:

- at the first entrance -

- to the second entrance -

Here U _1Gy (t) and U _2Gy (t) are the voltages of the first and second samples of the first group signal, respectively;

U _1Ci (t) is the voltage of the first sample of the i-th channel signal;

U _2Ci (t) is the voltage of the second sample of the i-th channel signal;

U _1Ci and ϕ _1Ci - the amplitude and phase of the voltage of the first sample of the i-th channel signal;

U _2Ci and ϕ _2Ci - the amplitude and phase of the voltage of the second sample of the i-th channel signal;

ω _Ci is the angular channel (carrier) voltage frequency of the i-th channel signal (ω _Ci = 2πf _i );

θ _Ci (t) is a function that determines the type of angular manipulation of the voltage of the i-th channel signal;

N is the number of channel signals in the received group signal. Simultaneously with the outputs of the linear paths of receiving the RPU 10 ₁ and 10 _2, the first and second inputs of the BCS 15 _1-i receive voltage samples of the second group signal. By analogy with expressions (1) and (2), samples of the second group signal can be represented as:

- at the first entrance -

- at the second entrance -

и

- соответственно напряжения первого и второго образцов второго группового сигнала;Here

and

- respectively, the voltage of the first and second samples of the second group signal;

- напряжение первого образца i-го канального сигнала;

- voltage of the first sample of the i-th channel signal;

- напряжение второго образца i-го канальный сигнала;

- voltage of the second sample of the i-th channel signal;

и

- амплитуда и фаза напряжения первого образца i-го канального сигнала;

and

- the amplitude and phase of the voltage of the first sample of the i-th channel signal;

и

- амплитуда и фаза напряжения второго образца i-го канального сигнала;

and

- the amplitude and phase of the voltage of the second sample of the i-th channel signal;

- угловая канальная (несущая) частота напряжения i-го канального сигнала

;

- angular channel (carrier) voltage frequency of the i-th channel signal

;

- функция, определяющая вид угловой манипуляции i-го канального сигнала;

- a function that determines the type of angular manipulation of the i-th channel signal;

N is the number of channel signals in the second group signal. To simplify the analysis of the operation of BCS 15 _1-i, we assume that the transmission coefficient of any of the filters (17, 20, 24), as well as the adder 22 in each BCS 15 _1-1 , ..., 15 _1-N and 15 _2-1 , ..., 15 _2-N is equal to one. In addition, due to the fact that the structure of each BCS is a closed loop system of self-regulation with feedback, signal delays or changes in their initial phases when passing through these BCS filters will not be taken into account.

We also assume that at the output of the channel filter 17 of each phasing unit 16 ₁ (16 ₂ ), the maximum voltage amplitude of the sample of the i-th channel signal is much higher than the maximum average voltage amplitude of the corresponding inter-channel interference sample, i.e. U _1Cimax >> U _1MPimax and U _2Cimax >> U _2MPimax . Such amplitude ratios make it possible not to take into account the negative effect of the voltage of the inter-channel interference samples created by the second group signal, which can occur in the following cases:

1. For example, when using OFT or DOPT signals for data transmission, when in each channel signal the average number of abrupt changes in the phase of the carrier wave occurring per unit time is much less than the maximum possible number of abrupt changes in phase determined by the speed of channel signal manipulation. In this case, when the code combinations transmitted by each channel signal, consisting of a different number of symbols of the same type (in the form of long “taps” and long “depressions”), alternate at a fairly low average speed, the spectral components of the channel signal are grouped near the carrier wave of the channel signal .

The bandwidth occupied by the spectrum of such a channel signal is less than the maximum possible value of the bandwidth occupied by the channel signal, equal to Δf (Fig. 2, a, b), and the negative effect of the voltage of the interchannel interference samples can be ignored (even in the absence of polarization fading, providing Significant reduction of the negative effect of the voltage of the inter-channel interference samples due to the implementation of weighted coherent addition of the voltage of the channel signal samples, at which the voltage suppression occurs of one of the samples of this interference in that phasing unit 16 ₁ (16 ₂ ), at the output of the channel filter 17 of which the voltage of the inter-channel interference sample exceeds the voltage level of the corresponding sample of the channel signal).

2. With an increase in the multiplicity k of the channel signal of the transmitted group signal, for example, when using phase difference manipulation, which allows to reduce the speed of channel signal manipulation to a certain value at which the channel signal bandwidth becomes significantly less than Δf.

In view of the foregoing, samples of the i-th channel signal of the first group signal filtered at the inputs of the normalizing amplifiers 18 of the i-th BCS 15n, filtered by channel filters 17 of the phasing units 16 ₁ and 16 _2, can be represented as follows:

- for the first node

- for the second node

It should be noted that in the proposed communication system, RPUs 10 ₁ and 10 ₂ should operate in the off mode of their own automatic gain control (AGC) system, since the RPU AGC can only adjust the level of a group signal received in the corresponding wide band, and not each channel signal with the reception band in N of the smaller reception band of the group signal.

The AGC system of each normalizing amplifier 18 and 24 of any BCS can be characterized by the coefficient of regulation of the AGC system. The AGC control coefficient shows how many times the range of the signal at the output of the normalizing amplifier is less than at its input [6]:

where U _{IN MIN} and U _{OUT MIN} are the minimum input and minimum output voltages, which are limited by the value of the real sensitivity of the normalizing amplifier 18 of the phasing unit 16 ₁ (16 ₂ ), and U _{IN MAX} and U _{OUT MAX} are limited by the maximum value of input oscillations at which the level combinational components at the output of the normalizing amplifier 18 does not exceed permissible.

For each of the identical normalizing amplifiers 18 and 24 of any BCS 15 _1-1 , ..., 15 _1-N and 15 _2-1 , ..., 15 _2-N, we consider acceptable, for example, a change in the signal at the input of the normalizing filtered by the corresponding channel filter 17 amplifier 18 by 100 dB when the signal at its output changes by no more than 3 dB. AGC systems with such parameters are implemented in modern RPUs [10].

At the output of the normalizing amplifiers 18 of each phasing unit 16 ₁ and 16 _{2, the} filtered samples of the i-th channel signal are equalized in level and fed to the inputs of the first multipliers 19, the other inputs of which come from the normalizing amplifier of the resulting oscillation 24, the resulting signal:

where U _Pi , ω _Pi , ϕ _Pi are the amplitude, angular frequency, and phase of the resulting signal, respectively.

The output product of the first multiplier 19 of the first phasing unit 16b at one input of which receives the filtered and normalized voltage of the first sample of the i-th channel signal, and at its other input, the resulting signal, can be represented as:

where K ₁ is the value of the transfer coefficient of the normalizing amplifier 18 of the first phasing unit 16 ₁ at which normalization of the first sample of the input channel signal with amplitude U _{1Ci is provided} .

The first term in braces is easily eliminated by the measuring filter 20 of the first phasing unit 16 ₁ , because its spectrum is much higher than the spectrum of the second term.

The second term in curly brackets of expression (7) is harmonic oscillation (without manipulation) at the difference circular frequency ω _Фi = ω _Ci -ω _Pi , which coincides with the central frequency of the measuring filter 20 phasing nodes 16 ₁ and 16 ₂ . Since this oscillation is directly proportional to the amplitude of the received signal U _1Ci , then in the absence of interference at the inputs of the i-th BCS 15 _{1-i considered} , at the output of the measuring filter 20, the amplitude of this oscillation will be maximum and correspond to the maximum “weight” of the voltage of the received first sample channel signal in the normalized oscillation at the output of the normalizing amplifier 18.

The output voltage of the measuring filter 20 of the first phasing unit 16 ₁ , taking into account the foregoing, can be represented as:

For a more accurate assessment in the phasing unit 16 ₁ (16 ₂ ) of the level or "weight" of the channel signal sample in the normalized mixture of signal and noise at the output of the normalizing amplifier 18, the passband of the measuring filter 20 of each phasing unit, on the one hand, should be extremely small and, on the other hand, it is necessary that this band provides the ability to "track" the signal level during its fading and changes in the frequency of the channel signal during its reception. In the practical implementation of the communication system, this band can be selected on the order of (20-25) Hz.

Similarly to (8), one can imagine the output voltage of the measuring filter 20 of the second phasing unit 16 ₂ corresponding in this case also to the maximum “weight” of the voltage of the received second sample of the ith channel signal:

where K ₂ is the value of the transfer coefficient of the normalizing amplifier 18 of the second phasing unit 16 ₂ , which ensures normalization of the second sample of the input channel signal.

The output of the second multiplier 21 of the first phasing unit 16i will be:

Similarly, the output of the second multiplier 21 of the second phasing unit 16 _{2 is} recorded:

The first terms in braces (10) and (11) are eliminated during further filtering of the output product of the adder 22 by the filter of the resulting oscillation 23 and can be ignored. Therefore, the voltage of the first sample of the channel signal at the output of the filter of the resulting oscillation 23, which must be taken into account when summing in the adder 22 (at its first input), can be represented as:

Similarly, you can imagine the voltage of the second sample of the channel signal at the output of the filter of the resulting oscillation 23, which must be taken into account when summing in the adder 22 (at its second input):

In this case, the output voltage of the filter of the resulting oscillation 23 is written in the form:

Given that in the phasing nodes 16 ₁ and 16 _2, the voltage amplitudes of the corresponding samples of the i-th channel signal are equalized by each of the normalizing amplifiers 18 to a certain normalized value U _CH , the maximum range of which does not exceed 3 dB when the input voltage amplitude changes to 100 dB , then the magnitude of the amplitude of the output normalized oscillation U _CH with a limited, for example, up to 40 dB range of changes in the amplitude of the input oscillations, can be considered constant:

In view of (15), expression (14) can be represented as:

Using the normalizing amplifier 24, the resulting voltage U _{P CiФ} (t) is normalized by level, i.e. reduced to the form (6).

с порядковым номером i второго группового сигнала с помощью i-го БКС 15_2-i из состава второй группы N других БКС 15_2-1, …, 15_2-N.Similarly, the selection and "weight" addition of two samples of each channel signal at a frequency

with serial number i of the second group signal using the i-th BCS 15 _2-i from the second group N of other BCS 15 _2-1 , ..., 15 _2-N .

In this case, the 17 phasing nodes 16 ₁ and 16 _{2 of the} i-BCS 15 filtered by channel filters _{2 2-i} samples of the i-th channel signal of the second group signal will have the form:

- for the first node

- for the second node

The demodulation of each i-th resulting channel signal of the first group signal from the output of the i-th BCS 15 _1-i (from the first group of N BCS) is performed by the corresponding channel demodulator of the first block of N channel demodulators 11 ₁ using one of the known methods [2], and the demodulation of each i-th resulting channel signal of the second group signal from the output of the i-th BCS 15 _2-i from the second group of N BCS is performed by the corresponding channel demodulator of the second block of N channel demodulators 11 _{2 in a} similar way m

As a result, the first kN binary channel sequences are formed at the outputs of the first block of N channel demodulators 11 ₁ , and the second kN binary channel sequences are formed at the outputs of the second block of N channel demodulators 11 ₂ .

Using a parallel-serial converter 12, all 2kN channel sequences are converted into one binary sequence similar to that transmitted from the output of encoder 3, which, after decoding in decoder 13, is then transmitted to the information receiver 14.

Thus, in accordance with the proposed method of decameter radio communication, the amplitudes of the received voltages of samples of any i-th channel signal of both the first and second group signals using the corresponding i-th BCS 15 _1-i (15 _2-i are squared and added in phase at the output of this BCS or at the input of the corresponding i-th channel demodulator of the block of N channel demodulators 11 ₁ (11 ₂ ) .In this case, the data transfer rate is increased by 2 times.

Based on the above analysis method of the operation of any BCS, it is not difficult to show that the amplitudes of the received RPU 10 ₁ and 10 ₂ voltages of two samples of the additive i-th concentrated in the interference spectrum, which is (when broadcasting) harmonic oscillation at a fixed frequency falling into the band frequencies occupied by the i-th channel signal, for example, the first group signal, in the adder 22 of the i-th BCS 15 _{1-i are} added geometrically. Moreover, the larger the voltage amplitude of the sample concentrated along the interference spectrum at the output of the channel filter 17, for example, the first sample and U _1Pi (the first phasing unit 16 ₁ ) with respect to the voltage amplitude of the first sample of the channel signal U _1Ci at the output of this filter, the smaller “Weight” receives the converted voltage of this sample concentrated along the interference spectrum from the output of the second multiplier 21 to the first input of the adder 22.

As a result, when using the present invention, an increase in the signal-to-noise ratio is provided at the output of any i-th BCS 15 _1-i (at the input of the i-th demodulator of the first block of N channel demodulators 11 ₁ ) and at the output of any i-th BCS 15 _{2 -i} (at the input of the i-th demodulator of the second block of N channel demodulators 11 ₂ ) with respect to the value of a similar signal to noise ratio at the input of the i-th demodulator of the prototype [5] under the same communication conditions.

In conclusion, it should be noted that the implementation of the present invention is a decameter radio communication method with high-speed data transmission will allow to achieve the following advantages with respect to the known methods [1], [4], [5]:

1. To increase the group data transfer rate by 2 times without actually expanding the frequency band when transmitting and receiving group signals.

2. To increase the noise immunity of the reception of binary information both under the interfering effect of spectrum-focused (sinusoidal) noise and fluctuation noise.

3. To expand the functionality in relation to the known method [5] in terms of transmission and reception of a wider class of channel signals with angular manipulation: both phase-difference and frequency-manipulated for any multiplicity of channel signal multiplexing.

Information sources

1. Klovsky D.D. Theory of signal transmission. Textbook for high schools. M .: Communication. 1973. 376 p.

2. Fink L.M. Theory of discrete message transmission. M .: Soviet radio. 1970.728 s.

3. Klovsky D. D., Nikolaev B. I. Engineering implementation of radio circuits (in discrete message transmission systems under conditions of intersymbol interference). M .: Communication. 1975.200 p.

4. Kiselev A.M., Mahotin V.V., Ryzhov N.Yu., Shatalova G.V. A method for implementing a high-speed parallel modem // Radio engineering. 2006. Issue. 11. S.5-15.

5. Ginsburg VV, Girshov B.C., Zayezny A.M., Kagan B.D. Kustov OV, Okunev Yu.B. et al. Equipment for the transmission of discrete information MS-5. / Edited by Arrival A.M. and Okuneva Yu.B. - M .: Communication. 1970.152 s.

6. N.A. Sartasov, V.M. Edvabny, V.V. Mushroom. Short-wave trunk radio receivers. M .: Communication. 1971.288 s.

7. G.Z. Eisenberg, S.P. Belousov, E.M. Zhurbenko, G.A. Kliger, A.G. Kurashov. Shortwave antennas. / Edited by G.Z. Eisenberg. - 2nd, rev. and add. - M .: Radio and communications, 1985 .-- 536 p.

8. M.V. Nazarov, B.I. Kuvshinov, O.V. Popov. Theory of signal transmission. M .: Communication. 1970 .-- 368.

9. M.P. Dolukhanov. Propagation of radio waves. M .: Communication. 1972. - 336 p.

10. Berezovsky V.A., Dulkeit I.V., Savitsky O.K. Modern decameter radio communication: equipment, systems and complexes. / Ed. V.A. Berezovsky. - M .: Radio engineering. 2011 .-- 444 p.

11. I.S. Gonorovsky Radio engineering circuits and signals. Textbook for high schools. Ed. 3rd, rev. and additional M .: Sov. radio. 1977 .-- 608 p.

12. Bronstein I.N., Semendyaev K.A. Math reference. M: Phys.-Math. literature. 1962 .-- 608 p.

Claims (1)

A decameter radio communication method with high-speed data transmission, in which the transmitted information data stream from the message source is encoded on the transmitting side, after which the binary sequence with a speed V _u = 1 / T _u determined by the duration of the binary symbol T _u is fed to the input of a serial-parallel converter by means of which the input sequence is distributed on kN of its outputs in such a way that the binary elements formed on these outputs of the corresponding kN channel sequences minutes followed parallel and in synchronism with the channel speed V _ch = 1 / T _ch, the duration T _ch = kNT _u binary element of each channel sequence is selected much greater than the magnitude of the average lag time rays on the radio path by selecting the required number N of parallel orthogonal channel signals at different channel frequencies in the transmitted group signal and multiplexing factor k of each channel signal, in the block of N channel manipulators, consisting of N of the same channel manipulators serial numbers from 1 to N, each arm with a sequence number i at the initial time of each clock period of duration T _CAD supplied in parallel and synchronously k symbols corresponding k channel sequences to generate a corresponding element of the transmitted channel signal duration T _kan on channel frequency f _i with by the same serial number i, after which the channel signals generated by the channel manipulators are summed at the output of the block of N channel manipulators and the N-channel group signal from the frequency interval between neighboring channel frequencies Δf = 1 / T _ch emit broadcast via a radio transmission apparatus and the transmission antenna emitting an electromagnetic field with a horizontal polarization, and on the reception side is carried out taking two samples reflected from the ionosphere radio signals via two antennas, the receiving electromagnetic waves polarized in mutually perpendicular directions, and two radio receivers, while demodulating each of the N channel signals transmitted on the channel frequency f _i with serial number i in the group signal, is carried out using the corresponding channel demodulator with the same serial number i of the block N channel demodulators, consisting of N of the same channel demodulators with serial numbers from 1 to N, the results of the demodulation of N channel signals in the form kN channel binary sequences, formed on respective unit outputs N-channel demodulators channel speed V _ch = 1 / T _ch is supplied to respective inputs of a parallel-serial etc. a generator whose output binary sequence is decoded at a speed of V _u = 1 / T _u , after which the binary stream is transmitted to the message recipient, characterized in that on the transmitting side, the speed of the information data stream from the message source is doubled, and the binary sequence received after encoding the input serial-to-parallel converter with data rate V _u = 2 / T _u, to further distribute additional kN its outputs so that formed n additional outputs binary elements of respective additional kN channel sequence followed in parallel and in synchronism with the binary elements of the channel sequence generated in kN it outputs a channel speed V _ch = 1 / T _ch, the additional block N channel manipulators, consisting of N single-type channel manipulators with sequence numbered from 1 to N, each arm with a sequence number i, at the initial time of each clock period of duration T _CAD supplied in parallel and synchronous k on k symbols corresponding additional channel sequences to generate a corresponding element of the transmitted channel signal on a channel frequency

with the same serial number i, after which the channel signals generated by the channel manipulators are summed up at the output of the additional block of N channel manipulators and the additional group N - channel group signal is broadcast using an additional radio transmitting device and an additional transmitting antenna emitting an electromagnetic field with vertical polarization, and on the receiving side, the output voltages of the radio receivers are fed to the corresponding inputs of each of the 2N coherent blocks I (BCS), wherein each BCS with serial number i from the first group of N BCS with serial numbers from 1 to N provides the optimum coherent summation of voltages of the two samples of channel signal transmitted on the channel frequency f _i with sequence number i in the composition of the baseband signal and then the output voltage of this BCS is fed to the input of the corresponding channel demodulator with serial number i of the block N of channel demodulators, and each BCS with serial number i from the second group of N BCS with serial numbers from 1 to N performs optimal coherent summation of the voltages of two samples of the channel signal transmitted at the channel frequency

with serial number i as part of an additional group signal, after which the output voltage of this BCS is fed to the input of the corresponding channel demodulator with serial number i of an additional block of N channel demodulators, consisting of N identical channel demodulators with serial numbers from 1 to N, each of which serial number i demodulates the channel signal transmitted at the channel frequency

with sequence number i in additional baseband, results demodulation N channel signals in a second group signal in the form kN channel binary sequences generated for respective outputs additional block N channel demodulators channel speed V _ch = 1 / T _ch is supplied to the corresponding additional inputs parallel-serial converter, in each BCS, the voltage of the input signal from the output of the corresponding radio receiver is fed to the input of the corresponding node and phasing, in each of which the voltage of the input signal is filtered by a channel filter, normalized by the level of a normalizing amplifier and fed to the first input of the second multiplier and the first input of the first multiplier, the output voltage of which is filtered by a measuring filter and applied to the second input of the second multiplier, the output voltage of the second multiplier each phasing node is fed to the corresponding input of the adder, the output voltage of which is filtered by the filter of the resulting oscillation and fed to the output of the BCS and h A normalizing amplifier of the resulting oscillation is fed to the second input of the first multiplier of each phasing unit.

Images