RU2572002C1 - Voltage converter control method - Google Patents

Abstract

FIELD: electricity.

SUBSTANCE: in voltage converter control method two sequences of paraphrase pulse signals are generated - the first sequence and the second sequence, at that the second sequence of paraphrase pulse signals is shifted in regard to the first sequence of paraphrase pulse signals per adjusted time period. The first and second transistors are controlled by paraphrase pulse signals of their first sequence while the third and fourth transistors are controlled by paraphrase pulse signals of their second sequence.

EFFECT: improving operating reliability of voltage converter.

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Description

A device in which the proposed method for controlling power transistors (power switches) is implemented relates to a power converter technique. The device is designed to convert and control the energy consumed from a direct current source, and transfer the converted energy to its receiver using transformer coupling between the source and receiver power circuits.

A known voltage converter, which contains power controlled keys connected by a bridge circuit, as well as a circuit that is formed by a winding of a magnetic energy storage device, a capacitor and a primary winding of a power transformer. These elements are connected in series. The primary winding of the transformer is magnetically connected to the secondary winding connected through a rectifier to the capacitor of the output filter. The load and the capacitor of the output filter are connected in parallel (V.I. Meleshin. Transistor converter technology. - M .: Technosphere. 2005., Fig. 13.7-a, p. 295).

In the known device for regulating the energy transmitted to the consumer, resonance phenomena occurring in an LC circuit formed by a winding of a magnetic energy storage device and a capacitor are used, which are connected to each other in series.

The control of power transistors forming a bridge circuit is carried out by alternately unlocking for a time equal to half the period of operation of the device (actually slightly less than half the period), two pairs of transistors, and in each pair the transistors belong to two opposite branches of the bridge circuit.

The energy transferred to the consumer is regulated by changing the switching frequency of the power transistors, i.e. the frequency of the device. In this case, two control modes are possible.

The first mode is implemented provided that within the frequency range used for regulation, the operating frequencies are lower than the resonant frequency of the LC circuit. In this case, the current pulses of the output circuit of the secondary rectifier current flowing into the capacitor of the output filter have the form of half waves separated by pauses. By reducing the operating frequency, a reduction in the average power transmitted to the load is achieved.

The disadvantage of this control mode is that the pulsed currents of power transistors and valve elements of the output rectifier significantly exceed the average currents, which are proportional to the power transmitted to the load. Because of this, there is a need to increase the installed power of transistors and valve elements at a given value of power in the load. In addition, the heat loss power in the circuit elements is proportional to the square of the instantaneous current values, and therefore it grows more sharply than the average power transmitted to the load. As a result, the energy conversion efficiency decreases (efficiency).

The second control mode of the power transistors of the bridge circuit is implemented provided that within the frequency range used for regulation, the operating frequencies are higher than the resonant frequency of the LC circuit.

In the second control mode, the rectified current of the secondary winding is a continuous sequence of current pulses, gradually increasing from zero and gradually falling to zero. By increasing the operating frequency, the area of these pulses (their amplitude and duration) is reduced and, as a result, the power transmitted to the load is reduced. The disadvantage of the second control mode is that a decrease in the output power with an increase in the operating frequency is accompanied by an increase in switching losses, since the number of switching per unit time increases. In addition, it is difficult to implement the idle mode, since for this the switching frequency must be increased many times, which is accompanied by a decrease in efficiency energy conversion.

A known method of controlling power transistors of a bridge circuit, called "phase control" ("phase-shift pulse wight modulation" - "phase-shift PWM", Eng.). The method consists in the fact that the first and second transistors connected in series in the bridge circuit are controlled by paraphase pulse signals of the first sequence. The third and fourth transistors, also connected in series in the bridge circuit, are controlled by paraphase pulse signals of their second sequence. In this case, the second sequence of paraphase pulse signals is shifted relative to the first sequence by an adjustable time. By changing the shear time between the pulse sequences, the output voltage is regulated.

Phase control is used in bridge circuits in which the primary winding of the power transformer is connected to the output circuit through the inductor winding, and the output filter of the device is equivalent to a series LC circuit. The filter is connected to the secondary winding of the power transformer (Meleshin V.I., Ovchinnikov D.A. Control of transistor power converters. - M .: Technosphere. 2011. - 576 p., Chapter 3).

The inductor in the primary circuit of the transformer is designed to store enough energy to ensure the resonant process of recharging the capacitors of power transistors and unlocking them at zero voltage. The inductance of the inductor winding in the primary winding of the transformer limits the slew rate of the primary winding current. Until the instantaneous value of the primary winding current reaches the level that is transformed from the secondary winding to the primary winding, the voltage on the transformer windings is zero, and it does not transmit energy. Thus, even with the maximum value of the delay time of the second sequence of control paraphase pulses relative to the first sequence (the maximum delay time is equal to half the period of the circuit), there is some minimum pause time in the energy transfer by the transformer. To reduce this time, the inductance of the inductor winding in the primary circuit of the transformer and, accordingly, the energy stored in the inductor, are reduced to a minimum, but so that this minimum would be sufficient to implement the unlocking mode of the power transistors at zero voltage.

Regulation is carried out by changing the pause in the energy transfer by the transformer. Its minimum value, as noted, is due to the presence of inductance in the primary circuit of the transformer. The converter as a whole with this regulation is equivalent to a voltage source with respect to the load. The average level of the equivalent voltage source is proportional to the product of the supply voltage and the duration of the energy transfer by the transformer in each cycle, related to the duration of the cycle. The current of the output circuit of the converter during phase control is determined by the load resistance, and with a low value of this voltage, the current of the output circuit can take unacceptably high values. This makes complicating both the power circuit the introduction of an output circuit current sensor into it, and the control algorithm, as well as a control device that implements this algorithm. The task of this complication is to prevent current overload of the converter output circuit. The need to complicate the control algorithm and the control device that implements the algorithm also arises when creating a system of converters, the output circuits of which are connected in parallel and connected to a common load. In this case, the purpose of complication is to prevent current overload of individual converters in the system and to ensure the alignment of currents in their output circuits.

The essence of the proposal contained in this application is the use of phase control of power transistors of a voltage converter, the power circuit of which is similar in topology to a bridge resonant DC / DC converter.

The object in which the control method is implemented is a voltage converter. The device contains transistors (power controlled keys) forming a bridge circuit and a two-terminal device for its load. The first and second transistors of the bridge circuit, connected in series, form the first transistor circuit, which is connected between the power buses. The third and fourth bridge transistors connected in series form a second transistor circuit that is connected between the power buses. The midpoints of the first and second transistor circuits are respectively the first and second terminals of the output circuit of the bridge circuit, and the first and second terminals of the bipolar load of the bridge circuit are connected to them. The bipolar is made in the form of inductive and capacitive energy storage devices connected in series, as well as the primary winding of the transformer (or the primary windings of several transformers). The secondary winding of the transformer through the rectifier is connected to the capacitor of the output filter. If several transformers are used, then in each of them the secondary winding is connected to the output filter capacitor through the corresponding rectifier

The method of controlling the voltage converter is that they form two sequences of paraphase pulse signals - the first and second, and the second sequence of paraphase pulse signals is shifted relative to the first sequence by an adjustable time. The first and second transistors are controlled by paraphase pulse signals of their first sequence, and the third and fourth transistors are controlled by paraphase pulse signals of their second sequence.

The voltage converter when using this control method with respect to the control method by changing the switching frequency of the power transistors discussed above, gets new properties. Namely:

, является ограниченной по величине функцией трех переменных. Первая переменная - напряжение питания Е, которому значение $${\overline{I}}_{out}$$

, пропорционально. Вторая переменная - выходное напряжения U_out. При его уменьшении значение $${\overline{I}}_{out}$$

возрастает, но остается ограниченным даже в режиме короткого замыкания, когда U_out=0. Третья переменная - регулирующий параметр D. Он равен длительности сдвига по времени между двумя последовательностями парафазных сигналов управления, отнесенной к длительности тактов работы DC/DC-преобразователя, т.е. к половине периода работы этого устройства.1. With respect to the load, the converter acts as a current source. Its average value $${\overline{I}}_{out}$$

is a function of three variables limited in size. The first variable is the supply voltage E, to which the value $${\overline{I}}_{out}$$

proportionally. The second variable is the output voltage U _out . When it decreases, the value $${\overline{I}}_{out}$$

increases, but remains limited even in short circuit mode when U _out = 0. The third variable is the control parameter D. It is equal to the duration of the time shift between two sequences of paraphase control signals related to the duration of the clock cycles of the DC / DC converter, i.e. by half the period of operation of this device.

монотонно нарастает от нуля до максимума. Этот максимум ограничен и зависит от величин Е и U_out. Ограниченными также являются амплитудные значения тока, поступающего в первичную обмотку трансформатора от выходной цепи транзисторной мостовой схемы, и, соответственно, тока вторичной обмотки, передаваемого через выпрямитель в нагрузку.2. When the regulatory parameter increases from zero to unity, the value $${\overline{I}}_{out}$$

monotonously grows from zero to maximum. This maximum is limited and depends on the values of E and U _out . Also limited are the amplitude values of the current entering the transformer primary winding from the output circuit of the transistor bridge circuit, and, accordingly, the secondary winding current transmitted through the rectifier to the load.

, выраженной в относительных (relative) единицах, где $$U{\text{'}}_2$$

, - напряжение, трансформируемое из вторичной обмотки в первичную обмотку; N_tr - число трансформаторов, первичные обмотки которых соединены последовательно, а вторичная обмотка каждого из трансформаторов через соответствующий выпрямитель подключена к конденсатору выходного фильтра, наряженному до напряжения U_out. Согласно результатам моделирования электрических процессов для диапазона изменения питающего и выходного напряжений, характеризуемого соотношением 0.45≤U_r≤0.8, максимум выходной мощности, достигаемый в режиме D→1, не выходит за пределы ±10% по отношению к усредненному уровню максимумов мощности, отвечающих указанному диапазону.3. In a wide range of changes in the supply and output voltages, there is a parametric stabilization of the maximum output power achieved in the D → 1 mode. The specified range can be characterized by a change in value. $$U_r=\frac{N_{tr}\cdot U{\text{'}\text{'}}_2}{E}$$

expressed in relative units, where $$U{\text{'}\text{'}}_2$$

, is the voltage transformed from the secondary winding to the primary winding; N _tr is the number of transformers whose primary windings are connected in series, and the secondary winding of each of the transformers is connected through an appropriate rectifier to the output filter capacitor dressed up to voltage U _out . According to the results of modeling electrical processes for the range of supply and output voltages characterized by the ratio 0.45≤U _r ≤0.8, the maximum output power achieved in D → 1 mode does not exceed ± 10% with respect to the average level of power maxima corresponding to the specified range.

4. The current flowing through the primary and secondary windings of the transformer is presented in the form of a sequence of pulses, gradually increasing from zero and gradually decreasing to zero. In operating modes close to the maximum output power mode, current pulses follow continuously (without pauses). The shape of these pulses is closer to rectangular than in the same converter, but in the frequency control mode of its output power.

The above new properties of the converter, which are acquired when using phase control transistors of a bridge circuit of a resonant type, lead to several advantages with respect to the same circuit, but provided that the output power is controlled by changing the switching frequency of the transistors (regulation by pulse frequency modulation - CHIM).

Property 1 means that the frequency of the device is constant. Because of this, there is no reason for the increase in switching loss power caused by the increase in the number of switching per unit time. In addition, the frequency spectrum of the output voltage ripple is reduced, which simplifies its filtering.

Property 2 means that for any value of the supply voltage, it is possible to stabilize the output voltage in idle mode. The converter does not have this property when using PFM.

Property 3 means that at a given level of output power of the converter, it is possible to regulate its output voltage over a wide range, provided that there is a slight change in the supply voltage, or it is possible to stabilize this value of the output voltage of the device, provided that the supply voltage changes significantly. The converter does not have this property when using PFM.

Properties 2 and 3 mean that it is possible to connect the output circuits of several converters in parallel to the total load without complicating the control system of these devices. There is no danger of overcurrent in current and power of each of the converters.

Property 4 means that for each given average value of the current flowing through the transformer windings, its amplitude value is reduced. Accordingly, the heat loss power in the transformer and output rectifier decreases.

The above-mentioned new properties of a resonator-type converter, which are acquired using phase control of bridge transistors to control the output power, lead to several advantages with respect to converter circuits, where phase control of bridge transistors is also used, but the circuit topology is different. In them, the primary winding of the power transformer is connected to the output circuit of the transistor bridge through the inductor winding, and the output filter of the device is equivalent to a serial LC circuit. The filter is connected to the secondary winding of the power transformer. (Meleshin V.I., Ovchinnikov D.A. Control of transistor converters of electricity. - M.: Technosphere. 2011. - 576 p., Chapter 3).

Properties 1, 2, and 3 are not characteristic of known circuits with phase control of bridge transistors. The converter as a whole with this regulation is equivalent to a voltage source with respect to the load. The average level of the equivalent voltage source is proportional to the product of the supply voltage and the duration of the energy transfer by the transformer in each cycle, related to the duration of the cycle. The current of the output circuit of the converter during phase control is determined by the load resistance, and with a low value of this voltage, the current of the output circuit can take unacceptably high values. The disadvantages of this property of the known devices have been considered previously.

The rise and fall rates of the transformer winding currents in phase-controlled converters of bridge transistors having the above topology are significantly higher than in a resonant-type converter with the same control method. Therefore, in a resonant-type converter, the switching processes in the rectifier valve elements are accompanied by a lower level of high-frequency interference, due to which noise-suppressing filters of lower energy consumption, mass and dimensions can be used.

The essence of the proposal contained in this application is the use of phase control of power transistors of an electric energy conversion device, the power circuit of which is similar in topology to a bridge resonant DC / DC converter.

The object in which the method of phase control of power transistors is implemented is a converter, the circuit of which is shown in FIG. one.

A power source 3 is connected to the power buses 1 and 2 of the voltage converter, which is, for example, a constant voltage source E. The converter contains transistors 4, 5, 6 and 7 (power controlled keys) forming a bridge circuit.

The first and second transistors 4 and 5 of the bridge circuit connected in series form a first transistor circuit that is connected between the power lines 1 and 2. The third and fourth transistors of the bridge circuit 6 and 7 connected in series form a second transistor circuit that is connected between the power lines 1 and 2. The midpoints of the first and second transistor circuits are respectively the first and second terminals of the output circuit of the transistor bridge circuit.

A two-terminal device is connected to the terminals of the output circuit of the transistor bridge circuit, which contains the winding of the inductive energy storage device 8 connected in series, the primary winding of the transformer 10 and the capacitor 11, which is a capacitive energy storage device.

The secondary winding 12 of the transformer 10 through the valve elements of the rectifier is connected to the capacitor of the output filter 17. In FIG. 1 valve elements are diodes 13, 14, 15 and 16, connected by a bridge circuit. A DC load 18 is connected in parallel with the output filter capacitor.

The design of the secondary winding, as well as the design of the rectifier, are not essential features of the device. So, for example, the secondary winding may contain two sections. In this case, the middle point of the secondary winding is connected directly to the first output of the output filter capacitor, and the extreme terminals of the two-section secondary winding are connected to the second output of the output filter capacitor through the rectifier valve elements. The only important thing is that the secondary winding through the rectifier is connected to the capacitor of the output filter.

Diagrams of two sequences of paraphase pulse control signals of power transistors 4, 5, 6 and 7, connected by a bridge circuit, are presented in FIG. 2.

, f - частота работы схемы.Pulse signals U _A and U _B refer to the first sequence of paraphase signals. There is a pause of duration T _p1 , during which the voltage of the signals U _A and U _B are simultaneously zero, and in reality T _p1 << T, where $$T=\frac{\mathrm{one}}{f}$$

, f is the frequency of the circuit.

Signals U _A and U _{B are} controlled by power transistors 4 and 5. A pause between these signals is necessary to exclude the possibility of simultaneous unlocking of transistors 4 and 5, which would lead to the release of significant heat loss power in them.

Pulse signals U _C and U _D form a second sequence of paraphase signals. There is a pause of duration T _p2 , during which the signal voltages U _C and U _D are simultaneously equal to zero, and in fact T _p2 << T.

Signals U _C and U _{D are} controlled by power transistors 6 and 7. A pause between these signals is necessary to exclude the possibility of simultaneous unlocking of transistors 6 and 7, which would lead to the release of significant heat loss power in them.

In the general case, it is possible to individually set pause intervals between pulse signals in their first and second sequences. The diagrams in FIG. 2 correspond to the case when T _p1 = T _p2 = T.

, где D - регулирующий параметр. Изменением его в пределах от нуля до единицы обеспечивают регулирование мощности в нагрузке.The second sequence of paraphase pulsed signals is delayed relative to the first sequence for a while $$T_d=D\cdot \frac{\mathrm{<figure-callout\; id="2"\; label="T"\; filenames="00000065.png"\; state="\{\{state\}\}">T</figure-callout>}}{2}$$

where D is the regulatory parameter. By changing it from zero to unity, power is regulated in the load.

The properties of the converter when using phase control of its transistors follow from the nature of the electrical processes occurring in the circuit, which are discussed below. The processes are illustrated by diagrams of the change in time of electrical quantities obtained as a result of modeling using the PSpice software package. The diagrams are shown in FIG. 3, 4 and 5. They correspond to the scheme in which the transformer is made with the same number of turns of the primary and secondary windings, i.e. K _tr = 1. This condition is not fundamental, but the illustration of the processes during its implementation is simplified.

The voltage converter shown in FIG. 1. works in one of two modes. The first of them is characterized by the continuity of energy transfer by the transformer 10. It corresponds to the absence of pauses in the current flow through the secondary winding 12. rectified with the help of diodes 13, 14, 15 and 16. The first mode is realized if the parameter D, which is changed during the regulation, meets the condition D _lim ≤D≤1.

If the current flows through the secondary winding in the direction from the end of the winding to its beginning, marked with a dot in FIG. 1, then the diodes 14 and 15 are in the conduction state. In this case, the voltage on the secondary winding has a positive polarity (plus - at the beginning of the winding, minus - at its end). The voltage value of positive polarity on the secondary winding 12 exceeds the voltage to which the capacitor 17 of the output filter is charged by an amount equal to the voltage drop across the rectifier valve elements in a state of direct conduction (in this case, they are diodes 14 and 15).

In real conditions, the voltage ripple across the capacitor 17 of the output filter is negligible. In addition, the voltage drops across the valve elements of the secondary rectifier are negligible compared to the average voltage level U _out on the capacitor 17. Therefore, without a significant error, we can assume that while the current flows through the secondary winding 12 in the direction from its end to the beginning, the voltage positive polarity on it does not change in time, and it is equal to U _out .

If the current flows through the secondary winding in the direction from the beginning of the winding to its end, then the diodes 16 and 13 are in the conduction state. The voltage on the secondary winding has a negative polarity (minus at the beginning of the winding, plus at its end). The absolute value of the voltage of negative polarity on the secondary winding 12 exceeds the voltage to which the capacitor 17 of the output filter is charged by an amount equal to the voltage drop across the rectifier valve elements in direct conduction (in this case, they are diodes 13 and 16).

By analogy with the voltage interval of positive polarity on the secondary winding 12, we can assume that while the current flows through the winding from the beginning to the end, the voltage of negative polarity on it does not change in time, and its absolute value is U _out .

Thus, if we take into account the magnetic coupling between the transformer windings, it is typical for the converter to operate in the first mode that rectangular voltage pulses of positive polarity on the windings without a pause are replaced by voltage pulses of negative polarity of the same amplitude. Due to the symmetry of the topology of the circuit, as well as the control signals for the clock cycles of the device, the positive and negative pulses are the same in duration, which is half the period of operation of the device.

, где $$K_{tr}=\frac{W_2}{W_1}$$

. С учетом изложенного выше: U₂≈U_out, $$U_1\approx \frac{U_{out}}{K_{tr}}$$

.The voltage U ₁ on the primary winding (9 in FIG. 1), the number of turns of which is equal to W ₁ , is related to the transformation ratio with the voltage U ₂ on the secondary winding (12 in FIG. 1) having the number of turns W ₂ . Namely, $$U_{\mathrm{one}}=\frac{U_2}{K_{tr}}$$

where $$K_{tr}=\frac{W_2}{W_{\mathrm{one}}}$$

. In view of the above: U ₂ ≈U _out , $$U_{\mathrm{one}}\approx \frac{U_{out}}{K_{tr}}$$

.

The electrical processes in the circuit during the implementation of the first mode of its operation can be considered only for the interval when the voltage on the transformer windings is positive. In the interval when these stresses are negative, the processes are similar.

. Поскольку $$U_1\approx \frac{U_{out}}{K_{tr}}=const$$

, справедливо выражение $$\Phi \left(t\right)=-{\Phi }_{\max }+\frac{U_1}{W_1}\cdot \left(t-t_0\right)$$

.Due to the symmetry of the positive and negative polarity voltages on the transformer windings in magnitude and duration, the magnetic flux Φ in the transformer varies symmetrically with time in relation to zero. Therefore, by the time t _{0 of the} beginning of the interval of positive polarity of the voltage across the windings, the magnetic flux reaches a negative amplitude value in sign, i.e. Φ (t ₀ ) = - Φ _max , and as a whole, the flow increases in this interval. Its increase in time occurs in accordance with the law of electromagnetic induction, i.e. $$\Phi \left(t\right)=\Phi \left(t_0\right)+\frac{\mathrm{one}}{W_{\mathrm{one}}}\cdot {\displaystyle \underset{t_0}{\overset{t}{\int }}{U}_{\mathrm{one}}\left(t\right)dt}$$

. Insofar as $$U_{\mathrm{one}}\approx \frac{U_{out}}{K_{tr}}=const$$

, fair expression $$\Phi \left(t\right)=-{\Phi }_{\max }+\frac{U_{\mathrm{one}}}{W_{\mathrm{one}}}\cdot \left(t-t_0\right)$$

.

, а также принимая во внимание отмеченную выше линейность изменения во времени знакопеременного магнитного потока, его амплитуда может быть представлена в виде $${\Phi }_{\max }=\frac{U_1\cdot T}{4\cdot W_1}$$

.By the end of the interval of positive voltage polarity on the transformer windings, the magnetic flux in it reaches a positive amplitude value + Φ _max . Given the duration of this interval, equal to $$\frac{\mathrm{<figure-callout\; id="2"\; label="T"\; filenames="00000065.png"\; state="\{\{state\}\}">T</figure-callout>}}{2}$$

, and also taking into account the linearity of the time-varying magnetic flux noted above, its amplitude can be represented as $${\Phi }_{\max }=\frac{U_{\mathrm{one}}\cdot T}{\mathrm{four}\cdot W_{\mathrm{one}}}$$

.

, где L_mag - параметр магнитопровода трансформатора. Физически он эквивалентен индуктивности одновитковой обмотки трансформатора, сцепленной с его магнитным потоком Φ.Actually, the magnetic characteristic of the transformer is almost linear, i.e. there is a proportional relationship between the magnetic flux Φ and the magnetomotive force F, which corresponds to the equality $$F=\frac{\Phi }{L_{mag}}$$

where L _mag is the transformer magnetic circuit parameter. Physically, it is equivalent to the inductance of a single-turn winding of a transformer coupled to its magnetic flux Φ.

, где $$F_{\max }=\frac{U_1\cdot T}{4\cdot L_{mag}\cdot W_1}$$

.Given the linearity of the magnetic characteristics of the transformer, as well as the law of the change in time of the magnetic flux coupled to its windings, the change in time of the magnetomotive force is described by the expression $$F\left(t\right)=-F_{\max }+\frac{U_{\mathrm{one}}}{L_{mag}\cdot W_{\mathrm{one}}}\cdot \left(t-t_0\right)$$

where $$F_{\max }=\frac{U_{\mathrm{one}}\cdot T}{\mathrm{four}\cdot L_{mag}\cdot W_{\mathrm{one}}}$$

.

The magnetomotive force (MDF) in the transformer is created by the currents of its windings, and it is equal to F (t) = W ₁ · I ₁ (t) + W ₂ · I ₂ (t). The current of the winding is positive, and it creates a positive component of the ISF, if the current is directed from the beginning of the winding to the end. The current of the winding is negative, and it creates a negative component of the ISF, if the current is directed from the end of the winding to the beginning.

. Ранее было показано, что положительной полярности напряжений на обмотках трансформатора отвечает отрицательный знак тока вторичной обмотки. Следовательно, на интервале положительной полярности напряжений справедливо соотношение $$I_1\left(t\right)\ge \frac{F\left(t\right)}{W_1}$$

. В момент начала этого интервала I₁(t₀)=I_µ,max, где $$I_{\mu ,\max }=\frac{U_1\cdot T}{4\cdot L_{mag}\cdot {\left(W_1\right)}^2}$$

. Через время, равное половине периода, когда заканчивается интервал положительной полярности напряжений на обмотках трансформатора, I₁(t₀+T/2)=+I_µ,max. В остальные моменты времени этого интервала $$I_1\left(t\right)>\frac{F\left(t\right)}{W_1}$$

.The change in time of the current of the secondary winding of the transformer is presented in the form $$I_2\left(t\right)=-\frac{\mathrm{one}}{K_{tr}}\cdot \left(I_{\mathrm{one}}\left(t\right)-\frac{F\left(t\right)}{W_{\mathrm{one}}}\right)$$

. It was previously shown that the positive polarity of the voltage on the transformer windings corresponds to the negative sign of the secondary current. Therefore, in the interval of positive voltage polarity, the relation $$I_{\mathrm{one}}\left(t\right)\ge \frac{F\left(t\right)}{W_{\mathrm{one}}}$$

. At the beginning of this interval, I ₁ (t ₀ ) = I _{µ, max} , where $$I_{\mu ,\max }=\frac{U_{\mathrm{one}}\cdot T}{\mathrm{four}\cdot L_{mag}\cdot {\left(W_{\mathrm{one}}\right)}^2}$$

. After a time equal to half the period when the interval of positive polarity of the voltage across the transformer windings ends, I ₁ (t ₀ + T / 2) = + I _{µ, max} . At other time points of this interval $$I_{\mathrm{one}}\left(t\right)>\frac{F\left(t\right)}{W_{\mathrm{one}}}$$

.

The current I _{1 of the} primary winding 9 of the transformer 10 flows through the winding of the inductor 8, the inductance of which is indicated by the symbol L ₁ , and also closes through the capacitor 11, the capacitance of which is indicated as C ₁ , since these elements are connected in series with the primary winding 9.

Note. Further in the text, the voltage applied to the winding of the inductor 8 is indicated by the symbol "U _L1 ". The voltage across the capacitor 11 is indicated by the symbol "U _C1 ". As for the current of these elements, it is the current I ₁ , common both for them and for the primary winding 9 of the transformer 10.

The presence of a capacitor 11 in the current loop I ₁ means that this current does not have a constant component. Accordingly, the voltage across the capacitor 11 is an alternating function symmetrical about zero.

At the beginning of the interval of positive polarity of the voltage across the transformer windings, the current I ₁ and voltage U _C1 , if displayed by vectors, coincide in direction. This means that the capacitor 11 is a source of energy. Part of this energy is transmitted to the inductor 8, and another part of it enters the primary winding 9 of the transformer 10. It should be noted that throughout the entire interval of positive polarity of the voltage across the windings of the transformer 10, its primary winding 9 is a receiver of energy that is carried by current I ₁ , since voltage U ₁ on this winding and current I ₁ , if they are displayed by vectors, are directed towards each other.

. Протекание тока I₁ через конденсатор 11 вызывает сначала плавное уменьшение напряжения положительной полярности на нем. При этом конденсатор разряжается током I₁, отдавая накопленную энергию. Затем напряжение на конденсаторе изменяет знак, и его абсолютное значение начинает нарастать (фиг. 3 и 4). Конденсатор заряжается током I₁, запасая энергию.At the first stage of the interval of positive polarity of the voltage on the transformer windings, current I ₁ closes along the circuit: power supply 3 - transistor 4 of the bridge circuit - winding of the inductor 8 - primary winding 9 of the transformer 10 - capacitor 11 - transistor 7 of the bridge circuit - power source 3. At the same time energy is stored in the inductor 8, and the current I ₁ increases at a speed determined by the expression $$\frac{d{I}_{\mathrm{one}}}{dt}=\frac{E+U_{C\mathrm{one}}-U_{\mathrm{one}}}{L_{\mathrm{one}}}$$

. The flow of current I ₁ through the capacitor 11 first causes a smooth decrease in voltage of positive polarity on it. In this case, the capacitor is discharged by the current I ₁ , giving up the stored energy. Then the voltage across the capacitor changes sign, and its absolute value begins to increase (Figs. 3 and 4). The capacitor is charged by current I ₁ , storing energy.

и U_C1, улучшает соотношение между средним значением $${\overline{I}}_1$$

и амплитудным значением I_1,max тока I₁ на интервале положительной полярности напряжений на обмотках трансформатора. Это улучшение состоит в том, что при данной мощности, передаваемой в первичную обмотку трансформатора, которая равна $$P_1={\overline{I}}_1\cdot U_1$$

, уменьшается значение I_1,max.The nature of the change in voltage across the capacitor 11, if we take into account the relationship between the quantities $$\frac{d{I}_{\mathrm{one}}}{dt}$$

and U _C1 , improves the relationship between the average $${\overline{I}}_{\mathrm{one}}$$

and the amplitude value of I _{1, max} current I ₁ on the interval of positive polarity of the voltage across the transformer windings. This improvement consists in the fact that at a given power transmitted to the primary winding of the transformer, which is equal to $$P_{\mathrm{one}}={\overline{I}}_{\mathrm{one}}\cdot U_{\mathrm{one}}$$

, decreases the value of I _{1, max} .

At time t ₁ , the second stage of the interval of positive polarity of the voltage across the transformer windings begins. At this moment, under the influence of control signals (Fig. 2), the transistor 7 of the bridge circuit is locked, and the transistor 6 is unlocked. This causes a change in the copier, along which the current I ₁ closes. The new (second) circuit is represented as: inductor winding 8 - primary winding 9 of transformer 10 - capacitor 11 - third bridge transistor 6 (in the state of inverse conduction) - first bridge transistor 4 (in the state of direct conduction) - inductor winding 8.

The second circuit does not include an energy source 3. Accordingly, its consumption from the source ceases. However, the circulation of the current I ₁ along the second circuit means that the energy is transmitted both to the primary winding 9 of the transformer 10 and to the capacitor 11. The negative voltage thereon continues to increase in absolute value (Fig. 3).

, причем U_C1<U.The source of energy in the second stage is the inductor 8. The output of energy previously stored in it means that the current I ₁ decreases (Fig. 3). The current decay rate is determined by the equality $$\frac{d{I}_{\mathrm{one}}}{dt}=-\frac{U_{\mathrm{one}}-U_{C\mathrm{one}}}{L_{\mathrm{one}}}$$

, and U _C1 <U.

At time t ₂ , the third stage of the interval of positive polarity of the voltage across the transformer windings begins. At this moment, under the influence of control signals (Fig. 2), the transistor 4 of the bridge circuit is locked and the transistor 5 is unlocked. This causes a change in the circuit along which the current I ₁ closes. The new (third) circuit is represented as: inductor winding 8 - primary winding 9 of transformer 10 - capacitor 11 - transistor 6 of the bridge circuit (in the state of inverse conduction) - energy source 3 - transistor 5 of the bridge circuit (in the state of inverse conduction) - inductor winding 8.

, причем U_C1<0. Она выше, чем на втором этапе. Поэтому на границе второго и третьего этапов наблюдается скачкообразное изменение скорости спада тока I₁ (фиг. 3).In the new (third) circuit, the voltage on the primary winding of the transformer (U ₁ ), the voltage on the capacitor (U _C1 ) and the voltage of the power supply (E), if these voltages are displayed by vectors, are directed towards the current I ₁ . This means that the current I ₁ transfers energy to these elements. The energy source in the third stage, as in the second, is the inductor 8. The output of energy previously stored in it means that the current I ₁ decreases (Fig. 3). The current decay rate is determined by the equality $$\frac{d{I}_{\mathrm{one}}}{dt}=-\frac{U_{\mathrm{one}}+E-U_{C\mathrm{one}}}{L_{\mathrm{one}}}$$

, and U _C1 <0. It is higher than in the second stage. Therefore, at the boundary of the second and third stages, an abrupt change in the current decay rate I ₁ is observed (Fig. 3).

At time t ₃ , which is shifted relative to time t ₀ by half the period, the current I ₁ decreasing in time reaches the level + I _{µ, max} . At this interval, the positive polarity of the voltage across the transformer windings ends (Fig. 3).

In control mode, in which D = 1, the switching of the first pair of transistors (4 and 5) occurs simultaneously with the switching of the second pair of transistors (6 and 7). Because of this, after the first stage of the interval of positive polarity of the voltage on the transformer windings, the third stage immediately begins, and the second stage is completely absent (Fig. 4).

, доходит до уровня +I_µ,max в момент, сдвинутый по отношению к моменту t₀ на половину периода. В этом случае отсутствует третий этап интервала положительной полярности напряжений на обмотках трансформатора.In the control mode, at which D <D _lim , the current I ₁ , decreasing after the moment t ₁ at a speed determined by the equality $$\frac{d{I}_{\mathrm{one}}}{dt}=-\frac{U_{\mathrm{one}}-U_{C\mathrm{one}}}{L_{\mathrm{one}}}$$

reaches the level + I _{µ, max} at the moment shifted with respect to the moment t ₀ by half the period. In this case, there is no third stage of the interval of positive polarity of the voltage on the transformer windings.

, доходит до уровня +I_µ,max раньше момента t₂, когда выключается транзистор 4 мостовой схемы и включается транзистор 5.In the control mode, at which D <D _lim , the current I ₁ , decreasing after the moment t ₁ at a speed determined by the equality $$\frac{d{I}_{\mathrm{one}}}{dt}=-\frac{U_{\mathrm{one}}-U_{C\mathrm{one}}}{L_{\mathrm{one}}}$$

reaches the level + I _{µ, max} before the moment t ₂ when the transistor 4 of the bridge circuit turns off and the transistor 5 turns on.

The current I ₁ , having decreased to the level + I _{µ, max} , continues to be closed along the same second circuit. It contains: inductor winding 8 - primary winding 9 of transformer 10 - capacitor 11 - third bridge transistor 6 (in the state of inverse conduction) - first bridge transistor 4 (in the state of direct conduction) - inductor winding 8.

The current I _{1 is} circulated due to the energy stored in the transformer 10 and inductor 8. The capacitor 11 continues to be charged with current I ₁ , and the voltage of negative polarity increases modulo it. In this case, the transformer 10 and the inductor 8 transmit a part of the stored energy to the capacitor 11, and the current I ₁ decreases.

.The considered processes corresponding to the mode D <D _lim are illustrated by the diagrams shown in FIG. 5. The energy transfer by the transformer to the load circuit is discrete. It is implemented only at intervals when $$I_{\mathrm{one}}\left(t\right)>\frac{F\left(t\right)}{W_{\mathrm{one}}}$$

.

, где T_trans - длительность интервала, во время которого выполняется неравенство $$I_1\left(t\right)>\frac{F\left(t\right)}{W_1}$$

, чему отвечает протекание тока по вторичной обмотке и осуществление процесса передачи (трансформации) энергии. На протяжении этого интервала, как было показано ранее, справедливо соотношение $$U_1\approx \frac{U_{out}}{K_{tr}}$$

. Соответственно $$P_1=U_1\cdot {\overline{I}}_1$$

, где $${\overline{I}}_1=\frac{1}{T/2}\cdot {\displaystyle \underset{t_0}{\overset{t_0+T_{trans}}{\int }}{U}_1\left(t\right)\cdot I_1\left(t\right)dt}$$

.The average value of the power supplied to the primary winding of the transformer and transmitted by it to the load circuit is determined by the equality $${\overline{I}}_{\mathrm{one}}=\frac{\mathrm{one}}{T/2}\cdot {\displaystyle \underset{t_0}{\overset{{\mathrm{<figure-callout\; id="0"\; label="t"\; filenames="00000065.png,00000069.png"\; state="\{\{state\}\}">t</figure-callout>}}_0+T_{trans}}{\int }}{U}_{\mathrm{one}}\left(t\right)\cdot I_{\mathrm{one}}\left(t\right)dt}$$

, where T _trans is the duration of the interval during which the inequality $$I_{\mathrm{one}}\left(t\right)>\frac{F\left(t\right)}{W_{\mathrm{one}}}$$

, which corresponds to the flow of current through the secondary winding and the implementation of the process of transfer (transformation) of energy. Throughout this interval, as was shown earlier, the relation $$U_{\mathrm{one}}\approx \frac{U_{out}}{K_{tr}}$$

. Respectively $$P_{\mathrm{one}}=U_{\mathrm{one}}\cdot {\overline{I}}_{\mathrm{one}}$$

where $${\overline{I}}_{\mathrm{one}}=\frac{\mathrm{one}}{T/2}\cdot {\displaystyle \underset{t_0}{\overset{{\mathrm{<figure-callout\; id="0"\; label="t"\; filenames="00000065.png,00000069.png"\; state="\{\{state\}\}">t</figure-callout>}}_0+T_{trans}}{\int }}{U}_{\mathrm{one}}\left(t\right)\cdot I_{\mathrm{one}}\left(t\right)dt}$$

.

. При условии 0≤D<D_lim, которое отвечает режиму дискретной передачи мощности трансформатором, справедливо неравенство $$D\cdot \frac{T}{2}<T_{trans}<\frac{T}{2}$$

(фиг. 3, 4, 5).Under the condition D _lim ≤D≤1, which corresponds to the continuous transmission of power by the transformer, $$T_{trans}=\frac{\mathrm{<figure-callout\; id="2"\; label="T"\; filenames="00000065.png"\; state="\{\{state\}\}">T</figure-callout>}}{2}$$

. Under the condition 0≤D <D _lim , which corresponds to the mode of discrete power transmission by the transformer, the inequality $$D\cdot \frac{\mathrm{<figure-callout\; id="2"\; label="T"\; filenames="00000065.png"\; state="\{\{state\}\}">T</figure-callout>}}{2}<T_{trans}<\frac{T}{2}$$

(Fig. 3, 4, 5).

выпрямленного тока вторичной обмотки связано со значением $${\overline{I}}_1$$

коэффициентом трансформации, т.е. $${\overline{I}}_{out}=\frac{{\overline{I}}_1}{K_{tr}}$$

.Average value $${\overline{I}}_{out}$$

the rectified current of the secondary winding is associated with the value $${\overline{I}}_{\mathrm{one}}$$

transformation ratio, i.e. $${\overline{I}}_{out}=\frac{{\overline{I}}_{\mathrm{one}}}{K_{tr}}$$

.

The properties of a circuit similar in topology to a bridge resonant DC / DC converter, using phase control of its transistors, which follow from the considered picture of electrical processes, are formulated below.

ограничена на интервале, когда в проводящем состоянии одновременно находятся два транзистора, принадлежащие противолежащим ветвям мостовой схемы (например, при положительной полярности напряжений на обмотках трансформатора такими транзисторами являются 4 и 7). Соответственно на указанном интервале, во время которого входная цепь преобразователя подключена к источнику питания 3, мгновенные значения тока I₁ оказываются ограниченными. Они возрастают при увеличении длительности интервала, которая равна $$D\cdot \frac{T}{2}$$

. Максимум мгновенных значений реализуется при условии D=1 (фиг. 3 и 4).- At each level U _{out of the} output voltage, with which the voltage U ₁ on the primary winding 9 of the transformer 10 is associated with a transformation coefficient, the value $$\frac{d{I}_{\mathrm{one}}}{dt}$$

is limited to the interval when two transistors belonging to the opposite branches of the bridge circuit are simultaneously in the conducting state (for example, when the transistor windings have positive polarity, these transistors are 4 and 7). Accordingly, at the indicated interval, during which the input circuit of the converter is connected to the power source 3, the instantaneous current values I ₁ are limited. They increase with increasing interval duration, which is equal to $$D\cdot \frac{\mathrm{<figure-callout\; id="2"\; label="T"\; filenames="00000065.png"\; state="\{\{state\}\}">T</figure-callout>}}{2}$$

. The maximum instantaneous values are realized under the condition D = 1 (Fig. 3 and 4).

и $${\overline{I}}_{out}$$

возрастают, если увеличивать регулирующий параметр D. В режиме непрерывной передачи мощности трансформатором это возрастание вызвано увеличением мгновенных значений тока I₁ и связанных с ними линейно мгновенных значений тока I₂. При этом неизменна длительность импульсов этих токов, которая равна половине периода работы схемы (фиг. 3 и 4). В режиме дискретной передачи мощности трансформатором рост $${\overline{I}}_1$$

и $${\overline{I}}_{out}$$

при увеличении регулирующего параметра D связан как с возрастанием мгновенных значений токов I₁ и I₂, так и с увеличением длительности импульсов этих токов (фиг. 5).- Values $${\overline{I}}_{\mathrm{one}}$$

and $${\overline{I}}_{out}$$

increase if the control parameter D is increased. In the continuous transfer of power by the transformer, this increase is caused by an increase in the instantaneous current values I ₁ and the linearly instantaneous current values I ₂ associated with them. In this case, the pulse duration of these currents, which is equal to half the period of operation of the circuit, is unchanged (Figs. 3 and 4). Discrete Power Transformer Growth $${\overline{I}}_{\mathrm{one}}$$

and $${\overline{I}}_{out}$$

with an increase in the control parameter, D is associated both with an increase in the instantaneous values of the currents I ₁ and I ₂ , and with an increase in the pulse duration of these currents (Fig. 5).

возрастает. Соответственно возрастают мгновенные и амплитудные значения токов I₁ и I₂, а также значения величин $${\overline{I}}_1$$

и $${\overline{I}}_{out}$$

. При этом все перечисленные величины остаются ограниченными при любом уровне U_out, включая U_out→0 (режим короткого замыкания на выходе).- When lowering the level U _{out of the} output voltage with which the voltage U ₁ on the primary winding 9 of the transformer 10 is associated with a transformation coefficient, the value $$\frac{d{I}_{\mathrm{one}}}{dt}$$

increasing. Accordingly, the instantaneous and amplitude values of the currents I ₁ and I ₂ increase, as well as the values of $${\overline{I}}_{\mathrm{one}}$$

and $${\overline{I}}_{out}$$

. Moreover, all the listed values remain limited at any level of U _out , including U _out → 0 (output short circuit mode).

- With respect to the capacitor of the output filter 17, shunted by the load 18, the rest of the converter circuit acts as an equivalent current source. The current of the equivalent source is represented as a sequence of unipolar current pulses, gradually increasing from zero and gradually decreasing to zero. In operating modes close to the maximum output power mode, current pulses follow continuously (without pauses). At low output power levels, the pulses are separated by pauses.

, каждая из которых отвечает определенному значению напряжения U₁, связанному с выходным напряжением U_out коэффициентом трансформации. Функции $${\overline{I}}_{out}\left(D\right)$$

монотонно нарастают при увеличении регулирующего параметра D. В семействе каждая из функций расположена "тем выше", чем меньше значение U₁, которому эта функция отвечает. В качестве примера на фиг. 6 дано семейство регулировочных характеристик по току, полученное в результате моделирования процессов в схеме. Электрические величины на фиг. 6 представлены в относительных единицах (relative). При этом приняты следующие обозначения: $$U_r=\frac{U_1}{E}$$

; $$I_r=\frac{{\overline{I}}_{out}\left(D\right)}{I_{norm}}$$

, где в качестве "нормирующей константы" I_norm принято среднее значение тока, передаваемого в цепь нагрузки, которое соответствует режиму работы с параметрами U_r=0.1, D=1.- The converter as a regulator of the average value of the current transmitted to the load circuit can be characterized by a family of current-regulating characteristics. It is represented as a set of functional dependencies $${\overline{I}}_{out}\left(D\right)$$

, each of which corresponds to a specific voltage value U ₁ associated with the output voltage U _{out by} the transformation coefficient. Functions $${\overline{I}}_{out}\left(D\right)$$

monotonously increase with increasing regulatory parameter D. In the family, each of the functions is located "the higher", the smaller the value of U ₁ to which this function corresponds. As an example in FIG. Figure 6 shows a family of current control characteristics obtained as a result of modeling processes in the circuit. The electrical quantities in FIG. 6 are presented in relative units. The following notation is accepted: $$U_r=\frac{U_{\mathrm{one}}}{E}$$

; $$I_r=\frac{{\overline{I}}_{out}\left(D\right)}{I_{norm}}$$

, where the average value of the current transmitted to the load circuit, which corresponds to the mode of operation with the parameters U _r = 0.1, D = 1, is taken as the “normalizing constant” I _norm .

, каждая из которых отвечает определенному значению напряжения U₁, связанному с выходным напряжением U_out коэффициентом трансформации. Каждая из функций $${\overline{P}}_{out}\left(D\right)$$

представляется в виде произведения $${\overline{P}}_{out}\left(D\right)=U_{out}\cdot {\overline{I}}_{out}\left(D\right)$$

. Его второй сомножитель монотонно нарастает при увеличении регулирующего параметра D, достигая максимального значения $${\overline{I}}_{out,\max }$$

, отвечающего условию D=1. Тем же свойством обладают функции $${\overline{P}}_{out}\left(D\right)$$

семейства регулировочных характеристик по мощности, причем максимум каждой из функций $${\overline{P}}_{out}\left(D\right)$$

представляется в виде $$P_{out,\max }=U_{out}\cdot {\overline{I}}_{out,\max }$$

. Второй сомножитель произведения $$U_{out}\cdot {\overline{I}}_{out,\max }$$

возрастает при уменьшении первого сомножителя. Поэтому произведение в целом обладает экстремумом, если величины U_out и U₁, которые связаны между собой коэффициентом трансформации, изменяются от нуля до максимума (максимум U₁ ограничен значением, равным Е). Вблизи экстремума значения P_out,max, отвечающие ряду уровней выходного напряжения, незначительно отличаются друг от друга. Это означает, что устройство обладает свойством параметрической стабилизации максимального уровня выходной мощности по отношению к изменению выходного напряжения (или напряжения питания). В качестве примера на фиг. 7 дано семейство характеристик регулирования выходной мощности, полученное в результате моделирования процессов в схеме. Электрические величины на фиг. 7 представлены в относительных единицах (relative). При этом приняты обозначения: $$U_r=\frac{U_1}{E}$$

; $$P_r=\frac{{\overline{P}}_{out}\left(D\right)}{P_{norm}}$$

, где P_norm=1000.- The converter as a regulator of the average value of the power transmitted to the load circuit can be characterized by a family of characteristics for regulating the output power. It is represented as a set of functional dependencies $${\overline{P}}_{out}\left(D\right)$$

, each of which corresponds to a specific voltage value U ₁ associated with the output voltage U _{out by} the transformation coefficient. Each function $${\overline{P}}_{out}\left(D\right)$$

submitted as a work $${\overline{P}}_{out}\left(D\right)=U_{out}\cdot {\overline{I}}_{out}\left(D\right)$$

. Its second factor monotonously increases with increasing control parameter D, reaching a maximum value $${\overline{I}}_{out,\max }$$

corresponding to the condition D = 1. Functions have the same property. $${\overline{P}}_{out}\left(D\right)$$

family of power control characteristics, with the maximum of each function $${\overline{P}}_{out}\left(D\right)$$

represented as $$P_{out,\max }=U_{out}\cdot {\overline{I}}_{out,\max }$$

. The second factor of the work $$U_{out}\cdot {\overline{I}}_{out,\max }$$

increases with decreasing first factor. Therefore, the product as a whole has an extremum if the values of U _out and U ₁ , which are connected by a transformation coefficient, vary from zero to a maximum (maximum U _{1 is} limited by a value equal to E). Near the extremum, the values of P _{out, max} corresponding to a number of output voltage levels differ slightly from each other. This means that the device has the property of parametric stabilization of the maximum level of output power with respect to a change in the output voltage (or supply voltage). As an example in FIG. Figure 7 shows a family of characteristics for regulating the output power obtained as a result of modeling processes in the circuit. The electrical quantities in FIG. 7 are presented in relative units. In this case, the notation: $$U_r=\frac{U_{\mathrm{one}}}{E}$$

; $$P_r=\frac{{\overline{P}}_{out}\left(D\right)}{P_{norm}}$$

where P _norm = 1000.

All that has been stated above regarding the picture of electrical processes and the properties of a DC / DC converter with one power transformer, the circuit of which is presented in FIG. 1 is true for a multi-transformer circuit. An example of a circuit comprising two transformers is given in FIG. 8.

The circuit of FIG. 8 with respect to the circuit of FIG. 1 is supplemented by a second transformer 19. Its primary winding 20 is connected in series to the primary winding circuit 9 of the transformer 10. The secondary winding 21 of the transformer 19 is connected to the output filter capacitor 17 through a rectifier formed by diodes 22, 23, 24 and 25.

The difference in the description of the processes in the circuit of FIG. 8 with respect to the circuit of FIG. 1 consists only in the fact that the voltage U ₁ is equal to the sum of the voltages on the primary windings 9 and 20. Otherwise, like electrical processes in the circuits of FIG. 1 and 8, and the properties of these schemes are identical.

Claims (1)

A method for controlling a bridge voltage converter containing transistors (power controlled keys) forming a bridge circuit and a bipolar load bridge bridge, the first and second bridge transistors connected in series form the first transistor circuit that is connected between the power buses, the third and fourth bridge transistors circuits connected in series form a second transistor circuit that is connected between the power buses, the midpoints of the first and second transistor circuits are respectively Accordingly, the first and second terminals of the output circuit of the bridge circuit, and the first and second terminals of a two-terminal network connected to them are made in the form of inductive and capacitive energy storage devices connected in series, as well as the primary winding of a transformer (or primary windings of several transformers) in which (for which) the secondary winding (secondary windings) through the rectifier (through the rectifiers) is connected (connected) to the capacitor of the output filter, characterized in that two sequences of paraphase pulses are formed waist signals - first and second, and second sequence paraphase pulse signals are shifted relative to the first sequence at adjustable time control of the first and second transistors is carried paraphase pulse signals of the first sequence, and control the third and fourth transistors - paraphase pulse signals of the second sequence.

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