RU2543493C1 - Radar sensor for approach speed of moving object with obstacle - Google Patents

Abstract

FIELD: radio engineering, communication.

SUBSTANCE: radar sensor is in the form of a system of two circuits, one of which is used in a phase tracking circuit and the other in a tracking failure detector circuit. Co-processing of information obtained from discriminators enables to monitor failure of phase tracking and correction input.

EFFECT: higher accuracy of determining the time of phase-locked-loop frequency control failure and enabling correction thereof.

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Description

The invention relates to the field of near radar and can be used in phase locked loop (FAP) systems in a radar sensor Doppler frequency shift.

Known radar sensors for determining the speed of approach of a moving object with an obstacle (see RF Patent No. 2501034, IPC G01S 13/58, 12/14/2011) The problem solved in this invention is to increase the likelihood of correctly measuring the speed of approaching a vehicle to a radar station ( Radar) and reducing the overall dimensions and cost characteristics of locators for the state traffic safety inspection (STSI). This result is achieved through the implementation of locators using a lower frequency signal emitted by the radar, and by measuring the speed of approach of the car to the radar at a shorter and predetermined distance interval. The traffic police locator contains a radar for measuring the initial speed of a moving object, a receiving and transmitting antenna, which is installed, if necessary, on the traffic police car and which emits a continuous signal with frequency modulation according to a one-sided ramp law, and has a traffic speed converter for the traffic police , as well as the outputs of the radar speed calculator, are connected, respectively, to the first and second inputs of the subtraction circuit.

The disadvantage of this sensor is the static data on disruptions that do not allow the adjustment of the system to avoid these disruptions.

Closest to the technical nature of the claimed invention is a radar speed sensor for detecting phase-outs, based on mathematical models of the phase-locked-in statistical dynamics of synchronization systems, selected as a prototype (see L. A. Belov. Microwave signal generation devices and their components: training allowance.M.: Publishing House MPEI, 2010. p.268), operating in the frequency range 24 GHz.

The sensor contains a highly stable generator, a phase-locked loop (PLL), the output of which is connected to a controlled voltage generator (VCO), the output of which is connected to the input of the transmitting channel, which includes a frequency multiplier, the output of which is connected through an attenuator to an amplifier connected through a duplexer to the receiver transmitting antenna, the second output of the control voltage generator is connected to the input of the receiving channel, including the amplifier of the receiving channel, connected through a frequency multiplier of the receiving channel to ervym input quadrature coherent receiver, a second input coupled through the duplexer to the antenna.

The sensor is made on serial chips. An unmodulated signal with a frequency stabilized by a highly stable (quartz) generator is formed in the channel of the radio transmitting device, and the signal reflected from the obstacle with a Doppler frequency offset is fed to the input of a coherent I / Q receiver, where the quadrature components I and Q are extracted for digital processing.

Amplifiers form a reference oscillation with a frequency of 24 GHz.

The phase-locked loop uses a correction filter, which is synthesized in such a way that the maximum dynamic error of tracking the phase of the signal does not exceed the specified limits. A decrease in the dynamic tracking error is possible due to the expansion of the PLL filter frequency band, and this, in turn, leads to an increase in the fluctuation error of the phase tracking.

The disadvantage of this method is the choice of a compromise between fluctuation and dynamic error, i.e. finding some optimal filter bandwidth.

The technical problem solved in the invention is to increase the accuracy of determining the moments of disruption of the PLL and the possibility of their correction. Achievable technical result of the invention is the tracking of moments of disruption of phase tracking and input signal correction in order to avoid these disruptions.

The task is realized in that in the radar proximity sensor of a moving object with an obstacle, containing a highly stable generator, a phase-locked loop (PLL), the output of which is connected to a controlled voltage generator (VCO), one of the outputs of which is connected to the input of the transmitting channel including the multiplier frequency, the output of which through an attenuator is connected to an amplifier connected through a duplexer to a transceiver antenna, the second output of the controlled voltage generator is connected to the input the receiving channel, including the amplifier of the receiving channel, connected through the frequency multiplier of the receiving channel to the first input of the coherent quadrature receiver, the second input of which is connected through a duplexer to the transceiver antenna, the PLL contains a first phase discriminator, a phase offset unit, a first integrator, an adder, a second phase discriminator, threshold device, feedback correction block, third phase discriminator, second integrator, fourth phase discriminator, inverter, filter, third integ an envelope detector, wherein the input of the first phase discriminator is connected to the first output of the highly stable generator, and the output is connected to the input of the first integrator, the output of which is connected to the first input of the fourth phase discriminator and the input of the adder, the output of the adder is connected to the first input of the second phase discriminator, the second output of the highly stable generator is connected to the input of the third phase discriminator connected to the input of the second integrator, the first output of which is connected to the second input of the sums ator, and the second output of the second integrator is connected to the second input of the fourth phase discriminator, the output of the fourth phase discriminator is connected to the inverter input, the first output of which is connected to the envelope detector connected to the second input of the second phase discriminator, the second output of the inverter is connected to the filter, the output of which is connected through the third integrator with the first input of the correction unit, the second input of the correction unit is connected to the first output of the threshold device, the output of the correction unit is connected respectively with the second input of the third phase discriminator and through the phase shift unit with the second input of the first phase discriminator, the output of the second phase discriminator is connected to the input of the threshold device, the second output of which is connected to the VCO.

Essentially, the radar sensor is a system of two loops, one of which is used in the phase tracking loop, and the other in the loop breakdown detector loop. Due to the joint processing of information received from both discriminators, it is possible to track the failures of phase tracking and input correction. This invention due to the joint processing of information received from both discriminators, it is possible to track the moments of failure of phase tracking and introduce correction so as to avoid these failures.

The invention is illustrated by drawings, where

- figure 1 shows a radar sensor of the speed of approach of a moving object with an obstacle,

- figure 2 shows the dependence of the suppression coefficient K _under on the total probability of error R _OS for different values of the phase φ ₀ (t). Curves 1-5 correspond to φ ₀ : 0, π / 20, π / 10, 3π / 20, π / 5,

- figure 3 presents the dependence of the phase of the input and the reference signal of the phase discriminator of the traditional phase-converters on time at T _f = 200 / f _n ,

- figure 4 presents the dependence of the phase of the input and the reference signal of the phase discriminator of the traditional phase-converting converter from time to time at T _f = 140 / f _n

- figure 5 shows the dependence of the phase of the input and reference signals of the phase discriminator of the two-discriminatory phase-difference converter at a filter time constant equal to T _f = 500 / f _n .

The radar sensor of the speed of approaching a moving object with an obstacle contains a generator 1, a phase-locked loop 2 phase-locked loop (PLL), the output of which is connected to a controlled voltage generator (VCO) 3, one of the outputs of which is connected to the input of a transmitting channel including a frequency multiplier 4, the output of which through an attenuator 5 connected to an amplifier 6, connected through a duplexer 7 with a transceiver antenna 8, the second output of the generator 3 of the control voltage is connected to the input of the receiving channel, including the amplifier 9 p the receiver channel connected through the frequency multiplier 10 of the receiving channel to the first input of the coherent quadrature receiver 11, the second input of the latter is connected through the duplexer 7 to the transmit-receive antenna 8. The PLL block 2 contains a first phase discriminator 12, a phase bias unit 13, a first integrator 14, adder 15, second phase discriminator 16, threshold device 17, feedback correction unit 18, third phase discriminator 19, second integrator 20, fourth phase discriminator 21, inverter 22, filter 23, third integrator 24, detector 25 envelope.

The input of the first phase discriminator 12 is connected to the first output of the highly stable generator 1, and the output is connected to the input of the first integrator 14, the output of which is connected to the first input of the fourth phase discriminator 21 and the input of the adder 15, the output of the adder is connected to the first input of the second phase discriminator 16 , the second output of the highly stable generator 1 is connected to the input of the third phase discriminator 19, connected to the input of the second integrator 20, the first output of which is connected to the second input of the adder, and W the second output of the second integrator 20 is connected to the second input of the fourth phase discriminator 21, the output of the fourth phase discriminator 21 is connected to the input of the inverter 22, the first output of which is connected to the envelope detector 25 connected to the second input of the second phase discriminator 16, the second output of the inverter is connected to the filter 23, the output of which is connected through the third integrator 24 to the first input of the correction unit 18, the second input of the correction unit 18 is connected to the first output of the threshold device 17, the output of the correction unit 18 is connected respectively GOVERNMENTAL third to the second input of the phase discriminator 19 and via the phase shift unit 13 with a second input of the first phase discriminator 12, the output of the second phase discriminator 16 is connected to the input of the threshold device 17, a second output connected to the VCO 3.

The invention consists in the following.

Let us consider the situation of phase tracking failure using the example of the Costas scheme (see (1) Shakhtarin B.I. Synchronization in radio communications and radio navigation: a training manual. M: Helios ARV, 2007. (2) GLONASS. Principles of construction and operation, ed. Petrova A.I., Kharisova V.N. M.: Radio Engineering, 2005).

The uniqueness interval of the phase discriminator for the Costas scheme is ± π / 4.

Let the signal X ₁ (t) be present at the FAP input

where A is the signal amplitude, ω ₀ is the cyclic frequency of the carrier oscillation, φ ₀ (t) is the mismatch with the phase of the reference signal.

If the value of φ ₀ (t) exceeds the value ± π / 4, then the system loses stability, and phase tracking is disrupted. Moreover, all the observable parameters of the phase response system behave as if there is a signal X ₂ (t) at the input of the system

Thus, according to the behavior of the FAP system, it is impossible to make an unambiguous conclusion about whether there is currently a breakdown in the phase monitoring of the signal X ₁ (t), or if the FAP system is stable, but there is a signal X ₂ (t) at the input. From which we can conclude that the optimal detector for disruption in the tracking of the PLL circuit is the optimal signal discriminator X ₁ (t) and X ₂ (t).

To synthesize an optimal signal discriminator, the input signal U (t) can be written in the following form:

U (t) = θX ₁ (t) + (1-θ) X ₂ (t) + n (t)

where n (t) is white Gaussian noise, θ is a random parameter taking value 1 with probability p and value 0 with probability 1-p.

The value θ = 1 corresponds to the correct phase tracking mode, and the value θ = 0 corresponds to the phase failure state when the loop of the PLL turns out to be unstable.

The probability p depends on the ratio of the FAP band and the phase fluctuation of the input signal.

The task is to synthesize a system that would distinguish the signals X ₁ (t) and X ₂ (t) with a minimum total probability of error.

2. The synthesis of the structural diagram of the optimal phase disruption detection detector for the phase of the PLL is carried out in two stages. First, on the basis of the criterion of the minimum of the total probability of error, a scheme for the optimal detector for the failure of oscillations is synthesized. Then the task is to optimize the circuit in terms of minimizing computational costs.

The optimal detector for stalling oscillations is actually the optimal discriminator of signals (1) and (2). The structural diagram of the optimal signal discriminator can be determined from the formula for calculating the likelihood ratio ((3) Shakhtarin B. I. Signal Detection: a training manual. M.: Helios ARV, 2006).

в виде [3]:Record the likelihood ratio $$l\left(\raisebox{1ex}{$U$}\!\left/ \!\raisebox{-1ex}{${\varphi }_0$}\right.\right)$$

in the form [3]:

where T is the observation interval, E is the energy of the useful signal, N ₀ is the spectral density of the noise power n (t).

We transform the numerator of the fraction (3) to the form:

Where

Similarly, we get the value for the denominator of expression (3)

Substituting expressions (4) and (7) in relation (3), we obtain:

The last expression can be rewritten as:

where l ₀ is the threshold value with which the decisive statistics are compared.

Prologarithm the right and left side of the expression (8), we obtain:

From expression (9), we can obtain the algorithm of the optimal signal discriminator for coherent reception:

where h is the threshold value after normalization.

(детектор огибающей), с блоком 16, также выход инвертора 22 соединен через фильтр 23 с третьим интегратором 24 и далее подается на блок 18 коррекции обратной связи.This algorithm leads to the structural diagram shown in figure 1, in which the comparison of the decisive statistics with a threshold occurs in block 17 of the threshold device (PU). In block 12, the first phase discriminator is used, in block 13, the signal is shifted by π / 2, and in block 14, integration is performed. From the output of the first integrator 14, the signal is supplied to the fourth phase discriminator 21 and to the adder 15, from the output of the adder the signal is fed to the second phase discriminator 16, fed to the threshold device 17 and then fed to the feedback correction block 18. From the output of block 18, the signal is fed to the third phase discriminator 19 and the phase bias unit 13. The output of the third phase discriminator is connected to the second integrator 20. The output of the second integrator 20 is connected to the adder 15 and the fourth phase discriminator 21, the fourth phase output the second discriminator 21 is connected to the input of the inverter 22, the output of which is connected through a block 25 using the function $$\sqrt{\mathrm{one}-\left(\mathrm{..}\right)}$$

(envelope detector), with block 16, also the output of the inverter 22 is connected through a filter 23 to a third integrator 24 and then fed to the feedback correction block 18.

The phase detection breakdown detector circuit uses an estimate of the phase φ ₀ from the FAP system and calculates cosφ ₀ -sinφ ₀ , which leads to additional computational costs.

In order to reduce computational costs, it is possible to transform the expression cosφ ₀ -sinφ ₀ so that to calculate it, use the signal from the output of the phase discriminator (PD) of the Costas circuit U _PD ~ sin 2φ ₀ . For this, it is necessary to express the difference cosφ ₀ -sinφ _{0 in} terms of sin 2φ ₀

, тоAs

then

.

.

An even greater reduction in computational costs can be achieved if we pay attention to the fact that expressions (5) and (6) completely coincide with the signal formulas in the quadrature and in-phase channel of the discriminator of the Costas circuit [1]. Thus, a phase tracking disruption detector for its work can use the signals from the output of the common-mode and quadrature channels of the Kostas circuit discriminator, and not perform their calculations on their own.

In order to obtain a block diagram of a two-discriminatory phase-response phase, it is necessary to include an optimal tracking failure detector in the Costas scheme and add block 18, a feedback correction block (BOS), into the feedback circuit. In block 15, 1 was used, in block 16, K _f (p), and in block 17, 1 / p.

BKOS, receiving information about the fact of tracking failure, shifts the phase of the reference signal by a predetermined value (for example, ± π / 4 for the Costas scheme).

3. The noise immunity analysis of the optimal phase detection breakdown detector

Under the noise immunity of the system understand its ability to work in conditions of external interference. As a characteristic of noise immunity, often take the boundary (largest) value of the ratio of the power of the interfering signal to the power of the useful signal K_under= P_pom/ P, at where the system can still solve the target problem with the given characteristics [2]. Here P is the power of the useful signal, P_pom - interference power in the frequency band of the useful signal, K_under - suppression coefficient [2].

In our case, the target of the detector is the correct detection of the fact of the failure of phase tracking with the total probability of error not exceeding the specified value.

The total probability of a circuit error consists of two components:

where R _OSH (X ₁ ) is the probability of false detection of a phase tracking failure; R _osh (X ₂ ) is the probability of missing the fact of failure of phase tracking, p is the probability of the arrival of signal X ₁ .

To determine R _OSH (X ₁ ), suppose that a signal is present at the input of the system

Substituting the last equation in the decisive statistics (9) and taking into account formulas (5) and (6), we obtain:

False detection of a phase tracking failure occurs when the value of the decisive statistics is less than a threshold. The probability of such an event is determined by the expression:

- условная плотность распределения вероятности величины q при условии, что на входе присутствует сигнал Х₁(t), Z₀=ln l₀ - величина порога в выражении (9).Where $$W\left(\raisebox{1ex}{$q$}\!\left/ \!\raisebox{-1ex}{$X_{\mathrm{one}}$}\right.\right)$$

is the conditional density of the probability distribution of q, provided that the signal X ₁ (t) is present at the input, Z ₀ = ln l ₀ is the threshold value in expression (9).

Analyzing equation (13), we conclude that q is distributed according to the Gaussian law, since n (t) is a Gaussian interference, and integration is a linear operation. Find its mathematical expectation and variance:

Similarly, we can obtain the mathematical expectation and variance of the decisive statistics q, when the signal U (t) = X ₂ (t) + n (t) is present at the input:

It can be shown that if the signals X ₁ (t) and X ₂ (t) are equally probable (p = 1/2), then the threshold value Z ₀ = 0. Then the probability of error P _OSH (X ₁ ) is determined by the expression

- интеграл вероятности.Where

is the probability integral.

Similarly, we can show that

Substituting expressions (19) and (20) into expression (11) and taking into account the condition of equiprobability of the signals X ₁ (t) and X ₂ (t) (p = 1/2), we obtain the expression for the total error probability:

через мощность полезного сигнала P и мощность помехи в полосе частот полезного сигнала Р_П:Express in the last expression the factor $$\frac{A^2\mathrm{<figure-callout\; id="0"\; label="T\; N"\; filenames="00000039.png,00000040.png"\; state="\{\{state\}\}">T\; </figure-callout>}}{N_{}}$$

through the power of the useful signal P and the interference power in the frequency band of the useful signal P _P :

In view of (22), expression (21) can be rewritten in the form:

Figure 5 shows graphs of the dependence of the suppression coefficient K _under on the total probability of an error P _Ош for different values of the mismatch of the phase of the received signal with the phase of the reference signal φ ₀ (t).

From the analysis of the graphs it follows that in order to obtain a lower total probability of error, it is necessary to increase the signal-to-noise ratio at the input of the tracking interruption detector.

Thus, the probability of error, as expected, depends on the signal-to-noise ratio.

, т.к. сигналы Х₁(t) и Х₂(t) оказываются коррелированны.From the expression (21), we can conclude that the optimal tracking breakdown detector cannot work when $${\mathrm{<figure-callout\; id="0"\; label="\phi "\; filenames="00000039.png,00000040.png"\; state="\{\{state\}\}">\varphi </figure-callout>}}_0=\pm \frac{\pi }{\mathrm{four}}$$

because the signals X ₁ (t) and X ₂ (t) are correlated.

To compare the characteristics of the traditional and dual-discriminatory FAP schemes, both schemes were simulated under the following operating conditions:

1) a signal with a carrier frequency f _n = 100 kHz and an initial phase varying according to a parabolic law was applied to the input of the system

y (t _k ) = cos (ω _н t + φ _k ),

,

,

where ω _n = 2πf _n - angular frequency of the carrier wave, φ _k - current value of the input signal phase, a and b - coefficients determined by the model of the object, and φ ₀ - random variable that varies in the interval [-π; π];

2) the signal-to-noise ratio at the input is -5 dB;

3) the transfer characteristic of the filter K _f (p) in the feedback loop of the FAP was chosen on the basis of ensuring the third order astatism of the tracking system [2]

,

,

where K _f - filter gain, T _f - filter time constant.

6 shows the phase φ of the input _Rin and implementation reference signal φ _op conventional PLL circuit at a signal / noise ratio at the input of -5 dB, the filter time constant is T _f = 200 / f _n.

During the simulation, it was found that due to the presence of noise at the input of the circuit, tracking failures occur with a probability of 52%. To eliminate this phenomenon, expand the filter band.

Figure 4 shows the operation of the traditional FAP scheme with a filter with a wider band (T _f = 140 / f _n ). The probability of failure of phase tracking is less than 1%.

During the simulation, it was found that the standard deviation of the phase estimate is 0.4 radians.

Then, under similar conditions, a two-discriminatory FAP scheme was simulated.

In FIG. 5 according to the invention, the phase of the input and reference signal of the dual-discrimination FAP circuit is presented at a filter time constant equal to T _f = 500 / f _n .

During the simulation, it was found that the mean square deviation of the phase estimate is 0.15 radians, i.e. Compared with the traditional FAP scheme, the accuracy of the phase measurement increased 2.7 times.

This invention due to the joint processing of information obtained from both discriminators, it is possible to track the moments of failure of phase tracking and introduce a correction so as to avoid these failures.

Thus, the invention improves the accuracy of determining the moments of disruption of the PLL, and it is possible to correct them. Achievable technical result of the invention is the tracking of moments of disruption of phase tracking and input signal correction in order to avoid these disruptions.

Claims (1)

A radar proximity sensor of a moving object with an obstacle, containing a highly stable generator, a phase-locked loop, the output of which is connected to a controlled voltage generator, one of the outputs of which is connected to the input of a transmitting channel including a frequency multiplier, the output of which is connected through an attenuator to an amplifier connected through a duplexer with a transmitting and receiving antenna, the second output of the controlled voltage generator is connected to the input of the receiving channel, including the receiving amplifier the ala connected through the frequency multiplier of the receiving channel to the first input of the coherent quadrature receiver, the second input of which is connected through a duplexer to the receiving and transmitting antenna, the phase locked loop contains a first phase discriminator, a phase biasing unit, a first integrator, adder, a second phase discriminator, a threshold device, feedback correction block, third phase discriminator, second integrator, fourth phase discriminator, inverter, filter, third integrator, envelope detector, pr and the input of the first phase discriminator is connected to the first output of the highly stable generator, and the output is connected to the input of the first integrator, the output of which is connected to the first input of the fourth phase discriminator and the input of the adder, the output of the adder is connected to the first input of the second phase discriminator, the second output of the highly stable generator is connected with the input of the third phase discriminator connected to the input of the second integrator, the first output of which is connected to the second input of the adder, and the second output is second about the integrator is connected to the second input of the fourth phase discriminator, the output of the fourth phase discriminator is connected to the input of the inverter, the first output of which is connected to the envelope detector connected to the second input of the second phase discriminator, the second output of the inverter is connected to the filter, the output of which is connected through the third integrator to the first the input of the correction unit, the second input of the correction unit is connected to the first output of the threshold device, the output of the correction unit is connected respectively to the second input of the third phase discriminator and through the phase displacement unit with the second input of the first phase discriminator, the output of the second phase discriminator is connected to the input of the threshold device, the second output of which is connected to the controlled voltage generator.

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