RU2519032C1 - Instrumentation amplifier - Google Patents

Abstract

FIELD: radio engineering, communication.

SUBSTANCE: instrumentation amplifier comprises: an input precision converter of (1) of a first (2) and a second (3) input voltage source connected to a common power supply bus (4), a first (9), a second (10) and a third (11) feedback resistor, an active adder (12) with an inverting (13) and a non-inverting (14) input.

EFFECT: high coefficient of attenuation of input in-phase voltage and eliminating the in-phase component in output signals of operational amplifiers, which improves efficiency of the amplitude characteristic thereof.

4 cl, 26 dwg

Description

The present invention relates to the field of measurement technology, radio engineering, communications and can be used as a device for the precision amplification of analog signals in microcircuits for various functional purposes (for example, for measuring and automatic systems, medical equipment, diagnostics, etc.).

The creation of analog and analog-to-digital interfaces of mixed systems on a chip (SoC), focused on interaction with sensitive elements (sensors) of the bridge type, always involves the use of instrumental amplifiers (DUTs) with both fixed and controlled parameters that perform the functions of suppressing common-mode voltage and gain differential voltage. These devices are the basis for both analog ports and a whole class of complex functional blocks (SF blocks) of SoC. A sufficiently large dynamic range of measured values and a relatively high conversion accuracy predetermined the use of precision operational amplifiers (op amps) in such interfaces. Most of the currently known solutions are associated with the use of the classical structure of constructing a DUT consisting of three op amps and a set of precision resistors [1-16].

The closest prototype of the claimed device is a tool amplifier, presented in patent US 2010/0259323 A1 of figure 1 by Paul L. Bugyik. It contains the input precision converter 1 of the first 2 and second 3 sources of input voltage associated with a common power bus 4, the first 5 and second 6 outputs of the input precision converter 1, the first 7 and second 8 device inputs associated with the first 2 and second 3 input sources voltages, the first 9, second 10 and third 11 feedback resistors, an active adder 12 with inverting 13 and non-inverting 14 inputs, the output of the device 15 associated with the output of the active adder 12, the first 5 output of the input precision conversion ator 1 is connected to the inverting input 13 of the active adder 12, a second inlet 6 outlet precision converter 1 is connected to the non-inverting input 14 of the adder 12 active.

Significant disadvantages of the known device, the architecture of which is also present in many other instrumentation amplifiers [1-16], are as follows:

1. The first flaw. Even when using strictly identical operational amplifiers in the structure of the input precision transducer 1, the limiting value of the attenuation coefficient of the input common-mode signal (K _sn ) will be determined by the following relation:

$$K_{\mathrm{from}n}=\frac{R_{\mathrm{four}}}{{\mathrm{<figure-callout\; id="3"\; label="R"\; filenames="00000022.png,00000023.png"\; state="\{\{state\}\}">R</figure-callout>}}_3+R_{\mathrm{four}}}(\mathrm{one}+\frac{R_{\mathrm{one}}}{R_2})-\frac{R_{\mathrm{one}}}{{\mathrm{<figure-callout\; id="2"\; label="R"\; filenames="00000023.png,00000029.png"\; state="\{\{state\}\}">R</figure-callout>}}_2}=\frac{R_{\mathrm{four}}-R_3\frac{R_{\mathrm{one}}}{R_2}}{{\mathrm{<figure-callout\; id="3"\; label="R"\; filenames="00000022.png,00000023.png"\; state="\{\{state\}\}">R</figure-callout>}}_3+R_{\mathrm{four}}},\text{(one)}$$

where R ₁ ÷ R ₄ - resistors included in the structure of the active adder (12) of figure 1.

Therefore, a deep attenuation of the input common-mode signal in the DUT of Fig. 1 is possible only with strictly agreed resistors R ₁ ÷ R ₄ .

It can be shown that even when the conditions of strict identity of the resistances of the resistors (R ₁ = R ₂ = R ₃ = R ₄ = R) K _{sn are} not better than

$$K_{\mathrm{from}n\min }=\Delta K_{\mathrm{from}n}={\Theta }_R,\text{(2)}$$

where Θ _R is the error of the resistances of the resistors of the active adder (12) (Fig. 1).

Thus, from the above relations (1) and (2), it can be seen that the maximum realized attenuation coefficient of the input common-mode signal of the DUT of Fig. 1 is limited by the error of the resistor resistance of the active adder Θ _R. As a result, even for precision technologies for which Θ _R = 0.1%, the attenuation coefficient of the input common-mode voltage of the DUT of Fig. 1 does not exceed 60 dB, which is clearly not enough to build even non-precision measuring and sensor systems. Therefore, in the production of such ICs, an expensive, precision laser tuning of resistors R ₁ ÷ R _{4 is used} , aimed at achieving the required quality indicators of resistors (for example, Θ _R = 0.01%), at which the attenuation coefficient of the input common-mode signal does not exceed 75 dB.

2. The second drawback. The inefficiency of using the pass-through characteristics of the first (A1) and second (A2) operational amplifiers in the structure of the DUT of FIG. 1. This is because (the main part of the amplitude characteristic of the first (A1) and second (A2) operational amplifiers contains a component of the input common-mode voltage

$$U_{A\mathrm{one}}=U_{\mathrm{from}n}-\frac{R_9}{R_{\mathrm{eleven}}}{U}_d,{\text{U}}_{\text{A2}}=U_{\mathrm{from}n}+\frac{R_{10}}{R_{\mathrm{eleven}}}{U}_d,\text{(3)}$$

where U _A1 and U _A2 are the voltage at the output of the first (A1) and second (A2) operational amplifiers, respectively, U _sn is the common-mode voltage at the inputs of the instrument amplifier 7 and 8, U _d is the differential voltage at the inputs of the instrument amplifier 7 and 8, R ₉ ÷ R ₁₁ - feedback circuit resistors included in the structure of the input precision Converter 1 of figure 1.

The main objectives of the invention are as follows. 1. To exclude the precision resistors R ₁ ÷ R ₄ from the structure of the active adder 12 and, therefore, the need for expensive precision laser tuning of these resistors. This will allow not only to increase the yield of suitable products during production, but also to obtain the highest possible attenuation coefficient of the input common-mode voltage in the DUT structure.

2. To exclude the common-mode component in the output signals of the first (A1) and second (A2) operational amplifiers, which will improve the efficiency of using their amplitude characteristics.

The problem is solved in that in the instrument amplifier of figure 1, containing the input precision Converter 1 of the first 2 and second 3 sources of input voltage associated with a common power bus 4, the first 5 and second 6 outputs of the input precision Converter 1, the first 7 and second 8 the inputs of the device associated with the first 2 and second 3 sources of input voltages, the first 9, second 10 and third 11 feedback resistors, an active adder 12 with inverting 13 and non-inverting 14 inputs, the output of the device 15 associated with the output is active the first adder 12, and the first 5 output of the input precision converter 1 is connected to the inverting input 13 of the active adder 12, the second 6 output of the input precision converter 1 is connected to the non-inverting input 14 of the active adder 12, new elements and communications are provided - the input precision converter 1 includes the first 15 a voltage-current converter, the inverting input of which is connected to the first 7 input of the device, and the non-inverting input is connected to the second 8 input of the device, the first 16 current output of the first 15 The voltage-current generator is connected to the first 17 current output of the second 18 voltage-current converter and is connected to a non-inverting current input of the first 19 current-voltage converter and to the inverting current input of the second 20 current-voltage converter, the second current output 21 of the first 15 voltage-current converter is connected to the second 22 current output of the second 18 voltage-current converter and is connected to the inverting current input of the first 19 current-voltage converter and to the non-inverting current input of the second 20 current-voltage converter, the signal at the first 16 current output of the first 15 voltage-current converters is out of phase with the signal at the second 21 current output of the first 15 voltage-current converters, and the signal is at the first The 17 current output of the second 18 voltage-current converter is out of phase with the signal at the second 22 current output of the second 18 voltage-current converter and is in phase with the signals at the first 16 current output of the first 15 voltage-current converter, output 23 ne of the 19 output current-voltage converter is connected to the first 5 output of the input precision converter 1, and is also connected to the inverting input of the second 18 voltage-current converter through a second 10 feedback resistor, the inverting input of the second 18 voltage-current converter "Is connected to the common bus of power supplies 4 through an additional feedback resistor 24, the output 25 of the second 20 of the output current-voltage converter is connected to the second 6 output of the input precision converter 1, and connected to the non-inverting input of the second 18 voltage-current converter through the first 9 feedback resistor, and the non-inverting input of the second 18 voltage-current converter connected to the common bus of power supplies 4 through the third 11 feedback resistor.

The circuit of the instrumental amplifier of the prototype is shown in the drawing of figure 1.

The drawing of figure 2 presents a diagram of the inventive device in accordance with claim 1, and in the drawing of figure 3 - in accordance with paragraph 2 of the claims. The drawing of figure 4 presents a simplified graphical representation of the input precision Converter 1, corresponding to claim 1 of the claims. The drawing of figure 5 presents a simplified graphical representation of the active adder 12 corresponding to claim 2 of the claims.

In the drawing of Fig.6 presents a diagram of the IUT corresponding to claim 3 of the claims, and in the drawing of Fig.7 is the corresponding claim 4 of the claims.

The drawing of Fig. 8 shows the connection diagram of the inventive instrumental amplifier of Fig. 2 in the PSpice environment on the models of components of the bipolar-field analog base matrix crystal ABMK_1_3, which was used to study its main characteristics.

The drawing of Fig.9 is a diagram of the inventive instrumental amplifier of Fig.6 in the PSpice environment on the models of the components of the bipolar-field analog base matrix crystal ABMK_1_3, which was used to study its main characteristics (Figs. 18-26).

The drawing of Fig. 10 and Fig. 11 shows the logarithmic amplitude and phase frequency characteristics of the differential voltage gain of the instrument amplifier of Fig. 8 for various values of the resistance of the feedback resistors in the structure of the input precision transducer (1) of Fig. 2, which determines the realized value of this coefficient.

On Fig shows the frequency dependence of the transfer coefficient of the input common-mode voltage of the instrument amplifier of Fig.8 (Fig.2) at various values of the differential voltage gain (K _d = 20 dB, 40 dB, 60 dB).

The drawing of Fig.13 shows graphs of the boundary stresses at the outputs (5) and (6) of the input precision transducer (1) when applying a differential signal to the inputs (7) and (8) of the device of Fig. 8 (Fig.2), at various values differential voltage gain (K _d = 20 dB, 40 dB, 60 dB).

The drawing of Fig. 14 shows the values of the zero drift voltage at the output (15) of the instrumentation amplifier of Fig. 8 (Fig. 2) when the temperature changes from -40 to +85 degrees Celsius.

The drawing of Fig. 15 shows a histogram showing the possible values of the zero drift voltage of the instrumentation amplifier of Fig. 8 (Fig. 2) as a result of applying the Monte Carlo method (Gaussian distribution, change in the ratio of resistor values 0.1%), and in the drawing of Fig. 17 - a histogram reflecting the possible values of the transfer coefficient of the input common-mode voltage under similar conditions.

The drawing of Fig.16 shows graphs of the deviation of the transfer coefficient of the input common-mode voltage of the instrumental amplifier of Fig.8 (Fig.2) using the Monte Carlo method (Gaussian distribution, change in the ratio of resistor ratings 0.1%).

On the drawing of Fig. 18 - Fig. 20 shows the logarithmic amplitude and phase frequency characteristics of the differential voltage gain of the instrument amplifier of Fig. 9 for various values of the resistance of the feedback resistors in the structure of the input precision transducer 1 of Fig. 6, which determines the real value of this coefficient. The drawing of Fig.19 shows the characteristic of the differential voltage gain of the instrumentation amplifier of Fig.9 in the passband.

In Fig.21 shows the frequency dependence of the transfer coefficient of the input common-mode voltage of the instrumental amplifier of Fig.9 (Fig.6) for various values of the differential voltage gain (K _d = 20 dB, 40 dB, 60 dB).

The drawing of Fig. 22 shows graphs of the boundary stresses at the outputs (5) and (6) of the input precision transducer (1) when applying a differential signal to the inputs (7) and (8) of the device of Fig. 9 (Fig. 6) for various values of the differential voltage gain (Cd = 20 dB, 40 dB, 60 dB).

The drawing of Fig.23 shows the voltage values of the zero drift at the output (15) of the instrumental amplifier of Fig.9 (Fig.6) when the temperature changes from -40 to +85 degrees Celsius.

On the drawing of Fig.24 is a histogram reflecting the possible values of the zero drift voltage of the instrumentation amplifier of Fig.9 (Fig.6) as a result of applying the Monte Carlo method (Gaussian distribution, change in the ratio of resistors 0.1%), and in the drawing of Fig.26 - a histogram reflecting the possible values of the transfer coefficient of the input common-mode voltage under similar conditions.

The drawing of Fig.25 shows graphs of the deviation of the transfer coefficient of the input common-mode voltage of the instrumental amplifier of Fig.9 (Fig.6) using the Monte Carlo method (Gaussian distribution, change in the ratio of resistor ratings 0.1%).

The graphs of FIG. 16 - FIG. 17 (DUT of FIG. 2) and FIG. 25 - FIG. 26 (DUT of FIG. 3) show that the inventive device, in contrast to the DUT of FIG. 1, is characterized by a high attenuation coefficient of the input common-mode signal, weakly dependent on the error of resistive elements. This will avoid expensive laser precision tuning of resistors and, therefore, increase the yield of products during production. In addition, the voltage of the zero drift of the instrumental amplifier, determined by the zero bias voltage of the active adder 12, as can be seen in FIG. 15 (DUT 2) and FIG. 24 (DUT 6), also has a low dependence on the error of the resistive elements.

The instrument amplifier of Fig. 2 contains an input precision converter 1 of the first 2 and second 3 input voltage sources connected to a common power bus 4, the first 5 and second 6 outputs of the input precision converter 1, the first 7 and second 8 device inputs connected to the first 2 and second 3 sources of input voltages, the first 9, second 10 and third 11 feedback resistors, an active adder 12 with inverting 13 and non-inverting 14 inputs, the output of the device 15 associated with the output of the active adder 12, and the first 5 output is input of the first precision transducer 1 is connected to the inverting input 13 of the active adder 12, the second 6 output of the input precision transducer 1 is connected to the non-inverting input 14 of the active adder 12. The input precision transducer 1 includes the first 15 voltage-current converter, the inverting input of which is connected to the first 7 the input of the device, and the non-inverting input is connected to the second 8 input of the device, the first 16 current output of the first 15 voltage-current converter is connected to the first 17 current output of the second 18 converter voltage-current interface and is connected to a non-inverting current input of the first 19 current-voltage output converter and to an inverting current input of the second 20 current-voltage output converter, the second current output 21 of the first 15 voltage-current converter the second 22 current output of the second 18 voltage-current converter and is connected to the inverting current input of the first 19 output current-voltage converter and to the non-inverting current input of the second 20 output current-n converter voltage ”, and the signal at the first 16 current output of the first 15 voltage-current converters is out of phase with the signal at the second 21 current output of the first 15 voltage-current converters, and the signal at the first 17 current output of the second 18 voltage-current converters the signal at the second 22 current output of the second 18 voltage-current converter and is in phase with the signals at the first 16 current output of the first 15 voltage-current converter, the output 23 of the first 19 output current-voltage converter is connected to the first 5 output one precision transducer 1, and is also connected to the inverting input of the second 18 voltage-current converter through the second 10 feedback resistor, and the inverting input of the second 18 voltage-current converter is connected to the common bus of power supplies 4 through an additional feedback resistor 24 , the output 25 of the second 20 output current-voltage converter is connected to the second 6 output of the input precision converter 1, and is also connected to the non-inverting input of the second 18 voltage-current converter h the first 9 feedback resistor, and the non-inverting input of the second 18 voltage-current converter is connected to the common bus of power supplies 4 through the third 11 feedback resistor.

In the drawing of FIG. 3, in accordance with claim 2, the active adder 12 comprises a third 26 voltage-current converter, the inverting input of which is connected to the output of the device 15, and the non-inverting input is connected to the common bus of the power sources 4, fourth 27 the voltage-current converter, the inverting input of which is connected to the inverting input 13 of the active adder 12, the non-inverting input is connected to the non-inverting input 14 of the active adder 12, the first 28 current output of the third 26 of the voltage-current converter is connected inen with the first 29 current output of the fourth 27 voltage-current converter and is connected to the non-inverting current input of the third 30 of the current-voltage converter, the second 31 current output of the third 26 of the voltage-current converter is connected to the second 32 current output of the fourth 27 the voltage-current converter and is connected to the inverting current input of the third 30 of the current-voltage converter, the signal at the first 28 current output of the third 26 of the voltage-current converter is out of phase with the signal the second 31 current output of the third 26 voltage-current converter, and the signal at the first 29 current output of the fourth 27 voltage-current converter is out of phase with the signal at the second 32 current output of the fourth 27 voltage-current converter and in phase with the signal at the first 28 current the output of the third 26 voltage-current converter.

In the drawing of figure 4, in accordance with claim 1, a simplified graphical representation of the input precision transducer 1, in which the elements 15, 18, 19, 20 are conventionally indicated by the active element 33.

In the drawing of Fig. 5, in accordance with claim 2, a simplified graphical representation of the active adder 12 is shown, in which the elements 26, 27, 30 are conventionally indicated by the active element 34.

In the drawing of FIG. 6, in accordance with claim 3, the first 5 output of the input precision transducer 1 is connected to the inverting input 13 of the active adder 12 through the first 35 low-pass filter having an input 37 and an output 38, and the second 6 output is an input precision the converter 1 is connected to a non-inverting input 14 of the active adder 12 through a second 36 low-pass filter having an input 39 and an output 40. Moreover, each of the low-pass filters 35, 36 contains series-connected resistors 41, 42, 43, an amplifier 44, as well as capacitors 45, 46, 47 .

In the drawing of Fig. 7, in accordance with claim 4, the first 5 and second 6 outputs of the input precision transducer 1 are connected to the corresponding inverting 13 and non-inverting 14 inputs of the active adder 12 through the differential input (antiphase inputs 49, 50) and differential output (antiphase outputs 51, 52) low-pass filter 48.

Consider the operation of the DUT figure 2.

(13) и неинвертирующий $$(Bx{.}_{\Sigma 1}^{(+)})$$

(14) входы активного сумматора (12) соответственно, где осуществляется вычитание поступающих сигналов, что позволяет значительно уменьшить погрешность, вносимую в итоговой результат напряжением дрейфа нуля входного преобразователя (1), и уменьшить коэффициент передачи входного синфазного напряжения (при условии реализации входного прецизионного преобразователя (1) в рамках одного кристалла и единого технологического процесса), а также суммирование усиленного дифференциального напряжения. Причем ослабленная синфазная составляющая входного сигнала и напряжение дрейфа нуля, поступающие с выходов (23) и (25) первого (19) и второго (20) входных преобразователей «ток-напряжение» на инвертирующий и пеивертирующий входы второго (18) преобразователя «напряжение-ток» соответственно, воспринимаются как синфазное напряжение, подавляемое за счет реализации высокого коэффициента ослабления синфазной составляющей входного сигнала во входных дифференциальных каскадах второго (18) преобразователя «напряжение-ток». В итоге на выходе активного сумматора (12) и выходе устройства (15) появляется усиленный дифференциальный сигнал с погрешностью, определяемой напряжением смещения нуля активного сумматора (12).A signal containing in-phase and differential components is fed to the inverting 7 (Vx.1) and non-inverting 8 (Vx.2) inputs of the input precision transducer 1 and fed to the inverting and non-inverting inputs of the first 15 voltage-current converters, respectively. Due to the implementation of a high attenuation coefficient of the common-mode component of the input signal in the input differential stages of the first 15 voltage-current converters, the common-mode component of the signal is suppressed, the amplitude of the differential component of the signal is amplified according to the selected ratios of the first (9) and third (11), as well as the second (10) and optional (24) feedback resistors. The amplified differential component of the input signal together with the common-mode component of the signal weakened with respect to the input, as well as the error introduced by the zero drift voltage from the output (23) of the first (19) output current-voltage converter and from the output (25) of the second (20) output converter "current-voltage", go to the output (5) and output (6) of the input precision transducer (1) and then to the inverting $$(Bx{.}_{\Sigma \mathrm{one}}^{(-)})$$

(13) and non-inverting $$(Bx{.}_{\Sigma \mathrm{one}}^{(+)})$$

(14) the inputs of the active adder (12), respectively, where the input signals are subtracted, which can significantly reduce the error introduced in the final result by the zero-drift voltage of the input transducer (1) and reduce the transfer coefficient of the input common-mode voltage (provided that the input precision transducer is implemented (1) within the framework of a single crystal and a single technological process), as well as the summation of the amplified differential voltage. Moreover, the weakened common-mode component of the input signal and the zero drift voltage coming from the outputs (23) and (25) of the first (19) and second (20) input current-voltage converters to the inverting and peaking inputs of the second (18) voltage-to-voltage converter current ”, respectively, are perceived as common-mode voltage, suppressed due to the implementation of a high attenuation coefficient of the common-mode component of the input signal in the input differential stages of the second (18) voltage-current converter. As a result, at the output of the active adder (12) and the output of the device (15), an amplified differential signal appears with an error determined by the zero bias voltage of the active adder (12).

Let us show analytically that the above properties of the instrumental amplifier are implemented in the inventive scheme of figure 2.

Indeed, using the methods of analysis of electronic circuits, it can be shown that the proposed instrumental amplifier (figure 2) is characterized by the following parameters:

Differential signal gain

$$K_d=\frac{2}{\frac{R_{24}}{{\mathrm{<figure-callout\; id="24"\; label="R"\; filenames="00000023.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{24}+{\mathrm{<figure-callout\; id="10"\; label="R"\; filenames="00000022.png,00000023.png"\; state="\{\{state\}\}">R</figure-callout>}}_{10}}+\frac{R_{\mathrm{eleven}}}{R_{\mathrm{eleven}}+{\mathrm{<figure-callout\; id="9"\; label="R"\; filenames="00000022.png,00000023.png"\; state="\{\{state\}\}">R</figure-callout>}}_9}},\text{(four)}$$

therefore, the choice of the ratio of the values of the first (9) R ₉ and third (11) R ₁₁ feedback resistors, as well as the second (10) R ₁₀ feedback resistor and additional (24) R ₂₄ feedback resistor, sets the value of the parameter K _{d of the} instrument amplifier ( figure 2). In particular, the equalities R ₁₁ = R ₂₄ = r and R ₉ = R ₁₀ = R can be used, then

$$K_d=\mathrm{one}+\frac{R}{r}.\text{(5)}$$

The common-mode voltage transfer coefficient of the instrument amplifier:

$$K_{\mathrm{from}n}=2\cdot K_d\cdot K_{\mathrm{about}\mathrm{from}\mathrm{from}n\mathrm{one}}\cdot K_{\mathrm{about}\mathrm{from}\mathrm{from}\mathrm{<figure-callout\; id="12"\; label="n"\; filenames="00000022.png,00000023.png"\; state="\{\{state\}\}">n</figure-callout>}12},\text{(6)}$$

where K _osn1 is the common-mode voltage attenuation coefficient realized in the input voltage-current converters (15, 18) of the input precision transducer (1), K _osn12 is the common-mode attenuation coefficient of the active adder (12) (Fig. 2).

The voltage of the zero drift of the instrumentation amplifier is determined by the ratio:

$$U_{dR.\mathrm{AND}\mathrm{At}}=\left(U_{dR.5}-U_{dR.6}\right)\cdot K_{\mathrm{about}\mathrm{from}\mathrm{from}\mathrm{<figure-callout\; id="12"\; label="n"\; filenames="00000022.png,00000023.png"\; state="\{\{state\}\}">n</figure-callout>}12}+U_{dR.12},\text{(7)}$$

where U _dr.IU - voltage zero drift of the instrument amplifier, U _dr.5 and U _dr.6 - voltage zero drift at the output (5) and output (6) of the input precision transducer (1), respectively, U _dr.12 - drift voltage zero of the active adder (12). Given the implementation of the input precision Converter (1) (figure 2) in a single technological process within the framework of a single crystal U _{dr. 5} = U _{dr. 6} :

$$U_{dR.\mathrm{AND}\mathrm{At}}=U_{dR.12}=K_{d.12}\cdot E_{\mathrm{from}m.12},\text{(8)}$$

where K _d.12 - gain of the differential signal of the active adder (12), E _{see 12} - EMF bias of the active adder (12) ( _figure 5). Since the active adder (12) is used as an adder of signals (K _d.12 = 1):

$$U_{dR.\mathrm{AND}\mathrm{At}}=E_{\mathrm{from}m.12}.\text{(9)}$$

Thus, the voltage of the zero drift of the instrumental amplifier is determined by the emf bias of the active adder (12).

The voltage at the outputs (5) and (6) of the input precision transducer (1):

$${\mathrm{<figure-callout\; id="5"\; label="U"\; filenames="00000022.png,00000023.png"\; state="\{\{state\}\}">U</figure-callout>}}_5=K_{\mathrm{about}\mathrm{from}\mathrm{from}n\mathrm{one}}\cdot U_{\mathrm{from}n}-\frac{R_{10}}{{\mathrm{<figure-callout\; id="24"\; label="R"\; filenames="00000023.png,00000024.png"\; state="\{\{state\}\}">R</figure-callout>}}_{24}}\cdot \frac{{\mathrm{<figure-callout\; id="2"\; label="U\; d"\; filenames="00000023.png,00000029.png"\; state="\{\{state\}\}">U\; </figure-callout>}}_d}{2},{\text{<figure-callout id="6" label="U" filenames="00000022.png,00000024.png" state="{{state}}">U</figure-callout>}}_{\text{6}}=K_{\mathrm{about}\mathrm{from}\mathrm{from}n\mathrm{one}}\cdot U_{\mathrm{from}n}+\frac{R_9}{R_{\mathrm{eleven}}}\cdot \frac{{\mathrm{<figure-callout\; id="2"\; label="U\; d"\; filenames="00000023.png,00000029.png"\; state="\{\{state\}\}">U\; </figure-callout>}}_d}{2}.\text{(10)}$$

Thus, when K _osn1 << 1, the efficiency of using the amplitude characteristic at the outputs (5) and (6) of the input precision transducer (1) increases (Fig. 2). Indeed, in the prototype device (figure 1):

$$U_{A\mathrm{one}}=U_{\mathrm{from}n}-\frac{R_9}{R_{\mathrm{eleven}}}{U}_d,{\text{U}}_{\text{A2}}=U_{\mathrm{from}n}+\frac{R_{10}}{R_{\mathrm{eleven}}}{U}_d,\text{(eleven)}$$

these voltages are determined by the common-mode voltage at the input of the instrument amplifier U _sn .

In addition, in the prototype (figure 1):

$$U_{dR.\mathrm{AND}\mathrm{At}}=E_{\mathrm{from}m.\mathrm{BUT}3}(\mathrm{one}+\frac{R_{\mathrm{one}}}{R_2})=2E_{\mathrm{from}m.\mathrm{BUT}3}\text{(12)}$$

zero drift is almost 2 times greater than the zero drift of the claimed device.

The decrease in K _sn (formula 6) due to K _OSSN1 << 1 is explained by the differential properties of the input main and additional differential stages of the input voltage-current converters (15, 18) of the input precision converter (1).

Thus, the influence of technological errors in the manufacture of resistors Θ _Ri applies only to the differential gain (4) and practically does not affect the transmission coefficient of the common mode signal (6) and its zero drift (9), which is also seen in Fig. 15 - Fig. 17 (IU of Fig.2) and in Fig.24 - Fig.26 (IU of Fig.6).

The error in the implementation of the differential transmission coefficient is determined by the influence of the static gain (µ):

$$S_{\mu }^{Kd}=\mathrm{one}-\frac{\mu }{\mu +K_d}=\frac{K_d}{\mu +K_d/2}\approx \frac{K_d}{\mu },\text{(12)}$$

$$\frac{\Delta K_d}{K_d}=S_{\mu }^{K_d}.\frac{\Delta \mu }{\mu },\text{(13)}$$

$$K_{d.\max }=\frac{\Delta K_d/K_d\cdot \mu }{\Delta \mu /\mu },\text{(fourteen)}$$

where Δμ / μ is the error of the static gain, determined by the error of the process, ΔK _d / K _d is the error of the differential gain, μ is the static gain, the increase of which, using effective circuitry, can reduce the error of the differential gain.

Consider the operation of the DUT 6.

The operation of the instrumentation amplifier of FIG. 6 differs from the operation of the instrumentation amplifier of FIG. 2 in that low-pass filters (35, 36) for each of the amplification channels are included in its structure between the outputs of the input precision transducer 1 and the inputs of the active adder (12). This implementation of the instrumental amplifier allows minimizing the error in the EMF of the low-pass filter bias introduced into the final result due to the implementation of a high common-mode signal attenuation coefficient in the input stages of the fourth (27) voltage-current converter of the active adder (12) (Fig. 5), providing its suppression.

Depending on the selected values of the capacitors in the RC circuit of the low-pass filters (35, 36), the cutoff frequency of the passband of the frequency characteristics of the instrument amplifier is determined. So each of the low-pass filters (35, 36) of Fig.6, implements the transfer function

$$F_F(p)=\frac{a_0}{{\mathrm{<figure-callout\; id="3"\; label="p"\; filenames="00000022.png,00000023.png"\; state="\{\{state\}\}">p</figure-callout>}}^3+p^2{a}_2+p{a}_{\mathrm{one}}+a_0},\text{(fifteen)}$$

the coefficients of which

$$\begin{array}{l}{a}_2=\frac{{\mathrm{<figure-callout\; id="41"\; label="g"\; filenames="00000027.png"\; state="\{\{state\}\}">g</figure-callout>}}_{41}+g_{42}}{{\mathrm{<figure-callout\; id="45"\; label="C"\; filenames="00000027.png,00000032.png"\; state="\{\{state\}\}">C</figure-callout>}}_{45}}+\frac{{\mathrm{<figure-callout\; id="42"\; label="g"\; filenames="00000027.png"\; state="\{\{state\}\}">g</figure-callout>}}_{42}+g_{43}}{{\mathrm{<figure-callout\; id="46"\; label="C"\; filenames="00000027.png"\; state="\{\{state\}\}">C</figure-callout>}}_{46}},{\text{a}}_{\text{one}}=\frac{g_{41}({\mathrm{<figure-callout\; id="42"\; label="g"\; filenames="00000027.png"\; state="\{\{state\}\}">g</figure-callout>}}_{42}+g_{43})+g_{42}{g}_{43}}{C_{45}{\mathrm{<figure-callout\; id="46"\; label="C"\; filenames="00000027.png"\; state="\{\{state\}\}">C</figure-callout>}}_{46}}+\frac{g_{42}{g}_{43}}{C_{45}{\mathrm{<figure-callout\; id="46"\; label="C"\; filenames="00000027.png"\; state="\{\{state\}\}">C</figure-callout>}}_{46}},\\ a_0=\frac{g_{41}{g}_{42}{g}_{43}}{C_{45}{C}_{46}{\mathrm{<figure-callout\; id="47"\; label="C"\; filenames="00000027.png"\; state="\{\{state\}\}">C</figure-callout>}}_{47}},{\text{<figure-callout id="41" label="g" filenames="00000027.png" state="{{state}}">g</figure-callout>}}_{\text{41}}=\raisebox{1ex}{$\mathrm{one}$}\!\left/ \!\raisebox{-1ex}{${\mathrm{<figure-callout\; id="41"\; label="R"\; filenames="00000027.png"\; state="\{\{state\}\}">R</figure-callout>}}_{41}$}\right.,{\mathrm{<figure-callout\; id="42"\; label="g"\; filenames="00000027.png"\; state="\{\{state\}\}">g</figure-callout>}}_{42}=\raisebox{1ex}{$\mathrm{one}$}\!\left/ \!\raisebox{-1ex}{$R_{42}$}\right.{\mathrm{<figure-callout\; id="43"\; label="g"\; filenames="00000027.png"\; state="\{\{state\}\}">g</figure-callout>}}_{43}=\raisebox{1ex}{$\mathrm{one}$}\!\left/ \!\raisebox{-1ex}{${\mathrm{<figure-callout\; id="43"\; label="R"\; filenames="00000027.png"\; state="\{\{state\}\}">R</figure-callout>}}_{43},$}\right.\text{(16)}\end{array}$$

where C ₄₅ , C ₄₆ , C ₄₇ are the capacitors in the RC circuit of the low-pass filters (35, 36), R ₄₁ , R ₄₂ , R ₄₃ are the resistors in the RC circuit of the low-pass filters (35, 36), are characterized by a low ( ≤1) elemental sensitivity to instability of parameters of resistors and capacitors. To implement a small non-uniformity of the amplitude-frequency characteristic (AFC) in the passband with a rational choice of the approximating function, the quality factor of the pole is insignificant, therefore, additional parametric conditions can be used:

$${\mathrm{<figure-callout\; id="41"\; label="R"\; filenames="00000027.png"\; state="\{\{state\}\}">R</figure-callout>}}_{41}={\mathrm{<figure-callout\; id="42"\; label="R"\; filenames="00000027.png"\; state="\{\{state\}\}">R</figure-callout>}}_{42}={\mathrm{<figure-callout\; id="43"\; label="R"\; filenames="00000027.png"\; state="\{\{state\}\}">R</figure-callout>}}_{43};{\text{<figure-callout id="45" label="C" filenames="00000027.png,00000032.png" state="{{state}}">C</figure-callout>}}_{\text{45}},={\mathrm{<figure-callout\; id="46"\; label="C"\; filenames="00000027.png"\; state="\{\{state\}\}">C</figure-callout>}}_{46}={\mathrm{<figure-callout\; id="47"\; label="C"\; filenames="00000027.png"\; state="\{\{state\}\}">C</figure-callout>}}_{47};{\text{<figure-callout id="45" label="C" filenames="00000027.png,00000032.png" state="{{state}}">C</figure-callout>}}_{\text{45}}=hC.\text{(17)}$$

Then

$$a_2=\frac{2(h+\mathrm{one})}{RCh},{\text{a}}_{\text{one}}=\frac{\mathrm{four}}{R^2{C}^{2}h},{\text{a}}_{\text{0}}=\frac{\mathrm{one}}{R^3{C}^{3}h},\text{(eighteen)}$$

which, up to the ratios of nominal values of the same type of elements, corresponds to the structure of the ladder filter.

Thus, the proposed instrumental amplifier of Fig.2 and Fig.6 compares favorably with the prototype and analogues in that it is characterized by lower power consumption and lower cost due to a decrease in the number of components used and their accuracy requirements, a higher (maximum achievable) attenuation coefficient in-phase signal, weakly dependent on the error of the resistive elements in the circuit, which avoids the costly precision laser tuning of these resistors.

Also, the proposed instrument amplifier is characterized by a more efficient use of the amplitude characteristic at the outputs (5) and (6) of the input precision transducer (1) and a low zero-drift voltage of the instrument amplifier, which has a low dependence on the error of non-identical resistive elements in the circuit due to the implementation of a high coefficient attenuation of the in-phase component of the input signal in the input stages of the input voltage-current converters (15, 18) of the input precision transducer (1) (Fig.2, Fig.6).

These theoretical conclusions confirm the graphs of figure 10 - figure 26

BIBLIOGRAPHIC LIST

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Claims (4)

1. An instrument amplifier comprising an input precision transducer (1) of the first (2) and second (3) input voltage sources connected to a common power bus (4), the first (5) and second (6) outputs of the input precision transducer (1) , the first (7) and second (8) device inputs associated with the first (2) and second (3) input voltage sources, the first (9), second (10) and third (11) feedback resistors, an active adder (12 ) with inverting (13) and non-inverting (14) inputs, the output of the device (15) associated with the output of the active adder (12), and the first (5) output of the input precision transducer (1) is connected to the inverting input (13) of the active adder (12), the second (6) output of the input precision transducer (1) is connected to the non-inverting input (14) of the active adder (12), characterized in that the input precision converter (1) includes the first (15) voltage-current converter, the inverting input of which is connected to the first (7) input of the device, and the non-inverting input is connected to the second (8) input of the device, the first (16) current output the first (15) voltage-t converter ok ”is connected to the first (17) current output of the second (18) voltage-current converter and is connected to the non-inverting current input of the first (19) current-voltage converter and to the inverting current input of the second (20) current-output converter -voltage ”, the second current output (21) of the first (15) voltage-current converter is connected to the second (22) current output of the second (18) voltage-current converter and is connected to the inverting current input of the first (19) output converter "Current-voltage" and to non-invert the current input of the second (20) current-voltage converter, the signal at the first (16) current output of the first (15) voltage-current converter is out of phase with the signal at the second (21) current output of the first (15) voltage converter -current ”, and the signal at the first (17) current output of the second (18) voltage-current converter is out of phase with the signal at the second (22) current output of the second (18) voltage-current converter and is in phase with the signals at the first (16) current output of the first (15) voltage-current converter, output (23) per the output (19) current-voltage converter is connected to the first (5) output of the input precision converter (1), and is also connected to the inverting input of the second (18) voltage-current converter through a second (10) feedback resistor, moreover, the inverting input of the second (18) voltage-current converter is connected to a common bus of power sources (4) through an additional feedback resistor (24), the output (25) of the second (20) output current-voltage converter is connected to the second ( 6) output precision input converter I (1), and is also connected to the non-inverting input of the second (18) voltage-current converter through the first (9) feedback resistor, and the non-inverting input of the second (18) voltage-current converter is connected to the common bus of power sources ( 4) through the third (11) feedback resistor. 2. The instrument amplifier according to claim 1, characterized in that the active adder (12) contains a third (26) voltage-current converter, the inverting input of which is connected to the output of the device (15), and the non-inverting input is connected to a common bus of power sources (4), the fourth (27) voltage-current converter, the inverting input of which is connected to the inverting input (13) of the active adder (12), the non-inverting input is connected to the non-inverting input (14) of the active adder (12), the first (28) current output of the third (26) voltage converter IE-current ”is connected to the first (29) current output of the fourth (27) voltage-current converter and connected to the non-inverting current input of the third (30) current-voltage converter, the second (31) current output of the third (26) the voltage-current converter is connected to the second (32) current output of the fourth (27) voltage-current converter and is connected to the inverting current input of the third (30) current-voltage converter, the signal at the first (28) current output of the third (26) voltage-current converter the signal at the second (31) current output of the third (26) voltage-current converter is out of phase, and the signal at the first (29) current output of the fourth (27) voltage-current converter is out of phase to the signal at the second (32) current output of the fourth ( 27) the voltage-current converter and is in phase with the signal at the first (28) current output of the third (26) voltage-current converter. 3. The instrument amplifier according to claim 1, characterized in that the first (5) output of the input precision transducer (1) is connected to the inverting input (13) of the active adder (12) through the first (35) low-pass filter, and the second (6) the output of the input precision transducer (1) is connected to the non-inverting input (14) of the active adder (12) through the second (36) low-pass filter. 4. The instrument amplifier according to claim 1, characterized in that the first (5) and second (6) outputs of the input precision transducer (1) are connected to the corresponding inverting (13) and non-inverting (14) inputs of the active adder (12) through the differential input and differential output filter frequency (48).

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