RU2303274C1 - Radio direction-finding method and radio direction finder for realization of said method - Google Patents

Abstract

FIELD: radio engineering, possible use in systems for detecting and determining location of radio radiation sources.

SUBSTANCE: method involves reducing methodic components of systematic direction-finding errors, random components of direction-finding errors resulting from distortion of space and time structure of electromagnetic waves of used radio signals because of electro-dynamic interaction between antennas of direction finding meter and influence of mast device by means of determining value of parameter, characterizing presence of quadrature component of interference signal and azimuth estimation error, on basis of which trustworthiness of results of determining of azimuth detection and inclination angle of radio signal source electromagnetic wave. Device achieves technical result due to introduction of additional components and connections between them.

EFFECT: increased precision, increased sensitivity of direction-finding.

2 cl, 28 dwg

Description

The invention relates to radio engineering, in particular to direction finding, and can be used in systems for detecting and determining the location of radio emission sources.

To unambiguously determine in the circular azimuthal sector the direction of propagation of electromagnetic waves (EMW) S, which coincides with the rectilinear propagation of EMW with the direction to the source of radio emission (IRI), phase-sensitive (FS) direction finding methods are widely used [1. Kukes I.S., Old Man M.E. Basics of direction finding. - M .: Owls. radio, 1964. - 640 p.]; [2. Saidov A.S., Tagilaev A.R., Aliev N.M., Aslanov G.K. Design of phase automatic direction finders. - M .: Radio and communications, 1997. - 160 p.]. The essence of the FS methods of direction finding consists in the need to assess the spatial-temporal structure of the electromagnetic field (EMF) created by the IRI in the direction-finding plane (azimuth plane) at least at three spatially separated points, which is most technically simple by using a flat three-element equidistant ring antenna array (CAR), consisting of identical antenna elements (AE) not directed in the azimuthal plane. As AEs with the indicated properties, axisymmetric antennas of the vibrator type are used, the symmetry axes of which are orthogonal to the CAR plane, and the phase centers are uniformly located on a circle of radius r _caw (caw is an abbreviation of the English word "circular arrays with" (ring antenna array)).

и случайной

составляющих в соответствии с выражением:The physical basis of the FS of direction finding methods, as is known [1], is the following EMW properties in the radiation zone: constant amplitude of the components of the electromagnetic field within the CAR aperture (base) and a flat EMF phase front, the normal of which coincides with the direction to the IRI. In the general case, direction finding accuracy is characterized by the operational angular error of the direction finder, which includes instrumental errors characterizing the accuracy of the direction finder-goniometer, which are usually systematic, and errors from other sources, which are manifested in the actual operation of the direction finder, which are, as a rule, random. In a number of cases, a fairly accurate separation of direction finding errors into random and systematic components is not possible. Therefore, almost always without much error, the operational accuracy of direction finding can be characterized by the resulting mean square error (SD) σ _θ in the azimuthal plane, determined through systematic variances

and random

components in accordance with the expression:

The systematic component of the _standard deviation σ _θs as applied to the FS direction finders, in turn, includes the methodological and structural components of errors. The methodological components of errors associated with the direction finding method are due to local heterogeneity of the EMF at the openings of the CAR, leading to distortions in the shapes of the radiation patterns of the antennas of the CAR. The structural components of errors associated with the technical implementation of direction finders are due to the non-identity of the direction finder channels, violation of the symmetry of the CAR structure, and installation errors.

The random component of the _standard deviation σ _θr is due to the influence of internal noise and external noise.

In this case, the main quality indicators of the developed methods of direction finding are the methodological component of the systematic standard deviation provided by the direction finding method and the limiting sensitivity, determined according to [3. GOST 23288-78. Direction finders. Terms and Definitions. - M .: Publishing house of standards, 1979. - 6 pp.], "The minimum electromagnetic field generated by the direction-finding object in the installation location of the radio direction finder antenna, which provides the indication of the direction finding with a given accuracy and probability." Therefore, to analyze the known methods of direction finding and devices that implement them, based on the use of flat three-element equidistant CARs, we assume that: firstly, the structural components of direction finding errors are eliminated by known methods (by calibrating the channels with the appropriate orientation in the CAR space, ensuring symmetry of the structure CAR, etc.) and the systematic _standard deviation σ _θs is determined only by the methodological component; secondly, the random component of the _standard deviation σ _θr , which determines the limiting sensitivity of the direction finder, is due to the influence of internal noise from the direction-finding meter, which are inevitably present during the implementation of direction-finding methods [1], [2]. Naturally, under the influence of external interfering signals, the resulting standard deviation σ _{θ of the} direction finder deteriorates in the general case, and the direction finding results become not reliable, which determines the need to analyze the possibility of known direction finding methods to assess the reliability of direction finding results, that is, the possibility of detecting direction finding in the presence of external jamming signals.

The potential reduction of the random _standard deviation σ _{θr of} direction finding methods based on the use of a flat three-element equidistant CAR of radius r _caw , with uniform measurements (at the same signal levels induced in each of the three CAR antennas under the influence of EMF direction finding IRI) and mutually uncorrelated, distributed over a centered the normal law of the internal noise of the direction-finding meter channels, reduced to the phase centers of the CAR antennas, are determined using the well-known Rao-Cramer inequality [4 . Levin B.R. Theoretical foundations of statistical radio engineering. The second book. 2nd edition, revised. and add. - M .: Owls. Radio, 1975, p. 89]; [5. Tikhonov V.I. Statistical radio engineering. 2nd ed., Revised. and add. - M .: Radio and communication, 1982. - 624 p.]. In this case, the lower boundary of the random _standard deviation σ _{θr of} azimuth estimation θ IRI using a three-element CAR, expressed in radians, can be represented as [2, p. 43]:

where λ is the wavelength of the radio signal, direction finding IRI;

β is the angle of inclination of the wave front (elevation angle) of the IRI;

q is the ratio of the effective voltage of the IRI signal to the effective value of the voltage of internal noise reduced to the phase centers of the CAR antennas.

From formula (2) it follows that the decrease in the random _standard deviation σ _{θr is} achieved both by increasing the CAR radius r _caw and by increasing the signal-to-noise ratio q by increasing (for a given effective value of the internal noise voltage) the reception efficiency of the electromagnetic radiation of each of the CAR antennas .

It is known [1] that direction finding in the circular azimuth sector is unambiguous by the formation and comparison of at least three identical mismatched unambiguous unambiguous phase antenna patterns. In addition, the limitation of the FS of direction finding methods is the ability to unambiguously measure phase differences between signals only within ± 180 °, which leads to the limitation of the spatial separation between the CAR antennas, not exceeding a value close to half the wavelength of the radio signal, which, accordingly, limits the possibility of increasing radius r _{caw of a} three-element equidistant CAR to a value close to the third part of the wavelength of the radio signal. In this regard, the most effective way to reduce σ _θr is to increase the efficiency of signal formation in CAR antennas under the influence of an electromagnetic field signal, in particular for vibrator-type antennas, to increase their effective length, which is achieved by increasing the electrodynamic dimensions of the antennas. In this case, an increase in the efficiency of receiving EMF by CAR antennas inevitably leads to an increase (due to EMF scattering) of the distortion of the structure of the EMF incident on the CAR, as a result of which the equiphase surface of the total EMF formed by the direction-finding IRI signal becomes non-flat, which is equivalent to distortion of both amplitude and and phase diagrams of the antenna patterns of the CARs and leads to the occurrence of methodological direction finding errors due to the mutual influence of the antennas, that is, to an increase in the systematic _standard deviation σ _θs .

In the practically important case of placing the geometric center of the CAR on the mast device, which is a conductive axisymmetric structural element, EMF scattering on the specified conductive element leads to an additional distortion of the equiphase surface of the total EMF formed by the signal from the direction-finding IRI, and, accordingly, to an additional increase in the systematic component of the standard deviation σ _θs . And finally, while receiving at the frequency of a radio signal of a direction-finding IRI an interfering radio signal of another purpose, the structure of the EMF flat front created by a direction-finding IRI can be significantly distorted, which leads to an increase in the random component of the _standard deviation σ _θr and, accordingly, to a deterioration in direction finding accuracy.

,

и

соответственно, которые описываются выражениями:Taking into account the foregoing, in the general case, under the influence of the EMF of the source of the direction-finding radio signal, characterized, firstly, by the amplitude Е _s and phase φ _so at the point О, which is the center of a plane three-element equidistant CAR of radius r _caw formed by the first, second, and third antennas A ₁ , A ₂ and A ₃ with an angular orientation in the direction-finding plane α ₁ , α ₂ and α _3, respectively, and interelement distance b; secondly, the propagation direction S, described by the angle θ between the projection of the S direction on the direction finding plane of the OP and the ON line (reference direction) and the angle β between the direction S and the projection of the S direction on the direction finding plane of the OP, and also under the influence of the electromagnetic field interference signal of another destination and taking into account internal noise brought to the phase centers of the antennas, signals are generated at the outputs of identical omnidirectional antennas A ₁ , A ₂ and A ₃

,

and

respectively, which are described by the expressions:

where i = 1, 2, 3 - number of the antenna CAR;

t is the time;

- мнимая единица;

- imaginary unit;

ω is the circular frequency of the radio signal;

E _h and φ _ho are the amplitude and phase, respectively, in the center of the CAR of the electromagnetic field of the interfering radio signal;

θ _h and β _h - respectively, the azimuth and elevation angle of the propagation direction of the interfering radio signal;

и

- комплексные ДН i-ой антенны в направлении распространения пеленгуемого и помехового радиосигналов соответственно;

and

- complex IDs of the i-th antenna in the direction of propagation of the direction-finding and jamming radio signals, respectively;

- составляющая радиосигнала в фазовом центре i-ой антенны, обусловленная внутренним шумом i-го канала пеленгационного измерителя, являющаяся стационарным гауссовским случайным процессом с нулевым математическим ожиданием.

- the component of the radio signal in the phase center of the i-th antenna, due to the internal noise of the i-th channel of the direction-finding meter, which is a stationary Gaussian random process with zero mathematical expectation.

,

и

антенн могут быть представлены в виде:Given the mutual influence of antennas and the central structural element of a three-element equidistant CAR, complex DNs

,

and

antennas can be represented as:

where W _i = ψcos (θ-α _i ) is the phase delay of the electromagnetic field in the phase center of the i-th antenna A _i relative to the phase of the electromagnetic field in the center of the CAR (i = 1, 2, 3);

h is the coefficient of signal generation efficiency in each identical CAR antenna under the influence of an electromagnetic field signal with a wavelength λ (in particular, the effective length of the antenna of the vibrator type);

- комплексный коэффициент передачи входной цепи антенны;

- the integrated gain of the input antenna circuit;

- комплексный коэффициент ослабления электромагнитной волны, рассеянной одной из идентичных антенн решетки, зависящей от эффективности приема радиосигнала антенной h, параметров согласования антенны и межэлементного расстояния антенн в решетке (так как антенны решетки идентичны, а структура антенной решетки симметричная, то указанные коэффициенты

являются одинаковыми для каждой из трех антенн решетки);

- the complex attenuation coefficient of the electromagnetic wave scattered by one of the identical antennas of the array, depending on the reception efficiency of the radio signal by the antenna h, the antenna matching parameters and the antenna distance between the antennas in the array (since the array antennas are identical and the structure of the antenna array is symmetrical, the indicated coefficients

are the same for each of the three antenna arrays);

- комплексный коэффициент ослабления электромагнитной волны, рассеянной проводящим центральным элементом конструкции антенной решетки (в частности - мачтового устройства), зависящий от рассеивающих свойств центрального элемента и радиуса r_caw решетки.

- the complex attenuation coefficient of the electromagnetic wave scattered by the conductive central structural element of the antenna array (in particular, the mast device), depending on the scattering properties of the central element and the radius r _{caw of the} array.

существенно зависит от ряда конструктивных факторов мачтовых устройств. В отличие от

параметр

может быть представлен через импедансы нагрузки, собственного и взаимного сопротивления антенн решетки. Так, в случае выполнения условия:It should be noted that the analytical representation of the scattering properties of the mast device in the general case is very difficult, since

significantly depends on a number of design factors of mast devices. Unlike

parameter

can be represented through the impedances of the load, self and mutual resistance of the array antennas. So, if the condition is met:

могут быть представлены в виде [6. Виноградов А.Д., Левашов П.А. Новые предельные ограничения на формы диаграмм направленности малобазовых фазо- и поляризационно-чувствительных радиопеленгаторов. - Радиотехника, 2004, №5, с.77-82]:complex nam

can be represented as [6. Vinogradov A.D., Levashov P.A. New limit restrictions on the shape of radiation patterns of low-base phase- and polarization-sensitive direction finders. - Radio engineering, 2004, No. 5, p.77-82]:

- параметр, определяющий степень искажения ДН из-за взаимного влияния антенн КАР;Where

- a parameter that determines the degree of distortion of the DN due to the mutual influence of the CAR antennas;

Z _R - load impedance in the terminal section of the array antennas;

и

- собственное и взаимное сопротивление антенн в составе решетки;

and

- intrinsic and mutual resistance of the antennas in the array;

i = 1, 2, 3.

и

и параметры

,

,

и

связаны соотношениями:In this case, the parameters

and

and parameters

,

,

and

are connected by the relations:

It follows from formulas (4) and (8) that the shapes of both the amplitude and phase ID of the array antennas, due to the electrodynamic interaction between the antennas and the mast device, are uneven in the azimuthal plane and depend, in the general case, on the direction of EMW propagation. A detailed study of the unevenness of complex antenna patterns in three-element CARs is given in [7. Vinogradov A.D., Krachkovsky A.B., Podshivalova G.V. Investigation of direction-finding characteristics of annular antenna arrays taking into account the mutual influence of antenna elements. - Radio engineering, No. 12, 2002, pp. 49-56], shows that for antenna arrays commensurate with the wavelength, the unevenness of the amplitude pattern is of the order of (10 ÷ 15) dB, and the deviation of the phase pattern from the phase of the electromagnetic field at the antenna location - up to (30 ÷ 40) °.

Consider the direction finding of radio signals under the above conditions using known methods of direction finding and devices that implement them.

и

радиан соответственно, одновременное или поочередное измерение трех разностей фаз φ_i между сигналами

и

, принятыми n-ой и k-ой антеннами по правилу:A known method of direction finding, including the reception of a radio signal using three identical omnidirectional antennas, forming an equidistant annular antenna array in the direction-finding plane, whose radius r _caw is chosen so that the distance between the antennas does not exceed half the wavelength λ of the radio signal, the position of the first, second and third antennas oriented relative to the reference direction in the direction finding plane passing through the center of the antenna array, at angles 0,

and

radian, respectively, simultaneous or sequential measurement of three phase differences φ _i between signals

and

adopted by the n-th and k-th antennas according to the rule:

where i = 1, 2, 3;

- символ Кронекера с параметром у, принимающем значения у=i или у=n соответственно, и однозначное определение азимута θ и угла β наклона фронта волны источника радиосигнала по формулам:

- Kronecker symbol with parameter y, taking values y = i or y = n, respectively, and an unambiguous determination of the azimuth θ and the angle β of the slope of the wave front of the radio signal source according to the formulas:

[8. Patent of the Russian Federation No. 2258241, cl. G01S 3/14, 3/74, publ. 2005].

A device that implements the aforementioned direction finding method comprises three identical omnidirectional antennas forming an equidistant annular array of radius r _caw in the direction-finding plane, at which the distance between the antennas does not exceed half the minimum wavelength of the radio signal, three identical radio receiver units made with a common local oscillator, three units measuring the phase difference and the azimuth calculator θ and elevation angle β of the radio signals [8].

The disadvantages of the known method of direction finding and the device that implements it are the low accuracy and sensitivity of direction finding, as well as the inability to assess the reliability of the results of direction finding. These disadvantages are due to the following reasons.

According to mathematical expressions (3) and (4), even in the absence of a disturbing signal for another purpose (E _h = 0), the mutual influence between identical non-directional array antennas and the influence of the mast device (if any) leads to a distortion of the electromagnetic field structure at the locations of the array antennas , which is manifested, firstly, in the non-uniformity of the amplitude antennas of the antennas in the azimuthal plane, reaching (10 ÷ 15) dB, and secondly, in the distortion of the phase IDs of each of the antennas, which leads to errors in the estimation of phase differences φ _i , determined according to formula (11). Let us evaluate the influence of each of these factors on the quality of direction finding of Iran.

According to [1, pp. 63-66], the random mean square error σ _{φ of} measuring the phase difference between two signals of different levels depends on the signal-to-noise ratios characterizing the first and second channels of the direction-finding meter. Therefore, for the indicated case of non-equal measurements, the random standard deviation σ _φ can be represented as:

where q is the signal-to-noise ratio for the direction-finding meter channel with the largest signal amplitude;

and - the coefficient of non-accuracy of measurements, equal to the ratio of the amplitude of the smallest signal to the amplitude of the largest signal (a≤1).

So, for example, with non-uniformity of the amplitude DNs of pairs of antennas equal to (10 ÷ 15) dB, the values of the coefficient of unevenness are a = (0.316 ÷ 0.178), and, according to (14), the random standard deviation σ _φ increases in (2.5 ÷ 4 ) times, respectively. This is equivalent to a corresponding deterioration in the signal-to-noise ratio of the direction-finding meter compared to the case of uniform measurements, which, according to formula (2), leads to an increase (2.5 ÷ 4) times in the random _standard deviation σ _{θr of} determining the IRI azimuth.

Errors of estimation of phase differences φ _i between pairs of signals received by antennas with "distorted" due to the mutual influence of phase patterns, according to [7], reach values of ± (50 ÷ 80) °, which leads to methodological components of systematic direction finding errors, the maximum values which reach (6 ÷ 10) ° in azimuth and tens of degrees in elevation.

To reduce the non-uniformity of the amplitude and distortions of the phase ID antennas in the CAR, the efficiency of reception and, accordingly, the scattering of electromagnetic waves by array antennas is limited (by reducing the electrodynamic dimensions of the antennas) to a certain allowable value at which the systematic error component σ _θs due to the mutual influence of the antennas does not exceed the set value, which, respectively, leads to a deterioration in the sensitivity of direction finding and an increase in the random error component σ _θr .

, которые при их обработке согласно рассматриваемых способа и устройства пеленгования по формулам (11-13) приводят к возникновению аномальных ошибок пеленгования. Из-за отсутствия признака наличия или отсутствия в момент измерений помехового радиосигнала достоверность результатов пеленгования снижается.If there is an interference signal of another purpose at the frequency of the direction-finding IRI signal, according to (3), the structure of the total interference electromagnetic field leads to the formation of signals at the antenna outputs

which, when processed according to the considered method and direction finding device according to formulas (11-13), lead to the occurrence of anomalous direction finding errors. Due to the absence of a sign of the presence or absence at the time of measurement of the interfering radio signal, the reliability of the direction finding results is reduced.

и

радиан соответственно, одновременное измерение трех разностей фаз φ_i между сигналами

и

, принятыми n-ой и k-ой антеннами по правилу:A known method of direction finding, including the reception of a radio signal using three identical omnidirectional antennas, forming an equidistant annular antenna array in the direction-finding plane, whose radius r _caw is chosen so that the distance between the antennas does not exceed half the wavelength λ of the radio signal, the position of the first, second and third antennas oriented relative to the reference direction in the direction finding plane passing through the center of the antenna array, at angles 0,

and

radian, respectively, simultaneous measurement of three phase differences φ _i between signals

and

adopted by the n-th and k-th antennas according to the rule:

where i = 1, 2, 3;

- символ Кронекера с параметром у, принимающем значения у=i или у=n соответственно;

- Kronecker symbol with parameter y, taking values y = i or y = n, respectively;

* - sign of complex conjugation, the simultaneous formation of three amplitude values of the difference signals R _i according to the rule:

- знаковая функция параметра X, принимающего значения

или Х=φ_i соответственно, и однозначное определение азимута θ источника радиосигнала по формуле:Where

- sign function of the parameter X taking values

or X = φ _i, respectively, and an unambiguous determination of the azimuth θ of the radio source by the formula:

[9. Patent of the Russian Federation No. 2262119, cl. G01S 3/14, 3/74, publ. 2005]

A device that implements the aforementioned direction finding method comprises three identical omnidirectional antennas forming an equidistant annular array of radius r _caw in the direction-finding plane, at which the distance between the antennas does not exceed half the minimum wavelength of the radio signal, three identical radio receiving units made with a common local oscillator, three units phase difference measurements, three differential signal generating units and an azimuth calculator θ of radio signals [9].

The disadvantages of the known method of direction finding and the device that implements it are the low accuracy and sensitivity of direction finding, as well as the inability to assess the reliability of the results of direction finding. These disadvantages are due to the following reasons.

в пределах от 0,3 до 0,5 максимальные ошибки "разноса" изменяются в пределах от 0,1° до 0,4° соответственно.Firstly, the compensation method used in the direction-finding method under consideration for eliminating the methodological components of the systematic direction-finding errors due to the mutual influence of the antennas and the mast device is based on the formation of difference radiation patterns of antenna pairs, which, as follows from (4), have a priori defined functional dependence on azimuth θ and elevation angle β of the direction-finding radio signal, not related to parameters due to electrodynamic interaction both between antennas, and mast device. However, in the indicated method and device of direction finding, another component of methodological errors appears, due to the frequency dependence of the shapes of the differential radiation patterns of pairs of antennas, called, as is known [1], the error of "separation". The physical essence of the “separation” errors consists in the mismatch of the slope of the difference pattern with the steepness of the relative phase pattern of the pair of antennas associated with an exact analytical relationship with the position of the front of the electromagnetic wave. The separation errors are proportional to the ratio of the base b (the distance between the antennas) to the wavelength of the radio signal λ. According to [6] for the considered method and direction finding device when changing the ratio

in the range from 0.3 to 0.5, the maximum “separation” errors vary in the range from 0.1 ° to 0.4 °, respectively.

Secondly, as follows from formula (16), the sign of the difference radiation patterns of pairs of antennas is determined by the sign of the phase difference of the signals received by the specified pair of antennas. As previously indicated, an unambiguous measurement of phase differences between signals is possible within ± 180 °. An error in determining the sign of the phase difference between the signals near 0 °, due, for example, to the influence of internal noise, respectively, according to (16), leads to an error in determining the sign of the difference radiation pattern, which in the case under consideration (for "in-phase" signals) is close to zero value, which, taking into account algorithm (17), leads to a random direction finding error, the potentially attainable value of which is determined by formula (2). A completely different situation occurs when the error in determining the phase difference between the signals, the average value of which is close to ± 180 °, occurs when the distances between the antennas are close to half the wavelength of the radio signal. In this case, an error in determining the sign of the phase difference, according to (16), leads to an error in determining the sign of the difference radiation pattern, which, naturally, is not equal to zero and, as a rule, is close to the maximum value, which, according to (17), leads to to the occurrence of abnormal direction finding errors, which can be tens of degrees, which significantly reduces the direction finding reliability. As noted earlier, under conditions of mutual influence between the antennas and the mast device, firstly, the systematic components of the errors in measuring the phase differences can reach ± (50 ÷ 80 °); secondly, significantly (up to 4 times) the random error components of the measurements of phase differences increase due to the non-uniformity of measurements due to a decrease in the level of the useful signal in one of the antennas relative to the other. In this regard, to reduce the likelihood of anomalous errors and thereby increase the reliability of direction finding, the radius r _{caw of the} CAR will be limited to a certain acceptable value at which the distance b between the antennas provides the maximum phase differences of the order of ± 100 °, which is achieved for the three-element equidistant CAR if the condition :

where λ _min is the minimum wavelength of the operating wavelength range of direction-finding radio signals. However, according to (2), this method of reducing anomalous errors leads to a deterioration in the sensitivity of direction finding and an increase in the random component of direction finding errors σ _θr .

, которые при их обработке согласно рассматриваемых способа и устройства радиопеленгования по формуле (17) приводят к возникновению аномальных ошибок пеленгования. Из-за отсутствия признака наличия в момент измерений помехового радиосигнала достоверность результатов пеленгования снижается.Thirdly, if there is an interference signal of another purpose at the frequency of the direction-finding IRI signal, according to (3), the structure of the total interference electromagnetic field leads to the formation of signals at the antenna outputs

which, when they are processed according to the considered method and radio direction finding device according to formula (17), lead to the occurrence of anomalous direction finding errors. Due to the absence of a sign of the presence of an interfering radio signal at the time of measurement, the reliability of the direction finding results is reduced.

In addition, an additional disadvantage of the known method and device of direction finding is the ability to determine the angle of arrival of the radio signal only in the azimuthal plane.

и

радиан соответственно, а геометрические размеры антенн по их осям симметрии соизмеримы с длиной волны λ радиосигнала, одновременное или поочередное измерение разностей фаз φ_i между сигналами

и

, принятыми n-ой и k-ой антеннами, по формулеThe closest in technical essence to the proposed method is a direction finding method, which includes receiving a radio signal using three identical non-directional axisymmetric vibrator-type antennas, forming an equidistant annular antenna array in the direction-finding plane, whose radius r _caw is less than the third part of the wavelength λ of the radio signal, and the position of the first, the second and third antennas are oriented relative to the reference direction in the direction-finding plane passing through the center of the antenna array, at angles 0

and

radians, respectively, and the geometric dimensions of the antennas along their symmetry axes are commensurate with the wavelength λ of the radio signal, the simultaneous or sequential measurement of phase differences φ _i between signals

and

adopted by the n-th and k-th antennas, according to the formula

where i = 1, 2, 3;

- символ Кронекера;

- Kronecker symbol;

- символ Кронекера,

- Kronecker symbol,

the choice of three phase differences φ ₁ , φ ₂ and φ ₃ m-th, the modulus of which is the maximum or one of the maximum values of the modules of the phase differences, the simultaneous or alternate formation of three amplitude values of the difference signals R _i according to the formula:

Where

m is the value of the index of the maximum phase difference;

- знаковая функция параметра F_i, и однозначное определение азимута θ и угла β наклона фронта волны источника радиосигнала по формулам:

- sign function of the parameter F _i , and an unambiguous determination of the azimuth θ and the angle β of the slope of the wave front of the radio signal source according to the formulas:

Where

[10. Patent of the Russian Federation No. 2158001, cl. G01S 3/00, publ. 2000].

It should be noted that through trigonometric transformations, formula (22), taking into account expressions (6), (24) and (25), coincides with formula (17), that is, it is presented in the form:

и

радиан соответственно, а геометрические размеры антенн по их осям симметрии соизмеримы с длиной волны λ радиосигнала, три идентичных радиоприемных блока, выполненных с общим гетеродином, входы которых соединены с выходами соответствующих антенн, три блока измерения разности фаз, три блока формирования разностных сигналов, компаратор, блок формирования однозначных амплитудных значений разностных сигналов, амплитудный вычислитель азимута, вычислитель угла места, датчик параметров вычислений, формирующий априорно известные значения расстояния b между антеннами, длины волны λ радиосигнала и углов α_i ориентации антенн, и генератор управляющих сигналов, причем пара выходов первого радиоприемного блока соединена соответственно с вторыми парами входов вторых блоков измерения разности фаз и формирования разностных сигналов и первыми парами входов третьих блоков измерения разности фаз и формирования разностных сигналов, пара выходов второго радиоприемного блока соединена соответственно с первыми парами входов первых блоков измерения разности фаз и формирования разностных сигналов и вторыми парами входов третьих блоков измерения разности фаз и формирования разностных сигналов, пара выходов третьего радиоприемного блока соединена соответственно с вторыми парами входов первых блоков измерения разности фаз и формирования разностных сигналов и первыми парами входов вторых блоков измерения разности фаз и формирования разностных сигналов, выходы первого, второго и третьего блоков измерения разности фаз, первого, второго и третьего блоков формирования разностных сигналов и первый, второй и третий выходы компаратора соединены соответственно с первым, вторым, третьим, четвертым, пятым, шестым, седьмым, восьмым и девятым входами блока формирования однозначных амплитудных значений разностных сигналов, первый, второй и третий выходы которого соединены соответственно с первым, вторым и третьим входами амплитудного вычислителя азимута, выход генератора управляющих сигналов соединен с управляющими входами первого, второго и третьего радиоприемных блоков и управляющим входом датчика параметров вычислений, первый и второй выходы которого соединены соответственно с первым и вторым входами вычислителя угла места, кроме того, выходы первого, второго и третьего блоков измерения разности фаз соединены с соответствующими входами компаратора и соответствующими входами вычислителя угла места, а первый, второй и третий выходы компаратора соединены, кроме того, с соответствующими входами вычислителя угла места, и, наконец, третий выход датчика параметров вычислений соединен с соответствующим входом амплитудного вычислителя азимута, причем выходы амплитудного вычислителя азимута и вычислителя угла места являются выходами значений соответственно азимута θ и угла β наклона фронта волны источника радиосигнала [10].A direction-finding device is also known, comprising three antennas made of identical non-directional axisymmetric vibrator types, forming in the direction-finding plane an equidistant annular antenna array of radius r _caw smaller than the third part of the wavelength λ of the radio signal, and the position of the phase centers of the first, second and third antennas is oriented relative to the reference direction in direction finding plane passing through the center of the antenna array, at angles 0,

and

radians, respectively, and the geometrical dimensions of the antennas along their symmetry axes are commensurate with the wavelength λ of the radio signal, three identical radio receiving units made with a common local oscillator, the inputs of which are connected to the outputs of the corresponding antennas, three phase difference measuring units, three differential signal generating units, a comparator, unit for generating unambiguous amplitude values of difference signals, an amplitude azimuth calculator, an elevation angle calculator, a calculation parameter sensor that generates a priori known values of the state b between the antennas, the wavelength λ of the radio signal, and the antenna orientation angles α _i , and a control signal generator, the pair of outputs of the first radio receiving unit being connected respectively to the second pairs of inputs of the second phase difference measuring units and generating differential signals and the first pairs of inputs of the third difference measuring units phases and the formation of differential signals, the pair of outputs of the second radio receiving unit is connected respectively to the first pairs of inputs of the first blocks of the measurement of the phase difference and the formation of the difference signals and second pairs of inputs of the third phase difference measurement units and generating differential signals, a pair of outputs of the third radio receiving unit are connected respectively to second pairs of inputs of the first phase difference measurement units and generating differential signals and the first pairs of inputs of the second phase difference measurement units and generating differential signals, the outputs of the first, second and third blocks for measuring the phase difference, the first, second and third blocks of the formation of differential signals and the first, second and third outputs the comparator is connected respectively to the first, second, third, fourth, fifth, sixth, seventh, eighth and ninth inputs of the unit for generating unique amplitude values of the difference signals, the first, second and third outputs of which are connected respectively to the first, second and third inputs of the amplitude azimuth calculator, the output of the control signal generator is connected to the control inputs of the first, second and third radio receiving units and the control input of the calculation parameter sensor, the first and second outputs of which are respectively, with the first and second inputs of the elevator calculator, in addition, the outputs of the first, second and third phase difference measurement units are connected to the corresponding inputs of the comparator and the corresponding inputs of the elevator, and the first, second and third outputs of the comparator are connected, in addition, the corresponding inputs of the elevator calculator, and finally, the third output of the calculation parameter sensor is connected to the corresponding input of the amplitude azimuth calculator, the outputs of the amplitude azimuth calculator and elevator calculators are outputs of values of azimuth θ and angle β of the wave front slope β of the radio signal source [10].

The disadvantages of the closest analogues of the method of direction finding and direction finding for its implementation are low accuracy and sensitivity of direction finding, as well as the inability to assess the reliability of the results of direction finding. These disadvantages are due to the following reasons.

в пределах от 0,3 до 0,575 максимальные ошибки "разноса" изменяются в пределах от 0,1° до 0,44° соответственно.First, the formation of difference radiation patterns in the closest analogs in accordance with formula (20) allows, under the conditions of a priori uncertainty of the distortion parameters of antenna patterns, to eliminate the methodological component of azimuth determination errors θ due to the mutual influence between the antennas and the mast device, however, as in the previously considered analogue [9], when determining the azimuth θ by formula (28), “separation” errors occur proportional to the ratio of the base (distance between antennas) b to the wavelength λ. According to [6] for the considered method and device of direction finding when changing the ratio

in the range from 0.3 to 0.575, the maximum “separation” errors vary from 0.1 ° to 0.44 °, respectively.

Secondly, as follows from formulas (20) and (21), the signs of the difference radiation patterns of all three antenna pairs are determined by the signs of the two smallest modulo phase differences between the selected pairs of signals, which are more than the third (maximum modulo) phase difference resistant to cases of sign change (when approaching the boundary value of an unambiguous measurement of phase differences within ± 180 °) due to systematic errors in determining phase differences due to distortion of phase diagrams of antenna patterns and random errors ok determination of phase differences associated with the distortion of the amplitude radiation patterns of the antennas resulting from the electrodynamic interaction between the antennas and the influence of the mast device. These factors lead to the fact that when the ratio of the base b to the wavelength exceeds a value of the order of (0.35 ÷ 0.4), the anomalous direction finding errors that can amount to tens of degrees become characteristic of the considered direction finding method and device, which significantly reduces the reliability direction finding. In this regard, to reduce the likelihood of anomalous errors and thereby increase the reliability of direction finding, the radius r _{caw of the} antenna array is limited to a certain acceptable value at which the distance b between the antennas provides the maximum values of two of the three measured phase differences of the order of ± 100 °, which for a three-element equidistant CAR is achieved when the following conditions are met:

where λ _min is the minimum wavelength of the working wavelength range. However, according to (2), this method of reducing anomalous errors leads to a deterioration in the sensitivity of direction finding and an increase in the random component of direction finding errors σ _θr .

Thirdly, as follows from formulas (23), (26), (27), (19) and (21), the angle β of the slope of the wave front of the radio signal is estimated from the measured phase differences φ _i between the pairs of signals received by antennas with "distorted" due to the mutual influence of phase radiation patterns. According to [6], the systematic errors in measuring the phase differences φ _i in a three-element equidistant CAR reach ± (50 ÷ 80) °, which leads to an unacceptably large, component of tens of degrees, methodical component of direction finding errors in elevation angle β. In addition, due to the non-evenness of measurements of phase differences between the signals received by the electrodynamic interacting antennas of the three-element CAR, the random component of direction finding errors in elevation angle β increases. This actually leads to obtaining in all cases only a qualitative estimate of the slope of the front of the electromagnetic wave.

, которые при их обработке согласно рассматриваемых способа и устройства радиопеленгования по формулам (22) и (23) приводят к возникновению аномальных ошибок пеленгования. Из-за отсутствия признака наличия в момент измерений помехового радиосигнала достоверность результатов пеленгования снижается.Fourthly, in the presence of a different purpose interference signal at the frequency of the direction-finding IRI signal, according to (3), the structure of the total interference electromagnetic field leads to the formation of signals at the antenna outputs

which, when they are processed according to the considered method and the direction finding device according to formulas (22) and (23), cause anomalous direction finding errors. Due to the absence of a sign of the presence of an interfering radio signal at the time of measurement, the reliability of the direction finding results is reduced.

In general, these shortcomings lead to a decrease in the quality of direction finding and to the limitation of the scope of the radio direction finder.

The problem solved by the invention is improving the quality of direction finding and expanding the arsenal of means in the direction finding of radio emission sources.

The technical result that can be obtained by implementing the method is to increase the accuracy and sensitivity of unambiguous direction finding by reducing the methodological components of systematic direction finding errors, random components of direction finding errors and eliminating anomalous direction finding errors due to distortion of the spatio-temporal structure of electromagnetic waves of direction finding radio signals due to the electrodynamic interaction between the antennas of the direction-finding meter and the influence of the mast triplets.

The technical result that can be obtained by performing the device is to increase the accuracy, sensitivity and reliability of direction finding results.

,

радиан соответственно, а геометрические размеры антенн по их осям симметрии соизмеримы с длиной волны λ радиосигнала, измерение разностей фаз φ_i между сигналами

и

, принятыми n-ой и k-ой антеннами, в соответствии с формулойThe problem is solved in that in the direction finding method, which includes receiving a radio signal using three antennas made by identical non-directional axisymmetric vibrator type, forming an equidistant annular antenna array in the direction-finding plane, and the position of the phase centers of the first, second and third antennas is oriented relative to the reference direction in the plane direction finding, passing through the center of the antenna array, at angles 0,

,

radians, respectively, and the geometric dimensions of the antennas along their symmetry axes are commensurate with the wavelength λ of the radio signal, the measurement of phase differences φ _i between the signals

and

adopted by the n-th and k-th antennas, in accordance with the formula

Where

- символ Кронекера;

- Kronecker symbol;

- символ Кронекера,

- Kronecker symbol,

, и их амплитудных значений r_i по формулеdifferential signal generation

, and their amplitude values r _i according to the formula

and measuring the azimuth θ _{R of the} radio source using three unique amplitude values of the difference signals R _i according to the formula

according to the invention, the amplitude values u _{i of the} signals received by the i-th array antennas are additionally measured in accordance with the expression

and form the coefficients P _i and K _{i the} irregularities of the antenna patterns of the array in the array according to the formulas

- знаковая функция,Where

- sign function,

choose from three values of the indices i serial numbers of the antennas i = 1, i = 2, i = 3 one value of the index ξ, one value of the index γ and one value of the index ν, not equal to each other, from the condition

moreover, the value of the index ξ is assigned the value of the index i at which the product r _i P _i is the minimum or one of the minimum if the above products r _i P _i are equal, and the value of the index ν is assigned the value of the index i at which the product r _i P _i is the maximum or one of the maximum in case of equality of the above products r _i P _i , and the value of index γ is assigned the remaining value of index i,

determine the coefficient p of the uniqueness of direction finding in accordance with the expression

where K _mid - a priori known average value of the coefficients K _{i the} unevenness of the antenna patterns, depending on the electrodynamic dimensions of the antennas, the design of the lattice and mast device,

form three unambiguous amplitude values of the difference signals R _i if the distance b between the antennas does not exceed three tenths of the wavelength λ of the radio signal, in accordance with the expression

Where

and if the distance b between the antennas exceeds three tenths of the wavelength λ of the radio signal, in accordance with the expression

Where

- символ Кронекера,

- Kronecker symbol,

- знаковая функция параметра Y, принимающего значения Y=φ_i или Y=R_γ соответственно,

- sign function of the parameter Y, taking the values Y = φ _i or Y = R _γ, respectively,

по формулеmeasure the phase difference φ _Ri between the difference signals

according to the formula

- знаковая функция параметра Y, принимающего значения Y=R_k или Y=R_n соответственно,Where

- sign function of the parameter Y, taking values Y = R _k or Y = R _n, respectively,

determine the value of the parameter μ, characterizing the presence of the quadrature component of the interfering signal, according to the formula

checking that the conditions for exceeding the minimum amplitude r _{ξ of the} difference signals relative to the a priori known minimum amplitude of the difference signal r _min are checked according to the formula

Where

q _min is the alriorically known minimum necessary signal-to-noise ratio, which provides direction finding of radio emission sources with given accuracy and probability;

U _eff - the effective value of the voltage of the internal noise of the channels of the formation of differential signals,

determine the azimuth θ _{φ of the} radio signal source using phase differences φR _i between the difference signals according to the formulas

where b is the distance between the antennas, not exceeding two third of the wavelength λ of the radio signal,

determine the error Δθ azimuth estimates θ by the formula

Where

determine the azimuth θ and evaluate the angle β of the slope of the wave front of the source of the radio signal according to the formulas

Where

β _sp is a sign of the presence at the receiving point of a radio signal propagating in the form of a spatial electromagnetic wave, the front slope of which cannot be determined and is within

Δθ _max - a priori known value of the maximum permissible error in determining the azimuth θ,

and the values of the parameter μ and the error Δθ judge the reliability of the results of determining the azimuth θ and the angle β of the slope of the wave front of the radio signal source, and this reliability is inversely proportional to the values of the parameter μ and the error Δθ of determining the azimuth of the radio signal.

и

радиан соответственно, а геометрические размеры антенн по их осям симметрии соизмеримы с длиной волны λ радиосигнала, три идентичных радиоприемных блока, выполненных с общим гетеродином, входы которых соединены с выходами соответствующих антенн, три блока измерения разности фаз, три блока формирования разностных сигналов, компаратор, блок формирования однозначных амплитудных значений разностных сигналов, амплитудный вычислитель азимута, вычислитель угла места, датчик параметров вычислений, формирующий априорно известные значения расстояния b между антеннами и длины волны λ радиосигнала, и генератор управляющих сигналов, причем пара выходов первого радиоприемного блока соединена соответственно с вторыми парами входов вторых блоков измерения разности фаз и формирования разностных сигналов и первыми парами входов третьих блоков измерения разности фаз и формирования разностных сигналов, пара выходов второго радиоприемного блока соединена соответственно с первыми парами входов первых блоков измерения разности фаз и формирования разностных сигналов и вторыми парами входов третьих блоков измерения разности фаз и формирования разностных сигналов, пара выходов третьего радиоприемного блока соединена соответственно с вторыми парами входов первых блоков измерения разности фаз и формирования разностных сигналов и первыми парами входов вторых блоков измерения разности фаз и формирования разностных сигналов, выходы первого, второго и третьего блоков измерения разности фаз, первые выходы первого, второго и третьего блоков формирования разностных сигналов и первый, второй, и третий выходы компаратора соединены соответственно с первым, вторым, третьим, четвертым, пятым, шестым, седьмым, восьмым и девятым входами блока формирования однозначных амплитудных значений разностных сигналов, первый, второй и третий выходы которого соединены соответственно с первым, вторым и третьим входами амплитудного вычислителя азимута, выход генератора управляющих сигналов соединен с управляющими входами первого, второго и третьего радиоприемных блоков и управляющим входом датчика параметров вычислений, первый и второй выходы которого соединены соответственно с первым и вторым входами вычислителя угла места, выход которого является выходом значения угла β наклона фронта волны источника радиосигнала, согласно изобретению расстояние между антеннами выбрано не превышающем двух третьих длины волны радиосигнала, компаратор выполнен с возможностью определения упорядоченной совокупности трех номеров антенн, через фазовые центры которых последовательно во времени проходит фронт электромагнитной волны источника радиосигнала, датчик параметров вычислений выполнен с возможностью формирования априорно известных среднего значения K_mid коэффициентов неравномерности диаграмм направленности антенн в составе решетки, минимальной амплитуды разностного сигнала r_min, при которой обеспечивается минимально необходимое отношение сигнал/шум q_min относительно действующего значения напряжения U_eff внутреннего шума каналов формирования разностных сигналов радиопеленгатора и максимально допустимой ошибки Δθ_max определения азимута, блок формирования однозначных амплитудных значений разностных сигналов выполнен с возможностью формирования амплитудных значений разностных сигналов с учетом знаков разностей фаз между сигналами, принятыми антеннами, в случае, если расстояние b между антеннами не превышает трех десятых частей длины волны λ радиосигнала, или с учетом результатов сравнения амплитуд сигналов, принятых антеннами трехэлементной решетки, в случае, если расстояние b между антеннами превышает три десятые части длины волны λ радиосигнала, вычислитель угла места выполнен с возможностью адаптивной оценки угла места с использованием разностей фаз между разностными сигналами, принятыми тремя различными парами антенн, в зависимости от отношения минимальной амплитуды разностных сигналов к действующему значению напряжения внутреннего шума каналов формирования разностных сигналов радиопеленгатора и погрешности оценки азимута радиосигнала, и дополнительно введены блок формирования коэффициентов неравномерности амплитудных диаграмм направленности антенн, первая, вторая и третья пары входов которого соединены с парами выходов первого, второго и третьего радиоприемных блоков соответственно, вычислитель коэффициента однозначности пеленгования, вычислитель шумового порогового коэффициента, блок определения разности фаз между разностными сигналами, вычислитель квадратурной составляющей помехового сигнала, фазовый вычислитель азимута, вычислитель погрешности оценки азимута, вычислитель азимутального порогового коэффициента и блок определения азимута, причем вторые и третьи выходы первого, второго и третьего блоков формирования разностных сигналов соединены соответственно с первым, вторым, третьим, четвертым, пятым и шестым входами блока определения разности фаз между разностными сигналами, а первые выходы первого, второго и третьего блоков формирования разностных сигналов соединены соответственно с объединенными первыми, объединенными вторыми и объединенными третьими входами компаратора и вычислителя шумового порогового коэффициента, первый выход компаратора соединен с объединенными четвертым входом вычислителя шумового порогового коэффициента и первыми входами вычислителя коэффициента однозначности пеленгования и фазового вычислителя азимута, первый, второй, третий, четвертый, пятый и шестой выходы блока формирования коэффициентов неравномерности амплитудных диаграмм направленности антенн соединены соответственно с четвертым, пятым и шестым входами компаратора и вторым, третьим и четвертым входами вычислителя коэффициента однозначности пеленгования, пятый вход и выход которого соединены соответственно с третьим выходом датчика параметров вычислений и объединенными десятым входом блока формирования однозначных амплитудных значений разностных сигналов и вторым входом фазового вычислителя азимута, первый, второй и третий выходы блока формирования однозначных амплитудных значений разностных сигналов соединены соответственно с объединенными первым входом вычислителя квадратурной составляющей помехового сигнала и седьмым входом блока определения разности фаз между разностными сигналами, с объединенными вторым входом вычислителя квадратурной составляющей помехового сигнала и восьмым входом блока определения разности фаз между разностными сигналами и с объединенными третьим входом вычислителя квадратурной составляющей помехового сигнала и девятым входом блока определения разности фаз между разностными сигналами, первый, второй и третий выходы которого соединены соответственно с объединенными третьими, объединенными четвертыми и объединенными пятыми входами вычислителя угла места и фазового вычислителя азимута, объединенные шестые входы фазового вычислителя азимута и вычислителя угла места и первый вход блока определения азимута соединены с выходом вычислителя шумового порогового коэффициента, пятый вход которого соединен с четвертым выходом датчика параметров вычислений, выход фазового вычислителя азимута соединен с объединенными первым входом вычислителя погрешности оценки азимута и вторым входом блока определения азимута, третий вход которого, объединенный с вторым входом вычислителя погрешности оценки азимута, соединен с выходом амплитудного вычислителя азимута, выход вычислителя погрешности оценки азимута и пятый выход датчика параметров вычислений соединены соответственно с первым и вторым входами вычислителя азимутального порогового коэффициента, выход которого соединен с объединенными седьмым входом вычислителя угла места и четвертым входом блока определения азимута, первый и второй выходы датчика параметров вычислений соединены соответственно с объединенными одиннадцатым входом блока формирования однозначных амплитудных значений разностных сигналов и седьмым входом фазового вычислителя азимута и с объединенными двенадцатым входом блока формирования однозначных амплитудных значений разностных сигналов и восьмым входом фазового вычислителя азимута, причем выход блока определения азимута является выходом значения азимута θ источника радиосигнала, а выходы вычислителя квадратурной составляющей помехового сигнала и вычислителя погрешности оценки азимута являются выходами параметров достоверности результатов пеленгования μ и Δθ соответственно.The problem is also solved by the fact that in the direction finder containing three antennas made by identical non-directional axisymmetric vibrator type, forming in the direction-finding plane an equidistant annular antenna array, and the position of the phase centers of the first, second and third antennas is oriented relative to the reference direction in the direction-finding plane passing through the center of the antenna array, at angles 0,

and

radians, respectively, and the geometrical dimensions of the antennas along their symmetry axes are commensurate with the wavelength λ of the radio signal, three identical radio receiving units made with a common local oscillator, the inputs of which are connected to the outputs of the corresponding antennas, three phase difference measuring units, three differential signal generating units, a comparator, unit for generating unambiguous amplitude values of difference signals, an amplitude azimuth calculator, an elevation angle calculator, a calculation parameter sensor that generates a priori known values of the state b between the antennas and the wavelength λ of the radio signal, and a control signal generator, wherein the pair of outputs of the first radio receiving unit is connected respectively to the second pairs of inputs of the second phase difference measurement and generating signals and the first pairs of inputs of the third phase difference measuring and generating of differential signals, a pair of outputs of the second radio receiving unit is connected respectively to the first pairs of inputs of the first blocks of the measurement of the phase difference and the formation of differential signals and the second pair and the inputs of the third blocks for measuring the phase difference and generating differential signals, the pair of outputs of the third radio receiving unit is connected respectively to the second pairs of inputs of the first blocks for measuring the phase difference and generating differential signals and the first pairs of inputs of the second blocks for measuring the phase difference and generating differential signals, the outputs of the first, second and third blocks for measuring the phase difference, the first outputs of the first, second and third blocks for generating differential signals and the first, second, and third outputs of the comparator connected respectively to the first, second, third, fourth, fifth, sixth, seventh, eighth and ninth inputs of a unit for generating unique amplitude values of difference signals, the first, second and third outputs of which are connected respectively to the first, second and third inputs of an amplitude azimuth calculator, output the control signal generator is connected to the control inputs of the first, second and third radio receiving units and the control input of the calculation parameter sensor, the first and second outputs of which are connected respectively According to the invention, the distance between the antennas is not exceeding two-thirds of the wavelength of the radio signal, the comparator is configured to determine an ordered set of three antenna numbers through phase the centers of which the front of the electromagnetic wave of the radio signal source passes sequentially in time, the calculation parameter sensor is configured to Hovhan priori known average value K _mid coefficients unevenness antenna pattern consisting of a lattice, the minimum difference signal amplitude r _min, which provides a minimum required signal / noise ratio q _min relative to the effective value U _eff voltage internal noise channels forming difference signals finder and the maximum errors Δθ _{max for} determining the azimuth, the unit for generating unique amplitude values of the difference signals is configured to the amplitude values of the difference signals taking into account the signs of the phase differences between the signals received by the antennas, if the distance b between the antennas does not exceed three tenths of the wavelength λ of the radio signal, or taking into account the results of comparing the amplitudes of the signals received by the antennas of the three-element grating, if the distance b between the antennas exceeds three tenths of the wavelength λ of the radio signal, the elevator calculator is adapted to adaptively estimate the elevation angle using phase differences between the spacing signals received by three different pairs of antennas, depending on the ratio of the minimum amplitude of the differential signals to the effective value of the internal noise voltage of the channels for generating the differential signals of the direction finder and the error in estimating the azimuth of the radio signal, and an additional unit for generating the unevenness coefficients of the amplitude antenna radiation patterns, the first, second and the third pair of inputs of which are connected to pairs of outputs of the first, second and third radio receiving units, respectively, direction finding coefficient calculator, noise threshold coefficient calculator, phase difference determination unit between difference signals, interfering signal quadrature component calculator, phase azimuth calculator, azimuth estimation error calculator, azimuth threshold coefficient calculator and azimuth determination unit, the second and third outputs of the first, second and the third difference signal generating units are connected respectively to the first, second, third, fourth, fifth and the first inputs of the comparator and calculator of the noise threshold coefficient, the first output of the comparator is connected to the combined fourth input of the calculator, and the first outputs of the first, second, and third blocks of generating the difference signals are connected respectively to the combined first, combined second and combined third inputs of the comparator and calculator noise threshold coefficient and the first inputs of the calculator of the uniqueness coefficient of direction finding and the phase calculator azim the first, second, third, fourth, fifth and sixth outputs of the unit for generating the unevenness coefficients of the amplitude radiation patterns of the antennas are connected respectively to the fourth, fifth and sixth inputs of the comparator and the second, third and fourth inputs of the calculator of the direction finding coefficient, the fifth input and output of which are connected respectively, with the third output of the calculation parameter sensor and the combined tenth input of the unit for generating unique amplitude values of the difference signals and the second input m of the phase azimuth calculator, the first, second and third outputs of the unit for generating unique amplitude values of the difference signals are connected respectively to the combined first input of the calculator of the quadrature component of the interference signal and the seventh input of the unit for determining the phase difference between the difference signals, with the combined second input of the calculator of the quadrature component of the interference signal and the eighth input of the phase difference determination unit between the difference signals and with the combined third input of the calculator the cadre component of the interfering signal and the ninth input of the phase difference determination unit between the difference signals, the first, second and third outputs of which are connected respectively to the combined third, combined fourth and combined fifth inputs of the elevation calculator and the phase azimuth calculator, the combined sixth inputs of the azimuth phase calculator and calculator elevation angle and the first input of the azimuth determination unit are connected to the output of the noise threshold factor computer, the fifth input of which is connected nen with the fourth output of the calculation parameters sensor, the output of the phase azimuth calculator is connected to the combined first input of the azimuth estimation error calculator and the second input of the azimuth determination unit, the third input of which, combined with the second input of the azimuth estimation error calculator, is connected to the output of the amplitude azimuth calculation calculator, azimuth estimation errors and the fifth output of the calculation parameter sensor are connected respectively to the first and second inputs of the azimuth threshold threshold calculator coefficient, the output of which is connected to the combined seventh input of the elevator calculator and the fourth input of the azimuth determination unit, the first and second outputs of the calculation parameter sensor are connected respectively to the combined eleventh input of the unit for generating unique amplitude values of difference signals and the seventh input of the phase azimuth calculator and to the combined twelfth input unit for generating unique amplitude values of the difference signals and the eighth input of the phase azimuth calculator, and you the course of the azimuth determination unit is the output of the azimuth value θ of the radio signal source, and the outputs of the quadrature component of the interfering signal and the azimuth estimation error calculator are the outputs of the reliability parameters of the direction finding results μ and Δθ, respectively.

The solution of the problem with the achievement of the technical result is due to the following.

In the proposed method of direction finding, in contrast to the known:

firstly, the bearing characteristic in azimuth is determined using both single-valued amplitude difference beam patterns of antenna pairs and single-phase difference difference radiation patterns of antenna pairs, each of these radiation patterns providing full compensation for the methodological components of direction finding errors due to the mutual influence of antennas and mast devices, and the joint use of these amplitude and phase radiation patterns allows t, while maintaining the unambiguity of direction finding in the circular azimuthal sector provided by the use of amplitude difference radiation patterns, halve the "separation" errors inherent in direction-finding characteristics using only amplitude difference radiation patterns;

secondly, to ensure the unambiguity of the formation of both amplitude and phase difference beam patterns of antenna pairs, the revealed regularity of the relationships between the amplitude and phase distortions of the antenna patterns of a three-element equidistant annular array antenna arising from the electrodynamic interaction between antennas and the influence of the mast device, which allows not only increase to two third of the radio wave the distance between the antennas of the three-element to face antenna array with the preservation of the uniqueness of direction finding and a corresponding reduction in random direction finding errors, but also significantly reduce the anomalous direction finding errors associated with distortion of the phase diagrams of the antennas due to the mutual influence of the antennas and the mast device and random errors in measuring the phase difference between signals received by antennas with significantly different amplitude radiation patterns, also due to the mutual influence between the antennas and the mast device;

, что в общем случае при пеленговании радиосигналов в круговом азимутальном секторе обеспечивает не только повышение точности, но и повышение достоверности оценок угла β наклона фронта волны источника радиосигнала;thirdly, the direction-finding characteristic by elevation angle β for the vast majority of directions of propagation of electromagnetic waves in the circular azimuthal sector is formed using unambiguous phase difference radiation patterns of pairs of antennas that are not susceptible to distortion due to the mutual influence of antennas and mast devices, which eliminates the corresponding methodological component of systematic errors in determining the angle β of the slope of the wave front of the radio signal source, and for azimuthal directions p the propagation of electromagnetic waves close to the azimuthal directions passing through the center of the antenna array and any of the three antennas, by comparing the amplitude θ _R and phase θ _{φ of the} azimuthal direction-finding characteristics, an unambiguous assessment of the belonging of the radio signal to the signal propagating in the form of surface electromagnetic waves (β = 0 ), or a signal propagating in the form of spatial electromagnetic waves

that in the general case when direction finding of radio signals in the circular azimuthal sector provides not only an increase in accuracy, but also an increase in the reliability of estimates of the angle β of the slope of the wave front of the radio signal source;

с азимутальными направлениями, близкими к азимутальным направлениям, проходящим через центр антенной решетки и любой из трех антенн, а при одновременном приеме на одной длине волны пеленгуемого и помехового радиосигналов - зависят от соотношения комплексных амплитуд пеленгуемого и помехового сигналов, приводящего к формированию суммарной пространственно-временной структуры интерференционного поля в фазовых центрах трехэлементной антенной решетки, которая различным образом искажает амплитудную θ_R и фазовую θ_φ азимутальные пеленгационные характеристики; совместное использование предложенных двух признаков наличия в момент измерений помехового радиосигнала позволяет проводить оценку достоверности результатов пеленгования в круговом азимутальном секторе при произвольных фазовых соотношениях пеленгуемого и помехового радиосигналов.fourthly, two rules are proposed for assessing the reliability of direction-finding results, the first of which is based on a comparison of single-valued amplitude difference radiation patterns of pairs of antennas and allows, under conditions of mutual influence of antennas and a mast device, to determine the level of the quadrature component of the jamming radio signal in the circular azimuth sector of the direction-finding radio signal in case of mismatch projections on the direction-finding plane of the directions of propagation of electromagnetic waves of the direction-finding and interference noise of the diosignals, and the second is based on a comparison of the amplitude azimuth direction finding characteristic θ _R obtained using unambiguous amplitude radiation patterns of pairs of antennas and the phase azimuth direction-finding characteristic θ _φ obtained using unambiguous phase difference radiation patterns of pairs of antennas, the differences between which are independent of the mutual influence of antennas and mast devices when receiving one direction-finding radio signal do not exceed the "separation" errors, the maximum value of of the other (for the largest distance between the antennas, equal to two-thirds of the wavelength of the radio signal) does not exceed 1 °, with the exception of cases of receiving one direction-finding radio signal propagating in the form of a spatial electromagnetic wave

with azimuthal directions close to the azimuthal directions passing through the center of the antenna array and any of the three antennas, and while simultaneously receiving direction-finding and jamming radio signals at the same wavelength, they depend on the ratio of the complex amplitudes of the direction-finding and jamming signals, leading to the formation of the total spatio-temporal the structure of the interference field in the phase centers of the three-element array antenna, which distorts different amplitude and phase θ _R θ _φ azimuthal bearing ionic characteristics; the joint use of the proposed two signs of the presence of an interfering radio signal at the time of measurement makes it possible to assess the reliability of direction finding results in the circular azimuth sector with arbitrary phase ratios of the direction-finding and interfering radio signals.

To implement the proposed method of direction finding, the direction finder, in contrast to the known one, includes a unit for generating unevenness coefficients of amplitude antenna radiation patterns, a calculator for the unambiguity coefficient of direction finding, a unit for generating unique amplitude values of difference signals, a unit for determining the phase difference between difference signals, a calculator for the quadrature component of the interference signal , noise threshold calculator, phase azimuth calculator, calculator azimuth estimation error calculator, azimuth threshold coefficient calculator and azimuth determination unit. In addition, the maximum distance between antennas allowed for unambiguous direction-finding was increased, which, as is known, helps to improve the accuracy and sensitivity of direction-finding, the comparator is able to determine an ordered set of antenna numbers through which the front of the electromagnetic wave of the radio signal passes through, not using phase differences between the signals received by the pairs of antennas, as is done in the closest analogue, and using amplitude values of both the difference signals received by the pairs of antennas and the signals received by each of the antennas of the three-element CAR, which makes it possible to eliminate anomalous direction finding errors, and the elevator calculator is made with the possibility of adaptive estimation of the elevation angle depending on the ratio of the minimum amplitude of the difference signals to the actual voltage noise of radio receiving blocks and errors in estimating the azimuth of the radio signal, which eliminates the methodological component of the systematic error in determining the slope of the front lektromagnitnyh waves due to mutual influences antenna and mast device and increases the accuracy of direction finding. And finally, the calculation parameter sensor is configured to generate a priori the known average values of the coefficients of the irregularity of the radiation patterns, the minimum necessary amplitude of the difference signal and the maximum permissible error in determining the azimuth, which makes it possible to operate both the input and the modified direction finder blocks.

These advantages, as well as features of the present invention are illustrated by the best option for its implementation with reference to the accompanying drawings.

Figure 1 depicts the location of the antennas in the direction-finding plane, explaining the essence of the proposed method of direction finding and direction finding for its implementation;

figure 2 - typical graphs of the functional dependence on the azimuth θ of the distortion values of one (first) phase difference Δφ ₁ (θ) between the signals received by a pair of antennas (second and third antennas) three-element CAR, due to the mutual influence of the antennas;

figure 3 - typical graphs of amplitude radiation patterns of one (first) antenna D ₁ (θ) in the azimuthal plane (β = 0) of a three-element CAR, due to the mutual influence of the antennas;

, изменяющихся в пределах от 0,3 до 0,66;figure 4 - experimental graphs of abnormal amplitude azimuthal radiation patterns D _{1 of} one (first) antenna of a three-element CAR with a ratio of the total length of a symmetric antenna of vibrator type to the distance b between the antennas in the array of 1.5, for various ratios

varying from 0.3 to 0.66;

трехэлементной КАР при β=0, свойственных правилу оценивания азимута с использованием однозначных амплитудных значений разностных сигналов согласно предложенному способу радиопеленгования (ошибок "разноса" |Δθ_Rmax|) и правилу оценивания азимута с использованием разностей фаз между парами сигналов согласно известным фазочувствительным способам пеленгования (ошибок, обусловленных взаимным влиянием между антенными элементами КАР);figure 5 - graphs of the dependencies of the modules of the maximum methodological errors on the ratio

three-element CAR at β = 0, characteristic of the rule of azimuth estimation using unique amplitude values of difference signals according to the proposed method of radio direction finding (separation errors | Δθ _Rmax |) and the rule of azimuth estimation using phase differences between signal pairs according to known phase-sensitive direction-finding methods (errors due to the mutual influence between the antenna elements of the CAR);

6 is a functional diagram of the claimed direction finder;

Fig. 7 is a functional diagram of an embodiment of a unit for generating non-uniformity coefficients of amplitude antenna patterns;

Fig. 8 is a functional diagram of an embodiment of a comparator;

Fig.9 is a functional diagram of a variant of implementation of the calculator of the coefficient of uniqueness of direction finding;

figure 10 is a functional diagram of a variant of implementation of the calculator noise threshold coefficient;

11 is a functional diagram of an embodiment of a unit for determining a phase difference between difference signals;

12 is a functional diagram of an embodiment of a phase difference calculator included in the phase difference determination unit between the difference signals and in the embodiment of the phase difference measurement unit;

Fig. 13 is a functional diagram of an embodiment of a unit for generating unique amplitude values of difference signals;

Fig. 14 is a functional diagram of an embodiment of a low-frequency calculator of unique amplitudes of difference signals included in a unit for generating unique amplitude values of difference signals;

FIG. 15 is a functional diagram of an embodiment of a high-frequency calculator of unique amplitudes of difference signals included in a unit for generating unique amplitude values of difference signals; FIG.

Fig. 16 is a functional diagram of an embodiment of a calculator for a parameter of the cyclic number of antennas included in a unit for generating unique amplitude values of difference signals;

17 is a functional diagram of an embodiment of an amplitude azimuth calculator;

Fig. 18 is a functional diagram of an embodiment of a calculator of a quadrature component of an interfering signal;

Fig. 19 is a functional diagram of an embodiment of a phase azimuth calculator;

Fig. 20 is a functional diagram of an embodiment of a calculator of an error in azimuth estimation;

Fig - functional diagram of a variant of implementation of the calculator elevation;

Fig. 22 is a functional diagram of an embodiment of an azimuth determination unit;

23 is a functional diagram of an embodiment of a differential signal generating unit;

Fig. 24 is an external view of an embodiment of an equidistant annular antenna array containing three identical rotationally axisymmetric vibrator-type antennas in a symmetrical design located on a mast device;

25 is an external view of an embodiment of a radio direction finder receiver including all direction finder blocks (except for the antenna array) and a monitor of an operator’s workstation;

Fig - experimental results of direction finding a broadband radio source having a true azimuth of 181 °, one of the samples of the direction finder that implements the claimed method of direction finding presented on the monitor screen of the operator’s workstation;

Fig.27 is the same as in Fig.26, for the true azimuth value of 196.8 °;

Fig.28 is the same as in Fig.26, for the true azimuth value of 211.7 °.

- расстояние между фазовыми центрами пар антенн эквидистантной КАР (база пеленгационных пар антенн); θ - азимут ИРИ, равный углу между проекцией направления S распространения ЭМВ на плоскость пеленгования ОР и линией ON (опорным направлением); β - угол наклона фронта волны (угол места) ИРИ, равный углу между направлением S распространения ЭМВ и проекцией направления S на плоскость пеленгования ОР.The spatial arrangement of the phase centers of the antennas A ₁ , A ₂ and A _{3 of a} flat three-element CAR of radius r _caw relative to the center O of the antenna array, the reference direction ON and the direction to IRI S is shown in Fig. 1, where, in addition, it is indicated: α ₁ , α ₂ and α ₃ are the angles between the reference direction ON and the lines passing through the center O CAR and the phase centers of the first A ₁ , second A ₂ and third A ₃ antennas, respectively, with α ₁ = 0 °;

- the distance between the phase centers of the pairs of antennas of the equidistant CAR (base of direction finding pairs of antennas); θ is the azimuth of the IRI, equal to the angle between the projection of the direction S of the propagation of electromagnetic waves on the direction-finding plane OP and the line ON (reference direction); β is the angle of inclination of the wave front (elevation angle) of the IRI, equal to the angle between the direction S of the EMW propagation and the projection of the direction S on the direction finding plane OR.

,

и

соответственно, которые при отсутствии электромагнитного поля помехового сигнала (Е_h=0) и пренебрежимо малых составляющих радиосигналов, обусловленных внутренним шумом каналов пеленгационного измерителя, с учетом обозначений параметров, приведенных в формуле (3), описываются выражениями:The electromagnetic field of the source of the direction-finding radio signal, characterized by: firstly, the amplitude E _s and the phase φ _so at point O (Fig. 1), which is the center of a flat three-element equidistant ring antenna array of radius r _caw formed by the first, second and third antennas A ₁ , A ₂ and A ₃ with an angular orientation in the direction-finding plane α ₁ , α ₂ and α _3, respectively, and interelement distance b; secondly, the propagation direction S, described by the angle θ between the projection of the S direction on the direction-finding plane of the OP and the ON line (reference direction) and the angle β between the direction S and the projection of the S direction on the direction-finding plane of the OP, forms at the inputs of identical non-directional axisymmetric antennas A ₁ A ₂ and A ₃ vibrator-type signals

,

and

respectively, which in the absence of an electromagnetic field of the interfering signal (Е _h = 0) and negligible components of the radio signals caused by the internal noise of the direction-finding meter channels, taking into account the designations of the parameters given in formula (3), are described by the expressions:

,

и

антенн трехэлементной антенной решетки с учетом электродинамического взаимодействия в общем виде описываются выражениями (4) или (8).moreover, complex radiation patterns

,

and

antennas of a three-element antenna array, taking into account the electrodynamic interaction, are generally described by expressions (4) or (8).

и

, принятых n-ой и k-ой антеннами КАР, так и в разностных сигналах

и их амплитудных значениях r_i (i=1, 2, 3), определяемых по правилам (30), (34) и (35) соответственно, которые с учетом (60), (4) - (10) описываются выражениями:Information on the direction of propagation of the electromagnetic wave of the IRI along the azimuth θ and elevation angle β for arbitrary electrodynamic dimensions of the axisymmetric antennas of the three-element equidistant CAR is contained in the phase differences φ _i between the signal pairs

and

received by the n-th and k-th CAR antenna, and in difference signals

and their amplitude values r _i (i = 1, 2, 3), determined by the rules (30), (34) and (35), respectively, which, taking into account (60), (4) - (10), are described by the expressions:

where θ _i = θ-α _i is the angle between the projection of the electromagnetic propagation direction and the direction of the antenna A _i from the center O of the antenna array;

Δφ _i = χ ₁ -χ ₂ is the distortion of the phase difference φ _i due to the electrodynamic interaction between the CAR antennas;

и

- действительная и мнимая части комплексного параметра

,

and

- real and imaginary parts of the complex parameter

,

- коэффициент передачи входной цепи при формировании разностного сигнала из пары сигналов, принятых парой антенн.

- transmission coefficient of the input circuit when generating a differential signal from a pair of signals received by a pair of antennas.

,

и

соответственно.Typical graphs of the functional dependence on the azimuth in the distortion values of one (first) phase difference Δφ _i (θ) between the signals received by a pair of antennas (second and third antennas) of a three-element CAR, characterized by the ratio of the total length 2l _{a of the} antennas along their axis of symmetry to the distance b between antennas (base) equal to 1.5, obtained by mathematical modeling using formulas (63) and (64), are presented in figure 2, where the curves 1, 2 and 3 are obtained for β = 0 and with the ratios

,

and

respectively.

является приближенным, а выражение (62), описывающее

является точным независимо от отсутствия

или наличия

электродинамически взаимосвязанного с антеннами мачтового устройства, расположенного в центре КАР, что обусловлено полной компенсацией искажений комплексных ДН антенн

,

и

описываемых выражениями (4), при формировании по правилу (34) разностных сигналов, принятых парами антенн. Вместе с тем разности фаз φ_i между сигналами, принятыми парами антенн, содержат "искаженную" (из-за взаимного влияния антенн и мачтового устройства) и однозначную (в пределах возможностей однозначного измерения разностей фаз ±π радиан) информацию об азимуте и угле места β (с учетом формулы (5)), а разностные сигналы

содержат "неискаженную" (из-за взаимного влияния антенн и мачтового устройства) и неоднозначную информацию об азимуте θ и угле места β ИРИ, которая содержится как в амплитудах r_i, так и в фазах разностных сигналов. Кроме того, в случае использования в качестве антенн трехэлементной эквидистантной КАР идентичных осесимметричных антенн вибраторного типа с геометрическими размерами по их осям симметрии, соизмеримыми с длиной волны λ радиосигнала, достаточно "грубая", но однозначная информация об азимуте θ и угле места β ИРИ также содержится в амплитудах u_i сигналов, принятых i-ыми антеннами трехэлементной КАР, которые с учетом (37), (60) и (8) описываются выражениями:It should be noted that expression (61) describing φ _i , in the presence of a mast device

is approximate, and expression (62) describing

is accurate regardless of absence

or availability

the mast device electrodynamically interconnected with the antennas located in the center of the CAR

,

and

described by expressions (4), when generating, according to rule (34), difference signals received by pairs of antennas. At the same time, the phase differences φ _i between the signals received by the antenna pairs contain “distorted” (due to the mutual influence of the antennas and mast devices) and unambiguous (within the limits of the possibility of unambiguous measurement of phase differences ± π radians) azimuth and elevation angle β (taking into account formula (5)), and difference signals

contain “undistorted” (due to the mutual influence of antennas and mast devices) and ambiguous information about the azimuth θ and elevation angle β of the IRI, which is contained both in the amplitudes r _i and in the phases of the difference signals. In addition, if three-element equidistant CARs are used as antennas, identical axisymmetric vibrator-type antennas with geometric dimensions along their symmetry axes commensurate with the wavelength λ of the radio signal are quite “rough”, but unequivocal information about the azimuth θ and elevation angle β of the IRI is also contained in the amplitudes u _{i of the} signals received by the i-th antennas of the three-element CAR, which, taking into account (37), (60) and (8), are described by the expressions:

where D _i is the amplitude radiation pattern of the i-th antenna in the composition of the three-element equidistant CAR determined by the formula:

, полученные методом математического моделирования с использованием формулы (66), представлены на фиг.3, где кривые 1, 2 и 3 получены при

;

и

соответственно. Кроме того, на фиг.3 соответствующими кривым 1, 2 и 3 символами отмечены результаты экспериментальных исследований D₁(θ) при соответствующих отношениях

. Необходимо отметить, что указанные амплитудные диаграммы направленности не нормированы в диапазоне изменения длин волн. Таким образом, для формирования однозначных пеленгационных характеристик, не подверженным методическим ошибкам пеленгования, обусловленным взаимным влиянием антенн и мачтового устройства, целесообразно использовать амплитуды и фазы разностных сигналов с устранением их неоднозначности на основе использования как разностей фаз φ_i между парами сигналов, так и соотношений амплитуд u_i сигналов, принятых антеннами решетки. Вместе с тем в зависимости от отношения расстояния b между антеннами (

) к длине волны λ радиосигнала правила и качество формирования однозначных пеленгационных характеристик (вероятность правильного устранения неоднозначности пеленгования) является различными в случаях использования разностей фаз φ_i или соотношений амплитуд u_i сигналов, принятых антеннами КАР.Typical graphs of amplitude radiation patterns of one (first) antenna D ₁ (θ) in the azimuthal plane (β = 0) for a three-element CAR, characterized by the ratio

obtained by mathematical modeling using formula (66) are presented in figure 3, where curves 1, 2 and 3 are obtained at

;

and

respectively. In addition, in figure 3, the corresponding curves 1, 2 and 3 symbols indicate the results of experimental studies of D ₁ (θ) with the corresponding ratios

. It should be noted that these amplitude radiation patterns are not normalized in the range of wavelengths. Thus, for the formation of unambiguous direction-finding characteristics that are not subject to methodological direction finding errors due to the mutual influence of antennas and mast devices, it is advisable to use the amplitudes and phases of the difference signals with the elimination of their ambiguity based on the use of both phase differences φ _i between signal pairs and amplitude ratios u _{i the} signals received by the antennas of the array. However, depending on the ratio of the distance b between the antennas (

) to the wavelength λ of the radio signal, the rules and the quality of formation of unambiguous direction-finding characteristics (the probability of correct elimination of direction finding ambiguity) is different in cases where phase differences φ _i or amplitude ratios u _i received by CAR antennas are used.

From the formula (30) for measuring the phase differences φ _i it follows that, in the case of an unambiguous measurement of all three phase differences φ _i , the condition is satisfied:

According to (67), any of the three phase differences φ _m (m = 1, 2 or 3) can be determined by summing the two other phase differences according to the formula:

Therefore, if at least two of the three phase differences φ _{i are} measured unambiguously, which is possible if their absolute values (modules) are less than π radians, then the third (including the maximum modulus) phase difference φ _m can be uniquely determined by the formula (68). Therefore, to reduce the likelihood of anomalous errors in determining phase differences between signals that are close to antiphase (the module of the phase differences between them is close to π radian) and distorted due to the mutual influence of the array antennas in accordance with (61) and the effect of noise in accordance with (14), it is necessary: firstly, to determine an ordered set of phase differences φ _ξ , φ _γ and φ _υ , determined by formula (30), for which the condition is satisfied:

where ξ, γ, and ν are indices that are not equal to each other and take one of three values 1, 2, or 3; secondly, using the measured values of φ _ξ and φ _{γ to} determine the value of φ _ν by formula (68). An ordered set of phase differences φ _ξ , φ _γ and φ _ν can be determined by direct comparison of the moduli of the indicated phase differences, as is done in the prototype [10], however, taking into account, according to (61), the presence of distortions Δφ _{i of the} measured phase differences reaching at sizes antennas commensurate with the wavelength, values of the order of ± 50 ° (see Fig. 2), and also, taking into account additional measurement errors of the phase differences σ _φ associated with the non-uniformity of the measurements under the influence of noise and determined by the formula (14), this ordered owl the completeness of phase differences is more preferably determined by comparing both the amplitude values of the difference signals r _i and the amplitude values u _{i of the} signals received by the antennas of the three-element CAR. This is due to the following features of the azimuthal dependences r _i and u _i .

The amplitude values of the difference signals r _i in accordance with (35) and (62) are determined by the expression:

Where

From a comparison of (70) and (61) it follows that, since r _i does not depend on Δφ _i , the ordered set of phase differences (69) can be determined more reliably based on the definition of an ordered set according to the rule:

для различных отношений

, изменяющихся в пределах от 0,3 до 0,66. Анализ формулы (66) и представленных теоретических (см. фиг.3) и экспериментальных (см. фиг.3 и фиг.4) результатов позволяет выявить следующие закономерности амплитудных ДН трехэлементной КАР: а) все амплитудные ДН D_i при

имеют явно выраженный минимум в азимутальных направлениях θ_min.i, определяемых условиями:The determination of the ordered set of phase difference (69) according to rule (71) allows one to eliminate errors caused by the mutual influence of antennas and mast devices, however, to reduce the likelihood of “confusing neighboring” indices ξ and γ or γ and ν under noise exposure, an additional comparison of the amplitude the values u _{i of the} signals received by the antennas, which, in turn, according to (65) and (66), are determined by the amplitude radiation patterns of D _i antennas in the three-element CAR. The rules for comparing the amplitude values u _{i of} signals received by antennas, designed to improve the quality of determining an ordered set of phase difference (69) and, ultimately, to resolve the direction finding ambiguity, follow from the analysis of amplitude radiation patterns D _i described by formula (66), explained graphs of the first antenna D ₁ (θ) presented in the above figure 3, as well as in figure 4, which shows the measurement results (in relative units (rel. units)) of the normalized amplitude azimuthal signals D ₁ od the first (first) antenna of the three-element CAR at

for various relationships

varying from 0.3 to 0.66. The analysis of formula (66) and the theoretical (see Fig. 3) and experimental (see Fig. 3 and Fig. 4) results presented allows us to identify the following regularities of the amplitude patterns of the three-element CAR: a) all amplitude patterns of D _i at

have a pronounced minimum in the azimuthal directions θ _min.i determined by the conditions:

неравномерность амплитудных ДН превышает 6 дБ; в) при

все амплитудные ДН D_i имеют максимум в направлениях θ_max.i, определяемых условиями:b) when

non-uniformity of amplitude DNs exceeds 6 dB; c) when

all amplitude MD D _i have a maximum in the directions θ _max.i determined by the conditions:

крутизна изменения амплитудных ДН в районах минимумов и максимумов не превышает (0,04÷0,05) дБ/град и при отклонении минимумов (максимумов) амплитудных ДН на угол

уровень принимаемого радиосигнала возрастает (уменьшается) не более чем на (1,2÷1,5) дБ.d) when

the steepness of the change in amplitude MDs in the regions of minima and maxima does not exceed (0.04 ÷ 0.05) dB / deg and when the minima (maxima) of the amplitude MDs deviate by an angle

the level of the received radio signal increases (decreases) by no more than (1.2 ÷ 1.5) dB.

The indicated regularities of the forms of the amplitude azimuthal DN D _i allows the formation of formulas (38) and (39) of the coefficients P _i and K _{i of the} irregularities of the radiation patterns necessary to improve the quality of eliminating direction finding ambiguity.

и

различаются между собой (то есть Р_i стремится к единице), тем меньше истинное значение модулей разностей фаз между указанными сигналами, что и определяет предпочтительность использования формулы (40) по сравнению с формулой (71) для формирования упорядоченной совокупности разности фаз (69).The coefficients P _i determined by the formula (38) satisfy the condition: P _i ≥1. Therefore, the smaller the compared amplitudes u _n and u _k signals

and

differ from each other (that is, P _i tends to unity), the lower the true value of the modulus of the phase difference between the indicated signals, which determines the preference for using formula (40) compared to formula (71) for forming an ordered set of phase difference (69).

позволяют осуществлять селекцию принимаемых электромагнитных волн по четырем зонам в азимутальной плоскости: зоне равноточных измерений и трем зонам неравноточных измерений, соответствующим азимутальным сектором, определяемым из условий:The coefficients K _i determined by the formula (39), in the range of wavelengths

allow the selection of received electromagnetic waves in four zones in the azimuthal plane: the zone of equal measurements and three zones of unequal measurements, the corresponding azimuthal sector, determined from the conditions:

и отношения длины мачтового устройства к длине волны радиосигнала и находится в пределах, близких к единице. Так, например, усредненное в пределах диапазона изменения длин волн

значение К_mid для трехэлементной КАР, амплитудные ДН которой представлены на фиг.4, составляет 0,84.where i = 1, 2, 3. Moreover, for the indices ξ, γ, and ν selected by formula (40), the antenna numbers A _ξ , A _γ, and A _ν are determined by formula (39), the coefficient K _ξ is compared with the a priori known average the value of K _{mid of the} coefficients K _{i of the} unevenness of the antenna patterns in the array and the direction finding coefficient p is determined by formula (41). The value of the parameter K _mid depends on the electrodynamic dimensions of the antennas (length 2l _a along the axis of symmetry and the diameter of the axisymmetric vibrator antenna, referred to the wavelength λ of the radio signal), the ratio

and the ratio of the length of the mast device to the wavelength of the radio signal and is in the range close to unity. So, for example, averaged over a range of wavelengths

the value of K _mid for a three-element CAR, the amplitude of which is presented in figure 4, is 0.84.

In connection with the aforementioned, if the distance b between the antennas does not exceed the third part of the wavelength λ of the radio signal, i.e., there is a small non-uniformity of the amplitude MDs, three unique phase differences are formed using the two measured phase differences φ _ξ and φ _γ F _i (i = 1, 2, 3).

, β=0, φ_ξ=0 и

однозначно формируются три разности фаз F_i, определяется с учетом (14), (61), (63) и (64) в общем виде из условия:The minimum required signal-to-noise ratio q _min , at which, for one of the most unfavorable (with the highest possible probability of anomalous errors to eliminate the unambiguity of direction finding by formula (43)) cases of receiving radio signals at

, β = 0, φ _ξ = 0 and

uniquely formed three phase differences F _i , is determined taking into account (14), (61), (63) and (64) in general form from the condition:

In particular, for the characteristics shown in Fig. 2 and Fig. 3 (curves 2) with the coefficient a of the unevenness of measurements equal to the ratio of the amplitude of the smallest signal to the amplitude of the largest signal and equal to 0.6, it follows from (75) that for the direction-finding meter channel with the largest signal amplitude, the signal-to-noise ratio q _min = 4 (the signal-to-noise ratio for the meter channel with the smallest signal amplitude is 0.6 q _min = 2.4, respectively).

Taking into account (43) and (61) for the unique phase differences F _i we get:

Since the condition is satisfied:

then, taking into account (70) and (76), the unambiguous amplitude values (taking into account the signs) of the difference signals R _i generated by the formula (42) can be represented as:

и

между которыми в соответствии с (30) измеряется разность фаз φ_ξ, используемая при формировании R_ξ по формуле (44), принимается соответствующей парой антенн А_γ и А_ν, расположенных на линии, наиболее близкой к фронту электромагнитной волны ИРИ. Это означает, что третья антенна А_ξ, не входящая в состав выбранной пары антенн, размещается либо со стороны прихода фронта ЭМВ (в этом случае К_ξ≥К_mid), либо со стороны, противоположной стороне прихода фронта ЭМВ (в этом случае К_ξ<К_mid). Поэтому, в зависимости от результатов оценки коэффициента р однозначности пеленгования по формуле (41) и результатов сравнения взаимного расположения γ-ой пары антенн А_ξ и А_ν (сигналы, принятые которыми, используются для формирования r_γ) и вышеупомянутой выбранной ξ-той пары антенн А_γ и А_ν, по формуле (45), приводящему к определению значения коэффициента l, связанного с априорно выбранным в соответствии с (30)-(35) правилом сравнения сигналов, принятых последовательными парами антенн, по формуле (44) производится определение второго однозначного амплитудного значения разностных сигналов R_γ. Учитывая то, что, с одной стороны, R_ξ, R_γ и Rv_ν связаны соотношением, аналогичным (67), а с другой стороны, следующим из условия (40) соотношением:If the distance b between the antennas exceeds a third of the wavelength λ of the radio signal, that is, there is a significant non-uniformity of the amplitude DNs, three unambiguous amplitude values of the difference signals R _i are formed by the formula (44) using the corresponding amplitude values of the difference signals r _i , one measured the phase difference φ _ξ and the values of the coefficient p of the uniqueness of direction finding obtained by comparing the amplitude values of the signals u _i in accordance with (39) and (41). In this case, a pair of signals selected by formula (40)

and

between which, in accordance with (30), the phase difference φ _ξ is measured, which is used in the formation of R _ξ according to formula (44), is taken by the corresponding pair of antennas A _γ and A _ν located on the line closest to the front of the electromagnetic wave of the IRI. This means that the third antenna A _ξ , which is not part of the selected pair of antennas, is placed either on the side of the arrival of the EMF front (in this case, K _ξ ≥K _mid ), or on the side opposite to the side of the arrival of the EMF front (in this case, K _ξ <K _mid ). Therefore, depending on the results of estimating the direction finding coefficient p by formula (41) and the results of comparing the relative positions of the γ-th pair of antennas A _ξ and A _ν (the signals received by which are used to form r _γ ) and the aforementioned selected ξ-th pair antennas A _γ and A _ν , according to the formula (45), which leads to the determination of the coefficient l associated with the a priori selected in accordance with (30) - (35) rule for comparing signals received by successive pairs of antennas, using the formula (44), second clearly th amplitude value of the difference signals R _γ . Taking into account the fact that, on the one hand, R _ξ , R _γ, and Rv _ν are related by a relation similar to (67), and on the other hand, by the relation follows from condition (40):

we find that the sign of the third single-valued amplitude value of the difference signals R _{γ is} opposite to the sign of the aforementioned second unambiguous amplitude value of the difference signals R _γ , which is realized when determining R _γ by the formula (44).

, при котором в круговом азимутальном секторе реализуется достоверное определение трех однозначных амплитудных значений разностных сигналов R_i в соответствии с (44) следует считать

в связи со следующим обстоятельством. При

для β=0 и направлений распространения ЭМВ вдоль линий, соединяющих любые пары антенн трехэлементной КАР, из (70) получаем:It should be noted that the maximum limit value of the ratio

in which in the circular azimuthal sector a reliable determination of three unique amplitude values of the difference signals R _i is implemented in accordance with (44) should be considered

in connection with the following circumstance. At

for β = 0 and EME propagation directions along the lines connecting any pairs of antennas of a three-element CAR, from (70) we obtain:

соответствие упорядоченной совокупности (71) и упорядоченной совокупности Р_ξ≤Р_γ≤Р_ν нарушается, что приводит к снижению качества определения упорядоченной совокупности индексов ξ, γ и ν номеров антенн по формуле (40) и, соответственно, к увеличению вероятности появления аномальных ошибок пеленгования.Moreover, the ordered set of indices ξ, γ, and ν of the serial numbers of the antennas at which condition (40) is satisfied is determined only by the coefficients P ₁ , P _2, and P _{3 of the} irregularity of the antenna patterns in the array. When

the correspondence between the ordered set (71) and the ordered set Р _ξ ≤Р _γ ≤Р _{ν is} violated, which leads to a decrease in the quality of determining the ordered set of indices ξ, γ, and ν of antenna numbers by formula (40) and, accordingly, to an increase in the probability of anomalous errors direction finding.

, β=0 и

однозначно формируется разность фаз φ_ξ (и, соответственно, R_ξ, R_γ и R_ν), определяется с учетом (14), (61), (63), (64), (38) и (40) в общем виде из условия:The minimum required signal-to-noise ratio q _min , at which, for one of the most unfavorable (with the highest possible probability of anomalous errors of eliminating direction-finding ambiguity according to formula (44)) radio signal reception

, β = 0 and

the phase difference φ _{ξ is} uniquely formed (and, accordingly, R _ξ , R _γ and R _ν ), it is determined taking into account (14), (61), (63), (64), (38) and (40) in the general form from the condition:

In particular, for the characteristics shown in Fig. 2 and Fig. 3 (curves 2) with the coefficient a of the unevenness of the measurements being 0.9 (since the signals U _γ and U _ν with amplitudes u _γ and u are used when measuring the phase difference φ _{ξ ν} , close in value and significantly exceeding the amplitude u _{ξ of the} signal not used to estimate the above phase difference), and taking into account the opposite signs of the measured phase difference and the error Δφ _ξ due to the mutual influence of the antennas (the mutual influence between the antennas leads to a "slowdown" speed propagation of the EMF front between the pair of antennas under consideration), it follows from formula (81) that q _min ≈2.3.

It should be noted that the formation of three unique amplitude values of the difference signals R _i (i = 1, 2, 3) according to the above formula (44), described by expression (78), is possible even if there is no signal in one of the three antennas of the three-element CAR caused by full compensation of the electromagnetic field at the location of the aforementioned antenna due to the electrodynamic interaction of the lattice antennas and the mast device. Moreover, due to the mentioned electrodynamic interaction, the signal levels in two other antennas increase, which leads to an additional improvement in direction-finding sensitivity and a decrease in the random components of direction-finding errors σ _θr determined by formula (2). At the same time, under the indicated conditions (the absence of a signal in one of the three channels of the direction-finding meter due to the electrodynamic interaction of antenna elements), none of the considered analogs [8–10] provides the possibility of determining the direction of propagation of electromagnetic waves.

To simplify the method of determining the azimuth θ _R using the unique amplitude values of the difference signals R _i , as in the prototype [10], in expressions (78) we replace the first sine functions with arguments. Moreover, expressions (78) with errors of "separation" [1] can be represented as:

where h _r is the effective effective length of the direction finding pair of antennas with a difference radiation pattern, determined taking into account (5) and (70) according to the expression:

By well-known trigonometric transformations, the numerator X ₁ and the denominator X _{2 of} expression (36), taking into account (82) and (6), can be represented as:

, подробно исследованы и проанализированы в ближайшем аналоге [10] и работе [7], где показано, что в наихудшем случае при

, β=0 и

, где m=0, 1, 2, ...11, абсолютная ошибка "разноса" не превышает 0,9°. График зависимости модуля максимальных ошибок "разноса" |Δθ_Rmax| от отношения

при β=0, свойственных формуле (36) оценивания азимута θ_R, представлен на фиг.5 в виде кривой 1. Необходимо отметить, что указанные методические ошибки оценки азимута θ_R по формуле (36) не зависят от электродинамического взаимодействия между антенными элементами решетки в отличие от способов пеленгования, основанных на сравнении разностей фаз между сигналами, принятыми парами антенн трехэлементной КАР. Так, в частности, на фиг.5 для сравнения представлены в виде кривых 2, 3 и 4 зависимости от отношения

максимальных систематических ошибок |θ_max| характерных для правила (12) определения азимута θ при различных отношениях общей длины 2l_a симметричных антенн к расстоянию b между антеннами в трехэлементной КАР.From (84) and (85) immediately follows the formula (36) for estimating the azimuth θ _{R of the} radio signal source using three unique amplitude values of the difference signals R _i . The absolute values and signs of errors of the “separation” Δθ _R characteristic of formula (36) for estimating the azimuth θ _R and due to the use of an approximation of the form sin (X) ≈ X when representing R in the form (82) depend on θ, β and

, are investigated and analyzed in detail in the closest analogue [10] and work [7], where it is shown that in the worst case for

, β = 0 and

, where m = 0, 1, 2, ... 11, the absolute error of the "separation" does not exceed 0.9 °. The graph of the dependence of the module of the maximum errors "separation" | Δθ _Rmax | from attitude

at β = 0, characteristic of formula (36) for estimating the azimuth θ _R , is shown in Fig. 5 in the form of curve 1. It should be noted that these methodological errors in estimating the azimuth θ _R according to formula (36) are independent of the electrodynamic interaction between the antenna elements of the array unlike direction finding methods, based on a comparison of phase differences between signals received by pairs of antennas of a three-element CAR. So, in particular, in Fig. 5 for comparison are presented in the form of curves 2, 3 and 4 depending on the ratio

maximum systematic errors | θ _max | characteristic of rule (12) for determining the azimuth θ for different ratios of the total length 2l _{a of} symmetrical antennas to the distance b between the antennas in a three-element CAR.

The determination of the parameter μ in accordance with expression (47) using the three unambiguous amplitude values of the difference signals R _i (i = 1, 2, 3) determines the presence of the quadrature component of the interfering signal. So, in the general case of simultaneous reception of EMF sources of direction-finding and interference signals described by formula (3) and negligible components of radio signals due to the internal noise of the direction-finding meter channels, taking into account the rule for obtaining R _i described by formulas (30) - (35) , (37) - (45), and approximations of the form sin (X) ≈X the parameter μ described by expression (47) can be represented as:

where q _h is a parameter that determines the ratio of amplitudes and directions of propagation of electromagnetic waves of interfering and direction-finding radio signals, described by the expression:

| q _h | ≤1.

которое соответствует наличию квадратурной составляющей помехового сигнала μ=q_h, а в случаях отсутствия квадратурной составляющей помехового сигнала (φ_ho-φ_so=0 или φ_ho-φ_so=π), совпадения линий распространения пеленгуемого и помехового сигналов (θ_h-θ=0 или θ_h-θ=π) или отсутствия помехового сигнала (Е_h≪E_s) μ=0. Необходимо также отметить, что при наличии составляющих радиосигналов, обусловленных внутренним шумом каналов пеленгационного измерителя, которыми по сравнению с уровнем пеленгуемого радиосигнала нельзя пренебречь (то есть при низких отношениях сигнал/шум q) значения параметра μ могут изменяться в пределах от 0 до 1 при изменении от q≫1 до q=q_min≈(1÷2) соответственно.It follows from formula (86) that, under the condition

which corresponds to the presence of the quadrature component of the interfering signal μ = q _h , and in the absence of the quadrature component of the interfering signal (φ _ho -φ _so = 0 or φ _ho -φ _so = π), the propagation lines of the direction-finding and interference signals (θ _h -θ = 0 or θ _h -θ = π) or the absence of an interfering signal (Е _h ≪E _s ) μ = 0. It should also be noted that in the presence of components of the radio signals caused by the internal noise of the direction-finding meter channels, which cannot be neglected compared to the level of the direction-finding radio signal (i.e., at low signal-to-noise ratios q), the values of the parameter μ can vary from 0 to 1 when changing from q≫1 to q = q _min ≈ (1 ÷ 2), respectively.

Thus, estimating the value of the parameter μ by the formula (47) allows us to judge the reliability of the results of direction finding, and the reliability of the results of direction finding is inversely proportional to the value of the parameter μ.

и

от знаков лепестков их амплитудных диаграмм направленности определение вышеупомянутых разностей фаз φ_Ri производится по формуле (46), которая с учетом (62), (70), (78), (5), (6) и (31)-(33) после известных тригонометрических преобразований представляется в виде:To eliminate the dependence of the phase differences φ _Ri between the difference signals

and

from the signs of the petals of their amplitude radiation patterns, the aforementioned phase differences φ _{Ri are} determined by the formula (46), which, taking into account (62), (70), (78), (5), (6) and (31) - (33) after the known trigonometric transformations it is represented in the form:

Formula (88), taking into account the relationships between the indices i, n and k defined by formulas (31) - (33), essentially describes the phase differences φ _Ri between the kth and nth signals with amplitudes r _k and r _n, respectively, adopted equivalent directional electrodynamically non-interacting antennas, the phase centers of which are located at points C _k and C _n indicated in Fig. 1 and located in the middle of the line segments connecting the phase centers of the kth pair of antennas A _i and A _n and n pairs of antennas A _i and A _n . Moreover, the aforementioned equivalent directional antennas form in the direction-finding plane a three-element equidistant annular antenna array with a radius less than 2 times smaller than the antenna array formed by antennas A ₁ , A ₂ and A ₃ (see figure 1), however, two of the three amplitudes of the signals generated by the equivalent directional antennas according to (8), (65) and (70), on average (1.5 ÷ 2) times the maximum of the three amplitudes of the signals received by the antennas A ₁ , A ₂ and A ₃ , which leads to an improvement in the signal-to-noise ratio q and compensates for the neg The active effect of decreasing the radius r _{caw of the} lattice on the boundary of the potentially attainable direction finding accuracy determined by formula (2).

и

имеющими амплитуды r_γ и r_ν, превышающими амплитуду r_ξ. Значение минимальной амплитуды r_min разностного сигнала, описываемое формулой (49), является априорно известной величиной, определяемой минимально необходимым отношением сигнал/шум q_min, обеспечивающем в соответствии с (2) пеленгование источников радиоизлучения с требуемыми точностью и вероятностью, и действующим значением напряжения внутреннего шума U_eff каналов формирования разностных сигналов пеленгационного измерителя, зависящим от аппаратной реализации радиоприемных каналов.A comparison of formulas (61) and (88) shows that the phase differences φ _Ri between the difference signals, unlike the phase differences φ _i between the signals received by the CAR antennas, are not distorted due to the electrodynamic interaction between the antennas and the mast device, which makes it possible determination of azimuth θ and elevation angle β using φ _Ri without methodological errors. However, for azimuthal directions that coincide or are close to the lines passing through the center O CAR and any of the three antennas A ₁ , A ₂ or A ₃ (see figure 1), one of the three amplitudes r ₁ , r ₂ or r ₃ difference signals, namely r ₃ , due to the "sinus" radiation pattern defined by formula (70), can decrease to a value less than the minimum required level r _{min of} difference signals, at which, according to (14), the phase difference between the signals is measured with random mean square error σ _φ , which provides direction finding of radio sources exercises with the required accuracy and probability. In this case, only one of the three phase differences φ _R1 , φ _R2 or φ _R3 can be reliably determined, namely φ _Rξ , which is determined, according to (46), between the difference signals

and

having amplitudes r _γ and r _ν exceeding the amplitude r _ξ . The value of the minimum amplitude r _{min of the} difference signal described by formula (49) is a priori known value determined by the minimum necessary signal-to-noise ratio q _min , which ensures, in accordance with (2), direction finding of radio emission sources with the required accuracy and probability, and the effective value of the internal voltage noise U _{eff of the} channels for generating the difference signals of the direction-finding meter, depending on the hardware implementation of the radio receiving channels.

In connection with the foregoing, in order to select the rule of unambiguous determination of the azimuth θ _{φ of the} radio signal source using phase differences φ _Ri between the difference signals, condition (48) is checked, under which the azimuth θ _φ is determined by the formula (50), and otherwise, by formula (51).

So, in the case r _ξ ≥r _{min, the} numerator X ₁ and the denominator X _{2 of} expression (50), taking into account (88) and (6) by known trigonometric transformations, can be represented as:

Formulas (89) and (90) directly imply rule (50) for unambiguous estimation of the azimuth θ _{φ of the} radio source using the three phase differences φ _R1 , φ _R2 and φ _R3 between the difference signals.

In the case where r _ξ ≥ r _min , the azimuth θ _{φ is} estimated, firstly, using only one phase difference φ _Rξ , described in view of (88) and (6) by the expression:

, принимающего в зависимости от значения коэффициента р одно из двух значений 0 или π, и выбора знака функции arcsin(X) путем ее умножения на коэффициент р, что обеспечивает устранение ошибок "зеркального" оценивания азимута в случае добавления вышеупомянутого слагаемого, равного π, в-третьих, с учетом предположения, что пеленгуемый радиосигнал распространяется в виде поверхностной электромагнитной волны, при которой β=0 и, соответственно, cosβ=1. Из формулы (91) с учетом вышеупомянутых факторов непосредственно следует формула (51) однозначного оценивания азимута θ_φ источника радиосигнала, которая в случае приема поверхностных сигналов обеспечивает получение несмещенной оценки азимута θ источника радиосигнала, а в случае приема пространственных сигналов позволяет оценить азимут с ошибками Δθ_sp, зависящими от углов β прихода пространственных волн, называемых "высотными" ошибками азимутальных радиопеленгаторов [1]. Для выбора правил однозначного определения азимута θ и оценивания угла β наклона фронта волны источника радиосигнала и суждении о достоверности результатов пеленгования по формуле (52) с учетом (53) производят сравнение результатов оценивания азимута θ_R и θ_φ с использованием соответственно амплитудных значений разностных сигналов R_i и разностей фаз φ_Ri между разностными сигналами (то есть с использованием амплитудных и фазовых пеленгационных характеристик), определяя при этом погрешность Δθ однозначной оценки азимута θ. Причем определение значения параметра m по формуле (53) позволяет исключить аномальные ошибки определения погрешности Δθ для азимутов ИРИ, близких к 0°. Так, например, в случае, если θ_R=359°, а θ_φ=1°, в соответствии с (53) получаем: |θ_R-θ_φ|=|359°-1°|=358°>180°=π и, соответственно, m=1 и Δθ=|358°-360°|=|2°|=2°.secondly, with the use of information on the value of the direction finding coefficient p, determined by formula (41), which allows one to resolve the ambiguity of the function arcsin (X) by adding the term

taking, depending on the value of the coefficient p, one of the two values 0 or π, and choosing the sign of the function arcsin (X) by multiplying it by the coefficient p, which eliminates the errors of "mirror" azimuth estimation in the case of adding the above term equal to π in thirdly, taking into account the assumption that the direction-finding radio signal propagates in the form of a surface electromagnetic wave, in which β = 0 and, accordingly, cosβ = 1. From formula (91), taking into account the above factors, directly follows formula (51) for an unambiguous estimation of the azimuth θ _{φ of the} radio signal source, which, in the case of receiving surface signals, provides an unbiased estimate of the azimuth θ of the radio signal source, and in the case of receiving spatial signals, it allows to estimate the azimuth with errors Δθ _sp , depending on the angles β of the arrival of spatial waves, called "high-altitude" errors of azimuth direction finders [1]. To select the rules for the unambiguous determination of the azimuth θ and estimation of the angle β of the slope of the wave front of the radio signal source and judging the reliability of the direction finding results using formula (52), taking into account (53), the azimuth estimation results θ _R and θ _φ are compared using the amplitude values of the difference signals R _i and phase differences φ _Ri between the difference signals (that is, using amplitude and phase direction-finding characteristics), while determining the error Δθ of an unambiguous estimate of the azimuth θ. Moreover, the determination of the value of the parameter m by the formula (53) allows us to exclude anomalous errors in determining the error Δθ for the IRI azimuths close to 0 °. So, for example, if θ _R = 359 °, and θ _φ = 1 °, in accordance with (53) we obtain: | θ _R -θ _φ | = | 359 ° -1 ° | = 358 °> 180 ° = π and, accordingly, m = 1 and Δθ = | 358 ° -360 ° | = | 2 ° | = 2 °.

; во-вторых, наличием систематической средней квадратической ошибки σ_θs2 пеленгования, обусловленной неидентичностью каналов пеленгационного измерителя, которая при современных способах калибровки каналов пеленгационного измерителя [2] составляет: σ_θs2≈0,5°; в-третьих, требуемым в соответствии с [3] предельно допустимым значением случайной средней квадратической ошибки σ_θr, обусловленной воздействием внутренних шумов пеленгационного измерителя при минимальной напряженности электромагнитного поля пеленгуемого радиосигнала, характеризующей, в соответствии с (2), предельную чувствительность радиопеленгатора, которую целесообразно выбирать из условия:The obtained error value Δθ is compared with the a priori known value of the maximum permissible error Δθ _{max for} determining the azimuth, which is determined: firstly, by the presence of the aforementioned "separation" errors characteristic of the rule for estimating the azimuth θ _R according to formula (36), the maximum value of which for the worst cases not more than 0,9 ° and that can be attributed to systematic error of direction finding, characterized systematic mean square error σ _θs1, wherein the value determined by the formula "three sigma" [4], comp S THE:

; secondly, the presence of a systematic mean square error σ _{θs2 of} direction-finding, due to the non-identity of the direction-finding meter channels, which with modern methods of calibrating direction-finding meter channels [2] is: σ _θs2 ≈0.5 °; thirdly, the maximum permissible value of the random mean square error σ _θr , required in accordance with [3], due to the influence of internal noise from the direction-finding meter at the minimum electromagnetic field strength of the direction-finding radio signal, which characterizes, in accordance with (2), the maximum sensitivity of the direction-finder, which is appropriate choose from the condition:

Moreover, taking into account (1) and (92) using the criterion of “three sigma” [4], the value of the maximum permissible error Δθ _max can be obtained by the formula:

According to the results of the verification of the conditions:

and conditions (48) select the rules for unambiguous determination of the azimuth θ and estimation of the angle β of the slope of the wave front of the radio signal source. So, in cases where r _ξ ≥ r _min or r _ξ <r _min and Δθ≤Δθ _max , the azimuth θ _φ is estimated and the azimuth θ is determined by formula (54) by averaging the results of the azimuth θ _R and θ _φ s taking into account the value of the parameter m, determined by the formula (53), which allows to reduce the methodological component of direction finding errors inherent in the formula (36) by half. If r _ξ <r _min and Δθ> Δθ _max , the azimuth estimate θ _φ obtained by formula (51) is biased, and therefore, the azimuth θ is determined in accordance with (55), only using the azimuth estimate θ _R. It should be noted that under the condition r _ξ <r _min , that is, in cases where the propagation line of an electromagnetic wave is close in angular orientation with a line passing through the center of the CAR and the phase center of one of the CAR antennas, the “separation” errors inherent in the formula (36) do not reach their maximum values.

In the case where r _ξ ≥ r _min , taking into account the relationships between the measured values of φ _R1 , φ _R2 , φ _R3 , on the one hand, and parameters b, λ, θ and β, on the other hand, described by expressions (89) and (90 ), we can obtain the following equality:

Using known trigonometric transformations and limiting the value of the function f (x) modulo and level in accordance with (59), which is necessary to obtain real values of the function arccos (X) in case of errors in determining φ _Ri , due to the influence of noise and interference, from (95) we obtain the formula (56) for estimating the angle β of the slope of the wave front of the radio signal source without methodological errors due to the electrodynamic interaction of the antennas and the mast device.

If r _ξ <r _min and Δθ≤Δθ _max , the azimuth estimates θ _R and θ _φ obtained by formulas (36) and (51) coincide, which confirms the validity of the assumption cosβ = 1 used in the definition of formula (51 ) taking into account (91), that is, the direction-finding signal propagates in the form of a surface electromagnetic wave at which β = 0, which corresponds to formula (57) for estimating the angle β of the slope of the wave front of the radio signal source.

(причем cosβ_sp<1), что соответствует формуле (58) оценивания угла β наклона фронта волны источника радиосигнала.And in the case where r _ξ <r _min and Δθ> Δθ _max , the azimuth estimates θ _R and θ _φ obtained by formulas (36) and (51) do not coincide, which indicates the illegality of the assumption cosβ = 1 used in the determination formulas (51) taking into account (91), that is, the direction-finding signal propagates in the form of a spatial electromagnetic wave, the slope of the front β _{sp of} which is not known and can be within

(and cosβ _sp <1), which corresponds to formula (58) for estimating the angle β of the slope of the wave front of the radio signal source.

The error Δθ of an unambiguous azimuth estimate, determined by formula (52), in the case of simultaneous reception of direction-finding and jamming radio signals, is characterized by an inverse proportion to the ratio of the direction-finding field strength to the field strength of the jamming radio signals, i.e., it is inversely proportional to the reliability of the direction finding results. It should be noted that in the vast majority of practically important cases, while receiving direction-finding and uncorrelated with it in phase and direction of propagation of interfering radio signals, condition (48) is fulfilled, which leads to the determination of θ and β using formulas (54) and (56), respectively, and the value of Δθ can be used to judge the reliability of the results of direction finding.

The combination of the obtained values of the error Δθ of the azimuth estimation and the aforementioned parameter μ, inversely proportional to the quality of the direction finding results, makes it possible to increase the probability of a correct judgment on the reliability of the results of determining the azimuth θ and the angle β of the slope of the wave front of the radio source.

The direction finder that implements the proposed method of direction finding (Fig.6) contains three antennas 1.1, 1.2 and 1.3 (identical non-directional axisymmetric vibrator type) forming an equidistant annular array in the direction-finding plane, and the geometric dimensions of the antennas 1.1, 1.2 and 1.3 along their symmetry axes are comparable with a wavelength of the radio signal, and the distance between the antennas 1.1, 1.2 and 1.3 does not exceed two-thirds of the wavelength of the radio signal. The device has three identical radio receiving units (BPM) 2.1, 2.2 and 2.3, made with a common local oscillator, three phase difference measurement units (BIRF) 3.1, 3.2 and 3.3, three differential signal generating units (BFRS) 4.1, 4.2 and 4.3, a generating unit the coefficients of non-uniformity of the amplitude antenna radiation patterns (BFKN) 5, comparator 6, the calculator of the uniqueness of the direction finding coefficient (VKOP) 7, the unit for the formation of unique amplitude values of the difference signals (BFOA) 8, the calculator of the noise threshold coefficient (VSHPK) 9, the unit for determining different the phases between the difference signals (BORF) 10, the amplitude azimuth calculator (ABA) 11, the quadrature component of the interfering signal (VKSP) 12, the phase azimuth calculator (FVA) 13, the azimuth estimation error calculator (VPOA) 14, the azimuth threshold coefficient calculator ( VAPK) 15, elevation angle calculator (VUM) 16, azimuth determination unit (BOA) 17, calculation parameters sensor (DPV) 18 and control signal generator (HUS) 19.

The outputs of the antennas 1.1, 1.2 and 1.3 are connected to the inputs of the corresponding RPM 2.1, 2.2 and 2.3. The pair of outputs of the first BPM 2.1 is connected respectively to the second pairs of inputs of the second BIRF 3.2 and BFRS 4.2 and the first pairs of inputs of the BFKN 5 and third BIRF 3.3 and BFRS 4.3. The pair of outputs of the second BPM 2.2 is connected respectively to the first pairs of inputs of the first BIRF 3.1 and BFRS 4.1 and the second pairs of inputs of the BFKN 5 and third BIRF 3.3 and BFRS 4.3. A pair of outputs of the third BPM 2.3 is connected respectively to the second pairs of inputs of the first BIRF 3.1 and BFRS 4.1, the third pair of inputs of the BFKN 5 and the first pairs of inputs of the second BIRF 3.2 and BFRS 4.2. The outputs of BIRF 3.1, BIRF 3.2 and BIRF 3.3 are connected respectively to the first, second and third inputs of BFOA 8. The first output of the first BFRS 4.1 is connected to the combined fourth input of BFOA 8 and the first inputs of the comparator 6 and VFSK 9. The first output of the second BFRS 4.2 is connected to the combined the fifth input of BFFA 8 and the second inputs of the comparator 6 and VSHPK 9. The first output of the third BFRS 4.3 is connected to the combined sixth input of the BFFA 8 and the third inputs of the comparator 6 and VSHPK 9. The second and third outputs of the BFRS 4.2 and 4.3 are connected respectively to the first, second, third , even The fourth, fifth, and sixth inputs of BORF 10. The first, second, and third outputs of comparator 6 are connected to the seventh, eighth, and ninth inputs of BFOA 8, respectively, and the first output of comparator 6 is also connected to the combined fourth input of VSHPK 9 and the first inputs of VKOP 7 and FVA 13. The first, second, third, fourth, fifth and sixth outputs of BFKN 5 are connected respectively to the fourth, fifth and sixth inputs of comparator 6 and the second, third and fourth inputs of VKOP 7. The output of VKOP 7 is connected to the combined tenth input of BFOA 8 and second entrance VA 13. The first, second and third outputs of BFOA 8 are connected respectively to the combined seventh input of BORF 10 and the first inputs of ABA 11 and VKSP 12, to the combined eighth input of BORF 10 and the second inputs of ABA 11 and VKSP 12 and to the combined ninth input of BORF 10 and the third inputs of ABA 11 and VKSP 12. The first, second and third outputs of BORF 10 are connected respectively to the combined third inputs of the FVA 13 and VUM 16, to the combined fourth inputs of the FVA 13 and VUM 16 and to the combined fifth inputs of the FVA 13 and VUM 16. The output of the VSHPK 9 is connected to the combined first input of the BOA 17 and the sixth input am FVA 13 and VUM 16. The output of FVA 13 is connected to the combined first input of VPOA 14 and the second input of BOA 17. The output of ABA 11 is connected to the combined second input of VPOA 14 and the third input of BOA 17. The output of VPOA 14 is connected to the first input of VAPK 15. The output VAKP 15 is connected to the combined seventh input of the VUM 16 and the fourth input of the BOA 17. The first output of the DPV 18 is connected to the combined eleventh input of the BFOA 8, the seventh input of the PVA 13 and the first input of the VUM 16. The second output of the DPV 18 is connected to the combined twelfth input of the BFOA 8, the eighth FVA 13 input and second VUM 16 input The third, fourth and fifth outputs of the DPV 18 are connected respectively to the fifth input of VKOP 7, the fifth input of VSHPK 9 and the second input of VAPK 15. The output of the HUS 19 is connected to the control inputs of the DPV 18 and RPB 2.1, 2.2 and 2.3. The outputs of BOA 17 and VUM 16 are the outputs of the azimuth θ and angle β of the wave slope of the wave source of the radio signal, respectively, and the outputs of the VKSP 12 and VPOA 14 are the outputs of the reliability parameters of the direction finding results μ and Δθ, respectively.

The direction finder (Fig.6), which implements the proposed method of direction finding, works as follows.

, поступившие на входы РПБ 2.1, 2.2 и 2.3, подвергаются типовым для современных радиоприемных блоков преобразованиям: синхронной фильтрации в полосе частот ΔF, усилению, преобразованию на промежуточную частоту с идентичными общими комплексными коэффициентами передачи, получаемыми либо с использованием идентичных РПБ, либо с использованием результатов калибровки неидентичности общих комплексных коэффициентов передачи РПБ, синхронному преобразованию сигналов промежуточной частоты в цифровые сигналы

путем дискретизации по уровню и времени принимаемых сигналов U_i(m_dΔt) и осуществления дискретного преобразования Фурье (в частности - быстрого преобразования Фурье (БПФ)) с получением действительной

и мнимой

составляющих сигналов, являющихся цифровыми эквивалентами принимаемых радиосигналов [11. Марпл мл. С.Л. Цифровой спектральный анализ и его приложения: Пер. с англ. - М.: Мир, 1990. - 584 с.]:The radio signals received by the identical antennas 1.1, 1.2 and 1.3, described by the above expressions (3), from their outputs go to the inputs of the corresponding RPMs 2.1, 2.2 and 2.3. On command from the output of the GUS 19, the signals

received at the inputs of the RPM 2.1, 2.2, and 2.3 are subjected to transformations typical of modern radio receiver units: synchronous filtering in the ΔF frequency band, amplification, conversion to an intermediate frequency with identical common complex transmission coefficients, obtained either using identical RPMs or using the results Calibration of the non-identity of the overall complex transmission coefficients of the BPM, the synchronous conversion of intermediate frequency signals into digital signals

by sampling by the level and time of the received signals U _i (m _d Δt) and performing a discrete Fourier transform (in particular, a fast Fourier transform (FFT)) to obtain a valid

and imaginary

component signals, which are digital equivalents of the received radio signals [11. Marple ml S.L. Digital spectral analysis and its applications: Per. from English - M .: Mir, 1990. - 584 p.]:

where Δt is the interval of uniform sampling of signals over time;

- объем обрабатываемого массива временных отсчетов радиосигналов;

- volume of the processed array of time samples of radio signals;

ΔT is the required (acceptable when filtering signals in the ΔF frequency band) time of observation of radio signals;

m _d = 0, 1, ... M-1 - serial numbers of time samples of radio signals;

k _ω = 0, 1, ... M-1 is the number of spectral components of the radio signals.

. В дальнейшем для упрощения обработки используют только М/2 комплексных спектральных составляющих, а остальные М/2 спектральных составляющих, соответствующих отрицательным частотам, полагают равными нулю. Таким образом, прием совокупности радиосигналов, отфильтрованных РПБ 2.1, 2.2 и 2.3 в широкой полосе частот ΔF с последующим разделением по частоте методом преобразования Фурье эквивалентен одновременному приему сигналов в соответствующих элементарных частотных каналах (ЭЧК) с шириной полосы

и общим количеством М/2. Указанное обстоятельство обеспечивает получение высокой чувствительности РПБ (за счет уменьшения действующего значения напряжения U_eff внутреннего шума, обратно пропорционального величине

) при одновременном увеличении быстродействия радиопеленгатора и повышении частотной избирательности РПБ. Необходимо отметить, что для реализации предложенного способа пеленгования возможно также осуществление поочередной синхронной фильтрации и преобразования сигналов в последовательных парах РПБ 2.1 и РПБ 2.2, РПБ 2.2 и РПБ 2.3 и, наконец, РПБ 2.3 и РПБ 2.1 последовательно в интервалах ΔT₁, ΔT₂ и ΔT₃ времени наблюдения сигналов соответственно. В этом случае совокупность трех РПБ 2.1, 2.2 и 2.3 может быть реализована с использованием соответствующей коммутации входов и выходов только двух физически существующих РПБ, однако, при этом минимально необходимое для осуществления пеленгования общее время ΔT_Σ наблюдения сигналов по сравнению с временем ΔT наблюдения сигналов при реализации способа пеленгования с синхронными фильтрацией и преобразованием сигналов одновременно в трех РПБ 2.1, 2.2 и 2.3 увеличивается в три раза: ΔT_Σ=ΔT₁+ΔT₂+ΔT₃=3ΔT.Moreover, after the Fourier transform is carried out according to the above expression, the received radio signals are represented as M complex spectral components, the frequency distance between which, that is, in fact the frequency resolution, is

. In the future, to simplify the processing, only M / 2 complex spectral components are used, and the remaining M / 2 spectral components corresponding to negative frequencies are assumed to be zero. Thus, the reception of a set of radio signals filtered by RPB 2.1, 2.2 and 2.3 in a wide frequency band ΔF, followed by frequency separation by the Fourier transform method, is equivalent to the simultaneous reception of signals in the corresponding elementary frequency channels (ECH) with a bandwidth

and the total number of M / 2. This circumstance ensures the high sensitivity of the RPM (by reducing the effective value of the voltage U _{eff of} internal noise inversely proportional to

) while increasing the speed of the direction finder and increasing the frequency selectivity of the RPM. It should be noted that for the implementation of the proposed method of direction finding, it is also possible to carry out alternate synchronous filtering and signal conversion in serial pairs of RPM 2.1 and RPM 2.2, RPM 2.2 and RPM 2.3 and, finally, RPM 2.3 and RPM 2.1 sequentially in the intervals ΔT ₁ , ΔT ₂ and ΔT ₃ signal observation time, respectively. In this case, the combination of three RPMs 2.1, 2.2, and 2.3 can be implemented using the corresponding switching inputs and outputs of only two physically existing RPMs, however, the minimum signal observation time ΔT _Σ required for direction finding as compared with the signal observation time ΔT at the implementation of the direction finding method with synchronous filtering and signal conversion simultaneously in three RPMs 2.1, 2.2 and 2.3 increases three times: ΔT _Σ = ΔT ₁ + ΔT ₂ + ΔT ₃ = 3ΔT.

и мнимые

спектральные составляющие сигналов с пары выходов РПБ 2.1 поступают одновременно на вторые пары входов БИРФ 3.2 и БФРС 4.2 и первые пары входов БФКН 5, БИРФ 3.3 и БФРС 4.3. Действительные

и мнимые

спектральные составляющие сигналов с пары выходов РПБ 2.2 поступают одновременно на первые пары входов БИРФ 3.1 и БФРС 4.1 и вторые пары входов БФКН 5, БИРФ 3.3 и БФРС 4.3. Действительные

и мнимые

спектральные составляющие сигналов с пары выходов РПБ 2.3 поступают одновременно на вторые пары входов БИРФ 3.1 и БФРС 4.1, третью пару входов БФКН 5 и первые входов БИРФ 3.2 и БФРС 4.2.Valid

and imaginary

the spectral components of the signals from the pair of outputs of the RPB 2.1 arrive simultaneously to the second pairs of inputs of the BIRF 3.2 and BFRS 4.2 and the first pairs of inputs of the BFKN 5, BIRF 3.3 and BFRS 4.3. Valid

and imaginary

the spectral components of the signals from the pair of outputs RPM 2.2 are fed simultaneously to the first pairs of inputs BIRF 3.1 and BFRS 4.1 and the second pairs of inputs BFKN 5, BIRF 3.3 and BFRS 4.3. Valid

and imaginary

the spectral components of the signals from the pair of outputs of RPM 2.3 arrive simultaneously to the second pair of inputs of BIRF 3.1 and BFRS 4.1, the third pair of inputs of BFKN 5 and the first inputs of BIRF 3.2 and BFRS 4.2.

In BIRF 3.1, 3.2 and 3.3, simultaneously or alternately by formula (30), the phase differences are determined, respectively, φ ₁ , φ ₂ and φ ₃ between pairs of signals simultaneously received respectively at their first and second pairs of inputs.

,

и

и их амплитудных значений r₁, r₂ и r₃ с использованием пар сигналов, одновременно поступившим соответственно на их первые и вторые пары входов.In BFRS 4.1, 4.2 and 4.3, at the same time or alternately according to formulas (34) and (35), the formation of difference signals is performed, respectively

,

and

and their amplitude values r ₁ , r ₂ and r ₃ using pairs of signals simultaneously received respectively at their first and second pairs of inputs.

In BFKN 5 according to formulas (37), (38) and (39), the amplitude values of u ₁ , u ₂ and u _{3 of the} signals received by antennas 1.1, 1.2 and 1.3 are determined and the coefficients P ₁ , P ₂ , P _{3 are formed} , K ₁ , K ₂ , K ₃ irregularities of the antenna patterns using pairs of signals simultaneously or alternately received at three pairs of inputs BFKN5.

и мнимым

спектральным составляющим разностных сигналов

(i=1, 2, 3) соответственно с вторых и третьих выходов БФРС 4.1, 4.2 и 4.3 поступают соответственно на первый, второй, третий, четвертый, пятый и шестой входы БОРФ 10.The signals corresponding to the measured phase differences φ ₁ , φ ₂ and φ ₃ from the outputs of BIRF 3.1, 3.2 and 3.3, respectively, are supplied to the first, second and third inputs of BFOA 8. The signals corresponding to the generated amplitude values of r ₁ , r ₂ and r ₃ difference signals from the first outputs of the BFRS 4.1, 4.2 and 4.3 are received respectively at the combined fourth input of the BFOA 8 and the first inputs of the comparator 6 and VSHPK 9, at the combined fifth input of the BFOA 8 and the second inputs of the comparator 6 and VSHPK 9 and at the combined sixth input of the BFOA 8 and the third inputs of the comparator 6 and VSHPK 9. And the signals corresponding real

and imaginary

spectral components of difference signals

(i = 1, 2, 3) respectively from the second and third outputs of the BFRS 4.1, 4.2 and 4.3 are received respectively at the first, second, third, fourth, fifth and sixth inputs of BORF 10.

The signals corresponding to the generated coefficients P ₁ , P ₂ and P _{3 of the} irregularity of the antenna patterns from the first, second and third outputs of the BFKN 5 are received respectively at the fourth, fifth and sixth inputs of the comparator 6, where of the three indices 1,2 or 3 are ordinal of antenna numbers, one value of ξ, one value of γ, and one value of ν, which are not equal to each other, are selected for which condition (40) is satisfied. The signal corresponding to the selected value ξ of the index number of the antenna, from the first output of the comparator 6 is fed to the combined seventh input of the BFOA 8, the fourth input of the VSHPK 9 and the first inputs of VKOP 7 and FVA 13, and the signals corresponding to the selected values of γ and ν of the index numbers of the antennas, s the second and third outputs of the comparator 6 are supplied respectively to the eighth and ninth inputs of the BFOA 8.

The signals corresponding to the generated coefficients K ₁ , K ₂ and K _{3 of the} irregularity of the antenna patterns from the fourth, fifth and sixth outputs of the BFKN 5 are supplied respectively to the second, third and fourth inputs of VKOP 7, the fifth input of which from the third output of the DPV 18 receives a signal corresponding to the a priori known average value of K _mid the unevenness of the radiation patterns of the antennas in the array. In VKOP 7, using the formula (41), the determination of the coefficient p of the uniqueness of direction finding is carried out. The signal corresponding to the value of the direction finding coefficient p from the output of VKOP 7 is fed to the combined tenth input of the BFOA 8 and the second input of the PVA 13. The signal corresponding to the value of the distance b between the antennas from the first output of the DPV 18 is fed to the combined eleventh input of the BFOA 8, the seventh input FVA 13 and the first input of VUM 16.

By the command from the output of the ГУС 19 to the control input of the ДПВ 18, a signal corresponding to the wavelength λ of the radio signal is generated in the ДПВ 18, which from the second output of the ДПВ 18 is fed to the combined twelfth input of the BFOA 8, the eighth input of the FVA 13 and the second input of the VUM 16.

In BFOA 8, three unambiguous amplitude values of the difference signals R ₁ , R ₂ and R ₃ are generated by the formula (42), if b≤0,3λ, or by the formula (44), if b> 0,3λ. The signals corresponding to the three unambiguous amplitude values of the difference signals R ₁ R ₂ and R ₃ generated from the first, second and third outputs of BFOA 8 are received respectively at the combined seventh input of BORF 10 and the first inputs of ABA 11 and VKSP 12, at the combined eighth input of BORF 10 and the second inputs of ABA 11 and VKSP 12 and the combined ninth input of BORF 10 and the third inputs of ABA 11 and VKSP 12.

In BORF 10, using the formula (46), the phase differences φ _R1 , φ _R2 and φ _R3 between the difference signals are determined, and the signals corresponding to the generated phase differences φ _R1 , φ _R2 and φ _R3 from the first, second and third outputs of BORF 10 are received, respectively to the combined third, to the combined fourth and combined fifth inputs of FVA 13 and VUM 16.

The signal corresponding to the a priori known minimum amplitude r _{min of} difference signals from the fourth output of the DPA 18 is fed to the fifth input of the VSHPK 9, where according to formula (48), the condition for exceeding the minimum amplitude r _{ξ of} difference signals relative to the a priori known minimum amplitude r _{min of} difference signals is checked . The signal corresponding to the result of checking the fulfillment of condition (48) in the form of a logical unit, if condition (48) is satisfied, or a logical zero, if condition (48) is not fulfilled, from the output of VSHPK 9 is fed to the combined sixth inputs of FVA 13 and VUM 16 and the first entrance BOA 17.

In FVA 13, the azimuth θ _{φ of the} radio signal is determined using the phase differences φ _R1 , φ _R2 and φ _R3 between the difference signals according to formula (50) if a logical unit signal is received at the sixth input of FVA 13, or according to formula (51) in the event that the sixth input of the FVA 13 receives a logic zero signal.

The signal corresponding to the azimuth θ of the radio source from the output of the FVA 13 is fed to the combined first input VPOA 14 and the second input BOA 17.

In ABA 11, the azimuth θ _{R of the} radio signal is determined using the unique amplitude values of the difference signals R ₁ , R ₂ and R ₃ according to the formula (36). The signal corresponding to the azimuth θ _{R of the} radio signal source, from the output of ABA 11 is fed to the combined second input of VPOA 14 and the third input of BOA 17.

In VPOA 14 by the formula (52), the error Δθ of the azimuth estimate is determined. The signal corresponding to the error Δθ of the azimuth estimation, from the output of the VPOA 14 is fed to the first input of the VAPK 15, the second input of which from the fifth output of the DPA 18 receives a signal corresponding to the a priori known value of the maximum permissible error Δθ _{max of} determining the azimuth.

In VAPK 15, a signal is generated in the form of a logical unit if Δθ≤Δθ _max , or a signal in the form of a logical zero if Δθ> Δθ _max , which from the output of VAPK 15 is fed to the combined seventh input of the VUM 16 and the fourth input of the BOA 17 .

In BOA 17, the azimuth θ of a radio signal source is determined by formula (54) if at least one of its first or fourth inputs receives a signal of a logical unit, or by formula (55) if its first and second inputs are logical zero signals.

In VUM 16, the angle β of the slope of the wave front of the radio signal source is estimated by formula (56) in the case where a logical unit signal is received at its sixth input, by formula (57) if a logical zero signal is received at its sixth input, and at the seventh input - a signal of a logical unit, or according to formula (58) in the event that logic six signals are received at its sixth and seventh inputs.

In VKSP 12, using the formula (47), the value of the parameter μ characterizing the presence of the quadrature component of the interfering signal is determined.

The signals from the inputs of the BOA 17 and VUM 16 correspond to the values of the azimuth θ and angle β of the slope of the wave front of the radio signal received by the direction finder, and the signals from the outputs of the VKSP 12 and VPOA 14 correspond to the values of the parameter μ and the errors Δθ of determining the azimuth of the radio signal, according to which the reliability of the results of direction finding sources of radio emission.

In the direction finder that implements the proposed method of direction finding, the known typical blocks for radio receivers are used, various embodiments of which are described in a number of scientific and technical information sources. Specific functional diagrams of individual blocks may differ in functional diagrams of their execution, structural and elemental bases, relationships between functional elements, however, the generalized functional diagram (Fig. 6), which describes the claimed device by an independent claim, is preserved.

, где l_a - длина "плеча" для симметричных антенн, размещаемых на мачтовом устройстве, и общая длина для несимметричных антенн вибраторного типа, размещаемых на плоской подстилающей поверхности. При вышеупомянутом отношении

и выборе отношения диаметра d_a осесимметричных антенн вибраторного типа к геометрическому размеру 2l_a антенн по их осям симметрии в пределах

рекомендуемые средние значения К_mid диаграмм направленности антенн в составе решетки выбираются в пределах 0,8≤К_mid≤1,1 соответственно, которые для дополнительного улучшения качества функционирования радиопеленгатора могут быть уточнены в пределах (5÷10)% в зависимости от отношения высоты мачтового устройства к длине волны λ радиосигналов, от отношения

и от характеристик конкретных известных типовых схем согласующе-симметрирующих устройств, применяемых в антенной технике.The ratio of the geometric dimensions 2l _a along the axis of symmetry to the distance b between the antennas in a flat equidistant three-element CAR is recommended for the design of axisymmetric antennas 1.1, 1.2 and 1.3 of the vibrator type.

where l _a is the length of the "shoulder" for symmetrical antennas placed on the mast device, and the total length for asymmetric antennas of a vibrator type placed on a flat underlying surface. With the above relation

and choosing the ratio of the diameter d _{a of the} axisymmetric antennas of the vibrator type to the geometric size 2l _{a of the} antennas along their symmetry axes within

recommended average values of K _mid radiation patterns of the antennas in the array are selected in the range of 0.8 ≤ K _mid ≤ 1.1, respectively, which, to further improve the quality of the operation of the direction finder, can be refined within (5 ÷ 10)% depending on the ratio of the mast height devices to the wavelength λ of radio signals, from the ratio

and from the characteristics of specific known typical patterns of matching-balancing devices used in antenna technology.

Functional diagrams of RPB 2.1, 2.2 and 2.3, BIRF 3.1, 3.2 and 3.3, BFRS 4.1, 4.2 and 4.3, GUS 19 and DPV 18, the relationships between the above blocks and the timing diagrams of their operation do not differ from the diagrams, relationships and timing diagrams of similar blocks the closest analogue and other aforementioned analogs.

причем значения λ должны удовлетворять условию λ≤1,5b. На третьем выходе ДПВ 18 формируется сигнал, соответствующий вышеупомянутому рекомендуемому среднему значению К_mid. На четвертом выходе ДПВ 18 формируется сигнал, соответствующий минимальной амплитуде r_min, определяемой по формуле (49). При этом, с учетом (2) и (14), в зависимости от отношения b/λ, изменяющегося, например, в пределах

при случайной СКО σ_θr=1° значение q_min выбирается в пределах 40≤q_min≤10 соответственно, а при σ_θr=2° - в пределах 20≤q_min≤5 соответственно. Действующее значение напряжения внутреннего шума U_eff каналов формирования разностных сигналов, как отмечалось ранее, зависит от технической реализации РПБ 2.1, 2.2 и 2.3 и определяется с учетом известной формулы Найквиста для одного радиоприемного канала и с учетом увеличения в

раз напряжения внутреннего шума из-за использования для формирования разностных сигналов двух радиоприемных каналов, шумы в которых некоррелированы, в соответствии с выражением:At the same time, at the first output of the DPA 18, a signal is formed corresponding to the distance b between the antennas 1.1, 1.2 and 1.3 of the equidistant three-element CAR. At the second output of the DPA 18, depending on the frequency f, which the RPM 2.1, 2.2 and 2.3 are tuned to, a signal corresponding to the wavelength is generated by a command from the output of the GUS 19 to the control input of the DPA 18

moreover, the values of λ must satisfy the condition λ≤1.5b. At the third output of the DPA 18, a signal is generated corresponding to the aforementioned recommended average value of K _mid . At the fourth output of the DPA 18, a signal is generated corresponding to the minimum amplitude r _min determined by the formula (49). Moreover, taking into account (2) and (14), depending on the ratio b / λ, varying, for example, within

for random _standard deviations, σ _θr = 1 °, the q _min value is selected within 40≤q _min ≤10, respectively, and for σ _θr = 2 °, it is selected within 20≤q _min ≤5, respectively. The effective value of the internal noise voltage U _eff of the difference signal generation channels, as noted earlier, depends on the technical implementation of the RPM 2.1, 2.2, and 2.3 and is determined taking into account the well-known Nyquist formula for one radio receiving channel and taking into account the increase in

times the voltage of internal noise due to the use for the formation of differential signals of two radio receiving channels in which the noise is uncorrelated, in accordance with the expression:

where k ₀ = 1.38 · 10 ^-23 W / (Hz · K) is the Boltzmann constant;

T ₀ = 288 K - standard absolute temperature;

N _W - noise figure RPB 2.1,2.2 and 2.3;

R _I - input impedance RPB 2.1, 2.2 and 2.3;

Δf is the band of the elementary frequency channel RPB 2.1, 2.2 and 2.3.

So, with typical values of radio receivers N _w = 10, R _in = 50 Ohm and Δf = 2 kHz from (97) we get:

, изменяющегося, в частности, в пределах

рекомендуемые значения r_min составляют:In view of the foregoing, for example, at σ _θr = 1 ° depending on the ratio

varying, in particular, within

Recommended r _min values are:

At the fifth output of the DPA 18, a signal is generated corresponding to the a priori known value of the maximum permissible error Δθ _{max for} determining the azimuth, the recommended value in accordance with (93) is (2 ÷ 2.5) °.

пары входов которых являются соответственно первой (Вх.11 и Вх.12), второй (Вх.21 и Вх.22) и третьей (Вх.31 и Вх.32) парами входов БФКН 5, три элемента сравнения (ЭС) 21.1, 21.2 и 21.3, три умножителя 22.1, 22.2 и 22.3, три сумматора 23.1.23.2 и 23.3, три коммутатора 24.1, 24.2 и 24.3, шесть делителей 25.1. 25.2, 25.3, 25.4, 25.5 и 25.6, выходы которых являются соответственно первым (Вых.1), вторым (Вых.2), третьим (Вых.3), четвертым (Вых.4), пятым (Вых.5) и шестым (Вых.6) выходами БФКН 5, и датчик 26 числа 2. Причем выход ФП 20.1 подсоединен к объединенным первым входам коммутатора 24.3, ЭС 21.3, умножителя 22.1 и сумматора 23.3 и вторым входам коммутатора 24.2, ЭС 21.2 и сумматора 23.2. Выход ФП 20.2 подсоединен к объединенным первым входам коммутатора 24.1, ЭС 21.1, сумматора 23.1 и умножителя 22.2 и вторым входам коммутатора 24.3, ЭС 21.3 и сумматора 23.3. Выход ФП 20.3 подсоединен к объединенным первым входам коммутатора 24.2, ЭС 21.2, сумматора 23.2 и умножителя 22.3 и вторым входам коммутатора 24.1, ЭС 21.1 и сумматора 23.1. Выходы ЭС 21.1, 21.2 и 21.3 подсоединены к управляющим входам коммутаторов 24.1, 24.2 и 24.3 соответственно, пары выходов которых подсоединены соответственно к парам входов делителей 25.1, 25.2 и 25.3. Выход датчика 26 подсоединен к объединенным вторым входам умножителей 22.1, 22.2 и 22.3, выходы которых подсоединены к первым входам делителей 25.4, 25.5 и 25.6 соответственно, вторые входы которых подсоединены к выходам сумматоров 23.1, 23.2 и 23.3 соответственно. Необходимо отметить, что параметрами X1 и X2 обозначена пара сигналов, поступающих на соответствующую пару входов ФП 20.1, 20.2 и 20.3 вида

Functional diagram of an embodiment of BFKN 5 (Fig. 7) contains three functional converters (FP) 20.1, 20.2 and 20.3 of the form

pairs of inputs of which are respectively the first (Vkh.11 and Vkh.12), the second (Vkh.21 and Vkh.22) and the third (Vkh.31 and Vkh.32) pairs of inputs BFKN 5, three elements of comparison (ES) 21.1, 21.2 and 21.3, three multipliers 22.1, 22.2 and 22.3, three adders 23.1.23.2 and 23.3, three switches 24.1, 24.2 and 24.3, six dividers 25.1. 25.2, 25.3, 25.4, 25.5 and 25.6, the outputs of which are respectively the first (Out.1), second (Out.2), third (Out.3), fourth (Out.4), fifth (Out.5) and sixth (Output 6) the outputs of BFKN 5, and the sensor 26 is number 2. Moreover, the output of the FP 20.1 is connected to the combined first inputs of the switch 24.3, ES 21.3, multiplier 22.1 and adder 23.3 and second inputs of the switch 24.2, ES 21.2 and adder 23.2. The output of the FP 20.2 is connected to the combined first inputs of the switch 24.1, ES 21.1, the adder 23.1 and the multiplier 22.2 and the second inputs of the switch 24.3, ES 21.3 and the adder 23.3. The output of the FP 20.3 is connected to the combined first inputs of the switch 24.2, ES 21.2, the adder 23.2 and the multiplier 22.3 and the second inputs of the switch 24.1, ES 21.1 and the adder 23.1. The outputs of ES 21.1, 21.2 and 21.3 are connected to the control inputs of the switches 24.1, 24.2 and 24.3, respectively, the pairs of outputs of which are connected respectively to the pairs of inputs of the dividers 25.1, 25.2 and 25.3. The output of the sensor 26 is connected to the combined second inputs of the multipliers 22.1, 22.2 and 22.3, the outputs of which are connected to the first inputs of the dividers 25.4, 25.5 and 25.6, respectively, the second inputs of which are connected to the outputs of the adders 23.1, 23.2 and 23.3, respectively. It should be noted that the parameters X1 and X2 indicate a pair of signals arriving at the corresponding pair of inputs of the FP 20.1, 20.2 and 20.3 of the form

и мнимой

составляющим первого (i=1), второго (i=2) и третьего (i=3) сигналов

и

с пар входов Вх.11 и Вх.12, Вх.21 и Вх.22, Вх.31 и Вх.32 БФКН 5 поступают на пары входов ФП 20.1, 20.2 и 20.3 соответственно, где производится формирование сигналов, соответствующих их амплитудным значениям u₁, u₂ и u₃ соответственно. Сигнал с выхода ФП 20.1 поступает на объединенные первые входы коммутатора 24.3, ЭС 21.3, умножителя 22.1, сумматора 23.3 и вторые входы коммутатора 24.2, ЭС 21.2 и сумматора 23.2. Сигнал с выхода ФП 20.2 поступает на объединенные первые входы коммутатора 24.1. ЭС 21.1, сумматора 23.1, умножителя 22.2 и вторые входы коммутатора 24.3, ЭС 21.3 и сумматора 23.3. Сигнал с выхода ФП 20.3 поступает на объединенные первые входы коммутатора 24.2, ЭС 21.2, сумматора 23.2, умножителя 22.3 и вторые входы коммутатора 24.1, ЭС 21.1 и сумматора 23.1. В ЭС 21.1, 21.2 и 21.3 производится сравнение амплитуд сигналов, поступивших на их пары входов, и в случаях, если амплитуды сигналов, поступивших на их первые входы, не меньше амплитуд сигналов, поступивших на их вторые входы, на выходах соответствующих ЭС формируются сигналы логических единиц, а в противном случае - сигналы логических нулей. Сигналы с выходов ЭС 21.1. 21.2 и 21.3 поступают на управляющие входы соответственно коммутаторов 24.1. 24.2 и 24.3, в каждом из которых производится соединение первого и второго входа соответственно с первым и вторым выходом в случае, если на его управляющий вход поступает сигнал логической единицы, или соединение первого и второго входа соответственно с вторым и первым выходом в случае, если на его управляющий вход поступает сигнал логического нуля. Сигналы, поступившие на пары входов коммутаторов 24.1, 24.2 и 24.3, с их пар выходов поступают на пары входов делителей 25.1, 25.2 и 25.3 соответственно, в каждом из которых производится формирование сигнала, равного отношению амплитуды сигнала, поступившего на первый вход, к амплитуде сигнала, поступившего на его второй вход. Кроме того, сигнал с выхода датчика 26 поступает на объединенные вторые входы умножителей 22.1. 22.2 и 22.3, в каждом из которых формируется сигнал, соответствующий произведению сигналов, поступивших на их пары входов. Сигналы с выходов умножителей 22.1, 22.2 и 22.3 поступают на первые входы делителей 25.4. 25.5 и 25.6. В сумматорах 23.1. 23.2 и 23.3 производится суммирование сигналов, поступивших на их пары входов и, далее, просуммированные сигналы с выходов сумматоров 23.1, 23.2 и 23.3 поступают на вторые входы соответственно делителей 25.4, 25.5 и 25.6, в каждом из которых производится формирование сигнала, равного отношению амплитуды сигнала, поступившего на первый вход, к амплитуде сигнала, поступившего на его второй вход. На выходах делителей 25.1, 25.2, 25.3, 25.4, 25.5 и 25.6, являющихся первым (Вых.1), вторым (Вых.2), третьим (Вых.3), четвертым (Вых.4), пятым (Вых.5) и шестым (Вых.6) выходами БФКН 5, формируются сигналы, соответствующие коэффициентам P₁, P₁, Р₃, K₁, K₂ и К₃ неравномерности диаграмм направленности антенн в составе решетки соответственно.BFKN 5 works as follows. Signals corresponding to the actual

and imaginary

components of the first (i = 1), second (i = 2) and third (i = 3) signals

and

from pairs of inputs Vkh.11 and Vkh.12, Vkh.21 and Vkh.22, Vkh.31 and Vkh.32 BFKN 5 arrive at pairs of inputs FP 20.1, 20.2 and 20.3 respectively, where the formation of signals corresponding to their amplitude values u ₁ , u ₂ and u _3, respectively. The signal from the output of the FP 20.1 is fed to the combined first inputs of the switch 24.3, ES 21.3, the multiplier 22.1, the adder 23.3 and the second inputs of the switch 24.2, ES 21.2 and the adder 23.2. The signal from the output of the FP 20.2 is fed to the combined first inputs of the switch 24.1. ES 21.1, adder 23.1, multiplier 22.2 and second inputs of the switch 24.3, ES 21.3 and adder 23.3. The signal from the output of the FP 20.3 is fed to the combined first inputs of the switch 24.2, ES 21.2, the adder 23.2, the multiplier 22.3 and the second inputs of the switch 24.1, ES 21.1 and the adder 23.1. In ES 21.1, 21.2 and 21.3, the amplitudes of the signals received at their input pairs are compared, and in cases where the amplitudes of the signals received at their first inputs are not less than the amplitudes of the signals received at their second inputs, logical signals are generated at the outputs of the corresponding ES units, and otherwise - signals of logical zeros. Signals from the outputs of ES 21.1. 21.2 and 21.3 arrive at the control inputs of the switches 24.1, respectively. 24.2 and 24.3, in each of which the connection of the first and second inputs, respectively, with the first and second output, if a logical unit signal is received at its control input, or the connection of the first and second inputs, respectively, with the second and first output, if its control input receives a logic zero signal. The signals received at the pairs of inputs of the switches 24.1, 24.2 and 24.3, from their pairs of outputs are fed to the pairs of inputs of the dividers 25.1, 25.2 and 25.3, respectively, in each of which a signal is generated that is equal to the ratio of the amplitude of the signal received at the first input to the signal amplitude entering his second entrance. In addition, the signal from the output of the sensor 26 is fed to the combined second inputs of the multipliers 22.1. 22.2 and 22.3, in each of which a signal is formed corresponding to the product of the signals received at their input pairs. The signals from the outputs of the multipliers 22.1, 22.2 and 22.3 are fed to the first inputs of the dividers 25.4. 25.5 and 25.6. In the adders 23.1. 23.2 and 23.3, the signals received at their input pairs are summed up, and further, the summed signals from the outputs of the adders 23.1, 23.2 and 23.3 are fed to the second inputs of the dividers 25.4, 25.5 and 25.6, respectively, in each of which a signal is formed equal to the signal amplitude ratio received at the first input to the amplitude of the signal received at its second input. At the outputs of the dividers 25.1, 25.2, 25.3, 25.4, 25.5 and 25.6, which are the first (Out.1), second (Out.2), third (Out.3), fourth (Out.4), fifth (Out.5) and the sixth (Out.6) outputs of the BFCH 5, signals are generated corresponding to the coefficients P ₁ , P ₁ , P ₃ , K ₁ , K ₂ and K _{3 of the} irregularity of the antenna patterns in the array, respectively.

The functional diagram of an embodiment of the comparator 6 (Fig. 8) contains three multipliers 27.1, 27.2 and 27.3, the first and second inputs of which are respectively the first (Bx.1) and fourth (Bx.4), second (Bx.2) and fifth ( Vkh.5), the third (Vkh.3) and the sixth (Vkh.6) inputs of the comparator 6, three sensors 28, 29 and 30 of numbers 1.2 and 3 respectively, three elements of comparison (ES) 31.1. 31.2 and 31.3, three switches 32.1, 32.2 and 32.3, six elements "And" (EI) 33.1, 33.2, 33.3, 33.4, 33.5 and 33.6, and the combined outputs of EI 33.1, 33.2 and 33.3 and the combined outputs of EI 33.4, 33.5 and 33.6 are respectively the first (Out.1) and third (Out.3) outputs of the comparator 6, six elements "NOT" (ENE) 34.1, 34.2, 34.3, 34.4, 34.5 and 34.6 and three additional elements "And" (EI) 35.1, 35.2 and 35.3, the combined outputs of which are the second output (Out.2) of the comparator 6. Moreover, the outputs of the multipliers 27.1. 27.2 and 27.3 are connected respectively to the combined first inputs of ES 31.1 and 31.2, to the combined second input of ES 31.1 and the first input of ES 31.3 and to the combined second inputs of ES 31.2 and ES 31.3. The output of the sensor 28 is connected to the combined first inputs of the switches 32.1 and 32.2 and ENE 34.1 and 34.2. The output of the sensor 29 is connected to the combined second input of the switch 32.1 and the first inputs of the switch 32.3 and ENE 34.3 and 34.4. The output of the sensor 30 is connected to the combined first inputs of the ENE 34.5 and 34.6 and the second inputs of the switches 32.2 and 32.3. The outputs of the ES 31.1, 31.2 and 31.3 are connected to the control inputs of the switches 32.1, 32.2 and 32.3, respectively. The first and second outputs of the switch 32.1 are connected respectively to the combined first inputs of the EI 33.1 and 33.2 and to the combined first inputs of the EI 33.4 and 33.5. The first and second outputs of the switch 32.2 are connected respectively to the combined second input of the EM 33.1 and the first input of the EM 33.3 and to the combined second input of the EM 33.4 and the first input of the EM 33.6. The first and second outputs of the switch 32.2 are connected respectively to the combined second inputs of the EI 33.2 and 33.3 and to the combined second inputs of the EI 33.5 and 33.6. In addition, the combined outputs of EI 33.4, 33.5 and 33.6 are connected to the combined second inputs of ENE 34.2, 34.4 and 34.6. And finally, the outputs of the EHE 34.1 and 34.2, 34.3 and 34.4, 34.5 and 34.6 are connected respectively to the pairs of inputs of the EI 35.1, 35.2 and 35.3.

Comparator 6 operates as follows. Signals corresponding to the amplitudes r ₁ , r ₂ and r _{3 of the} difference signals and the coefficients P ₁ , P ₂ and P _{3 of the} irregularity of the radiation patterns of the antennas, as part of the array from the inputs Вх.1, Вх.2, Вх.3, Вх.4, Input 5 and input 6 of the comparator 6 are received respectively at the first inputs of the multipliers 27.1, 27.2 and 27.3 and the second inputs of the multipliers 27.1. 27.2 and 27.3, in which the multiplication of the signals received at their input pairs is performed. Signals from the outputs of the multipliers 27.1. 27.2 and 27.3 arrive respectively at the combined first inputs of ES 31.1 and 31.2, at the combined second input of ES 31.1 and the first input of ES 31.3, and at the combined second inputs of ES 31.2 and ES 31.3. In each of the ES 31.1, 31.2 and 31.3, a signal of a logical unit is formed if the amplitude of the signal received at its first input does not exceed the amplitude of the signal received at its second input, or a logic zero signal otherwise. The signals from the outputs of the ES 31.1, 31.2 and 31.3 arrive at the control inputs of the switches 32.1. 32.2 and 32.3 respectively. The signal corresponding to the number 1 from the output of the sensor 28 is fed to the combined first inputs of the switches 32.1 and 32.2 and ENE 34.1 and 34.2. The signal corresponding to the number 2 from the output of the sensor 29 is fed to the combined second input of the switch 32.1 and the first inputs of the switch 32.3 and ENE 34.3 and 34.4. The signal corresponding to the number 3 from the output of the sensor 30 is fed to the combined first inputs of the ENE 34.5 and 34.6 and the second inputs of the switches 32.2 and 32.3. In each of the switches 32.1, 32.2 and 32.3, if a logical unit signal is received at its control input, its first and second inputs are connected to the first and second outputs, respectively. Further, the signals from the first and second outputs of the switch 32.1 are received respectively at the combined first inputs of the EI 33.1 and 33.2 and at the combined first inputs of the EI 33.4 and 33.5, the signals from the first and second outputs of the switch 32.2 are received respectively at the combined second input of the EI 33.1 and the first input of EI 33.3 and to the combined second input of the EM 33.4 and the first input of the EM 33.6, and the signals from the first and second outputs of the switch 32.3 are respectively fed to the combined second inputs of the EM 33.2 and 33.3 and the combined second inputs of the EM 33.5 and 33.6. In each of the EI 33.1, 33.2. 33.3, 33.4, 33.5 and 33.6, one of its inputs (for example, the first) is connected to the output if the signals received at its inputs are equal to each other, otherwise, none of the inputs is connected to its output. The signal from the output of one of the three EIs 33.1, 33.2 and 33.3, corresponding to the selected value ξ of the antenna serial number, is fed to the first output (Output 1) of the comparator 6 and simultaneously to the combined second inputs of the ENE 34.1. 34.3 and 34.5. The signal from the output of one of the EI 33.4. 33.5 or 33.6, corresponding to the selected value ν of the serial number of the antennas, is fed to the third output (Out.3) of the comparator 6 and simultaneously to the combined second inputs of ENE 34.2, 34.4 and 34.6. In each of EHE 34.1, 34.2, 34.3, 34.4, 34.5 and 34.6, the first of its inputs is connected to the output if the signals received at its inputs are not equal to each other, otherwise, a signal is generated at the output that does not correspond None of the antenna index numbers (for example, a signal corresponding to zero). Signals from the outputs of ENE 34.1, 34.2, 34.3. 34.4, 34.5 and 34.6 respectively enter the first and second inputs of EI 35.1, 35.2 and 35.3, the principle of operation of which coincides with the principle of functioning of EI 33.1, 33.2, 33.3, 33.4, 33.5, 33.6. The signal from the output of one of the three EIs 35.1, 35.2 or 35.3, corresponding to the selected value γ of the serial number of the antennas, is fed to the second output (Out.2) of the comparator 6.

Functional diagram of a variant of the implementation of VKOP 7 (Fig. 9) contains a switch 36, a control input, the first, second and third inputs of which are respectively the first (Vkh.1), the second (Vkh.2), the third (Vkh.3) and the fourth ( Input 4) the inputs of VKOP 7, and a series-connected subtractor 37 and functional converter (FP) 38 of the form sgn (X), and the output of the switch 36 is connected to the first input of the subtractor 37, the second input of which is the fifth input (Input 5) VKOP 7 , and the output of the FP 38 is the output (Out.) VKOP 7.

VKOP 7 works as follows. The signals corresponding to the selected value ξ of the index number of the antenna, the coefficients K ₁ , K ₂ and K _{3 of the} irregularity of the antenna patterns and the average value of K _{mid of} the uneven coefficients of the antenna patterns, from inputs Bx.1, Bx.2, Bx.3, Bx. 4 and Vkh.5 VKOP 7 respectively received at the control, first, second and third inputs of the switch 36 and the second input of the subtractor 37. In the switch on the signal received at its control input, one of the inputs is connected, the value of the serial number of which coincides with the value ξ of the index number of the antenna to its output. The signal from the output of the switch 36 is fed to the first input of the subtractor 37, where the difference between the amplitudes of the signals received respectively at its first and second inputs is determined. The signal from the output of the subtractor 37 is fed to the input of the FI 38, where the signal corresponding to +1 is generated if the signal received at the input of the FI 38 does not have a negative sign, or equal to -1 otherwise. The signal from the output of the FP 38, corresponding to the value of the direction finding coefficient p, is fed to the output (Out) of the VKOP 7.

The functional diagram of the embodiment of VSHPK 9 (FIG. 10) contains a switch 39, the principle of operation of which coincides with the principle of operation of the switch 36, and the first, second, third and control inputs are respectively the first (Vkh.1), the second (Vkh.2), the third (Vkh.3) and the fourth (Vkh.4) inputs of the VSHPK 9, and the subtractor 40 and the control signal shaper (FCS) 41 connected in series, the output of the switch 39 being connected to the first input of the subtractor 40, the second input of which is the fifth input (Bx .5) VSHPK 9, and the output of FUS 41 is in travel (O). VSHPK 9.

VSHPK 9 works as follows. The signals corresponding to the amplitudes r ₁ , r ₂ and r _{3 of the} difference signals, the selected value ξ of the index number of the antenna and the minimum amplitude r _{min of the} difference signals, from inputs Input 1, Input 2, Input 3, Input 4 and Input 5 VSHPK 9 respectively received at the first, second, third and control inputs of the switch 39 and the second input of the subtractor 40. In the switch on the signal received at its control input, one of the inputs is connected, the value of the serial number of which coincides with the value ξ of the index number of the antenna, to his exit. The signal from the output of the switch 39 is fed to the first input of the subtractor 40, where the difference between the amplitudes of the signals received respectively at its first and second inputs is determined. The signal from the output of the subtractor 40 is fed to the input of the FSF 41, where the signal corresponding to the logical unit is generated if the signal received at the input of the FSF 41 does not have a negative sign, or the signal corresponding to logical zero, otherwise. The signal from the output of the FUS 41, corresponding to a logical unit or zero, is fed to the output (Out) of VSHPK 9.

The functional diagram of an embodiment of BORF 10 (Fig. 11) contains three AF 42.1, 42.2 and 42.3 of the form sgn (X), six multipliers 43.1, 43.2, 43.3, 43.4, 43.5 and 43.6 and three phase difference calculators (VRF) 44.1, 44.2 and 44.3, the outputs of which are respectively the first (Out.1), second (Out.2) and third (Out.3) outputs of BORF 10, and the first inputs of the multipliers 43.1, 43.2, 43.3, 43.4, 43.5 and 43.6 and the inputs of the FP 42.1, 42.2 and 42.3 are respectively the first (Vkh.1), the second (Vkh.2), the third (Vkh.3), the fourth (Vkh.4), the fifth (Vkh.5), the sixth (Vkh.6), the seventh (Vkh) .7), the eighth (Bx.8) and ninth (Bx.9) inputs of BORF 10. Moreover, the outputs of the FP 42.1, 42.2 and 42.3 are connected respectively, to the combined second inputs of the multipliers 43.1 and 43.2, to the combined second inputs of the multipliers 43.3 and 43.4 and to the combined second inputs of the multipliers 43.5 and 43.6. The output of the multiplier 43.1 is connected to the combined first inputs of the second pair of inputs of the VRF 44.2 and the first pair of inputs of the VRF 44.3. The output of the multiplier 43.2 is connected to the combined second inputs of the second pair of inputs of the VRF 44.2 and the first pair of inputs of the VRF 44.3. The output of the multiplier 43.3 is connected to the combined first inputs of the first pair of inputs of VRF 44.1 and the second pair of inputs of VRF 44.3. The output of the multiplier 43.4 is connected to the combined second inputs of the first pair of inputs of VRF 44.1 and the second pair of inputs of VRF 44.3. The output of the multiplier 43.5 is connected to the combined first inputs of the second pair of inputs of VRF 44.1 and the first pair of inputs of VRF 44.2. The output of the multiplier 43.6 is connected to the combined second inputs of the second pair of inputs of VRF 44.1 and the first pair of inputs of VRF 44.2.

и мнимой

составляющим первого (i=1), второго (i=2) и третьего (i=1) разностных сигналов

с входов Вх.1, Вх.2, Вх.3, Вх.4, Вх.5 и Вх.6 БОРФ 10 поступают на первые входы умножителей 43.1, 43.2, 43.3, 43.4, 43.5 и 43.6 соответственно, а сигналы, соответствующие однозначным амплитудным значениям R₁, R₂ и R₃ разностных сигналов, с входов Вх.7, Вх.3 и Вх.9 БОРФ 10 поступают соответственно на входы ФП 42.1, 42.2 и 42.3. В каждом из ФП 42.1, 42.2 и 42.3 производится формирование сигнала, соответствующего +1 в случае, если знак сигнала на его входе не отрицательный, или формирование сигнала, соответствующего -1 в противном случае. Сигналы с выходов ФП 42.1, 42.2 и 42.3 поступают соответственно на объединенные вторые входы умножителей 43.3 и 43.3 и на объединенные вторые входы умножителей 43.5 и 43.6. В каждом умножителе 43.1, 43.2, 43.3, 43.4, 43.5 и 43.6 производится перемножение сигналов, поступивших на их пары входов. Сигнал с выхода умножителя 43.1 поступает на объединенные первые входы второй пары входов ВРФ 44.2 и первой пары входов ВРФ 44.3. Сигнал с выхода умножителя 43.2 поступает на объединенные вторые входы второй пары входов ВРФ 44.2 и первой пары входов ВРФ 44.3. Сигнал с выхода умножителя 43.3 поступает на объединенные первые входы первой пары входов ВРФ 44.1 и второй пары входов ВРФ 44.3. Сигнал с выхода умножителя 43.4 поступает на объединенные вторые входы первой пары входов ВРФ 44.1 и второй пары входов ВРФ 44.3. Сигнал с выхода умножителя 43.5 поступает на объединенные первые входы второй пары входов ВРФ 44.1 и первой пары входов ВРФ 44.2. И, наконец, сигнал с выхода умножителя 43.6 поступает на объединенные вторые входы второй пары входов ВРФ 44.1 и первой пары входов ВРФ 44.2. В каждом из ВРФ 44.1, 44.2 и 44.3 производится определение разности фаз между сигналом, действительная и мнимая составляющие которого поступают соответственно на первый и второй входы первой пары входов, и сигналом, действительная и мнимая составляющие которого поступают соответственно на первый и второй входы второй пары входов. Сигналы с выходов ВРФ 44.1, 44.2 и 44.3, соответствующие разностям фаз φ_R1, φ_R2 и φ_R3 между разностными сигналами, поступают соответственно на первый (Вых.1), второй (Вых.2) и третий (Вых.3) БОРФ 10.BORF 10 works as follows. Signals corresponding to the actual

and imaginary

components of the first (i = 1), second (i = 2) and third (i = 1) difference signals

from inputs Input 1, Input 2, Input 3, Input 4, Input 5 and Input 6 BORF 10 are supplied to the first inputs of the multipliers 43.1, 43.2, 43.3, 43.4, 43.5 and 43.6, respectively, and the signals corresponding to unique the amplitude values R ₁ , R ₂ and R _{3 of the} difference signals from inputs Вх.7, Вх.3 and Вх.9 BORF 10 respectively received at the inputs of the FP 42.1, 42.2 and 42.3. In each of the FP 42.1, 42.2 and 42.3, a signal is generated corresponding to +1 if the signal sign at its input is not negative, or a signal corresponding to -1 is generated otherwise. The signals from the outputs of the FP 42.1, 42.2 and 42.3 are respectively supplied to the combined second inputs of the multipliers 43.3 and 43.3 and to the combined second inputs of the multipliers 43.5 and 43.6. In each multiplier 43.1, 43.2, 43.3, 43.4, 43.5 and 43.6, the signals received at their input pairs are multiplied. The signal from the output of the multiplier 43.1 is fed to the combined first inputs of the second pair of inputs of the VRF 44.2 and the first pair of inputs of the VRF 44.3. The signal from the output of the multiplier 43.2 is fed to the combined second inputs of the second pair of inputs of the VRF 44.2 and the first pair of inputs of the VRF 44.3. The signal from the output of the multiplier 43.3 goes to the combined first inputs of the first pair of inputs of the VRF 44.1 and the second pair of inputs of the VRF 44.3. The signal from the output of the multiplier 43.4 goes to the combined second inputs of the first pair of inputs of the VRF 44.1 and the second pair of inputs of the VRF 44.3. The signal from the output of the multiplier 43.5 is fed to the combined first inputs of the second pair of inputs of the VRF 44.1 and the first pair of inputs of the VRF 44.2. And finally, the signal from the output of the multiplier 43.6 goes to the combined second inputs of the second pair of inputs of the VRF 44.1 and the first pair of inputs of the VRF 44.2. In each of the VRF 44.1, 44.2, and 44.3, the phase difference is determined between the signal, the real and imaginary components of which arrive at the first and second inputs of the first pair of inputs, respectively, and the signal, the real and imaginary components of which arrive at the first and second inputs of the second pair of inputs, respectively . The signals from the outputs of the VRF 44.1, 44.2, and 44.3, corresponding to the phase differences φ _R1 , φ _R2, and φ _R3 between the difference signals, respectively arrive at the first (Out.1), second (Out.2) and third (Out.3) BORF 10 .

, причем объединенные первые входы умножителей 45.2 и 45.3 и объединенные первые входы умножителей 45.1 и 45.4 являются соответственно первым (Вх.11) и вторым (Вх.12) входами первой пары входов ВРФ 44.1, 44.2 и 44.3, объединенные вторые входы умножителей 45.1 и 45.3 и объединенные вторые входы умножителей 45.2 и 45.4 являются соответственно первым (Вх.21) и вторым (Вх.22) входами второй пары входов ВРФ 44.1, 44.2 и 44.3, а выход ФП 48 является выходом (Вых.) ВРФ 44.1, 44.2 и 44.3. Кроме того, выходы умножителей 45.1 и 45.2 подсоединены соответственно к первому и второму входу вычитателя 46, выход которого подсоединен к первому входу ФП 48, а выходы умножителей 45.3 и 45.4 подсоединены соответственно к первому и второму входам сумматора 47, выход которого подсоединен ко второму входу ФП 48. При поступлении действительной и мнимой составляющих первого сигнала соответственно на Вх.11 и Вх.12 ВРФ 44.1, 44.2 и 44.3 и поступлении действительной и мнимой составляющих второго сигнала соответственно на Вх.21 и Вх.2 ВРФ 44.1, 44.2 и 44.3 на выходе (Вых.) ВРФ 44.1, 44.2 и 44.3 формируется сигнал, соответствующий разности фаз между вышеупомянутой парой первого и второго сигналов. Необходимо отметить, что параметрами Х₁ и Х₂ обозначена пара сигналов, поступающих соответственно на первый и второй входы ФП 48, вида

Functional diagram of an implementation option VRF 44.1, 44.2 and 44.3 (Fig. 12), which are part of BORF 10, and which can be used as BIRF 3.1, 3.2 and 3.3, contains four multipliers 45.1, 45.2, 45.3 and 45.4, a subtractor 46, an adder 47 and AF 48 species

moreover, the combined first inputs of the multipliers 45.2 and 45.3 and the combined first inputs of the multipliers 45.1 and 45.4 are the first (Bx.11) and second (Bx.12) inputs of the first pair of inputs of the VRF 44.1, 44.2 and 44.3, the combined second inputs of the multipliers 45.1 and 45.3 and the combined second inputs of the multipliers 45.2 and 45.4 are respectively the first (Bx.21) and second (Bx.22) inputs of the second pair of inputs of the VRF 44.1, 44.2 and 44.3, and the output of the FP 48 is the output (Output) of the VRF 44.1, 44.2 and 44.3 . In addition, the outputs of the multipliers 45.1 and 45.2 are connected respectively to the first and second inputs of the subtractor 46, the output of which is connected to the first input of the FP 48, and the outputs of the multipliers 45.3 and 45.4 are connected to the first and second inputs of the adder 47, the output of which is connected to the second input of the FP 48. Upon receipt of the real and imaginary components of the first signal, respectively, at Vx.11 and Vkh.12, VRF 44.1, 44.2 and 44.3 and receipt of the real and imaginary components of the second signal, respectively, at Vkh.21 and Vkh.2 VRF 44.1, 44.2 and 44.3 at the output (Ex.) BP 44.1, 44.2 and 44.3 forming a signal corresponding to the phase difference between the above pair of first and second signals. It should be noted that the parameters X ₁ and X ₂ indicate a pair of signals arriving respectively at the first and second inputs of the FP 48, of the form

Functional diagram of a variant of the implementation of the BFOA 8 (Fig.13) contains an input switch 49, the first, second, third, fourth, fifth and sixth inputs of which are respectively the first (Vkh.1), the second (Vkh.2), the third (Vkh.3 ), fourth (Vkh.4), fifth (Vkh.5) and sixth (Vkh.6) inputs of BFOA 8, a low-frequency calculator of unique amplitudes of difference signals (NVOA) 50, a high-frequency calculator of unique amplitudes of difference signals (VVOA) 51, an output switch 52, the calculator of the parameter of the cyclical number of antennas (WPC) 53, the first and second inputs of which are are respectively the seventh (Bx.7) and eighth (Bx.8) inputs of BFOA 8, the divider 54, the first and second inputs of which are the eleventh (Bx.11) and twelfth (Bx.12) inputs of BFOA 8, respectively, connected in series by the subtractor 55 and a driver of control signals (FUS) 56, and, finally, a sensor 57 of the number 0.3. In this case, the first, second, third, fourth, fifth and sixth outputs of the first and second groups of outputs of the switch 49 are connected respectively to the first, second, third, fourth, fifth and sixth inputs of the NVOA 50 and VVOA 51, the first, second and third outputs of which are connected respectively, to the first, second, and third inputs of the first and second groups of inputs of the output switch 52, the first, second, and third outputs of which are the first (Out.1), second (Out.2), and third (Out.3) outputs of BFOA 8. In addition, the seventh and eighth inputs of VVOA 51 are combined respectively, with the first and second inputs of the VPNs 53, the combined seventh input of the HBOA 50 and the ninth input of the BBOA 51 are the ninth (Bx.9) input of the BFOA 8, the tenth input of the BBOA 51 is the tenth (Bx.10) input of the BFOA 8. The output of the BBOC 53 is connected to the eleventh input of VVOA 51. And finally, the output of the divider 54 together with the output of the sensor 57 are connected respectively to the first and second inputs of the subtractor 55, and the output of the FSF 56 is connected to the combined control inputs of the switches 49 and 52.

BFOA 8 works as follows. The signals corresponding to the phase differences φ ₁ , φ ₂ and φ ₃ between the signals received by the corresponding antenna pairs, the amplitudes r ₁ , r ₂ and r _{3 of the} difference signals, the selected values of the antenna number indices ξ, γ and ν, the direction finding coefficient p and the values of the distance b between the antennas and the wavelength λ of the radio signal from inputs Вх.1, Вх.2, Вх.3, Вх.4, Вх.5, Вх.6, Вх.7, Вх.8, Вх.9, Вх .10, Bx.11 and Bx.12 of BFOA 8 are supplied respectively to the first, second, third, fourth, fifth and sixth inputs of the switch 49, to the combined first input of the VPNs 53 and the seventh input VVOA 51, to the combined second input of the VPNs 53 and the eighth input of the VVOA 51, to the combined ninth input of the VVOA 51 and the seventh input of the VVOA 50, the tenth input of the VVOA 51 and the first and second inputs of the divider 54. In the divider 54, the ratio of the amplitude of the signal received at its first input to the amplitude of the signal received at its second input. The signal from the output of the divider 54 together with the signal from the output of the sensor 57 are respectively supplied to the first and second inputs of the subtractor 55, where the difference between the amplitude of the signal received at its first input and the amplitude of the signal received at its second input is determined. The signal from the output of the subtractor 55 is fed to the input of the FSF 56, the output of which forms a signal of a logical unit if the signal received at its input is not equal to zero and has a positive sign or a logic zero signal otherwise. The signal from the output of the FSF 56 is fed to the combined control inputs of the switches 49 and 52. In the RSCN 53, the values of ξ and γ of the selected antenna numbers, according to rule (45), form the parameter l, which takes one of two values +1 or -1, and the signal corresponding to one of the aforementioned generated values of the parameter l, from the output of the VPNs 53 goes to the eleventh input of VVOA 51. In the case of the logic zero signals arriving at the combined control inputs of switches 49 and 52, six inputs of switch 49 are connected to the corresponding six the outputs of its first group of outputs, and the three inputs of the first group of inputs of the switch 52 are connected respectively to its three outputs. In this case, the signals from the six inputs of the switch 49 are received respectively at the six outputs of its first group of outputs and, further, are supplied respectively to the first six inputs of the IEEE 50, where, taking into account the value ν of the index number of the antenna received at its seventh input, according to rule (42) the formation of three unique amplitude values of R ₁ , R ₂ and R ₃ difference signals. The signals corresponding to the three unique amplitude values R ₁ , R ₂ and R _{3 of the} difference signals from the three outputs of the IEEA 50 are respectively supplied to the three inputs of the first group of inputs of the switch 52 connected to its three outputs, and then from the three outputs of the switch 52 are respectively the first (Out.1), second (Out.2) and third (Out.3) outputs of the BFOA 8. In the case of a logic unit receiving the combined control inputs of the switches 49 and 52, six inputs of the switch 49 are connected to the corresponding six outputs of its second group outputs, and the three inputs of the second group of inputs of the switch 52 are connected respectively to its three outputs. In this case, the signals from the six inputs of the switch 49 are respectively supplied to the six outputs of its second group of outputs and then fed respectively to the first six inputs of the VVOA 51, where, taking into account the values of ξ, γ and ν of the indices of the antenna numbers, the values of the direction finding coefficient p and the value of the parameter l cyclical numbers of antennas received at its seventh, eighth, ninth, tenth and eleventh inputs, respectively, according to rule (44), the formation of three unique amplitude values of R ₁ , R ₂ and R ₃ difference signals. The signals corresponding to the three values of R ₁ , R _2, and R ₃ from the three outputs of VVOA 51 are respectively supplied to the three inputs of the second group of inputs of the switch 52 connected to its three outputs, and then from the three outputs of the switch 52 are respectively supplied to the outputs of the Outputs 1 Exit 2 and Exit 3 of BFOA 8.

Functional diagram of the implementation option of the IEEA 50 (Fig. 14), which is part of BFOA 8, contains three adders 58.1, 58.2 and 58.3, three inverters 59.1, 59.2 and 59.3, sensors 60, 61 and 62 of numbers 1, 2 and 3, respectively, three “I” element (EI) 63.1, 63.2 and 63.3, three switches 64.1, 64.2 and 64.3, three FPs 65.1, 65.2 and 65.3 of the form sgn (X) and three multipliers 66.1, 66.2 and 66.3, the outputs of which are the first (Out.1 ), the second (Out.2) and third (Out.3) outputs of the IEEE 50. Moreover, the combined first inputs of the adders 58.2 and 58.3 and the switch 64.1 are the first input (In1) of the IEEE 50, the combined second input of the adder 58.3 and the first inputs of the total and 58.1 and switch 64.2 are the second input (Input 2) of IEEE 50, the combined first input of switch 64.3 and the second inputs of adders 58.1 and 58.2 are the third input (Input 3) of IEA 50, the second inputs of multipliers 66.1, 66.2 and 66.3 are the fourth (Input 4), fifth (Input 5) and sixth (Input 6) inputs of IEEE 50, and the combined second inputs of EI 63.1, 63.2 and 63.3 are the seventh input (Input 7) of IEA 50. In addition, the outputs of adders 58.1 , 58.2 and 58.3 are connected respectively to the inputs of inverters 59.1, 59.2 and 59.3, the outputs of which are connected to the second inputs of the switches 64.1, 64.2 and 64.3, respectively but. The outputs of the sensors 60, 61 and 62 are connected to the first inputs of EI 63.1, 63.2 and 63.3, respectively, the outputs of which are connected to the control inputs of the switches 64.1, 64.2 and 64.3, respectively. And finally, the outputs of the switches 64.1, 64.2 and 64.3 are connected to the inputs of the FP 65.1, 65.2 and 65.3, respectively, the outputs of which are connected to the first inputs of the multipliers 66.1, 66.2 and 66.3, respectively.

IEEA 50 works as follows. The signals corresponding to the phase differences φ ₁ , φ ₂ and φ ₃ , the amplitudes r ₁ , r ₂ and r _{3 of the} difference signals and the selected value ν of the antenna number, from inputs Вх.1, Вх.2, Вх.3, Вх.4, Vkh.5, Vkh.6 and Vkh.7 NVOA 50 arrive respectively on the combined first inputs of the adders 58.2 and 58.3 and the switch 64.1, on the combined second input of the adder 58.3 and the first inputs of the adder 58.1 and switch 64.2, on the combined first input of the switch 64.3 and second the inputs of adders 58.1 and 58.2, to the second inputs of the multipliers 66.1, 66.2 and 66.3 and to the combined second inputs of the EI 63.1, 63.2 and 63.3. The signals from their outputs, summed in each of the adders 58.1, 58.2, and 58.3, respectively, go to the inputs of inverters 59.1, 59.2, and 59.3, where their signs are reversed, and then the signals from the inverters 59.1, 59.2, and 59.3 go to the second inputs of switches 64.1 , 64.2 and 64.3 respectively. The signals from the outputs of the sensors 60, 61 and 62 are fed to the first inputs of EI 63.1, 63.2 and 63.3, respectively. In each of EI 63.1, 63.2 or 63.3, a signal of a logical unit is generated if the signals received at its pair of inputs coincide, or a signal of a logical zero otherwise. The signals of a logical unit or zero from the outputs of EI 63.1, 63.2 and 63.3 are fed to the control inputs of the switches 64.1, 64.2 and 64.3, respectively. In each of the switches 64.1, 64.2 and 64.3, by the command received at its control input as a logic zero signal, its first input is connected to the output, and by the command received at its control input as a logical unit, its second input is connected to the exit. The signals from the corresponding inputs of the switches 64.1, 64.2 and 64.3 go to their outputs and then to the inputs of the FP 65.1, 65.2 and 65.3, respectively. The signals corresponding to the unit sign functions from the outputs of the FP 65.1, 65.2 and 65.3 are fed to the first inputs of the multipliers 66.1, 66.2 and 66.3. The signals pairwise multiplied in each of the multipliers 66.1, 66.2 and 66.3, corresponding to the three single-valued amplitude values R ₁ , R ₂ and R _{3 of the} difference signals, from the outputs of the multipliers 66.1, 66.2 and 66.3 are received respectively at the outputs Output 1, Output 2 and Output .3 IEE 50.

Functional diagram of an embodiment of VVOA 51 (FIG. 15), which is part of BFOA 8, contains four input switches 67.1, 67.2, 67.3 and 67.4, four multipliers 68.1, 68.2, 68.3 and 68.4, two phase transitions 69.1 and 69.2 of the form sgn (X) , an inverter 70, and three output switches 71.1, 71.2, and 71.3, the first, second, and third inputs of switch 67.1 being the first (Input 1), second (Input 2), and third (Input 3) inputs of VVOA 51, the combined first , the combined second and combined third inputs of the switches 67.2, 67.3 and 67.4 are the fourth (Bx.4), fifth (Bx.5) and sixth (Bx.6) inputs of VVOA 51, respectively the combined control inputs of switches 67.1, 67.2 and 71.1 are the seventh (Input 7) input of VVOA 51, the combined control inputs of switches 67.3 and 71.2 are the eighth (Input 8) input of VVOA 51, the combined control inputs of switches 67.4 and 71.3 are the ninth (Input. 9) VVOA 51 input, the first and second inputs of the multiplier 68.1 are the tenth (Vkh.10) and eleventh (Vkh.11) inputs of VVOA 51, respectively, and the combined first, combined second and combined third outputs of switches 71.1, 71.2 and 71.3 are respectively the first (Exit 1), the second (Exit 2) and the third (Exit 3) VVOA outputs 51. In addition, the output of switch 67.1 is connected to the input of the FP 69.1, the output of which is connected to the first input of the multiplier 68.2, the second input and output of which is connected respectively to the output of the switch 67.2 and the input of the switch 71.1. The output of switch 67.3 is connected to the first input of the multiplier 68.3, the second input and output of which are connected respectively to the output of the multiplier 68.1 and to the combined inputs of the FP 69.2 and the switch 71.2, and the output of the FP 69.2 is connected to the first input of the multiplier 68.4. Finally, the output of the switch 67.4 is connected to the input of the inverter 70, the output of which is connected to the second input of the multiplier 68.4, the output of which is connected to the input of the switch 71.3.

VVOA 51 works as follows. The signals corresponding to the phase differences φ ₁ , φ ₂ and φ ₃ , the amplitudes r ₁ , r ₂ and r _{3 of the} difference signals, the selected values of the antenna numbers ξ, γ and ν, the direction finding coefficient p and the value of the parameter l of the antenna number cyclicity, s inputs I.1, I.2, I.3, I.4, I.5, I.6, I.7, I.8, I.9, I.10 and I.11 VVOA 51 are received respectively at the first , the second and third inputs of switch 67.1, and the combined first, combined second and combined third inputs of switches 67.2, 67.3 and 67.4 - to the combined control inputs of switches 67.1, 67.2 and 71.1, to integrate nennye control inputs of the switches 67.3 and 71.2, in the combined control switch inputs 67.4 and 71.3 and the first and second inputs of the multiplier 68.1. The signals multiplied in the multiplier 68.1 from its output go to the second input of the multiplier 68.3. In each of the switches 67.1, 67.2, 67.3 and 67.4, by a signal supplied to its control input, one of its inputs is connected to its output, the serial number of which coincides with the antenna number of the corresponding control signal. In the same way, in each of the switches 71.1, 71.2 and 71.3, a signal supplied to its control input connects one of its outputs to its input, the serial number of which coincides with the antenna number of the corresponding control signal. The signals from the above corresponding control signals of the inputs of the switches 67.1, 67.2, 67.3 and 67.4 are fed to their outputs and then from their outputs respectively to the input of the FP 69.1, the second input of the multiplier 68.2, the first input of the multiplier 68.3 and the input of the inverter 70. The signal from the output of the FP 69.1 is received to the first input of the multiplier 68.2, where the pair of signals multiplied by its pair of inputs is multiplied. Next, the signal from the output of the multiplier 68.2 goes to the input of the switch 71.1. After the multiplication of the signals received at the pair of inputs of the multiplier 68.3, the signal from its output goes to the combined inputs of the switch 71.2 and FP 69.2. After converting the signal to FP 69.2, the signal from its output goes to the first input of the multiplier 68.4. After converting the signal in the inverter 70, the signal generated at its output is fed to the second input of the multiplier 68.4, where the signals received at its pair of inputs are multiplied, and then the generated signal from the output of the multiplier 68.4 goes to the input of the switch 71.3. The signals corresponding to the three unique amplitude values R _ξ , R _γ and R _{ν of the} difference signals from the inputs of switches 71.1, 71.2 and 71.3, respectively, are fed to one of the outputs corresponding to each of the switches 71.1, 71.2 and 71.3, as a result of which an ordered set of three signals corresponding to three unique amplitude values of R ₁ , R ₂ and R ₃ difference signals, respectively, is supplied to the first (Out.1), second (Out.2) and third (Out.3) outputs of VVOA 51.

Functional diagram of a variant of the implementation of the WPCN 53 (FIG. 16), which is part of the BFOA 8, contains the element “I” (EI) 72, the comparison element (ES) 73, the subtractor 74, the multiplier 75, the adder 76 and the sensors 77 and 78 of the numbers 1 and 3, respectively, wherein the output of the sensor 77 is connected to the combined first input of the EI 72 and the second input of the subtractor 74, the combined second input of the EI 72 and the first input of the subtractor 74 are the first input (Input 1) of the WRC 53, the outputs of the EI 72 and the sensor 78 are connected respectively to the pair of inputs of the multiplier 75, the outputs of the subtractor 74 and the multiplier 75 are connected respectively to a pair of inputs of the adder 76, whose output is connected to the first input of the ES 73, and a second input and an output ES 73 are respectively a second input (INP2) and the output (O). VPTSN 53.

VPTSN 53 works as follows. The signals corresponding to the selected values of the antenna numbers ξ and γ from the inputs Вх.1 and Вх.2 of the WRC 53 are supplied respectively to the combined second input of the EI and the first input of the subtractor 74 and to the second input of the ES 73. The signal corresponding to the number 1 from the sensor output 77 arrives at the combined first input of EI 72 and the second input of subtractor 74. When the signals received at the pair of inputs of EI 72 coincide, a signal corresponding to 1 is generated at its output, otherwise, a signal corresponding to 0. A pair of signals from the outputs of EI 72 and sensor 78 goes to a pair of inputs smartly Itel 75 where their multiplication is performed. At the output of the subtractor 74, a signal is generated corresponding to the difference between the signals received at its first and second inputs. A pair of signals from the outputs of the subtractor 74 and the multiplier 75 is supplied respectively to the pair of inputs of the adder 76, where they are summed, and the total signal from the output of the adder 76 goes to the first input of the ES 73. In the case of equality of the signals received at the pair of inputs of the ES 73 at its output a signal is generated corresponding to a value of +1, otherwise, a signal corresponding to a value of -1 is generated. The signal corresponding to the value of the parameter l of the cyclic number of the antennas, from the output of the ES 73 is fed to the output (Output) of the VPNs 53.

, выход которого является выходом (Вых.) АВА 11, и датчики 82 и 83 чисел 2 и

соответственно, причем выход датчика 82 подсоединен к первому входу умножителя 79.1, второй вход которого является первым входом (Вх.1) АВА 11, а выход подсоединен к первому входу вычитателя 80.1, второй вход которого, объединенный с вторым входом вычитателя 80.3, являются вторым входом (Вх.2) АВА 11, а выход подсоединен к первому входу вычитателя 80.2. Объединенные второй вход вычитателя 80.2 и первый вход вычитателя 80.3 являются третьим входом (Вх.3) АВА 11, а выход вычитателя 80.2 подсоединен к первому входу ФП 81. И наконец, выходы датчика 83 и вычитателя 80.3 подсоединены соответственно к паре входов умножителя 79.2, выход которого подсоединен к второму входу ФП 81.Functional diagram of the implementation option ABA 11 (Fig. 17) contains two multiply inhabitants 79.1 and 79.2, three subtractors 80.1, 80.2 and 80.3, type 81 FP

, the output of which is the output (Output) ABA 11, and sensors 82 and 83 of the numbers 2 and

accordingly, the output of the sensor 82 is connected to the first input of the multiplier 79.1, the second input of which is the first input (Bx.1) of ABA 11, and the output is connected to the first input of the subtractor 80.1, the second input of which, combined with the second input of the subtractor 80.3, is the second input (Input 2) ABA 11, and the output is connected to the first input of the subtractor 80.2. The combined second input of the subtractor 80.2 and the first input of the subtractor 80.3 are the third input (Vkh.3) ABA 11, and the output of the subtractor 80.2 is connected to the first input of the FI 81. Finally, the outputs of the sensor 83 and the subtractor 80.3 are connected respectively to the pair of inputs of the multiplier 79.2, the output which is connected to the second input of the FP 81.

обозначены сигналы, поступающие на его первый и второй входы соответственно.ABA 11 works as follows. The signals corresponding to the unambiguous amplitude values R ₁ , R ₂ and R _{3 of the} difference signals from inputs Вх.1, Вх.2 and Вх.3 АВА 11 are supplied respectively to the second input of the multiplier 79.1, to the combined second inputs of the subtractors 80.1 and 80.3 and to combined the first input of the subtractor 80.3 and the second input of the subtractor 80.2. The signal from the output of the sensor 82 is fed to the first input of the multiplier 79.1, in which the multiplication of the signals received at its pair of inputs is performed. The signal from the output of the multiplier 79.1 is fed to the first input of the subtractor 80.1, where the signal received at its second input is subtracted from it. The signal from the output of the subtractor 80.1 is fed to the first input of the subtractor 80.2, where the signal received at its second input is subtracted from it. Next, the signal from the output of the subtractor 80.2 is fed to the first input of the FI 81. In the subtractor 80.3, the signal received at its second input is subtracted from the signal received at its first input. The signal from the output of the sensor 83, together with the signal from the output of the subtractor 80.3, are respectively supplied to a pair of multipliers 79.2, where they are multiplied. And, finally, the signal from the output of the multiplier 79.2 goes to the second input of the FC 81. In the FC 81, the azimuth θ _{R of the} radio signal is calculated, and the signal corresponding to the obtained value θ _R from the output of the PC 81 goes to the output (Output) ABA 11. It should be noted that the parameters X ₁ and X ₂ in the FC 81 species

the signals arriving at its first and second inputs, respectively, are indicated.

, причем объединенные вход ФП 84.1 и первые входы умножителей 85.1 и 85.2 являются первым входом (Вх.1) ВКСП 12, объединенные вход ФП 84.2, второй вход умножителя 85.1 и первый вход умножителя 85,3 являются вторым входом (Вх.2) ВКСП 12, объединенные вход ФП 84.3 и вторые входы умножителей 85.2 и 85.3 являются третьим входом (Вх.3) ВКСП 12, а выход ФП 91 является выходом (Вых.) ВКСП 12. Кроме того, выходы ФП 84.1, 84.2 и 84.3 подсоединены соответственно к трем входам сумматора 86.1, выход которого подсоединен к объединенным первым входам сумматоров 89.1 и 89.2, выходы которых подсоединены соответственно к первому и второму входам делителя 90. Выходы умножителей 85.1. 85.2 и 85.3 подсоединены соответственно к трем входам сумматора 86.2, выход которого подсоединен к объединенным второму входу умножителя 85.4 и входу инвертора 88, выход которого подсоединен к второму входу сумматора 89.2. И наконец, выход датчика 87 подсоединен к первому входу умножителя 85.4, выход которого подсоединен к второму входу сумматора 89.1.Functional diagram of an embodiment of VKSS 12 (Fig. 18) contains three AF 84.1, 84.2 and 84.3 of the type X ² , four multipliers 85.1, 85.2, 85.3 and 85.4, two three-input adders 86.1 and 86.2, a number 87 sensor 87, an inverter 88, two adders 89.1 and 89.2 and series-connected divider 90 and FP 91 types

moreover, the combined input of the FP 84.1 and the first inputs of the multipliers 85.1 and 85.2 are the first input (Bx.1) of the VKSP 12, the combined input of the FP 84.2, the second input of the multiplier 85.1 and the first input of the multiplier 85.3 are the second input (Bx.2) of the VKSP 12 , the combined input of the FP 84.3 and the second inputs of the multipliers 85.2 and 85.3 are the third input (Bx.3) of the VKSP 12, and the output of the FP 91 is the output (Output) of the VKSP 12. In addition, the outputs of the FP 84.1, 84.2 and 84.3 are connected respectively to three the inputs of the adder 86.1, the output of which is connected to the combined first inputs of the adders 89.1 and 89.2, the outputs of which are connected respectively to first and second inputs of the divider 90. The outputs of the multipliers 85.1. 85.2 and 85.3 are connected respectively to the three inputs of the adder 86.2, the output of which is connected to the combined second input of the multiplier 85.4 and the input of the inverter 88, the output of which is connected to the second input of the adder 89.2. And finally, the output of the sensor 87 is connected to the first input of the multiplier 85.4, the output of which is connected to the second input of the adder 89.1.

. После преобразования сигнала в ФП 91 сигнал, соответствующий значению параметра μ, характеризующего наличие квадратурной составляющей помехового сигнала, с выхода ФП 91 поступает на выход (Вых.) ВКСП 12.VKSP 12 works as follows. The signals corresponding to the unambiguous amplitude values R ₁ , R ₂ and R _{3 of the} difference signals from the inputs Вх.1, Вх.2 and Вх.3 VKSP 12 are received respectively at the combined input of the FP 84.1 and the first inputs of the multipliers 85.1 and 85.2, at the combined input FP 84.2, the second input of the multiplier 85.1 and the first input of the multiplier 85.3 and the combined input of the FP 84.3 and the second inputs of the multipliers 85.2 and 85.3. The signals converted from FP 84.1, 84.2 and 84.3 from their outputs are respectively supplied to the three inputs of the adder 86.1, where they are summed. The signals summed in the adder 86.1 in the form of a total signal from the output of the adder 86.1 go to the combined first inputs of the adders 89.1 and 89.2. The signals in pairs multiplied in multipliers 85.1, 85.2 and 85.3 from their outputs go respectively to the three inputs of the adder 86.2, where they are summed. The summed signal from the output of the adder 86.2 goes to the combined input of the inverter 88 and the second input of the multiplier 85.4, the first input of which receives the signal from the output of the sensor 87. The signal from the output of the multiplier 85.4 goes to the second input of the adder 89.1, where it is summed with the signal received at its first entrance. The signal from the output of the inverter 88 is fed to the second input of the adder 89.2, where it is summed with the signal received at its first input. The signals from the outputs of the adders 89.1 and 89.2 respectively received at the first and second inputs of the divider 90. In the divider 90, the amplitude of the signal received at its first input is divided by the amplitude of the signal received at its second input. The signal from the output of the divider 90 goes to the input of the FP 91 type

. After converting the signal to FP 91, the signal corresponding to the value of the parameter μ characterizing the presence of the quadrature component of the interfering signal is fed from the output of the FP 91 to the output (Out) of VKSP 12.

Functional diagram of an embodiment of FVA 13 (Fig. 19) contains an input switch 92, switch 93, six multipliers 94.1, 94.2, 94.3, 94.4, 94.5 and 94.6, three dividers 95.1, 95.2 and 95.3, two subtractors 96.1 and 96.2, type 97 FP arcsin (X), adder 98, output switch 99, the output of which is the output (Out) of the PVA 13, sensors 100, 101, 102 and 103 of the numbers 1, 2, 3 and π, respectively, and, in addition, a functional converter (FP ) 104, which coincides in the scheme with ABA 11 (Fig. 17), moreover, the combined first input of the subtractor 96.1 and the control input of the switch 93 are the first input (Bx.1) of the FVA 13, combined the second inputs of the multiplier 94.4 and subtractor 96.2 are the second input (Vkh.2) FVA 13, the first, second and third inputs of the switch 92 are the third (Vkh.3), fourth (Vkh.4) and fifth (Vkh.5) inputs of the FVA 13, the combined control inputs of the switches 92 and 99 are the sixth input (Bx.6) of the FVA 13, and the first input of the multiplier 94.1 and the second input of the multiplier 94.2 are the seventh (Bx.7) and eighth (Bx.8) inputs of the FVA 13. Moreover, the first, second and third outputs of the first and second groups of outputs of the switch 92 are connected respectively to the first, second and the network inputs of the FP 104 and the switch 93, the outputs of which are connected to the first inputs of the switch 99 and the multiplier 94.2, respectively. The output of the sensor 103 is connected to the combined second inputs of the multipliers 94.1, 94.3 and 94.5. The outputs of the multipliers 94.2 and 94.1 are connected respectively to the first and second inputs of the divider 95.1, the output of which is connected to the input of the FP 97, the output of which is connected to the first input of the multiplier 94.4. The output of the sensor 100 is connected to the combined second input of the subtractor 96.1 and the first input of the subtractor 96.2, the output of which is connected to the first input of the multiplier 94.5, the output of which is connected to the first input of the divider 95.3. The output of the subtractor 96.1 is connected to the first input of the multiplier 94.3, the output of which is connected to the first input of the multiplier 94.6. The output of the sensor 101 is connected to the combined second inputs of the divider 95.3 and the multiplier 94.6, the output of which together with the output of the sensor 102 are connected respectively to the first and second inputs of the divider 95.2. The outputs of the divider 95.2, the multiplier 94.4 and the divider 95.3 are connected respectively to the three inputs of the adder 98, the output of which is connected to the second input of the switch 99.

FVA 13 works as follows. The signals corresponding to the selected value ξ of the antenna number, the value of the direction finding coefficient p, the phase differences φ ₁ , φ ₂ and φ ₃ between the difference signals, a logical unit or logical zero, and the values of the distance b between the antennas and the wavelength λ of the radio signal from the inputs Вх. 1, Input 2, Input 3, Input 4, Input 5, Input 6, Input 7 and Input 8 are received respectively at the combined first input of the subtractor 96.1 and the control input of the switch 93, at the combined second inputs of the multiplier 94.4 and subtractor 96.2, at the first, second and third inputs of the switch 92, at nennye control inputs of switches 92 and 99, the first input of the multiplier 94.1, and a second input of multiplier 94.2. In the case of input from Vh.6 FVA 13 to the combined control inputs of the switches 92 and 99 of the logical unit signal, the first, second, and third inputs of the switch 92 are connected respectively to the first, second, and third outputs of the first group of its outputs, and the first input of the switch 99 is connected to its exit. In this case, the signals from inputs Vkh.3, Vkh.4 and Vkh.3 FVA 13 come from the first, second and third inputs of the switch 92, respectively, to the first, second and third outputs of its first group of outputs and then from the above outputs of the switch 92 are received respectively the first, second and third inputs of the FP 104. In the FP 104, the signals corresponding to the phase differences φ _R1 , φ _R2 φ _R3 are processed in full accordance with the signal processing rules in the aforementioned ABA 11. The signal corresponding to the obtained azimuth value θ _φ from the output of the FP 104 goes to the first input com utatora 99 and further to the output of the switch 99. The signal corresponding to the obtained value of the azimuth θ _φ, the output switch 99 is supplied to the output (O). 13. In the case of PVA proceeds with Vh.6 PVA 13 on the combined control inputs of switches 92 and 99 of the signal logical zero, the first, second and third inputs of the switch 92 are connected respectively to the first, second and third outputs of the second group of its outputs, and the second input of the switch 99 is connected to its output. In this case, the signals from inputs Vh.3, Vkh.4 and Vkh.5 FVA 13 come from the first, second and third inputs of the switch 92, respectively, to the first, second and third outputs of its second group of outputs and then from the above outputs of the switch 92 are received respectively the first, second and third inputs of the switch 93. In the switch 93, by the command received at its control input and corresponding to the selected value ξ of the antenna number, its ξth input is connected to its output. Next, the signal corresponding to the selected value of the phase difference φ _Rξ from the output of the switch 93 is fed to the first input of the multiplier 94.2, in which it is multiplied with the signal received at its second input. The signal from the output of the sensor 103 is supplied to the combined second inputs of the multipliers 94.1, 94.3 and 94.5. The multiplier 94.1 multiplies the pair of signals received at its pair of inputs. The signals from the outputs of the multipliers 94.2 and 94.1 are respectively supplied to the first and second inputs of the divider 95.1, where the signal received at its first input is divided by the signal received at its second input. The signal from the output of the divider 95.1 goes to the input of the FP 97, where it is converted in accordance with the functional dependence arcsin (X) and from the output of the FP 97 goes to the first input of the multiplier 94.4, where it is multiplied with the signal received at the second input of the multiplier 94.4. The signal from the output of the sensor 100 is fed to the combined second input of the subtractor 96.1 and the first input of the subtractor 96.2. In subtractors 96.1 and 96.2, the signals received at their second inputs are subtracted from the signals received at their first inputs. The signals from the outputs of the subtractors 96.1 and 96.2 are fed to the first inputs of the multipliers 94.3 and 94.5, respectively. The signals multiplied in multipliers 94.3 and 94.5 from their outputs are respectively supplied to the first inputs of the multiplier 94.6 and the divider 95.3. The signal from the output of the sensor 101 is fed to the combined second inputs of the multiplier 94.6 and the divider 95.3. A pair of signals received at the inputs of the multiplier 94.6 is multiplied, and the resulting signal from the output of the multiplier 94.6, together with the signal from the output of the sensor 102, are respectively supplied to the first and second inputs of the divider 95.2. In the dividers 95.2 and 95.3, the signals received at their first inputs are divided into the signals received at their second inputs. The signals from the outputs of the divider 95.2, the multiplier 94.4 and the divider 95.3 are fed to the corresponding inputs of the adder 98, where they are summed. The summed signal corresponding to the azimuth value θ _φ from the output of the adder 98 is fed to the second input of the switch 99, which is connected to its output. The signal corresponding to the azimuth θ _φ from the output of the switch 99 is fed to the output (Out) of the PVA 13.

The functional diagram of the embodiment of VPOA 14 (Fig. 20) contains two subtractors 105.1 and 105.2, the first and second inputs of the subtractor 105.1 being the first (Bx.1) and second (Bx.2) inputs of VPOA 14, two types of FP 106.1 and 106.2 of the form | X |, and the output of FP 106.2 is the output (Out) of HLPA 14, a comparison element (ES) 107, two multipliers 108.1 and 108.2, a sensor 109 of the angular parameter π and a sensor 110 of number 2, and the output of the subtractor 105.1 is connected to the input of the FP 106.1 the output of which is connected to the combined first inputs of the subtractor 105.2 and the ES 107, the output of the sensor 109 is connected to the combined the second input of the ES 107 and the first input of the multiplier 108.1, the output of the sensor 110 is connected to the second input of the multiplier 108.1, the outputs of the ES 107 and the multiplier 108.1 are connected respectively to the first and second inputs of the multiplier 108.2, the output of which is connected to the second input of the subtractor 105.2, the output of which is connected to the input FP 106.2.

VPOA 14 works as follows. The signals corresponding to the values θ _φ and θ _{R of the} azimuth of the radio source from the inputs Bx.1 and Bx.2 respectively received at the first and second inputs of the subtractor 105.1. The signal corresponding to the difference between the above values of azimuths from the output of the subtractor 105.1 is fed to the input of the FP 106.1, where the module of the signal received at its input is determined. The signal from the output of the FP 106.1 is fed to the combined first inputs of the subtractor 105.2 and the ES 107. The signal from the output of the sensor 109, corresponding to the angular value of the parameter π, is fed to the combined second input of the ES and the first input of the multiplier 108.1. ES 107 compares the values of the signals received at its pair of inputs. If the signal received at the first input 107 is not less than the signal received at its second input, a signal corresponding to unity is generated at its output, otherwise, a signal corresponding to zero. The signal from the output of the sensor 110 is fed to the second input of the multiplier 108.1, where the multiplication of the signals received at its pair of inputs is performed.

The signals from the outputs of the ES 107 and the multiplier 108.1 are respectively supplied to the pair of inputs of the multiplier 108.2, where they are multiplied. The signal from the output of the multiplier 108.2 is fed to the second input of the subtractor 105.2. The subtractor 105.2 determines the difference between the signals received at its first and second inputs, respectively. The signal from the output of the subtractor 105.2 goes to the input of the FP 106.2, where its module is determined. The signal corresponding to the error Δθ of the azimuth estimate, from the output of the FP 106.2 goes to the output (Output) VPOA 14.

An implementation option of VAPK 15 is a comparison element that generates a signal in the form of a logical unit at the output, if the signal received at its first input is not less than the signal received at its second input, or the signal is a logical zero otherwise.

, ФП 121 вида arccos(X), датчики 122, 123, 124 и 125 чисел π, 2, 3 и 1 соответственно, датчик 126 нулевого значения угла наклона фронта волны источника радиосигнала и датчик 127 символа β_sp, соответствующего наличию в точке приема радиосигнала, распространяющегося в виде пространственной электромагнитной волны, наклон фронта которой не определен. При этом второй вход умножителя 112.4 и первый вход умножителя 112.5 является соответственно первым (Вх.1) и вторым (Вх.2) входами ВУМ 16. Объединенные вход ФП 111.1 и первые входы умножителей 112.1 и 112.2 являются третьим входом (Вх.3) ВУМ 16. Объединенные вход ФП 111.2, второй вход умножителя 112.1 и первый вход умножителя 112.3 являются четвертым входом (Вх.4) ВУМ 16. Объединенные вход ФП 111.3 и вторые входы умножителей 112.2 и 112.3 являются пятым входом (Вх.5) ВУМ 16. Управляющие входы коммутаторов 118.2 и 119 являются соответственно шестым (Вх.6) и седьмым (Вх.7) входами ВУМ 16, а выход коммутатора 118.2 является выходом (Вых.) ВУМ 16. Кроме того, выходы ФП 111.1, 111.2 и 111.3 и умножителей 112.1, 112.2 и 112.3 подсоединены соответственно к первому, второму и третьему входам сумматоров 113.1 и 113.2, выходы которых подсоединены соответственно к первому и второму входам вычитателя 114, выход которого подсоединен к входу ФП 115, выход которого подсоединен к входу ФП 120, выход которого подсоединен к первому входу умножителя 112.7. Выход умножителя 112.7 подсоединен к объединенным первым входам ЭС 116 и коммутатора 118.1. Выход датчика 125 подсоединен к объединенным вторым входам коммутатора 118.1 и ЭС 116, выход которого подсоединен к управляющему входу коммутатора 118.1, выход которого подсоединен к входу ФП 121, выход которого подсоединен к первому входу коммутатора 118.2. Выход датчика 123 подсоединен к второму входу умножителя 112.5, выход которого подсоединен к первому входу делителя 117. Выход датчика 122 подсоединен к первому входу умножителя 112.4, выход которого совместно с выходом датчика 124 подсоединены соответственно к паре входов умножителя 112.6, выход которого подсоединен к второму входу делителя 117. Выход делителя 117 подсоединен к второму входу умножителя 112.7. И наконец, выходы датчиков 126 и 127 подсоединены соответственно к первому и второму входам коммутатора 119, выход которого подсоединен ко второму входу коммутатора 118.2.The functional diagram of the embodiment of the VUM 16 (Fig. 21) contains three phase transitions 111.1, 111.2 and 111.3 of the form X ² , seven multipliers 112.1, 112.2, 112.3, 112.4, 112.5, 112.6 and 112.7, two adders 113.1 and 113.2, a subtractor 114, FP 115 of the form | X |, a comparison element (ES) 116, a divider 117, three switches 118.1, 118.2 and 119, FP 120 of the form

, FI 121 of the arccos (X) type, π, 2, 3, and 1 sensors 122, 123, 124, and 125, respectively, a sensor 126 of a zero value of the wave front slope angle of the radio signal source and a sensor 127 of the β _sp symbol corresponding to the presence of the radio signal at the receiving point propagating in the form of a spatial electromagnetic wave, the front slope of which is not defined. In this case, the second input of the multiplier 112.4 and the first input of the multiplier 112.5 is the first (Vkh.1) and the second (Vkh.2) inputs of the VUM 16. The combined input of the FP 111.1 and the first inputs of the multipliers 112.1 and 112.2 are the third input (Vkh.3) of the VUM 16. The combined input of the FP 111.2, the second input of the multiplier 112.1 and the first input of the multiplier 112.3 are the fourth input (Vkh.4) VUM 16. The combined input of the FP 111.3 and the second inputs of the multipliers 112.2 and 112.3 are the fifth input (Vkh.5) VUM 16. The control the inputs of the switches 118.2 and 119 are respectively the sixth (Bx.6) and seventh (Bx.7) inputs of the VUM 16, and the outputs One switch 118.2 is the output (Output) of the VUM 16. In addition, the outputs of the FP 111.1, 111.2, and 111.3 and the multipliers 112.1, 112.2, and 112.3 are connected respectively to the first, second, and third inputs of the adders 113.1 and 113.2, the outputs of which are connected respectively to the first and the second inputs of the subtractor 114, the output of which is connected to the input of the FP 115, the output of which is connected to the input of the FP 120, the output of which is connected to the first input of the multiplier 112.7. The output of the multiplier 112.7 is connected to the combined first inputs of the ES 116 and the switch 118.1. The output of the sensor 125 is connected to the combined second inputs of the switch 118.1 and the ES 116, the output of which is connected to the control input of the switch 118.1, the output of which is connected to the input of the FP 121, the output of which is connected to the first input of the switch 118.2. The output of the sensor 123 is connected to the second input of the multiplier 112.5, the output of which is connected to the first input of the divider 117. The output of the sensor 122 is connected to the first input of the multiplier 112.4, the output of which together with the output of the sensor 124 are connected respectively to the pair of inputs of the multiplier 112.6, the output of which is connected to the second input the divider 117. The output of the divider 117 is connected to the second input of the multiplier 112.7. Finally, the outputs of the sensors 126 and 127 are connected respectively to the first and second inputs of the switch 119, the output of which is connected to the second input of the switch 118.2.

VUM 16 works as follows. The signals corresponding to the distance b between the antennas, the wavelength λ of the radio signal, the phase differences φ _R1 , φ _R2 and φ _R3 between the difference signals and the first and second signals of logical units or zeros from the inputs Вх.1, Вх.2, Вх.3, Vkh.4, Vkh.5, Vkh.6 and Vkh.7 VUM 16 arrive respectively on the second input of the multiplier 112.4, on the first input of the multiplier 112.5, on the combined input of the FC 111.1 and the first inputs of the multipliers 112.1 and 112.2, on the combined input of the FC 111.2, the second input of the multiplier 112.1 and the first input of the multiplier 112.3, to the combined input of the FP 111.3 and the second inputs of the multipliers 112.2 and 112.3, to 118.2-governing the switch input and to a control input of the switch 119.

и с выхода ФП 120 поступает на первый вход умножителя 112.7. Сигнал с выхода датчика 123 поступает на второй вход умножителя 112.5. После перемножения сигналов в умножителе 112.5 сигнал с его выхода поступает на первый вход делителя 117. Сигнал с выхода датчика 122 поступает на первый вход умножителя 112.4, где производится его перемножение с сигналом, поступившим на его второй вход. Сигналы с выходов умножителя 112.4 и датчика 124 поступают соответственно на пару входов умножителя 112.6, где производится их перемножение. Сигнал с выхода умножителя 112.6 поступает на второй вход делителя 117. В делителе 117 производится деление сигнала, поступившего на его первый вход, на сигнал, поступивший на его второй вход. Сигнал с выхода делителя 117 поступает на второй вход умножителя 112.7, где производится его перемножение с сигналом, поступившим на его первый вход.The signals converted to FP 111.1, 111.2 and 111.3, from their outputs are fed to the corresponding inputs of the adder 113.1. The pairwise multiplied signals from the outputs of the multipliers 112.1, 112.2 and 112.3 are fed to the corresponding inputs of the adder 113.2. The signals summed in each of the adders 113.1 and 113.2, from their outputs, respectively arrive at the first and second inputs of the subtractor 114. After subtracting from the signal received at the first input of the subtractor 114, the signal received at its second input, the received differential signal from the output of the subtractor 114 is fed to the input of the FP 115, where its module is determined. The signal from the output of the FP 115 is fed to the input of the FP 120, where it is converted in accordance with the functional dependence

and from the output of the FI 120 goes to the first input of the multiplier 112.7. The signal from the output of the sensor 123 is supplied to the second input of the multiplier 112.5. After multiplying the signals in the multiplier 112.5, the signal from its output goes to the first input of the divider 117. The signal from the output of the sensor 122 goes to the first input of the multiplier 112.4, where it is multiplied with the signal received at its second input. The signals from the outputs of the multiplier 112.4 and the sensor 124 are respectively supplied to the pair of inputs of the multiplier 112.6, where they are multiplied. The signal from the output of the multiplier 112.6 is supplied to the second input of the divider 117. In the divider 117, the signal received at its first input is divided into the signal received at its second input. The signal from the output of the divider 117 is fed to the second input of the multiplier 112.7, where it is multiplied with the signal received at its first input.

The signal from the output of the multiplier 112.7 is fed to the combined first inputs of the ES 116 and the switch 118.1. The signal from the output of the sensor 125 is fed to the combined second inputs of the ES 116 and the switch 118.1. In ES 116, a logical unit signal is generated if the signal received at its first input is less than the signal received at its second input, or a logic zero signal otherwise. The signal of a logical unit or zero from the output of the ES 116 is fed to the control input of the switch 118.1. If a logical unit signal arrives at the control input of the switch 118.1, its first input is connected to its output and, accordingly, the signal from its first input goes to its output, and if a logical zero signal arrives at the control input of the switch 118.1, its second input is connected to its output and, accordingly, the signal from its second input goes to its output. The signal from the output of the switch 118.1 is fed to the input of the FP 121, after the conversion in which the signal from the output of the FP 121 is fed to the first input of the switch 118.2.

The signals from the outputs of the sensors 126 and 127 are respectively supplied to the first and second inputs of the switch 119. If a logical unit signal arrives at the control input of the switch 119, its first input is connected to its output and, accordingly, the signal from its first input goes to its output, and if a logical zero signal arrives at the control input of the switch 119, its second input is connected to its output and, accordingly, the signal from its second input goes to its output. The signal from the output of the switch 119 is fed to the second input of the switch 118.2. If a logical unit signal arrives at the control input of the switch 118.2, its first input is connected to its output and, accordingly, the signal from its first input goes to its output, and if a logical zero is received at the control input of the switch 118.2, its second input is connected to its output and, accordingly, the signal from its second input goes to its output. The signal corresponding to the value of the angle β of the slope of the front of the IRI wave, from the output of the switch 118.2 goes to the output (Out) of the VUM 16.

Functional diagram of an embodiment of BOA 17 (FIG. 22) comprises a subtractor 128, two adders 129.1 and 129.2, a divider 130, an FP 131 of the form | X |, a comparison element (ES) 132, a multiplier 133, two switches 134.1 and 134.2 and sensors 135 and 136 values of π and number 2, respectively, with the control input of the switch 134.1 being the first input (In1) of the BOA 17, and the output of the switch 134.1 is the output (Out) of the BOA 17, the combined second inputs of the subtractor 128 and the adder 129.1 are the second input (Bx .2) BOA 17, the combined first inputs of the subtractor 128 and the adder 129.1 and the second input of the switch 134.2 i are the third input (Vkh.3) BOA 17, and the control input of the switch 134.2 is the fourth input (Vkh.4) BOA 17. In addition, the output of the subtractor 128 is connected to the input of the FP 131, the output of which is connected to the first input of the ES 132. The sensor output 135 is connected to the combined second inputs of the ES 132 and the multiplier 133. The output of the ES 132 is connected to the first input of the multiplier 133. The output of the adder 129.1 together with the output of the sensor 136 are connected respectively to the first and second inputs of the divider 130, the output of which together with the output of the multiplier 133 are connected respectively to pa re inputs of adder 129.2, the output of which is connected to the combined first inputs of switches 134.1 and 134.2.

BOA 17 works as follows. The signals corresponding to the first signal of a logical unit or zero, the azimuth value θ _φ , the azimuth value θ _R and the second signal of a logical unit or zero, from the inputs Vx.1, Vkh.2, Vkh.3 and Vkh.4 BOA 17 are respectively supplied to the control the input of the switch 134.1, to the combined inputs of the subtractor 128 and the adder 129.1, to the combined first inputs of the subtractor 128 and the adder 129.1 and the second input of the switch 134.2 and to the control input of the switch 134.2. The signal from the output of the subtractor 128 goes to the input of the FI 131 and then, from its output it goes to the first input of the ES 132. The total signal from the output of the adder 129.1 and the signal from the output of the sensor 136 go respectively to the first and second inputs of the divider 130, where the signal is divided, received at its first input, the signal received at its second input. The signal from the output of the sensor 135 is fed to the combined second inputs of the ES 132 and the multiplier 133. The signal corresponding to unity is generated at the output of the ES 132 if the signal received at its first input is not less than the signal received at its second input, or the signal corresponding to zero, otherwise. The signal from the output of the ES 132 is fed to the first input of the multiplier 133, where it is multiplied with the signal received at its second input. The signals from the outputs of the multiplier 133 and the divider 130 are respectively supplied to the pair of inputs of the adder 129.2, where they are summed. The signal from the output of adder 129.2 is fed to the combined first inputs of switches 134.1 and 134.2. If a logical unit signal arrives at the control input of the switch 134.2, its first input is connected to its output, and, accordingly, the signal from its first input goes to its output. If a logical zero signal arrives at the control input of the switch 134.2, its second input is connected to its output, and, accordingly, the signal from its second input is sent to its output. If a logical unit signal arrives at the control input of the switch 134.1, its first input is connected to its output, and, accordingly, the signal from its first input goes to its output. And if a logical zero signal arrives at the control input of the switch 134.1, its second input is connected to its output, and, accordingly, the signal from its second input goes to its output. The signal corresponding to the unambiguous azimuth θ from the output of the switch 134.1 goes to the output (Out) of the BOA 17.

, причем первые входы вычитателей 137.1 и 137.2 являются соответственно первым (Вх.11) и вторым (Вх.12) входами первой пары входов БФРС 4.1 (4.2, 4.3), вторые входы вычитателей 137.1 и 137.2 являются соответственно первым (Вх.21) и вторым (Вх.22) входами второй пары входов БФРС 4.1 (4.2, 4.3), выход вычитателя 137.1 подсоединен к входу ФП 138.1 и одновременно является вторым выходом (Вых.2) БФРС 4.1 (4.2, 4.3), выход вычитателя 137.2 подсоединен к входу ФП 138.2 и одновременно является третьим выходом (Вых.3) БФРС 4.1 (4.2, 4.3), выходы ФП 138.1 и 138.2 подсоединены соответственно к паре входов сумматора 139, а выход ФП 140 является первым выходом (Вых.1) БФРС 4.1 (4.2, 4.3).The functional diagram of an implementation option for each of the BFRS 4.1, 4.2 and 4.3 (Fig. 23) contains two subtractors 137.1 and 137.2, two AF 138.1 and 138.2 of the form X ² and series-connected adder 139 and the AF 140 of the form

moreover, the first inputs of the subtractors 137.1 and 137.2 are the first (Bx.11) and second (Bx.12) inputs of the first pair of inputs of the BFRS 4.1 (4.2, 4.3), the second inputs of the subtractors 137.1 and 137.2 are the first (Bx.21) and the second (Vkh.22) inputs of the second pair of inputs BFRS 4.1 (4.2, 4.3), the output of the subtractor 137.1 is connected to the input of the FI 138.1 and at the same time is the second output (Vykh.2) BFRS 4.1 (4.2, 4.3), the output of the subtractor 137.2 is connected to the input FP 138.2 and at the same time is the third output (Output 3) of BFRS 4.1 (4.2, 4.3), the outputs of FP 138.1 and 138.2 are connected respectively to the pair of inputs of the adder 139, 140 and the output OP is the first output (OUT 1) BFRS 4.1 (4.2, 4.3).

BFRS 4.1 (4.2, 4.3) works as follows. The signals corresponding to the real and imaginary components of the first signal, the real and imaginary components of the second signal, from inputs Вх.11, Вх.12, Вх.21 and Вх.22 BFRS 4.1 (4.2, 4.3) are received respectively at the first inputs of the subtractors 137.1 and 137.2 and to the second inputs of the subtractors 137.1 and 137.2. In each of the subtractors 137.1 and 137.2, the signal received at its second input is subtracted from the signal received at its first input. The signals from the outputs of the subtractors 137.1 and 137.2, corresponding to the real and imaginary components of the difference signal, are supplied to the inputs of the FI 138.1 and 138.2, respectively, and, in addition, to the second (Out.2) and third (Out.3) outputs of the BFRS 4.1 (4.2 , 4.3). After converting the signals to FP 138.1 and 138.2, the signals from their outputs are respectively sent to the pair of inputs of the adder 139, where they are summed, and then the summed signal from the output of the adder 139 is fed to the input of the FI 140. After converting the signal to FP 140, the signal corresponding to the amplitude value of the difference signal, from the output of the FP 140 enters the first output (Output 1) of the BFRS 4.1 (4.2, 4.3).

It will be appreciated by those skilled in the art that embodiments of various blocks of the direction finder functional diagram (FIG. 6) may have various design differences that are not the subject of the present invention. So, the implementation options for antennas 1.1, 1.2 and 1.3 are given, for example, in [12. Drabkin A.L., Zuzenko V.L., Kislov A.G. Antenna feeder devices. - M .: Owls. radio, 1974], radio receiving units 2.1, 2.2 and 2.3 - in [13. Red at. T. Circuitry of radio receivers. Practical Guide: Trans. with him. - M .: Mir, 1989]; [fourteen. Ashikhmin A.V. Sergeev V.B., Sergienko A.R. Radio receiving tracts of automated radio monitoring complexes: features, solutions and prospects. - Special equipment, special issue, 2002, p. 57-64]; [fifteen. Weissblat A.V. Microwave switching devices on semiconductor diodes. - M .: Radio and communications, 1987]; [16. Nefedov E.I., Saidov A.S., Tagilaev A.R. Microwave Broadband Microstrip Controls. - M .: Radio and communications, 1994]. The implementation of other device units and their functional elements based on digital signal processing is described in a number of works [17. Rabiner L., Gould B. Theory and application of digital signal processing. - M .: Mir, 1978]; [eighteen. Horowitz P., Hill W. Art of circuitry: Per. from English - M .: Mir, 1986]; [19. Ugryumov E.P. Digital circuitry. - SPb .: BHV - Petersburg, 2001]; [twenty. Kupriyanov M.S., Matyushkin B.D. Digital signal processing: processors, algorithms, design tools. - St. Petersburg: Polytechnic, 1999].

(λ_max и λ_min - максимальная и минимальная длина волны рабочего диапазона длин волн) представлен на фиг.24, а внешний вид приемоиндикатора радиопеленгатора, включающего остальные блоки радиопеленгатора и монитор автоматизированного рабочего места (АРМ) оператора, представлен на фиг.25. На фиг.26, 27 и 28 приведены экспериментальные результаты пеленгования широкополосного источника радиоизлучения одним из образцов вышеупомянутого радиопеленгатора при следующих условиях: а) спектральные составляющие пеленгуемого сигнала равномерно распределены в полосе длин волн λ_max≤λ≤λ_min с коэффициентом перекрытия k_λ=4; б) расстояние b между антеннами решетки и длина 2l_a симметричных антенн решетки удовлетворяет условиям

и

соответственно; в) отношение сигнал/шум на частотах измерений составляло в среднем 30 дБ; г) сигналы от пеленгуемого источника радиоизлучения до радиопеленгатора распространялись в виде поверхностных электромагнитных волн (β=0); д) истинные значения азимутов θ ИРИ составляли 181° (фиг.26), 196,8° (фиг.27) и 211,7° (фиг.28) соответственно; е) результаты пеленгования отмечались на экране монитора АРМ оператора в виде точек на частотной оси (оси длин волн λ радиосигнала) в полярной системе координат (частота составляющих радиосигнала соответствует и пропорциональна радиальной координате, а значение азимута соответствует угловой координате) и, кроме того, в виде фрагмента таблицы текущих измеренных значений азимутов частотных составляющих широкополосного источника радиоизлучения. Усредненная по всем азимутальным направлениям и частотным составляющим систематическая средняя квадратическая ошибка σ_θs определения азимута вышеупомянутым радиопеленгатором, включающая как методическую, так и структурную (обусловленную погрешностями калибровки каналов радиопеленгатора и неидентичностью конструкции антенной решетки) составляющие, при проведении серии повторяющихся экспериментов в различных условиях составляет (0,5÷0,7)°. Из сравнения результатов экспериментальных исследований, приведенных на фиг.26, 27 и 28 для различных азимутов θ, следует, что методическая составляющая ошибок пеленгования, устранение которой является целью изобретения, проявляющаяся согласно предложенному способу пеленгования в максимальной степени для азимутального направления θ=196,8°, не превышает структурную составляющую ошибок пеленгования, проявляющуюся для всех азимутальных направлений и полностью определяющую ошибки пеленгования для азимутальных направлений θ=181° и θ=211,7°. Необходимо отметить, что согласно фиг.5 для указанных электродинамических размеров трехэлементной решетки (кривая 3 на фиг.5) методические составляющие ошибок пеленгования, обусловленные взаимным влиянием антенн и мачтового устройство, свойственные аналогу [8], на порядок превышают общие систематические ошибки, характерные для радиопеленгатора, реализующего предложенный способ радиопеленгования.The proposed method of direction finding and direction finding for its implementation are implemented during the modernization of a small series of electronic systems for detecting and determining the location of radio emission sources. Appearance of an equidistant annular antenna array containing three identical axisless symmetric vibrator-type antennas in a symmetrical design with the aforementioned electrodynamic dimensions, mounted on a mast device, providing the operation of the direction finder in accordance with the proposed direction finding method in the operating range of the wavelength λ of the radio signal with an overlap coefficient

(λ _max and λ _min are the maximum and minimum wavelengths of the operating wavelength range) is shown in FIG. 24, and the appearance of the radio direction finder including the other direction finder blocks and the operator’s automated workstation monitor (AWP) is shown in FIG. 25. FIGS. 26, 27 and 28 show the experimental results of direction finding of a broadband radio emission source by one of the samples of the aforementioned direction finder under the following conditions: a) the spectral components of the direction-finding signal are uniformly distributed in the wavelength band λ _max ≤λ≤λ _min with an overlap coefficient k _λ = 4 ; b) the distance b between the array antennas and the length 2l _{a of the} symmetrical array antennas satisfies the conditions

and

respectively; c) the signal-to-noise ratio at measurement frequencies averaged 30 dB; d) the signals from the direction-finding source of radio emission to the direction finder were propagated in the form of surface electromagnetic waves (β = 0); e) the true azimuths θ of the IRI were 181 ° (Fig. 26), 196.8 ° (Fig. 27) and 211.7 ° (Fig. 28), respectively; f) direction finding results were recorded on the operator’s workstation monitor screen in the form of points on the frequency axis (wavelength axis λ of the radio signal) in the polar coordinate system (the frequency of the radio signal components corresponds and is proportional to the radial coordinate, and the azimuth value corresponds to the angular coordinate) and, in addition, in the form of a fragment of the table of the current measured azimuths of the frequency components of the broadband source of radio emission. The systematic mean square error σ _θs averaged over all azimuthal directions and frequency components of the azimuth of the aforementioned direction finder, including both the methodological and structural (due to the errors in the calibration of the direction finder channels and the identity of the antenna array design), constitutes (during a series of repeated experiments under various conditions, 0.5 ÷ 0.7) °. From a comparison of the results of experimental studies shown in Figs. 26, 27 and 28 for different azimuths θ, it follows that the methodological component of direction finding errors, the elimination of which is the aim of the invention, manifested according to the proposed direction finding method to the maximum extent for the azimuth direction θ = 196.8 ° does not exceed the structural component of direction finding errors, which manifests itself for all azimuthal directions and completely determines direction finding errors for the azimuthal directions θ = 181 ° and θ = 211.7 °. It should be noted that according to FIG. 5, for the indicated electrodynamic dimensions of the three-element grating (curve 3 in FIG. 5), the methodological components of direction-finding errors due to the mutual influence of antennas and the mast device, typical of the analogue [8], exceed the general systematic errors typical for direction finder implementing the proposed method of direction finding.

The proposed method of direction finding in comparison with the closest analogue [10] provides:

firstly, with the same ratio of the distance b between the antennas of the three-element array to the wavelength of the radio signal, a 2-fold reduction in the "separation" errors;

secondly, with the same maximum permissible probability of occurrence of anomalous direction finding errors associated with distortion of the phase patterns of antennas due to the mutual influence of antennas and a mast device, and random errors in measuring phase differences between signals received by antennas with significantly different amplitude radiation patterns, the possibility increase (1.9 ÷ 2.2) times the distance b between the antennas of the three-element array, which provides a corresponding improvement in the limiting sensitivity p direction finding;

thirdly, the elimination of the methodological components of errors in determining the angle β of the slope of the wave front of the radio signal due to the mutual influence between the antennas and the influence of the mast device.

In addition, the proposed method of direction finding and direction finding for its implementation in comparison with the closest analogue [10] and other analogs [8], [9] allow us to evaluate the reliability of the results of direction finding (determining the presence of a jamming radio signal at the time of measurement or evaluating the signal / (noise ratio) + interference)) without statistical processing of the measurement results, usually used in the above cases, which improves the efficiency of the operation of the direction finder in complex electronic environments and enables the creation of high-speed automatic direction finders.

, отсутствие зависимости погрешности определения азимута θ и угла β наклона фронта волны от степени электродинамического взаимодействия между антеннами трехэлементной решетки и влияния мачтового устройства, возможность оценивания отношения сигнал/(помеха + шум) без необходимости статистической обработки результатов измерений в течение увеличенного промежутка времени, позволяет повысить точность и угловую чувствительность малобазовых радиопеленгаторов и обеспечить возможность оперативного оценивания достоверности результатов пеленгования.Small "separation" errors typical of the proposed direction finding method and the direction finder for its implementation do not exceed 0.45 ° at the maximum relative size of the grating base for unambiguous direction finding

, the absence of dependence of the error in determining the azimuth θ and the angle β of the wavefront inclination on the degree of electrodynamic interaction between the antennas of the three-element array and the influence of the mast device, the possibility of estimating the signal / (noise + noise) ratio without the need for statistical processing of measurement results over an extended period of time, allows accuracy and angular sensitivity of low-base direction finders and provide the ability to quickly evaluate the reliability of the results of langaniya.

The most successfully claimed radio direction finding method and direction finder for its implementation can be used in broadband high-speed mobile complexes for detecting and determining the location of radio emission sources, which are intended, inter alia, for functioning in a complex electronic environment.

Claims (2)

1. A radio direction finding method, including receiving a radio signal using three antennas made by identical non-directional axisymmetric vibrator types, forming an equidistant annular antenna array in the direction-finding plane, wherein the position of the phase centers of the first, second and third antennas is oriented relative to the reference direction in the direction-finding plane passing through the center antenna array, at angles

radians, respectively, and the geometric dimensions of the antennas along their symmetry axes are commensurate with the wavelength λ of the radio signal, the measurement of phase differences φ _i between the signals

and

adopted by the n-th and k-th antennas, in accordance with the formula

where i = 1, 2, 3;

- Kronecker symbol;

- Kronecker symbol, differential signal generation

and their amplitude values r _i , according to the formula

and measuring the azimuth θ _{R of the} radio source using three unique amplitude values of the difference signals R _i according to the formula

characterized in that it further measures the amplitude values of u _i , signals received by the i-th array antennas, in accordance with the expression

and form the coefficients P _i and K _{i the} irregularities of the antenna patterns of the array in the array according to the formulas

Where

- sign function, choose from three values of the indices i serial numbers of the antennas i = 1, i = 2, i = 3 one value of the index ξ, one value of the index γ and one value of the index v, not equal to each other, from the condition r _ξ P _ξ ≤r _γ P _γ ≤r _v P _v , moreover, the value of the index ξ is assigned the value of the index i at which the product r _i P _i is the minimum or one of the minimum if the above products r _i P _i are equal, and the value of the index v is assigned the value of the index i at which the product r _i P _i is the maximum or one of the maximum if the above products are equal, r _i P _i , and the remaining value of index i is assigned to the index value γ, the direction finding coefficient p is determined in accordance with the expression

where K _mid - a priori known average value of the coefficients K _{i the} unevenness of the antenna patterns, depending on the electrodynamic dimensions of the antennas, the design of the lattice and mast device, form three unambiguous amplitude values of the difference signals R _i if the distance b between the antennas does not exceed three tenths of the wavelength λ of the radio signal, in accordance with the expression R _i = r _i sgn {F _i }, Where

and if the distance b between the antennas exceeds three tenths of the wavelength λ of the radio signal, in accordance with the expression

Where

- Kronecker symbol,

- sign function of the parameter Y, taking the values Y = φ _i , or Y = R _γ, respectively, measure the phase difference φ _Ri between the difference signals

according to the formula

Where

- sign function of the parameter Y, taking values Y = R _k or Y = R _n, respectively, determine the value of the parameter μ, characterizing the presence of the quadrature component of the interfering signal, according to the formula

checking that the conditions for exceeding the minimum amplitude r _{ξ of the} difference signals relative to the a priori known minimum amplitude of the difference signal r _min are checked according to the formula r _ξ ≥r _min , r _min = q _min U _eff ; q _min - a priori known minimum necessary signal-to-noise ratio, providing direction finding of radio emission sources with given accuracy and probability; U _eff - the effective value of the voltage of the internal noise of the channels of the formation of differential signals, determine the azimuth θ _{φ of the} radio signal source using phase differences φ _Ri between the difference signals according to the formulas

where b is the distance between the antennas, not exceeding two third of the wavelength λ of the radio signal, determine the error Δθ azimuth estimates θ by the formula

Where

determine the azimuth θ and evaluate the angle β of the slope of the wave front of the source of the radio signal according to the formulas

Where

β _sp is a sign of the presence at the receiving point of a radio signal propagating in the form of a spatial electromagnetic wave, the front slope of which cannot be determined and is within

Δθ _max - a priori known value of the maximum permissible error in determining the azimuth θ, and the values of the parameter μ and the error Δθ judge the reliability of the results of determining the azimuth θ and the angle β of the slope of the wave front of the radio signal source, and this reliability is inversely proportional to the values of the parameter μ and the error Δθ of determining the azimuth of the radio signal. 2. A direction finder comprising three antennas made of identical non-directional axisymmetric vibrator types forming an equidistant annular antenna array in the direction-finding plane, wherein the position of the phase centers of the first, second and third antennas is oriented relative to the reference direction in the direction-finding plane passing through the center of the antenna array, at angles

radians, respectively, and the geometrical dimensions of the antennas along their symmetry axes are commensurate with the wavelength λ of the radio signal, three identical radio receiving units made with a common local oscillator, the inputs of which are connected to the outputs of the corresponding antennas, three phase difference measuring units, three differential signal generating units, a comparator, a unit for generating unambiguous amplitude values of the difference signals, an amplitude azimuth calculator, an elevation angle calculator, a calculation parameter sensor and a control signal generator, wherein and the outputs of the first radio receiving unit are connected respectively to the second pairs of inputs of the second phase difference measuring units and generating differential signals and the first pairs of inputs of the third phase difference measuring units and generating differential signals, the pair of outputs of the second radio receiving unit is connected respectively to the first pairs of inputs of the first phase difference measuring units and the formation of differential signals and second pairs of inputs of the third blocks for measuring the phase difference and the formation of differential signals, a pair of outputs tr the third radio receiving unit is connected respectively to the second pairs of inputs of the first phase difference measuring units and generating differential signals and the first pairs of inputs of the second phase difference measuring units and generating differential signals, the outputs of the first, second and third phase difference measuring units, the first outputs of the first, second and third blocks of the formation of differential signals and the first, second, and third outputs of the comparator are connected respectively to the first, second, third, fourth, fifth, sixth, seventh, eighth and nine the input inputs of the unit for generating unique amplitude values of difference signals, the first, second and third outputs of which are connected respectively to the first, second and third inputs of the amplitude azimuth calculator, the output of the control signal generator is connected to the control inputs of the first, second and third radio receiving blocks and the control input of the parameter sensor calculations, the first and second outputs of which are the outputs of the signal corresponding to the a priori known distance b between the antennas, and the signal, respectively which corresponds to the value of the wavelength of the radio signal λ, are connected respectively to the first and second inputs of the elevator calculator, the output of which is the output of the value of the angle β of the slope of the wave front of the radio signal source, characterized in that the distance between the antennas is chosen not exceeding two-thirds of the wavelength of the radio signal, the comparator is made with the ability to determine an ordered combination of three antenna numbers, through the phase centers of which the front of the electromagnetic wave of the radio signal source passes sequentially in time and, a sensor computing parameters configured to generate a priori known average value K _mid coefficients unevenness antenna pattern consisting of a lattice, the minimum difference signal amplitude r _min, which provides a minimum required signal / noise ratio q _min relative to the current voltage value U _eff internal noise channels for generating differential signals of the direction finder, and the maximum allowable error Δθ _{max for} determining the azimuth specific values of the difference signals is configured to generate the amplitude values of the difference signals taking into account the signs of the phase difference between the signals received by the antennas, if the distance b between the antennas does not exceed three tenths of the wavelength λ of the radio signal, or taking into account the results of comparing the amplitudes of the signals received antennas of a three-element array, if the distance b between the antennas exceeds three tenths of the wavelength λ of the radio signal, the elevation angle calculator is adapted to adaptively estimates of the elevation angle using phase differences between the difference signals received by three different pairs of antennas, depending on the ratio of the minimum amplitude of the difference signals to the effective value of the internal noise voltage of the channels for generating the differential signals of the direction finder and the error in estimating the azimuth of the radio signal, and an additional block for generating amplitude non-uniformity coefficients antenna patterns, the first, second and third pairs of inputs of which are connected to pairs of outputs ne of the first, second and third radio receiving units, respectively, a direction finding coefficient calculator, a noise threshold coefficient calculator, a phase difference determination unit between difference signals, a quadrature component of an interfering signal, a phase azimuth calculator, an azimuth estimation error calculator, an azimuth threshold coefficient calculator and an azimuth determination unit wherein the second and third outputs of the first, second, and third difference signal generating units are connected respectively, with the first, second, third, fourth, fifth and sixth inputs of the phase difference determination unit between the difference signals, and the first outputs of the first, second and third difference signal generation units are connected respectively to the combined first, combined second and combined third inputs of the noise comparator and calculator threshold coefficient, the first output of the comparator is connected to the combined fourth input of the noise threshold factor calculator and the first inputs of the coefficient calculator direction finding direction and phase azimuth calculator uniqueness, the first, second, third, fourth, fifth and sixth outputs of the unit for generating the unevenness of the amplitude antenna radiation patterns are connected to the fourth, fifth and sixth inputs of the comparator and the second, third and fourth inputs of the direction finding coefficient calculator, the fifth input and output of which are connected respectively to the third output of the calculation parameter sensor, which is the output of the signal corresponding a priori the known average value of the coefficient of non-uniformity of the antenna radiation patterns in the antenna array, and the combined tenth input of the unit for generating unique amplitude values of the difference signals and the second input of the phase azimuth calculator, the first, second and third outputs of the unit for generating the unique amplitude values of difference signals are connected respectively to the combined first input calculator of the quadrature component of the interfering signal and the seventh input of the phase difference determination unit two difference signals, with the combined second input of the calculator of the quadrature component of the interfering signal and the eighth input of the unit for determining the phase difference between the difference signals and with the combined third input of the calculator of the quadrature component of the interfering signal and the ninth input of the unit for determining the phase difference between the difference signals, the first, second and third outputs which are connected respectively to the combined third, combined fourth and combined fifth inputs of the elevator and phase calculator of the azimuth calculator, the combined sixth inputs of the phase azimuth calculator and elevator calculator and the first input of the azimuth determination unit are connected to the output of the noise threshold coefficient calculator, the fifth input of which is connected to the fourth output of the calculation parameter sensor, which is the output of the signal corresponding to the a priori known value of the minimum difference amplitude signal at which the minimum necessary signal-to-noise ratio is provided relative to the effective value of the internal voltage of the noise of the channels for forming the differential signals of the direction finder, the output of the phase azimuth calculator is connected to the combined first input of the azimuth estimation error calculator and the second input of the azimuth determination unit, the third input of which, combined with the second input of the azimuth estimation error calculator, is connected to the output of the amplitude azimuth calculator, the output of the calculator azimuth estimation errors and the fifth output of the calculation parameter sensor, which is the output of the signal corresponding to the a priori known value of m the maximum permissible azimuth determination errors are connected respectively to the first and second inputs of the azimuth threshold coefficient calculator, the output of which is connected to the combined seventh input of the elevation calculator and the fourth input of the azimuth determination unit, the first and second outputs of the calculation parameter sensor are connected respectively to the combined eleventh input of the formation unit unambiguous amplitude values of the difference signals and the seventh input of the phase azimuth calculator and with the combined twelve the input of the unit for generating unique amplitude values of difference signals and the eighth input of the azimuth phase calculator, the output of the azimuth determination unit being the output of the azimuth value 9 of the radio signal source, and the outputs of the quadrature component of the interfering signal and the azimuth estimation error calculator being the outputs of the reliability parameters of direction finding results μ and Δθ respectively.

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