RU2097785C1 - Phase parametric sonar - Google Patents

Abstract

FIELD: echo-sounding measuring systems, intended for classifying of bottom deposits and for detection and classifying of bottom and bottomside objects according to the acoustic impedance. SUBSTANCE: phase parametric sonar uses series-connected synchronizer 1, RF pulse generator 2, power amplifier 3, radiator 4, series-connected phase-shift meter 10 and indicator 11, two selective amplifiers 8 and 9, whose outputs are connected to the respective inputs of phase-shifter 10. The sonar has control signal generator 5 and series-connected audio-frequency antenna 6 and difference frequency signal amplifier 7, whose output is connected to the inputs of selective amplifiers 8 and 9, the control input of the first of them is connected to the first control output of RF pulse generator 2, and the control input of the other - to the output of control signal generator 5, whose two inputs are connected respectively to the two control outputs of RF pulse generator 2. The phase parametric sonar features a wide operating frequency band (more than an octave). EFFECT: enhanced accuracy and truth of the results of echo-signal phase-shift measurement. 3 cl, 11 dwg

Description

 The invention relates to hydroacoustic measuring systems and is intended for the classification of bottom sediments, as well as for the detection and classification of bottom and bottom objects by acoustic impedance. The classification of types of bottom soils and location objects by acoustic impedance is carried out using a biharmonic signal with multiple frequencies [1, 2]. Receiving multi-frequency signals that are rigidly coupled in phase is ensured by the use of parametric excitation of the emitter.

 There are a number of devices for determining the characteristics and remote classification of underwater objects using phase-frequency characteristics (PFC) [1] The use of conventional linear antennas in devices of this type (for example, the Federal Republic of Germany patent No. 2006153) is associated with a number of disadvantages: the radiating converter must be very broadband (not less than an octave), the generator blocks must have identical phase response, in addition, as a result of inhomogeneities in the water, the phase synchronism of the emitted signals is broken.

 The known device [3] adopted as a prototype, contains a serially connected synchronizer, a radio pulse generator, a power amplifier and an acoustic emitter, two echo signal amplification channels, each of which contains a serially connected receiving transducer, a selective (resonant) amplifier and an amplitude limiter, as well as a doubler frequency, phase detector and indicator connected to the output of the phase detector, the first input of which is connected through the frequency doubler to the output of the first channel eniya echo signal, and the second with the output of the second channel gain.

In this device, the acoustic emitter operates in the mode of emission of pulsed harmonic oscillations [3] In the medium, due to its nonlinearity, harmonics are formed with frequencies that are multiples of the frequency of the emitted signal f ₀ , ie 2f ₀ , 3f ₀ , etc. When reflected from an underwater object or bottom soil, a change (shift) in the phase of harmonics occurs due to the difference in the acoustic impedance of the object from the water impedance. The first and second harmonics of the echo signals are used to determine this phase shift.

 The prototype device has a number of significant drawbacks, the main of which is the small frequency range for measuring the phase response of location objects. The latter is limited by the broadband of both the transmitting and receiving antennas, and in this device, it is no more than a few percent of the carrier frequency (within the antenna bandwidth).

 It must also be borne in mind that high-frequency oscillations do not penetrate into the soil and object and therefore cannot be used to classify bottom sediments, as well as to detect and classify silted objects.

 Other significant disadvantages of the prototype are low accuracy and reliability of the measurement of phase shift. This is due to the fact that the phase shift meter is based on a phase detector with preliminary doubling of the frequency of oscillations of the first harmonic of the echo signal. As is known [5], the value of the signal at the output of the phase detector, which in this case is an estimate of the phase shift, depends not only on the phase difference of the input signals, but also on their amplitudes. The influence of the amplitudes of the input signals on the measurement result is reduced by amplitude limiters at the input of the phase detector. Naturally, the error in the operation of the amplitude limiters, especially with a large dynamic range of input signals, significantly affects the measurement result. The quality of the phase detector also strongly depends on the symmetry of the circuit and the stability of the parameters of its elements.

 In addition, when measuring the phase difference of pulsed signals (with a duration of 10 15 periods of the carrier frequency), a problem arises in realizing high speed of the phase meter with a low level of ripple of the output signal. In this case, the load time constant of the phase detector must satisfy conflicting requirements. On the one hand, its value should be less than 1 -2 periods of the carrier frequency, and on the other, significantly exceed this period.

 The objective of the invention is the creation of a phase parametric sonar operating in a wide frequency range, as well as with increased accuracy and reliability of the measurement of phase response objects of the location. This is achieved by the fact that bi-harmonic echo signals of difference frequency waves (VChF) are used to measure the phase response, arising in the medium when a signal is emitted with in-pulse amplitude modulation, as well as amplification of the echo signals of the VChF is carried out by double frequency conversion and the measurement of the actual phase shift is carried out on the basis of fixing the transition moments through the “zero” echo of the RF: second harmonic in the positive direction, and the first in both directions.

 The specified technical effect is achieved by the fact that a shaper is introduced into the well-known sonar, which contains a serially connected synchronizer, a radio pulse generator, a power amplifier and an emitter, a serially connected phase shift meter and an indicator, and two selective amplifiers, the outputs of each of which are connected to the corresponding inputs of the phase shift meter a control signal and a series-connected low-frequency antenna and an amplifier of signals of difference frequencies, the output of which is sub li ne selective amplifiers to the inputs, a first control input of which is connected to the first output of the control radio pulse generator, and the second control input to the output driver control signal, two inputs of which are connected respectively with two control outputs radio pulse generator.

 In the proposed sonar, selective amplifiers can be performed according to the double frequency conversion scheme and contain in series the first signal multiplier, a fixed-bandpass filter, a second signal multiplier and a low-pass filter (LPF), the output of which is the output of the amplifier, the input of which is a signal input the first signal multiplier, the control input of which is connected in parallel with the control input of the second signal multiplier and is the control input ohm selective amplifier.

 In the proposed sonar, the phase shift meter can contain two zero-level detectors, an RS trigger and a series-connected scale generator, a counter, a register and a digital-to-analog converter, the output of which is the output of the phase shift meter, the pair of inputs of which are respectively the inputs of the zero level latches, the first of which which is connected to the input S of the RS-flip-flop, and the output of the second to the input R of the RS-flip-flop, the output of which is connected to the counter resolution enable input and the register entry input.

In FIG. 1 is a structural diagram of the proposed sonar; in FIG. 2
5 structural diagrams of its individual blocks: FIG. 2 radio pulse generators, FIG. 3 shaper control signal, FIG. 4 of a selective amplifier (one of the harmonics of the RF echo), FIG. 5 phase shift meters; in FIG. 6 time diagrams of the operation of the radio pulse generator 2; in FIG. 7 - oscillograms at the output of blocks 3, 6 9; in FIG. 8 and 9 are spectral diagrams (spectrograms) explaining the principle of operation of the selective amplifier and driver of the control signal; in FIG. 10 timing diagrams of the phase shift meter; in FIG. 11 oscillograms of signals obtained in a full-scale experiment.

 The phase parametric sonar (Fig. 1) contains a synchronizer 1, a radio pulse generator (GRI) 2, a power amplifier 3, an emitter 4, a driver of the control signal (FUS) 5, a receiving low-frequency antenna 6, an amplifier of signals of difference frequencies (UDMF) 7, selective amplifiers 8 and 9, phase shift meter (IFS) 10, indicator 11.

 The synchronizer 1, the radio pulse generator 2, the power amplifier 3 and the emitter 4 are connected in series. Two control outputs of the radio pulse generator 2 are connected to the corresponding inputs of the driver of the control signal 5. The receiving low-frequency antenna 6 is connected to the input of the difference signal amplifier 7, the output of which is connected to the inputs of the selective amplifiers 8 and 9, the control inputs of which are connected respectively to the first control output of the radio pulse generator 2 and the output of the driver of the control signal 5. The output of each selective amplifier 8 and 9 is connected to the corresponding input of the phase shift meter 10, whose output is connected to the input 11 of the indicator.

 The synchronizer 1 can be made in the form of series-connected master oscillator and counter. As the master oscillator, a multivibrator on the timer 1006VI1 [4] is used and the counter is run on standard microcircuits of 176 or 561 series (for example, K176IE8).

Radio pulse generator (GRI) 2, an example of implementation of which is shown in FIG. 2 contains two synchronized generators 12 and 13, respectively, with frequencies f _n and f _g1 , the first signal multiplier 14, the low-pass filter (LPF) 15, the second signal multiplier 16, the electronic key 17 and the standby multivibrator 18, the input of which is connected to the output, are connected in series synchronizer 1, and the output is connected to the control input of the electronic key 17, the output of which is the output of the GRI 2. The outputs of the generators 12 and 13, which are the first and second outputs of the control of the GRI 2, respectively, are connected to the first and the second inputs of the first multiplier 14, and the output of the generator 12 is also connected to the second input of the multiplier 16. The synchronization inputs of the generators 12 and 13 are interconnected and are the input of the radio pulse generator 2. The multipliers are implemented on microcircuits (such as K140MA1, K525PS1-K525PS3, etc.) . The low-pass filter is designed as a traditional Butterworth filter based on LC elements and has a passband (of the order of 30 kHz) sufficient to transmit the difference frequency F f _n f _n1 . Generators 12 and 13, an electronic key and a standby multivibrator are built according to standard schemes.

 Power amplifier 3 is performed according to a typical circuit of a push-pull power amplifier based on KT847A transistors.

The emitter 4 can be made on the basis of piezoceramics. In the current model of the phase parametric sonar, a round radiator with a diameter of 100 mm based on piezoceramics of the T5K type is used. The resonant frequency of 300 kHz, the radiation pattern (LH) of the antenna at the pump frequencies in both planes has a width of the order of 4 ^o .

 The control signal generator (FCS) 5 (Fig. 3) contains two signal multipliers 19 and 20 and the low-pass filter 21 connected in series, the two inputs of the first multiplier 19 being combined and being the first input of the FSF, the second input of the second multiplier 20 being the second input of the FSF, and the output Low-pass filter 21 by the output of the driver of the control signal.

 As a receiving low-frequency antenna 6, a standard 1P-2G hydrophone with a working frequency range from 0.5 Hz to 20 kHz and a sensitivity of 180, 150 and 120 μV / Pa, respectively, at frequencies of 2, 10 and 20 kHz can be used.

 The differential frequency signal amplifier (UDMF) 7 is a series-connected low-pass filter with a passband sufficient to transmit the echo signals of the difference frequency waves (F and 2F), and a low-frequency amplifier made, for example, on the basis of a standard operational amplifier based on the K548UN1 chip.

 The selective amplifiers 8 and 9 are made according to the double frequency conversion scheme (Fig. 4) and contain in series the first signal multiplier 22, a fixed-bandpass filter 23, the second signal multiplier 24 and the low-pass filter 25, the output of which is the output of the selective amplifier, the input of which serves the signal input of the first signal multiplier 22, the control input of which is connected in parallel with the control input of the second signal multiplier 24 and is the control input of the selective amplifier.

 The phase shift meter (IFS) 10, the implementation diagram of which is shown in FIG. 5, contains two zero-level latches 26 and 27, an RS flip-flop 28, and a scale generator 29, a counter 30, a register 31, and a digital-to-analog converter (DAC) 32, the output of which is the output of a phase shift meter, the two inputs of which are inputs of the first, respectively, in series and the second zero-level latches 26 and 27, the output of the first of which is connected to the input S of the RS-flip-flop 28, and the output of the second to the input R of the RS-flip-flop 28, the output of which is connected to the counter resolution input of the counter 30 and the register write input 31.

The sonar designed to classify bottom sediments and location objects is based on the study of their phase-frequency characteristics (PFC) using biharmonic phase-coupled signals generated in the medium when signals are emitted with intrapulse amplitude modulation. The carrier oscillation is modulated by a harmonic signal with a frequency F. The spectrum of such an oscillation contains three frequency components (f _n , f _n F, f _n + F), between which there is a nonlinear interaction in the medium. As a result, oscillations with combination frequencies (f _comb kf _n + m • (f _n + F) + n • (f _n F)) are formed, including waves of difference frequencies (F, 2F). The result of the interaction of the carrier frequency fluctuation (f _n ) with each of the side components (f _n F, f _n + F) is the formation of the first harmonic of the VGF with the frequency F. The interaction of the side components of the AM vibration with each other ensures the formation of the second harmonic of the VGF with the frequency 2F. The study of the phase structure of the echo signals of the RF allows the classification of objects according to their acoustic rigidity. In contrast to the known in the proposed sonar, the study of the phase response of location objects is simply implemented in a wide frequency range (more than two octaves) by changing the modulation frequency of the emitted signal. In this case, the reception of echo signals of the RF is provided by one low-frequency antenna.

 The proposed sonar operates as follows.

Synchronizer 1 sets the time mode of the sonar and generates a pulsed synchronization signal (U ₁ in Fig. 6, a), which provides periodically binding phase oscillations of the generators 12 and 13 of the GRI 2.

где k коэффициент пропорциональности (определяется типом перемножителя);
U = kU_oU_ω амплитуда несущего колебания;
M = U_Ω/U_o коэффициент модуляции, очевидно, что значение М может изменяться в широких пределах регулировкой как амплитуды U_Ω сигнала с выхода ФНЧ 15, так и смещения U₀.GRI 2 provides the formation of a pulse with the required characteristics and can be performed, for example, according to the circuit of FIG. 2. Generators 12 and 13 form harmonic oscillations U ₁₂ and U ₁₃ (Fig. 6, b, c) with frequencies f _n and f _r1, respectively. Generators 12 and 13 operate in synchronization mode — oscillations of U ₁₂ and U _{13 are} forced to fail for the duration of the synchronization signal U _I (Fig. 6, a, b, c). The duration of the synchronization pulse τ _sync is chosen so that during its operation all the transients in the generators after the breakdown of oscillations are over. At the end of the synchronization pulse, the generators 12 and 13 are started. Moreover, the oscillations of these generators after starting always begin with the same phase. In the multiplier 14 is an analogue multiplication of the oscillations U ₁₂ and U ₁₃ (respectively, with frequencies f _n and f _g1 ). The resulting oscillation U ₁₄ (plot r in Fig. 6) represents the sum of two harmonic oscillations with frequencies f _n + f _g1 and F f _n f _g1 . The filter 15 selects the oscillation U _{15 of the} differential frequency F (plot d), which is fed to the first input (input X1) of the multiplier 16. The bias voltage U ₀ is applied to the second input (input X2), and the oscillation U _{12 of the} carrier frequency is applied to the third input (input X1). f _n At the output of block 16, an oscillation U _{16 is formed} (plot e)

where k is the coefficient of proportionality (determined by the type of multiplier);
U = kU _o U _{ω the} amplitude of the carrier oscillation;
M = U _Ω / U _o the modulation coefficient, it is obvious that the value of M can vary widely by adjusting both the amplitude U _{Ω of the} signal from the output of the low-pass filter 15 and the offset U ₀ .

The waiting multivibrator 18 is started periodically by the trailing edge of the synchronization pulses U ₁ and generates pulses U ₁₈ (diagram w), closing the key 17. As long as the key is closed to its output, the oscillation U ₁₆ passes. In this way, a U ₁₇ radio pulse (curve h) is formed with intrapulse amplitude modulation
U ₁₇ = U (1 + MsinΩt) • sinω _n t 0 <t <τ _and .
The radio pulses are repeated periodically with a period T of the synchronization signal U ₁ . In this case, the initial phases of the carrier frequency (f _n ) and frequency (F) envelope are constant from pulse to pulse.

The pulse signal U ₁₇ with GRI 2 is amplified in a power amplifier 3 (U ₃ at M> 1 in Fig. 7, a). A feature of power amplifier 3 is the high linearity of the amplitude characteristic, eliminating the appearance of a difference frequency signal directly in the transmission path.

The amplified signal is supplied to the emitter 4 and is radiated into the water. When powerful acoustic waves are emitted, water exhibits nonlinear properties. The interaction region of these waves is a section with a length of 1 _WZ inversely proportional to the attenuation coefficient (1 _WZ 1 / β). The width of the interacting wave beams is determined by the directivity characteristic of the initial high-frequency components, called pump waves. Thus, the primary emitter 4 and the interaction region 1 are two main elements of a radiating parametric antenna (IPA).

Nonlinear interaction of pump waves in water with a carrier frequency f _n , upper and lower side frequencies f _WB = f _n + f and f _NB f _n F gives waves of combination frequencies
f _comb kf _n + mf _WB + nf _NB , (2)
where k, m, n are natural numbers: 0, 1, 2, 3. i.e. second harmonics 2f _n , 2f _WB 2 (f _n + F), 2f _NB 2 (f _n - F), oscillations of the total frequencies f _n + f _WB 2f _n + F, f _n + f _NB 2f _n F, difference frequency waves F f _n - f _NB , 2F f _WB f _NB . Due to the power-law frequency dependence of the attenuation (β ~ f ⁿ ), where n 1 2), the waves of the fundamental frequencies, harmonics, and waves of the total frequencies decay faster than the waves of difference frequencies (F, 2F). Therefore, the source waves and the VLF will propagate beyond the interaction region 1 of the _{VZ of the} initial waves, while the latter propagate over long distances due to small attenuation.

где S_Э1 и S_Э2 амплитуды на частотах F и 2F, зависящие от эффективной площади рассеяния, АЧХ объекта и расстояния D до него;
Φ_o(Ω) и Φ_o(2Ω) фазовые сдвиги при отражении на частотах F и 2F, зависящие от акустической "жесткости" объекта.So, when an IPA is excited by a three-component signal in water, a two-component phase-coupled signal of the VLF is formed:
S (t) = S ₁ sinΩ ₁ t + S ₂ sinΩ ₂ t, 0 ≅ t ≅ τ _and , (3)
where Ω ₁ = Ω = 2πF, Ω ₂ = 2πF ₂ , F ₂ = 2F.
The echo signal that forms when the signal (3) is reflected from objects has the form:

where S _E1 and S _E2 amplitudes at frequencies F and 2F, depending on the effective scattering area, the frequency response of the object and the distance D to it;
Φ _o (Ω) and Φ _{o (2Ω)} phase shifts upon reflection at frequencies F and 2F, depending on the acoustic "rigidity" of the object.

 Antenna 6 receives the RF echo (4), noise, and high frequency echo. The noise and the RF echo signal are partially suppressed in the antenna 6. The antenna output signal is shown in FIG. 7, b of the photograph obtained during field tests of the current layout of the sonar. The amplifier 7 provides a linear amplification of the echo of the RF and additionally suppresses high-frequency oscillations that might not have been suppressed by the low-frequency antenna 6 (Fig. 7, c). A feature of amplifier 7 is the presence of a low-pass filter at its input, which suppresses the RF signal by 50-60 dB.

 Pre-amplified in block 7, the signals of the RF are fed to the selective amplifiers 8 and 9, made according to the scheme (Fig. 4). Amplifiers 8 and 9 use double frequency conversion, which allows using the bandpass filter with a fixed tuning to filter the desired harmonic in a given frequency band.

 The principle of operation of the selective amplifier 8 or 9 is explained using spectrograms in FIG. 8. For simplicity of explanation, the spectrograms are given without taking into account the pulse nature of the signals.

At the input of each of the selective amplifiers, the signal U ₇ acts, which is an additive mixture of the oscillations of the first and second harmonics of the RF echo signal, as well as noise residues. The waveform of this signal is shown in FIG. 7c, and the spectrogram in FIG. 8 a.

The multiplier 22 (Fig. 4) transfers the spectrum of the signal U ₇ using the oscillation U ₁₃ (the spectrum is given in Fig. 8, b) to the high-frequency oscillation region (Fig. 8, c). The fixed-bandpass filter 23 has a center frequency f _p matching the carrier frequency f _{n of the} generator 12 (Fig. 8d). One of the components of the output signal U ₂₂ , namely the component with a frequency f _g1 + F, corresponding to the first harmonic (F) of the echo signal, falls into the passband of the filter 23, and the remaining components are suppressed (Fig. 8, e). In this case, regardless of the frequency F of the signal component U ₂₂ with a frequency f _g1 + F will always be in the band of the filter 23, since in the generator of the radio pulse 2 the relation f _g1 + F f _n takes place. The output of the multiplier 24 is a two-component signal U ₂₄ (t). Next, with the help of the low-pass filter 25, the first harmonic (F) of the echo of the RF frequency is extracted, the oscillogram of which is shown in FIG. 7d, and the spectrogram in FIG. 8, e.

The echo signal undergoes similar transformations when filtering the second harmonic of the (2F) HFC with the only difference that the forward and backward transfers of the spectrum of the signal are performed using U ₅ oscillations with a frequency f _g2 . An oscillogram of the second harmonic of the frequency response from the output of amplifier 9 is given in FIG. 7, d.

 The peculiarity and advantage of the selective amplifier is that due to the use of double frequency conversion, the required harmonic of the RF echo signal is extracted in a given range of its change by a bandpass filter with a fixed setting, and not a tunable filter.

To generate the signal U ₅ with a frequency f _r2 , which provides the second harmonic of the RF frequency, a block 5 is used, the functional diagram of which is shown in FIG. 3, and the explanatory spectrograms in FIG. 9. The scheme is synthesized in accordance with the equality
f _g2 2f _g1 f _n (5).

The multiplier 19 doubles the frequency f _g1 oscillations U ₁₃ (Fig. 9, a) by multiplying the input signal itself (kU ₁₃ • U ₁₃ = kU $\genfrac{}{}{0ex}{}{\text{2}}{\text{g1}}$ sin ² ωt = 0.5kU $\genfrac{}{}{0ex}{}{\text{2}}{\text{g1}}$ + 0.5U $\genfrac{}{}{0ex}{}{\text{2}}{\text{g1}}$ sin2ωt The signal U ₁₉ is supplied to the first input (X1) of the analog multiplier 20, to the second input (Y1) of which the signal U _{12 of the} carrier frequency f _n is received (Fig. 9, b). As a result of multiplication, the spectrum of the output oscillation U ₂₀ will contain components with a total frequency of 2f _g1 + f _n and a difference frequency f _g2 2f _g1 f _n (Fig. 9, c). The low pass filter 21, the amplitude-frequency characteristic of which is shown by the dashed line in FIG. 9c, only components with a difference frequency f _g2 2f _g1 f _{n are selected} .

 The tonic harmonics filtered in amplifiers 8 and 9 are fed to the corresponding inputs of the phase shift meter 10, the functional diagram of which is shown in FIG. 5.

In the known device, the measurement of the phase shift is provided by measuring the phase difference, reduced to the same frequency by the oscillations of both harmonics of the VLF. For this, the oscillation frequency of the first harmonic is doubled in the frequency multiplier, and then using the phase detector, the oscillation phases of the converted first and second harmonics of the echo signal are compared. This method of obtaining information about the phase shift has a number of disadvantages. The detection result depends not only on the phase difference of the studied signals, but also on their amplitudes. In the prototype, amplitude limiters are used to reduce the influence of the signal level on the measurement result. However, with a large dynamic range of echo signals, the error in the operation of the limiters significantly affects the accuracy of the obtained estimate of the phase shift. The quality of the phase detector to a large extent depends on the symmetry of the circuit and the stability of the parameters of its elements. It should also be noted certain difficulties in coordinating a low level of ripple and a sufficiently high speed. On the one hand, to obtain small ripples, the load time constant of the phase detector must satisfy the condition τ _n >> 1 / F, and on the other hand, its value should not be more than (1 - 2) / F.

 The noted disadvantages are eliminated in the proposed scheme of the phase shift meter. The operation of the circuit is based on measuring the interval between the moments corresponding to the intersection of the zero level with the first harmonic signal and the intersection of the zero level in the positive direction with the second harmonic signal.

In FIG. 10 shows stress plots at individual points of the phase shift meter. FIG. 10a corresponds to the reflection of the VLF from an acoustically “rigid” object (the phase shift is zero), FIG. 10, b from an acoustically "soft" object (phase shift is 180 ^o ).

где
k 0, 1, 2, Откуда

В моменты времени t_1К на выходе фиксатора 26 формируются короткие импульсы U₂₆(t) (фиг. 6,б).It is convenient to consider the operation of block 10 using the example of the situation illustrated in FIG. 10 b Filtered signals of the first U ₁ (t) U ₈ (t) and the second U ₂ (t) U ₉ (t) harmonics of the VCH are fed to the input of the corresponding zero-level clamp 26 and 27. Comparators of the КР554САЗ type can be used as zero-level clamps. In the latch 26, all moments of the intersection of the zero level with the first harmonic are fixed. They are determined from the condition:

Where
k 0, 1, 2, where

At time t _1K , short pulses U ₂₆ (t) are formed at the output of the latch 26 (Fig. 6, b).

или
2Ω(t_2k-2D/c)+Φ_o = 2kπ,
где k 0, 1, 2.In the latch 27, the moments of crossing the zero level by the oscillation of the second harmonic of the LFD are recorded only in the positive direction. These time points are determined from the condition:

or
2Ω (t _2k -2D / c) + Φ _o = 2kπ,
where k 0, 1, 2.

Where from
t _2k = (1 / 2Ω) (2kπ-Φ _o + 4D / c). (7)
At time t at the output of the latch 27 also produces short pulses U ₂₇ (t) (Fig. 10, b).

где ΔΦ дискретность измерения фазы,
T_W 1/F период колебаний первой гармоники ВРЧ,
Dt_max = Φ_o.max/2Ω = 1/2F, Φ_o.max = 2π.
Начальная установка счетчика 30 осуществляется передним фронтом импульса U₂₈(t). Накопленное счетчиком число, соответствующее количеству масштабных меток, в момент окончания импульса U₂₈(t) переписывается в регистр 31 и сохраняется в нем до момента окончания следующего импульса U₂₈(t).Based on (6) and (7), the time interval between t _1K and t _2K will be:
Δt = t _2k -t _1k = Φ _o / 2Ω. (eight)
That is, the phase shift measurement can be performed by measuring Δt
The signals U ₂₆ (t) and U ₂₇ (t) are respectively supplied to the inputs S and R of the trigger 28, at the output of which a sequence of rectangular pulses U ₂₈ (t) is formed (Fig. 6, b). The duration of these pulses Δt is proportional to the phase shift that occurred in the echo signal (formula (8)). Further, in the circuit, the duration Δt is measured by counting with a counter 30 marks of the scale frequency generated by the generator 29 (Fig. 5), during the duration of the pulse U ₂₈ (t). The period for following scale marks is selected from the condition:

where ΔΦ is the phase resolution
T _W 1 / F the oscillation period of the first harmonic of the VLF,
Dt _max = Φ _o.max / 2Ω = 1 / 2F, Φ _o.max = 2π.
The initial installation of the counter 30 is carried out by the leading edge of the pulse U ₂₈ (t). The number accumulated by the counter corresponding to the number of scale marks at the moment of the end of the pulse U ₂₈ (t) is written to the register 31 and stored in it until the end of the next pulse U ₂₈ (t).

 Thus, the value of the measured phase shift at the output of the register 31 is presented in digital form. To obtain the measurement results in analog form, the information stored in the register 31 is converted using the DAC 32.

 As follows from the above description, the proposed scheme of the phase shift meter has, in comparison with the corresponding circuit of the prototype, high accuracy and reliability of measuring the phase response of location objects.

In FIG. Figure 11 shows the oscillograms of the first harmonic voltages of the RF (a) and the output signal of the phase shift meter (b) obtained in the field experiment using an experimental model of a phase parametric sonar. As a location object, a hollow duralumin spherical cylinder filled with air was used. The sounding was carried out in a horizontal plane, perpendicular to the axis of the spherical cylinder. A hydrophone was installed in front of the spherical cylinder, which made it possible to record both the incident and reflected waves. In FIG. 11, and a more powerful signal corresponds to an incident wave. The phase difference in the incident (region 1) and reflected (region 2) waves has different values, which is naturally determined by the acoustic properties of the object. In region 3, where there are no signals and only sonar noise is active, the phase has a random value uniformly distributed in the range from 0 to 360 ^o .

 The use of a parametric antenna in the mode of radiation of a signal with in-pulse amplitude modulation, the introduction of a driver of the control signal, the implementation of selective amplifiers according to the double conversion scheme, and a phase-shift meter in the form of a phase-time interval converter allowed: firstly, to expand the relative frequency range of sonar operation ( not less than an octave), and secondly, to increase the accuracy and reliability of the obtained measurement results.

 In addition, the proposed design of selective amplifiers with double frequency conversion provides filtering of harmonics of the RF in a given frequency range using a filter with a fixed (rather than tunable) setting. This is crucial to simplify the sonar receiving path. Solving a similar problem in the prototype device complicates the design of the receiver due to the need for a coordinated tuning of the selective amplifiers when the oscillation frequency of the pump signal changes.

Sources of information
1. Telyatnikov V. I. Methods and devices for the classification of hydroacoustic signals, // Foreign Radio Electronics, 1979, N 9, p. 19 38.

 2. Voloshchenko V.Yu. Maksimov V.N. Experimental studies of a parametric locator for the classification of underwater objects, In the book. Applied Acoustics, Taganrog, TRTI, 1985, no. XII. from. 36 39.

 3. Voloshchenko V.Yu. Maksimov V.N. Timoshenko V.I. Parametric acoustic system for classifying objects of location. // Acoustics and ultrasound equipment, Kiev, 1986, N 21, p. 63 65.

 4. Colombet EA Microelectronic means for processing analog signals, M. Radio and communications, 1991, 376 p.

 5. Bobrov N.V. Maximov G.V. Michurin V.I. Nikolaev D.P. Calculation of radios, M. Military Publishing, 1971, 496 p.

Claims (3)

 1. Phase parametric sonar, containing a serially connected synchronizer, a radio pulse generator, a power amplifier and an emitter, a series-connected phase shift meter and an indicator, two selective amplifiers, the outputs of each of which are connected to the corresponding inputs of the phase shift meter, characterized in that driver of the control signal and series-connected low-frequency antenna and amplifier of signals of difference frequencies, the output of which is connected to odes selective amplifiers, the first control input of which is connected to the first output of the control radio pulse generator, and the second control input to the output driver control signal, two inputs of which are connected respectively with two control outputs radio pulse generator.  2. The sonar according to claim 1, characterized in that the selective amplifiers comprise a series-connected first signal multiplier, a fixed-bandpass filter, a second signal multiplier and a low-pass filter, the output of which is the amplifier output, the input of which is the signal input of the first signal multiplier, the control input of which is connected in parallel with the control input of the second signal multiplier and is the control input of the selective amplifier.  3. The sonar according to claim 1, characterized in that the phase shift meter contains two zero-level detectors, an RS trigger and a series-connected scale generator, counter, register and digital-to-analog converter, the output of which is the output of the phase shift meter, the pair of inputs of which are respectively the inputs of the zero level latches, the output of the first of which is connected to the S-input of the RS-flip-flop, and the output of the second to the R-input of the RS-flip-flop, the output of which is connected to the counter resolution input of the counter counter and the recording input histra.

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